JP3353586B2 - Drive device for brushless DC motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a drive device for a brushless DC motor.

[0002]

2. Description of the Related Art A conventional device for driving a brushless DC motor without a position sensor will be described. FIG. 1 is a block diagram showing a conventional driving device for a position sensorless brushless DC motor.

As shown in FIG. 1, a conventional position sensorless brushless DC motor driving device includes a rectifier circuit 2, a 120-degree conduction type voltage type inverter (inverter) 3 composed of six semiconductor switching elements, and a brushless brushless DC motor. DC
Motor 4, rotor position detecting means 5, and drive control means 7
It is composed of

The input terminal of the inverter 3 is supplied with DC power rectified and smoothed by a rectifier circuit 2 from a commercial power supply 1. An output end of the inverter 3 is connected to a stator armature winding of the brushless DC motor 4 to conduct and cut off the DC power, thereby providing a brushless DC motor.
The C motor 4 is rotated.

[0005] The brushless DC motor 4 is composed of a stator in which polyphase armature windings are star-connected, and a rotor which forms a magnetic pole pair by permanent magnets. Back electromotive force is generated.

[0006] The back electromotive force is input to the rotor position detecting means 5 from the stator armature winding end 6, and the rotor position detecting means 5 detects the position of the rotor. Is input to

The drive control means 7 includes an inverter drive circuit 8, an output pattern generation circuit 9, and a PWM control circuit 1.
0.

The output pattern generation circuit 9 determines a pattern for driving the gate of each switching element of the inverter 3 in accordance with the detection timing input from the rotor position detection means 5. Further, the output pattern generation circuit 9
A signal is generated by synthesizing this pattern with the output of the PWM control circuit 10 that determines the duty ratio of the PWM chopper control for conduction and cutoff in accordance with the rotation speed command 11, and the signal is input to the inverter drive circuit 8. Then, each gate of the inverter 3 is driven by the inverter drive circuit 8.

Next, the rotor position detecting means 5 of the conventional position sensorless brushless DC motor driving device will be described. FIG. 2 is a circuit diagram showing one phase of the rotor position detecting means 5.

As shown in FIG.
Is a voltage dividing circuit 20 composed of resistors 21 and 22, a capacitor 23 for removing a DC component, and a first-order lag filter circuit 24 composed of a resistor 25 and a capacitor 26.
And a comparison circuit 27 including resistors 28, 29, 30 and a comparator 31.

A terminal voltage of the stator armature winding end 6 is input from an input terminal 32 to a voltage dividing circuit 20, and the terminal voltage is a voltage dividing ratio of the resistors 21 and 22 based on 0 V of the DC power supply. Is detected by

The detected terminal voltage includes, in addition to the fundamental wave component of the back electromotive force, a high frequency component by the PWM chopper control, a spike voltage generated in the return mode after commutation, and the resistance of the resistors 21 and 22 of the voltage dividing circuit 20. The offset includes a DC offset due to variation. To alleviate these,
Through the connection of the capacitor 23 for cutting the DC component, the harmonic component is reduced by the first-order lag filter 24 and the phase is shifted by 90 degrees.

The comparison circuit 27 compares the output of the first-order lag filter 24 with the reference voltage 34, and
3 outputs a rotor position detection signal as a pulse signal.
As the reference voltage 34, a neutral point voltage directly obtained by synthesizing a neutral point voltage of the stator armature winding or a terminal voltage after the first-order lag filter 24 is used.

Since the detection signal shifted by 90 degrees is used as a rotor position detection signal of a phase delayed by 120 degrees from the detected phase, the current phase angle is driven at a position of almost 0 degrees. For example, when driven in three phases U, V, and W, the detection signal detected at the terminal voltage of the U phase is W
It is used as a phase rotor position detection signal.

As shown in FIG. 3, the rotor of the brushless DC motor 4 generally has a yoke 12 for forming a magnetic path and a permanent magnet 13 for a field-like magnetic field.
As shown in FIG. 4, a surface magnet type in which the permanent magnet 13 for the field is adhered to the outer peripheral surface of the yoke 12, a yoke 15 in which the rotor forms a magnetic path, and a permanent magnet for the field. An embedded magnet type having a magnet 16 and being formed by inserting the field permanent magnet 16 into a slot provided in the yoke 15 is used. Optimal methods have been proposed for these control methods (Morimoto, Ueno,
Takeda "Wide-range variable speed control of PM motor with embedded magnet structure"
1994 IEEJ Transactions D, Vol. 114, No. 6, p662-
p667).

In general, the torque equation of a brushless DC motor using a permanent magnet represented by dq coordinates is given by the following equation.

T = p · {φmag · iq + (Ld−Lq) · id · iq}. . . (1) where T is the torque, p is the number of pole pairs, φmag is the armature interlinkage flux by the permanent magnet, id and iq are the d-axis component and q-axis component of the armature current, and Ld and Lq are the d-axis components, respectively. Inductance and q-axis inductance are shown.

When the amplitude of the armature current is I and the phase angle (current phase angle) of the armature current with respect to the q axis is θ, i
d and iq are respectively defined as follows.

Id = −I · sin θ. . . (2) iq = I · cos θ. . . (3) Here, I indicates the armature current amplitude, and θ indicates the current phase angle viewed from the q axis.

From the equations (2) and (3), the torque equation (1) can be converted into the following equation.

[0021]

(Equation 1)

Here, L1 is as follows.

L1 = (Lq−Ld) / 2. . . (5) In the case of the surface magnet type rotor shown in FIG. 3, non-salient polarity (Ld = Lq) where the d-axis inductance and the q-axis inductance are equal.
), The second term in the torque equation (1) becomes 0, and the motor torque is generated in proportion to the q-axis current iq. Further, similarly, in the equation (4), the second term is 0, and it can be seen that the maximum torque is obtained by setting the current phase angle θ to 0 degree. That is, from the equations (2) and (3), setting the current phase angle θ to 0 ° and id = 0 is an optimal operation control method for a brushless DC motor having a surface magnet type rotor structure having a non-salient polarity. is there. This is a control method for keeping the d-axis current at 0, which is generally called id = 0 control.

The surface magnet type rotor is, as shown in FIG.
The outer peripheral surface is covered with a non-magnetic sleeve 14,
Prevents permanent magnets from scattering due to high rotation.

On the other hand, in the embedded magnet type rotor, as shown in FIG. 4, a magnetic pole generated by a field permanent magnet 16 is provided on the outer periphery of a yoke 15 formed by laminating many silicon steel plates. There is no fear of scattering or the like, and it is possible to achieve particularly high rotation. This embedded magnet type rotor is
Since the magnetic equivalent air gap in the q-axis direction is smaller than the magnetic equivalent air gap in the d-axis direction, a reverse saliency (Ld <Lq) in which the q-axis inductance is larger than the d-axis inductance is shown.

The torque of the motor thus generated is obtained from the torque equation (1) by the torque by the magnet proportional to the q-axis current iq of the first term and the reluctance torque generated by the reverse saliency of the second term. I have. Therefore, according to the torque equation (4), the current phase angle θ at which the combination of the torque by the magnet and the reluctance torque is maximized (hereinafter, referred to as advance angle θ)
Is an optimal operation control method for a brushless DC motor having an embedded magnet type rotor structure having reverse saliency. This is generally called a maximum torque control, which is a control method that effectively utilizes reluctance torque. FIG.
9 shows the relationship between the advance angle θ and the motor torque T when the maximum torque control is performed.

Further, as a control method of the embedded magnet type, a method has been proposed in which a high-speed rotation range is expanded by equivalent field-weakening control that positively utilizes reluctance torque. In this method, an optimal operation is performed by the maximum torque control up to an operation region in which the maximum applied voltage applied to the motor is equal to the back electromotive voltage of the motor. Normally, in a high-speed rotation region higher than this, operation cannot be performed because the maximum applied voltage is equal to the back electromotive voltage of the motor. However, by further advancing the advance angle θ, the d-axis current can flow positively, and the armature linkage flux of the permanent magnet can be equivalently weakened by the d-axis armature reaction. If this equivalent field-weakening control is controlled so that the motor output becomes a constant output, the operating range of the high-speed rotation range can be expanded.
This is generally called maximum output control.

[0028]

In the conventional rotor position detecting means 5 described above, the first-order lag filter 24
Since the high-frequency component is cut from the terminal voltage of the stator armature winding end 6 and the terminal voltage is shifted by 90 degrees, the cutoff frequency of the first-order lag filter 24 is set to several Hz to several tens Hz or less. ing.

On the other hand, the fundamental wave component of the back electromotive force generated at the stator armature winding end 6 due to the rotation of the rotor is generally varied to about several hundred Hz depending on the motor rotation range. The phase of the rotor position detection signal that has passed through 24 causes a detection delay due to an increase in the rotation frequency of the brushless DC motor 4,
An accurate commutation phase cannot be obtained, for example, a spike voltage is generated in the reflux mode after commutation, and the pulse width increases depending on the load current condition, so that detection proceeds. Particularly, when the motor is driven at a high rotation speed, the operation is performed in a delayed phase, so that the operation range is restricted and the motor efficiency is reduced.

Therefore, it is difficult to perform optimal operation control of the brushless DC motor 4 with the configuration of the conventional rotor position detecting means 5, and particularly in the case of the embedded magnet type rotor structure, the motor torque is reduced as shown in FIG. On the other hand, it is necessary to control the advance angle θ to advance, and there is a problem that the motor performance cannot be fully utilized.

The maximum output control method described above is used to extend the operating range of a brushless DC motor without a position sensor having an embedded magnet rotor structure.

However, a high-resolution encoder is required to precisely control the advance angle θ, which is expensive, and many brushless DC motors without a position sensor are used under special environments. There is a problem that a sensor for detecting a rotor position such as a high-resolution encoder or a Hall element cannot be used.

An object of the present invention is to provide a rotor position detecting means capable of always detecting an accurate rotor position without being affected by changes in the rotation frequency or load of a brushless DC motor, It is an object of the present invention to provide a brushless DC motor drive device capable of easily expanding the operating range of a brushless DC motor having a rotor structure.

[0034]

This and other objects are achieved by the present invention which is defined below as (1) to (12).

(1) A stator in which three-phase armature windings U, V, and W are star-connected, a rotor that forms a magnetic pole pair by a permanent magnet embedded in a yoke that forms a magnetic path, and a plurality of semiconductors A 120-degree conducting inverter having a switching element, rotor position detecting means for detecting a terminal voltage generated at an armature winding end of the stator and generating a signal corresponding to a magnetic pole position of the rotor; A brushless DC motor driving device comprising: a drive control unit that performs speed adjustment by PWM chopper control with the inverter based on a signal from the detection unit; wherein the rotor position detection unit is configured to detect an end of an armature winding of the stator. First line voltage generating means for generating a line voltage Vw-u between the armature windings W-U based on the terminal voltage, and an armature winding UV A second line voltage generating unit for generating a line voltage Vu-v between the first and second lines, a third line voltage generating unit for generating a line voltage Vv-w between the armature windings V and W, First amplifying means for amplifying a signal related to the line voltage Vw-u output from the first line voltage generating means;
The line voltage V output from the second line voltage generating means
second amplification means for amplifying a signal relating to uv, and the line voltage Vv- output from the third line voltage generation means.
third amplification means for amplifying a signal regarding w, first comparison means for comparing a signal regarding the line voltage Vw-u with a signal output from the second amplification means, and the line voltage Vu. Second comparing means for comparing a signal related to −v with a signal output from the third amplifying means, and a signal related to the line voltage Vv−w and a signal output from the first amplifying means. Third comparing means for comparing, wherein the rotor position detecting means is determined by setting an amplification factor of the first, second and third amplifying means,
A brushless DC motor driving device, wherein the rotor position is detected at a current phase angle θ at which the combination of the torque by the magnet of the rotor and the reluctance torque is maximized.

(2) The brushless device according to (1), wherein the rotor position detecting means is configured to detect a predetermined magnetic pole position of the rotor at which the current phase angle θ is advanced by 30 electrical degrees or more. Drive device for DC motor.

(3) The first comparing means outputs a high-level signal when the signal related to the line voltage Vw-u is larger than the signal output from the second amplifying means. The second comparing means outputs a high-level signal when the signal related to the line voltage Vu-v is larger than the signal output from the third amplifying means, and the third comparing means outputs the high-level signal. The signal related to the inter-voltage Vv-w is the first signal.
The brushless DC motor driving device according to the above (1) or (2), which is configured to output a high-level signal when the signal is larger than the signal output from the amplifying means.

(4) The rotor position detecting means includes an applied voltage determining means for determining a motor applied voltage, and the first, second, and third amplifying means based on a signal from the applied voltage determining means. The brushless DC motor drive device according to any one of (1) to (3), further including first, second, and third gain switching means for changing a gain.

(5) The rotor position detecting means includes n (n is an integer of 2 or more) applied voltage determining means for determining a motor applied voltage, and the first position detecting means based on a signal from the applied voltage determining means. The brushless DC according to any one of (1) to (3), further comprising n gain switching means for changing gains of the second and third amplification means.
Motor drive.

(6) The brushless DC motor drive device according to (4) or (5), wherein a hysteresis circuit is added to the applied voltage determination means.

(7) The drive control means includes a chop-on detection means for detecting a signal from the rotor position detection means at the time of chop-on of PWM chopper control, and an open-phase signal from the chop-on detection means. The brushless DC according to any one of the above (1) to (6), further comprising: an open-phase selecting means for selecting the edge in accordance with the timing, and an edge detecting means for detecting a predetermined edge based on a signal from the open-phase selecting means. Motor drive.

(8) When a predetermined magnetic pole position of the rotor is detected by the rotor position detecting means, a commutation signal is output in synchronization with the detection. Brushless D according to any of the above)
Drive device for C motor.

(9) A commutation signal is output after the rotor has been rotated by a predetermined phase shift amount from when a predetermined magnetic pole position of the rotor is detected by the rotor position detection means. A drive device for a brushless DC motor according to any one of (1) to (7).

(10) The brushless D according to (9), further comprising a shift amount setting means for setting the phase shift amount.
Drive device for C motor.

(11) The apparatus for driving a brushless DC motor according to the above (10), wherein the setting of the phase shift amount by the shift amount setting means is changed at least in accordance with the rotation speed of the rotor.

(12) The setting of the phase shift amount by the shift amount setting means is changed according to the rotation speed of the rotor and the motor current.
Drive device for C motor.

[0047]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, a driving apparatus for a brushless DC motor (position sensorless brushless DC motor) according to the present invention will be described in detail based on a preferred embodiment shown in the accompanying drawings.

Embodiment 1 Embodiment 1 of the present invention will be described below with reference to the accompanying drawings.

FIG. 6 is a block diagram showing an example of the configuration of a brushless DC motor driving device according to the present invention.

As shown in the drawing, the brushless DC motor driving device includes a stator 41 in which three-phase armature windings (excitation coils) are star-connected, and a rotor 42 which forms magnetic poles (magnetic pole pairs) by permanent magnets. And a 120-degree conducting voltage-type inverter (hereinafter, referred to as a motor) that adjusts speed by chopper control.
An inverter 43), a rotor position detecting means 44 utilizing a back electromotive force at the time of chop-on, and a drive control means 45 for driving and controlling the inverter 43 by an output (synonymous with a signal) of the rotor position detecting means 44. It is configured.

Each of the phase excitation coils U, V, W of the stator 41 in which the three-phase armature windings are star-connected is connected to the inverter 4
3 is connected to the output side. A DC power supply Ed is applied to the input side of the inverter 43. The P-side of each transistor has Ed + of the DC power supply Ed, and the N-side has Ed- of Ed-.
Is connected. The inverter 43 is connected to the P-side return diodes Da +, Db +, and Dc +.
Side transistor (semiconductor switching element) Ta +,
Tb +, Tc +, and N-side freewheeling diodes Da-,
It is composed of N-side transistors (semiconductor switching elements) Ta-, Tb-, and Tc- to which Db- and Dc- are connected.

The drive control means 45, as shown in FIG.
Two windings are selected from the excitation coils of each phase, and the P-side transistor and the N-side transistor are sequentially turned on in a combined excitation pattern, thereby forming a rotating magnetic field in the stator 41 and rotating the rotor 42. . Further, for either the P-side transistor or the N-side transistor of the excitation pattern, conduction and interruption (hereinafter, conduction is referred to as chop-on and interruption is referred to as chop-off) are alternately repeated by PWM chopper control, and chop-on,
The speed is adjusted by adjusting the input power by varying the duty ratio of chop off. At this time, the terminal voltages Vu, Vv, Vw are obtained for the phase excitation coils U, V, W of the stator 41 with reference to the Ed− side of the DC power supply Ed (the reference is Ed + side of the DC power supply Ed or Ed−). Either side may be used, but in this embodiment, the Ed- side is used as a reference).

Next, these terminal voltage waveforms will be described. FIG. 8 shows the case where the advance angle of the brushless DC motor 40 is 0.
Back electromotive force (back electromotive voltage) of U phase, V phase, W phase
FIG. 3 is a diagram showing a relationship between a, eb, ec and a drive signal.
In this case, in any phase, only two transistors of a predetermined one phase on the P side and a different predetermined one phase on the N side among the transistors are operating. Therefore, the terminal voltage of each phase is applied twice (6 electrical degrees) in one cycle.
There is a period in which both the P-side and N-side transistors do not operate. Hereinafter, this period is referred to as “open period”, and the phase in this state is referred to as “open phase”.

The "advance angle" refers to the phase angle (current phase angle) of the current flowing through the armature winding (excitation coil) of the stator with respect to the q axis in the dq coordinate system.

Here, the terminal voltage at the time of chop-on during the open period will be examined. FIG. 9 shows that Ta +
FIG. 9 is a circuit diagram showing an equivalent circuit when -Tb- is conductive. As shown in the figure, drive signals are input to Ta + and Tb-,
It is assumed that a current i is conducted from Ta + to Tb-, that is, a so-called U-phase and V-phase are conducted and chopped. At this time, the terminal voltage Vw of the W phase which is an open phase can be expressed as follows.
In FIG. 9, L is an inductance, r is a resistance value,
VCE is the collector-emitter saturation voltage of the transistor.

First, the following equation is established from the loop through which the current i flows.

Ed = 2 (L · di / dt + ri) + ea−eb + 2 · VCE. . . (6) L · di / dt + ri = Ed / 2− (ea−eb) / 2−VCE. . . (7) At this time, the terminal voltage Vw of the W phase which is an open phase is represented by the following equation.

Vw = VCE + Ldi / dt + ri-eb + ec. . . (8) By substituting the above equation (7) into the equation (8), the following equation is obtained.

Vw = Ed / 2 + ec− (ea + eb) / 2. . . (9) Here, as shown in FIG. 10A, assuming that the back electromotive force has a completely symmetric waveform, near the point P where the back electromotive force ec of the open phase W phase becomes 0V, Since ea = −eb, the equation (9) is as follows.

Vw = Ed / 2 + ec. . . (10) This relationship is established irrespective of the PWM chopper control method such as the P-side chopper or the N-side chopper.
An open phase terminal voltage that satisfies the equation is obtained.

Further, from the above equation (10), it can be seen that the open-phase terminal voltage Vw changes (sways up and down) in accordance with the back electromotive force ec with reference to a half of the potential of the DC power supply Ed. That is, at the time of chop-on, the time when the open-phase terminal voltage Vw becomes Ed / 2 (the time when ec = 0)
However, this is the detection timing advanced by 30 degrees in electrical angle from the commutation point of the advance angle of 0 degrees.

Next, a method for detecting the point P at which the back electromotive force ec of the open phase terminal voltage Vw becomes 0 V will be described.
In this example, during the open period of the W phase, the other U and V phases are chopped on, that is, the U phase terminal voltage Vu is Vu = Ed−VCE. . . (11), and the V-phase terminal voltage Vv is Vv = 0 + VCE. . . (12) As shown in FIG. 10B, the terminal voltage of the V phase with respect to the W phase is Vv−Vw = −ec− (Ed / 2−VCE). . . (13), and the terminal voltage of the W phase with respect to the U phase is: Vw−Vu = ec− (Ed / 2−VCE). . . (14) In the above equations (13) and (14), ec
= 0 and the point P of the back electromotive force ec is obtained,
/ 2−VCE), Vv−Vw and Vw−Vu match. In general, VCE is a small value.
Since it is included in both equations (4), it does not affect the detection of the P point of ec.

Next, the rotor position detecting means 44 utilizing the above-described counter electromotive force at the time of chop-on will be described. FIG. 11 is a circuit diagram showing a configuration example of the rotor position detection means 44.

As shown in the figure, the terminal voltages Vu, Vv, Vw at the end of each armature winding of the stator 41 are respectively
The first line voltage generating unit 50a, the second line voltage generating unit 50b, and the third line voltage generating unit 50c
It is converted into line voltages Vw-u, Vu-v, Vv-w.

The line voltages are supplied to the first amplifying means 80a, the second amplifying means 80b, and the third amplifying means 8, respectively.
0c, and these amplification means 80a, 80b, 80
c (output) and the line voltage generation means 50
The signals (outputs) from a, 50b, and 50c are output to first comparing means 66a, second comparing means 66b, and third comparing means 6, respectively.
6c. The comparing means 66a, 66b, 66
In step c, these are directly compared with each other to detect the magnetic pole position of the rotor 42 advanced by 30 degrees or more in electrical angle from the commutation point of the advance angle of 0 degree, and to detect a signal corresponding to the magnetic pole position of the rotor 42 (magnetic pole position). Signal) Up, Vp and Wp are generated.

The magnetic pole position signals Up and Vp of the rotor 42
, Wp are input from the rotor position detection means 44 to the drive control means 45, and the drive control means 45 controls the drive of the inverter 43 based on the magnetic pole position signal.

Next, the configuration and operation of each line voltage generating means will be described in detail.

The first line voltage generating means 50a includes a resistor 5
1, 52, 53, and 54, and an amplifier 55.

The U-phase terminal voltage Vu and the W-phase terminal voltage Vw are input (applied) to the first line voltage generating means 50a.

That is, the terminal voltage Vw is input to the + input terminal of the amplifier 55 via the resistor 51, and the terminal voltage Vu
Is input to the-input terminal of the amplifier 55 via the resistor 52, and the terminal voltage of the W phase with respect to the U phase is output.

In the first line voltage generating means 50a, the resistor 53 is connected between the output terminal and the-input terminal of the amplifier 55, and the resistor 54 is connected from the + input terminal of the amplifier 55 to the negative side of the DC power supply Ed. This is a so-called differential amplifier connected to the ground. Here, resistance 51 = resistance 52 = R1,
If the resistance 53 = the resistance 54 = R2, the output (voltage) Vw-u of the first line voltage generation means 50a is as follows: Vw-u = R2 / R1. (Vw-Vu). . . (15a). As a result, a line voltage between the W phase and the U phase amplified at an amplification rate determined by the ratio of R2 / R1 is obtained.

Similarly, the second line voltage generating means 50b
Is composed of resistors 56, 57, 58, 59 and an amplifier 60.

Then, the V-phase terminal voltage Vv and the U-phase terminal voltage Vu are input to the second line voltage generating means 50b.

That is, the terminal voltage Vu is input to the + input terminal of the amplifier 60 via the resistor 56, and the terminal voltage Vv
Is input to the negative input terminal of the amplifier 60 via the resistor 57, and the terminal voltage of the U phase with respect to the V phase is output.

In the second line voltage generating means 50b, the resistor 58 is connected between the output terminal and the − input terminal of the amplifier 60, and the resistor 59 is connected from the + input terminal of the amplifier 60 to the negative side of the DC power supply Ed. This is a so-called differential amplifier connected to the ground. Here, resistance 56 = resistance 57 = R1,
If the resistance 58 = the resistance 59 = R2, the output Vu-v of the second line voltage generating means 50b is as follows: Vu-v = R2 / R1. (Vu-Vv). . . (15b) As a result, a line voltage between the U-phase and the V-phase amplified at the amplification rate determined by the ratio of R2 / R1 is obtained.

Similarly, the third line voltage generating means 50c
Is composed of resistors 61, 62, 63, 64 and an amplifier 65.

The W-phase terminal voltage Vw and the V-phase terminal voltage Vv are input to the third line voltage generating means 50c.

That is, the terminal voltage Vv is input to the + input terminal of the amplifier 65 via the resistor 61, and the terminal voltage Vw
Is input to the negative input terminal of the amplifier 65 via the resistor 62, and a V-phase terminal voltage with respect to the W-phase is output.

In the third line voltage generating means 50c, the resistor 63 is connected between the output terminal and the − input terminal of the amplifier 65, and the resistor 64 is connected from the + input terminal of the amplifier 65 to the negative side of the DC power supply Ed. This is a so-called differential amplifier connected to the ground. Here, resistance 61 = resistance 62 = R1,
If the resistance 63 = the resistance 64 = R2, the output Vv-w of the third line voltage generation means 50c is as follows: Vv-w = R2 / R1２ (Vv-Vw). . . (15c). As a result, a line voltage between the V phase and the W phase amplified at an amplification rate determined by the ratio of R2 / R1 is obtained.

Next, the configuration and operation of each amplifying means will be described in detail.

The output Vw- of the first line voltage generating means 50a
u is input to the first amplifying means 80a. The first amplifying unit 80a includes resistors 81 and 82 and an amplifier 83. That is, in the first amplifying unit 80a, the output side of the first line voltage generating unit 50a is connected to the + input terminal of the amplifier 83, and the resistor 81 is connected from the − input terminal of the amplifier 83 to the negative side of the DC power supply Ed. This is a so-called common-mode amplifier that is connected so as to be grounded and the resistor 82 is connected between the output terminal and the − input terminal of the amplifier 83. here,
Assuming that the resistance 81 = R3 and the resistance 82 = R4, the output Vw-u (gain) of the first amplifying means 80a is Vw-u (gain) = (1 + R4 / R3) .Vw-u. . . (16a). As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + R4 / R3), and the line voltage satisfying Vw-u <Vw-u (gain) is obtained.

Similarly, the second line voltage generating means 50
The output Vu-v of b is input to the second amplifying means 80b. The second amplifying unit 80b includes resistors 84 and 85 and an amplifier 86. That is, in the second amplifying unit 80b, the output side of the second line voltage generating unit 50b is connected to the + input terminal of the amplifier 86, and the resistor 84 is connected from the − input terminal of the amplifier 86 to the negative side of the DC power supply Ed. The resistor 85 is connected to the ground, and the resistor 85 is connected to the output terminal of the
This is a so-called common-mode amplifier connected between the input terminal. Here, assuming that the resistance 84 = R3 and the resistance 85 = R4, the output Vu-v (gain) of the second amplifying means 80b is Vu-v (gain) = (1 + R4 / R3) .Vu-v. . . (16b) As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + R4 / R3), and the line voltage satisfying Vu-v <Vu-v (gain) is obtained.

Similarly, the third line voltage generating means 50
The output Vv-w of c is input to the third amplifying means 80c. The third amplifying unit 80c includes resistors 87 and 88 and an amplifier 89. That is, in the third amplifying unit 80c, the output side of the third line voltage generating unit 50c is connected to the + input terminal of the amplifier 89, and the resistor 87 is connected from the − input terminal of the amplifier 89 to the negative side of the DC power supply Ed. The resistor 88 is connected to the ground, and the resistor 88 is connected to the output terminal of the amplifier 89-
This is a so-called common-mode amplifier connected between the input terminal. Here, assuming that the resistance 87 = R3 and the resistance 88 = R4, the output Vv-w (gain) of the third amplifying means 80c is Vv-w (gain) = (1 + R4 / R3) · Vv-w. . . (16c). As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + R4 / R3), and the line voltage satisfying Vv-w <Vv-w (gain) is obtained.

Next, the operation of each comparing means will be described in detail.

In the first comparing means 66a, the output Vu-v (gain) of the second amplifying means 80b and the output Vw-u of the first line voltage generating means 50a are compared, and Vu-v (gain) is compared. ) ≧ Vw-
At the time of u, a low voltage (low level voltage) is output,
When Vu-v (gain) <Vw-u, a High voltage (high-level voltage) is output, thereby generating a pulse signal (magnetic pole position signal) Up.

Similarly, in the second comparing means 66b,
The output Vv-w (gain) of the third amplifying means 80c is compared with the output Vu-v of the second line voltage generating means 50b.
When (gain) ≧ Vu-v, a low voltage is output and Vv-w (g
ain) A high voltage is output when <Vu-v, thereby generating a pulse signal (magnetic pole position signal) Vp.

Similarly, in the third comparing means 66c,
The output Vw-u (gain) of the first amplifying means 80a is compared with the output Vv-w of the third line voltage generating means 50c.
When (gain) ≧ Vv-w, a low voltage is output and Vw-u (g
ain) When <Vv-w, a High voltage is output, thereby generating a pulse signal (magnetic pole position signal) Wp.

The comparing means 66a, 66b, 66c
It is preferable to use a comparator such as a comparator (hereinafter, the comparing means is also referred to as a comparator).

FIG. 12 shows the first comparison means 66a.
The output Vw-u of the first line voltage generation means 50a and (1+
R4 / R3) The second amplifying means 80 amplified with the amplification factor
FIG. 9 is a diagram showing the detection timing of rotor position detection when comparing the output with the output Vu-v (gain) of FIG.

As shown in the figure, the period from point H to point J (electrical angle 60 degrees) is the open period. A point J at which Vu-v (gain) shows a peak indicates a lead angle of 0 degree. Assuming that the direction of point H is the advance angle θ based on this point J, the advance angle θ determined by the intersection of Vw-u and Vu-v (gain) can be expressed by the following equation.

Θ = {A / (1 + A)} · 60. . . (17) Here, A in the expression (17) indicates an amplification factor. The above (1
It can be seen from equation (7) that the advance angle θ is determined by the amplification factor A. For example, when A = 1, it is detected at a lead angle θ = 30 degrees. If A = 2, the lead angle θ = 40 degrees,
The rotor position detection phase can be easily advanced (to an electrical angle of 30 degrees or more).

In this rotor position detection, the advance angle θ
Is preferably set to advance 30 degrees or more, more preferably to advance 40 degrees or more, and even more preferably to advance 50 degrees or more.

By advancing the rotor position detection phase as described above, in particular, the operating range of the motor is widened, thereby improving the degree of freedom in system design using, for example, a brushless DC motor driving device.

FIG. 13 is a timing chart showing signal waveforms at various parts in the circuit diagram shown in FIG. In addition,
In FIG. 13, Vw-u (gain), Vu-v (gain), Vv-w (gain)
Indicates a case where the amplification factor A is about 4 times.

As shown in FIG.
4, the magnetic pole position signals Up, Vp, and Wp are set so that the advance angle θ advances by 30 degrees or more.

By shifting the lead angle θ in accordance with the characteristics of the motor on the basis of the signals Up, Vp, Wp advanced by 30 degrees or more (for example, the timing of the commutation is set to a predetermined value). By delaying to the advance angle θ), optimal operation control becomes possible.

The method of shifting is not particularly limited.
For example, using a detection signal as a trigger, a method using a counter that counts reference pulses, or using a microcomputer, store the desired shift amount in a storage device such as a ROM, and read and add the shift amount corresponding to the instantaneous rotation speed. Any method may be used.

Further, the shift amount may be varied to an optimum value for the motor characteristics by comparing with only one of the rotation speed, only the motor current value, the rotation speed and the motor current value, or the like. For example,
The shift amount can be varied according to the motor current value so as to satisfy the relationship of FIG. In any case, by using the rotor position detecting means 44 of the present invention, the detection signal can be set so as to advance exactly 30 degrees or more, so that the advance angle setting range is widened and contributes to the expansion of the operation range.

(Embodiment 2) Embodiment 2 of the present invention will be described below with reference to the accompanying drawings.

FIG. 14 is a circuit diagram showing a main part of another configuration example of the rotor position detecting means of the brushless DC motor driving device according to the present invention.

As shown in the figure, the rotor position detecting means 44 is the same as the rotor position detecting means 44 of the first embodiment.
First amplifying means 80a, second amplifying means 80b,
Are connected to a first gain switching means 90a, a second gain switching means 90b, and a third gain switching means 90c, respectively. Each of the gain switching means 90a, 90b, and 90c performs the switching at a predetermined timing based on the output L1 of the applied voltage determining means 101 for determining the motor applied voltage.

Next, the operation of the applied voltage determining means 101 for determining the motor applied voltage will be described in detail. FIG.
FIG. 3 is a circuit diagram showing a configuration example of an applied voltage determination unit 101.

Since the applied voltage of the motor is adjusted by changing the duty ratio of chop-on and chop-off of the transistor by PWM chopper control, in this embodiment, it is determined by the duty ratio of chop-on and chop-off. , And determine the voltage applied to the motor.

As shown in FIG. 15, the triangular wave generation circuit 20
4 outputs a chopping frequency triangular wave or a sawtooth wave. Then, a command voltage V0 for determining a desired duty ratio and an output of the triangular wave generating circuit 204 are input to a comparator 205, and the comparator 205 compares the command voltage V0 with an output of the triangular wave generating circuit 204. Then, a rectangular wave corresponding to a desired duty ratio is output.

In this embodiment, the command voltage V0 for determining the desired duty ratio is also input to the comparator 208. Then, the comparator 208 compares the command voltage V0 with a reference voltage V1 for setting the switching timing of each of the gain switching means 90a, 90b, 90c, and Hig.
The output signal L1 of the h voltage or the Low voltage is output.

The method for determining the desired duty ratio is not particularly limited. In the present invention, in addition to the above-described method,
For example, a method using a reference oscillator and a timer or the like inside the microcomputer may be adopted.

Next, the configuration and operation of each gain switching means will be described in detail.

As shown in FIG. 14, the first gain switching means 90a has a resistor 91 and an analog switch 92 connected in series, and the second gain switching means 90b has a resistor 93 and an analog switch 94 connected in series. And the third
The gain switching means 90c has a resistor 95 and an analog switch 96 connected in series. in this case,
The analog switches 92, 94, and 96 are simultaneously turned on and off by the output signal L1 of the applied voltage determining means 101 for determining the motor applied voltage.
The resistances 91, 93,
95 conducts and non-conducts.

As the analog switches 92, 94, and 96, it is preferable to use a switching element such as a MOSFET having a low on-resistance or a photo MOS switch as shown in FIG.

The photo MOS switch can directly switch the MOSFET 207 electrically insulated by the light of the LED 206, and can directly trigger the LED 202 from a port of the microcomputer. Therefore, the applied voltage determining means 101 is constituted by the microcomputer. For example, it is convenient in an application in which the ground line is different from the rotor position detecting means 44.

As shown in FIG. 14, the first amplifying means 80
“a” is composed of resistors 81 and 82 and an amplifier 83. In this case, the output side of the first line voltage generating means 50a is connected to the + input terminal of the amplifier 83, and the resistor 81 is connected to the negative side of the DC power supply Ed from the-input terminal of the amplifier 83, A resistor 82 is connected between the output terminal and the − input terminal of the amplifier 83, and a resistor 9 connected in series is connected.
1 and an analog switch 92 are connected in parallel with the resistor 82 between the output terminal and the − input terminal of the amplifier 83, forming a so-called common-mode amplifier. Here, resistance 81 = R
3, the resistance 82 = R4 and the resistance 91 = R5, the output Vw-u (gain) of the first amplifying means 80a is Vw-u (gain) = (1 + Rz / R3) .Vw-u. . . (18a) Rz = (R4.R5) / (R4 + R5). . . (19) As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + Rz / R3), and the line voltage satisfying Vw-u <Vw-u (gain) is obtained. Therefore, by switching the resistance 91 = R5 between conduction and non-conduction by the analog switch 92, the combined resistance Rz of the equation (19) is varied, and (1 + Rz)
/ R3) can be changed.

Similarly, the second amplifying means 80b includes resistors 84 and 85 and an amplifier 86. In this case, the output side of the second line voltage generating means 50b is connected to the + input terminal of the amplifier 86, and the resistor 84 is connected to the amplifier 8
6, the negative input terminal is connected to the negative side of the DC power supply Ed, the resistor 85 is connected between the output terminal of the amplifier 86 and the negative input terminal, and the resistor 93 and the analog switch 94 connected in series are connected. , A so-called common-mode amplifier connected between the output terminal and the − input terminal of the amplifier 86 in parallel with the resistor 85. Here, resistance 84 = R3, resistance 8
Assuming that 5 = R4 and the resistance 93 = R5, the second amplifying means 8
0b is Vu-v (gain) = (1 + Rz / R3) .Vu-v. . . (18b) As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + Rz / R3) and Vu-v <Vu-v (gain). Therefore, by switching the resistance 93 = R5 between conducting and non-conducting by the analog switch 94, the combined resistance Rz of the equation (19) is varied, and (1 + Rz)
/ R3) can be changed.

Similarly, the third amplifying means 80c is composed of resistors 87 and 88 and an amplifier 89. In this case, the output side of the third line voltage generating means 50c is connected to the + input terminal of the amplifier 89, and the resistor 87 is connected to the amplifier 8
9 is connected to the negative side of the DC power supply Ed so as to be grounded, a resistor 88 is connected between the output terminal and the − input terminal of the amplifier 89, and a resistor 95 and an analog switch 96 connected in series are connected. , A so-called common-mode amplifier connected between the output terminal and the − input terminal of the amplifier 89 in parallel with the resistor 88. Here, resistance 87 = R3, resistance 8
Assuming that 8 = R4 and the resistor 95 = R5, the third amplifying means 8
0c output Vv-w (gain) is: Vv-w (gain) = (1 + Rz / R3) .Vv-w. . . (18c). As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + Rz / R3), and the line voltage satisfying Vv-w <Vv-w (gain) is obtained. Therefore, by switching the resistance 93 = R5 between conducting and non-conducting by the analog switch 94, the combined resistance Rz of the equation (19) is varied, and (1 + Rz)
/ R3) can be changed.

The first, second, and third gain switching means 90a, 90b, and 90c are connected to the first, second, and third amplification means 80a, 80b, and 80c, respectively.
The gain can be similarly changed by connecting in parallel to 1, 84 and 87.

In this case, the output V of the first amplifying means 80a is
wu (gain) is: Vw-u (gain) = (1 + R4 / Rz1) · Vw-u. . . (20) Rz1 = (R3 · R5) / (R3 + R5). . . (21) As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + R4 / Rz1), and the line voltage satisfying Vw-u <Vw-u (gain) is obtained. Therefore, by switching the resistance 91 = R5 between conducting and non-conducting by the analog switch 92, the combined resistance Rz1 of the equation (21) is varied, and (1 + R4
/ Rz1) can be changed.

FIG. 17 shows a change in the motor nakedness characteristic when the gain was actually varied in the configuration of the present invention using the brushless DC motor having the embedded magnet rotor structure having the characteristics shown in FIG. It is a graph.

As shown in the figure, the nakedness characteristic S1 has a lead angle of 0
Degrees, S2 is a lead angle of 20 degrees, S3 is a lead angle of 30 degrees, S4
Is the result of measuring the advance angle of 40 degrees. However, the bare characteristics S1 and S2 having an advance angle of less than 30 degrees are obtained by shifting the output of the rotor position detecting means 44 of the present invention.

The bare characteristic S1 is obtained by setting the output of the rotor position detecting means 44 to an advance angle of 30 degrees and adding a fixed shift amount of -30 degrees (delayed by 30 degrees). The bare characteristic S2 is obtained by setting the output of the rotor position detecting means 44 to an advance angle of 30 degrees and adding a fixed shift amount of -10 degrees.

In the experiment, the output L1 of the applied voltage discriminating means 101 was manually operated, and the bare characteristics S3 and S4
Are measured individually.

From the results shown in FIG. 17, the number of revolutions has increased by 1500 rpm from the maximum value of 8000 rpm at the advance of 0 degree to 9500 rpm at the advance of 40 degrees. Also, when the torque is compared at 7800 rpm, when the maximum torque at the advance of 0 degree is 1 (0.4 Nm), at the advance of 40 degrees, the torque of about 5 times (1.98 Nm) is obtained. It has gained.

FIG. 18 is a graph showing the motor efficiency when the advance angle θ is changed at 6000 rpm, and FIG.
It is a graph which shows the inverter efficiency at the time of changing advance angle (theta) at 6000 rpm.

In this case, E1 in FIG. 18 is a motor load torque of 1 N · m, and E2 is a motor load torque of 2 N · m.
And E3 is the motor load torque 3N · m,
This is the motor efficiency when the advance angle is changed from 20 degrees to 40 degrees.

In FIG. 19, F1 is a motor load torque of 1 N · m, F2 is a motor load torque of 2 N · m,
F3 is the inverter efficiency when the lead angle is varied from 20 degrees to 40 degrees at a motor load torque of 3 N · m.

From these results, it was found that the lead angle was changed from 20 degrees to 40 degrees.
It can be seen that the motor efficiency at the time of moving forward is reduced by about 1.5%, and the inverter efficiency is reduced by about 0.5%, so that the total efficiency is reduced by about 2%.

As can be seen from the above description, in normal operation, in consideration of the efficiency and the like, the advance angle θ at which the motor current per output torque is reduced at the rated point, that is, the output torque of the motor is maximized at the rated torque. It is preferable to operate at the advance angle θ, but even if efficiency is somewhat sacrificed during motor operation,
When the rotation speed and the torque are needed at one point, the operation range can be expanded by switching the advance angle to the advance side by the set motor applied voltage.

In this embodiment, the gain switching means is configured by using an analog switch and a resistor. In addition, for example, an FET is used instead of the analog switch and the resistor, and the FET is activated. A method of operating in a region and continuously varying the resistance value can be adopted.

Although the gain is switched by the voltage applied to the motor, the gain can be switched also by the motor speed and the motor current.

(Embodiment 3) Embodiment 3 of the present invention will be described below with reference to the accompanying drawings.

FIG. 20 is a circuit diagram showing another configuration example of the amplifying means (first amplifying means) 80a for one phase in the present invention. Since the second and third amplifying units 80b and 80c are configured in the same manner as the first amplifying unit 80a, description thereof will be omitted.

As shown in the figure, the rotor position detecting means 44 in the present embodiment comprises n resistors in parallel with the resistor 82 of the first amplifying means 80a of the rotor position detecting means 44 of the first embodiment. (N is an integer of 2 or more) gain switching means 901, 902. . . 90n are connected. These gain switching means 901, 902. . . 9
0n, the switching is performed by n individually applied voltage discriminating means 101, 102,. . . 10n outputs L1, L2. . . This is performed at a predetermined timing based on Ln.

Next, n applied voltage discriminating means 101, 102. . . 10n will be described. FIG. 21 is a circuit diagram showing a configuration example of the applied voltage determining means.

As shown in the figure, a command voltage V0 for determining a desired duty ratio is applied to n comparators 301 and 30.
2. . . 30n. And n comparators 3
01, 302. . . 30n, the command voltage V0
And n gain switching means 901, 902. . . 9
N reference voltages V for setting the switching timing of 0n
1, V2. . . Vn is compared with the output signals L1, L2... Of the High voltage or the Low voltage. . . Ln is output.

As shown in FIG. 20, the gain switching means 901 comprises a resistor 911 and an analog switch 921 connected in series. Similarly, other gain switching means 902. . 90n are also the resistors 91
2. . 91n and analog switches 922. . 92n are connected in series. Analog switch 92
1. . 92n are applied voltage discriminating means 101. . 10n output signals L1. .
Ln, which is turned on and off by the resistance 911. .
91n conducts and non-conducts. Here, the resistance 81 = R3,
Resistor 82 = R4, n resistors 911. . 91n = R51
1. . Assuming that R51n, the output Vw-u (gain) of the first amplifying means 80a is Vw-u (gain) = (1 + Rz2 / R3) .Vw-u. . . (22) Rz2 = 1 / (1 / R4 + 1 / R511 +... + 1 / R51n). . . (23) As a result, the line voltage is amplified at the amplification rate determined by the ratio of (1 + Rz2 / R3), and the line voltage satisfying Vw-u <Vw-u (gain) is obtained. Therefore, n resistors 911. . 91n =
R511. . R51n are connected to the analog switches 921. .
By switching between conduction and non-conduction by 92n, the combined resistance Rz2 of the equation (23) is varied to obtain (1 + Rz2 / R
The amplification factor determined by the ratio 3) can be continuously changed.

Further, the n gain switching means 90
1. . The gain can be similarly changed by connecting 90n in parallel with the resistor 81 of the first amplifying means 80a.

In this case, the output V of the first amplifying means 80a is
wu (gain) is: Vw-u (gain) = (1 + R4 / Rz3) · Vw-u. . . (24) Rz3 = 1 / (1 / R3 + 1 / R511 +... + 1 / R5n). . . (25) As a result, the line voltage is amplified at an amplification rate determined by the ratio of (1 + R4 / Rz3), and a line voltage satisfying Vw-u <Vw-u (gain) is obtained. Therefore, n resistors 911. . 91n =
R511. . R51n are connected to the analog switches 921. .
By switching between conducting and non-conducting by 92n, the combined resistance Rz3 of the equation (25) is varied, and (1 + R4 / R
The amplification rate determined by the ratio of z3) can be changed. In addition,
This may be used in combination with the configuration of FIG. This makes it possible to gradually increase the operation range during motor operation.

(Embodiment 4) Embodiment 4 of the present invention will be described below with reference to the accompanying drawings.

FIG. 22 is a circuit diagram showing another example of the configuration of the applied voltage discriminating means of the present invention and the circuits in the vicinity thereof.

As shown in the figure, in this embodiment, a hysteresis circuit 110 is added to the applied voltage discriminating means 101 of the second embodiment.

This hysteresis circuit 110 mainly includes a comparator 402 and a flip-flop 501.

Command voltage V0 for determining a desired duty ratio is input to the + input terminal of comparator 402, and reference voltage VH1 for setting the hysteresis level is input to the-input terminal.
Is entered. In the comparator 402, the command voltage V
0 is compared with the reference voltage VH1, and when VH1 <V0, Hi
gh voltage is output, and a low voltage is output when VH1 ≧ V0.

The flip-flop 501 is a D-type flip-flop with preset and clear, and has independent data (D), preset (PR), clear (CL), clock (CLK) input, and complementary outputs Q and NOT.
I have a Q.

Data (D) and clock (CLK) are represented by G
It is input to ND and is fixed at a low voltage. The preset (PR) includes the comparator 3 of the applied voltage determination unit 101.
01, the output C2 of the comparator 402 is input to clear (CL), and the output NOTQ is output to the first, second, and third gain switching means 90a, 90 at the next stage.
b, 90c. Each voltage value V0, V1, VH1
Are set so that V0 (maximum value)>V1> VH1.

FIG. 23 is a diagram showing the relationship between the inputs and outputs of the comparators 301 and 401 and the flip-flop 501.

As shown in the figure, when the command voltage V0 is Low
When rising from the level and lower than V1 and VH1 (State 1)
, The output of the comparator 301 becomes a high voltage (H), the output of the comparator 402 becomes a low voltage (L), and the output NOTQ of the flip-flop 501 becomes a high voltage.

Next, when the command voltage V0 becomes higher than VH1 (state 2), the output of the comparator 301 becomes High voltage, the output of the comparator 402 becomes High voltage, and the output NOTQ of the flip-flop 501 becomes High. Voltage.

Here, the case where NOTQ is High voltage is defined as the first gain (gain before switching),
The case of voltage is defined as a second gain (switched gain).

The command voltage V0 further rises, and V1, VH1
When the voltage becomes higher (state 3), the output of the comparator 301 becomes Low voltage, and the output of the comparator 402 becomes High.
Voltage, and the output NOTQ of the flip-flop 501 becomes a low voltage. At this time, the gain is switched to the second gain by the gain switching means of the next stage, and the operation is performed with the second gain up to the maximum value of V0.

Next, when the command voltage V0 rises from the High level and is lower than V1 (state 4), the comparator 30
1 becomes a high voltage, the output of the comparator 402 becomes a high voltage, and the output N of the flip-flop 501 becomes
OTQ is latched at a low voltage.

The command voltage V0 further rises, and V1, VH1
When the voltage becomes lower (state 5), the output of the comparator 301 becomes High voltage, and the output of the comparator 402 becomes Low.
And the output NOTQ of the flip-flop 501 becomes a High voltage. At this time, the gain is switched to the first gain by the gain switching means of the next stage, and the operation is performed with the first gain up to the minimum value of V0.

FIG. 24 is a graph showing the TN characteristics of this embodiment. In this graph, assuming that the maximum value of the command voltage V0 is a duty ratio of 100%, the reference voltage V1 = 90% for setting the switching timing of the first gain switching means 90a, and the reference voltage VH1 for setting the hysteresis level. = 80%, and 30
The case where the advance angle is advanced from 40 degrees to 40 degrees is shown.

In FIG. 24, A1 is a duty ratio of 90%,
A2 is a motor nakedness characteristic at a lead angle of 30 degrees, A2 is a duty ratio 90%, a motor nakedness characteristic at a lead angle of 40 degrees, and A3 is
A4 is a duty ratio of 80%, a motor nakedness at a lead angle of 40 degrees, and A5 is a duty ratio of 80%, a lead angle of 30.
Motor nakedness in degrees.

As shown in the figure, when the duty ratio is increased, the motor is operated at an advance angle of 30 degrees until the duty ratio changes from 0% to 90%. That is, when the duty ratio is less than 90%, the motor
The operation is performed in a region inside (at the origin of the graph).

When the duty ratio becomes 90%, the advance angle becomes 3
The angle is switched from 0 degrees to 40 degrees, whereby the rotation speed of the motor increases and shifts to the A2 naked characteristic while the duty ratio remains 90%. That is, when the duty ratio is 90%, the motor is operated on A2.

When the duty ratio is 90% to 100%
Up to%, the motor is operated at 40 degrees advance. In this case, the motor is operated in the region between A2 and A3.

To decrease the duty ratio,
The hysteresis circuit 110 changes the timing of switching the advance angle from 90% at the time of rising to 80%.

Therefore, when decreasing the duty ratio, the motor is operated at the advance angle of 40 degrees until the duty ratio changes from 100% to 80%. That is, when the duty ratio exceeds 80%, the motor is operated in a region between A3 and A4.

When the duty ratio becomes 80%, the advance angle becomes 4
The angle is switched from 0 degrees to 30 degrees, whereby the rotation speed of the motor decreases and shifts to the A5 naked characteristic while the duty ratio remains at 80%. That is, when the duty ratio is 80%, the motor is operated on A5.

Then, when the duty ratio is from 80% to 0%, the motor is operated at an advance angle of 30 degrees. In this case, the motor is operated in a region inside A5 (on the origin side of the graph).

As described above, in the present embodiment, the command voltage V0
By giving hysteresis to the output timing of the applied voltage discriminating means when the voltage rises and falls, the operating range can be expanded and the expanded region can be operated arbitrarily.

In the present embodiment, the hysteresis circuit 110 is added to the applied voltage discriminating means 101 of the second embodiment. However, the present invention is not limited to the configuration shown in FIG. Applied voltage discriminating means 101. . 1
For example, n hysteresis circuits may be added to 0n.

(Embodiment 5) Embodiment 5 of the present invention will be described below with reference to the accompanying drawings.

FIG. 25 is a block diagram showing a configuration example of the drive control means 45 of the brushless DC motor drive device of the present invention.

As shown in the figure, the drive control means 45
Inverter drive circuit 8 for driving inverter 43
An output pattern generating circuit 9 for outputting a drive signal pattern of the brushless DC motor 40 in accordance with the detection timing of the rotor position detecting means 44, and setting a chop-on / chop-off ratio of the PWM chopper control in accordance with the rotational speed command 11. (The ratio of chop-on and chop-off is variable) A PWM control circuit (PWM
M generating circuit) 10, chop-on detecting means 71, open phase selecting means 72, and edge detecting means 73.

In this embodiment, the outputs (magnetic pole position signals) Up, Vp and Wp of the rotor position detecting means 44 are respectively
It is detected by the chop-on detecting means 71 at the time of the chop-on of the PWM chopper control.

The open phase of the output of the chop-on detecting means 71 is selected by the open phase selecting means 72 in accordance with the current drive signal pattern output of the output pattern generating circuit 9.

The edge detecting means 73 holds (detects) an edge at any desired point in the output (open phase signal) P2 of the open phase selecting means 72. The outputs (detection signals) P4 and PWM of the edge detecting means 73
The output (PWM signal) P1 of the control circuit 10 is
It is input to the output pattern generation circuit 9.

Then, the next drive signal pattern is input from the output pattern generation circuit 9 to the inverter drive circuit 8, and based on the drive signal pattern, the inverter drive circuit 8 causes each of the transistors Ta +, Tb + of the inverter 43 to operate. , Tc +, Ta-, Tb-, and Tc- are respectively driven.

FIG. 26 is a circuit diagram showing a configuration example of the chop-on detecting means 71, the open phase selecting means 72 and the edge detecting means 73, and FIG. 27 is a timing chart showing the operation of the circuit shown in FIG.

As shown in these figures, the chop-on detecting means 71 outputs the output Up of the rotor position detecting means 44.
, Vp, Wp are isolated from the DC power supply Ed by the detection signal Up
Insulation couplers 70a, 7 for converting to s, Vps, Wps
0b, 70c and three AND circuits 101, 102, 1
03 AND gate 100 and three EX-ORs
And an EX-OR gate 110 composed of circuits 111, 112 and 113. The detection signals Ups, Vps, Wps and the PWM signal P1 are input to the AND gate 100. From the AND gate 100, the PWM signal P1
Data signals (pulse signals) Ups +, Vps + and Wps + are output as signal components during the chop-on period. The logical formula is as shown in the following formulas (26a), (26b) and (26c).

Ups + = Ups · P1. . . (26a) Vps + = Vps · P1. . . (26b) Wps + = Wps · P1. . . (26c) The EX-OR gate 110 has the data signal Ups +,
Vps + and Wps + and the PWM signal P1 are input.
The EX-OR gate 110 outputs the data signal Ups-, as a signal component in the chop-on period delayed by 180 degrees in electrical angle with respect to the signal components Ups +, Vps +, and Wps + in the chop-on period of the PWM signal P1. Vps-
, Wps- are output. The logical expression is as follows (27
a), (27b) and (27c).

[0171]

(Equation 2)

Here, NOT (Ups +), NOT (Vps +)
+), NOT (Wps +), NOT (P1) represent the negation of Ups +, Vps +, Wps +, P1, respectively.

The data signals Ups +, Vps +, Wps +
, Ups-, Vps-, and Wps- are signals corresponding to the back electromotive voltage appearing during the chop-on period and the spike voltage.

The open phase selecting means 72 is provided with the data selector 12
1, an AND gate 200 including three AND circuits 201, 202, and 203, and an OR gate 210.

Data signals Ups +, Vps +, Wps +, Ups
, Vps− and Wps− are input to the data selector 121 as data signals of the data selector 121.

On the other hand, in the open-phase selecting means 72, the drive signal pattern T
ad +, Tbd +, Tcd +, Tad-, Tbd-, T
Open phase selection signals S1, S2, S3 are generated based on cd-.

In this case, the drive signal patterns Tad +, Tbd +, Tcd + output from the output pattern generation circuit 9
, Tad-, Tbd-, Tcd- are AND gate 2
00, and the AND gate 200 generates signals K1, K2, and K3. These signals K1, K
2, K3 are input to the OR gate 210, and the OR gate 210 generates the open phase selection signal S1. The drive signal pattern Tb is used as the open phase selection signal S2.
d + is used, and Tcd + is used as the open phase selection signal S3. The logical expressions of K1, K2, K3 and S1 are as shown in Expressions (28a), (28b), (28c) and (29), respectively.

K1 = Tad + .Tcd-. . . (28a) K2 = Tbd +. Tad-. . . (28b) K3 = Tcd + .Tbd-. . . (28c) S1 = K1 + K2 + K3. . . (29) These open phase selection signals S1, S2, S3 are respectively
Input to the data selector 121, and the data selector 121 selects only the open periods of the data signals Ups +, Vps +, Wps +, Ups-, Vps-, and Wps- according to the truth table shown in FIG. The open phase signal P2 is output. This open phase signal P2 is input to the edge detecting means 73.

The edge detecting means 73 mainly includes a mono-multi 311 and a mono-multi 312.

The open phase signal P2 is first input to the mono multi 311. The mono-multi 311 is triggered in synchronization with the edge (rising edge) of the first pulse of the open-phase signal P2, and is retriggered in synchronization with the edge of the sequentially input pulse. And a pulse signal P3 having a pulse width determined by the time constant t1 of the external capacitor C1.

The output of the mono multi 311, that is,
The pulse signal P3 is input to the mono multi 312. The mono-multi 312 is triggered in synchronization with the edge of the first pulse of the pulse signal P3, whereby the pulse signal having the pulse width determined by the time constant t2 of the external resistor R2 and the external capacitor C2 (detection signal ) Output P4.

Here, the open phase signal P2 is detected as a signal synchronized with the ON period of the chopping cycle of the PWM signal P1. The open phase signal P2 includes a detected edge and an edge due to a spike voltage. This spike voltage also appears in synchronization with the ON period of the chopping cycle,
With the load operation of the brushless DC motor, as the motor current increases, the time width increases within the ON period of the chopping cycle, and the pulse edge also increases.

Since the generation timing of the spike voltage coincides with the commutation timing, there is a time difference between the adjacent detection edge and the edge due to the spike voltage that is equal to or shorter than the chopping cycle in the generation time. Therefore, the mono multi 31
1, the waveform shaping is performed so that the open-phase signal P2 has a pulse width of the time constant t1. The setting condition of the time constant t1 of the mono-multi 311 is preferably as shown in the following equation (30).

1.5T <t1 (= C1 · R1) <2.0T. . . (30) Here, T is the chopping frequency fc of the PWM signal P1.
(T = 1 / fc)
).

As a result, a pulse signal P3 is obtained. The rising edge of the pulse signal P3 corresponds to the rotor position detection signal in which the advance angle θ has advanced by 30 degrees or more.

Here, assuming that a commutation in which the detection position where the advance angle θ is advanced by 30 degrees is shifted by 30 degrees is performed, a period from the rising edge of the pulse signal P3 to the next falling edge (pulse signal P3 Is determined by the detection edge advanced by 30 degrees and the effect of the spike voltage appearing in the reflux mode after commutation.

The mono multi 312 detects only the rising edge of the pulse signal P3 and outputs a detection signal P4 having a pulse width determined by the time constant t2.

As a result, only the rising edge of the first pulse of the open-phase signal P2, that is, only the first detection edge is detected, and the operation of the brushless DC motor always proceeds exactly 30 degrees or more regardless of the presence or absence of a load. A rotor position detection signal, that is, a detection signal P4 is obtained.

As described above, in the present embodiment, even if the output of the rotor position detecting means 44 includes the spike voltage appearing in the recirculation mode after commutation in addition to the signal of the rotor magnetic pole position to be detected, The position of the magnetic pole can be properly and reliably detected, whereby a normal operation can always be performed.

In the present embodiment, the open phase is selected by the logic element, but the present invention is not limited to the illustrated configuration. For example, when a microcomputer or the like is used, the open phase is uniquely determined by the output drive signal pattern. Therefore, the open phase selection signal is output from the microcomputer simultaneously with the output of the drive pattern, and based on the open phase selection signal, ,
The data selector 121 outputs data signals Ups +, Vps +, W
Only the open periods of ps +, Ups-, Vps-, and Wps- may be selected to output the open-phase signal P2.

Further, the data signal may be directly taken into the microcomputer and processed entirely in the microcomputer.

Embodiment 6 Hereinafter, Embodiment 6 of the present invention will be described with reference to the accompanying drawings.

FIG. 29 is a block diagram showing another example of the configuration of the brushless DC motor driving device according to the present invention. Note that description of common points with the above-described fifth embodiment will be omitted, and differences will be described.

As shown in the figure, the brushless DC motor driving apparatus of the present embodiment uses a brushless DC motor based on a detection signal P4 generated based on the magnetic pole position signals Up, Vp, Wp from the rotor position detection means 44. By detecting the number of rotations of the motor 40 and setting an appropriate phase shift amount based on the number of rotations, a wide range of operation can be performed with high efficiency.

In this embodiment, a control section (shift amount setting means) 400 is connected to the output side of the edge detecting means 73,
The output side of the control section 400 is connected to the output pattern generation circuit 9. In this case, the detection signal P4 is input to the control unit 400 from the edge detection unit 73, and the control unit 400
Based on the detection signal P4, the output pattern generation circuit 9
Controls the operation of

The control unit 400 includes, for example, an A / D converter for converting each input signal into a digital signal, an I / O port, a timer counter (timer), a CPU, and a microcontroller including a memory such as a ROM or an EEPROM. An appropriate phase shift amount is set based on a detection signal P4 and the like from the edge detection means 73.

In this case, a commutation signal is output at a timing determined by the phase shift amount finally corrected from the control unit 400, and the output pattern generation circuit 9 outputs the commutation signal based on the commutation signal. Drive signal pattern Ta
d +, Tbd +, Tcd +, Tad-, Tbd-, Tc
d- is output.

Specifically, when the brushless DC motor having the surface magnet type rotor structure is driven (id = 0 control), for example, the detection point (detection time point) of the rotor position detection means 44 is set to 30 electrical degrees. Control unit 4
00 shifts (delays) 30 degrees. Thus, driving (commutation) at the position where the advance angle is always 0 degrees becomes possible.

In driving the brushless DC motor having the magnet-embedded rotor structure, when the operating range in the high rotation region is expanded (field weakening control), for example, the following may be performed.

The detection point of the rotor position detection means 44 is set so as to advance by an electrical angle of 40 degrees or more. In the low / medium rotation range, the control unit 400 shifts the detection point by a predetermined angle. Drive and perform efficient operation utilizing reluctance torque. On the other hand, in the high rotation region,
The control unit 400 shifts by a predetermined angle according to the rotation speed. Thereby, the operating range is expanded to the shift 0, that is, the position where the advance angle advances by 40 degrees or more.

Note that the phase shift amount (shift amount) in the low / medium rotation region must originally be determined based on the motor current, load torque, and the like. The amount of phase shift that improves the efficiency is set in the low / medium rotation region.

Next, the control of the commutation timing in the drive control of the brushless DC motor 40 will be described.

FIG. 30 is a flowchart showing the operation of control section 400 in one commutation cycle. Hereinafter, description will be made based on this flowchart.

First, the rotational speed of the brushless DC motor 40 (the rotational speed of the rotor 42) is measured from the detection signal P4 generated based on the magnetic pole position signals Up, Vp, Wp from the rotor position detecting means 44, and The rotation speed is read (step S101).

In this step S101, the interval between two adjacent detection signals P4, that is, the time from the rise of the detection signal P4 to the rise of the next detection signal P4 is measured, and based on the measured value, the brushless DC motor is measured. The rotation speed of 40 is calculated.

Next, based on the rotation speed, the control unit 4
The data of the amount of phase shift is read from the memory built in 00 (step S102).

In this case, the memory previously stores data of the phase shift amount corresponding to the rotational speed collected in an experiment or the like in the form of a table.
At 102, data of an appropriate phase shift amount corresponding to the rotation speed is read from the memory. The data of the phase shift amount is stored as time converted data.

Next, a timer time is set based on the read phase shift amount (step S10).
3).

Next, timer counting is started (step S104).

Next, it is determined whether or not the timer time has elapsed (time-up) (step S105).

If it is determined in step S105 that the timer time has not elapsed, the process returns to step S104 to continue counting the timer (step S104), and determines again whether the timer time has elapsed (step S105). ).

When it is determined in step S105 that the timer time has elapsed, a commutation signal is output (step S106).

Thus, the program (control operation) for one commutation cycle is completed.

As described above, in this embodiment, since the detection point by the rotor position detection means 44 can be set so as to advance by 30 electrical degrees or more, the operating range of the brushless DC motor 40 is widened.

Further, the point of commutation can be shifted (delayed) by a predetermined angle by the control unit 400.
Operation efficiency in the low / medium rotation range can be improved.

Since the amount of phase shift (time to delay commutation) is automatically set according to the number of revolutions, more precise operation control can be performed easily and reliably.

In the present embodiment, the control section 400 mainly performs reading of the number of revolutions, storage of phase shift data, timer counting, and the like. In the present invention, in addition to these, the PWM control circuit 10, the operation of the output pattern generation circuit 9, the reading of the rotational speed command 11, the processing of other digital signals, and the like may be performed by the control unit 400 collectively.

(Embodiment 7) Embodiment 7 of the present invention will be described below with reference to the accompanying drawings.

FIG. 31 is a block diagram showing another example of the configuration of the brushless DC motor driving apparatus according to the present invention. The description of the common points with the above-described sixth embodiment will be omitted, and different points will be described.

As shown in the figure, the brushless DC motor driving apparatus according to the present embodiment has the same structure as that of the sixth embodiment.
It has a current detecting means 401 for detecting a motor current. In this case, the motor current is detected by the current detecting unit 401, and the detected value of the motor current (motor current value) is input from the current detecting unit 401 to the control unit 400.

Next, control of commutation timing in drive control of the brushless DC motor 40 will be described.

FIG. 32 is a flowchart showing the operation of control section 400 in one commutation cycle. Hereinafter, description will be made based on this flowchart.

First, the rotation speed of the brushless DC motor 40 (the rotation speed of the rotor 42) is measured from the detection signal P4 generated based on the magnetic pole position signals Up, Vp, Wp from the rotor position detection means 44, and The rotation speed is read (step P101).

In step P101, the interval between two adjacent detection signals P4, that is, the time from the rise of the detection signal P4 to the rise of the next detection signal P4 is measured, and based on the measured value, the brushless DC motor The rotation speed of 40 is calculated.

Next, the motor current value is read (step P102).

Next, based on the rotation speed and the motor current value, data of the phase shift amount is read from the memory built in the control section 400 (step P103).

In this case, the memory previously stores data of the number of rotations and the amount of phase shift corresponding to the motor current value collected in an experiment or the like in the form of a table. , Data of an appropriate phase shift amount corresponding to the rotation speed and the motor current value is read. The data of the phase shift amount is stored as time converted data.

Next, a timer time is set based on the read phase shift amount (step P10).
4).

Next, timer counting is started (step P105).

Next, it is determined whether or not the timer time has elapsed (time-up) (step P106).

If it is determined in step P106 that the timer time has not elapsed, the process returns to step P105 to continue counting the timer (step P105), and again determines whether the timer time has elapsed (step P106). ).

If it is determined in step P106 that the timer time has elapsed, a commutation signal is output (step P107).

Thus, the program (control operation) for one commutation cycle is completed.

As described above, in this embodiment, similarly to the above-described sixth embodiment, the detection point by the rotor position detection means 44 can be set so as to advance by an electrical angle of 30 degrees or more. Spreads and shifts (delays) the commutation point by a predetermined angle by the control unit 400.
Therefore, the operation efficiency in the low / medium rotation range can be improved.

Further, in this embodiment, since the amount of phase shift (time for delaying commutation) is automatically set according to the number of revolutions and the motor current value, more precise operation control can be easily and reliably performed. It can be carried out.

Accordingly, the operating range in the high rotation region is expanded, and the operation efficiency particularly in the low / middle rotation region is improved. Further, over the entire operation range (when the field-weakening operation is not performed), high-efficiency operation becomes possible (in particular, it is advantageous in the case of a motor having an embedded magnet type rotor structure).

In this embodiment, the control section 400 mainly reads the number of revolutions, stores the data of the phase shift, counts the timer, reads the motor current value (A / D conversion).
In the present invention, the control unit 400 collectively performs the operations of the PWM control circuit 10, the output pattern generation circuit 9, the reading of the rotational speed command 11, the processing of other digital signals, and the like. It may be configured as follows.

[0238]

As described above, according to the brushless DC motor driving apparatus of the present invention, it is possible to set the rotor position detection signal so as to accurately advance the electrical angle by 30 degrees or more. Therefore, the range in which the advance angle can be set (angle range)
And the operating range of the brushless DC motor can be expanded, and the efficiency can be improved.

The applied voltage discriminating means includes first, second,
When the third gain switching means is provided, the first,
Since the gain of the line voltage by the second and third amplifying means can be changed, the operation range can be further expanded.

When n applied voltage discriminating means (n is an integer of 2 or more) and n gain switching means are provided, the operating range can be expanded stepwise during motor operation. It becomes possible.

When a hysteresis circuit is added to the applied voltage discriminating means, the operating range can be expanded and the expanded area can be operated arbitrarily.

Also, the signal from the rotor position detecting means is
Chop-on detecting means for detecting when the chop is on in the PWM chopper control, open-phase selecting means for selecting a signal from the chop-on detecting means in accordance with the open phase,
In particular, in the case of having an edge detecting means for detecting a predetermined edge based on a signal from the open phase selecting means, the magnetic pole position of the rotor can be properly and reliably detected, and thus, a normal operation can be performed. be able to.

In the case where the rotor position detecting means detects the magnetic pole position of a predetermined rotor and outputs a commutation signal after the rotor rotates by a predetermined phase shift amount, the brushless motor can easily be used. The operating range of the DC motor can be expanded, and the efficiency can be improved.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a driving device for a conventional sensorless brushless DC motor.

FIG. 2 is a circuit diagram showing a phase component of a rotor position detecting means of a conventional sensorless brushless DC motor driving device.

FIG. 3 is a plan view schematically showing a surface magnet type rotor.

FIG. 4 is a plan view schematically showing an embedded magnet type rotor.

FIG. 5 is a graph showing an advance angle θ-torque T characteristic when maximum torque control is performed.

FIG. 6 is a block diagram illustrating a configuration example of a drive device for a brushless DC motor according to the present invention.

FIG. 7 is a diagram showing an excitation pattern of the brushless DC motor according to the present invention.

FIG. 8 is a diagram showing a U-phase, a V-phase, and a W-phase back electromotive force ea when the advance angle is 0 degree in a brushless DC motor according to the present invention;
FIG. 9 is a diagram showing a relationship between eb and ec and a drive signal.

FIG. 9 shows Ta + −Tb− in chopper control according to the present invention.
FIG. 6 is a circuit diagram showing an equivalent circuit when is conducting.

FIG. 10 is a diagram showing detection timing of back electromotive force in the present invention.

FIG. 11 is a circuit diagram illustrating a configuration example of a rotor position detection unit according to the present invention.

FIG. 12 is a diagram showing detection timing of rotor position detection in the present invention.

FIG. 13 is a timing chart showing signal waveforms at various points in the circuit diagram shown in FIG. 11;

FIG. 14 is a block diagram showing a main part of another configuration example of the rotor position detecting means in the present invention.

FIG. 15 is a circuit diagram showing a configuration example of an applied voltage determination unit in the present invention.

FIG. 16 is a circuit diagram illustrating a configuration example of a photo MOS switch according to the present invention.

FIG. 17 is a graph showing motor nakedness characteristics in the present invention.

FIG. 18 is a graph showing motor efficiency in the present invention.

FIG. 19 is a graph showing inverter efficiency according to the present invention.

FIG. 20 is a circuit diagram showing another configuration example of the first amplifying means in the present invention.

FIG. 21 is a circuit diagram showing another configuration example of the applied voltage discriminating means in the present invention.

FIG. 22 is a circuit diagram showing another configuration example of the applied voltage discriminating means and the circuit in the vicinity thereof according to the present invention.

FIG. 23 is a diagram showing a relationship between input and output of a comparator and a flip-flop according to the present invention.

FIG. 24 is a graph showing TN characteristics in the present invention.

FIG. 25 is a block diagram illustrating a configuration example of a drive control unit according to the present invention.

FIG. 26 is a circuit diagram illustrating a configuration example of a chop-on detection unit, an open phase selection unit, and an edge detection unit according to the present invention.

FIG. 27 is a timing chart showing the operation of the circuit shown in FIG. 26;

FIG. 28 is a diagram showing a truth table according to the present invention.

FIG. 29 is a block diagram showing another configuration example of the brushless DC motor driving device of the present invention.

FIG. 30 is a flowchart showing the operation of the control unit in one commutation cycle.

FIG. 31 is a block diagram showing another configuration example of the brushless DC motor driving device of the present invention.

FIG. 32 is a flowchart showing the operation of the control unit in one commutation cycle.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Commercial power supply 2 Rectifier circuit 3 Voltage-type inverter 4 Brushless DC motor 5 Rotor position detecting means 6 Stator armature winding end 7 Drive control means 8 Inverter drive circuit 9 Output pattern generation circuit 10 PWM control circuit 11 Revolution number command 12 Yoke 13 Field permanent magnet 14 Sleeve 15 Yoke 16 Field permanent magnet 20 Voltage dividing circuit 21, 22 Resistor 23 Capacitor 24 First-order lag filter circuit 25 Resistance 26 Capacitor 27 Comparison circuit 28-30 Resistance 31 Comparator 32 Input terminal 33 Output terminal 34 reference voltage 35 reference voltage 40 brushless DC motor 41 stator 42 rotor 43 voltage type inverter 44 rotor position detecting means 45 drive control means 50a first line voltage generating means 50b second line voltage generating means 50c third line Voltage generator 51 to 54 resistor 55 amplifier 56 to 59 resistor 60, 65 amplifier 61 to 64 resistor 61 to 64 resistor 66a first comparing means 66b second comparing means 66c third comparing means 70a, 70b, 70c insulating coupler 71 chop-on Detecting means 72 open phase selecting means 73 edge detecting means 80a first amplifying means 80b second amplifying means 80c third amplifying means 81, 82 resistor 83 amplifier 84, 85 resistor 86 amplifier 87, 88 resistor 89 amplifier 90a first Gain switching means 90b second gain switching means 90c third gain switching means 91, 93, 95 resistors 92, 94, 96 analog switches 101-10n applied voltage discriminating means 100 AND gates 101-103 AND circuit 110 EX-OR Gates 111-113 EX- R circuit 121 Data selector 200 AND gate 201 to 203 AND circuit 204 Triangular wave generation circuit 205 Comparator 206 LED 207 MOSFET 208 Comparator 210 OR gate 301 to 30n Comparator 311 312 Mono multi 400 Control section 401 Current detection means 402 Comparison Device 501 flip-flop S101 to S106 step P101 to P107 step

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02P 6/08 H02P 6/18

Claims (12)

(57) [Claims] 1. A stator in which three-phase armature windings U, V, and W are star-connected, a rotor that forms a magnetic pole pair by a permanent magnet embedded in a yoke that forms a magnetic path, and a plurality of semiconductor switching devices. A 120-degree conduction type inverter having an element, rotor position detecting means for detecting a terminal voltage generated at an armature winding end of the stator and generating a signal corresponding to a magnetic pole position of the rotor, and the rotor position detection Based on the signal from the means,
In a brushless DC motor drive device having a drive control unit that performs speed adjustment by chopper control, the rotor position detection unit detects armature winding WU based on a terminal voltage of an armature winding end of the stator. A first line voltage generating unit for generating a line voltage Vw-u between the first and second lines; a second line voltage generating unit for generating a line voltage Vu-v between the armature windings U and V; Third line voltage generating means for generating a line voltage Vv-w between the child windings V and W, and a line voltage V output from the first line voltage generating means
a first amplifying means for amplifying a signal relating to w-u, and a line voltage Vu- output from the second line voltage generating means.
a second amplifying means for amplifying a signal relating to v, and a line voltage V output from the third line voltage generating means.
a third amplifying means for amplifying a signal related to vw, a first comparing means for comparing a signal related to the line voltage Vw-u and a signal output from the second amplifying means, A second comparing unit that compares a signal related to a voltage Vu-v with a signal output from the third amplifying unit; a signal related to the line voltage Vv-w and a signal output from the first amplifying unit And a third comparing means for comparing the first, second and third rotor positions with each other.
A brushless DC motor characterized in that the position of the rotor is detected at a current phase angle θ at which the combination of the torque and the reluctance torque by the magnet of the rotor is determined by setting the amplification factor of the amplifying means. Drive. 2. The brushless DC motor according to claim 1, wherein the rotor position detecting means is configured to detect a predetermined magnetic pole position of the rotor at which the current phase angle θ is advanced by 30 degrees or more in electrical angle. Drive. 3. The method according to claim 1, wherein the first comparing unit is configured to output the line voltage V
When the signal related to wu is larger than the signal output from the second amplifying means, a high-level signal is output, and the second comparing means determines that the signal related to the line voltage Vu-v A high-level signal is output when the signal is larger than the signal output from the third amplifying means.
3. The brushless DC motor according to claim 1, wherein a high-level signal is output when a signal related to the line voltage Vv-w is larger than a signal output from the first amplifying unit. 4. Drive. 4. An applied voltage discriminating means for discriminating a motor applied voltage, and said first, second, and third rotor position detecting means based on a signal from said applied voltage discriminating means.
4. The brushless DC motor drive device according to claim 1, further comprising first, second, and third gain switching means for changing a gain of said amplification means. 5. The rotor position detecting means includes: n (n is an integer of 2 or more) applied voltage determining means for determining a motor applied voltage; and 4. The brushless DC motor driving device according to claim 1, further comprising n gain switching means for changing the gains of the second and third amplification means. 6. The brushless DC motor driving device according to claim 4, wherein a hysteresis circuit is additionally provided to said applied voltage determining means. 7. The drive control means includes: a chop-on detection means for detecting a signal from the rotor position detection means at the time of chop-on of PWM chopper control; and a signal from the chop-on detection means in an open phase. 7. The brushless DC motor drive device according to claim 1, further comprising: an open phase selecting unit that selects the combination, and an edge detecting unit that detects a predetermined edge based on a signal from the open phase selecting unit. . 8. When a predetermined magnetic pole position of the rotor is detected by the rotor position detecting means, a commutation signal is output in synchronization with the detection.
8. The driving device for a brushless DC motor according to any one of claims 7 to 7. 9. A commutation signal is output after the rotor has rotated by a predetermined phase shift amount from when a predetermined magnetic pole position of the rotor is detected by the rotor position detection means. 8. The driving device for a brushless DC motor according to any one of 1 to 7. 10. The brushless DC motor driving device according to claim 9, further comprising a shift amount setting means for setting the phase shift amount. 11. The brushless DC motor drive device according to claim 10, wherein the setting of the phase shift amount by the shift amount setting means is changed at least in accordance with the rotation speed of the rotor. 12. The brushless DC motor driving device according to claim 10, wherein the setting of the phase shift amount by the shift amount setting means is changed according to the rotation speed of the rotor and a motor current.