JPH08331883A - Driver of brushless dc motor - Google Patents

Abstract

PURPOSE: To detect a rotor position always accurately without being influenced by the variation of a rotation frequency and a load by a method wherein the rotor position is detected in accordance with line voltages between armature windings. CONSTITUTION: Line voltage generators 50a-50c generate line voltages between armature windings in accordance with terminal voltages Vu , Vv and Vw . The line voltages are amplified by amplifiers 80a-80c. A comparator 66a compares the output of the line voltage generator 50a with the output of the amplifier 80b. A comparator 66b compares the output of the line voltage generator 50b with the output of time amplifier 80c. A comparator 66c compares the output of the line voltage generator 50c with the output of the amplifier 80a. The outputs of the respective comparators 66a-66c are used as rotor position detection outputs. With this constitution, the rotor position can be detected always accurately without being influenced by the variation of a rotation frequency and a load.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a brushless DC motor driving device.

[0002]

2. Description of the Related Art A conventional position sensorless brushless DC motor drive device will be described. FIG. 1 is a block diagram showing a conventional position sensorless brushless DC motor drive device.

As shown in FIG. 1, a conventional position sensorless brushless DC motor drive device includes a rectifier circuit 2, a 120-degree conduction type voltage inverter (inverter) 3 composed of six semiconductor switching elements, and a brushless. DC
Motor 4, rotor position detection means 5 and drive control means 7
It is composed of

A DC power source rectified and smoothed by the rectifier circuit 2 is supplied from the commercial power source 1 to the input terminal of the inverter 3. The output end of the inverter 3 is connected to the stator armature winding of the brushless DC motor 4, and the brushless D is obtained by connecting and disconnecting the DC power supply.
The C motor 4 is adapted to rotate.

The brushless DC motor 4 is composed of a stator in which polyphase armature windings are star-connected and a rotor forming a magnetic pole pair by permanent magnets, and the stator armature winding end 6 is formed by rotating the rotor. Back electromotive force is generated.

The counter electromotive force is input to the rotor position detecting means 5 from the stator armature winding end 6, the rotor position detecting means 5 detects the rotor position, and the drive control means 7 is provided as a pulse signal. Entered in.

The drive control means 7 includes an inverter drive circuit 8, an output pattern generation circuit 9 and a PWM control circuit 1.
It consists of 0.

The output pattern generation circuit 9 determines a pattern for driving the gate of each switching element of the inverter 3 in accordance with the detection timing input from the rotor position detection means 5. Furthermore, the output pattern generation circuit 9
This pattern is combined with the output of the PWM control circuit 10 that determines the duty ratio of PWM chopper control conduction / interruption according to the rotation speed command 11 to generate a signal, and the signal is input to the inverter drive circuit 8. Then, each gate of the inverter 3 is driven by the inverter drive circuit 8.

Next, the rotor position detecting means 5 of the conventional position sensorless brushless DC motor driving device will be described. FIG. 2 is a circuit diagram showing one phase of the rotor position detecting means 5.

As shown in the figure, the rotor position detecting means 5
Is a voltage dividing circuit 20 including resistors 21 and 22, a capacitor 23 for removing a DC component, and a first-order lag filter circuit 24 including a resistor 25 and a capacitor 26.
And a comparison circuit 27 including resistors 28, 29, 30 and a comparator 31.

The terminal voltage of the stator armature winding end 6 is input from the input terminal 32 to the voltage dividing circuit 20, and the terminal voltage is divided by the resistors 21 and 22 with 0 V of the DC power source as a reference. Detected by.

In addition to the fundamental wave component of the back electromotive force, the detected terminal voltage has a high frequency component under the PWM chopper control, a spike voltage generated in the return mode after commutation, and resistors 21 and 22 of the voltage dividing circuit 20. DC offset due to variations is included. To mitigate these,
In order to cut the DC component, the first-order lag filter 24 is used to reduce the harmonic component through the connection of the capacitor 23, and the phase is shifted by 90 degrees.

The comparison circuit 27 compares the output of the first-order lag filter 24 with the reference voltage 34 to output the output terminal 3
A rotor position detection signal is output to 3 as a pulse signal.
As the reference voltage 34, a neutral point voltage directly used for the stator armature winding or a neutral point voltage obtained by combining the terminal voltage after the primary delay filter 24 is used.

Since the detection signal which is phase-shifted by 90 degrees is used as the rotor position detection signal of the phase which is delayed by 120 degrees from the detected own phase, the current phase angle is driven at a position of approximately 0 degrees. For example, when driving in three phases of U, V and W, the detection signal detected by the terminal voltage of U phase is W
It is used as a phase rotor position detection signal.

As shown in FIG. 3, the rotor of the brushless DC motor 4 generally has a yoke 12 forming a magnetic path and a rocker-shaped permanent magnet 13 for field magnetism.
A surface magnet type formed by sticking the above-mentioned permanent magnet 13 for field magnet on the outer peripheral surface of the yoke 12, a yoke 15 in which a rotor forms a magnetic path, and a permanent magnet for field magnet as shown in FIG. An embedded magnet type having a magnet 16 and having the field permanent magnet 16 inserted into a slot provided in the yoke 15 is used. Optimal methods have been proposed for these control methods (Morimoto, Ueno,
Takeda "Wide range variable speed control of embedded magnet type PM motor"
1994 IEEJ Transactions, Vol. 114, No. 6, p662-
p667).

Generally, the torque equation of a brushless DC motor using permanent magnets represented by dq coordinates is given by the following equation.

T = p · {φmag · iq + (Ld−Lq) · id · iq}. ． ． (1) where T is the torque, p is the number of pole pairs, φmag is the armature flux linkage by the permanent magnets, id and iq are the d-axis component and q-axis component of the armature current, and Ld and Lq are the d-axis, respectively. Inductance and q-axis inductance are shown.

When the amplitude of the armature current is I and the phase angle (current phase angle) of the armature current with respect to the q axis is θ, i
d and iq are defined as follows, respectively.

I d = −I · sin θ. ． ． (2) iq = I · cos θ. ． ． (3) Here, I represents the amplitude of the armature current, and θ represents the current phase angle viewed from the q axis.

The torque equation (1) can be converted from the equations (2) and (3) into the following equation.

[0021]

[Equation 1]

Here, L1 is as follows.

L1 = (Lq-Ld) / 2. ． ． (5) In the case of the surface magnet type rotor shown in FIG. 3, the d-axis inductance and the q-axis inductance are equal and non-salient (Ld = Lq
), The second term in the torque equation (1) becomes 0, and the motor torque is generated in proportion to the q-axis current iq. Further, also in the equation (4), similarly, it is understood that the second term becomes 0 and the maximum torque is obtained when the current phase angle θ is 0 degree. That is, according to the equations (2) and (3), setting the current phase angle θ to 0 degree and id = 0 is an optimum operation control method for the brushless DC motor having the non-salient surface magnet type rotor structure. is there. This is a control method that generally keeps the d-axis current at 0, and is called id = 0 control.

The surface magnet type rotor, as shown in FIG.
The outer peripheral surface is covered with a non-magnetic sleeve 14,
Prevents scattering of permanent magnets due to high rotation.

On the other hand, in the embedded magnet type rotor, as shown in FIG. 4, the magnetic pole generated by the field permanent magnet 16 is provided on the outer periphery of the yoke 15 formed by laminating a large number of silicon steel plates. There is no concern about scattering, etc., and particularly high rotation speeds are possible. This embedded magnet rotor
Since the magnetically equivalent air gap in the q-axis direction is smaller than the magnetically equivalent air gap in the d-axis direction, the q-axis inductance is larger than the d-axis inductance, and the reverse saliency (Ld <Lq) is shown.

Therefore, the torque of the motor generated is obtained from the torque equation (1) by the torque by the magnet proportional to the q-axis current iq of the first term and the reluctance torque generated by the reverse saliency of the second term. There is. Therefore, from the torque equation (4), the current phase angle θ (hereinafter referred to as the advance angle θ) that maximizes the combination of the torque by the magnet and the reluctance torque.
Is the optimum operation control method for the brushless DC motor having the embedded magnet type rotor structure having the reverse salient polarity. This is a control method that effectively utilizes reluctance torque and is called maximum torque control. Figure 5
The relationship between the advance angle θ and the motor torque T when the maximum torque control is performed is shown in FIG.

Further, as an embedded magnet type control method, there has been proposed a method of expanding the high speed rotation range by equivalent field weakening control which positively utilizes reluctance torque. In this method, the optimum torque is controlled by the maximum torque control up to an operation range where the maximum applied voltage applied to the motor is equal to the back electromotive force of the motor. Normally, in a high speed rotation region higher than this, the maximum applied voltage and the back electromotive force of the motor become equal, so that the motor cannot be operated. However, by advancing the advance angle θ further, it is possible to positively flow the d-axis current and equivalently weaken the armature flux linkage of the permanent magnet by the d-axis armature reaction. If this equivalent field weakening control is controlled so that the motor output becomes a constant output, the operating range in the high speed rotation range can be expanded.
This is generally called maximum output control.

[0028]

In the above-mentioned conventional rotor position detecting means 5, the first-order lag filter 24 is used.
Since the high frequency component is cut from the terminal voltage of the stator armature winding end 6 and the terminal voltage is phase shifted by 90 degrees, the cutoff frequency of the first-order lag filter 24 is set to several Hz to several tens Hz or less. ing.

On the other hand, the fundamental wave component of the back electromotive force generated at the stator armature winding end 6 due to the rotation of the rotor is generally variable up to about several hundred Hz depending on the motor rotation range, so that the first-order lag filter is used. The phase of the rotor position detection signal that has passed through 24 causes a detection delay due to an increase in the rotation frequency of the brushless DC motor 4, and
An accurate commutation phase cannot be obtained because a spike voltage is generated in the recirculation mode after commutation, and the pulse width increases depending on the condition of the load current, leading to advance in detection. In particular, when driving at high rotation, the operation is performed in a lag phase, so that the operating range is limited and the motor efficiency is reduced.

Therefore, it is difficult to optimally control the operation of the brushless DC motor 4 with the conventional rotor position detecting means 5. Especially, in the case of the embedded magnet type rotor structure, as shown in FIG. On the other hand, it is necessary to control the advance angle θ so as to advance, and there is a problem that the motor performance cannot be fully utilized.

Further, in the position sensorless brushless DC motor having the embedded magnet type rotor structure, the maximum output control method as described above is used to extend the operating range.

However, since a high-resolution encoder is required to precisely control the advance angle θ, the cost is high, and many position sensorless brushless DC motors are used under a special environment. However, there is a problem that a sensor for detecting the rotor position, such as a high resolution encoder or a Hall element, cannot be used.

An object of the present invention is to provide a rotor position detecting means capable of always detecting an accurate rotor position without being influenced by a change in a rotation frequency or a load of a brushless DC motor. It is an object of the present invention to provide a brushless DC motor drive device capable of simply expanding the operating range of a brushless DC motor having a die rotor structure.

[0034]

This and other objects are achieved by the present invention which is defined below as (1) to (12).

(1) A 120-degree conduction type equipped with a stator in which three-phase armature windings U, V, and W are star-connected, a rotor forming a magnetic pole pair with permanent magnets, and a plurality of semiconductor switching elements. An inverter, rotor position detection means for detecting a terminal voltage generated at the armature winding end of the stator and generating a signal corresponding to the magnetic pole position of the rotor, and based on the signal from the rotor position detection means,
In a drive device for a brushless DC motor having drive control means for performing speed adjustment by PWM chopper control by the inverter, the rotor position detection means includes armature winding based on a terminal voltage of an armature winding end of the stator. A first line voltage generation means for generating a line voltage Vw-u between the lines W-U and a line voltage Vu between the armature windings U-V.
Second line voltage generating means for generating -v, third line voltage generating means for generating a line voltage Vv-w between the armature windings V-W, and the first line voltage First amplification means for amplifying a signal related to the line voltage Vw-u output from the generation means, and second amplification means for amplifying a signal related to the line voltage Vu-v output from the second line voltage generation means. Amplifying means, third amplifying means for amplifying a signal relating to the line voltage Vv-w output from the third line voltage generating means, a signal relating to the line voltage Vw-u, and the second First comparing means for comparing the signal output from the amplifying means, and second comparing means for comparing the signal related to the line voltage Vu-v with the signal output from the third amplifying means, The signal related to the line voltage Vv-w and the signal from the first amplifying means are output. Brushless DC motor driving device, characterized in that a third comparison means for comparing the signal.

(2) When the phase angle of the current flowing through the armature winding of the stator with respect to the q axis in the dq coordinate system is the current phase angle θ, the rotor position detecting means is:
The brushless DC motor drive device according to (1), which is configured to detect a predetermined magnetic pole position of the rotor in which the current phase angle θ advances by an electrical angle of 30 degrees or more.

(3) The first comparing means outputs a high level signal when the signal relating to the line voltage Vw-u is larger than the signal output from the second amplifying means, and the first comparing means outputs the high level signal. The second comparing means outputs a high level signal when the signal related to the line voltage Vu-v is larger than the signal output from the third amplifying means, and the third comparing means outputs the high level signal. The signal related to the inter-voltage Vv-w is the first signal.
The brushless DC motor drive device according to (1) or (2), which is configured to output a high-level signal when the signal is larger than the signal output from the amplification means.

(4) The rotor position detecting means determines an applied voltage determining means for determining a motor applied voltage, and the first, second and third amplifying means based on a signal from the applied voltage determining means. The brushless DC motor drive device according to any one of (1) to (3), further including first, second, and third gain switching means for changing a gain.

(5) The rotor position detecting means determines the motor applied voltage by n (n is an integer of 2 or more) applied voltage determining means, and the first voltage based on the signal from the applied voltage determining means. The brushless DC according to any one of (1) to (3) above, including n gain switching means for changing the gains of the second and third amplifying means.
Motor drive device.

(6) The brushless DC motor drive device according to (4) or (5), wherein a hysteresis circuit is attached to the applied voltage determining means.

(7) The drive control means detects the signal from the rotor position detection means at the same time as the chop-on of the PWM chopper control, and the signal from the chop-on detection means in the open phase. Brushless DC according to any one of the above (1) to (6), further comprising open phase selection means for selecting in accordance with the above, and edge detection means for detecting a predetermined edge based on a signal from the open phase selection means. Motor drive device.

(8) When the rotor position detecting means detects a predetermined magnetic pole position of the rotor, the commutation signal is output in synchronization with the detection. ) Brushless D according to any one of
C motor drive device.

(9) The commutation signal is output after the rotor has rotated a predetermined amount of phase shift since the predetermined magnetic pole position of the rotor was detected by the rotor position detecting means. The brushless DC motor drive device according to any one of (1) to (7).

(10) The brushless D described in (9) above, which has shift amount setting means for setting the phase shift amount.
C motor drive device.

(11) The brushless DC motor drive device according to (10), wherein the setting of the phase shift amount by the shift amount setting means is changed at least according to the rotation speed of the rotor.

(12) The setting of the phase shift amount by the shift amount setting means is changed according to the rotation speed of the rotor and the motor current.
C motor drive device.

[0047]

BEST MODE FOR CARRYING OUT THE INVENTION A brushless DC motor (position sensorless brushless DC motor) drive device according to the present invention will be described below in detail with reference to the preferred embodiments shown in the accompanying drawings.

(Embodiment 1) Embodiment 1 of the present invention will be described below with reference to the accompanying drawings.

FIG. 6 is a block diagram showing an example of the configuration of a brushless DC motor drive device according to the present invention.

As shown in the drawing, the brushless DC motor drive device includes a stator 41 in which three-phase armature windings (excitation coils) are star-connected, and a rotor 42 which forms magnetic poles (magnetic pole pairs) by permanent magnets. A brushless DC motor (motor) 40 configured as described above, and a 120-degree conduction type voltage inverter (hereinafter, referred to as
43), the rotor position detecting means 44 utilizing the back electromotive force at the time of chop-on, and the drive control means 45 for controlling the drive of the inverter 43 by the output (synonymous with a signal) of the rotor position detecting means 44. It is configured.

The exciting coils U, V, W of each phase of the stator 41 in which the three-phase armature windings are star-connected are the inverters 4
3 is connected to the output side. A direct current power source Ed is applied to the input side of the inverter 43, and the P side of each transistor has Ed + of the direct current power source Ed and the N side has Ed− of Ed.
Is connected. The inverter 43 has a P-side feedback diode Da +, Db +, Pc
Side transistor (semiconductor switching element) Ta +,
Tb +, Tc +, and N-side freewheeling diode Da-, respectively.
It is composed of N-side transistors (semiconductor switching elements) Ta-, Tb-, and Tc- connected to Db- and Dc-.

The drive control means 45, as shown in FIG.
Two windings are selected from the exciting coils of each phase, and the P-side transistor and the N-side transistor are sequentially connected in an exciting pattern in a combined manner, thereby forming a rotating magnetic field in the stator 41 and rotating the rotor 42. . Further, for either one of the P-side transistor and the N-side transistor of the excitation pattern, conduction and interruption (hereinafter, conduction is referred to as chop-on and interruption is referred to as chop-off) are alternately repeated by PWM chopper control, and chop-on,
The input power is adjusted by varying the chop-off duty ratio to adjust the speed. At this time, the terminal voltages Vu, Vv, and Vw are obtained in the excitation coils U, V, and W of the stator 41 on the basis of the Ed- side of the DC power source Ed (the reference is the Ed + side or Ed- side of the DC power source Ed. Either side may be used, but in the present embodiment, the Ed− side is used as a reference.

Next, the waveforms of these terminal voltages will be described. FIG. 8 shows the advance angle 0 in the brushless DC motor 40.
Degree, U-phase, V-phase, W-phase counter electromotive force (back electromotive force) e
It is a figure which shows the relationship between a, eb, ec, and a drive signal.
In this case, in any of the phases, only two transistors of a predetermined one phase on the P side and a predetermined one phase on the N side, which are different from that, are operating among the respective transistors. Therefore, the terminal voltage of each phase is twice in one cycle (6 in electrical angle).
There is a period in which neither the P-side transistor nor the N-side transistor operates at the same time. Hereinafter, this period is referred to as an "open period" and the phase in this state is referred to as an "open phase".

The term "advance angle" means a phase angle (current phase angle) of the current flowing through the armature winding (excitation coil) of the stator with respect to the q axis in the dq coordinate system.

Here, the terminal voltage at the time of chop-on in the open period will be examined. Figure 9 shows Ta + with chopper control
It is a circuit diagram showing an equivalent circuit when -Tb- is conductive. As shown in the figure, drive signals are input to Ta + and Tb-,
It is assumed that the current i conducts from Ta + to Tb-, that is, the so-called U-phase and V-phase conduct, and the chop is on. At this time, the terminal voltage Vw of the W phase which is the open phase can be expressed as follows.
In FIG. 9, L is the inductance, r is the resistance value,
VCE is the collector-emitter saturation voltage of the transistor.

First, the following equation holds from the loop through which the current i flows.

Ed = 2 (L · di / dt + ri) + ea−eb + 2 · VCE. ． ． (6) L.di / dt + ri = Ed / 2- (ea-eb) / 2-VCE. ． ． (7) At this time, the terminal voltage Vw of the W phase which is an open phase is expressed by the following formula.

Vw = VCE + L.di / dt + ri-eb + ec. ． ． (8) Substituting the equation (7) into the equation (8), the following equation is obtained.

Vw = Ed / 2 + ec- (ea + eb) / 2. ． ． (9) Here, as shown in FIG. 10A, assuming that the counter electromotive force has a completely symmetrical waveform, near the point P where the counter electromotive force ec of the open phase W phase is 0V, Since ea = -eb, the equation (9) becomes the following equation.

Vw = Ed / 2 + ec. ． ． (10) This relationship is established regardless of the PWM chopper control method such as the P-side chopper or the N-side chopper.
An open phase terminal voltage that satisfies the formula is obtained.

Further, from the equation (10), it is understood that the open phase terminal voltage Vw changes (swings up and down) according to the counter electromotive force ec with reference to the potential of 1/2 of the DC power source Ed. That is, at the time of chop-on, the open phase terminal voltage Vw becomes Ed / 2 (the time when ec = 0).
However, the detection timing is 30 degrees ahead of the commutation point where the advance angle is 0 degrees.

Next, a method for detecting the point P at which the back electromotive force ec of the open phase terminal voltage Vw becomes 0V will be described.
In the case of this example, during the open period of the W phase, the state of the other U and V phases at the time of chop-on, that is, the U phase terminal voltage Vu is Vu = Ed-VCE. ． ． (11) and the V-phase terminal voltage Vv is Vv = 0 + VCE. ． ． (12) As shown in FIG. 10B, the terminal voltage of the V phase with respect to the W phase is Vv-Vw = -ec- (Ed / 2-VCE). ． ． (13), and the terminal voltage of the W phase with respect to the U phase is Vw-Vu = ec- (Ed / 2-VCE). ． ． (14) In equations (13) and (14) above, ec
If the point P of the counter electromotive force ec is found by substituting = 0,-(Ed
/ 2-VCE), Vv-Vw and Vw-Vu coincide. Generally, VCE is a small value, and equation (13), (1
Since it is included in both equations (4), it does not affect the detection of the point P of ec.

Next, the rotor position detecting means 44 utilizing the above-mentioned counter electromotive force at the time of chop-on will be described. FIG. 11 is a circuit diagram showing a configuration example of the rotor position detecting means 44.

As shown in the figure, the terminal voltages Vu, Vv, Vw at each armature winding end of the stator 41 are
The voltage is input to the first line voltage generation means 50a, the second line voltage generation means 50b, and the third line voltage generation means 50c,
It is converted into line voltages Vw-u, Vu-v, Vv-w.

The respective line voltages are respectively amplified by the first amplifying means 80a, the second amplifying means 80b and the third amplifying means 8.
0c, and these amplification means 80a, 80b, 80
The signal (output) from c and the line voltage generating means 50.
The signals (outputs) from a, 50b, and 50c are the first comparison means 66a, the second comparison means 66b, and the third comparison means 6.
6c is input. The comparison means 66a, 66b, 66
In c, these are directly compared with each other to detect the magnetic pole position of the rotor 42 which is advanced by 30 degrees or more in electrical angle from the commutation point of 0 degree advance angle, and a signal corresponding to the magnetic pole position of the rotor 42 (magnetic pole position Signal) Up, Vp, Wp.

Magnetic pole position signals Up and Vp of the rotor 42
, Wp are input to the drive control means 45 from the rotor position detection means 44, and the drive control means 45 controls the drive of the inverter 43 based on the magnetic pole position signal.

Next, the configuration and operation of each line voltage generating means will be described in detail.

The first line voltage generating means 50a includes a resistor 5
1, 52, 53, 54 and an amplifier 55.

The U-phase terminal voltage Vu and the W-phase terminal voltage Vw are input (applied) to the first line voltage generating means 50a.

That is, the terminal voltage Vw is input to the + input terminal of the amplifier 55 via the resistor 51, and the terminal voltage Vu
Is input to the-input terminal of the amplifier 55 via the resistor 52, and the W-phase terminal voltage with respect to the U-phase is output.

In the first line voltage generating means 50a, the resistor 53 is connected between the output terminal and the-input terminal of the amplifier 55, and the resistor 54 is connected from the + input terminal of the amplifier 55 to the negative side of the DC power source Ed. It is a so-called differential amplifier connected to ground. Here, resistor 51 = resistor 52 = R1,
Assuming that the resistance 53 = the resistance 54 = R2, the output (voltage) Vw-u of the first line voltage generation means 50a is Vw-u = R2 / R1. (Vw-Vu). ． ． (15a). As a result, the line voltage between the W phase and the U phase amplified with the amplification factor determined by the ratio of R2 / R1 is obtained.

Similarly, the second line voltage generating means 50b.
Is composed of resistors 56, 57, 58 and 59, and an amplifier 60.

Then, the V-phase terminal voltage Vv and the U-phase terminal voltage Vu are input to the second line voltage generating means 50b.

That is, the terminal voltage Vu is input to the + input terminal of the amplifier 60 via the resistor 56, and the terminal voltage Vv
Is input to the-input terminal of the amplifier 60 via the resistor 57, and the U-phase terminal voltage with respect to the V-phase is output.

In the second line voltage generating means 50b, the resistor 58 is connected between the output terminal and the-input terminal of the amplifier 60, and the resistor 59 is connected from the + input terminal of the amplifier 60 to the negative side of the DC power source Ed. It is a so-called differential amplifier connected to ground. Here, resistor 56 = resistor 57 = R1,
Assuming that the resistance 58 = the resistance 59 = R2, the output Vu-v of the second line voltage generating means 50b is Vu-v = R2 / R1. (Vu-Vv). ． ． (15b) As a result, the line voltage between the U-phase and the V-phase amplified with the amplification factor determined by the ratio of R2 / R1 is obtained.

Similarly, the third line voltage generating means 50c.
Is composed of resistors 61, 62, 63, 64 and an amplifier 65.

Then, the W-phase terminal voltage Vw and the V-phase terminal voltage Vv are input to the third line voltage generating means 50c.

That is, the terminal voltage Vv is input to the + input terminal of the amplifier 65 via the resistor 61, and the terminal voltage Vw
Is input to the-input terminal of the amplifier 65 via the resistor 62, and the V-phase terminal voltage with respect to the W-phase is output.

In the third line voltage generating means 50c, the resistor 63 is connected between the output terminal and the-input terminal of the amplifier 65, and the resistor 64 is connected from the + input terminal of the amplifier 65 to the negative side of the DC power source Ed. It is a so-called differential amplifier connected to ground. Here, resistor 61 = resistor 62 = R1,
When the resistance 63 = the resistance 64 = R2, the output Vv-w of the third line voltage generating means 50c is Vv-w = R2 / R1. (Vv-Vw). ． ． (15c). As a result, the line voltage between the V phase and the W phase amplified with the amplification factor determined by the ratio of R2 / R1 is obtained.

Next, the configuration and operation of each amplifying means will be described in detail.

Output Vw- of first line voltage generating means 50a
u is input to the first amplification means 80a. The first amplifying means 80a is composed of resistors 81 and 82 and an amplifier 83. That is, in the first amplification means 80a, the output side of the first line voltage generation means 50a is connected to the + input terminal of the amplifier 83, and the resistor 81 is connected from the − input terminal of the amplifier 83 to the negative side of the DC power supply Ed. It is a so-called common-mode amplifier which is connected so as to be grounded and in which the resistor 82 is connected between the output terminal and the-input terminal of the amplifier 83. here,
Assuming that the resistor 81 = R3 and the resistor 82 = R4, the output Vw-u (gain) of the first amplifying means 80a is Vw-u (gain) = (1 + R4 / R3) .Vw-u. ． ． (16a) is obtained. As a result, the line voltage that is amplified by the amplification factor determined by the ratio of (1 + R4 / R3) and satisfies Vw-u <Vw-u (gain) is obtained.

Similarly, the second line voltage generating means 50.
The output Vu-v of b is input to the second amplification means 80b. The second amplifying means 80b is composed of resistors 84 and 85 and an amplifier 86. That is, in the second amplification means 80b, the output side of the second line voltage generation means 50b is connected to the + input terminal of the amplifier 86, and the resistor 84 is connected from the − input terminal of the amplifier 86 to the negative side of the DC power supply Ed. The resistor 85 is connected to the ground, and the resistor 85 is connected to the output terminal of the amplifier 86.
It is a so-called in-phase amplifier connected between the input terminal and the input terminal. Here, assuming that the resistor 84 = R3 and the resistor 85 = R4, the output Vu-v (gain) of the second amplifying means 80b is Vu-v (gain) = (1 + R4 / R3) .Vu-v. ． ． (16b) is obtained. As a result, the line voltage which is amplified by the amplification factor determined by the ratio of (1 + R4 / R3) and satisfies Vu-v <Vu-v (gain) is obtained.

Similarly, the third line voltage generating means 50
The output Vv-w of c is input to the third amplifying means 80c. The third amplification means 80c is composed of resistors 87 and 88 and an amplifier 89. That is, in the third amplifying means 80c, the output side of the third line voltage generating means 50c is connected to the + input terminal of the amplifier 89, and the resistor 87 is connected from the − input terminal of the amplifier 89 to the negative side of the DC power source Ed. The resistor 88 is connected to the ground, and the resistor 88 is connected to the output terminal of the amplifier 89.
It is a so-called in-phase amplifier connected between the input terminal and the input terminal. Here, assuming that the resistor 87 = R3 and the resistor 88 = R4, the output Vv-w (gain) of the third amplifying means 80c is Vv-w (gain) = (1 + R4 / R3) .Vv-w. ． ． (16c). As a result, a line voltage that is amplified by an amplification factor determined by the ratio of (1 + R4 / R3) and satisfies Vv-w <Vv-w (gain) is obtained.

Next, the operation of each comparison means will be described in detail.

In the first comparing means 66a, the output Vu-v (gain) of the second amplifying means 80b and the output Vw-u of the first line voltage generating means 50a are compared and Vu-v (gain ) ≧ Vw-
When u, Low voltage (low level voltage) is output,
When Vu-v (gain) <Vw-u, a high voltage (high level voltage) is output, and a pulse signal (magnetic pole position signal) Up is thereby generated.

Similarly, in the second comparing means 66b,
The output Vv-w (gain) of the third amplifying means 80c and the output Vu-v of the second line voltage generating means 50b are compared to obtain Vv-w.
When (gain) ≥ Vu-v, Low voltage is output, and Vv-w (g
When ain) <Vu-v, the High voltage is output, and thereby the pulse signal (magnetic pole position signal) Vp is generated.

Similarly, in the third comparing means 66c,
The output Vw-u (gain) of the first amplification means 80a and the output Vv-w of the third line voltage generation means 50c are compared, and Vw-u
When (gain) ≥ Vv-w, Low voltage is output and Vw-u (g
When ain) <Vv-w, the High voltage is output, and thereby the pulse signal (magnetic pole position signal) Wp is generated.

The comparison means 66a, 66b, 66c
It is preferable to mainly use a comparator such as a comparator (hereinafter, the comparing means is also referred to as a comparator).

FIG. 12 shows that the first comparing means 66a
The output Vw-u of the first line voltage generating means 50a and (1+
R4 / R3) second amplification means 80 amplified by the amplification factor
It is a figure which shows the detection timing of rotor position detection when comparing with output Vu-v (gain) of b.

As shown in the figure, the period from the point H to the point J (60 electrical degrees) is the open period. A point J at which Vu-v (gain) shows a peak indicates a lead angle of 0 degree. Assuming that the direction of the point H is the advance angle θ with reference to this point J, the advance angle θ determined by the intersection of Vw-u and Vu-v (gain) can be expressed by the following equation.

Θ = {A / (1 + A)} · 60. ． ． (17) Here, A in the equation (17) indicates an amplification factor. Above (1
From equation (7), it can be seen that the advance angle θ is determined by the amplification factor A. For example, when A = 1, the lead angle θ is detected at 30 °. If A = 2, the advance angle θ = 40 degrees,
The phase of rotor position detection can be easily advanced (advanced angle of 30 degrees or more).

In this rotor position detection, the electrical angle is the advance angle θ.
Is preferably set to advance 30 degrees or more, more preferably set to 40 degrees or more, and further preferably set to advance 50 degrees or more.

By advancing the rotor position detection phase as described above, the operating range of the motor is particularly widened, which improves the degree of freedom in system design using, for example, a drive device of a brushless DC motor.

FIG. 13 is a timing chart showing the signal waveform of each part in the circuit diagram shown in FIG. In addition,
In FIG. 13, Vw-u (gain), Vu-v (gain), Vv-w (gain)
Shows the case where the amplification factor A is set to about 4 times, respectively.

As shown in the figure, the rotor position detecting means 4
4, the magnetic pole position signals Up, Vp, Wp set so that the advance angle θ advances by 30 degrees or more are generated.

The signals Up, Vp, and Wp that have advanced by 30 degrees or more obtained here are used as a reference to shift to an advance angle θ according to the characteristics of the motor (for example, the commutation timing is set to a predetermined value). Optimum operation control becomes possible by delaying to the advance angle θ.

The shifting method is not particularly limited,
For example, a method using a counter that counts reference pulses using a detection signal as a trigger, or a microcomputer is used to store a desired shift amount in a storage device such as a ROM, and the shift amount corresponding to the instantaneous rotation speed is read and added. Any method such as a method of doing may be used.

Further, the shift amount may be varied to the optimum value for the motor characteristics by comparing the number of revolutions only, the motor current value only, the number of revolutions and the motor current value, or the like. For example,
The shift amount can be changed corresponding to the motor current value so as to satisfy the relationship of FIG. In any case, since the detection signal can be set to advance accurately by 30 degrees or more by using the rotor position detecting means 44 in the present invention, the advance angle setting range is widened and contributes to the expansion of the operating range.

(Second Embodiment) A second embodiment of the present invention will be described below with reference to the accompanying drawings.

FIG. 14 is a circuit diagram showing a main part of another configuration example of the rotor position detecting means of the brushless DC motor driving device of the present invention.

As shown in the figure, the rotor position detecting means 44 is the rotor position detecting means 44 of the first embodiment described above.
First amplifying means 80a, second amplifying means 80b, third
The first gain switching means 90a, the second gain switching means 90b, and the third gain switching means 90c are connected to the amplification means 80c. These gain switching means 90a, 90b, 90c are each switched at a predetermined timing based on the output L1 of the applied voltage determination means 101 for determining the motor applied voltage.

Next, the operation of the applied voltage discriminating means 101 for discriminating the motor applied voltage will be described in detail. FIG.
FIG. 3 is a circuit diagram showing a configuration example of applied voltage discriminating means 101.

The voltage applied to the motor is adjusted by varying the duty ratio of chop-on and chop-off of the transistor by PWM chopper control. Therefore, in the present embodiment, the applied voltage of the chop-on and chop-off is changed. , Determine the motor applied voltage.

As shown in FIG. 15, the triangular wave generation circuit 20
From 4, the triangular wave of the chopping frequency or the sawtooth wave is output. Then, the command voltage V0 that determines the desired duty ratio and the output of the triangular wave generation circuit 204 are input to the comparator 205, and the comparator 205 compares the command voltage V0 with the output of the triangular wave generation circuit 204. Then, a rectangular wave corresponding to the desired duty ratio is output.

In this embodiment, the command voltage V0 that determines the desired duty ratio is also input to the comparator 208. Then, the comparator 208 compares the command voltage V0 with the reference voltage V1 that sets the switching timing of each of the gain switching means 90a, 90b, 90c, and becomes High.
The output signal L1 of the h voltage or the low voltage is output.

The method of determining the desired duty ratio is not particularly limited, and in the present invention, in addition to the method described above,
For example, a method using a reference oscillator and a timer inside the microcomputer may be adopted.

Next, the configuration and operation of each gain switching means will be described in detail.

As shown in FIG. 14, the first gain switching means 90a has a resistor 91 and an analog switch 92 connected in series, and the second gain switching means 90b has a resistor 93 and an analog switch 94 connected in series. And the third
The gain switching means 90c has a resistor 95 and an analog switch 96 connected in series. in this case,
The analog switches 92, 94 and 96 are turned on and off at the same time by the output signal L1 of the applied voltage determination means 101 that determines the motor applied voltage, and the analog switches 9
By turning on and off 2, 94 and 96, resistors 91, 93,
95 conducts and does not conduct.

As the analog switches 92, 94 and 96, it is preferable to use a switching element such as MOSFET having a low on-resistance or a photo-moss switch as shown in FIG.

The photoMOS switch can directly switch the MOSFET 207 electrically insulated by the light of the LED 206, and can directly trigger the LED 202 from the port of the microcomputer. Therefore, the applied voltage discriminating means 101 is composed of the microcomputer. This is convenient in applications where the rotor position detection means 44 and the ground line are different, such as when.

As shown in FIG. 14, the first amplification means 80
A is composed of resistors 81 and 82 and an amplifier 83. In this case, the output side of the first line voltage generating means 50a is connected to the + input terminal of the amplifier 83, and the resistor 81 is connected from the − input terminal of the amplifier 83 to the negative side of the DC power source Ed so as to be grounded. The resistor 82 is connected between the output terminal and the-input terminal of the amplifier 83, and the resistor 9 is connected in series.
1 and the analog switch 92 are connected in parallel with the resistor 82 between the output terminal and the-input terminal of the amplifier 83, forming a so-called common-mode amplifier. Here, the resistor 81 = R
3, the resistor 82 = R4, and the resistor 91 = R5, the output Vw-u (gain) of the first amplifying means 80a is Vw-u (gain) = (1 + Rz / R3) .Vw-u. ． ． (18a) Rz = (R4.R5) / (R4 + R5). ． ． (19) As a result, the line voltage which is amplified by the amplification factor determined by the ratio of (1 + Rz / R3) and satisfies Vw-u <Vw-u (gain) is obtained. Therefore, the resistance 91 = R5 is switched to conductive or non-conductive by the analog switch 92 to change the combined resistance Rz of the equation (19) to obtain (1 + Rz
The amplification factor determined by the ratio of / R3) can be changed.

Similarly, the second amplifying means 80b is composed of resistors 84 and 85 and an amplifier 86. In this case, the output side of the second line voltage generating means 50b is connected to the + input terminal of the amplifier 86, and the resistor 84 is connected to the amplifier 8
6 is connected to the negative side of the DC power supply Ed so as to be grounded, the resistor 85 is connected between the output terminal of the amplifier 86 and the − input terminal, and the resistor 93 and the analog switch 94 connected in series are connected. , A so-called common-mode amplifier is connected in parallel with the resistor 85 between the output terminal and the-input terminal of the amplifier 86. Here, resistor 84 = R3, resistor 8
5 = R4 and resistor 93 = R5, the second amplification means 8
The output Vu-v (gain) of 0b is Vu-v (gain) = (1 + Rz / R3) .Vu-v. ． ． (18b). As a result, the line voltage which is amplified by the amplification factor determined by the ratio of (1 + Rz / R3) and satisfies Vu-v <Vu-v (gain) is obtained. Therefore, the resistance 93 = R5 is switched between conductive and non-conductive by the analog switch 94 to change the combined resistance Rz of the equation (19) to obtain (1 + Rz
The amplification factor determined by the ratio of / R3) can be changed.

Similarly, the third amplifying means 80c is composed of resistors 87 and 88 and an amplifier 89. In this case, the output side of the third line voltage generating means 50c is connected to the + input terminal of the amplifier 89, and the resistor 87 is connected to the amplifier 8
9 is connected to the negative side of the DC power supply Ed so as to be grounded, the resistor 88 is connected between the output terminal of the amplifier 89 and the − input terminal, and the resistor 95 and the analog switch 96 connected in series are connected. , So-called common-mode amplifier connected in parallel with the resistor 88 between the output terminal and the-input terminal of the amplifier 89. Here, resistor 87 = R3, resistor 8
8 = R4 and resistance 95 = R5, the third amplifying means 8
The output Vv-w (gain) of 0c is Vv-w (gain) = (1 + Rz / R3) .Vv-w. ． ． (18c). As a result, the line voltage which is amplified by the amplification factor determined by the ratio of (1 + Rz / R3) and satisfies Vv-w <Vv-w (gain) is obtained. Therefore, the resistance 93 = R5 is switched between conductive and non-conductive by the analog switch 94 to change the combined resistance Rz of the equation (19) to obtain (1 + Rz
The amplification factor determined by the ratio of / R3) can be changed.

The first, second and third gain switching means 90a, 90b and 90c are respectively connected to the resistor 8 of the first, second and third amplifying means 80a, 80b and 80c.
The gain can be similarly changed by connecting in parallel to 1, 84, and 87.

In this case, the output V of the first amplifying means 80a
wu (gain) is Vw-u (gain) = (1 + R4 / Rz1) .Vw-u. ． ． (20) Rz1 = (R3 · R5) / (R3 + R5). ． ． (21) As a result, the line voltage which is amplified by the amplification factor determined by the ratio of (1 + R4 / Rz1) and satisfies Vw-u <Vw-u (gain) is obtained. Therefore, the resistance 91 = R5 is switched between conductive and non-conductive by the analog switch 92 to change the combined resistance Rz1 of the equation (21), and (1 + R4
The amplification factor determined by the ratio / Rz1) can be changed.

FIG. 17 shows a change in bare motor characteristics when the gain of the brushless DC motor having the embedded magnet type rotor structure having the characteristics shown in FIG. 5 is actually changed with the configuration of the present invention. It is a graph.

As shown in the figure, the nakedness characteristic S1 shows that the advance angle is 0.
Degrees, S2 is 20 degrees advance, S3 is 30 degrees advance, S4
Is the result of measuring an advance angle of 40 degrees. However, the bare characteristics S1 and S2 with an advance angle of less than 30 degrees are obtained by shifting the output of the rotor position detecting means 44 of the present invention.

The bare characteristic S1 is obtained by setting the output of the rotor position detecting means 44 to an advance angle of 30 degrees and adding a fixed shift amount of -30 degrees (delayed by 30 degrees). The bare characteristic S2 is obtained by setting the output of the rotor position detecting means 44 to an advance angle of 30 degrees and adding a fixed shift amount of -10 degrees.

Further, in the experiment, the output L1 of the applied voltage discriminating means 101 is manually operated and the naked characteristics S3 and S4 are obtained.
Are measured individually.

From the result of FIG. 17, the number of revolutions is increased by 1500 rpm from the maximum value of 8000 rpm at 0 degree of advance to 9500 rpm at 40 degree of advance. As for the torque, when comparing at 7800 rpm, assuming that the maximum torque at an advancing angle of 0 degree is 1 (0.4 N · m), a torque of about 5 times (1.98 N · m) at an advancing angle of 40 degree is obtained. It has gained.

FIG. 18 is a graph showing the motor efficiency when the advance angle θ is changed at 6000 rpm, and FIG. 19 is
It is a graph which shows the inverter efficiency when advancing angle (theta) is changed at 6000 rpm.

In this case, E1 in FIG. 18 is the motor load torque of 1 N · m, and E2 is the motor load torque of 2 N · m.
E3 is the motor load torque of 3 Nm,
It is the motor efficiency when the advance angle is changed from 20 degrees to 40 degrees.

Further, in FIG. 19, F1 is a motor load torque of 1 N · m, F2 is a motor load torque of 2 N · m,
F3 is the inverter efficiency when the motor load torque is 3 N · m and the advance angle is swung from 20 degrees to 40 degrees.

From these results, the advance angle is 20 degrees to 40 degrees.
It can be seen that the motor efficiency at the time of advancing to about 1% decreases by about 1.5%, and the inverter efficiency decreases by about 0.5%, so that the total efficiency decreases by about 2%.

As can be seen from the above, in normal operation, in consideration of efficiency and the like, the output angle of the motor becomes maximum at the advance angle θ at which the motor current per output torque decreases at the rated point, that is, at the rated torque. It is preferable to drive at an advance angle θ, but even if the efficiency is sacrificed to some extent during motor operation,
When a little more rotation speed and torque are required at one point, the operating range can be expanded by switching to the advance side by the set motor applied voltage.

In this embodiment, the gain switching means is composed of an analog switch and a resistor. However, in addition to this, for example, an FET is used instead of the analog switch and the resistor, and the FET is activated. It is possible to employ a method of operating in a region and continuously changing the resistance value.

Further, although the gain is switched by the voltage applied to the motor, the gain can also be switched by the motor rotation speed and the motor current.

(Embodiment 3) Embodiment 3 of the present invention will be described below with reference to the accompanying drawings.

FIG. 20 is a circuit diagram showing another configuration example of the amplifying means (first amplifying means) 80a for one phase in the present invention. Note that the second and third amplifying means 80b and 80c are configured in the same manner as the first amplifying means 80a, and therefore description thereof will be omitted hereinafter.

As shown in the figure, the rotor position detecting means 44 in the present embodiment is n in parallel with the resistor 82 of the first amplifying means 80a of the rotor position detecting means 44 of the first embodiment. (N is an integer of 2 or more) gain switching means 901, 902. ． ． 90n are connected. These gain switching means 901, 902. ． ． 9
0n, the switching of each of the n applied voltage determination means 101, 102. ． ． 10n outputs L1, L2. ． ． It is designed to be performed at a predetermined timing based on Ln.

Next, n applied voltage discriminating means 101, 102. ． ． 10n will be described. FIG. 21 is a circuit diagram showing a configuration example of the applied voltage determining means.

As shown in the figure, the command voltage V0 for determining the desired duty ratio is n comparators 301, 30.
2. ． ． It is input to 30n. And n comparators 3
01, 302. ． ． The command voltage V0 by 30n
, And n gain switching means 901, 902. ． ． 9
N reference voltages V for setting the switching timing of 0n
1, V2. ． ． Vn are compared, and the output signals L1, L2. ． ． Ln is output.

As shown in FIG. 20, the gain switching means 901 has a resistor 911 and an analog switch 921 connected in series. Similarly, other gain switching means 902. ． 90n also has a resistance 91
2. ． 91n and analog switches 922. ． 92n are connected in series. Analog switch 92
1. ． 92n are applied voltage determining means 101. ． 10n output signal L1. ．
It is turned on and off by Ln, which causes resistors 911. ．
91n is conductive and non-conductive. Here, the resistor 81 = R3,
Resistor 82 = R4, n resistors 911. ． 91n = R51
1. ． If R51n, the output Vw-u (gain) of the first amplifying means 80a is Vw-u (gain) = (1 + Rz2 / R3) .Vw-u. ． ． (22) Rz2 = 1 / (1 / R4 + 1 / R511 + ... + 1 / R51n). ． ． (23) As a result, the line voltage that is amplified by the amplification factor determined by the ratio of (1 + Rz2 / R3) and satisfies Vw-u <Vw-u (gain) is obtained. Therefore, the n resistors 911. ． 91n =
R511. ． R51n to the analog switch 921. ．
By switching between conducting and non-conducting by 92n, the combined resistance Rz2 of the formula (23) is changed to (1 + Rz2 / R).
The amplification factor determined by the ratio of 3) can be continuously changed.

Also, n gain switching means 90
1. ． Even if 90n is connected in parallel to the resistor 81 of the first amplifying means 80a, the amplification factor can be similarly changed.

In this case, the output V of the first amplifying means 80a
wu (gain) is Vw-u (gain) = (1 + R4 / Rz3) .Vw-u. ． ． (24) Rz3 = 1 / (1 / R3 + 1 / R511 + ... + 1 / R5n). ． ． (25) As a result, the line voltage that is amplified by the amplification factor determined by the ratio of (1 + R4 / Rz3) and satisfies Vw-u <Vw-u (gain) is obtained. Therefore, the n resistors 911. ． 91n =
R511. ． R51n to the analog switch 921. ．
The combined resistance Rz3 of the formula (25) is changed by switching the conductive state and the non-conductive state by 92n, and (1 + R4 / R
The amplification factor, which is determined by the ratio of z3), can be changed. In addition,
This may be used in combination with the configuration of FIG. This makes it possible to gradually expand the operation range during the operation of the motor.

(Fourth Embodiment) A fourth embodiment of the present invention will be described below with reference to the accompanying drawings.

FIG. 22 is a circuit diagram showing another example of the structure of the applied voltage discriminating means and the circuit in the vicinity thereof in the present invention.

As shown in the figure, in this embodiment, a hysteresis circuit 110 is added to the applied voltage discriminating means 101 of the second embodiment described above.

The hysteresis circuit 110 is mainly composed of a comparator 402 and a flip-flop 501.

The command voltage V0 for determining the desired duty ratio is input to the + input terminal of the comparator 402, and the reference voltage VH1 for setting the hysteresis level is input to the-input terminal.
Is entered. Then, in the comparator 402, the command voltage V
0 and the reference voltage VH1 are compared, and when VH1 <V0, Hi
The gh voltage is output, and when VH1 ≧ V0, the Low voltage is output.

The flip-flop 501 is a D-type flip-flop with preset and clear, and has independent data (D), preset (PR), clear (CL), clock (CLK) inputs, and complementary outputs Q and NOT.
I have a Q.

Data (D) and clock (CLK) are G
It is input to ND and fixed at the low voltage. For the preset (PR), the comparator 3 of the applied voltage discrimination means 101 is used.
The output C1 of 01 is input, the output C2 of the comparator 402 is input to the clear (CL), and the output NOTQ is the first, second, and third gain switching means 90a, 90 of the next stage.
b, 90c. Each voltage value V0, V1, VH1
Is set so that V0 (maximum value)>V1> VH1.

FIG. 23 is a diagram showing an input / output relationship between the comparators 301 and 401 and the flip-flop 501.

As shown in the figure, the command voltage V0 is Low.
When rising from the level and lower than V1 and VH1 (state 1)
In addition, the output of the comparator 301 becomes the high voltage (H), the output of the comparator 402 becomes the low voltage (L), and the output NOTQ of the flip-flop 501 becomes the high voltage.

Next, when the command voltage V0 becomes higher than VH1 (state 2), the output of the comparator 301 becomes a high voltage, the output of the comparator 402 also becomes a high voltage, and the output NOTQ of the flip-flop 501 becomes high. It becomes a voltage.

Here, the case where NOTQ is the high voltage is the first gain (gain before switching), and Low
The case of voltage is the second gain (switched gain).

Further, the command voltage V0 increases and V1, VH1
When it becomes higher (state 3), the output of the comparator 301 becomes a Low voltage and the output of the comparator 402 becomes High.
And the output NOTQ of the flip-flop 501 becomes a low voltage. At this time, the gain is switched to the second gain by the gain switching means in the next stage, and the operation is performed at the second gain up to the maximum value of V0.

Next, when the command voltage V0 rises from the high level and is lower than V1 (state 4), the comparator 30
The output of 1 becomes the high voltage, the output of the comparator 402 becomes the high voltage, and the output N of the flip-flop 501 becomes
OTQ is latched at the Low voltage.

Further, the command voltage V0 rises to V1, VH1
When it becomes lower (state 5), the output of the comparator 301 becomes a high voltage and the output of the comparator 402 becomes low.
Then, the output NOTQ of the flip-flop 501 becomes a high voltage. At this time, the gain is switched to the first gain by the gain switching means in the next stage, and the operation is performed at the first gain up to the minimum value of V0.

FIG. 24 is a graph showing the TN characteristics of this embodiment. In the graph, when the maximum value of the command voltage V0 is 100%, the reference voltage V1 = 90% for setting the switching timing of the first gain switching means 90a, and the reference voltage VH1 for setting the hysteresis level. = 80%, 30 by changing the amplification factor A
The case where the advance angle is advanced from 40 degrees to 40 degrees is shown.

In FIG. 24, A1 is a duty ratio of 90%,
Motor bare characteristics at an advancing angle of 30 degrees, A2 is a duty ratio of 90%, motor bare characteristics at an advancing angle of 40 degrees, and A3 is
Motor bare characteristics at a duty ratio of 100% and an advance angle of 40 degrees; A4 is a duty ratio of 80%; motor bare characteristics at an advance angle of 40 degrees; A5 is a duty ratio of 80% and an advance angle of 30.
It is a motor naked characteristic in degrees.

As shown in the figure, when the duty ratio is increased, the motor is operated at an advance angle of 30 degrees until the duty ratio changes from 0% to 90%. That is, when the duty ratio is less than 90%, the motor is
It is operated in the area inside 1 (the origin side of the graph).

When the duty ratio becomes 90%, the advance angle becomes 3
Switching from 0 degree to 40 degrees, whereby the rotation speed of the motor rises and shifts to the bare characteristic of A2 while the duty ratio remains 90%. That is, when the duty ratio is 90%, the motor operates on A2.

Then, the duty ratio is 90% to 100.
Up to%, the motor is operated with an advance angle of 40 degrees. In this case, the motor is operated in the region between A2 and A3.

When decreasing the duty ratio,
The hysteresis circuit 110 changes the timing for switching the advance angle from 90% to 80% when the duty ratio is increased.

Therefore, when the duty ratio is decreased, the motor is operated at the advance angle of 40 degrees until the duty ratio changes from 100% to 80%. That is, when the duty ratio exceeds 80%, the motor is operated in the region between A3 and A4.

When the duty ratio reaches 80%, the advance angle becomes 4
Switching from 0 degrees to 30 degrees, whereby the rotation speed of the motor decreases and the characteristics shift to the bare characteristic of A5 while the duty ratio remains 80%. That is, when the duty ratio is 80%, the motor operates on A5.

Then, from the duty ratio of 80% to 0%, the motor is operated with the advance angle of 30 degrees. In this case, the motor is operated in the area inside A5 (on the origin side of the graph).

As described above, in this embodiment, the command voltage V0
By giving a hysteresis to the output timing of the applied voltage determination means by the rise and fall of, the operating range can be expanded and the expanded region can be operated arbitrarily.

Although the hysteresis circuit 110 is added to the applied voltage discriminating means 101 of the second embodiment in the present embodiment, the present invention is not limited to the illustrated configuration, and for example, n of the above-mentioned third embodiment is used. Individual applied voltage determining means 101. ． 1
You may add n hysteresis circuits to 0n.

(Fifth Embodiment) The fifth embodiment of the present invention will be described below with reference to the accompanying drawings.

FIG. 25 is a block diagram showing an example of the structure of the drive control means 45 of the brushless DC motor drive device of the present invention.

As shown in the figure, the drive control means 45 is
Inverter drive circuit 8 for driving the inverter 43
And an output pattern generation circuit 9 for outputting a drive signal pattern of the brushless DC motor 40 in accordance with the detection timing of the rotor position detection means 44, and a chop-on / chop-off ratio of PWM chopper control is set in accordance with the rotation speed command 11. (The ratio of chop-on and chop-off is variable) PWM control circuit (PW that outputs PWM signal P1
M generation circuit) 10, chop-on detection means 71, open phase selection means 72, and edge detection means 73.

In this embodiment, the outputs (magnetic pole position signals) Up, Vp, Wp of the rotor position detecting means 44 are respectively
It is detected by the chop-on detecting means 71 at the time of chop-on in the PWM chopper control.

Then, the open phase selecting means 72 selects the open phase of the output of the chop-on detecting means 71 in accordance with the current drive signal pattern output of the output pattern generating circuit 9.

The edge detecting means 73 holds (detects) the edge at any one desired point of the output (open phase signal) P2 of the open phase selecting means 72. The output (detection signal) P4 and PWM of this edge detection means 73
The output (PWM signal) P1 of the control circuit 10 is
It is input to the output pattern generation circuit 9.

Then, the next drive signal pattern is input from the output pattern generation circuit 9 to the inverter drive circuit 8, and based on the drive signal pattern, the inverter drive circuit 8 causes the respective transistors Ta + and Tb + of the inverter 43 to be driven. , Tc +, Ta-, Tb-, and Tc- are driven.

FIG. 26 is a circuit diagram showing a configuration example of the chop-on detecting means 71, the open phase selecting means 72 and the edge detecting means 73, and FIG. 27 is a timing chart showing the operation of the circuit shown in FIG.

As shown in these figures, the chop-on detecting means 71 has the output Up of the rotor position detecting means 44.
, Up, Vp, Wp isolated from DC power source Ed
Isolation couplers 70a, 7 for converting to s, Vps, Wps
0b, 70c and three AND circuits 101, 102, 1
AND gate 100 consisting of 03 and three EX-ORs
The EX-OR gate 110 is composed of circuits 111, 112, and 113. The detection signals Ups, Vps, Wps and the PWM signal P1 are input to the AND gate 100. From the AND gate 100, the PWM signal P1
The data signals (pulse signals) Ups +, Vps +, Wps + are output as the signal components in the chop-on period. The logical formulas are as shown in the following formulas (26a), (26b) and (26c).

Ups + = Ups · P1. ． ． (26a) Vps + = Vps.P1. ． ． (26b) Wps + = Wps · P1. ． ． (26c) The EX-OR gate 110 has the data signal Ups +,
Vps +, Wps + and the PWM signal P1 are input.
From the EX-OR gate 110, a data signal Ups- is output as a signal component in the chop-on period that is delayed by 180 degrees in electrical angle from the signal components Ups +, Vps +, and Wps + in the chop-on period of the PWM signal P1. Vps-
, Wps- are output. The logical formula is (27)
This is as shown in equations (a), (27b) and (27c).

[0171]

[Equation 2]

Here, NOT (Ups +), NOT (Vps
+), NOT (Wps +), and NOT (P1) represent negation of Ups +, Vps +, Wps +, and P1, respectively.

These data signals Ups +, Vps +, Wps +
, Ups-, Vps-, Wps- are signals corresponding to the counter electromotive voltage appearing during the chop-on period and the spike voltage.

The open phase selecting means 72 is the data selector 12
1, an AND gate 200 including three AND circuits 201, 202, and 203, and an OR gate 210.

Data signals Ups +, Vps +, Wps +, Ups
-, Vps-, and Wps- are input to the data selector 121 as data signals of the data selector 121.

On the other hand, in the open phase selecting means 72, the drive signal pattern T output from the output pattern generating circuit 9 is output.
ad +, Tbd +, Tcd +, Tad-, Tbd-, T
The open phase selection signals S1, S2, S3 are generated based on cd-.

In this case, the drive signal patterns Tad +, Tbd +, Tcd + output from the output pattern generating circuit 9 are output.
, Tad-, Tbd-, Tcd- are AND gates 2
00, and the AND gate 200 generates signals K1, K2, and K3. These signals K1, K
2, K3 are input to the OR gate 210, and the OR gate 210 generates the open phase selection signal S1. Further, as the open phase selection signal S2, the drive signal pattern Tb
d + is used, and Tcd + is used as the open phase selection signal S3. The logical expressions of K1, K2, K3 and S1 are as shown in expressions (28a), (28b), (28c) and (29), respectively.

K1 = Tad + .Tcd-. ． ． (28a) K2 = Tbd + .Tad-. ． ． (28b) K3 = Tcd + .Tbd-. ． ． (28c) S1 = K1 + K2 + K3. ． ． (29) These open phase selection signals S1, S2, S3 are respectively
The data selector 121 inputs the data signals Ups +, Vps +, Wps +, Ups-, Vps-, and Wps- to each open period according to the truth table shown in FIG. 28. The open phase signal P2 is output. The open phase signal P2 is input to the edge detecting means 73.

The edge detecting means 73 is mainly composed of a mono-multi 311 and a mono-multi 312.

The open phase signal P2 is first input to the monomulti 311. The mono-multi 311 is triggered in synchronization with the edge (rising edge) of the first pulse of the open phase signal P2, and is further retriggered in synchronization with the edges of the pulses that are sequentially input, whereby the external resistor R1 is connected. And a pulse signal P3 having a pulse width determined by the time constant t1 of the external capacitor C1.

The output of this monomulti 311, that is,
The pulse signal P3 is input to the monomulti 312. The monomulti 312 is triggered in synchronization with the edge of the first pulse of the pulse signal P3, whereby a pulse signal (detection signal) having a pulse width determined by the time constant t2 of the external resistor R2 and the external capacitor C2. ) Output P4.

Here, the open phase signal P2 is detected as a signal synchronized with the ON period of the chopping cycle of the PWM signal P1. The open phase signal P2 includes a detection edge and an edge due to a spike voltage. This spike voltage also appears in synchronization with the on period of the chopping cycle,
Due to the load operation of the brushless DC motor, the time width increases within the ON period of the chopping cycle as the motor current increases, and the pulse edge also increases.

Since the generation timing of this spike voltage coincides with the commutation timing, there is a time difference of less than the chopping cycle in the generation time between the adjacent detection edge and the edge due to the spike voltage. Therefore, Mono Multi 31
By 1, waveform shaping is performed with the open phase signal P2 having a pulse width of time constant t1. It is preferable that the setting condition of the time constant t1 of the monomulti 311 is as shown in the following formula (30).

1.5T <t1 (= C1 · R1) <2.0T. ． ． (30) where T is the chopping frequency fc of the PWM signal P1
Chopping cycle T (T = 1 / fc
) Represents.

As a result, the pulse signal P3 is obtained. The rising edge of the pulse signal P3 corresponds to the rotor position detection signal with the advance angle θ advanced by 30 degrees or more.

Here, assuming that the detection position where the advance angle θ advances by 30 degrees is shifted by 30 degrees, commutation is performed, the period from the rising edge of the pulse signal P3 to the next falling edge (pulse signal P3 Pulse width) is determined by the detection edge advanced by 30 degrees and the effect of the spike voltage appearing in the return mode after commutation.

The monomulti 312 detects only the rising edge of the pulse signal P3 and outputs a detection signal P4 having a pulse width determined by the time constant t2.

As a result, only the rising edge of the first pulse of the open phase signal P2, that is, only the first detection edge is detected, and the brushless DC motor operation always advances accurately by 30 degrees or more regardless of the presence or absence of the load. The rotor position detection signal, that is, the detection signal P4 is obtained.

As described above, in this embodiment, even if the output of the rotor position detecting means 44 includes the signal of the rotor magnetic pole position to be detected and the spike voltage appearing in the recirculation mode after commutation, the rotor is detected. It is possible to detect the magnetic pole position properly and surely, so that normal operation can always be performed.

Although the open phase is selected by the logic element in the present embodiment, the present invention is not limited to the illustrated configuration. For example, when using a microcomputer or the like, the open phase is uniquely determined by the output drive signal pattern, so the open phase selection signal is output from the microcomputer at the same time as the output of the drive pattern, and based on this open phase selection signal. ,
The data selector 121 outputs the data signals Ups +, Vps +, W
The open phase signal P2 may be output by selecting only the open period of each of ps +, Ups−, Vps−, and Wps−.

Further, the data signal may be directly taken in by the microcomputer and processed entirely in the microcomputer.

(Sixth Embodiment) A sixth embodiment of the present invention will be described below with reference to the accompanying drawings.

FIG. 29 is a block diagram showing another example of the configuration of the brushless DC motor drive device of the present invention. The description of the common points with the above-described fifth embodiment will be omitted, and the different points will be described.

As shown in the figure, in the brushless DC motor drive device of this embodiment, the brushless DC is generated from the detection signal P4 generated based on the magnetic pole position signals Up, Vp and Wp from the rotor position detecting means 44. By detecting the rotation speed of the motor 40 and setting an appropriate phase shift amount based on this rotation speed, a highly efficient and wide-range operation is possible.

In this embodiment, the control section (shift amount setting means) 400 is connected to the output side of the edge detecting means 73,
The output pattern generation circuit 9 is connected to the output side of the control unit 400. In this case, the detection signal P4 is input from the edge detection unit 73 to the control unit 400, and the control unit 400
Based on this detection signal P4, the output pattern generation circuit 9
Control the operation of.

The control unit 400 is provided with, for example, an A / D converter for converting each input signal into a digital signal, an I / O port, a timer counter (timer), a CPU, and a micro memory including a memory such as a ROM and an EEPROM. It is composed of a computer and is adapted to set an appropriate phase shift amount based on the detection signal P4 from the edge detection means 73 and the like.

In this case, the commutation signal is output at the timing determined by the finally corrected phase shift amount from the control unit 400, and based on this commutation signal, the output pattern generation circuit 9 outputs the next signal. Drive signal pattern Ta
d +, Tbd +, Tcd +, Tad-, Tbd-, Tc
d- is output.

Specifically, when the brushless DC motor having the surface magnet type rotor structure is driven (id = 0 control), for example, the detection point (detection time point) by the rotor position detection means 44 is 30 in electrical angle. Set to advance once, control unit 4
00 shifts 30 degrees (delays). As a result, driving (commutation) at a position where the advance angle is 0 degrees is always possible.

Further, in driving the brushless DC motor having the magnet embedded rotor structure, when the operating range in the high rotation region is expanded (field weakening control), for example, the following may be performed.

The detection point by the rotor position detecting means 44 is set to advance by 40 degrees or more in electrical angle, and in the low / medium rotation region, the control section 400 shifts by a predetermined angle, for example, at the position where the advance angle advances by 20 degrees. Drive and perform efficient operation utilizing reluctance torque. On the other hand, in the high rotation range,
The control unit 400 performs a predetermined angle shift according to the rotation speed. As a result, the operating range is expanded to shift 0, that is, to a position where the advance angle advances by 40 degrees or more.

The amount of phase shift (shift amount) in the low / medium rotation range should be determined based on the motor current, load torque, etc. The amount of phase shift that improves efficiency is set in the low / medium rotation range.

Next, control of commutation timing in drive control of the brushless DC motor 40 will be described.

FIG. 30 is a flow chart showing the operation of the control section 400 in one commutation cycle. Hereinafter, description will be given based on this flowchart.

First, the rotational speed of the brushless DC motor 40 (rotational speed of the rotor 42) is measured from the detection signal P4 generated based on the magnetic pole position signals Up, Vp and Wp from the rotor position detecting means 44, and The number of rotations is read (step S101).

In step S101, the interval between two adjacent detection signals P4, that is, the time from the rise of the detection signal P4 to the rise of the next detection signal P4 is measured, and the brushless DC motor is measured based on the measured value. The number of rotations of 40 is calculated.

Then, based on this rotation speed, the control unit 4
The data of the phase shift amount is read out from the memory built in 00 (step S102).

In this case, in the memory, the data of the phase shift amount corresponding to the rotation speed collected by the experiment or the like is tabulated and stored in advance, and this step S
At 102, data of an appropriate phase shift amount corresponding to the rotation speed is read from the memory. The phase shift amount data is stored as time-converted data.

Next, the timer time is set based on the read phase shift amount (step S10).
3).

Then, the timer count is started (step S104).

Next, it is determined whether or not the timer time has elapsed (time-up) (step S105).

When it is determined in step S105 that the timer time has not yet elapsed, the process returns to step S104, the timer count is continued (step S104), and it is determined again whether the timer time has elapsed (step S105). ).

If it is determined in step S105 that the timer time has elapsed, a commutation signal is output (step S106).

This completes the program (control operation) for this one commutation cycle.

As described above, in this embodiment, since the detection point by the rotor position detecting means 44 can be set so as to advance by an electrical angle of 30 degrees or more, the operating range of the brushless DC motor 40 is expanded.

Further, since the commutation point can be shifted (delayed) by a predetermined angle by the control unit 400,
It is possible to improve the operating efficiency in the low / medium speed range.

Since the phase shift amount (time for delaying commutation) is automatically set according to the number of revolutions, more precise operation control can be performed easily and reliably.

In the present embodiment, the control unit 400 mainly reads the number of revolutions, stores the data of the phase shift, counts the timer, etc., but in the present invention, in addition to these, the PWM control circuit. 10. The control unit 400 may collectively perform the operation of the output pattern generation circuit 9, the reading of the rotation speed command 11, and the processing of other digital signals.

(Embodiment 7) Embodiment 7 of the present invention will be described below with reference to the accompanying drawings.

FIG. 31 is a block diagram showing another structural example of the drive device for the brushless DC motor of the present invention. The description of the common points with the above-described sixth embodiment will be omitted, and the different points will be described.

As shown in the figure, the brushless DC motor drive device of the present embodiment has the configuration of the sixth embodiment,
It has a current detection means 401 for detecting the motor current. In this case, the motor current is detected by the current detection unit 401, and the detected value of the motor current (motor current value) is input from the current detection unit 401 to the control unit 400.

Next, commutation timing control in the drive control of the brushless DC motor 40 will be described.

FIG. 32 is a flow chart showing the operation of the control section 400 in one commutation cycle. Hereinafter, description will be given based on this flowchart.

First, the rotation speed of the brushless DC motor 40 (rotation speed of the rotor 42) is measured from the detection signal P4 generated based on the magnetic pole position signals Up, Vp and Wp from the rotor position detecting means 44, and The number of rotations is read (step P101).

In step P101, the interval between two adjacent detection signals P4, that is, the time from the rise of the detection signal P4 to the rise of the next detection signal P4 is measured, and the brushless DC motor is measured based on the measured value. The number of rotations of 40 is calculated.

Then, the motor current value is read (step P102).

Next, based on the rotation speed and the motor current value, the data of the phase shift amount is read from the memory built in the control section 400 (step P103).

In this case, in the memory, the data of the phase shift amount corresponding to the rotation speed and the motor current value collected by the experiment or the like is tabulated and stored in advance, and in step P103, the memory is stored. Data of an appropriate phase shift amount corresponding to the rotation speed and the motor current value is read from. The phase shift amount data is stored as time-converted data.

Then, the timer time is set based on the read phase shift amount (step P10).
4).

Then, the timer count is started (step P105).

Then, it is determined whether or not the timer time has elapsed (time-up) (step P106).

If it is determined in step P106 that the timer time has not elapsed, the process returns to step P105, the timer count is continued (step P105), and it is again determined whether the timer time has elapsed (step P106). ).

When it is determined in step P106 that the timer time has elapsed, a commutation signal is output (step P107).

This completes the program (control operation) for this one commutation cycle.

As described above, also in this embodiment, as in the case of the above-described sixth embodiment, the detection point by the rotor position detecting means 44 can be set so as to advance by 30 electrical degrees or more, so that the operating range of the brushless DC motor 40 is increased. Spread, and the control unit 400 shifts (delays) the commutation point by a predetermined angle.
Therefore, the operating efficiency in the low / medium speed range can be improved.

Further, in this embodiment, since the phase shift amount (time to delay commutation) is automatically set according to the rotation speed and the motor current value, more precise operation control can be performed easily and reliably. It can be carried out.

Therefore, the operating range in the high rotation speed range is expanded, and the operating efficiency is improved especially in the low / medium speed rotation speed range. In addition, high efficiency operation becomes possible over the entire operating range (when field weakening operation is not performed) (particularly advantageous in the case of a motor having an embedded magnet type rotor structure).

In this embodiment, the control unit 400 mainly reads the rotation speed, stores the phase shift data, the timer count, and reads the motor current value (A / D conversion).
However, in the present invention, in addition to the above, the control unit 400 collectively performs operations of the PWM control circuit 10 and the output pattern generation circuit 9, reading of the rotation speed command 11, and other processing of digital signals. It may be configured as follows.

[0238]

As described above, according to the brushless DC motor drive device of the present invention, the rotor position detection signal can be set so as to accurately advance by 30 degrees or more in electrical angle. For this reason, the range in which the lead angle can be set (angle range)
, Thereby expanding the operating range of the brushless DC motor and improving the efficiency.

Further, the applied voltage discriminating means and the first, second,
If the third gain switching means is included, the first,
Since the amplification factor of the line voltage by the second and third amplifying means can be changed, the operating range can be further expanded.

Further, in the case of having n applied voltage discriminating means (n is an integer of 2 or more) and n gain switching means, the operating range can be expanded stepwise during motor operation. It will be possible.

When a hysteresis circuit is attached to the applied voltage determining means, the operating range can be expanded and the expanded area can be operated arbitrarily.

Further, the signal from the rotor position detecting means is
A chop-on detecting means for detecting at the time of chop-on of PWM chopper control, and an open-phase selecting means for selecting a signal from the chop-on detecting means according to an open phase,
In the case of having the edge detecting means for detecting a predetermined edge based on the signal from the open phase selecting means, in particular, the magnetic pole position of the rotor can be detected properly and surely, thus performing normal operation. be able to.

If the rotor position detecting means detects the magnetic pole position of a predetermined rotor and outputs the commutation signal after the rotor has rotated a predetermined amount of phase shift, brushless operation can be easily performed. The operating range of the DC motor can be expanded and the efficiency can be improved.

[Brief description of drawings]

FIG. 1 is a block diagram showing a conventional sensorless brushless DC motor drive device.

FIG. 2 is a circuit diagram showing phases of a rotor position detecting means of a conventional sensorless brushless DC motor driving device.

FIG. 3 is a plan view schematically showing a surface magnet type rotor.

FIG. 4 is a plan view schematically showing an embedded magnet type rotor.

FIG. 5 is a graph showing an advance angle θ-torque T characteristic when maximum torque control is performed.

FIG. 6 is a block diagram showing a configuration example of a brushless DC motor drive device of the present invention.

FIG. 7 is a diagram showing an excitation pattern of a brushless DC motor according to the present invention.

FIG. 8 is a back electromotive force ea of U phase, V phase, and W phase when the advance angle is 0 degree in the brushless DC motor according to the present invention;
It is a figure which shows the relationship between eb and ec, and a drive signal.

FIG. 9 shows Ta + -Tb- in the chopper control of the present invention.
FIG. 6 is a circuit diagram showing an equivalent circuit when is conductive.

FIG. 10 is a diagram showing a detection timing of a back electromotive force in the present invention.

FIG. 11 is a circuit diagram showing a configuration example of rotor position detecting means in the present invention.

FIG. 12 is a diagram showing the detection timing of rotor position detection in the present invention.

13 is a timing chart showing signal waveforms of various parts in the circuit diagram shown in FIG.

FIG. 14 is a block diagram showing a main part of another configuration example of the rotor position detecting means in the present invention.

FIG. 15 is a circuit diagram showing a configuration example of an applied voltage determining means in the present invention.

FIG. 16 is a circuit diagram showing a configuration example of a photo MOS switch according to the present invention.

FIG. 17 is a graph showing a motor bare characteristic according to the present invention.

FIG. 18 is a graph showing motor efficiency in the present invention.

FIG. 19 is a graph showing the inverter efficiency in the present invention.

FIG. 20 is a circuit diagram showing another configuration example of the first amplifying means in the present invention.

FIG. 21 is a circuit diagram showing another configuration example of the applied voltage determining means in the present invention.

FIG. 22 is a circuit diagram showing another configuration example of the applied voltage determining means and circuits in the vicinity thereof in the present invention.

FIG. 23 is a diagram showing a relationship between inputs and outputs of a comparator and a flip-flop in the present invention.

FIG. 24 is a graph showing TN characteristics in the present invention.

FIG. 25 is a block diagram showing a configuration example of drive control means in the present invention.

FIG. 26 is a circuit diagram showing a configuration example of chop-on detection means, open-phase selection means, and edge detection means in the present invention.

27 is a timing chart showing the operation of the circuit shown in FIG.

FIG. 28 is a diagram showing a truth table according to the present invention.

FIG. 29 is a block diagram showing another configuration example of the drive device for the brushless DC motor of the present invention.

FIG. 30 is a flowchart showing the operation of the control unit in one commutation cycle.

FIG. 31 is a block diagram showing another configuration example of the drive device for the brushless DC motor of the present invention.

FIG. 32 is a flowchart showing the operation of the control unit in one commutation cycle.

[Explanation of symbols]

 1 Commercial Power Supply 2 Rectifier Circuit 3 Voltage Inverter 4 Brushless DC Motor 5 Rotor Position Detection Means 6 Stator Armature Winding End 7 Drive Control Means 8 Inverter Drive Circuit 9 Output Pattern Generation Circuit 10 PWM Control Circuit 11 Rotation Speed Command 12 Yoke 13 Field permanent magnet 14 Sleeve 15 Yoke 16 Field permanent magnet 20 Voltage dividing circuit 21, 22 Resistor 23 Capacitor 24 First-order lag filter circuit 25 Resistor 26 Capacitor 27 Comparison circuit 28-30 Resistor 31 Comparator 32 Input end 33 Output end 34 reference voltage 35 reference voltage 40 brushless DC motor 41 stator 42 rotor 43 voltage type inverter 44 rotor position detecting means 45 drive control means 50a first line voltage generating means 50b second line voltage generating means 50c third line Voltage generator 51-54 resistance 55 amplifier 56-59 resistance 60, 65 amplifier 61-64 resistance 61-64 resistance 66a 1st comparison means 66b 2nd comparison means 66c 3rd comparison means 70a, 70b, 70c isolation coupler 71 chop-on Detecting means 72 Open phase selecting means 73 Edge detecting means 80a First amplifying means 80b Second amplifying means 80c Third amplifying means 81,82 Resistor 83 Amplifier 84,85 Resistor 86 Amplifier 87,88 Resistor 89 Amplifier 90a First Gain switching means 90b Second gain switching means 90c Third gain switching means 91, 93, 95 Resistors 92, 94, 96 Analog switch 101-10n Applied voltage discriminating means 100 AND gates 101-103 AND circuit 110 EX-OR Gate 111-113 EX- R circuit 121 Data selector 200 AND gates 201 to 203 AND circuit 204 Triangular wave generation circuit 205 Comparator 206 LED 207 MOSFET 208 Comparator 210 OR gates 301 to 30n Comparator 311, 312 Monomulti 400 Control unit 401 Current detection means 402 Comparison Device 501 flip-flop S101 to S106 steps P101 to P107 steps

Claims (12)

[Claims] 1. A 120-degree conduction type inverter comprising a stator in which three-phase armature windings U, V, W are star-connected, a rotor forming a magnetic pole pair by permanent magnets, and a plurality of semiconductor switching elements. And rotor position detection means for detecting a terminal voltage generated at the armature winding end of the stator and generating a signal corresponding to the magnetic pole position of the rotor,
A drive device for a brushless DC motor, comprising: a drive control unit that performs speed adjustment by PWM chopper control by the inverter based on a signal from the rotor position detection unit, wherein the rotor position detection unit is an armature winding of the stator. First line voltage generating means for generating a line voltage Vw-u between the armature windings W and U based on the terminal voltage at the line ends, and a line voltage Vu between the armature windings U and V. Second line voltage generating means for generating -v, third line voltage generating means for generating a line voltage Vv-w between the armature windings V-W, and the first line voltage Line voltage V output from the generation means
a first amplifying means for amplifying a signal relating to w-u, and a line voltage V outputted from the second line voltage generating means.
second amplifying means for amplifying a signal relating to u-v, and the line voltage V outputted from the third line voltage generating means.
third amplifying means for amplifying a signal regarding vw, first comparing means for comparing a signal regarding the line voltage Vw-u with a signal output from the second amplifying means, Second comparing means for comparing a signal relating to the voltage Vu-v and a signal outputted from the third amplifying means, a signal relating to the line voltage Vv-w and a signal outputted from the first amplifying means And a third comparing means for comparing the following. 2. When the phase angle of the current flowing through the armature winding of the stator with respect to the q axis in the dq coordinate system is the current phase angle θ, the rotor position detecting means determines that the current phase angle θ is electric. The brushless DC motor drive device according to claim 1, wherein the drive device is configured to detect a predetermined magnetic pole position of the rotor advanced by 30 degrees or more in angle. 3. The first comparison means includes the line voltage V
When the signal relating to w-u is larger than the signal output from the second amplifying means, a high level signal is output, and the second comparing means outputs the signal relating to the line voltage Vu-v to the first signal. And outputs a high level signal when the signal is larger than the signal output from the amplifying means of No. 3, and the third comparing means is
The brushless DC motor according to claim 1 or 2, which is configured to output a high-level signal when a signal related to the line voltage Vv-w is larger than a signal output from the first amplifying means. Drive. 4. The rotor position detecting means includes an applied voltage determining means for determining a motor applied voltage, and the first, second and third means based on a signal from the applied voltage determining means.
4. The brushless DC motor drive device according to claim 1, further comprising first, second and third gain switching means for changing the gain of the amplifying means. 5. The rotor position detecting means includes n (n is an integer of 2 or more) applied voltage determining means for determining a motor applied voltage, and the first, based on a signal from the applied voltage determining means. 4. The brushless DC motor drive device according to claim 1, further comprising n number of gain switching means for changing the gains of the second and third amplifying means. 6. The brushless DC motor drive device according to claim 4, wherein a hysteresis circuit is attached to the applied voltage determining means. 7. The drive control means detects a signal from the rotor position detection means at the time of chop-on of PWM chopper control, and a signal from the chop-on detection means in an open phase. 7. The brushless DC motor drive device according to claim 1, further comprising: an open phase selecting means for selecting in combination and edge detecting means for detecting a predetermined edge based on a signal from the open phase selecting means. . 8. A commutation signal is output in synchronization with the detection of a predetermined magnetic pole position of the rotor by the rotor position detecting means.
8. The brushless DC motor drive device according to any one of 1 to 7. 9. A commutation signal is output after the rotor has rotated a predetermined amount of phase shift after the predetermined magnetic pole position of the rotor is detected by the rotor position detecting means. 8. The brushless DC motor drive device according to any one of 1 to 7. 10. The drive device for the brushless DC motor according to claim 9, further comprising shift amount setting means for setting the phase shift amount. 11. The brushless DC motor drive device according to claim 10, wherein the setting of the phase shift amount by the shift amount setting means is changed at least according to the rotation speed of the rotor. 12. The brushless DC motor drive device according to claim 10, wherein the setting of the phase shift amount by the shift amount setting means is changed according to the rotation speed of the rotor and the motor current.