JPH0767302B2 - Brushless motor drive circuit - Google Patents

Description

The present invention relates to a drive circuit for a brushless motor.

(Prior Art) A drive circuit of a brushless motor that employs a current drive system is conventionally known. Since this current-driven brushless motor is driven by a constant current, there is a problem that torque ripple occurs. Therefore, as an example of a drive circuit for a brushless motor capable of correcting this torque ripple, a circuit as shown in FIG. 8 has been proposed (Japanese Patent Laid-Open No. 59-76192).

The drive circuit of the brushless motor supplies the command signal from the command signal generating means 1 to the current supplying means 2 and the command current generating means 3. The current value flowing through the drive coil 4 is detected by the coil current detecting means 5 and given to the synthesizing means 6. The synthesizing means 6 compares the current signal from the command current generating means 3 with the current value flowing through the drive coil 4 detected by the coil current detecting means 5, and outputs a current error signal such that the current value matches the current signal. Is forming. The current error signal formed by the synthesizing means 6 is supplied to the means 7 for changing the supply current.

Due to such a current feedback function, the means 7 for changing the supply current is such that the drive coil is always driven with a constant current value when the motor command signal from the command signal generating means 1 is constant. Become.

Due to the function of this current feedback function, particularly in the case of a brushless motor, even if there is an interphase imbalance or the like on the means 7 side that changes the supply current, it is possible to suppress torque fluctuations due to this. However, as a fate when driving a motor with a constant current, a torque ripple of the least common multiple of the number of stator driving coil poles and the number of rotor magnetic poles is about 14% of the total torque per motor revolution. To do.

Therefore, in the drive circuit of this brushless motor, the state of the motor movable portion 8 is detected by utilizing the hall element of the modulation signal generating means 9 to form a signal having a phase substantially opposite to that of the torque ripple, and the command current generating means is generated. Modulation means for current value from 3
The torque ripple is corrected by superimposing at 10 at an appropriate ratio and giving it to the synthesizing means 6 as a modulated current signal, and performing current feedback while eventually pulsating the motor current in the direction in which the torque ripple is canceled. Is going. still,
Other conventional brushless motor drive circuits that can correct torque ripple include Japanese Patent Publication No. 56-34551 and Japanese Patent Laid-Open No. 58-1.
92490 and JP-A-59-35585.

(Problems to be Solved by the Invention) The drive circuit of each brushless motor is provided as an integrated circuit. However, the addition of a torque ripple correction circuit and the like increases the number of elements inside the integrated circuit, and It has the drawback of becoming complicated and large.

Further, since the drive circuit of the conventional brushless motor forms the torque ripple correction signal by utilizing the waveform of the output signal of the hall element that is the rotor position detecting means, variations in the hall element output waveform, distortion, It is easily affected by temperature characteristics and the like, and the torque ripple correction effect is affected by the accuracy of the Hall element.

It is an object of the present invention to provide a drive circuit for a brushless motor that does not complicate the circuit configuration and that is not affected by the accuracy of the detection means for forming the torque ripple correction signal.

(Means for Solving the Problem) In order to achieve such an object, a drive circuit of a brushless motor according to the present invention includes a stator having an m-phase drive coil, a rotor having magnetic poles, the stator and the rotor. Position detecting means for obtaining an m-phase sine wave-like output signal according to the relative positional relationship with and a rectangular wave pulse-like waveform obtained by logarithmically compressing the output signal of the position detecting means to round the inflection point. Signal processing means for molding and synthesizing m-phase soft switching signals, positive side / negative side switching element groups for switching energization to the m-phase drive coil according to the output signal of the signal processing means, and the positive side. Current control means for controlling the energization amount to the drive coil by the negative side switch element group, and only the both positive side / negative side switch element groups without flowing into the drive coil together with the current flowing in the drive coil And a current detection unit connected to the current control unit so as to give negative feedback to the current control unit, and the signal processing unit includes a current supplied to the drive coil and a reactive current in the current detection unit. It is designed to output a signal that causes the flow.

Further, the signal processing means of the present invention conducts m-phase 120-degree energization, and can simultaneously energize the positive-side and negative-side switching element groups at the energization timing of each non-energized area to the drive coil of each phase. A reactive current is passed through the current detecting means by outputting a signal.

Furthermore, the present invention is characterized in that, in the above configuration, the fifth harmonic component is superimposed on the counter electromotive voltage waveform generated in the drive coil.

(Operation) With the configuration as described above, the drive circuit of the brushless motor of the present invention drives the current detection means by operating the positive side / negative side switch element group with the output signal from the signal processing means. A reactive current is made to flow in addition to the current passed through the coil. The current detection unit detects the current flowing through the drive coil and also detects the reactive current that does not flow through the drive coil and flows only through the positive and negative side switch element groups. The current supplied to the drive coil and the reactive current detected by the current detection means are negatively fed back to the current control means, and the current control means controls the drive of the positive side / negative side switch element groups so that a constant current flows. Here, since a constant current flows through the current detecting means, the total torque of the motor is reduced by the reactive current that does not flow through the drive coil. This phenomenon is the same for all m phases. Therefore,
The reduction of the torque due to the reactive current suppresses the peak of the combined torque, so that the torque ripple is reduced.

Further, the signal processing means of the present invention performs m-phase 120-degree energization, and at the same time the energization timing becomes a non-energized region to the drive coil of each phase as a constant energization section, the positive side and negative side switching element groups. It is possible to obtain an output signal that simultaneously turns on the power. Therefore, the positive-side and negative-side switching element groups of the same phase are simultaneously energized at the energization timing in the non-energized area to the drive coil of each phase according to the output signal from the signal processing means. As a result, a current flows through the path of power source positive electrode-in-phase positive side switching element-in-phase negative side switching element-power source negative electrode, and no current flows in the drive coil.

Further, the torque ripple is further reduced by superimposing the fifth harmonic component on the back electromotive force waveform generated in the drive coil while flowing the reactive current as described above.

(Example) Hereinafter, the structure of the present invention will be described in detail based on an example shown in the drawings. In the present specification, the positive side and the negative side are focused on the current flowing through the drive coil, and the one entering the coil side is the positive side and the one flowing out from the coil side is the negative side.

FIG. 1 is a block diagram showing an embodiment of the present invention. The embodiment shown in FIG. 1 has a three-phase 120 having three position detecting means.
Is an example of a soft switching energization method. Also, the second
(I) and (II) are waveform charts showing a signal detected by the same position detecting means and a signal processed by the same signal processing means.

In FIG. 1, the drive circuit of the brushless motor includes a stator (not shown), a rotor (not shown), Hall elements 11u, 11v, 11w as position detecting means, and Hall amplifier circuits 16u, 1
6v, 16w and the signal processing means 20 consisting of the signal synthesis circuit 12,
A switching element group 13a on the positive side and a switching element group 13b on the negative side, a current detection resistor 14 which is a current detection means,
At least a current control means 15 is provided.

The above elements will be described below.

In this embodiment, the stator is m as shown in FIG.
It has drive coils Lu, Lv, and Lw of phases (= 3, the same applies hereinafter). Although not shown, the rotor has magnetic poles. The Hall elements 11u, 11v, 11w are the second ones according to the relative positional relationship between the stator having the three-phase drive coils Lu, Lv, Lw and the rotor.
It is an element that outputs three-phase sinusoidal signals Vu, Vv, Vw as shown in FIG.
It is input to u, 16v, 16w. Hall amplifier circuit 16u, 16v, 1
Each signal logarithmically compressed with 6w
Is being supplied to. The signal processing means 20 including the hall amplifier circuits 16u, 16v, 16w and the signal synthesizing circuit 12 includes a hall element 11
Output signals Vu, Vv, Vw from u, 11v, 11w are logarithmically compressed to form a rectangular wave pulse with blunted inflection points, and three-phase soft switching as shown in Fig. 2 (II) Can be combined into a signal. The output signal of the signal synthesis circuit 12 is a signal circuit.
17a and the signal circuit 17b. Signal circuit 17a, 17b
Is a circuit that obtains the signals Sda and Sdb by adjusting the amplitudes of the three-phase soft switching signals according to the current error signal Sc. The output signal Sda from the signal circuit 17a is supplied to the positive side switching element group 13a, and the signal Sdb from the signal circuit 17b is supplied to the negative side switching element group 13b. The switching element group 13a on the positive side includes the signal synthesis circuit 12 to the signal circuit 1
The signal Sda output via 7a switches energization to the three-phase drive coils Lu, Lv, Lw. Further, the switching element group 13b on the negative side includes the signal synthesis circuit 12 to the signal circuit 1
The signal Sdb output via 7b switches the energization to the three-phase drive coils Lu, Lv, Lw. Positive side switching element group 13a, in this embodiment, the transistors Q ₃₁ ~
It consists of Q ₃₃ . The switching element group 13b on the negative side is
In this embodiment, it is constituted by transistors Q ₃₄ to Q _36. In the positive-side switching element group 13a and the negative side switching element group 13b, in the above embodiment is constituted by the transistors Q ₃₁ to Q _36, field effect transistors, may be constituted by a thyristor and the like.

The current detection resistor 14 does not flow in the drive coils Lu, Lv, Lw together with the current flowing in the drive coils Lu, Lv, Lw, and flows only in both the positive side and negative side switch element groups 13a, 13b to generate torque. A current detection unit that detects a reactive current that does not contribute is configured. The current control means 15 uses the positive side switching element group 13a and the negative side switching element group 13b to drive the drive coils Lu, Lv, L
It is configured so that the amount of current supplied to w can be controlled. For example, a current feedback amplifier is adopted. In this embodiment, the current control means 15 compares the detection signal from the current detection resistor 14 with the current reference value that should be passed through the drive coils Lu, Lv, Lw and energizes the drive coils Lu, Lv, Lw. Although the current error signal Sc for controlling the amount is obtained and given to the signal circuit 17a and the signal circuit 17b, the configuration is not limited to such a feedback system configuration, and the drive coils Lu and L
Any configuration may be used as long as it can control the amount of electricity supplied to v and Lw.

Since it is configured as described above, it operates as follows.

Three-phase sinusoidal output signals Vu, Vv, Vw as shown in FIG. 2 (I) are generated from the Hall elements 11u, 11v, 11w according to the relative positional relationship between the stator and the rotor. . Hall element 11u,
The output signals Vu, Vv, Vw from 11v, 11w are the Hall amplifier circuit 16
Input to u, 16v, 16w. The signals from the Hall amplifier circuits 16u, 16v, 16w are input to the signal combining circuit 12. As a result, the output signals Vu, Vv, Vw from the Hall element are logarithmically compressed and shaped like a rectangular wave pulse with an inflection point blunted, and three-phase software as shown in FIG. It is combined with the switching signals (V _U , V _V , V _W ). Further, the combined three-phase soft switching signal is independently adjusted in amplitude in accordance with the current error signal Sc in the signal circuit 17a to be the positive signal Sda, and at the same time in the signal circuit 17b to be the current error signal Sc. The signal on the negative side is independently adjusted according to the amplitude.
Will be Sdb. The positive side signal Sda from the signal circuit 17a is given to the positive side switching element group 13a, the negative side signal Sdb from the signal circuit 17b is given to the negative side switching element group 13b, respectively, and the three-phase drive coil Lu, Switch the power supply to Lv and Lw. The current flowing through the drive coils Lu, Lv, Lw is detected by the current detection resistor 14.

In the current control means 15, the detection signal at the current detection resistor 14 and the current reference value V _CTL to be passed through the drive coil Lu, Lv, Lw are compared to determine the amount of electricity supplied to the drive coil Lu, Lv, Lw. A current error signal Sc for controlling is formed. This current error signal Sc
Is supplied to the signal circuit 17a and the signal circuit 17b.

In this embodiment, the reactive current flows through the current detection resistor 14 while operating as described above. This reactive current is applied to the positive side switching element group 13a and the negative side switching element group 13b in a certain conduction section, that is, the timing region.
It is obtained by energizing both at the same time. And
In this embodiment, as described above, a reactive current is passed through the current detection resistor 14, and a part of the current flowing through the drive coils Lu, Lv, Lw is cut off, without using a special circuit configuration, and a Hall element. Etc. without being affected by the accuracy of position detection means such as
The torque ripple can be reduced.

Further, the present invention will be described in more detail with reference to the circuit diagram of FIG. In addition, FIG. 4 to FIG.
It is a waveform diagram for explaining the operation of the same embodiment.

In Figure 3, holes amplifier circuit 16u, 16v, 16w and the signal synthesizing circuit 12 includes a transistor Q ₁ to Q _6, the current source circuit I ₀
And a resistor R ₀ and diodes D ₂₁ and D ₂₂ . Signal circuit 17a is composed of a transistor Q ₇ to Q ₁₂ and the variable current source circuit I _CTL. Signal circuit 17b, the transistor Q ₁₃ to Q ₁₈ and the variable current source circuit I _CTL and a diode D ₁₃ to D
_It consists of ₁₅ and a resistor R ₁₅ . The switching element group 13A on the positive side is a transistor Q ₃₁ connected in Darlington.
~ Q ₃₃ , Q ₃₇ ~ Q ₃₉ , the negative side switching element group 13b is a Darlington-connected transistor Q ₃₄ ~
It consists of Q ₃₆ and Q _{40 to} Q ₄₂ .

As shown in FIG. 2 (I), the hall elements 11u, 11v, and 11w temporarily output sine wave signals having a phase difference of 120 degrees. When the output signal of the hall element 11u is Vu, the output signal of the hall element 11v is Vv, and the output signal of the hall element 11w is Vw, Vu = 0.2sinθ (1) Vv = 0.2sin (θ-120 °) (2 ) Vw = 0.2sin (θ-240 °) ... (3) , and the three-phase Hall amplifier 16u they consist each transistor Q ₁ to Q _6, 16v, is input to 16w.

Further, in the signal synthesizing circuit 12, the collector currents of the transistors Q ₁ and Q ₄ , the collector currents of the transistors Q ₂ and Q ₅ , and the collector currents of the transistors Q ₃ and Q ₆ are synthesized with each other, and the voltage is converted by the resistor R _0. To do. The voltage waveform at this time is as shown in Fig. 2 (II), and the voltages V _U , V _V ,
V _W is expressed by the following equations (4) to (6).

Where e is the base of the natural logarithm, K is the Boltzmann constant [J / °
K] and T are absolute temperatures [° K], q is electron charge [C], and the same applies hereinafter.

These voltages V _U , V _V , and V _W are supplied to the differential amplifier circuit including the NPN transistors Q _{7 to} Q ₉ in the signal circuit 17a and the P amplifier in the signal circuit 17b.
And a differential amplifier circuit composed of NP transistors Q _{16 to} Q ₁₈ . The signal circuit 17a supplies the currents I'u, I'v, I'w proportional to the variable current source circuit I _CTL from the collectors of the NPN transistors Q _{7 to} Q ₉ , and the signal circuit 17b proportional to the variable current source circuit I _CTL . The currents Iu, Iv, and Iw are output from the collectors of PNP transistors Q _{16 to} Q ₁₈ . Current proportional to this variable current source circuit I _CTL
Iu, Iv, Iw are expressed by the following equations (7) to (9), and the currents I′u, I ′
v and I′w are expressed by the following equations (10) to (12).

Current I'u, I'v, I'w, the transistor Q ₃₁ ~Q _33, Q ₃₇
To Q ₃₉ are inputted to the switching element group 13a on the positive side of the current amplification factor α, and multiplied by α, respectively, the current αI′u,
αI′v, αI′w are made to flow in as source currents of the three-phase drive coils Lu, Lv, Lw.

Current Iu, Iv, Iw also transistor _{Q 34 ~Q 36, Q 40 ~Q} 42 formed by the input to the negative side of the switching element group 13b of the current amplification factor α from the current is α multiplied respectively αIu, αIv, αIw Out as a sink current from the three-phase drive coils Lu, Lv, Lw.

Of the source current and the sink current of each phase thus obtained, for example, the U-phase currents αI′u and αIu are shown in the waveform diagram of FIG. 4 (note that the V-phase current and the W-phase current are, respectively). It becomes the same waveform with a phase difference of 120 degrees). As can be seen from the waveform diagram of FIG. 4, there is an energization timing (indicated by t ₀ in FIG. 4) in the non-energized region at regular intervals.
The point that there is an energization timing that is this non-energized area is
The same applies to the V phase and the W phase. In this way, the rotor rotates by sequentially switching the energization of the three-phase drive coils Lu, Lv, Lw according to the output signals Vu, Vv, Vw of the Hall elements 11u, 11v, 11w as the rotational position detecting means. .

On the other hand, the currents flowing through the three-phase drive coils Lu, Lv, Lw are collected in the current detection resistor 14 and converted into the voltage E ₁ . This voltage
E ₁ is used as a current detection signal and is supplied to the current control means 15. The current control means 15, the voltage E _1, compared with the current reference value V _CTL, the signal circuit 17a so that the voltage E ₁ is always equal to the current reference value V _CTL, 17b variable current source circuit I of
Control _CTL . Then, when the current reference value V _CTL maintains a constant value, the voltage E ₁ generated in the current detection resistor 14 is also fixed, and a constant current is always supplied to the drive coils Lu, Lv, Lw. In this way, the brushless motor will rotate.

Here, assuming that the three-phase back electromotive force generated in the drive coils Lu, Lv, Lw is a sine wave, the torques T _u , T _v , T _w generated in each phase can be expressed as follows.

T _U ∝ (αI′u−αIu) sin (θ + 30 °) T _V ∝ (αI′v−αIv) sin (θ−90 °) T _w ∝ (αI′w−αIw) sin (θ−210 °) ... (13) Therefore, the combined torque T is T = T _u + T _v + T _w (14) The combined torque T is as shown in FIG. 4, and the torque ripple at this time is about 14.3%. At this time, the voltages V _U , V _V , and V output from the signal synthesis circuit 12
_W is represented by the equations (4) to (6), and the amplitudes of the voltages V _U , V _V , and V _W were 400 [mV _PP ].

Next, in the signal synthesis circuit 12, adjust the resistance R ₀ or
Alternatively, by adjusting the value of the current source circuit I ₀ , the amplitudes of the voltages V _U , V _V , and V _{W in} the expressions (4) to (6) can be adjusted to 100 [mV
_PP ], it operates as follows.

In this way, when the amplitudes of the output voltages V _U , V _V , and V _W of the signal synthesis circuit 12 are set to 100 [mV _PP ], the current I ′ from the signal circuit 17a is changed.
The waveforms of u, I'v, I'w and the currents Iu, Iv, Iw from the signal circuit 17b are further rounded. Therefore, the source current αI′u, which flows in due to the switching element group 13a on the positive side,
The waveforms of αI′v, αI′w and the sink currents αIu, αIv, αIw flowing out by the negative side switching element group 13b change. Source current αI'u, αI'v, αI '
FIG. 5 shows the waveforms of w and the sink currents αIu, αIv, αIw focusing on the U phase. As can be seen from FIG. 5, there is a timing region t _{0 in} which the source current αI′u and the sink current αIu occur simultaneously. Before and after the center point of the timing region t ₀ , αI′u≈αIu, and the U-phase current at this time hardly flows through the drive coil Lu and the power supply V _CC passes through the transistors Q ₃₁ and Q _34. It will flow into the current detection resistor 14. This is
This means that a relatively large amount of reactive current that does not contribute to torque generation is generated. Further, if the current feedback that causes a constant current to flow is applied, the current that flows through the current detection resistor 14 is constant, so the total torque of the motor decreases by the amount of this reactive current. This phenomenon is the same for the V phase and the W phase, and occurs once every 60 degrees, and as can be seen from FIG. 5, it is synchronized with the peak period of the combined torque. Therefore, the reduction of the torque due to the reactive current suppresses the peak of the combined torque, and the torque ripple is reduced. In this case, the torque ripple is reduced to 9.7%, which is about 32% better than the conventional one.

FIG. 6 is a characteristic diagram showing the relationship of the torque ripple [%] with respect to the amplitudes [mV _PP ] of the voltages V _U , V _V , and V _{W in} the expressions (4) to (6).

By changing the value of the resistor R ₀ of the signal synthesizing circuit 12 or the current source circuit I ₀ , the voltage V _{U of} the equations (4) to (6),
When the amplitude [mV _PP ] of V _V and V _W is changed, the torque ripple [%] changes as shown in FIG. As can be seen from the figure, when changing from 20 [mV _PP ] to 600 [mV _PP ], the torque ripple improves at 300 [mV _PP ] or less, reaching the minimum value around 80 [mV _PP ], and thereafter. Turn to rise. Therefore, by changing the value of the resistor R ₀ of the signal synthesis circuit 12 or the current source circuit I ₀ ,
By adjusting the amplitudes of the voltages V _U , V _V , and V _{W in} the equation (6) to around 80 [mV _PP ], the torque ripple will be greatly improved.

As can be seen from the above, this embodiment does not require a new additional circuit for correcting the torque ripple,
It is only necessary to adjust the resistance R ₀ of the signal synthesis circuit 12 or the current source circuit I ₀ . The Hall elements 11u, 11v, 11w use the Hall amplifier circuits 16u, 16v, 16w in the saturation region, so they are irrelevant to the torque ripple correction function and affect element variations and Hall output accuracy. Do not receive

As described above, in the present embodiment, since an additional circuit is not newly required for torque ripple correction, the circuit scale does not increase and the cost does not increase. Further, in this embodiment,
Since the torque ripple correction signal is not formed by the hall elements 11u, 11v, 11w, it is not affected by the accuracy of the hall elements 11u, 11v, 11w. In this embodiment, the current feedback function itself is basically unchanged, and the circuit is stable. Although not shown, in the case of a large brushless motor, a pre-driver may be provided in front of each switching element group 13a, 13b.

FIG. 7 is a characteristic diagram shown to explain another embodiment of the present invention, in which the horizontal axis shows the fifth harmonic component n [%] of the back electromotive voltage waveform, and the vertical axis shows the torque ripple. [%] Is shown respectively.

5th in improving torque ripple of brushless motor
It has already been proposed to generate a counter electromotive voltage Ea containing a harmonic component (n / 100) sin5θ in the drive coils Lu, Lv, Lw. The back electromotive force Ea is expressed as follows.

Ea = sin θ + (n / 100) sin5 θ (15) By including the fifth harmonic component in the counter electromotive voltage sin θ, the torque ripple is reduced. Further, it will be specifically described with reference to FIG. FIG. 7 shows the state of the torque ripple when the fifth harmonic component n [%] is changed. As can be seen from this figure, by changing the ratio of the fifth harmonic component n contained in the counter electromotive voltage Ea, the torque ripple is reduced as shown in Graph A.

Therefore, when the present invention is applied to the brushless motor,
As shown in the graph B, the minimum value of the torque ripple is cut by 1%. Moreover, the fifth harmonic distortion applied to the back electromotive voltage waveform may be small.

As described above, in the present embodiment, the minimum value of the torque ripple may be smaller than that of the conventional one, and the fifth value given to the counter electromotive voltage waveform may be used.
Since the harmonic distortion is small, it is not necessary to force the magnets of the rotor to be corrected and magnetized, and the efficiency of the motor is improved. In particular, a multi-pole circumferentially opposed motor is effective because there is no room for correction magnetization.

(Effect of the Invention) As is apparent from the above description, according to the present invention, the current flowing in the drive coil does not flow into the drive coil and only the positive-side and negative-side switching element groups of the same phase flow only by the operation of the signal synthesis circuit output. Since the torque generated in the rotor is reduced by the reactive current that does not flow in the drive coil, it is possible to suppress the peak of the combined torque and reduce the torque ripple. it can. Moreover, it is not necessary to enlarge the circuit scale in order to reduce the torque ripple.

According to the present invention, since the torque ripple correction signal is not formed by the detection signal from the rotational position detecting means, it is not affected by the accuracy of the rotational position detecting means.

Further, according to the present invention, the positive side and negative side in-phase switching elements are simultaneously energized at the energization timing in the non-energized region. The reactive current can be surely passed through the detection means.

Furthermore, according to the present invention, the drive of the brushless motor in which the positive and negative switching elements of the same phase are simultaneously energized in a constant energizing section by the signal synthesizing circuit to cause the current detecting means to pass the reactive current. Since the fifth harmonic is superimposed on the counter electromotive voltage generated in the drive coil in the circuit, the torque ripple can be remarkably improved.

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of the present invention, FIGS. 2 (I) and (II) are waveform diagrams for explaining the same embodiment, and FIG. 3 is a circuit showing a concrete example of the same embodiment. FIG. 4, FIG. 4 to FIG. 6 are waveform diagrams and characteristic diagrams for explaining the operation of the same example, FIG. 7 is a characteristic diagram for explaining another embodiment of the present invention, and FIG. FIG. 3 is a block diagram showing a drive circuit of FIG. 11u, 11v, 11w ... Hall element (rotational position detection means), 12
...... Signal synthesis circuit, 13a ...... Positive side switching element group, 13b ...... Negative side switching element group, 14 ...... Current detection resistor (current detection means), 15 ...... Current control means, 20 ......
Signal processing means.

Claims (3)

[Claims] 1. A stator having an m-phase drive coil, a rotor having magnetic poles, and a position for obtaining an m-phase sine wave-like output signal according to a relative positional relationship between the stator and the rotor. The detection means, signal processing means for logarithmically compressing the output signal of the position detection means to shape the waveform into a rectangular wave pulse having an inflection point blunted, and to synthesize it into an m-phase soft switching signal; and the signal processing means. A positive side / negative side switching element group that switches energization to the m-phase drive coil according to the output signal of, and a current control unit that controls the energization amount to the drive coil by the positive side / negative side switching element group. A current detection connected to the current control means so as to detect a reactive current flowing only in the positive and negative side switching element groups without flowing in the drive coil together with the current flowing in the drive coil, and to give negative feedback to the current control means. And a stage, the signal processing means,
A drive circuit for a brushless motor, wherein a signal for causing a current to be applied to the drive coil and a reactive current to flow to the current detection means is output. 2. The signal processing means energizes m-phase 120 degrees, and simultaneously energizes the positive-side and negative-side switching element groups at the energization timing of each non-energized area to the drive coil of each phase. 2. The drive circuit for the brushless motor according to claim 1, wherein a reactive current is supplied to the current detecting means by outputting a signal that can be generated. 3. A drive circuit for a brushless motor according to claim 1, wherein a fifth harmonic component is superimposed on the back electromotive voltage waveform generated in the drive coil.