JPH0452646B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a current amplifier that obtains an output current corresponding to a voltage difference between input signals applied to a pair of differential input terminals, and is particularly suitable for integration into an integrated circuit.

For example, it is possible to realize an instrumentation amplifier that removes the common mode component from a signal transmitted as a balanced signal and converts it into an unbalanced signal, a voltage-to-current converter whose gain can be electrically controlled, a voltage amplifier, or a variable impedance circuit. As a means for achieving this, a means for obtaining a current output by combining a differential input-differential output voltage-current conversion circuit and a multiplication circuit is required.

This type of current amplifier is essentially suitable for an integrated circuit configuration because it provides better characteristic matching and thermal coupling between elements when integrated. However, since the absolute value of the transconductance depends on the absolute value of the diffused resistance inside the integrated circuit, there is a problem that a sufficient absolute value cannot be secured. That is, the diffused resistance within the integrated circuit is ±20%
Variation in absolute value of degree and approximately 1500ppm/deg
The absolute value of the transconductance of an integrated circuit current amplifier is also directly affected by this temperature dependence.

However, the above-mentioned variations in absolute values and temperature dependence do not pose a problem in the case of a current amplifier that is part of a circuit device that is entirely configured inside an integrated circuit. In contrast, the current output from the integrated circuit is connected to external L (coil), C (capacitor), and R (resistance).
Voltage amplifiers that convert into voltage using elements, etc.
Alternatively, in a circuit that uses a current amplifier as a variable impedance circuit to realize programmable frequency characteristics or filter characteristics, it is strongly desired to realize the absolute value of transconductance with high precision.

FIG. 1 is a circuit diagram showing an example of such a current amplifier. In FIG. 1, 1 is a positive power supply terminal, and 2 is a negative power supply terminal. The differential input-differential output voltage-current conversion circuit 3 includes a resistor 5 having a resistance value R ₀ connected between each output terminal of a current source 4 that supplies a pair of mutually equal currents I _x ;
A PNP transistor 6 whose emitter is connected to each connection point between the current source 4 and the resistor 5,
It consists of 7. The differential input voltage supplied to the differential input terminals 8, 9, which are the respective paces of these transistors 6, 7, is converted into a current, which is then applied to the differential output terminal 10, which is the respective collector of these transistors 6, 7. , 11 as currents I ₁ and I ₂ . These output currents I ₁ and I ₂ are supplied to the NPN transistors 13 and 13 forming the PN junction pair 12.
14 respectively. transistor 13,
14, each base is connected to each collector to form a PN junction anode, and these diode-connected transistors 13, 14
The emitters, which serve as cathodes, are commonly connected and connected to a constant voltage source. Furthermore, PN junction pair 1
The anodes of the transistors 14 and 13 constituting the transistor 2 are respectively connected to the bases of NPN transistors 16 and 17 constituting the common emitter transistor pair 15. These PN junction pair 12 and common emitter transistor pair 15 constitute a multiplier 18. Current source 19 connected to the common emitter of common emitter transistor pair 15
is used to flow current 2I _Y and determine the multiplication coefficient of the multiplier 18. Output terminal 2 of multiplier 18
Current inversion (current mirror) circuit 2 is installed at 0 and 21.
2 are connected, common emitter transistor pair 15
The difference current between the respective collector output currents I ₃ and I ₄ of the transistors 16 and 17 is taken out to the output terminal 23 as an output current i ₀ . That is, terminal 20
The polarity of the output current I ₄ is inverted by the current inversion circuit 22, and the difference component between the output currents I ₄ and I ₃ of the terminals 21 and 20 becomes the output current i ₀ and appears at the output terminal 23. Note that the current reversal (current mirror) circuit 22
is composed of PNP type transistors 24 and 25.

In the configuration shown in FIG. 1, the output currents I ₁ and I ₂ of the voltage-current inversion circuit 3 are as follows: I ₁ ≒I _x -v/R ₀ ... I ₂ ≒ I _x -v/R ₀ ... Therefore, the output currents I ₃ and I ₄ of the multiplier 18 are as follows: I ₃ =I ₁ (I _Y /I _X ) . . . I ₄ = I ₂ (I _Y /I _X ) . From these formulas, the output current i ₀ is i ₀ =I ₄ −I ₃ =2vI _Y /I _X R ₀ . Therefore, this circuit converts the potential difference between the terminals 8 and 9 into a current, and the conversion coefficient to the output current can be controlled by the current value I _Y of the current source 19. That is, this circuit has a transconductance of 2I _Y /I _X R ₀ , and this transconductance is determined by the current ratio I _Y /I _X and the resistance value R ₀ .

By the way, the absolute value accuracy of the diffused resistor in the integrated circuit is about ±20% as described above, but the relative value accuracy is about ±2%, which is a good characteristic. Therefore, when the circuit shown in Figure 1 is integrated, the current ratio I _Y / _I
R ₀ has a variation of about ±20%, and as a result,
The accuracy of the transconductance is almost uniquely determined by the accuracy of the diffused resistance. Therefore, when such a current amplifier is configured as a current output type integrated circuit, the performance of the circuit is dominated by the accuracy of the diffused resistance and is not fully satisfactory.

In view of the above points, the present invention makes it possible to set the transconductance with high accuracy without depending on the absolute value of the diffused resistance inside the integrated circuit when it is integrated into an integrated circuit, thereby improving the characteristics of the circuit device to which it is applied. An object of the present invention is to provide a current amplifier that can reduce manufacturing conditions of integrated circuits and improve yields.

That is, the characteristics of the current amplifier according to the present invention are as follows:
In an integrated circuit current amplifier that obtains an output current corresponding to a differential voltage applied between two input terminals, a first bias current source connected to the emitter of a differential transistor and a voltage-current conversion coefficient are configured. a differential input-differential output voltage-current conversion circuit including a first resistor formed in a defined integrated circuit; a PN junction pair to which differential output current of the voltage-current conversion circuit is supplied; a multiplier circuit comprising at least one pair of common-emitter transistors each having a base connected to one end of each of the pair of PN junctions; an output terminal for taking out an output from the common-emitter transistor pair of the multiplier circuit; A second bias current source connected to the common emitter of the transistor pair, a reference voltage generating means, and a resistance value inversely proportional to the reference voltage generated by the reference voltage generating means and a second resistor formed in the integrated circuit. means for generating a first reference current determined by a relationship, and means for generating a second reference current determined by an inversely proportional relationship between the reference voltage and the resistance value of a third resistor connected outside the integrated circuit. The first reference current generating means generates the first reference current.
and controlling the second bias current source by the second reference current generating means.

Preferred embodiments of the present invention will be described below with reference to the drawings.

FIG. 2 is a circuit diagram showing a current amplifier as a first embodiment of the present invention. In this figure 2,
Components similar to those in FIG. 1 showing the prior art of the present invention are given the same reference numerals and their explanation will be omitted.

First, the reference voltage source 31 is obtained by using, for example, a forward voltage drop of a PN junction inside an integrated circuit.
Based on this, an internal reference current I _RI and an external reference current I _RE are created. The internal reference current I _RI is determined by the reference voltage source 31 and an internal reference resistor 32 formed by a diffused resistor inside the integrated circuit, and the absolute value of the resistance value R _I of this internal reference resistor 32 is approximately ±20%. However, the relative value accuracy with respect to other diffused resistors inside the integrated circuit is approximately ±2%. The external reference current I _RE is determined by the reference voltage source 31 and an external reference resistor 35 external to the integrated circuit connected between the reference voltage terminal 33 and the reference current terminal 34 , and the resistance of the external reference resistor 35 The value R _E is kept highly accurate, with an error of ±several percent or less, for example.
One ends of the internal reference resistor 32 and the external reference resistor 35 are connected to the reference voltage terminal 33, and the other ends of each of these resistors 32 and 35 are connected to the current inversion circuit 3.
6 and 37, respectively. The NPN transistors 38 and 39 and the current source 40 are circuits for canceling the forward voltage drop of the diode that occurs at the input end of a current inversion (current mirror) circuit. Generates a voltage that is the voltage plus the forward voltage drop of the diode. As a result, the electromotive voltage of the reference voltage source 31 is substantially applied to the internal and external reference resistors 32 and 33.

The current inversion circuit 36 includes an NPN transistor 41,
42, and the collector of the transistor 42, which is its output terminal, is connected to the input terminal of a current inversion (current mirror) circuit 43. This current inversion circuit 43 includes PNP type transistors 44, 45,
46, and outputs a current I _RI (corresponding to the above-mentioned current _I has been done. Therefore, the steady current ( _IX ) of the transistors 6 and 7 constituting the differential input/differential output voltage-current conversion circuit 3 described above with reference to FIG.
becomes the internal reference current _IRI . Next, the current inversion (current mirror) circuit 37 is composed of NPN transistors 47, 48, and 49, and the collector of the transistor 47 is connected to the base to serve as a current input terminal, and
The collectors of are connected in common and serve as output terminals, allowing a current 2I _RE that is twice the external reference current I _RE to flow. This output terminal is connected to the common emitter of the common emitter transistor pair 15.

The output current i ₀ in the first embodiment of such a configuration is equal to I _X , I _Y in the prior art circuit of FIG.
are replaced with I _RI and I _RE , respectively, and from the above formula, i ₀ =2vI _RE /I _RI R ₀ . . . is obtained. Therefore, the transconductance of this first embodiment circuit is 2I _RE /I _RI R ₀ . I _RE /I _RI in this transconductance is
It is given as the resistance ratio between the internal reference resistor 32 and the external reference resistor 33, and when expressed using this resistance ratio, it becomes i ₀ = 2vR _I /R _E R ₀ ..., and the transconductance is 2R _I /R _E R becomes ₀ . Here, R _I and R ₀ in the formula are both resistances 5 formed by diffusion etc. inside the integrated circuit.
The absolute values of these resistance values greatly depend on the manufacturing conditions of the integrated circuit, the operating temperature, etc., but the relative accuracy between R _I and R ₀ is extremely high and a good value is achieved. It is something to do.
When the ratio of these resistance values R ₀ and R _I is K (= R _I / R ₀ ), the formula is expressed as i ₀ = 2vK/R _E ……, and the resistance value R _E of the external reference resistor 35 and,
It is determined by the ratio of the diffusion resistance inside the integrated circuit. In other words, the transconductance 2K/R _E can be determined with high precision without depending on the absolute value of the diffused resistance, which has low precision.As a result, the performance of circuit characteristics can be improved, and the manufacturing conditions of integrated circuits can be relaxed and yields can be improved. can be expected.

Next, FIG. 3 shows a second embodiment of the present invention,
Portions corresponding to those in FIG. 2 are given the same reference numerals and their explanations will be omitted. The second embodiment shown in FIG. 3 differs from the first embodiment shown in FIG. The second difference in means has two major differences.

First, regarding the first difference, the configurations of the voltage-current conversion circuit and the multiplication circuit will be explained. That is, in FIG. 3, first and second NPN transistors 101 and 102 whose bases are connected to first and second input terminals 8 and 9, respectively,
Each base is the first and second transistor 10
The third PNP type is connected to each emitter of 1,102, and a resistor 5 is connected between each emitter.
A voltage-current conversion circuit is constituted by the fourth transistors 6 and 7, and each collector is connected to the first and second transistors.
NPN type fifth and sixth transistors 105, each of which is connected to each emitter of the transistors 101 and 102, and whose base is connected to each collector of the third and fourth transistors 6 and 7, respectively;
106, each collector is connected to each base of the fifth and sixth transistors 105 and 106, each base is connected to each emitter of the fifth and sixth transistors 105 and 106, and each emitter is connected to each base of the fifth and sixth transistors 105 and 106, respectively. Commonly connected NPN type 7th,
The eighth transistors 13 and 14 have their bases and collectors connected to the seventh and eighth transistors 13 and 1.
4, and each emitter of the NPN type is connected to each emitter of the seventh and eighth transistors 13 and 14, respectively.
A multiplication circuit is constituted by ten transistors 103 and 104 and at least one common emitter transistor pair 15 whose bases are connected to the bases of the seventh and eighth transistors 13 and 14, respectively.

In the above configuration, the transistors 101,1
02 operates as an emitter follower and drives transistors 6 and 7, respectively. PN junction pair 12
The transistors 13 and 14 constituting the circuit are not diode-connected as shown in FIG. 2, but by connecting diode-connected transistors 103 and 104 between their respective bases and emitters,
Each constitutes a current inversion (current mirror) circuit. Here, transistors 105 and 106
are the respective collector currents I ₁ and I ₂ of transistors 6 and 7
A feedback circuit is formed to maintain balance between the collector currents of the transistors 13 and 14, and the operating currents of the transistors 101 and 102 are determined.
Further, as described above, the transistors 13 and 103
and 14 and 104 respectively constitute a current inversion (current mirror) circuit, so transistors 6 and 101 and transistors 7 and 102
A current that is always kept at a constant ratio flows through each of them. This current ratio is between transistors 13 and 103,
and the base-emitter saturation current ratio of 14 and 104, that is, the emitter area ratio. For example, if this current ratio is 1, transistors 6 and 101 and 7 and 102
operate with approximately equal current. As a result, the same change in the base-emitter voltage of transistors 6 and 7 caused by the change in I ₁ and I ₂ based on the input signal v occurs between the base and emitter voltages of transistors 101 and 102.
This also occurs in the emitter voltage, and these cancel each other out. Therefore, the voltage changes at the input terminals 8 and 9 are faithfully transmitted to both ends of the resistor 5, and the above-mentioned
The currents I ₁ and I ₂ expressed by the formulas have higher accuracy.

Note that the base-emitter voltage of a transistor is determined as a logarithmic relationship of the emitter current, so
The above effect is not limited to the saturation current ratio of transistors 13, 103 and 14, 104 being 1.

Furthermore, the base current of the common emitter transistor pair 15 is the same as that of the transistors 13, 14, 103,
It has no effect on the current of 104. Base current is provided by transistors 105 and 106. Therefore, by setting the conversion coefficients such as I _{Y 〓I}

Next, the above-mentioned second embodiment in the second embodiment of FIG.
The difference in reference current generation means, which is the difference between the two, will be explained. Here, when the first embodiment shown in FIG. 2 is integrated into an integrated circuit, in order to generate the external reference current I _RE , two of the reference voltage terminals 33 and reference current terminals 34 for connecting the external resistor 35 are required. However, in the case of the second embodiment shown in FIG. 3, only one terminal 107 is provided as a dedicated pin terminal for external resistance connection. That is, in this second embodiment, the reference voltage source 3
Connect one end of 1 to positive power supply terminal 1, and connect the other end to
By connecting the emitter of the PNP transistor 111, the positive power supply terminal 1 and the terminal 10
7, a reference voltage is generated by adding the electromotive voltage of the reference voltage source 31 and the base-emitter voltage of the transistor 111, and this reference voltage is applied to the external reference resistor 3.
The external reference current generated by being applied to 5
I _RE is flowing into the integrated circuit from the terminal 107. Here, PNP type transistors 111, 11
2, 113 and the current source 114 are the first
Similarly to the transistors 38, 39 and current source 40 of the embodiment, this circuit is for canceling the forward voltage drop of the diode occurring at the input terminal of the current inverting circuit 37. Further, the internal reference current circuit shown in FIG. 3 is also provided with NPN transistors 116, 117, 118 and a current source 119, similar to the external reference current circuit. According to the configuration for generating the reference current of the second embodiment, only one external connection pin terminal (terminal 107) is required for generating the external reference current I _RE , and the integrated circuit package The degree of freedom increases.

In the first embodiment shown in FIG. 2 and the second embodiment shown in FIG. It is possible to use it in combination with.

Next, the reference voltage generating means inside the integrated circuit mainly has the following three methods.

That is, the first method is to use the forward drop of the PN junction (for example, V _BE of a transistor), which has a large temperature coefficient and is used as a simple means.

The second method uses the breakdown voltage of a PN junction such as a Zener diode.

Thirdly, there is a band gear preference method using thermal voltage V _T (=kT/q) and V _BE .

All of these methods are applicable to the present invention. Also, the reference voltage terminal (terminal 33 in Figure 2)
A reference voltage generated outside the integrated circuit may be applied to the integrated circuit. Furthermore, if the voltage supplied to the integrated circuit is stable, the power supply voltage or a voltage obtained by dividing it can be used as the reference voltage.

Next, the current amplifier according to the present invention has a multiplication circuit.
By providing a plurality of common emitter transistor pairs 15 connected to the PN junction pair 12, a plurality of outputs can be obtained.

By the way, when providing current amplifiers for multiple channels (for example, left and right channels in audio stereo) on the same integrated circuit chip, only one reference voltage generating means of the first and second embodiments is provided and the reference current is generated. It may be configured such that a plurality of them are taken out by a current inversion (current mirror) circuit or the like and supplied to each current amplifier.

That is, FIG. 4 shows 1 as an application example of the present invention.
Two current amplification circuit sections 5 in one integrated circuit 51
2,53 is shown. In this FIG. 4, current amplification circuit sections 52 and 53 have two input terminals 54, 55 and 56, 5, respectively.
7 and are connected to differential input terminal pairs 58 and 59, respectively. Current amplification circuit section 5
The outputs of 2 and 53 are output terminals 60 and 61, respectively.
It is also supplied to each inverting input terminal of operational amplifiers 62 and 63. Each output terminal 64, 65 of these operational amplifiers 62, 63 is connected to a signal output terminal 66, 67, respectively, and each output terminal 64, 65 is connected to each terminal 60, 61 via a feedback network 68, 69, respectively. It is connected to the. One reference current generation circuit section 70 is provided inside the integrated circuit 51, and this circuit section 7
The internal reference current I _RE and external reference current I _RE outputted from
53 respectively.

With such a configuration, the differential input terminal pair 58,
The two channels of differential input voltages applied to the terminals 59 and 59 are converted into current outputs by the current amplification circuits 52 and 53, and are further converted into current outputs by the operational amplifiers 62 and 63.
Then, feedback circuit networks 68 and 69 take out voltage signals weighted with arbitrary frequencies to signal output terminals 66 and 67, respectively. The voltage gain of this circuit is completely independent of the absolute value of the diffusion resistance inside the integrated circuit. Furthermore, the gain can be arbitrarily set by changing the external reference resistor 35.

Next, it is also preferable to apply the current amplifier according to the present invention to an audio noise reduction device, for example, a sliding band type noise reduction device as shown in FIG.

In the noise reduction device shown in FIG. 5, an input signal supplied to an input terminal 71 is transmitted through a main signal path 72 with a gain of 1 and a flat frequency characteristic, and a sub-signal path 72 whose main part is a high-pass filter with a variable cut-off frequency. The outputs from the main signal path 72 and the sub signal path 73 are added together by an adder 74 and sent to an output terminal 75. Here, the cut-off frequency of the variable cut-off frequency high-pass filter provided in the sub-signal path 73 is controlled in accordance with a control signal from a control circuit provided with a level detection circuit for detecting a signal level. This high-pass filter with a variable cut-off frequency is generally composed of a variable resistance circuit and a filter network, and the current amplifier of the present invention can be applied to this variable resistance circuit.

That is, FIG. 6 shows an example of the configuration of a variable resistance circuit using a current amplifier, in which the output terminal 23 of the current amplifier 81 of the present invention is connected to the inverting input terminal 9 to form a negative feedback path. Input terminal 82
is connected to the terminal 8, and the output terminal 83 is connected to the terminal 23, thereby forming a two-terminal circuit.
The equivalent circuit viewed from the input/output terminals 82 and 83 of this variable resistance circuit is a series connection circuit of a buffer amplifier 84 and a variable resistance 85, as shown in FIG.

Here, FIG. 8 shows an application example in which the current amplifier of the first embodiment is used as the variable resistance circuit of FIG. 6 and applied to the aforementioned sliding band type noise reduction device.

In this FIG. 8, a fifth
The main part of the noise reduction device shown in the figure is formed, and noise reduction is performed on the input signal supplied to the input terminal 71, and the output terminal 75
is output from.

That is, the input signal of the input terminal 71 is supplied to the base of one transistor 6 constituting the voltage-current conversion circuit of the above-mentioned current amplifier, and
The input signal is supplied via a main signal path 72 to a non-inverting input terminal of an operational amplifier 92 which operates as a summing circuit. Two pairs of common emitter transistors 93 and 94 are connected to the PN junction pair 12 provided at the input stage of the multiplier circuit of the current amplifier described above, so that two current outputs can be obtained. The output terminal of the common emitter transistor pair 93 is connected to the base of the other transistor 7 of the voltage-current conversion circuit, and is also connected to one end of a capacitor 96 via a terminal 95 of the integrated circuit 91. The other end of the capacitor 96 is grounded.

As shown in FIG. 6, the current amplification circuit section consisting of a part of the above voltage-current conversion circuit and multiplication circuit (PN junction 12 and common emitter transistor pair 93) has an output from the common emitter transistor pair 93 connected to the input stage. Since negative feedback is provided to the base of the transistor 7, the equivalent circuit viewed from the terminal 95 becomes a variable resistance circuit as shown in FIG. This variable resistance circuit and capacitor 96
This constitutes a substantially linear high-pass filter circuit, and by changing the resistance value of the variable resistance circuit, the cutoff frequency of this filter circuit is changed. Here, the resistance value of the above variable resistance circuit is
This can be controlled by controlling the common emitter current of the common emitter transistor pair 93. That is, the common emitter of the common emitter transistor pair 93 is connected to a current inverting (current mirror) circuit 97 whose input terminal is connected to the reference current terminal 34.
A current source 99 whose current value is controlled in accordance with a control signal from a control input terminal 98 is connected in parallel with the output side transistor of the current inversion circuit 97 . A control circuit for obtaining a control voltage from the output of the sub-signal path is connected to the control input terminal 98, but is omitted in FIG. 8 to simplify the explanation.

Next, the common emitter transistor pair 94 of the multiplier circuit produces a current output corresponding to the potential difference between the terminals 71 and 91 in order to obtain a sub-signal output with high-pass characteristics. The output of this common-emitter transistor pair 94 is supplied to an inverting input terminal of an operational amplifier 92 which acts as the adder circuit. In this case, the operational amplifier 92
2 as a voltage follower, and since the sub-signal path output is a current source, it acts as an inverting amplifier for the output. Incidentally, the gain of the sub-signal path is determined by the product of the transconductance from the input terminal 71 to the output of the common-emitter transistor pair 94 and the resistance value of the negative feedback resistor 100 of the operational amplifier 92. Since this resistor 100 is formed as a diffused resistor inside the integrated circuit, the transconductance should be configured to be inversely proportional to the absolute value of the diffused resistor. Therefore, while the common emitter current of the common emitter transistor pair 93 is given by the external reference current I _RE , the common emitter current of the common emitter transistor pair 94 is given by the internal reference current I _RI . is given.

In the noise reduction device having such a configuration, when there is no input signal to the input terminal 71 (no signal), the cutoff frequency of the sub-signal path is in the lowest state, and at this time the current of the current source 99 is It becomes approximately 0. Therefore, the common emitter current of the common emitter transistor pair 93 becomes approximately the external reference current _IRE , and as is clear from the above description of the present invention, the variable resistance circuit depends on the absolute value of the diffused resistance within the integrated circuit. An independent equivalent resistance value is obtained,
The cutoff frequency of the sub-signal path is not affected by the diffusion resistance.

Next, when an input signal is applied, a current flows through the current source 99 under the action of a control circuit (not shown). Therefore, the cut-off frequency of the auxiliary signal path increases and is determined primarily by the current source 99.
In this case, by adopting a configuration in which the current of the current source 99 with respect to the control circuit input does not depend on the absolute value of the diffused resistance, the cutoff frequency of the sub-signal path when the input signal is applied also changes to the absolute value of the diffused resistance. Can be controlled to avoid dependence.

Note that the present invention is not limited to the above-mentioned embodiments. For example, the conductivity type of each of the above-mentioned transistors may be exchanged between PNP type and NPN type, or two parallel current sources may be used. Instead of using this, the output of one current source may be divided and used.

As explained above, according to the current amplifier according to the present invention, when it is integrated into an integrated circuit, from the same reference voltage, an internal reference current that is inversely proportional to the diffused resistance within the integrated circuit and a reference resistance connected to the outside of the integrated circuit are generated. By using an external reference current that is inversely proportional to , it is possible to make the transconductance of the current amplifier independent of the absolute value of the diffusion resistance. Therefore,
In applications such as certain voltage amplifiers, programmable filter circuits, and sliding band type noise reduction devices that are advanced versions of these, the accuracy of gain, frequency characteristics, etc. does not depend on the absolute value of the diffused resistance, and the accuracy is high. Accuracy can be improved. Furthermore, as a result, improvements in circuit characteristics, relaxation of integrated circuit manufacturing conditions, and improvement in yield can be expected.

[Brief explanation of drawings]

Figure 1 is a circuit diagram showing the prior art of a current amplifier;
FIG. 2 is a circuit diagram showing a first embodiment of a current amplifier according to the present invention, FIG. 3 is a circuit diagram showing a second embodiment, and FIG. 4 is a circuit diagram showing an application example of the current amplifier according to the present invention. 5 is a block diagram showing the basic configuration of a sliding band noise reduction device, FIG. 6 is a block diagram showing another application example of the current amplifier according to the present invention, and FIG. 7 is a block diagram showing the basic configuration of a sliding band type noise reduction device. 6
8 is a circuit diagram showing a specific example in which the current amplifier according to the present invention is applied to the variable resistance circuit of FIG. 6 to construct the noise reduction device of FIG. 5. 3... Voltage-current conversion circuit, 5... First resistor, 8, 9... Input terminal, 12... PN junction,
15, 93, 94... common emitter transistor pair, 31... reference voltage source, 32... internal reference resistor (second resistor), 35... external reference resistor (third resistor).

Claims (1)

[Claims] 1. In an integrated circuit current amplifier that obtains an output voltage corresponding to a differential voltage applied to two input terminals, a first transistor connected to the emitter of a differential transistor;
a differential input-differential output voltage-to-current conversion circuit comprising a bias current source and a first resistor formed in an integrated circuit that defines a voltage-to-current conversion coefficient; The PN junction pair and their respective bases to which the output current is supplied are
a multiplier circuit comprising at least one common-emitter transistor pair connected to each end of the PN junction pair; an output terminal for taking out an output from the common-emitter transistor pair of the multiplier circuit; and a common emitter terminal of the common-emitter transistor pair. a second bias current source connected to the integrated circuit; a reference voltage generating means; means for generating a reference current; and a third circuit connected to the reference voltage and external to the integrated circuit.
means for generating a second reference current determined in inverse proportion by the resistance value of the resistor; the first reference current generating means controls the first bias current source; A current amplifier characterized in that the second bias current source is controlled by reference current generating means.