JPS596605A - Current amplifier - Google Patents

Abstract

PURPOSE:To improve the chracteristics of an applied circuit device by preventing transconductance from dependence upon the absolute value of a diffused resistor in an integrated circuit when the circuit device is used as the integrated circuit. CONSTITUTION:Internal reference current IRI and external reference current IRE are formed on the basis of a reference voltage source 31. The current IRI and IRE are determined by the voltage source 31 and an internal reference resistor 32 and an external reference resistor 35 which are formed by respective diffused resistors. One end of the reference resistors 32, 35 is connected to a reference voltage terminal 33 and the other ends of respective resistors 32, 35 are connected to an npn transistors (TR) 36, 37. The current inversion circuit 36 consists of npn TRs 41, 42 and its collector is connected to the input terminal of a current inversion circuit 43. The internal reference current IRI of the circuit 43 is the steady state current of TRs 6, 7 constituting a voltage-current converting circuit 3 of differential input and output.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a current amplifier that obtains an output current corresponding to a voltage difference between input signals applied to a pair of differential input terminals, and is particularly suitable for integration into an integrated circuit.

For example, it is possible to realize an instrumentation amplifier that removes the common mode component from a signal transmitted as a balanced signal and converts it into an unbalanced signal, a voltage-to-current converter whose gain can be electrically controlled, a voltage amplifier, or a variable impedance circuit. Therefore, a means for obtaining a current output by combining a differential input-differential output voltage-current conversion circuit and a multiplication circuit is required.

This type of current amplifier is essentially suitable for an integrated circuit configuration because it provides better characteristic matching and thermal coupling between elements when integrated. However, since the absolute value of the transconductance depends on the absolute value of the diffused resistance inside the integrated circuit, there is a problem that a sufficient absolute value cannot be secured.

In other words, the diffused resistance in an integrated circuit has a variation in absolute value of about ±20% and a temperature dependence of about 1500 m/del, and the absolute value of the transconductance of a current amplifier integrated in an integrated circuit also has a variation in absolute value of about ±20% and a temperature dependence of about 1500 m/del. be affected as is.

However, the above-mentioned variations in absolute values and temperature dependence do not pose a problem in the case of a current amplifier that is part of a circuit device that is entirely configured inside an integrated circuit. On the other hand, a voltage amplifier that converts the current output from an integrated circuit into voltage using an external L (coil), C (capacitor), R (resistance) element, etc., or a current amplifier as a variable impedance circuit. It is strongly desired to realize the absolute value of transconductance with high precision in circuits that realize programmable frequency characteristics and filter characteristics.

FIG. 1 is a circuit diagram showing an example of such a current amplifier. In FIG. 1, 1 is a positive power supply terminal, and 2 is a negative power supply terminal. A differential input-differential output voltage-current conversion circuit 3 has a resistance value R connected between each output terminal of a current source 4 that supplies a pair of mutually equal currents Ix.
The resistor 5 is composed of a PNPN resistor 6.0 and a PNPN transistor 6.7 having an emitter connected to the connection point between the current source 4 and the resistor 5, respectively. The differential input voltage supplied to the differential input terminals 8, 9, which are the bases of each of these transistors 6.7, is converted into a current, and the differential output terminal, which is the collector of each of these transistors 6. 10.
11, the currents L and I2 are output. These output currents L and I2 are respectively supplied to NPN transistors 13 and 14 forming the PN junction pair 12. The bases of the transistors 13 and 14 are connected to their respective collectors to form a PN junction anode, and the emitters, which serve as cathodes of these diode-connected transistors 13 and 14, are commonly connected to form a constant current source. It is connected to the. Further, the anodes of the transistors 14 and 13 forming the PN junction pair 12 are connected to the N l) N forming the common emitter transistor pair 15.
The bases of the transistors 16 and 17 are connected to each other. These PN junction pair 12 and common emitter transistor pair 15 constitute a multiplier 18. A current source 19 connected to the common emitters of the common emitter transistor pair 15 supplies a current 2) to determine the multiplication coefficient of the multiplier 18. Output terminal 20 of multiplier 18.
A current inversion (current mirror) circuit 22 is connected to 21, and the cap output current I3.14 of each transistor 16.17 of the common emitter transistor pair 15 is connected to
The difference current is outputted to the output terminal 23 with an output current of 1°. That is, the output current 4 of the terminal 20 is 1) inverted in stiffness by the current inverting circuit 22, and becomes the difference component of the output current I4゜I3 of the terminal 21.20 or the output current 1o, which appears at the output terminal 23. . Note that the current inversion (current mirror) circuit 22 is an I) NP type transistor 24.
25.

In the configuration shown in FIG. 1, the output current L, I2 of the voltage-current inverting circuit 3 is: ■1,000 I x v/'RO...Song ■I2
+ I x + v, 'Ro song,...
, ■, and the output currents of the multiplier 18 I3, I4
I3 = L (IY/IX)...
・・・・・・・・・・・・Q, I4 = I2 (I
v/Ix) ・・・・・・・・・It becomes a song. From the 0 to 0 formula, the output current 1o is 'O''
l4I3 = 2 ~ "IY/IXI (0... curve ■. Therefore, this circuit converts the electrical difference between terminals 8 and 9 into a current, and the conversion coefficient to the output current is the current The current value of the source 19 can be controlled by the current value 1. That is, this circuit
It has a transconductance of IY/IxRo, and this transconductance is determined by the current ratio IY/IX and the resistance value Ro.

Incidentally, the absolute value accuracy of the diffused resistor in the integrated circuit is about ±20% as described above, but the relative value accuracy has good characteristics of about ±2 factor. Therefore, when the circuit shown in Fig. 1 is integrated, the current ratio Iy/Ix can be set with high accuracy because it is determined by the voltage ratio and resistance ratio. As a result, the accuracy of the transconductance is almost uniquely determined by the accuracy of the diffused resistance.

Therefore, when such a current amplifier is configured as a current output type integrated circuit, the performance of the circuit is dominated by the accuracy of the diffused resistance, and is not fully satisfactory.

In view of the above-mentioned points, the present invention enables the transconductance to be set with high accuracy without depending on the absolute value C of the diffused resistance inside the integrated circuit when integrated circuit, and the characteristics of the applied circuit device. It is an object of the present invention to provide a current amplifier that can be improved, ease manufacturing conditions for integrated circuits, and improve yield.

That is, the characteristics of the current amplifier according to the present invention are that, in an integrated circuit current amplifier that obtains an output current corresponding to a differential voltage applied between two input terminals, a first bias current source and a voltage-current a differential input-differential output voltage-to-current conversion circuit with a first resistor formed in an integrated circuit that defines a conversion coefficient; and a PN junction to which the differential output current of the voltage-to-current conversion circuit is supplied. a multiplier circuit comprising at least one common emitter transistor pair and a respective base connected across the pair of PN junction transistors; a second bias current source connected to the common emitters of the common emitter transistor pair; , a reference voltage generating means, a means for generating a first reference current determined by the reference voltage generated by the reference voltage generating means and a resistance value of a second resistor formed in the integrated circuit, means for generating a second reference voltage determined by the resistance value of a third resistor connected to the outside of the circuit, and the first bias current source is controlled by the first reference current generating means. Hereinafter, preferred embodiments of the second reference current generating means according to the present invention will be described with reference to the drawings.

FIG. 2 is a circuit diagram showing a current amplifier as a first embodiment of the present invention. In FIG. 2, the same parts as in FIG. 1 showing the prior art of the present invention are given the same reference numerals, and their explanation will be omitted.

First, the reference voltage source 31 is, for example, a PN inside the integrated circuit.
This is obtained by utilizing the forward voltage of the junction, and based on this reference voltage source 31, an internal reference current IRI and an external reference current IRE are generated. The internal reference current IRT is determined by the reference voltage source 31 and an internal reference resistor 32 formed by a diffused resistor inside the integrated circuit, and the absolute value of the resistance value RH of this internal reference resistor 32 is ±2.
Although it includes an error of about 0%, the relative value accuracy with respect to other diffused resistors inside the integrated circuit is about ±2 inches.

The external reference current 111g is determined by the reference voltage source 31 and an external reference resistor 35 external to the integrated circuit connected between the reference voltage terminal 33 and the reference current terminal 34, and the resistance value RE of the external reference resistor 35 is , for example, the error is maintained at a high accuracy of less than a third.

One ends of the internal reference resistor 32 and the external reference resistor 35 are connected to the reference voltage terminal 33, respectively.
．． The other end of the valley 35 is connected to current inversion circuits 36 and 37, respectively. The NPN transistors 38 and 39 and the current source 40 are circuits for canceling the forward voltage drop of the diode that occurs at the input end of a current inversion (current mirror) circuit. Generates a voltage that is the voltage plus the forward voltage drop of the diode. As a result, internal and external reference resistances 32.
33, substantially the electromotive voltage of the reference voltage source 31 is applied.

The current inversion circuit 36 is constituted by NPN transistors 41 and 42, and the collector of the transistor 42, which is an output terminal thereof, is connected to the input terminal of a current inversion (current mirror) circuit 43. This current reversing circuit 43 is composed of PNP transistors 44, 45 and 46, and is of a two-output type that outputs currents (corresponding to the above-mentioned current Ix) from the respective collectors of transistors 45 and 46. Each output terminal is connected to both ends of the resistor 5 mentioned above. Therefore, the steady current (Ix) of the transistor 6.7 constituting the differential input/differential output voltage-current conversion circuit 3 described above with reference to FIG. 1 becomes the internal reference current IrB. Next, the current inversion (current mirror) circuit 37 includes NPN transistors 47, 48 .
49, the collector of transistor 47 is connected to the base and serves as a current input terminal, and the collectors of transistors 48 and 49 are commonly connected and serve as an output terminal, and a current 2 twice the external reference current 111E is generated. Run the IRE. This output terminal is a common emitter 1-transistor pair 15
connected to a common emitter.

The output current Io in a first embodiment of such a configuration is equal to Ix in the prior art circuit of FIG.

Iy is replaced with I+u and IRH, respectively, and from the above formula 0, io = 2 V IRE / IRIRO...
・・・・・・・・・・・・■ is obtained. Therefore, the transconductance of this first embodiment circuit is 2 I
It becomes RE/InIRQ.

InE/IRr in this transconductance is given as the resistance ratio between the internal reference resistor 32 and the external reference resistor 33. Using this resistance ratio, the formula 0 is expressed in Table 4 as follows: 1o=2■RI/RERo... ·river·
Song...■ and eggplant, transconductance is 2 R1
/RE RO becomes. Here, ■RI and Ro in the formula
are the respective resistance values of the resistors 5 and 32 formed by diffusion etc. inside the integrated circuit, and the absolute values of these resistance values largely depend on the manufacturing conditions of the integrated circuit, the operating temperature, etc., but R1 , Ro is extremely high and achieves a good value. These resistance values R
When the ratio of o and R+ is K (= Rr/Ro), ■
The formula is i o = 2 v K/ RE ,,,
The resistance value of the external reference resistor 35 is 1' (
It is determined by the ratio of E and the diffusion resistance inside the integrated circuit. In other words, the transconductance 2K/RE can be determined with high precision without depending on the absolute value of the diffused resistance, which has low precision, and as a result, the performance of circuit characteristics can be improved, the manufacturing conditions of integrated circuits can be eased, and the yield can be improved. You can expect it.

Next, FIG. 3 shows a second embodiment of the present invention, and parts corresponding to those in FIG. 2 are given the same reference numerals and their explanations will be omitted.

The second embodiment shown in FIG. 3 differs from the first embodiment shown in FIG. second means
There are two major differences.

First, regarding the first difference, the configurations of the voltage-current conversion circuit and the multiplication circuit will be explained. That is, in FIG. 3, first and second NPN transistors 1 whose respective bases are connected to the first and second input terminals 8.9
01, 102, and the bases of the first and second transistors io1. Each is connected to the i02 valley emic,
A voltage-current conversion circuit is constituted by third and fourth transistors 6.7 of PNP type with a resistor 5 connected between the valley and emitter, and each collector is connected to each of the first and second transistors 101, 102. NPN type fifth and sixth transistors 105 and 106 whose emitters are respectively connected and whose valley bases are connected to the collectors of the third and fourth transistors 6.7, respectively; 6th
are connected to the valley bases of the transistors 105 and 106, respectively, and each base is connected to the valley bases of the fifth and sixth transistors 10.
5. The seventh and eighth NPN transistors 13 and 14 (!:, G base, :'L' transistors are connected to the respective emitters of NPs connected to the respective bases of the transistors 13.14 and having their emitters connected to the emitters of the seventh and eighth transistors 13.14, respectively.
N-type ninth and tenth transistors 103 and 104;
A multiplier circuit is constituted by at least one pair of common emitter transistors 15 whose bases are respectively connected to the bases of the seventh and eighth transistors 13 and 14.

In the above configuration, transistors 101 and 102 operate as emitter followers and drive transistors 6 and 7, respectively. The transistors 13 and 14 constituting the P-N junction pair 12 are not diode-connected as shown in FIG. 2, but are diode-connected between their emitters. By connecting them, a current inversion (current mirror) circuit is constructed. Here, the transistor 105,
106 is each collector current of the transistor 6.7.

Form a feedback circuit with I2 and the valley collector current of transistors 13 and 14, and further 1.
The operating currents of the 7-sisters 101 and 102 are determined. Further, as described above, the transistors 13, 103 and 14, 104 are current inverters (current mirrors), respectively.
Since they constitute a circuit, currents that are always maintained at a constant ratio flow through transistors 6 and 101 and transistors 7 and 102, respectively. This current ratio is set by the base-to-emitter saturation current ratios, ie, the emitter area ratios of transistors 13 and 103 and 14 and 104. For example, when this current ratio is 1, transistors 6 and 101 and transistors 7 and 102 operate with approximately equal currents. As a result, based on the input signal ■ (II
, I2 causes a change in the base-emitter voltage of transistor 6.7, and the same change occurs in the voltage between the emitters of transistors 101 and 102, and these cancel each other out. Therefore, the voltage change at the input terminal 8.9 is accurately transmitted to both ends of the resistor 5, and the currents I+ and I2 expressed by the equations (1) and (2) become more accurate.

Note that since the base-emitter voltage of a transistor is determined as a logarithmic function of the emitter current, the above effect is
~The saturation current ratio of transistors 13, 103, and 14, 104 is not limited to 1.

Furthermore, the base current of the common emitter transistor pair 15 has no effect on the currents of the transistors 13, 14, 103, 104. The base current is transistor 1
Powered by 05.106. Therefore, IY
> By setting a conversion coefficient such as IX, a decrease in accuracy of the conversion coefficient, generation of nonlinear components, abnormal operation, etc. will not occur.

Next, the second difference in the reference current generating means in the second embodiment shown in FIG. 3 will be explained. Here, when the first embodiment shown in FIG. 2 is integrated into an integrated circuit, in order to generate an external reference current IRE, a reference voltage terminal 33 for connecting an external resistor 35 and a reference current terminal 3 are provided.
However, in the case of the second embodiment shown in FIG. 3, only one terminal 107 is provided as a dedicated pin terminal for connection of a large external resistance. That is,
In this second embodiment, one end of the reference voltage source 31 is connected to the positive power supply terminal 1, and the other end is connected to the emitter of the PNPN upper type transistor 111, thereby connecting the positive power supply terminal 1 and the terminal 107. A reference voltage is generated by adding the electromotive voltage of the reference voltage source 31 and the base-emitter voltage of the transistor 111, and this reference voltage is applied to the external reference resistor 35 to generate an external reference current IR.
E flows into the integrated circuit from the terminal 107. Here, the PNPN upper type transistors 111, 112, 113 and the current source 114 are arranged in the order of diodes generated at the input terminal of the current inverting circuit 37, similar to the transistors 38, 39 and the current source 40 of the first embodiment described above. This is a circuit to cancel directional voltage drop. Also, regarding the internal reference current circuit in FIG. 3, the NpN transistors 116, 117, 118 and the current source 11
9 is provided. According to the configuration for generating the reference current of the second embodiment, the external connection pin terminal for generating the external reference current RE is pin 1 (terminal 107).
), which increases the degree of freedom in packaging the integrated circuit.

In the first embodiment shown in FIG. 2 and the second embodiment shown in FIG. It is possible to use it in combination with.

Next, the reference voltage generating means inside the integrated circuit mainly has the following three methods.

That is, the first method is to use the forward drop of the PN junction (for example, VBE of a transistor), which has a large temperature coefficient and is used as a simple means.

The second method is to use the breakdown voltage of a PN junction such as a Zener diode.

The third method is a band gap reference method using thermal voltage VT (=kT/Q) and VBE.

All of these methods are applicable to the present invention. Further, a reference voltage generated outside the integrated circuit may be applied to the reference voltage terminal (terminal 33 in FIG. 2). Furthermore, if the voltage supplied to the integrated circuit is stable, the power supply voltage or a voltage obtained by dividing it can be used as the reference voltage.

Next, the current amplifier according to the present invention can obtain a plurality of outputs by including a plurality of common emitter transistor pairs 15 connected to the PN junction pair 12 of the multiplication circuit.

By the way, in the case where current amplifiers for multiple channels (for example, left and right channels in audio stereo) are provided on the same integrated circuit chip, only one reference current generating means of the eleventh and second embodiments is provided and the reference current is generated. It may be configured such that a plurality of them are taken out by a current inversion (karen)/mirror circuit or the like and supplied to each current amplifier.

That is, FIG. 4 shows an example in which the present invention is applied, in which two current amplification circuit sections 52 and 53 are provided in one integrated circuit 51. In FIG. 4, current amplifier circuit sections 52 and 53 each have two input terminals 54.
55 and 56.57, and are connected to differential input terminal pairs 58 and 59, respectively. The outputs of the current amplification circuit sections 52 and 53 are supplied to output terminals 60 and 61, respectively, and to respective inverting input terminals of operational amplifiers 62 and 63. These operational amplifiers 62.
Valley output terminal 64.65 of 63 is signal output terminal 66.67
and each output terminal 64.65 is connected to each terminal 60.61 via a feedback network 68, 69, respectively.
It is connected to the.

Inside the integrated circuit 51, one reference current generation circuit section 70 is provided.
The internal reference current IRE and the external reference current IRE output from this circuit section 70 are supplied to two current amplification circuit sections 52 and 53, respectively.

With such a configuration, the two channels of differential input voltage applied to the differential input terminal pair 58 and 59 are converted into current output by the current amplifier circuit section 52.53, and further, the operational amplifier 62.63 and Feedback network 68.69
The voltage signals weighted with arbitrary frequencies are outputted to signal output terminals 66 and 67, respectively. The voltage gain of this circuit is completely independent of the absolute value of the diffusion resistance inside the integrated circuit. Furthermore, the gain can be arbitrarily set by changing the external reference resistor 35.

Next, it is also preferable to apply the current amplifier according to the present invention to an audio noise reduction device, for example, a sliding band type noise reduction device as shown in FIG.

In the noise reduction device of FIG. 5, the input signal supplied to the input terminal 71 is a main signal path 72 with a gain of 1 and a flat frequency characteristic, and a sub signal path 72 whose main part is a high-pass filter with a variable cut-off frequency. The outputs from the main signal path 72 and the sub signal path 73 are summed by an adder 74 and sent to an output terminal 75. Here, the cut-off frequency of the high-pass filter with a variable cut-off frequency provided in the sub-signal path 73 is controlled in accordance with a control signal from a control circuit provided with a level detection circuit for detecting a signal level. This high-pass filter with a variable cut-off frequency is generally composed of a variable resistance circuit and a filter network, and the current amplifier of the present invention can be applied to this variable resistance circuit.

That is, FIG. 6 shows an example of the configuration of a variable resistance circuit using a current amplifier, in which the output terminal 23 of the current amplifier 81 of the present invention is connected to the inverting input terminal 9 to form a negative feedback path. The input terminal 82 is connected to the terminal 8, and the output terminal 83 is connected to the terminal 23, thereby forming a two-terminal circuit.

The equivalent circuit viewed from the input/output terminals 82 and 83 of this variable resistance circuit is a series connection circuit of a buffer amplifier 84 and a variable resistance 85, as shown in FIG.

Here, FIG. 8 shows an application example in which the current amplifier of the first embodiment is used as the variable resistance circuit of FIG. 6 and applied to the aforementioned sliding band type noise reduction device.

In this FIG. 8, the main part of the noise reduction device shown in FIG. Ru.

That is, the input signal of the input terminal 71 is supplied to the base of one of the transistors 6 constituting the voltage-current conversion circuit of the current amplifier mentioned above, and the input signal is supplied to the base of the transistor 6 as an adder circuit via the main signal path 72. Operating operational amplifier 9
2 non-inverting input terminal. Two common emitter transistor pairs 93 and 94 are connected to the PN junction pair 12 provided at the input stage of the multiplier circuit of the current amplifier described above, so that two current outputs can be obtained. The output terminal of the common emitter transistor pair 93 is connected to the base of the other transistor 7 of the voltage-current conversion circuit, and is also connected to one end of a capacitor 96 via a terminal 95 of the integrated circuit 91. The other end of capacitor 96 is grounded.

As shown in FIG. 6, the current amplification circuit section consisting of a part of the above voltage-current conversion circuit and multiplication circuit (PN junction 12 and common emitter transistor pair 93) is configured such that the output of the common emitter transistor pair 93 is connected to the input stage. transistor 7
Since negative feedback is provided to the base of the terminal 95, the equivalent circuit seen from the terminal 95 becomes a variable resistance circuit as shown in FIG.

This variable resistance circuit and the capacitor 96 constitute a substantially linear high-pass filter circuit, and by changing the resistance value of the variable resistance circuit, the cutoff frequency of this filter circuit is changed.

Here, the resistance value of the variable resistance circuit can be controlled by controlling the common emitter current of the common emitter transistor pair 93. That is, the common emitter of the common emitter transistor pair 93 is connected to the output end of a current inversion (current mirror) circuit 97 whose input end is connected to the reference current terminal 34, and the output side transistor of this current inversion circuit 97 is connected to the common emitter of the common emitter transistor pair 93. A current source 99 whose current value is controlled in accordance with a control signal from a control input terminal 98 is connected in parallel. Note that a control circuit for obtaining a control voltage from the sub-signal path output is connected to the control input terminal 98;
It is omitted in FIG. 8 to simplify the explanation.

Next, the emitter common transistor pair 94 of the multiplication circuit
is connected to terminal 7 in order to obtain an auxiliary signal path output with high-pass characteristics.
It produces a current output corresponding to a potential difference between 1.91 and 1.91. The output of this common emitter transistor pair 94 is supplied to an inverting input terminal of an operational amplifier 92 which acts as the adder circuit. In this case, the operational amplifier 92 acts as a voltage follower for the main signal path 72, and since the output of the sub signal path is a current source, it acts as an inverting amplifier for the output. By the way, the gain of the sub-signal path is
It is determined by the product of the transconductance from the input terminal 71 to the output of the common emitter transistor pair 94 and the resistance value of the negative feedback resistor 100 of the operational amplifier 92. Since this resistor 100 is formed as a diffused resistor inside the integrated circuit, the transconductance should be configured to be inversely proportional to the absolute value of the diffused resistor. therefore,
The common emitter current of the common emitter transistor pair 93 is given by the external reference current IRE, while the common emitter current of the common emitter transistor pair 94 is given the internal reference current IRI.

In the noise reduction device having such a configuration, when there is no input signal to the input terminal 71 (no signal)
At this point, the cut-off frequency of the sub-signal path is at its lowest, and
The current of this switching current source 99 becomes approximately zero. therefore,
The common emitter current of the common emitter transistor pair 93 becomes approximately the external reference current IRE, and as is clear from the above description of the present invention, the variable resistance circuit has an equivalent resistance that does not depend on the absolute value of the diffused resistance in the integrated circuit. is obtained, and the cutoff frequency of the sub-signal path is not affected by the diffusion resistance.

Then, when an input signal is applied, a control circuit (not shown)
A current flows through the current source 99 due to the action of . Therefore, the cutoff frequency of the auxiliary signal path increases and is determined primarily by the current source 99. In this case, by adopting a configuration in which the current of the current source 99 with respect to the control circuit input does not depend on the absolute value of the diffused resistance, the cutoff frequency of the sub-signal path when the input signal is applied also changes to the absolute value of the diffused resistance. Can be controlled to avoid dependence.

Note that the present invention is not limited to the above-mentioned embodiments. For example, the conductivity types of the valley transistors may be exchanged between PNP type and NPN type, and two parallel current sources may be used. Instead of using this, the output of one current source may be divided and used.

As explained above, when the current amplifier according to the present invention is integrated into an integrated circuit, from the same reference voltage, an internal reference current that is inversely proportional to the diffused resistance within the integrated circuit and a reference resistor connected to the outside of the integrated circuit are generated. By using an external reference current that is inversely proportional to , it is possible to make the transconductance of the current amplifier independent of the absolute value of the diffusion resistance. Therefore, in applications such as certain voltage amplifiers, programmer full filter circuits, and advanced slide ink band noise reduction devices, the accuracy of gain, frequency characteristics, etc. does not depend on the absolute value of the diffused resistance. , high precision can be achieved. Furthermore, as a result, improvements in circuit characteristics, relaxation of integrated circuit manufacturing conditions, and improvement in yield can be expected.

[Brief explanation of the drawing]

FIG. 1 is a circuit diagram showing a prior art of a current amplifier, FIG. 2 is a circuit diagram showing a first embodiment of a current amplifier according to the present invention,
FIG. 3 is a circuit diagram showing a second embodiment, FIG. 4 is a block circuit diagram showing an application example of the current amplifier according to the present invention, and FIG. 5 is a basic configuration of a noise reduction device using a slide ink bundle method. 6 is a block diagram showing another application example of the current amplifier according to the present invention, FIG. 7 is an equivalent circuit diagram of FIG. 6, and FIG. 8 is a block diagram showing another application example of the current amplifier according to the present invention. 6 is a circuit diagram showing a specific example of the noise reduction device of FIG. 5 applied to a variable resistance circuit. FIG. 3・・・・・・・・・・・・・・・・・・Voltage-current conversion circuit 5・・・・・・・・・・・・・・・・・・First resistor 8.9 ......Input terminal 12...
・・・・・・・・・・・・PN conjugate 15.93.9
4...Common emitter transistor pair 31...
......Reference voltage source 32...
...Internal reference resistance (second resistance) 35...
......External reference resistance (third resistance) 1st m

Claims (1)

[Claims] In an integrated circuit current amplifier that obtains an output current corresponding to a differential voltage applied between two input terminals, a first
a differential input-differential output voltage-to-current conversion circuit comprising a bias current source and a first resistor formed in an integrated circuit that defines a voltage-to-current conversion coefficient; a multiplier circuit comprising a pair of PN junctions to which an output current is supplied and at least one pair of common emitter transistors each having a base connected to both ends of the pair of common emitter transistors; a second bias current source, a reference voltage generating means, and a first reference current determined by the reference voltage generated by the reference voltage generating means and the resistance value of a second resistor formed in the integrated circuit. and means for generating a second reference voltage determined by the reference voltage and the resistance value of a third resistor connected to the outside of the integrated circuit, wherein the first reference current generating means generates the second reference voltage. 1 bias current source, and the second reference current generating means controls the second bias current source.
A current amplifier characterized in that it controls a bias current source.