JPH0446582A - Controller for ac servomotor - Google Patents

Abstract

PURPOSE:To share IM and SM only by simple changeover by obtaining the angle of rotation at the position of magnetic flux corresponding to sliding velocity and selectively adding the angle of the sliding rotation to the position of magnetic flux detected. CONSTITUTION:An inverter 2 is switching-driven according to PWM gate pulses given from a control board 101, and converts input DC power 3 into the ACs of fixed frequency and fixed voltage and outputs them. The control of an AC servomotor 1 variably controls the speed of the motor 1 at a desired value fundamentally, the actual speed of the motor 1 is detected for the control, and motor currents are controlled so as to bring the deviation of the actual speed of the motor 1 and a speed command value to zero. When the motor currents are controlled, motor currents are divided into a (d) axis component and a (q) axis component, and a voltage vector is selected in response to these components and the inverter 2 is driven. Accordingly, the position of the magnetic flux of the motor 1 is detected, and the current components of (d) and (q) axis are controlled in conformity with the detection.

Description

DETAILED DESCRIPTION OF THE INVENTION [Industrial Application Field] The present invention relates to a control device for an AC servo motor, and can control an induction motor (hereinafter referred to as IM) or a synchronous motor (hereinafter referred to as IM) by simple switching. The present invention relates to a control device that can be applied to control both types of SM (referred to as SM).

(Prior Art) Conventionally, as a control device for an AC servo motor, it has been common to configure an optimal system according to the applied IM or SM.

However, many of the elements that make up the control device are
Since it can be applied to both IM and SM motors, a common control device has been proposed so that it can be applied to both IM and SM motors by simple switching (Japanese Patent Application No. 1983-
No. 181839).

According to this, since the design and manufacture of the control device can be standardized, there is an advantage that the number of design steps can be reduced, and the manufacturing efficiency can be improved and costs can be reduced.

[Problem to be solved by the invention]

However, the above-mentioned common use of the conventional control device has a problem in that sufficient consideration is not given to the common use of the magnetic flux position detection (control) means. Especially IM related to magnetic flux position detection
If the output pulse per revolution of the rotation detector (rotation encoder) of the SM or SM is different, design and m
There was a problem that had to be addressed from the production stage.

Further, when implementing a control device exclusively for IM or 8M using a microcomputer or the like, there is a problem in that sufficient consideration is not given to improving control accuracy and response speed.

The first object of the present invention is to connect IM and S by simple switching.
An object of the present invention is to provide a control device for an AC servo motor that can be shared by M.

A second object of the present invention is to provide an AC servo motor control device that improves the response speed of layer control and has good control accuracy when realizing an IM or SM control device using a microcomputer. It is in.

[Means to solve the problem]

In order to achieve the above-mentioned first object, the present invention includes speed detection means for detecting the speed of an AC motor based on pulses input from a rotation detector attached to an AC motor to be controlled; magnetic flux position detection means that detects the magnetic flux position of the AC motor by counting with a pull-down counter; and speed control that receives as input a deviation between the detected speed and a given speed command and outputs a q-axis current command according to this deviation. and a q-axis current control means that receives the q-axis current command as input and generates and outputs a q-axis voltage command.

d-axis current command generation means that inputs the detected speed and outputs a d-axis current command and a magnitude of magnetic flux according to the detected speed; and switching means that selects either the d-axis current command or the d-axis current zero command; d-axis current control means that receives the d-axis current command or d-axis current zero command as input through the switching means and generates and outputs a d-axis voltage command according to the input command;
A gate signal for an inverter that receives the q-axis voltage command, the d-axis voltage command, and the detected magnetic flux position as input, selects a combination of voltage vectors to be applied to the AC motor, and operates the AC motor based on this selection. a PWM control means that generates and outputs a PWM control means that inputs the detected current of the AC motor and the detected magnetic flux position, converts this detected current into a detected current of the q-axis and the d-axis, and generates and outputs the current of the q-axis and the d-axis. detecting current phase conversion means that provides negative feedback to each current control means, and calculating a sliding speed using the magnitude of the magnetic flux and the q-axis current command as input;
A control device for an AC servo motor, comprising: a slip rotation angle calculation means for calculating a rotation angle of a magnetic flux position corresponding to the slip velocity; and a switching means for selectively adding this slip rotation angle to the detected magnetic flux position. That's what I did.

Furthermore, in order to achieve the second object, the present invention obtains a q-axis current command and a d-axis current command of the AC motor in order to match the detected speed of the AC motor to be controlled with the given speed command, Speed control in which PWM data consisting of a combination of voltage vectors to be generated by the inverter and a time width is determined by PWM control based on these current commands, and this PWM data is output to a PWM signal generation means that generates a gate signal for the inverter. and current control means for converting the detected current of the AC motor into a q-axis component and a d-axis component to respectively correct the 1-axis current command and the d-axis current command. The control device includes a host microcomputer using a general-purpose microprocessor and a slave microcomputer using a digital signal processor, and the slave microcomputer controls the speed control means and the current control means. The host microcomputer performs sampling of measurement data such as detected speed and speed command necessary for the control calculation, and the above-mentioned 1.
) WM data is output to the PWM signal generating means.

[Effect]

With this configuration, according to the present invention, the above object is achieved through the following actions.

First, the speed detection means and the magnetic pole position detection means are necessary for both the IM and the SM, and can be used in common.

Furthermore, both IM and SM are the same in that the speed is controlled by controlling the q-axis current based on the speed deviation, and the speed control means for generating the q-axis current command can be shared.

On the other hand, since the d-axis current is a magnetic flux component, IM
In the case of , the magnetic flux must always be provided as a constant magnetic flux current component, but in the case of the SM, the magnetic flux is provided by a permanent magnet of the rotor, so it does not necessarily have to be provided. Therefore,
The d-axis current command generating means is dedicated to IM, and can be shared by setting the command value to zero in the case of SM using the switching means.

The PWM control means for controlling the output voltage of the inverter based on the d+q-axis current command set in this way is the same for both IM and SM, so they can be used in common.

In addition, the motor current is fed back as described in d.

The current control system that corrects and controls the q-axis current command is IM.

Both SM and SM are the same, so they can be used in common. However, when converting the motor detection current into d+q-axis current components, the magnetic flux position of the motor is required, and especially in the case of IM, the detected magnetic flux position must be corrected in consideration of rotor slippage. Therefore, the full rotation angle calculated by the full rotation angle calculating means is added to the detected magnetic flux position. This addition can be selected by the switching means, so in the case of SM, it is sufficient to avoid addition.

In addition, when the AC motor is a synchronous motor, magnetic pole position detection corrects the count value of the up/down counter based on a signal representing a specific magnetic pole position input from a magnetic pole position detector attached to the synchronous motor. Set up a circuit,
It is desirable that this magnetic pole position detection circuit can be selectively connected to the up/down counter by a switching means. According to this, the accuracy of the detected magnetic flux position is improved,
Appropriate SM control becomes possible.

Further, the magnetic flux position detection means may be provided with rotation angle conversion means for multiplying the count value of the up-down counter by a conversion coefficient that makes the number of pulses output per rotation from the rotation detector correspond to 360°. desirable. According to this, even if the types of rotation detectors are different and the number of pulses per rotation is different, adaptation can be easily made by simply changing the coefficients.

On the other hand, according to another invention of the present invention, the role of control calculations is shared by a slave microcomputer consisting of a digital signal processor, so that the calculation speed is faster than when configured with a general-purpose microprocessor alone. , response speed and control accuracy are improved.

In that case, the sampling period of the control calculation is set to a number of times the carrier period of the PWM control and the ASR sampling period of the speed control means, respectively, based on the ACR sampling period of the current control means. It is desirable to do so. That is, A.S.
Although the sampling period of R control becomes longer, since motor speed fluctuations are slower than current fluctuations, it is possible to reduce the calculation processing related to ASR and shorten the data acquisition and transfer time required for it. Since the control calculation cycle (sampling cycle) of ACR can be made faster, -layer response speed and accuracy can be improved.

In that case, data transmission between the host microcomputer and the slave microcomputer, except for measurement data related to the speed control means, shall be performed at each ACR sampling period, and control calculations related to the speed control means shall be performed as described above. The AC is divided into integer parts.
It is preferable that the measurement data is executed every R sampling period, and the measurement data necessary for the contents of the divided control calculations is divided into parts of the integer and transmitted from the host microcomputer every ACR sampling period.

Further, the PWM signal generating means includes a timer that is reset every carrier period twice the ACR sampling period, a pair of registers, and a pair of comparators for each phase, and the PWM data is It consists of a pair of time data for each phase that defines the time width for turning on the switching elements of each phase of the inverter from the starting point of the carrier cycle, and each pair of data is transferred to the register each time the timer is reset. It can be done.

Note that the smaller time data of each pair of data is transferred to the register each time the timer is reset, and the larger time data of each pair of time data is transferred to the ACR sampling. It is also possible to execute the process every cycle when the time data is larger than the value of the timer at the time of transfer, thereby further improving control accuracy and response speed.

Further, the present invention can be applied to the IM/SM common control device, and thereby the response speed and control accuracy of the IM/SM common control device can be improved.

〔Example〕

Hereinafter, the present invention will be explained based on examples.

FIG. 1 shows a functional configuration diagram of an entire AC servo motor drive control system including an AC servo motor control device according to an embodiment of the present invention. In the figure, an AC servo motor 1 is driven at a desired speed by variable frequency, variable voltage AC supplied from an inverter 2. The inverter 2 is driven by switching according to the PWM gate pulse given from the control board 101.
It converts the input DC power 3 into AC of a predetermined frequency and voltage and outputs it. Here, the control board 101 corresponds to the control device according to the present invention, and the AC servo motor 1 can be applied to an IM or an SM. Although FIG. 1 shows the switching state when applied to IM, it can also be applied to SM as is with simple switching as described later.

AC servo motor 1 (hereinafter referred to as "AC servo motor") in this embodiment.

The control of the motor (simply referred to as a motor) basically involves variable control of the speed of the motor 1 to a desired value.
The actual speed of the motor is detected, and the motor current is controlled to make the deviation between this and the speed command value zero. To control this motor current, the motor current is divided into a d-axis component and a q-axis component, a voltage vector is selected according to these components, and the inverter 2
For this purpose, the magnetic flux position of the motor 1 is detected and the current components of the dr and q axes are controlled accordingly. This will be explained in detail below.

A rotary encoder 4 attached to the shaft of the motor 1 generates pulses in response to changes in the rotational position of the motor 1.
A constant number of pulses Pr are output per rotation of the motor 1. This Pels is up and down (U/D
) is input to the counter 7 and the speed detection means 13, and the speed detection means 13 detects the speed ωr of the motor 1 from the number of pulses and the time width.

A speed command 01 for the motor 1 is given to the control board from a speed command means 14 such as a host control device.

This speed command ωr* is input to an adder, where the deviation from the detected speed value ωr is determined. This speed deviation is input to the speed control means 15, which generates a q-axis current command ■Q* of the q-axis current component orthogonal to the rotating magnetic flux φ to make the deviation zero.

On the other hand, in the case of IM, a d-axis current component is required to generate the magnetic flux φ, and this d-axis current command IIi is generated by the d-axis current command generation means 16 and outputted via the changeover switch 17. This changeover switch 17 is for switching according to IM and SM, and the magnetic flux φ in the case of SM is
Since is given by the permanent magnet of the rotor, the d-axis current command ID rate may be zero, so when applied to SM, the changeover switch 17 is switched to the ground side.

Here, FIG. 2 shows a detailed configuration diagram of the d-axis current command generating means 16. As shown in the figure, the means 16 includes a magnetic flux command generating means 16a, a magnetic flux controlling means 16b, and a magnetic flux calculating means 16.
Contains c. The magnetic flux command generating means 16a uses the detected speed ωr detected by the speed detecting means 13 as a human power, and outputs a corresponding magnetic flux command φ wheel based on a preset function. The magnetic flux command is the magnetic flux control means 1
．． 6b, where a d-axis current command ID rate is generated and outputted to the changeover switch 17.

On the other hand, the magnetic flux calculation means 16c calculates the magnetic flux φ based on the following equation (1), provides negative feedback to the input 1 of the magnetic flux control means +6b, and also outputs the negative feedback to the input 1 of the magnetic flux control means +6b.
Output to 2. In equation (1), ■ is the excitation inductance of the TM, 'I'2 is the second-order time constant of the IM, and S is the Laplace operator.

In the above example, the d-axis current command ID rate is changed according to the speed ωr, but it may be changed according to the torque, or both can be used together. Furthermore, the ID book may be set to a constant value.

The q-axis current command IQ set in this way and the d-axis current command ID
It is input to the axis current control means 28 and the d-axis current control means 29. Detection currents IQ and ID obtained by detecting the current flowing through the motor 1 and converting it into d+q-axis components are negatively fed back to the adders 18 and 19, respectively. Therefore, the current control means 28 and 29 for each axis operate according to the deviation between the command and the detected value, and control the q-axis voltage command vQ and the d-axis voltage command V, respectively.
D copies are generated and output to the PWM control means 30.

Here, current commands IQ*, ID* are defined as magnetic flux potential I of motor 1.
Detection current I that is adjusted to Q and fed back
u''w must also be converted into DC into IQ and ID according to the magnetic flux position 0.

Therefore, magnetic flux position detection and detection current conversion will be described in detail next.

First, the detected currents Iu and Iv of each phase of the motor 1.

A procedure for converting Iw into q-axis and d-axis detection currents IQ and ID will be explained. The three-phase currents Iu to Iw are detected as analog quantities by current detection means 24 and 25 such as Hall CT,
This is converted into a digital quantity by the A/D converter 26 provided on the control board 101. Next, in the 3/2 phase conversion means 27, the 2-phase current converter is converted based on the following equations (2) and (3).
, IQ, and further based on equation (4), IQ and I
Convert to D.

Iu+Iv+Iw=0 (3) By using equation (3), as shown in FIG. 1, the current detection for one phase (Iv) can be omitted.

sin θ, θ of QOBθ used in the calculation of equation (4) above
is the magnetic flux position of the motor 1, which is detected by the following procedure. Furthermore, in order to reduce the time required to calculate 5inO and cosθ, a sinθ/cosO table 9 is provided in which the values of sinθ and CO5θ are stored using θ as an address. Then, the 3/2 phase conversion means 27 calculates the above equation (4) using the values of sin θ and cos θ read from the sin O/cos θ table 9 in correspondence with the detected magnetic flux position θ. It is supposed to be done.

First, the U/D counter 7 counts input pulses to detect changes in the rotational position of the rotor, thereby detecting the magnetic flux position. The U/D counter 7 operates up and down according to the forward and reverse rotation of the motor 1.

Further, it is reset every reference position M (usually one rotation). Therefore, the count value Pc of the U/D counter 7 becomes a value corresponding to the rotational position of the motor 1 through one rotation (360°). However, since the rotary encoder 4 may have a different number of output pulses Pr per rotation, the magnetic flux that always makes the count value of the U/D counter 7 correspond to the rotation angle M (for example, 360') is used. It must be detected as the position θ. This correspondence is performed by multiplying the count value PC by a conversion coefficient K by the rotation angle conversion means A8. This coefficient is set to 1 so as to satisfy the following equation (5), and is changeable.

Incidentally, the magnetic flux position θ detected by the U/D counter 7 as described above may be set to an initial value of 0'' in the case of IM, but in the case of SM, it must be adjusted to the position of the rotor magnetic poles made of permanent magnets, etc. This will be explained later.

On the other hand, the number of areas N required for a 5 in O or cos θ table is N=2n (where n is an integer), and the larger n (the smaller the pitch of O), the better the calculation accuracy. Further, a table covering the entire range from 0=a to 360 degrees may be used, but this is not preferable because the table becomes enormous. Therefore, by using the symmetry of sin θ or cos O, one range obtained by dividing θ = 0 to 360° into 172 or 1/4 is set as a table, and the other ranges are essentially divided by converting the read address. It can be the same as having a full range table. In this embodiment, as shown in FIG. 3, the range of θ=O to 360@ is divided into 90 units, and four modes MDI to MD4 are set. And s i n
As shown in Fig. 4, the O/cos θ table 9 is created by assigning the number of areas N to θ=O~90°.
In addition to forming a table of inθ and cosO, for the mode, the address conversion means 10 identifies which mode MDI to MD4 the value of the input magnetic flux position O belongs to, and inverts the address order according to the mode to which it belongs. ,si
The sign of nθ or cosO is inverted and read out.

Using sin θ and cosa read in this way, the 3/2 phase converting means 27 converts the detected current into IQ and ID, respectively, by calculations shown in the following equations.

By the way, in magnetic flux position control in the case of IM, slippage must be taken into consideration. Therefore, as shown in FIG. 1, the slip calculation means 22 calculates the following equation from the magnetic flux φ output from the d-axis current command generation means 16 and the q-axis current command IQ* output from the speed control means 15. Obtain the sliding speed command 05 by (6).

The Veri speed command ωS obtained in this way is converted into a rotating angle O8 by the rotation angle conversion means B23, and θS is added to the detected magnetic flux position θ by an adder 21 via a changeover switch 20. . As a result, the slip ω
Control is performed based on the magnetic flux position in consideration of S. Note that the unit of ωS* is the number of rotations, and the conversion coefficient of the rotation angle conversion B23 is set to 2 so as to satisfy the following equation (7).

Note that M has the same dimension as in equation (5) above.

Based on the magnetic flux position O obtained in consideration of slip as described above, the detected current I in the 3/2 phase conversion means 27 is
v~W is converted into q, d-axis current IQ, ID, which is q
, d*tll current command I Q*.

It is given to adders 18 and 19 as the feedback amount of the ID book. The qrd-axis current control means 28.29 is operated according to the deviation of the current output from the adder 18.19, and generates the q+d-axis voltage command VQ necessary to make the deviation zero.
*, VD* are generated and output to the I) WM control means 30. This PWM control means 30 uses a well-known pulse width modulation (PWM) method to select a voltage vector according to the voltage commands IQ* and ID*, determines the output time width of the voltage vector, and based on these, A gate signal for each switching element of the inverter 2 is generated and output. Its detailed functions will be explained next.

FIG. 5 shows a detailed top view of the PWM control means 3o. Although only the U-phase voltage is shown in the figure, the same applies to the other (2) and W-phases. As shown, applied voltage generating means 3
0a, vector time conversion means 30b, register conversion means 30c, PWM signal generation means 30d, gate circuit 30e
It is formed with The applied voltage generating means 30a calculates the following equation (8) based on the input Q* and D cars.
The voltage VTυ to be applied to the U phase of the motor 1 and its phase δ are determined and output.

Vtυ= l V D* l sinδ+ IVQ
"1 cos δ The vector diagram related to this equation (8) is shown in Figure 6 (
Shown in b). Further, FIG. 3A shows the vector relationship between the induced voltages eou, eov, and eow of each phase of the motor 1, the motor current, and the terminal voltage VTU.

As shown in FIG. 4B, the magnetic flux φ1 is located at a position delayed by 90° with respect to the q-axis voltage command ■Q* related to the torque command, and this position coincides with the above-mentioned θ. Further, the phase α of the magnetic flux φ2 to be applied to the inverter 2 is at the position α=(θ+δ). Therefore, the conduction width of the switching element of the inverter 2 is controlled so that the voltage Vtu is applied to the U phase of the motor 1 at a phase 906 ahead of the magnetic flux φ2. This control is executed by the vector time conversion means 30b.

Here, the voltage vector that can be generated by the inverter 2 having a three-phase bridge configuration and the method of generating a desired voltage vector VTU by combining the voltage vectors are well-known techniques. I will explain using 7th
As shown in Figure (a), the inverter 2 has switching elements U, V, W, X, Y, and Z in three paired arms each, upper and lower.

Now, when the switching element U is on and the switching elements v and W are off, the switching elements X, Y, and Z of the paired arms are turned off, on, and on in a complementary manner, respectively. At this time, the current flowing in the motor 1 flows from the U terminal and flows out from the W terminal.As a result, the voltage vector generated in the motor 1 is set to 1, and Vl
It is written as (100). Note that “l” and “’0” in parentheses
corresponds to "on" and "off" of the switching elements U, V, and W. Similarly, the switching elements U, V correspond to "on" and "off" of the switching elements U, V, and W.

By changing the on/off combination of W, X, Y, and Z, as shown in Figure 7(b), 6
voltage vectors v1 to ■6 and two zero vectors Vo
,V, is obtained. and.

In order to generate the voltage vector vTυ, FIG.
), Vl (100) and ■2 (110)
All you have to do is combine them. In addition, which voltage vectors should be combined to synthesize a voltage vector having a desired phase is determined by α
It is determined by For example, in the interval α=O to 60°, ■1 and ■2 may be combined. This specific example will be further explained with reference to FIG. Figure 8 shows how to select and output the voltage vectors V1 and v2 when the magnetic flux φ2 is in phase α.The output time width vector of vl is set to Vi, and the output time width of The vector is expressed as Vj. That is, it shows the concept of changing the magnitudes of voltage vectors Vi and 2 by adjusting the time widths of vl and Vj. These Vi and Vj can be determined by the following equation (9). At this time, the value of 5 inches (60 to α) is obtained from the sia/cosO table 9. In addition, VZ
are zero vectors Vo, V7, and Tc is the carrier period of PMW.

Vi=VTuXKvXsiri (60-a), Vj
=vruXKvX S 1 n αVz =Tc-2(
Vi−Vj) ・−(9) Also, the constant Kv is α=
When the maximum voltage VTU required by the motor 1 is 0, Vi is set to be equal to 172 of the carrier period Tc.

Vi and Vj obtained in this way are input to the register conversion means 30c, and the PW signal for generating the PWM signal is
It is converted into time data to be stored in each register of the M signal generating means 30d. For the sake of explanation, the PWM signal generating means 30d will first be explained along the time charts shown in FIGS. 5 and 9. As shown in FIG. 5, the PWM signal generating means 30d is divided into U-phase, ■-phase, and W-phase space vector units 301, 302, and 303. Since each of these space vector units has a similar configuration, the description will be made based on the U-phase space vector unit 30]. The U-phase space vector unit 301 has a timer U and a register U.
It includes a register REGA, a register UREGB, and comparators A and B. Timers U, V, and W of each space vector section 3.01 to 303 are set and reset in synchronization, respectively.
The timing is synchronized with the carrier cycle Tc. Comparators A and B are registers UREGA and UR, respectively.
It is a conveyor match type that operates to invert I]''l, 111.I+high output when the time data of EGB and the contents of timer U match, and each output is a comparison signal U1...U2 (Vl ,V2゜Wl,W,2
) to the gate circuit 30e.

The gate circuit 30e receives, for example, the comparison signal tJ1. U2's EN
A PWM signal U is generated and output by OR logic processing.

Similarly, PWM signals V and W are generated and output, and their inverted signals x, y, and z are output to each switching element of the inverter 2. That is, registers UREGA, B
The gate pulse width is determined by the time data stored in , VREGA,B, and WREGA,B. Therefore,
The register conversion means 30c converts the time vector ■i.

Based on vj, time data to be stored in each register is determined according to the following equation (10), and is sequentially stored in the corresponding register.

UREGB=R○.

VREGB=RO+Vi.

WREGB=RO+Vi+Vj.

WREGA=R○+Vi+Vj+RO.

VREGA=RO+Vi+Vj+RO+Vl.

UREGA=RO+Vi+Vj+RO+Vi+Di...
・(10) Here, RO is an offset value, and the calculation result is T c
/ It is determined by adjusting the value to a value not greater than 2.

Further, the calculation result of the register group on the B side is selected so as to be shorter than that of the register group on the A side. Furthermore, in the calculation of the B side jister group, after adding the offset RO at the beginning, Vi
Or add vj. In calculating the A-side jister, add the offset RO to the B-side jister value, and then calculate V
Add i or vj. By doing this, it becomes unnecessary to find Vz in equation (9). In addition,
Rewriting of each register UREGA, B, etc. is performed at the timing when each timer U, V, W is cleared. In the case of the example shown in Figure 8, Δ
The period where the mark is offset RO is Vo (000) and V7
(111) is selected, Vi selects the Vl (100) vector, vj selects the 2(110) vector, and the rest selects the Vo (000) vector.

FIG. 9 is a time chart of this state, and RO and Vi in the row marked as the calculation formula (9) used in FIG.

V J, RO, V j+ V i+ V z represent the time during which these nodes are selected. P.W.
View the MU, PWMV, and PWMW signals vertically.

They match the above vectors, respectively (000) (1
00), (110), (111), (110), (10
0), (000).

I) WM signals U, V thus formed.

W, X, Y, and Z are amplified and applied to the gates of each switching element of the inverter 2, and are variable frequencies and variable voltages necessary to move the motor 1, which is IM, in accordance with the speed command ωr tree. AC is output to motor 1.

On the other hand, if the motor 1 is an SM, this can be easily handled by switching the changeover switches 12, 17, 20 to the SM side as described below. In SM and IM speed control, in order to make the deviation between the speed command ωr* and the detected speed ωr zero, the motor current is controlled separately into each component of the d and q axes.

The voltage vector is selected by PWM control according to the q-axis current, and the motor current is fed back to control d, q!
There are common control processes such as the need to correspond to the magnetic flux position of the motor when converting to 111 current and when selecting the voltage vector. On the other hand, due to the basic configuration differences between IM and SM, in the case of SM (1) the magnetic flux position θ is determined mechanically from the rotor's magnetic pole position, so the magnetic flux position O used for control is adjusted to the rotor's magnetic pole position. (2) The rotating magnetic flux and the rotation of the rotor must be synchronized.
Since there is no slip, there is no need to consider slip as in IM regarding the magnetic flux position O. (3) In IM, the d-axis current component creates the magnetic flux of the motor, but in SM, the magnetic flux is generated by the rotor. Since it is given by a permanent magnet etc., d!
This differs from the IM case in that the 1ill current command may be zero. Therefore, regarding (2) and (3) above, the first
By switching the changeover switches 2° and 17 shown in the figure to the ground side, the dIIIffl flow command ID tree is made zero, and the correction by the slip speed ωS tree is made zero. Regarding (1) above, switch 12 to the magnetic pole position detection circuit 6 side in the changeover switch shown in FIG. I have to.

The magnetic pole position detection circuit 6 has the configuration shown in FIG. 10, and receives output signals φU and φV from the magnetic pole position detector 5.

It operates according to φ.

The magnetic pole position detector 5 has a rotor whose magnetic pole position matches the position of the motor 1, and electromagnetically detects the rotation of this rotor to output rectangular wave output signals φυ, φV, as shown in FIG. φW
These signals are inverted every 180 degrees.

Also, the positions are shifted by 120 degrees.

Therefore, the “H” level and “L” level of φU to φ
If we associate 1" and 110" and consider 3-bit addresses based on them, we can obtain 6 addresses for each 60°. Now, with the U phase as a reference, a count value Pco=P corresponding to the magnetic flux position θ is placed in each area of the above address of the ROM6a.
c, is written in advance. And read/write (
R/W) The control means 6b controls 6 based on φU to φW.
The R/W signal is sent to ROM6a and U/D counter 7 every 0°.
Output to . Thereby, the count value Pc of the U/D counter 7 is always corrected to match the magnetic pole position O of the SM.

As described above, according to this embodiment, the means relating to magnetic flux position detection and the means relating to current feedback control are shared to the maximum extent possible, and only the different means can be selected with a simple changeover switch. Design of SM control device,
Manufacturing can be largely standardized, resulting in practical effects such as reduction in design man-hours, improvement in manufacturing efficiency, and cost reduction.

In addition, since the rotation angle converting means A is provided to detect the magnetic flux position θ, it is possible to
Even if the number of output pulses per rotation Pr changes, it can be easily adjusted by simply changing the coefficient, and the control device can also satisfy versatility.

In the embodiment of FIG. 1, each means is shown as a hardware image, but it goes without saying that it can be realized by software using a microcomputer.

Here, a preferred embodiment realized using a microcomputer is shown in FIG. This embodiment uses the speed control ASR and current control AC of the AC servo motor of the embodiment shown in FIG.
Calculation and control related to R are performed by DSP (Digital
By dividing the roles between a slave microcomputer 40 using a signal processor (Signal Processor) and a host microcomputer 41 using a general-purpose microcomputer, it is possible to perform faster arithmetic processing, increase control response speed, and improve control speed. This is intended to improve accuracy. That is, as is well known, DSP has the advantage of being able to simultaneously process addition and multiplication at high speed, but it is not suitable for general data input/output processing or transmission processing.

The host microcomputer 41 is a CI that performs calculations and control) U42
Centered around the measurement data area 43 and speed command area 44.
, the transmission flag area 45, and the PWM signal generating means 30d are connected via a data bus 46. Measuring means 4
7 is the magnetic pole position detection means 6, U/D counter 7, shown in FIG.
The changeover switch 12 and the speed detection means 13 are collectively represented, and the output signals of the rotary encoder 4 and the magnetic pole position detector 5 are inputted, and the detected speed value ωr and the U/D counter 7 are calculated based on the input signals. The count value Pc is stored in the measurement data area 43 of the host microcomputer 41. Further, speed commands ωr from the speed command means 14 are stored in the speed command area 44.

The slave microcomputer 40 has a trigger board (49, reception data area 51, reception flag area 52, current board 5).
8. A calculation means 54, a data table 55, and a register data area 53 are connected to a data bus 50, and the data bus 50 is connected to a data bus 46 of a host microcomputer 41.

The calculation means 54 has a reception data area 51, a register data area 53 . It is connected to the data table 55 by an internal bus, and data is further distributed by an area 56 and a soft counter 57.
connected by an internal bus.

The calculation timings of the slave microcomputer 40 and the host microcomputer 41 are synchronized by a trigger signal TMO output from a sampling timer 48 provided in the host microcomputer 41. The signal is input to the trigger boat 49. In addition, the current boat 58 of the slave microcomputer 40 has
The values obtained by digitally converting the motor currents Iυ and Iw by the A/D converter 26 are input. Note that the portion surrounded by a dashed line in FIG. 12 corresponds to the control board 101 in FIG. 1.

The detailed functional configuration of the embodiment configured as described above will be explained in the first section.
The operation will be explained with reference to the time chart shown in FIG. The CPU 42 of the host microcomputer 41 takes in the count i[Pc of the U/D counter from the measuring means 47 at every predetermined sampling period TACR, and the detected speed ωr at every sampling period TASR, and stores them in the measurement data area 43. Further, 02 speed commands are fetched from the speed command means 14 for each sampling period TASR and stored in the speed command area 44. At the timing when the trigger signal TMO is input to the trigger boat 49, the slave microcomputer 40 transfers each of the measurement data area 43 and speed command area 44 of the host microcomputer 41 to the data bus 4.
6 and 50 and stored in the received data area 51. In addition, in accordance with this, the transmission flag area 45
The data is taken in and stored in the reception flag area 52.

Further, the sampling period T AC is output from the A/D converter 26.
The motor currents Iu and Iw are taken in for each R via the current port 58, and the calculation means 54 uses the reception data area 51, the current data Iu and Iw, and the data in the data table 55, as described in the embodiment in FIG. Performs control processing. That is, first, in order to make the deviation between the detected speed ωr and the speed command 02 zero, the q-axis current command IQ and the d-axis current command ID* are adjusted, and based on this, the inverter 2 is set to PWM.
Speed control (ASR) consisting of a control loop is performed. Second, current control (ACR) is performed, which consists of a loop that detects the motor current and performs feedback correction on the above-mentioned Q* and DQ trees. This current control includes magnetic flux position detection including the U/D counter 7, correction of the magnetic flux position by sliding, and s i
n (j/cos θ table).

By the way, in the above-mentioned control loop, since the speed fluctuation of the motor is slower than the current fluctuation, "A"
The sampling period TAsn of the SR system may be longer than the sampling period TACR of the ACR system, and TAsu=n
Set to A X TACR (where nA is an integer).

This makes it possible to reduce the load on the slave microcomputer 40 related to ASR-related arithmetic processing and pig transfer, and further improve high-speed processing. Furthermore, regarding the carrier period Tc related to PWM control, T c = nuX
Set to TACR (where nB is an integer). FIG. 1:3 shows an example where n^=3 and jl 11=2. In line with this, the transfer of ASR-related data sent from the host microcomputer 41 to the slave microcomputer 40 is
It is not necessary to transfer all the data every sampling period of R, and it is sufficient to divide it into n6 times and transfer it. For this reason, the quantity and order of transmission data are determined, the order is stored in the transmission flag area 45, and the transfer is controlled accordingly. The ASR data that has been sent in pieces is decoded by the data content in the reception flag area 52.
In accordance with this, the pig allocations are stored in the area 56 in order.

Then, the software counter 57 counts the TASR based on the TACR, and executes the ASR arithmetic processing in accordance with the time-up. In addition, the PWM obtained by the ACR calculation process executed every 'rAcrt
The data of registers UREGA, B, etc. related to signal generation are as follows:
- Single transfer (9 After being stored in the data area 53, it is transferred to each register of the PWM signal generating means 30d via the data buses 50 and 46.

The gate circuit 30e is a PWM signal generating means 30d.
is output from. I) WM signal ut, u2°Vl, V2
．． Goo 1-signal U based on Wl, W2.

v, w, x, y, z are generated and output to the inverter 2. Although FIG. 13 shows only the U phase, the same applies to the other phases.

As described above, according to the embodiment in FIG. 12, the measurement data input processing (43) and the PWM signal generation means (30d) related to control are executed by a general-purpose microcomputer, and the main control calculations related to ASR and ACR are performed. Since this is executed by the DSP, in addition to the effects of the embodiment shown in FIG. 1, the calculation speed can be improved, the high-speed responsiveness of the control can be enhanced, and highly accurate servo motor control can be realized.

Further, as shown in FIG. 13, in consideration of the control characteristics of ASR and ACR, the ASR control period is set to be an integral multiple of ACR, so that the calculation load on the slave microcomputer 40 is reduced, and Since the data transfer is performed in parts, the data transfer time can be shortened and the negative layer response speed can be improved.

Note that the embodiment in FIG. 12 is based on the IM and S according to the embodiment in FIG.
Although it was explained as realizing a shared control device for M,
The technical concept of this embodiment is not limited to a shared device, but can be applied to a control device dedicated to IM or 8M to achieve the same effect as described above.

Next, a modification of the embodiment shown in FIGS. 12 and 13 will be described.

The magnetic flux position used as a reference by the PWM control means is shown in Fig. 6 (
As explained in b), the magnetic flux φ2 corrected by the phase angle δ determined by VQlk and ID* was used. However, the first
As in the embodiment shown in FIG. 2, the rotational position is detected by the rotary encoder 4 and then transferred to the slave microcomputer 40 via the measuring means 47 and the host microcomputer 41, where the calculation of the magnetic pole position θ and the calculation of PWM control are performed. The time TD is spent until the result is transferred to the host microcomputer 41 and the gate is output to the inverter 2. Even during this period, since the motor 1 is rotating, the magnetic flux position α will have an error. Therefore, this error can be expressed as DTHD by the following formula (1
1), and correct α using equation (12).

DTHD:KDXTDX(,1r ・= (11
) α = θ + δ + DTHD (12) Note that Ko is a coefficient for converting the rotational speed into an angle. formula(
Figure 6(c) shows the position of magnetic flux φ3 corrected by 12).
Shown below. As can be seen from the figure, the voltage vector to be applied to the U phase of motor 1 is VTU3. By performing such correction, the control accuracy of the motor is further improved.

Next, a modified example of the rewriting timing of UREGA, B, etc. related to PWM signal generation will be described with reference to FIG. The data for UREGA, B, etc. is the PWM in Figure 5.
It has been explained that the timer U (or V, W) of the signal generating means is rewritten every Tc cycle when it is cleared. On the other hand, as shown in FIG. 14, since data such as UREGA and B are calculated in the TACR cycle, rewriting in the TACR cycle shorter than the Tc cycle improves control responsiveness. However, since the comparators A and B in FIG. 5 are of the so-called conveyor match type, they are of a type in which the output is inverted only when -M occurs. Therefore, Fig. 14 (
, ), if the contents of UREGA are rewritten to the calculation result at timing (3), the PWM signal U2 that appropriately reflects the latest data is output. However, if the content to be rewritten is smaller than the value of timer U and the relationship is as shown in FIG. 14(b), there will be no so-called conveyor match because they will cross at timing (3). Therefore, the PWM signal U2 is not inverted at time point (3), and as a result, the gate signal U also becomes as shown by the solid line, and an error occurs. As a result, in the period TE shown in FIG. 14(b), the Vl (100) vector is selected even though the zero vector should have been selected, causing an overcurrent to flow in the motor 1 and causing unstable rotation. Become. Note that the above-mentioned problem becomes a problem only in relation to the A-side register.

Therefore, as shown in FIG.
REGA; VREGA, WREGA (7) The rewritten data is the timer U, V at that time.

The above problem can be solved by determining whether the value is larger or smaller than the value of W and rewriting only when it is larger. That is, when the value of the rewritten data is small, rewriting is not performed. As a result, the same carrier period Tc
The PWM signal U2 is inverted at timing 2 in FIG. 15.

〔Effect of the invention〕

As described above, according to the present invention, it is possible to provide an AC servo motor control device that can be used commonly by IM and SM by simple switching. As a result, it is possible to reduce the man-hours for designing and manufacturing the device, and also to improve manufacturing efficiency and reduce costs.Furthermore, when the AC motor is a synchronous motor, it is possible to A magnetic pole position detection circuit is provided for correcting the counted value of the up/down counter based on a signal representing a specific magnetic pole position inputted from a magnetic pole position detector, and this magnetic pole position detection circuit selectively adjusts the up/down counter by a switching means. According to what was made possible to connect to the down counter,
By improving the accuracy of the detected magnetic flux position, appropriate control of the SM becomes possible.

Further, the magnetic flux position detection means is provided with rotation angle conversion means for multiplying the count value of the up-down counter by a conversion coefficient that makes the number of pulses output per rotation from the rotation detector correspond to 360 degrees. According to , even if the number of pulses per rotation is different due to a different type of rotation detector, it can be easily adapted by simply changing the coefficients.

On the other hand, according to another invention of the present invention, the role of control calculations is shared by a slave microcomputer consisting of a digital signal processor, so that the calculation speed is faster than when configured with a general-purpose microprocessor alone. , response speed and control accuracy are improved. In that case, the PWM signal generating means also
Each phase has a timer that is reset every carrier period that is twice the R sampling period, a pair of registers, and a pair of comparators, and the PWM data is the time to turn on the switching elements of each phase of the inverter. Each phase consists of a pair of time data whose width is defined from the starting point of the carrier cycle, and the smaller time data of each pair of time data is transferred to the register each time the timer is reset. According to the control method, the larger time data of each pair of time data is executed when the time data is larger than the width of the timer at the time of transfer in each ACR sampling period. Accuracy and response speed are further improved.

Further, the present invention can be applied to the IM/SM common control device, and thereby the response speed and control accuracy of the IM/SM common control device can be improved.

[Brief explanation of the drawing]

FIG. 1 is an overall configuration diagram of an embodiment of the present invention, and FIG.
Detailed configuration diagram of the d-axis current command generating means shown in Figures 3 and 4.
The figure is a diagram for explaining the sin fl / cos & table in Figure 1, Figure 5 is a detailed configuration diagram of the PWM control means in Figure 1, and Figures 6, 7, and 8. Figure 1 is a vector diagram to explain the operation of the embodiment, Figure 9 is a time chart to explain the operation of PWM control, and Figures 10 and 11 are the magnetic pole position detection means shown in Figure 1. 12 is an overall configuration diagram of another embodiment of the present invention, and FIG. 13 is a time chart 1 to 1 explaining the operation of the embodiment shown in FIG. 12. ,
FIG. 14 and FIG. 15 are time charts for explaining a modified example. 6... Magnetic pole position detection means, 7... Up/down counter, 8... Rotation angle conversion means A, 9... sin θ/c
osO pull, 10... Address conversion means for mode, 12, 17. 20... Switching means, 15... Speed control means, 16... dIIll current command generation means, 22 Slip calculation means, 23. ...Rotation angle conversion means B, 27...
・3/2 phase conversion means, 28...q-axis current control means, 2
9...d-axis current control means, 3o...PWM control means,
30d: PWM signal generating means, 30e: gate circuit, 40: slave microcomputer, 4 host microcomputer.

Claims (1)

[Claims] 1. Speed detection means for detecting the speed of the AC motor based on pulses input from a rotation detector attached to the AC motor to be controlled, and counting the pulses by an up/down counter. a magnetic flux position detection means for detecting the magnetic flux position of the AC motor; a speed control means for inputting a deviation between the detected speed and a given speed command and outputting a q-axis current command according to this deviation; q-axis current control means that takes the current command as input and generates and outputs a q-axis voltage command; and d-axis current command generation means that takes the detected speed as input and outputs a d-axis current command and magnetic flux size in accordance therewith. and,
A switching means for selecting one of the d-axis current command and the d-axis current zero command, and inputting the d-axis current command or the d-axis current zero command through this switching means, and adjusting the d-axis voltage according to the input command. A d-axis current control means that generates and outputs a command, the q-axis voltage command, the d-axis voltage command, and the detected magnetic flux position as inputs, selects a combination of voltage vectors to be applied to the AC motor, and selects the combination of the voltage vector to be applied to the AC motor. a PWM control means that generates and outputs a gate signal for an inverter that drives the AC motor based on the AC motor; a detection current phase conversion means that converts the current into a current and provides negative feedback to each of the current control means of the q-axis and the d-axis, and calculates a sliding speed by inputting the magnitude of the magnetic flux and the q-axis current command, and calculates the sliding speed. A control device for an AC servo motor, comprising a slip rotation angle calculation means for determining a rotation angle of a magnetic flux position corresponding to the rotation angle, and a switching means for selectively adding this slip rotation angle to the detected magnetic flux position. 2. The magnetic flux position detection means includes rotation angle conversion means for multiplying the count value of the up-down counter by a conversion factor that makes the number of pulses output per rotation from the rotation detector correspond to 360 degrees. The control device for an AC servo motor according to claim 1, characterized in that: 3. When the AC motor is a synchronous motor, magnetic pole position detection for correcting the count value of the up/down counter based on a signal representing a specific magnetic pole position input from a magnetic pole position detector attached to the synchronous motor. 3. The control device for an AC servo motor according to claim 1, wherein a circuit is provided, and the magnetic pole position detection circuit can be selectively connected to the up/down counter by a switching means. 4. In order to match the detected speed of the AC motor to be controlled with the given speed command, obtain the q-axis current command and d-axis current command of the AC motor, and calculate P based on these current commands.
PWM data consisting of the combination of voltage vectors and time width that the inverter should generate through WM control is obtained, and this PWM data is
PW that generates the gate signal of the inverter using the M data.
a speed control means for outputting to the M signal generation means; and a current control means for converting the detected current of the AC motor into a q-axis component and a d-axis component to respectively correct the q-axis current command and the d-axis current command. A control device for an AC servo motor comprising: a host microcomputer using a general-purpose microprocessor; and a slave microcomputer using a digital signal processor, the slave microcomputer controlling the speed control means. and performs control calculations related to the current control means,
The host microcomputer samples measurement data such as detected speed and speed command necessary for the control calculation,
A control device for an AC servo motor, characterized in that the PWM data is output to the PWM control signal generating means. 5. The sampling period of the control calculation is set to an integer multiple of the carrier period of the PWM control and the ASR sampling period of the speed control means, respectively, based on the ACR sampling period of the current control means. The control device for an AC servo motor according to claim 4. 6. Data transmission between the host microcomputer and the slave microcomputer, except for measurement data applied to the speed control means, shall be carried out every ACR sampling period, and control calculations applied to the speed control means shall be performed using the integer number. The measured data necessary for the content of the divided control calculations is divided into 1/1 integer parts for each ACR sampling period from the host microcomputer. 5. The control device for an AC servo motor according to claim 4, wherein the control device transmits the information. 7. The PWM signal generating means includes a timer that is reset every carrier period twice the ACR sampling period, a pair of registers, and a pair of comparators for each phase, and the PWM data is Consisting of a pair of time data for each phase that defines the time width for turning on the switching elements of each phase of the inverter from the starting point of the carrier cycle,
Claims 4, 5, and 6, wherein each pair of data is transferred to the register each time the timer is reset.
A control device for an AC servo motor according to any one of the above. 8. The PWM signal generating means includes a timer that is reset every carrier period twice the ACR sampling period, a pair of registers, and a pair of comparators for each phase, and the PWM data is Consisting of a pair of time data for each phase that defines the time width for turning on the switching elements of each phase of the inverter from the starting point of the carrier cycle,
The smaller time data of each pair of time data is transferred to the register each time the timer is reset, and the larger time data of each pair of time data is transferred to the register at each ACR sampling period. 7. The control device for an AC servo motor according to claim 4, wherein the control device executes the transfer when the value is larger than the value of the timer at the time of transfer. 9. Speed detection means for detecting the speed of the AC motor based on pulses input from a rotation detector attached to the AC motor to be controlled; and counting the pulses with an up/down counter to determine the magnetic flux position of the AC motor. a magnetic flux position detection means for detecting the q-axis current command; a speed control means for inputting the deviation between the detected speed and the given speed command and outputting a q-axis current command according to this deviation; q-axis current control means for generating and outputting an axis voltage command; d-axis current command generation means for receiving the detected speed as input and outputting a d-axis current command and magnetic flux size in accordance therewith;
A switching means for selecting one of the d-axis current command and the d-axis current zero command, and inputting the d-axis current command or the d-axis current zero command through this switching means, and adjusting the d-axis voltage according to the input command. A d-axis current control means that generates and outputs a command, the q-axis voltage command, the d-axis voltage command, and the detected magnetic flux position as inputs, selects a combination of voltage vectors to be applied to the AC motor, and selects the combination of the voltage vector to be applied to the AC motor. a PWM control means that generates and outputs a gate signal for an inverter that drives the AC motor based on the AC motor; a detection current phase conversion means that converts the current into a current and provides negative feedback to each of the current control means of the q-axis and the d-axis, and calculates a sliding speed by inputting the magnitude of the magnetic flux and the q-axis current command, and calculates the sliding speed. A control device for an AC servo motor, comprising a slip rotation angle calculation means for calculating a rotation angle of a magnetic flux position corresponding to the magnetic flux position, and a switching means for selectively adding this slip rotation angle to the detected magnetic flux position, the control device comprising: speed control means, the q-axis current control means, and the d
an axis current command generation means, the d-axis current control means, and a portion of the PWM control means that selects a voltage vector;
The detection current phase conversion means, the slip rotation angle calculation means, and each switching means are formed using a digital signal processor, and the detection data from the speed detection means and the magnetic flux position detection means are taken in, and the 1. A control device for an AC servo motor, characterized in that a general-purpose microprocessor outputs voltage vector selection data to a gate circuit that generates a PWM gate signal. 10. When the PWM control selects the combination of the voltage vectors and generates the gate signal, the detected magnetic flux position is set as a reference at least from the time when the magnetic flux position is detected to when the gate signal is output. The control device for an AC servo motor according to any one of claims 1, 2, 3, 4, 5, 6, 7, 8, and 9, characterized in that the correction is performed by adding a phase corresponding to a time delay.