JP3890907B2 - Electric motor drive control device - Google Patents

Description

[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a drive control device for an electric motor, and more particularly to a drive control device for driving a plurality of AC motors by outputting a variable frequency, variable voltage AC from a DC power source using digital servo control.
[0002]
[Prior art]
A drive control device that drives an AC motor by outputting a variable frequency, variable voltage AC from a DC power source using digital servo control generally uses a microcomputer or the like as a current feedback calculation means to perform discrete calculation processing. Is going. In order to drive the electric motor with high controllability up to the high-speed rotation region in the conventional drive control device, as described in, for example, Japanese Patent Application Laid-Open No. 9-47065, a high-speed operation of about 100 μsec is generally used as a calculation cycle of discrete system calculation processing. In addition to applying arithmetic processing, the delay time is minimized as much as possible by compensating for the effect of control dead time caused by discretization. Furthermore, in order to independently drive and control a plurality of motors with high controllability up to the high-speed rotation region, it is necessary to perform high-speed current feedback calculation of about 100 μsec for each motor for the above reasons.
[0003]
[Problems to be solved by the invention]
In the current servo device that performs current feedback control of a plurality of motors by the conventional technology as described above, for example, a current control calculation that performs a next command voltage calculation and an angle correction calculation for each of the plurality of motors every calculation cycle of about 100 μsec. Since time is required, the calculation load becomes very heavy. For this reason, it is necessary to arrange at least one digital arithmetic unit such as a microcomputer having a suitable processing capacity for each of the plurality of electric motors. As an example, when a plurality of three-phase synchronous motors are controlled by vector current feedback, calculation using a 32-bit RISC microcomputer occupies about 70% of the calculation load when the assembler is used. When drive control is performed by one microcomputer, the current feedback calculation of the microcomputer does not fit within one control cycle, and the calculation fails. Therefore, there is a problem that the number of computers increases because it is necessary to provide a dedicated microcomputer for each electric motor. As described above, in the drive control device for driving a plurality of electric motors according to the prior art, when a general microcomputer for driving an electric motor is used, it has been an obstacle to downsizing and cost reduction of the driving device. .
[0004]
In addition, even when the current high-speed motor control microcomputer is applied to the drive control devices for a plurality of electric motors and the drive control of at most two electric motors is performed, the calculation cycle of 100 μsec is required to cope with the high-speed rotation range. If set, there is a problem that the calculation time of the current feedback calculation is not in time within the period and the calculation fails. For example, even when a high-performance 32-bit RISC microcomputer is used, the calculation load per motor is about 50%, and when two motors are driven and controlled, they cannot be controlled within 100 μsec and cannot be controlled.
[0005]
In addition, in order to ensure the synchronism of current acquisition time even when driving multiple m motors with a single microcomputer after limiting the performance and current response characteristics in the high-speed rotation range and extending the calculation cycle. , Out of m motors, it is necessary to simultaneously capture the currents of any two phases of the three-phase currents iu (k), iv (k), iw (k) flowing through the k-th motor. Therefore, if analog / digital (hereinafter abbreviated as A / D) capture timing for each control cycle is defined, the A / D converter needs 2 × m inputs, and the A / D input port is insufficient. was there.
[0006]
The present invention has been made to solve the above problems, and one object of the present invention is to provide a drive control device for an electric motor that can control a plurality of electric motors even with an arithmetic device having a low processing capacity by reducing the calculation load. Therefore, another object of the present invention is to provide a drive control device for an electric motor that does not require many A / D input ports.
[0007]
[Means for Solving the Problems]
  In order to achieve the above object, the present invention is configured as described in the claims. That is, in the first aspect of the present invention, in the drive control apparatus for the motor that commonly performs current feedback control of a plurality of m motors using one digital arithmetic means (for example, a microcomputer),A current feedback control calculation for feedback control of the current flowing through each motor is performed at a predetermined cycle at a different timing for each motor, and a digital output for outputting a calculation result of the current feedback control calculation is performed at a predetermined cycle for each motor. For each of the n digital outputs, the calculation period of the current feedback control is an integer n times the output period of the digital output, and the n digital outputs are output within the same time as the current feedback control calculation period for each motor. Angle correction calculations that perform correction equivalent to the delay time from the current angle reading timing to each digital output in the current feedback calculation cycle of each motorConfigured to do.
[0008]
Further, according to a second aspect of the present invention, in the first aspect, a series of processes of reading current, next (next calculation cycle) voltage command calculation, and digital output for each of the m motors are executed with being shifted from each other. It is composed.
[0010]
【The invention's effect】
  In claim 1, the current feedback calculation means performs heavy load current feedback calculation with a calculation cycle that is n times longer than a digital output cycle such as PWM, and the calculation load is greatly reduced, and calculation is performed at different timings. By doing so, it is possible to disperse the calculation load over time. As a result, even if a microcomputer having a lower processing capability is used, the calculation can be executed without failing, so that the price of the microcomputer can be reduced.
  Furthermore, for each of n digital outputs such as PWM output within the same time as the current feedback control calculation cycle for each motor, this corresponds to the delay time from the current angle reading timing to each digital output in each motor. By performing angle correction calculations for correction within the current feedback calculation cycle of each motor, smooth digital output such as PWM as if the current feedback calculation was performed every output cycle of digital output such as PWM There is an effect that becomes possible.
[0011]
Further, according to claim 2, by executing the current taking times for the m motors while being shifted from each other, the currents to be taken in simultaneously are the three-phase currents iu (k) and iv (k) flowing through the kth motor. ) And iw (k), the current is only for any two phases, and the processing can be performed by the two-channel A / D input ports that can simultaneously capture. As a result, the microcomputer does not need a multi-channel simultaneous analog port, and an effect is obtained that processing can be performed only with the two-channel simultaneous analog port of a normal motor control microcomputer.
[0013]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 is a block diagram showing a configuration of an embodiment of the present invention, and shows a case where two motors A and B are controlled.
In FIG. 1, an electric motor A101 is a synchronous motor, for example, and is driven by an output of a PWM power converter A104 (for example, an inverter circuit). A rotation angle detector A102 (for example, a resolver or an encoder) detects the rotation angle of the electric motor A101 and sends the result to the current angle θre calculation block A111. The current sensor A103 detects a current that flows from the PWM power converter A104 to the u, v, and w phases of the electric motor A101. Similarly, also on the motor B121 side, the rotation angle detector B122, the current sensor B123, the PWM power conversion unit B124, and the current angle θre calculation block B125 perform detection and each calculation for the motor B121 in the same manner as described above.
[0014]
The current control calculation block 105 has a current command value iq * A, id * A and iq * B, id * B (for example, a value determined according to the opening and rotation speed of the accelerator pedal) and an actual value given from the outside. Voltage command values vd ′ * A, vq ′ * A and vd ′ * B, vq ′ * B are calculated based on deviations from the two-phase currents iqA, idA and iqB, idB obtained from the operation. The non-interference calculation block 106 corrects the interference term and calculates the two-phase voltage command values vd * A, vq * A and vd * B, vq * B. The two-phase → three-phase conversion block 107 (in the figure, the two-phase → three-phase is abbreviated as 2φ → 3φ) performs axis conversion and performs three-phase voltage command values vu * A, vv * A, vw * A and vu. * B, vv * B, vw * B are output. With that output, the PWM power converter A104 and the PWM power converter B124 are driven. The PWM power conversion unit A104 and the PWM power conversion unit B124 are composed of, for example, a PWM signal generation unit and an inverter circuit, and create a PWM signal by comparing the triangular wave and the three-phase voltage command value, thereby changing the inverter circuit. Drive to supply necessary current to the motor A101 and the motor B121.
[0015]
The current value A / D conversion unit 109 converts the analog current value detected by the current sensor A 103 and the current sensor B 123 into a digital value. The three-phase → two-phase conversion block 110 performs axis conversion to convert three-phase current values into two-phase currents iqA, idA, iqB, and idB, and sends them to the current control calculation block 105. The current angle θre calculation block A111 and the current angle θre calculation block B125 calculate the current angle θreA and the current angle θreB necessary for the axis conversion based on the outputs of the rotation angle detector A102 and the rotation angle detector B122. The θre correction calculation block A112 and the θre correction calculation block B126 perform advance angle correction for the control dead time for the current angle θreA and the current angle θreB. Further, the sin / cos reference blocks 113 and 114 obtain numerical values necessary for the axis conversion from the current angle θreA and the current angle θreB, and send them to the two-phase → three-phase conversion block 107 and the three-phase → two-phase conversion block 110.
Note that each of the above-described operations and conversion blocks are illustrated separately in the drawing, but can actually be configured by a common computer or the like. Details of the calculation will be described later.
[0016]
(Example)
As an embodiment of the present invention, a system that uses two motors represented by a synchronous motor and reads the current angle using a resolver, an encoder, or the like as the rotation angle detector A102 and rotation angle detector B122 of FIG. 1, or an induction motor In a system that uses two motors typified by and reads the current angle using the same rotation angle sensor as above or calculates the current angle by means of calculation, the current rotation angle of the motor is detected as described above. Then, the phase current value of the motor is read using the current sensor 103 and the current sensor B123, current feedback control calculation processing is performed using digital calculation means typified by a microcomputer or DSP, and PWM is calculated based on the obtained control command value. Control of the converted power converter A104 and the PWM power converter B124 Ri, the case of applying the present invention to the drive control unit supplies electric power to the electric motor.
[0017]
2 to 4 are flowcharts showing the processing procedure in the embodiment of the present invention, FIG. 5 is a waveform diagram for explaining the operation of the embodiment of the present invention, and FIG. 6 shows the processing contents for each interrupt timing of the embodiment. It is a chart. 3 and 4 are connected at the portion {circle around (1)}.
In this embodiment, it is assumed that two 10 kHz PWM timers (triangular wave generation) indicated by 320 in FIG. 5 are operated, and for example, an interrupt occurs every 100 μsec at the timing of the valley of the timer waveform.
[0018]
An arbitrary two-phase current value of the three-phase current flowing through the motor A101 at the first interruption (i = 0 interval) within the k calculation cycle time, for example, iuA, ivA is taken into the current value A / D converter ( At the same time, the current angle θreA is read (steps 201 and 202 in FIG. 3).
[0019]
  Next, in the interrupt period of i = 0, two angle correction calculations are performed together until the output timing of each PWM output that is output twice within the k control period (step 203 in FIG. 3). Two-phase current values iqA and idA are calculated by performing axis conversion of the three-phase current using the current angle θreA (step 204 in FIG. 3), and a current command value iq *A, id *A current control calculation (step 205 in FIG. 3) and a non-interference calculation (step 206 in FIG. 3) are executed using the deviation from A, and the obtained voltage command values vq * A and vd * A are calculated first. The three-phase voltage command values vu * A, vv * A, and vw * A obtained after the axis conversion using the corrected angles up to the two output timings are temporarily stored (step of FIG. 3). 207: The calculation time is 310 in FIG.
[0020]
The non-interference control described above is a vector current control in which current control is performed by separating the motor into a q-axis component orthogonal to the secondary magnetic flux and a d-axis component parallel to the secondary magnetic flux. This is a control in which the d-axis current is used as the q-axis voltage and the q-axis current is used as the d-axis voltage due to the effects of the inductance and the rotation speed, and the interferences are mutually subtracted by calculation in advance.
[0021]
  Also, current value A / D conversion is performed on any two-phase current values, for example, iuB and ivB, among the three-phase currents flowing through the motor B121 in the second interruption (i = 1 section) within the k control cycle time. (312 in FIG. 5) and at the same time the current angle θr at this timeeB(Steps 208 and 209 in FIG. 3).
[0022]
  Next, in the interrupt period of i = 0, two angle correction calculations are performed together until the output timing of each PWM output that is output twice within the k control period (step 210 in FIG. 3). The three-phase current is the current angle θreBA two-phase current value i by performing axis conversion usingqB, IdB(Step 211 in FIG. 3) and the current command value iq *B, id *The current control calculation (step 212 in FIG. 3) and the non-interference calculation (step 213 in FIG. 3) are executed using the deviation from B, and the obtained voltage command values vq * B and vd * B are calculated first. The three-phase voltage command values vu * B, vv * B, and vw * B obtained after the axis conversion using the corrected angles up to the two output timings previously stored are temporarily stored (step of FIG. 3). 214: The calculation time is 313 in FIG.
[0023]
On the other hand, as shown in FIG. 2, the counters i and j are integer values that can take values from 0 to 1, respectively, so as to correspond to the control of two electric motors. Set to.
In the previous interrupt cycle of i = 0, the three-phase voltage command values vu * A (0), vv * A (0), vw * A (0) to the motor A101 and the three-phase voltage command values to the motor B121. By loading vu * B (1), vv * B (1), and vw * B (1) (step 215 in FIG. 4), the three phases calculated within the i = 0 interrupt period within the k control period Voltage command values vu * A (0), vv * A (0), vw * A (0) and a three-phase voltage command value vu * B (calculated within i = 1 interrupt cycle in the k-1 control cycle) 1), vv * B (1) and vw * B (1) are taken in (step 216 in FIG. 4).
By writing this value to the PWM compare register or the like (311 in FIG. 5), processing such as comparing the PWM timer with the PWM compare register value is performed at the timing of the next interrupt cycle i = 1, and the process is sent to the motor A101. PWM output (315 in FIG. 5) and PWM output (317 in FIG. 5) to the motor B121 can be obtained.
[0024]
In the interrupt cycle of i = 1, the three-phase voltage command values vu * A (1), vv * A (1), vw * A (1) to the motor A101 and the three-phase voltage command values to the motor B121. By loading vu * B (0), vv * B (0), vw * B (0), the three-phase voltage command value vu * A ( 1), vv * A (1), vw * A (1) and i = 1 in the k control period, and three-phase voltage command values vu * B (0) and vv * B (0) calculated in the interrupt period , Vw * B (0) is taken in.
By writing this value to the PWM compare register or the like (314 in FIG. 5), processing such as comparing the PWM timer with the PWM conveyor register value at the timing of the next interrupt cycle i = 1 is performed. A PWM output (316 in FIG. 5) and a PWM output (318 in FIG. 5) to the motor B121 can be obtained.
[0025]
When two electric motors are controlled simultaneously as described above, the current calculation cycle is set to twice the PWM interrupt cycle, the calculation is performed with one interrupt cycle shifted from each other, and output at each interrupt cycle. By performing the control dead time correction up to the PWM output timing for the control calculation of each electric motor, the control in the order as shown in FIG. 6 becomes possible. As a result, smooth and stable drive control is possible as if each motor was independently controlled and calculated every interruption period of 100 μsec, and simultaneous input of four inputs is 2 Since each input can be distributed and captured, the number of A / D input ports can be reduced, and the calculation load can be reduced to ½ or less.
[0026]
The present invention can be applied to, for example, control of a traveling motor and a power generation motor in a series hybrid electric vehicle, but is not limited thereto.
Moreover, in the Example, although the case where two electric motors were controlled was illustrated, three or more plurality may be sufficient. In this case, if the number of motors is m, the current feedback calculation cycle is an integer n times the output cycle of the digital output (n ≧ m) and the current is supplied to each motor, as described above. What is necessary is just to comprise so that a feedback calculation may be performed at the timing which does not mutually overlap.
[0027]
Next, details of control in each of the two motors A101 and B121 will be described.
7 to 10 are flowcharts showing a processing procedure in one of the electric motors, and FIG. 11 is a waveform diagram for explaining the operation. The control in each motor is the same except that the timing is shifted from each other, so only one will be described.
First, a schematic procedure is shown in FIG. 7 and FIG. 8 as a means for realizing specific arithmetic processing of the digital servo means. 7 and 8 are connected at the portions {circle around (1)} and {circle around (2)}. At this time, as an example for explanation, the conditions of the current feedback control are a PWM carrier frequency of 10 kHz (cycle = 100 μsec) and a current control cycle of 400 μsec. In the present invention, the PWM carrier frequency and the current control period can be arbitrarily set independently of each other.
[0028]
It is assumed that a 10 kHz PWM timer (triangular wave generation) indicated by 1204 in FIG. 11 is operated and, for example, an interrupt occurs every 100 μsec at the timing of the valley of the timer waveform. In the present embodiment, when the interruption 1205 of the k calculation cycle is generated, first, the phase current read values iu (k) and iv (k) of the motor are set to id (k) using the current angle θre (k) (1228 in FIG. 11). ) And iq (k) (step 1106 in FIG. 7), and current control calculation is performed based on the calculated current two-phase current value and current command value (step 1107 in FIG. 7). Accordingly, after performing non-interference control (step 1108 in FIG. 7), the PWM output command value in the k control period is calculated (step 1109 in FIG. 8). Then, output command value calculation of four PWM pulses to be output in the k control period is started. That is, immediately after the first interrupt 1205 in the k control cycle, the control cycle counter i = 0 is set, and iu (k) and iv (k) are captured and A / D conversion is performed (step 1101 in FIG. 7). Further, the current angle θre (k) is read (step 1102 in FIG. 7), and the correction angle θ′re (k, j) at the output timing of each of the four PWMs output in the k control period is calculated at a time. (Step 1103 in FIG. 7).
[0029]
Hereinafter, the correction angle calculation procedure (steps 1104 and 1105) in step 1103 of FIG. 7 will be described in detail with reference to FIG. First, the current angle θre is captured (step 1121 in FIG. 9). Since the kth current feedback calculation cycle and the four PWM output cycles by the kth calculation are delayed by 1 PWM cycle (100 μsec at 10 kHz), the PWM cycle counter j adds 1 to the control cycle counter i. J = i + 1. This means that the output command value calculation for the kth cycle is always completed one PWM cycle before the output for the kth cycle starts.
[0030]
Further, in order to calculate each of the four PWM output times, from the time average value PWMout (k) (1222 in FIG. 11) of the four PWM output timings to the output timing PWMout (k, j) of each PWM pulse. The delay (advance) time DELTA (k, j) (corresponding to Δτd1 to Δτd4 in FIG. 11) is calculated by the following formula (1) (1122 in FIG. 9).
DELTA (k, j) = (10^-6/ 2) (2 × j−n−1) × (dθre / dt) (Equation 1)
Further, by adding the delay times DELTA (k, j) from the time average value (1222) and adding 1 PWM period 100 μsec for calculation, the four times of the k control period from the timing of reading the current angle θre. The delay times Δτd1 to Δτd4 up to the respective PWM outputs are obtained, multiplied by the average angular velocity dθre / dt during the k control period, and added to the current angle θre, thereby correcting each PWM output time. The subsequent angles θ′re (k, 1) (1234 in FIG. 11) to θ′re (k, 4) (1238 in FIG. 11) are calculated (step 1123 in FIG. 9).
[0031]
At this time, the delay (advance) time from the time average value PWMout (k) of PWM output timing (1222 in FIG. 11) to each PWM pulse output time has a delay direction (positive sign) and an advance direction (negative sign). Since there is a pair, only one of the DELTA (k, j) calculations (1122 in FIG. 9) is performed, and the paired time is the sign inversion of the calculation formula of the angle correction calculation (step 1123 in FIG. 9). Based on θ ′ obtained, sin (θ ′) and cos (θ ′) can be obtained by referring to the map (step 1124 in FIG. 9).
With the above calculation processing, the angle correction calculation can be completed in j = 1 to n / 2 without performing j = 1 to n (1125 in FIG. 9).
[0032]
The non-interference control described above is a vector current control in which current control is performed by separating the motor into a q-axis component orthogonal to the secondary magnetic flux and a d-axis component parallel to the secondary magnetic flux. This is a control in which the d-axis current is used as the q-axis voltage and the q-axis current is used as the d-axis voltage due to the effects of the inductance and the rotation speed, and the interferences are mutually subtracted by calculation in advance.
[0033]
Hereinafter, the calculation of the PWM output command value in step 1109 of FIG. 8 will be described based on FIG.
First, axis conversion is performed on each of the four PWM pulses from j = 1 to 4 using the previously calculated correction angle θ ′ (step 1131 in FIG. 10), and the obtained u, v, w phases are obtained. The voltage command is stored (step 1132 in FIG. 10).
[0034]
In the first 100 μsec interrupt job in the kth control cycle, the command voltage values of the PWM pulses output in the k control cycle as described above are calculated together, and thereafter the PWM compare register every time a periodic interrupt occurs The stored command voltage values are written in order (step 1133 in FIG. 10). A PWM signal (such as 1216 in FIG. 11) can be generated by comparing the value written in the PWM compare register (such as rcu (k, 1) in FIG. 11) with the triangular wave 1204. Thus, an accurate and high-resolution output can be obtained as if the calculation is performed every 100 μsec, even though the current feedback calculation is performed every 400 μsec.
[0035]
Since it is generally considered that the speed change of the motor in the current feedback calculation cycle is small, the calculation of the correction amount of each of the n digital outputs output simultaneously with one cycle of the current feedback calculation is performed by one current feedback calculation. By performing collectively within the period, the current feedback calculation period A and the output period B of the digital output can be made different. As a result, the load on the digital computing means can be reduced without deteriorating the controllability in the case of high speed rotation.
[0036]
The control as described above is performed in each motor, and as described with reference to FIGS. 2 to 6, the current feedback calculations for the respective motors may be performed at a timing that does not overlap each other. That is, the calculation period of current feedback in a digital calculation means such as a computer is an integer n times the output period of digital output, and n digital outputs that are output within the time of one calculation period of the current feedback calculation. Each correction amount is calculated within one calculation cycle of the current feedback calculation, and each of the n digital outputs output within the same time as the current feedback calculation cycle of each motor. What is necessary is just to comprise so that a phase angle correction calculation may be collectively performed within the electric current feedback calculation period of each electric motor.
[0037]
In this embodiment, for example, if n = 4 and the PWM carrier frequency is 10 kHz (cycle = 100 μsec), the PWM waveform follows the command value every 100 μsec even though the current feedback calculation is performed with a control cycle of 400 μsec. The three-phase voltage command value close to the sine wave can be output. This performance becomes more remarkable in the high-speed rotation region.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a configuration of an embodiment of the present invention.
FIG. 2 is a part of a flowchart showing a processing procedure in an embodiment of the present invention.
FIG. 3 is another part of the flowchart showing the processing procedure in the embodiment of the present invention.
FIG. 4 is another part of the flowchart showing the processing procedure in the embodiment of the present invention.
FIG. 5 is a waveform diagram for explaining the operation of the embodiment of the present invention.
FIG. 6 is a table showing processing details for each interrupt timing according to the embodiment.
FIG. 7 is a part of a flowchart showing a processing procedure in each electric motor.
FIG. 8 is another part of a flowchart showing a processing procedure in each electric motor.
FIG. 9 is a flowchart showing details of correction angle calculation contents in step 1103 of FIG. 7;
10 is a flowchart showing details of a calculation content of a PWM output command value in step 1109 of FIG.
11 is a waveform diagram for explaining the operation in FIGS. 7 to 10; FIG.
[Explanation of symbols]
101 ... Electric motor A 102 ... Rotation angle detector A
103 ... Current sensor A 104 ... PWM power conversion unit A
105 ... Current control calculation block 106 ... Non-interference calculation block
107 ... 2 phase → 3 phase conversion block 109 ... Current value A / D converter
110 ... 3 phase → 2 phase conversion block 111 ... Current angle θre calculation block A
112... Θre correction calculation block A
113, 114 ... sin / cos reference block
121 ... Electric motor B 122 ... Rotation angle detector B
123 ... Current sensor B 124 ... PWM power converter B
125 ... Current angle θre calculation block B 126 ... θre correction calculation block B
iq * A, id * A, iq * B, id * B ... Current command value
vd '* A, vq' * A, vd '* B, vq' * B ... Voltage command value
vd * A, vq * A, vd * B, vq * B ... 2-phase voltage command value
vu * A, vv * A, vw * A ... 3-phase voltage command value on the A side
vu * B, vv * B, vw * B ... B-side three-phase voltage command value
iqA, idA, iqB, idB ... Two-phase current
θreA, θreB ... Current angle

Claims (2)

In an electric motor drive control device that commonly performs current feedback control of a plurality of m electric motors using one digital arithmetic means,
A current feedback control calculation for feedback control of the current flowing through each motor is performed at a predetermined cycle at a different timing for each motor,
Digital output for outputting the calculation result of the current feedback control calculation is performed for each motor at a predetermined cycle,
The calculation period of the current feedback control is an integer n times the output period of the digital output,
Further, for each of the n digital outputs that are output within the same time as the current feedback control calculation cycle for each motor, a correction corresponding to the delay time from the current angle reading timing to each digital output in each motor is performed. An electric motor drive control device characterized in that angle correction calculation is performed collectively within a current feedback calculation period of each electric motor .   2. The motor drive control device according to claim 1, wherein a series of processes of current reading, next voltage command calculation, and digital output for each of the m motors are executed while being shifted from each other.

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