JP5473071B2 - Load control device - Google Patents

Description

  The present invention relates to a load control device that performs PWM control of an inverter using a two-phase modulation method.

  A drive load system for driving a plurality of loads such as an electric motor smoothes a DC power supply, inverters for converting DC power to AC power, the same number of loads as the load, and a DC voltage provided between the DC power supply and the inverter. And a plurality of loads. When the smoothing capacitor and the inverter are modularized in the driving load system, the current from each inverter to the smoothing capacitor affects each other, and the ripple current flowing through the smoothing capacitor increases. However, in view of the life of the smoothing capacitor, the loss generated in the smoothing capacitor, the size of the module, etc., it is desirable that the ripple current is low.

  Patent Document 1 discloses an inverter device that can reduce the ripple current of a smoothing capacitor connected to a DC power supply. The inverter device compares a command value of one phase of the first motor with a triangular wave as a carrier signal, and simultaneously compares a command value of another phase with a triangular wave as another carrier signal that is an inverted signal of the triangular wave. To calculate a PWM signal. Similarly, the inverter device compares the command value of one phase of the second motor with a triangular wave that is a carrier signal, and simultaneously, the command value of another phase and the triangular wave as another carrier signal that is an inverted signal of the triangular wave. Are compared to calculate the PWM signal. The inverter device reduces the ripple current of the smoothing capacitor by PWM control of the switching timing of the two inverter bridges corresponding to each of the first motor and the second motor by the two-phase modulation method using these PWM signals.

JP 2008-148395 A JP 2002-300800 A JP 2006-352951 A

  The inverter device described above uses two types of carrier signals (triangular waves) for each motor. In the case of a motor having a plurality of armatures, the inverter device uses two types of carrier signals for each armature. This complicates the circuit that generates the two types of carrier signals. However, the complexity of the circuit is not preferable in terms of a simple system. That is, an apparatus with a simple configuration that can reduce the ripple current is desirable.

  An object of the present invention is to provide a load control device having a simple configuration capable of reducing a ripple current.

  In order to solve the above problems and achieve the object, a load control device according to a first aspect of the present invention includes a power supply unit that outputs a DC voltage (for example, the converter 101 in the embodiment), and the DC The AC is applied to a rotary inductive load (for example, the electric motor 10 in the embodiment) having a plurality of armatures (for example, the armatures A1 and A2 in the embodiment) by converting the voltage into a three-phase AC voltage. A plurality of power conversion units (for example, inverters 103a and 103b in the embodiment) for applying a voltage, and provided in parallel between the power supply unit and the plurality of power conversion units, smooth the DC voltage. A control unit (e.g., PWM control of each of the smoothing unit (e.g., the smoothing capacitor C in the embodiment) and each of the plurality of power conversion units using a two-phase modulation method based on individual carrier signals having the same period) In the embodiment ECU 105), wherein the control unit detects an electrical angle of a rotor (for example, the rotor R in the embodiment) of the rotary inductive load. Voltage phase calculation for calculating the phase on the dq axis of each output voltage based on the voltage on the dq axis output from the unit (for example, the resolver 113 in the embodiment) and the plurality of power conversion units Unit (for example, the output voltage phase calculation unit 121 in the embodiment) and the output voltage phase on the dq axis of the plurality of power conversion units and the electrical angle of the rotor of the rotary inductive load, And a phase setting unit (for example, a carrier phase offset command generation unit 123 in the embodiment) that sets a phase difference of the carrier signal.

  Furthermore, in the load control device according to the second aspect of the present invention, the phase setting unit includes the output voltage phase on the dq axis of the arbitrary one of the plurality of power conversion units and the rotary induction. Output voltage reference electrical angle deriving unit (for example, in the embodiment) for deriving an output voltage reference electrical angle that is an electrical angle of the rotor based on the output voltage phase based on the electrical angle of the rotor of the capacitive load Ripple switching determination processing unit 145) and an output voltage phase difference calculation unit (for example, output in the embodiment) that calculates an output voltage phase difference that is a difference between output voltage phases on the dq axis of the plurality of power conversion units. A voltage phase difference calculating unit 147) and a count value generating unit (for example, converting the electrical angle of the rotor of the rotary inductive load into a count value having the same period as the carrier signal based on the output voltage reference electrical angle The implementation of A difference between the phase of the carrier signal for the one arbitrary power conversion unit and the phase of the carrier signal for the other power conversion unit is the output voltage phase difference. And it changes based on the said count value, It is characterized by the above-mentioned.

  Furthermore, in the load control device according to the third aspect of the present invention, when the phase setting unit changes the phase difference of the carrier signal, the output voltage reference electric power is equal to the number of armatures included in the rotary inductive load. It is characterized by offsetting a phase that is an integral multiple of the angle divided by the period of the angle.

  According to the load control device of the first to third aspects of the invention, the ripple current can be reduced with a simple configuration.

The figure which shows the structure of the electric motor 10 The figure which shows the system containing the load control apparatus of one Embodiment. (A) Graph showing each phase voltage and each interphase voltage when PWM control is performed on the inverters 103a and 103b by the three-phase modulation method, and (b) PWM control is performed on the inverters 103a and 103b by the two-phase modulation method. Graph showing each phase voltage and interphase voltage When the inverters 103a and 103b perform PWM control by the two-phase modulation method, the PWM signal of each phase obtained from each phase command voltage with respect to the inverter carrier signal, the ripple current flowing through the smoothing capacitor C, and the duty of any phase Graph showing signal The figure which shows the timing of the ripple current which arises when the phase of the carrier signal with respect to each of inverter 103a, 103b corresponds, and its countermeasure The block diagram which shows the internal structure of the inverter control part 105I The block diagram which shows the internal structure of the control part 125 The block diagram which shows the internal structure of the output voltage phase calculation part 121 The figure which shows the relationship between the output voltage phase of d axis reference | standard, and the output voltage phase of q axis | shaft reference | standard which the output voltage phase calculation part 121 outputs The figure which shows the timing of the ripple current which generate | occur | produces when the phase of the carrier signal with respect to each of inverter 103a, 103b has shifted | deviated 30 degree | times by an electrical angle. The block diagram which shows the internal structure of the carrier phase offset command generation part 123 The figure which shows an example of the relationship between output voltage reference electrical angle (theta) vsa, output voltage reference electrical angle (theta) vsb, output voltage phase difference (theta) c, and the relationship between ripple phase switching count value CA, CB. The figure which shows each phase relationship of ripple phase switching count value CA and CB when output voltage phase difference (theta) c is 0 degree | times, 30 degree | times, 60 degree | times, 90 degree | times, and 120 degree | times. Flowchart showing operation of carrier phase difference selection processing unit 149 The figure which shows the phase relationship of the ripple phase switching count value CA in the case of (theta) c = 40 degree | times and output voltage phase (theta) va, (theta) vb. The figure which shows the phase relationship between ripple phase switching count value CA in case of (theta) c = 80 degree | times, and output voltage phase (theta) va, (theta) vb.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

  FIG. 2 is a diagram illustrating a system including a load control device according to an embodiment. The load control device shown in FIG. 2 is a device that controls the operation of a rotary inductive load (hereinafter referred to as “motor”) 10 such as a permanent magnet synchronous motor that operates as a motor during power running and operates as a generator during regenerative operation. It is. The electric motor 10 includes one rotor R and two armatures A1 and A2 corresponding to the rotor R. FIG. 1 is a diagram illustrating a configuration of the electric motor 10. As shown in FIG. 1, the armatures A <b> 1 and A <b> 2 are provided on the rotation axis of the rotor R with the rotor R interposed therebetween. In the present embodiment, the electric motor 10 having the two armatures A1 and A2 will be described as an example, but an electric motor in which two windings W1 and W2 are provided in parallel on one armature may be used. In this case, the armature A1 is read as the winding W1, and the armature A2 is read as the winding W2.

  As shown in FIG. 2, the load control device includes a converter 101, a smoothing capacitor C, an inverter 103a corresponding to the armature A1 of the electric motor 10, an inverter 103b corresponding to the armature A2 of the electric motor 10, an ECU 105, Voltage sensor 109, phase current sensors 111ua and 111wa corresponding to armature A1, phase current sensors 111ub and 111wb corresponding to armature A2, and resolver 113 are provided. The converter 101, the smoothing capacitor C, and the inverters 103a and 103b are provided between a DC power source B such as a capacitor and the electric motor 10. Inverters 103 a and 103 b are provided in parallel with converter 101. The smoothing capacitor C is provided in parallel with the converter 101 between the converter 101 and the inverters 103a and 103b.

  Converter 101 steps up or steps down output voltage V1 of DC power supply B. The smoothing capacitor C smoothes the DC voltage. Inverters 103a and 103b have transistors corresponding to respective phases (U phase, V phase, W phase) connected in series in two upper and lower stages, and a free wheel diode connected in parallel with each transistor. Inverters 103a and 103b convert the output voltage V2 of converter 101 into a three-phase alternating current by a transistor switching operation using a PWM signal corresponding to a carrier signal having a set frequency. Inverters 103a and 103b are PWM controlled by a two-phase modulation method. Further, when the electric motor 10 is regenerated, the inverters 103a and 103b convert the three-phase AC voltage generated by the electric motor 10 into DC by the switching operation of the transistors.

  Hereinafter, two-phase modulation performed on the inverters 103a and 103b will be briefly described. FIG. 3 (a) is a graph showing each phase voltage and each inter-phase voltage when PWM control is performed on the inverters 103a and 103b by the three-phase modulation method, and FIG. 3 (b) shows two inverters 103a and 103b. It is a graph which shows each phase voltage at the time of performing PWM control by a phase modulation system, and each interphase voltage. As shown in FIG. 3, since the voltage between the phases does not change between the three-phase modulation and the two-phase modulation, the output of the inverters 103a and 103b to the load (electric motor 10) does not change. However, at the time of two-phase modulation, as shown in FIG. 3B, the duty of any one of the three phases remains 0% or 100%, and such a state is alternated between the phases. Repeat. In the example shown in FIG. 3B, the duty of the V phase is 100% → the duty of the U phase is 0% → the duty of the W phase is 100% → the duty of the V phase is 0% at every electrical angle of 60 degrees with reference to the output voltage phase. → U phase duty 100% → W phase duty 0%. Switching is not performed for the phase where the duty does not change.

  FIG. 4 shows a PWM signal of each phase obtained from each phase command voltage with respect to the inverter carrier signal and a ripple current flowing through the smoothing capacitor C when the inverters 103a and 103b perform PWM control by the two-phase modulation method. It is a graph which shows the duty signal of this phase. As shown in FIG. 4, the duty signal of any phase has periodicity every 60 degrees of electrical angle, and the smoothing capacitor C has peaks and valleys of an inverter carrier signal (hereinafter simply referred to as “carrier signal”). Occurs at the turn. That is, in the example shown in FIG. 4, during the phase A, a ripple current is generated when the carrier signal is a trough, and during the phase B, a ripple current is generated when the carrier signal is a peak.

  FIG. 5 is a diagram showing the timing of the ripple current generated when the phase of the carrier signal for each of the inverters 103a and 103b coincides with the countermeasure. In the example shown in FIG. 5, the phase of the carrier signal for the inverter 103 a and the phase of the carrier signal for the inverter 103 b match, so that the ripple current generated at the same timing is superimposed. However, by offsetting the phase of one of the carrier signals by 180 degrees, the ripple currents corresponding to each of the inverters 103a and 103b do not interfere with each other.

  Voltage sensor 109 provided in the load control device shown in FIG. 2 detects output voltage V2 of converter 101. Phase current sensors 111ua and 111wa detect u-phase current Iua and w-phase current Iwb output from inverter 103a, respectively. Similarly, phase current sensors 111ub and 111wb detect u-phase current Iua and w-phase current Iwb output from inverter 103b, respectively. The resolver 113 detects the electrical angle of the rotor R of the electric motor 10. Signals indicating values detected by the voltage sensor 109, the phase current sensors 111ua, 111wa, 111ub, 111wb and the resolver 113 are sent to the ECU 105. The torque command value Ta for the armature A1 side of the motor 10 and the torque command value Tb for the armature A2 side of the motor 10 are also input to the ECU 105 from the outside.

  The ECU 105 is, for example, a CPU, and performs PWM control of switching of each transistor constituting the converter 101 and the inverters 103a and 103b by a two-phase modulation method. The ECU 105 includes a converter control unit 105C and an inverter control unit 105I. Converter control unit 105 </ b> C performs PWM control of switching of each transistor constituting converter 101. Further, the inverter control unit 105I performs PWM control of switching of each transistor constituting the inverters 103a and 103b by a two-phase modulation method.

  FIG. 6 is a block diagram showing the internal configuration of the inverter control unit 105I. As illustrated in FIG. 6, the inverter control unit 105 </ b> I includes an output voltage phase calculation unit 121, a carrier phase offset command generation unit 123, and a control unit 125. Hereinafter, each component will be described in detail.

  FIG. 7 is a block diagram illustrating an internal configuration of the control unit 125. As shown in FIG. 7, the control unit 125 includes a first control unit 125a and a second control unit 125b. The first control unit 125a and the second control unit 125b include an angular velocity calculation unit 151, a current command calculation unit 153, a three-phase-dq conversion unit 155, a current FB control unit 157, and a dq-3 phase conversion unit 159. , And a PWM control unit 161.

  The first control unit 125a includes the torque command value Ta, the detected value of the electrical angle θ of the rotor R of the electric motor 10 detected by the resolver 113, the u-phase current Iua and w-phase detected by the phase current sensors 111ua and 111wa. Each value of the current Iwa, the value of the output voltage V2 of the converter 101 detected by the voltage sensor 109, and the carrier phase offset command generated by the carrier phase offset command generation unit 123 are input. Similarly, the second control unit 125b includes the torque command value Tb, the detected value of the electrical angle θ of the rotor R of the electric motor 10, the u-phase current Iub and the w-phase current Iwb detected by the phase current sensors 111ub and 111wb. The value, the value of the output voltage V2 of the converter 101 detected by the voltage sensor 109, and the carrier phase offset command generated by the carrier phase offset command generation unit 123 are input.

  Hereinafter, each component provided in each of the first control unit 125a and the second control unit 125b will be described. Note that “a” attached to the reference numeral in FIG. 7 corresponds to the first control unit 125a, and “b” corresponds to the second control unit 125b, but the first control unit 125a and the second control unit 125b Since the operations are common, “a” and “b” are omitted in the following description.

  The angular velocity calculation unit 151 calculates the angular velocity ω of the rotor R of the electric motor 10 by time-differentiating the detected value of the electric angle θ of the rotor R of the electric motor 10. The angular velocity ω calculated by the angular velocity calculator 151 is input to the current command calculator 153. Based on the torque command value T and the angular velocity ω of the rotor R of the electric motor 10, the current command calculation unit 153 supplies a current (hereinafter referred to as “d-axis armature”) to the d-axis side armature. The command value Id_c of the d-axis current ”) and the command value Iq_c of the current (hereinafter referred to as“ q-axis current ”) that flows through the q-axis side armature (hereinafter referred to as“ q-axis armature ”) are calculated.

  The three-phase-dq conversion unit 155 is configured based on each value of the u-phase current Iu and the w-phase current Iw detected by the phase current sensors 111u and 111w and the detected value of the electrical angle θ of the rotor R of the electric motor 10. Phase-dq conversion is performed to calculate a detected value Id_s of the d-axis current and a detected value Iq_s of the q-axis current. The current FB control unit 157 is configured to reduce the deviation ΔId between the d-axis current command value Id_c and the detection value Id_s and the deviation between the q-axis current command value Iq_c and the detection value Iq_s ΔIq. Hereinafter, a command value Vd_c of “d-axis voltage”) and a command value Vq_c of a terminal voltage of the q-axis armature (hereinafter referred to as “q-axis voltage”) are determined.

  The dq-3 phase conversion unit 159 uses the d-axis voltage command value Vd_c and the q-axis voltage command value Vq_c determined by the current FB control unit 157 and the detected value of the electrical angle θ of the rotor R of the electric motor 10. Based on this, dq-3 phase conversion is performed to derive the command values of the three-phase voltages Vu, Vv, Vw. The PWM control unit 161 performs PWM switching of the transistors constituting the inverters 103a and 103b based on the command values of the three-phase voltages Vu, Vv, and Vw derived from the dq-3 phase conversion unit 159 and the carrier phase offset command. Control.

  The output voltage phase calculation unit 121 included in the inverter control unit 105I calculates the output voltage phase θva based on the d-axis voltage command value Vda_c and the q-axis voltage command value Vqa_c for the armature A1 of the electric motor 10. Similarly, the output voltage phase calculation unit 121 calculates the output voltage phase θvb based on the command value Vdb_c of the d-axis voltage and the command value Vqb_c of the q-axis voltage for the armature A2 of the electric motor 10. FIG. 8 is a block diagram showing an internal configuration of the output voltage phase calculation unit 121. As shown in FIG. 8, the output voltage phase calculation unit 121 includes a means 131 for calculating the d-axis reference output voltage phase θva, a means 133 for calculating the d-axis reference output voltage phase θvb, a 90-degree shifter 135, including. The 90-degree shifter 135 delays the phases of the d-axis reference output voltage phase θva and the d-axis reference output voltage phase θvb by 90 degrees, respectively. Therefore, as shown in FIG. 9, the output voltage phases θva and θvb output from the output voltage phase calculation unit 121 are phases of the output voltage with reference to the q axis.

  The carrier phase offset command generation unit 123 included in the inverter control unit 105I derives a carrier phase offset command input to the control unit 125. FIG. 10 is a diagram showing the timing of the ripple current generated when the phase of the carrier signal for each of the inverters 103a and 103b is shifted by 30 degrees in electrical angle. In the example shown in FIG. 10, there are four combinations of ripple signal generation timings (1) to (4), and each combination comes in a cycle of an electrical angle of 30 degrees. In the phase (1) and the phase (3), a ripple current is generated when one carrier signal becomes a peak and a ripple current is generated when the other carrier signal becomes a valley. The timing of occurrence does not overlap. On the other hand, in phase (2), a ripple current is generated when one carrier signal has a peak, and a ripple current is generated when the other carrier signal also has a peak. Therefore, the ripple current generation timings coincide. Further, in phase (4), a ripple current is generated when one carrier signal is a trough, and a ripple current is generated when the other carrier signal is also a trough, so that the generation timing of the ripple current coincides.

  In the example shown in FIG. 10, the overlap of ripple current is prevented by offsetting the phase of one of the carrier signals by 180 degrees in the period of phase (2) and phase (4) every 30 degrees of electrical angle. Can do. The offset phase angle at the time of offset is a number obtained by dividing 360 degrees by the number of armatures included in the electric motor 10. In the present embodiment, since the electric motor 10 has two armatures A1 and A2, 360/2 = 180 degrees is the offset phase angle.

  The carrier phase offset command generation unit 123 determines which phase period is to be offset in accordance with the relationship between the phase of the carrier signal and the electrical angle for each of the inverters 103a and 103b. That is, according to the determination, the carrier phase offset command generation unit 123 outputs a carrier phase offset command. FIG. 11 is a block diagram illustrating an internal configuration of the carrier phase offset command generation unit 123. As shown in FIG. 11, the carrier phase offset command generation unit 123 includes an output voltage reference electrical angle derivation unit 141, a ripple switching determination processing unit 145, an output voltage phase difference calculation unit 147, and a carrier phase difference selection processing unit 149. Including. Hereinafter, each component will be described in detail with reference to FIGS. FIG. 12 is a diagram illustrating an example of the relationship between the output voltage reference electrical angle θvsa and the ripple phase switching count value CA.

  The output voltage reference electrical angle deriving unit 141 derives the output voltage reference electrical angle θvsa by adding the electrical angle θ of the rotor R of the motor 10 to the output voltage phase θva. The output voltage reference electrical angle θvsa is the electrical angle θ of the rotor R of the electric motor 10 with respect to the output voltage phase θva. Note that the cycle of the output voltage reference electrical angle θvsa is 360 degrees.

  The ripple switching determination processing unit 145 converts the electrical angle θ of the rotor R of the electric motor 10 into a count value (ripple phase switching count value CA) having a period of 60 degrees based on the output voltage reference electrical angle θvsa. The range of the ripple phase switching count value CA is 0 to 60 degrees. The output voltage phase difference calculator 147 calculates a difference θc between the output voltage phase θva and the output voltage phase θvb (hereinafter referred to as “output voltage phase difference”).

  As shown in FIG. 4, the generation timing of the ripple current with respect to the carrier signal is different for each electrical angle of 60 degrees. Accordingly, the ripple current generation timing varies every 60 degrees even on the output voltage phases θva and θvb, and the cycle thereof is an electrical angle of 120 degrees. For this reason, the range of the output voltage phase difference θc is 0 degree to 120 degrees. FIG. 13 shows the phase relationships of the output voltage phases θva and θvb when the output voltage phase difference θc is 0 degree, 30 degrees, 60 degrees, 90 degrees and 120 degrees.

  The carrier phase difference selection processing unit 149 commands the carrier signal phase offset to each of the inverters 103a and 103b according to the ripple phase switching count value CA and the output voltage phase difference θc derived by the ripple switching determination processing unit 145. (Carrier phase offset command) is generated. FIG. 14 is a flowchart showing the operation of the carrier phase difference selection processing unit 149.

  As shown in FIG. 14, the carrier phase difference selection processing unit 149 determines whether or not the output voltage phase difference θc is 60 degrees (θc = 60 degrees) (step S103). If θc = 60 degrees, step S105 is performed. If θc = 60 degrees (θc ≠ 60 degrees), the process proceeds to step S107. When θc = 60 degrees, as shown in phase (1) or phase (3) of FIG. 10, the generation timing of the ripple current in the phase of the two carrier signals is shifted by 180 degrees on the carrier phase. Therefore, in step S105, the carrier phase difference selection processing unit 149 generates a carrier phase offset command (carrier offset 0) that sets the phase difference between the two carrier signals to 0 degrees.

  In step S107, the carrier phase difference selection processing unit 149 determines whether or not the output voltage phase difference θc is less than 60 degrees (θc <60 degrees). If θc <60 degrees, the process proceeds to step S109, and θc> 60. If so, the process proceeds to step S115. In step S109, the carrier phase difference selection processing unit 149 determines whether or not the ripple phase switching count value CA is less than the output voltage phase difference θc (CA <θc). If CA <θc, the process proceeds to step S111. If CA> θc, the process proceeds to step S113. FIG. 15 is a diagram showing the phase relationship between the ripple phase switching count value CA and the output voltage phases θva and θvb when θc = 40 degrees. In the phase (1) shown in FIG. 15, CA <θc, and in the phase (2), CA> θc.

  When θc <60 degrees and CA <θc, the ripple current generation timing in the phase of the two carrier signals is shifted by 180 degrees on the carrier phase. Accordingly, in step S111, the carrier phase difference selection processing unit 149 generates a carrier phase offset command (carrier offset 0) that sets the phase difference between the two carrier signals to 0 degrees. On the other hand, when θc <60 degrees and CA> θc, as shown in phase (2) or phase (4) in FIG. 10, the generation timing of the ripple current in the phase of the two carrier signals is the same. Therefore, in step S113, the carrier phase difference selection processing unit 149 generates a carrier phase offset command (carrier offset 180) that prevents the ripple current from overlapping by offsetting the phase of one of the carrier signals by 180 degrees.

  In step S115, the carrier phase difference selection processing unit 149 determines whether or not the ripple phase switching count value CA is less than “output voltage phase difference θc−60” (CA <θc−60), and CA <θc−60. If so, the process proceeds to step S117, and if CA> θc-60, the process proceeds to step S119. FIG. 16 is a diagram showing a phase relationship between the ripple phase switching count value CA and the output voltage phases θva and θvb when θc = 80 degrees. In phase (3) shown in FIG. 16, CA <θc-60, and in phase (4), CA> θc-60.

  When θc> 60 degrees and CA <θc−60, the ripple current generation timing in the phase of the two carrier signals is the same. Therefore, in step S117, the carrier phase difference selection processing unit 149 generates a carrier phase offset command (carrier offset 180) that prevents the ripple current from overlapping by offsetting the phase of any one of the carrier signals by 180 degrees. On the other hand, when θc> 60 degrees and CA> θc−60, the ripple current generation timing in the phase of the two carrier signals is shifted by 180 degrees on the carrier phase. Accordingly, in step S119, the carrier phase difference selection processing unit 149 generates a carrier phase offset command (carrier offset 0) that sets the phase difference between the two carrier signals to 0 degrees.

  The carrier phase offset command generated by the carrier phase difference selection processing unit 149 is output from the carrier phase offset command generating unit 123 and input to the control unit 125 of the inverter control unit 105I.

  As described above, the generation timing of the ripple current is switched every 60 degrees on the phase of the carrier signal for each of the inverters 103a and 103b PWM-controlled by the two-phase modulation method. In the present embodiment, in view of such characteristics, it is determined whether or not the ripple current generation timings in the phases of the two carrier signals overlap according to the output voltage phase difference θc and the ripple phase switching count value CA. The As a result of the determination, if the ripple current generation timing overlaps, the ripple current overlap is prevented by offsetting the phase of one of the carrier signals. Thus, by preventing duplication of ripple current, the ripple current can be reduced without increasing the types of carrier signals.

  In this embodiment, since the duty switching period (duty period of 0% or 100%) for the inverters 103a and 103b is 60 degrees in electrical angle, the cycle in which the ripple current generation timing is switched is 60 in electrical angle. However, the duty switching period may be any value between 0 and 120 degrees.

  In the present embodiment, the number of armatures provided in the electric motor 10 is two. However, the number of armatures is not limited to two and may be three or more.

10 Rotary inductive load (electric motor)
A1, A2 Armature R Rotor B DC power supply 101 Converter C Smoothing capacitors 103a, 103b Inverter 105 ECU
109 Voltage sensor 111ua, 111wa, 111ub, 111wb Phase current sensor 113 Resolver 105C Converter control unit 105I Inverter control unit 121 Output voltage phase calculation unit 123 Carrier phase offset command generation unit 125 Control unit 125a First control unit 125b Second control unit 151 Angular velocity calculation unit 153 Current command calculation unit 155 3-phase-dq conversion unit 157 Current FB control unit 159 dq-3 phase conversion unit 161 PWM control unit 131 Means for calculating d-axis reference output voltage phase θva 133 d-axis reference output Means 135 for calculating voltage phase θvb 135 90-degree shifter 141 Output voltage reference electrical angle derivation unit 145 Ripple switching determination processing unit 147 Output voltage phase difference calculation unit 149 Carrier phase difference selection processing unit

Claims (3)

A power supply unit that outputs a DC voltage;
A plurality of power converters for converting the DC voltage into a three-phase AC voltage and applying the AC voltage to a rotary inductive load having a plurality of armatures;
A smoothing unit that is provided in parallel between the power supply unit and the plurality of power conversion units, and smoothes the DC voltage;
Each of the plurality of power conversion units is a load control device including a control unit that performs PWM control using a two-phase modulation method based on individual carrier signals having the same period,
The controller is
A rotor angle detector for detecting an electrical angle of the rotor of the rotary inductive load;
An output voltage phase calculation unit that calculates a phase of each output voltage on the dq axis based on each voltage on the dq axis output by the plurality of power conversion units;
A phase setting unit that sets a phase difference of the carrier signal according to an output voltage phase on a dq axis of the plurality of power conversion units and an electrical angle of a rotor of the rotary inductive load;
A load control device comprising: The load control device according to claim 1,
The phase setting unit includes:
Based on the output voltage phase on the dq axis of any one of the plurality of power conversion units and the electrical angle of the rotor of the rotary inductive load, the output voltage phase as a reference An output voltage reference electrical angle deriving unit for deriving an output voltage reference electrical angle that is an electrical angle of the rotor;
An output voltage phase difference calculation unit that calculates an output voltage phase difference that is a difference in output voltage phase on the dq axis of the plurality of power conversion units;
A count value generation unit that converts the electrical angle of the rotor of the rotary inductive load into a count value having the same cycle as the carrier signal, based on the output voltage reference electrical angle;
Load control, wherein a difference between a phase of a carrier signal for the arbitrary one power conversion unit and a phase of a carrier signal for the other power conversion unit is changed based on the output voltage phase difference and the count value apparatus. The load control device according to claim 1 or 2,
When changing the phase difference of the carrier signal, the phase setting unit offsets a phase that is an integral multiple of an angle obtained by dividing the period of the output voltage reference electrical angle by the number of armatures included in the rotary inductive load. A load control device.

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