JP7221424B2 - Inverter control device and motor drive device - Google Patents

Description

The present disclosure relates to an inverter control device that controls an inverter, and an electric motor drive device that includes the inverter control device.

An electric motor that provides propulsion to the electric vehicle is driven by an inverter. Beacons, which are receivers for various traffic signals, are placed on the track on which the electric train runs. In order to prevent these ground coils from malfunctioning, electric vehicles are regulated against leakage noise. Patent Document 1 below describes that a hollow core made of a ferromagnetic material such as ferrite or amorphous metal is arranged around wiring that connects an inverter and an electric motor in order to suppress common mode noise. It is

JP-A-2004-187368

However, the space under the floor of an electric vehicle is limited. Therefore, in some cases, there is not enough space for additionally installing a filter component such as the core described in Patent Document 1 above. In such a case, it may be necessary to review the specifications of the inverter control device in order to secure space for additionally installing the filter component.

Also, the filter characteristics of the filter component must be determined according to the impedance including the motor. However, the manufacturer of the electric motor and the manufacturer of the inverter control device are not always the same. In this case, relying on the additional installation of the filter component increases the design man-hours of the manufacturer of the inverter control device and increases the adjustment work on the actual vehicle. For this reason, it is desirable to suppress leakage noise without relying on additional filter components.

The present disclosure has been made in view of the above, and an object of the present disclosure is to obtain an inverter control device capable of suppressing leakage noise without relying on additionally installing a filter component.

In order to solve the above-described problems and achieve an object, the present disclosure is an inverter control device that controls an inverter that drives a three-phase electric motor by pulse width modulation. The inverter control device includes a carrier generation section that generates a three-phase carrier wave, and a pulse width modulation control section that controls the switching state of the inverter by comparing the carrier wave and the modulated wave. The carrier wave generator includes a phase calculator that calculates first to third carrier wave phases based on the carrier wave frequency command and the motor frequency, and a carrier wave that outputs a three-phase carrier wave based on the first to third carrier wave phases. and an output unit. The first to third carrier wave phases have a phase difference with each other, and the phase difference also changes continuously as the motor frequency changes.

According to the inverter control device according to the present disclosure, it is possible to suppress leakage noise without relying on additionally installing a filter component.

1 is a diagram showing a configuration example of an electric motor drive device including an inverter control device according to Embodiment 1; FIG. Diagram showing a configuration example of a general inverter main circuit A diagram showing the equivalent circuit of FIG. 1 used to explain the leakage current A diagram used for explaining a method of generating a pulse width modulation (PWM) control signal given to the semiconductor element of each arm shown in FIG. A diagram showing a PWM control signal generated by each phase voltage command shown in FIG. FIG. 6 shows a common mode voltage caused by the PWM control signal shown in FIG. 5; A diagram showing an example of each phase voltage command with a modulation rate different from that in FIG. A diagram showing a PWM control signal generated by each phase voltage command shown in FIG. 9 is a diagram showing common mode voltages caused by the PWM control signals shown in FIG. 8; FIG. A diagram showing an example of waveforms when the carrier wave phase of each phase is shifted by 120° from each other in each phase voltage command and carrier wave shown in FIG. A diagram showing a PWM control signal generated by each phase voltage command and each phase carrier wave shown in FIG. FIG. 12 shows an example of a common mode voltage waveform generated by the PWM control signal shown in FIG. 11; A diagram showing an example of the harmonic distribution of the motor current according to the conventional control when the carrier wave of each phase is the same. A diagram showing an example of the harmonic distribution of the motor current according to the conventional control when the carrier wave phases of the phases are mutually shifted by 120°. A diagram showing an example of common-mode voltage waveforms according to conventional control when carrier wave phases of respective phases are shifted from each other by 120°. FIG. 3 is a diagram used for explaining the concept of the control method in Embodiment 1. FIG. FIG. 3 is a diagram showing a configuration example of a carrier generation unit according to Embodiment 1; FIG. 10 is a diagram showing a configuration example of a carrier generation unit according to a modification of Embodiment 1; FIG. 2 is a block diagram showing an example of a hardware configuration that implements the functions of the inverter control device according to Embodiment 1; FIG. 4 is a block diagram showing another example of a hardware configuration that implements the functions of the inverter control device according to Embodiment 1; FIG. 10 is a diagram showing a configuration example of a carrier generation unit according to Embodiment 2; FIG. 11 is a diagram showing a configuration example of a carrier generation unit according to Embodiment 3; A diagram showing the detailed configuration of the frequency modulation section shown in FIG. FIG. 23 is a diagram showing an output image of the frequency modulation amount output from the frequency modulation section shown in FIG.

An inverter control device and an electric motor drive device according to embodiments of the present disclosure will be described below in detail with reference to the accompanying drawings. In the following embodiments, an electric motor drive device applied to a railway system will be described as an example, but application to other uses is not excluded.

Embodiment 1.
FIG. 1 is a diagram showing a configuration example of an electric motor drive device 1 including an inverter control device 5 according to Embodiment 1. As shown in FIG. An electric motor drive device 1 according to Embodiment 1 includes a filter capacitor 3, an inverter 4, and an inverter control device 5, as shown in FIG. Inverter 4 is connected to electric motor 105 . The electric motor 105 is a three-phase electric motor mounted on the electric vehicle.

DC power supplied from the overhead wire 101 is supplied to the electric motor driving device 1 via the current collector 102 and the filter reactor 2 . There is a substation (not shown) at the end of the overhead wire 101 , and the overhead wire 101 is positioned as an external power source when viewed from the electric motor drive device 1 . The overhead wire voltage, which is the voltage of the overhead wire 101 applied to the current collector 102, and each conversion capacity of the inverter 4 differ depending on the driving method. The range of overhead line voltage is approximately 600 to 3000 [V]. Also, the range of conversion capacity is several tens to several hundred [kVA].

A positive terminal P of the electric motor drive device 1 is connected to the filter reactor 2 . A negative terminal N of the electric motor drive device 1 is connected to the rail 104 via the wheel 103 . As a result, a DC current generated by the DC power supplied from the overhead wire 101 flows through the filter reactor 2, the motor driving device 1, the motor 105, the wheels 103 and the rails 104, and returns to the substation.

Note that FIG. 1 shows an overhead wire as the overhead wire 101 and a pantograph-like current collector as the current collector 102, but the present invention is not limited to these. The overhead wire 101 may be a third rail used in a subway or the like, and the current collector 102 may be a third rail current collector. Moreover, although FIG. 1 shows the case where the overhead wire 101 is a DC overhead wire, the overhead wire 101 may be an AC overhead wire. When the overhead wire 101 is an AC overhead wire, a transformer for stepping down the AC voltage to be received is provided instead of the filter reactor 2, and the AC voltage output from the transformer is converted to a DC voltage in the subsequent stage of the transformer. A converter is provided for converting to

Moreover, although the filter reactor 2 is shown as an external component of the electric motor drive device 1 in FIG. 1 , it is not limited to this. Filter reactor 2 may be provided inside electric motor drive device 1 .

The filter capacitor 3 is connected between the positive terminal P and the negative terminal N inside the motor drive device 1 . Thereby, the filter capacitor 3 is connected in parallel across each end of the inverter 4 on the input side of the inverter 4 .

Filter capacitor 3 smoothes the applied DC voltage. Also, the filter capacitor 3 is connected to the filter reactor 2 and forms an LC filter circuit together with the filter reactor 2 . This LC filter circuit suppresses a surge voltage flowing from the overhead line 101 side. Also, the LC filter circuit suppresses the magnitude of the ripple component of the current flowing through the inverter 4 . The inverter 4 is a power conversion circuit that supplies electric power to the electric motor 105 .

The operation of inverter 4 is controlled by inverter control device 5 . Inverter 4 converts the DC voltage of filter capacitor 3 into an AC voltage having an arbitrary voltage value and an arbitrary frequency, and applies the AC voltage to electric motor 105 .

The inverter control device 5 includes a voltage command calculation section 6 , a PWM control section 7 and a carrier wave generation section 8 . Information about the torque command and information about the electric motor frequency are input to the inverter control device 5 . A voltage command calculator 6 calculates a voltage command based on the torque command and the motor frequency. A carrier wave generator 8 generates a carrier wave based on the motor frequency. PWM control unit 7 generates a PWM control signal for controlling the switching elements of inverter 4 based on the voltage command and the carrier wave, and outputs the generated PWM control signal to inverter 4 . A PWM control signal is generated by comparing the amplitude of the voltage command and the carrier wave. A method of generating the PWM control signal will be described later.

FIG. 2 is a diagram showing a configuration example of a general inverter main circuit. The inverter main circuit includes upper arm semiconductor elements UPI, VPI, WPI and lower arm semiconductor elements UNI, VNI, WNI.

Semiconductor element UPI and semiconductor element UNI are connected in series to form a U-phase leg. Semiconductor element VPI and semiconductor element VNI are connected in series to form a V-phase leg. Semiconductor element WPI and semiconductor element WNI are connected in series to form a W-phase leg. The U-phase, V-phase and W-phase legs are connected in parallel to form a three-phase bridge circuit.

FIG. 3 is a diagram showing an equivalent circuit of FIG. 1 used for explanation of leakage current. In FIG. 3, the electric motor 105 is represented by a circuit symbol of inductance. Regardless of the type of motor, when a three-phase motor is driven by a three-phase inverter, the neutral point potential of the three-phase motor fluctuates. Therefore, as shown in FIG. 3, an equivalent circuit is formed in which the stray capacitance 34 is connected between the neutral point potential 32 of the electric motor 105 and the reference potential 31, which is the ground potential. The neutral point potential 32 contains high frequency components resulting from PWM control. Therefore, when a U-phase voltage 33U, a V-phase voltage 33V, and a W-phase voltage 33W are applied to the electric motor 105, a leakage current 35 flows through the stray capacitance .

A potential difference between the neutral point potential 32 and the reference potential 31 is called a "common mode voltage". For a three-phase inverter, the common mode voltage is calculated as (Vu+Vv+Vw)/3. However, Vu is the U-phase voltage 33U, Vv is the V-phase voltage 33V, and Vw is the W-phase voltage 33W.

FIG. 4 is a diagram used for explaining a method of generating PWM control signals to be applied to the semiconductor elements of each arm shown in FIG. FIG. 4 shows a U-phase voltage command 26U, a V-phase voltage command 26V, and a W-phase voltage command 26W, which are sinusoidal waves, and a carrier wave 28, which is a triangular wave. The horizontal axis represents the phase angle, and the vertical axis represents the amplitude value. 1 [Vpu] on the vertical axis corresponds to 1/2 of the voltage amplitude applied to the inverter 4 , in other words, 1/2 of the DC voltage, which is the voltage of the filter capacitor 3 .

FIG. 5 is a diagram showing PWM control signals generated by each phase voltage command shown in FIG. The U-phase PWM control signal, the V-phase PWM control signal, and the W-phase PWM control signal are shown in this order from the top. The horizontal axis in FIG. 5 is the same phase angle as in FIG. FIG. 5 shows the waveform of the PWM control signal when the modulation factor is 0.8. In this specification, the modulation factor is defined as the ratio of 1/2 of the DC voltage applied to the inverter 4 to the amplitude of the AC phase voltage applied to the motor 105 by the inverter 4 . Note that the definition here is just an example, and the modulation rate may be defined in any way.

The PWM control unit 7 compares the U-phase voltage command 26U and the carrier wave 28 which is a triangular wave signal. When the U-phase voltage command 26U is greater than the carrier wave 28, it is "ON", and when the U-phase voltage command 26U is less than the carrier wave 28, it is "OFF". Thus, the U-phase PWM control signal shown in the upper part of FIG. 5 is generated. The V-phase PWM control signal and the W-phase PWM control signal are generated by comparing each of the V-phase voltage command 26V and the W-phase voltage command 26W with the carrier wave 28, similarly to the U-phase PWM control signal. The V-phase PWM control signal and W-phase PWM control signal generated at this time are shown in the middle part and the lower part of FIG. 5, respectively.

FIG. 6 is a diagram showing common mode voltages produced by the PWM control signals shown in FIG. In FIG. 6, the horizontal axis represents the phase angle, and the vertical axis represents the amplitude of the common mode voltage. As shown in FIG. 6, the common mode voltage when the modulation factor is relatively large has a waveform that changes step by step between levels of ±1 [Vpu] or ±1/3 [Vpu].

A leakage current flows whenever there is a change in the common mode voltage. Further, the amplitude of the leakage current increases as the amount of change (voltage change rate) per unit time of the common mode voltage increases. Therefore, in order to suppress the leakage current 35 flowing through the stray capacitance 34, it is important to reduce the rate of change of the common mode.

FIG. 7 is a diagram showing an example of each phase voltage command with a modulation rate different from that in FIG. FIG. 7 shows each phase voltage command when the modulation factor is 0.1. The distinction between solid lines, broken lines and dashed-dotted lines is the same as in FIG. That is, the solid line indicates the U-phase voltage command 26U, the dashed line indicates the V-phase voltage command 26V, and the dashed line indicates the W-phase voltage command 26W.

FIG. 8 is a diagram showing PWM control signals generated by each phase voltage command shown in FIG. FIG. 8 shows the waveform of the PWM control signal when the modulation factor is 0.1. The arrangement of the PWM control signals is the same as in FIG. 5, and is shown in the order of the U-phase PWM control signal, the V-phase PWM control signal, and the W-phase PWM control signal from the top side.

When the modulation factor is a value close to 0, as shown in FIG. 8, the difference between the PWM control signals of each phase becomes small, and the output of the inverter 4 is mostly in the state of zero voltage vector. Here, the zero voltage vector state means a state in which all the upper arm side elements are turned on or all the lower arm side elements are turned on.

FIG. 9 is a diagram showing common mode voltages produced by the PWM control signals shown in FIG. As shown in FIG. 9, when the modulation factor is close to 0, the common mode voltage has a waveform that alternates between levels of ±1 [Vpu] in a short period of time. As a result, the voltage change rate is increased and the leakage current 35 flowing through the stray capacitance 34 is increased as compared with the case where the modulation factor is 0.8, which is relatively large.

FIG. 10 is a diagram showing an example of waveforms when the phase voltage commands and carrier waves shown in FIG. 7 are shifted from each other by 120 degrees. In the example shown in FIG. 10, the V-phase carrier wave 28V indicated by the dashed line is 120° behind the U-phase carrier wave 28U indicated by the solid line. Also, the W-phase carrier wave 28W indicated by the dashed line leads the U-phase carrier wave 28U by 120° in phase. It should be noted that the phase leading by 120° is equivalent to the phase lagging by 240°.

11 is a diagram showing a PWM control signal generated by each phase voltage command and each phase carrier wave shown in FIG. 10. FIG. The waveforms shown in FIG. 11 indicate that the timings of turning on or off the PWM control signals for each phase are shifted from each other. Therefore, when the carrier wave phases of the respective phases are shifted from each other by 120°, unlike FIG. 8, even if the modulation factor takes a value close to 0, it can be understood that a zero voltage vector does not occur.

12 is a diagram showing an example of common mode voltage waveforms generated by the PWM control signal shown in FIG. 11. FIG. According to FIG. 12, even when the modulation factor is close to 0, the common mode voltage has a waveform that alternates between levels of ±1/3 [Vpu]. Therefore, compared to the example of FIG. 9 in which each phase has the same carrier wave phase, the voltage change rate is smaller, so the leakage current 35 flowing through the stray capacitance 34 is reduced.

FIG. 13 is a diagram showing an example of the harmonic distribution of the motor current according to the conventional control when the carrier wave of each phase is the same. The vertical axis in FIG. 13 represents the current amplitude. Control parameters for the waveform in FIG. 13 are a carrier frequency of 800 Hz, a drive frequency of 20 Hz, and a modulation rate of 0.2. According to FIG. 13, the 20 Hz component of the drive frequency appears large, and the 2f component (1600 Hz) of the carrier frequency appears although the amplitude is relatively small.

FIG. 14 is a diagram showing an example of the harmonic distribution of the motor current according to the conventional control when the carrier wave phases of the respective phases are shifted from each other by 120°. Control parameters other than the carrier phase are the same as in the example of FIG. FIG. 14 shows that the 1f component (800 Hz) of the carrier wave frequency has significantly increased. As can be understood from this result, when the carrier wave phases of the respective phases are shifted from each other by 120°, there arises a problem that the electromagnetic noise emitted from the motor increases.

In FIG. 12, it has been explained that the leakage current 35 flowing through the stray capacitance 34 is reduced when the carrier wave phases of the respective phases are shifted from each other by 120°. However, when the carrier wave phases of the respective phases are shifted from each other by 120°, if the value of the modulation factor is relatively large, the effect of reducing the leakage current 35 flowing through the stray capacitance 34 is reduced. Found by the Disclosers. This point will be described below.

FIG. 15 is a diagram showing an example of common-mode voltage waveforms according to conventional control when carrier wave phases of respective phases are shifted from each other by 120°. Control parameters other than the carrier wave phase in the waveform of FIG. 15 are a carrier wave frequency of 800 Hz, a drive frequency of 20 Hz, and a modulation factor of 0.8. According to FIG. 15, even when the carrier wave phase is shifted by 120°, the common mode voltage waveform changes to a level of ±1 [Vpu] as the modulation factor increases, and the leakage current 35 is effectively reduced. can be understood to be smaller.

FIG. 16 is a diagram used for explaining the concept of the control method in Embodiment 1. FIG. The horizontal axis represents the modulation rate, and the vertical axis represents the noise level. Since the modulation rate increases as the speed of the motor increases, the modulation rate on the horizontal axis may be read as the speed of the motor. The noise level on the vertical axis means the noise level caused by the leakage current 35 . A thick solid line K1 represents the noise level when there is a carrier wave phase difference, that is, when there is a phase difference between the carrier wave phases of the respective phases. A thick dashed line K2 represents the noise level when there is no carrier wave phase difference, that is, when there is no phase difference between the carrier wave phases of the respective phases. A thin dashed line K3 drawn parallel to the horizontal axis represents the limit value of the noise level. Note that the limit value is a value that varies depending on the vehicle type and route of the electric vehicle.

According to FIG. 16, when there is no carrier wave phase difference, the noise level caused by the leakage current 35 decreases as the modulation rate increases. On the other hand, when there is a carrier wave phase difference, it is shown that the noise level caused by the leakage current 35 increases as the modulation rate increases. Therefore, it can be said that it is desirable from the standpoint of noise to select the conventional modulation method without carrier wave phase difference within the range where the noise level limit value can be satisfied without carrier wave phase difference.

Based on the matters explained above, in the inverter control device 5 according to Embodiment 1, the carrier wave generating section 8 is configured as shown in FIG. FIG. 17 is a diagram showing a configuration example of the carrier generator 8 according to the first embodiment. The carrier generation unit 8 according to Embodiment 1 includes a phase calculation unit 81 and a carrier wave output unit 82, as shown in FIG.

Phase calculator 81 includes integrator 811 , phase difference calculator 812 , subtractor 813 , and adder 814 . The integrator 811 calculates the first carrier phase θ _cu based on the carrier frequency command _fc . A phase difference calculator 812 calculates a phase difference Δθ _c based on the motor frequency. A subtractor 813 calculates a second carrier phase θ _cv by subtracting the phase difference Δθ _c from the first carrier phase θ _cu . _The adder 814 calculates the third carrier phase _θcw by adding the phase difference _Δθc to the first carrier phase θcu. In addition, in FIG. 17, the input signal of the phase difference calculation unit 812 is the electric motor frequency, but it is not limited to this. Instead of the motor frequency, the modulation factor or the speed information of the electric car may be used. Further, the phase difference calculator 812 may be configured to output the phase difference Δθ _c according to a reference table, or may be configured to output the phase difference Δθ _c by functional calculation.

In the processing of FIG. 17, the phase difference _Δθc means the shift width of the carrier wave phase. As described with reference to FIG. 16, the carrier wave phase shift width between phases causes a noise problem in relation to the modulation factor. Therefore, it is desirable to minimize the shift width of the carrier wave phase. Also, a configuration is conceivable in which an operation mode is provided and the shift width of the carrier wave phase is controlled by switching the operation mode. However, there is a concern that a torque shock may occur in the configuration that switches the operation mode. Therefore, in Embodiment 1, the carrier wave phase shift width is continuously changed, and the carrier wave phase shift width is gradually decreased as the motor frequency increases.

The first carrier phase θ _cu , the second carrier phase θ _cv and the third carrier phase θ _cw calculated by the phase calculator 81 are input to the carrier wave output unit 82 . A carrier wave output unit 82 generates and outputs a U-phase carrier wave _cu using the first carrier wave phase θ _cu . A carrier wave output unit 82 generates and outputs a V-phase carrier wave _cv using the second carrier wave phase _θcv . A carrier wave output unit 82 generates and outputs a W-phase carrier wave _cw using the third carrier wave phase _θcw .

As described above, the phase calculator 81 calculates the first carrier phase θ _cu , the second carrier phase θ _cv and the third carrier phase θ _cw based on the carrier frequency command _fc and the motor frequency. Further, the carrier wave output unit 82 outputs three-phase carrier waves, U-phase carrier c _u and V-phase carrier c, based on the first carrier phase θ _cu , the second carrier phase θ _cv and the third carrier phase θ _cw. Output _{the v} and W phase carriers _cw . According to the carrier wave generator 8 configured in this manner, it is possible to achieve both common mode noise suppression in the low speed range and noise suppression in the medium to high speed range. Also, it is possible to realize control that does not cause a torque shock in mode transition.

Note that in FIG. 17, the phase difference between the first carrier phase θ _cu and the second carrier phase θ _cv and the phase difference between the first carrier phase θ _cu and the third carrier phase θ _cw are equal to each other, but their phase differences may be different. The essential point is that the first carrier phase θ _cu , the second carrier phase θ _cv and the third carrier phase θ _cw have a phase difference with each other, and this phase difference changes as the motor frequency changes. It is in. If this point is ensured, it is possible to obtain the effects of the first embodiment described here.

Further, the carrier generation section 8 shown in FIG. 17 may be configured as shown in FIG. FIG. 18 is a diagram showing a configuration example of the carrier generator 8A according to the modification of the first embodiment. In the carrier generation section 8A shown in FIG. 18, the phase calculation section 81 in the configuration of the carrier generation section 8 shown in FIG. 17 is replaced with a phase calculation section 81A. Integrators 815 and 816 are added to the configuration of phase calculator 81 shown in FIG. 17 in phase calculator 81A. The rest of the configuration is the same as the configuration of the phase calculator 81 shown in FIG. 17, and the same components are denoted by the same reference numerals, and overlapping descriptions are omitted.

In FIG. 18, the input carrier wave frequency command f _c is converted into the carrier wave frequency command of each phase, that is, the U-phase carrier wave frequency command f _cu , the V-phase carrier wave frequency command f _cv , and the W-phase carrier wave frequency command inside the phase calculation unit 81A. It is a configuration in which _fcw is branched into three and is input to an individual integrator provided for each phase. Since the same signal is input to each of the integrators 811, 815 and 816, an operation equivalent to that of FIG. 17 is performed.

As described above, the inverter control device according to Embodiment 1 includes a carrier wave generating section that generates a three-phase carrier wave, and the carrier wave generating section generates first to third carrier waves based on the carrier wave frequency command and the motor frequency. Calculates the carrier phase of The first through third carrier wave phases have a phase difference with each other, and this phase difference varies continuously as the motor frequency changes. This can reduce the common mode voltage caused by the PWM control signal. As a result, the leakage current that can flow through the stray capacitance can be reduced, so it is possible to suppress leakage noise without relying on additional filter components.

Preferably, the second and third carrier phases each have a phase difference of opposite sign and equal magnitude with respect to the first carrier phase. By doing so, it is possible to enhance the effect of reducing common mode noise.

Also, when the motor frequency is zero, the mutual phase difference between the first to third carrier phases is preferably 120°. By doing so, it is possible to further enhance the effect of reducing common mode noise.

Next, a hardware configuration for realizing the functions of the inverter control device 5 according to Embodiment 1 will be described with reference to FIGS. 19 and 20. FIG. FIG. 19 is a block diagram showing an example of a hardware configuration that implements the functions of inverter control device 5 according to the first embodiment. FIG. 20 is a block diagram showing another example of a hardware configuration that implements the functions of inverter control device 5 according to the first embodiment.

When realizing part or all of the functions of the inverter control device 5 in Embodiment 1, as shown in FIG. , and an interface 204 for inputting and outputting signals.

The processor 200 may be arithmetic means such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). The memory 202 includes non-volatile or volatile semiconductor memories such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable ROM), EEPROM (registered trademark) (Electrically EPROM), Examples include magnetic discs, flexible discs, optical discs, compact discs, mini discs, and DVDs (Digital Versatile Discs).

Memory 202 stores a program that executes the functions of inverter control device 5 in the first embodiment. Processor 200 performs the above-described processing by exchanging necessary information via interface 204, executing programs stored in memory 202, and referring to tables stored in memory 202. It can be carried out. Results of operations by processor 200 may be stored in memory 202 .

Further, when realizing part of the functions of the inverter control device 5 in Embodiment 1, the processing circuit 203 shown in FIG. 24 can also be used. The processing circuit 203 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. Information to be input to the processing circuit 203 and information to be output from the processing circuit 203 can be obtained via the interface 204 .

Part of the processing in the inverter control device 5 may be performed by the processing circuit 203 , and the processing not performed by the processing circuit 203 may be performed by the processor 200 and the memory 202 .

Embodiment 2.
FIG. 21 is a diagram showing a configuration example of the carrier generation section 8B according to the second embodiment. 21 is provided with a frequency modulation section 83 and an adder 84 in the preceding stage of the phase calculation section 81 in the configuration of the carrier generation section 8 shown in FIG. The frequency modulation section 83 is provided for noise reduction.

The frequency modulation section 83 has a random number generator 831 and an amplifier 832 . The adder 84 is configured to receive the basic carrier frequency command f _c0 and the output of the frequency modulation section 83 . The rest of the configuration is the same as the configuration of the phase calculator 81 shown in FIG. 17, and the same components are denoted by the same reference numerals, and overlapping descriptions are omitted.

In FIG. 21 , in the frequency modulation section 83 , a gain is given by the amplifier 832 to the output of the random number generator 831 . The output of amplifier 832 is input to adder 84 as frequency modulation amount _Δfc . The frequency modulation section 83 is a component that performs so-called random modulation, and calculates a frequency modulation amount _Δfc based on the output of the random number generator 831 . The adder 84 adds the carrier wave frequency command _fc and the frequency modulation amount _Δfc , and outputs the addition result to the phase calculator 81 as a new carrier wave frequency command _fc1 .

According to the carrier generator 8B shown in FIG. 21, the frequency modulation amount _Δfc is generated based on the random number generated by the random number generator 831. FIG. Then, the generated frequency modulation amount _Δfc is added to the carrier wave frequency command _fc common to each phase. Therefore, even if a phase difference is provided between each carrier, all carriers are frequency modulated based on random numbers generated from a unique seed. As a result, the carrier waves of each phase are frequency-modulated while maintaining the phase difference between the waveforms. Therefore, it is possible to suppress the deterioration of the common mode noise reduction effect.

As described above, according to the inverter control device according to the second embodiment, the carrier generation unit calculates the frequency modulation amount based on the output of the random number generator, and inputs the calculated frequency modulation amount to the phase calculation unit. is added to all carrier frequency commands. This makes it possible to reduce noise while suppressing deterioration of the common mode noise reduction effect.

Embodiment 3.
FIG. 22 is a diagram showing a configuration example of the carrier generation section 8C according to the third embodiment. In the carrier generation section 8C shown in FIG. 22, the configuration of the carrier generation section 8A shown in FIG. 18 is provided with a frequency modulation section 85 in front of the phase calculation section 81A, and the phase calculation section 81A is replaced with the phase calculation section 81B. It is In the phase calculation section 81B shown in FIG. 22, adders 817a, 817b and 817c are added before the integrators 811, 815 and 816 in the configuration of the phase calculation section 81A shown in FIG. The frequency modulation section 85 is provided to reduce noise more effectively than the frequency modulation section 83 of the second embodiment shown in FIG. The rest of the configuration is the same as the configuration of the carrier generation section 8A shown in FIG. 18, and the same reference numerals are given to the equivalent configuration sections to omit redundant description.

FIG. 23 is a diagram showing the detailed configuration of the frequency modulating section 85 shown in FIG. 22. As shown in FIG. The frequency modulation section 85 includes a list 851, an order control section 852, and switching sections 853a, 853b, and 853c. A list 851 stores frequency values of a plurality of data points. More specifically, the list ₈₅₁ contains n+ ₁ arbitrarily selected frequency values f _i ( _{f_0} , _{f_1 ,} . is stored. n+1 is the number of data points. n, f _L and f _H are arbitrary set values. In addition, it is desirable to be _fL = _fH . Moreover, it is desirable that the frequency values f _i at the n+1 points are equally spaced, that is, that when the frequency values f _i are arranged in ascending or descending order, all the differences between adjacent frequency values f _i are equal. Also, in FIG. 23 , n, f _L , and f _H are given from the outside of the frequency modulating section 85 , but they may be set inside the frequency modulating section 85 .

The order control unit 852 controls the switching units 853a, 853b, and 853c to randomly select one element from the elements stored in the list 851 without duplication, and selects the selected element as the first frequency modulation amount Δf _cu . , a second frequency modulation amount Δf _cv , and a third frequency modulation amount Δf _cw . Also, after all the elements in the list 851 have been selected, the elements in the list are selected in a random order different from the previous one, and this selection operation is repeated.

From the frequency modulation unit 85, elements selected one by one in a random first order without duplication from the elements stored in the list 851 are output as the first frequency modulation amount Δf _cu and sent to the adder 817a. is entered. Further, from the frequency modulation unit 85, the elements selected one by one in the second random order without duplication from the elements stored in the list 851 are output as the second frequency modulation amount Δf _cv , and the adder 817b. Furthermore, from the frequency modulation unit 85, elements selected one by one in a random third order without duplication from the elements stored in the list 851 are output as the third frequency modulation amount _Δfcw , and the adder 817c.

In FIG. 22, the adder 817a adds the carrier frequency command _fc and the first frequency modulation amount Δf _cu , and the addition result is input to the integrator 811 as the U-phase carrier frequency command f _cu . The adder 817b adds the carrier wave frequency command _fc and the second frequency modulation amount _Δfcv , and the addition result is input to the integrator 815 as the V-phase carrier wave frequency command _fcv . The adder 817c adds the carrier wave frequency command _fc and the third frequency modulation amount _Δfcw , and the addition result is input to the integrator 816 as the W-phase carrier wave frequency command _fcw . The subsequent operation is as described above, and the description here is omitted.

FIG. 24 is a diagram showing an output image of the frequency modulation amount _Δfc output from the frequency modulating section 85 shown in FIG. The frequency modulation amount Δf _c represents any one of the first frequency modulation amount Δf _cu , the second frequency modulation amount Δf _cv and the third frequency modulation amount Δf _cw . In the upper part of FIG. 24, the frequency modulation amount Δf _c output from the frequency modulation unit 85 is represented by the sequence {f _xi }={f _x0 , f _x1 , f _x2 , . . . , f _xn } (i=0, 1 , 2, . . . , n). Also, the lower portion of FIG. 24 shows the time at which each element of the sequence {f _xi } is output. The output time of each element may be rephrased as the time required for switching the output. According to FIG. 24, the output time of element f _xi is the reciprocal of element f _xi . That is, the time from the selection of the i-th element to the switching to the (i+1)-th element is the reciprocal of the i-th element. In the list 851, when the time from the output of the first element to the output of the last element is T _cyc , the time T _cyc can be expressed by the following equation (1).

The frequency modulation amount Δf _c is different for each phase, but only the order in which the elements are selected from the list 851 is different. Therefore, regardless of the order in which the elements stored in the list 851 are selected, the time T _cyc required to select all the elements in the list 851 is always the same. That is, the carrier wave frequency of each phase is modulated by a different random number, but the carrier wave phase difference returns to the set value every time T _cyc . Therefore, by using the frequency modulation section 85 according to the third embodiment, it is possible to achieve both reduction of common mode noise and improvement of audibility. Specifically, the smaller the number of data points n+1 and the narrower the range of frequency values stored in the list 851, the more the common mode noise reduction effect is prioritized. Conversely, the greater the number of data points n+1 and the wider the range of frequency values stored in the list 851, the more priority is given to the improvement of hearing.

As described above, according to the inverter control device according to Embodiment 3, the carrier generation unit includes the first frequency modulation amount Δf _cu , the second frequency modulation amount Δf _cv and the third frequency modulation amount Δf _cw and a frequency modulation unit that outputs The first frequency modulation amount Δf _cu is an element selected one by one in a random first order without duplication from each element stored in the list of the first phase. Also, the second frequency modulation amount Δf _cv is an element selected one by one in a random second order without duplication from each element stored in the list of the second phase. The third frequency modulation amount Δf _cw is an element selected one by one in a random third order without duplication from each element stored in the third phase list. These first frequency modulation amount Δf _cu , second frequency modulation amount Δf _cv and third frequency modulation amount Δf _cw are added one-to-one to the carrier wave frequency command. Also, when the first frequency modulation amount Δf _cu , the second frequency modulation amount Δf _cv and the third frequency modulation amount Δf _cw are selected from the list for each phase and output, the i-th element is selected The time from switching to the i+1th element after switching to the i+1th element is the reciprocal of the ith element. As a result, it is possible to achieve both reduction of common mode noise and improvement of audibility.

Note that the number of data points in the list for each phase may be changed according to the motor frequency. Further, when the minimum value of the elements stored in the list for each phase is set as the first value and the maximum value is set as the second value, at least one of these first and second values is the motor frequency may be changed according to By changing the number of data points, the first value, or the second value according to the motor frequency, it is possible to perform control according to the priority between the common mode noise reduction effect and the degree of improvement in hearing.

It is also desirable that the first and second values in each phase list have equal absolute values and opposite signs, and that each element in the list has equally spaced values. By setting in this way, it is possible to perform control according to the priority between reduction of common mode noise and improvement of audibility without changing the total amount of switching loss generated in the inverter 4 .

It should be noted that the configurations shown in the above embodiments are merely examples, and can be combined with another known technique, or can be combined with other embodiments, or deviate from the gist of the invention. It is also possible to omit or change part of the configuration as long as it is not necessary.

For example, although the inverter control device has been described as a device included inside the electric motor drive device, it is not limited to this. It is sufficient that the inverter control device and the inverter are electrically connected, and the inverter control device may be configured as a device other than the electric motor drive device.

1 electric motor driving device 2 filter reactor 3 filter capacitor 4 inverter 5 inverter control device 6 voltage command calculation unit 7 PWM control unit 8, 8A, 8B, 8C carrier wave generation unit 26U U-phase voltage command, 26V V-phase voltage command, 26W W-phase voltage command, 28 Carrier wave, 28U U-phase carrier wave, 28V V-phase carrier wave, 28W W-phase carrier wave, 31 Reference potential, 32 Neutral point potential, 33U U-phase voltage, 33V V-phase voltage, 33W W-phase voltage 34 stray capacitance 35 leakage current 81, 81A, 81B phase calculator 82 carrier wave output section 83, 85 frequency modulation section 84, 814, 817a, 817b, 817c adder 101 overhead wire 102 collection Electrical Device 103 Wheel 104 Rail 105 Electric Motor 200 Processor 202 Memory 203 Processing Circuit 204 Interface 811, 815, 816 Integrator 812 Phase Difference Calculator 813 Subtractor 831 Random Number Generator 832 Amplifier , 851 list, 852 sequence control section, 853a, 853b, 853c switching section, N negative side terminal, P positive side terminal, UNI, UPI, VNI, VPI, WNI, WPI semiconductor element.

Claims (10)

An inverter control device that controls an inverter that drives a three-phase electric motor by pulse width modulation,
a carrier wave generation unit that generates a three-phase carrier wave;
A pulse width modulation control unit that controls the switching state of the inverter by comparing the carrier wave and the modulated wave,
The carrier generation unit
a phase calculator that calculates first to third carrier wave phases based on the carrier wave frequency command and the motor frequency;
a carrier wave output unit that outputs the three-phase carrier wave based on the first to third carrier wave phases,
the first to third carrier wave phases have a phase difference with each other;
The phase difference continuously changes as the motor frequency changes , and the phase difference gradually decreases as the motor frequency increases.
An inverter control device characterized by: 2. The inverter control device according to claim 1, wherein the second and third carrier wave phases each have a phase difference of opposite sign and equal magnitude with respect to the first carrier wave phase. The inverter control device according to claim 2, wherein the phase difference is 120° when the motor frequency is zero. The carrier wave generator continuously changes the phase difference when the electric motor is driven.
The inverter control device according to any one of claims 1 to 3, characterized in that: The carrier generation unit includes a frequency modulation unit that calculates a frequency modulation amount based on the output of a random number generator, and adds the frequency modulation amount to all the carrier wave frequency commands that are input to the phase calculation unit. The inverter control device according to any one of claims 1 to 4 , characterized by: The carrier generation unit includes a frequency modulation unit that calculates first to third frequency modulation amounts,
The frequency modulation unit includes a list in which frequency values of a plurality of data points are stored,
Each element stored in the list is greater than or equal to a first value and less than or equal to a second value;
An element selected one by one in a random first order without duplication from each element stored in the list is output as the first frequency modulation amount,
An element selected one by one in a random second order without duplication from each element stored in the list is output as the second frequency modulation amount,
An element selected one by one in a random third order without duplication from each element stored in the list is output as the third frequency modulation amount,
When the number of data points is n+1 and i is an integer of 1 or more and n or less,
the time from the i-th element being selected to switching to the i+1-th element is the reciprocal of the i-th element,
The inverter control device according to any one of claims 1 to 4 , wherein the first to third frequency modulation amounts are added one-to-one to the carrier wave frequency command. The inverter control device according to claim 6 , wherein the number of data points in the list is changed according to the electric motor frequency. The inverter control device according to claim 6 or 7 , wherein at least one of the first value and the second value is changed according to the motor frequency. 9. The first value and the second value in the list are equal in absolute value and opposite in sign, and each element in the list has equally spaced values. 1. The inverter control device according to claim 1. an inverter control device according to any one of claims 1 to 9 ;
an inverter controlled by the inverter control device;
An electric motor drive with

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