JPH10323055A - Power converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To attain continuous and smooth control of an inverter output voltage, by computing the rising and falling and the timing of the rising and falling of pulses for the positive-side and negative-side switching elements of a power converter interchangeably. SOLUTION: A multiple pulse generating means 1 outputs a dipolar modulated waveform, or unipolar modulated waveform, or overmodulated waveform according to output voltage related information and shift control information. A one- pulse generating means 2 outputs (one pulse mode) a one-pulse waveform according to the output voltage related information. A shift controlling means 3 shifts each PWM mode continuously. Gate signals to be the outputs of the shift controlling means 3 are given to the switching elements in each-phase (U-phase, V- phase, W-phase) switching unit 7a, 7b, 7c through gate amplifiers, and on-off control is performed. Consequently, it becomes possible to adjust the output voltage of the inverter from zero to a maximum voltage continuously and smoothly.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an improvement in a power converter for converting direct current to alternating current or alternating current to direct current.
The present invention relates to control of an output voltage of a power converter.

[0002]

2. Description of the Related Art A three-level inverter divides a DC power supply voltage (an overhead line voltage) into two DC voltages by a capacitor connected in series, thereby creating three voltage levels of a high potential, an intermediate potential and a low potential. These three levels of voltages are selectively derived to the inverter output terminal by the on / off operation of the main circuit switching element, and have the following features.

That is, as the number of steps of the output voltage pulse increases, the apparent switching frequency is increased, and an output with less distortion can be obtained. Since the voltage applied to the element is reduced by about half compared to the two levels, a switching element having a relatively low withstand voltage can be used. For example, the loss generated around the element can be reduced as the voltage applied to the element decreases. Incidentally, there is the following method as a method of controlling the generation of the output voltage pulse of the three-level inverter.

(1) New Developments of
3rd Level Pied Brueem Strategies "New
Developments of 3-Level PWM Strategies "(EPE '
89 Record, 1989), page 412, FIG. 1 shows a modulation method called unipolar modulation (expressing the output voltage by alternately outputting pulses positive and negative through zero voltage within a half cycle of the output voltage), and unipolar modulation (output). A modulation method called an output voltage by outputting a pulse of a single polarity during a half cycle of the voltage, and a modulation method in which the dipolar modulation and the unipolar modulation are mixed in one cycle (hereinafter, in this specification, a partial (Referred to as dipolar modulation).

[0005] (2) PDB system in power converters: unextension
Of the subharmonic method "PWM Systems
in Power Converters: An Extension of the “Subharm
onic ”method” (IEEETrasaction on Industrial Elector
onics and Control Instrumentation, Vol.IECI-
28, No. 4, November 1981), page 316, FIG.
FIG. 3B shows a modulation method (hereinafter, referred to as a modulation method for expressing the output voltage by reducing the number of pulses so as to fill a slit between the pulses from the central portion, in which a half cycle of the output voltage is composed of a plurality of pulses of a single polarity. , Referred to herein as overmodulation).

(3) Study of 2 and 3
Level Precalculated Modulati "Study of 2 and 3-Level Precalculated Modulati
ons ”(EPE'91 Record, 1991)
FIG. 16 proposes an output voltage pulse generation control method for covering the output voltage from 0 to 100%.

[0007]

When a three-level inverter is used for an application such as a railway vehicle, the maximum voltage (output voltage) at which the voltage utilization rate reaches 100% from zero voltage in order to realize speed control over a wide range. Is a voltage region in which only a single pulse exists within a half cycle of
(Referred to as one pulse), it is required that the fundamental wave of the inverter output voltage can be continuously controlled and the harmonics of the inverter output voltage can be smoothly controlled.

The prior art (1) switches between dipolar modulation capable of controlling a minute voltage including zero, unipolar modulation means covering a medium speed region (medium voltage), and one pulse covering a maximum voltage. Therefore, the maximum voltage can be output from the zero voltage, and the continuity of the fundamental wave can be maintained.However, when switching between unipolar modulation and one pulse, the harmonics of the output voltage become discontinuous, and the frequency sharply increases. There was a problem that noise was generated due to the change.

[0009] Further, in the technique shown in the above-mentioned prior art (2), there is a problem that the maximum voltage cannot be expressed from the zero voltage.

In the prior art (1), in order to continuously control the fundamental wave of the output voltage, pulse data corresponding to the phase and voltage of the fundamental wave is stored in a memory, and based on the data, each modulation is performed. Since the pulse train is output, the control is complicated. Furthermore, the above-mentioned prior art (3) has a problem that in unipolar modulation, the modulation method switches the number of pulses existing in a half cycle of the fundamental wave, thereby complicating the control.

Further, the above-mentioned prior art has a problem that an unpleasant discontinuous sound is generated when the modulation method and the number of pulses are switched.

An object of the present invention is to realize a three-level pulse generation control capable of controlling the output voltage of a three-level inverter from zero to a maximum and continuously and smoothly controlling the output voltage of the inverter.

[0013]

SUMMARY OF THE INVENTION An object of the present invention is to provide a power converter for controlling a plurality of switching elements to convert a direct current into an alternating phase voltage having three levels of potentials. In a power conversion device comprising pulse generation means of a dipolar modulation mode for generating a positive / negative pulse train in a half cycle of the fundamental wave and a pulse train having a zero potential between the pulses in a phase of the power converter, the pulse generating means comprises: The first interrupt processing calculates the rising timing of the pulse of the positive switching element and the falling timing of the pulse of the negative switching element in the power converter in a first interrupt processing. In the interruption process of 2, the fall timing of the pulse of the positive switching element and the pulse of the negative switching element in the power converter are determined. It calculates the timing of raising is achieved than that which is adapted to perform these first and second arithmetic processing alternately.

According to the above-mentioned pulse generating means, there is no wrapping of the pulse pattern for raising the positive and negative switching elements in the same arithmetic processing, so that the output voltage can be continuously and smoothly controlled particularly at low speed. .

[0015]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, an outline of the present invention will be described with reference to Table 1 and FIGS. 1 to 3, and an embodiment will be described with reference to FIGS. 1 and 4 to 13. FIG.

A three-level inverter (also called an NPC inverter) is a DC power supply voltage (an overhead wire voltage in the case of an electric vehicle).
Is divided into two DC voltages by a capacitor connected in series to create three voltage levels of a high potential, an intermediate potential and a low potential. These three levels of voltages are turned on and off by a main circuit switching element. It is selectively derived to the inverter output terminal.

As an example of the main circuit configuration, FIG. 1 shows a basic configuration (in the case of three phases) when applied to a railway electric car.

In FIG. 1, reference numeral 4 denotes a DC overhead line (traffic line) which is a DC voltage source; 50, a DC reactor;
Is a clamp capacitor divided and arranged to generate an intermediate potential point O (hereinafter referred to as a neutral point) from the voltage of the DC voltage source 4. Reference numerals 7a, 7b and 7c each comprise a switching element capable of self-extinguishing, and a high-potential point voltage (P-point voltage), a neutral point voltage (O-point voltage) and a low-potential point according to a gate signal applied to the switching element. A switching unit that selectively outputs a voltage (N-point voltage). In this example, the switching unit 7a includes 70 to 73 self-extinguishing switching elements (here, IGBTs, but GTOs, transistors, or the like), and 74 to 77.
And the auxiliary rectifying elements 78 and 79. Also, the load is shown for the case of the induction motor 6. The switching units 7b and 7c have the same configuration as the switching unit 7a.

First, the basic operation of the U-phase switching unit 7a will be described with reference to Table 1 by way of example.

[0020]

[Table 1]

In the following, the clamp capacitor 51 will be described.
And the voltages vcp and vcn of 52 are perfectly smoothed DC voltages divided to Ed / 2, and the neutral point (point 0) is virtually grounded. Unless otherwise noted, the output voltage indicates the inverter output phase voltage.

The switching elements 70 to 73 constituting the switching unit 7a are turned on and off according to three types of conduction patterns as shown in Table 1. That is,
In the output mode P that outputs the P-point potential on the DC side, 70,
71 is on, 72 and 73 are off, and the output voltage is Ed / 2
In the output mode O for outputting the neutral point potential, 7
An output mode N in which 1, 72 are on, 70, 73 are off, a zero potential is output as an output voltage, and an N-point potential is output.
In this case, 70 and 71 are off, 72 and 73 are on, and the output voltage is -Ed / 2.

Table 1 shows an equivalent circuit of one phase of the main circuit (switching unit and clamp capacitor) in each output mode. The switching unit is equivalent to 3
It can be considered as a direction change switch. Here, switching functions Sp, S representing the conduction state of the element by two values of 1, 0
If n is used, it can be expressed as Sp = 1, Sn = 0 in the output mode P, Sp = 0, Sn = 0 in the output mode O, Sp = 0, Sn = 1 in the output mode N. At this time, the switching functions Sp, Sn
And the gate signals Gpu, Gpx, Gnx, Gnu (the off signal is 0 and the on signal is 1) given to the switching elements 70, 71, 72, 73 can be expressed by the following equation.

[0024]

(Equation 1)

Therefore, by preparing two switching functions Sp and Sn for each phase, the conduction state of the switching element can be determined. The switching functions Sp and Sn are determined by pulse width modulation (PWM) control.
The output voltage eu is determined to be sinusoidal.

The details of the main circuit of the three-level inverter are described in JP-A-51-47848 and JP-A-56-74088.

When a wide range of speed control from a variable voltage variable frequency (VVVF) region to a constant voltage variable frequency (CVVF) region is performed with a limited power supply voltage as in an electric car, a solid line in FIG. 2 is used. Such output voltage characteristics are required. That is, in the low-speed region, the output voltage is adjusted substantially in proportion to the inverter frequency (this region is referred to as a VVVF control region), whereby the magnetic flux in the motor is maintained substantially constant, a predetermined torque is secured, and In the high speed region, the inverter frequency is continuously increased while maintaining the maximum output voltage of the inverter (this region is referred to as a CVVF control region), thereby realizing high speed operation by maximizing the voltage utilization rate with a limited voltage. is there. However, in a conventionally known unipolar modulation method, in a region where the inverter frequency is low and a minute output voltage control is required (near the starting point of the VVVF control region), the minimum output pulse determined by the minimum on-time of the switching element. Voltage pulses smaller than the width cannot be realized,
As shown by the broken line in FIG. 2, a voltage higher than the command is output.

For example, assuming that all the voltage pulses of the inverter output voltage have the minimum pulse width determined by the minimum on-time Ton of the switching element, the output voltage effective value E at this time is

[0029]

E ≒ 2FcTonEmax Here, Fc: carrier frequency (Equation 2), and a voltage smaller than this cannot be controlled. Here, Emax is the effective value of the 180 ° conduction square wave voltage,

[0030]

(Equation 3)

The maximum output voltage of the three-level inverter also substantially matches this Emax. According to the above (Equation 2), when Fc = 500 kHz and Ton = 100 μs,
E = 0.1Emax, in which case the maximum output voltage Emax
Voltage of 10% or less cannot be controlled. Therefore, there is a problem that the lower limit of the controllable output voltage is limited only by the unipolar modulation, and it is difficult to continuously control the voltage.

In order to solve this problem, dipolar modulation (dipolar mode) is effective.
Care must be taken when shifting from dipolar modulation to unipolar modulation (unipolar mode).

On the other hand, the maximum voltage E that can be output by unipolar modulation is the limit point of ideal sine wave modulation (modulation rate A = 1).
so

[0034]

E = (π / 4) Emax ≒ 0.785Emax (Equation 4) In consideration of the minimum off time Toff of the switching element,

[0035]

E ≒ (π / 4) (1-FcToff) Emax where Fc: carrier frequency (Equation 5). For example, Fc = 500 Hz, Toff = 200
In μs, E = 0.707Emax, and in this case, it is possible to cover only about 70% of the maximum output voltage Emax. At this time, if the pulse width in the one-pulse mode cannot be adjusted, the fundamental wave becomes discontinuous, and
If the pulse width of the one-pulse mode is adjustable, the pulse width is reduced to maintain continuity.
Harmonic continuity is lost.

There are various modulation schemes that cover this voltage range. However, overmodulation (overmodulation) is considered from the viewpoint of easiness of pulse generation control, consistency with unipolar modulation, continuity of harmonics included in the output voltage, and the like. Mode) is the most effective. In the overmodulation area, the output voltage can be expanded to about one pulse by gradually filling the narrow slit between pulses in the center part (near the peak of the fundamental instantaneous value) of the voltage pulse train of a half cycle of the output voltage. And

In the limit of overmodulation control, that is, in a region where the modulation rate is extremely large, the operation shifts to a so-called one-pulse mode in which only one pulse exists in a half cycle of the output voltage, and the output voltage at this time almost reaches Emax. . However, as it is, the transition timing from overmodulation to one pulse or from one pulse to overmodulation depends on the modulation rate and the carrier frequency, so that this timing cannot be set arbitrarily. Voltage continuity is lost.

Therefore, from the overmodulation control, it is possible to control the voltage by the pulse width control which is not the extension of the overmodulation (that is, the one pulse mode which does not make the modulation rate infinite).
Shift to pulse control. This enables a transition at a predetermined timing between overmodulation and one-pulse control,
A continuous transition of the fundamental voltage is realized.

By continuously performing these series of transition controls, a stable and accurate output voltage is obtained from zero voltage to the maximum voltage continuously while selecting a pulse mode corresponding to the required output voltage. .

That is, as shown in FIG. 2, when the induction motor 6 is controlled with V / F = constant as shown, dipolar modulation is used from the start-up to F1, and when the inverter frequency reaches F1, unipolar modulation is performed. Move to the area, F2
To sequentially shift to an overmodulation area, and further to F1 to a one-pulse area.

FIG. 3 shows an example of a modulated wave capable of realizing the above idea based on a unified voltage command.

The basic modulated wave a proportional to the fundamental wave component of the output voltage is created from the following equation based on the inverter frequency command Fi * and the output voltage command E * from the higher-level current control means.

[0043]

Where: A: modulation rate, t: time, θ: phase (= 2πFi * t) (6) where modulation rate A (0 ≦ A ≦ 1) in the sine wave modulation area. )
Is given by the following equation.

[0044]

(Equation 7)

The basic modulation wave a is exactly the same for both dipolar modulation and unipolar modulation, and is the same for overmodulation except that the method of calculating the modulation factor A is different as described later.

In order to enable continuous transition between dipolar modulation and unipolar modulation, positive and negative bias modulation waves abp and abn represented by the following equations are provided here.

[0047]

(Equation 8)

In the dipolar modulation control, the above abp, abn
Become the positive-side modulated wave ap and the negative-side modulated wave an as they are.

[0049]

(Equation 9)

Here, the switching functions Sp, S
In order to simplify the creation of n, both ap and an are set to be positive. Finally, the pulse width of the output voltage is
It is set in proportion to the magnitudes of ap and an. In the case of dipolar modulation, the positive and negative pulses are controlled by being shifted by approximately 180 °.

In the unipolar modulation, the positive and negative modulated waves ap, a
n is

[0052]

(Equation 10)

[0053]

[Equation 11]

Is given by

If the minimum off time of the switching element is so small that it can be ignored, the maximum pulse is output when the instantaneous values of ap and an are 1 or more (overmodulation described later).

Here, it is understood that the setting of the bias B is extremely important in the transition control. Transition control between the dipolar modulation region and the unipolar modulation region is realized by the value of B. (a) When A / 2 ≦ B <0.5 Dipolar modulation (b) When B = 0, unipolar modulation is performed.

On the other hand, in the overmodulation control, the modulation rate A is increased to 1 or more, the slit between the pulses in the center part of the half cycle of the output voltage (zero voltage output period) is suppressed, and the output voltage is improved.

When the voltage command is further increased, the mode shifts from the overmodulation mode to the one-pulse mode. This operation will be described in the following embodiment.

Thus, dipolar modulation, unipolar modulation and overmodulation are realized based on a unified voltage command.
Continuous transition control up to one pulse which is the maximum output is possible. Hereinafter, a configuration of an embodiment that realizes the above concept will be described.

FIG. 1 shows an example of a pulse width modulator for controlling the above-mentioned switching unit and outputting an AC voltage having three levels of potential.

In FIG. 1, reference numeral 1 denotes a multi-pulse generating means for outputting a dipolar modulation waveform, a unipolar modulation waveform, or an overmodulation waveform in accordance with output voltage-related information and transition control information, and 2 denotes a one-pulse waveform in accordance with output voltage-related information. 1-pulse generating means for output (1-pulse mode), 3
Is a transition control means for continuously transiting each PWM mode. A gate signal, which is an output of the transition control means 3, is supplied to a switching element in a switching unit of each phase via a gate amplifier (not shown) to be turned on / off. The pulse width modulating means composed of the multi-pulse generating means 1, the one-pulse generating means 2, and the transition control means 3 is a feature of the present invention.

In this example, the output voltage-related information taken into the pulse width modulation means is based on the higher-order current control means 8.
Given by The current control means 8 creates a slip frequency command Fs * of the induction motor 6 from the current command by the current adjusting means 81 (based on the deviation between the current command value and the actual motor current), and sets the rotation frequency attached to the induction motor 6. The rotation frequency Fr of the induction motor detected by the detection means 61 and the above-mentioned Fs * are added to the inverter frequency command F.
Create i *.

Further, based on this Fi * and the DC voltage Ed of the three-level inverter (the voltage between PNs, which is equal to the sum of the clamp capacitor voltages vcp + vcn), the output voltage setting means 82 creates an output voltage command E *.

The output voltage setting means 82 sets the slope to be large when Ed is low (Ed = Ed1), and to be small when Ed is high (Ed = Ed3). Thus, the output voltage characteristics shown in FIG. 2 are realized. These current control means may output an instantaneous value of the output voltage.

The configuration and operation of the pulse width modulation means will be described in detail with reference to FIGS.

FIG. 4 shows an example of the overall configuration of the pulse width modulation means. Here, the multi-pulse generating means 1 includes a basic modulated wave generating means 11, a bias superimposing means 12, a positive / negative distribution means 13, a reference signal generating means 14, and a pulse generating means 15.

The fundamental modulation wave generating means 11 obtains the phase θ by time-integrating the inverter frequency command Fi * received as the output voltage related information by the phase calculating means 112, and obtains the sine value sin θ at this θ. On the other hand, from the voltage command E *, which is one of the output voltage related information, the amplitude A of the basic modulated wave (modulation rate) by the amplitude setting means 111.
Is calculated, and after being halved, multiplied by sin θ to create and output an instantaneous basic modulated wave a / 2 having an amplitude of ２. The bias superimposing means 12 adds and subtracts the bias B from the multi-pulse transition control means 31 of the transition control means 3 to this a / 2, and performs two positive and negative bias modulation waves abp and abn.
And output.

Here, the continuous transition between the dipolar modulation and the unipolar modulation depends on the setting of the bias B. FIG.
An example of the configuration of the dipolar / unipolar transition control means 311 performed by setting the bias B is shown in FIG. The dipolar / unipolar transition control means 311 multiplies the output voltage command E * by 4 / π by 311a to obtain the modulation rate A.
, And the bias generation means 311b determines the bias B according to the modulation rate A. That is, B = Bo (where Bo ≧ A / 2) when the modulation factor A is small and a minute output voltage is required, and B = 0 when A = A1. If A1 is determined in advance so that the output voltage at the time of A = A1 is higher than the voltage shown in the equation (Equation 2), voltage control from a minute voltage including zero can be performed.

Further, the positive and negative bias modulation waves abp, abp
The positive and negative distribution means 13 distributes and combines the positive part of abp and abn into ap and the negative part of abp and abn into an, so that the output voltage fundamental wave from dipolar modulation to unipolar modulation is obtained. Positive / negative modulated waves ap and an that maintain the continuity of the components are created.

On the basis of the positive and negative modulated waves ap and an, the pulse generator 15 generates switching functions Sp and Sn whose pulse generation period is 2To. Reference signal generating means 1
4 determines the pulse generation period To according to the switching frequency command Fsw *. Here, the relationship between Fsw * and To can be expressed by the following equation.

[0071]

(Equation 12) To = 1 / (2Fsw *) (Equation 12) The pulse generation operation of the pulse generation means 15 will be described with reference to FIG.

In FIG. 6, the pulse timing setting means 151 calculates the rising timing Tpup of Sp and the falling timing Tndn of Sn based on ap, an, aoff, and To (an and aoff will be described later) according to the following equation. (Process 1).

[0073]

(Equation 13)

[0074]

[Equation 14]

In the next cycle, the falling timing Tpdn of Sp and the rising timing Tnup of Sn are determined in the same manner as in the processing 1 (processing 2).

[0076]

(Equation 15)

[0077]

(Equation 16)

The switching functions Sp and Sn are created by alternately performing the processing 1 and the processing 2 described above.

Here, aon and aoff are values determined from the minimum on-time Ton and the minimum off-time Toff of the switching element.

[0080]

[Equation 17]

Is given by That is, as shown in FIG. 7 (example of Sp), the ON pulse width Twon and the OFF pulse width Twoff are

[0082]

(Equation 18)

And has the characteristic indicated by the broken line in FIG. Here, the solid line in FIG. 8 is used so that the on-pulse width Twon does not become shorter than the minimum on-time Ton determined by the switching element, and the off-pulse width Twoff does not become shorter than the minimum off-time Toff determined by the switching element. The characteristics shown below. To achieve this, FIG.
The function of the pulse timing setting means 151 is added.
Since the discontinuity of the output voltage fundamental wave component generated thereby is extremely small, it can be ignored.

Note that aoff can be varied within a range where the discontinuity of the output voltage fundamental wave component can be neglected, and is given from the unipolar / overmodulation transition control means 312 as the transition timing from unipolar modulation to overmodulation. . If aoff is set to be constant, pulse generation can be further simplified.

That is, the pulse timing setting means 15
1 automatically shifts from unipolar to overmodulation.
Unipolar / overmodulation transition control means 312 that outputs off
There is no need to provide

The switching function generating means 152 generates a reference signal having a period To, and synchronizes with the above-mentioned Tpup,
Sp and Sn are set based on Tndn or Tpdn and Tnup. FIG. 9 shows an example of a switching function at the time of overmodulation. When the instantaneous value Ap of ap exceeds aoff, the slit between the pulses of the switching function Sp (the hatched portion in FIG. 9C) is filled. This filled slit width is smaller than the minimum off time Toff of the switching element,
Since it gradually disappears by about one or two pieces, it hardly affects the fundamental wave of the output voltage.

FIG. 10 shows a flowchart when the pulse timing setting means 151 is realized by software.

By the way, in the overmodulation control, the maximum voltage state is maintained by filling the slit between the pulses in the central part of the half cycle of the output voltage, and only in the vicinity of the zero cross of the modulated wave.
Performs PWM control. Therefore, in this region, the modulation factor A and the actually output voltage become non-linear, and even if the modulation factor A is linearly increased, the output voltage does not linearly increase in accordance with this.

Therefore, the output voltage at the time of overmodulation is linearized by making the setting of the modulation factor A non-linear. That is, if the switching frequency in the PWM control part is sufficiently high, the fundamental wave effective value E of the output voltage and the modulation factor A
Can be expressed by the following equation.

[0090]

[Equation 19]

Therefore, the relationship between E * and A is calculated in advance from the relationship of the above equation, and the amplitude setting means 1 shown in FIG.
By configuring 11, the output voltage can be linearly adjusted with respect to E *. As a result, it is possible to improve the voltage controllability particularly in a high voltage region close to one pulse.

When the voltage command is further increased, the mode is switched from the overmodulation mode to the one-pulse mode by the operation of the changeover switch 32 of the transition control means 3. Changeover switch 3
2 is one of the outputs of the multi-pulse shift control means 31
_{When PM} is _SPM = 0, multi-pulse side When _SPM = 1, it is switched to one pulse side. FIG. 12 shows an example of the one-pulse / multi-pulse switching control means 313. In this example, the voltage command E
A hysteresis is provided so that when * exceeds E1P, the mode shifts from the multi-pulse mode to the one-pulse mode, and when E * becomes smaller than EMP, the mode shifts from the one-pulse mode to the multi-pulse mode. As a result, careless transition of the PWM mode is suppressed, and a stable output voltage with less transient fluctuation is obtained.

The one-pulse generation means 2 includes a phase calculation means 2
1 and pulse generating means 22. The operation of the phase calculation means 21 may be exactly the same as 111, and the output of 111 may be used by omitting 21.

FIG. 11 shows an example of the configuration of the pulse generating means 22. In the three-level PWM, unlike the two-level PWM, the output voltage can be adjusted by controlling the pulse width during one-pulse control. Therefore, from the voltage command E *, the rising timing phase α and the falling timing phase β of the pulse are

[0095]

(Equation 20)

Is obtained by By setting α and β with reference to phase θ, and creating and outputting Sp and Sn,
A one-pulse waveform is realized.

As described above, dipolar modulation, unipolar modulation, and overmodulation are realized based on a unified voltage command.
Continuous transition control up to one pulse which is the maximum output is possible. In this embodiment, the output voltage can be continuously and smoothly adjusted from the zero voltage to the maximum voltage, and further, there is an effect that a stable output voltage with high accuracy can be provided.

In the first embodiment shown in FIG. 4, the output pulse train of the multi-pulse generator is generated asynchronously with the inverter frequency, and the output pulse of the one-pulse generator is controlled in synchronization with the inverter frequency. ing.

The reason for this is that, in the above-mentioned prior art which employs the synchronous method in the multi-pulse region, firstly, control for managing the phase is complicated, and secondly, the output voltage command is sine based on a request for some control. When it is necessary to distort the wave (in FIG. 1, the inverter frequency Fi * and the output voltage command E
* Is adjusted according to electric vehicle control requirements)
There is a problem that the output voltage command cannot be faithfully reproduced.

That is, the first problem is that the synchronous type has a table having a relationship between a phase and a generated pulse for each pulse mode in order to output a pulse of an integral multiple of the inverter frequency. The pulse generation phase is read from the obtained phase and output. The amount of calculation required for managing the phase and the memory for each pulse mode are enormous, which complicates the control.

The second problem is that the synchronous type shown in the prior art has pulse data of 90 °, but since the data is created so that the output voltage becomes a sine wave, the output voltage is reduced. There is a problem that it cannot be expressed exactly as instructed.

Therefore, in the present embodiment, these problems are solved by making the generation of pulses in the multi-pulse mode asynchronous with the inverter frequency.

That is, with respect to the first problem, a pulse can be generated independently without being restricted by the inverter frequency for generating the pulse. That is, in FIG. 4, the switching frequency command Fsw * can be set independently of the inverter frequency command Fi * (FIG. 4,
Reference generation 14 is independent of inverter frequency). Therefore, the control can be simplified without requiring a complicated control procedure for pulse generation.

As for the second problem, the asynchronous system eliminates the necessity of having data for each phase, and can output a pulse corresponding to an instantaneous voltage command. Even a distorted sine wave can be faithfully represented. Further, as described above, since the control regarding the phase calculation and the like is simplified, the calculation for outputting the pulse corresponding to the sequential voltage command can be performed, and the calculation cycle can be shortened. You can increase the degree.

Further, in the case of the asynchronous type, since the switching frequency does not depend on the inverter frequency, the change in the switching frequency can be minimized. , Unpleasant sound) can be minimized.

Although the above embodiment has been described by taking a three-level inverter as an example, the same applies to a two-level inverter and a multi-level inverter of three or more levels.

By the way, in the case of a switching element such as a GTO thyristor which performs switching at a relatively low frequency, of the output voltage harmonics, a sideband component generated depending on the switching frequency and a fundamental component of the inverter frequency are generated. Interference may occur. In order to avoid this, among the PWM modes of the multi-pulse generating means, the dipolar modulation mode and the unipolar modulation mode are asynchronous with respect to the inverter frequency, and the overmodulation mode and the one-pulse mode are synchronized (FIG. 14).

With such a configuration, a more stable voltage can be supplied even during overmodulation.

FIG. 15 shows another embodiment of the multi-pulse shift control. FIG. 15 shows only the multi-pulse shift control means 31. This shifts the four PWM modes depending on both the inverter frequency command Fi * and the voltage command E *. That is, dipolar modulation when Fi * <F1 and E * <E1, unipolar modulation when Fi * ≧ F1 and E1 ≦ E * <E2, overmodulation when E2 ≦ E * <E3,
When E * ≧ E3, one pulse is set. As a result, the transition condition of dipolar modulation → unipolar modulation → overmodulation → one pulse is satisfied even in the case where the output voltage is soft-started in a high-frequency region where the frequency is high, for example, at the time of regenerative start or re-powering, and stable. Voltage rise is possible. In addition, since dipolar modulation control is always performed in the low frequency region, current concentration on a specific switching element as in the case of unipolar modulation can be avoided.

Next, a second embodiment will be described.

If the first embodiment is extended to introduce partial dipolar modulation in which both modulation waveforms coexist between dipolar modulation and unipolar modulation as shown in FIG. 16, the output voltage and the switching frequency can be further increased. Can be increased in smoothness.

An example of the output voltage command waveform is shown in FIG.
17 is exactly the same as FIG. 3 except for (b).
Hereinafter, the partial dipolar will be described.

Due to the effects of bias superimposition and positive / negative distribution, even if the bias B is set in a range that is neither dipolar modulation nor unipolar modulation (0 <B <A / 2), the voltage required for the fundamental modulated wave is excessively or insufficiently set. It is possible to reproduce without. In this case, the vicinity of the peak of the output voltage is unipolar modulation, and the base of the output voltage is partial dipolar modulation, which is dipolar modulation. At this time, the positive modulation wave ap and the negative modulation wave an

[0114]

(Equation 21)

[0115]

(Equation 22)

Is obtained. It can be seen that (ap-an) always coincides with the fundamental modulation wave a, and the continuity of the instantaneous value of the output voltage fundamental wave is also maintained.

If the bias B is gradually reduced in accordance with the increase of the modulation factor A by utilizing the above-mentioned properties, the transition from the dipolar modulation to the unipolar modulation can be continuously performed via the partial dipolar modulation. Of course, the reverse is also possible.

FIG. 18 shows an example of the dipolar / unipolar transition control means. If the bias B is set as shown by the solid line in FIG. 18, dipolar modulation is performed in the region of 0 ≦ A ≦ A1, partial dipolar modulation is performed in the region of A1 <A <A2,
Unipolar modulation is performed in the region of A ≧ A2. in this case,
Since no abnormal noise is generated from the electric motor when switching between dipolar modulation and unipolar modulation, it is effective in reducing the noise of the device.

When FIG. 18 is applied, the PWM mode can be managed for each area as shown in FIG. FIG. 19 shows only the multi-pulse shift control means 31. This is five kinds of PW
In the M mode, the inverter frequency command Fi * and the voltage command E *
The transition depends on both. That is, F
Dipolar modulation when i * <Fo and E * <Eo, Fo
Partial dipolar modulation when ≦ Fi * <F1 and Eo ≦ E * <E1, unipolar modulation when Fi * ≧ F1 and E1 ≦ E * <E2, overmodulation when E2 ≦ E * <E3, E * ≧
In the case of E3, one pulse is used. Thus, even when the output voltage is soft-started in a high-speed region where the frequency is high, for example, at the time of regenerative start or re-powering, dipolar modulation → partial dipolar modulation → unipolar modulation → overmodulation → 1
The transition condition of a pulse is satisfied, and a stable voltage rise becomes possible. In addition, the same effect as at the time of regenerative activation can be obtained at the time of idling re-adhesion. Further, in any operating state, there is an effect that generation of abnormal noise from the electric motor at the time of pulse mode switching can be minimized.

By the way, in the inverter used in the electric vehicle control device for railway vehicles, the variable range of the inverter frequency Fi * is about 0 to 300 Hz. The inverter frequency Fcv at which the output voltage becomes maximum is 1/5 to 1/3 of the inverter frequency variable upper limit, and the upper limit of Fcv is about 100 Hz. To avoid fluctuations in output current due to interference between the harmonics generated around the switching frequency and the fundamental wave of the inverter frequency when generating pulses asynchronously, use Fcv
A switching frequency that is about 10 times that of the above, that is, a switching frequency of 1 kHz or more is required.

Further, in order to reduce noise (abnormal noise and the like described above),
It is effective to minimize the fluctuation of the switching frequency, and the fluctuation of the switching frequency in the multi-pulse region can be made within 1 to 2Fi by introducing overmodulation.

Of course, if a microprocessor or the like is used, a part or all of the pulse width modulation means can be programmed and realized in software. FIG. 21 shows an example of a flowchart for realizing the processing of the rise and fall timing of the pulse in the pulse width modulation means of FIG. 4 by software. Although the above description has been made with reference to the case of an induction motor load as an example, the present invention is not limited to this, and similar effects can be expected with other AC motors. Although all of the above descriptions have been directed to inverters, the output terminals of these inverters can be connected to an AC power supply via a reactance element to operate as a self-excited converter that converts AC to DC. is there. In this case, the same effect as that of the inverter can be expected.

Although the above description has been made of the case of a three-level inverter, the concept of the present invention can be applied to a multi-level inverter of three or more levels.

[0124]

According to the present invention, it is possible to continuously and smoothly adjust the inverter output voltage from zero voltage to the maximum voltage. Further, there is an effect that the pulse generation control system is simplified.

Further, when applied to an electric vehicle, an electric vehicle with low noise can be provided.

[Brief description of the drawings]

FIG. 1 is a configuration diagram showing one embodiment of the present invention.

FIG. 2 is a diagram illustrating a relationship between output voltage characteristics and a PWM mode.

FIG. 3 is an explanatory diagram of a modulated wave for continuous transition to a PWM mode in a multi-pulse region.

FIG. 4 is a detailed explanatory diagram of the configuration in FIG. 1;

FIG. 5 is a diagram showing an example of dipolar / unipolar transition control means.

FIG. 6 is a diagram showing an example of a pulse generating means in the multi-pulse generating means.

FIG. 7 is a waveform diagram showing a relationship between on / off pulse widths.

FIG. 8 is a view showing characteristics of an on / off pulse width.

FIG. 9 is a diagram showing an example of an overmodulation waveform.

FIG. 10 is a view showing a flowchart of pulse timing setting means by software.

FIG. 11 is a diagram illustrating a configuration example of an amplitude setting unit.

FIG. 12 is a diagram showing an example of a multi-pulse / 1-pulse switching control means.

FIG. 13 is a diagram showing an example of one-pulse generation means.

FIG. 14 is a diagram showing a configuration example of another embodiment.

FIG. 15 is a diagram illustrating an example of a transition control unit.

FIG. 16 is a relationship diagram between an output voltage characteristic and a PWM mode when another PWM mode is included.

FIG. 17 is a diagram for explaining another PWM mode modulated wave.

FIG. 18 is a configuration diagram of a transition control unit that realizes another PWM mode.

FIG. 19 is a diagram illustrating an example of a transition control unit.

FIG. 20 is a diagram illustrating a relationship between an inverter frequency and a switching frequency.

FIG. 21 is a view showing a flowchart of pulse width modulation means by software.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... Multi-pulse generation means, 2 ... 1 pulse generation means, 3 ... Transition control means, 4 ... DC overhead wire, 6 ... Induction motor, 7a, 7
b, 7c: switching unit, 8: current control means,
11: basic modulated wave generating means, 12: bias superimposing means,
13: positive / negative distribution means, 14: reference signal generation means, 15 ...
Pulse generating means, 21: phase calculating means, 22: pulse generating means, 31: multi-pulse shift control means, 32: changeover switch, 50: DC reactor, 51, 52: clamp capacitor, 61: rotational frequency detecting means.

Continued on the front page (72) Inventor Mutsumi Terunuma 4026 Kuji-cho, Hitachi City, Ibaraki Prefecture Inside Hitachi Research Laboratory, Hitachi, Ltd. (72) Inventor Yuto Suzuki 4026 Kuji-machi, Hitachi City, Ibaraki Prefecture Hitachi Research, Hitachi, Ltd. (72) Inventor Yoshio Tsutsui 4026 Kuji-cho, Hitachi City, Ibaraki Prefecture Inside Hitachi Research Laboratory, Hitachi, Ltd. (72) Inventor Eiichi Toyoda 1070 Mo, Hitachinaka City, Ibaraki Prefecture Mito Plant, Hitachi, Ltd.

Claims (1)

[Claims] 1. A power converter for controlling a plurality of switching elements to convert a direct current to an alternating phase voltage having three levels of potentials, wherein the power converter outputs positive and negative half-cycles of a fundamental wave of an output phase voltage. In a power conversion device comprising pulse generation means of a dipolar modulation mode for generating a pulse train having a zero potential between the pulses in the phase of the power converter, the pulse generation means is performed by processing of a microcomputer, In a first interrupt processing, a rising timing of a pulse of a positive switching element and a falling timing of a pulse of a negative switching element in the power converter are calculated. Calculate the falling timing of the pulse of the positive switching element and the rising timing of the pulse of the negative switching element in the converter, A power converter characterized in that the first and second calculation processes are performed alternately.