JP3677552B2 - Electric vehicle power converter - Google Patents

Description

The present invention relates to an improvement of a power converter that converts direct current to alternating current or alternating current to direct current,
In particular, the present invention relates to control of the output voltage of a power converter for an electric vehicle .

The three-level inverter generates three voltage levels of high potential, intermediate potential, and low potential by dividing the DC power supply voltage (overhead voltage) into two DC voltages with a series-connected capacitor. These three levels of voltages are selectively derived to the inverter output terminal by the on / off operation, and have the following characteristics.
That is, by increasing the number of steps of the output voltage pulse, the apparent switching frequency is increased and an output with less distortion can be obtained. Since the voltage applied to the element is halved compared to the two levels, a switching element having a relatively low breakdown voltage can be used. The loss generated around the element can be reduced as the applied voltage of the element decreases.
By the way, as a method for controlling generation of output voltage pulses of the three-level inverter, there are the following methods.
(1) New Developments of Three Levels, P4B Strategies “New Developments of 3-Level PWM Strategies” (EPE '89 Record, 1989), page 412, FIG. 1 (Non-Patent Document 1) shows dipolar modulation (output) Modulation method called unipolar modulation (by outputting a pulse of a single polarity during the half cycle of the output voltage), which expresses the output voltage by alternately outputting positive and negative pulses via a zero voltage within a half cycle of the voltage A modulation method called “representing output voltage” and a modulation method in which the dipolar modulation and the unipolar modulation are mixed in one cycle (hereinafter referred to as partial dipolar modulation) have been proposed.
(2) Pdw system in power converters: Unextension of the subharmonic method “PWM Systems in Power Converters: An Extension of the“ Subharmonic ”method of Instrumental Industr. -28, No. 4, November 1981), page 316, Fig. 2 (b) (non-patent document 2), the half cycle of the output voltage is composed of a plurality of single-polarity pulses. A modulation method that expresses the output voltage by reducing the number of pulses so as to fill the slits between them (hereinafter referred to as excessive Tone referred to as) has been proposed.
(3) Study of 2 and 3 Level Precalculated Modulations “Study of 2 and 3-Level Precalculated Modulations” (EPE '91 Record, 1991), page 411, FIG. 16 (Non-patent Document 3) An output voltage pulse generation control method for covering the output voltage from 0 to 100% has been proposed.

"New Developments of 3-Level PWM Strategies" (EPE'89 Record, 1989), page 412, FIG. “PWM Systems in Power Converters: An Extension of the“ Subharmonic ”method” (IEEE Transactions and Industrial Instruments, Vol. 3, Vol. 3, No. 3, Vol. Page 411 of "Study of 2 and 3-Level Precalculated Modulations" (EPE'91 Record, 1991), FIG.

By the way, in the case of using a three-level inverter for an application such as a railway vehicle, in order to realize speed control over a wide range, the maximum voltage from which zero voltage reaches 100% (a single output voltage within a half cycle of the output voltage). It is required that the fundamental wave of the inverter output voltage can be continuously controlled and the harmonics of the inverter output voltage can be controlled smoothly until the voltage range where only the pulse exists, hereinafter referred to as one pulse.
By the way, in the above prior art (1), dipolar modulation that can control a minute voltage including zero, unipolar modulation means that covers a medium speed region (medium voltage), and 1 pulse that covers the maximum voltage are switched. The maximum voltage can be output from zero voltage, and the continuity of the fundamental wave can be maintained, but the harmonics of the output voltage become discontinuous when switching between unipolar modulation and one pulse, and noise due to sudden and large changes in frequency There was a problem that occurred.
Further, the technique shown in the above prior art (2) has a problem that the maximum voltage cannot be expressed from zero voltage.
In the prior art (1), in order to continuously control the fundamental wave of the output voltage, pulse data corresponding to the phase and voltage of the fundamental wave is stored in a memory, and a pulse train corresponding to each modulation is generated based on this data. Since it is an output, the control is complicated. Furthermore, since the prior art (3) is a modulation method that switches the number of pulses existing in a half cycle of the fundamental wave in unipolar modulation, there is a problem in that control is complicated.
Furthermore, the above-described prior art has a problem that an unpleasant discontinuous sound is generated when the modulation method and the number of pulses are switched.
The object of the present invention is to control the output voltage of the three-level inverter from zero to the maximum, realize three-level pulse generation control that can continuously and smoothly control the inverter output voltage, and mount the three-level inverter on an electric vehicle. It is to prevent the discontinuous sound that occurs.

In order to solve the above problems, a power converter that converts a direct current into an AC phase voltage having a three-level potential that is a positive, negative, and intermediate voltage by switching control of a plurality of switching elements, and a switching element of the power converter controls, operating range of the power conversion device for an electric vehicle and a control device for acceleration and deceleration control the connected AC motor that drive the electric vehicle on the output side of the power converter, at a low speed of an electric vehicle In the power converter
Within a half cycle of the output voltage fundamental wave, multiple voltage pulses with the same polarity are output continuously.
A positive voltage pulse and a negative voltage pulse before and after the first period via an intermediate voltage.
First pulse width modulation control means having a second period of alternating output, and an electric vehicle
In the high-speed operation region than in the low-speed operation region, the fundamental wave of the output phase voltage of the power converter
Second pulse width for generating a plurality of single polarity pulse trains in the phase of the power converter in a half cycle
Shifts modulation control means and first and second pulse width modulation control means according to the operation region
Means .

According to the present invention, the inverter output voltage can be adjusted continuously and smoothly. In the variable speed control of an AC motor using a three-level inverter, in particular acceleration / deceleration
Inverter output voltage continuity over time can be obtained, preventing discontinuous noise and abnormal noise
An electric vehicle that can be stopped and that is low in noise can be provided.

Hereinafter, the outline of the present invention will be described with reference to Table 1 and FIGS. 1 to 3, and then Example 1 will be described with reference to FIGS. 1 and 4 to 13, Example 2 will be described with reference to FIGS. 14 and 15 to 19. It explains using.
A three-level inverter (also referred to as an NPC inverter) divides a DC power supply voltage (in the case of an electric vehicle, an overhead wire voltage) into two DC voltages by a capacitor connected in series, thereby generating a high potential, an intermediate potential, and a low potential 3 One voltage level is created, and these three levels of voltages are selectively derived to the inverter output terminal by the on / off operation of the main circuit switching element.

As an example of the main circuit configuration, FIG. 1 shows a basic configuration (in the case of three phases) when applied to a railway electric vehicle.
In FIG. 1, 4 is a DC overhead line (train line) which is a DC voltage source, 50 is a DC reactor, 51 and 52 are for generating an intermediate potential point O (hereinafter referred to as a neutral point) from the voltage of the DC voltage source 4. This is a clamp capacitor that is divided. 7a, 7b and 7c are constituted by switching elements capable of self-extinguishing, and a high potential point voltage (P point voltage), a neutral point voltage (O point voltage) and a low potential point according to a gate signal applied to the switching element. This is a switching unit that selectively outputs a voltage (N-point voltage). In this example, the switching unit 7a has 70 to 73 self-extinguishing switching elements (in this case, IGBTs, but may be GTOs, transistors, etc.), 74 to 77 reflux rectifiers, and 78 and 79 auxiliary rectifiers. It consists of elements. Further, the load is shown for the induction motor 6. The switching units 7b and 7c have the same configuration as 7a.

  Here, first, the basic operation of the U-phase switching unit 7a will be described with reference to Table 1.

なお、以下では、クランプコンデンサ５１および５２の電圧ｖcp，ｖcnは完全平滑でＥｄ／２に分圧された直流電圧とし、中性点（０点）は仮想的に接地されているものとする。また、ことわりのない限り、出力電圧はインバータ出力相電圧を指すものとする。
スイッチングユニット７ａを構成するスイッチング素子７０から７３は、表１に示すように３通りの導通パターンに従いオン・オフ動作する。すなわち、直流側のＰ点電位を出力する出力モードＰでは、７０，７１がオン，７２，７３がオフで、出力電圧はＥｄ／２となり、中性点電位を出力する出力モードＯでは、７１，７２がオン，７０，７３がオフで、出力電圧としてゼロ電位が出力され、Ｎ点電位を出力する出力モードＮでは、７０，７１がオフ，７２，７３がオンで、出力電圧は−Ｅｄ／２となる。
表１中に各出力モードにおける主回路１相分（スイッチングユニットとクランプコンデンサ）の等価回路を示した。スイッチングユニットは、等価的に３方向の切換えスイッチと見なせる。ここで、素子の導通状態を１，０の２値で表わすスイッチング関数Ｓｐ，Ｓｎを用いると、
出力モードＰのとき Ｓｐ＝１，Ｓｎ＝０
出力モードＯのとき Ｓｐ＝０，Ｓｎ＝０
出力モードＮのとき Ｓｐ＝０，Ｓｎ＝１
と表現できる。このとき、スイッチング関数Ｓｐ，Ｓｎと、スイッチング素子７０，７１，７２，７３に与えるゲート信号Ｇpu，Ｇpx，Ｇnx，Ｇnu（オフ信号を０，オン信号を１とする）の関係は、次式で表せる。

In the following description, it is assumed that the voltages vcp and vcn of the clamp capacitors 51 and 52 are DC voltages that are completely smooth and divided by Ed / 2, and the neutral point (0 point) is virtually grounded. Unless otherwise noted, the output voltage indicates the inverter output phase voltage.
As shown in Table 1, the switching elements 70 to 73 constituting the switching unit 7a are turned on / off according to three conduction patterns. That is, in the output mode P that outputs the P-point potential on the DC side, 70 and 71 are on, 72 and 73 are off, the output voltage is Ed / 2, and in the output mode O that outputs the neutral point potential, 71 , 72 are on, 70, 73 are off, a zero potential is output as an output voltage, and in an output mode N that outputs an N-point potential, 70, 71 are off, 72, 73 are on, and the output voltage is -Ed / 2.
Table 1 shows an equivalent circuit of one phase of the main circuit (switching unit and clamp capacitor) in each output mode. The switching unit can be equivalently regarded as a three-way changeover switch. Here, when switching functions Sp and Sn representing the conduction state of the element by binary values of 1 and 0 are used,
When output mode P: Sp = 1, Sn = 0
In the output mode O, Sp = 0, Sn = 0
When the output mode is N: Sp = 0, Sn = 1
Can be expressed as At this time, the relationship between the switching functions Sp, Sn and the gate signals Gpu, Gpx, Gnx, Gnu (off signal is 0 and on signal is 1) given to the switching elements 70, 71, 72, 73 is as follows. I can express.

従って、各相毎に２つのスイッチング関数Ｓｐ，Ｓｎを用意することにより、スイッチング素子の導通状態を決定することができる。このスイッチング関数Ｓｐ，Ｓｎは、パルス幅変調（ＰＷＭ）制御により、出力電圧ｅｕが正弦波状になるように決定される。
なお、３レベルインバータの主回路の詳細は、特開昭５１−４７８４８号公報，特開昭５６−７４０８８号公報などに記載されている。

Therefore, by preparing two switching functions Sp and Sn for each phase, the conduction state of the switching element can be determined. The switching functions Sp and Sn are determined so that the output voltage eu becomes sinusoidal by pulse width modulation (PWM) control.
Details of the main circuit of the three-level inverter are described in JP-A-51-47848, JP-A-56-74088, and the like.

By the way, in the case of performing speed control over a wide range from the variable voltage variable frequency (VVVF) region to the constant voltage variable frequency (CVVF) region with a limited power supply voltage as in an electric vehicle, the output shown by the solid line in FIG. Voltage characteristics are required. That is, by adjusting the output voltage in proportion to the inverter frequency in the low speed region (this region is referred to as the VVVF control region), the magnetic flux in the motor is kept substantially constant, a predetermined torque is ensured, In the high-speed region, the inverter frequency is continuously increased while maintaining the maximum output voltage of the inverter (this region is called the CVVF control region), thereby realizing high-speed operation with the maximum voltage utilization rate with a limited voltage. is there.
However, in the conventionally known unipolar modulation method, the minimum output pulse determined by the minimum on-time of the switching element in a region where the inverter frequency is low and a minute output voltage control is required (near the starting point of the VVVF control region). A voltage pulse smaller than the width cannot be realized, and a voltage larger than the command is output as shown by a broken line in FIG.
For example, when considering the case where the voltage pulses of the inverter output voltage all have a minimum pulse width determined by the minimum on-time Ton of the switching element, the output voltage effective value E at this time is

ここに、Ｆｃ：キャリア周波数
で与えられ、これよりも小さな電圧は制御できない。ここで、Ｅmaxは１８０゜通流の方形波電圧の実効値であり、

Here, Fc is given by the carrier frequency, and a voltage smaller than this cannot be controlled. Here, Emax is the effective value of the square wave voltage with 180 ° flow,

で与えられ、３レベルインバータの最大出力電圧もほぼこのＥmaxに一致する。
上記（数２）によれば、Ｆｃ＝５００ｋＨｚ，Ｔon＝１００μｓのとき、Ｅ＝０.１Ｅmaxであり、この場合、最大出力電圧Ｅmaxの１０％以下の電圧は制御できないことになる。そのため、ユニポーラ変調だけでは制御可能な出力電圧の下限値が制限され、連続的な電圧制御が困難であるという問題があった。

The maximum output voltage of the three-level inverter is almost equal to this Emax.
According to the above (Equation 2), when Fc = 500 kHz and Ton = 100 μs, E = 0.1Emax. In this case, a voltage of 10% or less of the maximum output voltage Emax cannot be controlled. For this reason, the lower limit value of the controllable output voltage is limited only by unipolar modulation, and there is a problem that continuous voltage control is difficult.

In order to solve this problem, dipolar modulation (dipolar mode) is effective. However, in the prior art, attention is required when shifting from this dipolar modulation to unipolar modulation (unipolar mode).
On the other hand, the maximum voltage E that can be output by unipolar modulation is the limit point of ideal sine wave modulation (modulation factor A = 1).

であり、スイッチング素子の最小オフ時間Ｔoffを考慮した場合には、

In consideration of the minimum OFF time Toff of the switching element,

ここに、Ｆｃ：キャリア周波数
となる。例えば、Ｆｃ＝５００Ｈｚ，Ｔoff＝２００μｓのとき、Ｅ＝０.７０７Ｅmaxであり、この場合には、最大出力電圧Ｅmaxの約７０％までしかカバーできないことになる。この時、１パルスモードのパルス幅を調整できないとすると、基本波が不連続となり、また、１パルスモードのパルス幅が調整可能とすると、パルスの幅を小さくして連続性を保とうとするため、今度は、高調波の連続性が失われてしまう。
この電圧範囲をカバーする変調方式は種々考えられるが、パルス発生制御の容易さ，ユニポーラ変調との整合性，出力電圧に含まれる高調波の連続性等の観点から過変調（過変調モード）が最も効果的であるといえる。過変調領域では、出力電圧半周期の電圧パルス列の中央部分（基本波瞬時値のピーク付近）におけるパルス間の狭幅スリットを徐々に埋めることにより、出力電圧を１パルス付近まで拡大することを可能としている。

Here, Fc is the carrier frequency. For example, when Fc = 500 Hz and Toff = 200 μs, E = 0.707 Emax. In this case, only about 70% of the maximum output voltage Emax can be covered. At this time, if the pulse width of the 1-pulse mode cannot be adjusted, the fundamental wave becomes discontinuous. If the pulse width of the 1-pulse mode can be adjusted, the pulse width is reduced to maintain continuity. This time, the continuity of the harmonics is lost.
Various modulation methods that cover this voltage range are conceivable, but overmodulation (overmodulation mode) is considered from the viewpoints of ease of pulse generation control, consistency with unipolar modulation, continuity of harmonics contained in the output voltage, etc. It can be said that it is the most effective. In the overmodulation region, it is possible to expand the output voltage to near one pulse by gradually filling the narrow slit between pulses in the middle part of the voltage pulse train of the output voltage half cycle (near the peak of the fundamental wave instantaneous value). It is said.

In the limit of overmodulation control, that is, in a region where the modulation rate is extremely large, the mode shifts to a so-called one-pulse mode in which only one pulse exists in a half cycle of the output voltage, and the output voltage at this time reaches approximately Emax. However, as it is, the transition timing from overmodulation to one pulse or from one pulse to overmodulation depends on the modulation rate and the carrier frequency. Therefore, this timing cannot be arbitrarily set. Voltage continuity is impaired.
Therefore, the overmodulation control is shifted to one-pulse control capable of voltage control by pulse width control that is not an extension of overmodulation (that is, how to create a one-pulse mode in which the modulation rate is not infinite). This enables transition at a predetermined timing between overmodulation and one-pulse control, and realizes continuous transition of the fundamental voltage.

By continuously performing these series of transition controls, a stable and stable output voltage can be obtained continuously from zero voltage to the maximum voltage while selecting a pulse mode corresponding to the required output voltage.
That is, as shown in FIG. 2, when the induction motor 6 is controlled at V / F = constant as shown in the figure, dipolar modulation is used from the start to F1, and when the inverter frequency reaches F1, the unipolar modulation region is entered. Then, F2 is sequentially shifted to the overmodulation region, and further F3 is shifted to one pulse region.

FIG. 3 shows an example of a modulated wave that can realize the above idea based on a unified voltage command.
The fundamental modulation wave a proportional to the fundamental wave component of the output voltage is created from the following equation based on the inverter frequency command Fi * and the output voltage command E * from the higher-level current control means.

ここに、Ａ：変調率，ｔ＝時間，θ：位相（＝２πＦｉ*ｔ）
ここで、正弦波変調領域における変調率Ａ(０≦Ａ≦１)は、次式で与えられる。

Where A: modulation factor, t = time, θ: phase (= 2πFi * t)
Here, the modulation factor A (0 ≦ A ≦ 1) in the sine wave modulation region is given by the following equation.

この基本変調波ａは、ダイポーラ変調，ユニポーラ変調とも全く同一であり、過変調では後で説明するように変調率Ａの算出方法が異なる以外は、やはり同じである。

The fundamental modulation wave a is exactly the same for dipolar modulation and unipolar modulation, and is the same in overmodulation except that the calculation method of the modulation factor A is different as will be described later.

  In order to enable continuous transition between dipolar modulation and unipolar modulation, here, positive and negative bias modulated waves abp and abn shown in the following equations are provided.

ダイポーラ変調制御では、上記ａbp，ａbnがそのまま正側変調波ａｐと負側変調波ａｎとなる。

In the dipolar modulation control, the above abp and abn become the positive modulation wave ap and the negative modulation wave an as they are.

なお、ここではスイッチング関数Ｓｐ，Ｓｎの作成を簡便化するため、ａｐ，ａｎとも正となるように設定している。最終的に、出力電圧のパルス幅は、ａｐ，ａｎの大きさに比例して設定され、ダイポーラ変調の場合には、正負パルスをほぼ１８０゜ずつずらして制御する。

Here, in order to simplify the creation of the switching functions Sp and Sn, both ap and an are set to be positive. Finally, the pulse width of the output voltage is set in proportion to the magnitudes of ap and an, and in the case of dipolar modulation, control is performed by shifting the positive and negative pulses by approximately 180 °.

  In unipolar modulation, the positive and negative modulation waves ap, an are

で与えられる。
スイッチング素子の最小オフ時間が無視できるほど小さい場合には、ａｐ，ａｎの瞬時値が１以上のとき最大のパルスを出力する（後述の過変調）。
ここで、バイアスＢの設定は移行制御において極めて重要であることがわかる。Ｂの値によりダイポーラ変調領域とユニポーラ変調領域との移行制御が実現され、
（ａ）Ａ／２≦Ｂ＜０.５のとき ダイポーラ変調
（ｂ）Ｂ＝０ のとき ユニポーラ変調
となる。
一方、過変調制御では、変調率Ａを１以上まで高め、出力電圧の半周期の中央部分のパルス間のスリット（ゼロ電圧出力期間）を抑制して、出力電圧を向上させる。
さらに電圧指令を高めた場合には、過変調モードから１パルスモードに移行する。この動作については、以下の実施例の中で説明する。
このように、ダイポーラ変調，ユニポーラ変調及び過変調を統一した電圧指令に基づいて実現し、最大出力となる１パルスまでの連続移行制御が可能となる。

Given in.
When the minimum OFF time of the switching element is negligibly small, the maximum pulse is output when the instantaneous value of ap and an is 1 or more (overmodulation described later).
Here, it can be seen that the setting of the bias B is extremely important in the transition control. The transition control between the dipolar modulation region and the unipolar modulation region is realized by the value of B,
(A) When A / 2 ≦ B <0.5 Dipolar modulation (b) When B = 0 Unipolar modulation.
On the other hand, in the overmodulation control, the modulation factor A is increased to 1 or more, and a slit (zero voltage output period) between pulses in the central portion of the half cycle of the output voltage is suppressed to improve the output voltage.
When the voltage command is further increased, the overmodulation mode is shifted to the 1 pulse mode. This operation will be described in the following embodiments.
In this way, dipolar modulation, unipolar modulation, and overmodulation are realized based on a unified voltage command, and continuous transition control up to one pulse that becomes the maximum output becomes possible.

The configuration of the first embodiment that realizes the above concept will be described below.
FIG. 1 shows an example of a pulse width modulation device that controls the above-described switching unit and outputs an AC voltage having a three-level potential.
In FIG. 1, 1 is a multi-pulse generating means for outputting a dipolar modulation waveform, a unipolar modulation waveform, or an overmodulation waveform in accordance with output voltage related information and transition control information, and 2 outputs a 1 pulse waveform in accordance with output voltage related information (1 1-pulse generating means (pulse mode), and 3 is a transition control means for continuously shifting each PWM mode. The gate signal which is the output of the transition control means 3 is given to the switching element in the switching unit of each phase via a gate amplifier (not shown), and is turned on / off. The pulse width modulation means composed of the multi-pulse generation means 1, the 1-pulse generation means 2 and the transition control means 3 is a characteristic part of the present invention.
In this example, the output voltage related information taken into the pulse width modulation means is given from the higher-level current control means 8. The current control means 8 creates a slip frequency command Fs * of the induction motor 6 from the current command by the current adjustment means 81 (by the deviation between the current command value and the actual motor current), and the rotational frequency attached to the induction motor 6. The inverter frequency command Fi * is created by adding the rotation frequency Fr of the induction motor detected by the detecting means 61 and the Fs *.
Further, the output voltage setting means 82 creates an output voltage command E * based on this Fi * and the DC voltage Ed of the three-level inverter (a voltage between PNs and equal to the sum of the clamp capacitor voltages vcp + vcn).
The output voltage setting means 82 sets a large slope when Ed is low (Ed = Ed1) and a small slope when Ed is high (Ed = Ed3) so that the output voltage always becomes as required. Thus, the output voltage characteristic shown in FIG. 2 is realized. These current control means may output an instantaneous value of the output voltage.

The configuration and operation of the pulse width modulation means will be described in detail with reference to FIGS.
FIG. 4 shows an example of the overall configuration of the pulse width modulation means. Here, the multi-pulse generation unit 1 includes a fundamental modulation wave generation unit 11, a bias superimposition unit 12, a positive / negative distribution unit 13, a reference signal generation unit 14, and a pulse generation unit 15.
The fundamental modulation wave generating means 11 obtains the phase θ by time integrating the inverter frequency command Fi * received as the output voltage related information by the phase calculating means 112, and obtains the sine value sin θ at this θ. On the other hand, the amplitude setting means 111 calculates and outputs the amplitude A (modulation factor) of the fundamental modulation wave from the voltage command E *, which is one of the output voltage related information, and halves and then multiplies it by sin θ to obtain an amplitude of 1 / Two instantaneous basic modulation waves a / 2 are generated and output. The bias superimposing means 12 adds and subtracts the bias B from the multi-pulse transition control means 31 of the transition control means 3 to this a / 2 to create and output two positive and negative bias modulated waves abp and abn.

Here, the continuous transition between the dipolar modulation and the unipolar modulation depends on the setting of the bias B. FIG. 5 shows a configuration example of the dipolar / unipolar transition control means 311 performed by setting the bias B. The dipolar / unipolar transition control means 311 obtains the modulation rate A by multiplying the output voltage command E * by 4 / π by 311a, and the bias generation means 311b determines the bias B corresponding to the modulation rate A. That is, B = Bo (where Bo ≧ A / 2) is set where the modulation factor A is small and a minute output voltage is required, and B = 0 is set when A = A1 is reached. If A1 is determined in advance so that the output voltage when A = A1 is larger than the voltage shown in Expression (2), voltage control from a minute voltage including zero becomes possible.
Further, the positive and negative bias modulated waves abp and abn are distributed and synthesized by the positive / negative distribution means 13 to the positive part of abp and abn to ap and the negative part of abp and abn to an. The positive and negative modulation waves ap, an maintaining the continuity of the fundamental component of the output voltage from unipolar modulation to unipolar modulation are generated.
Based on the positive and negative modulation waves ap, an, the pulse generator 15 generates switching functions Sp, Sn having a pulse generation period of 2 To. The reference signal generation means 14 determines the pulse generation period To according to the switching frequency command Fsw *. Here, the relationship between Fsw * and To can be expressed by the following equation.

The pulse generation operation of the pulse generation means 15 will be described with reference to FIG.
In FIG. 6, the pulse timing setting means 151 obtains the rising timing Tpup of Sp and the falling timing Tndn of Sn from the following equations based on ap, an, aoff, and To (an and aoff will be described later). Process 1).

次の周期では、Ｓｐの立下がりのタイミングＴpdn及びＳｎの立上がりのタイミングＴnupを処理１と同様に求める（処理２）。

In the next cycle, the fall timing Tpdn of Sp and the rise timing Tnup of Sn are obtained in the same manner as in process 1 (process 2).

上記の処理１と処理２を交互に行うことにより、スイッチング関数Ｓｐ，Ｓｎが作成される。
ここで、ａon，ａoffは、スイッチング素子の最小オン時間Ｔon及び最小オフ時間Ｔoffから定まる値であり、

The switching functions Sp and Sn are created by alternately performing the processing 1 and the processing 2 described above.
Here, aon and aoff are values determined from the minimum on-time Ton and the minimum off-time Toff of the switching element,

で与えられる。すなわち、図７（Ｓｐの例）に示すように、オンパルス幅Ｔwon、及びオフパルス幅Ｔｗoffは、

Given in. That is, as shown in FIG. 7 (example of Sp), the on-pulse width Twon and the off-pulse width Twoff are

となり、図８の破線で示す特性を持つ。ここで、オンパルス幅Ｔwonがスイッチング素子によって定められた最小オン時間Ｔon以下とならないように、また、オフパルス幅Ｔｗoffがスイッチング素子によって定められた最小オフ時間Ｔoff以下とならないように、図８の実線で示す特性とする。これを実現するため、図６のパルスタイミング設定手段１５１の機能を付加した。これによって発生する出力電圧基本波成分の不連続は極めて小さいため、無視しても差し支えない。

And has the characteristics shown by the broken line in FIG. Here, the solid line in FIG. 8 prevents the on-pulse width Twon from being equal to or less than the minimum on-time Ton determined by the switching element and the off-pulse width Twoff to be equal to or less than the minimum off-time Toff defined by the switching element. The characteristics shown are as follows. In order to realize this, the function of the pulse timing setting means 151 of FIG. 6 is added. Since the discontinuity of the fundamental component of the output voltage generated by this is extremely small, it can be ignored.

Note that aoff can be varied within a range in which the discontinuity of the fundamental component of the output voltage can be ignored, and is given from the unipolar / overmodulation transition control means 312 as the transition timing from unipolar modulation to overmodulation. If aoff is set constant, pulse generation can be simplified.
That is, since the pulse timing setting unit 151 automatically shifts from unipolar to overmodulation, it is not necessary to provide the unipolar / overmodulation shift control unit 312 that outputs aoff.
The switching function generating means 152 generates a reference signal having a period To and sets Sp and Sn based on the above Tpup, Tndn or Tpdn, Tnup in synchronization therewith.

  An example of the switching function during overmodulation is shown in FIG. When the instantaneous value Ap of ap exceeds aoff, the slits between the pulses of the switching function Sp (hatched portions in FIG. 9C) are filled. The buried slit width is smaller than the minimum OFF time Toff of the switching element, and gradually disappears by about 1 to 2 at a time.

FIG. 10 shows a flowchart when the pulse timing setting means 151 is realized by software.
By the way, in overmodulation control, the maximum voltage state is maintained by filling a slit between pulses in the central portion of the output voltage half cycle, and PWM control is performed only in the vicinity of the zero cross of the modulated wave. Therefore, in this region, the modulation factor A and the output voltage that is actually output are nonlinear, and even if the modulation factor A is increased linearly, the output voltage does not increase linearly following this.
Therefore, by linearizing the setting of the modulation factor A, the output voltage at the time of overmodulation is linearized. That is, if the switching frequency in the PWM control portion is sufficiently high, the relationship between the fundamental value E of the output voltage and the modulation factor A can be expressed by the following equation.

従って、上式の関係からあらかじめＥ*とＡの関係を算出しておき、図１１に示す振幅設定手段１１１を構成することにより、出力電圧をＥ*に対して直線的に調整できる。その結果、特に１パルスに近い高電圧域での電圧制御性を向上できる。

Therefore, the output voltage can be linearly adjusted with respect to E * by calculating the relationship between E * and A in advance from the relationship of the above equation and configuring the amplitude setting means 111 shown in FIG. As a result, it is possible to improve voltage controllability particularly in a high voltage range close to one pulse.

When the voltage command is further increased, the overmodulation mode is shifted to the one-pulse mode by the action of the changeover switch 32 of the transition control means 3. Changeover switch 32, which is one _{S PM} of the output of the multi-pulse transfer controlling means 31
Multi-pulse side when S _PM = 0
When S _PM = 1, it is switched to the 1 pulse side. FIG. 12 shows an example of the 1-pulse / multi-pulse switching control means 313. In this example, hysteresis is provided so that when the voltage command E * exceeds E1P, the multi-pulse mode is shifted to the 1-pulse mode, and when E * is smaller than EMP, the mode is shifted from the 1-pulse mode to the multi-pulse mode. Yes. This suppresses inadvertent PWM mode transition and obtains a stable output voltage with little transient fluctuation.
The one-pulse generating unit 2 includes a phase calculating unit 21 and a pulse generating unit 22. The operation of the phase calculation means 21 may be exactly the same as 111, and 21 may be omitted and the output of 111 may be used.

  A configuration example of the pulse generating means 22 is shown in FIG. Unlike the two-level PWM, the three-level PWM can adjust the output voltage by controlling the pulse width during one-pulse control. Therefore, from the voltage command E *, the rise timing phase α and the fall timing phase β of the pulse are calculated.

で求める。このα，βを位相θを基準にしてセットし、Ｓｐ，Ｓｎを作成，出力することにより、１パルス波形を実現する。

Ask for. By setting α and β with reference to the phase θ, and creating and outputting Sp and Sn, a one-pulse waveform is realized.

In this way, dipolar modulation, unipolar modulation, and overmodulation are realized based on a unified voltage command, and continuous transition control up to one pulse that becomes the maximum output becomes possible.
In this embodiment, the output voltage can be continuously and smoothly adjusted from the zero voltage to the maximum voltage, and there is an effect that it is possible to provide a highly accurate and stable output voltage.

In the first embodiment shown in FIG. 4, the output pulse train of the multi-pulse generator is generated asynchronously with the inverter frequency, and the output pulse of the one-pulse generator is controlled in synchronization with the inverter frequency.
The reason for this is that, in the above-described conventional technique that employs the synchronous system in the multi-pulse region, first, control for phase management is complicated, and second, the output voltage command is distorted from a sine wave due to some control request. When there is a need to correct the output voltage command (in FIG. 1, the inverter frequency Fi * and the output voltage command E * are adjusted according to the requirements for electric vehicle control), there is a problem that the output voltage command cannot be faithfully reproduced.
In other words, the first problem is that the synchronous type outputs a pulse that is an integer multiple of the inverter frequency, and therefore has a table having a relationship between the phase and the generated pulse for each pulse mode, and the phase obtained from the pulse mode and the inverter frequency. From this, the pulse generation phase is read out and output. The amount of calculation required for the management of the phase and the memory for each pulse mode become enormous, resulting in complicated control.
The second problem is that the synchronous type shown in the prior art has pulse data for 90 °, but the data is created so that the output voltage is a sine wave. There is a problem that it cannot be expressed accurately.

Therefore, in this embodiment, these solutions are achieved by making the generation of pulses in the multi-pulse mode asynchronous with the inverter frequency.
That is, for the first problem, the pulses can be generated independently without being constrained by the inverter frequency for generating the pulses. That is, in FIG. 4, the switching frequency command Fsw * can be set independently of the inverter frequency command Fi * (FIG. 4, the reference generation 14 is independent of the inverter frequency).
For this reason, a complicated control procedure for generating a pulse is not required, and the control can be simplified.
Also, for the second problem, the asynchronous method eliminates the need for data for each phase and can output a pulse corresponding to an instantaneous voltage command. Even so, it can be expressed faithfully. In addition, since the control related to the phase calculation and the like is simplified as described above, the calculation for outputting the pulse corresponding to the sequential voltage command can be performed, and the calculation cycle can be shortened. The degree can be increased.
In addition, since the switching frequency does not depend on the inverter frequency when the asynchronous method is used, the change in the switching frequency can be minimized, and the change in sound quality before and after the pulse mode switching (abnormal noise, unpleasant sound) seen in the synchronous method There is also an effect that can be minimized.

  Although the above embodiment has been described by taking a three-level inverter as an example, the same applies to a two-level inverter or a multi-level inverter having three or more levels.

By the way, in the case of a switching element such as a GTO thyristor that performs switching at a relatively low frequency, interference between the sideband component generated depending on the switching frequency and the fundamental component of the inverter frequency is present in the output voltage harmonics. May occur. In order to avoid this, the dipolar modulation mode and the unipolar modulation mode are made asynchronous with respect to the inverter frequency in the PWM mode of the multi-pulse generating means, and the overmodulation mode and the one-pulse mode are made synchronous (FIG. 14).
With this configuration, a more stable voltage can be supplied even during overmodulation.
FIG. 14 shows a second embodiment of the present invention.

  FIG. 15 shows an example of multipulse transition control. FIG. 15 shows only the multipulse transition control means 31. This shifts the four PWM modes depending on both the inverter frequency command Fi * and the voltage command E *. That is, dipolar modulation when Fi * <F1 and E * <E1, unipolar modulation when Fi * ≧ F1 and E1 ≦ E * <E2, overmodulation when E2 ≦ E * <E3, and 1 when E * ≧ E3. Pulse. As a result, even when the output voltage is soft-started in a high-speed region where the frequency is high, such as during regenerative start-up or repowering, the transition condition of dipolar modulation → unipolar modulation → overmodulation → one pulse is satisfied and stable. The voltage can be raised. In addition, since dipolar modulation control is always performed in a low frequency region, current concentration on a specific switching element as in unipolar modulation can be avoided.

Next, Example 2 will be described.
If the first embodiment is expanded and partial dipolar modulation in which both modulation waveforms are mixed is introduced between dipolar modulation and unipolar modulation as shown in FIG. 16, the smoothness of the output voltage and the switching frequency is further increased. be able to.

An example of the output voltage command waveform is shown in FIG. 17 is exactly the same as FIG. 3 except for (B). Hereinafter, this partial dipolar will be described.
Due to the effect of bias superposition and positive / negative distribution, even if the bias B is set in a range where neither dipolar modulation nor unipolar modulation is set (0 <B <A / 2), the voltage as required by the fundamental modulation wave is reproduced without excess or deficiency. It is possible. In this case, the vicinity of the peak of the output voltage is unipolar modulation, and the base is partial dipolar modulation which is dipolar modulation. The positive side modulation wave ap and the negative side modulation wave an at this time are

となる。（ａｐ−ａｎ）が常に基本変調波ａに一致し、出力電圧基本波の瞬時値の連続性も維持されることがわかる。
上記性質を利用して、変調率Ａの増加に従ってバイアスＢを徐々に減少させれば、ダイポーラ変調からユニポーラ変調まで部分ダイポーラ変調を介して連続的に移行できる。当然ながら、その逆も可能である。

It becomes. It can be seen that (ap-an) always coincides with the fundamental modulation wave a, and the continuity of the instantaneous value of the output voltage fundamental wave is also maintained.
If the bias B is gradually decreased as the modulation factor A increases by using the above property, it is possible to continuously shift from dipolar modulation to unipolar modulation through partial dipolar modulation. Of course, the reverse is also possible.

  An example of the dipolar / unipolar transition control means is shown in FIG. If the bias B is set as shown by the solid line in FIG. 18, dipolar modulation is performed in the region of 0 ≦ A ≦ A1, partial dipolar modulation is performed in the region of A1 <A <A2, and unipolar modulation is performed in the region of A ≧ A2. In this case, no abnormal noise is generated from the electric motor when switching between dipolar modulation and unipolar modulation, which is effective in reducing the noise of the apparatus.

When FIG. 18 is applied, the PWM mode can be managed for each region as shown in FIG. FIG. 19 shows only the multipulse transition control means 31.
This shifts the five PWM modes depending on both the inverter frequency command Fi * and the voltage command E *. That is, dipolar modulation when Fi * <Fo and E * <Eo, partial dipolar modulation when Fo ≦ Fi * <F1 and Eo ≦ E * <E1, and unipolar modulation when Fi * ≧ F1 and E1 ≦ E * <E2. , E2 ≦ E * <E3 is overmodulated, and E * ≧ E3 is one pulse. As a result, even when the output voltage is soft-started at a high speed with a high frequency, such as during regenerative start-up or repowering, the transition condition of dipolar modulation → partial dipolar modulation → unipolar modulation → overmodulation → one pulse is used. Satisfactory and stable voltage rise is possible. In addition, the same effect as that at the time of regenerative activation can be given at the time of re-adhesion. Furthermore, in any operating state, there is an effect that generation of abnormal noise from the motor at the time of pulse mode switching can be minimized.

FIG. 20 shows the relationship between the inverter frequency and the switching frequency.
By the way, in the inverter used for the electric vehicle control apparatus for railway vehicles, the variable range of the inverter frequency Fi * is about 0 to 300 Hz. The inverter frequency Fcv at which the output voltage is maximum is 1/5 to 1/3 of the upper limit of the inverter frequency variable, and the upper limit of Fcv is about 100 Hz. In order to avoid fluctuations in the output current due to interference between the harmonics generated around the switching frequency and the fundamental frequency of the inverter frequency when generating pulses asynchronously, the switching frequency is about 10 times Fcv, that is, 1 kHz or more. A switching frequency is required.
Furthermore, to reduce noise (such as the above-mentioned abnormal noise), it is effective to minimize fluctuations in the switching frequency. By introducing overmodulation, fluctuations in the switching frequency in the multi-pulse region are within 1 to 2 Fi. Can be.
Of course, if a microprocessor or the like is used, a part or all of the pulse width modulation means can be programmed and realized by software.

FIG. 21 shows an example of a flowchart for realizing by software the rise and fall timing calculations in the pulse width modulation means of FIG.
In the above description, the induction motor load is described as an example. However, the present invention is not limited to this. In addition, all the above explanations are for inverters, but the output terminals of these inverters can be connected to an AC power source via a reactance element to operate as a self-excited converter that converts AC to DC. is there. In this case, the same effect as that of the inverter can be expected.

  Although the above has described the case of a three-level inverter, the concept of the present invention can be applied to a multi-level inverter having three or more levels.

  The present invention makes it possible to continuously and smoothly adjust the inverter output voltage from zero voltage to the maximum voltage. Further, the pulse generation control system can be simplified. Car can be provided.

The block diagram which shows Example 1 of this invention. The figure explaining the relationship between an output voltage characteristic and PWM mode. Explanatory drawing of the modulation wave for PWM mode continuous transition in a multipulse area | region. FIG. 2 is a detailed explanatory diagram of the configuration of FIG. 1. The figure which shows an example of a dipolar / unipolar transfer control means. The figure which shows an example of the pulse generation means in a multipulse generation means. The wave form diagram which shows the relationship of on-off pulse width. The figure which shows the characteristic of an on-off pulse width. The figure which shows an example of an overmodulation waveform. The figure which shows the flowchart of the pulse timing setting means by software. The figure which shows the example of 1 structure of an amplitude setting means. The figure which shows an example of a multi-pulse / 1 pulse switching control means. The figure which shows an example of 1 pulse generation means. The figure which shows the example of 1 structure of Example 2 of this invention. The figure which shows an example of a transfer control means. The output voltage characteristic in the case of including another PWM mode and the relationship diagram of PWM mode. The figure explaining the modulation wave of other PWM modes. The block diagram of the transfer control means which implement | achieves another PWM mode. The figure which shows an example of a transfer control means. The figure explaining the relationship between an inverter frequency and a switching frequency. The figure which shows the flowchart of the pulse width modulation means by software.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Multi pulse generation means, 2 ... 1 pulse generation means, 3 ... Transition control means, 4 ... DC overhead wire, 6 ... Induction motor, 7a, 7b, 7c ... Switching unit, 8 ... Current control means, 11 ... Fundamental modulation wave Generating means, 12 ... bias superimposing means, 13 ... positive / negative distributing means, 14 ... reference signal generating means, 15 ... pulse generating means, 21 ... phase calculating means, 22 ... pulse generating means, 31 ... multi-pulse shift control means, 32 ... Changeover switch, 50 ... DC reactor, 51, 52 ... Clamp capacitor, 61 ... Rotational frequency detection means

Claims (1)

A power converter that converts a direct current into an AC phase voltage having a three-level potential that is positive, negative, and intermediate voltage by switching control of a plurality of switching elements, and controls the switching element of the power converter, and the power converter Connected to the output side of an electric car
In an electric vehicle power conversion device comprising a control device for controlling acceleration / deceleration of an alternating current motor,
In the driving range of the electric vehicle at a low speed, a half cycle of the fundamental voltage of the output voltage of the power converter
A first period for continuously outputting a plurality of voltage pulses of the same polarity within the period;
The positive and negative voltage pulses are alternately output via the intermediate voltage before and after one period.
First pulse width modulation control means having a second period,
In the high-speed driving range than the low-speed driving range of the electric vehicle, the power converter
A plurality of single polarity pulse trains are applied to the phase of the power converter in the half cycle of the fundamental wave of the power phase voltage.
Second pulse width modulation control means for generating;
Means for shifting the first and second pulse width modulation control means according to the operation region
When
Power conversion device for an electric vehicle characterized by comprising a.

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