JP3755089B2 - Electric vehicle control device - Google Patents

Description

The present invention relates to a control device for an electric vehicle, and more particularly to a control technique for a PWM (pulse width modulation) inverter that supplies an AC output to an electric motor that drives the electric vehicle.

An example of an inverter modulation method is described in Non-Patent Document 1.
As shown in FIG. 2, in the inverter of an electric railway vehicle, when the fundamental frequency of the output voltage is low, control is performed to keep the ratio between the magnitude of the output voltage and the fundamental frequency constant (the region where this control is performed is variable voltage). When the fundamental frequency of the output voltage rises and the magnitude of the output voltage becomes maximum, frequency control is performed while maintaining the maximum voltage (the region where this control is performed can be changed to a constant voltage). Called the frequency domain). In the variable voltage variable frequency region, in order to adjust the output voltage by pulse width modulation control, a multi-pulse mode in which a half cycle of the output voltage is composed of a plurality of voltage pulses is used. On the other hand, in the constant voltage variable frequency region, in order to increase the voltage utilization rate to the maximum and reduce the size of the apparatus, a one-pulse mode in which a half cycle of the output voltage is configured by a single pulse is used.

Page 14 of "Evaluation of Recent Inverter Control Technology", April 1993, "Science of Electric Vehicles" published by Electric Vehicle Research Society, Fig. 1

Conventional inverters using GTO thyristors as switching elements (hereinafter referred to as
As shown in FIG. 3, the GTO inverter uses a multi-pulse mode of a pulse number switching method in which the number of pulses included in one cycle is gradually decreased as the output voltage fundamental wave frequency increases. It was. This is because the upper limit of the switching frequency of the GTO thyristor is several hundred Hz. In this method, since the switching frequency becomes discontinuous when the number of pulses is switched, the timbre change of the magnetic noise occurs with the switching of the number of pulses, which is problematic.
In the GTO inverter, between the output voltage of the three-pulse mode and the one-pulse mode including three voltage pulses in a half cycle of the output voltage, about 10% depending on the limit of the minimum off time of the GTO thyristor. There is a jump, 3 pulse mode and 1
There has been a problem that the torque generated by the motor fluctuates when the pulse mode is switched.

An object of the present invention is to control an electric motor that drives an electric vehicle by using an AC output of a two-level inverter that controls the magnitude of an output voltage from zero to a maximum voltage by a combination of a multi-pulse mode and a one-pulse mode. While eliminating the discontinuity of the switching frequency of the level inverter and eliminating the harsh magnetic noise timbre,
The purpose is to reduce the gap between the output voltages of the multi-pulse mode and the one-pulse mode, and to control the entire output voltage almost continuously.

In order to solve the above-mentioned problem, a plurality of switching elements are controlled by a plurality of control modes to convert a DC voltage into a three-phase AC phase voltage of variable voltage and variable frequency, and output a positive / negative binary pulse as the phase voltage. In a control device for an electric vehicle provided with a two-level power converter and an electric motor that is supplied with an AC output from the power converter and drives the electric vehicle, the control mode includes a half cycle of the output voltage fundamental wave. An asynchronous overmodulation control mode is interposed between a pulse width modulation control mode (bipolar mode) that outputs a plurality of voltage pulses and a single pulse control mode that outputs a single voltage pulse in the half cycle of the output voltage fundamental wave. , the asynchronous over-modulation control mode, as the waveform of the half cycle of the output voltage, equally spaced pulses and the output voltage pulse interval of the zero cross vicinity of the fundamental wave of the output voltage becomes uniform Has a wide pulse having a center peak of the wave, so the start of the one-pulse control mode gradually widening the width of the wide pulse, when driving the electric vehicle from a low speed range to the high speed range, the power converter Voltage command to control output voltage
Depending on the size of the pulse width modulation control mode, asynchronous overmodulation control mode,
The pulse control mode is shifted sequentially.

As described above, according to the present invention, a motor that drives an electric vehicle is controlled by using an AC output of an inverter that controls the magnitude of an output voltage from zero to the maximum voltage by a combination of a multi-pulse mode and a one-pulse mode. In this case, the discontinuous change in the magnetic noise of the two-level inverter can be eliminated, and the entire output voltage range can be controlled almost continuously.
Also, by adopting the overmodulation mode as the control mode, the multi-pulse mode and 1
It is possible to configure a two-level inverter that can be switched smoothly without changing current and torque generated by the electric motor that drives the electric vehicle when the pulse mode is switched.

  The best mode of the present invention will be described below with reference to FIGS.

The configuration of the PWM mode of the inverter of the present invention is as shown in FIG. It operates in bipolar mode (pulse width modulation control mode) in the low output voltage range, overmodulation mode (overmodulation control mode) in the high output voltage range, and 1 pulse mode (1 pulse control mode) in the maximum output voltage range. Here, the multi-pulse mode has a bipolar mode and an overmodulation mode.
FIG. 1 is a block diagram showing an embodiment of the present invention, in which a voltage type two-level inverter is used as a converter for controlling an induction motor for driving an electric vehicle. In the figure,
6 is an induction motor, 5 is a two-level three-phase PWM inverter for driving the induction motor, 9 is a DC overhead line serving as a power source for the inverter, and 7 and 8 are filter reactors and capacitors on the inverter DC input side.
The multi-pulse generating means 2, the one-pulse generating means 3, and the PWM mode selecting means 4 in FIG.
The phase θx of the output voltage fundamental wave of each phase obtained by integrating the output voltage command E * of the inverter and the frequency command Fi * by the integrator 1 (subscript x is a generic name for the phase). That is, the inverter control signal is generated based on any one of the phases u, v, and w. Of the inverter control signals, S1x, S2x, Sx
Is called a switching function, and is defined as 1 when the positive arm of the inverter is on and 0 when the negative arm is on.

First, an inverter control signal generation method will be described. An example (for one phase) of the multi-pulse generating means 2 of FIG. 1 is shown in FIG. Here, the switching functions of the bipolar mode and the overmodulation mode are generated by the same means. The output voltage command → modulation rate conversion means 21 obtains the modulation rate A, that is, the amplitude of the modulation wave from the output voltage command E *. When the carrier wave amplitude is 1, 0 ≦ A ≦ 1 in the bipolar mode and A in the overmodulation mode
> 1. In order to make the magnitude of the output voltage fundamental wave coincide with the voltage command, E * and A are made to correspond in (Equation 1) in the bipolar mode and (Equation 2) in the overmodulation mode.

In the function y = sin (x) 22, the phase of the output voltage fundamental wave (equivalent to the phase of the modulation wave) θx
Find the sin of Multiplying this by the modulation factor A is the modulated wave ax. Modulated wave a
x and the carrier wave frequency (equivalent to the switching frequency in the bipolar mode) Fc are given to the switching function calculating means 24 to obtain the switching function S1x. The switching function calculation means 24 generates a carrier wave that is a triangular wave having an amplitude of 1 and a frequency Fc,
The switching function is generated by comparing the value of the modulated wave with that. Further, the switching function may be obtained by calculation from the modulated wave ax and the pulse interval without using the triangular wave.

An example of the waveform of the switching function in the bipolar mode and the overmodulation mode obtained by the triangular wave comparison is shown in FIGS. 6 and 7, respectively.
In the inverter device of the present invention, devices capable of switching at several kHz, such as IGBTs and large-capacity power transistors, are used as switching elements (hereinafter collectively referred to as IGBT inverters), and in the multi-pulse mode, modulated waves are used. And the carrier wave is asynchronous.
In the bipolar mode shown in FIG. 6, since 0 ≦ A ≦ 1, the carrier wave 242
And the switching function 243 correspond to each other, and the carrier wave 242 and the modulated wave 241 are not synchronized. Further, since A> 1 in the overmodulation mode shown in FIG.
The switching function 246 of the wide pulse is obtained in the part exceeding, and the switching function 246 according to the comparison between the carrier wave 245 and the modulated wave 244 is obtained in the other part. Also in this overmodulation mode, the carrier wave 245 and the modulated wave 244 are generated asynchronously. As described above, in the figure, for the sake of understanding, a method of obtaining a switching function by comparing a carrier wave and a modulated wave is shown, but the switching function can also be obtained by calculation from the modulated wave ax and the pulse interval.

This makes the switching frequency constant in bipolar mode,
In the overmodulation mode, the switching frequency in the 1-pulse mode described below can be gradually approached. In this multi-pulse mode, since the modulated wave and the carrier wave are asynchronous, it is necessary to make the carrier wave frequency sufficiently higher than the modulated wave frequency.

FIG. 8 shows an example of the waveform of the switching function generated by the one-pulse generating means of FIG. The value of the switching function S2x is 1 when the sign of the fundamental wave 31 (which can have any amplitude) of the output voltage is positive, and the value of S2x is 0 when the sign is negative.

Next, the combination of the multi-pulse mode and the one-pulse mode for controlling the high output voltage range will be described. As a document written on overmodulation methods,
Proceedings of the Annual Meeting of the Institute of Electrical Engineers of Japan Annual Report No. 106 “Overmodulation Control Method for Voltage Type Three-Phase PWM Inverter”. According to this, it is stated that the operation of the 6-step inverter, that is, the operation of the 1 pulse mode, has a very large modulation rate in the overmodulation mode. However, when the 1-pulse mode is realized in the form of extension of the overmodulation mode (the 1-pulse mode is realized by extremely increasing the modulation rate), the following disadvantages occur.
First, the point at which the overmodulation mode and the one-pulse mode are switched depends on the switching frequency and cannot be arbitrarily set. Second, when the modulation wave in the overmodulation mode and the carrier wave are asynchronous (hereinafter referred to as asynchronous PWM), the modulation wave is near the boundary between the overmodulation mode and the one-pulse mode due to the effect of the turn-on and turn-off times of the element. Pulses near the zero cross may or may not come out. As a result, an imbalance occurs between the positive and negative of the output voltage, and a beat phenomenon occurs in which low-frequency pulsations are superimposed on the load current of the inverter. Third, in the overmodulation mode, as shown in FIG. 7, the output voltage waveform (equivalent to the waveform of a switching function described later) has a uniform pulse interval near the modulation wave (equivalent to the output voltage fundamental wave) zero cross. In other words, it is divided into a portion where the pulse generation period is uniform (equal interval pulse) and a wide pulse portion centering on the modulation wave peak, and the overmodulation mode and the 1 pulse mode in the equal interval pulse portion of the overmodulation mode. Switching can occur. In this case, the load current of the inverter may be disturbed, and the switching element may be destroyed due to overcurrent, or the generated torque of the motor may be significantly changed.

In order to solve these problems, the phase for switching the overmodulation mode and the one-pulse mode (hereinafter referred to as the transition voltage) and the phase of the output voltage fundamental wave (hereinafter referred to as the switching voltage).
(Referred to as transition phase).
First, management of the transition voltage will be described. When the transition voltage is set and the overmodulation mode and the 1-pulse mode are switched using the transition voltage as a boundary, the transition voltage is set to 1 as much as possible.
The output voltage in the pulse mode, that is, a value close to 100% is desirable. This is because the smaller the difference from the maximum value of the output voltage in the overmodulation mode, the smaller the variation in the torque generated by the motor at the time of switching.
However, in asynchronous PWM, the width of each voltage pulse included in one period of the fundamental wave of the output voltage is different for each period, and the output voltage is 10 in the overmodulation mode.
As the number of pulses near the zero cross of the output voltage fundamental wave decreases as it approaches 0%,
This effect becomes apparent and an imbalance occurs between the positive and negative output voltages, and a beat phenomenon occurs in the load current of the inverter. An example of this state is shown in FIG.

FIG. 10 is an example of the relationship between the average pulse number near the output voltage fundamental wave zero cross and the current pulsation due to the beat phenomenon. As shown in FIG. 7, since the portion where the absolute value of the modulated wave is 1.0 or less corresponds to an equally-spaced pulse, the average number of pulses is given by the equation shown in (Formula 3). The current pulsation rate was defined by (Equation 4).

From FIG. 10, if at least one pulse is not secured near the output voltage fundamental wave zero cross, the low frequency pulsation of the inverter load current due to the beat phenomenon becomes extremely large.
Therefore, the set value of the transition voltage is set to a value that ensures at least one voltage pulse in the vicinity of the output voltage fundamental wave zero cross. This value is the output voltage fundamental frequency F
Since it depends on i * and the carrier frequency Fc of the multi-pulse mode, means for calculating from these values may be provided, or it may be calculated and set in advance from the upper limit of the output voltage fundamental frequency Fi *. But you can.

Subsequently, management of the transition phase will be described. Depending on the phase of the output voltage fundamental wave when switching between the overmodulation mode and the one-pulse mode, the state of transient fluctuations in the load current of the inverter and the torque generated by the motor immediately after switching is different. An example of current fluctuation is shown in FIG.
It is shown in 1. FIG. 12A shows a case where the phase of the U-phase output voltage fundamental wave is switched at the same time for three phases at 0 °, as shown in FIG.
On the other hand, as shown in FIG. 13B, FIG. 13B shows a case where the phase of the U-phase output voltage fundamental wave is switched at 90 ° in three phases, and there is almost no transient fluctuation of the current. Absent.

FIG. 14 is an example of the relationship between the phase of the fundamental wave of the output voltage (U-phase reference) and the transient current fluctuation when switching from the overmodulation mode to the 1-pulse mode at once. here,
The current fluctuation rate is defined by (Equation 5).

In FIG. 14, the current fluctuation rate increases every 60 ° in the phase of the fundamental voltage of the output voltage. This is a case where the overmodulation mode and the 1-pulse mode are switched when one of the three phases is an equidistant pulse in the overmodulation mode. In this case, the temporary three-phase output voltage due to the mixture of both modes is changed. As the imbalance increases, the transient current fluctuation also increases. Therefore, as shown in FIG. 15, transient current and torque fluctuations can be suppressed by setting a transition phase in a portion where all phases become wide pulses in the overmodulation mode.

Here, in order to switch the overmodulation mode and the 1-pulse mode at the same time for the three phases, there must be a section where the output voltage of the overmodulation mode in all three phases becomes a wide pulse.
For this purpose, the intersection of two phase modulation waves among the three phases (based on the phase of the U phase modulation wave,
At 30 °, 90 °, 150 °, 210 °, 270 °, 330 °), the absolute value of the modulated wave must be greater than 1.0. Considering the case of 30 °,
Since au = Asin 30 °> 1.0, A> 2, in the overmodulation mode, the modulation factor A and the output voltage E
Since the correspondence of * is given by (Equation 2), it must be E *> 95.6%. Therefore, the transition voltage is larger than 95.6% to switch overmodulation mode and 1-pulse mode in three phases at a time, and a value that secures at least one voltage pulse near the output voltage fundamental wave zero cross by overmodulation. It becomes.

FIG. 16 shows PWM mode selection means 4 for realizing the management of the transition voltage and transition phase.
This is an example of the configuration. The mode selection command generating means 42 compares the transition voltage Ec set in the transition voltage means 41 with the voltage command E *, and generates a mode selection command Mc that indicates whether the multi-pulse mode or the one-pulse mode should be selected.
Here, the mode selection command Mc is determined based on the output voltage command E *. However, since the output voltage command E * uniquely corresponds to the modulation factor A, the modulation factor Ac corresponding to the transition voltage is set in advance. It may be set and the mode selection command Mc may be generated by comparing this with the modulation factor A.
In the variable voltage variable frequency region, the output voltage command and the fundamental voltage of the output voltage also correspond to each other. Therefore, the fundamental frequency Fic of the output voltage corresponding to the transition voltage is set in advance, and the frequency command Fi * May be generated to generate a mode selection command Mc.
The transition phase management means 44 refers to Mc, and when the mode needs to be switched, the phase θx of the output voltage fundamental wave is compared with the transition phase θc set in the transition phase setting means 43, and θx has reached θc. For example, the mode selection signal M is switched. In the mode selection switches 45, 46 and 47, either the output S1x of the multi-pulse generation means or the output S2x of the one-pulse generation means is selected according to the mode selection signal M, and the switching function Sx is determined.

The management of the transition phase may be performed by the following method. If the absolute value of the modulated wave of each phase is taken and all three phases are greater than 1.0, then all phases are in the overmodulated wide pulse portion. Accordingly, at such time, the output of the multi-pulse generating means and the one-pulse generating means is switched.

As described above, the output voltage gap between the multi-pulse mode and the one-pulse mode is reduced from about 10% in the conventional GTO inverter to about 1-2%, and the magnitude of the output voltage is substantially continuous from zero to the maximum voltage. In addition, it is possible to configure a two-level inverter that can be switched smoothly without variation in current and generated torque of the motor when switching between the multi-pulse mode and the one-pulse mode.
The relationship between the fundamental frequency of the output voltage and the switching frequency in the present invention is shown in FIG.
Thus, there is no large discontinuity as in the conventional inverter modulation method of FIG. 3, and the discontinuous timbre change of the magnetic noise can be eliminated.

The present invention controls an electric motor that drives an electric vehicle using an AC output of an inverter that controls the magnitude of an output voltage from zero to the maximum voltage by a combination of a multi-pulse mode and a one-pulse mode, and controls the driving of the electric vehicle. Especially available.

The block diagram which shows one Embodiment of this invention The figure which shows the driving characteristic of the inverter for vehicles The figure which shows the example of the modulation system of the conventional inverter The figure which shows the driving | running characteristic of the inverter by this invention The figure which shows an example of a structure of a multipulse generation means Diagram showing modulated wave, carrier wave, and switching function in bipolar mode Diagram showing modulation wave, carrier wave, and switching function in overmodulation mode The figure which shows the switching function of the fundamental wave of output voltage and 1 pulse mode Diagram showing how the beat phenomenon occurs Relationship between average pulse number and current pulsation near the output voltage fundamental wave zero cross Diagram showing that the state of transient current fluctuation immediately after mode switching differs depending on the transition phase The figure which shows the switching timing of Fig.11 (a) The figure which shows the switching timing of FIG.11 (b) Diagram showing the relationship between transition phase and transient current fluctuation immediately after mode switching Diagram showing transition phase settable section The figure which shows an example of a structure of a PWM mode selection means The figure which shows the relationship between the output voltage fundamental wave frequency and switching frequency of the inverter in this invention

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Integrator, 2 ... Multi pulse generation means, 3 ... 1 pulse generation means, 4 ... PWM mode selection means, 5 ... 2 level three phase PWM inverter, 6 ... Induction motor, 7 ... Filter reactor, 8 ... Smoothing capacitor, 9 ... DC overhead wire, 21 ... frequency command → modulation wave amplitude reference conversion means, 22 ... function y = sin (x), 23 ... switching frequency, 24 ...
Switching function computing means, 241, 244 ... modulated wave ax, 242, 245 ... carrier wave c, 243, 246 ... switching function S1x, 31 ... fundamental voltage of output voltage, 4
DESCRIPTION OF SYMBOLS 1 ... Transition voltage setting means, 42 ... Mode selection command generation means, 43 ... Transition phase setting means, 44 ... Transition phase management means, 45, 46, 47 ... Mode selection switch

Claims (1)

A two-level power converter that controls a plurality of switching elements in a plurality of control modes to convert a DC voltage into a three-phase AC phase voltage of variable voltage and variable frequency, and outputs positive and negative binary pulses as the phase voltage; AC output from the power converter is supplied,
In a control device for an electric car equipped with an electric motor for driving the electric car,
The control mode includes a pulse width modulation control mode (bipolar mode) for outputting a plurality of voltage pulses in a half cycle of the output voltage fundamental wave, and a single pulse for outputting a single voltage pulse in the half cycle of the output voltage fundamental wave. Asynchronous overmodulation control mode is interposed between the control mode and
The asynchronous over-modulation control mode, as the waveform of the half cycle of the output voltage, around the peaks of the fundamental wave of equally spaced pulses and the output voltage pulse interval of the zero cross vicinity of the fundamental wave of the output voltage becomes uniform Having a wide pulse and gradually increasing the width of the wide pulse until the start of the one-pulse control mode;
When driving the electric vehicle from a low speed range to a high speed range , the output voltage of the power converter
Depending on the magnitude of the voltage command to control the pulse width modulation control mode and the non-
A control device for an electric vehicle, wherein a synchronous overmodulation control mode and the one-pulse control mode are sequentially shifted.

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