JP6525732B2 - Power conversion circuit and switching power supply using the same - Google Patents

Description

  The present invention relates to a power conversion circuit and a switching power supply using the same, and more particularly to a power conversion circuit including an isolation transformer and a switching element, and a switching power supply using the same.

  Patent Document 1 discloses a transformer including a primary winding, a secondary winding, and an auxiliary winding, a switching element connected in series to the primary winding, and rectifying and smoothing a voltage between terminals of the secondary winding. An output voltage generation circuit that generates an output voltage, an auxiliary power supply circuit that rectifies and smoothes a voltage across terminals of an auxiliary winding to generate an auxiliary power supply voltage, and is driven by the auxiliary power supply voltage to turn on / off switching elements SUMMARY A switching power supply device is disclosed that includes a power control IC (Integrated Circuit). During overload, the output voltage and the auxiliary power supply voltage decrease. When the auxiliary power supply voltage falls below the set value, the overcurrent protection reference voltage is lowered in the power supply control IC according to the decrease. As a result, the output current is reduced as the output voltage decreases, and highly safe output characteristics are realized. Such an output characteristic is called “F” character because it is in the form of “F” when graphed (see FIGS. 3A and 3B).

  Further, as a conventional switching power supply device, there is one that converts an output current into a voltage using a resistance element and determines whether or not it is in an overcurrent state based on the voltage.

Unexamined-Japanese-Patent No. 2006-14573

  However, Patent Document 1 requires a dedicated power supply control IC having a function of reducing the overcurrent protection reference voltage according to the reduction of the auxiliary power supply voltage, and the cost is higher than when a general purpose power supply control IC is used. There was a problem of becoming

  Further, in the conventional switching device described above, when the ambient temperature changes, the resistance value of the resistance element changes, and the voltage between the terminals of the resistance element changes. There was a problem that it could not be determined correctly. If it continues to supply power without being able to determine that an overcurrent condition has occurred, the electronic components may overheat and be damaged.

  SUMMARY OF THE INVENTION Therefore, the main object of the present invention is to provide a highly safe and low cost power conversion circuit and a switching power supply using the same.

  A power conversion circuit according to the present invention includes first and second input terminals receiving a first DC voltage, an isolation transformer including a primary winding and a secondary winding, and on / off in response to a PWM signal. And an output voltage generation circuit for rectifying and smoothing a voltage between terminals of a secondary winding to generate a second DC voltage and applying it to a load, and a primary winding between the first and second input terminals. A first resistance element connected in series with the line and the switching element, and when the voltage across the first resistance element is lower than the reference voltage, the overcurrent detection signal is brought to the inactive level, and the first resistance element is An overcurrent detection circuit that brings the overcurrent detection signal to an activation level when the inter-terminal voltage is higher than the reference voltage, and brings the output control signal to a first logic level when the overcurrent detection signal is an activation level, Whether the overcurrent detection signal is at the activation level The timer circuit which sets the output control signal to the second logic level after a lapse of the first time since the change to the inactivation level, and the frequency of the PWM signal when the output control signal is the second logic level. The frequency setting circuit sets the first value and sets the frequency of the PWM signal to a second value lower than the first value when the output control signal is at the first logic level, and the frequency setting circuit Control the pulse width of the PWM signal such that the second DC voltage becomes the target voltage when the output control signal is at the second logic level, and the output control signal And a control circuit which narrows the pulse width of the PWM signal in response to the logic level of 1. The resistance value of the first resistance element changes in accordance with the ambient temperature of the power conversion circuit. The overcurrent detection circuit changes the reference voltage in response to the change in the ambient temperature, and compensates for the change in the resistance value of the first resistance element with the change in the ambient temperature.

  In the power conversion circuit according to the present invention, an overcurrent detection circuit which brings an overcurrent detection signal to an activation level when a voltage across terminals of a first resistance element connected in series with a primary winding exceeds a reference voltage; A timer circuit for generating an output control signal based on the overcurrent detection signal; and a control circuit for reducing the frequency and pulse width of the PWM signal when the output control signal is at the first logic level. The current detection circuit changes the reference voltage in accordance with the ambient temperature to compensate for the change in the resistance value of the first resistance element with the change in the ambient temperature. Therefore, it is possible to realize a low temperature dependent character curve characteristic with a simple configuration, and to realize a highly safe and low cost power conversion circuit.

It is a circuit block diagram showing composition of a switching power supply which forms the basis of the present invention. It is a figure which shows the character characteristic of a switching power supply device. It is a figure which shows the character of a switching power supply device. FIG. 1 is a circuit block diagram showing a configuration of a switching power supply device according to Embodiment 1 of the present invention. FIG. 5 is a circuit block diagram showing configurations of an overcurrent detection circuit and a voltage holding circuit shown in FIG. 4. It is a circuit block diagram which shows the structure of the frequency setting circuit shown in FIG. 5 is a time chart showing operations of the frequency setting circuit and the power supply control IC shown in FIG. 4; It is a circuit block diagram which shows the principal part of the switching power supply device by Embodiment 2 of this invention. It is a time chart which shows operation | movement of the blanking circuit shown in FIG. It is a circuit block diagram which shows the principal part of the switching power supply device by Embodiment 3 of this invention.

  In order to facilitate understanding of the present invention, a switching power supply as a basis of the present invention will be described first before describing the embodiments. FIG. 1 is a circuit block diagram showing a configuration of a switching power supply 1 as a basis of the present invention. In FIG. 1, the switching power supply device 1 includes a rectifier circuit 2, a capacitor 3, and a power conversion circuit 4.

  The rectifier circuit 2 rectifies an AC voltage of a commercial frequency supplied from the commercial AC power supply 5 and converts it into a DC voltage (pulse voltage). The capacitor 3 smoothes the DC voltage (pulsating current voltage) generated by the rectifier circuit 2. The negative terminals of rectifier circuit 2 and capacitor 3 receive ground voltage GND. The power conversion circuit 4 converts the DC voltage (first DC voltage) VI smoothed by the capacitor 3 into a desired DC voltage (second DC voltage) VO and applies the voltage to the load 6.

  Thus, the switching power supply 1 rectifies and smoothes the AC voltage supplied from the commercial AC power supply 5 to convert it into the DC voltage VI, converts the DC voltage VI into the desired DC voltage VO, and converts it into the load 6. Supply. Assuming that the load current is IO, the output of the switching power supply 1 is VO × IO.

  In order to supply desired power (maximum design power) from commercial AC power supply 5 to load 6, rectifier circuit 2, capacitor 3 and power conversion circuit 4 are required, and DC voltage VI is received from DC power supply. When the desired power (maximum design power) is supplied to the load 6, only the power conversion circuit 4 is a necessary component. Therefore, the designed maximum power value of switching power supply 1 and the designed maximum power value of power conversion circuit 4 are the same value.

  Here, the maximum current value that can be output by the switching power supply device 1 is set as the rated current value IR. For example, when the design maximum power of power conversion circuit 4 and switching power supply device 1 is 50 W and the rated voltage value is 5 V, rated current value IR is 10 A.

  A state in which the output current IO exceeds the rated current value IR is referred to as an overcurrent state. That is, the overcurrent state is a state in which the switching power supply 1 outputs a current exceeding the rated current value IR, in other words, a state in which the load 6 tries to flow the output current IO larger than the rated current value IR. It is. For example, when the rated current value IR is 10 A, a state where an output current IO (= 10.5 A) of 105% of the rated current value IR is to be supplied, an output current IO of 130% of the rated current value IR (= A state where it is intended to flow 130 A), or a state where it is intended to flow an output current IO (= 20 A) of 200% of the rated current value IR is an overcurrent state.

  The load short-circuited state is a state in which a circuit, wiring, load 6 and the like after the output terminal of the switching power supply 1 are shorted (electrically shorted). In this case, since the impedance on the load 6 side is indeterminate, it is not known how many times the current IO flows as the rated current value IR.

  In the overcurrent state, the power supply of the switching power supply device 1 is restricted, and the output voltage VO falls below the rated voltage value (= 5 V). This is called a drop in the output voltage VO. The fact that the output voltage VO becomes an overvoltage, that is, the output voltage VO becomes a voltage (6 V, 20 V, etc.) larger than the rated voltage value (= 5 V) can not normally occur unless the switching power supply 1 fails. So I will omit the explanation.

  Such an output current IO-output voltage VO characteristic of the switching power supply device 1 has a so-called letter characteristic. In the figure, the output voltage VO gradually decreases with an increase in the output current IO when the output current IO exceeds the rated current value IR, and is also called a constant power control voltage drooping characteristic. When such output characteristics are graphed, they are in the shape of "字", so they are called "特性" characteristics. For example, in the switching power supply 1 having a maximum power of 50 W, a rated voltage value of 5 V and a rated current value of 10 A, when the output current IO is gradually increased, VO / IO is 5 V / 10 A, 4. It changes to 5V / 14.4A, 3.5V / 18.6A, and so on, and the output voltage VO gradually decreases with the increase of the output current IO.

  FIGS. 2 (a) and 2 (b) are diagrams showing the letter-like characteristic. In FIG. 2A, when the output current IO exceeds the rated current value IR, the case where the output voltage VO decreases as the output current IO increases is shown. In an actual flyback converter power supply, as shown in FIG. 2B, when the output current IO exceeds 130% of the rated current value IR, the output voltage VO decreases as the output current IO increases.

  The characteristic is generated because the switching power supply 1 can not take out an output power larger than the design maximum power, and is not a characteristic that can be accurately controlled, but is due to the mechanism of the flyback converter power supply . In the switching power supply 1 having the following characteristic, when the output current IO is in the overcurrent state exceeding the rated current value IR, the electronic components may be overheated and damaged.

  As a method of protecting the electronic component in the case of an overcurrent state, there is a method of providing the switching power supply device 1 with a drooping characteristic. The drooping characteristic is a characteristic that sharply reduces the output voltage VO when the output current IO exceeds the upper limit value IH which is equal to or higher than the rated current value IR, and is also called a constant current control voltage drooping characteristic. If the switching power supply unit 1 has a drooping characteristic, the output voltage VO can be rapidly reduced when the output current IO exceeds the upper limit value IH, and the electronic component may be overheated and damaged in the overcurrent state. It can be prevented.

  As another method of protecting the electronic component in the case of an overcurrent state, there is a method of providing the switching power supply device 1 with a fold-back characteristic. The curve characteristic is a characteristic that reduces both the output voltage VO and the output current IO when the output current IO exceeds the upper limit value IH which is equal to or higher than the rated current value IR, and is also called a foldback characteristic. When such output characteristics are graphed, they are in the form of "f", so they are called f-characters.

  In the drooping characteristic, when the output current IO exceeds the upper limit value IH, only the output voltage VO is reduced, while in the curve characteristic, when the output current IO exceeds the upper limit value IH, both the output voltage VO and the output current IO Reduce Therefore, the effect of preventing overheating and breakage of the electronic component in the overcurrent state is larger in the fold characteristic than in the sag characteristic.

  For example, if output limitation is applied when output current IO exceeds upper limit value IH (= 1.3 × IR) of 130% of rated current value IR, when output current IO is gradually increased, VO / IO VO is maintained at 5 V until it reaches 5.0 V / 13 A, and thereafter, the output power itself decreases sharply to 4.8 V / 12 A, 4.5 V / 10 A, 3.5 V / 7 A, and so on. This makes it possible to prevent abnormal overheating and damage of the electronic components that make up the switching power supply 1 even in an overcurrent state and a load short state where a current larger than the overcurrent state may flow.

  FIGS. 3 (a) and 3 (b) are diagrams showing letter character characteristics. In FIG. 3A, when the output current IO exceeds the upper limit value IH (= 1.3 × IR) of 130% of the rated current value IR, the output is limited, and both the output voltage VO and the output current IO decrease. The case is shown. In FIG. 3B, when the output current IO exceeds the upper limit value IH (= 1.05 × IR) of 105% of the rated current value IR, the output is limited, and both the output voltage VO and the output current IO decrease. The case is shown. As a result, it is possible to prevent abnormal overheating and damage of the electronic components that make up the switching power supply device 1.

  In order to realize such a curve characteristic, it is necessary to provide the switching power supply 1 with an additional circuit. In general, when power conversion circuit 4 includes an isolation transformer, a resistive element for converting output current IO into a voltage is provided on the secondary winding side of the isolation transformer, and the voltage is transmitted to the control circuit on the primary winding side. It is necessary to have a photocoupler, etc. In this prior art, there is a problem that the number of electronic parts is increased, resulting in an increase in size and cost of the apparatus.

  For example, in a switching power supply for vehicles, it is required to operate normally in a wide temperature range from -40 ° C to + 85 ° C. However, when the ambient temperature changes, the resistance value of the resistance element changes, and the voltage between the terminals of the resistance element changes. Therefore, under an environment where the ambient temperature changes significantly, it is not possible to accurately determine whether the overcurrent state is present. is there. If it is not possible to accurately determine whether or not it is in the over current state, although the over current state is already in progress, the power limit is not performed and the excessive power is output, or the over current state is in progress. It will not be possible to output the maximum power. For this reason, it is necessary to allow a large margin in the output characteristics, resulting in an increase in size and cost of the apparatus.

  Further, Patent Document 1 discloses an auxiliary power supply circuit that rectifies and smoothes a voltage across terminals of an auxiliary winding of an isolation transformer to generate an auxiliary power supply voltage, and a power supply driven by the auxiliary power supply voltage to turn on / off switching elements. A switching power supply comprising a control IC is disclosed. When the auxiliary power supply voltage falls below the set value in the overcurrent state, the overcurrent protection reference voltage in the power supply control IC is lowered according to the decrease, and the curve characteristic is realized.

  However, in Patent Document 1, it is necessary to use a dedicated power supply control IC having a function of reducing the overcurrent protection reference voltage according to the decrease of the auxiliary power supply voltage, compared to the case where a general purpose power supply control IC is used. There is a problem that the cost is high. Further, the upper limit value IH of the output current IO is set in the power supply control IC, and the upper limit value IH can not be freely changed. In addition, the degree of reduction of the auxiliary power supply voltage in the overcurrent state is affected by the degree of coupling between the primary winding and the auxiliary winding of the transformer, and the character variation varies depending on the device. The present invention aims to solve these problems.

First Embodiment
FIG. 4 is a circuit block diagram showing a configuration of switching power supply device 10 according to the first embodiment of the present invention, which is to be compared with FIG. The switching power supply 10 has an isolated flyback converter power supply with a loop characteristic. The switching power supply 10 differs from the switching power supply 1 in that the power conversion circuit 4 is replaced by a power conversion circuit 11.

  Power conversion circuit 11 includes input terminals T1 and T2, output terminals T3 and T4, isolation transformer 12, N channel MOS transistor 13 (switching element), resistance element 14, diode 15, capacitor C1, feedback circuit (F / B) 16 , An overcurrent detection circuit 17, a voltage holding circuit 18, a frequency setting circuit 19, and a power control IC 20.

  The input terminal T1 is connected to the positive terminal of the capacitor 3, the input terminal T2 is connected to the negative terminal (line of the ground voltage GND) of the capacitor 3, and the DC voltage VI is applied between the input terminals T1 and T2. A DC voltage VO is output between the output terminals T3 and T4, and a load 6 is connected.

  Isolation transformer 12 includes a primary winding 12a and a secondary winding 12b. The primary winding 12a and the secondary winding 12b are magnetically coupled but are electrically isolated. The isolation transformer 12 electrically isolates the input terminals T1 and T2 from the output terminals T3 and T4. One terminal (a terminal with a black circle) of primary winding 12a is connected to input terminal T2 through N channel MOS transistor 13 and resistance element 14, and the other terminal of primary winding 12a is input terminal T1. Connected to

  The gate of N channel MOS transistor 13 receives a PWM (pulse width modulation) signal φ 20 from power supply control IC 20. When the PWM signal φ20 is set to the “H” level, the N channel MOS transistor 13 is turned on, and when the PWM signal φ20 is set to the “L” level, the N channel MOS transistor 13 is turned off.

  A voltage V14 of a value corresponding to the current I1 flowing through the primary winding 12a and the N channel MOS transistor 13 is generated between the terminals of the resistance element 14. Assuming that the resistance value of the resistance element 14 is R14, V14 = I1 × R14.

  One terminal (the terminal with the black circle) of the secondary winding 12b is connected to the anode of the diode 15, and the cathode of the diode 15 is connected to the output terminal T3. The diode 15 rectifies the voltage between the terminals of the secondary winding 12b and converts it to a direct current (pulse current) voltage. The capacitor C1 is connected between the output terminals T3 and T4 to smooth the output voltage VO. The diode 15 and the capacitor C1 constitute an output voltage generation circuit.

  Feedback circuit 16 detects output voltage VO, and generates control signal φ16 for controlling the on time of N channel MOS transistor 13 such that the detected value matches target voltage VT (5 V). The power supply control IC 20 is supplied with φ16 via the built-in photo coupler.

  Feedback circuit 16 provides control signal φ 16 to power supply control IC 20 to lengthen the on time of N channel MOS transistor 13 when output voltage VO is lower than target voltage VT, and output voltage VO is higher than target voltage VT. When it is high, the on time of the N channel MOS transistor 13 is shortened. The control signal φ 16 is transmitted to the power control IC 20 through the photocoupler, so the circuit on the primary winding 12 a side and the circuit on the secondary winding 12 b side are electrically isolated.

  The overcurrent detection circuit 17 compares the terminal voltage V14 of the resistance element 14 with the reference voltage VR. If the terminal voltage V14 of the resistance element 14 is lower than the reference voltage VR (V14 <VR), the overcurrent detection signal is detected. When OC is set to the inactive level "L" level and the voltage V14 across the terminals of resistance element 14 exceeds the reference voltage VR (V14> VR), the overcurrent detection signal OC is set to the active level "H" level. Do. Overcurrent detection circuit 17 includes a linear temperature coefficient resistor for changing reference voltage VR in accordance with the ambient temperature of switching power supply device 10 to compensate for the change in the resistance value of resistance element 14 accompanying the change in ambient temperature. . This will be described later.

  The resistance value of the resistance element 14 and the reference voltage VR are set so that the terminal voltage V14 of the resistance element 14 reaches the reference voltage VR when the output current IO of the switching power supply 10 reaches the upper limit value IH. There is. Overcurrent detection signal OC is applied to voltage holding circuit 18.

  Voltage holding circuit 18 sets output control signal CNT to "H" level when overcurrent detection signal OC is raised from the "L" level of the inactivation level to the "H" level of the activation level, at least in advance. The set time Tc maintains the output control signal CNT at "H" level. The time Tc is set to a time sufficiently longer than one cycle of the PWM signal φ20.

  When output control signal CNT is at "H" level and overcurrent detection signal OC falls from "H" level at activation level to "L" level at deactivation level, from that time on, it is made in advance. After a predetermined time Tc, output control signal CNT falls to "L" level. When the output control signal CNT is at the “H” level and the overcurrent detection signal OC changes a plurality of times between the “H” level and the “L” level, the overcurrent detection signal OC is finally changed to “L The output control signal CNT falls to "L" level after lapse of a predetermined time Tc from being set to the "level". Output control signal CNT is applied to frequency setting circuit 19 and power supply control IC 20.

  Frequency setting circuit 19 is connected to power supply control IC 20 and sets frequency f of PWM signal φ 20 to a first value f 1 when output control signal CNT is at “L” level, and output control signal CNT is “H”. If it is a level, the frequency f of the PWM signal φ 20 is set to a second value f 2 lower than the first value f 1. For example, f1 is 100 kHz and f2 is 25 kHz. The frequency setting circuit 19 includes a variable capacitance capacitor for setting the frequencies f1 and f2, and a linear temperature coefficient resistor for compensating for a change in capacitance value of the variable capacitance capacitor due to a change in ambient temperature. This will be described later.

  The power supply control IC 20 is a general-purpose power supply control IC, and generates a PWM signal φ 20 of the frequency set by the frequency setting circuit 19. When output control signal CNT is at the “L” level, power supply control IC 20 generates PWM signal φ 20 having frequency f of f 1 and increases the pulse width of PWM signal φ 20 according to control signal φ 16 from feedback circuit 16. Reduce. The time for which the PWM signal φ20 is set to the “H” level in one cycle (1 / f) of the PWM signal φ20 is a pulse width. The ratio of the pulse width of the PWM signal φ20 to one cycle of the PWM signal φ20 is called a duty ratio.

  Further, when the output control signal CNT is set to the “H” level, the power supply control IC 20 generates the PWM signal φ20 with the frequency f of f2 and narrows the pulse width of the PWM signal φ20. Therefore, when output control signal CNT is set to “H” level, the frequency of PWM signal φ 20 is reduced and the pulse width is shortened as compared with the case where output control signal CNT is “L” level. As a result, the number and time required for N channel MOS transistor 13 to be turned on per unit time are reduced, and both output voltage VO and output current IO are reduced to realize the letter-like characteristic.

  FIG. 5 is a circuit block diagram showing configurations of the overcurrent detection circuit 17 and the voltage holding circuit 18. In FIG. 5, the overcurrent detection circuit 17 is an external circuit of the power control IC 20 and includes resistance elements 21 and 22 and a comparator 23. Resistive element 21 includes a linear temperature coefficient resistor for changing reference voltage VR in response to a change in ambient temperature to compensate for the change in resistance of resistive element 14 caused by the change in ambient temperature. Resistance elements 21 and 22 are connected in series between the line of DC power supply voltage VCC and the line of ground voltage GND, and divide DC power supply voltage VCC to generate reference voltage VR.

  The reference voltage VR can be set to a desired value by using the resistive element 21 having a desired resistance value and the resistive element 22 having a desired resistance value. At least one of the resistive elements 21 and 22 may be a variable resistive element. Assuming that the resistance value of the resistance element 21 is R21 and the resistance value of the resistance element 22 is R22, VR = VCC × R22 / (R21 + R22). R22 / (R21 + R22) is a partial pressure ratio.

  The comparator 23 compares the terminal voltage V14 of the resistance element 14 with the reference voltage VR, and when the terminal voltage V14 of the resistance element 14 is lower than the reference voltage VR (V14 <VR), the overcurrent detection signal OC When the voltage V14 across the terminals of the resistance element 14 exceeds the reference voltage VR (V14> VR), the overcurrent detection signal OC is activated to H level. Level (DC power supply voltage VCC).

  When the output current IO of the switching power supply device 10 reaches the upper limit value IH, the current I1 flowing through the primary winding 12a of the isolation transformer 12 reaches the upper limit value, and the inter-terminal voltage V14 of the resistance element 14 becomes the reference voltage VR. The resistance values of the resistance elements 14, 21 and 22 are set so as to reach the above. Further, the ratio of the upper limit value IH of the output current IO to the rated current value IR can be freely set to 101%, 105%, 130% or the like by changing the resistance value of the resistance elements 14, 21 and 22. .

  The DC power supply voltage VCC may be a DC voltage supplied from another power supply circuit, or a DC voltage obtained by rectifying and smoothing the voltage across the auxiliary winding (not shown) of the isolation transformer 12. It does not matter. When the DC power supply voltage VCC is generated from the voltage between the terminals of the auxiliary winding of the isolation transformer 12, the DC power supply voltage VCC may fluctuate depending on the structure of the isolation transformer 12. That is, when the degree of coupling of the isolation transformer 12 is high, that is, the coupling between the windings is good, the DC power supply voltage VCC is lowered as the output voltage VO is lowered due to an overcurrent or a load short. If the degree of coupling of the isolation transformer 12 is low, that is, if the coupling between the windings is bad, the DC power supply voltage VCC does not decrease even if the output voltage VO decreases due to an overcurrent or a load short.

  In the patent document 1, since the auxiliary winding voltage is lowered in the overcurrent state, an insulation transformer having a high degree of coupling, that is, a good coupling between the windings is used. On the other hand, the first embodiment is based on the premise that the DC power supply voltage VCC does not decrease even when the output voltage VO decreases, so that the isolation transformer 12 has a low degree of coupling, that is, poor coupling between the windings. used. The DC power supply voltage VCC is also used as a power supply voltage of the power supply control IC 20.

  Further, in the overcurrent detection circuit 17 of FIG. 5, the DC power supply voltage VCC is divided by the resistance elements 21 and 22 to generate the reference voltage VR. However, the present invention is not limited to this. The reference voltage VR may be generated using a semiconductor element or circuit that can generate a constant voltage, such as a shunt regulator or a zener diode.

  Voltage holding circuit 18 includes a diode D1 and a capacitor C2. The anode of diode D1 receives overcurrent detection signal OC, and the cathode is connected to output node 18a of voltage holding circuit 18. Capacitor C2 is connected between the cathode of diode D1 and the line of ground voltage GND. The signal appearing at output node 18a is output control signal CNT.

  When overcurrent detection signal OC is at the inactive level "L" level (ground voltage GND), capacitor C2 is not charged, and output control signal CNT is set to "L" level (ground voltage GND). When overcurrent detection signal OC is raised from the inactive level "L" level to the active level "H" level (DC power supply voltage VCC), the output terminal of comparator 23 receives capacitor C2 via diode D1. The current flows through the capacitor C2, the capacitor C2 is charged to the DC power supply voltage VCC, and the output control signal CNT is set to the “H” level (DC power supply voltage VCC).

  Even when the overcurrent detection signal OC falls to the inactive level "L" level (ground voltage GND) with the capacitor C2 charged to the DC power supply voltage VCC, comparison is made from the capacitor C2 through the diode D1 No current flows to the output terminal of the unit 23, and the output control signal CNT is maintained at "H" level (DC power supply voltage VCC). However, current flows from the capacitor C2 to the frequency setting circuit 19, and the output control signal CNT becomes “L” level after a certain time Tc. This will be described later.

Here, a method of compensating for the change of the resistance value of the resistance element 14 accompanying the change of the ambient temperature will be described. Consider the case where the switching power supply 10 is operated in a wide temperature range, for example, in the range of -40 ° C to + 85 ° C. Generally, electronic components have temperature characteristics. For example, the resistance value of the resistance element 14 may decrease at a low temperature (-40 ° C.). In general, the magnitude of temperature change of a resistive element is expressed using a temperature coefficient of resistance (ppm / ° C. = 10 ⁻⁶ / ° C.).

  The resistance temperature coefficient of the resistance element 14 is α (ppm / ° C.), the resistance value of the resistance element 14 at + 20 ° C. is R (+ 20 ° C.), and the resistance value of the resistance element 14 at x ° C. is R (x C), R (x.degree. C.) = | 20-x | .times..alpha. + R (20.degree. C.). For example, if resistance temperature coefficient α of resistance element 14 is −300 ppm / ° C. and resistance value R (20 ° C.) of resistance element 14 is 1.0 Ω when + 20 ° C., the resistance at −40 ° C. The resistance value R (−40 ° C.) of the element 14 is 0.982 Ω. Assuming that the current I1 flowing through the resistance element 14 is 1 A, the voltage V14 across the terminals of the resistance element 14 when the ambient temperature is −40 ° C. is 1 A × 0.982 Ω = 0.982.

  Note that, in a normal resistance element (except for linear temperature coefficient resistors), the temperature coefficient of resistance α is not a specific constant value, but includes an error, and is described, for example, within ± 100 (ppm / ° C.). Therefore, it is unclear how the resistance value of the resistance element changes according to the change of ambient temperature, that is, whether it decreases at low temperature, decreases at high temperature, rises at low temperature, or rises at high temperature is there.

  Next, the operation of the overcurrent detection circuit 17 with respect to changes in ambient temperature will be described. For example, the DC power supply voltage VCC is 12 V, the resistance value of the resistance element 21 is 11 kΩ, and the resistance value of the resistance element 22 is 1 kΩ. When the ambient temperature is a normal temperature (+ 20 ° C.), the DC power supply voltage VCC is divided by the resistance elements 21 and 22 and the reference voltage VR is 1.0V.

  If the reference voltage VR remains at 1.0 V when the ambient temperature reaches -40 ° C., the current I1 flowing through the resistance element 14 does not reach 1.0 V / 0.982 Ω = 1.0183 A. The circuit 23 will not bring the overcurrent detection signal OC to the activation level "H". Therefore, assuming that the upper limit value IH of the output current IO is 100%, the overcurrent detection signal OC is not set to the “H” level when the output current IO of about 102% does not flow, and the overcurrent protection operation is not performed. A problem arises.

  Therefore, in the first embodiment, for example, a linear temperature coefficient resistor whose resistance temperature coefficient is defined in advance is used as resistance element 21. Assuming that the resistance temperature coefficient α of the resistance element 21 (that is, linear temperature coefficient resistance) is +330 ppm / ° C., and the resistance value of the resistance element 21 is 11 kΩ = 11000 Ω when the ambient temperature is 20 ° C., the ambient temperature is − The resistance value of the resistive element 21 at a temperature of 40 ° C. rises to 11,218 Ω.

  Assuming that the resistance value of the resistive element 22 remains 1 kΩ when the ambient temperature reaches −40 ° C., a voltage division is performed between the resistive element 21 having a DC power supply voltage VCC of 1218 Ω and the resistive element 22 having 1 kΩ. And the reference voltage VR is set to 0.982V. Therefore, even if the ambient temperature becomes low (−40 ° C.) and the resistance value of resistance element 14 decreases, the reference voltage VR can be lowered accordingly, so the output exceeds the design maximum power. Sometimes, the overcurrent detection signal OC can be brought to the "H" level of the activation level.

  In the first embodiment, although the case where the resistance value of resistance element 14 decreases at low temperature has been described, the present invention is not limited to this. When increasing at low temperature, decreasing at high temperature, increasing at high temperature In any case, by using a linear temperature coefficient resistor having a temperature coefficient of resistance corresponding to each case as the resistance element 21, the change of the resistance value of the resistance element 14 accompanying the change of the ambient temperature is compensated. Can. Further, although the case where only the resistive element 21 of the resistive elements 21 and 22 includes the linear resistive temperature coefficient resistor has been described, the present invention is not limited to this. Even if only the resistive element 22 includes the linear resistive temperature coefficient resistor Alternatively, both of the resistance elements 21 and 22 may include linear resistance temperature coefficient resistors.

  FIG. 6 is a circuit block diagram showing a configuration of frequency setting circuit 19. Referring to FIG. 6, frequency setting circuit 19 is an external circuit of power supply control IC 20, and includes resistance elements 24 and 25, capacitors 26 and 27, and NPN bipolar transistor 28. Resistive element 25 includes a linear temperature coefficient resistor for compensating for the change in capacitance value of each of capacitors 26 and 27 due to the change in ambient temperature. The resistance element 25 is connected between the resistance terminal RT and the capacitance terminal CT of the power supply control IC 20. Capacitor 26 is connected between capacitance terminal CT and the line of ground voltage GND. One terminal of the capacitor 27 is connected to the capacitance terminal CT.

  The collector of NPN bipolar transistor 28 is connected to the other terminal of capacitor 27, and the emitter is connected to the line of ground voltage GND. One terminal of resistance element 24 receives output control signal CNT, and the other terminal is connected to the base of NPN bipolar transistor 28. Resistor element 24 limits the base current of NPN bipolar transistor 24 and also limits the current flowing out of capacitor C 2 of voltage holding circuit 18. Resistance element 25 constitutes a current source. Capacitors 26, 27 and NPN bipolar transistor 28 constitute a variable capacitance capacitor.

  When output control signal CNT is at "L" level, NPN bipolar transistor 28 is turned off, and only capacitor 26 is connected between capacitance terminal CT and the line of ground voltage GND. The power supply control IC 20 charges the capacitor 26 through the resistance element 25. The voltage across the terminals of the capacitor 26 rises gradually. In response to the voltage across the terminals of the capacitor 26 exceeding the threshold voltage VTH, the power supply control IC 20 instantly discharges the charge of the capacitor 26 and immediately starts charging the capacitor 26. At this time, the frequency at which the charge of the capacitor 26 is charged and discharged is the frequency f1 of the PWM signal φ20.

  When output control signal CNT is at "H" level, NPN bipolar transistor 28 is turned on, and capacitors 26, 27 are connected in parallel between capacitance terminal CT and the line of ground voltage GND. The power supply control IC 20 charges the capacitors 26 and 27 via the resistance element 25. The voltage across the terminals of the capacitors 26, 27 gradually rises. The power supply control IC 20 instantly discharges the charges of the capacitors 26 and 27 in response to the voltage across the terminals of the capacitors 26 and 27 exceeding the threshold voltage VTH, and immediately starts charging the capacitors 26 and 27. . At this time, the frequency for charging and discharging the charges of the capacitors 26 and 27 is the frequency f2 of the PWM signal φ20.

  The larger the capacitance value of the capacitor between the capacitance terminal CT and the line of the ground voltage GND, the longer the time required to charge the capacitor and the lower the frequency of the PWM signal φ20. Therefore, f1> f2. Assuming that the capacitance value of the capacitor 26 is C26 and the capacitance value of the capacitor 27 is C27, f1 / f2 = (C26 + C27) / C26. By replacing the capacitor 26 with a first capacitor having an arbitrary capacitance value and replacing the capacitor 27 with a second capacitor having an arbitrary capacitance value, f1 / f2 can be set to an arbitrary value. It has become. In other words, the first capacitance value C26 of the variable capacitance capacitor when the output control signal CNT is at the "L" level, and the second capacitance value of the variable capacitance capacitor when the output control signal CNT is at the "H" level Each of (C26 + C27) is changeable. Each of the capacitors 26 and 27 may be a variable capacitance capacitor.

  When overcurrent detection signal OC falls from "H" level to "L" level, the charge of capacitor C2 of voltage holding circuit 18 is supplied to ground voltage GND through resistance element 24 and the base and emitter of NPN bipolar transistor 28. The voltage of the output control signal CNT gradually decreases. Therefore, when the overcurrent detection signal OC falls from the “H” level to the “L” level, the output control signal CNT falls from the “H” level to the “L” level after a certain time Tc. The capacitance value of the capacitor C2 and the resistance value of the resistive element 24 are set such that the time Tc is a desired time. Voltage holding circuit 18, resistance element 24, and NPN bipolar transistor 28 constitute a timer circuit.

  FIGS. 7A to 7D are time charts showing the operation of the power control IC 20. FIG. In particular, FIG. 7 (a) shows the voltage of the capacitive terminal CT when the output control signal CNT is at the “L” level, and FIG. 7 (b) shows the waveform of the PWM signal φ20 in that case. FIG. 7 (c) shows the voltage of the capacitance terminal CT when the output control signal CNT is at the “H” level, and FIG. 7 (d) shows the waveform of the PWM signal φ20 in that case.

  When the output control signal CNT is at the "L" level, as shown in FIGS. 7A and 7B, the NPN bipolar transistor 28 is turned off, and the capacitor 26 is charged / discharged at a relatively high frequency f1. The frequency of φ20 is f1. At this time, the time during which PWM signal φ20 is set to the “H” level within one cycle λ of PWM signal φ20, that is, the pulse width TP is controlled such that output voltage VO of switching power supply 10 matches target voltage VT. Ru. FIG. 7B shows a state in which the pulse width TP of the PWM signal φ20 is maintained at a constant value.

  When the output control signal CNT is at the “H” level, as shown in FIGS. 7C and 7D, the NPN bipolar transistor 28 is turned on, and the capacitors 26 and 27 are charged and discharged at a relatively low frequency f2, The frequency of the PWM signal φ20 is f2. FIG. 7C shows the case where f2 = f1 / 4. At this time, pulse width TP of PWM signal φ 20 is greater than pulse width TP in a period immediately before output control signal CNT is set to the “H” level (ie, a period during which output control signal CNT is set to the “L” level). It is fixed for a short time. FIG. 7D shows a state in which the pulse width TP of the PWM signal φ20 is shortened to about 1⁄3.

  Therefore, when output control signal CNT is set to "H" level, the on time and on number of times of N channel MOS transistor 13 per unit time are reduced, and it is 1 compared to the design maximum power that can be originally output. The power handled by the secondary circuit is reduced, and the power transmitted to the secondary side is also reduced. As a result, both of the output voltage VO and the output current IO are reduced, and a fold-back characteristic is realized.

  In FIGS. 7A to 7D, the case of f2 / f1 = 1/4 has been described. However, the present invention is not limited to this, and if f2 / f1 is a value larger than 0 and smaller than 1 Any value may be used. For example, f2 / f1 = 1/5 or f2 / f1 = 1/10 may be used.

  Next, the operation of the frequency setting circuit 19 with respect to changes in ambient temperature will be described. Generally, when the ambient temperature changes, the capacitance value of the capacitor changes. For example, assuming that the capacitance value of each of capacitors 26 and 27 at normal temperature (+ 20 ° C.) is 100%, the capacitance value of each of capacitors 26 and 27 at low temperature (-40 ° C.) decreases to 80%. Think. Assuming that the resistance value of resistance element 25 is constant regardless of the ambient temperature, and the current flowing to capacitors 26 and 27 through resistance element 25 is constant regardless of the ambient temperature, the temperature is low (-40 ° C.) , The speed at which the capacitors 26, 27 are charged becomes faster, and the frequency of the PWM signal .phi. 20 deviates from 100 kHz (or 25 kHz).

  Therefore, in the first embodiment, when the resistance value at normal temperature (+ 20 ° C.) is 100%, the temperature coefficient such that the resistance value at low temperature (-40 ° C.) increases to 100/80 = 125%. A linear temperature coefficient resistor is used as the resistive element 25. Thereby, it is possible to compensate for the change in capacitance value of capacitors 25 and 27 caused by the change in ambient temperature, and to suppress the change in frequency of PWM signal φ20 caused by the change in ambient temperature.

  Next, the operation of the switching power supply device 10 will be described. In FIG. 4, an AC voltage of a commercial frequency supplied from the commercial AC power supply 5 is rectified by the rectifier circuit 2, smoothed by the capacitor 3, and converted into a DC voltage VI. DC voltage VI is applied between input terminals T1 and T2 of power conversion circuit 11.

  When N channel MOS transistor 13 is turned on, current flows from the positive terminal of capacitor 3 through the primary winding 12a of isolation transformer 12, N channel MOS transistor 13, and resistance element 14 in the negative terminal of capacitor 3. I1 flows and electromagnetic energy is stored in the insulating transformer 12. The current I1 is converted to a voltage V14 by the resistance element 14, and the voltage V14 is monitored by the overcurrent detection circuit 17.

  When N channel MOS transistor 13 is turned off, the electromagnetic energy stored in isolation transformer 12 is released, and a current is supplied from the secondary winding 12b to the parallel connection of capacitor C1 and load 6 through diode 15 . In other words, the voltage across the terminals of the secondary winding 12 b is rectified by the diode 15, smoothed by the capacitor C 1, converted to the DC voltage VO, and supplied to the load 6.

  Control signal φ 16 is generated by feedback circuit 16 so that output voltage VO matches target voltage VT, and power control IC 20 controls pulse width TP of PWM signal φ 20 based on control signal φ 16. When the PWM signal φ20 is at the “H” level, the N-channel MOS transistor 13 is turned on, and when the PWM signal φ20 is at the “L” level, the N-channel MOS transistor 13 is turned off.

  Current I1 flowing through primary winding 12a is converted to voltage V14 by resistance element 14. When output current IO is smaller than upper limit value IH, inter-terminal voltage V14 of resistance element 14 becomes smaller than reference voltage VR, and overcurrent detection circuit 17 causes overcurrent detection signal OC to be at the inactive level "L". The voltage holding circuit 18 brings the output control signal CNT to "L" level. When output control signal CNT is at "L" level, only capacitor 26 of frequency setting circuit 19 is connected to capacitance terminal CT of power supply control IC 20, and power supply control IC 20 generates PWM signal φ20 of constant frequency f1.

  When VO <VT, the pulse width TP of the PWM signal φ20 is expanded, and when VO> VT, the pulse width TP of the PWM signal φ20 is narrowed. In other words, when VO <VT, the duty ratio of the PWM signal φ20 is increased, and when VO> VT, the duty ratio of the PWM signal φ20 is decreased. Thus, the current IO equal to or less than the rated current value IR can be supplied to the load 6 while maintaining the output voltage VO at the target voltage VT.

  When output current IO exceeds upper limit value IH, voltage V14 across terminals of resistance element 14 becomes higher than reference voltage VR, and overcurrent detection signal OC is set to "H" level of activation level by overcurrent detection circuit 17 The voltage control circuit 18 sets the output control signal CNT to the “H” level for a predetermined time Tc. When output control signal CNT is set to “H” level, capacitors 26 and 27 of frequency setting circuit 19 are connected to capacitance terminal CT of power supply control IC 20, and the frequency of PWM signal φ20 is reduced from f1 to f2, and PWM signal The pulse width TP of φ20 is shortened. As a result, both the output voltage VO and the output current IO of the switching power supply device 10 are reduced, the loop characteristic is realized, and the switching power supply device 10 is protected.

  When the ambient temperature changes, the resistance value of resistance element 21 of overcurrent detection circuit 17 changes accordingly to change reference voltage VR, and the resistance value of resistance element 14 changes with the change of ambient temperature (ie, the resistance element The change in voltage V14 across the fourteen terminals is compensated. When the ambient temperature changes, the resistance value of resistance element 25 of frequency setting circuit 19 changes accordingly, and the capacitance value of capacitors 26 and 27 changes with the change of ambient temperature (that is, the change of the frequency of PWM signal φ20). ) Is compensated. Therefore, a small letter-like character with temperature dependency is realized.

  Next, the case where the output of the switching power supply device 10 is reduced to restore from the over-current protection state to the normal output state will be described. In the overcurrent state, the impedance of the load 6 is lowered, and the load 6 tries to flow the output current IO exceeding the upper limit value IH.

  In general, there are three methods for releasing the overcurrent protection state of the switching power supply 10. The first method is a method in which the power supply from the commercial AC power supply 5 is shut off and then turned on again. The second method is to disconnect the load 6 once. In the third method, the switching power supply device 1 can output the normal output voltage VO and the output current IO by raising the output impedance, that is, reducing the output current IO while receiving AC power from the commercial AC power supply 5 It is a method of returning to the state. The third method is called automatic return.

  In general, the first or second method is employed in a switching power supply provided with a latch circuit that self-holds an operation stop state at the time of overcurrent or load short circuit. When adopting the first or second method, it is necessary to temporarily stop the operation of the product equipped with the switching power supply device.

  On the other hand, in the third method, the AC power from the commercial AC power supply 5 is improved by improving the state of the load 6, that is, eliminating the overload or load shorting state to reduce the current consumption of the load 6. While being received, the output voltage VO of the switching power supply 10 can be returned to the rated voltage value, and the normal state in which the output current IO can be taken out up to the design maximum power can be restored.

  The switching power supply 10 of the first embodiment employs the third method. That is, at the time of an overcurrent or a load short circuit, the overcurrent detection signal OC is set to the activation level "H" level, the output control signal CNT is set to the "H" level for a predetermined time Tc, and the output voltage VO of the switching power supply 10 Reduce the output current IO. This lowers the voltage VO and the current IO supplied to the load 6 and improves the overcurrent condition and the load short condition. After the predetermined time Tc has elapsed, the output voltage VO and the output current IO are returned to their original states and are automatically restored.

  As described above, in the first embodiment, the current I1 flowing through the primary winding 12a of the isolation transformer 12 is converted to the voltage V14 by the resistor element 14, and the PWM is generated when the voltage V14 exceeds the reference voltage VR. The frequency of the signal φ20 is lowered from f1 to f2 and the pulse width TP of the PWM signal φ20 is narrowed. Further, the resistance element 21 compensates for the change in the resistance value of the resistance element 14 due to the change in the ambient temperature. Therefore, it is possible to easily realize the switching power supply device 10 having high safety, which has a small temperature-dependent letter shape characteristic.

  Since the letter-characteristic can be realized using the general-purpose power supply control IC 20, the overcurrent detection circuit 17, the voltage holding circuit 18, and the frequency setting circuit 19, compared to Patent Document 1 using a dedicated power supply control IC. The cost of the apparatus can be reduced.

  Since the change in the voltage between the terminals of the auxiliary winding of the insulating transformer is not used as in Patent Document 1, it is possible to obtain stable V-shaped characteristics with little variation without being affected by the degree of coupling of the insulating transformer. By changing the resistance value of resistance elements 14, 21 and 22, the ratio between upper limit value IH of output current IO and rated current value IR can be changed, and the letter-characteristic with a high degree of freedom can be realized. Can.

  The output current IO is detected, and when the detected value exceeds the upper limit value IH, the overcurrent detection signal OC is set to the “H” level, which is isolated and sent to the power control IC on the primary side. The letter character can be realized with a simple circuit with little

  Furthermore, since the state automatically recovers to the original state when the overcurrent state or the load short-circuit state is eliminated, it is not necessary to shut off the AC power from the commercial AC power supply 5 or disconnect the load 6.

Second Embodiment
The impedance of the primary winding 12a of the isolation transformer 12 includes an inductance component and a stray capacitance component. Therefore, even when the overcurrent state is not obtained, a large current I1 flows through resistance element 14 via the stray capacitance component of primary winding 12a at the moment when N channel MOS transistor 13 is turned on, and the overcurrent detection signal There is a possibility that OC is brought to "H" level to reduce the output voltage VO and the output current IO of the switching power supply. In the second embodiment, the occurrence of such a malfunction is prevented.

  8 is a circuit block diagram showing a main part of a switching power supply according to a second embodiment of the present invention, which is to be compared with FIG. Referring to FIG. 8, this switching power supply differs from switching power supply 10 of the first embodiment in that blanking circuit 30 is added between overcurrent detection circuit 17 and voltage holding circuit 18. is there.

  Blanking circuit 30 is an external circuit of power supply control IC 20, and includes an NPN bipolar transistor 31, a diode 32, a capacitor 33, and a resistance element 34. The collector of the NPN bipolar transistor 31 is connected to the output terminal of the comparator 23, and the emitter is connected to the line of the ground voltage GND. The anode of diode 32 is connected to the line of ground voltage GND, and the cathode is connected to the base of NPN bipolar transistor 31. One terminal of resistance element 34 receives PWM signal φ 20, and the other terminal is connected to the base of NPN bipolar transistor 31 through capacitor 33.

  Resistance element 34 includes a linear temperature coefficient resistor to compensate for the change in threshold voltage Vth of NPN bipolar transistor 31 with the change in ambient temperature. Capacitor 33 operates as a differentiating circuit, differentiates PWM signal φ 20 and applies it to the base of NPN bipolar transistor 31. The diode 32 prevents the base voltage of the NPN bipolar transistor 31 from becoming a negative voltage. Resistor element 34, capacitor 33 and diode 32 constitute a drive circuit.

  FIGS. 9A to 9D are time charts showing the operation of the blanking circuit 30. FIG. 9 (a) shows the waveform of the PWM signal φ 20, FIG. 9 (b) shows the waveform of the current I1 flowing in the resistance element 14, and FIG. 9 (c) shows the waveform of the current I33 flowing in the capacitor 33. FIG. 9D shows the waveform of the base current I31 of the NPN bipolar transistor 31.

  As shown in FIG. 9A, the PWM signal φ20 is raised from the “L” level to the “H” level at time t1, and is raised from the “H” level to the “L” level at time t3. Do. In response to PWM signal φ20, N channel MOS transistor 13 is turned on at time t1 and turned off at time t2.

  When the impedance of primary winding 12a of isolation transformer 12 includes only an inductance component, current I1 flowing through primary winding 12a from when N channel MOS transistor 13 is turned on to when it is turned off (time t1 to t3). Rises linearly. However, since the impedance of primary winding 12a of isolation transformer 12 actually includes an inductance component and a stray capacitance component, when N channel MOS transistor 13 is turned on at time t1, the stray capacitance component of primary winding 12a is As a result, spike-like positive current flows through the resistance element 14 (see a portion surrounded by a circular dotted line A).

  Therefore, the current I1 is the sum of the current flowing in the inductance component of the primary winding 12a and rising linearly, and the spike-like current flowing through the stray capacitance component of the primary winding 12a, as shown in FIG. The waveform is as shown in b). The terminal voltage V14 of the resistance element 14 has the same waveform as the current I1. When voltage V14 corresponding to current I1 in a portion surrounded by dotted line A exceeds reference voltage VR, overcurrent detection signal OC is at the activation level "H" even though overcurrent does not flow in load 6. The output voltage VO and the output current IO are reduced. The blanking circuit 30 prevents such a malfunction.

  Since the PWM signal φ 20 is a pulse signal, each of the rising edge and the falling edge of the PWM signal φ 20 transiently becomes an AC voltage and passes through the capacitor 33. In other words, capacitor 33 operates as a differentiating circuit, and as shown in FIG. 9C, a spike-like positive current flows in response to the rising edge of PWM signal φ20 (time t1), and the falling of PWM signal φ20 A spike-like negative current flows in response to the edge (time t3).

  The base of the NPN bipolar transistor 31 is maintained at 0 V or more by the diode 32. In other words, the spike-like negative current (the portion surrounded by the circular dotted line B) flows through the diode 22 to the line of the ground voltage GND. Therefore, only the spiked positive current of the current I33 flowing through the capacitor 33 flows through the base of the NPN bipolar transistor 31, as shown in FIG. 9 (d). When a spike-like positive current I31 flows to the base, a spike-like current flows between the collector and the emitter of NPN bipolar transistor 31, and the output terminal of comparator 23 is forced to "L" level (ground voltage GND). Ru.

  Therefore, even if voltage V14 generated by spike-like positive current I1 (portion surrounded by circular dotted line A) flowing to the stray capacitance component between terminals of primary winding 12a exceeds reference voltage VR, At (t1 to t2), the output terminal of the comparator 23 is forcibly set to “L” level, and the overcurrent detection signal OC is never set to “H” level. In other words, in response to the rising edge of the PWM signal φ20, the overcurrent detection signal OC is blanked for a predetermined time Tb. The time Tb is set to a time shorter than one cycle of the PWM signal φ20.

  Therefore, in the second embodiment, the switching power supply device is prevented from malfunctioning due to the spike-like positive current flowing in the stray capacitance component between the terminals of primary winding 12a at the moment N channel MOS transistor 13 is turned on. be able to.

  Next, the operation of the blanking circuit 30 with respect to changes in ambient temperature will be described. When the ambient temperature decreases, the threshold voltage Vth of the NPN bipolar transistor 31 increases. Therefore, in the second embodiment, a linear temperature coefficient resistor whose resistance value decreases when the ambient temperature decreases is used as the resistance element 34. When the ambient temperature decreases and the threshold voltage Vth of the NPN bipolar transistor 31 increases, the resistance value of the resistance element 34 decreases, and the reduction of the base current of the NPN bipolar transistor 31 is suppressed. Therefore, it is possible to prevent the NPN bipolar transistor 31 from becoming difficult to turn on when the ambient temperature decreases.

Third Embodiment
FIG. 10 is a circuit block diagram showing a main part of a switching power supply according to a third embodiment of the present invention, which is to be compared with FIG. Referring to FIG. 10, this switching power supply differs from switching power supply 10 of the first embodiment in that overcurrent detection circuit 17 is replaced by overcurrent detection circuit 40.

  Overcurrent detection circuit 40 includes NPN bipolar transistors 41 and 42 and resistance elements 43-45. The collectors of NPN bipolar transistors 41 and 42 are connected to the line of DC power supply voltage VCC via resistance elements 43 and 44 respectively, and their emitters both receive ground voltage GND (reference voltage). The base of the NPN bipolar transistor 41 receives the voltage V14 across the terminals of the resistive element 14 through the resistive element 45, and the collector is connected to the base of the NPN bipolar transistor 42. The signal appearing at the collector of the NPN bipolar transistor 42 becomes the overcurrent detection signal OC.

  The reference voltage VR in the overcurrent detection circuit 40 is represented by a function using the threshold voltage Vth of the NPN bipolar transistor 41, the ambient temperature, and the resistance value of the resistance element 45 as parameters. For example, when the output current IO reaches the upper limit value IH at room temperature (+ 20 ° C.), the base-emitter voltage VBE of the NPN bipolar transistor 41 reaches the threshold voltage Vth. Each resistance value is set. Resistive elements 43 and 44 limit the current flowing to the collectors of NPN bipolar transistors 41 and 42, respectively. Resistance element 45 limits the current flowing to the base of NPN bipolar transistor 41. When the ambient temperature changes and the resistance value of resistance element 14 changes, reference voltage VR also changes accordingly, and resistance element 45 compensates for the change in resistance value of resistance element 14 accompanying the change in ambient temperature.

  When the output current IO is smaller than the upper limit value IH, the base-emitter voltage VBE of the NPN bipolar transistor 41 becomes smaller than the threshold voltage Vth, and the NPN bipolar transistor 41 is turned off. As a result, the collector of NPN bipolar transistor 41 attains "H" level (DC power supply voltage VCC), the base of NPN bipolar transistor 42 attains "H" level, and NPN bipolar transistor 42 is turned on, and overcurrent detection signal OC. Becomes the inactive level "L" level.

  When output current IO exceeds upper limit value IH, base-emitter voltage VBE of NPN bipolar transistor 41 becomes larger than threshold voltage Vth, and NPN bipolar transistor 41 is turned on. As a result, the collector of NPN bipolar transistor 41 attains to the “L” level (ground voltage GND), the base of NPN bipolar transistor 42 attains to the “L” level, and NPN bipolar transistor 42 is turned off. It becomes "H" level of activation level.

  Next, the operation of the overcurrent detection circuit 40 with respect to changes in ambient temperature will be described. The base-emitter of the NPN bipolar transistor 41 has diode characteristics. The diode has a temperature characteristic that the forward voltage VF increases as the temperature decreases and decreases as the temperature increases. The base-emitter voltage VBE required to turn on the NPN bipolar transistor 41 is called a threshold voltage Vth. Assuming that the threshold voltage Vth of the NPN bipolar transistor 41 is 0.6 V when the ambient temperature is normal temperature (+ 20.degree. C.), the threshold voltage Vth is lower than 0.6 V at a low temperature (-40.degree. C.). The threshold voltage Vth becomes larger than 0.6 V at high temperature (+ 85 ° C.).

  In addition, the resistance value of the resistance element 14 also changes according to the ambient temperature. Therefore, this overcurrent detection circuit 40 compensates for the change in threshold voltage Vth of NPN bipolar transistor 41 with the change in ambient temperature and the change in resistance value of resistance element 14 with the change in ambient temperature. A linear temperature coefficient resistor with a temperature coefficient is used as the resistive element 45.

  For example, a linear temperature coefficient resistor is used as the resistive element 45, which has a small resistance at low temperature (-40.degree. C.) and a large resistance at high temperature (+ 85.degree. C.). Therefore, the resistance value of resistance element 45 is lowered even when the ambient temperature is lowered and the resistance value of resistance element 14 is lowered and the threshold voltage Vth of NPN bipolar transistor 41 is also raised, so that the base of NPN bipolar transistor 41 A reduction in current can be suppressed, and overcurrent protection can be performed accurately even at low temperatures. The other configuration and operation are the same as in the first embodiment, and therefore the description will not be repeated.

  In the third embodiment, the overcurrent detection circuit 40 includes only the NPN bipolar transistors 41 and 42 and the resistance elements 43 to 45. Therefore, the size and cost of the device can be reduced as compared with the first embodiment. it can. Moreover, since the resistance element 45 compensates for the change in the resistance value of the resistance element 14 and the change in the threshold voltage Vth of the NPN bipolar transistor 41 due to the change in ambient temperature, it is possible to realize a response characteristic with small temperature dependence. it can.

  When the inactivation level of the overcurrent detection signal OC is set to the “H” level and the activation level is set to the “L” level, the NPN bipolar transistor 42 and the resistance element 44 are removed. The signal appearing at the collector may be the over current detection signal OC. In this case, the size and cost of the apparatus can be further reduced.

  It should be understood that the embodiments disclosed herein are illustrative and non-restrictive in every respect. The scope of the present invention is indicated not by the above description but by the claims, and is intended to include all the modifications within the meaning and scope equivalent to the claims.

  1, 10 switching power supply, 2 rectification circuits, 3, 26, 27, 33, C1, C2 capacitors, 4, 11 power conversion circuits, 5 commercial AC power supplies, 6 loads, T1, T2 input terminals, T3, T4 output terminals , 12 isolation transformer, 12a primary winding, 12b secondary winding, 13 N channel MOS transistors, 14, 21, 22, 25, 25, 34, 43-45 resistive elements, 15, 32, D1 diodes, 16 feedback circuits, 17, 40 Overcurrent Detection Circuit, 18 Voltage Holding Circuit, 19 Frequency Setting Circuit, 20 Power Supply Control IC, 23 Comparator, 28, 31, 41, 42 NPN Bipolar Transistor, 30 Blanking Circuit.

Claims (15)

A power conversion circuit,
First and second input terminals receiving a first DC voltage;
An isolation transformer including a primary winding and a secondary winding;
A switching element that turns on / off in response to the PWM signal;
An output voltage generation circuit that rectifies and smoothes a voltage across terminals of the secondary winding and generates a second DC voltage to be applied to a load;
A first resistance element connected in series with the primary winding and the switching element between the first and second input terminals;
When the voltage across terminals of the first resistance element is lower than the reference voltage, the overcurrent detection signal is set to the inactivation level, and when the voltage across terminals of the first resistance element is higher than the reference voltage, the over An overcurrent detection circuit for setting the current detection signal to an activation level;
When the overcurrent detection signal is at the activation level, the output control signal is set to the first logic level, and the first time has elapsed since the overcurrent detection signal changed from the activation level to the deactivation level. A timer circuit to bring the output control signal to the second logic level later;
The frequency of the PWM signal is set to a first value when the output control signal is a second logic level, and the frequency of the PWM signal is set to a first value when the output control signal is a first logic level. A frequency setting circuit configured to set a second value lower than the value of 1,
The PWM signal of the frequency set by the frequency setting circuit is generated, and when the output control signal is at the second logic level, the pulse of the PWM signal is set such that the second DC voltage becomes the target voltage. Control circuitry for controlling the width and narrowing the pulse width of the PWM signal in response to the output control signal being brought to the first logic level;
The resistance value of the first resistance element changes in accordance with the ambient temperature of the power conversion circuit,
The power conversion circuit, wherein the overcurrent detection circuit changes the reference voltage in response to a change in the ambient temperature, and compensates for a change in the resistance value of the first resistive element with the change in the ambient temperature.   The power conversion circuit according to claim 1, wherein the first time is longer than one cycle of the PWM signal.   The power conversion circuit according to claim 1 or 2, wherein when the output current of the power conversion circuit reaches an upper limit value equal to or higher than a rated current value, a voltage between terminals of the first resistance element reaches the reference voltage. .   The output voltage and the output current of the power conversion circuit are both reduced when the voltage across terminals of the first resistive element exceeds the reference voltage. Power conversion circuit as described. The overcurrent detection circuit is
A voltage dividing circuit that divides a DC power supply voltage to generate the reference voltage;
Comparing the voltage between terminals of the first resistive element with the reference voltage, and setting the overcurrent detection signal to an inactive level if the voltage between terminals of the first resistive element is lower than the reference voltage; A comparator for setting the overcurrent detection signal to an activation level when a voltage across terminals of the first resistance element is higher than the reference voltage;
The voltage dividing circuit includes second and third resistance elements connected in series between the line of the DC power supply voltage and the line of a reference voltage,
At least one of the second and third resistance elements includes a first temperature coefficient resistor whose resistance value changes according to the ambient temperature,
The first temperature coefficient resistor changes the reference voltage in response to a change in the ambient temperature, and compensates for a change in the resistance value of the first resistive element with the change in the ambient temperature. The power conversion circuit according to any one of claims 1 to 4.   The power conversion circuit according to claim 5, wherein a division ratio of the voltage dividing circuit is changeable. The overcurrent detection circuit includes second, third and fourth resistance elements and first and second transistors,
One terminal of the second resistance element receives a voltage between terminals of the first resistance element,
One terminal of the third and fourth resistance elements receives a DC power supply voltage,
The first electrodes of the first and second transistors are connected to the other terminals of the second and third resistance elements, respectively, and their second electrodes both receive a reference voltage,
The control electrode of the first transistor is connected to the other terminal of the second resistance element,
The control electrode of the second transistor is connected to the first electrode of the first transistor,
The threshold voltage of the first transistor varies with the ambient temperature,
The reference voltage is determined by the resistance value of the second resistance element and the threshold voltage of the first transistor.
The second resistive element includes a first temperature coefficient resistor whose resistance value changes according to the ambient temperature,
The first temperature coefficient resistor changes the reference voltage in response to a change in the ambient temperature, and compensates for the change in the resistance value of the first resistive element with the change in the ambient temperature.
The signal appearing at the first electrode of the second transistor is the overcurrent detection signal,
The DC power supply voltage is an activation level of the overcurrent detection signal, and the reference voltage is an inactivation level of the overcurrent detection signal.
When the voltage across the terminals of the first resistance element is lower than the reference voltage, the first transistor is turned off and the second transistor is turned on to bring the overcurrent detection signal to an inactivation level.
When the voltage across terminals of the first resistive element is higher than the reference voltage, the first transistor is turned on and the second transistor is turned off, and the overcurrent detection signal becomes an activation level. The power conversion circuit according to any one of claims 1 to 4. The overcurrent detection circuit includes second and third resistance elements and a first transistor,
One terminal of the second resistance element receives a voltage between terminals of the first resistance element,
One terminal of the third resistance element receives a DC power supply voltage,
The first electrode of the first transistor is connected to the other terminal of the second resistive element, the second electrode receives a reference voltage, and the control electrode is connected to the other terminal of the second resistive element. And
The threshold voltage of the first transistor varies with the ambient temperature,
The reference voltage is determined by the resistance value of the second resistance element and the threshold voltage of the first transistor.
The second resistive element includes a first temperature coefficient resistor whose resistance value changes according to the ambient temperature,
The first temperature coefficient resistor changes the reference voltage in response to a change in the ambient temperature, and compensates for the change in the resistance value of the first resistive element with the change in the ambient temperature.
The signal appearing at the first electrode of the first transistor is the overcurrent detection signal,
The DC power supply voltage is an inactivation level of the overcurrent detection signal, and the reference voltage is an activation level of the overcurrent detection signal.
When the voltage across terminals of the first resistance element is lower than the reference voltage, the first transistor is turned off and the overcurrent detection signal is at the inactivation level.
5. The device according to claim 1, wherein when the voltage across terminals of the first resistive element is higher than the reference voltage, the first transistor is turned on and the overcurrent detection signal becomes an activation level. The power conversion circuit according to any one of the items. The activation level of the overcurrent detection signal is a DC power supply voltage, and the deactivation level of the overcurrent detection signal is a reference voltage.
The timer circuit is
A diode whose anode receives the overcurrent detection signal;
A first capacitor having one terminal connected to the cathode of the diode and the other terminal receiving the reference voltage;
A second resistance element connected between one terminal of the first capacitor and the line of the reference voltage,
The power conversion circuit according to any one of claims 1 to 4, wherein a signal appearing at one electrode of the first capacitor is the output control signal.   The circuit further includes a blanking circuit that fixes the overcurrent detection signal at the inactivation level for a second time shorter than one cycle of the PWM signal in response to the switching element being turned on. The power converter circuit according to any one of claims 9 to 10. When the PWM signal is a reference voltage, the switching element is turned off, and when the PWM signal is a DC power supply voltage, the switching element is turned on,
The activation level of the overcurrent detection signal is the DC power supply voltage, and the deactivation level of the overcurrent detection signal is the reference voltage.
The overcurrent detection circuit includes an output terminal for outputting the overcurrent detection signal.
The blanking circuit is
A third transistor connected between the output terminal and the line of the reference voltage;
11. The power conversion circuit according to claim 10, further comprising: a drive circuit that turns on said third transistor for said second time in response to said PWM signal changing from said reference voltage to said DC power supply voltage. The drive circuit is
A third resistance element having one terminal receiving the PWM signal;
A second capacitor connected to the other terminal of the third resistor element and having the other terminal connected to the control electrode of the third transistor;
The power conversion according to claim 11, wherein said third resistive element includes a second temperature coefficient resistor for compensating for a change in threshold voltage of said third transistor accompanying a change in said ambient temperature. circuit. The frequency setting circuit is
A second capacitance having a first capacitance value when the output control signal is at a second logic level, and a second capacitance larger than the first capacitance value when the output control signal is at a first logic level A variable capacitance capacitor having a value
Current source for charging the variable capacitance capacitor,
The current source supplies a current having a value corresponding to the ambient temperature to the variable capacitance capacitor, and a third temperature coefficient resistor for compensating for a change in capacitance value of the variable capacitance capacitor accompanying a change in the ambient temperature. Including
When the voltage across terminals of the variable capacitor reaches a predetermined voltage value, the variable capacitor is discharged,
The power conversion circuit according to any one of claims 1 to 12, wherein the frequency at which the variable capacitance capacitor is charged and discharged is the frequency of the PWM signal.   The power conversion circuit according to claim 13, wherein each of the first and second capacitance values is changeable. A power conversion circuit according to any one of claims 1 to 14.
And a DC voltage generation circuit that rectifies and smoothes an AC voltage supplied from an AC power supply to generate the first DC voltage.

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