JP2009136105A5 - - Google Patents

Description

Switching power supply device and driving method thereof

The present invention relates to a switching power supply that controls a switching operation by a pulse width modulation signal, converts a DC voltage into a desired voltage, and supplies power to an electronic device or the like, and more particularly, a switching power supply that controls an output current by digital control. The present invention relates to an apparatus and a driving method thereof.

  A general switching power supply device is provided with an overcurrent protection circuit that limits its supply capability so that an output current exceeding a predetermined value does not flow. This is because when an electronic device, etc., which is a load connected to the output, has a low impedance failure, excessive current continues to flow, preventing the electronic device from overheating and burning, and failure of the switching power supply device itself. Its purpose is to prevent overheating and burning.

  On the other hand, various electronic devices serving as loads of the switching power supply device are devices equipped with electromagnetic drive system parts such as motors and relays, and electronic circuits equipped with digital elements such as CPUs, DSPs, FPGAs, etc. In the case of a device in which a current flows, there is a device in which a large current (hereinafter, peak current) flows for a very short time even if the device does not fail. In such a case, such a peak current can be safely supplied even in a switching power supply device composed of small and small capacity power components for a short time. Then, for example, a predetermined set value (hereinafter referred to as an overcurrent set value) that exceeds the peak current value necessary for normal operation of the load and is allowed as the upper limit value of the output current is set, and the capacitor, resistance, etc. A peak overcurrent protection circuit that monitors the duration of the peak current with the timer circuit configured in the above, and changes the overcurrent setting value to a smaller value when that time exceeds a predetermined setting time has been put to practical use. ing.

  The overcurrent protection circuit as described above is generally an analog control circuit composed of discrete components. Therefore, for example, in order to provide a control characteristic such as an overcurrent set value with a function of adjusting the control characteristic in accordance with the operating state of the electronic device or the like, a complicated circuit for performing various signal processing is necessary. It must be composed of many discrete components, which is not only troublesome in terms of circuit operation evaluation and other management, but also because of the increase in the number of components, which can prevent downsizing and cost reduction of switching power supply devices. It is one.

  On the other hand, an overcurrent protection circuit based on digital control can easily adjust or switch control characteristics by incorporating complicated signal processing into an arithmetic program, and has excellent intelligence. For example, as disclosed in Patent Document 1, a current detection resistor that converts an output current supplied to a load into a voltage signal, and an analog / digital converter (hereinafter referred to as A / D) that converts the voltage signal into a digital value. Converter), calculation means for determining whether or not the output current is in an overcurrent state exceeding the overcurrent set value based on the digital value, and a pulse of the switching element when the overcurrent state is determined. There has been proposed an overcurrent protection circuit including pulse width varying means for performing processing for forcibly shortening the width. Note that the arithmetic means and the pulse width varying means are configured using a digital processor. This digital processor includes an addition / subtraction function, multiplication / division function, comparison function, timer function, storage function, etc., and stores each set value or target value set from the outside by a storage function such as a memory or a register. It can be left.

Further, the switching power supply device is provided with a first set value as an overcurrent set value and a second set value lower than the first set value, and the output current is initially in an overcurrent state. Is limited based on the first set value, and is limited based on the second set value after a predetermined time has elapsed. Specifically, the comparison function of the digital processor compares the digital value of the output current with the first set value and the second set value, and as a result of the calculation, the digital value exceeds the first set value. The pulse width is forcibly narrowed when it is determined that it is present, and even if this digital value does not exceed the first set value, it is determined that the state exceeding the second set value has continued for a predetermined time or more. In this case, the pulse width is controlled to be forcibly narrowed.
JP 2001-119933 A

  However, in the overcurrent protection circuit disclosed in Patent Document 1, when mounted on a current practical switching power supply device, a control characteristic capable of high-speed response to a steep increase in output current is obtained. Therefore, there is a problem that an expensive digital processor that operates with a high-speed clock is required.

  Generally, in order to avoid a failure of a power semiconductor or the like inside the switching power supply device, the control of the overcurrent protection circuit must respond in a short time to one cycle of the switching operation. Current general-purpose switching power supply apparatuses are often set to a switching frequency of 500 kHz or more from the viewpoint of miniaturization of magnetic parts and the like. For example, when the switching frequency is set to 500 kHz, the response delay time of the overcurrent protection circuit of a typical analog control IC is set to 130 nsec. In order to obtain a response speed equivalent to this, 1 of the switching operation is required. Control characteristics capable of responding at 130 nsec or less with respect to the period of 2 μsec are required.

  However, in the flow of digital control of the overcurrent protection circuit disclosed in Patent Document 1, the process of fetching the digital value of the output current obtained through the current detection resistor and the A / D converter into the digital processor 50 is performed. Whether or not the overcurrent state exceeds the overcurrent set value by about the clock is determined by the arithmetic processing based on the digital value, and the overcurrent state is determined and the switching element is controlled to forcibly turn off. It is considered that the processing man-hours of about 50 clocks and about 100 clocks are necessary for the processing. As described above, in order to complete the processing of 100 clocks in a period of 130 nsec, the clock frequency of the digital processor needs to be about 769 MHz, which is a value obtained by dividing the processing man-hour of 100 clocks by 130 nsec. Further, since the digital processor performs various arithmetic processes in addition to the above-described overcurrent protection control, a higher clock frequency is actually required. In other words, in order to obtain an overcurrent protection circuit with high-speed response equivalent to analog control, a digital processor capable of high-speed signal processing with a clock frequency several orders of magnitude greater than the switching frequency of the switching power supply circuit is required. I need it.

  Such a high-performance digital processor operating at a high speed is very expensive. Therefore, adopting digital control in a general-purpose switching power supply device has a cost problem, and prevents the digital control switching power supply device excellent in intelligentness from spreading.

  Control characteristics such as overcurrent set value and peak overcurrent protection circuit setting time depend on individual characteristics of each component of the power supply circuit, variation in component characteristics due to temperature, variation in input voltage and output voltage, etc. It fluctuates due to a change in output current detection accuracy.

  If the overcurrent setting value (output current limit value) is set higher or the time limit is set longer, the electronic device, etc. will generate heat and burn out when it becomes overcurrent due to a failure of the load electronic device, etc. The protection function for preventing the failure cannot be sufficiently exhibited, and at the same time, the risk of failure of the switching power supply increases. For example, in order to ensure the safety of the switching power supply itself, large power components with sufficient current rating and temperature rating should be used to avoid failure, overheating and burning against the actual limit value of the output current. Must be used to construct the power circuit. Therefore, this is one factor that hinders downsizing and cost reduction of the switching power supply device.

The present invention has been made in view of the above-mentioned background art. It has excellent intelligentness, has a high-speed response equal to or higher than analog control while using a relatively low-cost general-purpose digital processor, etc., and has high accuracy. It is an object of the present invention to provide a switching power supply device having a digitally controlled overcurrent protection function capable of obtaining an overcurrent protection characteristic and a driving method thereof.

  According to the first aspect of the present invention, a drive pulse generation circuit that outputs a drive pulse that has been pulse-width modulated at a predetermined switching frequency, and a DC input voltage is intermittently generated by the drive pulse from the drive pulse generation circuit to generate an AC voltage. A switching power supply device comprising an inverter circuit having a switching element such as a MOS-FET for generating a rectifying and smoothing circuit that rectifies and smoothes the alternating voltage to generate an output voltage and supplies an output current to a load, A target value setting unit that outputs a changeable value that is a predetermined target value related to the control of the output current, a calculation unit that performs a calculation process related to the control of the output current based on the target value, and a calculation unit thereof Current control pulse generating means comprising a pulse generator for generating a current control pulse voltage for controlling the output current based on the calculation result It is as it has. Furthermore, a current detection circuit for detecting an output current from the rectifying / smoothing circuit or a current flowing through the switching element, and a current detected by the current detection circuit are set based on an output of the current control pulse generating means. A current limit signal generating circuit for outputting a current limit signal for limiting the current when a reference value is exceeded, and the drive pulse generating circuit drives the switching element when the current limit signal is output This is a switching power supply device that operates so as to stop or narrow the on-duty of the drive pulse.

  The current control pulse generating means may comprise a digital processor configured to output the current control pulse voltage as a pulse width modulated signal, and smoothes the current control pulse voltage by smoothing the current control pulse voltage. A pulse smoothing circuit for generating a voltage is connected, the current detection circuit detects the output current or the current flowing through the switching element, and outputs a current detection voltage based on the detected current, the current limit signal generation circuit A comparison circuit that compares the current detection voltage with a reference voltage determined based on the smoothing voltage, and outputs the current limit signal when the current detection voltage exceeds the reference voltage; When the current limit signal is output, the pulse generation circuit stops the on-duty of the drive pulse that drives the switching element from increasing. Properly is the switching power supply device operable to narrow.

  In the invention according to claim 3 of the present application, the current limiting signal generation circuit includes a first resistor having one end connected to the output of the pulse smoothing circuit and one end connected to the other end of the first resistor. A second resistor and a collector terminal are connected to the other end of the second resistor, a base terminal is connected to a midpoint of the first resistor and the second resistor, and an emitter terminal of the current detection circuit output A first NPN transistor connected to one end, a base terminal is connected to the collector of the first NPN transistor, an emitter terminal is connected to the other end of the current detection circuit output, and a collector terminal generates the current limit signal A second NPN transistor that is an output of the circuit, one of the first and second NPN transistors is connected to a ground potential, and the current detection circuit includes the rectifying and smoothing circuit When the output current of the second NPN transistor flows or when the current flows through the switching element, the emitter terminal of the second NPN transistor has a lower potential than the emitter terminal of the first NPN transistor. A detection voltage is output.

  In the invention according to claim 4 of the present application, when the current limit signal is continuously generated by the current limit signal generation circuit or repeatedly generated at each switching frequency for a predetermined time or more, the time is exceeded. Current limit operation time monitoring means for outputting a signal, the target value setting unit of the current control pulse generation means is based on the time excess signal of the current limit operation time monitoring means, the current limit signal generation circuit of the current limit signal generation circuit The target value is determined and output to the calculation unit so as to further limit the current detected by the current detection circuit by a current limit signal, and the current calculated by the calculation unit based on the target value A control pulse voltage is output from the pulse generator.

  In the invention according to claim 5 of the present application, the calculation unit of the current control pulse generating unit calculates a time ratio of the current control pulse voltage based on a target value output from the target value setting unit, and the pulse generation unit Is configured to generate a rectangular wave with a constant frequency having a high-level and low-level voltage of the duty ratio calculated by the arithmetic unit, and the pulse smoothing circuit has one end connected to the output of the current control pulse generating means And a capacitor connected between the other end of the resistor and the ground, and a voltage across the capacitor serves as an output of the pulse smoothing circuit.

  According to the sixth aspect of the present invention, the calculation unit of the current control pulse generating unit calculates a time ratio of the current control pulse voltage based on a target value output from the target value setting unit, and the pulse generation unit Is configured to repeat the high-level and floating-level output states calculated by the arithmetic unit within a certain period, and one end of the pulse smoothing circuit is connected to the output of the current control pulse generating means. And a capacitor connected between the other end of the resistor and the ground, and a discharge path connected in parallel to the capacitor, the voltage across the capacitor serving as the output of the pulse smoothing circuit It is. The discharge path connected in parallel to the capacitor is an input impedance of the current signal generation circuit.

  In the invention of claim 7, the pulse smoothing circuit is connected between a first resistor having one end connected to the output of the current control pulse generating means, and the other end of the resistor and the ground. A capacitor for generating an output voltage of the pulse smoothing circuit at both ends, a discharge path connected in parallel to the capacitor, and a current from the connection point of the capacitor and the first resistor toward the output of the current control pulse generating means A series circuit of a diode and a second resistor connected so as to be able to flow, and the arithmetic unit of the current control pulse generating means is given a target value determined based on the operating time excess signal The output state of the pulse generator is controlled to a low level for a predetermined time, and the voltage of the capacitor of the pulse smoothing circuit is controlled via a series circuit of the diode and a second resistor. Are those electricity.

  The invention according to claim 8 of the present application includes an error amplification circuit that compares the output voltage of the switching power supply device with a predetermined target value and outputs an inverted and amplified output voltage control signal, and the current limit signal generation circuit includes: A comparison circuit that compares the reference voltage determined based on the smoothed voltage with the current detection voltage, and outputs a current limit signal that is different between when the current detection voltage exceeds the reference voltage and otherwise. The drive pulse generation circuit includes a sawtooth wave generation circuit that generates a sawtooth wave voltage driven at the switching frequency, and the sawtooth wave voltage is input to a first input terminal and input to a second input terminal. A comparator that compares the generated output voltage control signal with the sawtooth voltage, and outputs a different signal between a period when the sawtooth voltage is lower and a period other than the sawtooth voltage, and the current limiter When the signal selection circuit is in a state where the current limit signal is continuously generated or repeatedly generated for each switching frequency by the current limit signal generation circuit, A duty ratio determined based on the current limit signal is output from the drive pulse generation circuit and a time ratio determined based on the output voltage control signal when the current limit signal is not output. The operation of selecting the output signal is performed so that the drive pulse is output from the drive pulse generation circuit.

  The invention according to claim 9 of the present application includes temperature detection means for sensing the environmental temperature of the switching power supply device by a temperature sensing element and outputting a temperature signal based thereon, and the target value setting unit of the current control pulse generation means is The target value is determined based on the temperature signal, and the determined target value is output to the calculation unit.

  The invention according to claim 10 of the present application includes input voltage detection means for detecting an input voltage of the switching power supply device and outputting an input voltage signal based on the input voltage, and the target value setting unit of the current control pulse generating means includes The target value is determined based on an input voltage signal, and the determined target value is given to the arithmetic unit.

  The invention according to claim 11 of the present application includes output voltage detection means for detecting an output voltage of the switching power supply device and outputting an output voltage signal based thereon, and the target value setting unit of the current control pulse generating means The target value is determined based on an output voltage signal, and the determined target value is given to the arithmetic unit.

The invention according to claim 12 of the present application is a drive pulse generation circuit that outputs a drive pulse that is pulse-width modulated at a predetermined switching frequency, and an alternating current by intermittently applying a DC input voltage by the drive pulse from the drive pulse generation circuit. An inverter circuit having a switching element for generating a voltage, a rectifying / smoothing circuit for rectifying and smoothing the AC voltage to generate an output voltage and supplying an output current to a load, and a predetermined target value related to the control of the output current. A target value setting unit that outputs a changeable value, a calculation unit that performs calculation processing related to output current control based on the target value and outputs a calculation result, and controls the output current based on the calculation result And a pulse generator for generating a current control pulse voltage, and an output current of the rectifying and smoothing circuit or a current flowing through the switching element is detected. The flow detection circuit, wherein is detected current by the current detecting circuit, a current limit signal to output a current limit signal for limiting the current when it exceeds a predetermined reference value set by the current control pulse voltage A switching power supply device driving method comprising a generation circuit, wherein at the beginning of power supply operation, a predetermined DC voltage is supplied to the input of the switching power supply device to start it, and a load is connected to the output. Performing a load setting process for outputting an output current equal to a desired overcurrent set value, adjusting a predetermined fixed coefficient set in a calculation program of the target value setting unit provided in the current control pulse generating means, A target value variable process for continuously changing the target value output to the calculation unit is performed, and the output current is in the state of the overcurrent set value in response to the change of the target value. Boundary value extraction processing for extracting a boundary value of the fixed coefficient at a boundary point where the output voltage is continuously changed and the output voltage is held at a constant value with respect to the change of the target value A boundary value storage process for storing the boundary value of the fixed coefficient as a set value of the fixed coefficient by the arithmetic program, and the target value setting unit stores the fixed coefficient stored by the boundary value storage process. As a fixed coefficient set in the calculation program of the target value setting unit, a new target value to be output to the calculation unit is obtained, and the new target value is output to the calculation unit. Then, an arithmetic process related to the control of the output current is performed based on the new target value, and the calculation result is output to the pulse generation unit, and the pulse generation unit is based on the calculation result. Generating a current control pulse voltage for controlling the output current, and the current limit signal generation circuit is configured such that the current detected by the current detection circuit is set based on the current control pulse voltage. When the current exceeds the value, a current limit signal for limiting the current is output, and the drive pulse generation circuit operates to stop or narrow the on-duty of the drive pulse by the current limit signal. This is a method of driving the switching power supply apparatus .

  According to the switching power supply device of the present invention, a digitally controlled overcurrent protection circuit that is excellent in intelligence and capable of high-speed response can be realized by a low-cost digital processor with a relatively low-speed clock. In particular, optimal overcurrent protection control can be performed even with respect to fluctuations in the external temperature, input voltage, output voltage, and the like. In addition, the overcurrent protection characteristics can be automatically adjusted according to individual differences in characteristics of discrete components, etc. constituting the switching power supply device and the usage state of the switching power supply device. Can be secured. A small and low-cost switching power supply device can be provided.

In addition, according to the switching power supply driving method of the present invention, the overcurrent protection characteristics are adjusted programmatically and set to an optimum state in accordance with individual differences in characteristics of discrete components and the like constituting the switching power supply. Therefore, it is possible to reduce processes such as adjustment of resistance values and replacement of parts, and it is possible to ensure overcurrent protection and safety of load electronic devices and switching power supply devices themselves with a simple circuit configuration.

Hereinafter, a first embodiment of a switching power supply device according to the present invention will be described with reference to FIGS. First, an outline will be described based on the overall circuit diagram shown in FIG. The switching power supply device 10 is connected to a DC input power supply Ein, and generates an output voltage Vout by rectifying and smoothing the AC voltage V2 and an inverter circuit 12 that generates the AC voltage V2 by the on / off operation of the switching element TR1. A rectifying / smoothing circuit 14 is provided, and an output voltage Vout terminal is connected to the load 22. The output voltage Vout is connected to an error amplification circuit 16 including an inverting amplifier that outputs an output voltage control signal V (vol) obtained by amplifying the difference between the output voltage Vout and a predetermined reference voltage Vref. Further, a drive pulse generation circuit 18 that performs pulse width modulation according to the output voltage control signal V (vol) output from the error amplifier circuit 16 and outputs the drive pulse Vg toward the drive terminal of the switching element TR1 is provided. .

In addition, the temperature detection means 24 that detects the environmental temperature of the switching power supply 10 and outputs a temperature signal, the input voltage detection means 26 that detects the input voltage and outputs the input voltage signal, and detects and outputs the output voltage Output voltage detecting means 28 for outputting a voltage signal is connected to the current control pulse generating means 32, respectively.

  The current control pulse generating means 32 comprises a digital processor or the like, and performs a calculation process based on the target value setting unit 32a that determines a target value based on the temperature signal, the input voltage signal, and the output voltage signal. The calculating part 32b and the pulse generation part 32c which outputs the current control pulse voltage Vc for controlling an output current based on the calculation result are provided. The output of the pulse generator 32c is connected to the input of a pulse smoothing circuit 34, which is an integrating circuit composed of a resistor R1 and a capacitor C1. Then, both ends of the capacitor C1 where the smoothed voltage Vb is generated are connected as an output of the pulse smoothing circuit 34 to a current limit signal generating circuit 36 described later.

  A current detection circuit 38, which is a current detection resistor R0, is inserted between the source terminal of the switching element TR1 and the negative terminal of the input power source Ein, and a voltage corresponding to the switching current Isw flowing through the current detection resistor R0 is generated at both ends. . The both ends of the current detection resistor R0 are connected to the current limit signal generation circuit 36 as outputs of the current detection circuit 38.

The current limit signal generation circuit 36 is an overcurrent protection circuit for the switching power supply device 10, and includes a reference voltage Vr determined based on the smoothed voltage Vb, and a voltage V generated across the reference voltage Vr and the current detection resistor R0. (R0) and a comparator CP1 that is a comparison circuit that outputs a current limiting signal V (cur). The output of the comparator CP1 is a reference value in which the voltage detected corresponding to the current of the current detection resistor R0 of the current detection circuit 38 is set based on the output of the pulse generator 32c of the current control pulse generator 32. When a certain reference voltage Vr is exceeded, it is output as a current limiting signal for limiting the current. The output is connected to the drive pulse generation circuit 18 as an output of the current limit signal generation circuit 36. The drive pulse generation circuit 18 is a circuit that performs pulse width modulation based on the input current limiting signal V (cur) and outputs the drive pulse Vg of the switching element TR1, and its output is supplied to the drive terminal of the switching element TR1. It is connected.

  Next, the configuration and operation of each circuit block of the switching power supply device 10 will be described in detail with reference to FIGS. As shown in FIG. 1, the inverter circuit 12 includes a primary side winding T1a of a transformer T1 and a switching element TR1 connected in series with an input power source Ein, and a secondary side of the transformer T1 by an on / off operation of the switching element TR1. An AC voltage V2 is generated in the winding T1b. The polarity of each winding of the transformer T1 is configured in a well-known single forward system.

  A rectifying / smoothing circuit 14 is connected to the secondary side winding T1b of the transformer T1, and when the switching element TR1 is turned on, a forward side rectifying element TR2 that conducts and flows a pulse current, and a forward side rectifying element TR2 The flywheel side rectifying element TR3 that is complementarily turned on / off, the synchronous rectification driving circuit 14a that drives the rectifying elements TR2 and TR3 in synchronization with the on / off operation of the switch element TR1, and the choke coil Lo and the capacitor Co This is a smoothing circuit. The rectifying / smoothing circuit 14 rectifies and smoothes the voltage induced in the secondary winding T1b and outputs the output voltage Vout. A load 22 is connected to both ends of the capacitor Co, and an output voltage Vout and an output current Iout are supplied.

  As shown in FIG. 1, the error amplifying circuit 16 includes an operational amplifier OP1 whose output voltage analog signal is input to its inverting input terminal, a predetermined reference voltage Vref connected to its non-inverting input, gain adjustment and phase compensation. And an inverting amplifier circuit that outputs an output voltage control signal V (vol). Therefore, the output voltage control signal V (vol) is obtained by amplifying the difference between the output voltage and the reference voltage Vref. When the output voltage becomes higher than the reference voltage Vref, the output voltage control signal V (vol) continuously decreases, and vice versa. Will rise continuously.

  As shown in FIG. 2A, the temperature detecting means 24 includes a thermistor 24a that is a resistance value change type temperature sensing element, an amplifier 24b that converts the resistance value of the thermistor 24a into an analog voltage signal, and the analog voltage signal. And an A / D conversion circuit 24c that converts the signal into a temperature signal that is a digital signal and outputs the temperature signal. The thermistor 24a is mounted on a printed board inside the switching power supply device and measures the temperature on the printed board.

  As shown in FIG. 2B, the input voltage detecting means 26 observes the input voltage Vin, converts the input voltage Vin into a predetermined analog voltage, and inputs the analog voltage as a digital signal. And an A / D conversion circuit 26b that converts the voltage signal and outputs the voltage signal. The voltage conversion circuit 26a is set as appropriate, such as a voltage dividing circuit for resistance-dividing the input voltage Vin, or a circuit for converting a voltage generated in the third winding provided in the transformer T1 into a voltage proportional to the input voltage. Is. Here, as in the latter example, the input voltage Vin may be indirectly observed.

  As shown in FIG. 2C, the output voltage detection means 28 observes the output voltage Vout, converts the output voltage Vout into a predetermined analog voltage, and outputs the analog voltage as a digital signal. And an A / D conversion circuit 28b that converts the voltage signal and outputs the voltage signal. The voltage conversion circuit 28a is set as appropriate, such as a voltage dividing circuit for resistance-dividing the output voltage Vout, or a circuit for converting the voltage generated in the third winding provided in the transformer T1 into a voltage proportional to the output voltage. Is. Here, as in the latter example, the output voltage Vout may be observed indirectly.

The current control pulse generating means 32 is configured using, for example, a general-purpose digital processor (microcomputer). As shown in FIG. 1, the current control pulse generation means 32 includes a target value setting unit 32a, a calculation unit 32b, and a pulse generation unit 32c. The target value setting unit 32a outputs the target value C toward the calculation unit 32b based on the temperature signal, the input voltage signal, and the output voltage signal. When the temperature signal, input voltage signal, and output voltage signal are not input from the outside, the temperature signal default value, input voltage signal default value, and output voltage signal default value stored in advance are fixed. Based on this, the target value C is determined.

  The calculation unit 32b performs calculation processing based on the target value C to determine the set value B, and outputs the set value B and the set value A that is fixedly set in advance toward the pulse generation unit 32c. .

  As shown in FIG. 3, the pulse generation unit 32c includes a clock circuit c1 that generates a clock signal, counters c2 and c3, and an output waveform generation unit c4. As shown in the time chart of FIG. 4, the initial values of the counters c2 and c3 are both zero, and the count-up operation is performed in synchronization with the clock signal of the clock circuit c1. The counter c2 is given a set value A. When the count number reaches the set value A, the count number is reset to zero, starts counting up again from zero, and repeats this series of operations. The counter c3 is given a set value B smaller than the set value A. When the count number reaches the set value B, the count number is reset to zero and the count-up is stopped. When the counter c2 is reset, the count-up operation starts again from zero in synchronization with the counter c2, and this series of operations is repeated. The output waveform generator c4 monitors the operations of the counters c2 and c3, and outputs a high level voltage when the counter c3 is counting up, and outputs a low level voltage when the counting up of the counter c3 is stopped. To output a pulse-width modulated waveform. For example, if the clock frequency of the clock circuit c1 is Fck, the set value A = 1000, and the set value B = 500, the output waveform generator generates a rectangular wave with a cycle T = 1000 / Fck and a high level time ratio of 0.5. The pulse width is digitally set by the set value A and the set value B, and pulse width modulation is performed by controlling this.

  In the switching power supply device 10, the switching frequency is set to 500 kHz, and the current control pulse generating means 32 is set so as to perform pulse width modulation at a constant frequency of, for example, about 10 kHz from the viewpoint of preventing frequency interference with the switching frequency. Accordingly, a constant set value A corresponding to about 10 kHz is set from the calculation unit 32b. Further, the calculation unit 32b determines the set value B based on the target value C that changes according to information from the temperature detection unit 24, the input voltage detection unit 26, the output voltage detection unit 28, etc., and outputs the set value B to the pulse generation unit 32c. The pulse generation unit 32c changes the count number for resetting the counter c3 based on the set value B. In this way, the current control pulse generating means 32 generates a predetermined current control pulse voltage Vc subjected to pulse width modulation, and outputs it to the subsequent pulse smoothing circuit 34.

  As shown in FIG. 1, the pulse smoothing circuit 34 charges the capacitor C1 through the resistor R1 when the current control pulse voltage Vc is at a high level, and charges the capacitor C1 through the resistor R1 when the current control pulse voltage Vc is at a low level. To discharge. As a result, the smoothed voltage Vb generated at both ends of the capacitor C1 is substantially a DC voltage, and increases when the high-level time ratio (on duty) of the current control pulse voltage Vc increases, and decreases when the high-level time ratio decreases. descend.

  As shown in FIG. 1, the current detection circuit 38 is a current detection resistor R0 inserted in a path through which the switching current Isw flows. In the switching power supply device 10, the switching current Isw is a pulse current that repeats a substantially trapezoidal shape, and the relationship between the peak value Iswp and the output current Iout uses the primary winding number N1 and the secondary winding number N2 of the transformer T1. In this case, Iswp≈ (N2 / N1) × Iout. That is, the peak value of the current detection voltage V (R0) generated in the current detection resistor R0 is a value approximately proportional to the output current Iout. Thus, in the switching power supply device 10, the output current Iout is detected by observing the switching current Isw through the current detection circuit 38. When the output current Iout flows, the current detection voltage V (R0) is generated in a direction in which the side connected to the source terminal of the switching element TR1 becomes a high potential. The output of the current detection circuit 38 is connected to the ground of the pulse averaging circuit 34 at its high potential terminal.

  As shown in FIG. 1, the current limit signal generation circuit 36 compares the reference voltage Vr, which changes according to the smoothed voltage Vb, with the current detection voltage V (R0) by the comparator CP1, and the current detection voltage V (R0). Is lower than the reference voltage Vr, that is, when the switching current Isw is smaller than a predetermined value, a high level is output. Conversely, when the current detection voltage V (R0) is higher than the reference voltage Vr, that is, when the switching current Isw is larger than a predetermined value, a low level is output. Further, the relationship between the smoothing voltage Vb and the reference voltage Vr is configured such that when the smoothing voltage Vb increases, the reference voltage Vr also increases, and conversely, when the smoothing voltage Vb decreases, the reference voltage Vr also decreases. .

As shown in FIG. 5, the drive pulse generation circuit 18 has a sawtooth wave generation circuit 18 b that generates a sawtooth wave voltage V (osc) and a sawtooth wave voltage V (osc) input to an inverting terminal. The output voltage control signal V (vol) input from the terminal a1 which is an output is provided with a comparator CP11 input to the non-inverting terminal. Further, the output signal of the comparator CP11, a NAND circuit NAND11 in which the current limit signal is input from the terminal a2 V (cur) is input, the output signal of the NAND circuit NAND11 is inputted to the reset terminal R, transmitter A trigger signal generated by the OS 11 is input to the set terminal S, and a known set-reset flip-flop FF11 (hereinafter referred to as RS-FF11) in which the output terminal Q is connected to the terminal a0 of the drive pulse generation circuit 18. A signal selection circuit 18a is provided. The signal output from the output terminal Q of the RS-FF 11 becomes a driving pulse Vg for driving the switching element TR1, and is input to the gate which is the driving terminal of the switching element TR1 via the terminal a0.

  The sawtooth wave generation circuit 18b includes a constant voltage DC power supply Vcc11, a charging resistor R11 having one end connected to the DC power supply Vcc11, a timer capacitor C11 connected between the other end of the charging resistor R11 and the ground, and a timer capacitor. A reset element S11 connected to both ends of C11 and an oscillator OS11 for controlling the reset element S11 are provided, and a voltage V (osc) generated at both ends of the timer capacitor C11 is output.

  The oscillator OS11 generates an impulse trigger pulse. The trigger pulse is a repetitive pulse having a constant period, and determines the switching frequency of the switching element TR1 and the turn-on timing of the switching element TR1. Further, the reset element S11 short-circuits both ends of the timer capacitor C11 when a trigger pulse is input, and is instantaneously opened, and continues to be open until the next trigger pulse is input.

  The drive pulse generation circuit 18 configured as described above operates as follows. First, the sawtooth wave generation circuit 18b receives a trigger pulse from the oscillator OS11, the reset element S11 short-circuits both ends of the timer capacitor C11, and the electric charge is discharged to V (osc) ≈0. Further, the reset element S11 is instantaneously opened, the timer capacitor C11 is charged by the charging current supplied through the charging resistor R11, and V (osc) increases. At this time, the time constant of the series circuit of the charging resistor R11 and the timer capacitor C11 is set to a sufficiently large value as compared with the period of the trigger pulse, so V (osc) increases linearly with a substantially constant slope. To do. Thereafter, when the next trigger pulse is input, the reset element S11 is short-circuited and the above operation is repeated. By such an operation, the sawtooth voltage V (osc) is generated at both ends of the timer capacitor C11.

  The comparator CP11 outputs a high level when the output voltage control signal V (vol) is higher than the sawtooth voltage V (osc), and outputs a low level in the opposite case. The current limiting signal V (cur) input from the terminal a2 is input to one input terminal of the NAND circuit NAND11, and the output signal of the comparator CP11 is input to the other input terminal. As a result, when the signal from the comparator CP11 is at a low level or one of the current limit signals V (cur) is at a low level, the output of the NAND circuit NAND11 becomes a high level. The output of the output terminal Q becomes high level by the impulse-like trigger pulse of the oscillator OS11 input to the set terminal S, and the RS-FF 11 is output when a high level signal is input from the NAND circuit NAND11 to the reset terminal R. The output of terminal Q is inverted to low level. A signal output from the output terminal Q of the RS-FF 11 is a drive pulse Vg input to the drive terminal of the switching element TR1. That is, the switching element TR1 is turned on while the output terminal Q is outputting a high level, and the switching element TR1 is turned off while the output terminal Q is outputting a low level.

Next, a series of operations of the switching power supply device 10 configured as described above will be described based on the time chart of FIG. First, period 1 is a period in which the overcurrent protection circuit is not operating. The overcurrent set value is controlled by the target value determined by the target value setting unit 32a of the current control pulse generating means 32. The target value setting unit 32a has an arithmetic expression using the temperature signal output from the temperature detection unit 24, the input voltage signal output from the input voltage detection unit 26, and the output voltage signal output from the output voltage detection unit 28 as parameters. The target value is determined based on the arithmetic expression. For example, if the temperature dependence of the voltage VBE generated between the base and emitter of a bipolar transistor, which is one of the circuit components, affects the overcurrent setting value, the temperature coefficient (about -2 mV / ° C) is expressed in the calculation formula. It is included. Further, when the fluctuation of the input voltage Vin has a non-negligible influence on the substantially proportional relationship between the peak value Iswp of the switching current and the output current Iout, a coefficient related to the input voltage Vin is included in the arithmetic expression. When the overcurrent protection circuit operates, the output voltage decreases as will be described later. When such a change in the output voltage has a non-negligible influence on the substantially proportional relationship between the peak value Iswp of the switching current and the output current Iout, a coefficient related to the output voltage Vout is included in the arithmetic expression. In this manner, the target value is appropriately changed based on the usage state and the operating state of the switching power supply device 10, and the current control pulse voltage Vc, the smoothing voltage Vb, and the reference voltage Vr are changed based on the target value. As a result, the overcurrent set value is automatically adjusted.

  In period 1, since the output current Iout is a small value equal to or less than the overcurrent set value, the current limit signal V (cur) output from the current limit signal generation circuit 36 is maintained at a high level, and the NAND of the drive pulse generation circuit 18 is maintained. A high level signal is continuously input to one input terminal of the circuit NAND11. Therefore, the drive pulse Vg generated at the output terminal Q of the RS-FF 11 has the same logic as the output signal of the comparator CP11, and the drive pulse Vg is exclusively generated by the error amplification circuit 16 as shown in the time chart of FIG. The pulse width is controlled based on the output voltage control signal V (vol) to be output.

Thus, in the switching power supply device 10, the on-duty of the drive pulse Vg of the switching element TR1 is controlled so that the output voltage Vout becomes equal to the reference voltage Vref in the error amplifier circuit 16, and the output voltage Vout is kept constant. It is. At this time, the output voltage Vout is determined as shown in Equation (1).

  Here, N1 is the number of primary turns of the transformer T1, N2 is the number of secondary turns of the transformer T1, T is the switching period, and duty is the on-duty of the switching element TR1.

  In the next period 2, for example, the load electronic device malfunctions to a low impedance, the output current Iout increases, the overcurrent protection circuit operates, and the switching current Isw is set so that the output current Iout does not exceed the overcurrent set value. This is a state that is restricted.

  In this state, the output voltage Vout decreases and the output voltage control signal V (vol) increases. As a result, the period during which the output voltage control signal V (vol) is higher than the sawtooth voltage V (osc) becomes longer, and the output voltage control signal V (vol) is always higher than the sawtooth voltage V (osc). Then, the output of the comparator CP11 is always at a low level, the switching element TR1 is always turned on, and the switching current Isw can always flow.

  On the other hand, as the switching current Isw increases, the current detection voltage V (R0) generated in the current detection resistor R0 of the current detection circuit 38 increases, and the current limit signal V that is the output of the comparator CP1 of the current limit signal generation circuit 36. (Cur) goes low. Then, the input of the NAND circuit NAND11 becomes low level, and the output of the NAND circuit NAND11 changes from low level to high level. Accordingly, a reset signal is output from the NAND circuit NAND11 to the reset terminal R at the timing when the current limit signal V (cur) is inverted from the high level to the low level. The drive pulse Vg generated by the output terminal Q of the RS-FF 11 is inverted from the high level to the low level at the timing when the reset signal is input, thereby turning off the switching element TR1. As a result, the output current Iout is limited so as not to exceed the overcurrent set value, and the on-duty duty of the switching element TR1 becomes small, so that the output voltage Vout also decreases based on the equation (1).

In this state, the RS-FF 11 has a property of holding the state of the output terminal Q until the next set signal is input to the set terminal S. Therefore, when the output terminal Q becomes low level and the switching element TR1 is turned off, the switching current Isw does not flow and the current limit signal V (cur) is inverted to high level, but until the next period starts, the output terminal The state of Q (low level) is maintained, and the switching element TR1 maintains the off state. On the other hand, when the output voltage Vout decreases as described above, the error amplifier circuit 16 as an inverting amplifier is saturated to a high level, and the output voltage control signal V (vol) maintains the high level saturation voltage value.

  Accordingly, the drive pulse Vg generated at the output terminal Q of the RS-FF 11 is inverted to the low level in synchronization with the timing at which the current limit signal V (cur) is inverted to the low level, as shown in the time chart of FIG. The pulse width of the drive pulse Vg is controlled exclusively based on the current limit signal V (cur) output from the current limit signal generation circuit 36.

As described above, in the switching power supply device 10, the digital processor constituting the current control pulse generation means 32 may be anything that operates at a low frequency of about 10 kHz, and a low-cost general-purpose digital processor can be used. It is possible to realize a switching power supply device with excellent intelligence.

  In addition, the use state and operation state (environmental temperature, input voltage, output voltage state) in which the switching power supply device 10 is operating are monitored in real time, and the overcurrent set value is always automatically adjusted to a predetermined value based on the monitored state. As a result, even when an electronic device with a load is in an abnormal state, a burnout accident can be prevented and high safety can be provided. Furthermore, the safety of the switching power supply itself can be ensured without selecting large parts that have an excessive margin for the rated current and temperature as the power components that make up the power supply circuit. It can contribute to the conversion.

Next, a second embodiment of the switching power supply device according to the present invention will be described with reference to FIGS. In addition, the same structure as the said switching power supply apparatus 10 attaches | subjects the same code | symbol, and abbreviate | omits description. First, an outline will be described based on the overall circuit diagram of FIG. The switching power supply device 50 is connected to a DC input power supply Ein, and generates an AC voltage V2 by ON / OFF operation of the switching element TR1, and rectifies and smoothes the AC voltage V2 to obtain an output voltage Vout. A smoothing circuit 14 is provided, and an output voltage Vout terminal is connected to the load 22. The output voltage Vout is connected to an error amplifier circuit 16 of an inverting amplifier that outputs an output voltage control signal V (vol) obtained by amplifying the difference between the output voltage Vout and a predetermined reference voltage Vref. Further, a drive pulse generation circuit 18 that performs pulse width modulation according to the output voltage control signal V (vol) output from the error amplifier circuit 16 and outputs the drive pulse Vg toward the drive terminal of the switching element TR1 is provided. .

  In addition, a target value setting unit 32a that determines a target value based on an operation time excess signal output from a current limit operation time monitoring unit 52 described later, a calculation unit 32b that performs calculation processing based on the target value, and its calculation Based on the result, there is provided a current control pulse generator 54 including a pulse generator 54c that generates a current control pulse voltage Vc for controlling the output current. The output of the current control pulse generator 54 is connected to the input of the pulse smoothing circuit 56. In the pulse generator 54c, the switch element S41 and the switch element S42 are connected in series between the power supply voltage Vcc41 and the ground, and the middle point of the two switch elements is connected to the pulse smoothing circuit 56. The two switch elements are controlled by signals output from the calculation unit 32b. The pulse generator 54c performs an operation different from that of the pulse generator 32c in the switching power supply device 10 described above. Details will be described later.

  The pulse smoothing circuit 56 forms an integrating circuit with the first resistor R21 and the capacitor C21, and a series circuit of a diode D21 and a second resistor R22 is connected in parallel with the first resistor R21. At that time, the cathode of the diode D21 is connected to the output terminal e0 of the current control pulse generating means 54. The outputs of the pulse smoothing circuit 56 are both ends of a capacitor C21 that generates a smoothed voltage Vb, and is connected to a current limit signal generating circuit 58 described later.

  A current detection circuit 38, which is a current detection resistor R0, is inserted between the source terminal of the switching element TR1 and the negative terminal of the input power source Ein, and a voltage corresponding to the switching current Isw flowing through the current detection resistor R0 is generated at both ends. . Both ends of the current detection resistor R0 are connected to the current limit signal generation circuit 58 as outputs of the current detection circuit 38.

  The current limit signal generation circuit 58 has a resistor R31 having one end connected to the output of the pulse smoothing circuit 56, a resistor R32 having one end connected to the other end of the resistor R31, and a collector terminal connected to the other end of the resistor R32. The transistor TR31 has an emitter terminal connected to the ground of the pulse smoothing circuit 56, and the base terminal of the transistor TR31 is connected to the middle point of the resistor R31 and the resistor R32. Further, a transistor TR32 having a base terminal connected to the collector terminal of the transistor TR31, an emitter terminal connected to the negative terminal side of the input power source Ein of the current detection resistor R0 is provided, and the collector terminal serves as an open collector to generate a current limiting signal. The output of the circuit 58 is formed. Transistors TR31 and TR32 are NPN transistors.

  The drive pulse generation circuit 18 is a circuit that performs pulse width modulation according to the input current limit signal V (cur) and outputs the drive pulse Vg of the switching element TR1, and its output is supplied to the drive terminal of the switching element TR1. It is connected.

The current limit operation time monitoring means 52 includes first comparison means 52a including a comparator to which the output of the error amplifier circuit 16 and the output of the current limit signal generation circuit 58 are input. Further, a timer circuit 52b whose operation is controlled by an output signal of the comparison means 52a and a second comparison means 52c composed of a comparator are provided. When the output of the comparison means 52a continues at a low level for a predetermined time, the comparison means 52c generates an operation time excess signal instead of the signal output before that, and the target value setting unit 32a of the current control pulse generation means 54 It is configured to output to

  Next, the configuration and operation details of each circuit block of the switching power supply device 50 will be individually described based on FIGS. The inverter circuit 12, the rectifying / smoothing circuit 14, the error amplifying circuit 16, the drive pulse generating circuit 18, and the current detecting circuit 38 are the same as those of the switching power supply device 10 described above, and thus description thereof is omitted.

  The current control pulse generation means 54 is configured using, for example, a general-purpose digital processor (microcomputer), and includes a target value setting unit 32a, a calculation unit 32b, and a pulse generation unit 54c as shown in FIG. When the operation time excess signal is input from the current limit operation time monitoring unit 52, the target value setting unit 32a performs arithmetic processing based on the operation time excess signal, and reduces the overcurrent set value to a predetermined low value. In this manner, the target value is determined and output to the calculation unit 32b. When the operation time excess signal is not input from the outside, the target value is determined based on a predetermined default value that is fixedly stored in advance.

  The calculation unit 32b performs calculation processing based on the target value input from the target value setting unit 32a, and outputs a control signal to each of the switch elements S41 and S42 of the pulse generation unit 54c. The switch element S42 on the ground side of the pulse generation unit 54c is turned on for a predetermined period from when the operation time excess signal is output from the current limiting operation time monitoring unit 52, and is always in an off state during other periods. When the switch element S42 is on, the switch element S41 is off. On the other hand, when the switch element S42 is off, the switch element S41 repeats on / off operations.

  Therefore, in the period when the switch element S42 is off, when the switch element S41 is on, a high level voltage (Vcc41) is generated at the output terminal e0. When the switch element S41 is off, the output terminal e0 is in the floating state. And electrically disconnected from the pulse generator 54c. Further, during the period in which the switch element S42 is on, the switch element S41 is in the off state, and the output terminal e0 is at the ground potential (low level).

  When the output terminal e0 of the pulse generator 54c is at a high level, the pulse smoothing circuit 56 slowly charges the capacitor C21 through the resistor R21, and the smoothing voltage Vb gradually increases. When the output terminal e0 is floating, the charge of the capacitor C21 is gently discharged through the discharge path of the circuit portion connected in parallel to the capacitor C21, and the smoothing voltage Vb is gradually decreased. Therefore, when the time ratio at which the output terminal e0 of the current control pulse generator 54 becomes high level increases, the smoothing voltage Vb increases. Conversely, when the time ratio at which the output terminal e0 becomes high level decreases, the smoothing voltage Vb increases. Vb decreases. In the switching power supply device 50 of this embodiment, since the input side impedance of the current limiting signal generation circuit 58 also serves as the discharge path, a dedicated resistor or the like that forms the discharge path is not particularly connected. .

On the other hand, when the output terminal e0 is at the ground potential, the charge of the capacitor C21 is suddenly discharged through the series circuit of the resistor R22 and the diode D21 in addition to the resistor R21, and the smoothing voltage Vb is sharply reduced. That is, when the operation time excess signal is output from the current limit operation time monitoring means 52, the smoothing voltage Vb can be sharply lowered, thereby enabling the overcurrent set value described later to be switched at high speed. It becomes possible.

The operation of the current limit signal generation circuit 58 will be described with reference to FIG. The transistor TR31 operates in an active state, and a voltage VBE1 generated by a base-emitter current is generated between the base and the emitter. Therefore, the current I31 flowing through the resistor R31 is expressed as in Expression (2).

Here, if the current amplification factor hFE of the transistor TR31 is sufficiently large, the current I32 flowing through the resistor R32 is equal to the current I31, and the voltage Vr generated at the resistor R32 is expressed as shown in Expression (3).

As can be seen from Equation (3), in the current limit signal generation circuit 58, the voltage Vr generated in the resistor R32 can be controlled by the smoothed voltage Vb, and the role of the reference voltage Vr in the current limit signal generation circuit 36 described above. Fulfill.

Further, the voltage Vce1 generated between the collector and the emitter of the transistor TR31 is expressed as in Expression (4).

Also, assuming that the voltage applied between the base and emitter of the transistor TR32 is Vbe2, the switching current Isw flows through the current detection resistor R0, and the voltage V (R0) is generated, the voltage applied to Vbe2 is It is expressed as equation (5).

Here, when the voltages VBE1 and VBE2 generated by the base-emitter currents of the transistors TR31 and TR32 exhibit the same characteristics, and the voltages at which the transistors TR31 and TR32 can be turned on are equal to VBE, the current detection voltage V (R0 ) Becomes equal to the reference voltage Vr, Vbe2 = VBE, and the transistor TR32 can be turned on. For example, when the switching current Isw = 0, the voltage Vbe2 applied between the base and emitter of the transistor TR32 cannot be turned on because it is lower than the base-emitter voltage VBE as shown in the equation (5). However, when the switching current Isw increases and the current detection voltage V (R0) reaches the reference voltage Vr, the voltage Vbe2 applied between the base and the emitter of the transistor TR32 is a voltage between the base and the emitter at which the TR32 can be turned on. VBE can be reached and turned on.

  Thus, the current limit signal generation circuit 58 includes the reference voltage Vr that changes according to the smoothed voltage Vb. Further, the current detection voltage V (R0) and the reference voltage Vr are compared via the base-emitter voltage when the transistors TR31 and TR32 are turned on. When the current detection voltage V (R0) is lower than the reference voltage Vr, That is, when the output current Iout is small, the collector terminal of the transistor TR32 outputs a high level. Conversely, when the current detection voltage V (R0) reaches the reference voltage Vr, that is, when the output current Iout reaches a predetermined value, a low level is output.

  Further, the relationship between the smoothing voltage Vb and the reference voltage Vr is configured such that when the smoothing voltage Vb increases, the reference voltage Vr also increases, and conversely, when the smoothing voltage Vb decreases, the reference voltage Vr also decreases. As shown in FIG. 5, in the drive pulse generation circuit 18, a pull-up resistor R12 is connected between the terminal a2 to which the output of the current limit signal generation circuit 58 is connected and the power supply voltage Vcc11. The configuration is compatible with collector type output.

The comparison means 52a provided in the current limit operation time monitoring means 52 includes an output voltage control signal V (vol) output from the error amplifier circuit 16, and a current limit signal V (cur) output from the current limit signal generation circuit 58. In normal operation, the current limit signal V (cur) is higher and outputs a high level, and when the current limit signal V (cur) is lower, it outputs a low level.

  In the timer circuit 52b, when the comparing means 52a outputs a low level, the reset element TR51 is turned off, and charging of the timer capacitor C51 is started. When the voltage across the timer capacitor C51 becomes higher than the predetermined reference voltage Vrr, the comparison means 52c changes from the high level signal output before that to the low level, and this is the current control pulse as the operation time excess signal. Output toward the target value setting unit 32a of the generating means 54.

  When the comparison means 52a outputs a high level, the reset element TR51 is on, so that the timer capacitor C51 is not charged and the timer circuit 52b does not operate.

Next, a series of operations of the switching power supply device 50 configured as described above will be described based on the time chart of FIG. Period 1 is a period in which the overcurrent protection circuit is not operating. Period 2 is a period in which the overcurrent protection circuit is operating according to the first overcurrent set value. Period 3 is a period in which the overcurrent protection circuit operates according to the second overcurrent setting value. Here, the operations in the period 1 and the period 2 are the same as the operation of the switching power supply device 10 described above, and thus the description thereof is omitted.

As described above, when the period 2 starts, the current limit signal V (cur) generated at the output of the current signal generation circuit 58 is lower than the output voltage control signal V (vol) generated at the output of the error amplifier circuit 16. Will continue. These two signals are input to the comparison means 52a of the current limit operation time monitoring means 52, and the comparison means 52a continuously outputs a low level. Then, the reset element TR51 is switched from an on state to an off state by a low level signal output from the comparison unit 52a. As a result, the timer capacitor C51 is charged from the DC power supply Vcc51 via the charging resistor R51, and the voltage across the timer capacitor C51 rises. The voltage across the timer capacitor C51 is compared with a predetermined reference voltage Vrr by the comparator 52c. When the voltage across the timer capacitor C51 becomes higher than the reference voltage Vrr, the voltage is output to the output of the comparator 52c before that. An operation time excess signal is generated instead of the digital signal. The length of the period 2 is set to an allowable upper limit value Tmax of the time during which the switching power supply device 50 can supply the peak current to the load. The power supply voltage Vcc51, the charging resistor R51, the timer capacitor C51, and the reference voltage It can be set as appropriate according to Vrr. In this way, the current limit operation time monitoring unit 52 reaches the allowable upper limit value Tmax for the overcurrent protection operation time by the first overcurrent set value toward the target value setting unit 32a of the current control pulse generation unit 54. A signal is sent informing you of the fact.

  The current control pulse generator 54 is controlled by the target value determined by the target value setting unit 32a. When the operating time excess signal is input, the target value setting unit 32a changes the overcurrent setting value from the first setting value to the second setting value, so that the second target value is set based on the operating time excess signal. Determine by calculating. Since the second overcurrent set value is set in consideration of the safety of a failed electronic device or the like, the second set value is set smaller than the first set value. When the second target value is input to the calculation unit 32b, the high-level time ratio of the current control pulse voltage Vc is reduced based on the second target value, the smoothing voltage Vb is reduced, and the reference voltage Vr is reduced. As a result, the overcurrent set value is changed to a second set value that is smaller than the first set value, and the output current Iout is limited so as not to exceed the second set value.

  Further, in the case of the switching power supply device 50, in order to switch the first overcurrent set value to the second set value at high speed, it is necessary to sharply lower the smoothing voltage Vb. Therefore, a series circuit of a resistor R22 and a diode D21 is used for the switch element S42 of the pulse generator 54c and the pulse smoothing circuit 56. When the switch element S42 is not used in the pulse generation unit 54c, the smoothing voltage Vb decreases only at a moderate speed because the resistance value of the discharge path of the capacitor C21 is large. Therefore, the switch element S42 is turned on for a very short time to switch the first overcurrent set value to the second set value, and a discharge path is provided by the resistor R22 and the diode D21 having a small resistance value. As a result, as shown in FIG. 9, the smoothing voltage Vb can be sharply lowered, and the overcurrent set value can be switched at high speed.

  As described above, the current control pulse generating means 54 of the switching power supply device 50 can achieve sufficient performance even if it is constituted by a general-purpose digital processor with a low-speed clock. Further, the current limit operation time monitoring means 52 can also be improved by using a general-purpose digital processor. As described above, in the switching power supply 50 of this embodiment, the switching power supply provided with an overcurrent protection circuit with excellent intelligence and high accuracy can be obtained without using a high-performance and expensive digital processor. Can be provided.

  In addition, when a load electronic device is in an abnormal state by controlling the first and second overcurrent setting values and the overcurrent protection operation time with precision corresponding to the device through which the peak current flows However, it can provide a high level of safety to prevent burning accidents. Furthermore, the safety of the switching power supply itself can be ensured without selecting large parts that have an excessive margin for the rated current and temperature as the power components that make up the power supply circuit. It can contribute to the conversion.

Next, a modification of the switching power supply device 10 according to the first embodiment of the present invention will be described with reference to FIG. Here, the current detection circuit 38 which is the current detection resistor R 0 is inserted on the negative side of the capacitor Co and the negative side of the load 22 of the rectifying and smoothing circuit 14. In this modification, since the output current Iout flows through the current detection resistor R0, the output current Iout is detected more accurately than when the output current Iout is detected using the switching current Isw flowing through the switching element TR1 as a substitute characteristic. be able to. The switching current Isw is a substantially trapezoidal pulse current repeated at the switching frequency, whereas the output current Iout is a direct current or a current that fluctuates at a relatively low speed. Since this is the same as that of the switching device 10, the description thereof is omitted.

  In addition, this modification is a form suitable for a non-insulated switching power supply device or the like. However, even in the case of an insulation type switching power supply device, there is no problem in applying this modification as long as the insulation between the primary circuit and the secondary circuit of the power supply device can be secured even in the overcurrent protection circuit.

Next, a modification of the switching power supply device 50 according to the second embodiment of the present invention will be described with reference to FIG. Here, the current detection circuit 38, which is the current detection resistor R0, is replaced with a current detection circuit 60 that observes the switching current Isw using the current transformer T2. In the current detection circuit 60, the primary side winding T2a of the current transformer T2 is connected in series with the switching element TR1, and the series circuit of the rectifier diode D61 and the current detection resistor R0 is connected in parallel to the secondary side winding. Both ends of the detection resistor R0 are connected to the current limit signal generation circuit 58 as outputs of the current detection circuit 60. In this case, a current obtained by multiplying the switching current Isw by the turn ratio of the current transformer T2 flows through the current detection resistor R0, and a current detection voltage V (R0) having a value obtained by further multiplying the current by R0 is output. In this modification, the ground of the pulse smoothing circuit that outputs the smoothed voltage Vb may be connected to the emitter terminal side of TR32. In this case, the current detection voltage V (R0) is set to a sufficiently small value as compared with the smoothing voltage Vb, and thus operates in the same manner as the switching power supply device 50 described above. Even in a circuit using the current transformer T2, a circuit in which the ground of the pulse smoothing circuit is connected to the emitter terminal side of the TR31 can be configured as in the switching power supply device of the second embodiment. .

Next, a modified example of the drive pulse generation circuit 18 of the switching power supply device of the present invention will be described with reference to FIG. In the drive pulse generation circuit 62 of this modification, since the charging resistor R11 of the sawtooth wave generation circuit 62b is connected to the input voltage Vin, the slope at which the sawtooth wave voltage V (osc) rises is proportional to the input voltage. Change. As a result, feedforward control corresponding to the fluctuation of the input voltage is added to the feedback control of the output voltage, and the high-speed response of the output voltage stabilization control is improved.

  Further, a voltage generating element Rb that generates a voltage proportional to the input voltage is inserted between the timer capacitor C11 and the charging resistor, and the total value of the voltage generated in the timer capacitor C11 and the voltage generated in the voltage generating element Rb is calculated. A sawtooth voltage V (osc) is input to the comparator CP11. As a result, a DC voltage corresponding to the input voltage Vin is superimposed on the sawtooth voltage V (osc), the delay time of the comparator CP11 is corrected according to the input voltage Vin, and the feedforward control characteristics are improved, The high speed response of the voltage stabilization control is further improved.

The signal selecting circuit 62a includes a terminal a1 to the output voltage control signal V (vol) is input, directly and a2 terminal current limit signal V (cur) is input, and to simplify the circuit configuration . The signal selection circuit 62a only needs to select a signal so that the pulse width modulation of the drive pulse Vg is performed based on the movement of the current control signal V (cur) when an overcurrent is detected. Even when the signal selection circuit 62a is used, the same operation as that of the drive pulse generation circuit 18 described above is performed.

However, since the signal selection circuit 62a directly connects the output of the error amplifier circuit 16 and the output of the current limit signal generation circuit 36 of the first embodiment or the current limit signal generation circuit 58 of the second embodiment, switching is performed. The present invention is not applicable to the configuration provided with the current limiting operation time monitoring means 52 of the power supply device 50. This is because the comparison means 52a of the current limiting operation time monitoring means 52 cannot detect the operating state of overcurrent protection. Therefore, when the signal selection circuit 62a is used, for example, the comparison unit 52a of the current limiting operation monitoring unit 52 monitors the output voltage Vout and outputs a low level when the output voltage Vout drops below a predetermined voltage. A method of configuring the above may be used. This is because the state in which the output voltage Vout drops below a predetermined voltage is a state in which the current limit signal V (cur) repeatedly generates a low level for each switching frequency, or a state in which the low level continues. Since it can be regarded as being, it is a method of monitoring as a substitute characteristic.

Furthermore, a modified example of the error amplifier circuit 16 of the switching power supply device of the present invention will be described with reference to FIG. The error amplifier 64 includes an A / D converter 64a that converts the output voltage Vout into a digital signal, control pulse generation means 64b that generates a control pulse based on the digital signal, and output voltage control by smoothing the control pulse. A pulse smoothing circuit 64c that outputs a signal V (vol) is provided. According to the error amplifier 64, the error amplification circuit 16 can be configured using a low-cost general-purpose digital processor with a low speed and a low clock frequency, and a low-cost and intelligent switching power supply device can be realized. can do.

Next, an embodiment of an initial setting method in the initial driving of the switching power supply device of the present invention will be described with reference to FIGS. First, each step shown in the flow of FIG. 14 will be described.

  Step S81 is a step of starting by supplying an input voltage to the switching power supply device 10. Thus, the switching power supply device 10 is placed in a standby state in which the switching element performs a switching operation to output a predetermined output voltage Vout and can supply the output current Iout.

Step S82 is a step of connecting the load device to the output of the switching power supply device 10 and setting the load device so that the output current Iout equal to the desired overcurrent set value is output. The load device used here is preferably a current control type electronic load device, for example. According to the electronic load, the output voltage regardless of the state of Vout, it is possible to constant output current Iout is equal to the overcurrent setting value to control the load state to be output.

  Step S83 is a step of changing a predetermined fixed coefficient K set in the calculation program of the target value setting unit 32a provided in the current control pulse generating means 32 and continuously changing the target value output to the calculation unit 32b. It is. Note that when the fixed coefficient K is continuously varied, a region where the output voltage Vout is kept constant and a region where the output voltage Vout continuously changes are generated.

  Step S84 is a step of extracting a fixed coefficient boundary value K (lim) that is a boundary point between the two regions.

  Step S85 is a step of storing the extracted fixed coefficient boundary value K (lim) in the arithmetic program as a set value of the fixed coefficient.

  The operation of the initial setting method of the present embodiment described above will be described with reference to FIGS. FIG. 15A is an example of overcurrent protection characteristics of the switching power supply device 10. The output voltage Vout is controlled to a predetermined constant value by the error amplifier circuit 16 when the output current Iout is small. When the output current Iout increases and reaches the overcurrent set value, the output current Iout is limited by the current limit signal generation circuit 36, and the output voltage Vout also decreases at the same time. Here, the change in the fixed coefficient K is a change in the overcurrent set value. When the fixed coefficient K changes, the target value determined by the target value calculation circuit changes, the reference voltage Vr of the current limit signal generation circuit 36 changes, and as a result, the overcurrent set value changes. The initial setting method of the present embodiment is a method for setting the overcurrent set value to a desired Iocp using the fixed coefficient K.

In step S81, the switching power supply device enters a standby state in which output current can be supplied. Next, in step S82, the current setting of the electronic load device is set to a desired overcurrent setting value Iocp. At this time, since the fixed coefficient K is an arbitrary value, as shown in FIG. 15B, what value the output voltage Vout indicates is undefined.
The arithmetic expression set in the target value setting unit can be expressed as, for example, Expression (6).

  C is a target value, T24 is an output signal of the temperature detection means 24, V26 is an output signal of the input voltage detection means 26, V28 is an output signal of the output voltage detection means 28, K is a fixed coefficient, and f is a predetermined parameter. Is a function of The fixed coefficient K is normally fixed to a predetermined value when the switching power supply device 10 is used. In step S83, the fixed coefficient K is changed, and thereby the target value is continuously changed. Use as a means.

  When the fixed coefficient K is continuously varied in step S83, as shown in FIG. 15B, there are a region where the output voltage Vout maintains a constant value and a region where it continuously changes. Then, a process of extracting a fixed coefficient boundary value K (lim) that is a boundary point between the two regions is performed.

In step S84, the fixed coefficient K of the arithmetic expression of the target value setting unit is fixedly set and stored in K (lim). Thus, the overcurrent characteristic of the switching power supply 10 is fixed to the characteristic when K = K (lim) shown in FIG. 15A, and the initial setting is completed.

As described above, according to the initial setting method of the control characteristics of the switching power supply device of this embodiment , for example, the inductance value of the transformer T1, the resistance value of the current detection resistor R0 of the current detection circuit 38, and the current limit signal generation circuit Even if the overcurrent set value varies due to individual differences in the characteristics of the electronic components constituting the circuit, such as the accuracy of the comparison, all can be adjusted to the desired overcurrent set value. At that time, complicated hardware adjustment operations such as component replacement and semi-fixed resistor adjustment are not necessary, adjustment is possible only by software processing, and automation is also easy.

  In addition, this invention is not limited to the said embodiment, It can apply also to a push pull system, a bridge system, a flyback system, various chopper systems, etc.

The current limiting operation time monitoring means may be a method of monitoring the output voltage Vout using a digital processor constituting the current control pulse generating means. The A / D conversion circuit used for the temperature detection means, the input voltage detection means, and the output voltage detection means may also use a built-in function in the digital processor. In addition, the arithmetic expression set for each arithmetic processing may be any expression that accurately formulates the operating characteristics, properties, etc. of the switching power supply device, and is derived by engineering circuit network analysis or the like. Any of a theoretical formula, an empirical formula derived by a test / experiment, etc., or an empirical formula combining these and an empirical rule may be used.

  Further, in the above initial setting method, a plurality of fixed coefficients K may be set, and a series of steps may be performed a plurality of times under different conditions. For example, the first fixed coefficient K may be adjusted in an environment at an ambient temperature of 25 ° C., and the second fixed coefficient K may be adjusted in an environment at an ambient temperature of 50 ° C. The first fixed coefficient K may be adjusted by setting the input voltage to the lowest value in the allowable range, and the second fixed coefficient K may be adjusted by setting the input voltage to the highest value in the allowable range. Good. As a result, the influence of individual differences in the characteristics of the electronic components constituting the circuit can be eliminated in various ways under different conditions, and the overcurrent characteristics of the switching power supply device can be more accurately matched to the desired characteristics. .

1 is a block diagram showing a first embodiment of a switching power supply device of the present invention. It is a functional block diagram which shows embodiment of the temperature detection means (a) , input voltage detection means (b) , and output voltage detection means (c) with which this 1st embodiment is provided. It is a functional block diagram of the pulse generation part with which this first embodiment is provided. It is a time chart which shows the actual operation | movement of the pulse generation part with which this 1st embodiment is provided. It is a functional block diagram which shows the drive pulse generation circuit with which this 1st embodiment is provided. It is a time chart which shows the actual operation | movement of the drive pulse generation circuit with which this 1st embodiment is provided. It is a block diagram which shows 2nd embodiment of the switching power supply device of this invention. It is a circuit diagram which shows the current limiting signal generation circuit with which this 2nd embodiment is provided. It is a time chart which shows the actual operation | movement of the current limiting signal generation circuit with which this 2nd embodiment is provided. It is a circuit diagram which shows other embodiment of the current detection circuit with which embodiment of the switching power supply device of this invention is provided. It is a circuit diagram which shows other embodiment of the current detection circuit with which embodiment of the switching power supply device of this invention is provided. It is a functional block diagram which shows other embodiment of the drive pulse generation circuit with which embodiment of the switching power supply device of this invention is provided. It is a functional block diagram of an error amplification circuit provided in an embodiment of a switching power supply device of the present invention. It is a flowchart which shows the initial setting method of embodiment of the drive method of the switching power supply device of this invention. It is a graph explaining the behavior of the output voltage of the switching power supply device in the target value variable processing and the boundary value extraction processing of the initial setting method which is the driving method of the switching power supply device of the present invention.

10, 50 Switching power supply device 12 Inverter circuit 16, 64 Error amplification circuit 18, 62 Drive pulse generation circuit 24 Temperature detection means 26 Input voltage detection means 28 Output voltage detection means 32, 54 Current control pulse generation means 32a Target value setting unit 32b Arithmetic units 32c and 54c Pulse generation units 34 and 56 Pulse smoothing circuits 36 and 58 Current limit signal generation circuits 38 and 60 Current detection circuit

Claims (12)

A drive pulse generation circuit that outputs a drive pulse that is pulse width modulated at a predetermined switching frequency, and an inverter circuit having a switching element that generates an AC voltage by intermittently applying a DC input voltage using the drive pulse from the drive pulse generation circuit And a rectifying and smoothing circuit that rectifies and smoothes the AC voltage to generate an output voltage and supplies an output current to the load.
A target value setting unit that outputs a changeable value that is a predetermined target value related to the control of the output current, a calculation unit that performs a calculation process related to the control of the output current based on the target value, and a calculation unit thereof Current control pulse generating means comprising a pulse generator for generating a current control pulse voltage for controlling the output current based on the calculation result;
A current detection circuit for detecting an output current from the rectifying / smoothing circuit or a current flowing through the switching element;
When the current detected by the current detection circuit exceeds a reference value set based on the output of the current control pulse generating means, a current limit signal is generated for outputting a current limit signal for limiting the current. With circuit,
The drive power generation circuit operates so as to stop or narrow the on-duty of the drive pulse for driving the switching element when the current limit signal is output. The current control pulse generating means comprises a digital processor configured to be able to output the current control pulse voltage as a pulse width modulated signal,
A pulse smoothing circuit for smoothing the current control pulse voltage to generate a smoothed voltage is connected,
The current detection circuit detects the output current or the current flowing through the switching element and outputs a current detection voltage based on the detected current.
The current limit signal generation circuit includes a comparison circuit that compares a reference voltage determined based on the smoothed voltage with the current detection voltage, and the current limit signal is output when the current detection voltage exceeds the reference voltage. Output signal,
2. The drive pulse generation circuit operates to stop or narrow an on-duty of a drive pulse for driving the switching element when the current limit signal is output. Switching power supply. The current limiting signal generation circuit includes a first resistor having one end connected to the output of the pulse smoothing circuit, a second resistor having one end connected to the other end of the first resistor, and a collector terminal A first NPN connected to the other end of the second resistor, a base terminal connected to a midpoint of the first resistor and the second resistor, and an emitter terminal connected to one end of the current detection circuit output. A transistor, a base terminal is connected to the collector of the first NPN transistor, an emitter terminal is connected to the other end of the current detection circuit output, and a collector terminal is a second NPN that is the output of the current limit signal generation circuit The first and second NPN transistors, one of the emitter terminals is connected to the ground potential,
When the output current of the rectifying / smoothing circuit flows or when the current flows through the switching element, the current detection circuit is configured such that the emitter terminal of the second NPN transistor is the emitter of the first NPN transistor. 3. The switching power supply device according to claim 2, wherein the current detection voltage is output in a direction in which the potential becomes lower than a terminal. Current limit operation time monitoring that outputs an overtime signal when the current limit signal is continuously generated by the current limit signal generation circuit or repeatedly generated for each switching frequency for a predetermined time or longer. With means,
The target value setting unit of the current control pulse generation means is detected by the current detection circuit by the current limit signal of the current limit signal generation circuit based on the time excess signal of the current limit operation time monitoring means. The target value is determined and output to the calculation unit so as to further limit the current, and the current control pulse voltage calculated by the calculation unit based on the target value is output from the pulse generation unit. The switching power supply device according to claim 1, 2, or 3. The calculation unit of the current control pulse generating means calculates a time ratio of the current control pulse voltage based on a target value output by the target value setting unit, and the pulse generation unit is calculated when the calculation unit calculates Configured to generate a constant frequency square wave having a ratio of high and low level voltages;
The pulse smoothing circuit includes a resistor having one end connected to the output of the current control pulse generating means, and a capacitor connected between the other end of the resistor and the ground, and the voltage across the capacitor is the pulse. 5. The switching power supply device according to claim 2, which is an output of a smoothing circuit. The calculation unit of the current control pulse generating means calculates a time ratio of the current control pulse voltage based on a target value output by the target value setting unit, and the pulse generation unit is calculated when the calculation unit calculates It is configured to repeat the high-level and floating-level output states within a certain period,
The pulse smoothing circuit includes a resistor having one end connected to the output of the current control pulse generating means, a capacitor connected between the other end of the resistor and the ground, and a discharge path connected in parallel to the capacitor. 5. The switching power supply device according to claim 2, wherein a voltage across the capacitor serves as an output of the pulse smoothing circuit. The pulse smoothing circuit is connected between a first resistor having one end connected to the output of the current control pulse generating means, the other end of the resistor and the ground, and an output voltage of the pulse smoothing circuit at both ends. And a discharge path connected in parallel to the capacitor, and connected to allow a current to flow from the connection point between the capacitor and the first resistor toward the output of the current control pulse generating means. A series circuit of a diode and a second resistor;
When the target value determined based on the operation time excess signal is given, the calculation unit of the current control pulse generation unit controls the output state of the pulse generation unit to a low level for a predetermined time, and the pulse 7. The switching power supply device according to claim 6, wherein the voltage of the capacitor of the smoothing circuit is discharged through a series circuit of the diode and the second resistor. An error amplification circuit that compares the output voltage of the switching power supply device with a predetermined target value and outputs an inverted and amplified output voltage control signal,
The current limit signal generation circuit compares a reference voltage determined based on the smoothed voltage with the current detection voltage, and differs depending on whether the current detection voltage exceeds the reference voltage or otherwise. Comparing circuit that outputs current limit signal
The drive pulse generation circuit includes a sawtooth wave generation circuit that generates a sawtooth wave voltage driven at the switching frequency, and the sawtooth wave voltage input to a first input terminal and the input to a second input terminal. A comparator that compares an output voltage control signal with the sawtooth voltage and outputs a different signal between a period when the sawtooth voltage is lower and a period other than that, and a signal selection circuit to which the current limit signal is input With
The signal selection circuit is determined based on the current limit signal when the current limit signal is continuously generated by the current limit signal generation circuit or repeatedly generated at each switching frequency. When a drive pulse of a ratio is output from the drive pulse generation circuit and the current limit signal is not output, a drive pulse of a time ratio determined based on the output voltage control signal is output from the drive pulse generation circuit 3. The switching power supply device according to claim 1, wherein an operation of selecting an output signal is performed. A temperature detection unit that senses the environmental temperature of the switching power supply device with a temperature sensing element and outputs a temperature signal based thereon is provided.
The target value setting unit of the current control pulse generating unit determines the target value based on the temperature signal, and outputs the determined target value to the calculation unit. The switching power supply device according to 3 or 4. An input voltage detecting means for detecting an input voltage of the switching power supply device and outputting an input voltage signal based on the detected input voltage,
The target value setting unit of the current control pulse generating means determines the target value based on the input voltage signal, and gives the determined target value to the arithmetic unit. The switching power supply device according to 3 or 4. An output voltage detecting means for detecting an output voltage of the switching power supply device and outputting an output voltage signal based on the detected output voltage;
The target value setting unit of the current control pulse generating means determines the target value based on the output voltage signal and gives the determined target value to the arithmetic unit. The switching power supply device according to 3 or 4. A drive pulse generation circuit that outputs a drive pulse that is pulse width modulated at a predetermined switching frequency, and an inverter circuit having a switching element that generates an AC voltage by intermittently applying a DC input voltage using the drive pulse from the drive pulse generation circuit A rectifying / smoothing circuit that rectifies and smoothes the AC voltage to generate an output voltage and supplies an output current to the load; and a target value that outputs a predetermined target value that can be changed with respect to the control of the output current A setting unit, a calculation unit that performs calculation processing related to output current control based on the target value and outputs a calculation result, and a pulse generator that generates a current control pulse voltage for controlling the output current based on the calculation result A current detection circuit that detects an output current of the rectifying / smoothing circuit or a current flowing through the switching element, and the current detection circuit More sensed current, the current control pulse voltage given switching power supply device including a current limit signal generating circuit for outputting a current limit signal for limiting the current when it exceeds the reference value set by In the driving method of
At the beginning of power supply operation, power supply start processing is performed by supplying a predetermined DC voltage to the input of the switching power supply device and starting the power supply,
Connect the load to the output and perform the load setting process to output the output current equal to the desired overcurrent setting value,
A target value variable process is performed in which a predetermined fixed coefficient set in a calculation program of the target value setting unit provided in the current control pulse generating means is adjusted, and the target value output to the calculation unit is continuously changed. ,
In the state where the output current is the overcurrent set value, the output voltage continuously changes in response to the change of the target value,
In response to a change in the target value, a boundary value extraction process is performed to extract a boundary value of the fixed coefficient at a boundary point where the output voltage is maintained at a constant value.
A boundary value storage process for storing the boundary value of the fixed coefficient as a setting value of the fixed coefficient is stored by the arithmetic program,
In the target value setting unit, the set value of the fixed coefficient stored by the boundary value storing process is used as a fixed coefficient set in the calculation program of the target value setting unit, and a new target value to be output to the calculation unit Obtaining the new target value to the calculation unit,
In the calculation unit, calculation processing related to the control of the output current is performed based on the new target value, and the calculation result is output to the pulse generation unit,
The pulse generator generates a current control pulse voltage for controlling the output current based on the calculation result,
The current limit signal generation circuit outputs a current limit signal for limiting the current when the current detected by the current detection circuit exceeds the reference value set based on the current control pulse voltage. Output,
The driving pulse generation circuit operates so as to stop or narrow the on-duty of the driving pulse according to the current limiting signal, or to narrow it.