JP6509190B2 - Transposition modulation system, method and apparatus - Google Patents

Description

  This application is a continuation-in-part of copending US patent application Ser. The present application is also directed to U.S. Provisional Patent Application No. 2003/01998, filed on March 15, 2013, Patent No. 3, filed on March 15, 2013, Patent No. 3 filed on March 15, 2013 4, the patent document 5 filed on March 15, 2013 and the patent document 6 filed on March 15, 2013 claim the priority of these, the contents of which are incorporated herein by reference. It has been incorporated.

  The present disclosure relates primarily to signal processing, and more specifically, to transpose and modulate input signals to increase the information bandwidth of defined communication channels and to perform time delay shifting of input signals according to input control signals. System, method, and apparatus for transmitting, receiving and demodulating a signal

  Existing transmission schemes, whether voice, video or data, have band limitations imposed by national and global regulatory agencies that control the use of the frequency spectrum. Carrier modulation schemes have evolved from initial amplitude modulation to current schemes that combine two or more carriers with various combinations of amplitude modulation, frequency modulation, or phase modulation. A number of advanced carrier modulation schemes have been developed to maximize energy across the allocated channel bandwidth and maximize available information bandwidth of the allocated communication channel.

  One new basic carrier modulation has been developed and first patented (see, for example, Vokac et al., U.S. Pat. No. 5,648,798, the entire contents of which are incorporated herein by reference), which are simultaneously present in the same carrier signal It applies a new type of carrier modulation that does not interfere with amplitude modulation, frequency modulation, and / or phase modulation.

  The concept of transposition (TM) modulation is based on the initial concept of how to add information to a carrier signal without affecting the amplitude, frequency or phase of the carrier signal (e.g., the entire contents of which are incorporated herein by reference). See Vokac et al., US Pat. By generating the inflection as will be shown later, it is possible to carry information in the carrier signal. This scheme is not detected with existing demodulators of amplitude modulation, frequency modulation or phase modulation.

  Using the previously patented generation scheme, the following time domain waveforms are generated (the inflection is exaggerated for clarity): When applied in practice, inflection is invisible.

  Early approaches to generating this type of waveform were imperfect in that there was a small amplitude change to be removed by the adjustment circuit. For example, FIG. 1 shows a TM modulated signal 100 generated in accordance with the prior art taught in US Pat. As can be seen in the figure, an amplitude change error exists between the negative peak 101 and the negative peak 102.

  Thus, there is still an unmet need in the industry to address the aforementioned deficiencies and deficiencies.

U.S. Patent Application Publication No. 2014/0269969 (U.S. Patent Application No. 13 / 841,889) US Provisional Patent Application No. 61 / 798,437 US Provisional Patent Application No. 61 / 794,786 US Provisional Patent Application No. 61 / 798,120 US Provisional Patent Application No. 61 / 794,942 US Provisional Patent Application No. 61 / 794,642 U.S. Pat. No. 4,613,974

  Embodiments of the present disclosure transmit and receive and demodulate transposed modulated signals to increase the information bandwidth of defined communication channels and to perform time delay shifting of input signals according to input control signals. Systems, methods, and apparatus for performing. In one embodiment, a method is provided for increasing the information bandwidth of a defined communication channel, the method comprising the steps of: receiving a first modulated signal having a first carrier signal frequency; Receiving a second modulated signal having a signal frequency, wherein the second modulated signal is modulated with information independent of the information modulating the first carrier signal, the second carrier signal The signal frequency comprises the steps of: receiving a second modulated signal harmonically or subharmonically related to the first carrier signal frequency; combining the first signal and the second signal; ,including.

  In another embodiment, a time shift modulator is provided which time shifts the input signal according to the input control signal. The time shift modulator includes an all pass filter that is corrected by voltage controlled time delay.

  In another embodiment, a method is provided for increasing the communication bandwidth in a fixed communication channel, the method comprising adding a second transposed modulated signal to a combined signal, the combined signal being a first The second transposed modulated signal, which includes the transposed modulated signal and the first fundamental carrier signal, has the same frequency as the first fundamental carrier signal and a phase angle relative to the first fundamental carrier signal. The adding step is added to the combined signal using a second base carrier signal that is 90 degrees.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasound communication, the method comprising the step of applying transpose modulation to the ultrasound communication signal by direct amplitude modulation of a single ultrasound transducer.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasonic communication, the method including ultrasonic communication by directly amplitude modulating the first ultrasonic transducer with the fundamental carrier signal component of transposition modulation. Applying transpose modulation to the signal, and directly amplitude modulating the second ultrasound transducer with the third harmonic carrier signal component of the transpose modulation.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasonic communication, the method comprising: transposing modulated base carrier signal and And adding the three harmonic carrier signal component to the ultrasound communication signal.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasound communication, the method comprising: angle modulating the first ultrasound transducer directly with the fundamental carrier signal component of transpose modulation; Applying transpose modulation to the ultrasound communication signal by directly angle modulating the transducer with the third harmonic carrier signal component of the transpose modulation.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasonic communication, the method comprising: transposing modulated base carrier signal and And adding the three harmonic carrier signal component to the ultrasound communication signal.

  In yet another embodiment, a system for increasing optical information communication bandwidth is provided. The system includes a light beam and a light modulator. The system is configured to modulate the light beam directly with the transposed modulation signal.

  In another embodiment, a method is provided for increasing the optical information communication bandwidth, the method comprising directly modulating a light beam of a first frequency with a transposed modulation base carrier frequency component.

  In yet another embodiment, a method is provided for increasing the optical information communication bandwidth, the method comprising directly modulating a light beam of a second frequency with a transpose modulated third harmonic component signal.

  Other systems, methods, features, and advantages of the present disclosure will become apparent to one with skill in the art upon review of the following drawings and detailed description. It is intended that all such additional systems, methods, features and advantages be included within this description, be within the scope of the present disclosure, and be protected by the accompanying claims.

  Many aspects of the disclosure will be better understood with reference to the following drawings. Elements in the drawings are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the present disclosure. Further, in the drawings, like reference numerals refer to corresponding elements throughout the several views.

FIG. 2 shows a TM modulation signal generated according to the prior art. 3 is a flow chart illustrating a method of modulating a carrier signal according to a first exemplary embodiment of the present disclosure. FIG. 7 illustrates a signal generated as a quarter period according to an embodiment of the present disclosure. FIG. 4 illustrates the signals shown in FIG. 3 after summing quarter cycles according to one embodiment of the present disclosure. FIG. 5 illustrates an input modulation signal that can be used in the embodiments provided by the present disclosure to generate the signal shown in FIG. 4; Fig. 5 is a plot showing the frequency spectrum of the signal shown in Fig. 4; FIG. 7 is a plot showing the resulting frequency spectrum as a result of heterodyned the third harmonic component of the signal shown in FIG. 6 with the second harmonic, according to one embodiment provided by the present disclosure. FIG. 6 illustrates a filter applicable to one embodiment provided by the present disclosure. FIG. 1 is a block diagram illustrating a software-based direct spectrum system that generates signals according to an embodiment provided by the present disclosure. FIG. 7 is a block diagram illustrating a sub-period calibration system that demodulates a signal, according to one embodiment provided by the present disclosure. FIG. 5 is a block diagram illustrating a third harmonic phase detection system for demodulating a signal, according to one embodiment provided by the present disclosure. FIG. 1 is a block diagram illustrating a fast Fourier transform based system for demodulating a signal according to one embodiment provided by the present disclosure. FIG. 5 is a block diagram illustrating a TM transmitter that generates and transmits a signal that consists of a TM signal added to an existing signal, according to one embodiment provided by the present disclosure. FIG. 7 is a block diagram illustrating an example implementation of a carrier signal generation portion of a TM transmitter, according to one embodiment provided by the present disclosure. FIG. 7 is a block diagram illustrating an example implementation of a TM modulated signal processing portion of a TM transmitter, according to an embodiment provided by the present disclosure. FIG. 5 is a block diagram illustrating a TM receiver that receives a signal having a TM signal added to an existing signal, and extracts and demodulates the TM signal, according to one embodiment provided by the present disclosure. FIG. 7 is a block diagram illustrating an example implementation of the carrier signal and harmonic recovery portion of a TM receiver, according to an embodiment provided by the present disclosure. FIG. 7 is a block diagram illustrating an example implementation of the TM separation and demodulation portion of a TM receiver, according to one embodiment provided by the present disclosure. FIG. 7 is a graph illustrating the frequency response behavior of a time delay based filter circuit of separation function according to one embodiment provided by the present disclosure.

  Many embodiments of the present disclosure may take the form of computer executable instructions, including programmable computer or microprocessor implemented algorithms. However, the present disclosure may be practiced with other computer system configurations as well. Some aspects of the present disclosure may be implemented in a dedicated computer or data processor programmed, configured, or constructed with the limitation of performing one or more of the methods or algorithms described below.

  The following aspects of the present disclosure include a removable computer disk that can read magnetic or optical data, a fixed magnetic disk, a floppy (registered trademark) disk drive, an optical disk drive, a magneto-optical disk drive, a magnetic tape, a hard disk drive (HDD May be stored or distributed on a computer readable medium including solid state drive (SSD), Compact Flash or non-volatile memory, and distributed electronically via a network including a cloud You may Data structures and data transmissions unique to aspects of the present disclosure are also within the scope of the present disclosure.

  FIG. 2 is a flow chart 200 illustrating a method of modulating a carrier signal according to a first exemplary embodiment of the present disclosure. It should be noted that any process description or block in the flowchart may be understood as representing a module, segment, code portion or step including one or more instructions for performing a specific logical function of the process. Alternative implementations may be performed in an order different from that shown or described, including that the functions are performed substantially simultaneously in the reverse order, depending on the functionality required Embodiments are also included within the scope of the present invention, which would be understood by one skilled in the art of the present invention. The method solves the problem of amplitude variation in the prior art (eg, as shown in FIG. 1 above) and may be implemented in hardware or software, or any combination thereof. The method illustrated in FIG. 2, sometimes referred to as a “quarter cycle collection” (QC) method, provides a look-up table (LUT) 210 as a quick way to obtain results without the need for continuous performance of operations. (Not based on the LUT 210, the result can be generated using arithmetic functions). The QC scheme is based on the time domain.

  Referring to FIG. 3, the modulated output signal 300 from the method shown in FIG. 2 includes four separate quarter period segments in each full signal period. FIG. 3 shows three full periods (eg, periods a, b and c), which may be output according to the quarter period scheme shown in FIG. Each cycle consists of four quarter cycle segments (eg, 301, 302, 303, and 304). Although a gap is shown between the quarter period segments, this is for illustrative purposes only. Furthermore, the inflection amplitude positions (a1, a2, b1, b2, c1, c2) are exaggerated for the purpose of illustration. These inflections are formed between adjacent quarter period segments as shown.

As shown in FIG. 3, the "first" quarter cycles (301a, 301b, and 301c) of each cycle may differ in amplitude depending on the modulation value applied. The same applies to the other quarters in each illustrated period. That is, the second quarter cycle (302a, 302b, 302c), the third quarter cycle (303a, 303b, 303c), and the fourth quarter cycle (304a, 304b, 304c) of each cycle are applied modulations. Depending on the value, the amplitude may be different. If the amplitude of the "first" quarter cycle (eg, 301a, 301b, 301c) in a cycle is low, then the "second" quarter cycle (eg, 302a, 302b, 302c) in that same cycle has complementary amplitudes. The reason is that a constant amplitude always exists between the negative peak value (Pk ₋ ) in the entire cycle and the positive peak value (Pk ₊ ) in the cycle. The same applies to the "third" and "fourth" quarter cycles in each cycle. Thereby, the positive peak value (Pk ₊ ) in each period is always the same. By making the negative peak value (Pk ₋ ) the same, the amplitude change due to the applied modulation value is eliminated.

  As further shown in FIG. 3, the “first” quarter period (301a, 301b, 301c) and the “third” quarter period (303a, 303b, 303c) in each period have the same amplitude. Similarly, the “second” quarter cycle (302a, 302b, 302c) and the “fourth” quarter cycle (304a, 304b, 304c) in each cycle have the same amplitude. This is done to make the area under the curve of each period the same regardless of the applied modulation value. This ensures that the average value of each period is zero and that no "DC" value shift occurs in the carrier signal due to the applied modulation value.

  However, it is noted that in some applications DC shifts may be tolerated, so that the area under the curves may not have to be the same, ie there may be no symmetry between periods. I want to. In such cases, information or "symbols" may be conveyed at a rate of 2 symbols per period. That is, two different inflection points may exist for each cycle (for example, one is located in the middle of the rising half cycle between the negative peak and the positive peak, and the other is the positive peak and the negative peak). Located in the middle of the falling half cycle).

  Each quarter period may be generated by a constant clock or time step, so that the frequency does not change from one period to the next as a result of the applied modulation value. Each inflection (a1, a2, b1, b2, c1, c2) occurs at an angle corresponding to 180 degree separation from just one half cycle to the next half cycle. This ensures that there is no phase change due to the applied modulation value.

  Summation of these quarter cycles (eg, as shown in FIG. 3) results in a smooth continuous waveform 300, as shown in FIG.

  FIG. 5 shows a TM modulation signal 500, which is used to generate the modulated signal 300 shown in FIG. As shown in FIGS. 4 and 5, there is one TM modulation value 500 per carrier cycle. However, as noted above, if the area under the curve may differ from period to period, ie there may be no symmetry between periods so that two symbols can be carried in each period, 1 There may be two TM modulation values per carrier period. In such a case, it is possible to represent two different symbols (ie, information) in one carrier cycle by the presence of two TM modulation values per carrier cycle. This approach may be suitable for transmission over optical fiber, for example, although there is no other signal occupying the transmission bandwidth, DC shift is typically for transmission over other media. It is because it is unsuitable.

The variable called the TM modulation period _tTMM is the time during which the TM modulation value is held, which is an integer multiple of the carrier period. This would mean that in such a case the maximum TM modulation frequency f _TMM is half of the carrier frequency f _C. That is, the modulation bandwidth is limited to 1⁄2 of f _C , which means that the lower limit of the sampling rate of the Nyquist rate, ie, the signal sampling without aliasing, is twice the bandwidth of the band-limited signal. As it is known to be. However, if there are two TM modulation values per carrier period, then the maximum TM modulation frequency f _TMM is equal to the carrier frequency f _C. There is no minimum value of f _TMM , including the DC response.

Referring back to FIG. 2, the LUT 210 stores quarter cycles specific to each value of TM modulation. There are four quarter cycles in each carrier cycle (eg, as shown in FIG. 3). If one digital bit (N = 1) is allocated for each TM modulation period, only two unique TM modulation levels, ie, two unique sets of one set consisting of two quarter cycles, are stored in LUT 210. The first level represents logic "0" and the second level represents logic "1". If two digital bits (N = 2) are assigned to each _tTMM , there will be four potential TM modulation levels. Similarly, if 3 bits (N = 3) are assigned to each _tTMM , then 8 TM modulation levels will be present (and so on).

The LUT 210 accommodates 2 ^N different quarter-cycle waveforms, and since all the waveforms are composed of four quarter-cycle waveforms, a total of 4 × 2 ^N waveforms are accommodated. The number of time steps or clock periods per quarter cycle (e.g., processor clock or CPU clock to read the LUT 210) will depend on the allowable waveform fluctuation that the electronic device implementing the method can tolerate . This may require sub-nanosecond time steps if the carrier frequency is in the 300 MHz range. Lower carrier frequencies may be preferred by both TM schemes (e.g., LUT branches and "operational branches" described herein) and may be heterodyned to carrier frequencies.

At block 202, a TM modulation signal is input to the LUT 210. The TM modulation signal may be a signal including any number of digital bits or a signal represented by this (eg, a signal of N bits width). The LUT 210 contains values or representations corresponding to each quarter period. This may be generated by the operation branch 220 if it is not accommodated in the LUT 210. For example, for each TM modulation value that may be represented in row 210a (eg, 1 to 2 ^N ), a quarter cycle is associated with the TM modulation value and in column 210b (eg, from initial time to 1⁄4 cycle) It may be expressed and stored as coordinate data (e.g., x, y) over time. At block 204, a carrier signal having a carrier frequency f _C is input. The carrier signal may be an RF signal and may operate as a clock signal. At block 206, it is determined whether to use the LUT 210 or the arithmetic branch 220 to perform the modulation. Either LUT 210 or operation branch 220 may be utilized to generate the modulated output signal. If LUT 210 is utilized, the quarter period associated with the received TM modulation value is output from LUT 210 to analog gate 208.

When the operation branch 220 is used, for example, when the operation branch 210 is selected in block 206, the TM modulation signal is input to the operation block 220. Arithmetic block 220 outputs a quarter periodic waveform that is approximately the same as the waveform that would be output by LUT block 210 if the same TM modulation value was received. However, operation block 220 does not store quarter period values associated with each TM modulation value, but generates a quarter period corresponding to each received TM modulation value. Operation block 220 produces a modulated quarter period. This is initially a cosine segment of length 180 °, at a frequency twice the carrier frequency (2 f _C ), 0 ° to 90 °, 90 ° to 180 °, 180 ° to 270 °, and 270 ° to 270 ° This is done by generating at each equivalent carrier frequency quadrant of 360 °. Thus, these generated cosine segments constitute quarter period segments at the carrier frequency. The amplitude settings are: received TM modulation values corresponding to 0 ° to 90 ° quarters and 180 ° to 270 ° quarters (ie "first" and "third" quarter cycles), and 90 ° to With complementary modulation values corresponding to 180 ° and 270 ° to 360 °. As those skilled in the art will readily appreciate, any sine wave signal may be generated using known mathematical relationships, which may be implemented in circuitry and / or software. Thus, the cosine segment of operation branch 220 may be so generated, having an amplitude set by the received TM modulation value.

  Arithmetic branch 220 performs mathematical operations to generate quarter-period segments using a processor with a clock that is more than the carrier frequency. This is to execute software code or to drive a hardware based waveform generator, which may be any known waveform generator. Operation branch 220 is likely to require a higher clock frequency than LUT branch 210. The output from the LUT 210 or operation branch 220 is directed to an analog gate 208, which collects quarter cycles into one continuous signal and passes it to the next heterodyne block 212.

Aspects of the present disclosure are considered from the frequency domain for transmission and heterodyne purposes. FIG. 6 is a plot of the frequency spectrum of the TM modulated signal 300 shown in FIG. 4, where f _C is the carrier signal frequency, 2 f _c , 3 f _c etc. are the second harmonic of the carrier frequency, the second Three harmonics etc. Signal 300 has the spectrum shown in FIG. 6 at the origin, possibly with visible inflection.

In addition to the fundamental carrier frequency component 610, there is a third harmonic component 620 of the signal 300, which includes phase modulation. The TM modulation component is present only at the third harmonic. That is, the TM modulation component is the third harmonic component 620. There is no second harmonic signal. Two output frequencies (3f _c -2f _c ) and (3f _c + 2f _c ) by generating the second harmonic signal as a local oscillator in block 214 and heterodyne the third harmonic component using a mixer circuit Is generated. This is shown in FIG. The TM modulation component, ie, the third harmonic component 620, is downshifted to the fundamental carrier frequency (signal 710). The additive component of the heterodyne process, ie, the fifth harmonic component 730, may be removed at block 214 (eg, by the filter 810 shown in FIG. 8) and may be output for transmission on the designated communication channel. It can be filtered to match.

  Unlike the known modulation techniques, as given by the present disclosure, the third harmonic is phase shifted, but this phase shift is a phase shift with respect to the fundamental carrier rather than the third harmonic. In the case of normal FM or PM transmission, it is the carrier itself that is phase shifted. In TM, the fundamental does not change, and the phase of the third harmonic is associated only with the fundamental.

  This difference is important for several reasons. For each half period of the base carrier (i.e., each TM modulation symbol), there are 1.5 periods of the third harmonic that is not modulated. The third harmonic changes only when the data changes (i.e., when the TM modulation signal 500 changes). Thus, there is little impact on power and spectrum, and another reason why the present invention is transparent to conventional modulation is, in most practical applications, the communication channel as well as AM and FM broadcasts of radio. Even so, there may be more than 100 carrier periods per TM symbol, between which there is no change in the third harmonic (ie no change in modulation). The third harmonic is only shifted in phase (time) with respect to the fundamental wave.

In order to implement the QC scheme, the analog bandwidth needs to be at least three times the carrier frequency, since the third harmonic (eg, 3f _C ) is used. Furthermore, the QC scheme requires that the clock frequency be 16 times the carrier signal frequency, simply because there are 4 time steps per quarter period. The QC may be generated at the lower carrier signal and then heterodyned upward to the desired carrier frequency. This lower carrier frequency defines the upper frequency limit of the TM modulation value.

  9a and 9b are block diagrams illustrating a direct spectrum (DS) generation system and scheme in a further embodiment of the present disclosure. The DS generation method can be said to simplify the implementation of TM modulation. In the DS scheme, the sideband spectrum is generated directly and its energy is added to all the other present in the bandwidth of the communication channel. The DS scheme is based on the frequency domain.

  Referring to FIG. 6, existing transmitters have some complex modulation format. QAM, QPSK, OFDM, etc. are representative types of complex modulations utilized. The sideband energy in the existing modulation is represented by component 610 in FIG. The application of TM modulation produces the third harmonic and the TM sideband energy is represented by component 620. Note that the second harmonic component may also be present, but this does not include modulation.

  The second harmonic signal is useful in that it can be used to shift the TM sideband energy 620 down to the fundamental carrier frequency 610. This is done by heterodyne processing using a mixer function that multiplies the two sinusoidal input signals to produce a subtractive frequency output and a summed frequency output. Referring to FIG. 7, hatching represents energy converted from the third harmonic 720 into the fundamental wave 710 and the fifth harmonic 730.

  The use of the second harmonic is optional. A stable second harmonic can be obtained by means of a phase lock loop known in the art. Also, non-linearities may exist, and if so, some of the sideband energy may actually be downconverted, which is also stable as a downconversion scheme. It may not be reliable.

  Communication regulations require that every transmitter use an output filter to ensure that there is no energy radiated out of the designated communication channel. As shown in FIG. 8, an output filter 810 that filters out harmonics may be utilized for transmission on a designated communication channel. The filter may include a passband 812.

  Figures 9a and 9b show two systems and schemes for direct spectrum generation, utilizing the concepts described above. FIG. 9a shows a software based system and scheme for direct spectrum generation and FIG. 9b shows a hardware based system and scheme for direct spectrum generation. In FIG. 9 a, clock signal 910 and digital modulation signal 920 are input to microprocessor 901. In FIG. 9 b, carrier signal 915 and analog modulation signal 925 are input to non-linear analog circuit 902. Third harmonic sidebands (e.g., TM modulation component 620) are generated directly by microprocessor 901 and / or circuit 902 based on the input signal. The microprocessor 901 and / or the circuit 902 further directly heterodynes the third harmonic sideband 620 with the input clock 910 (FIG. 9a) or the carrier 915 (FIG. 9b) to obtain sideband energy (e.g. 710). It is possible to generate directly at the fundamental frequency. The DS scheme relies on software generation of the entire equation or non-linear analog circuitry to implement the equation. That is, the microprocessor 901 (FIG. 9a) and / or the circuit 902 (FIG. 9b) directly calculates and generates the third harmonic sideband 620 based on the input signal, using known mathematical relationships. . The third harmonic sideband 620 is then heterodyned by the microprocessor 901 and / or the non-linear analog circuit 902 to shift the third harmonic sideband 620 to the fundamental frequency.

  In the following, systems and methods for receiving and demodulating transpose modulation are disclosed. FIG. 10 is a block diagram illustrating a system and method for demodulating a TM modulated signal, which may be referred to as "sub-period calibration" (SCC). The TM modulation SCC demodulation scheme operates in the time domain, for example, by reconstruction of the waveform (eg, signal 300 of FIG. 4) as shown in the QC scheme section.

  In the SCC scheme, the third harmonic is added to the received signal 1001 in a broadband environment. The phase lock loop 1010 generates a strict, unmodulated third harmonic signal that is added or multiplied to the received signal 1001 at element 1020. Thereafter, voltage levels of each positive peak and negative peak are detected by the positive peak detector 1030 and / or the negative peak detector 1040 and a reference having a matching negative peak value and a positive peak value (by the reference ramp generator 1050) Used to generate lamps. Thus, every half cycle of the received signal 1001, the system (ie, the occurrence of each peak) is calibrated when a new reference ramp is generated. The ramps are regenerated every half cycle of the carrier signal 1001. The timing of the peaks is used by the peak timing element 1060 to set the timing of the reference ramp. The inflection is detected by the detectors 1030 and 1040, and the timing of the inflection is used to sample the reference ramp output from the reference ramp generator 1050 and to hold the sampled ramp value. The voltage is an analog value of TM modulation, is output from the sample and hold element 1070, may be used directly, and may be converted to digital. The reference ramp has a positive slope from the negative half cycle to the positive half cycle of the carrier wave. At the next half cycle of the carrier (ie, from positive half cycle to negative half cycle), the reference ramp has a negative slope.

  One advantage of SCC demodulation systems and schemes is to provide a robust demodulation scheme. This is because SCC demodulation relates only to the occurrence of negative and positive peaks and the presence of inflection between these peaks. Thus, SCC demodulation is much less susceptible to noise induced errors than other demodulation techniques.

  FIG. 11 is a block diagram illustrating a demodulation system and method according to a further embodiment of the present disclosure, which may be referred to as “third harmonic phase detection” (3PD). The TM modulation third harmonic phase detection (3PD) demodulation scheme works by regenerating the third harmonic component and demodulating the phase modulation present in that component.

  As shown in FIG. 11, the received TM modulated signal 1101 is used by phase lock loop 1110 to generate a stable, unmodulated base carrier signal, which is received by subtractor element 1120 as received signal 1101. Is subtracted from The output from the subtracting element 1120 can be filtered by the fundamental notch filter 1130 to remove any spurious emissions at the fundamental carrier frequency. Thus, the remaining signal is sideband energy (eg, a TM modulation component) that drives the third harmonic phase detector 1140. The third harmonic phase detector 1140 may be any known or conventional phase detector. The resulting output 1150 is a TM modulated analog value.

  FIG. 12 is a block diagram illustrating further demodulation systems and methods according to another embodiment of the present disclosure. The demodulation system and method shown in FIG. 12 is a fast Fourier transform (TMFFT) demodulation scheme of TM modulation and operates by analysis of sideband spectra.

  The TMFFT scheme can be said to provide the simplest hardware implementation, but it can also be said to be the most complex in terms of signal processing. The TM-modulated received signal 1201 is quantized by an analog-to-digital converter and then analyzed by an FFT function 1210. After the receiver amplifies the signal to a level suitable for conversion to digital bits, this signal is output to element 1210. Element 1210 may be a processor such as a computer CPU or a more specialized processor such as a field programmable gate array specifically designed for Fourier transform calculations or some custom integrated circuit. The output of FFT element 1210 is a number of data values that represent the signal strength of received TM signal 1201 at multiple discrete frequencies. The TM spectrum is known as it relates to the TM operating mode, ie the number of bits per symbol (ie the number of allocated bits per TM modulation period) and the symbol rate.

  The symbol rate is equal to the carrier frequency divided by the number of carrier periods per symbol. Mathematically speaking, in one easy-to-understand example, dividing 1 MHz of the carrier frequency by 10 of the number of carrier cycles per symbol results in 100,000 symbols per second.

  The frequency of the symbols is 50,000 symbol periods per second, dividing 100,000 symbols per second by two.

  Thus, the frequency of interest in this example is 50 kHz, ie, the symbol period frequency is 50 kHz above and 50 kHz below the carrier frequency. When there are many modulation levels such as 6 bits or 64 modulation levels per symbol, 100kHz or 150kHz is also included in the field of view to additionally include Bessel-related sidebands to make the FFT demodulation process more accurate. It will come in. Also, if the number of carrier periods per symbol is small, increasing the number of sideband frequencies will reduce the demodulation error rate. Depending on the receiver, the carrier frequency may be heterodyned to an intermediate frequency (IF) for amplification or heterodyned to baseband to zero.

  The 50 kHz FFT output value will have a value that follows TM modulation. If the TM modulation has 4 bits per symbol, then the numerical value of the FFT output is batched into 16 levels and converted to 4 binary bits to generate a TM modulation value.

  FIG. 13 is a block diagram illustrating a TM transmitter 1300 that generates and transmits a signal consisting of a TM signal added to an existing signal (eg, a modulated RF signal). TM transmitter 1300 includes a carrier signal generation portion 1310 and a TM modulation signal processing portion 1320. One exemplary implementation of the carrier signal generation portion 1310 is shown in the block diagram of FIG. 14 and one exemplary implementation of the TM modulation signal processing portion 1320 is shown in the block diagram of FIG.

  The carrier signal generation portion 1310 obtains low level samples of the existing signal 1301 (which may or may not be modulated) (eg, by the directional coupler 1312), and the existing conventional modulation (eg, It is operative to obtain a single fundamental frequency carrier signal (FC, or fundamental carrier), removing AM, FM, or any other conventional modulation type). The existing conventional modulation may be removed from the samples of the existing signal 1301 by the band pass filter stage 1314, which may be a narrow band pass selected to remove the conventional modulation from the frequency carrier signal. It may have an area. A second harmonic generator 1316 generates a second harmonic signal (H2, or second harmonic), for example, by multiplying itself with the FC signal. Similarly, a third harmonic generator 1318 generates a third harmonic signal (H3), for example, by multiplying the FC signal by H2. It will be appreciated that the first and second harmonic generators 1316, 1318 may be any known scheme or circuit that generates harmonics (e.g. including a phase lock loop) or that scheme or A circuit may be included.

  As shown in FIG. 14, the band pass filter stage 1314 includes a first SAW filter 1313 having a very narrow bandwidth, a gain stage 1315, a comparator 1317 for stabilizing amplification, and a second narrow bandwidth. And the SAW filter 1310 of FIG. A phase adjustment stage 1311 may be included in the carrier signal generation portion 1310 to phase lock and lock the TM output fundamental carrier with the existing fundamental signal (ie, FT). The second and third harmonic generators may be implemented as signal multipliers 1316, 1318.

  As shown in FIGS. 14 and 15, the TM base carrier and the existing signal base carrier are noted as being phase locked (eg, by phase adjustment stage 1311). This may be accomplished by a feedback loop that places the TM modulation circuit inside a large phase locked loop to perform phase matching of the TM output base carrier signal with the base carrier of the existing signal in combiner 1338.

  The circuit shown in FIG. 14 is important in that it generates a signal at a frequency that is totally dependent on the exact existing signal frequency. The off frequency signal does not affect the TM operation.

  The TM modulation signal is processed through TM modulation signal processing portion 1320. TM modulation is performed on the third harmonic, and the frequency is converted to a fundamental (FC) to be combined with the existing signal 1301.

  The TM modulation signal input 1302 to the TM modulation signal processing portion 1320 of the TM transmitter 1300 is analog in nature and is bandlimited (eg, by a low pass modulation Nyquist limiting filter 1322), thereby providing communication channel bandwidth and Consistent sideband energy is generated. The TM modulation signal is then processed by inverse optimizer 1324 and TM modulator (or time shift modulator) 1326. As shown in FIG. 15, a gain stage 1321 may be included, and the invert optimizer 1324 may include a sample and hold function in addition to the optimization function. The low pass filter 1322, gain stage 1321, and inverse optimizer 1324 operate to limit the bandwidth of the TM modulation to the bandwidth of the communication channel. Optimization may be turned on or off depending on the presence or absence of an input signal to the inverse optimizer 1324.

  The third harmonic signal (H3) drives the TM modulator (or time shift modulator) 1326 by time shifting the third harmonic. This produces a set of Bessel function sidebands. Only one set of upper and lower sidebands is required for TM demodulation. These sidebands are bandwidth limited with respect to the third harmonic, and this limitation is achieved by filtering the TM modulation signal prior to TM modulation processing to match the bandwidth of the communication channel. It will be.

  The inventors have confirmed in simulation that only one pair of sidebands is generated in the time shift modulation disclosed herein, unlike phase shift modulation. This is confirmed in the laboratory with a measuring instrument such as an oscilloscope or a spectrum analyzer. Phase modulation produces Bessel series sidebands as expected. On the other hand, in time shift modulation, only one set of upper sideband and lower sideband is generated.

  The TM modulator (or time shift modulator) 1326 can perform time shift modulation with an all pass filter corrected by voltage controlled time delay. The control voltage is provided by the conditional TM modulation signal (with or without optimization). A time shift is performed on the third harmonic (H3) signal. Although TM modulator 1326 is described herein primarily with respect to time shift modulation, as one skilled in the art will appreciate, TM modulator 1326 may be a phase shift modulator as well.

  In the following, the circuits, principles and functionality of the time delay shifts available in TM modulator (or time shift modulator) 1326 will be described in more detail. For single frequency input signals, such as sine waves, the time delay is similar to phase shift.

  The all-pass filter may include an operational amplifier, and the operational amplifier has a feedback resistor connected between the output of the operational amplifier and the negative (inverting) input of the operational amplifier, and a second resistor of the same value operates The amplifier is connected between the negative (inverted) input and the signal input, and the positive (non-inverted) input is connected to the midpoint of the series network of capacitors and resistors, which is one end of the signal It is connected to the input and the other end is grounded.

  The value of the capacitor or resistor in series is modifiable by the control signal by using a four quadrant multiplier, the output of the four quadrant multiplier replaces the ground connection of the series RC network, the four quadrant multiplier One input of is connected to the intermediate connection of the series network and to the second input as control signal input (ie, the TM modulation signal).

  The phase shift may be generated as a time shift by time delay shifting of the input signal according to the input control signal.

  A TM on / off selector 1328 may be included, which may be an unmodulated third harmonic (H3) signal (eg, the output of the third harmonic generator 1318) or a TM modulated third harmonic signal (eg, For example, one of the outputs of the time shift modulator 1326 is selected. This function holds the full power supplied to transmit antenna 1340, regardless of whether a TM is used.

  The TM modulated H3 signal is downed by multiplying the TM modulated H3 signal by the second harmonic (H2) signal in the downconverter 1330 (or “heterodyne frequency conversion” block 1330 shown in FIG. 15). It is converted. As a result, the sideband energy of the TM-modulated H3 signal is shifted to the FC frequency and then subjected to modulation band-pass filtering by the band-pass filter 1332 at FC.

  The resulting FC-based TM signal is then passed through an amplifier 1334 which implements conventional amplification to establish usable power levels. The amplifier 1334 may be, for example, an RF power amplifier. The FC based TM signal travels through the final bandlimited band pass filter 1338 and the combiner 1338, and the TM output signal is added to the existing signal 1301. The resulting combined signal is connected to transmit antenna 1340 for transmission. The phase is locked between the TM output signal and the existing signal 1301.

  FIG. 16 is a block diagram illustrating a TM receiver 1600 that receives the combined signal 1641 (eg, the TM signal is added to the existing signal), and extracts and demodulates the TM signal.

  TM receiver 1600 is as close as possible to antenna 1640 or is the output of a first generated IF (intermediate frequency) (e.g. an existing receiver, the intermediate frequency output being more common depending on the communication facility) To get the received combined signal as close as possible.

  TM receiver 1600 includes a carrier signal and harmonic recovery portion 1610 and a TM separation and demodulation portion 1620. An exemplary implementation of the carrier signal and harmonic recovery portion 1610 is shown in the block diagram of FIG. 17 and an exemplary implementation of the TM separation and demodulation portion 1620 is shown in the block diagram of FIG.

  The carrier signal and harmonic recovery portion 1610 of the TM receiver 1600 (a) recovers the existing fundamental carrier signal (FC) as an unmodulated signal, (b) the second harmonic of the recovered fundamental signal And (c) circuitry for performing the generation of the third harmonic signal of the recovered fundamental signal. These all act as local oscillator signals, except that they are derived exactly from the received signal. The circuit to do this is similar to that used in the transmitter 1300.

  In the carrier signal and harmonic recovery portion 1610, the received signal 1641 (eg, existing signal) from the receiving antenna 1640 or from the IF (intermediate frequency) output of an existing receiver (which is common for some communication facilities) (A combined RF signal) in which the TM signal is added is filtered by a very narrow band pass filter stage 1614 to remove all existing modulation to obtain a pure base carrier signal (FC, or base carrier) . The FC is squared in the second harmonic generator 1616 to generate a second harmonic signal (H2). The FC and second harmonic signal (H2) are multiplied together in a third harmonic generator 1618 to produce a third harmonic signal (H3).

  As shown in FIG. 17, the front end of receiver 1600 may have AGC (automatic gain control) controlled gain and may include SAW filter 1613 and gain stage 1615. Similar to the carrier signal generation portion 1310 of the transmitter 1300, the carrier signal and harmonic recovery portion 1610 of the receiver 1600 may include a comparator 1617 and a second SAW filter 1619. The carrier signal and harmonic recovery portion 1610 may include a phase adjustment stage 1611 that compensates for the phase shift of the SAW filter. The second and third harmonic generators may be implemented as signal multipliers 1616, 1618.

  In the TM separation and demodulation portion 1620, the received signal 1641 (after being processed by the receiver front end with AGC control gain) is bandpass filtered by the bandpass filter 1636 with a bandwidth equal to the bandwidth of the communication channel Ru.

  Thereafter, separation and extraction processing is performed on the broadband received signal. The first function is differential processing between the received signal (implemented by the time delay amplifier 1634) and the delayed version of that signal. The delay is equal to 1⁄4 of the period of the third harmonic. The time delay amplifier 1634 uses a time delay based filter circuit (eg, delay stage 1633 and differential amplifier stage 1635) that separates the received TM energy.

  The separated signal (for example, the difference signal of the fundamental frequency) is upconverted by being loaded with the (H2) signal in the upconverter 1630. That is, the separated signal is heterodyned to the frequency of the third harmonic of the received fundamental signal (FC). The result is that the signal of the third harmonic frequency (after being filtered by the band pass filter 1632 to remove the product term of the fundamental wave) is subjected to TM modulation. At this point, amplitude variations can occur due to the effects of existing carrier modulation and transmission media. Thus, this signal is provided to the analog comparator 1650 along with the signal common reference, which generates a signal without amplitude variations. By filtering with the band pass filter 1631, the third harmonic carrier frequency subjected to TM modulation is selected, and other harmonics are removed.

  The output signal from the band pass filter 1632 is not only directed to the analog comparator 1650 as described above, but also used as an input to the TM signal detector 1628, the TM signal detector 1628 having a correlation function (ie, Comparison or phase between the signal received as output from the band pass filter 1632 and the third harmonic signal (H3) received as output from the third harmonic generator 1618) To detect The TM signal detector 1628 outputs a signal indicating whether a TM is used, ie, whether a TM signal is present in the received signal 1641.

  In the extraction process, separation and extraction of the TM signal from the received signal 1641 are completed. The extracted signal (eg, the output from the band pass filter 1631) is a recovered reference third harmonic signal (eg, used as a reference, received existing carrier signal) necessary for demodulation of TM information Time-shifted TM modulation in comparison with (H3)) derived from.

  The TM demodulator 1626 is TM modulated (received as a reference input from the third harmonic generator 1618) and the third harmonic signal (H3) (received as an input from the band pass filter 1631) The TM signal is demodulated by detecting the time shift between it and the third harmonic signal. The TM demodulator 1626 can detect a time shift between input signals by multiplying the two signals into a correlation function. Alternatively, the TM demodulator 1626 detects the timing difference between the (H3) reference and the TM modulated signal from the separation and extraction process using an exclusive OR function.

  The separation and demodulation portion 1620 of the TM may optionally include an inversion optimizer 1624 that recovers the signal transmitted from the transmitter with the inversion optimizer 1324 described above.

  The demodulated signal passes through a modulation low pass filter 1622 where all carrier and other noise sources are removed to obtain a TM modulated output signal 1602.

FIG. 18 shows a basic set of functions for extracting the received TM signal from the combined signal. The time delay differentiator or filter (i.e., delay stage 1633) may have the optimally set delay time (.25) / (3 * f _fc ). Deviations from this value only reduce the level of the separated TM signal, but in practice the specific value cancels the level of the separated TM signal. The extraction function only removes the amplitude fluctuations of the separated signal.

  The time delay based filter circuit of FIG. 18 has the natural frequency response behavior shown in FIG. There is a periodic signal that cancels out at frequencies such as DC, sixth harmonic.

  Existing signal receivers do not respond to the sideband energy of the TM signal. Adding TM to an existing signal has the effect of reducing the signal-to-noise ratio (SNR) of the received existing signal. Similarly, there is also a noise contribution from the existing signal to the received TM signal.

  The TM receiver 1600 relies on the second and third harmonic signals associated with the received fundamental signal. Frequency shifts due to Doppler effects or variations in signal path length (moving receivers or transmitters) do not affect the demodulation of the TM signal. This is because the entire process is based on the frequency of the existing signal.

  As described above with respect to FIGS. 13-19, TM modulation may be provided for the carrier signal and may be derived from any existing transmitter signal. The modulated TM signal may then be combined with the existing transmitter signal (whether modulated or not), thereby increasing the information bandwidth of the predefined communication channel. Although FIGS. 13-19 have been specifically described with respect to the TM signal, this is not intended to be limiting, and the same function or principle is applicable to any modulated signal. The information may be conveyed as a time difference or phase angle difference between two signals of different frequencies.

  There, by summing a second carrier signal having a frequency that is harmonically related to the first carrier signal frequency, the frequency being modulated with information unrelated to the information modulating the first carrier signal. A method is provided for increasing the information bandwidth of any defined communication channel. The second signal and the modulation sidebands may be heterodyned to the frequency of the first carrier signal, and the modulation sidebands may be less than or equal to the bandwidth of the communication channel.

  The first carrier signal and the second modulated carrier signal may be transmitted without any modulation.

  The first carrier signal may or may not be modulated (with any modulation scheme), and the second carrier signal may be time shift modulated or angle modulated. The second carrier signal may have a frequency that is in a known relationship with the frequency of the first carrier signal. Similarly, the second carrier signal may have a phase angle relationship or timing relationship with the first carrier signal. The information to be conveyed may cause a time shift or angular modulation of the second carrier signal, and the means of modulation of the second carrier signal vary the temporal or phase angular relationship with the first carrier signal. Good.

  Sideband energy or communication frequency shifted to occupy the same frequency range as the first carrier signal, if modulated by time shift or angle modulation of the second carrier signal Sideband energy is generated that is located within the communication channel bandwidth that will be used for The combination of the first carrier signal and the sidebands of the second carrier signal may be transmitted together within the frequency limits of the communication channel and may be received by the receiving device. Furthermore, the combination of these two carrier signals may be transmitted without band limiting and may be received by the receiving device.

  The receiver may demodulate the modulation information of the second carrier signal using the first carrier signal as a demodulation reference signal.

  Also described herein is the quadrature transposed modulation method, ie, the information bandwidth in the fixed communication channel by adding the second transposed modulated (TM) signal to the existing transposed modulated signal. , Provide a way to increase the information bandwidth provided by transposition modulation as previously described.

  In one embodiment, the second transposed modulation signal is summed with the existing transposed modulation signal by using a fundamental carrier signal frequency that is the same frequency as the existing transposed modulation signal and differs in phase by 90 degrees or quadrature. May be done. This maintains the mutual non-influence of transposition modulation and conventional amplitude modulation, frequency modulation and phase modulation. Adding the quadrature transpose modulation signal also has mutual non-influence between the two transpose modulated carriers, both of which are non-influence with the existing conventional modulation signal Have.

  Transposition modulation may be utilized in various manners, including, for example, for optical communications. Provided herein is a method for placing a wideband transpose modulated signal directly onto the optical frequency beam to increase the data bandwidth of the communication.

  A light beam is used because of its broadband characteristics. They are modulated by various means. Transposition modulation may be placed on the light beam for various modes of information transmission, all such modes being considered by the present disclosure and within the scope of the present disclosure.

  One such example can be described as follows. A carrier signal is required for any modulation scheme that carries information. In transposition modulation, an existing modulated carrier signal may be used as a transposed modulation carrier. Transposition modulation may also provide a carrier signal in the absence of a carrier signal. The transposed carrier signal is used to drive the light modulator, regardless of whether the existing signal is used.

  Light modulators range from light emitting diode drivers and laser diode drivers to light beam modulators that change the opacity and phase of the light beam. According to the present disclosure, it is possible to apply a transposed carrier signal to a single modulator. This makes it possible to increase the existing information bandwidth using conventional amplitude modulation, frequency modulation or phase modulation.

  A band-limited transposed modulation signal is transmitted, which will include the third harmonic component of the transposed modulation if no other modulation is present. This places the fundamental frequency components at lower frequencies, typically with less attenuation. Transposition demodulation relies on this component as a reference. The higher the frequency of the third harmonic component, the wider the bandwidth enabling the maximum information modulation bandwidth.

  In another embodiment, two separate light beams may be utilized, in which case the lower frequency beam is modulated in a conventional manner (e.g. amplitude modulation or phase modulation) and for transposition modulation Used as a reference carrier for The higher frequency light beam is used for the third harmonic component of the transpose modulation.

  Another mode in which transposition modulation can be used is ultrasound communication, for example, it can be used for underwater wireless communication. For example, in one embodiment, a transposed modulation signal can be applied to the ultrasound transducer to generate an acoustic signal, and the acoustic signal is received and demodulated to recover the original modulation information. Good. This modulation imposes a near zero impedance drive on the transducer to force operation above the transducer's natural resonant frequency.

  The receiver may use natural broadband transducers based on technology that converts acoustic energy into electrical energy without resonant peak response in addition to the modulation broadband. Such receive transducers are manufactured using MEMS (Micro-Electro-Mechanical Systems) technology that provides broadband response with high sensitivity.

One transposed modulation carrier frequency may be used. Two separate ultrasound frequencies may be used to separately transport the base carrier signal component and the third harmonic component signal.
The present invention has the following means.
The method of the means 1 of the present invention is
A method of generating a signal,
Providing a look-up table having a plurality of quarter-periodic waveforms, wherein each of the quarter-periodic waveforms is associated with a respective input level;
Receiving an input signal;
Outputting a quarter period waveform associated with the level of the received input signal;
It is characterized by including.
The method of means 2 of the present invention is the method described in means 1;
(A) the input level includes voltage swing;
(B) the input signal is an N-bit wide signal, the look-up table contains 2 ^N unique quarter-period waveforms, each waveform being associated with a respective input level,
(C) receiving a carrier signal, the output quarter period waveform further comprising the step of a frequency being the same as the carrier signal;
(D) collecting the output quarter-cycle waveforms into one continuous output signal, optionally,
Frequency shifting a third harmonic component of the output signal to the carrier signal frequency, wherein the third harmonic component generates a second harmonic and mixes the second and third harmonics Preferably frequency shifted to the carrier signal frequency by further comprising the step of frequency shifting;
Characterized by one or more of the above steps,
It is characterized by
The method of means 3 of the present invention is
A method of generating a signal,
Receiving a carrier signal;
Receiving an input signal;
Generating a quarter period waveform based on the input signal;
Collecting the generated quarter-cycle waveforms into a continuous output signal, the output signal forming a segment between a first negative peak and a positive peak of the output signal, The output signal includes a first inflection during one quarter period, the output signal forming a segment between the positive peak and the second negative peak of the output signal, between adjacent quarter periods Including a second inflection, wherein the first and second inflections represent levels of the input signal,
It is characterized by
The method of means 4 of the present invention is the method described in means 3;
(A) the first and second inflections represent different levels of the input signal;
(B) The quarter-cycle waveform is generated as a cosine segment with a length of 180 ° at a frequency twice the carrier signal frequency, and the first quarter-cycle waveform is at an equivalent carrier frequency quarter of 0 ° to 90 °. A second quarter wave waveform is generated at an equivalent carrier frequency quadrant of 90 ° to 180 °, and a third quarter wave waveform is generated at an equivalent carrier frequency quadrant of 180 ° to 270 °, the fourth A quarter period waveform is generated at an equivalent carrier frequency quarter of 270 ° to 360 °, optionally, the quarter period waveform is generated by a waveform generator or software,
(C) frequency shifting a third harmonic component of the output signal to the carrier signal frequency, wherein the third harmonic component generates a second harmonic to generate the second and third harmonics. Preferably frequency shifted to the carrier signal frequency by mixing
Characterized by one or more of the above steps,
It is characterized by
The method of means 5 of the present invention is
A method of generating a modulated signal, comprising
Receiving a carrier signal;
Receiving an input signal;
Generating a third harmonic sideband based on the received signal;
Shifting the third harmonic sideband to the frequency of the received carrier signal, or
Adding a third harmonic to the modulated signal;
Detecting a peak amplitude of a sum of the modulated signal and the third harmonic;
Generating a reference ramp when the peak amplitude is detected;
Detecting an inflection of a sum of the modulated signal and the third harmonic;
Sampling the reference ramp when an inflection is detected;
Holding and outputting the sample value, wherein the third harmonic is preferably generated by a phase locked loop.
It is characterized by
The method of means 6 of the present invention is
A method of demodulating a signal,
Filtering out all but the third harmonic component from the modulated signal;
Detecting the phase of the third harmonic component, or
Converting the modulated signal to a digital signal;
Performing a fast Fourier transform on the digital signal.
It is characterized by
The system of means 7 of the present invention is
A system for generating a signal,
A look-up table comprising a plurality of quarter-cycle waveforms, each of said quarter-cycle waveforms being associated with a respective input level, said look-up table being configured to receive an input signal and being output The quarter period waveform is associated with the level of the received input signal,
It is characterized by
The system of means 8 of the present invention is the system described in means 7;
(A) the input level includes voltage swing;
(B) the input signal is an N-bit wide signal, the look-up table contains 2 ^N unique quarter-period waveforms, each waveform being associated with a respective input level,
(C) the look-up table is further configured to receive a carrier signal, and the output quarter period waveform is the same in frequency as the carrier signal;
(D) further comprising an analog gate that collects the output quarter period waveform into one continuous output signal,
(E) A frequency shift element for frequency shifting the third harmonic component of the output signal to the carrier signal frequency, comprising a mixer for mixing the third harmonic component and the second harmonic component. Further comprising the frequency shifting element, preferably
Characterized by one or more of said features,
It is characterized by
The system of means 9 of the present invention is
A system for generating a signal,
A waveform generator configured to generate a quarter cycle waveform based on the received carrier signal and the received input signal;
An analog gate that collects the generated quarter-cycle waveforms into a continuous output signal, the output signal forming a segment between a first negative peak and a positive peak of the output signal The output signal includes a first inflection during one quarter period, the output signal forming a segment between the positive peak and the second negative peak of the output signal, between adjacent quarter periods The analog gate including a second inflection, the first and second inflections representing the level of the input signal;
Equipped with
It is characterized by
The system of means 10 of the present invention is the system according to means 9;
(A) the first and second inflections represent different levels of the input signal;
(B) The quarter-cycle waveform is generated as a cosine segment with a length of 180 ° at a frequency twice the carrier signal frequency, and the first quarter-cycle waveform is at an equivalent carrier frequency quarter of 0 ° to 90 °. A second quarter wave waveform is generated at an equivalent carrier frequency quadrant of 90 ° to 180 °, and a third quarter wave waveform is generated at an equivalent carrier frequency quadrant of 180 ° to 270 °, the fourth A quarter period waveform is generated at an equivalent carrier frequency quarter of 270 ° to 360 °, optionally
Optionally further comprising a frequency shifter for frequency shifting the third harmonic component of the output signal to the carrier signal frequency,
A phase locked loop for generating a second harmonic component;
A mixer, the third harmonic component being frequency shifted to the carrier signal frequency by mixing the second and third harmonics,
Characterized by one or more of said features,
It is characterized by
The system of the means 11 of the present invention is
A system for generating a modulated signal,
Generating a third harmonic sideband based on the received carrier signal and the received input signal; frequency shifting the third harmonic sideband to the frequency of the received carrier signal; , A processor configured to do
Equipped with
It is characterized by
The system of means 12 of the present invention is
A system for demodulating a signal,
A signal adder for adding a third harmonic to the modulated signal;
A peak detector for detecting a peak amplitude of a sum of the modulated signal and the third harmonic;
A reference ramp generator that generates a reference ramp when a peak amplitude is detected;
An inflection detector for detecting an inflection of the sum of the modulated signal and the third harmonic;
A sample and hold element that samples the reference ramp when an inflection is detected and holds and outputs the sample value;
Optionally, further comprising a phase locked loop generating said third harmonic,
It is characterized by
The system of the means 13 of the present invention is
A system for demodulating a signal,
A subtraction element that subtracts the base carrier signal from the received modulated signal;
A fundamental notch filter that leaves only the third harmonic component from the modulated signal by filtering out the fundamental carrier signal;
A third harmonic phase detector for detecting the phase of the third harmonic component;
Equipped with
It is characterized by
The system of the means 14 of the present invention is
A system for demodulating a signal,
An analog-to-digital converter that converts the modulated signal into a digital signal;
A processor configured to perform a fast Fourier transform on the digital signal;
Equipped with
It is characterized by
The method of the means 15 of the present invention is
A method of increasing the information bandwidth of a defined communication channel, comprising
Receiving a first modulated signal having a first carrier signal frequency;
Receiving a second modulated signal having a second carrier signal frequency, wherein the second modulated signal is modulated with information unrelated to modulating the first carrier signal Receiving the second modulated signal, wherein the second carrier signal frequency is harmonically or subharmonically related to the first carrier signal frequency;
Combining the first signal and the second signal;
including
It is characterized by
The method of the means 16 of the present invention is the method described in the means 15, and
The second signal is heterodyned to the frequency of the first carrier signal,
It is characterized by
The method of the means 17 of the present invention is the method described in the means 15 or 16;
Transmitting the combined signal by a transmitting device;
Receiving the combined signal by a receiving device;
Demodulating the second modulated signal from the combined signal using the first carrier signal as a reference;
Further include,
It is characterized by
The time shift modulator of the means 18 of the present invention
A time shift modulator for time delay shifting an input signal according to an input control signal, comprising an all pass filter corrected by voltage control time delay
It is characterized by
The time-cissing modulator according to the means 19 of the present invention is the time-cissing modulator according to the means 18.
The all pass filter is
An operational amplifier, wherein a feedback resistor is connected between the output of the operational amplifier and the inverting input of the operational amplifier;
A second resistor having a resistance value substantially equal to that of the feedback resistor, the second resistor being connected between the inverting input and the signal input of the operational amplifier;
A series network of a capacitor and a resistor, with a first end connected to the signal input, a second end connected to ground, and an intermediate connection connected to the non-inverting input of the operational amplifier A series network of said capacitors and resistors being
It is characterized by
The time shift modulator of the means 20 of the present invention is the time-cised modulator described in the means 19.
The value of the series capacitor or series resistor is corrected by the control signal by using a four quadrant multiplier, the output of the multiplier replacing the ground connection of the series network of capacitors and resistors, said 4 One input of a quadrant multiplier is connected to the intermediate connection of the series network of capacitors and resistors, and a second input of the four quadrant multiplier is connected to the control signal input.
It is characterized by
The method of means 21 of the present invention.
A method of increasing communication bandwidth in a fixed communication channel, comprising:
Adding a second transposed modulated signal to a combined signal, the combined signal including a first transposed modulated signal and a first base carrier signal, the second transposed modulated signal The signal is summed with the combined signal using a second fundamental carrier signal having the same frequency as the first fundamental carrier signal and a phase angle of 90 degrees with respect to the first fundamental carrier signal. To be
It is characterized by
The method of the means 22 of the present invention is a method of increasing the communication bandwidth as described in the means 21;
(A) a function of doubling the information bandwidth without increasing the communication channel bandwidth;
(B) Function to improve the spectral efficiency of the transposed modulation
Characterized by one or both of
It is characterized by
The method of means 23 of the present invention is
A method of increasing the information bandwidth of ultrasonic communication,
Ultrasonic wave communication by applying transpose modulation to the ultrasonic communication signal by direct amplitude modulation of a single ultrasonic transducer, or by directly amplitude modulating the first ultrasonic transducer with the fundamental carrier signal component of the transpose modulation Applying transpose modulation to the signal;
Directly amplitude modulating the second ultrasound transducer with the third harmonic carrier signal component of the transpose modulation;
including
It is characterized by
The method of means 24 of the present invention is
A method of increasing the information bandwidth of ultrasonic communication,
Adding the transposed modulated base carrier signal and the third harmonic carrier signal component to the ultrasound communication signal by directly amplitude modulating the single ultrasound transducer using a wideband modulation technique, or
Ultrasonic communication by directly angle modulating the first ultrasound transducer with the fundamental carrier signal component of transposition modulation and directly angle modulating the second ultrasound transducer with the third harmonic carrier signal component of transposition modulation Applying transpose modulation to the signal,
including
It is characterized by
The method of the means 25 of the present invention is
A method of increasing the information bandwidth of ultrasonic communication,
Adding the transposed modulated base carrier signal and the third harmonic carrier signal component to the ultrasound communication signal by direct angle modulating the single ultrasound transducer using a wideband modulation technique;
including
It is characterized by
The system of the means 26 of the present invention is
A system for increasing optical information communication bandwidth,
With a light beam,
And an optical modulator,
Configured to directly modulate the light beam with a transposed modulation signal
It is characterized by
The system of the means 27 of the present invention is the system described in the means 26;
The light beam is configured to be amplitude modulated with a transposed modulation signal, or the light beam is configured to be phase modulated with the transposed modulation signal.
It is characterized by
The method of means 28 of the present invention is
A method of increasing the optical information communication bandwidth,
Directly modulating the light beam of the first frequency with a transpose modulation fundamental carrier frequency component, or
Directly modulating the light beam of the second frequency with the transposed modulation third harmonic component signal,
including
It is characterized by

  It is emphasized that the above-described embodiments of the present disclosure, and in particular any " preferred " embodiments, are only possible examples and have been described so that the principles of the present disclosure may be clearly understood. I want to keep it. Various changes and modifications may be made to the above-described embodiments of the present disclosure without departing substantially from the spirit and principles of the present disclosure. All such modifications and variations are intended to be included herein within the scope of the present disclosure and protected by the following claims.

Claims (4)

A method of generating a carrier signal, comprising
Providing a look-up table having a plurality of quarter-period waveforms, wherein the full waveform is comprised of four quarter-period waveforms, each of said quarter-period waveforms being associated with a respective input level Providing said steps;
Receiving an input signal represented by digital bits conveying modulation levels;
Outputting a quarter period waveform associated with one or more digital bits of the received input signal;
Collecting the output quarter-cycle waveforms into one continuous output signal to generate a modulated carrier signal;
Including
The continuous output signal comprises a first inflection between adjacent quarter-cycles forming a segment of the continuous output signal between a first negative peak value and a positive peak value, the continuous output signal being A second inflection between adjacent quarter-cycles forming a segment of the continuous output signal between a positive peak value and a second negative peak value, the first inflection and the second inflection Represents the modulation level,
How the first quarter period and the third quarter cycle the amplitude are the same, and the second quarter period and the fourth quarter cycle is the same amplitude. (A) the input level includes voltage swing;
(B) the input signal is an N-bit wide signal, the look-up table contains 2 ^N unique quarter-period waveforms, each waveform being associated with a respective input level,
(C) receiving a carrier signal, the output quarter period waveform further comprising the step of a frequency being the same as the carrier signal;
(D) frequency shifting a third harmonic component of the continuous output signal to the carrier signal frequency, wherein the third harmonic component generates a second harmonic to generate the second and third harmonics. Frequency-shifting to the carrier signal frequency by mixing waves, further comprising the step of frequency-shifting,
The method according to claim 1, characterized by satisfying at least one of the (a) to (d). A system for generating a carrier signal,
A look-up table comprising a plurality of quarter-period waveforms, the entire waveform being composed of four quarter-period waveforms, each of said quarter-period waveforms being associated with a respective input level, said look-up table comprising: modulation levels And a quarter period waveform that is output is associated with one or more digital bits of the received input signal, and is configured to receive an input signal represented by a digital bit carrying
It further comprises an analog gate that collects the output quarter-cycle waveforms into one continuous output signal to generate a modulated carrier signal,
The continuous output signal comprises a first inflection between adjacent quarter-cycles forming a segment of the continuous output signal between a first negative peak value and a positive peak value, the continuous output signal being A second inflection between adjacent quarter-cycles forming a segment of the continuous output signal between a positive peak value and a second negative peak value, the first inflection and the second inflection Represents the modulation level,
System first quarter period and the third quarter cycle the amplitude are the same, and the second quarter period and the fourth quarter cycle is the same amplitude. (A) the input level includes voltage swing;
(B) the input signal is an N-bit wide signal, the look-up table contains 2 ^N unique quarter-period waveforms, each waveform being associated with a respective input level,
(C) the look-up table is further configured to receive a carrier signal, and the output quarter period waveform is the same in frequency as the carrier signal;
(D) A frequency shift element for frequency shifting the third harmonic component of the continuous output signal to the carrier signal frequency, comprising a mixer for mixing the third harmonic component and the second harmonic component. Further comprising the frequency shift element,
The system of claim 3, characterized by satisfying at least one of the (a)-(d).

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