JP2016519472A - Transposed modulation system, method and apparatus - Google Patents

Abstract

The present invention relates to a system for transmitting, receiving and demodulating a transposed modulated signal and for time delay shifting an input signal in response to an input control signal in order to increase the information bandwidth of a defined communication channel. The present invention relates to a method and an apparatus. A method for increasing the information bandwidth of a defined communication channel includes receiving a first modulated signal having a first carrier signal frequency and a second modulated signal having a second carrier signal frequency. Receiving the signal, wherein the second modulated signal is modulated with information unrelated to information modulating the first carrier signal, and the second carrier signal frequency is Receiving the second modulated signal that is harmonically or sub-harmonically related to a carrier signal frequency of one, combining the first signal and the second signal; including. [Selection] Figure 13

Description

  This application is a continuation-in-part of Patent Document 1 filed on March 15, 2013, pending. This application is also a US Patent Provisional Application filed on March 15, 2013, Patent Document 2 filed on March 15, 2013, Patent Document filed on March 15, 2013, and Patent Document filed on March 15, 2013. 4, and claims the priority of Patent Document 5 filed on March 15, 2013 and Patent Document 6 filed on March 15, 2013, the contents of which are hereby incorporated by reference. It has been incorporated.

  The present disclosure mainly relates to signal processing, and more specifically, transposed and modulated to increase the information bandwidth of a defined communication channel and to perform time delay shifting of the input signal according to the input control signal. The present invention relates to a system, method and apparatus for transmitting, receiving and demodulating signals.

  Existing transmission schemes have bandwidth limitations imposed by national and global regulatory agencies that manage the use of frequency spectrum, whether the carrier is voice, video or data. Carrier modulation schemes have evolved from initial amplitude modulation to current schemes that combine two or more carriers in various combinations of amplitude modulation, frequency modulation, or phase modulation. A number of advanced carrier modulation schemes have been developed to maximize energy across the allocated channel bandwidth and maximize the available information bandwidth of the allocated communication channel.

  A new basic carrier modulation has been developed and is initially patented (see, for example, Vokac et al., US Pat. No. 6,057,086, the entire contents of which are incorporated herein), which are simultaneously present on the same carrier signal. It applies a new type of carrier modulation that does not interfere with amplitude modulation, frequency modulation, and / or phase modulation.

  The concept of transpose (TM) modulation is based on the initial concept of how to add information to a carrier signal without affecting the amplitude, frequency, or phase of the carrier signal (for example, the entire contents are described herein). See incorporated Vokac et al., US Pat. By generating inflections as will be shown later, it is possible to carry information on the carrier signal. This scheme is not detected by existing demodulators with amplitude, frequency, or phase modulation.

  Using the previously patented generation method, the following time domain waveforms are generated (inflections are exaggerated for clarity): When actually applied, the inflection is invisible.

  Early schemes for generating this type of waveform were incomplete in that there was a small amplitude change to be removed by the adjustment circuit. For example, FIG. 1 shows a TM modulated signal 100 generated according to the prior art taught in US Pat. As can be seen in the figure, there is an amplitude change error between the negative peak 101 and the negative peak 102.

  Thus, there remains an unmet need in the industry to address the aforementioned imperfections and deficiencies.

US Patent Application Publication No. 2014/0269969 (US Patent Application No. 13 / 841,889) US Provisional Patent Application No. 61 / 798,437 US Provisional Patent Application No. 61 / 794,786 US Provisional Patent Application No. 61 / 798,120 US Provisional Patent Application No. 61 / 794,942 US Provisional Patent Application No. 61 / 794,642 U.S. Pat. No. 4,613,974

  Embodiments of the present disclosure provide for transmission, reception and demodulation of transposed modulated signals in order to increase the information bandwidth of a defined communication channel and to perform a time delay shift of the input signal according to the input control signal. Systems, methods, and apparatus for performing are provided. In one embodiment, a method is provided for increasing the information bandwidth of a defined communication channel, the method comprising: receiving a first modulated signal having a first carrier signal frequency; and a second carrier Receiving a second modulated signal having a signal frequency, wherein the second modulated signal is modulated with information independent of the information modulating the first carrier signal; Receiving a second modulated signal, the signal frequency being harmonically or sub-harmonically related to the first carrier signal frequency, and combining the first signal and the second signal; ,including.

  In another embodiment, a time shift modulator is provided that time delay shifts an input signal in accordance with an input control signal. The time shift modulator includes an all-pass filter that is modified by a voltage controlled time delay.

  In another embodiment, a method for increasing communication bandwidth in a fixed communication channel is provided, the method comprising adding a second transposed modulated signal to a combination signal, wherein the combination signal is a first signal. The transposed and modulated signal and the first basic carrier signal, and the second transposed and modulated signal has the same frequency as the first basic carrier signal and a phase angle with respect to the first basic carrier signal. Adding to the combined signal using a second basic carrier signal that is 90 degrees.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasound communication, the method comprising applying transposition modulation to the ultrasound communication signal by direct amplitude modulation of a single ultrasound transducer.

  In another embodiment, a method for increasing the information bandwidth of ultrasonic communication is provided, the method comprising directly ultrasonically modulating a first ultrasonic transducer with a fundamental carrier signal component of transposition modulation. Applying transposition modulation to the signal and directly amplitude modulating the second ultrasonic transducer with a third harmonic carrier signal component of the transposition modulation.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasonic communications, which directly amplitude modulates a single ultrasonic transducer using a broadband modulation technique, thereby providing a transposed modulated fundamental carrier signal and a second carrier signal. Adding a third harmonic carrier signal component to the ultrasonic communication signal.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasonic communication, wherein the first ultrasonic transducer is directly angularly modulated with a fundamental carrier signal component of transposition modulation to provide a second ultrasonic wave. Applying transposition modulation to the ultrasonic communication signal by directly angularly modulating the transducer with a third harmonic carrier signal component of transposition modulation.

  In another embodiment, a method is provided for increasing the information bandwidth of ultrasonic communications, which method directly angularly modulates a single ultrasonic transducer using a wideband modulation technique, thereby providing a transposed modulated fundamental carrier signal and a second carrier signal. Adding a third harmonic carrier signal component to the ultrasonic communication signal.

  In yet another embodiment, a system for increasing optical information communication bandwidth is provided. The system includes a light beam and a light modulator. The system is configured to directly modulate the light beam with a transposed modulation signal.

  In another embodiment, a method for increasing optical information communication bandwidth is provided, the method comprising directly modulating a light beam of a first frequency with a transposed modulated fundamental carrier frequency component.

  In yet another embodiment, a method for increasing optical information communication bandwidth is provided, the method comprising directly modulating a light beam of a second frequency with a transposed modulated third harmonic component signal.

  Other systems, methods, features, and advantages of the present disclosure will become apparent to those skilled in the art upon review of the following drawings and detailed description. All such additional systems, methods, features, and advantages are intended to be included herein, within the scope of this disclosure, and protected by the accompanying claims.

  Many aspects of the disclosure will be better understood with reference to the following drawings. The elements in the drawings are not necessarily to scale, but rather focus on clearly illustrating the principles of the present disclosure. Moreover, in the drawings, like reference numerals designate corresponding elements throughout the several views.

It is a figure which shows TM modulation signal produced | generated according to a prior art. 2 is a flowchart illustrating a method for modulating a carrier signal according to a first exemplary embodiment of the present disclosure. FIG. 4 illustrates a signal generated as a quarter period, according to one embodiment of the present disclosure. FIG. 4 illustrates a sum of quarter periods of the signal shown in FIG. 3 according to one embodiment of the present disclosure. FIG. 5 illustrates an input modulation signal that can be used in an embodiment provided by the present disclosure to generate the signal shown in FIG. 4. 5 is a plot showing the frequency spectrum of the signal shown in FIG. 7 is a plot illustrating a frequency spectrum obtained as a result of heterodyning the third harmonic component of the signal shown in FIG. 6 with respect to the second harmonic, according to one embodiment provided by the present disclosure. FIG. 6 illustrates a filter applicable to one embodiment provided by the present disclosure. FIG. 2 is a block diagram illustrating a software-based direct spectrum system that generates signals according to one embodiment provided by the present disclosure. 1 is a block diagram illustrating a sub-period calibration system that demodulates a signal, according to one embodiment provided by the present disclosure. FIG. FIG. 3 is a block diagram illustrating a third harmonic phase detection system that demodulates a signal according to one embodiment provided by the present disclosure. FIG. 2 is a block diagram illustrating a Fast Fourier Transform based system for demodulating a signal according to one embodiment provided by the present disclosure. FIG. 2 is a block diagram illustrating a TM transmitter that generates and transmits a signal consisting of a TM signal added to an existing signal, according to one embodiment provided by the present disclosure. FIG. 4 is a block diagram illustrating an example implementation of a carrier signal generation portion of a TM transmitter, according to one embodiment provided by the present disclosure. FIG. 4 is a block diagram illustrating an example implementation of a TM modulated signal processing portion of a TM transmitter, according to one embodiment provided by the present disclosure. 2 is a block diagram illustrating a TM receiver that receives a signal having a TM signal added to an existing signal, extracts and demodulates the TM signal, according to one embodiment provided by the present disclosure. FIG. FIG. 6 is a block diagram illustrating an example implementation of a carrier signal and harmonic recovery portion of a TM receiver, according to one embodiment provided by the present disclosure. FIG. 3 is a block diagram illustrating an example implementation of a TM separation and demodulation portion of a TM receiver, according to one embodiment provided by the present disclosure. 7 is a graph illustrating the frequency response behavior of a time delay based filter circuit of a separation function, according to one embodiment provided by the present disclosure.

  Many embodiments of the present disclosure may take the form of computer-executable instructions, including algorithms executed by programmable computers or microprocessors. However, the present disclosure may be similarly implemented by other computer system configurations. Certain aspects of the present disclosure may be implemented in a dedicated computer or data processor that is programmed, configured, or constructed solely to implement one or more of the methods or algorithms described below.

  The following aspects of the present disclosure include a removable computer disk capable of magnetic or optical reading, a fixed magnetic disk, a floppy (registered trademark) disk drive, an optical disk drive, a magneto-optical disk drive, a magnetic tape, and a hard disk drive (HDD). ), A solid state drive (SSD), a compact flash or non-volatile memory, and may be stored and distributed on a computer readable medium, as well as electronically distributed over a network including the cloud. It's okay. Data structures and data transmissions specific to aspects of the present disclosure are also encompassed within the scope of the present disclosure.

  FIG. 2 is a flowchart 200 illustrating a method for modulating a carrier signal, according to a first exemplary embodiment of the present disclosure. It should be noted that any process description or block in the flowchart is understood to represent a module, segment, code portion, or step that includes one or more instructions for performing a particular logic function of the process. Alternative implementations that may be performed in an order different from that shown or described, including the functions being performed substantially simultaneously in the reverse order, depending on the functionality required. Embodiments are also encompassed within the scope of the present invention and will be understood by those skilled in the art. The method solves the problem of amplitude variation in the prior art (eg, as shown in FIG. 1 above) and may be implemented in hardware or software, or any combination thereof. The method shown in FIG. 2 is sometimes referred to as a “quarter cycle collection” (QC) scheme, and a look-up table (LUT) 210 is a quick way to obtain results without the need for continuous execution of operations. (If not relying on LUT 210, the result can be generated using an arithmetic function). The QC scheme is based on the time domain.

  Referring to FIG. 3, the modulated output signal 300 from the method shown in FIG. 2 includes four separate quarter-period segments in each full signal period. FIG. 3 shows three full periods (eg, periods a, b, and c), which may be output by the quarter period scheme shown in FIG. Each period consists of four quarter-period segments (eg, 301, 302, 303, and 304). Gaps are shown between the quarter-period segments, but this is for illustration purposes only. Further, the amplitude positions (a1, a2, b1, b2, c1, c2) of the inflection are exaggerated for the purpose of illustration. These inflections are formed between adjacent quarter-period segments as shown.

As shown in FIG. 3, the amplitude of the “first” quarter period (301a, 301b, and 301c) of each period may be different depending on the applied modulation value. The same is true for each other quarter period in each period shown. That is, the second quarter period (302a, 302b, 302c), the third quarter period (303a, 303b, 303c), and the fourth quarter period (304a, 304b, 304c) of each period are applied modulation. The amplitude may be different depending on the value. If the amplitude of the “first” quarter period (eg, 301a, 301b, 301c) in a period is low, the amplitude of the “second” quarter period (eg, 302a, 302b, 302c) in that same period is complementary This is because a constant amplitude always exists between the negative peak value (Pk ₋ ) in the entire period and the positive peak value (Pk ₊ ) in the period. The same applies to the “third” and “fourth” quarter periods in each period. Thereby, the positive peak value (Pk ₊ ) in each cycle is always the same. Since the negative peak value (Pk ₋ ) is also the same, the amplitude change due to the applied modulation value is eliminated.

  As further shown in FIG. 3, the “first” quarter period (301a, 301b, 301c) and the “third” quarter period (303a, 303b, 303c) in each period have the same amplitude. Similarly, the “second” quarter period (302a, 302b, 302c) and the “fourth” quarter period (304a, 304b, 304c) in each period have the same amplitude. This is to make the area under the curve of each period the same regardless of the applied modulation value. This ensures that the average value of each period is zero and that no “DC” value shift due to the applied modulation value occurs in the carrier signal.

  However, it is noted that depending on the application, a DC shift may be tolerated and thus the area under the curve may not be the same, i.e., there may be no symmetry between periods. I want. In such cases, information or “symbols” may be conveyed at a rate of 2 symbols per period. That is, there may be two different inflection points for each period (for example, one is located in the middle of the rising half period between the negative peak and the positive peak, and the other is between the positive peak and the negative peak. In the middle of the half cycle of falling between).

  Each quarter period may be generated by a constant clock or time step, so that the frequency does not change from one period to the next as a result of the applied modulation value. Each inflection (a1, a2, b1, b2, c1, c2) occurs at an angle corresponding to a 180 degree separation from exactly one half cycle to the next half cycle. This ensures that there is no phase change due to the applied modulation value.

  When these quarter periods (for example, those shown in FIG. 3) are added together, a smooth and continuous waveform 300 is obtained as shown in FIG.

  FIG. 5 shows a TM modulated signal 500, which is used to generate the modulated signal 300 shown in FIG. As shown in FIGS. 4 and 5, there is one TM modulation value 500 per carrier period. However, as described above, if the area under the curve may be different for each period, i.e., if there is no symmetry between the periods so that two symbols can be carried in each period, 1 There may be two TM modulation values per carrier period. In such a case, two different symbols (that is, information) can be expressed in one carrier cycle by having two TM modulation values per carrier cycle. This approach may be suitable for transmission over optical fiber, for example, while there is no other signal that occupies the transmission bandwidth, but DC shifts are typically limited to transmission over other media. This is because it is inappropriate.

A variable called TM modulation period t _TMM is the time during which the TM modulation value is held, which is an integer multiple of the carrier period. This means that in such a case, the maximum TM modulation frequency f _TMM is half of the carrier frequency f _C. That is, the modulation bandwidth is limited to ½ of f _C , which means that the lower limit of the sampling rate of Nyquist rate, ie, signal sampling without aliasing, is twice the bandwidth of the band-limited signal. As it is known. However, if there are two TM modulation values per carrier period, the maximum TM modulation frequency f _TMM is equal to the carrier frequency f _C. There is no minimum value of f _TMM that includes a DC response.

Referring back to FIG. 2, LUT 210 stores a quarter period that is unique to each value of TM modulation. There are four quarter periods in each carrier period (eg, as shown in FIG. 3). If one digital bit (N = 1) is allocated for each TM modulation period, only two unique TM modulation levels are stored in the LUT 210, ie two unique sets, one set consisting of two quarter periods. The first level represents a logic “0” and the second level represents a logic “1”. If 2 digital bits (N = 2) are assigned to each _tTMM , there will be 4 potential TM modulation levels. Similarly, when 3 bits (N = 3) are assigned to each t _TMM , there are 8 TM modulation levels (the same applies hereinafter).

The LUT 210 accommodates 2 ^N different quarter-period waveforms, and each total waveform is composed of four quarter-period waveforms, so a total of 4 × 2 ^N waveforms are accommodated. The number of time steps or clock periods per quarter cycle (eg, processor clock or CPU clock for reading the LUT 210) will depend on the allowable waveform fluctuations that the electronic device implementing the method can tolerate. . This may require nanosecond time steps if the carrier frequency is in the 300 MHz range. Lower carrier frequencies may be preferred by both TM schemes (eg, the LUT branch and “computation branch” described herein) and can be heterodyned to the carrier frequency.

In block 202, the TM modulated signal is input to the LUT 210. The TM modulation signal may be a signal including an arbitrary number of digital bits or a signal expressed thereby (for example, a signal having an N bit width). The LUT 210 contains a value or representation corresponding to each quarter period. This may be generated by the arithmetic branch 220 if it is not accommodated in the LUT 210. For example, for each TM modulation value that can be represented in row 210a (eg, 1 to 2 ^N ), a quarter period is associated with the TM modulation value and in column 210b (eg, from the initial time to a quarter period). It may be expressed and stored as coordinate data (eg, x, y) over a period of time. At block 204, a carrier signal having a carrier frequency f _C is input. The carrier signal may be an RF signal and may operate as a clock signal. At block 206, it is determined whether to use the LUT 210 or the arithmetic branch 220 to perform the modulation. Either the LUT 210 or the arithmetic branch 220 may be used to generate the modulated output signal. When the LUT 210 is used, the quarter period associated with the received TM modulation value is output from the LUT 210 to the analog gate 208.

When the arithmetic branch 220 is used, for example, when the arithmetic branch 210 is selected in the block 206, the TM modulation signal is input to the arithmetic block 220. The arithmetic block 220 outputs a quarter-cycle waveform that is substantially the same as the waveform that would be output by the LUT block 210 when the same TM modulation value was received. However, the calculation block 220 does not store the quarter period value associated with each TM modulation value, but generates a quarter period corresponding to each received TM modulation value. The arithmetic block 220 generates a modulated quarter period. This involves first cosine segments of 180 ° length at 0 ° to 90 °, 90 ° to 180 °, 180 ° to 270 °, and 270 ° to 2 times the carrier frequency (2f _C ). This is done by generating at each equivalent carrier frequency quadrant of 360 °. Therefore, these generated cosine segments constitute a quarter-period segment at the carrier frequency. Amplitude settings are received TM modulation values corresponding to 0 ° -90 ° quarter and 180 ° -270 ° quarter (ie, “first” and “third” quarter periods), and 90 °- This is done with complementary modulation values corresponding to 180 ° quadrants and 270 ° to 360 ° quadrants. As will be readily appreciated by those skilled in the art, any sinusoidal signal may be generated using known mathematical relationships, and this generation may be implemented in circuitry and / or software. Accordingly, a cosine segment of the operational branch 220 having an amplitude set by the received TM modulation value may be generated as such.

  Arithmetic branch 220 performs numerical operations to generate quarter-period segments using a processor having a clock multiple of the carrier frequency. This is to execute software code or to drive a hardware-based waveform generator, which may be any known waveform generator. The arithmetic branch 220 is likely to require a higher clock frequency than the LUT branch 210. The output from LUT 210 or compute branch 220 is directed to analog gate 208, which collects quarter periods into one continuous signal and passes it to the next heterodyne block 212.

For purposes of transmission and heterodyne, aspects of the present disclosure are considered from the frequency domain. 6 is a plot of the frequency spectrum of the TM modulation signal 300 shown in FIG. 4, where f _C is the carrier signal frequency, 2f _C , 3f _C, etc. are the second harmonic of the carrier frequency, the second 3 harmonics. The signal 300 has the spectrum shown in FIG. 6 at the origin, possibly with visible inflections.

In addition to the fundamental carrier frequency component 610, there is a third harmonic component 620 of the signal 300, which includes phase modulation. The TM modulation component exists only in the third harmonic. That is, the TM modulation component is the third harmonic component 620. There is no second harmonic signal. At block 214, the second harmonic signal is generated as a local oscillator, and the third harmonic component is heterodyned using a mixer circuit to produce two output frequencies (3f _C -2f _C ) and (3f _C + 2f _C ). Is generated. This is shown in FIG. The TM modulation component, ie the third harmonic component 620, is shifted down to the fundamental carrier frequency (signal 710). The summed component of the heterodyne processing, ie, the fifth harmonic component 730, can be removed at block 214 (eg, by the filter 810 shown in FIG. 8) and is transmitted to the output for transmission of the specified communication channel. It can be filtered to match.

  Unlike known modulation techniques, as provided by the present disclosure, the third harmonic is phase shifted, but this phase shift is a phase shift relative to the base carrier rather than the third harmonic. In the case of normal FM or PM transmission, it is the carrier wave itself that is phase-shifted. In TM, the fundamental does not change and the phase of the third harmonic is associated only with the fundamental.

  This difference is important for several reasons. For each half period of the basic carrier (ie, each TM modulation symbol), there are 1.5 periods of the third harmonic that is not modulated. The third harmonic changes only when the data changes (ie, when the TM modulation signal 500 changes). Thus, there is little impact on power and spectrum, and another reason why the present invention is transparent with conventional modulation is that in most practical applications, as with radio AM and FM broadcasts, the communication channel Alone, there may be more than 100 carrier periods per TM symbol, during which the third harmonic does not change (ie, there is no modulation change). The third harmonic is only shifted in phase (time) with respect to the fundamental wave.

In order to implement the QC method, it is necessary to widen the analog bandwidth at least three times the carrier frequency because the third harmonic (for example, 3f _C ) is used. Furthermore, the QC scheme requires the clock frequency to be 16 times the carrier signal frequency, simply because there are four time steps per quarter period. The QC may be heterodyne up to the desired carrier frequency after being generated in the lower carrier signal. This lower carrier frequency defines the upper limit frequency of the TM modulation value.

  9a and 9b are block diagrams illustrating a direct spectrum (DS) generation system and scheme in a further embodiment of the present disclosure. The DS generation method can be said to be a simpler implementation of TM modulation. In the DS system, the sideband spectrum is generated directly and its energy is added to everything else that exists within the bandwidth of the communication channel. The DS method is based on the frequency domain.

  Referring to FIG. 6, the existing transmitter has some complex modulation format. Typical types of complex modulation used include QAM, QPSK, OFDM and the like. The sideband energy in the existing modulation is represented by component 610 in FIG. When TM modulation is applied, a third harmonic is generated, and TM sideband energy is represented by component 620. Note that a second harmonic component may also be present, but this does not include modulation.

  The second harmonic signal is useful in that it can be used to shift the TM sideband energy 620 down to the fundamental carrier frequency 610. This is done by heterodyne processing using a mixer function that multiplies two sine wave input signals to produce a subtracted frequency output and an added frequency output. Referring to FIG. 7, hatching represents energy converted from the third harmonic 720 to the fundamental 710 and the fifth harmonic 730.

  The use of the second harmonic is optional. A stable second harmonic can be obtained by a phase-locked loop known in the art. Also, there may be non-linearity, and if so, some of the sideband energy may actually be down-converted, but this is also a stability as a down-conversion scheme. It may not be reliable.

  As required by communications regulations, every transmitter must use an output filter to ensure that no energy is radiated outside the designated communications channel. As shown in FIG. 8, an output filter 810 that eliminates harmonics may be utilized for transmission on a designated communication channel. This filter may include a passband 812.

  Figures 9a and 9b show two systems and schemes for direct spectrum generation utilizing the concepts described above. FIG. 9a shows a software-based system and scheme for direct spectrum generation, and FIG. 9b shows a hardware-based system and scheme for direct spectrum generation. In FIG. 9 a, a clock signal 910 and a digital modulation signal 920 are input to the microprocessor 901. In FIG. 9 b, the carrier signal 915 and the analog modulation signal 925 are input to the nonlinear analog circuit 902. A third harmonic sideband (eg, TM modulation component 620) is generated directly by microprocessor 901 and / or circuit 902 based on the input signal. Microprocessor 901 and / or circuit 902 further heterodynes third harmonic sideband 620 directly with input clock 910 (FIG. 9a) or carrier 915 (FIG. 9b) to produce sideband energy (eg, 710). It can be generated directly at the fundamental frequency. The DS method relies on software generation of the entire arithmetic expression or a non-linear analog circuit that executes the arithmetic expression. That is, the microprocessor 901 (FIG. 9a) and / or the circuit 902 (FIG. 9b) directly calculates and generates the third harmonic sideband 620 using known mathematical relationships based on the input signal. . The third harmonic sideband 620 is then heterodyned by the microprocessor 901 and / or the non-linear analog circuit 902 to shift the third harmonic sideband 620 to the fundamental frequency.

  In the following, systems and methods for receiving and demodulating transposed modulation are disclosed. FIG. 10 is a block diagram illustrating a system and method for demodulating a TM modulated signal, which may be referred to as “sub-cycle calibration” (SCC). The TM-modulated SCC demodulation scheme operates in the time domain, for example, by reconstructing a waveform (eg, signal 300 in FIG. 4) as shown in the QC scheme section.

  In the SCC method, the third harmonic is added to the received signal 1001 in a broadband environment. A phase-locked loop 1010 generates a strict, unmodulated third harmonic signal that is added or multiplied to the received signal 1001 at element 1020. Thereafter, each positive peak and negative peak voltage level is detected by a positive peak detector 1030 and / or a negative peak detector 1040, and a reference having a matching negative peak value and positive peak value (by a reference ramp generator 1050). Used for ramp generation. Thus, once every half cycle of the received signal 1001, a new reference ramp is generated, the system (ie, the occurrence of each peak) is calibrated. The ramp is regenerated every half cycle of the carrier signal 1001. The peak timing is used by the peak timing element 1060 to set the timing of the reference ramp. Inflection is detected by detectors 1030 and 1040, and the timing of the inflection is used for sampling the reference ramp output from reference ramp generator 1050 and holding the sampled ramp value. The voltage is an analog value of TM modulation and is output from the sample and hold element 1070 and may be used directly or converted to digital. The reference ramp has a positive slope from the negative half cycle of the carrier to the positive half cycle. In the next half cycle of the carrier (ie, from the positive half cycle to the negative half cycle), the reference ramp has a negative slope.

  One advantage of the SCC demodulation system and scheme is that it provides a robust demodulation technique. This is because SCC demodulation is only concerned with the occurrence of negative and positive peaks and the presence of inflections between these peaks. Therefore, SCC demodulation is much less susceptible to errors caused by noise than other demodulation techniques.

  FIG. 11 is a block diagram illustrating a demodulation system and method according to further embodiments of the present disclosure, which may be referred to as “third harmonic phase detection” (3PD). The TM modulation third harmonic phase detection (3PD) demodulation scheme operates by regenerating the third harmonic component and demodulating the phase modulation present in that component.

  As shown in FIG. 11, the received TM modulated signal 1101 is used by a phase-locked loop 1110 to generate a stable, unmodulated fundamental carrier signal, which is received by a subtracting element 1120 and received signal 1101. Is subtracted from. The output from the subtraction element 1120 can be filtered by a fundamental notch filter 1130 to remove any spurious emissions at the fundamental carrier frequency. Therefore, the remaining signal is sideband energy (eg, TM modulation component) that drives the third harmonic phase detector 1140. Third harmonic phase detector 1140 may be any known or conventional phase detector. The resulting output 1150 is a TM modulated analog value.

  FIG. 12 is a block diagram illustrating additional demodulation systems and methods according to another embodiment of the present disclosure. The demodulation system and method shown in FIG. 12 is a TM-modulated Fast Fourier Transform (TMFFT) demodulation scheme that operates by sideband spectrum analysis.

  Although it can be said that the TMFFT scheme provides the simplest hardware implementation, it can also be said to be the most complex in terms of signal processing. The TM-modulated received signal 1201 is quantized by an analog-digital converter and then analyzed by an FFT function 1210. After the receiver amplifies the signal to a level suitable for conversion to digital bits, the signal is output to element 1210. Element 1210 may be a processor such as a computer CPU or a more specialized processor such as a field programmable gate array or some custom integrated circuit designed specifically for the computation of Fourier transforms. The output of the FFT element 1210 is a number of data values representing the signal strength of the received TM signal 1201 at a plurality of discrete frequencies. The TM spectrum is known because it is related to the TM operating mode, i.e. the number of bits per symbol (i.e. the number of bits allocated per TM modulation period) and the symbol rate.

  The symbol rate is equal to the carrier frequency divided by the number of carrier periods per symbol. Mathematically speaking, in an easy-to-understand example, dividing the carrier frequency of 1 MHz by the number of carrier periods per symbol of 10 gives 100,000 symbols per second.

  The symbol frequency is 50,000 symbol periods per second, dividing 100,000 symbols per second by two.

  Therefore, the frequency of interest in this example is 50 kHz, that is, the symbol period frequency is 50 kHz above and 50 kHz below the carrier frequency. If there are many modulation levels, such as 6 bits per symbol or 64 modulation levels, 100 kHz and 150 kHz are also considered to include additional Bessel related sidebands to make the FFT demodulation process more accurate. Will come in. If the number of carrier periods per symbol is small, the demodulation error rate is reduced by increasing the number of sideband frequencies. Depending on the receiver, the carrier frequency may be heterodyned to an intermediate frequency (IF) for amplification, or it may be zeroed by being heterodyned to baseband.

  The 50 kHz FFT output value has a value that follows TM modulation. When TM modulation has 4 bits per symbol, the numerical values of the FFT output are batched to 16 levels and converted to 4 binary bits to generate a TM modulation value.

  FIG. 13 is a block diagram illustrating a TM transmitter 1300 that generates and transmits a signal consisting of a TM signal added to an existing signal (eg, a modulated RF signal). TM transmitter 1300 includes a carrier signal generation portion 1310 and a TM modulated signal processing portion 1320. An example implementation of the carrier signal generation portion 1310 is shown in the block diagram of FIG. 14, and an example implementation of the TM modulation signal processing portion 1320 is shown in the block diagram of FIG.

  The carrier signal generation portion 1310 obtains a low level sample of the existing signal 1301 (which may or may not be modulated) (eg, by a directional coupler 1312) and provides an existing conventional modulation (eg, It operates to remove any AM, FM, or any other conventional modulation format) to obtain a single fundamental frequency carrier signal (FC, or fundamental carrier). The existing conventional modulation may be removed from the sample of the existing signal 1301 by the bandpass filter stage 1314, which is selected to remove the conventional modulation from the frequency carrier signal. It may have a region. The second harmonic generator 1316 generates the second harmonic signal (H2, that is, the second harmonic) by, for example, multiplying the FC signal by itself. Similarly, the third harmonic generator 1318 generates the third harmonic signal (H3) by, for example, multiplying the FC signal by H2. Of course, the first and second harmonic generators 1316, 1318 may be any known scheme or circuit that generates harmonics (eg, including a phase-locked loop), or the scheme or A circuit may be included.

  As shown in FIG. 14, the bandpass filter stage 1314 includes a first SAW filter 1313 having a very narrow bandwidth, a gain stage 1315, a comparator 1317 for stabilizing the amplification, and a second narrow bandwidth. And a SAW filter 1310. A phase adjustment stage 1311 may be included in the carrier signal generation portion 1310 to lock the TM output fundamental carrier in phase matching with the existing fundamental signal (ie, FT). The second and third harmonic generators may be implemented as signal multipliers 1316, 1318.

  As shown in FIGS. 14 and 15, the TM base carrier and the existing signal base carrier are noted to be phase locked (eg, by the phase adjustment stage 1311). This may be accomplished by a feedback loop that places a TM modulation circuit inside a large phase-locked loop to perform phase matching between the TM output fundamental carrier signal and the existing signal fundamental carrier at the combiner 1338.

  The circuit shown in FIG. 14 is important in that it generates a signal at a frequency that depends entirely on the exact existing signal frequency. The off-frequency signal does not affect the operation of TM.

  The TM modulated signal is processed through a TM modulated signal processing portion 1320. TM modulation is performed on the third harmonic, and the frequency is converted to a fundamental wave (FC) and combined with the existing signal 1301.

  The TM modulated signal input 1302 to the TM modulated signal processing portion 1320 of the TM transmitter 1300 is analog in nature and band limited (eg, by a low pass modulation Nyquist limiting filter 1322), thereby reducing the communication channel bandwidth and Consistent sideband energy is generated. The TM modulated signal is then processed by an inverse optimizer 1324 and a TM modulator (or time shift modulator) 1326. As shown in FIG. 15, a gain stage 1321 may be included and the inverting optimizer 1324 may include a sample and hold function in addition to the optimization function. The low pass filter 1322, the gain stage 1321, and the inverting optimizer 1324 operate to limit the TM modulation bandwidth to the communication channel bandwidth. Optimization may be turned on or off depending on the presence or absence of an input signal to the inverting optimizer 1324.

  The third harmonic signal (H3) drives the TM modulator (or time shift modulator) 1326 by time shifting the third harmonic. This generates a set of Bessel function sidebands. Only one set of upper and lower sidebands is required for TM demodulation. These sidebands are limited in bandwidth relative to the third harmonic, and this limitation is achieved by filtering the TM modulated signal to match the bandwidth of the communication channel prior to TM modulation processing. Is called.

  The present inventor has confirmed in simulation that only one pair of sidebands is generated in the time shift modulation disclosed in the present specification, unlike the phase shift modulation. This is confirmed in the laboratory with an instrument such as an oscilloscope or spectrum analyzer. Phase modulation produces Bessel series sidebands as expected. On the other hand, in time shift modulation, only one set of upper sideband and lower sideband is generated.

  The TM modulator (or time shift modulator) 1326 can perform time shift modulation with an all-pass filter that is modified by a voltage controlled time delay. The control voltage is given by a conditional TM modulation signal (with or without optimization). A time shift is performed on the third harmonic (H3) signal. Although the TM modulator 1326 is primarily described herein with respect to time shift modulation, as will be appreciated by those skilled in the art, the TM modulator 1326 may be a phase shift modulator as well.

  In the following, the circuit, principle and functionality of the time delay shift available in the TM modulator (or time shift modulator) 1326 will be described in more detail. For a single frequency input signal such as a sine wave, the time delay is similar to a phase shift.

  The all-pass filter may include an operational amplifier, where the feedback resistor is connected between the output of the operational amplifier and the negative (inverting) input of the operational amplifier, and a second resistor of the same value is operated on. The amplifier is connected between the negative (inverting) input and the signal input, and the positive (non-inverting) input is connected to the middle connection point of the series network of capacitors and resistors. Connected to the input and grounded at the other end.

  The value of the series capacitor or resistor can be modified by the control signal by using a 4-quadrant multiplier, and the output of the 4-quadrant multiplier replaces the ground connection of the series RC network, and the 4-quadrant multiplier. Is connected to an intermediate connection point of the series network and a second input (ie, a TM modulated signal) as a control signal input.

  A phase shift may be generated as a time shift by a time delay shift of the input signal according to the input control signal.

  A TM on / off selector 1328 may be included, which may be an unmodulated third harmonic (H3) signal (eg, the output of the third harmonic generator 1318) or a TM modulated third harmonic signal ( For example, one of the outputs of the time shift modulator 1326 is selected. This function preserves the total power supplied to the transmit antenna 1340 regardless of whether TM is used.

  The TM-modulated H3 signal is reduced by multiplying the TM-modulated H3 signal and the second harmonic (H2) signal in a down-converter 1330 (or “heterodyne frequency conversion” block 1330 shown in FIG. 15). Converted. As a result, the sideband energy of the TM-modulated H3 signal is shifted to the FC frequency, and then modulated and bandpass filtered by the bandpass filter 1332 in the FC.

  The resulting FC-based TM signal then passes through an amplifier 1334 that performs conventional amplification to establish a usable power level. The amplifier 1334 may be an RF power amplifier, for example. The FC-based TM signal travels through the final band-limited bandpass filter 1338 and combiner 1338, and the TM output signal is added to the existing signal 1301. The resulting combined signal is connected to transmit antenna 1340 for transmission. The phase is locked between the TM output signal and the existing signal 1301.

  FIG. 16 is a block diagram illustrating a TM receiver 1600 that receives a combination signal 1641 (eg, a TM signal added to an existing signal), extracts the TM signal, and demodulates it.

  The TM receiver 1600 is as close as possible to the antenna 1640 or the first generated IF (intermediate frequency) (eg, the output of an existing receiver, where the intermediate frequency output is common for some communication equipment Obtain the received combined signal as close as possible.

  TM receiver 1600 includes a carrier signal and harmonic recovery portion 1610 and a TM separation and demodulation portion 1620. An example implementation of the carrier signal and harmonic reconstruction portion 1610 is shown in the block diagram of FIG. 17, and an example implementation of the TM separation and demodulation portion 1620 is shown in the block diagram of FIG.

  The carrier signal and harmonic restoration portion 1610 of the TM receiver 1600 includes (a) restoring an existing fundamental carrier signal (FC) as an unmodulated signal, and (b) a second harmonic of the restored fundamental wave signal. Circuitry for generating a signal and (c) generating a third harmonic signal of the recovered fundamental signal. All of these operate as local oscillator signals except that they are accurately derived from the received signal. The circuitry to do this is similar to that used in transmitter 1300.

  In the carrier signal and harmonic recovery portion 1610, a received signal 1641 (eg, an existing signal) from the receive antenna 1640 or from the IF (intermediate frequency) output of an existing receiver (which is common for some communication equipment). The combined RF signal) is filtered by a very narrow bandpass filter stage 1614 to remove any existing modulation, resulting in a pure fundamental carrier signal (FC or fundamental carrier). . FC is squared by second harmonic generator 1616 to generate second harmonic signal (H2). The FC and the second harmonic signal (H2) are multiplied by a third harmonic generator 1618 to generate a third harmonic signal (H3).

  As shown in FIG. 17, the front end of the receiver 1600 may have AGC (automatic gain control) controlled gain and may include a SAW filter 1613 and a gain stage 1615. Similar to the carrier signal generation portion 1310 of the transmitter 1300, the carrier signal and harmonic recovery portion 1610 of the receiver 1600 may include a comparator 1617 and a second SAW filter 1619. The carrier signal and harmonic restoration portion 1610 may include a phase adjustment stage 1611 that compensates for the phase shift of the SAW filter. The second and third harmonic generators may be implemented as signal multipliers 1616, 1618.

  In the TM separation and demodulation portion 1620, the received signal 1641 (after being processed with AGC control gain by the receiver front end) is bandpass filtered by a bandpass filter 1636 with a bandwidth equal to the bandwidth of the communication channel. The

  Thereafter, separation and extraction processing is performed on the wideband received signal. The first function is the difference processing between the received signal (implemented by time delay amplifier 1634) and the delayed version of that signal. The delay is equal to 1/4 of the period of the third harmonic. Time delay amplifier 1634 uses a time delay based filter circuit (eg, delay stage 1633 and differential amplification stage 1635) that separates the received TM energy.

  The separated signal (for example, a fundamental frequency difference signal) is up-converted in the up-converter 1630 by applying the (H2) signal. That is, the separated signal is heterodyned to the third harmonic frequency of the received fundamental wave signal (FC). This result is obtained by performing TM modulation on the signal of the third harmonic frequency (after being filtered by the bandpass filter 1632 and removing the product term of the fundamental wave). At this point, amplitude variations can occur due to the effects of existing carrier modulation and transmission media. Therefore, this signal is provided to the analog comparator 1650 along with the signal common reference, and the comparator 1650 generates a signal with no amplitude variation. By filtering with the bandpass filter 1631, the third harmonic carrier frequency subjected to TM modulation is selected, and other harmonics are removed.

  The output signal from the bandpass filter 1632 is not only directed to the analog comparator 1650 as described above, but is also used as an input to the TM signal detector 1628, where the TM signal detector 1628 is a correlation function (ie, TM present based on comparison or phase between signal received as output from bandpass filter 1632 and third harmonic signal (H3) received as output from third harmonic generator 1618 Is detected. The TM signal detector 1628 outputs a signal indicating whether TM is used, that is, whether a TM signal is present in the received signal 1641.

  In the extraction process, separation and extraction of the TM signal from the received signal 1641 is completed. The extracted signal (eg, the output from bandpass filter 1631) is the recovered reference third harmonic signal (eg, the received existing carrier signal used as a reference) that is required for TM information demodulation. (TM) derived from (H3)) and time-shifted TM modulation.

  The TM demodulator 1626 is subjected to TM modulation (received as input from the bandpass filter 1631) and the third harmonic signal (received as reference input from the third harmonic generator 1618). The TM signal is demodulated by detecting a time shift with respect to the third harmonic signal. The TM demodulator 1626 can detect a time shift between input signals by multiplying two signals to obtain a correlation function. Alternatively, the TM demodulator 1626 uses an exclusive OR function to detect the timing difference between the (H3) reference and the TM modulated signal from the separation and extraction process.

  The TM separation and demodulation portion 1620 may optionally include an inverse optimizer 1624 that recovers the signal transmitted from the transmitter having the inverse optimizer 1324 described above.

  The demodulated signal passes through the modulation low pass filter 1622 and all of the carrier and other noise sources are removed to obtain a TM modulated output signal 1602.

FIG. 18 shows a basic function set for extracting the received TM signal from the combination signal. The time delay differentiator or filter (ie, delay stage 1633) may have an optimally set delay time (.25) / (3 × f _fC ). Deviation from this value only reduces the level of the separated TM signal, but in practice, the level of the separated TM signal is canceled by a specific value. The extraction function only removes the amplitude variation of the separated signal.

  The time delay based filter circuit of the separation function of FIG. 18 has the natural frequency response behavior shown in FIG. There are periodic signals that cancel at frequencies such as DC and sixth harmonics.

  Existing signal receivers do not respond to the sideband energy of the TM signal. Adding TM to an existing signal has the effect of reducing the signal-to-noise ratio (SNR) of the received existing signal. Similarly, there is also noise contribution from the existing signal to the received TM signal.

  The TM receiver 1600 relies on the second and third harmonic signals associated with the received fundamental signal. Frequency shifts due to Doppler effects or signal path length variations (moving receiver or transmitter) do not affect the demodulation of the TM signal. This is because the entire process is based on the frequency of the existing signal.

  As described above with respect to FIGS. 13-19, TM modulation may be applied to a carrier signal and can be derived from any existing transmitter signal. The modulated TM signal may then be combined with an existing transmitter signal (whether modulated or not), thereby increasing the information bandwidth of the predefined communication channel. Although FIGS. 13-19 have been described with particular reference to TM signals, this is not intended to be limiting, and the same functions and principles can be applied to any modulated signal. Information may be conveyed as a time difference or phase angle difference between two signals of different frequencies.

  Thus, by adding a second carrier signal having a frequency that is harmonically related to the first carrier signal frequency and that is modulated with information that is unrelated to the information that modulates the first carrier signal. A method is provided for increasing the information bandwidth of any defined communication channel. This second signal and modulation sideband may be heterodyned to the frequency of the first carrier signal, and the modulation sideband may be less than or equal to the bandwidth of the communication channel.

  The first carrier signal and the second modulated carrier signal may be transmitted without any modulation.

  The first carrier signal may or may not be modulated (in any modulation scheme) and the second carrier signal may be time shift modulated or angle modulated. The second carrier signal may have a frequency that is in a known relationship with the frequency of the first carrier signal. Similarly, the second carrier signal may have a phase angle relationship or a timing relationship with the first carrier signal. The information carried may cause a time shift or angle modulation of the second carrier signal, and the means for modulating the second carrier signal may vary the time relationship or phase angle relationship with the first carrier signal. Good.

  If the first carrier signal is modulated by time-shift modulation or angle modulation of the second carrier signal, the sideband energy frequency-shifted so as to occupy the same frequency range as that modulation, or communication Sideband energy disposed within the communication channel bandwidth that would be used for The combination of the first carrier signal and the sideband of the second carrier signal may be transmitted together within the frequency limit of the communication channel and may be received by a receiver. Furthermore, the combination of these two carrier signals may be transmitted without band limitation and received by the receiving device.

  The receiving apparatus may demodulate the modulation information of the second carrier signal using the first carrier signal as a demodulation reference signal.

  The present specification also describes a quadrature transposition modulation method, i.e., adding a second transposition modulated (TM) signal to an existing transposition modulated signal to reduce the information bandwidth in a fixed communication channel. Provides a way to increase the information bandwidth given by the transposition modulation described previously.

  In one embodiment, the second transposed modulation signal is added to the existing transposed modulation signal by using a fundamental carrier signal frequency that is the same frequency as the existing transposed modulation signal and that is 90 degrees out of phase or quadrature in phase. May be. Thereby, the mutual inefficiency of transposition modulation and conventional amplitude modulation, frequency modulation, and phase modulation is maintained. Adding a quadrature transposed modulation signal also has a mutual inefficiency between the two transposed modulated carriers, both of which have no effect on the existing conventional modulated signal. Have

  Transpose modulation may be used in various ways, including for optical communications, for example. The present specification provides a method for placing a wideband transposed modulation signal directly on an optical frequency beam to increase the data bandwidth of the communication.

  The light beam is used because of its broadband characteristics. They are modulated by various means. Transpose modulation may be placed on the light beam for various modes of information transmission, all of which are contemplated by the present disclosure and are within the scope of the present disclosure.

  One such example can be described as follows. Any modulation scheme that carries information requires a carrier signal. In transposed modulation, an existing modulated carrier signal may be used as a transposed modulated carrier. Transposed modulation may also provide a carrier signal when no carrier signal is present. The transposed modulated carrier signal is used to drive the optical modulator regardless of whether an existing signal is used.

  Optical modulators vary from light emitting diode drivers and laser diode drivers to light beam modulators that change the opacity and phase of a light beam. The present disclosure provides that a transposed modulated carrier signal can be applied to a single modulator. This makes it possible to increase the existing information bandwidth using conventional amplitude modulation, frequency modulation, or phase modulation.

  If no other modulation is present, a transposed modulated signal with no band limitation is transmitted that will contain the third harmonic component of the transposed modulation. This places the fundamental frequency component at a lower frequency, which typically has less attenuation. Transposition demodulation relies on this component as a reference. The higher the frequency of the third harmonic component, the wider the bandwidth that allows the maximum information modulation bandwidth.

  In another embodiment, two separate light beams may be utilized, in which case the lower frequency beam is modulated in a conventional manner (eg, amplitude modulation or phase modulation) and for transposition modulation. Used as a reference carrier. The higher frequency light beam is used for the third harmonic component of the transposition modulation.

  Another mode in which transposition modulation can be used is ultrasonic communication, which can be used for underwater wireless communication, for example. For example, in one embodiment, a transposed modulation signal can be applied to an ultrasonic transducer to generate an acoustic signal, and the acoustic signal is received and demodulated to restore the original modulation information. Good. This modulation process imposes a near zero impedance drive on the transducer to force operation above the natural resonant frequency of the transducer.

  The receiver may use a natural broadband transducer based on a technique that converts acoustic energy into electrical energy without a resonant peak response in addition to the modulation broadband. Such receiving transducers are manufactured using MEMS (microelectromechanical system) technology that provides a broadband response with high sensitivity.

  One transposed modulated carrier frequency may be used. Two separate ultrasonic frequencies may be used to carry the fundamental carrier signal component and the third harmonic component signal separately.

  It is emphasized that the above-described embodiments of the present disclosure, and in particular, any “preferred” embodiments, are only possible examples and are described so that the principles of the present disclosure are clearly understood. I want to do it. Various changes and modifications may be made to the above-described embodiments of the present disclosure without departing substantially from the spirit and principle of the present disclosure. All such modifications and variations are intended to be included herein within the scope of this disclosure and protected by the following claims.

Claims (17)

A method for increasing the information bandwidth of a specified communication channel,
Receiving a first modulated signal having a first carrier signal frequency;
Receiving a second modulated signal having a second carrier signal frequency, wherein the second modulated signal is modulated with information independent of the information modulating the first carrier signal; Receiving the second modulated signal, wherein the second carrier signal frequency is harmonically or sub-harmonically related to the first carrier signal frequency;
Combining the first signal and the second signal;
Including methods.   The method of claim 1, wherein the second signal is heterodyned to a frequency of the first carrier signal. Transmitting the combined signal by a transmitting device;
Receiving the combined signal by a receiving device;
Demodulating the second modulated signal from the combined signal using the first carrier signal as a reference;
The method according to claim 1, further comprising:   A time shift modulator for time delay shifting an input signal in accordance with an input control signal, comprising an all pass filter modified by a voltage control time delay. The all-pass filter is
An operational amplifier comprising a feedback resistor connected between an output of the operational amplifier and an inverting input of the operational amplifier;
A second resistor having a resistance value substantially equal to a resistance value of the feedback resistor, the second resistor connected between the inverting input and the signal input of the operational amplifier;
A series network of capacitors and resistors, with a first end connected to the signal input, a second end connected to ground, and an intermediate connection point connected to the non-inverting input of the operational amplifier A series network of said capacitors and resistors,
The time shift modulator according to claim 4.   The value of the series capacitor or the series resistor is modified by a control signal by using a 4-quadrant multiplier, the output of the multiplier replaces the ground connection of the series network of the capacitor and resistor, and the 4 6. An input of a quadrant multiplier is connected to the intermediate connection point of the series network of capacitors and resistors, and a second input of the 4-quadrant multiplier is connected to the control signal input. The described time shift modulator. A method for increasing the communication bandwidth in a fixed communication channel,
Adding a second transposed modulated signal to a combined signal, the combined signal including a first transposed modulated signal and a first fundamental carrier signal, wherein the second transposed modulated signal The signal is added to the combined signal using a second basic carrier signal having the same frequency as the first basic carrier signal and a phase angle of 90 degrees with respect to the first basic carrier signal. Said adding step,
Including methods. A feature characterized by one or both of: (a) a function of doubling the information bandwidth without increasing the communication channel bandwidth; and (b) a function of improving the spectral efficiency of the transposition modulation. A method for increasing the communication bandwidth according to claim 7. A method for increasing the information bandwidth of ultrasonic communication,
Applying transposition modulation to the ultrasonic communication signal by direct amplitude modulation of a single ultrasonic transducer;
Including methods. A method for increasing the information bandwidth of ultrasonic communication,
Applying transposition modulation to the ultrasonic communication signal by directly amplitude modulating the first ultrasonic transducer with a fundamental carrier signal component of transposition modulation;
Direct amplitude modulating a second ultrasonic transducer with a third harmonic carrier signal component of transposition modulation;
Including methods. A method for increasing the information bandwidth of ultrasonic communication,
Adding a transposition modulated fundamental carrier signal and a third harmonic carrier signal component to the ultrasound communication signal by directly amplitude modulating a single ultrasound transducer using a broadband modulation technique;
Including methods. A method for increasing the information bandwidth of ultrasonic communication,
By directly angle-modulating the first ultrasonic transducer with the fundamental carrier signal component of transposition modulation and directly angle-modulating the second ultrasonic transducer with the third harmonic carrier signal component of transposition modulation, ultrasonic communication Applying transposition modulation to the signal;
Including methods. A method for increasing the information bandwidth of ultrasonic communication,
Adding a transposition modulated fundamental carrier signal and a third harmonic carrier signal component to the ultrasound communication signal by directly angularly modulating a single ultrasound transducer using a broadband modulation technique;
Including methods. A system for increasing optical information communication bandwidth,
A light beam,
An optical modulator,
A system configured to directly modulate the light beam with a transposed modulation signal.   The system of claim 14, wherein the system is configured to amplitude modulate the light beam with a transposed modulation signal, or to phase modulate the light beam with the transposed modulation signal. A method of increasing optical information communication bandwidth,
Directly modulating a light beam of a first frequency with a transposed modulated fundamental carrier frequency component;
Including methods. A method of increasing optical information communication bandwidth,
Directly modulating a light beam of a second frequency with a transposed modulated third harmonic component signal;
Including methods.

Images