JP6439460B2 - Drive device - Google Patents

Description

  The present invention relates to a driving device that drives a control electrode such as a gate electrode or a base electrode.

  Since supply of overcurrent to the switching element due to a short circuit leads to thermal destruction of the element, fail-safe control such as stopping the switching element when a short circuit is detected is performed. In detecting a short circuit, it is necessary to distinguish between overcurrent and pulse noise due to a short circuit. In order to determine that the current is an overcurrent, for example, the drive circuit described in Patent Document 1 has a predetermined filter time. That is, when the output current flowing through the switching element remains above a predetermined threshold during the set filter time, it is determined that the current is due to a short circuit.

  In other words, the state in which the output current flows through the switching element is maintained during the filter time. The output current at the time of the short circuit is larger than that at the time of normal driving, and power that is not originally required is consumed within the filter time, which may lead to overheating of the switching element.

  As a method of suppressing the heat generation amount of the switching element at the time of a short circuit, it is conceivable to reduce the saturation current of the output current, increase the heat capacity of the switching element, and improve the response speed of the short circuit protection. Among them, as a method for suppressing the output current, there are techniques described in Patent Document 2 and Patent Document 3, for example.

  The semiconductor device described in Patent Document 2 improves the load short-circuit tolerance while realizing a low on-voltage by optimizing the size of the effective region through which the output current flows.

  In addition, the load driving device described in Patent Document 3 includes a gate drive circuit and a clamp circuit. At the time of a short circuit, the gate of the IGBT is controlled using a clamp voltage smaller than the gate voltage at the time of normal driving where no short circuit has occurred.

JP 2014-23342 A JP 2013-84922 A JP 2012-204985 A

  However, since the semiconductor device described in Patent Document 2 has a configuration in which the output current is smaller than the conventional configuration, the current capability of the switching element is reduced. Since the current capability and the on-voltage are in a trade-off relationship, as a result of reducing the current capability, the on-voltage during normal driving increases and the loss increases.

  Further, in the load driving device described in Patent Document 3, in addition to an increase in circuit scale due to the need for two power supply circuits, this technique alone ensures sufficient load short-circuit withstand capability required in recent years. I can't.

  The present invention has been made in view of the above-described problems, and an object of the present invention is to provide a drive device that can ensure a short-circuit tolerance while suppressing loss during normal drive.

  The invention disclosed herein employs the following technical means to achieve the above object. Note that the reference numerals in parentheses described in the claims and in this section indicate a corresponding relationship with specific means described in the embodiments described later as one aspect, and limit the technical scope of the invention. Not what you want.

In order to achieve the above object, the present invention provides a drive device for driving a power switching element (200) having a control electrode (210) for controlling an output current supplied to a load when a voltage is applied. And a control unit (10) for controlling the timing of voltage application to the control electrode. The control electrode is divided into two or more divided electrodes including a first electrode (211) and a second electrode (212) for control. The unit drives the power switching element in a safe mode in which a voltage is applied to one of the first electrode and the second electrode in a short circuit detection period for detecting a short circuit of the load, and in a normal drive period for normally driving the load. The power switching element is driven in a normal mode in which a voltage is applied to both the first electrode and the second electrode.

  According to this, in the safe mode, which is a period for detecting a short circuit, a voltage is applied to either the first electrode or the second electrode constituting a part of the control electrode. For this reason, the area of the control electrode to which the voltage is applied is smaller than the entire control electrode. Therefore, the channel formation region in the switching element is smaller than when a voltage is applied to the entire control electrode, and the output current capability can be reduced. Therefore, the power consumption at the time of the short circuit at the time of safe mode can be suppressed, and short circuit tolerance can be improved by suppressing the heat_generation | fever of a switching element.

In addition, in the normal mode, the control unit, that to drive the power switching device by applying a voltage to both the first electrode and the second electrode. According to this, in the normal mode, a channel is formed by applying a voltage to the second electrode in addition to the first electrode. For this reason, in the normal mode, a larger amount of output current can be output than in the safe mode, and the conduction resistance is reduced, so that loss during normal driving can be reduced.

It is a block diagram which shows schematic structure of the drive device and IGBT in 1st Embodiment. It is a circuit diagram which shows schematic structure of a drive circuit and a short circuit detection circuit. It is a front view which shows the electrode structure of IGBT. It is a flowchart which shows the control flow by a control part. It is a flowchart which shows the control flow by a control part. It is a flowchart which shows the control flow by a control part. It is a flowchart which shows the control flow by a control part. It is the schematic which shows the voltage-current characteristic in IGBT. It is a figure which shows the concept of the heat transfer distance. It is a conceptual diagram which shows the temperature change in case a heat source exists uniformly on one surface. It is a conceptual diagram which shows the temperature change in the conditions of L> = 2W. It is a conceptual diagram which shows the temperature change in the conditions of L <2W. It is a conceptual diagram which shows the temperature change in the conditions which made D small from the state shown in FIG. It is a conceptual diagram which shows the temperature change in the conditions which made D small from the state shown in FIG. It is a conceptual diagram which shows the temperature change in the conditions which made L small from the state shown in FIG. It is a figure which shows simulation conditions collectively. It is the simulation result which shows the stripe width dependence of the temperature rise of IGBT in the semiconductor substrate which has silicon as a main component. It is a simulation result which shows the stripe width dependence of the temperature rise of IGBT in the semiconductor substrate which has a silicon carbide as a main component. It is a front view which shows the area | region structure of IGBT in the modification 1. It is a front view which shows the area | region structure of IGBT in the modification 1. It is a front view which shows the area | region structure of IGBT in the modification 1. It is a circuit diagram which shows schematic structure of the drive circuit in 2nd Embodiment. It is a flowchart which shows the control flow by a control part. It is a circuit diagram which shows schematic structure of the drive circuit in 3rd Embodiment. It is a flowchart which shows the control flow by a control part. It is a circuit diagram which shows schematic structure of the drive circuit and IGBT in other embodiment. It is a circuit diagram which shows the cell structure of IGBT in other embodiment. It is a circuit diagram which shows the cell structure of IGBT in other embodiment. It is a front view which shows the area | region structure of IGBT in other embodiment.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the following drawings, the same reference numerals are given to the same or equivalent parts.

(First embodiment)
First, a schematic configuration of a drive device according to the present embodiment and a switching element that is a drive target of the drive device will be described with reference to FIGS. FIG. 3 is a diagram for explaining the concept, and the scale is not strict.

  The drive device in the present embodiment is a circuit that controls on / off of a load such as a motor, for example. As shown in FIG. 1, the supply of current to the load 300 is controlled by turning on and off a switching element 200 such as an IGBT, MOSFET, or other transistor. The drive device 100 is a device that controls the on / off of the switching element 200 to control the supply of current to the load 300.

  In this embodiment, an IGBT is described as an example of the switching element 200. Therefore, the switching element 200 is expressed as IGBT200. Further, the output current described in the claims corresponds to a collector current flowing between the collector and the emitter of the IGBT 200. Furthermore, the control electrode described in the claims corresponds to the gate electrode 210 of the IGBT 200. The divided first electrode as the gate electrode 210 is referred to as a first gate electrode 211, and the second electrode is referred to as a second gate electrode 212. That is, the divided electrodes described in the claims correspond to the first gate electrode 211 and the second gate electrode 212 in the present embodiment.

  First, the driving device 100 will be described. As shown in FIG. 1, the drive device 100 in this embodiment includes a control unit 10, a drive circuit 20, and a short circuit detection circuit 30.

  The control unit 10 outputs a control signal to the drive circuit 20 and controls application of a voltage to the gate electrode 210. The control unit 10 determines which of the gate electrodes 210 the voltage is applied to under a predetermined condition. That is, voltage application timing to the first gate electrode 211 and the second gate electrode 212 is controlled. Detailed description of the control performed by the control unit 10 will be made later.

  The drive circuit 20 is a circuit for applying a voltage to the gate electrode 210 based on a control signal input from the control unit 10. The drive circuit 20 includes a first drive circuit 21 and a second drive circuit 22, and is a first gate electrode 211 that is a divided part of the gate electrode 210 and another divided part. A voltage can be applied independently to the second gate electrode 212.

  Specifically, as shown in FIG. 2, the drive circuit 20 includes a first drive circuit 21 and a second drive circuit 22.

  The first drive circuit 21 includes a switch S1 and a switch S2, and a resistor R1 and a resistor R2. The first gate electrode 211 is connected to the gate potential Vg via the resistor R1 and the switch S1. The first gate electrode 211 is connected to the reference potential VSS of the drive circuit 20 through the resistor R2 and the switch S2. Note that the switch S1 and the switch S2 respectively correspond to a first on-side circuit and a first off-side circuit described in the claims. In FIG. 2, each of the switches S1 and S2 is expressed as one switching device, but each of the switches S1 and S2 may include a temperature special adjustment circuit that relaxes the temperature characteristics of the gate potential Vg, A switch having a function of supplying a constant current to the gate electrode 210 may be used. Further, the resistors R1 and R2 may be configured to be interposed between the drive circuit 20 and the gate electrode 210 of the IGBT 200, like a resistor R3 of the second drive circuit 22 described later.

  The second drive circuit 22 includes a switch S3, a switch S4, and a resistor R3. The second gate electrode 212 is connected to the gate potential Vg via the resistor R3 and the switch S3. The second gate electrode 212 is connected to the reference potential VSS via the resistor R3 and the switch S4. Note that the switch S3 and the switch S4 correspond to a second on-side circuit and a second off-side circuit described in the claims, respectively. Each of the switches S3 and S4 may include a temperature adjustment circuit or the like, or may be a switch having a function of supplying a constant current to the gate electrode 210. Further, the resistor R3 may be divided into an on-side circuit and an off-side circuit, like the resistors R1 and R2 of the first drive circuit 21.

  The reference potential VSS is often set to the same potential as the emitter potential of the IGBT 200. However, a lower potential can be used for the purpose of preventing erroneous turn-on. It is not limited.

  When applying a voltage to the first gate electrode 211 to turn on the IGBT 200, the switch S2 is turned off and then the switch S1 is turned on to supply charges to the first gate electrode 211 from the gate potential Vg side. On the other hand, when the first gate electrode 211 is turned off, the switch S1 is turned off and then the switch S2 is turned on to draw charges from the first gate electrode 211 to the reference potential VSS side.

  When the voltage is applied to the second gate electrode 212 to turn on the IGBT 200, the switch S4 is turned off and the switch S3 is turned on to supply charges to the second gate electrode 212 from the gate potential Vg side. On the other hand, when the second gate electrode 212 is turned off, the switch S3 is turned off and then the switch S4 is turned on to draw charges from the second gate electrode 212 to the reference potential VSS side.

  On / off of each of the switches S <b> 1 to S <b> 4 is controlled based on a control signal output from the control unit 10. That is, the control unit 10 controls the application of voltage to the first gate electrode 211 and the second gate electrode 212 by controlling on / off of the switches S1 to S4.

  Note that the switching speed when turning on the IGBT 200 depends on the resistance value of the resistor R1. Further, the switching speed when turning off depends on the resistance value of the resistor R2. Resistor R3 affects the transition time between the normal mode and the safe mode described in detail later.

  The short circuit detection circuit 30 is connected to the sense emitter terminal of the IGBT 200 as shown in FIG. The sense emitter terminal is a terminal for detecting a sense current having a correlation with a collector current as an output current. The short circuit detection circuit 30 includes a shunt resistor R4, a subtraction circuit 31, a comparison circuit 32, and a power source that generates a reference potential Vref.

  The shunt resistor R4 is connected between the sense emitter terminal and the reference potential VSS. The sense current flows from the sense emitter terminal of the IGBT 200 to the reference potential VSS side via the shunt resistor R4.

  The subtraction circuit 31 is a circuit that takes the potentials at both ends of the shunt resistor R4 as inputs and outputs the difference. That is, the subtraction circuit 31 outputs the amount of voltage drop due to the shunt resistor R4. Since this voltage drop amount is proportional to the current value of the sense current, the sense current can be detected by detecting the voltage drop amount by the shunt resistor R4. Since the sense current has a correlation with the collector current, the current value of the collector current as the output current can be detected as a result.

  The comparison circuit 32 is a circuit that compares the voltage drop amount output from the subtraction circuit 31 with the reference potential Vref. The comparison circuit 32 outputs a detection signal to the control unit 10 when the voltage drop amount is equal to or higher than the reference potential Vref. The reference potential Vref is set to a voltage value corresponding to a threshold current when determining a short circuit. That is, the comparison circuit 32 outputs a detection signal to the control unit 10 when the collector current becomes equal to or higher than the threshold current defined by the reference potential Vref.

  Note that the reference potential Vref is variable. The control unit 10 can change the threshold current by controlling the reference potential Vref.

  Next, the IGBT 200 that is a switching element will be described. As shown in FIG. 3, the IGBT 200 in this embodiment is a generally known vertical insulated gate bipolar transistor, and is formed on a semiconductor substrate 290 having silicon as a main component, for example. The IGBT 200 has a gate electrode 210 as a control electrode on the surface layer of the main surface 290a of the semiconductor substrate 290, and a pad 220 that is electrically connected to the gate electrode 210 and applies a voltage. Although not shown, the semiconductor region has an emitter region and a collector region. And an emitter electrode formed on the main surface 290a and in contact with the emitter region; and a collector electrode formed on the back surface opposite to the main surface 290a and in contact with the collector region. ing. The emitter electrode is connected to the reference potential VSS, and the collector electrode is connected to VCC, which is a power supply voltage, through a parallel circuit of the load 300 and the freewheeling diode 400.

  The gate electrode 210 in the present embodiment includes a first gate electrode 211 and a second gate electrode 212, and is formed to extend in a predetermined direction along the main surface 290a. The first gate electrode 211 and the second gate electrode 212 are alternately formed side by side.

  The pad 220 includes a first pad 221 connected to the first gate electrode 211 via a wiring, and a second pad 222 connected to the second gate electrode 212 via the wiring. That is, when applying a potential to the first gate electrode 211, the control unit 10 instructs the first drive circuit 21 to apply a voltage to the first pad 221. As a result, the first drive circuit 21 applies a predetermined voltage to the first pad 221, and accordingly, a potential is applied to the first gate electrode 211. In addition, when applying a potential to the second gate electrode 212, the control unit 10 instructs the second drive circuit 22 to apply a voltage to the second pad 222. As a result, the second drive circuit 22 applies a predetermined voltage to the second pad 222, and accordingly, a potential is applied to the second gate electrode 212.

  The emitter region is in contact with the gate electrode 210 through an insulating film, and carriers are moved between the emitter region and the collector region when a voltage is applied to the gate electrode 210. A collector current as an output current flows between the emitter electrode and the collector electrode by the movement of the carriers.

  Since the IGBT 200 in the present embodiment has the first gate electrode 211 and the second gate electrode 212 as the gate electrode 210, the emitter region in contact with each gate electrode 210 is used to apply a voltage to the gate electrode 210. In contrast, a carrier movement path is generated independently. That is, as shown in FIG. 3, when a voltage is applied to the first gate electrode 211, the carrier moves when the voltage is applied to the first region 231 that becomes a carrier movement path and the second gate electrode 212. A second region 232 serving as a path exists. Although FIG. 3 is a top view, different hatching is applied to each region in order to clarify the region.

  As described above, the first gate electrode 211 and the second gate electrode 212 are extended in a predetermined direction while being alternately arranged. Therefore, the first region 231 and the second region 232 are adjacent to each other and are formed in a stripe shape along the extending direction of the gate electrode 210. In the present embodiment, as shown in FIG. 3, the first gate electrode 211 and the second gate electrode 212 are formed to be spaced apart from each other at the same interval, and the first region 231 and the second region 232 are Except for the formation position, they are substantially equivalent. If the width of the first region 231 in the direction orthogonal to the extending direction (alignment direction of the gate electrodes 210) is L and the distance between adjacent first regions 231 in the alignment direction is D, the IGBT 200 in the present embodiment. Then, L = D.

  In addition, it is preferable that at least one of the first region 231 and the second region 232 is formed so that the area of the output region is such that the output current can flow more than the rated current defined for the switching element. In other words, the first gate electrode 211, the second gate electrode 212, and the emitter region are configured such that when a predetermined voltage is applied to the first gate electrode 211 or the second gate electrode 212, a current exceeding the rated current flows. The semiconductor substrate 290 is preferably laid out.

  In the present embodiment, since the first region 231 and the second region 232 are equivalent to each other, the rated current defined for the switching element is applied regardless of the voltage applied to either the first gate electrode 211 or the second gate electrode 212. The above collector current flows.

  Next, the operation of the driving apparatus 100 according to the present embodiment will be described with reference to FIGS.

  Hereinafter, the operation when the IGBT 200 is turned on, the operation when the IGBT 200 is turned on, and the operation when the IGBT 200 is turned off will be described separately. 4 to 7, the detection signal is a signal output to the control unit 10 when the short circuit detection circuit 30 detects a short circuit. The short circuit detection circuit 30 outputs an H signal to the control unit 10 when a short circuit is detected, and an L signal when the short circuit is not detected. In addition, the operation mode is a mode in which the control unit 10 controls the IGBT 200, a voltage is not applied to the gate electrode 210, a stop mode in which the operation of the IGBT 200 is stopped, and a voltage is applied to some of the gate electrodes 210. Thus, there are a safe mode for driving the IGBT 200 and a normal mode controlled so that the current capability is higher than that in the safe mode.

<At turn-on>
First, the turn-on operation of the IGBT 200 in a state where no short circuit has occurred will be described with reference to FIG.

  Prior to time t1 shown in FIG. 4, control unit 10 is in the stop mode, and IGBT 200 is in the off state. No voltage is applied to both the first gate electrode 211 and the second gate electrode 212. Therefore, the collector current is not flowing.

  It is assumed that the control unit 10 receives a control signal from the outside so as to turn on the IGBT 200 at time t1. The controller 10 drives the IGBT 200 in the safe mode for the purpose of confirming whether or not there is a short circuit. Specifically, the control unit 10 outputs a control signal so as to apply a voltage to the first gate electrode 211 to the first drive circuit 21 at time t1. Thereby, a voltage is applied to the first gate electrode 211. Along with this, the current value of the collector current starts to rise. The collector current becomes a constant value after the overshoot occurs. Note that the control unit 10 starts counting the filter time from the start of the safe mode. This filter time corresponds to the short-circuit detection period described in the claims. The filter time may be counted by any circuit as long as the predetermined time can be counted, and any method of an analog circuit and a digital circuit can be employed. Further, the filter time is not limited to the method determined by the time from the time t1, for example, the voltage applied to the first gate electrode 211 is monitored, and the period until the voltage reaches a certain value is determined. It may be a filter time. Further, it may be after a certain time has elapsed since the voltage has reached a certain value.

  On the condition that the control unit 10 does not detect the detection signal from the short circuit detection circuit 30 at the time t2 after a predetermined filter time has elapsed from the time t1, the control unit 10 A control signal is output so that a voltage is applied to the two gate electrodes 212. In the present embodiment, as shown in FIG. 4, in the safe mode, the collector current does not exceed the threshold current, and therefore no detection signal is input to the control unit 10. Therefore, the control unit 10 shifts to the normal mode, and the second drive circuit 22 applies a voltage to the second gate electrode 212. Note that a voltage is continuously applied to the first gate electrode 211.

  At time t2, since voltage is applied to both the first gate electrode 211 and the second gate electrode 212, the current mode is improved in the normal mode compared to the safe mode. Since the load 300 of the present embodiment is a motor as described above and is a load having a relatively high inductance, the value of the collector current hardly changes even when the mode is shifted from the safe mode to the normal mode. However, since the area acting as a channel in the semiconductor substrate 290 increases, the on-resistance decreases. A period in which voltage is applied to both the first gate electrode 211 and the second gate electrode 212 and the IGBT 200 is normally driving the load 300 corresponds to a normal drive period described in the claims.

  Next, a turn-on operation of the IGBT 200 in a state where a short circuit of the load 300 has occurred will be described with reference to FIG.

  Prior to time t3 shown in FIG. 5, control unit 10 is in the stop mode, and IGBT 200 is in the off state. No voltage is applied to both the first gate electrode 211 and the second gate electrode 212. Therefore, the collector current is not flowing.

  It is assumed that the control unit 10 receives a control signal from the outside so as to turn on the IGBT 200 at time t3. The controller 10 drives the IGBT 200 in the safe mode for the purpose of confirming whether or not there is a short circuit. Since the specific driving is as described in the turn-on operation in which no short circuit occurs, detailed description is omitted. At time t3, a voltage is applied to the first gate electrode 211, and the current value of the collector current starts to increase.

  Here, since the short circuit of the load 300 is assumed, the collector current exceeds the threshold current and rises with the saturation current as the upper limit. At time t <b> 4 when the collector current reaches the threshold current, the short circuit detection circuit 30 outputs a detection signal to the control unit 10. The control unit 10 determines that the detection signal is continuously input for a predetermined period of time, determines that the short-circuit state is present at time t5 in FIG. 5, and stops the application of the voltage to the first gate electrode 211. After time t5, both the first gate electrode 211 and the second gate electrode 212 are off, and the control unit 10 is in the stop mode.

  In this example, since a short circuit is detected during the safe mode, a voltage is applied to the second gate electrode 212 and the IGBT 200 is stopped before it is fully turned on. The saturation current of the IGBT 200 is smaller when only the first gate electrode 211 is applied than when the voltage is applied to both the first gate electrode 211 and the gate electrode 212. For this reason, the power consumption of the IGBT 200 within the filter time can be reduced. Therefore, the amount of heat generated by the IGBT 200 can be reduced, and thermal destruction can be prevented.

<On state>
First, a state where no short circuit occurs while the IGBT 200 is operating in an on state will be described with reference to FIG. It is assumed that pulse noise is superimposed on the detection signal of the short circuit detection circuit 30. Here, the on state is a state in which a voltage is applied to both the first gate electrode 211 and the second gate electrode 212, and is a full on state in the present embodiment.

  Prior to time t7 shown in FIG. 6, the IGBT 200 is driven in a full-on state in which a voltage is applied to both the first gate electrode 211 and the second gate electrode 212. At time t7, pulse noise is superimposed on the detection signal. When the detection signal is input to the control unit 10, the control unit 10 shifts to the safe mode. That is, the control unit 10 stops applying the voltage to the second gate electrode 212 and starts counting the filter time. As described above, the method for counting the filter time does not matter. Also in the transition from the normal mode to the safe mode, the value of the collector current does not change because the rated current is secured only by applying the voltage to the first gate electrode 211.

  Since the duration of the pulse noise is sufficiently shorter than the filter time, the pulse noise does not continue between time t7 and time t8 set as the filter time. Therefore, the control unit 10 determines that the load 300 is not in a short-circuited state, and again shifts to the normal mode and applies a voltage to the second gate electrode 212 at time t8. Note that a filter that is sufficiently shorter than the above filter time may be set separately during the transition to the safe mode after pulse noise is superimposed on the detection signal at time t7. Thereby, unnecessary switching between the safe mode and the normal mode can be prevented.

  Next, with reference to FIG. 7, an operation when a short circuit occurs in the load 300 in the ON state will be described.

  Assume that a short circuit has occurred in the load 300 at time t9 when the IGBT 200 is operating at full on. Since the voltage drop due to the load 300 is eliminated, the power supply voltage VCC is directly applied between the collector electrode and the emitter electrode of the IGBT 200. As a result, the collector current of the IGBT 200 starts to rise.

  The collector current exceeds the threshold current and rises up to the saturation current. At time t <b> 10 when the collector current reaches the threshold current, the short circuit detection circuit 30 outputs a detection signal to the control unit 10. The control unit 10 turns off the application of the voltage to the second gate electrode 212 in consideration of the possibility of a short circuit of the load 300 using the detection signal input as a trigger. That is, the normal mode is shifted to the safe mode. In this example, since it is assumed that a short circuit occurs at time t9, the collector current is maintained to be equal to or higher than the threshold current, and therefore the detection signal is continuously input to the control unit 10. The control unit 10 determines that the detection signal is continuously input for a predetermined period of time, determines that the short-circuit state exists at time t11 in FIG. 7, and stops the application of the voltage to the first gate electrode 211. After time t11, both the first gate electrode 211 and the second gate electrode 212 are in the off state, and the control unit 10 has shifted to the stop mode.

  In this example, since a short circuit is detected during the normal mode, a voltage is applied to both the first gate electrode 211 and the second gate electrode 212 immediately after detection. However, when the collector current becomes equal to or greater than the threshold current and the mode is shifted to the safe mode, the application of the voltage to the second gate electrode 212 is released. The saturation current of the IGBT 200 is smaller when only the first gate electrode 211 is applied than when the voltage is applied to both the first gate electrode 211 and the gate electrode 212. For this reason, the power consumption of IGBT200 at the time of safe mode can be reduced. Therefore, the amount of heat generated by the IGBT 200 can be reduced, and thermal destruction can be prevented. And when it determines with it being in a short circuit state after transfer to safe mode, IGBT200 is turned off as it is, and the drive by the drive device 100 is stopped safely.

<At turn-off>
Regarding the turn-off of the IGBT 200, the application of voltage to the first gate electrode 211 and the second gate electrode 212 is turned off from the normal mode state. As the timing of voltage application off, the first gate electrode 211 and the second gate electrode 212 may be turned off in synchronization with each other, or may be turned off by shifting each other.

  Next, with reference to FIGS. 8-18, the effect of the drive device 100 which concerns on this embodiment is demonstrated. A shown in FIG. 8 shows the characteristics of the normal mode in the present embodiment, and B shows the characteristics of the safe mode. C indicates the characteristics of the voltage clamp method described in Patent Document 3, for example. FIG. 8 is a schematic diagram, and the scale of each axis is not strict.

  The control unit 10 in the drive device 100 has a normal mode and a safe mode as operation modes of the IGBT 200. As described above, in this embodiment, the state in which the voltage is applied only to the first gate electrode 211 is the safe mode, and the state in which the voltage is applied to both the first gate electrode 211 and the second gate electrode 212 is the normal mode. is there.

  When the load 300 is short-circuited, the collector-emitter voltage is larger than when no short-circuit is occurring, so that the collector current as an output current is saturated as shown in FIG. Since this saturation current is smaller in the safe mode than in the normal mode, when the IGBT 200 is driven in the safe mode when a short circuit occurs, the saturation current of the IGBT 200 is suppressed, and the collector current and the heat generation amount of the IGBT 200 can be suppressed. In the normal mode, the IGBT 200 can be driven with a larger current capability and lower on-voltage than in the safe mode. That is, the loss can be suppressed due to the low on-voltage in the normal mode, and the short-circuit withstand capability can be ensured by the low saturation current in the safe mode.

  Although the saturation current can be suppressed even when the voltage applied to the gate electrode 210 is clamped to a predetermined voltage smaller than that during full-on as in the conventional configuration shown in FIG. 8C, two power supply circuits are required. As a result, the circuit scale increases. On the other hand, in the configuration as in the present embodiment, the saturation current at the time of short circuit can be suppressed without adding a power supply circuit. Further, when the saturation current is not sufficiently suppressed only by the clamp method, for example, the saturation current can be suppressed by combining the techniques of Patent Document 2, but in that case, the ON voltage is sacrificed. If the driving device 100 in the present embodiment is employed, there is no such contradiction.

  By the way, in general, in a drive circuit for driving a switching element, as an output stage (portions indicated by switches S1 to S4 in FIG. 3), current capability is relatively high because of high-speed switching for the purpose of reducing switching loss. Requires a large output stage. Furthermore, for the purpose of reducing the loss, there may be provided a temperature adjusting circuit for the gate potential Vg, a function for making the current charged / discharged to the gate electrode 210 constant, or the like. On the other hand, by dividing the gate electrode 210 into a plurality as in the present embodiment, the gate electrode 210 that turns on the output stage having a relatively large current capability and a high function first, that is, the first gate electrode 211. Only the output stage related to can be made. In addition, it has been described above that the first gate electrode 211 and the second gate electrode 212 may be turned off at the time of turn-off, but in such a configuration, only one of the off-side output stages is also provided. The output stage has a relatively large current capability and a high function.

  Further, as in the present embodiment, when the main surface 290a of the semiconductor substrate 290 is divided into the first region 231 and the second region 232, and these have substantially the same area, the voltage to the first gate electrode 211 is increased. The switches S1 and S2 that control the application of the current can have a current capability that is half that of the embodiment in which the gate electrode 210 is not divided. Thereby, for example, the physique of the driver IC can be reduced.

  In addition, the IGBT 200 in the present embodiment includes a first region 231 and a second region 232 as regions serving as a current path for the collector current. Since the first region 231 and the second region 232 extend in a stripe shape adjacent to each other, for example, when the first region 231 and the second region 232 bisect the main surface 290a of the semiconductor substrate 290, In comparison, the temperature increase of the IGBT 200 can be reduced. Hereinafter, a specific description will be given with reference to FIGS.

  The inventor conducted thermal simulation on the IGBT 200 in which the insulated gate bipolar transistor is formed on the semiconductor substrate 290. The simulation is a one-dimensional simulation for calculating heat transfer in the direction in which the regions 231 and 232 are arranged.

  FIG. 9 is a diagram illustrating a distribution of temperature change ΔT after a predetermined time has elapsed when a heat source is present at a certain point. A broken line in FIG. 9 indicates ΔT with respect to a position on the main surface 290a. As shown in FIG. 9, the distance from the heat source to the place where the temperature change occurs is W. This distance W is a distance that is affected by the temperature rise due to heat transfer from the heat source, and is hereinafter referred to as a heat transfer distance W. The heat transfer distance W depends on the thermal property value of the substance constituting the semiconductor substrate 290.

  FIG. 10 shows a temperature change ΔT after a predetermined time when a heat source that generates a predetermined amount of heat per unit time is uniformly distributed on the main surface 290a of the semiconductor substrate 290 with respect to a position on the main surface 290a. FIG. The broken line shown in FIG. 10 is synonymous with the broken line in FIG. 9, and the superposition of the temperature changes caused by the heat source existing on the main surface 290a becomes the temperature change ΔT of the entire main surface 290a. The solid line in FIG. 10 indicates the overall temperature change ΔT. In the structure in which the main surface 290a of the semiconductor substrate 290 is not divided into the first region 231 and the second region 232 as in this embodiment, that is, in the conventional configuration, the temperature change distribution shown by the solid line in FIG.

  FIG. 11 shows a case where the IGBT 200 has the first region 231 and the second region 232 described above, a voltage is applied only to the first gate electrode 211 and the first region 231 generates heat, and L ≧ 2W It is a figure which shows temperature change (DELTA) T after progress for the predetermined time when satisfy | filling. As already described, the width in the direction orthogonal to the extending direction of the first regions 231 (the arrangement direction of the gate electrodes 210) is L, and the separation distance between adjacent first regions 231 in the arrangement direction is D. . Further, although D = L in the present embodiment, hereinafter, other than D = L will be described in order to explain the operational effect of the IGBT 200 having the first region 231 and the second region 232.

  As shown in FIG. 11, under the condition of L ≧ 2W, the peak of ΔT shows the same value as the condition of FIG. That is, even when the IGBT 200 includes the first region 231 and the second region 232 described above, it is equivalent to the conventional one in terms of temperature rise. Since the heat transfer distance W is sufficiently small with respect to the size of the semiconductor substrate 290, the condition of L ≧ 2W is, for example, that the first region 231 and the second region 232 bisect the main surface 290a of the semiconductor substrate 290. It is in a state.

  FIG. 12 is a diagram showing a temperature change ΔT after elapse of a predetermined time when the D / L ratio is increased with respect to FIG. 11 and L <2W is satisfied. Under this condition, since the width L of the first region 231 serving as the heat source is smaller than the condition of FIG. 11, the peak due to the superposition of temperature changes by the heat source is smaller than the condition shown in FIG. That is, the temperature increase of the IGBT 200 can be reduced as compared with the case where the first region 231 and the second region 232 bisect the main surface 290a of the semiconductor substrate 290. Note that the temperature change distribution in FIG. 12 is a distribution under conditions that satisfy the relationship D> W−L / 2.

  In order to ensure a sufficient collector current even in a state where a voltage is applied only to the first gate electrode 211 such as in the safe mode, the ratio of the first region 231 to the main surface 290a is preferably as large as possible. . In other words, it is preferable to reduce D while satisfying L <2W. FIG. 13 is a diagram showing a temperature change ΔT after a predetermined time elapses when D is reduced from the condition of FIG. 12 while L <2W and D> W−L / 2 are satisfied. Since D> W−L / 2 is satisfied, heat is not transferred from the adjacent first region 231 and the peak value of the temperature change ΔT is equivalent to the condition in FIG.

  FIG. 14 is a diagram showing a temperature change ΔT after a predetermined time has elapsed under the condition that D is further reduced from the condition of FIG. 13 and satisfies D ≦ W−L / 2. Since D> W−L / 2, which is a dimensional condition in FIG. 13, is not satisfied, heat transfer from an adjacent first region 231 reaches one first region 231 and causes thermal interference. For this reason, the peak value of the temperature change ΔT becomes larger than the condition of FIG. However, since L <2W is satisfied, the temperature increase of the IGBT 200 can be reduced as compared with the case where the first region 231 and the second region 232 bisect the main surface 290a of the semiconductor substrate 290. Furthermore, since the D / L can be reduced as compared with the conditions of FIG. 13, even when a voltage is applied only to the first gate electrode 211, a rated collector current can be secured even with a smaller chip area. Can do. If there is a demand to suppress the temperature rise compared to the conditions of FIG. 14, L may be designed to be small. FIG. 15 is a diagram showing a temperature change ΔT under the condition where L is reduced from the condition of FIG. As shown in FIG. 15, by reducing the width L of the first region 231, the peak value of the temperature change ΔT can be made smaller than the condition of FIG.

  The above-described simulation results under the respective conditions regarding the dimensions of L, D, and W will be comprehensively described with reference to FIG. FIG. 16 is a diagram in which L and D dimensional conditions obtained by obtaining the simulation results shown in FIGS. 11 to 15 are plotted with the horizontal axis representing L and the vertical axis representing D. Each point of a to e shown in FIG. 16 is a dimensional condition of L and D from which the simulation results shown in FIGS. 11 to 15 were obtained.

  In the region I defined by L ≧ 2W, ΔT does not depend on L and D, and is equal to ΔT in the related art that generates heat uniformly. In the region II defined by L <2W and D> W−L / 2, ΔT decreases as L decreases, but does not depend on D. In region III defined by D ≦ W−L / 2, ΔT decreases as L decreases, but ΔT increases as D decreases. That is, by setting the dimensions of L, D, and W as the conditions relating to the region II and the region III, the temperature increase of the IGBT 200 can be reduced. Furthermore, by setting the dimensions of L, D, and W as conditions related to region III, a sufficient collector current can be reduced even in a state where a voltage is applied only to the first gate electrode 211, such as in a safe mode. The temperature increase of the IGBT 200 can be suppressed while ensuring the area.

  Subsequently, the heat transfer distance W will be discussed with reference to FIGS. 17 and 18.

  FIG. 17 shows the result of simulation in the case where a voltage is applied only to the first gate electrode 211 using a semiconductor substrate 290 mainly composed of silicon. The horizontal axis is the stripe width a (= L = D), and the vertical axis is the temperature change ΔT at the highest temperature rise, that is, at the center of the first region 231. A series of plots is an elapsed time t from voltage application to the first gate electrode 211. Also, the plot is made at 5 μs, which is generally required from when the short circuit occurs until the protection function operates. The vertical axis is normalized assuming that the magnitude of the temperature change ΔT when a = 1000 μm and t = 5 μs is 1. According to FIG. 17, a smaller stripe width a can suppress the temperature rise of the IGBT 200, but there is almost no variation in the temperature change ΔT until it falls below a = 200 μm. As described above with reference to FIGS. 10 to 12, the heat transfer distance W in silicon is approximately 100 μm in view of the fact that ΔT does not change until L falls below 2 W.

FIG. 18 shows the result of simulation in the case where a semiconductor substrate 290 containing silicon carbide (SiC) as a main component is used and a voltage is applied only to the first gate electrode 211. The method for taking the vertical and horizontal axes is the same as in FIG. According to FIG. 18, the result when the main component of the semiconductor substrate 290 is silicon and the result when the main component is SiC are substantially similar to each other in the shape of the plot. In other words, when the plot showing the result when the main component is SiC is enlarged or reduced in the horizontal axis direction, the result substantially coincides with the result when the main component is silicon. The enlargement ratio or reduction ratio in the horizontal axis direction depends on the thermal property value of the substance constituting the semiconductor substrate 290. Specifically, the coefficient r expressed by Equation 1 is a factor of the enlargement ratio in the horizontal axis direction.

Here, c is specific heat, ρ is density, and k is thermal conductivity. As shown in FIG. 17, when the main substance constituting the semiconductor substrate 290 is silicon, the heat transfer distance W is about 100 μm, so that the specific heat, density, and thermal conductivity in silicon are c _Si and ρ _Si respectively. , K _Si , the heat transfer distance W in a general material can be expressed by Equation 2. For example, when the semiconductor substrate 290 is composed of SiC as a main component, the heat transfer distance W obtained by Expression 2 is approximately 158 μm.

  Further, it is preferable to satisfy the relationship of L = D as in the present embodiment. In the present embodiment, since the first gate electrode 211 and the second gate electrode 212 are formed substantially equivalent to each other, the first region 231 and the second region 232 are substantially equivalent to each other and L = D is satisfied. In such a configuration, since the first gate electrode 211 and the second gate electrode 212 are formed substantially equivalent to each other, the manufacturing process can be simplified.

(Modification 1)
In the above-described embodiment, as illustrated in FIG. 3, the first region 231 and the second region 232 are illustrated as extending in one direction while satisfying L = D. However, it is not limited to the said form.

  The width of the first region 231 and the separation distance D of the first region 231 need not be constant for all the first regions 231. For example, as shown in FIG. 19, D may be set relatively large in some sections of the main surface 290a, and L may be set relatively small in other sections. Also good.

  As shown in FIG. 20, a plurality of structures in which the first region 231 and the second region 232 are repeatedly formed may be formed.

  Further, the stripe shape formed by the first region 231 and the second region 232 does not have to be in one direction for all the first regions 231 and the second regions 232. For example, as shown in FIG. 21, the direction in which the stripes face may be orthogonal to a part of the main surface 290a and another section. In such a case, it is preferable that the first regions 231 that are energized in both the normal mode and the safe mode and that generate a relatively large amount of heat are not adjacent to each other at the boundary between both sections. Thereby, the temperature rise in the boundary of both divisions can be suppressed.

  Note that the first region 231 and the second region 232 are formed in a stripe shape includes the case where the regions 231 and 232 are repeatedly arranged in the longitudinal direction.

(Second Embodiment)
The second off-side circuit in the first embodiment corresponds to the switch S4 of the second drive circuit 22. On the other hand, as shown in FIG. 22, the second drive circuit 22 in the present embodiment has a second soft cutoff circuit 24 in addition to the switch S4 as the second off-side circuit 23. The second soft cutoff circuit 24 is connected to the second gate electrode 212 in parallel with the resistor R3. The second soft cutoff circuit 24 includes a resistor R5 and a switch S5, which are connected in series between the second gate electrode 212 and the reference potential VSS. The switch S5 is turned on when a detection signal is input by the short circuit detection circuit 30.

  The operation of the second soft cutoff circuit 24 will be described below with reference to FIG. 23, taking as an example a case where a short circuit is detected while the IGBT 200 is in the ON state.

  As shown in FIG. 23, it is assumed that a short circuit has occurred in the load 300 at time t13 when the IGBT 200 is operating at full on. The power supply voltage VCC is directly applied between the collector electrode and the emitter electrode of the IGBT 200, and the collector current of the IGBT 200 starts to rise.

  The collector current exceeds the threshold current and rises up to the saturation current. At time t14 when the collector current reaches the threshold current, the short circuit detection circuit 30 outputs a detection signal to the control unit 10. The control unit 10 turns off the application of the voltage to the second gate electrode 212 in consideration of the possibility of a short circuit of the load 300 using the detection signal input as a trigger. That is, the normal mode is shifted to the safe mode.

  By the way, at the time of transition from the normal mode to the safe mode, in the configuration without the second soft cutoff circuit 24, the second gate electrode is connected via the resistor R3 by turning on the switch S4 substantially simultaneously with turning off the switch S3. The electric charge 212 is extracted. On the other hand, in the embodiment in which the second off-side circuit 23 includes the second soft cutoff circuit 24 as in the present embodiment, the control unit 10 keeps the switch S4 off as shown in FIG. Then, instead of the switch S4, the switch S5 is turned on by the input of the detection signal. As a result, the charge accumulated in the second gate electrode 212 is released to the reference potential VSS side via the resistor R5.

  In this example, since it is assumed that a short circuit occurs at time t13, the collector current is maintained at or above the threshold current. For this reason, the detection signal continues to be input to the control unit 10. The control unit 10 determines that the load 300 is in a short-circuited state at time t15 on condition that the input of the detection signal continues for a predetermined time, and stops the voltage application to the first gate electrode 211. After time t15, both the first gate electrode 211 and the second gate electrode 212 are in an off state, and the control unit 10 is in the stop mode.

  The effects of the driving device 100 in this embodiment will be described below.

  When a short circuit occurs in the normal mode and the collector current rises quickly, an overcurrent flows before shifting to the safe mode. Thereafter, the voltage application to the second gate electrode 212 is turned off for the transition to the safe mode. However, if the off-speed is too high, the collector current rapidly decreases, which is caused by the induced electromotive force by the load 300. Excessive surge may occur.

  In the driving device 100 according to this embodiment, since the second drive circuit 22 includes the second soft cutoff circuit 24, the resistance value of the resistor R5 is appropriately set to be applied to the second gate electrode 212. The voltage slew rate can be increased, and an excessive surge can be suppressed.

  It should be noted that the charge extraction via the second soft cutoff circuit 24 is performed only when the load 300 is short-circuited, and the charge is preferably extracted via the switch S4 in a normal turn-off operation. According to this, the slew rate of the voltage at the time of short circuit depends only on the resistance value of the resistor R5, and the resistance value of the resistor R3 can be set independently. In other words, the mutual transition speed between the normal mode and the safe mode and the transition speed from the normal mode to the safe mode at the time of a short circuit can be set independently. That is, a surge due to a large current change can be suppressed without sacrificing the mutual transition speed between the normal mode and the safe mode.

(Third embodiment)
The first off-side circuit in the first embodiment and the second embodiment corresponds to the switch S2 of the first drive circuit 21. On the other hand, as shown in FIG. 24, the first drive circuit 21 in the present embodiment has a first soft cutoff circuit 26 as a first off-side circuit 25 in addition to the switch S2. The first soft cutoff circuit 26 is connected to the first gate electrode 211 in parallel with the resistor R2. The first soft cutoff circuit 26 includes a resistor R6 and a switch S6, which are connected in series between the first gate electrode 211 and the reference potential VSS. The switch S6 is turned on when a detection signal is input by the short circuit detection circuit 30.

  The operation of the first soft cutoff circuit 26 will be described below with reference to FIG. 25, taking as an example the case where a short circuit is detected while the IGBT 200 is in the ON state.

  As shown in FIG. 25, it is assumed that a short circuit has occurred in the load 300 at time t17 when the IGBT 200 is operating at full on. The power supply voltage VCC is directly applied between the collector electrode and the emitter electrode of the IGBT 200, and the collector current of the IGBT 200 starts to rise.

  The collector current exceeds the threshold current and reaches the saturation current. At time t18 when the collector current reaches the threshold current, the short circuit detection circuit 30 outputs a detection signal to the control unit 10. The control unit 10 turns off the application of the voltage to the second gate electrode 212 in consideration of the possibility of a short circuit of the load 300 using the detection signal input as a trigger. That is, the normal mode is shifted to the safe mode. The control unit 10 turns on the switch S4 substantially simultaneously with turning off the switch S3. Thereby, the charge of the second gate electrode 212 is extracted.

  In this example, since it is assumed that a short circuit occurs at time t17, the collector current maintains the saturation region. For this reason, the detection signal continues to be input to the control unit 10. The control unit 10 determines that the load 300 is in a short-circuit state at time t19 on condition that the input of the detection signal continues for a predetermined time, and stops the application of the voltage to the first gate electrode 211. That is, the mode shifts to the stop mode.

  By the way, when the transition from the safe mode to the stop mode is performed, in the configuration without the first soft cutoff circuit 26, the first gate electrode is connected via the resistor R2 by turning on the switch S2 substantially at the same time as turning off the switch S1. The electric charge 211 is extracted. On the other hand, in the form in which the first off-side circuit 25 includes the first soft cutoff circuit 26 as in the present embodiment, the control unit 10 keeps the switch S2 off as shown in FIG. Then, instead of the switch S2, the switch S6 is turned on by the input of the detection signal. As a result, the charge accumulated in the first gate electrode 211 passes through the resistor R6 to the reference potential VSS side.

  After time t19, both the first gate electrode 211 and the second gate electrode 212 are in an off state, and the control unit 10 is in a state of shifting to the stop mode.

  The effects of the driving device 100 in this embodiment will be described below.

  In the driving device 100 in this embodiment, since the first drive circuit 21 includes the first soft cutoff circuit 26, the first drive circuit 21 is applied to the first gate electrode 211 by appropriately setting the resistance value of the resistor R6. The voltage slew rate can be increased, and an excessive surge can be suppressed.

  It should be noted that the charge extraction via the first soft cutoff circuit 26 is performed only when the load 300 is short-circuited, and the charge is preferably extracted via the switch S2 in a normal turn-off operation. According to this, the slew rate of the voltage at the time of short circuit depends only on the resistance value of the resistor R6, and the resistance value of the resistor R2 can be set independently. In other words, the turn-off speed at the time of normal driving where no short circuit occurs and the turn-off speed at the time of short circuit can be set independently. Therefore, a surge due to a large current change can be suppressed during a short circuit.

(Other embodiments)
The preferred embodiments of the present invention have been described above. However, the present invention is not limited to the above-described embodiments, and various modifications can be made without departing from the spirit of the present invention.

  For example, as shown in FIG. 26, a voltage adjustment circuit 27 such as a clamp circuit may be constructed in the drive circuit 20 in each of the above-described embodiments. The voltage adjustment circuit 27 suppresses the voltage applied to the gate electrode 210 to be lower than that during normal driving when a short circuit occurs. By adjusting the safe mode by the control unit 10 and the gate voltage by the voltage adjustment circuit 27, the saturation current at the time of short circuit can be further suppressed as compared with the above-described embodiments. FIG. 26 shows an example in which the voltage adjustment circuit 27 is connected only to the first gate electrode 211 to which a voltage is applied even in the safe mode, but both the first gate electrode 211 and the second gate electrode 212 are connected. It may be configured to be connected.

  Moreover, it is preferable that the areas of the effective regions in the first region 231 and the second region 232 are appropriately set. Here, the effective region is a portion that becomes a carrier injection source when a voltage is applied to the gate electrode 210, and corresponds to an emitter region.

  When the main surface 290a of the semiconductor substrate 290 is viewed from the front, the area ratio of the effective region in the first region 231 is α1, the area ratio of the effective region in the second region 232 is α2, and functions as an insulated gate bipolar transistor in the main surface 290a. An area ratio of the first region 231 in the region to be performed is β. That is, the area ratio of the effective region of the first region is α1 × β with respect to the region functioning as the insulated gate bipolar transistor in the main surface 290a, and the area ratio of the effective region of the first region is α2 × (1−β ).

  In general, the saturation current monotonously increases with respect to the area of the effective region, and the ON voltage monotonously decreases with respect to the area of the effective region. In safe mode, it is necessary to pass a current higher than the rated current, and to ensure short circuit tolerance. It is necessary to make it smaller than a predetermined current value. That is, α1 × β is determined by the requirements of the entire system including the connected load 300 and specifications such as standards.

  On the other hand, in the normal mode, reducing the ON voltage as much as possible leads to a reduction in loss. Therefore, it is preferable that α1 × β + α2 × (1−β), which is the area ratio occupied by the effective region, be as large as possible. Since α1 × β of the first term is determined by the system specifications as described above, it is required to increase the second term. For example, α2 = 1, which is the maximum value, is adopted for α2 that is not related to short-circuit tolerance, and α1> β is satisfied by satisfying α1> β while keeping the restrictions due to the rated current and short-circuit tolerance. β) can be set as large as possible. As a result, it is possible to ensure a short-circuit tolerance while realizing a low on-voltage.

  Further, in each of the above-described embodiments and modification examples, the common gate potential Vg is applied to the first gate electrode 211 and the second gate electrode 212 through the resistors R1 and R3. However, separate power sources may be prepared for the first gate electrode 211 and the second gate electrode 212, respectively.

  For example, when a negative voltage is applied to the gate electrode 210 to be turned off, a storage layer is formed in a region adjacent to the insulating film of the gate electrode 210, so that carriers are discharged from the emitter electrode. It can be made easier.

  For the threshold currents described in the above embodiments and modifications, a first threshold value in the normal mode and a second threshold value in the safe mode are prepared, and the first threshold value and the second threshold value are different from each other. It is preferably set.

  Specifically, the first threshold value in the normal mode is set larger than the second threshold value in the safe mode. In the normal mode, the current capability is larger than in the safe mode, and the saturation current at the time of short circuit is large. By setting the first threshold value to be larger than the second threshold value, it is possible to ensure a noise margin as well as to detect a short circuit in the normal mode.

  In the above-described embodiments and modifications, the sense emitter terminal to which the short-circuit detection circuit 30 is connected has been simply described. Specifically, as shown in FIG. 27, the IGBT 200 includes a main cell 200a that extracts an output current and a sense cell 200b that has a sense emitter terminal and extracts a sense current.

  The main cell 200 a has a first gate electrode 211 a and a second gate electrode 212 a as the gate electrode 210, and the sense cell 200 b has a first gate electrode 211 b and a second gate electrode 212 b as the gate electrode 210. The gate electrodes 211b and 212b divided on the sense cell 200b side correspond to the sense cell side divided electrodes described in the claims. The first gate electrodes 211 a and 211 b are connected to the first drive circuit 21. The second gate electrodes 212 a and 212 b are connected to the second drive circuit 22.

  When a voltage is output from the first drive circuit 21 or the second drive circuit 22 to the gate electrode 210, a collector current that is an output current flows through the main cell 200a and a sense current flows through the sense cell 200b. The sense current has a correlation with the output current. In the short circuit detection circuit 30, the shunt resistor R4 is connected between the sense cell 200b and the reference potential VSS. The short circuit detection circuit 30 detects the short circuit by converting the sense current output from the sense cell 200b into a voltage by the shunt resistor R4.

  The sense cell 200b does not necessarily have two gate electrodes 210, the first gate electrode 211b and the second gate electrode 212b. As shown in FIG. 28, the sense cell 200b may be composed of only one gate electrode 210, for example, the first gate electrode 211b. However, a voltage must be applied to the gate electrode 210 of the sense cell 200b in both the normal mode and the safe mode. According to this, since the sense current does not fluctuate when switching between the normal mode and the safe mode, stable sense current detection can be performed. In addition, in the short circuit detection circuit 30 shown in FIG. 27 and FIG. 28, the circuit diagram downstream from the subtraction circuit 31 is omitted.

  Further, in each of the above-described embodiments and modifications, an example in which the gate electrode 210 of the IGBT 200 is divided into two divided electrodes is shown, but the number of divisions may be three or more. In FIG. 29, the gate electrode 210 is divided into three as divided electrodes, a first gate electrode 211, a second gate electrode 212, and a third gate electrode (the third gate electrode is not shown). In the example, a first region 231, a second region 232, and a third region 233 through which an output current flows when voltage is applied are shown.

  In this example, the first region 231, the second region 232, and the third region 233 are repeatedly formed in this order, forming a stripe shape. Also in this example, if a voltage is applied only to the first gate electrode 211 in the safe mode, the separation distance D described in the claims is the total value of the widths of the second region 232 and the third region 233. It corresponds to.

  In such a form, for example, a voltage can be applied to the first gate electrode 211 during the safe mode, and a voltage can be applied to the second gate electrode 212 in addition to the first gate electrode 211 during the normal mode. It is also possible to apply a voltage to the third gate electrode in addition to the first gate electrode 211 in the normal mode. Furthermore, a voltage can be applied to the second gate electrode 212 and the third gate electrode in addition to the first gate electrode 211 in the normal mode. In addition, in the safe mode, in addition to the first gate electrode 211, a voltage can be applied to one of the second gate electrode 212 and the third gate electrode. The combination of the gate electrodes 210 to which the voltage is applied in each mode is arbitrary, and a combination in accordance with the rated current for driving the load 300 and the specification of the saturation current at the time of short circuit can be selected. For example, since the current flowing through the load 300 does not need a current rating that is equal to or greater than the command value, a method of making the region driven in the safe mode variable according to the command value is effective. Alternatively, since the power at the time of the short circuit depends on the power supply voltage VCC, when this voltage is high, it is also effective to reduce the area of the region where current flows in the safe mode.

  In each of the above embodiments and modifications, the IGBT 200 is described as an example of the switching element, but the switching element is not limited to an insulated gate bipolar transistor. For example, MOSFETs, bipolar transistors, and other generally known elements may be employed. In addition, when adopting MOSFET, the collector in the description of each of the above embodiments and modifications may be rephrased as the drain, and the emitter may be rephrased as the source. When the bipolar transistor is employed, it may be rephrased based on the gate.

  Further, in each of the above-described embodiments and modifications, as shown in FIG. 1, the method of driving the load 300 with one switching element is described as an example. However, a method of configuring the upper and lower arms with a plurality of switching elements is adopted. You can also.

  Further, in each of the above-described embodiments and modifications, the example in which the short-circuit detection circuit 30 for detecting the short circuit of the load 300 is connected to the sense emitter terminal is shown, but the sense emitter terminal, in other words, the sense cell 200b is not necessarily included. There is no need to detect through. For example, the short circuit detection circuit 30 may be constructed at the output terminal of the main cell 200a. As an example of the short circuit detection circuit 30, a method of inserting a shunt resistor or a method of inserting a current sensor can be adopted. In addition, a short circuit may be detected based on the collector-emitter voltage Vce of the IGBT 200 and its time change dVce / dt. In this case, a short circuit can be determined if Vce = VCC when the IGBT 200 is on. Alternatively, the potential Vge of the gate electrode 210 may be detected and a short circuit may be detected based on the presence or absence of the mirror period.

  In the above description, the short circuit of the load 300 has been described as a short circuit. However, the present invention is not limited to the short circuit of the load, and all short circuits that significantly increase the current flowing through the switching element, such as an arm short circuit due to a short circuit of the diode 400. Can be applied to the mode.

DESCRIPTION OF SYMBOLS 100 ... Drive apparatus, 10 ... Control part, 20 ... Drive circuit, 30 short circuit detection circuit,
200 ... IGBT (switching element), 210 ... gate electrode, 211 ... first gate electrode (first electrode), 212 ... second gate electrode (second electrode)
300 ... Load

Claims (12)

A drive device for driving a power switching element (200) having a control electrode (210) for controlling an output current supplied to a load by applying a voltage,
A control unit (10) for controlling the timing of voltage application to the control electrode;
The control electrode is divided into two or more divided electrodes including a first electrode (211) and a second electrode (212),
The controller is
In the short circuit detection period for detecting a short circuit of the load, the power switching element is driven in a safe mode in which a voltage is applied to one of the first electrode and the second electrode,
In normal driving period for normally driving the load, said power switching device is driven in the normal mode for applying a voltage to both of the first electrode and the second electrode,
The divided electrodes are extended in a predetermined direction orthogonal to the depth direction on at least a part of the main surface (290a) orthogonal to the depth direction of the semiconductor substrate (290) on which the power switching element is formed. ,
When the main surface is viewed from the front, the main surface has a first region (231) through which an output current flows when a voltage is applied to the first electrode, and a voltage is applied to the second electrode. A second region (232) through which an output current flows,
The first region and the second region are formed in a stripe shape along the extending direction of the divided electrodes,
The control unit drives the power switching element by applying a voltage to the first electrode in the safe mode,
The first electrode is arranged in a direction perpendicular to the extending direction.
When the specific heat of the material constituting the semiconductor substrate is c, the density is ρ, the thermal conductivity is k, the specific heat of silicon is c _Si , the density is ρ _Si , and the thermal conductivity is k _Si , it is defined by Equation 3. The driving device is formed so as to satisfy a relationship of L <2W between the heat transfer distance W and the width L of the first region .

The first electrode is formed so as to satisfy a relationship of D ≦ W−L / 2 among the width L of the first region, the separation distance D of the first region, and the heat transfer distance W. The drive device according to claim 1 , wherein: A drive device for driving a power switching element (200) having a control electrode (210) for controlling an output current supplied to a load by applying a voltage,
A control unit (10) for controlling the timing of voltage application to the control electrode;
The control electrode is divided into two or more divided electrodes including a first electrode (211) and a second electrode (212),
The controller is
In the short circuit detection period for detecting a short circuit of the load, the power switching element is driven in a safe mode in which a voltage is applied to one of the first electrode and the second electrode,
Driving the power switching element in a normal mode in which a voltage is applied to both the first electrode and the second electrode in a normal driving period for normally driving the load;
A first drive circuit (21) for applying a voltage to the first electrode; a second drive circuit (22) for applying a voltage to the second electrode;
A short circuit detection circuit (30) that outputs a detection signal to the control unit when a short circuit of the power switching element is detected, and
The first drive circuit includes a first on-side circuit (S1) for turning on the application of the voltage to the first electrode, and a first off-side circuit for turning off the application of the voltage to the first electrode. (S2, 25)
The second drive circuit includes a second on-side circuit (S3) for turning on the application of the voltage to the second electrode, and a second off-side circuit for turning off the application of the voltage to the second electrode. (S4, 23)
When turning on the power switching element,
The control unit turns on only the application of voltage to the first electrode and drives in the safe mode,
When the detection signal is not input within the short-circuit detection period, the application of the voltage to the second electrode is turned on to shift to the normal mode,
When the detection signal is input within the short-circuit detection period, if the detection signal continues for the short-circuit detection period, the application of the voltage to the second electrode is not turned on. drive braking system shall be the control means controls so as to turn off the application of voltage. A drive device for driving a power switching element (200) having a control electrode (210) for controlling an output current supplied to a load by applying a voltage,
A control unit (10) for controlling the timing of voltage application to the control electrode;
The control electrode is divided into two or more divided electrodes including a first electrode (211) and a second electrode (212),
The controller is
In the short circuit detection period for detecting a short circuit of the load, the power switching element is driven in a safe mode in which a voltage is applied to one of the first electrode and the second electrode,
Driving the power switching element in a normal mode in which a voltage is applied to both the first electrode and the second electrode in a normal driving period for normally driving the load;
A first drive circuit (21) for applying a voltage to the first electrode; a second drive circuit (22) for applying a voltage to the second electrode;
A short circuit detection circuit (30) that outputs a detection signal to the control unit when a short circuit of the power switching element is detected, and
The first drive circuit includes a first on-side circuit (S1) for turning on the application of the voltage to the first electrode, and a first off-side circuit for turning off the application of the voltage to the first electrode. (S2, 25)
The second drive circuit includes a second on-side circuit (S3) for turning on the application of the voltage to the second electrode, and a second off-side circuit for turning off the application of the voltage to the second electrode. (S4, 23)
The normal driving period that the power switching element current is supplied to the first electrode and the second electrode is driven, in a case where the detection signal is inputted to the control unit,
The controller turns off the application of voltage to the second electrode and shifts to the safe mode,
If the detection signal continues for the short-circuit detection period, turn off the application of voltage to the first electrode;
The detection signal is controlled drive operated device you characterized in that as between turns on the voltage application to continue above again unless a second electrode shifts to the normal mode of the short detection period. The second off-side circuit according to claim 3, characterized in that it comprises a second soft cutoff circuit to slow (24) the time change of the output current when the voltage application to the second electrode is turned off Or the drive device of Claim 4 . The first off-side circuit according to claim 3, characterized in that it comprises a first soft cutoff circuit to slow (26) the time change of the output current when the voltage application to the first electrode is turned off The drive device of any one of -5 . At least one of the first drive circuit and the second drive circuit has a voltage adjustment circuit (27) for adjusting a voltage applied to the first electrode and the second electrode, respectively. The drive device according to any one of claims 3 to 6 . A drive device for driving a power switching element (200) having a control electrode (210) for controlling an output current supplied to a load by applying a voltage,
A control unit (10) for controlling the timing of voltage application to the control electrode;
The control electrode is divided into two or more divided electrodes including a first electrode (211) and a second electrode (212),
The controller is
In the short circuit detection period for detecting a short circuit of the load, the power switching element is driven in a safe mode in which a voltage is applied to one of the first electrode and the second electrode,
Driving the power switching element in a normal mode in which a voltage is applied to both the first electrode and the second electrode in a normal driving period for normally driving the load;
Furthermore, when a short circuit of the power switching element is detected, a short circuit detection circuit (30) that outputs a detection signal to the control unit,
The short circuit detection circuit outputs the detection signal when the current value of the input current is equal to or greater than a predetermined threshold current,
The threshold current has a first threshold value in a normal mode and a second threshold value in a safe mode,
The second threshold value, driving a kinematic device you characterized in that it is set to a value smaller than the first threshold value. The plurality of control electrodes are formed on a main surface (290a) orthogonal to the depth direction of the semiconductor substrate (290) on which the power switching element is formed,
When the main surface is viewed from the front, the main surface has a first region (231) through which an output current flows when a voltage is applied to the first electrode, and a voltage is applied to the second electrode. A second region (232) through which an output current flows,
2. At least one of the first region and the second region is set such that an area occupying the main surface is such that a current equal to or higher than a predetermined rated current flows as an output current. drive device according to any one of 1-8. The first region and the second region, the area occupied on each of the main surface, as the output current, to claim 9, characterized in that it is set to a predetermined rated current or more current flows The drive device described. A short-circuit detection circuit (30) that outputs a detection signal to the control unit when a short circuit of the power switching element is detected;
The power switching element has a main cell (200a) that generates the output current when a voltage is applied to the control electrode, and a sense cell (200b) for monitoring the output current of the power switching element,
The current input to the short-circuit detection circuit, a driving apparatus according to any one of claims 1 to 10, characterized in that the sense current output of the sense cells. The control electrode of the sense cell has a sense cell side divided electrode (211b, 212b) divided corresponding to the divided electrode in the main cell, and the sense cell side divided electrode corresponds to the divided electrode in the main cell. The driving device according to claim 11 , wherein the driving device is driven as described above.

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