JP6425621B2 - Power converter - Google Patents

Description

  The present invention relates to a power conversion device capable of reducing switching loss of a switching circuit formed of semiconductor elements and reducing winding loss of a transformer during switching.

The conventional power conversion device adjusts the inductance using, for example, an orthogonal transformer, and has a power supply configuration that eliminates output fluctuation (for example, Patent Document 1).
In addition, a loss is reduced by providing a gap in the core of the transformer and performing a soft switching by a leakage inductance (for example, Patent Document 2).

JP-A-7-335456 JP-A-5-67536

  In such a power conversion device, it is necessary to form a magnetic path orthogonal to the transformer, to provide a gap in the core in order to obtain a leakage flux, and so on. Problems such as increased loss of

  The present invention has been made to solve the above-mentioned problems, and it is possible to provide switching without providing a gap in the core of the transformer and without adopting a configuration in which the magnetic paths of the core are orthogonal to each other. Power that can reduce the winding loss of the transformer by reducing the switching loss by suppressing the inrush current at turn-on of the semiconductor element of the circuit, and attenuating the oscillating current generated in the transformer winding during the switching off period It aims at providing a converter.

In the power converter according to the present invention, a first switching circuit for converting a DC voltage to an AC voltage, a transformer connected to the first switching circuit, and a voltage rectified by a rectifying circuit connected to the transformer are output. The third winding magnetically coupled to the first winding on the input side of the transformer and the second winding on the output side, and the current flowing from the third winding to the load is turned on and off. A second switching circuit, and command means for controlling the second switching circuit such that current flows from the third winding to the load when the first switching circuit is off, the command means comprising: A power converter characterized in that the second switching circuit is turned on so that current flows from the third winding to the load before the switching circuit is turned off .

  The present invention can reduce switching loss by controlling the current flowing in the third winding of the transformer in synchronization with the switching circuit.

BRIEF DESCRIPTION OF THE DRAWINGS It is a circuit schematic of the power converter device which shows Embodiment 1 of this invention. FIG. 3 is a winding state diagram of a transformer showing Embodiment 1 of the present invention. It is explanatory drawing of the leakage magnetic field intensity of the transformer which shows Embodiment 1 of this invention. It is explanatory drawing of the leakage magnetic field intensity of the transformer which shows Embodiment 1 of this invention. It is a wave form diagram of the leakage inductance and the switching loss which show Embodiment 1 of this invention. It is a voltage waveform figure of the transformer which shows Embodiment 1 of this invention. It is an equivalent circuit schematic of the transformer which shows Embodiment 1 of this invention. It is a voltage waveform figure of the transformer which shows Embodiment 1 of this invention. It is a switch timing waveform chart which shows Embodiment 2 of this invention. It is a voltage waveform figure of the transformer which shows Embodiment 2 of this invention. It is the circuit schematic of the power converter device which shows Embodiment 3 of this invention. It is an equivalent circuit schematic of the transformer which shows Embodiment 3 of this invention. It is a voltage waveform figure of the transformer which shows Embodiment 3 of this invention. It is the circuit schematic of the power converter device which shows Embodiment 4 of this invention. It is a winding state figure of the transformer which shows Embodiment 4 of this invention. It is explanatory drawing of the leakage magnetic field intensity of the transformer which shows Embodiment 4 of this invention. It is explanatory drawing of the leakage magnetic field intensity of the transformer which shows Embodiment 4 of this invention. It is a switching timing waveform figure which shows Embodiment 4 of this invention.

Embodiment 1
Hereinafter, power converter 100 according to Embodiment 1 of the present invention will be described. FIG. 1 is a circuit schematic diagram of a power conversion device 100 according to a first embodiment of the present invention.
As shown in FIG. 1, a DC voltage source 1 and a first switching circuit 2 connected to the DC voltage source 1 for converting a DC voltage to an AC voltage, the first switching circuit 2 is a semiconductor element such as a MOSFET (Metal (for example) -Oxide-semiconductor field-effect transistor) etc. A transformer 3 connected to the first switching circuit 2 for boosting or stepping down an AC voltage, in which the windings 3a, 3b and 3c are wound around the core at a ratio of n1: n3: n2, respectively, and 3e is The leakage inductance of the transformer 3 is shown. A rectifier circuit 4 connected to the winding 3b to rectify a transformed alternating current to a direct current, connected to the rectifier circuit 4 and connected to a load 5 operating with the rectified DC voltage and a winding 3c to damp a resonant voltage Damping resistor 7, second switching circuit 6 for turning on and off the connection between winding 3c and damping resistor 7, command means 200 for commanding first switching circuit 2 and second switching circuit 6 to turn on and off Configured The command means 200 is configured by a CPU or a logic IC. The current flowing in the winding 3 c is controlled by the command means 200 and the second switching circuit 6.

The operation of power converter 100 configured as described above will be described.
When the first switching circuit 2 is turned on, the second switching circuit 6 is controlled to be turned off based on the command from the command means 200. As a result, no current flows in the winding 3c, the degree of coupling of the transformer 3 decreases, and the leakage inductance 3e increases, so that the current flowing into the first switching circuit 2 is current limited, and the first switching circuit 2 is turned on. Switching loss can be reduced.
When the first switching circuit 2 is turned off, the second switching circuit 6 is controlled to be turned on based on the command from the command means 200. Thereby, the damping resistance 7 acts to suppress the resonance voltage of the winding 3a, and the winding loss of the transformer 3 can be reduced.

First, an operation capable of reducing the switching loss when the first switching circuit 2 is on will be described in detail.
2, 3 and 4 are diagrams for explaining that the leakage inductance 3e of the transformer 3 can be adjusted by the on / off control of the second switching circuit 6. FIG.
In general, the leakage inductance of the transformer is expressed by the following equation, using magnetic energy W stored in the magnetic circuit, and assuming that L1 is the leakage inductance converted to primary and the current is i1.

Further, assuming that the space of the leakage flux is S, the magnetic field strength of the leakage flux is H, the magnetic flux density is B, and the magnetic permeability is μ, the magnetic energy W is expressed by the following equation.

  From this relationship, it can be seen that if the magnetic field strength H of the leakage flux is reduced, the magnetic energy W stored in the space is reduced, and the leakage inductance can be reduced.

FIG. 2 shows the state of the windings of the transformer 3, which are concentrically wound around the core 3f in the order of the windings 3a, 3c and 3b. FIG. 3 is an enlarged view of a part of the winding arrangement of FIG. 2 (a portion surrounded by a broken line in FIG. 3), starting from the core 3f of the transformer 3 and the current flowing in the winding according to the distance d It is explained that the magnetic field strength of the leakage flux changes depending on the direction.
In FIG. 3, when the second switching circuit 6 is on, for example, a current of 2I flows through the winding 3a from the back to the front of the drawing, and each of the winding 3c and the winding 3b is from the front to the back A current of I flows toward.
Assuming that the current I flows and the magnetic field H0 of the leakage flux is generated, the current becomes 2I at a distance d1 where all the windings 3a are included, and a magnetic field of 2H0 is generated, and at a distance d2 including all the windings 3c, the windings 3c Since the current I flows in the opposite direction to the winding 3a, the magnetic field is reduced by H0 to H0, and the current I flows in the opposite direction to the winding 3a at the distance d3 including all the windings 3b, so the magnetic field is further reduced H0. It becomes 0. Thus, taking the magnetic field strength H of the leakage flux on the vertical axis and the distance d on the horizontal axis, the area X1 surrounded by the graph of the magnetic field of the leakage flux generated by the current flowing in the winding described above is the leakage inductance 3e of the transformer 3 Represents the size of the

Similarly, when the second switching circuit 6 is turned off in FIG. 4, a current of 2I flows in the winding 3a from the back to the front of the drawing, so a magnetic field of 2H0 is generated at the distance d1. Since no current flows in winding 3c, there is no change in the magnetic field at distance d2 and a magnetic field of 2H0 is generated, and current 2I flows in the winding 3b from the front to the back of the drawing sheet, so the magnetic field is reduced by 2H0 at distance d3. It becomes 0.
Therefore, the magnitude X2 of the leakage inductance indicated by the area surrounded by the magnetic flux graph of the leakage flux is larger than X1 in FIG. 3 in which the second switching circuit 6 described above is on. As described above, when the second switching circuit 6 is turned on and off, the energy (area surrounded by the graph) stored in the magnetic circuit changes depending on whether current flows through the winding 3c or not. , Adjustment of the leakage inductance 3e of the transformer 3 becomes possible.

FIG. 5 is a waveform diagram illustrating that the leakage loss 3e can be adjusted as described above to reduce the switching loss of the first switching circuit 2 formed of a semiconductor element.
In the figure, the upper part shows the relationship between the voltage and current between the drain and the source of the semiconductor element, and the lower part shows the change in switching loss due to the change in current. The vertical axis represents the size, and the horizontal axis represents the passage of time.
When the first switching circuit 2 is turned on at time t1, in the state in which the second switching circuit 6 is turned off, the leakage inductance can be made larger than in the state where the second switching circuit 6 is turned on. The current between the drain and the source is limited, and the switching loss can be reduced.

Next, an operation capable of reducing the winding loss of the transformer 3 when the first switching circuit 2 is off will be described in detail.
FIG. 6 is a voltage waveform diagram of the transformer 3 when the first switching circuit 2 performs the on / off switching operation. At this time, the second switching circuit 6 is off.
FIG. 7 shows an equivalent circuit obtained by converting the primary side of the transformer 3 when the first switching circuit 2 is off. The transformer 3 is represented by a leakage inductance 3 e and a winding resistance 3 g of the transformer. The parasitic capacitor component 2a constitutes a resonant circuit, and a resonant voltage as shown in the t2-t3 and t4-t5 regions of FIG. 6 is generated.
As shown in FIG. 8, when the second switching circuit 6 is turned on at timings t2 and t4 at the timing when the resonance voltage is generated (the first switching circuit 2 is turned off), the damping resistor 7 is connected. Assuming that the resistance value of the damping resistor 7 shown in FIG. 7 is R, a resistor represented by the following equation is connected in parallel to the transformer 3 using the number n1 of turns of the winding 3a and the number n2 of turns of the winding 3b. Is dumped.

  As described above, in the configuration of the first embodiment, by turning off the second switching circuit by the command means 200, control can be performed so that current does not flow to the winding 3c of the transformer, and the leakage inductance can be made large. Since the slope of the current is reduced, it is possible to reduce the switching loss that occurs when the first switching circuit 2 is turned on. Further, by turning on the second switching circuit by the command means 200, it is possible to reduce the winding loss of the transformer 3 by performing control so that current flows to the winding 3c of the transformer 3.

Embodiment 2
In the first embodiment, as described in FIG. 8, the first switching circuit 2 and the second switching circuit 6 are switched at the same timing. As described in the first embodiment, the leakage inductance 3e is small when the second switching circuit 6 is on, and the leakage inductance 3e is large when the second switching circuit 6 is off. The vibration energy generated during the period in which the first switching circuit 2 is off increases as the leakage inductance 3 e increases. Therefore, it is desirable to reduce the vibration energy as much as possible.
That is, in the voltage waveform of the transformer 3 at the time of control of the switching circuit of FIG. 9, when the second switching circuit 6 is operated at timing A as in FIG. 8, the first one is performed at timings t2 and t4 where the leakage inductance 3e is large. The switching circuit 2 is turned off, and the resonant voltage generated in the winding 3a immediately after the turning off is increased.
On the other hand, as in the timing B, the second switching circuit 6 is turned on by Δt1 before the first switching circuit 2 is turned off. As a result, the first switching circuit 2 is turned off in a state where the leakage inductance 3e of the transformer is reduced, so that the resonance voltage immediately after the turn-off generated in the winding 3a is reduced and the winding loss of the transformer 3 is further reduced. it can.
FIG. 10 is a waveform diagram showing the convergence state of the transformer voltage Vt at the timing A and the timing B described in FIG. The broken line indicates timing A and the solid line indicates timing B. The transformer voltage converges faster in the operation at timing B.
9 and 10, the second switching circuit is off at the timing when the first switching circuit is turned on, as in FIG. 8, and the leakage inductance is controlled by controlling so that no current flows to the winding 3c of the transformer. Therefore, the switching loss generated when the first switching circuit 2 is turned on can be reduced.

  As described above, in the configuration of the second embodiment, the second switching circuit 6 is turned on by the command means 200 before the first switching circuit 2 is turned off so that a current flows to the winding 3c. The resonant voltage generated in the winding 3a immediately after the first switching circuit 2 is turned off can be reduced, and the winding loss of the transformer 3 can be further reduced.

この共振回路のアドミタンスＹは、共振回路の容量をＣ、漏れインダクタンス３ｅの値をＬ、トランス３の巻線抵抗３ｇの抵抗値をＲａｃとすると、次式で表される。

これより、振動させないための条件が次式で表される。

一例であるが、仮に共振している周波数でのRac＝1Ω、漏れインダクタンスL＝1μH、巻線比n1/n2＝5の時、C1に40μFを接続すると、上記式の左辺は1.6となり、１より大きくなるため、共振が成立しないことになる。
以上のように、容量性負荷８を共振が生じない値に設定することで、図１３に示すように、第１のスイッチング回路２がオフになるタイミングで第２のスイッチング回路６をオンすることにより、巻線３ａに共振電圧が生じず、トランス３の巻線損失低減が図れる。 Third Embodiment
In the first and second embodiments, the damping resistor 7 is used to reduce the resonance voltage. However, in this embodiment, the capacitive load 8 is connected to the winding 3 c via the second switching circuit 6.
FIG. 11 is a schematic circuit diagram of a power conversion device 110 according to a third embodiment of the present invention, in which the same reference numerals as in FIG. 1 denote the same or corresponding parts.
Next, the operation will be described. When the first switching circuit 2 is turned off based on the command from the command means 200, the second switching circuit 6 is turned on, and the capacitive load 8 connected to the winding 3c generates a resonant voltage in the winding 3a. The winding loss of the transformer 3 can be reduced so as not to occur.
As shown in FIG. 12 as an equivalent circuit obtained by primary conversion of the transformer 3, the transformer 3 is represented by the leakage inductance 3e and the winding resistance 3g, and when the first switching circuit 2 is off, the first switching circuit 2 is The parasitic capacitor component 2a forms a resonant circuit to generate a resonant voltage (see t2-t3 and t4-t5 regions in FIG. 6).
On the other hand, when the second switching circuit 6 is turned on, assuming that the capacitance value of the capacitive load 8 is C1, using the number n1 of turns of the winding 3a and the number n2 of turns of the winding 3b, the capacitance Are connected in parallel to the transformer 3, and the resonant voltage fluctuates.

The admittance Y of this resonant circuit is expressed by the following equation, where C is the capacitance of the resonant circuit, L is the value of the leakage inductance 3e, and Rac is the resistance value of the winding resistance 3g of the transformer 3.

From this, the condition for preventing vibration is expressed by the following equation.

As an example, if 40 μF is connected to C1 when Rac = 1Ω, leakage inductance L = 1 μH, and winding ratio n1 / n2 = 5 at a resonating frequency, the left side of the above equation becomes 1.6, 1 As this becomes larger, resonance will not be established.
As described above, by setting the capacitive load 8 to a value at which resonance does not occur, as shown in FIG. 13, the second switching circuit 6 is turned on at the timing when the first switching circuit 2 is turned off. As a result, no resonance voltage is generated in the winding 3a, and the winding loss of the transformer 3 can be reduced.

  Thus, in the configuration of the third embodiment, the first switching circuit 2 is controlled by the command means 200 by connecting the capacitive load 8 set to a value at which no resonance occurs instead of the damping resistor 7 to the winding 3a. By controlling to turn on the second switching when turning off, it is possible to further reduce the winding loss of the transformer 3 without generating a resonant voltage.

Fourth Embodiment
FIG. 14 is a circuit schematic diagram of power converter 120 capable of further reducing the leakage inductance of the transformer of the first embodiment. In the figure, the same reference numerals as those in Embodiment 1 denote the same or corresponding components.
In the transformer 31, the windings 3a, 3h, 3b and 3c are wound around the core at a ratio of n1: n1: n3: n2, respectively. 31e indicates the leakage inductance of the transformer 31. The winding 3a and the winding 3h of the transformer 31 are connected in parallel, the winding 3b is connected to the rectifier circuit 4, and the load 5 is operated by the rectified DC voltage. The winding 3 c is connected to the rectifier circuit 11 via the third switching circuit 9 and operates the load 12 with the rectified voltage. The winding 3 c is connected to the damping resistor 7 via the fourth switching circuit 10. The third switching circuit 9 turns on and off the connection from the rectifier circuit 11 to the load 12 based on the command from the command means 210, and the fourth switching circuit 10 receives the damping resistor 7 based on the command from the command means 210. Switch on and off the connection.
The power conversion device 120 configured in this way turns on the third switching circuit 9 when the first switching circuit 2 is on based on the command from the command unit 210, and the fourth switching circuit 10 is turned on. Turn off. As a result, when a current flows through the winding 3c, the leakage magnetic field is canceled, so the degree of coupling of the transformer 31 is increased, the leakage inductance 31e is decreased, and the increase in winding resistance due to magnetic interference is suppressed, and the loss can be reduced. .

The operation will be described in detail below.
FIG. 15 shows the state of the windings of the transformer 31, which is concentrically wound around the core 3i in the order of the windings 3a, 3c, 3b and 3h, and the windings 3a and 3h are wound. It is the composition which inserted 3b and winding 3c. FIG. 16 is an enlarged view of a part of the winding of FIG. 15, starting from the core 3i of the transformer 31, and the magnetic field strength of the leakage flux changes according to the current flowing in the winding and its direction according to the distance d from the starting point. Explain that.
In FIG. 16, when the third switching circuit 9 is turned on and the fourth switching circuit 10 is turned off, for example, a current of I flows from the back to the front of the drawing through the windings 3a and 3h. The current I flows from the front to the back in 3b and 3b respectively.
Assuming that the current I flows and the magnetic field H0 of the leakage flux is generated, the current becomes I at a distance d4 at which all the windings 3a are included, and a magnetic field of H0 is generated, and at a distance d5 including all the windings 3c Since the current I flows in the opposite direction to the winding 3a, the magnetic field is reduced by H0 to 0, and the current I flows in the opposite direction to the winding 3a at a distance d6 including all the windings 3b. It becomes -H0. In d7 including all the windings 3h, the current I flows in the same direction as that of the windings 3a, so that a magnetic field of H0 is generated to be zero. Thus, since the leakage magnetic field is canceled, the leakage inductance can be sufficiently reduced.
FIG. 17 shows the magnitude of leakage inductance when the third switching circuit 9 is turned off and the fourth switching circuit 10 is also turned off. For example, since a current I flows in the winding 3a from the back to the front of the drawing, a magnetic field of H0 is generated at the distance d4, but no current flows in the winding 3c, and there is no change in the magnetic field at the distance d5 H0 The magnetic field is reduced by 2H0 and becomes -H0 at a distance d6. At the distance d7, a current of I flows through the winding 3h from the back to the front of the drawing surface, so that a magnetic field of H0 is generated and becomes 0. Therefore, the magnitude of the leakage inductance is larger than that shown in FIG.

  As described above, in the configuration of the fourth embodiment, when the third switching circuit 9 is turned on and the fourth switching circuit 10 is turned off by the command from the command unit 210 when the first switching circuit 2 is on, The leakage magnetic field is canceled, the increase in winding resistance due to magnetic interference is suppressed, and the loss is reduced.

Fifth Embodiment
An example of on / off control of the first switching circuit 2, the third switching circuit 9, and the fourth switching circuit 10 of the power conversion device 120 described in the fourth embodiment is illustrated in FIG. 18. FIG. 18 is a waveform diagram showing switching control and transformer voltage.
The third switching circuit 9 is turned on at t7 after Δt2 when the first switching circuit 2 is turned on (t6). Thus, when the first switching circuit 2 is turned on (t6), the third switching circuit 9 and the fourth switching circuit 10 are both turned off, as described with reference to FIG. Since the leakage inductance 31 e becomes larger than when 9 is on, and the current of the first switching circuit 2 during on (t 6) is current-limited, switching loss can be reduced.
When the third switching circuit 9 is on (t7), the degree of coupling of the transformer 31 is increased, and the magnetic field is canceled as described in FIG. 16, so the leakage inductance is reduced, and the high frequency resistance of the winding due to magnetic interference The increase is suppressed, and the increase in the winding loss of the transformer 31 is suppressed.
Further, when the first switching circuit 2 is turned off (t8), the third switching circuit 9 is turned off and the fourth switching circuit 10 is turned on. As a result, the damping resistor 7 is connected to the winding 3c, the resonant voltage is damped, and the reduction of the winding loss can be realized.

As described above, in the configuration of the fifth embodiment, the first, third, and fourth switching circuits are synchronized as described above in synchronization with the on / off of the first switching circuit 2 by the command from the command unit 210. By turning on and off, the winding loss can be reduced.
In the present invention, within the scope of the invention, each embodiment can be freely combined, or each embodiment can be appropriately modified or omitted.

  1 DC voltage source, 2 first switching circuit, 2a parasitic capacitor component 3 transformer, 3a winding, 3b winding, 3c winding, 3e leakage inductance, 3f core, 3g winding resistance, 3h winding, 3i core, 4 rectifier circuit 5 load 6 second switching circuit 7 damping resistance 8 capacitive load 9 third switching circuit 10 fourth switching circuit 31 transformer 31 e leakage inductance 100 power converter 110 Power converter, 120 Power converter, 200 command means, 210 command means

Claims (4)

A first switching circuit for converting a DC voltage to an AC voltage, a transformer connected to the first switching circuit, and a power conversion device for outputting a voltage rectified by a rectifier circuit connected to the transformer, the transformer A third winding magnetically coupled to a first winding on the input side and a second winding on the output side; a second switching circuit for turning on / off a current flowing from the third winding to the load Path, command means for controlling the second switching circuit such that current flows from the third winding to the load when the first switching circuit is off,
The power conversion device according to claim 1, wherein the command means turns on the second switching circuit so that current flows from the third winding to the load before the first switching circuit is turned off . The power converter according to claim 1, wherein the load is a dumping resistor or a load having a capacitive property.   In a power conversion device for outputting a voltage rectified by a first switching circuit for converting a DC voltage to an AC voltage, a transformer connected to the first switching circuit, and a first rectifier circuit connected to the transformer, First and fourth windings connected in parallel with the first switching circuit on the input side of the transformer, and a second winding connected to the first rectifier circuit on the output side of the transformer; A third winding magnetically coupled to the first, second, and fourth windings; a second switching circuit connected to the third winding; parallel to the second switching circuit , A second rectifier circuit connected to the third switching circuit, a load connected to the second switching circuit, the first, second, and third Switching Command means for controlling on / off of the path, and when the first switching circuit is on, the third switching circuit is turned on and the second switching circuit is turned off to supply current to the third winding And the third switching circuit is turned off and the second switching circuit is turned on when the first switching circuit is turned off. After said first switching circuit is turned on, the power converter according to claim 3, characterized in that on the third switching circuit.

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