JP2014168355A - Forward converter of active clamp system and driving method thereof - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a forward converter of active clamp system in which efficiency is enhanced by lowering the magnetic flux density of a main transformer and reducing the switching frequency, thereby reducing the switching loss, and to provide a driving method thereof.SOLUTION: On the primary of a main transformer, the primary winding of the main transformer arranged between a pair of main switching elements, an active clamp circuit arranged on the secondary of the main transformer, and a feedback circuit for lowering the switching frequency of the pair of main switching elements when a load current is large, are provided, and a forward converter of active clamp system turns the pair of main switching elements and the switching element of the active clamp circuit alternately.

Description

  The present invention relates to an active clamp type forward converter that improves efficiency by reducing the switching frequency by lowering the magnetic flux density of a main transformer and a driving method thereof.

  In Patent Document 1 below, a switching power source that supplies DC power from a DC voltage source to a load via a transformer and a semiconductor switch, more specifically, a semiconductor switch having a low withstand voltage by reducing the voltage applied to the semiconductor switch. An invention relating to a switching power supply in which the circuit used is configured on the primary side of a transformer is described.

  According to Patent Document 1, the voltage applied to the semiconductor switch is reduced and the cooling body for cooling the semiconductor switch is miniaturized, and the primary winding side of the transformer is interposed via the semiconductor switch. In a switching power supply that outputs a DC voltage that is insulated by turning on and off the semiconductor switch to the secondary winding side of the transformer, the first semiconductor switch and the primary winding of the transformer are connected to the DC power supply. A series circuit with a line is connected in parallel, a rectifier circuit is connected to the secondary winding of the transformer, and a series circuit of the second semiconductor switch, the tertiary winding of the transformer and the first capacitor is connected to the transformer 1 A switching power supply connected in parallel with the next winding.

  As a result, the withstand voltage of the second semiconductor switch can be reduced, and the withstand voltage of the first and second semiconductor switches can be reduced, whereby the on-resistance of the semiconductor switch can be kept lower. Become. Therefore, the conduction loss can be reduced, the conversion efficiency can be improved, the heat radiation due to the conventional conduction loss can be reduced, and the semiconductor switch cooling body and the like can be miniaturized. Further, it is described that the conversion efficiency can be further improved by suppressing the switching loss.

JP 2000-262055 A

  Conventionally, the main switching element has been operated in the same phase with a duty ratio of 0.5 or less, and since the main transformer was a single excitation system, the frequency variable freedom was narrow. For this reason, only a relatively high frequency can be used, resulting in a large switching loss due to the high frequency.

  The present invention has been made in view of the above-described problems, and is an active clamp system that reduces switching loss and improves efficiency by reducing the magnetic flux density of the main transformer and reducing the switching frequency. It aims at realizing a forward converter and its driving method.

  An active clamp type forward converter of the present invention includes, on the primary side of a main transformer, a primary winding of a main transformer disposed between a pair of main switching elements, an active clamp circuit disposed on a secondary side of the main transformer, A feedback circuit that reduces the switching frequency of the pair of main switching elements when the load current is large, and the pair of main switching elements and the switching elements of the active clamp circuit are alternately turned on. .

  The active clamp type forward converter driving method of the present invention is arranged on the primary side of the main transformer on the primary winding of the main transformer arranged between the pair of main switching elements and on the secondary side of the main transformer. An active clamp circuit, turning on the pair of main switching elements and turning off the switching elements of the active clamp circuit, and turning off the pair of main switching elements and switching elements of the active clamp circuit A feedback circuit that changes the switching frequency corresponding to the load current, and reduces the switching frequency of the pair of main switching elements when the load current is large.

  By reducing the magnetic flux density of the main transformer and reducing the switching frequency, it is possible to realize an active clamp type forward converter and its driving method in which the switching loss is reduced and the efficiency is improved.

It is a conceptual diagram explaining the circuit structure outline | summary of the active clamp type DC-DC converter concerning 1st embodiment. It is a figure explaining output capacity (Coss), (a) explains FET, and (b) is a figure explaining the equivalent circuit of the FET. For explanation of the circuit operation of the DC-DC converter in the active clamp system according to the first embodiment, a diagram illustrating a time chart of current and voltage of each part, (a) the main switching element (Q ₂ -A) and the main switching element _(Q 2 -B) and the gate voltage for operating the indicates _{(G 2),} shows the (b) a third switching element _{(Q 1)} a gate voltage for operating the _{(G 1)} the drain of (c) the main switching element _(Q 2 -A) and the main switching element _(Q 2 -B) - shows the source voltage, a main switching (d) the main switching element _(Q 2 -A) The drain-source current (I _Q2 ) with the element (Q ₂ -B) is shown, (e) shows the drain-source current (I _Q1 ) of the third switching element (Q ₁ ), and (f) Is the main transformer ( Shows the excitation voltage of ₁₎ the primary coil _{(L 1) 400 (1)} (V T1), shows the excitation voltage _{(V T3)} of (g) other primary coil _{(L 3),} (h) Indicates the drain-source voltage (V _D-S1 ) of the third switching element (Q ₁ ). 5 is a flowchart for sequentially explaining the outline of the operation of the active clamp type DC-DC converter according to the first embodiment. It is a figure explaining the example of a conventional circuit of one stone in the case of input low voltage. It is a chart which shows operation | movement description of the example of a conventional circuit in the case of an input low voltage with the current voltage of each site | part. It is a figure explaining the example of a conventional circuit of 2 stones in the case of input high voltage. It is a conceptual diagram explaining the structure outline | summary of the DC-DC converter of the active clamp system concerning 2nd embodiment. It is a figure explaining each line capacity | capacitance of the gate of a FET, a drain, and a source. It is a figure explaining the outline | summary of the forward converter of an active clamp system of this embodiment. It is a figure explaining the relationship between the magnetic field strength (H) and magnetic flux density (B) of the main transformer (T) of an active clamp type forward converter. It is a figure explaining the magnetic field strength (H) and magnetic flux density (B) of the main transformer (T) of the conventional forward converter. A chart for explaining the outline of the operation status of the forward converter of the active clamp scheme of FIG. 10, a (a) the main switching element (Q ₁₎ and the gate signal of the main switching element (Q ₂₎ (node G ₁₎ is an explanatory timing chart, the drain of the (b) is a timing chart for explaining the gate signal (node G ₂₎ of the switching elements of the active clamp circuit (Q _3), (c) the main switching element (Q ₂₎ (D) is a timing chart for explaining a voltage (V _D- S2) between the source (S2), and (d) is a current (IQ ₂ ) between the drain (D) and the source (S2) of the main switching element (Q ₂ ). (E) shows the current (I _Q3 ) of the switching element (Q ₃ ) of the active clamp circuit. It is a figure explaining. It is a circuit schematic diagram for demonstrating the forward converter operating method of an active clamp system. In order to explain the operation method of the active clamp type forward converter shown in FIG. 14, in the circuit, the case where the frequency of the forward converter is changed and the case where the frequency of the forward converter is not changed (a constant case) will be explained. FIG.

  The present embodiment proposes a DC-DC converter that reduces switching loss by changing the switching frequency according to the magnitude of the load current in an active clamp type forward converter. That is, the present embodiment is a proposal related to an active clamp frequency variable forward converter.

  Conventionally, the forward converter is a single-excitation operation in its main transformer and has to consider the saturation of the magnetic flux density, so that it is impossible to substantially cope with the variable switching frequency. In this embodiment, an active clamp system is employed in the forward converter to make the main transformer excitation positive and negative, thereby reducing the maximum value of the main transformer magnetic flux density, thereby avoiding saturation of the magnetic flux density.

  As a result, the main transformer can cope with variable switching frequency of a pair of main switching elements arranged on the primary side, and at heavy load, that is, when the load current is large, the switching frequency is reduced to reduce the switching loss. It becomes possible to make it. The operator may appropriately set the load current at which the switching frequency is reduced in relation to the overall efficiency.

  Here, since the switching loss occurs for each switching operation, it generally varies depending on the switching frequency, and it is known that the switching loss increases as the switching frequency increases. Therefore, it is possible to increase the efficiency of the DC-DC converter by reducing the switching loss by reducing the switching frequency under heavy load.

  A specific active clamp type forward converter driving method varies the switching frequency of a pair of main switching elements arranged on the primary side of the main transformer in accordance with the magnitude of the load current (output current). That is, when the load current is large, the switching frequency is reduced, and when the load current is not large, the switching frequency is returned to the initial value. Such switching frequency reduction operation may be performed when the load current is compared with a threshold value and the threshold value is exceeded.

  Conventionally, the forward converter operates the main switching elements (Q1, Q2) in the same phase with a duty ratio of 0.5 or less, and the magnetic flux of the main transformer (T) is a single excitation operation. There were severe restrictions from the viewpoint of saturation, and practically it was not possible.

In the active clamp system of the present embodiment, the pair of main switching elements (Q1, Q2) and the switching element (Q ₃ ) of the active clamp circuit provided on the secondary side of the main transformer (T) are alternately switched. . As a result, the excitation of the main transformer can be positive and negative, and the maximum value of the magnetic flux density of the transformer can be reduced so as not to exceed the saturation magnetic flux density. As the magnetic flux is determined by the product of voltage and time, even if the magnetic flux increases by lowering the frequency, it does not exceed the saturation magnetic flux density.

  Therefore, the switching frequency can be lowered without being restricted by saturation of the magnetic flux density. Further, by reducing the switching frequency, it is possible to reduce the switching loss that occurs at each switching, and therefore the efficiency of the DC-DC converter can be improved. Therefore, this embodiment will be described below with reference to the drawings.

FIG. 10 is a diagram for explaining the outline of the configuration of the active clamp type forward converter 100 of the present embodiment. As shown in FIG. 10, the primary side of the active clamp type forward converter 100 includes a pair of main switching elements (Q ₁ ) 20 (1) and main switching elements (Q ₂ ) 20 (2) connected to the DC power supply 10. ), A main transformer (T) 40, and an active clamp circuit 30 provided separately from the secondary winding for output on the secondary side of the main transformer (T) 40.

  As shown in FIG. 10, the secondary side of the active clamp type forward converter 100 has a rectifier diode (D) 70 (1) connected in series with the output secondary winding of the main transformer (T) 40. ), A rectifier diode (D) 70 (2) connected in parallel with the output secondary winding of the main transformer (T) 40, an inductor (L) 80, and a smoothing capacitor (C) 85. . Further, the load 90 may be a DC motor or an arbitrary load that operates with a DC power source.

  As is clear from FIG. 10, the active clamp type forward converter 100 according to the embodiment is active on the secondary side of the main transformer (T) 40 and the output secondary side winding connected to the load 90. The windings of the clamp circuit 30 are individually provided.

In FIG. 10, the active clamp circuit 30 includes a capacitor (C) 31 and a switching element (Q ₃ ) 32 connected in series with the winding. The output supplied to the load 90 is detected by the output voltage detection unit 1 and input to the comparator 3. Further, the primary current of the main transformer (T) 40 is detected by the current detection unit 2, and the oscillation frequency is adjusted by the oscillation frequency adjustment unit 4 corresponding to the detected primary side current.

The output of the oscillation frequency adjusting unit 4 is input to the comparator 3, and the output of the comparator 3 is the gate signal of the main switching element (Q ₁ ) 20 (1) and the main switching element (Q ₂ ) 20 (2). It becomes. Further, the inverted output of the comparator 3 becomes a gate signal of the switching element (Q ₃ ) 32 of the active clamp circuit 30.

With the circuit configuration described above, the active clamp type forward converter 100 has a main switching element (Q ₁ ) 20 (1) and a main switching element (Q ₂ ) 20 (2) when the load current to the load 90 is large. The operation of reducing the on / off switching drive frequency can be performed. Thereby, switching loss can be reduced. In this case, since the magnetic flux density of the main transformer (T) 40 is positive and negative excitation, there is no restriction on saturation of the magnetic flux density.

  FIG. 11 is a diagram for explaining the relationship between the magnetic field strength (H) of the main transformer (T) 40 of the active clamp type forward converter 100 and the magnetic flux density (B). FIG. 12 is a diagram for explaining the magnetic field strength (H) and magnetic flux density (B) of the main transformer (T) of the conventional forward converter.

As shown in FIG. 11 and FIG. 12, conventionally, only the positive side is so-called single excitation, and the magnetic flux density has reached the maximum value (B ₁ ). Further, the fluctuation range (ΔB ₁ ) of the magnetic flux density in the conventional single excitation shown in FIG. 11 is equal to the fluctuation range (ΔB ₂ ) of the magnetic flux density in the positive and negative excitation of the present embodiment shown in FIG. Therefore, the maximum value (B ₂ ) of the magnetic flux density is about ½ of (B ₁ ).

Therefore, it is not necessary to consider the restriction of the saturation magnetic flux density (B _m ), and the switching frequency of the main switching element can be reduced.

FIG. 13 is a timing chart for explaining the outline of the operation state of the active clamp type forward converter 100 of FIG. FIG. 13A is a timing chart for explaining gate signals (node G ₁ ) of the main switching element (Q ₁ ) 20 (1) and the main switching element (Q ₂ ) 20 (2), and FIG. FIG. 6 is a timing chart for explaining a gate signal (node G ₂ ) of the switching element (Q ₃ ) 32 of the active clamp circuit 30.

FIG. 13C is a timing chart for explaining the drain (D) -source (S ₂ ) voltage (V _D−S2 ) of the main switching element (Q ₂ ) 20 (2). ) Is a timing chart for explaining the current (IQ ₂ ) between the drain (D) and the source (S ₂ ) of the main switching element (Q ₂ ) 20 (2).

FIG. _13E is a diagram for explaining the current (I _Q3 ) of the switching element (Q ₃ ) 32 of the active clamp circuit 30. FIG. 13 illustrates a state in which the load current exceeds the threshold at time (t) and the operation for reducing the main switching drive frequency is performed.

Therefore, as shown in FIGS. 13A and 13B, the gate signals of the main switching element (Q ₁ ) 20 (1) and the main switching element (Q ₂ ) 20 (2) before time (t). While the switching frequency of the gate signal (node G ₂ ) of the (node G ₁ ) and the switching element (Q ₃ ) 32 of the active clamp circuit 30 is (f), the switching frequency is after the time (t). Is reduced to (1 / 2f).

  Thereby, switching loss is reduced and the efficiency of the DC-DC converter can be increased. FIG. 14 is a schematic circuit diagram for explaining an active clamp type forward converter operation method. Further, FIG. 15 is a circuit for explaining the active clamp type forward converter operation method shown in FIG. 14, in the case where the frequency of the forward converter is changed, and in the case where the frequency of the forward converter is not changed (a constant case). It is a figure demonstrated in comparison.

Referring to FIGS. 14 and 15, when an _L current (I _L ) flows with a certain inclination with respect to the inductor of the main transformer in accordance with on / off of the main switching element, if the frequency does not change, the _L current (I _L The slope of _L ) is always constant. Since the output current (I _o ) is the center value of the _L current (I _L ), the center value gradually approaches zero when the output current (I _o ) decreases as shown in FIG.

In this case, since the span (interval) is the same unless the frequency is changed, the lower limit of the _L current (I _L ) is stuck to zero (the choke coil L is cut off). Here, since the output voltage (V _o ) rises with a slope as a ripple corresponding to the el current (I _L ), the ripple is an el current (I) with respect to the center value of the output voltage (V _o ). _L ) and is proportional (correlated).

For this reason, when a situation occurs in which the lower limit of the _L current (I _L ) is stuck to zero (the choke coil L is in the cut-off state), the output voltage (V _o ) becomes a so-called uncontrolled state. It becomes unstable.

On the other hand, when the frequency is changed, although the slope of the EL current (I _L ) is the same, the span (interval) is shortened, so that the lower limit of the EL current (I _L ) is stuck to zero ( The situation where the choke coil L is cut off is less likely to occur. In addition, if a situation occurs in which the lower limit of the _L current (I _L ) is stuck to zero (the choke coil L is in a cut-off state), the dummy load resistance is considered for that portion, A cut-off state can be avoided.

The above active clamp type forward converter and its driving method can also be applied to an active clamp type DC-DC converter described later. An active clamp type DC-DC converter to which the above-described active clamp type forward converter and its driving method can be applied will be described below. In the following description, for the sake of convenience, the active clamp circuit is described on the primary side instead of the secondary side, but the circuit may be provided on either the primary side or the secondary side. Good (only the description direction on the drawing is different).
(Active clamp type DC-DC converter)

  In the active clamp type DC-DC converter described in the embodiment, an additional active clamp circuit is provided by using another primary winding added to the main transformer. As a result, the withstand voltage of the active element (switching element) can be lowered, the output capacitance (Coss) of the main switching element can be discharged and then turned on (conducted), and zero voltage switching can be performed to reduce loss.

  It is known that an active element having a FET as a typical example increases in resistance exponentially as the withstand voltage increases, so that the loss increases exponentially as the withstand voltage increases. Therefore, the loss can be greatly reduced if the voltage can be lowered so that the withstand voltage can be lowered and an active element having a low withstand voltage can be used.

If zero voltage switching is not used, a loss of (1/2) × Coss × (V _d−s ) ² × f occurs. In the active clamp type DC-DC converter according to the present embodiment, this loss is generated. It can be reduced. Further, the other added primary winding may be provided with a tap provided on the primary side of the main transformer. Here, f is the frequency, Coss is the output capacitance of the switching element, and V _d−s is the drain-source voltage of the switching element.

(First embodiment)
FIG. 1 is a conceptual diagram illustrating an outline of a circuit configuration of an active clamp type DC-DC converter 1000 according to the first embodiment. As shown in FIG. 1, the primary side of the active clamp type DC-DC converter 1000 includes a main switching element (Q ₂ -A) 200 (1) connected to a DC power supply (E ₁ ) 100 and a main switching element ( Q ₂ -B) 200 (2), the primary side coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400, the primary side choke coil 500 (1) of the coupling coil (T ₂ ) 500, Are connected in series to form an active clamp circuit.

Further, as shown in FIG. 1, the secondary side of the active clamp type DC-DC converter 1000 is connected in series with the secondary side coil (L ₂ ) 400 (3) of the main transformer (T ₁ ) 400. A rectifier diode (D ₁ ) 700 (1), a rectifier diode (D ₂ ) 700 (2) connected in parallel with the secondary coil (L ₂ ) 400 (3) of the main transformer (T ₁ ) 400, An inductance (L ₄ ) 800 and a smoothing capacitor (C ₂ ) 850 are provided. Further, the load (R ₀ ) 900 may be a DC motor or an arbitrary load that operates with a DC power supply.

As is clear from FIG. 1, the active clamp type DC-DC converter 1000 according to the first embodiment is separate from the primary coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400. Another primary coil (L ₃ ) 400 (2) provided on the primary side is provided.

The other primary side coil (L ₃ ) 400 (2) includes a capacitor (C ₁ ) 600, a secondary side choke coil 500 (2) of the coupling coil (T ₂ ) 500, and a third switching element (Q ₁ ) 300 are sequentially connected.

In FIG. 1, in the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2), an apparent so-called Coss is formed between the drain and the source. Output capacity exists. 2A and 2B are diagrams for explaining the output capacitance (Coss). FIG. 2A is a diagram for explaining an FET, and FIG. 2B is a diagram for explaining an equivalent circuit of the FET.

  As shown in FIG. 2, the switching element such as an FET has an apparent capacitance between any two of the three terminals of the gate (G), drain (D), and source (S). It is known. That is, the feedback capacitance (Crss) apparently exists between the gate and the drain, the output capacitance (Ciss) apparently exists between the gate and the source, and the output capacitance between the drain and the source. (Coss) apparently exists.

In the active clamp type DC-DC converter 1000 according to the first embodiment, the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) among the three capacitors described above. The charge stored in the output capacitor (Coss) of 200 (2) is reduced and typically the stored charge is reduced to zero, and then the main switching element (Q ₂ -A) 200 (1) and the main switching element ( by Q 2 _-B) 200 (2) is the on conduction, preferable are reduced power loss can be made to zero-voltage switching.

  FIG. 3 is a diagram illustrating a current / voltage time chart of each part for explaining the circuit operation of the active clamp type DC-DC converter 1000 according to the first embodiment.

FIG. 3A shows a gate voltage (G ₂ ) for operating the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2), and FIG. the third switching element _(Q 1) shows the 300 gate voltage for operating the _{(G 1),} (c) the main switching element _{(Q 2 -A) 200 (1} ) and the main switching element _(Q 2 -B) 200 (2) shows the drain-source voltage, and (d) shows the drain-source voltage between the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2). Current (I _Q2 ), (e) the drain-source current (I _Q1 ) of the third switching element (Q ₁ ) 300, and (f) the primary coil of the main transformer (T ₁ ) 400 (L ₁ ) 400 (1) excitation voltage (V _T1 ), (g) shows the excitation voltage (V _T3 ) of the other primary coil (L ₃ ) 400 (2), and (h) shows the drain of the third switching element (Q ₁ ) 300. -Indicates a source-to-source voltage (V _D-S1 ).

In FIG. 3A, the on-off cycle of the gate voltage (G ₂ ) for operating the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) is expressed as “ “T” and the on period of these gate voltages (G ₂ ) are shown as “D”.

Further, in the period “t” shown in FIG. 3C, the output capacitance (Coss) of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) is increased. Since the accumulated charge is released, after the charge is discharged, as shown in FIG. 3A, the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2 Even if the gate voltage (G ₂ ) that operates the above is turned on and they are turned on, zero voltage switching can be performed, so that power loss can be reduced.

Further, as shown in FIG. 3C, the drain-source voltage (Q ₂ ) between the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 ( ₂ ). The peak value of is (1/2) (E + V _F ). Here, “V _F ” is the minimum value (negative voltage value) of the excitation voltage (V _T1 ) of the primary side coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400, and “E” Is the maximum value (positive voltage value) of the excitation voltage (V _T1 ) of the primary coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400 (see FIG. 3F). ).

Further, the broken line shown in FIG. 3 (d) is a broken line that compares and explains a state in which the current (I _Q2 ) gradually decreases and gradually decreases in the conventional circuit example described later (FIG. 6). (See the _IT1 time chart in (e)). In the active clamp type DC-DC converter 1000 according to the first embodiment, the gate voltage (G ₁ ) for operating the third switching element (Q ₁ ) 300 is turned on (conducted), whereby coupling is performed. Since energy is transmitted through the coil (T ₂ ) 500 according to the ampere-turn law, the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) At the same time when the gate voltage (G ₂ ) to be operated is turned off, the current (I _Q2 ) becomes zero quickly and instantaneously as shown by the solid line in FIG.

Further, in FIG. 3F, the gate voltage (G ₂ ) for operating the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) is constant from the on state. The excitation voltage (V _T1 ) of the primary coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400 does not rise and changes to zero for the time, but this is the coupling coil (T ₂ ). This is because the primary side choke coil 500 (1) 500 takes in the rising voltage.

Further, as shown in FIG. 3G, the maximum value (positive value) of the excitation voltage (V _T3 ) of the other primary coil (L ₃ ) 400 (2) is (N ₃ / N _1). ) E, and its minimum value (negative value) is (N ₃ / N ₁ ) V _F. Here, “N ₁ ” is the number of turns of the primary side coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400, and “N ₃ ” is the other primary side coil (L ₃ ) 400 (2). Is the number of turns. In FIG. 1, the number of turns of the secondary coil (L ₂ ) 400 (3) of the main transformer (T ₁ ) 400 is indicated as “N ₂ ”.

Further, as shown in FIG. 3H, the maximum value (positive side) of the drain-source voltage (V _D-S1 ) of the third switching element (Q ₁ ) 300 is (N ₃ / N ₁ ) ( E + V _F ), and its minimum value is zero. Further, the number of turns of _{"N A"} coupling coil _(T 2) 500 of the primary side choke coil 500 (1), _{"N B"} coupling coil _(T 2) 500 of the secondary side choke coil 500 (2 ) Is assumed to be N _A : N _B = N ₁ : N ₃ .

Further, when the output capacitance (Coss) the accumulated charge like is large is large, a relatively small structure to N _1, thereby to increase the excitation current, be released more accumulated charge It becomes possible.

Further, the active clamp type DC-DC converter 1000 according to the first embodiment shown in FIG. 1 can be designed to have a low breakdown voltage of the third switching element (Q ₁ ) 300. For example, when the withstand voltage of the main switching element (Q ₂ -A) 200 (1) is 400 V and the withstand voltage of the main switching element (Q ₂ -B) 200 (2) is 400 V, conventionally, When a switching element having a withstand voltage design of 800 V is necessary, in the active clamp type DC-DC converter 1000 according to the first embodiment, the voltage can be adjusted by the turn ratio as described above. By appropriately adjusting the ratio, the withstand voltage of the third switching element (Q ₁ ) 300 can be designed to be low.

As a result, the circuit configuration is easy, small and light, and a low-loss circuit can be realized. In general, when the breakdown voltage of the switching element increases, the on-resistance increases and the on-loss increases exponentially. In the active clamp type DC-DC converter 1000 according to the first embodiment, a switching element having a relatively small on-resistance can be used as the third switching element (Q ₁ ) 300. Loss can be reduced.

  FIG. 4 is a flowchart for sequentially explaining the operation outline of the active clamp type DC-DC converter 1000 according to the first embodiment. Therefore, the operation outline of the active clamp type DC-DC converter 1000 will be sequentially described below for each step shown in FIG.

In the following description, the main switching element (Q ₂ -A) 200 (1), the main switching element (Q ₂ -B) 200 (2), and the third switching element (Q ₁ ) 300 are each represented by a gate voltage. (G ₂ ) and the gate voltage (G ₁ ) are switched on / off, and a control unit (not shown) that generates and outputs the gate voltage (G ₂ ) and the gate voltage (G ₁ ) is provided. You may have.

(Step S410)
The main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) of the active clamp type DC-DC converter 1000 are turned off (non-conducting). Further, the third switching element (Q ₁ ) 300 is turned on (conductive). As shown in FIG. 3, the off (non-conducting) operation of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2), and the third switching element (Q ₁ ) The on (conduction) operation of 300 is performed simultaneously.

(Step S420)
The third switching element (Q ₁ ) 300 is turned off. At this time, the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) remain in the OFF state due to the operation of step S 410. Accordingly, the primary switching element (Q ₂ -A) 200 (1) and the primary switching element (Q ₂ -B) 200 (2) and the third switching element (Q ₁ ) 300 are configured as primary active clamp circuits. All the three switching elements to be turned off are turned off.

Thus, trying to keep the magnetic field coupling coil _(T 2) 500 of the secondary side choke coil 500 (2), the current flows through the coupling coil _(T 2) 500 of the primary side choke coil 500 (1) Typically, a charge is generated in the output capacitance (Coss) of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) by causing a resonance phenomenon. Is discharged. That is, energy is transmitted from the secondary side choke coil 500 (2) to the primary side choke coil 500 (1) by the process of step S420.

(Step S430)
A period “t” sufficient for discharging the charge accumulated in the output capacitance (Coss) of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2). Is determined.

As the period “t”, a value set in advance in the control unit or the like of the DC-DC converter 1000 can be used. However, the period “t” may be configured to be changed and adjusted according to the driving situation. In the case of setting in advance, the electric charge accumulated in the output capacitance (Coss) of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) is discharged. It is assumed that the time period “t” is sufficient. Further, when the charge accumulated in the output capacitance (Coss) is large, the number of turns N ₁ of the primary coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400 is reduced, and the accumulated charge is further increased. In order to discharge a large amount, the number of turns N ₁ of the primary coil (L ₁ ) 400 (1) of the main transformer (T ₁ ) 400 may be configured as a variable number of turns, for example.

A period “t” sufficient for discharging the charge accumulated in the output capacitance (Coss) of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2). ”Has passed, the process proceeds to step S440 and is stored in the output capacitance (Coss) of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2). If the period “t” sufficient for discharging the electric charge has not elapsed, the process waits in step S430.

(Step S440)
The main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) are turned on to conduct. In this case, the output capacity (Coss) of the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) by the discharge process in the period “t” in step S 430 is The accumulated charge is typically zero.

Therefore, when the main switching element (Q ₂ -A) 200 (1) and the main switching element (Q ₂ -B) 200 (2) are turned on and conducted, so-called zero voltage switching can be performed. As described above, the loss can be reduced by the zero voltage switching.

(Step S450)
When the driving operation of the active clamp type DC-DC converter 1000 is ended, the steps described in this flow are ended. If the driving operation of the active clamp type DC-DC converter 1000 is not terminated, the process returns to step S410.

(Comparative example)
Here, in order to facilitate understanding of the active clamp type DC-DC converter 1000 according to the first embodiment, a conventional active clamp type DC-DC converter is illustrated in FIGS. explain.

  FIG. 5 is a diagram for explaining an example of a conventional circuit for one stone in the case of an input low voltage, and FIG. 6 is a chart showing the operation of the conventional circuit example of one stone in the case of an input low voltage by the current voltage of each part. FIG. 7 is a diagram for explaining a conventional circuit example of two stones in the case of an input high voltage.

  As can be understood from FIGS. 5 to 7, none of the conventional circuit examples can lower the withstand voltage of the switching element. In particular, in the circuit example of the input high voltage shown in FIG. Unless the withstand voltage is set to an extremely high value, it cannot be practically used.

Further, as shown in FIGS. 5 and 6, conventional, and insert the capacitor C ₁ and the switching element Q ₁ on the primary side of the transformer T _1, it turns on the switching element Q ₁ at the moment when the switching element Q ₂ is turned off By storing the energy of the transformer T _{1 in} the capacitor C ₁ and releasing it after a certain time, the time area of the positive voltage and the negative voltage on the primary side of the transformer T ₁ is the same as shown in FIG. Drive to become.

Further, as apparent from FIG. 5, conventionally since it is connected in a floating state switching element Q _1, it is necessary to use a MOS-FET or the like of the high voltage, inevitable that on loss increases It was. In FIG. 6, the relationship E · D = V _F (1−D) is established. Further, a condition that enables zero voltage switching is Lm · Im ² = Coss · Vds ² . Here, Im is the magnetizing current in the primary side of the main transformer T _1, Lm is the inductance of the main transformer T _1, Vds is the drain of the main switching element Q ₂ - source voltage.

Also shows FIGS. 6 (a) is the gate voltage _{G 2} for operating the main switching element _{Q 2,} (b) indicates a gate voltage _{G 1} for operating the switching element _{Q 1,} (c) the main switching element _{Q 2} drain - shows the source voltage, (d) a drain of the main switching element _{Q 2} - shows the source current _{I Q2,} (e) indicates the current _{I T1} of the primary side coil of the main transformer _{T 1,} (f) There drain of the switching element _{Q 1} - shows the source current _{I Q1,} (g) indicates the voltage _{V T1} of the primary side coil of the main transformer _{T 1,} (h) is the drain of the switching element _{Q 1} - source voltage (V _D-S1 ).

(Second embodiment)
FIG. 8 is a conceptual diagram illustrating an outline of a configuration of an active clamp type DC-DC converter 8000 according to the second embodiment. Second active clamp DC-DC Converter 8000 according to the embodiment of the, by the other winding is connected to the third switching element Q _1, is provided a tap from a desired position in the primary side of the main transformer, A part of the primary winding is used. Therefore, by adjusting the position of the tap, since the turns ratio is suitably adjustable, relatively low adjustable a third withstand voltage of the switching element Q _1.

As shown in FIG. 8, the primary side of the active clamp type DC-DC converter 8000 includes a main switching element (Q ₂ -A) 8200 (1) connected to a DC power supply (E ₁ ) 8100 and a main switching element ( Q and ₂ -B) 8200 (2), a main transformer _(T 1) 8400 of the primary coil _(L 1) 8400 (1), the primary side choke coil 8500 of the coupling coil _(T 2) 8500 (1) Are connected in series to form an active clamp circuit.

Further, as shown in FIG. 8, the secondary side of the active clamp type DC-DC converter 8000 is connected in series with the secondary coil (L ₂ ) 8400 (3) of the main transformer (T ₁ ) 8400. A rectifier diode (D ₁ ) 8700 (1), a rectifier diode (D ₂ ) 8700 (2) connected in parallel with the secondary coil (L ₂ ) 8400 (3) of the main transformer (T ₁ ) 8400, An inductance (L ₄ ) 8800 and a smoothing capacitor (C ₂ ) 8850 are provided. Further, the load (R ₀ ) 8900 may be a DC motor or an arbitrary load that operates with a DC power supply.

As is apparent from FIG. 8, the active clamp type DC-DC converter 8000 according to the second embodiment is a part of the primary coil (L ₁ ) 8400 (1) of the primary transformer (T ₁ ) 8400. Is provided with another primary side coil (L ₃ ) 8400 (2) provided with a tap.

The other primary side coil (L ₃ ) 8400 (2) includes a capacitor (C ₁ ) 8600, a secondary side choke coil 8500 (2) of the coupling coil (T ₂ ) 8500, and a third switching element (Q ₁ ) 8300 are sequentially connected.

As shown in FIG. 8, in the main switching element (Q ₂ -A) 8200 (1) and the main switching element (Q ₂ -B) 8200 (2), the so-called Coss is called between the drain and the source. This produces an apparent output capacity. FIG. 9 is a diagram illustrating the line capacitances of the gate, drain, and source of the FET.

  As shown in FIG. 9, a switching element such as an FET has a capacitance, Cgd, Cds, Cgs between any two of the three terminals of the gate (G), drain (D), and source (S). Is known to exist. Here, the apparent input capacitance Ciss = Cgd + Cgs, the apparent output capacitance Coss = Cds + Cgd, and the apparent feedback procedure Crss = Cgd.

In the active clamp type DC-DC converter 8000 according to the second embodiment, the main switching element (Q ₂ -A) 8200 (1) and the main switching element (Q ₂ -B) among the three capacitors described above. The charge accumulated in the output capacitance (Coss) of the 8200 (2) is reduced and typically the accumulated charge is reduced to zero, and then the main switching element (Q ₂ -A) 8200 (1) and the main switching element ( by Q ₂ -B) 8200 ₍₂₎ is the on conduction, preferable because it is possible to zero-voltage switching.

Here, “N ₁ ” is the number of turns of the primary coil (L ₁ ) 8400 (1) of the main transformer (T ₁ ) 8400, and “N ₃ ” is another primary coil (L ₃ ) formed by a tap. The number of turns is 8400 (2). In FIG. 8, the number of turns of the secondary coil (L ₂ ) 8400 (3) of the main transformer (T ₁ ) 8400 is shown as “N ₂ ”.

Further, _{"N A"} and the number of turns of the coupling coil _(T 2) 8500 of the primary choke coil 8500 (1), _{"N B"} coupling coil _(T 2) 8500 of the secondary side choke coil 8500 (2 ) Is assumed to be N _A : N _B = N ₁ : N ₃ .

Further, when the accumulated charge is large, such as when the output capacitance (Coss) of the main switching element (Q ₂ -A) 8200 (1) and the main switching element (Q ₂ -B) 8200 (2) is large, N ₁ It is possible to release a larger amount of accumulated charge by increasing the excitation current with a relatively small configuration. Further, the primary side of the main transformer (T ₁ ) 8400 is such that the number of turns N ₁ of the primary side coil (L ₁ ) 8400 (1) of the main transformer (T ₁ ) 8400 is reduced to discharge more accumulated charge. the number of turns _{N 1} of the coil _(L 1) 8400 ₍₁₎ may be variable number of turns.

  The active clamp type DC-DC converter of the present invention includes a first primary winding provided between two main switching elements and a first primary winding connected in series with the first primary winding on the primary side of the main transformer. A first choke coil, a primary side of the main transformer, a second primary winding connected in parallel with the third switching element, and a second choke coil connected in parallel with the second primary winding, The one choke coil and the second choke coil form a coupling transformer.

  The active clamp type DC-DC converter of the present invention is preferably characterized in that the coupling transformer transmits the energy of the output capacitance (Coss) of the main switching element.

  In the active clamp type DC-DC converter according to the present invention, the winding ratio of the first primary winding to the second primary winding is more preferably the first choke coil constituting the coupling transformer and the second primary winding. The winding ratio is the same as that of the two choke coils.

  The active clamp type DC-DC converter of the present invention includes a primary winding provided between two main switching elements on a primary side of a main transformer, and a first choke coil connected in series with the primary winding. A primary choke comprising a second choke coil connected in series with a third switching element, and a capacitor connected to the second choke coil and connected at multiple ends by forming a tap on the primary winding. The first choke coil and the second choke coil form a coupling transformer.

  The active clamp type DC-DC converter of the present invention is preferably characterized in that the coupling transformer transmits the energy of the output capacitance (Coss) of the main switching element.

  Further, the active clamp type DC-DC converter of the present invention more preferably has a winding ratio of a primary winding and a portion where the tap of the primary winding is formed and connected in parallel with the third switching element, The winding ratio of the first choke coil and the second choke coil constituting the coupling transformer is the same.

  Moreover, the switching power supply device of this invention is equipped with the DC-DC converter in any one of the above-mentioned.

  A motor driver according to the present invention includes any one of the above-described DC-DC converters.

  The active clamp type DC-DC converter driving method according to the present invention includes a first primary winding provided between two main switching elements on the primary side of a main transformer, and a first primary winding in series. A first choke coil connected to the main transformer, a second primary winding connected in parallel to the third switching element on the primary side of the main transformer, and a second choke coil connected in parallel to the second primary winding The first choke coil and the second choke coil form a coupling transformer, turning off the two main switching elements and simultaneously turning on the third switching element, and two main switching elements When turning on, after the third switching element is turned off first, the charge of the output capacitance (Coss) of the two main switching elements is discharged, and then the two main switching elements are discharged. A step of turning on the switching element, characterized by having a.

  Also, the driving method of the active clamp type DC-DC converter of the present invention is preferably characterized in that the coupling transformer transmits the energy of the output capacitance (Coss) of the main switching element.

  Further, in the driving method of the active clamp type DC-DC converter according to the present invention, the winding ratio of the first primary winding and the second primary winding is more preferably the first choke constituting the coupling transformer. The winding ratio of the coil and the second choke coil is the same.

  The active clamp type DC-DC converter driving method according to the present invention includes a primary winding provided between two main switching elements on a primary side of a main transformer, and a first winding connected in series with the primary winding. A choke coil, a second choke coil connected in series with a third switching element on the primary side of the main transformer, and a capacitor connected to the second choke coil and connected at multiple ends by forming taps on the primary winding The first choke coil and the second choke coil form a coupling transformer, turning off the two main switching elements and simultaneously turning on the third switching element, and two main switching elements When turning on, the third switching element is turned off first and the output capacitance (Coss) of the two main switching elements is After releasing, and having a step of turning on the two main switching elements.

  Also, the driving method of the active clamp type DC-DC converter of the present invention is preferably characterized in that the coupling transformer transmits the energy of the output capacitance (Coss) of the main switching element.

  In the active clamp type DC-DC converter driving method of the present invention, it is more preferable that the primary winding and the tap of the primary winding are formed and the winding with the portion connected in parallel with the third switching element. The ratio is the same as the winding ratio of the first choke coil and the second choke coil constituting the coupling transformer.

  The DC-DC converter according to the present invention is characterized in that, in any of the above-described active clamp type DC-DC converters, two main switching elements perform zero voltage switching.

  According to another aspect of the present invention, there is provided a driving method for a DC-DC converter according to any one of the above-described active-clamp DC-DC converters, wherein two main switching elements perform zero voltage switching. To do.

  With the above-described configuration, an active clamp type DC that discharges the charge of the output capacitance (Coss) of the main switching element and lowers the withstand voltage of the main switching element, reduces the on-resistance and reduces the on-loss. -A DC converter or the like can be realized.

  The active clamp type DC-DC converters 1000, 8000 and the like exemplified in the above embodiments are not limited to the description in each embodiment, and are within the scope of the technical idea described in each embodiment and are obvious. Within the scope, the configuration, operation, operation method, and the like can be changed as appropriate. In addition, for convenience of explanation, each embodiment has been described individually. However, the configurations of the embodiments may be applied in an appropriate combination, and operations thereof may be arranged in an appropriate combination.

  The active clamp type DC-DC converter of the present invention can be widely applied as a circuit configuration of various switching power supply devices, motor drive devices, inverters, and the like.

100 ... DC power supply _(E 1), 200 ... main switching element, 300 ... third switching element, 400 ... main transformer, 500 ... coupling coil _(T 2), 600 ... capacitor _{(C 1} ), 700 ... rectifier diode, 800 ... inductance _(L 4), 900 ... load _{(R 0), 1000 ·· DC} -DC converter in the active clamp method.

Claims (6)

In the active clamp type forward converter,
On the primary side of the main transformer, the primary winding of the main transformer disposed between a pair of main switching elements;
An active clamp circuit disposed on the secondary side of the main transformer;
A feedback circuit for reducing the switching frequency of the pair of main switching elements when the load current is large;
The active clamp type forward converter, wherein the pair of main switching elements and the switching elements of the active clamp circuit are alternately turned on. In the active clamp type forward converter according to claim 1,
The active clamp circuit is:
An active clamp type forward converter comprising a secondary winding of the main transformer, a capacitor, and the switching element. In the active clamp type forward converter according to claim 1 or 2,
An active clamp type forward converter characterized in that the excitation method of the main transformer is positive and negative excitation. In the drive method of the active clamp type forward converter,
On the primary side of the main transformer, the primary winding of the main transformer disposed between a pair of main switching elements;
An active clamp circuit disposed on the secondary side of the main transformer,
Turning on the pair of main switching elements and turning off the switching elements of the active clamp circuit;
Turning off the pair of main switching elements and turning on the switching elements of the active clamp circuit,
An active clamp type forward converter driving method, wherein a feedback circuit that changes a switching frequency in response to a load current reduces the switching frequency of the pair of main switching elements when the load current is large. In the driving method of the forward clamp type forward converter according to claim 4,
The active clamp circuit is:
A drive method for an active clamp type forward converter, comprising: a secondary winding of the main transformer; a capacitor; and the switching element. In the driving method of the active clamp type forward converter according to claim 4 or 5,
An active clamp type forward converter driving method, wherein the main transformer excitation method is positive or negative excitation.

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