JP5741673B2 - Rotor phase estimation device for synchronous motor - Google Patents

Description

The present invention relates to a synchronous motor (for example, a permanent magnet synchronous motor having a permanent magnet in the rotor, winding-type synchronization) in which the rotor exhibits salient pole characteristics with respect to application of a signal having a frequency higher than the drive fundamental frequency (for example, high frequency voltage). The phase (synonymous with position) and speed of the rotor used in the drive control device for an electric motor, a synchronous reluctance motor, a hybrid field synchronous motor having a permanent magnet and a field winding on the rotor, etc. The present invention relates to a rotor phase estimation device for estimation without using a position velocity sensor, that is, sensorless.

High-performance control of the synchronous motor can be achieved by a so-called vector control method. The vector control method requires information on the rotor phase, and conventionally a position speed sensor such as an encoder has been used. However, the use of this type of position / velocity sensor is not preferable in terms of reliability, axial volume, sensor cable routing, cost, and the like, so-called sensorless vector control method that does not require a position / velocity sensor. Research and development has been conducted for many years.

A powerful sensorless vector control method is a high-frequency signal application method in which a high-frequency signal with a frequency higher than the drive fundamental frequency is forcibly applied to the motor, and the response high-frequency signal that is the response is detected and processed to estimate the rotor phase. Various developments and reports have been made. The high-frequency signal application method applies a high-frequency voltage to detect and process a high-frequency current that is a response to obtain a rotor phase estimate (in a broad sense), and a high-frequency voltage that is a response by applying a high-frequency current. (In a broad sense) a high-frequency current application method for obtaining and estimating the rotor phase estimation value. The practicality is the former (broad sense) high-frequency voltage application method. Considering this point, hereinafter, the description will focus on a high-practical high-frequency voltage application method. The high-frequency current application method will be supplementarily described as necessary.

The rotor phase to be estimated may be determined at an arbitrary position of the rotor, but generally, either the negative salient pole phase or the positive salient pole phase of the rotor is selected as the rotor phase. As is well known to those skilled in the art, there is only an electrical phase difference of ± π / 2 (rad) between the negative salient pole phase and the positive salient pole phase. The other phases are naturally found. Considering the above, hereinafter, the negative salient pole phase of the rotor will be referred to as the rotor phase unless otherwise specified.

The high-frequency voltage application method in a broad sense includes a narrow-sense high-frequency voltage application method that determines the method of generating the high-frequency voltage to be applied, and a phase estimation that determines the generation of the rotor phase estimate by processing the high-frequency current that is the response of the applied high-frequency voltage Composed as a combination with law. Narrowly defined high-frequency voltage application methods that apply sinusoidal high-frequency voltage include generalized elliptical high-frequency voltage application method, constant circular high-frequency voltage application method (constant-amplitude high-frequency voltage application method), and linear high-frequency voltage application method. Is at least known.

また、一定真円形高周波電圧印加法による正弦形状高周波電圧は次式で表現される。

また、直線形高周波電圧印加法による正弦形状高周波電圧は次式で表現される。

The sinusoidal high-frequency voltage can be applied on the uvw coordinate system, on the αβ fixed coordinate system, or on the γδ quasi-synchronous coordinate system aiming at synchronization without phase difference to the dq synchronous coordinate system with the rotor phase as the d-axis phase. Application is also possible on the coordinate system. When applying voltage on a γδ general coordinate system that rotates at a coordinate speed _ωγ , including αβ fixed coordinate system, dq synchronous coordinate system, and γδ quasi-synchronous coordinate system as special cases, a sine by a generalized elliptical high-frequency voltage application method The shape high frequency voltage is expressed by the following equation.

Further, the sinusoidal high-frequency voltage by the constant true high-frequency voltage application method is expressed by the following equation.

The sinusoidal high-frequency voltage obtained by the linear high-frequency voltage application method is expressed by the following equation.

As clearly shown in an unquestionable form using equations (1) to (3) above, the signal of each high-frequency voltage vector element is a sine shape (a shape expressed by a cosine function or a sine function). ). In order to apply the sine-shaped high-frequency voltage, a sine-shaped high-frequency voltage command value is given from the outside of the current control loop, and this is used as an input to the power converter (inverter). Thereby, a sinusoidal high frequency voltage can be generated and applied. The narrowly defined high frequency voltage application method handled by the present invention applies the sinusoidal high frequency voltage as described above. In particular, the amplitude of the γ-axis element (first element, cosine function notation) and the δ-axis element (second element, sine function notation) of the applied high-frequency voltage are the same, and the locus of the applied high-frequency voltage is a perfect circle locus. It will be.

There are various types of phase estimation methods for applying a sinusoidal high-frequency voltage (in a narrow sense). The phase estimation methods related to the present invention include vector heterodyne method, scalar heterodyne method (hereinafter referred to as heterodyne method). Generic name). With respect to the heterodyne method, there are prior inventions listed in the “prior art document” column below.

The vector heterodyne method was developed for a constant true circular high-frequency voltage application method (Non-Patent Documents 1 to 3), and other narrowly defined high-frequency voltage application methods (generalized elliptical high-frequency voltage application method, linear shape). There are no examples showing application to high-frequency voltage application methods. On the other hand, the scalar heterodyne method was developed for a linear high-frequency voltage application method (Patent Document 1, Non-Patent Documents 4 to 6), and other narrowly defined voltage application methods (generalized elliptical high-frequency voltage application method, There is no example showing application to a constant circular high-frequency voltage application method. These heterodyne methods were originally developed as a phase estimation method in a very low speed region near zero speed, and there is no example showing stable use in a medium to high speed region.

In general, application of a high-frequency voltage for rotor phase estimation can be performed on any coordinate system of the uvw coordinate system, the αβ fixed coordinate system, and the γδ quasi-synchronous coordinate system. Further, the processing of the high-frequency current that is the response of the applied high-frequency voltage can be performed on any coordinate system. When high-frequency voltage application and high-frequency current processing are performed on the γδ quasi-synchronous coordinate system, the frequency of the drive current on the coordinate system is constantly zero (ie, direct current). The current has a frequency difference ωh substantially equal to the high frequency ωh at an arbitrary speed. In general, in the processing of a high-frequency current having rotor phase information, it is preferable to eliminate the influence of a driving current that does not have rotor phase information. The large frequency difference ωh between the two currents is advantageous for the separation of both currents and the selective suppression of high-frequency residual disturbance. In consideration of the comprehensive simplicity of rotor phase estimation, the present invention mainly applies a high frequency voltage on the γδ quasi-synchronous coordinate system and processes the high frequency current on the same coordinate system.

Non-Patent Document 3 is a conventional vector heterodyne method for estimating a rotor phase on a γδ quasi-synchronous coordinate system. FIG. 10 is a reproduction of the drawing presented in this document. In this document, first, a high-frequency current is extracted from the stator current, and then in the reverse-phase order with respect to the extracted high-frequency current and the applied high-frequency voltage (expressed by the equation (2) with a positive phase and a constant amplitude). A high-frequency product signal is generated through multiplication with a vector carrier signal, and then the high-frequency product signal is processed by a low-pass filter (LPF) to extract a DC component from which the high-frequency component has been removed. A rotational phase estimation value is generated as an input signal to a secondary DC PLL block including a primary PI type phase controller. As is clear from the figure and the above description, the feature of this estimation method is that the high-frequency product signal is first processed by a low-pass filter for high-frequency component removal to extract only the DC component, and then this DC component is converted to a DC PLL block. It is in the point to be an input signal to.

As a conventional scalar heterodyne method for estimating a rotor phase on a γδ quasi-synchronous coordinate system, there are Patent Document 1 and Non-Patent Documents 4-6. FIG. 11 is a reproduction of the drawing presented in Patent Document 1. The phase estimation procedure based on this is explained as follows, as understood from the figure. On the γδ quasi-synchronous coordinate system, the linear high-frequency voltage command value of equation (3) is given from outside the current control loop. Then, first, this high frequency component (that is, the δ axis element of the high frequency current) is extracted from the δ axis element of the stator current on the γδ quasi-synchronous coordinate system. Next, a product of the extracted high-frequency current δ-axis element and a scalar carrier signal having a phase difference of −π / 2 (rad) with respect to the applied high-frequency voltage is taken to generate a high-frequency product signal. Subsequently, the high-frequency product signal is subjected to high-frequency component removal processing using a moving average low-pass filter (FIR digital filter) to extract a DC component. Finally, the DC component is input to a secondary DC PLL block having a primary PI type phase controller to generate a rotational phase estimation value. As is clear from the figure and the above description, the feature of this estimation method is that the high-frequency product signal is first processed by a high-frequency component removing low-pass filter to extract the DC component from which the high-frequency component has been removed, and then the DC component. Is the input signal to the DC PLL block. Similarly, in Non-Patent Documents 4 to 6, first, a high-frequency product signal is filtered by a high-frequency component removing low-pass filter to extract a DC component from which the high-frequency component has been removed, and then this DC component is represented by a DC PLL block. As an input signal to the phase estimation block.

However, it is known that the PLL estimation system is easily destabilized by the conventional phase estimation method in which a high-frequency component removing low-pass filter is placed in front of the DC PLL block. The high-frequency component removal low-pass filter placed in front of the DC PLL block results in introducing dynamics into the PLL estimation system, which is a phase estimation loop, and this greatly destabilizes the PLL estimation system. It became a factor. However, the prior art cannot allow the input signal to the DC PLL block to have a high frequency component, and although it may be a destabilizing factor for the first purpose, such as phase estimation, it eliminates the high frequency component. The introduction and pre-installation of low-pass filters for use was inevitable.

As will be clarified later in the description of the present invention, the magnitude of the DC component obtained by processing the high-frequency product signal with the high-frequency component removing low-pass filter is at least a function of the amplitude of the high-frequency voltage, the frequency, and the stator inductance of the motor. It becomes. For this reason, it is very difficult to design a primary PI type phase controller for the PLL estimation system ignoring these, and every time the amplitude and frequency of the high frequency voltage are changed, the primary PI type phase controller. Has been forced to redesign. Non-Patent Document 2 repeatedly points out the above-mentioned problem in designing a controller in a phase estimation system. However, the design method of the controller that guarantees the stability of the phase estimation system (corresponding to the phase controller in the DC PLL) remains unestablished, including the structure that should be.

Yasuhiro Yamamoto: “PM motor control device”, published patent publication, JP 2003-348896 (2002-5-24)

L.Wang and R.D.Lorenz: “RotorPosition Estimation for Permanent-Magnet Synchronous Motor UsingSaliency-Tracking Self-Sensing Method”, ConferenceRecord of 2000 IEEE Industry Applications Conference (IAS 2000), pp. 445-450, (2000-10) Y. Chen, L. Wang, and LKong: “Research of Position SensorlessControl of PMSM Based on High Frequency Signal Injection”, Proc. Of International Conference of Electrical Machines and Systems (ICEMS 2008), pp.3973-3977 (2008-10) Junichiro Kondo, Takashi Yoneyama, Shun Taniguchi, Shinsuke Mochizuki, Shinji Wakao: “Study on rotational angular velocity sensorless control of permanent magnet synchronous motor for railway drive, simple and high performance control system”, IEEJ Technical Committee Materials, SPC-06- 185, LD-06-87, pp. 37-42 (2006) JHJang, SKSul, JIHa, K.Ide, and M. Sawamura: “Sensorless Driveof SMPM Motor by High-Frequency Signal Injection Based on Magnet Saliency”, Proc. Of 17th IEEE Applied Power Electronics Conference and Exposition (APEC 2002), Vol. 1, pp. 279-285 (2002-3). JHJang, SKSul, JIHa, K.Ide, and M. Sawamura: “Sensorless Driveof Surface-Mounted Permanent-Magnet Motor by High-Frequency Signal Injection Based on Magnet Saliency”, IEEE Trans. On IndustryApplications, Vol. 39, No .4, pp. 1031-1039 (2003-7 / 8) Y. Nakano, H. Sugiyama, Y. Yamamoto, and T. Ashikaga: “Sensor-less Vector Control System Using Concentrated Winding Permanent Magnet Motor”, Proc. Of 22nd International Battery, Hybrid and Fuel Cell Electric Vehicle Symposium & Exposition (EVS22), pp 677-686 (2006-10) Shinnaka Shinji: “Vector control technology of permanent magnet synchronous motor, first volume (from principle to the latest)”, Denpa Shimbun (2008-12) Shinnaka Shinji: “Vector control technology of permanent magnet synchronous motor, second volume (the essence of sensorless drive control)”, Denpa Shimbun (2008-12)

As is apparent from the above description, the conventional heterodyne method as a phase estimation method for the narrowly defined high-frequency voltage application method has the following problems .
2) A high-frequency component removing low-pass filter is placed in front of the direct-current PLL block to remove the high-frequency component from the high-frequency product signal, thereby generating a direct-current PLL block input signal consisting only of the direct-current component. However, the stability of the PLL estimation system is often impaired and destabilized due to the low-pass filter for removing high-frequency components.
3) The structure of the PLL estimation system for ensuring the stability of the PLL estimation system and the controller design method for the PLL estimation system have not been established, and the controller coefficients have been designed on a trial and error basis.
4) The DC PLL block input signal consisting only of the DC component is at least a function of the amplitude, frequency, and stator inductance of the high-frequency voltage, and the PLL estimation system has become unstable due to these changes or fluctuations. Since the controller design method has not been established, a great deal of effort and trial and error has been imposed on the redesign of the phase controller in the PLL estimation system as the high frequency voltage is changed.

The present invention has been made under the above-mentioned background. As a method for applying a high-frequency voltage in a narrow sense, a generalized elliptical high-frequency voltage application method (with elliptic coefficient K in equation (1) set to 1), or a constant perfect circular high-frequency voltage. It is an object of the present invention to provide a phase estimation device having the following characteristics with respect to a phase estimation device using an application method (see equation (2)).
2) To provide a phase estimation device that is structurally easy to ensure stability and that does not require the introduction or preposition of a high-frequency component removal low-pass filter that has become a major factor in the instability of a PLL estimation system.
3) A phase estimation apparatus having a design method capable of ensuring the stability of a PLL estimation system in the structure shown in the second) objective is provided.
4) Provided is a phase estimation device that does not require any redesign of a phase controller or the like in an estimation system even when there is a change in the amplitude, frequency, and stator inductance of a high-frequency voltage.

で表現するとき、次形式の３次以上の有理関数(sは微分演算子またはラプラス演算子、ｃ_ｄｉ,ｃ_ｎｉは係数)

として記述され、高周波積信号自体を直接入力として回転子位相推定値を生成出力する高周波PLL手段と、を備えることが記載されている。 In order to achieve the above object, the present specification uses a rotor for a drive control device for a synchronous motor in which salient pole characteristics are exhibited with respect to application of a high-frequency signal having a frequency higher than the drive fundamental frequency, and A rotor phase estimation device that generates and outputs a rotor phase estimation value through detection and processing of a response high frequency signal, which is a response of an applied high frequency signal, to a power converter in a drive control device for high frequency voltage application A high-frequency voltage command value that is included in the voltage command value and has a sinusoidal shape on a two-axis orthogonal γδ quasi-synchronous coordinate system with the rotor phase estimation value as the base axis (γ-axis) phase is external to the stator current control loop. Generating means and a carrier signal having a δ-axis element as a signal having a phase difference of about −π / 2 (rad) or about + π / 2 (rad) with respect to the γ-axis element of the high-frequency voltage command value; Due to carrier signal and high frequency voltage command value Through multiplication process using a high-frequency current, or through multiplication process using a stator current containing carrier signal and high frequency current, means for generating a high-frequency product signal which may contain high frequency components, the high frequency product signal u _{In the PLL} , the rotor phase estimate is

Is expressed by a rational function of the third or higher order (where s is a differential operator or Laplace operator, and c _di and c _ni are coefficients)

And a high-frequency PLL means for generating and outputting a rotor phase estimation value using the high-frequency product signal itself as a direct input.

が区間０＜Ｋ_ｈ≦２の任意の振動係数Κ_ｈに対してフルビッツ多項式（安定多項式）となるように、定めることが記載されている。 Polynomial this specification, having a rotor phase estimation apparatus, the coefficients of the organic physical function that describes the high-frequency PLL means, the amplitude of the indicia pressurized high-frequency voltage command value, a signal coefficient kappa _theta determined from frequency, etc.

Is defined to be a Hurwitz polynomial (stable polynomial) for an arbitrary vibration coefficient Κ _{h in the} interval 0 <K _h ≦ 2.

The specification of the present application describes the above-described rotor phase estimation device that performs generation of the high-frequency product signal on a γδ quasi-synchronous coordinate system.

The present specification describes the above-described rotor phase estimation device, in which the γ-axis element of the carrier signal is always set to zero, and the carrier signal is substantially a scalar signal.

The invention according to claim 1 is used in a drive control device for a synchronous motor in which a rotor exhibits salient pole characteristics with respect to application of a high-frequency signal having a frequency higher than the drive fundamental frequency, and the response of the applied high-frequency signal. A rotor phase estimation device that generates and outputs a rotor phase estimation value through detection and processing of a response high-frequency signal, and is included in a final signal command value to a power converter in the drive control device for applying a high-frequency signal And a means for generating a high-frequency signal command value that draws a perfect circular locus and a means for extracting a negative-phase component that is out of phase with the applied high-frequency signal from the response high-frequency signal resulting from the high-frequency signal command value And means for capturing the extracted anti-phase component as a vector, normalizing the norm to a value typified by 1, and generating a normalized anti-phase component, and a carrier signal having an anti-phase order with respect to the high-frequency signal command value Generated using the generated carrier signal to process the normalized inverse phase component, characterized by comprising: means for generating and outputting a rotor phase estimate value.

The invention according to claim 2 is the rotor phase estimation device according to claim 1, wherein the generated high-frequency signal command value is set as a high-frequency voltage command value, and the response high-frequency signal resulting therefrom is set as a high-frequency current. It is characterized by doing.

A third aspect of the present invention is the rotor phase estimation device according to the first aspect, wherein the processing of the normalized antiphase component using the carrier signal is performed by using the rotor phase estimation value as a base axis (γ axis). It is carried out on a biaxial orthogonal γδ quasi-synchronous coordinate system as a phase.

The effects described in the present specification will be described using mathematical expressions. As shown in FIG. 1, a γδ general coordinate system rotating at an arbitrary speed ω _γ is considered (see Non-Patent Document 7). The rotation from the main axis (γ axis) to the sub axis (δ axis) is defined as the positive direction. Further, it is assumed that the rotor phase of the synchronous motor having salient pole characteristics has a phase θ _γ instantaneously with respect to the γ axis. Unless otherwise specified, all 2x1 vector signals that represent the physical quantities of the synchronous motor to be handled below are defined on this coordinate system. As a special case, the γδ general coordinate system includes an αβ fixed coordinate system, a dq synchronous coordinate system, and a γδ quasi-synchronous coordinate system. That is, the γδ quasi-synchronous coordinate system is a special case of the γδ general coordinate system, and in consideration of this point, both coordinate systems adopt the same expression.

In the following description, without losing generality, the motor rotates in the positive direction including zero speed, and the speed of the γδ general coordinate system is also positive including zero. In addition, for the rotor phase estimation, the constant frequency ω _h of the high frequency voltage superimposed on the driving voltage is positive. This premise is for distinguishing the high-frequency voltage and the positive-phase and negative-phase components of the high-frequency current, so that the generality of the explanation and the essence of the principle of the present invention are not lost (non-patent document). 8). The above assumptions are naturally applied to the high-frequency voltages shown in the equations (1) to (3). The high-frequency voltage having a sine shape can be expressed using either a cosine function (cos function) or a sine function (sin function), but in the description of the present invention, it is expressed by equations (1) to (3). Like the high-frequency voltage, the γ-axis element (first element) of the applied voltage is expressed by a cosine function. In addition, the phase of other signals is expressed based on the γ-axis element (first element) of the applied voltage (see Non-Patent Document 8).

また、2x2行列であるベクトル回転器Ｒ（θ_γ）を以下のように定義する（非特許文献８）。

For the sake of simplicity in the following explanation, some signals, functions, etc. are defined. A normal phase signal u _p (ω _h t) and a negative phase signal u _n (ω _h t) that exist on the γδ general coordinate system and have a constant high frequency ω _h and a unit norm are defined as follows (non-patent document) Reference 8).

Also, a vector rotator R (θ _γ ) that is a 2 × 2 matrix is defined as follows (Non-patent Document 8).

(9)、(10)式に示した正相信号、逆相信号においては、そのδ軸要素（すなわち、第２要素、sinω_ht）は印加高周波電圧のγ軸要素（すなわち第１要素、cosω_ht、(1)〜(3)式参照）に対して、高周波数ωhが正の場合には−π/2(rad)の位相差を有する点には、反対に高周波数ωhが負の場合に約＋π/2(rad)の位相差を有する点には、特に注意されたい。本発明では、このような信号を、高周波積信号生成のためのキャリア信号として利用する。このため、これらの信号をキャリア正相信号、キャリア逆相信号と呼称する。 Furthermore, (6), (7) a positive-phase signal type, the positive phase signal having a [pi / 2 (rad) phase lead with respect to the phase-inverted signal, respectively u _uk reverse phase signal _{(omega h} t), u _It is defined as _jn (ω _h t). That is,

In the positive phase signal and the negative phase signal shown in the equations (9) and (10), the δ-axis element (that is, the second element, sinω _h t) is the γ-axis element (that is, the first element, cos ω _h t (see equations (1) to (3)), when the high frequency ωh is positive, on the contrary, the high frequency ωh is negative when it has a phase difference of −π / 2 (rad). In particular, it should be noted that the phase difference is approximately + π / 2 (rad). In the present invention, such a signal is used as a carrier signal for generating a high-frequency product signal. For this reason, these signals are called a carrier positive phase signal and a carrier negative phase signal.

(11)式が明示しているように、一般には、高周波電流の正相成分i_hp、逆相成分i_hnのいずれにも、回転子位相情報がＲ(２θ_γ)ｕ_ｐ（ω_ｈｔ）,Ｒ（２θ_γ）ｕ_ｎ（ω_ｈｔ）という形式で含まれている。真円軌跡を描く真円形高周波電圧を印加する場合に限り、次の(12)式が成立し、回転子位相情報は高周波電流の逆相成分のみに含まれる（非特許文献８参照）。

When a high-frequency voltage such as equations (1) to (3) having a sine shape is applied to a synchronous motor having saliency on the γδ general coordinate system, the high-frequency current i _1h corresponding to this response is On the coordinate system, the following expression is obtained (see Non-Patent Document 8).

As (11) is indicated, generally, the positive phase component i _hp of the high-frequency current, in any of the anti-phase component i _hn, the rotor phase information _{R (2θ γ) u p (} ω h t ), it is included in the form of _{R (2θ γ) u n (} ω h t). Only when a perfect circular high-frequency voltage that draws a perfect circular locus is applied, the following equation (12) is established, and the rotor phase information is included only in the anti-phase component of the high-frequency current (see Non-Patent Document 8).

上式における脚符ｆ、ｈは、それぞれ駆動周波数、高周波数の成分であることを示している。 (11) of the high-frequency current is included as a high frequency component of the stator current i _1. This can be expressed as follows using mathematical formulas.

The symbols f and h in the above formula indicate that they are components of a driving frequency and a high frequency, respectively.

(14)式のキャリア和信号も、本信号の元となった(9)、(10)式のキャリア正相信号、キャリア逆相信号と同様に、そのδ軸要素（すなわち、第２要素、sinω_ｈｔ）は、印加高周波電圧のγ軸要素（すなわち第１要素、cosω_ht、(1)〜(3)式参照）に対して±π/2(rad )の位相差を有する点には、特に注意されたい。 In the first aspect of the present invention, the carrier positive phase signal, the carrier negative phase signal of Equations (9) and (10), or a signal obtained by vector synthesis of both is used as the carrier signal for generating the high frequency product signal. This representative is the next carrier sum signal u _j (ω _h t).

Similarly to the carrier positive phase signal and carrier negative phase signal of (9) and (10), the carrier sum signal of equation (14) is also the δ-axis element (that is, the second element, sin .omega _{h t)} is, gamma longitudinal element (i.e. the first element, cos .omega _h t, (1) of the applied high frequency voltage to a point having a phase difference of - (3) see formula) ± [pi / 2 with respect to (rad) Please be especially careful.

ここに、ｉ_δｆは駆動用電流（固定子電流の駆動周波数成分）ｉ_１ｆのδ軸要素を意味する。 In the present specification, a high-frequency product signal that can contain a high-frequency component is generated through multiplication processing using a carrier signal and a high-frequency current or a stator current including a high-frequency current. For example, high-frequency product signal generated through multiplication of the carrier sum signal u _j and _{(ω h} t) and the stator current i ₁ as a carrier signal becomes the following equation.

Here, i _δf means the δ-axis element of the drive current (drive frequency component of the stator current) i _1f .

(15) the first term on the right side s _h of the high frequency product signal in the equation has a rotor phase and positive correlation range, the most important signal on the rotor phase detection. In the present invention, the first term is referred to as a high-frequency positive correlation signal. RF positive correlation signal is composed of high-frequency component of the DC component and frequency 2 [omega _h, becomes zero only if θ γ = _0. As understood from the equation (11), out of the coefficient (g _pm + g _nm ) governing the amplitude of the high-frequency positive correlation signal, the first coefficient is from the positive phase component of the high-frequency current, and the second coefficient is the inverse of the high-frequency current. Comes from phase components. The phase information of the high-frequency current positive phase component is obtained by multiplication processing (inner product processing) with the carrier positive phase signal. Further, the phase information of the high-frequency current anti-phase component is obtained by multiplication processing (inner product processing) with the carrier anti-phase signal. Since the carrier sum signal is a composite signal of both the positive and negative phases of the carrier positive phase signal and the carrier negative phase signal, it is possible to extract both phase information contained in the positive phase component and the negative phase component of the high-frequency current. Naturally, when a high-frequency product signal is generated by multiplying only the carrier positive-phase signal and the stator current, only phase information of the positive-phase component of the high-frequency current can be extracted. Similarly, when a high-frequency product signal is generated by multiplying only the carrier reverse-phase signal and the stator current, only phase information possessed by the reverse-phase component of the high-frequency current can be extracted. In the present specification, such a high-frequency product signal is expressed as an input signal u _PLL to the high-frequency PLL means (see equation (4)).

上式の右辺第2項は、駆動用電流ｉ_１ｆのδ軸要素ｉ_δｆに起因する項であり、固定子電流に代わって高周波電流とキャリア信号との乗算処理による場合には、本項は存在しない。換言するならば、固定子電流から駆動用電流を除去することにより、本項は除去できる。一方、右辺第1項は、印加高周波電圧によって支配される項であり、高周波積信号に常時残留し、回転子位相推定上の外乱として作用する。高周波積信号に含まれる残留成分(本発明では、高周波残留外乱と呼称している)に対する位相推定上の対策の良し悪しが、ヘテロダイン法の性能を支配する。 The second term n _h on the right-hand side of the high-frequency product signal in equation (15) indicates the high-frequency component of the frequencies 2ω _h and ω _h , which is non-zero regardless of the rotor phase. This high frequency component is arranged as follows when θ _γ = 0 where the rotor phase is zero.

The second term on the right side of the above equation is a term resulting from the δ-axis element i _δf of the driving current i _1f , and this term is obtained when the high-frequency current is multiplied by the carrier signal instead of the stator current. not exist. In other words, this term can be removed by removing the driving current from the stator current. On the other hand, the first term on the right side is a term governed by the applied high-frequency voltage, and always remains in the high-frequency product signal and acts as a disturbance in estimating the rotor phase. The quality of the phase estimation measure against the residual component contained in the high frequency product signal (referred to as high frequency residual disturbance in the present invention) dominates the performance of the heterodyne method.

この場合には、位相積分器の出力が高周波PLLブロックの出力となり、回転子位相推定値（γδ準同期座標系の位相）

となる。同時に、位相積分器の入力（高周波位相制御器Ｃ（ｓ）の出力）が、γδ準同期座標系の速度ω_γとなる。なお、同図では、固定子電流の駆動用成分のδ成分の分離除去が可能である点を考慮して、これに起因する高周波残留外乱は破線で示している。(4)式、(17)式、更には図２により明白なように、高周波積信号は、ローパスフィルタ処理されることなく、直接的に、高周波PLLブロックに入力されている。この点には特に注意されたい。 An example is shown. Consider the use of the equation (15) obtained through the multiplication of the carrier sum signal and the stator current as the direct input (ie, the high frequency product signal) of the high frequency PLL means shown in equation (4). Furthermore, the γδ general coordinate system is a γδ quasi-synchronous coordinate system in this special case. In this case, the PLL estimation system based on the equations (15) and (4) can be expressed as shown in FIG. In the example shown in the figure, the rational function of the equation (4) is realized by separating the high-frequency phase controller C (s) and the phase integrator 1 / s as the following equation (17).

In this case, the output of the phase integrator becomes the output of the high-frequency PLL block, and the rotor phase estimation value (phase of the γδ quasi-synchronous coordinate system)

It becomes. At the same time, the input of the phase integrator (the output of the high-frequency phase controller C (s)) becomes the velocity ω _γ of the γδ quasi-synchronous coordinate system. In the figure, in consideration of the fact that the δ component of the driving component of the stator current can be separated and removed, the high-frequency residual disturbance resulting from this is indicated by a broken line. As is clear from the equations (4), (17), and FIG. 2, the high frequency product signal is directly input to the high frequency PLL block without being subjected to the low pass filter processing. Special attention should be paid to this point.

本願明細書に記載されている効果の説明に関連して、具体例を交えながら改めて説明するが、高周波PLL手段（高周波PLLブロック）を定めた有理関数の次数を３次以上とする場合には、（すなわち、高周波位相制御器の次数を２次以上とする場合には）、(18)式の減衰率を十分に小さくできる。 The PLL estimation system of FIG. 2 has a high degree of design freedom because the high-frequency PLL block is configured with a rational function C (s) / = C _N (s) / (sC _D (s)) of the third or higher order. . By using this degree of freedom, it is possible to ensure the stability of the PLL estimation system (design of a high-frequency phase controller for stabilization is performed in connection with the explanation of the effects described in the present specification). . In this case, the high-frequency residual disturbance having a high frequency of 2ωh appears regularly after being amplified or reduced by multiplying the phase estimation value by the rate of the following equation.

In connection with the description of the effect described in the present specification, a description will be given again with specific examples. When the order of the rational function that defines the high-frequency PLL means (high-frequency PLL block) is set to the third order or higher, (I.e., when the order of the high-frequency phase controller is second or higher), the attenuation factor of the equation (18) can be made sufficiently small.

As explained using equation (15), according to the description of the present specification, a narrowly defined high-frequency voltage application method for applying a sinusoidal high-frequency voltage (generalized elliptical high-frequency voltage application method, constant circular high-frequency voltage application method) Method, linear phase high-frequency voltage application method, etc.) can be incorporated into the high-frequency product signal. Further, as described with reference to equations (17) and (18) and FIG. 2, the PLL estimation system having a high-order high-frequency phase controller can be stabilized (details of the stabilization are described later in the invention). The rotor phase estimation value can be obtained by substantially eliminating the influence of the high-frequency residual disturbance, which will be described in relation to the effect). These mean that the following effects can be obtained according to the description of the present specification.
1) Versatile, widely applicable to narrowly defined high-frequency voltage application methods (including generalized elliptical high-frequency voltage application method, constant true high-frequency voltage application method, linear high-frequency voltage application method, etc.) for applying sinusoidal high-frequency voltage It is possible to realize a phase estimation device rich in.
2) It is possible to realize a phase estimation device that does not require the introduction / prefix of a high-frequency component removal low-pass filter, which has become a major factor in destabilizing the PLL estimation system, and is structurally easy to ensure stability.

が回転子位相真値θ_αに概ね収斂した状態では、回転子位相真値から同推定値に至る伝達特性Ｆ_Ｃ（ｓ）は、以下のように近似表現される。

Then, the effect described in this specification is demonstrated. In the PLL estimation system described above, the rotor phase estimation value

There in the state where almost converges to the rotor phase true value theta _alpha, the transfer characteristic F _C leading to the estimate from the rotor phase true value _(s) is approximately represented as follows.

信号係数Ｋ_θは、印加高周波電圧の振幅、周波数、インダクタンス等によって定まる係数である。信号係数は、例えば、(1)〜(3)式に示した一般化楕円形高周波電圧、一定真円形高周波電圧、直線形高周波電圧では、各々次式となる。

上式におけるLmは、ｄ軸インダクタンスLd、ｑ軸インダクタンスLqから一意に定まる鏡相インダクタンスである（非特許文献７参照）。また振動係数Khは、(20)式に明示しているように、０〜２の間で高周波振動するゲイン相当値を係数化したものである。 Transfer characteristics of the above, in FIG. 2, to remove the high-frequency residual disturbance, it can also be obtained by approximating as follows RF positive correlation signal s _h of (15).

The signal coefficient _Kθ is a coefficient determined by the amplitude, frequency, inductance, and the like of the applied high frequency voltage. For example, the signal coefficients for the generalized elliptical high-frequency voltage, the constant true high-frequency voltage, and the linear high-frequency voltage shown in the equations (1) to (3) are as follows.

Lm in the above equation is a mirror phase inductance uniquely determined from the d-axis inductance Ld and the q-axis inductance Lq (see Non-Patent Document 7). The vibration coefficient Kh is obtained by converting a gain equivalent value that vibrates at a high frequency between 0 and 2 as specified in the equation (20).

According to the present specification, the rational function of the polynomial F _D (s) defined by the equation (5) is such that it becomes a Hurwitz polynomial (that is, a stable polynomial) for an arbitrary vibration coefficient 0 <K _h ≦ 2. The coefficient (same as the coefficient of the high frequency controller coefficient) is determined. The polynomial defined by equation (5) is the denominator polynomial itself having the transfer characteristic of equation (19). As a result, the denominator polynomial is a Hurwitz polynomial despite the high frequency vibration of the vibration coefficient. As a result, the PLL estimation system having the transfer characteristic of Equation (19) operates stably. That is, according to the description of the present specification, the effect that the rotor phase estimation apparatus including the above-described means can operate stably can be obtained.

３次分母多項式が安定な３重実根p0をもつようにするには、有理関数係数（高周波位相制御器係数）は以下のように定めればよい。

この場合、(18)式で記述した高周波残留外乱の低減率は次式となり、一般に十分な低減が確保される。

When the high-frequency phase controller is quadratic in the form of the following equation and the rational function is cubic, the rational function coefficient (high-frequency phase control) is used to stabilize the PLL estimation system when the vibration coefficient Kh is 1. If the design coefficient is designed, the PLL estimation system can be stabilized against any vibration coefficient.

In order for the third-order denominator polynomial to have a stable triple real root p0, the rational function coefficient (high-frequency phase controller coefficient) may be determined as follows.

In this case, the reduction rate of the high-frequency residual disturbance described by the equation (18) is as follows, and generally a sufficient reduction is ensured.

４次分母多項式が安定な４重実根p0をもつようにするには、有理関数係数（高周波位相制御器係数）は以下のように定めればよい。

この場合、(18)式で記述した高周波残留外乱の低減率は次式となり、一般に十分な低減が確保される。

If the high-frequency phase controller is cubic of the following form and the rational function is quaternary, the rational function coefficient (high-frequency phase control) is used so that the PLL estimation system stabilizes when the vibration coefficient Kh is 2. If the design coefficient is designed, the PLL estimation system can be stabilized against any vibration coefficient.

In order for the fourth-order denominator polynomial to have a stable quadruple real root p0, the rational function coefficient (high-frequency phase controller coefficient) may be determined as follows.

In this case, the reduction rate of the high-frequency residual disturbance described by the equation (18) is as follows, and generally a sufficient reduction is ensured.

４次分母多項式が安定な４重実根p0をもつようにするには、有理関数係数（高周波位相制御器係数）は以下のように定めればよい。

この場合、(18)式で記述した高周波残留外乱の低減率は次式のようにゼロとなり、理論上は完全な外乱排除が実施される。

If the high-frequency phase controller is cubic of the following form and the rational function is quaternary, the rational function coefficient (high-frequency phase control) is used so that the PLL estimation system stabilizes when the vibration coefficient Kh is 2. If the design coefficient is designed, the PLL estimation system can be stabilized against any vibration coefficient.

In order for the fourth-order denominator polynomial to have a stable quadruple real root p0, the rational function coefficient (high-frequency phase controller coefficient) may be determined as follows.

In this case, the reduction rate of the high-frequency residual disturbance described by the equation (18) becomes zero as shown in the following equation, and theoretically complete disturbance rejection is performed.

As can be understood from the design method presented above, the rational function coefficient (coefficient of the high-frequency phase controller) for carrying out the invention described in the specification is unique through the specification of the root of the denominator polynomial of the transfer characteristic. Can be determined. Thus, by providing another configuration in addition to the configuration described in the present specification, the effect that “a phase estimation device having a design method for guaranteeing the stability of the PLL estimation system can be realized” can be achieved. Can be obtained. As a result, the effect of the invention described in the specification can be enhanced.

Subsequently, other effects described in the specification will be described. As already described, the processing for the high-frequency current that is the response of the applied high-frequency voltage can be performed on any coordinate system of the uvw coordinate system, the αβ fixed coordinate system, and the γδ quasi-synchronous coordinate system. When high-frequency voltage application and high-frequency current processing are performed on the same γδ quasi-synchronous coordinate system, the frequency of the drive current on the same coordinate system is constantly zero. At an arbitrary speed, the frequency difference ωh is substantially equal to the high frequency ωh. As a result, the frequency of the high-frequency residual disturbance included in the high-frequency product signal is 2ωh. A large frequency difference of the high-frequency residual disturbance with respect to the direct current is advantageous for this elimination. In fact, equations (24) and (27) show how the reduction rate of high-frequency residual disturbance is determined according to the power of 2ωh. As understood from the above description, according to the configuration described in the present specification, “high-frequency residual disturbance can be easily eliminated even in the above-described PLL estimation system that does not introduce / prepend a low-frequency filter for removing high-frequency components. The effect of “being able to do this” is obtained. As a result, the effect mentioned above can be heightened.

(31)式は、スカラキャリア信号と高周波電流のδ軸要素あるいは高周波電流を含む固定子電流のδ軸要素とのスカラ乗算処理で、高周波積信号が生成できることを意味している。ベクトル乗算と比較するならば、高周波積信号生成のための演算をおおよそ半減できる。以上の説明より明白なように、本願明細書に記載の構成によれば、「高周波電流の正相成分、逆相成分の両成分に含まれる回転子位相情報を、少ない演算量で、高周波積信号に含有させることができる」と言う効果が得られる。ひいては、上述の本願明細書に記載の構成の効果を高めることができる。 Subsequently, another effect described in the present specification will be described. As described using equation (15), when generating a high-frequency product signal using the carrier sum signal of equation (14), rotor phase information included in the positive-phase component and the negative-phase component of the high-frequency current Can be incorporated into a high-frequency product signal. The carrier sum signal at this time, as clearly shown on the right side of equation (14), means that the γ-axis element of the carrier signal is always zero and the carrier signal is substantially a scalar signal. In this case, the multiplication processing of the equation (15) showing the generation of the high frequency product signal can be rearranged as the following equation (31).

Equation (31) means that a high-frequency product signal can be generated by scalar multiplication of the scalar carrier signal and the δ-axis element of the high-frequency current or the δ-axis element of the stator current including the high-frequency current. Compared with vector multiplication, the operation for generating a high-frequency product signal can be roughly halved. As is clear from the above description, according to the configuration described in the present specification, “the rotor phase information contained in both the positive-phase component and the negative-phase component of the high-frequency current is reduced with a small amount of computation and the high-frequency product. The effect of “can be included in the signal” is obtained. As a result, the effect of the structure as described in this-application specification can be heightened.

Then, the effect of the invention of Claim 1 is demonstrated. According to the first aspect of the present invention, a high frequency signal that draws a perfect circular locus is applied, a response high frequency signal that is this response is detected, and a reverse phase component is extracted therefrom. Here, for the sake of simplicity, the effect of the present invention will be described assuming that the applied high-frequency signal is a high-frequency voltage and the response high-frequency signal is a high-frequency current. Further, as a high-frequency signal for drawing a perfect circle locus, a response type and a constant type are known, but here, the effect of the present invention is taken by taking the case of application of a constant perfect circular high-frequency voltage of equation (2) as an example. explain. The high-frequency current corresponding to equation (2) is described by the following equation as a special case of equation (11c) (see equation (12)).

(32)式の逆相成分は、印加高周波電圧の振幅、周波数等の影響を受けているが、(33)式の正規化逆相成分は、これらの影響を全て排除した形で、回転子位相情報のみを有している。 As clearly shown in the above equation, the component corresponding to the first term on the right side of the equation (11c) does not exist in the antiphase component, and is composed only of the component corresponding to the second term. According to the first aspect of the present invention, when the antiphase component of the equation (32) is normalized to the norm 1, for example, the following normalized antiphase component is obtained.

The anti-phase component of equation (32) is affected by the amplitude and frequency of the applied high-frequency voltage, but the normalized anti-phase component of equation (33) eliminates all these effects, and the rotor It has only phase information.

(34)式は、積信号は、γδ一般座標系のγ軸から見た回転子位相の正弦値そのものとなってること示している。特に、本積信号は、高周波残留外乱の影響も、印加高周波電圧の振幅、周波数の影響も、更には、電動機のインダクタンスの影響も排除した形となっている。すなわち、直流の積信号となっている。本積信号を、直流PLLブロック、トラッキングオブザーバ等に用いることにより、これらの影響を排除した状態で、αβ固定座標系のα軸からみた回転子位相推定値を得ることができる。上記は、印加高周波信号として高周波電圧を用いた場合の説明であるが、印加高周波信号として高周波電流を用いた場合にも同様の説明が可能である。 According to the first aspect of the present invention, a product signal is generated through multiplication processing (inner product processing) of the normalized antiphase component of Equation (33) and the carrier antiphase signal defined by Equation (10). That is,

Equation (34) indicates that the product signal is the sine value of the rotor phase as viewed from the γ-axis of the γδ general coordinate system. In particular, the product signal has a form that excludes the influence of high-frequency residual disturbance, the influence of the amplitude and frequency of the applied high-frequency voltage, and the influence of the inductance of the motor. That is, it is a direct current product signal. By using this product signal for a DC PLL block, a tracking observer, etc., it is possible to obtain a rotor phase estimation value viewed from the α axis of the αβ fixed coordinate system in a state where these influences are eliminated. The above is a description when a high-frequency voltage is used as the applied high-frequency signal, but the same description can be made when a high-frequency current is used as the applied high-frequency signal.

As is clear from the above description, according to the invention described in claim 1, “the influence of the high-frequency residual disturbance, the influence of the amplitude and frequency of the applied high-frequency voltage, and the influence of the inductance of the motor are excluded. In this way, the rotor phase estimation value can be obtained. ” Furthermore, “a phase estimation device that does not require any redesign of the phase controller or the like in the PLL estimation system can be configured even when the amplitude and frequency of the applied high-frequency voltage and the inductance of the motor are changed and changed.” The effect is said.

Then, the effect of the invention of Claim 2 is demonstrated. Application of a high frequency signal to the electric motor is performed via a power converter (inverter). There are a voltage type and a current type as power converters, but the voltage type is the mainstream today. When it is assumed that a voltage source power converter is used, it is easier to generate and apply a high frequency voltage as a high frequency signal having a perfect circular locus. As can be seen from the above, according to the second aspect of the present invention, an effect that “a high-frequency signal having a true circular locus can be easily applied to the motor” can be obtained. As a result, the effect which improves the effect of the invention of Claim 1 comes to be acquired.

Then, the effect of the invention of Claim 3 is demonstrated. According to the third aspect of the present invention, if the frequency of the applied high-frequency signal is ωh, this frequency can be the frequency difference between the negative-phase component and the driving component of the response high-frequency signal, Extraction of the reverse phase component becomes easier. As a result, according to the third aspect of the invention, an effect is obtained that the normalized antiphase component and the rotor phase estimation value can be obtained more easily. As a result, the effect which improves the effect of the invention of Claim 1 comes to be acquired.

"Figure showing an example of the relationship between the three coordinate systems and the rotor phase" "Figure showing the principle of rotor phase estimation according to the present invention" "Block diagram showing the basic configuration of the drive control apparatus in one embodiment" “Block diagram showing basic configuration of phase velocity estimator in one embodiment” "Test data diagram showing validity of invention in one embodiment" "Test data diagram showing validity of invention in one embodiment" "Test data diagram showing validity of invention in one embodiment" "Block diagram showing basic configuration of phase velocity generator in one embodiment" "Block diagram showing basic configuration of phase velocity generator in one embodiment" "Block diagram showing conventional vector heterodyne method" "Block diagram showing the conventional scalar heterodyne method"

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 3 shows an example of a drive control device provided with the rotor phase estimation device of the present invention for a permanent magnet synchronous motor which is a typical synchronous motor. Although the main point of the present invention is in the rotor phase estimation device, the entire motor drive control system including the drive control device will be described in order to clarify the positioning of the rotor phase estimation device in the entire motor drive control system. 1 is a synchronous motor, 2 is a power converter (voltage type), 3 is a current detector, 4a and 4b are 3 phase 2 phase converters, 2 phase 3 phase converters, 5a and 5b are both vectors Rotator, 6 Current controller, 7 Command converter, 8 Speed controller, 9 Band stop filter, 10 Phase speed estimator using the present invention, 11 Coefficient unit , 12 respectively indicate cosine sine signal generators. In FIG. 3, various devices from 2 to 12 except for one electric motor constitute a drive control device. In this figure, a 2 × 1 vector signal is represented by one thick signal line to ensure simplicity. The following block diagram expression follows this.

The three-phase stator current detected by the current detector 3 is converted into a two-phase current on the αβ fixed coordinate system by the three-phase two-phase converter 4a, and then moved to the rotor phase by the vector rotator 5a. The phase difference is converted into a two-phase current in a γδ quasi-synchronous coordinate system aimed at phase synchronization. A drive current is extracted from the converted current through the band stop filter 9 and sent to the current controller 6. The current controller 6 generates a driving two-phase voltage command value on the γδ quasi-synchronous coordinate system so that the driving two-phase current on the γδ quasi-synchronous coordinate system follows the current command value of each phase. Here, the two-phase high-frequency voltage command value received from the phase velocity estimator 10 is superimposed on the driving two-phase voltage command value, and the two-phase voltage command value superimposed and synthesized is sent to the vector rotator 5b. In 5b, the voltage command value for superposition and synthesis on the γδ quasi-synchronous coordinate system is converted into a two-phase voltage command value in the αβ fixed coordinate system, and sent to the two-phase three-phase converter 4b. In 4b, the two-phase voltage command value is converted into a three-phase voltage command value and output as a final command value to the power converter 2. The power converter 2 generates electric power according to the command value, applies it to the synchronous motor 1 and drives it.

The phase speed estimator 10 receives the stator current that is the output of the vector rotator 5a (and receives the command value of the driving current as necessary), and estimates the rotor phase estimated value and the electric speed of the rotor. Value and high-frequency voltage command value are output. The rotor phase estimation value is converted into a cosine / sine signal by the cosine sine signal generator 12, and then passed to the vector rotators 5a and 5b that determine the γδ quasi-synchronous coordinate system. This means that the rotor phase estimation value is the phase of the γδ quasi-synchronous coordinate system.

As is well known to those skilled in the art, the two-phase current command value on the γδ quasi-synchronous coordinate system is obtained by converting the torque command value through the command converter 7. In the speed controller 8, the rotor speed estimated value (electric speed estimated value), which is one of the output signals from the phase speed estimator 10, is sent through the coefficient unit 11 as the inverse of the pole pair number Np, which is a constant value. It is sent after being multiplied and converted to a machine speed estimate. In the present example of FIG. 3, an example in which a speed control system is configured is shown, and thus a torque command value is obtained as an output of the speed controller 8. As is well known to those skilled in the art, the speed controller 8 is unnecessary when the control purpose is torque control and the speed control system is not configured. In this case, the torque command value is directly applied from the outside.

The core of the present invention resides in the phase velocity estimator 10 which is substantially synonymous with the rotor phase estimation device. In any of the speed control and the torque control, the phase speed estimator 10 does not require any change. Even when the motor to be driven is another synchronous motor, the phase speed estimator 10 does not require any change. In the following, an embodiment of the phase speed estimator 10 will be described without losing generality with respect to control modes such as speed control and torque control, and without losing generality with respect to the synchronous motor to be driven.

FIG. 4 shows an embodiment of the phase velocity estimator 10. The phase velocity estimator 10 basically includes three devices: a high-frequency voltage command device (displayed as HFVC) 10-1, a high-frequency product signal generator 10-2, and a high-frequency phase synchronizer 10-3.

(35)式の高周波電圧指令値生成では、γ軸速度に代わって、回転子電気速度推定値を利用している。図４に明示しているように、電気速度推定値はγ軸速度と本質的に等価であるが、必要に応じ、γ軸速度をローパスフィルタ処理して電気速度推定値を得るようにしている。なお、このときのローパスフィルタは、通常は、簡単な１次フィルタでよい。 The high-frequency voltage command device 10-1 generates a high-frequency voltage command value having a sine shape on a biaxial orthogonal γδ quasi-synchronous coordinate system having the rotor phase estimation value as a base axis (γ-axis) phase. As shown in FIG. 3, the high-frequency voltage command device 10-1 performs high-frequency voltage application together with the devices 5b, 4b, and 2 in the drive control device. As the high-frequency voltage command value generated by the high-frequency voltage command device, at least the following values based on the equations (1) to (3) are conceivable.

In the generation of the high-frequency voltage command value of equation (35), the estimated rotor electric speed is used instead of the γ-axis speed. As clearly shown in FIG. 4, the electrical speed estimation value is essentially equivalent to the γ-axis speed, but the γ-axis speed is low-pass filtered as necessary to obtain the electrical speed estimation value. . Note that the low-pass filter at this time may normally be a simple primary filter.

すなわち、固定子電流のδ軸要素から駆動用電流指令値のδ軸要素を減じ、実質的に高周波電流のδ軸要素を得て、高周波電流δ軸要素と印加高周波電圧指令値のγ軸要素に対して−π/2(rad)の位相差を有するキャリア信号sinω_ｈｔとの（スカラ）乗算処理を通じて、高周波積信号を生成している。本高周波信号の生成は、(38)式の第２式が示しているように、印加高周波電圧のγ軸要素に対して−π/2(rad)の位相差を有する信号をδ軸要素とするキャリア和信号（ベクトル）と高周波電流（ベクトル）の内積処理と等価である。高周波電圧指令値が(35)、(36)式のようにδ軸要素をもつ場合には、キャリア信号を本δ軸要素から直接生成するようにすると、演算量を低減できる。図４はこの例を示している。すなわち、高周波積信号生成器は、実質的にスカラ乗算器のみで実現できる。なお、(37)式のようなδ軸要素を持たない高周波電圧指令値を利用する場合には、高周波電圧指令値のγ軸要素に所定の位相差を付与して、キャリア信号を生成することになる。 The high-frequency voltage command value is superimposed on the driving voltage command value, and the high-frequency voltage is applied to the synchronous motor via the power converter, so that a high-frequency current flows. The high frequency current is included in this as a high frequency component of the stator current. The high-frequency product signal generator 10-2 that plays a role of generating a high-frequency product signal that can contain a high-frequency component faithfully realizes the final expression of the following expression in the embodiment of FIG.

That is, the δ-axis element of the driving current command value is subtracted from the δ-axis element of the stator current to substantially obtain the δ-axis element of the high-frequency current, and the γ-axis element of the high-frequency current δ-axis element and the applied high-frequency voltage command value through (scalar) multiplication process of the carrier signal sin .omega _{h t} having a phase difference of - [pi] / 2 (rad) with respect to, and generates a high-frequency product signal. This high-frequency signal is generated by using a signal having a phase difference of −π / 2 (rad) with respect to the γ-axis element of the applied high-frequency voltage as the δ-axis element, as shown in the second equation of equation (38). This is equivalent to the inner product processing of the carrier sum signal (vector) and the high-frequency current (vector). When the high-frequency voltage command value has a δ-axis element as in equations (35) and (36), the amount of calculation can be reduced by generating the carrier signal directly from the δ-axis element. FIG. 4 shows this example. That is, the high-frequency product signal generator can be realized substantially only by a scalar multiplier. When using a high-frequency voltage command value that does not have a δ-axis element as in equation (37), a predetermined phase difference is given to the γ-axis element of the high-frequency voltage command value to generate a carrier signal. become.

(39)式におけるＦ_ｄｃ(s)はＦ_ｄｃ（０）＝０の性質をもつ、直流成分のみの除去を目的とした広帯域幅をもつ直流成分除去フィルタである。(39)式は広帯域の直流成分除去フィルタのPLL推定系内への導入を意味するが、この種のフィルタの導入は、高周波位相同期器の設計に実質影響を与えない。換言するならば、この導入を無視して、PLL推定系を安定化するための高周波位相同期器を設計できる。 The high frequency product signal generator may be configured to generate the next high frequency product signal, instead of the equation (38).

F _dc (s) in the equation (39) is a DC component removal filter having a property of F _dc (0) = 0 and having a wide bandwidth for the purpose of removing only the DC component. Equation (39) means the introduction of a broadband DC component removal filter into the PLL estimation system, but the introduction of this type of filter does not substantially affect the design of the high-frequency phase synchronizer. In other words, this introduction can be ignored and a high-frequency phase synchronizer for stabilizing the PLL estimation system can be designed.

とを出力している。このときの高周波位相制御器Ｃ（Ｓ）の係数（有理関数の係数）は、既に説明した設計法に従い、PLL推定系が安定に動作するように設計されている。 The high-frequency phase synchronizer realizes a high-frequency PLL means, and in this example, is realized according to the expression (17). That is, the high frequency product signal is obtained as an input, and the speed ω _γ and the rotor phase estimation value of the γδ quasi-synchronous coordinate system

Is output. The coefficient (rational function coefficient) of the high-frequency phase controller C (S) at this time is designed so that the PLL estimation system operates stably in accordance with the design method already described.

Next, an example of a numerical experiment carried out in order to confirm the effects and usefulness of the present invention will be shown. The configuration of the drive control system is the same as that of the embodiment shown in FIGS. The high frequency voltage command device 10-1 was configured according to the equation (35) based on the generalized elliptical high frequency voltage method. At this time, in order to confirm the reduction characteristic of the high-frequency residual disturbance by the high-frequency phase synchronizer, the elliptic parameter K = 1 at which the maximum high-frequency residual disturbance occurs was selected. Table 1 shows an overview of the specifications of the test motor.

この場合、(21)式の第１式で定義した信号係数Ｋ_θは次の値をとる。

高周波位相制御器Ｃ（ｓ）の係数（有理関数の係数と同一）の設計は、PLL推定系の帯域幅が概ね150(rad/s)となるように、行うものとする。 The amplitude and frequency of the applied high-frequency voltage are as follows.

In this case, (21) the signal coefficient K _theta defined by the first equation of the following values.

The coefficient of the high-frequency phase controller C (s) (same as the coefficient of the rational function) is designed so that the bandwidth of the PLL estimation system is approximately 150 (rad / s).

また、(24)式に定めた高周波残留外乱低減率に関し、以下を得る。

上の高周波位相制御器係数は、試行錯誤的には選定が難しい、桁数の大きく異なる幅のある数値を示している。また、高周波残留外乱低減率は、実用上十分に小さい値を示している。 It is assumed that the high-frequency phase controller uses a quadratic one of the equation (22), and that the rational function uses a cubic one. The denominator polynomial of the transfer characteristic of the PLL estimation system is determined to have a stable triple root. In order to have the desired bandwidth, the root p _{0 at} this time is approximately p _o = −88. When the vibration coefficient _{Kh is set} to 1 and the high frequency phase controller coefficient is calculated using the equation (23), the following is immediately obtained.

Further, the following is obtained with respect to the high-frequency residual disturbance reduction rate defined in the equation (24).

The upper high-frequency phase controller coefficient indicates a numerical value with a large difference in the number of digits, which is difficult to select by trial and error. Further, the high-frequency residual disturbance reduction rate is a sufficiently small value for practical use.

(rad)として、回転子位相推定動作を開始させた。数値実験結果を図5に示す。図5(a)は、回転子位相推定の様子を示したものであり、上から、回転子位相真値θ_α、同推定値

、位相偏差

を示している。位相偏差の軸スケールは、位相真値、同推定値に比較し、5倍大きくしている。同図(b)は、これに対応した速度を示したものであり、上から、回転子の電気速度真値ω_２ｎ、座標系速度ω_γ、座標系速度を帯域幅150(rad/s)の1次ローパスフィルタで処理して得た速度推定値

である。同図(c)は、時刻1(s)近傍の定常状態における高周波積信号、すなわち高周波位相制御器への入力信号（高周波位相同期器への入力信号）ｕ_ＰＬＬである。図(a)より、回転子位相は約0.1(s)後には正しく推定されていることが確認される。一方、座標系速度には高い振幅(約9(rad/s))の高周波成分が出現しており、速度推定値として利用するには、追加的なフィルタが不可欠であることが確認される。図(c)より、高周波位相制御器への入力信号（高周波積信号）は回転子位相推定値が同真値へ実質的に収斂した後も高周波残留外乱を有していることを示している。しかしながら、この高周波残留外乱は、回転子位相推定値に対しては、(44)式の低減率で低減されており、回転子位相推定値上には実質的には出現していない。これら応答は、すでに解説した「本発明の効果」を裏付けるものである。 The permanent magnet synchronous motor is rotating at a constant speed at an electric speed of 30 (rad / s), and the PLL estimation system is the rotor phase true value θ _α = π / 6. The initial value of the rotor phase estimate is

The rotor phase estimation operation was started as (rad). The numerical experiment results are shown in FIG. Fig. 5 (a) shows the state of the rotor phase estimation. From the top, the rotor phase true value θ _α and the estimated value are shown.

, Phase deviation

Is shown. The axis scale of the phase deviation is 5 times larger than the true phase value and the estimated value. FIG. 4B shows the speed corresponding to this. From the top, the rotor electrical speed true value ω _2n , the coordinate system speed ω _γ , and the coordinate system speed are set to a bandwidth 150 (rad / s). Estimated velocity obtained by processing with the first-order low-pass filter

It is. FIG. 4C shows a high-frequency product signal in a steady state near time 1 (s), that is, an input signal to the high-frequency phase controller (input signal to the high-frequency phase synchronizer) u _PLL . From Fig. (A), it is confirmed that the rotor phase is correctly estimated after about 0.1 (s). On the other hand, a high-frequency component having a high amplitude (about 9 (rad / s)) appears in the coordinate system velocity, and it is confirmed that an additional filter is indispensable for use as a velocity estimation value. Figure (c) shows that the input signal (high-frequency product signal) to the high-frequency phase controller has high-frequency residual disturbance even after the rotor phase estimate has substantially converged to the same true value. . However, this high frequency residual disturbance is reduced with respect to the rotor phase estimated value by the reduction rate of the equation (44), and does not substantially appear on the rotor phase estimated value. These responses support the “effect of the present invention” already described.

また、(27)式に定めた高周波残留外乱低減率に関し、以下を得る。

上の高周波位相制御器係数は、試行錯誤的には選定が難しい桁数の大きく異なる幅のある数値を示している。また、高周波残留外乱低減率は、実用上十分に小さい値を示している。 It is assumed that the high-frequency phase controller uses the third order of the equation (25), and the rational function uses the fourth order. The denominator polynomial of the transfer characteristic of the PLL estimation system is determined to have a stable quadruple root. In order to have the desired bandwidth, the root p _{o at} this time is approximately p _o = −115. When the vibration coefficient _{Kh is set} to 2 and the high frequency phase controller coefficient is calculated using the equation (26), the following is immediately obtained.

Further, the following is obtained with respect to the high frequency residual disturbance reduction rate defined in the equation (27).

The upper high-frequency phase controller coefficient indicates a numerical value with a large difference in the number of digits that is difficult to select by trial and error. Further, the high-frequency residual disturbance reduction rate is a sufficiently small value for practical use.

から、約0.3(s)後には回転子位相は適切に推定されていることが確認される。また、座標系速度ω_γには、若干の高周波残留外乱が出現しているが(本図では、必ずしも明瞭でない)、その振幅は実用上の許容範囲内に収まっている。座標系速度とこのフィルタ処理後の信号である

との間には、大きな違いはない。なお、本数値実験における、定常状態での入力信号（高周波積信号）は図5(c)と同様であるので、この表示は避けた。これら応答は、すでに解説した「本発明の効果」を裏付けるものである。 The experiment was performed under the same conditions as in FIG. The result of the numerical experiment is shown in FIG. The meaning of the waveform in FIG. 6 is the same as that in FIG. Phase deviation in figure (a)

Thus, it is confirmed that the rotor phase is properly estimated after about 0.3 (s). In addition, a slight high-frequency residual disturbance appears in the coordinate system speed ω _γ (not necessarily clear in this figure), but the amplitude is within a practical allowable range. The coordinate system speed and the signal after this filter processing

There is no big difference between In this numerical experiment, the input signal (high-frequency product signal) in a steady state is the same as that shown in FIG. These responses support the “effect of the present invention” already described.

上の高周波位相制御器係数は、試行錯誤的には選定が難しい桁数の大きく異なる幅のある数値を示している。本高周波位相制御器による高周波残留外乱低減率は、(30)式に示したように、ゼロである。 It is assumed that the high-frequency phase controller uses the third order of equation (28), and the rational function uses the fourth order. The denominator polynomial of the transfer characteristic of the PLL estimation system is determined to have a stable quadruple root. To have the desired bandwidth, root root p _o at this time is generally a p _{o =} -100. When the vibration coefficient _{Kh is set} to 2 and the high frequency phase controller coefficient is calculated using the equation (29), the following is immediately obtained.

The upper high-frequency phase controller coefficient indicates a numerical value with a large difference in the number of digits that is difficult to select by trial and error. The high frequency residual disturbance reduction rate by this high frequency phase controller is zero as shown in the equation (30).

から、約0.3(s)後には回転子位相は適切に推定されていることが確認される。また、座標系速度ω_γには、高周波残留外乱が実質的に出現していないことも確認される。すなわち、(30)式の性質が確認される。高周波残留外乱の回転子推定値への出現有無を除けば、本高周波位相制御器による応答は、図6に示した高周波位相制御器による応答と、過渡応答においても、概ね同様である。これら応答は、すでに解説した「本発明の効果」を裏付けるものである。 Experiments were performed under the same conditions as in FIGS. The result of the numerical experiment is shown in FIG. The meaning of the waveform in FIG. 7 is the same as that in FIG. Phase deviation in figure (a)

Thus, it is confirmed that the rotor phase is properly estimated after about 0.3 (s). It is also confirmed that the high frequency residual disturbance does not substantially appear in the coordinate system speed ω _γ . That is, the property of equation (30) is confirmed. Except for the presence or absence of high-frequency residual disturbance in the estimated rotor value, the response by the high-frequency phase controller is substantially the same in the response by the high-frequency phase controller shown in FIG. 6 and the transient response. These responses support the “effect of the present invention” already described.

Subsequently, a second embodiment will be shown. FIG. 8 shows an embodiment of the phase velocity estimator 10. The high frequency voltage command device 10-1 and the high frequency phase synchronizer 10-3 are the same as those in the embodiment of FIG. The difference between FIG. 8 and FIG. 5 resides in the high-frequency product signal generator 10-2. First, a direct current component removal filter F _dc (s) is used to remove a drive current that is a direct current component from the stator current to obtain a high frequency current. Next, the inner product calculation of the high frequency current and the carrier reverse phase signal is performed to generate a high frequency product signal (input signal of the high frequency phase controller). At this time, the δ-axis element of the carrier reverse-phase signal has a phase difference of −π / 2 (rad) with respect to the γ-axis element of the high-frequency voltage command value, as shown in the second expression of equation (10). Have. The above inner product operation can also be expressed as follows using mathematical expressions.

As a signal used for multiplication processing (inner product processing) with a carrier reverse phase signal, the following equation may be used instead of equation (48).

In the case where the high-frequency voltage command value is a vector signal as expressed by equations (35) and (36), the carrier reverse phase signal can be obtained directly from these γ-axis and δ-axis elements. FIG. 8 shows this example, and the high-frequency product signal generator can be realized substantially only by the inner product. In addition, when using a high-frequency voltage command value having only a γ-axis element as in equation (37), the carrier reverse phase signal is generated by giving a predetermined phase difference to the γ-axis element of the high-frequency voltage command value. Will do.

The above is an embodiment using a carrier reverse phase signal, but the same applies to the case where a carrier positive phase signal is used, as can be easily understood by those skilled in the art.

Subsequently, one example (third example) using the invention according to claims 1 to 3 is shown. The typical configuration of the motor drive control system is substantially the same as that shown in FIG. A device configured based on the inventions of claims 1 to 3 is a phase velocity estimator 10. FIG. 8 shows an embodiment of the phase velocity estimator 10 using the inventions according to claims 1 to 3. This is composed of four devices: a high-frequency voltage command device (indicated as HFVC) 10-1, an anti-phase component extractor 10-4, a normalizer 10-5, and a final phase generator 10-6.

The high-frequency voltage command device 10-1 of the present embodiment generates a high-frequency voltage command value that draws a perfect circular locus based on the equation (35) or the equation (36), etc., with K = 1 as a condition. The negative phase component extractor 10-4 plays a role of extracting the negative phase component of the high frequency current from the stator current. The anti-phase component at this time is a component in the order of anti-phase with the applied high-frequency signal. For extraction of the reverse phase component, a D-factor filter described in Non-Patent Document 7 may be used. The extracted antiphase component is sent to the normalizer 10-5. The normalizer normalizes the anti-phase component and outputs the normalized anti-phase component to the final phase generator 10-6.

このときの積信号は、(51)式が示しているように、高周波成分を含有しない、直流的な成分である。最終位相生成器では、本積信号を位相同期器で処理して、回転子位相推定値とγδ準同期座標系の速度を出力している。直流的な位相偏差を入力として、回転子位相推定値等を生成出力する役割を担っている位相同期器は、通常の代表的な直流PLL法に基づき構成すればよい。この種の位相同期器の構成法に関しては、非特許文献７等に詳説されており、当業者には公知であるので、これ以上の説明は省略する。位相同期器の出力の１つである座標系速度は、必要に応じてローパスフィルタで処理して、回転子速度推定値として利用される。この点は、図５、８の実施例と同様であるので、これ以上の説明は、省略する。 The final phase generator in the present embodiment first performs the inner product processing shown in the equation (34) to generate a product signal of the following equation.

The product signal at this time is a direct current component that does not contain a high frequency component, as shown in equation (51). In the final phase generator, the product signal is processed by the phase synchronizer, and the rotor phase estimation value and the speed of the γδ quasi-synchronous coordinate system are output. A phase synchronizer having a role of generating and outputting a rotor phase estimation value or the like with a DC phase deviation as an input may be configured based on a typical typical DC PLL method. The configuration method of this type of phase synchronizer is described in detail in Non-Patent Document 7 and the like, and is well known to those skilled in the art, so further explanation is omitted. The coordinate system speed, which is one of the outputs of the phase synchronizer, is processed by a low-pass filter as necessary and used as a rotor speed estimation value. Since this point is the same as the embodiment of FIGS. 5 and 8, further description is omitted.

The embodiment of FIG. 8 is an example in which the phase velocity estimator is configured on a γδ quasi-synchronous coordinate system. The phase velocity estimator can also be configured on an αβ fixed coordinate system. The configuration of the phase velocity estimator according to the present invention on the αβ fixed coordinate system should be noted in the generation of the product signal. The points to be noted in generating the product signal are the same as in Non-Patent Documents 1 and 2. Since those skilled in the art can easily understand the points to be noted from these documents, further explanation will be omitted.

In the example of FIG. 8, the applied high frequency signal and the response high frequency signal are a high frequency voltage and a high frequency current, respectively. A person skilled in the art can easily understand the configuration when the applied high-frequency signal and the response high-frequency signal are a high-frequency current and a high-frequency voltage, respectively, and will not be described further.

In the embodiment of FIG. 8, the final phase generator is configured to use a product signal of a DC component and a phase synchronizer. However, the present invention does not limit the configuration of the final phase generator. Please point out.

The present invention is suitable for applications that require generation of high torque in a low speed range including zero speed, among applications in which the motor is sensorlessly driven for synchronization.

DESCRIPTION OF SYMBOLS 1 Synchronous motor 2 Power converter 3 Current detector 4a 3 phase 2 phase converter 4b 2 phase 3 phase converter 5a Vector rotator 5b Vector rotator 6 Current controller 7 Command converter 8 Speed controller 9 Band stop filter 10 Phase velocity estimator 10-1 High frequency voltage command device 10-2 High frequency product signal generator 10-3 High frequency phase synchronizer 10-4 Reverse phase component extractor 10-5 Normalizer 10-6 Final phase generator 11 Coefficient unit 12 Cosine sine signal generator

Claims (3)

Detecting and processing response high-frequency signals that are used in drive control devices for synchronous motors where the rotor shows salient pole characteristics when a high-frequency signal having a frequency higher than the drive fundamental frequency is applied, and that is a response to the applied high-frequency signal A rotor phase estimation device that generates and outputs a rotor phase estimation value through
Means for generating a high-frequency signal command value that is included in the final signal command value to the power converter in the drive control device for high-frequency signal application and that draws a perfect circle locus;
Means for extracting a negative phase component that is out of phase with respect to the applied high frequency signal from the response high frequency signal resulting from the high frequency signal command value;
Means for capturing the extracted antiphase component as a vector, normalizing the norm to a value represented by 1, and generating a normalized antiphase component;
Means for generating a carrier signal having a reverse phase order with respect to the high-frequency signal command value, processing the normalized anti-phase component using the generated carrier signal, and generating and outputting a rotor phase estimation value;
A rotor phase estimation apparatus comprising: The rotor phase estimation apparatus according to claim 1, wherein the generated high-frequency signal command value is a high-frequency voltage command value, and the response high-frequency signal resulting therefrom is a high-frequency current. The normalized antiphase component processing using the carrier signal is performed on a biaxial orthogonal γδ quasi-synchronous coordinate system having a rotor phase estimation value as a base axis (γ axis) phase. 1. The rotor phase estimation device according to 1.

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