JP5716617B2 - Signal processing circuit, signal processing method, and reception system - Google Patents

Description

  The present disclosure relates generally to a signal processing circuit, and more particularly to a signal processing circuit used in a receiving apparatus.

  In a receiver of a communication system, an interference wave signal (spurious signal) other than a transmission signal may be mixed in a received signal. The spurious signal may be a signal leaked from another device or a signal leaked from another signal source (for example, a clock signal system) of the receiver itself. Since the quality of the received signal deteriorates due to spurious, it is necessary to reduce the effect of spurious.

  Spurious is often an unmodulated wave or a narrow-band modulated wave. Therefore, if the spurious frequency position can be specified, the spurious can be reduced by suppressing the signal component at the specific frequency position. However, when the amplitude of the spurious is small, there is a problem that it is difficult to find the spurious and specify its frequency position.

  For example, there is a method of specifying a spurious frequency position by locking a PLL (Phase Locked Loop) circuit to an unmodulated or narrow-band modulated wave component in a received signal. In this method, when the spurious amplitude decreases, the difference between the loop filter output value in the PLL at the spurious frequency position and the loop filter output value at other frequency positions becomes small, and the spurious frequency position is specified. It becomes impossible. If the band of the loop filter is made sufficiently narrow, the PLL can be locked and the spurious frequency position can be specified, but there is a problem that the time required for pulling (locking operation) becomes long. Also, since the frequency range that the PLL can pull in becomes narrower, it is necessary to narrow the sweep step width when finding the spurious frequency position by sweeping the entire frequency region while changing the oscillation frequency of the PLL. Again, it will take longer. Similarly, regarding the loop filter used in the feedback loop for sweeping, it is impossible to determine whether or not the spurious frequency position can be detected unless the band is sufficiently narrowed. Further, if the band is sufficiently narrow, it takes time to determine whether or not the spurious frequency position can be detected, and there is a problem that the sweep operation becomes slow.

JP 2006-186421 A JP 2009-194481 A

  In view of the above, a signal processing circuit capable of accurately identifying the frequency position of spurious and reducing spurious is desired.

  The signal processing circuit includes an FFT unit that performs an FFT on an OFDM reception signal, a propagation channel estimation unit that obtains a propagation channel estimation value based on a pilot signal in reception data output from the FFT unit, and a propagation channel estimation value Based on the received signal output from the FFT unit, and the error vector obtained based on the data signal point in the received data output from the propagation channel compensation unit is saturated. If not, based on the error vector, if the magnitude of the error vector is saturated, based on the propagation path estimated value, a spurious frequency detection unit that detects a spurious frequency position, and And a spurious reduction unit that reduces the frequency component of the received signal at the spurious frequency position detected by the spurious frequency detection unit. .

  In the signal processing method, an OFDM reception signal is subjected to FFT, a propagation path estimation value is obtained based on a pilot signal in the reception data after the FFT, and the FFT reception signal is calculated based on the propagation path estimation value. If the error vector obtained based on the data signal point in the received data after the propagation path compensation is not saturated, the magnitude of the error vector is saturated based on the error vector. If it is, spurious frequency position is detected based on the propagation path estimated value, and each step of reducing the frequency component of the received signal at the detected spurious frequency position is included. .

  The reception system includes a tuner, a demodulation circuit that receives an output signal of the tuner and outputs a digital signal, a decoder that receives the digital signal and generates an output signal, a control circuit that controls the demodulation circuit and the decoder, An output device that outputs the output signal as at least one of video and audio, wherein the demodulation circuit performs FFT on an OFDM reception signal that is an output signal of the tuner, and reception output from the FFT unit A propagation path estimator that obtains a propagation path estimation value based on a pilot signal in the data; a propagation path compensation section that performs propagation path compensation on the received data output from the FFT section based on the propagation path estimation value; When the error vector obtained based on the data signal point in the received data output from the propagation path compensation unit is not saturated Is based on the error vector, and when the magnitude of the error vector is saturated, a spurious frequency detection unit that detects a spurious frequency position based on the propagation path estimation value, and the spurious frequency detection unit includes: And a spurious reduction unit that reduces the frequency component of the received signal at the detected spurious frequency position.

  According to at least one embodiment of the present disclosure, spurious can be reduced by accurately identifying the frequency position of the spurious.

It is a figure which shows an example of a structure of the signal processing circuit used in a receiver. It is a figure which shows the constellation when there is no propagation path fluctuation | variation and noise. It is a figure which shows the constellation in case there exists a spurious. It is a figure which shows propagation path estimated value distribution in case an unmodulated spurious with a small amplitude exists. It is a figure which shows propagation path estimated value distribution in case an unmodulated spurious with a large amplitude exists. It is a figure for demonstrating an error vector. It is a figure which shows distribution of the magnitude | size of an error vector in case the non-modulation spurious with small amplitude exists. It is a figure which shows distribution of the magnitude | size of an error vector in case an unmodulated spurious with a large amplitude exists. It is a figure which shows an example of a structure of the rotor of FIG. It is a figure which shows an example of a structure of a loop filter. It is a figure which shows the modification of a structure of a signal processing circuit. It is a figure which shows an example of a structure of the receiving system of OFDM.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.

  FIG. 1 is a diagram illustrating an example of a configuration of a signal processing circuit used in a receiving apparatus. The signal processing circuit 10 is a circuit that performs signal processing on a received signal of an orthogonal frequency division multiplexing (OFDM). The signal processing circuit 10 includes an RF circuit 11, an AD converter (ADC) 12, a PLL unit 13, a DC cut unit 14, an inverse rotor 15, an FFT unit 16, a propagation channel compensation unit 17, and a propagation channel estimation unit 18. The signal processing circuit 10 further includes a propagation path estimated value distribution measurement unit 19, a data subcarrier EVM measurement unit 20, and a spurious frequency detection unit 21.

  A signal received by the antenna 5 is input to the RF circuit 11. The RF circuit 11 converts the received signal from the antenna 5 into a baseband signal and outputs it. The AD converter 12 converts the baseband reception signal output from the RF circuit 11 from an analog signal to a digital signal. The PLL unit 13, the DC cut unit 14, and the reverse rotor 15 reduce spurious components in the digital reception signal output from the AD converter 12, and output a digital reception signal with reduced spurious components. The FFT unit 13 performs FFT (Fast Fourier Transform) on the OFDM digital reception signal and converts the time domain signal into a frequency domain signal. As a result, the orthogonal frequency division multiplexed signal is demodulated to obtain data for a plurality of subcarriers. The output of the FFT unit 13 includes a data signal and a pilot signal. The pilot signal is BSPK modulated, and the data signal is modulated by any one of QPSK, 16QAM, and 64QAM.

  The propagation path estimation unit 18 obtains a transmission path characteristic value for equalizing the transmission path characteristic with respect to the received data based on the reception data of the pilot signal output from the FFT section 16. The pilot signal is inserted between the data signals at regular intervals of the subcarriers when assigning the data signals to the plurality of subcarriers on the transmission side. That is, when a plurality of subcarriers arranged in the frequency axis direction are viewed in order, a predetermined number of consecutive subcarriers of the data signal and one subcarrier of the pilot signal are alternately arranged. The propagation path estimation unit 18 calculates the transmission path characteristic value at the position by dividing the received data of the subcarriers of the pilot signal by the original pilot signal value (the value of the known pilot signal inserted on the transmission side). To do. The propagation path estimation unit 18 further estimates the transmission path characteristic value at the subcarrier position of the data signal by interpolation based on the transmission path characteristic value at the subcarrier position of the pilot signal thus obtained, and transmits the transmission path characteristic at each subcarrier position. Obtain an estimate of the characteristic value. The channel compensator 17 equalizes the channel characteristics by dividing the value of the received data at the corresponding subcarrier position output from the FFT unit 16 by the estimated channel characteristic value at each subcarrier position. To do.

  The received data whose transmission path characteristics are equalized by the propagation path compensation unit 17 may be soft-decided by, for example, demapping processing and further corrected by error correction processing using Viterbi decoding processing or the like. The data after error correction may be supplied to a circuit such as a decoder, for example.

  As described above, the propagation path estimation unit 18 obtains a propagation path estimation value based on the pilot signal in the reception data output from the FFT unit 16. Further, the propagation path compensation unit 17 performs propagation path compensation on the reception data output from the FFT unit 16 based on the propagation path estimation value. In such a configuration, the signal processing circuit 10 detects the spurious frequency by the spurious frequency detector 21 in order to reduce spurious in the received signal. Specifically, the spurious frequency detection unit 21 determines that the error vector obtained based on the data signal point in the reception data output from the propagation path compensation unit 17 is not saturated based on the error vector. Detect spurious frequency position. The spurious frequency detection unit 21 detects the spurious frequency position based on the propagation path estimation value from the propagation path estimation unit 18 when the magnitude of the error vector is saturated. As will be described later, when the spurious amplitude is large, the error vector is saturated, and when the spurious amplitude is small, the error vector is not saturated. Therefore, the spurious frequency detection unit 21 detects the spurious frequency position based on the error vector when the spurious amplitude is small, and detects the spurious frequency based on the propagation path estimated value when the spurious amplitude is large. Detect position.

  The PLL unit 13 and the DC cut unit 14 function as a spurious reduction unit, and reduce the frequency component of the received signal at the spurious frequency position detected by the spurious frequency detection unit 21. The PLL unit 13 includes a rotor 25, an NCO (Numerically Controlled Oscillator) 26, and a loop filter 27. The DC cut unit 14 includes a loop filter 28 and a subtractor 29.

  The PLL unit 13 locks to a non-modulated wave or a narrow-band modulated wave signal in the vicinity of the detected spurious frequency position. Specifically, the NCO 26 first oscillates with the spurious frequency position detected by the spurious frequency detector 21 as the initial position. The rotor 25 shifts the entire frequency of the received signal so that the frequency position where the NCO 26 oscillates becomes a DC component. The loop filter 27 integrates the DC component of the received signal after this shift, and feeds back the integrated value to the NCO 26. The NCO 26 locks to an unmodulated wave or a narrow-band modulated wave (spurious) near the initial position by adjusting the oscillation frequency based on the feedback value. As a result, the rotor 25 shifts the entire frequency of the received signal so that the locked frequency becomes a DC component. The DC cut unit 14 reduces the signal component of the direct current component in the received signal after the shift. Specifically, the loop filter 28 integrates the direct current component of the shifted received signal and supplies the integrated value to the subtractor 29. The subtractor 29 reduces or removes the DC component in the shifted received signal by subtracting the integral value from the shifted received signal. Thereby, spurious in the received signal can be reduced or eliminated.

  The reverse rotor 15 returns the original frequency position of the received signal by shifting the entire frequency of the received signal after the shift in the reverse direction based on the frequency signal oscillated by the NCO 26. The rotor 25 and the reverse rotor 15 are basically mixers that multiply the target signal by the frequency signal oscillated by the NCO 26, and the frequency is shifted by this multiplication process.

  FIG. 2 is a diagram showing a constellation when there is no propagation path fluctuation or noise. In FIG. 2, the horizontal axis indicates the real number component (I component: in-phase component), and the vertical axis indicates the imaginary number component (Q component: quadrature component). A plurality of points 31 arranged on the complex plane shown in FIG. 2 indicate ideal data signal point positions. Although the data signal points 31 are originally arranged in a matrix by the same number in the vertical and horizontal directions, FIG. 2 shows the data positions in the simulation, and no data exists at positions where the data signal points are missing. The signal point 30 is the signal point position of the pilot signal. In this example, since the pilot signal is BPSK modulated, it is located on the leftmost side on the I axis.

  FIG. 3 is a diagram showing a constellation when there is a spurious. FIG. 3 shows data signal points of received data in a state where spurious values obtained by simulation are mixed. FIG. 3A shows a data signal point position of received data in a subcarrier having a frequency close to the spurious frequency. FIG. 3B shows a data signal point position of received data in a subcarrier having a frequency far from the spurious frequency. When the spurious is an unmodulated wave or a narrow-band modulated wave, the plot of the received data draws a circle around the ideal signal point as shown in FIGS. 3 (a) and 3 (b). The closer the subcarrier frequency is to the spurious frequency, the larger the radius of the circle, and the farther the subcarrier frequency is from the spurious frequency, the smaller the radius of the circle. Therefore, the spurious frequency position can be specified based on the size of the circle. That is, when the circle radii are compared between subcarriers, the frequency of the subcarrier having the largest circle radius may be detected as the spurious frequency position.

  In the signal processing circuit 10 shown in FIG. 1, the propagation path estimated value distribution measuring unit 19 obtains a first distribution in the frequency domain of the circle size obtained based on the pilot signal. In addition, the data subcarrier EVM measurement unit 20 obtains a second distribution in the frequency region of the circle size obtained based on the data signal (not including the pilot signal). The spurious frequency detection unit 21 detects the spurious frequency position based on at least one of the first distribution and the second distribution. Hereinafter, processing in the propagation path estimated value distribution measurement unit 19, the data subcarrier EVM measurement unit 20, and the spurious frequency detection unit 21 will be described in detail.

Let t be the symbol time and i be the subcarrier number. If the pilot signal which is a known signal is P (t, i) and the received data after FFT is R (t, i), the propagation path estimation value H (t, i) can be expressed by the following equation.
H (t, i) = R (t, i) / P (t, i)
However, t and i are variables that are valid only in the symbol time and the subcarrier number in which the pilot signal is inserted.

  By calculating the average value in the symbol time direction of this propagation path estimation value H (t, i) for each subcarrier, the distribution in the frequency domain of the propagation path estimation value (the vertical axis is the frequency of subcarriers on the horizontal axis) Distribution obtained by averaging the propagation path estimated values). Here, the calculation for obtaining the average value may be a moving average within a predetermined time frame, or may be a calculation using an IIR (Infinite Impulse Response) filter. For example, when the moving average is used, the propagation path estimated value distribution Hpa (t, i) is represented by the following expression.

ここでｔｎは移動平均を求める時間枠の幅である。スプリアスが影響しないサブキャリア位置においてＨｐａ（ｔ，ｉ）は１に近い値となるが、スプリアスが影響するサブキャリア位置においてＨｐａ（ｔ，ｉ）は１より大きな値となる。スプリアスの周波数にサブキャリアの周波数が近いほど、伝播路推定値分布Ｈｐａ（ｔ，ｉ）の値は大きくなる。従って、この伝播路推定値分布Ｈｐａ（ｔ，ｉ）のピーク位置に基づいて、スプリアスの周波数位置を特定することができる。

Here, tn is the width of the time frame for obtaining the moving average. Hpa (t, i) has a value close to 1 at a subcarrier position where spurious does not affect, but Hpa (t, i) has a value larger than 1 at a subcarrier position where spurious affects. The closer the subcarrier frequency is to the spurious frequency, the larger the value of the propagation path estimated value distribution Hpa (t, i). Therefore, the spurious frequency position can be specified based on the peak position of the propagation path estimated value distribution Hpa (t, i).

  Since the original value of the pilot signal is known and the position of the subcarrier of the pilot signal is also known, the propagation path estimated value distribution Hpa (t, i) obtained based on the known information is the amplitude of the spurious. Regardless of the size, it has a peak at the spurious frequency. However, as a result of the pilot signals being normally arranged at intervals in the frequency direction, the spurious frequency detection resolution is also equal to the frequency direction interval. That is, the spurious frequency cannot be specified with a resolution equal to the subcarrier interval.

  FIG. 4 is a diagram showing a propagation path estimated value distribution when there is an unmodulated spurious signal having a small amplitude. In FIG. 4, the horizontal axis indicates the subcarrier number, and the vertical axis indicates the value of the propagation path estimated value distribution Hpa (t, i). Since the subcarrier frequency increases as the subcarrier number increases, the horizontal axis may be considered as the subcarrier frequency. From the propagation path estimated value distribution in FIG. 4, it can be seen that there is a spurious frequency position near the data subcarrier number 1800, and the spurious frequency can be specified. However, as a result of the pilot signal being arranged for every predetermined number of subcarriers (every 12 subcarriers in this example), the spurious frequency can be specified only with the resolution of the predetermined number of subcarrier intervals.

  FIG. 5 is a diagram showing a propagation path estimated value distribution when there is an unmodulated spurious signal having a large amplitude. In FIG. 5, the horizontal axis indicates the subcarrier number, and the vertical axis indicates the value of the propagation path estimated value distribution Hpa (t, i). From the propagation path estimated value distribution of FIG. 5, it can be seen that a spurious frequency position exists in the vicinity of the data subcarrier number 1800, and the spurious frequency can be specified. However, as a result of the pilot signal being arranged for every predetermined number of subcarriers (every 12 subcarriers in this example), the spurious frequency can be specified only with the resolution of the predetermined number of subcarrier intervals.

  Returning to FIG. 1, the propagation path estimated value distribution measuring unit 19 takes the average value in the time symbol direction of the propagation path estimated value at each subcarrier position of the pilot signal supplied from the propagation path estimating unit 18, and A propagation path estimated value distribution Hpa (t, i) is obtained. The propagation path estimated value distribution Hpa (t, i) obtained by the propagation path estimated value distribution measurement unit 19 is supplied to the spurious frequency detection unit 21 as the first distribution.

  FIG. 6 is a diagram for explaining the error vector. FIG. 6 is a diagram showing a complex plane similar to that in FIG. 2, and shows a plurality of ideal data signal points arranged vertically and horizontally on the complex plane. A data signal point 36 indicates one received data of a certain subcarrier after the equalization processing is performed by the propagation path compensation unit 17. The error vector 37 is a vector having an ideal data signal point 35 closest to the data signal point 36 as a start point and a data signal point 36 as an end point. The magnitude of the error vector 37 corresponds to the radius of the circle based on the reception data described with reference to FIG.

Let t be the symbol time and i be the subcarrier number. When the ideal data signal point is I (t, i) and the received data signal point is R (t, i), the error vector E (t, i) can be expressed by the following equation.
E (t, i) = R (t, i) −I (t, i)
However, t and i are variables that are valid only in the symbol time and subcarrier number to which the data signal is allocated.

Further, EVM (Error Vector Magnitude) is expressed by the following equation.
EVM (t, i) = E ² (t, i)
By calculating the average value of the EVM (t, i) in the symbol time direction for each subcarrier, the distribution in the frequency domain of EVM (t, i) (the sub-axis frequency is the horizontal axis and the vertical axis is EVM ( distribution with the average value of t, i) can be obtained. Here, the calculation for obtaining the average value may be a moving average within a predetermined time frame, or may be a calculation using an IIR (Infinite Impulse Response) filter. For example, when the moving average is used, the EVM distribution EVMa (t, i) is expressed by the following equation.

ここでｔｎは移動平均を求める時間枠の幅である。スプリアスが影響しないサブキャリア位置においてＥＶＭａ（ｔ，ｉ）は０に近い値となるが、スプリアスが影響するサブキャリア位置においてＥＶＭａ（ｔ，ｉ）は０より大きな値となる。スプリアスの周波数にサブキャリアの周波数が近いほど、ＥＶＭ分布ＥＶＭａ（ｔ，ｉ）の値は大きくなる。従って、このＥＶＭ分布ＥＶＭａ（ｔ，ｉ）のピーク位置に基づいて、スプリアスの周波数位置を特定することができる。

Here, tn is the width of the time frame for obtaining the moving average. Although EVMa (t, i) has a value close to 0 at the subcarrier position where spurious does not influence, EVMa (t, i) becomes a value larger than 0 at the subcarrier position where spurious affects. The closer the subcarrier frequency is to the spurious frequency, the greater the value of the EVM distribution EVMa (t, i). Therefore, the spurious frequency position can be specified based on the peak position of the EVM distribution EVMa (t, i).

  The spurious frequency position can be specified as described above only when the spurious amplitude is small, as will be described below. First, unlike the pilot signal, the original value of the data signal is unknown. For example, in the case of the example shown in FIG. 6, the error vector 37 is obtained assuming that the ideal data signal point 35 closest to the data signal point 36 is the original data value of the data at the data signal point 36. . However, the assumption that the closest ideal data signal point 35 is the original data value is valid only when the spurious amplitude is small. When the spurious amplitude is small, the position of the received data signal point is relatively close to the original data signal point position, and the ideal data signal point closest to the received data signal point position is It matches the original data signal point. However, when the spurious amplitude is large, the position of the received data signal point is far from the original data signal point position, and the ideal data signal point closest to the received data signal point position is the original data signal point. It is completely different from the signal point. Therefore, in this case, the magnitude of the error vector is saturated at a magnitude corresponding to ½ of the interval between adjacent ideal data signal points, and does not reflect the magnitude of the spurious amplitude. At this time, if the error vector is saturated not only at the subcarrier frequency close to the spurious frequency but also at the subcarrier frequency far from the spurious frequency, the error vector size distribution is There is no clear peak. Accordingly, the spurious frequency position cannot be specified from the EVM distribution EVMa (t, i).

  FIG. 7 is a diagram showing a distribution of error vector magnitudes when there is an unmodulated spurious signal having a small amplitude. In FIG. 7, the horizontal axis indicates the subcarrier number, and the vertical axis indicates the value of the error vector magnitude distribution EVMa (t, i). Since the subcarrier frequency increases as the subcarrier number increases, the horizontal axis may be considered as the subcarrier frequency. From the distribution of the magnitude of the error vector in FIG. 7, it can be seen that there is a spurious frequency position near the data subcarrier number 1600, and the spurious frequency can be specified. In this case, since the data signal is basically allocated to each subcarrier, the spurious frequency can be specified with the resolution of the subcarrier interval. In general, if the time symbol position is different, the subcarrier position into which the pilot signal is inserted is also different, so that a data signal always appears at any time symbol position in each subcarrier. Therefore, the spurious frequency can be specified with the resolution of the subcarrier interval.

  FIG. 8 is a diagram showing a distribution of error vector magnitudes when there is an unmodulated spurious signal having a large amplitude. In FIG. 8, the horizontal axis indicates the subcarrier number, and the vertical axis indicates the value of the error vector magnitude distribution EVMa (t, i). In the EVM error vector magnitude distribution EVMa (t, i) shown in FIG. 8, the magnitude of the error vector is saturated, and there is no clear peak. Therefore, the frequency position of spurious cannot be specified with sufficient accuracy. If the shape of the entire distribution is analyzed, there is a possibility that the approximate position where the spurious frequency exists can be identified, but there is a large error, and there is no guarantee that the PLL unit 13 is within the pull-in range. Therefore, when the magnitude of the error vector is saturated in this way, it is not appropriate to detect the spurious frequency position based on the error vector.

  Returning to FIG. 1, the data subcarrier EVM measurement unit 20 obtains an error vector based on the data signal point in the received data after equalization supplied from the propagation path compensation unit 17, and based on this error vector, the above-described error vector is obtained. The EVM distribution EVMa (t, i) is obtained. The EVM distribution EVMa (t, i) obtained by the data subcarrier EVM measurement unit 20 is supplied to the spurious frequency detection unit 21 as the second distribution.

  The spurious frequency detector 21 detects a spurious frequency position based on at least one of the first distribution and the second distribution. Specifically, when the error vector is not saturated in the EVM distribution from the data subcarrier EVM measurement unit 20, the spurious frequency detection unit 21 detects the peak position of the EVM distribution as the spurious frequency position. Further, when the error vector is saturated in the EVM distribution from the data subcarrier EVM measurement unit 20, the spurious frequency detection unit 21 transmits the channel estimation value distribution Hpa (t, i) from the channel estimation value distribution measurement unit 19. Is detected as a spurious frequency position. Whether or not the error vector is saturated in the EVM distribution may be determined based on whether or not the variance of the EVM distribution is larger than a predetermined value, for example. Alternatively, the determination is made based on whether or not the EVM value equal to or greater than a value lower than the maximum value of the EVM distribution by a predetermined ratio (for example, 80% of the maximum value) is greater than a predetermined number or greater than a predetermined ratio. It's okay. Alternatively, whether or not the value of the EVM distribution reaching a known error vector saturation value (250 to 300 in the examples of FIGS. 7 and 8) is greater than a predetermined number or greater than a predetermined ratio. You may judge by whether or not. The determination as to whether or not the error vector is saturated in the EVM distribution is not particularly limited.

  As described above, when the spurious amplitude is large, the error vector is saturated, and when the spurious amplitude is small, the error vector is not saturated. Therefore, when the peak value of the propagation path estimated value distribution Hpa (t, i) from the propagation path estimated value distribution measuring section 19 is smaller than a predetermined value, the spurious frequency detection unit 21 determines the peak position of the EVM distribution as the spurious frequency. You may detect as a frequency position. Further, when the peak value of the propagation path estimated value distribution Hpa (t, i) from the propagation path estimated value distribution measuring section 19 is larger than a predetermined value, the spurious frequency detecting section 21 determines the propagation path estimated value distribution Hpa (t , I) may be detected as a spurious frequency position. For the predetermined value to be compared with the peak value of the propagation path estimated value distribution Hpa (t, i), a value corresponding to the condition that the magnitude of the error vector is saturated may be obtained in advance by calculation, simulation, or the like.

  When the amplitude of the spurious is large, spurious frequency detection with coarse resolution based on the propagation path estimated value distribution Hpa (t, i) is performed. However, when the amplitude of spurious is large, the frequency pull-in width by the PLL unit 13 (the maximum difference between the initial frequency of the PLL unit 13 and the lockable frequency) is large. Don't be.

FIG. 9 is a diagram showing an example of the configuration of the rotor 25 of FIG. The rotor 25 includes multipliers 41 to 45 and adders 46 and 47. As a result, the rotor 25 responds to the reception signal RI + jRQ and the NCO oscillation signal NI + jNQ.
(RI / NI + RQ / NQ) + j (RQ / NI-RI / NQ)
Calculate This corresponds to the product of the complex conjugate of the NCO oscillation signal and the received signal. The reverse rotor 15 of FIG. 1 may perform an operation of multiplying the shifted reception signal output from the DC cut unit 14 by 1 / (complex conjugate of the NCO oscillation signal). Only the in-phase component RI · NI + RQ · NQ of the output of the rotor 25 is supplied to the loop filter 27.

  FIG. 10 is a diagram illustrating an example of the configuration of the loop filter 27. The loop filter 27 includes a multiplier 51, an adder 52, and a delay element 53. The multiplier 51 multiplies the input value (the in-phase component RI · NI + RQ · NQ of the output of the rotor 25) by a predetermined coefficient α satisfying 0 <α <1. The pass band of the loop filter 27 is determined according to the value of the coefficient α. The adder 52 adds the output of the multiplier 51 and the output of the adder 52 one cycle before. The delay element 53 delays the output of the adder 52 by one cycle. The output of the delay element 53 becomes the output of the loop filter 27.

In the PLL unit 13 of FIG. 1, when the frequency ω ₂ of the NCO oscillation signal is equal to the spurious frequency ω ₁ and the phase difference between the two signals is 90 degrees, the in-phase component of the output of the rotor 25 is zero (the quadrature component is other than zero). ) At this time, the output of the loop filter 27 is also zero, the NCO 26 continues to oscillate at the current oscillation frequency, and the PLL unit 13 is locked. When the frequency ω ₂ of the NCO oscillation signal is higher than the spurious frequency ω ₁ , the output of the loop filter 27 becomes non-zero (for example, a negative value), and the NCO 26 has a frequency lower than the current oscillation frequency according to this output value. It oscillates at. As a result, the frequency ω ₂ of the NCO oscillation signal approaches the spurious frequency ω ₁ . When the frequency ω ₂ of the NCO oscillation signal is lower than the spurious frequency ω ₁ , the output of the loop filter 27 becomes non-zero (for example, a positive value), and the NCO 26 is higher than the current oscillation frequency according to this output value. Oscillates at a frequency. As a result, the frequency ω ₂ of the NCO oscillation signal approaches the spurious frequency ω ₁ .

  FIG. 11 is a diagram illustrating a modification of the configuration of the signal processing circuit. In FIG. 11, the same or corresponding elements as those of FIG. 1 are referred to by the same or corresponding numerals, and a description thereof will be omitted as appropriate. Compared with the signal processing circuit 10 shown in FIG. 1, the signal processing circuit 10 </ b> A shown in FIG. 11 is provided with a band elimination filter (BEF) 22 instead of the PLL unit 13, the DC cut unit 14, and the reverse rotor 15. Is different. Other configurations are the same between the signal processing circuit 10 of FIG. 1 and the signal processing circuit 10A of FIG.

  As in the case of FIG. 1, the spurious frequency detector 21 detects the frequency position of the spurious. A signal indicating the detected frequency position is supplied from the spurious frequency detection unit 21 to the band elimination filter 22. The band elimination filter 22 reduces or removes the component of the reception signal supplied from the AD converter 12 at the frequency position indicated by the signal supplied from the spurious frequency detection unit 21 by filtering. Thus, the band elimination filter 22 can reduce or remove spurious components from the received signal.

  FIG. 12 is a diagram illustrating an example of a configuration of an OFDM reception system. The reception system shown in FIG. 12 includes a tuner 111, an OFDM demodulation circuit 112, a decoder 113, a CPU 114, a display 115, and a speaker 116. Tuner 111 receives a reception signal received by an antenna and outputs a baseband signal. The OFDM demodulation circuit 112 receives a baseband signal from the tuner 111 and outputs a digital signal after OFDM demodulation as a transport stream TS. The OFDM demodulation circuit 112 includes the signal processing circuit 10 of FIG. 1 or the signal processing circuit 10A of FIG. The OFDM demodulating circuit 112 further includes a circuit for deinterleaving received data with equalized transmission path characteristics, a circuit for soft decision by demapping processing, a circuit for executing error correction processing using Viterbi decoding processing, etc. May include. The decoder 113 receives the transport stream TS from the OFDM demodulation circuit 112 and generates an output signal including a video signal and an audio signal by executing a decoding process. The CPU 114 controls operations of the OFDM demodulation circuit 112 and the decoder 113. The display 115 outputs a video based on the video signal. The speaker 116 outputs sound based on the sound signal.

  As mentioned above, although this invention was demonstrated based on the Example, this invention is not limited to the said Example, A various deformation | transformation is possible within the range as described in a claim.

10 signal processing circuit 11 RF circuit 12 AD converter (ADC)
13 PLL unit 14 DC cut unit 15 Inverse rotor 16 FFT unit 17 Propagation channel compensation unit 18 Propagation channel estimation unit 19 Propagation channel estimated value distribution measurement unit 20 Data subcarrier EVM measurement unit 21 Spurious frequency detection unit

Claims (8)

An FFT unit for FFT of an OFDM received signal;
A propagation path estimation unit for obtaining a propagation path estimation value based on a pilot signal in the reception data output from the FFT unit;
A propagation path compensator for performing propagation path compensation on the reception data output from the FFT unit based on the propagation path estimation value;
When the error vector obtained based on the data signal point in the received data output from the propagation path compensation unit is not saturated, the magnitude of the error vector is saturated based on the error vector Includes a spurious frequency detection unit that detects a spurious frequency position based on the propagation path estimation value, and a spurious reduction unit that reduces the frequency component of the received signal at the spurious frequency position detected by the spurious frequency detection unit. A signal processing circuit comprising: The spurious reduction unit is
A PLL circuit that locks to an unmodulated wave or a narrow-band modulated wave signal in the vicinity of the spurious frequency position;
The signal processing circuit according to claim 1, further comprising: a signal cut unit that reduces a frequency component of the reception signal at a position where the PLL circuit is locked.   3. The PLL circuit further includes a rotor that shifts the entire frequency of the received signal so that the locked frequency becomes a DC component, and the signal cut unit reduces the signal component of the DC component. The signal processing circuit described. A channel estimation value distribution measuring unit for obtaining a first distribution in the frequency domain of the channel estimation value;
An error vector estimation unit for obtaining the error vector based on data signal points in the reception data output from the propagation path compensation unit, and further obtaining a second distribution in the frequency domain of the error vector;
The signal processing according to claim 3, further comprising: the spurious frequency detection unit detecting a frequency position of the spurious based on at least one of the first distribution and the second distribution. circuit. FFT of OFDM received signal,
A propagation path estimated value is obtained based on a pilot signal in the received data after the FFT,
Based on the propagation path estimation value, perform propagation path compensation on the received data after the FFT,
If the error vector obtained based on the data signal point in the received data after the propagation path compensation is not saturated, based on the error vector, if the magnitude of the error vector is saturated, Based on the propagation path estimation value, detect the spurious frequency position,
A signal processing method comprising: reducing each frequency component of the received signal at the detected spurious frequency position. Obtaining a first distribution in the frequency domain of the propagation path estimated value;
The method further includes obtaining a second distribution in the frequency domain of the error vector, and detecting the spurious frequency position is based on at least one of the first distribution and the second distribution. 6. The signal processing method according to claim 5, wherein the frequency position of the spurious is detected. Tuner,
A demodulation circuit that receives an output signal of the tuner and outputs a digital signal;
A decoder that receives the digital signal and generates an output signal;
A control circuit for controlling the demodulation circuit and the decoder;
An output device that outputs the output signal as at least one of video and audio, and the demodulation circuit includes:
An FFT unit that performs an FFT on an OFDM reception signal that is an output signal of the tuner;
A propagation path estimation unit for obtaining a propagation path estimation value based on a pilot signal in the reception data output from the FFT unit;
A propagation path compensator for performing propagation path compensation on the reception data output from the FFT unit based on the propagation path estimation value;
When the error vector obtained based on the data signal point in the received data output from the propagation path compensation unit is not saturated, the magnitude of the error vector is saturated based on the error vector Includes a spurious frequency detection unit that detects a spurious frequency position based on the propagation path estimation value, and a spurious reduction unit that reduces the frequency component of the received signal at the spurious frequency position detected by the spurious frequency detection unit. A receiving system comprising: The spurious reduction unit is
A PLL circuit that locks to an unmodulated wave or a narrow-band modulated wave signal in the vicinity of the spurious frequency position;
8. The reception system according to claim 7, further comprising a signal cut unit that reduces a frequency component of the reception signal at a position where the PLL circuit is locked.

Images