JP5566223B2 - Diversity receiving apparatus and diversity receiving method - Google Patents

Description

  The present invention relates to a diversity receiving apparatus and a diversity receiving method for diversity receiving and demodulating a digital modulation signal based on an OFDM (Orthogonal Frequency Division Multiplexing) system.

  OFDM modulation is known as a modulation scheme with high resistance against multipath interference, and is adopted in terrestrial digital broadcasting standards such as DVB-T in Europe and ISDB-T in Japan.

  An OFDM modulated wave transmitted from a broadcast station may be received on a transmission path in a form in which various interference waves such as impulse noise, spurious interference, and co-channel NTSC interference are superimposed. In the configuration of the OFDM receiver, improvement of resistance to various interference waves is an important issue. In particular, in mobile receivers such as in-vehicle receivers and portable receivers that are expected to be used in various reception environments, time domain noise typified by impulse noise and frequency selection typified by spurious interference High immunity to both sexual noise is required.

  In the configuration of the in-vehicle receiver, it is essential to apply diversity reception technology. The diversity reception technique improves reception performance by combining signals received by a plurality of antennas. In particular, in OFDM modulation, a transmission data signal is distributed and transmitted over a large number of carriers, so that a significant improvement in performance can be achieved by appropriately combining these carriers on the receiving device side ( Patent Literatures 1 to 3).

  FIG. 1 schematically shows a configuration of a general diversity OFDM receiver. In the receiving apparatus of FIG. 1, the number of diversity receiving systems (hereinafter called branches) is two.

  FIG. 2 shows an example of the format of an OFDM signal that is a signal to be received by the receiving apparatus of FIG. As shown in FIG. 2, the OFDM signal has a data carrier (white circle in FIG. 2) and a pilot carrier (double in FIG. 2) expressed by a plurality of subcarrier positions in the carrier frequency direction and a plurality of symbol positions in the symbol time direction. Circle). The pilot carrier is a signal having a known value and is periodically inserted.

The diversity OFDM receiver of FIG. 1 includes detectors 1 and 2 for each branch, and further includes a demodulator 3 and a decoder 4. The detectors 1 and 2 in each branch sequentially convert the RF signals input from the antennas 1a and 2a into received OFDM symbols and output the received OFDM symbols. Each OFDM symbol is composed of received values Y ₁ and Y ₂ of a predetermined number of subcarriers.

  The detection units 1 and 2 generally perform processing such as RF filtering, frequency conversion to IF signals, IF filtering, A / D conversion, sampling frequency conversion, FFT window extraction, and FFT.

The received OFDM symbols output from the two detectors 1 and 2 are sequentially input to the demodulator 3. The demodulation unit 3 performs demodulation processing on the reception values Y ₁ and Y ₂ included in the reception OFDM symbol to calculate the demodulation value V and the reliability Q of the demodulation value V.

  The decoding unit 4 performs processing including Viterbi decoding, deinterleaving, and RS decoding on the demodulated value V to restore a transmission information sequence. The reliability Q described above is used in Viterbi decoding. By reducing the metric weight corresponding to the demodulated value V with low reliability Q, Viterbi decoding with high noise resistance is achieved.

  As shown in FIG. 3, the demodulation unit 3 includes transmission path estimation units 21 and 22, noise power estimation units 23 and 24, a synthesis equalization unit 25, and a transmission value estimation unit 26.

Each of the transmission path estimators 21 and 22 estimates the transfer characteristics of the received values Y ₁ and Y ₂ . Since the estimation accuracy of the transfer characteristics greatly affects the performance of the receiving device, various estimation methods have been devised. In the case of a receiving apparatus corresponding to the ISDB-T standard, it is common to perform transmission path estimation using a pilot carrier defined in the standard. Note that a specific method of channel estimation is not directly related to the present invention, and thus detailed description thereof is omitted.

The noise power estimation units 23 and 24 estimate the power of the noise component included in the received values Y ₁ and Y ₂ (hereinafter referred to as noise power). Details of the noise power estimation units 23 and 24 will be described later.

The synthesis equalization unit 25 calculates the demodulated value V by combining and equalizing the received values Y ₁ and Y ₂ of each branch based on the estimated transfer characteristic and the estimated noise power, and the reliability of the demodulated value V Q is calculated. Specifically, the demodulated value V and the corresponding reliability Q are calculated according to the following equations (1) and (2).

V = (Y ₁ H ₁ ^* / Z ₁ + Y ₂ H ₂ ^* / Z ₂ ) / Q (1)
Here, x ^* represents a complex conjugate of x.

Q = | H ₁ | ² / Z ₁ + | H ₂ | ² / Z ₂ (2)
In the above equations (1) and (2), H ₁ is the transfer characteristic of the received value Y ₁ estimated by the transmission path estimation unit 21, and H ₂ is the transfer characteristic of the reception value Y ₂ estimated by the transmission path estimation unit 22. , Z ₁ represents the noise power of the reception value Y ₁ estimated by the noise power estimation unit 23, and Z ₂ represents the noise power of the reception value Y ₂ estimated by the noise power estimation unit 24. The above synthesis equalization method is widely known as maximum ratio synthesis.

  In Equation (1), synthesis is performed in consideration of the estimated noise power of each branch. As is clear from the equation (1), the received value of the branch having a large estimated noise power can reduce the contribution to the synthesis result. Thereby, the influence of noise is suppressed. Each term of the formula (2) represents the CN ratio of each branch, and the reliability Q calculated as the sum is nothing but the combined CN ratio.

  Thus, since synthesis equalization and reliability calculation are performed based on estimated noise power, the estimation accuracy is very important. It can be said that the immunity to various types of noise is determined by the estimation accuracy of noise power.

  The transmission value estimation unit 26 estimates and outputs a transmission value based on the demodulated value V. A so-called slicer is often used for estimating the transmission value. In general, the transmission value is one of specified constellation points that characterize a digital modulation system such as QPSK or 16QAM. The slicer estimates a transmission value by selecting a point closest to the demodulated value from a set of constellation points. The estimated transmission value is supplied to the noise power estimation units 23 and 24. When the demodulated value V corresponds to a pilot signal, there is no need for estimation, and a known transmission value is output.

  Next, the noise power estimation units 23 and 24 will be described in detail. As shown in FIG. 4, each of the noise power estimation units 23 and 24 of each branch includes an instantaneous noise power calculation unit 31 and a symbol direction filter 32. The instantaneous noise power calculation unit 31 calculates an instantaneous power (hereinafter, instantaneous noise power) Eq of a noise component included in the received value Yq. The instantaneous noise power Eq is calculated according to the following equations (3) and (4). Note that q is a branch index (1 or 2).

Wq = Yq-D · Hq (3)
Eq = | Wq | ² … (4)
In Expression (3), D is an estimated transmission value output from the transmission value estimation unit 26. By multiplying the estimated transmission value D by the estimated transfer characteristic Hq, a received value in a state without noise is obtained, and this is obtained. By subtracting from the actual reception value Yq, the noise component Wq included in the reception value Yq is obtained. In equation (4), the instantaneous noise power Eq is calculated as the square of the absolute value of the noise component Wq.

  The instantaneous noise power Eq calculated in this way is merely a sample of the noise power and cannot be used for the synthesis equalization process as it is. In order to estimate the noise power, it is necessary to perform some averaging on the instantaneous noise power. The symbol direction filter 32 of FIG. 4 calculates the estimated noise power Zq by filtering the instantaneous noise power Eq in the symbol time direction.

  For example, the symbol direction filter 32 averages the instantaneous noise power Eq of the received value indicated by ◆ of the same subcarrier frequency in the format of the subcarrier modulation signal of FIG. Zq is calculated.

  Here, the estimated noise power Zq is calculated based on the instantaneous noise power Eq of the past symbol, and the instantaneous noise power Eq of the current symbol is not reflected in the estimated noise power Zq. This is because the estimated transmission value D is required to calculate the instantaneous noise power Eq, and the synthesis equalization process using the estimated noise power Zq is required to calculate the estimated transmission value D.

  FIGS. 6A and 6B are graphs showing estimated noise power calculated in an actual in-vehicle reception environment. 6 (a) shows the estimated noise power Z1 of the branch B1, and FIG. 6 (b) shows the estimated noise power Z2 of the branch B2. 6 (a) and 6 (b), it can be seen that the power distribution of frequency selective noise, which is different in each of the branches B1 and B2, is well estimated. As described above, by performing the synthesis equalization processing based on the estimated noise power calculated for each branch and for each subcarrier frequency, it is possible to exhibit extremely high resistance to frequency selective noise. .

  As described above, the conventional receiving apparatus has a very high resistance to frequency selective noise. However, in contrast, the tolerance to time domain noise represented by impulse noise is low. This is apparent when considering the case where a large impulse noise is superimposed only on a specific symbol. In the specific symbol, the estimated noise power should be calculated higher than in the normal case, but the conventional receiving apparatus does not have such a mechanism. At that particular symbol, a high value will be observed in the instantaneous noise power, but this is reflected in the next symbol as an increase in the estimated noise power, at which point the impact of the impulse noise is It has already disappeared.

Japanese Patent No. 4367276 Japanese Patent No. 4396423 Table 2005/109711

  In mobile reception and in-vehicle reception of OFDM signals, countermeasures against both frequency selective noise and time domain noise are required. However, the conventional diversity receiver has a problem in that no measures against time domain noise are taken and sufficient reception quality cannot be ensured.

  Therefore, the problem to be solved by the present invention includes the above-mentioned drawbacks as an example, and it is an object of the present invention to provide a diversity receiving apparatus and a diversity receiving method that are highly resistant to both frequency selective noise and time domain noise. Is the purpose.

  A diversity receiving apparatus according to claim 1 of the present application is a diversity receiving apparatus that receives and combines OFDM signals including a plurality of carriers including one or more pilot carriers for each symbol for each reception system. Detecting means for converting the OFDM signal into an OFDM signal in a frequency domain for each reception system and outputting a reception value of each of the plurality of carriers, and a transfer characteristic of an output reception value of the detection means for each reception system Estimating the noise power of the output reception value for each reception system based on the transmission path estimation means to be estimated, the output reception value of the detection means, the estimated transmission characteristic estimated by the transmission path estimation means and the estimated transmission value Noise power estimation means, output reception value of the detection means for each reception system, estimated transfer characteristics by the transmission path estimation means, and noise power Synthesis equalization means for calculating the demodulated value by synthesizing the estimated noise power by the determining means; decoding means for decoding the demodulated value to restore the transmission information sequence; and estimating the transmission value from the demodulated value Transmission value estimation means for outputting the estimated transmission value, the noise power estimation means for each reception system, the output reception value, the estimated transfer characteristic corresponding to the output reception value, and the output reception First instantaneous noise power calculating means for calculating the instantaneous noise power of the output reception value based on the estimated transmission value corresponding to the value, and the instantaneous noise power calculated by the first instantaneous noise power calculating means Symbol direction filter means for filtering in the direction and outputting first temporary noise power for each carrier frequency, and pilot reception corresponding to the pilot carrier of the output reception values Second instantaneous noise power calculating means for calculating an instantaneous noise power of the pilot reception value based on the estimated transmission characteristic corresponding to the pilot reception value and a known transmission value corresponding to the pilot reception value; Noise power correcting means for correcting one temporary noise power based on the instantaneous noise power of the pilot reception value calculated by the second instantaneous noise power calculating means and outputting the estimated noise power. Yes.

  The diversity reception method of the invention according to claim 10 of the present application is a diversity reception method of a diversity reception device that performs reception processing for each reception system and combines OFDM signals including a plurality of carriers including one or more pilot carriers for each symbol. A detection step of converting the received OFDM signal into a frequency domain OFDM signal for each reception system and outputting a reception value of each of the plurality of carriers, and a transfer characteristic of an output reception value of the detection step A transmission path estimation step for each reception system, an output reception value of the detection step, an estimated transmission characteristic estimated by the transmission path estimation step, and a noise power of the output reception value based on an estimated transmission value. Noise power estimation step for each estimation, received output value for each reception system, transmission path estimation A combined equalization step for calculating a demodulated value by combining and equalizing an estimated transfer characteristic by the step and an estimated noise power by the noise power estimating step; and a decoding step for decoding the demodulated value to restore a transmission information sequence; A transmission value estimation step of estimating a transmission value from the demodulated value and outputting the estimated transmission value, and the noise power estimation step corresponds to the output reception value and the output reception value for each reception system A first instantaneous noise power calculation step of calculating an instantaneous noise power of the output reception value based on the estimated transmission characteristic to be performed and the estimated transmission value corresponding to the output reception value, and the first instantaneous noise power calculation step. Symbol direction filtering that outputs the first temporary noise power for each carrier frequency by filtering the calculated instantaneous noise power in the symbol time direction. A pilot reception value corresponding to the pilot carrier of the output reception value, the estimated transfer characteristic corresponding to the pilot reception value, and a known transmission value corresponding to the pilot reception value. A second instantaneous noise power calculating step for calculating the instantaneous noise power of the value; and correcting the first temporary noise power based on the instantaneous noise power of the pilot reception value calculated by the second instantaneous noise power calculating step. And a noise power correction step for outputting the estimated noise power.

It is a block diagram which shows the general structure of a diversity receiver. It is a figure which shows the format of a received OFDM signal. FIG. 2 is a block diagram illustrating a configuration of a demodulator in the receiving apparatus of FIG. 1. FIG. 5 is a block diagram illustrating a configuration of a noise power estimation unit in the demodulation unit of FIG. 4. It is explanatory drawing for demonstrating operation | movement of a symbol direction filter. It is a figure which shows the estimated noise electric power for every branch. It is a block diagram which shows the structure of a noise power estimation part as 1st Example of this invention. It is explanatory drawing for demonstrating operation | movement of a carrier direction filter. It is a figure which illustrates the 1st and 2nd temporary noise electric power when time domain noise is not included. It is a figure which illustrates the 1st and 2nd temporary noise electric power in case the time domain noise is contained. It is a flowchart which shows operation | movement of the noise power correction | amendment part in the noise power estimation part of FIG. It is a figure which illustrates the relationship between the 1st and 2nd temporary noise electric power, and the threshold value in a 1st Example. It is a block diagram which shows the structure of a noise power estimation part as 2nd Example of this invention. It is a flowchart which shows operation | movement of the noise power correction | amendment part in the noise power estimation part of FIG. It is a figure which illustrates the threshold value in the 1st temporary noise power, 2nd temporary noise power, and a 1st Example in case a sine wave disturbance exists. It is a figure which illustrates the threshold value in the 1st temporary noise power, 2nd temporary noise power, and 1st Example in case sine wave disturbance and time domain noise exist. FIG. 16 is a diagram showing estimated noise power obtained in the first example in the case of the example of FIG. 15. FIG. 17 is a diagram illustrating estimated noise power obtained in the first example in the case of the example of FIG. 16. It is a figure which illustrates the threshold value in the 2nd temporary noise power, 3rd temporary noise power, and a 2nd Example in case sinusoidal interference exists. FIG. 20 is a diagram illustrating estimated noise power obtained in the second example in the case of the example of FIG. 19. It is a figure which illustrates the threshold value in the 2nd temporary noise power in the presence of sine wave interference and time domain noise, the 3rd temporary noise power, and the 2nd example. FIG. 22 is a diagram showing estimated noise power obtained in the second example in the case of the example of FIG. 21. It is a block diagram which shows the structure of a noise power estimation part as 3rd Example of this invention. It is a flowchart which shows operation | movement of the noise power correction | amendment part in the noise power estimation part of FIG. It is a figure which illustrates the threshold value in the 2nd temporary noise power, the 3rd temporary noise power, and a 2nd Example in case the time domain noise of low electric power exists. FIG. 26 is a diagram illustrating estimated noise power obtained in the second example in the case of the example of FIG. 25. It is a figure which illustrates the threshold value in the 1st and 2nd temporary noise electric power, the 1st and 2nd average power, and the 3rd example when time domain noise is not included. It is a figure which illustrates the threshold value in the 1st and 2nd temporary noise power, the 1st and 2nd average power, and the 3rd example when low-power time domain noise exists. FIG. 29 is a diagram illustrating estimated noise power obtained in the third example in the case of the example of FIG. 28. It is a block diagram which shows the structure of a noise power estimation part as 4th Example of this invention. It is a flowchart which shows operation | movement of the noise power correction | amendment part in the noise power estimation part of FIG.

  According to the diversity receiver and diversity reception method of the present invention, the demodulated value is obtained by combining and equalizing the output reception value of the detection means for each reception system, the estimated transfer characteristic by the transmission path estimation means, and the estimated noise power by the noise power estimation means. Synthesizing and equalizing means for calculating, a decoding means for decoding the demodulated value to restore the transmission information sequence, and a transmission value estimating means for estimating the transmission value from the demodulated value and outputting the estimated transmission value, The noise power estimation means calculates the instantaneous noise power of the output reception value for each reception system based on the output reception value, the estimated transfer characteristic corresponding to the output reception value, and the estimated transmission value corresponding to the output reception value. 1 instantaneous noise power calculation means, symbol direction filter means for filtering the instantaneous noise power in the symbol time direction and outputting first temporary noise power for each carrier frequency, and output reception Second instantaneous noise for calculating the instantaneous noise power of the pilot reception value based on the pilot reception value corresponding to the pilot carrier, the estimated transmission characteristic corresponding to the pilot reception value, and the known transmission value corresponding to the pilot reception value Power calculating means, and noise power correcting means for correcting the first temporary noise power based on the instantaneous noise power of the pilot reception value and outputting estimated noise power. Thus, the constant noise power is obtained by correcting the instantaneous noise power of the received value output from the detection means based on the instantaneous noise power of the pilot received value of the output received value, and the estimated noise power is Since the synthesis equalization processing is performed by the synthesis equalization means, good reception characteristics can be obtained not only for frequency selective noise but also for time domain noise. That is, a diversity receiving apparatus and a diversity receiving method that are highly resistant to both frequency selective noise and time domain noise are achieved.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  FIG. 7 shows the configuration of the noise power estimation unit of the diversity receiver as the first embodiment of the present invention. The first embodiment is applied to the demodulator 3 of the receiving apparatus shown in FIG. That is, the noise power estimator of the first embodiment is used as the noise power estimators 23 and 24 of the demodulator 3 shown in FIG.

  7 includes a pilot carrier extraction unit 43, a pilot instantaneous noise power calculation unit 44, a carrier direction filter 45, a noise power correction unit, in addition to the instantaneous noise power calculation unit 41 and the symbol direction filter 42. 46.

  Since the instantaneous noise power calculation unit 41 and the symbol direction filter 42 are the same as the instantaneous noise power calculation unit 31 and the symbol direction filter 32 shown in FIG. 4, description thereof is omitted here.

  The noise power estimated by the symbol direction filter 42 is referred to as a first temporary noise power Z1q.

  The pilot carrier extraction unit 43 extracts a reception value corresponding to the pilot carrier from the reception value Yq indicated by the output signal of the detection unit (detection unit 1 or 2 in FIG. 1).

  Only the reception value extracted by the pilot carrier extraction unit 43 is input to the pilot instantaneous noise power calculation unit 44. The pilot instantaneous noise power calculation unit 44 calculates the instantaneous noise power of these received values (hereinafter referred to as pilot instantaneous noise power). A specific calculation method is the same as the above-described equations (3) and (4). However, for the estimated transmission value D, since the target is a pilot signal, a known transmission value is used. This instantaneous noise power is calculated using a known transmission value, and is not calculated for data carriers other than the pilot carrier.

  The carrier direction filter 45 calculates the second temporary noise power Z2q for each carrier frequency by filtering the pilot instantaneous noise power output from the pilot instantaneous noise power calculation unit 44 in the carrier frequency direction. To give a simple example, the second temporary noise power at the position indicated by ● in the OFDM signal format shown in FIG. 8 is the simple average of the pilot instantaneous noise power at the four pilot carrier positions indicated by arrows A. Is calculated as In this example, the simple average value is also the second temporary noise power Z2q for the carrier positions indicated by ▲ and ■.

  Although simple filtering is illustrated here, it is also possible to obtain the second temporary noise power Z2q at the point ● as the weighted average value of the pilot instantaneous noise power at the four positions indicated by the arrow A, that is, the FIR calculation result. It is. It is also possible to use different weighting factors for the carrier positions of ●, ▲, and ■.

  Since the second temporary noise power Z2q calculated in this way is calculated based only on the current symbol, it strongly reflects the time domain noise included in the current symbol. As shown in FIG. 9, the second temporary noise power Z2q is substantially equal to the first temporary noise power Z1q during normal times, that is, when time domain noise is not included. However, it should be noted that the second temporary noise power Z2q is less accurate than the first temporary noise power Z1q. This is because the first temporary noise power Z1q is calculated with high accuracy by a long-term average of past symbols, whereas the second temporary noise power Z2q is calculated only from the pilot carrier of the current symbol. . On the other hand, when the current symbol includes time domain noise, the second temporary noise power Z2q is larger than the first temporary noise power Z1q, as shown in FIG.

  The noise power correction unit 46 detects the time domain noise by comparing the first temporary noise power Z1q and the second temporary noise power Z2q, and corrects the first temporary noise power Z1q when the time domain noise is detected. Estimated noise power Zq.

  As the simplest correction method of the noise power correction unit 46, a higher temporary noise power is selected from the first temporary noise power Z1q and the second temporary noise power Z2q for each carrier frequency, and is finally estimated. A method of using noise power is conceivable. In this way, for example, in FIG. 10, the second temporary noise power Z2q is selected, and the influence of time domain noise can be reduced.

  However, this method has a negative effect when there is no time domain noise. That is, the second temporary noise power Z2q with low accuracy is selected at the portion indicated by the arrow B in FIG. 9, which is not preferable. In normal times, it is desirable to select the first temporary noise power Z1q obtained with high accuracy as the final estimated noise power Zq.

  Therefore, in the preferred noise power correction unit 46, as shown in FIG. 11, the threshold Tq is set by adding a predetermined margin to the first temporary noise power Z1q for each carrier frequency (step S1). . For example, the threshold Tq is set as shown in FIG. Then, the second temporary noise power Z2q is compared with the threshold value Tq (step S2).

  The noise power correction unit 46 selects the second temporary noise power Z2q as the final estimated noise power Zq only when the second temporary noise power Z2q is equal to or greater than the threshold Tq (step S3). If the second temporary noise power Z2q is smaller than the threshold value Tq, the first temporary noise power Z1q is selected as the final estimated noise power Zq (step S4). By doing so, the second temporary noise power Z2q at the normal time becomes lower than the threshold value Tq, so that it is not selected as the final estimated noise power Zq.

  As described above, according to the first embodiment, when the time domain noise is included in the symbol, the second temporary noise power Z2q including the time domain noise is selected as the final estimated noise power Zq, and the time is included in the symbol. When the area noise is not included, the first estimated noise power Z1q with high accuracy is selected as the final estimated noise power Zq with respect to the frequency selective noise, so that both the frequency selective noise and the time domain noise are detected. It becomes possible to increase tolerance.

  However, it is not preferable to set the threshold Tq higher than necessary. This is because if the threshold value Tq is too high, the detection sensitivity of time domain noise is lowered. The threshold Tq is preferably set to the minimum necessary in view of the estimation accuracy of the second temporary noise power Z2q.

  Note that the threshold Tq can be set by multiplication in addition to the method by addition as described above. That is, when the second temporary noise power Z2q is larger than a predetermined multiple of the first temporary noise power Z1q, the second temporary noise power Z2q can be selected as the final estimated noise power Zq.

  FIG. 13 shows the configuration of the noise power estimation unit of the receiving apparatus as the second embodiment of the present invention. The noise power estimation unit in FIG. 13 includes the instantaneous noise power calculation unit 41, symbol direction filter 42, pilot carrier extraction unit 43, pilot instantaneous noise power calculation unit 44, carrier direction filter 45, and noise power correction unit 46 in FIG. In addition to the configuration described above, a carrier direction filter 47 is provided.

  The carrier direction filter 47 calculates the third temporary noise power Z3q for each carrier frequency by filtering the first temporary noise power Z1q output from the symbol direction filter 42 in the carrier frequency direction. Here, the carrier direction filter 47 desirably performs the same filtering as the carrier direction filter 45. That is, when the carrier direction filter 45 performs the above-described calculation, the carrier direction filter 47 similarly uses the third temporary noise power Z3q at the carrier position indicated by ▲, ●, and ■ in FIG. It is desirable to calculate as a simple average of the first temporary noise power at the four carrier positions indicated by the arrow A.

  The noise power correction unit 46 determines the final estimated noise power Zq based on the first to third temporary noise powers Z1q, Z2q, and Z3q for each carrier frequency. Specifically, as shown in FIG. 14, first, a threshold Tq is set by adding a predetermined margin to the third temporary noise power Z3q (step S11). Then, the second temporary noise power Z2q is compared with the threshold Tq (step S12), and if the second temporary noise power Z2q is lower than the threshold Tq, the first temporary noise power Z1q is selected as the final estimated noise power Zq. (Step S13). On the other hand, when the second temporary noise power Z2q is equal to or greater than the threshold Tq, a difference value obtained by subtracting the third temporary noise power Z3q from the second temporary noise power Z2q is added to the first temporary noise power Z1q. Thus, the final estimated noise power Zq is calculated (step S14).

  The second embodiment is effective in the case where there is narrow-band interference represented by sinusoidal interference. First, consider the movement of the noise power estimation unit of the first embodiment in such a sinusoidal disturbance environment.

  FIGS. 15 and 16 illustrate the first temporary noise power Z1q, the second temporary noise power Z2q, and the threshold value Tq in the first embodiment in the case where sinusoidal interference exists. FIG. 15 shows a case where time domain noise does not exist, and FIG. 16 shows a case where time domain noise exists. As shown in the drawing, a narrow band sinusoidal interference power spectrum is estimated with high resolution for the first temporary noise power Z1q. On the other hand, in the second temporary noise power Z2q, the spectrum of the sinusoidal interference that should be local spreads to the periphery due to the influence of the carrier direction filter. As a result, the estimated noise power Zq is calculated as shown in FIG. 17 in the case of FIG. 15 and as shown in FIG. 18 in the case of FIG. In either case, the estimated noise power Zq is estimated to be higher than the actual value around the sinusoidal disturbance, which is not preferable.

  Such a problem does not occur in the second embodiment. 19 and 21 illustrate the third temporary noise power Z3q, the second temporary noise power Z2q, and the threshold value Tq in the second embodiment when sinusoidal interference exists. FIG. 19 shows a case where time domain noise does not exist, and FIG. 21 shows a case where time domain noise exists. Due to the effect of the symbol direction filter 47, the spread of the spectrum around the sine wave interference is observed in the third temporary noise power Z3q and the threshold value Tq, similarly to the second temporary noise power Z2q. As a result, in the second embodiment, the estimated noise power is calculated as shown in FIG. 20 in the case of FIG. 19 and as shown in FIG. 22 in the case of FIG. The effectiveness of the second embodiment will be understood when compared with FIGS.

  Thus, in the second embodiment, the carrier resolution filter intentionally reduces the frequency resolution of the first temporary noise power to the same level as the second temporary noise power, and the difference between this result and the second temporary noise power. The final estimated noise power Zq is calculated by correcting the first temporary noise power based on the value. With this configuration, the estimated noise power Zq can be appropriately obtained even when narrow-band interference such as sinusoidal interference is present, and reception performance can be improved.

  FIG. 23 shows the configuration of the noise power estimation unit of the receiving apparatus as the third embodiment of the present invention. 23 includes an instantaneous noise power calculation unit 41, a symbol direction filter 42, a pilot carrier extraction unit 43, a pilot instantaneous noise power calculation unit 44, a carrier direction filter 45, a noise power correction unit 46, and a carrier. In addition to the configuration comprising the direction filter 47, first and second carrier direction averagers 48 and 49 are provided.

  The first carrier direction averager 48 calculates the first noise power average A1q by averaging the first temporary noise power Z1q for all pilot carrier frequencies. The second carrier direction averager 49 calculates the second noise power average A2q by averaging the pilot instantaneous noise power output from the pilot instantaneous noise power calculation unit 44 for all pilot carrier frequencies.

  As shown in FIG. 23, the noise power correction unit 46 includes a first temporary noise power Z1q, a second temporary noise power Z2q, a third temporary noise power Z3q, a first noise power average A1q, and a second noise power average A2q. Is supplied. The noise power correction unit 46 calculates the estimated noise power Zq based on these values Z1q, Z2q, Z3q, A1q, A2q. Specifically, as shown in FIG. 24, first, a threshold Tq is set by adding a predetermined margin to the first noise power average A1q (step S21). The second noise power average A2q and the threshold Tq are compared (step S22). If the second noise power average A2q is smaller than the threshold Tq, it is determined that the time domain noise is not included in the symbol, For the carrier frequency, the first temporary noise power Z1q is output as the estimated noise power Zq (step S23). On the other hand, when the second noise power average A2q is larger than the threshold value Tq, the same processing as the noise power correction unit in the second embodiment is performed. However, the threshold margin is zero. That is, the second temporary noise power Z2q and the third temporary noise power Z3q are compared (step S24). If the second temporary noise power Z2q is lower than the third temporary noise power Z3q, the first temporary noise power Z1q is It is selected as the estimated noise power Zq (step S25), and when the second temporary noise power Z2q is equal to or higher than the third temporary noise power Z3q, the third temporary noise power Z3q is subtracted from the second temporary noise power Z2q. The estimated noise power Zq is calculated by adding the obtained difference value to the first temporary noise power Z1q (step S26).

  In the first and second embodiments, the time domain noise detection, that is, the determination as to whether or not the time domain noise exists is determined by the second temporary noise power Z2q for each carrier frequency as in steps S2 and S12. Was done on the basis of. For this reason, it is necessary to set the threshold Tq margin in accordance with the low accuracy of the second temporary noise power Z2q. In contrast, in the third embodiment, time domain noise is detected based on the second noise power average A2q as in step S22. At this time, since the accuracy of the second noise power average A2q is higher than the accuracy of the second temporary noise power Z2q, the threshold Tq margin can be reduced. As a result, it becomes possible to increase the detection sensitivity of time domain noise.

  FIG. 25 exemplifies the second temporary noise power Z2q, the third temporary noise power Z3q, and the threshold value Tq in the second embodiment in the case where time domain noise having a lower power than those in FIGS. 16 and 21 is present. Note that FIG. 25 is enlarged in the vertical axis direction compared to FIG. 16 and FIG. As shown in FIG. 25, when the power of the time domain noise is small, the second temporary noise power Z2q falls within the threshold margin in most frequency regions. For this reason, in the case of the second embodiment, time domain noise is not detected, and the estimated noise power Zq is calculated as shown in FIG. 26, which is not preferable.

  FIG. 27 is a diagram illustrating setting of the first noise power average A1q, the second noise power average A2q, and the threshold value Tq in the third embodiment when no time domain noise exists. As shown in the figure, the first noise power average A1q and the second noise power average A2q in the absence of time domain noise are very close values. Therefore, in the third embodiment, the threshold margin can be set smaller than that in the second embodiment.

  FIG. 28 is a diagram illustrating the setting of the first noise power average A1q, the second noise power average A2q, and the threshold Tq in the same situation as in FIG. 25, that is, when low power time domain noise exists. In this way, in the third embodiment, the threshold margin can be set small, so that the noise power correction unit 46 can detect this even when the time domain noise is low power. As a result, the estimated noise power Zq is calculated as shown in FIG.

  As described above, according to the third embodiment, the time domain noise can be detected with high sensitivity based on the first noise power average A1q and the second noise power average A2q calculated with high accuracy by averaging. Therefore, it is possible to improve reception performance in a reception environment where time domain noise with small power exists.

  FIG. 30 shows the configuration of the noise power estimation unit of the receiving apparatus as the fourth embodiment of the present invention. 30 includes an instantaneous noise power calculation unit 41, a symbol direction filter 42, a pilot carrier extraction unit 43, a pilot instantaneous noise power calculation unit 44, a noise power correction unit 46, a first carrier direction averager 48, And a second carrier direction averager 49. Note that the carrier direction filter used in the previous embodiments is not provided.

  As shown in FIG. 30, the noise power correction unit 46 is supplied with the first temporary noise power Z1q, the first noise power average A1q, and the second noise power average A2q. As shown in FIG. 31, the noise power correction unit 46 first sets a threshold Tq by adding a predetermined margin to the first noise power average A1q (step S31). The second noise power average A2q is compared with this threshold Tq (step S32), and if the second noise power average A2q is smaller than the threshold Tq, it is determined that the time domain noise is not included in the symbol, The first temporary noise power Z1q is output as the estimated noise power Zq for all carrier frequencies (step S33). The processing so far is the same as in the third embodiment. On the other hand, when the second noise power average A2q is equal to or greater than the threshold Tq, the difference value obtained by subtracting the first noise power average A1q from the second noise power average A2q is set for all carrier frequencies. Estimated noise power Zq is calculated by adding to noise power Z1q (step S34).

  In the fourth embodiment, since no carrier direction filter is provided, there is an advantage that the calculation resources of the noise power correction unit 46 can be reduced. However, the frequency dependence of time domain noise is not considered at all. For this reason, a slight performance deterioration is expected as compared with the third embodiment. However, the amount of deterioration is slight, and the fourth embodiment is most preferable from the viewpoint of the cost to performance ratio.

  In each of the above-described embodiments, an example of a diversity receiving device having two branches (reception systems) each connected to an individual antenna is shown, but the present invention is not limited to this. Absent. Of course, the present invention can also be applied to a diversity receiver having three or more branches each connected to a separate antenna.

  Further, it may be configured as hardware, or all or a part of the configuration of each of the above embodiments may be formed by a computer reading and executing a program stored in a recording medium.

DESCRIPTION OF SYMBOLS 1, 2 Detection part 1a, 2a Antenna 3 Demodulation part 4 Decoding part 21, 22 Transmission path estimation part 23, 24 Noise power estimation part 31, 41 Instantaneous noise power calculation part 32, 42 Symbol direction filter 44 Pilot instantaneous noise power calculation part 45, 47 Carrier direction filter 46 Noise power correction unit 48, 49 Carrier direction averager

Claims (10)

A diversity receiver for receiving and combining OFDM signals composed of a plurality of carriers including one or more pilot carriers for each symbol for each reception system,
Detecting means for converting the received OFDM signal into an OFDM signal in a frequency domain for each reception system and outputting a reception value of each of the plurality of carriers;
Transmission path estimation means for estimating the transmission characteristic of the output reception value of the detection means for each reception system;
Noise power estimation means for estimating the noise power of the output reception value for each reception system based on the output reception value of the detection means, the estimated transmission characteristic estimated by the transmission path estimation means, and the estimated transmission value;
A combined equalization means for calculating a demodulated value by combining and equalizing an output reception value of the detection means for each reception system, an estimated transfer characteristic by the transmission path estimation means, and an estimated noise power by the noise power estimation means;
Decoding means for decoding the demodulated value to restore a transmission information sequence;
Transmission value estimation means for estimating a transmission value from the demodulated value and outputting the estimated transmission value,
The noise power estimation means is for each reception system,
First instantaneous noise power calculation means for calculating an instantaneous noise power of the output reception value based on the output reception value, the estimated transfer characteristic corresponding to the output reception value, and the estimated transmission value corresponding to the output reception value When,
Symbol direction filtering means for filtering the instantaneous noise power calculated by the first instantaneous noise power calculating means in a symbol time direction and outputting first temporary noise power for each carrier frequency;
The pilot reception value instantaneously based on the pilot reception value corresponding to the pilot carrier, the estimated transfer characteristic corresponding to the pilot reception value, and the known transmission value corresponding to the pilot reception value among the output reception values Second instantaneous noise power calculating means for calculating noise power;
Noise power correction means for correcting the first temporary noise power based on the instantaneous noise power of the pilot reception value calculated by the second instantaneous noise power calculation means and outputting the estimated noise power. A diversity receiving apparatus as a feature. The noise power correction means filters the instantaneous noise power calculated by the second instantaneous noise power calculation means in a carrier frequency direction and outputs a second temporary noise power for each carrier frequency; and
The diversity receiving apparatus according to claim 1, further comprising: a noise power correction unit that corrects the first temporary noise power based on the second temporary noise power and outputs the estimated noise power.   The noise power correction unit sets the first temporary noise power as the estimated noise power when a difference value obtained by subtracting the first temporary noise power from the second temporary noise power is smaller than a predetermined margin; The diversity receiving apparatus according to claim 2, wherein when the difference value is equal to or greater than the predetermined margin, the second temporary noise power is set as the estimated noise power. The noise power correction means includes a first carrier direction filter that filters the instantaneous noise power calculated by the second instantaneous noise power calculation means in a carrier frequency direction and outputs a second temporary noise power for each carrier frequency;
A second carrier direction filter that filters the first temporary noise power in a carrier frequency direction and outputs a third temporary noise power for each carrier frequency;
The diversity reception apparatus according to claim 1, further comprising: a noise power correction unit that corrects the first temporary noise power based on the second and third temporary noise powers and outputs the estimated noise power. .   The noise power correction unit sets the first temporary noise power as the estimated noise power when a difference value obtained by subtracting the second temporary noise power from the third temporary noise power is smaller than a predetermined margin; When the difference value is equal to or greater than the predetermined margin, a difference value obtained by subtracting the third temporary noise power from the second temporary noise power is added to the first temporary noise power to calculate the estimated noise power. The diversity receiver according to claim 4. The noise power correction means includes a first carrier direction filter that filters the instantaneous noise power calculated by the second instantaneous noise power calculation means in a carrier frequency direction and outputs a second temporary noise power for each carrier frequency;
A second carrier direction filter that filters the first temporary noise power in a carrier frequency direction and outputs a third temporary noise power for each carrier frequency;
A first carrier direction averager that averages the first temporary noise power over all pilot carrier frequencies and outputs a first noise power average;
A second carrier direction averager that averages the instantaneous noise power calculated by the second instantaneous noise power calculating means and outputs a second noise power average;
A noise power correction unit that corrects the first temporary noise power based on the second and third temporary noise powers and the average of the first and second noise powers and outputs the estimated noise power. The diversity receiver according to claim 1, wherein:   The noise power correction unit sets the first temporary noise power as the estimated noise power when a difference value obtained by subtracting the first noise power average from the second noise power average is smaller than a predetermined margin; When the difference value is greater than or equal to the predetermined margin and the second temporary noise power is smaller than the third temporary noise power, the first temporary noise power is set as the estimated noise power, and the difference value is the predetermined noise power. A difference value obtained by subtracting the third temporary noise power from the second temporary noise power when the second temporary noise power is not less than a margin and the second temporary noise power is not less than the third temporary noise power is obtained as the first temporary noise. The diversity receiver according to claim 6, wherein the estimated noise power is calculated by adding to power. The noise power correction means includes a first carrier direction averager that averages the first temporary noise power over all pilot carrier frequencies and outputs a first noise power average;
A second carrier direction averager that averages the instantaneous noise power calculated by the second instantaneous noise power calculating means and outputs a second noise power average;
The diversity reception according to claim 1, further comprising: a noise power correction unit that corrects the first temporary noise power based on the first and second noise power averages and outputs the estimated noise power. apparatus.   The noise power correction unit sets the first temporary noise power as the estimated noise power when a difference value obtained by subtracting the first noise power average from the second noise power average is smaller than a predetermined margin; When the difference value is equal to or greater than the predetermined margin, a difference value obtained by subtracting the first noise power average from the second noise power average is added to the first temporary noise power to calculate the estimated noise power. The diversity receiver according to claim 8. A diversity reception method of a diversity reception apparatus for receiving and combining OFDM signals composed of a plurality of carriers including one or more pilot carriers for each symbol for each reception system,
A detection step of converting the received OFDM signal into an OFDM signal in a frequency domain for each reception system and outputting a reception value of each of the plurality of carriers;
A transmission path estimation step for estimating the transmission characteristic of the output reception value of the detection step for each reception system;
A noise power estimation step for estimating the noise power of the output reception value for each reception system based on the output reception value of the detection step, the estimated transmission characteristic estimated by the transmission path estimation step, and the estimated transmission value;
A combined equalization step of calculating a demodulated value by combining and equalizing the output reception value for each reception system, the estimated transfer characteristic by the transmission path estimation step, and the estimated noise power by the noise power estimation step;
A decoding step of decoding the demodulated value to restore a transmission information sequence;
A transmission value estimation step of estimating a transmission value from the demodulated value and outputting the estimated transmission value,
In the noise power estimation step, for each reception system,
A first instantaneous noise power calculation step of calculating an instantaneous noise power of the output reception value based on the output reception value, the estimated transfer characteristic corresponding to the output reception value, and the estimated transmission value corresponding to the output reception value When,
A symbol direction filtering step of filtering the instantaneous noise power calculated in the first instantaneous noise power calculation step in a symbol time direction and outputting a first temporary noise power for each carrier frequency;
The pilot reception value instantaneously based on the pilot reception value corresponding to the pilot carrier, the estimated transfer characteristic corresponding to the pilot reception value, and the known transmission value corresponding to the pilot reception value among the output reception values A second instantaneous noise power calculating step for calculating noise power;
A noise power correcting step of correcting the first temporary noise power based on the instantaneous noise power of the pilot reception value calculated by the second instantaneous noise power calculating step and outputting the estimated noise power. A feature of diversity reception.

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