JP5547684B2 - Bandgap reference circuit - Google Patents

Description

  The present invention relates to a band gap reference circuit.

Conventionally, as a band gap reference circuit, for example, the one shown in FIG. 11 is known.
The bandgap reference circuit shown in FIG. 11 includes transistors Q1 and Q2 composed of PNP transistors, an operational amplifier OP1, transistors M1 and M2 composed of P-type MOS transistors functioning as current sources, resistors R1, R2, and R3. It consists of

The emitter size of the transistor Q2 is N times the emitter size of the transistor Q1 (N is a positive integer, N> 1), and the transistor sizes of the transistors M1 and M2 are the same.
The transistor M1, the resistor R3, and the transistor Q1 are connected in series and connected between the power supply VDD and the ground. Similarly, the transistor M2, the resistor R2, the resistor R1, and the transistor Q2 are connected in series, and the power supply Connected between VDD and ground.

The operational amplifier OP1 controls the gate voltages of the transistors M1 and M2 based on the potential at the junction of the resistors R1 and R2 and the potential of the emitter of the transistor Q1, and operates so as to control the current flowing through the transistors M1 and M2. To do.
Here, if the current flowing through the transistor M1 is I2, and the current flowing through the transistor M2 is I1, the transistors M1 and M2 have the same transistor size, so I1 = I2.

When the currents I1 and I2 are supplied, the voltage VN1 at the node N1, which is a connection point between the emitter of the transistor Q1 and the resistor R3, and the voltage VN2 at the node N2, which is a junction point between the resistors R1 and R2, are respectively Can be expressed as
VN1 = VBE (Q1) (1)
VN2 = VBE (Q2) + R1 × I1 (2)
Here, VBE (Q1) is the base-emitter voltage of the transistor Q1, and VBE (Q2) is the base-emitter voltage of the transistor Q2.

The operational amplifier OP1 controls the gate voltages of the transistors M1 and M2 so that the voltages at the nodes N1 and N2 are the same, that is, controls the currents I1 and I2, so that VN1 = VN2 is finally obtained.
From equation (1) and equation (2), when solving for current I1, I1 can be expressed by equation (3) below.
I1 = (VBE (Q1) −VBE (Q2)) / R1 (3)

As can be seen from FIG. 11, since the same current as I1 expressed by the equation (3) flows through the resistor R2, the output voltage Vout of the output terminal OUT can be expressed by the following equation (4).
Vout = VBE (Q2) + (R1 + R2) × I1 (4)
Here, if ΔVBE = (VBE (Q1) −VBE (Q2)), the output voltage Vout can be expressed by the following equation (5) from the equations (3) and (4).
Vout = VBE (Q2) + ΔVBE × (1 + R2 / R1) (5)

If the base current is ignored, the VBE (Q1) and VBE (Q2) voltages can be expressed by the following equations (6) and (7).
VBE (Q1) = k × T / q × ln (I1 / Is1) (6)
VBE (Q2) = k × T / q × ln (I2 / Is2) (7)
In Equations (6) and (7), k is a Boltzmann constant, T is an absolute temperature, q is an elementary charge, and Is1 and Is2 are saturation currents of the transistors Q1 and Q2.

Since the saturation current is proportional to the emitter area, the relationship between Is1 and Is2 is expressed by the following equation (8).
Is2 = N × Is1 (8)
From these formulas (6), (7), and (8), ΔVBE can be expressed by the following formula (9).
ΔVBE = VBE (Q1) −VBE (Q2)
= K * T / q * ln [(I1 / Is1) * (Is2 / I2)]
= K * T / q * ln (N) (9)
As can be seen from Equation (9), ΔVBE in Equation (5) is a voltage proportional to absolute temperature having a positive slope.

Moreover, VBE (Q2) in Formula (5) can be represented by following Formula (10).
VBE (Q2) = Vg0− (Vg0−VBE (TR)) / TR × T
-(Γ-1) × (k × T / q) × ln (T / TR) (10)
In the equation (10), Vg0 is a silicon band gap at a temperature of 0 [K], VBE (TR) is an emitter-base voltage at room temperature, TR is room temperature, and γ is a constant determined by a doping level. is there.
In Expression (10), the second and third terms depend on the temperature, and have a negative slope with respect to the temperature.

In the conventional bandgap reference circuit shown in FIG. 11, the output voltage Vout is the base-emitter voltage VBE (Q2) of the transistor Q2 shown in the equation (10) as shown in the equation (5). And the sum of voltages obtained by multiplying the voltage ΔVBE shown in the equation (9) by (1 + R2 / R1).
Therefore, by appropriately selecting the ratio between the resistors R2 and R1, the primary temperature characteristic is canceled by canceling the negative temperature characteristic of the base-emitter voltage VBE (Q2) of the transistor Q2 using the positive temperature characteristic of ΔVBE. Temperature compensation.

  In addition to the bandgap reference circuit that performs the primary temperature compensation, the secondary temperature compensation is performed by canceling the secondary component of the bandgap voltage using a current having a secondary temperature characteristic, for example. A bandgap reference circuit configured as described above has also been proposed (see, for example, Patent Document 2).

JP-A-9-260597 JP 2009-59149 A

  However, as shown in FIG. 11 described above, in the method of performing temperature compensation of the base-emitter voltage VBE (Q2) of the transistor Q2 using the positive temperature characteristic of ΔVBE, The third term has a high-order temperature characteristic that cannot be canceled by the first-order temperature compensation by ΔVBE. Therefore, even if an optimum value is selected as the ratio R2 / R1 between the resistors R2 and R1, a temperature drift of about several mV remains, and a highly accurate reference voltage cannot be realized.

Further, when the band gap reference circuit configured to perform the second-order temperature compensation described above is used, the second-order component of the band-gap voltage can be canceled, but the voltage error of the third-order or higher-order component, that is, the temperature Drift remains and a highly accurate reference voltage cannot be realized.
Therefore, a band gap reference circuit capable of performing temperature compensation of higher order components has been desired.
Therefore, the present invention has been made paying attention to the above-mentioned conventional unsolved problems, and an object thereof is to provide a band gap reference circuit capable of realizing a reference voltage with higher accuracy with respect to temperature. It is said.

A bandgap reference circuit according to a first aspect of the present invention includes a compensated signal generator that generates a compensated signal having temperature characteristics, and a first to n-th order (where n is the temperature characteristic of the compensated signal). An n-order temperature compensation unit that compensates so as to cancel the temperature characteristic up to an integer of 3 or more), and the compensated signal after the temperature characteristic is compensated by the n-order temperature compensation unit as a band gap reference signal An output bandgap reference circuit, wherein the n-th order temperature compensation unit has a temperature characteristic for canceling a signal component having a temperature characteristic in each order up to the n-th order included in the compensated signal. It has a separate temperature compensation signal generating unit for each order component that generates a compensation signal, wherein a sum of the respective temperature compensation signal of the n th order that the temperature compensation signal generator has generated It is characterized in that superimposed on the compensation signal.

A bandgap reference circuit according to a second aspect includes a compensated signal generator for generating a compensated signal having temperature characteristics, and a first to nth order (n is 3 or more) of the temperature characteristics of the compensated signal. A band gap reference circuit that outputs the compensated signal after the temperature characteristic is compensated by the n-th order temperature compensation unit as a band gap reference signal. The n-th order temperature compensation unit generates each temperature compensation signal having a temperature characteristic for canceling a signal component having a temperature characteristic in each order up to the n-th order included in the compensated signal, the has become the sum of the temperature compensation signal of n th order to be superimposed on the object to be compensated signal, the object compensation signal is a the compensated voltage signal having a temperature characteristic, of the order n Degrees compensation signal is characterized in that said included in the compensated voltage signal, a n-order temperature compensation voltage signal having a temperature characteristic that cancels the voltage signal component having said n-order temperature characteristic.
A bandgap reference circuit according to a third aspect is the bandgap reference circuit according to the first aspect, wherein the compensated signal is a compensated current signal having temperature characteristics, and the nth-order temperature compensated signal is the compensated current signal. It is an n-th order temperature compensation current signal having a temperature characteristic that cancels out the current signal component having the n-th order temperature characteristic contained in the compensation current signal.

A band gap reference circuit according to a fourth aspect includes a compensated signal generation unit that generates a compensated signal having temperature characteristics, and a first to nth order (n is 3 or more) of the temperature characteristics of the compensated signal. A band gap reference circuit that outputs the compensated signal after the temperature characteristic is compensated by the n-th order temperature compensation unit as a band gap reference signal. The n-th order temperature compensation unit generates each temperature compensation signal having a temperature characteristic for canceling a signal component having a temperature characteristic in each order up to the n-th order included in the compensated signal, The sum of the temperature compensation signals up to the nth order is superimposed on the compensated signal, and the compensated signal is a compensated current signal having temperature characteristics, and the nth order Degrees compensation signal, the contained in a compensation current signal, an n-th temperature compensating current signal having a temperature characteristic that cancels the current signal component having said n-order temperature characteristic, the n-th temperature compensating section, the A current-voltage conversion unit that converts an nth-order temperature compensation current signal into a voltage signal is provided.
The band gap reference circuit according to claim 5 is characterized in that the current-voltage conversion unit is a resistor.
The band gap reference circuit according to claim 6 is characterized in that the resistance value of the resistor is configured to be variable.
According to a seventh aspect of the present invention, there is provided a bandgap reference circuit comprising storage means for storing a resistance value of the resistor.
The band gap reference circuit according to claim 8, wherein the nth-order temperature compensation unit generates a primary compensation signal having a temperature characteristic that cancels a signal component having a first-order temperature characteristic included in the compensated signal. The temperature compensation signal generator is provided.

A bandgap reference circuit according to a ninth aspect includes a compensated signal generator that generates a compensated signal having temperature characteristics, and a first to n-th order (n is 3 or more) of the temperature characteristics of the compensated signal. A band gap reference circuit that outputs the compensated signal after the temperature characteristic is compensated by the n-th order temperature compensation unit as a band gap reference signal. The n-th order temperature compensation unit generates each temperature compensation signal having a temperature characteristic for canceling a signal component having a temperature characteristic in each order up to the n-th order included in the compensated signal, The sum of the temperature compensation signals up to the nth order is superimposed on the compensated signal, and the nth order temperature compensation unit has a first-order temperature characteristic included in the compensated signal. Comprising a first order temperature compensation signal generator for generating a primary compensation signal having a temperature characteristic of canceling a signal component, the first order temperature compensation signal generator, different connection properties in parallel between the power supply voltage pair The primary compensation signal is generated using the difference between the base-emitter voltages of the pair of bipolar transistors.
The band gap reference circuit according to claim 10 is characterized in that the compensated signal generator uses a base-emitter voltage of one of the pair of bipolar transistors as the compensated signal. .

  According to the present invention, the temperature compensation signal is superimposed on the compensated signal by the nth order temperature compensation unit, and this temperature compensation signal has the temperature characteristics in the respective orders up to the nth order among the temperature characteristics of the compensated signal. Since this is the sum of the temperature characteristics for canceling the signal components, temperature compensation up to the nth order can be performed on the compensated signal. Therefore, it is possible to realize a bandgap reference circuit capable of realizing a reference voltage with higher accuracy with respect to temperature.

It is a schematic block diagram which shows an example of the band gap reference circuit in this invention. It is a circuit diagram which shows an example of the band gap reference circuit in this invention. (A) is a characteristic diagram showing the relationship between absolute temperature and temperature compensation current, (b) is a characteristic diagram showing the relationship between absolute temperature and temperature compensation voltage. It is an example of the amount of temperature drift when temperature compensation is not performed and when temperature compensation is performed. It is an example of the calculation error result in the case where the high-order characteristic of the temperature drift is expanded by an n-th order polynomial. It is a circuit diagram which shows an example of the temperature compensation electric current generation circuit which has the characteristic of a quadratic function with respect to temperature. It is a circuit diagram which shows an example of the temperature compensation electric current generation circuit which has the characteristic of a cubic function with respect to temperature. It is an example of the electric current which flows into the transistor which comprises a differential pair. (A) is an example of the current flowing through the transistor PA, and (b) is an example of the current flowing through the transistor PB. It is a circuit diagram showing an example of a band gap reference circuit when transistors Q1 and Q2 made of NPN transistors are used as transistors Q1 and Q2 constituting a differential pair. It is a circuit diagram which shows an example of the conventional band gap reference circuit.

Embodiments of the present invention will be described below with reference to the drawings.
FIG. 1 is a configuration diagram showing an example of a bandgap reference circuit of the present invention.
The band gap reference circuit 1 in FIG. 1 is proportional to an absolute temperature, a constant current source 101 that generates a constant current I1, a transistor Q101 composed of a PNP transistor connected to the constant current source 101, and an adder circuit 102. A VT generation circuit 103 that generates a VT voltage, a multiplier 104 that multiplies the VT voltage generated by the VT generation circuit 103, and an Nth order temperature that generates an Nth order temperature compensation signal for performing Nth order temperature compensation. A compensation signal generation circuit 105.

  The transistor Q101 is a diode-connected transistor so that the base and collector are short-circuited. The emitter of the transistor Q101 is connected to the constant current source 101, and the collector is grounded. The voltage at the connection point between the constant current source 101 and the emitter of the transistor Q101 is input to the adder circuit 2 as the base-emitter voltage VBE of the transistor Q101.

The VT generation circuit 103 generates a VT (VT = k × T / q) voltage. Here, k is a Boltzmann constant, T is an absolute temperature, and q is an elementary charge.
The base-emitter voltage VBE of the transistor Q101, the M × VT voltage from the multiplier 104, and the higher-order temperature compensation voltage Vcomp from the N-th order temperature compensation signal generation circuit 105 are input to the adder circuit 102, and these The sum of the voltages is output as an output voltage, that is, a reference voltage Vout (= VBE + M × VT + Vcomp).

  That is, in the above-described conventional band gap reference circuit (FIG. 11), VT (VT = VT = proportional to absolute temperature) using a VBE voltage of diode-connected transistor Q2 and having a negative slope with respect to temperature. k × T / q) voltage is applied to reduce the temperature drift. However, in the band gap reference circuit 1 shown in FIG. 1, in addition to this, a temperature compensation signal generation circuit having a higher-order temperature coefficient ( An Nth-order temperature compensation signal generation circuit 105) is provided, and a temperature drift is further reduced by further adding a higher-order temperature compensation voltage Vcomp.

FIG. 2 is a circuit diagram showing a specific configuration of the bandgap reference circuit 1 of FIG.
As shown in FIG. 2, the bandgap reference circuit 1 is similar to the conventional bandgap reference circuit shown in FIG. 11 in that transistors Q1 and Q2 made of PNP transistors, an operational amplifier OP1, and a P functioning as a current source. In addition to transistors M1 and M2 formed of type MOS transistors, resistors R1, R2A, R2B, and R3 and a temperature compensation current generation circuit 10 are further provided. The resistors R2A and R2B are configured to have variable resistance values.

The emitter size of the transistor Q2 is N times the emitter size of the transistor Q1 (N is a positive integer, N> 1), and the transistor sizes of the transistors M1 and M2 are the same. These transistors Q1 and Q2 are diode-connected so that the base and collector are short-circuited.
The source of the transistor M1 is connected to the power supply voltage VDD, the drain is connected to the emitter of the transistor Q1 via the resistor R3, and the collector is grounded. Similarly, the source of the transistor M2 is connected to the power supply voltage VDD, the drain of the transistor M2 is connected to the emitter of the transistor Q2 via resistors R2B, R2A, and R1, and the collector of the transistor Q2 is grounded.

The voltage VN1 at the node N1, which is the connection point between the resistor R3 and the emitter of the transistor Q1, is input to the inverting input terminal of the operational amplifier OP1, and the voltage VN2 at the node N2, which is the junction point between the resistors R2A and R1, is non-inverted from the operational amplifier OP1. Input to the input terminal. The operational amplifier OP1 controls the gate voltages of the transistors M1 and M2 so that the voltage VN1 of the node N1 and the voltage VN2 of the node N2 are the same, and generates a current I1 ′ flowing through the transistor M2 and a current I2 ′ flowing through the transistor M1. Control.
The temperature compensated current generation circuit 10 is connected to a node N3 that is a junction point of the resistors R2A and R2B. The voltage at the connection point between the transistor M2 and the resistor R2B is output from the output terminal OUT as the output voltage Vout.

Here, the voltage change with respect to the current change at the node N1 is sufficiently small with respect to the voltage change with respect to the current change at the node N2. As described above, the operational amplifier OP1 controls the current flowing through the transistors M1 and M2 so that the VN1 voltage at the node N1 and the VN2 voltage at the node N2 are the same. When the compensation current Icomp is drawn, the current flowing through the transistors M1 and M2 is controlled so as not to change the current flowing through the resistor R1. Therefore, the current I2 ′ flowing through the transistor M1 and the current I1 ′ flowing through the transistor M2 can be expressed by the following equation (11).
I1 ′ = I2 ′ = I1 + Icomp = I2 + Icomp (11)
In the equation (11), I1 and I2 are currents flowing through the transistors M1 and M2 when the temperature compensation current generation circuit 10 shown in FIG. 11 is not connected.

From Expression (11), the output voltage Vout generated at the output terminal OUT can be expressed by the following Expression (12).
Vout
= VBE (Q2) + I1 * R1 + I1 * R2A + (I1 + Icomp) * R2B
= VBE (Q2) + I1 × (R1 + R2A + R2B) + Icomp × R2B
(12)
Here, assuming that R2A + R2B = R2, the sum of the first term and the second term in the equation (12) is the same as that in the equation (4), that is, the conventional bandgap reference circuit shown in FIG. It is the same as the output voltage Vout.

In the expression (12), the third term is a high-order temperature compensation voltage Vcomp newly added to the output voltage Vout.
The relationship between the temperature compensation current Icomp drawn by the temperature compensation current generation circuit 10 and the higher-order temperature compensation voltage Vcomp is, for example, as shown in FIG. 3 in the case of secondary temperature compensation. That is, as shown in the third term of Expression (12), Vcomp = Icomp × R2B.

3A shows the correspondence between the absolute temperature T and the temperature compensation current Icomp, where the horizontal axis represents the absolute temperature T and the vertical axis represents the temperature compensation current Icomp. 3B shows the correspondence between the absolute temperature T and the higher-order temperature compensation voltage Vcomp, where the horizontal axis represents the absolute temperature T and the vertical axis represents the higher-order temperature compensation voltage Vcomp.
FIG. 4 shows the calculated value of the temperature drift amount of the output voltage Vout after temperature compensation when the temperature compensation voltage “Icomp × R2B” shown in the third term in the equation (12) has the characteristic of a cubic function with respect to the temperature. It represents.

  Calculation of this temperature drift amount was performed as follows. That is, as described in the conventional bandgap reference circuit shown in FIG. 11, the current I1 flowing through the transistor M2 in FIG. 11 is ΔVBE (= VBE (Q1) −VBE as shown in the equation (3). (Q2)) divided by the resistance R1, and as shown in the equation (9), ΔVBE is proportional to the absolute temperature T. Therefore, the current I1 flowing through the transistor M2 is theoretically proportional to the temperature, It can be approximated by a linear function related to temperature. Moreover, since the second or higher order component of the temperature drift is caused by VBE (Q2) expressed in the first term of Expression (12), it was calculated using Expression (10).

  In FIG. 4, (a) represents the temperature drift amount of VBE (Q2) when there is no temperature compensation. (B) shows the temperature drift amount resulting from VBE (Q2) after temperature compensation, and shows the temperature drift amount after primary compensation, after secondary compensation, and after tertiary compensation. FIG. 4C is an enlarged view of the temperature drift amount after the secondary compensation in FIG. 4B, and FIG. 4D is an enlarged view of the temperature drift amount after the third compensation in FIG. 4A to 4D, the horizontal axis represents temperature [° C.]. The vertical axis indicates the amount of temperature drift, and the unit is voltage [V] in FIG. 4A and voltage [μV] in FIGS. 4B to 4D.

  As shown in FIG. 4, when temperature compensation is not performed, the VBE (Q2) voltage has a temperature drift of about 300 [mV] from −40 ° C. to 90 ° C. as shown in FIG. 4A. Yes. By performing the primary temperature compensation, the temperature drift can be suppressed to about 1200 [μV] as shown in FIG. By performing the second-order temperature compensation, a temperature drift of 50 [μV] or less can be achieved as shown in FIG. Further, by performing the third-order compensation, a temperature drift of 10 [μV] or less can be achieved as shown in FIG. Therefore, if the third-order compensation is performed, a temperature drift amount of 10 ppm or less can be calculated.

  Here, in an actual circuit, the temperature characteristics fluctuate due to process variations in resistance values and transistor characteristics, mismatch between elements, and the like. In order to correct this variation, the resistors R2A and R2B in FIG. 2 are configured to have variable resistance values and store the adjusted resistance values. For example, a non-volatile memory is mounted as storage means in an IC on which the bandgap reference circuit 1 is mounted, and the adjusted resistance values of the resistors R2A and R2B are held by the non-volatile memory. Thereafter, the resistance values stored in the nonvolatile memory are set as the resistance values of the resistors R2A and R2B, so that the resistors R2A and R2B can be easily adjusted.

FIG. 4 can be proved mathematically as follows.
In the temperature characteristic of VBE (Q2) shown in the equation (10), the third term having a higher-order temperature characteristic is modified as follows.
− (Γ−1) × (k × T / q) × ln (T / TR)
= − (Γ−1) × (k × TR / q) × (T / TR) × ln (T / TR)
(13)
At a temperature (about −40 to 100 ° C.) that is an actual use range, the formula (13) can be developed as follows.

  As described above, (T / TR) × ln (T / TR) having higher-order components can be developed as a power of temperature T. For example, when calculation is performed using γ = 3 and k × TR / q = 26 mV, it is as shown in FIG. 5 and it is clear that third-order or higher temperature correction is necessary to generate a high-precision reference voltage. . In FIG. 5, the horizontal axis represents temperature [° C.], and the vertical axis represents temperature drift [μV].

Next, a specific circuit of the temperature compensation current generation circuit 10 will be described.
FIG. 6 shows an example of a temperature compensation current generating circuit having a quadratic function characteristic with respect to temperature. 10a is a case where the temperature compensation current Icomp is pulled out, 10a ′ is a case where the temperature compensation current Icomp is added. Circuit. Note that the temperature compensation current generation circuit 10a is capable of performing the secondary temperature compensation of the temperature compensation target according to the secondary temperature characteristic of the temperature compensation target, or the temperature compensation current generation circuit 10a or 10a ′. Select. The secondary temperature characteristics to be compensated for temperature can be recognized at the time of design.

  This temperature compensation current generation circuit 10a includes transistors BIP1 to BIP4 made of NPN transistors, a current source 12 that generates a current Iptat proportional to temperature, and a constant current source that generates a constant current Ic regardless of temperature. 13, an operational amplifier OP 11, and transistors MP 1 and MP 2 composed of P-type MOS transistors operating as current sources.

  The transistors BIP1 to BIP4 have the same characteristics and are diode-connected so that the base and collector are short-circuited, the collector of the transistor BIP1 and the emitter of the transistor BIP2 are connected, and the collector of the transistor BIP2 is connected via the current source 12 Connected to the power supply voltage VDD, the emitter of the transistor BIP1 is grounded. Similarly, the collector of the transistor BIP3 and the emitter of the transistor BIP4 are connected, the collector of the transistor BIP4 is connected to the power supply voltage VDD via the constant current source 13, and the emitter of the transistor BIP3 is grounded.

  The transistor MP1 is connected between the power supply voltage VDD and the connection point of the transistors BIP3 and BIP4, and the source of the transistor MP2 is connected to the power supply voltage VDD. The drain of the transistor MP2 is connected to a pair of transistors constituting a mirror circuit. This mirror circuit is composed of a transistor MN1 and a transistor MN2 made of N-type MOS transistors, and the drain voltage of the transistor MN1 is supplied to the gates of the transistors MN1 and MN2. The drain of the transistor MP2 and the drain of the transistor MN1 constituting the mirror circuit are connected, and the sources of these transistors MN1 and MN2 are grounded. The drain of the transistor MN2 is connected to the node N3 in FIG.

That is, the current flowing through the transistor MP1 becomes the temperature compensation current Icomp11, and the drain current corresponding to the temperature compensation current Icomp11 flowing through the transistor MP2 flows through the transistor MN2 as Iout by the mirror circuit composed of the transistors MN1 and MN2, and the current corresponding to the temperature compensation current Icomp11. The current is drawn.
The node voltage Vnin of the collector node NIN, which is a connection point between the current source 12 and the transistor BIP2, is input to the inverting input terminal of the operational amplifier OP11, and the node of the collector node PIN, which is the connection point between the constant current source 13 and the transistor BIP4. The voltage Vpin is input to the non-inverting input terminal of the operational amplifier 11. The output side of the operational amplifier 11 is connected to the gates of the transistors MP1 and MP2.

The operational amplifier OP11 operates to adjust the current Icomp11 flowing through the transistor MP1 so that the node voltage Vnin of the collector node NIN of the transistor BIP2 and the node voltage Vpin of the collector node PIN of the transistor BIP4 are the same.
On the other hand, the temperature compensation current generation circuit 10a ′ for superimposing the temperature compensation current has a configuration in which the transistors MN1 and MN2 constituting the mirror circuit are removed from FIG. 6A, as shown in FIG. 6B. And a current corresponding to the temperature compensation current Icomp11 flowing through the transistor MP2 is added to the node N3 as Iout.

Here, the node voltage Vnin of the collector node NIN is expressed by the following equation (15).
Vnin = 2 × k × T / q × ln (Iptat / Is)
= K × T / q × ln (Iptat / Is) ² (15)
The node voltage Vpin of the collector node PIN is expressed by the following equation (16).
Vpin = k × T / q × ln (Ic / Is)
+ K × T / q × ln ((Ic + Icomp11) / Is)
= K × T / q × ln [Ic × (Ic + Icomp11) / Is ² ]
...... (16)
In the equations (15) and (16), Is is the saturation current of the transistors BIP1 to BIP4 made of NPN transistors.

These node voltages Vnin and Vpin are finally stabilized in a state where Vnin = Vpin by the operational amplifier OP11. Therefore, from the equations (15) and (16), Icomp11 is expressed by the following equation (17).
(Iptat / Is) ² = Ic × (Ic + Icomp11) / Is ²
Icomp11 = (Iptat ² / Ic) −Ic (17)
As can be seen from equation (17), Icomp11 is proportional to the square of Iptat. That is, since Iptat is a current proportional to temperature as described above, Icomp11 is a current proportional to the square of temperature.
That is, it means that the temperature compensation current Icomp11 having the characteristics of the quadratic function can be generated.

FIG. 7 shows an example of a temperature compensated current generation circuit 10b having a cubic function characteristic with respect to temperature.
This temperature compensation current generating circuit 1b is connected to constant current sources 21 to 23 for generating a predetermined current, three sets of differential pairs H1 to H3 composed of NPN transistors, and each NPN transistor constituting the differential pair. Resistors R11a to 13b, transistors PA and PB composed of P-type MOS transistors operating as current sources, and transistors MPa and MPb composed of P-type MOS transistors constituting current mirror circuits with the transistors PA and PB, respectively. , And a transistor pair MN21 and MN22 made of an N-type MOS transistor constituting a current mirror circuit.

The differential pair H1 includes transistors BIP21a and BIP21b made of NPN transistors, and the emitter of the transistor BIP21a is grounded via the resistor R11a and the constant current source 21, and similarly, the emitter of the transistor BIP21b is the resistor R11b and the constant current. Grounded via source 21.
The differential pair H2 includes transistors BIP22a and BIP22b made of NPN transistors, and the emitter of the transistor BIP22a is grounded via the resistor R12a and the constant current source 22, and similarly, the emitter of the transistor BIP22b is the resistor R12b and the constant current. Grounded via source 22.

  The differential pair H3 includes transistors BIP23a and BIP23b made of NPN transistors, and the emitter of the transistor BIP23a is grounded via the resistor R13a and the constant current source 23. Similarly, the emitter of the transistor BIP23b is the resistor R13b and the constant current. Grounded via source 23. These transistors BIP21a to BIP23b have the same characteristics.

The collectors of one of the transistors BIP21a, BIP22a, and BIP23a constituting each differential pair H1 to H3 are commonly connected to the drain of the transistor PA, and the source of the transistor PA is connected to the power supply voltage VDD.
The collectors of the other transistors BIP21b, BIP22b, BIP23b constituting each differential pair H1 to H3 are commonly connected to the drain of the transistor PB, and the source of the transistor PB is connected to the power supply voltage VDD.

The transistor MPa and the transistor MN22 are connected in series, the source of the transistor MPa is connected to the power supply voltage VDD, and the source of the transistor MN22 is grounded.
Similarly, the transistor MPb and the transistor MN21 are connected in series, the source of the transistor MPb is connected to the power supply voltage VDD, and the source of the transistor MN21 is grounded.

A constant voltage Vref is individually input to the bases of one of the transistors BIP21b, BIP22a, and BIP23b constituting each differential pair H1 to H3, and the base of the other transistors BIP21a, BIP22b, and BIP23a is 1 to the temperature. A common voltage VT having the following temperature characteristics is input.
The drain voltage of the transistor PA is input to the gates of the transistors PA and MPa constituting the current mirror circuit. The drain voltage of the transistor PB is input to the gates of the transistors PB and MPb constituting the current mirror circuit. The drain voltage of the transistor MN21 is input to the gates of the transistors MN21 and MN22 constituting the current mirror circuit.

Then, the current flowing through the connection point between the transistor MPa and the transistor MN22 is output or extracted as the temperature compensation current Icomp21. That is, the difference current between the current I _A and the current I _B flowing through the transistor PA is output or extracted as the temperature compensation current Icomp21.
Here, when the voltage VT and the constant voltage Vref are VT = Vref, the currents flowing through the transistors constituting the differential pairs H1 to H3 are equal, and the flowing current is the amount of current flowing through the constant current sources 21 to 23. It becomes half.

On the other hand, when the voltage VT is sufficiently larger than the constant voltage Vref, all the currents flowing through the constant current sources 21 to 23 flow through the transistors BIP21a, BIP22b, and BIP23a to which the voltage VT is input, and the transistor BIP21b to which the constant voltage Vref is input. No current flows through the BIP 22a and the BIP 23b.
Conversely, when the constant voltage Vref is sufficiently larger than the voltage VT, all the currents flowing through the constant current sources 21 to 23 flow through the transistors BIP21a, BIP22b, and BIP23a to which the constant voltage Vref is input, and the voltage VT is input. No current flows through the transistors BIP21b, BIP22a, and BIP23b.

  For example, the voltage VT has a negative temperature characteristic, and as shown in FIG. 7, a constant voltage VrefH corresponding to a high temperature is input to the base of the transistor BIP21b of the differential pair H1, and the base of the transistor BIP22a of the differential pair H2 is equivalent to room temperature. When a constant voltage VrefL corresponding to a low temperature is input to the base of the transistor BIP23b of the differential pair H3, currents flowing through the transistors BIP21a to BIP23b are as shown in FIG.

8A shows the relationship between the input voltage V _B to the base of each transistor in the differential pair H1, the currents I _1A and I _1B flowing through the transistors in the differential pair H1, and the absolute temperature T. FIG. Is. Similarly, (b) shows the relationship between the input voltage V _B to the base of each transistor in the differential pair H2, the currents I _2A and I _2B flowing through the transistors in the differential pair H2, and the absolute temperature T. It is. Similarly, (c) shows the relationship between the input voltage V _B to the base of each transistor in the differential pair H3, the currents I _3A and I _3B flowing through each transistor of the differential pair H3, and the absolute temperature T. It is.

Here, assuming that the currents flowing through the transistors PA and PB are I _A and I _B , these I _A and I _B can be expressed by the following equations (18) and (19), respectively.
I _A = I _1A + I _2A + I _3A (18)
I _B = I _1B + I _2B + I _3B (19)
Therefore, from the current characteristics of FIG. 8 and the equations (18) and (19), it can be seen that these I _A and I _B have the temperature characteristics shown in FIG. In FIG. 9A, the horizontal axis represents the absolute temperature T, and the vertical axis represents the current I _A flowing through the transistor PA. In FIG. 9 (b), the horizontal axis represents the absolute temperature T, and the vertical axis represents the current I _B through transistor PB.

That is, as shown in FIG. 9, the currents I _A and I _B flowing through the transistors PA and PB have a shape having a cubic function with respect to the temperature. Therefore, by adjusting each constant voltage Vref input to each differential pair H1 to H3, each resistance value of the resistors R11a to R13b, and each current amount flowing through the constant current sources 21 to 23, FIG. ), A current for compensating the temperature characteristic of the temperature drift amount after the secondary compensation shown in (c) can be generated.

  Therefore, in FIG. 2, if the temperature compensation current generation circuit 10a having the characteristics of the quadratic function of FIG. 6 is used as the temperature compensation current generation circuit 10, the temperature characteristics of the temperature drift amounts up to the primary and secondary are compensated. A band gap reference circuit can be configured. Further, in FIG. 2, the temperature compensated current generator circuit 10 includes the temperature compensated current generator circuit 10a having the characteristics of the quadratic function shown in FIG. 6 and the temperature compensated current generator circuit 10b shown in FIG. A band gap reference circuit that compensates for the temperature characteristic of the drift amount can be configured.

  That is, as shown in FIG. 4 described above, the temperature drift amount of the output voltage Vout after the primary temperature compensation has a characteristic represented by a quadratic function that is convex upward. Further, the temperature compensation current Icomp11 generated by the temperature compensation current generation circuits 10a and 10a 'having the characteristics of the quadratic function in FIG. 6 is a quadratic function that protrudes downward and has a quadratic function characteristic with respect to the temperature. It has the characteristic represented by. Therefore, 10a ′ is used as the temperature compensation current generating circuit, and the temperature compensation current Icomp11 is extracted from the node N3. As a result, the secondary temperature drift amount of the output voltage Vout is canceled out by the temperature compensation current Icomp11 having the opposite characteristic, and secondary temperature compensation can be performed.

  Further, as shown in FIG. 4 described above, the temperature drift amount of the output voltage Vout after the second-order temperature compensation has a cubic function characteristic. Further, the temperature compensation current Icomp21 generated by the temperature compensation current generating circuit 10b having the characteristic of the cubic function of FIG. 7 has a characteristic represented by the cubic function as shown in FIG. Therefore, third-order temperature compensation of the temperature drift amount of the output voltage Vout can be performed by applying the temperature compensation current Icomp21 to the node N3 in FIG.

Therefore, by connecting these temperature compensation current generation circuits 10a and 10b in parallel to the node N3, it is possible to configure a bandgap reference circuit that compensates for temperature characteristics up to the first, second and third order.
In addition, selection of the order and individual adjustment of the gain setting of the temperature compensation current in each order can be easily performed.

That is, as shown in the equation (11), the current I1 ′ flowing through the transistors M1 and M2 after adding the temperature compensation current generation circuit 10 is I1 ′ = I1 + Icomp, and I1 is the primary temperature compensation current. Yes, Icomp is a high-order temperature compensation current. When the voltage generated at the OUT terminal by the primary temperature compensation current is Vout1st and the voltage generated at the OUT terminal by the high-order temperature compensation current Icomp is Voutcomp, these voltages Vout1st and Voutcomp are respectively expressed by the following equations (20) and (21 ).
Vout1st = I1 × (R1 + R2A + R2B) (20)
Voutcomp = Icomp × R2B (21)
Therefore, it can be seen from Equation (21) that Voutcomp can be adjusted by changing the resistance value of the resistor R2B.

At this time, if the resistance values of R2A and R2B are adjusted so as to keep “R2A + R2B” constant, and Vout1st is adjusted so as not to change, only Voutcomp can be changed.
On the other hand, it can be seen from the equation (20) that Vout1st can be adjusted by changing the resistance value of the resistor R2A, and that only Vout1st can be made without changing Voutcomp.

  As described above, according to the band gap reference circuit 1 of the present embodiment, the temperature compensation current generation circuit 10 is further added to the conventional band gap reference circuit shown in FIG. A bandgap reference circuit capable of generating a large reference voltage can be realized, and the temperature compensated current generation circuit 10 can be realized with a relatively simple configuration shown in FIGS. This can be easily realized without adding additional elements.

  Further, as shown in FIG. 2 and the above equation (11), temperature compensation currents Icomp11 and Icomp21 for compensating the secondary and tertiary temperature characteristics are superimposed on the current I1 subjected to the primary temperature compensation. Therefore, it can be realized by simply connecting the temperature compensated current generation circuit of each order to the node N3 in parallel, which involves a significant change to the conventional band gap reference circuit. And can be realized easily.

  In the above embodiment, as shown in FIG. 2, the case where the bandgap reference circuit 1 is formed using the transistors Q1 and Q2 made of PNP type transistors has been described. However, the present invention is not limited to this. As shown in FIG. 10, NPN transistors can be used as the transistors Q1 and Q2.

The present invention is not limited to the bandgap reference circuit 1 shown in FIG. 2, but can be applied to any bandgap reference circuit that can be represented by the concept of FIG.
Here, in the above embodiment, the constant current source 101 and the transistor Q101 correspond to the compensated signal generation unit, the temperature compensation current generation circuits 10, 10a, 10a ′, 10b correspond to the nth-order temperature compensation unit, and VT The generation circuit 103 corresponds to the primary compensation unit.

DESCRIPTION OF SYMBOLS 1 Band gap reference circuit 10 Temperature compensation current generation circuit 10a Secondary temperature compensation current generation circuit 10b Third order temperature compensation current generation circuit 101 Constant current source 102 Addition circuit 103 VT generation 105 Nth order temperature compensation signal generation circuit

Claims (10)

A compensated signal generator for generating a compensated signal having temperature characteristics;
Wherein and a n-th temperature compensating section n-th (n is the integer of 3 or more) for compensating so as to cancel the temperature characteristics to the primary of the temperature characteristic which the compensation signal has,
A band gap reference circuit for outputting the compensated signal after the temperature characteristic is compensated by the n-th order temperature compensation unit as a band gap reference signal;
The n-th temperature compensating unit, wherein the compensated signal is included in, for each order component generating each temperature compensation signal having a temperature characteristic for canceling the signal component having a temperature characteristic of the respective orders to the order n have a separate temperature compensation signal generation unit,
A bandgap reference circuit, wherein a sum of each temperature compensation signal up to the nth order generated by each temperature compensation signal generation unit is superimposed on the compensated signal. A compensated signal generator for generating a compensated signal having temperature characteristics;
The n-th order from the primary of the temperature characteristics with a compensated signal (n is an integer of 3 or more) with the n-th temperature compensating section to compensate for the temperature characteristics of the up and,
A band gap reference circuit for outputting the compensated signal after the temperature characteristic is compensated by the n-th order temperature compensation unit as a band gap reference signal;
The n-order temperature compensation unit generates each temperature compensation signal having a temperature characteristic for canceling a signal component having a temperature characteristic in each order up to the n-th order included in the compensated signal;
The sum of the temperature compensation signals up to the nth order is superimposed on the compensated signal ,
The compensated signal is a compensated voltage signal having temperature characteristics,
The n-th temperature compensation signal is an n-th temperature compensation voltage signal having a temperature characteristic that cancels a voltage signal component having the n-th temperature characteristic included in the compensated voltage signal. Gap reference circuit. The compensated signal is a compensated current signal having temperature characteristics,
The n-th temperature compensation signal is an n-th temperature compensation current signal having a temperature characteristic that cancels out a current signal component having the n-th temperature characteristic included in the compensated current signal. The band gap reference circuit according to Item 1. A compensated signal generator for generating a compensated signal having temperature characteristics;
The n-th order from the primary of the temperature characteristics with a compensated signal (n is an integer of 3 or more) with the n-th temperature compensating section to compensate for the temperature characteristics of the up and,
A band gap reference circuit for outputting the compensated signal after the temperature characteristic is compensated by the n-th order temperature compensation unit as a band gap reference signal;
The n-order temperature compensation unit generates each temperature compensation signal having a temperature characteristic for canceling a signal component having a temperature characteristic in each order up to the n-th order included in the compensated signal;
The sum of the temperature compensation signals up to the nth order is superimposed on the compensated signal ,
The compensated signal is a compensated current signal having temperature characteristics,
The nth order temperature compensation signal is an nth order temperature compensation current signal having a temperature characteristic that cancels out a current signal component having the nth order temperature characteristic included in the compensated current signal,
The bandgap reference circuit, wherein the nth-order temperature compensation unit includes a current-voltage conversion unit that converts the nth-order temperature compensation current signal into a voltage signal .   5. The bandgap reference circuit according to claim 4, wherein the current-voltage conversion unit is a resistor.   6. The bandgap reference circuit according to claim 5, wherein the resistance value of the resistor is configured to be variable.   7. The band gap reference circuit according to claim 6, further comprising storage means for storing a resistance value of the resistor. The n-order temperature compensation unit includes a primary temperature compensation signal generation unit that generates a primary compensation signal having a temperature characteristic that cancels a signal component having a primary temperature characteristic included in the compensated signal. The band gap reference circuit according to any one of claims 1 to 7. A compensated signal generator for generating a compensated signal having temperature characteristics;
The n-th order from the primary of the temperature characteristics with a compensated signal (n is an integer of 3 or more) with the n-th temperature compensating section to compensate for the temperature characteristics of the up and,
A band gap reference circuit for outputting the compensated signal after the temperature characteristic is compensated by the n-th order temperature compensation unit as a band gap reference signal;
The n-order temperature compensation unit generates each temperature compensation signal having a temperature characteristic for canceling a signal component having a temperature characteristic in each order up to the n-th order included in the compensated signal;
The sum of the temperature compensation signals up to the nth order is superimposed on the compensated signal ,
The n-order temperature compensation unit includes a primary temperature compensation signal generation unit that generates a primary compensation signal having a temperature characteristic that cancels a signal component having a primary temperature characteristic included in the compensated signal,
The primary temperature compensation signal generator includes a pair of bipolar transistors having different characteristics connected in parallel between power supply voltages,
A band gap reference circuit, wherein the primary compensation signal is generated by utilizing a difference between a base-emitter voltage of the pair of bipolar transistors .   10. The band gap reference circuit according to claim 9, wherein the compensated signal generator uses a base-emitter voltage of one of the pair of bipolar transistors as the compensated signal.

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