JP4475528B2 - Synchronous motor control device and adjustment method thereof - Google Patents

Description

  The present invention relates to an AC synchronous motor control device and an adjustment method thereof, and more particularly, to a control device having a function of measuring / adjusting its electric constant without using a sensor that detects the rotational speed and rotor position of the synchronous motor, and It relates to the adjustment method.

  Patent Document 1 discloses a technique for automatically measuring an electric constant of an AC motor using a control device such as an inverter without using a motor rotation speed sensor and a rotor position sensor. In this technique, a DC voltage or an AC voltage is applied on the dq coordinate, and an electric constant of the AC motor is measured. Patent Document 2 discloses a technique for measuring an electrical constant by passing a direct current component to an arbitrary phase, fixing a rotor position, and superimposing an alternating current thereon, with respect to a synchronous motor. . Further, Patent Document 3 discloses a technique in which a pulsed voltage is applied to an electric motor for a minute period and an electric constant is measured from a transient response of current.

JP 60-183953 A JP 2000-50700 A JP 2001-67983 A

  In Patent Document 1, an induction motor is an object, and in particular, it cannot be applied to a synchronous motor having saliency.

  In Patent Document 2, an attempt is made to fix the rotor by flowing a direct current in the direction of the magnetic flux of the magnet. However, when alternating current is applied in the direction of the torque axis that is orthogonal to the direct current, the effect of direct current is not obtained. Vibrates and an error occurs in the measured value. Further, the direct current for fixing the position is a large current before and after the rating, and continuing to supply this places stress on the semiconductor device of the inverter and significantly reduces the life of the apparatus.

  Furthermore, in Patent Document 3, since a pulsed minute voltage is applied, the conditions are different from those in actual operation, and an error may be included in the measured value. Synchronous motors in recent years have a variety of rotor structures such as surface magnets, embedded magnets, or special shapes that take reluctance torque into account. There are various types of motors such as the number of slots and the number of poles. For this reason, there is a high possibility that the electromagnetic phenomenon at the time of applying the pulsed voltage is different from the phenomenon at the time of applying the sine wave voltage during normal driving, and there remains a problem in the accuracy of the measured value.

  In recent years, synchronous motors have been used for home appliances such as air conditioners and refrigerators, and automatic measurement / adjustment of electrical constants is also required for these devices. In this case, of course, neither a rotational speed sensor nor a position sensor is used, and the motor phase current sensor is often not used.

  SUMMARY OF THE INVENTION An object of the present invention is to provide a synchronous motor control device or an adjustment method thereof capable of measuring or automatically adjusting an electric constant with high accuracy in a short time without depending on the stator or rotor structure of the motor. is there.

  Another object of the present invention is to provide a synchronous motor control device or an adjustment method thereof capable of measuring an electric constant or adjusting automatically with high accuracy in a short time without mechanically fixing a rotor. .

  Still another object of the present invention is to provide a synchronous motor control device or an adjustment method thereof capable of automatically adjusting an electric constant without imposing a burden on an inverter.

  In one aspect of the present invention, the electric constant measurement or control system of the synchronous motor is supplied to the synchronous motor from the inverter while supplying a three-phase unbalanced alternating current having a frequency of 25% or more of the maximum drive frequency of the synchronous motor. It is characterized in that adjustment is performed.

  In the desirable embodiment, the frequency of the three-phase unbalanced alternating current is set to be equal to or higher than the maximum driving frequency of the synchronous motor.

  In another aspect of the present invention, the electric constant of a synchronous motor is measured or controlled while a three-phase unbalanced alternating current having a frequency of 25% or more of the maximum drive frequency of the synchronous motor is supplied from the inverter to the synchronous motor. In adjusting the system, there is provided means for setting the frequency of the three-phase unbalanced alternating current.

  In the desirable embodiment, it comprises an operation means for setting and inputting the frequency of the three-phase unbalanced alternating current from the outside, or a control means for converging to an appropriate frequency while changing the output frequency of the inverter.

  In still another aspect of the present invention, prior to the supply of the three-phase unbalanced alternating current, direct current is supplied from the inverter to the synchronous motor, and the rotor position is made to coincide with the magnetic pole axis.

  According to a preferred embodiment of the present invention, an electric motor control device capable of automatically adjusting an electric constant of an electric motor with a simple control configuration without using a position sensor for detecting a rotor position of an AC synchronous electric motor or an adjustment method thereof Can be realized.

  Other objects and features of the present invention will be clarified in the embodiments described below.

  Next, with reference to FIGS. 1-18, embodiment of the control apparatus of the synchronous motor by this invention is described. In the following embodiments, a permanent magnet type synchronous motor (hereinafter abbreviated as PM motor) will be described as an electric motor. However, the same applies to other synchronous motors such as a wound type synchronous motor and a reluctance motor. Is feasible.

Embodiment 1:
FIG. 1 is a block diagram of a synchronous motor control apparatus according to Embodiment 1 of the present invention. The control apparatus according to the first embodiment includes a rotation speed command generator 1 that generates a rotation speed command ωr * of an electric motor and an inverter controller 2. The inverter controller 2 calculates an AC voltage to be output by the inverter 3, converts it to a pulse width modulation (PWM) signal, and outputs the pulse width modulation (PWM) signal to the inverter 3. Next, as a main circuit, a DC power supply 4 for supplying power to the inverter 3 and a permanent magnet type synchronous motor 5 (hereinafter abbreviated as PM motor) to be controlled are provided. The DC power source 4 includes an AC power source 41, a diode bridge 42 constituting a rectifier circuit, and a smoothing capacitor 43, and generates a DC voltage V 0 at both ends of the smoothing capacitor 43. A direct current path from the direct current power source 4 to the inverter 3 is provided with a current detector 6 for detecting the direct current IDC.

  In the controller 2, the current reproducer 7 receives the detection current IDC and reproduces the three-phase alternating currents Iu, Iv, Iw flowing through the PM motor 5 by calculation. The reproduced three-phase AC currents Iuc, Ivc, and Iwc are coordinate-converted by the coordinate converter 8 into components Idc and Iqc on the axes d and q according to the phase angle θdc of the PM motor assumed inside the controller. Is done. The voltage command calculator 9 calculates voltage commands Vdc * and Vqc * for normally driving the PM motor 5 based on the speed command ωr * and the detected current values Idc and Iqc. The dq inverse converter 10 converts the voltage commands Vdc * and Vqc * into three-phase AC voltage commands Vu *, Vv * and Vw *. The PWM pulse generator 11 generates a pulse width modulation (PWM) signal for switching control of the inverter 3 based on the three-phase AC voltage commands Vu *, Vv *, Vw *. The voltage command switching unit 12 switches the voltage command between the normal drive mode and the PM motor automatic electric constant automatic measurement / adjustment mode. Similarly, the phase command switching unit 13 switches the phase command. The measurement frequency setting unit 14 sets the AC frequency ω1t for measurement in the electrical constant automatic measurement / adjustment mode. The motor constant automatic adjuster 15 receives the frequency ω1t and generates applied voltage commands Vdc * and Vqc * necessary for measuring the electric constant, and at the same time, the electric constants Ld and Lq of the PM motor 5 from the detected currents Idc and Iqc. Is calculated and output. The zero phase command generator 16 serves to fix the phase inside the controller to zero at the time of automatic measurement / adjustment of electrical constants. Finally, the mode switch 17 switches the operation mode of the controller 2 between the normal drive and the electrical constant automatic measurement mode in accordance with an external command.

  Next, the operation principle of the first embodiment will be described with reference to FIG.

  In this embodiment, the operation mode of the controller 2 includes two modes, a normal drive mode of the PM motor 5 and an electric constant automatic measurement mode, which are switched by a signal from the mode switch 17. During normal driving, the voltage command switching unit 12 and the phase command switching unit 13 are switched to the “0” side in the state shown in the figure, and are switched to the “1” side in the automatic measurement mode.

  First, the normal drive mode will be described. The rotation speed command generator 1 gives a motor rotation speed command ωr * to the voltage command calculator 9 by communication means such as digital or analog. In the voltage command calculator 9, the voltage commands Vdc * and Vqc * necessary for driving the PM motor and the AC phase θdc of the voltage applied to the PM motor 5 are determined based on the rotational speed command ωr * and the detected currents Idc and Iqc. Perform the operation.

  The current reproducer 7 reproduces the three-phase alternating current of the PM motor by calculation based on the power supply current IDC and the PWM signal detected by the current detector 6 by the method described in Japanese Patent Application Laid-Open No. 8-19263. Next, the coordinate converter 8 converts the reproduced alternating currents Iuc, Ivc, and Iwc into current components Idc and Iqc on the rotational coordinate axis (dq axis) that rotates at the angular frequency ω1 * based on the alternating phase θdc. . On the other hand, Vdc * and Vqc * are converted back into an AC amount by the dq coordinate inverse converter 10, and further converted into a pulse width modulated wave signal by the PWM pulse generator 11 and sent to the inverter 3. These basic operations are the same as those described in JP-A-2002-272194.

  Next, the operation in the automatic measurement mode, which is a feature of the present invention, will be described.

  In the automatic measurement mode, an alternating frequency that changes at the angular frequency ω1t set in the measurement frequency setting unit 14 is applied to Vdc * and Vqc *, and an alternating current flows through the PM motor 5. From this relationship between current and voltage, the d-axis inductance Ld and the q-axis inductance Lq of the PM motor 5 are calculated. These operations will be described using the processing flow of FIG. 2 and the phase current waveform of the PM motor of FIG.

  FIG. 2 is an automatic measurement processing flowchart of the synchronous motor control device according to the first embodiment. First, in step 201, the measurement frequency ω1t is set to zero, and in step 202, the DC voltage Vd is gradually applied to the Vdc * input terminal of the PM motor.

  FIG. 3 is a motor current waveform diagram in the synchronous motor control apparatus according to the first embodiment. First, as shown in the period (1) of FIG. 3, the voltage Vdc * is increased in a ramp shape so that the direct currents Iu to Iw are gradually increased. By doing so, the rotor position of the PM motor 5 coincides with the dc axis (magnet magnetic flux axis) for control.

  Next, in step 203, the measurement frequency ω1t is set to the electrical angular frequency ω1max that matches the maximum rotational speed used for normal driving of the PM motor 5. In step 204, the three-phase unbalanced AC voltage is applied to the PM motor 5 by setting the d-axis voltage command Vdc * to a simple sine wave. In step 205, the d-axis inductance Ld is calculated based on the current value detected at this time.

  Next, in step 206, the d-axis voltage command Vdc * is set to zero, the q-axis voltage command Vqc * is set as a sine wave, and a three-phase unbalanced AC voltage is applied to the PM motor. In step 207, similarly, the q-axis inductance Lq is calculated based on the detected current value.

  The above steps 204 to 207 are the motor constant automatic measurement mode period (2) in which the three-phase unbalanced alternating current in FIG. 3 is applied, and the current waveform of the PM motor 5 is the three-phase unbalanced as shown in the figure. AC current.

  Thereafter, the normal drive mode period (3) is entered, and three-phase balanced alternating current is supplied from the inverter 3 to the PM motor 5 to drive the PM motor 5.

  What is important in this embodiment is the setting of the measurement frequency ω1t. Even if an AC voltage is applied on the d-axis, no torque is generated in the PM motor 5, but basically a torque is generated for the current on the q-axis. However, by setting the measurement frequency ω1t to the same frequency (ω1t = ω1max) as the maximum speed frequency ω1max of the PM motor 5, the fluctuation of the rotor at the time of measuring the q-axis inductance Lq is suppressed. This principle will be described with reference to FIG.

  FIG. 4 is a block diagram showing a transfer function from the torque current to the rotor phase in the AC motor control apparatus according to Embodiment 1 of the present invention. FIG. 6A shows the transfer function from the torque current Iq to the phase change of the PM motor. The relationship between Iq and the phase angle θdc is expressed by a function such as the number of poles P of the motor, the magnetic flux Φ, and the inertia J of the mechanical system. Symbols in the figure mean Tm: generated torque, ωr: rotational speed of PM motor, ω1: electrical angular frequency. Here, assuming that Iq is an alternating component ΔIq that varies, the transfer function is as shown in FIG. It is assumed that the phase angle varies with ΔIq, and this magnitude is Δθ. Then, as shown in FIG. 5C, Δθ is inversely proportional to the square of ω1t.

  That is, it can be seen that in order to measure the inductance in a fixed state without changing the rotor as much as possible, it is better to increase the AC frequency during measurement as much as possible. However, at a frequency far from that during normal driving, the influence of the frequency characteristics of the PM motor appears, which may increase the measurement error, and it is not preferable to increase it unnecessarily. In the case of a commercially available motor for an air conditioner, the motor constant could be measured with satisfactory accuracy even at a measurement frequency of 25% of the maximum rotation frequency of the motor.

  In the first embodiment, the measurement frequency ω1t is set to the maximum drive frequency ω1max of the PM motor. As described above, when the measurement frequency ω1t is equal to or higher than the maximum drive frequency ω1max of the PM motor, the motor constant can be measured / adjusted with sufficient accuracy unless the motor is a special purpose motor. Considering the magnetic circuit characteristics of the PM motor, there is no big problem even if it is higher by about a few percent than the maximum speed frequency.

  Further, when the use of the PM motor is limited to a certain extent, such as a refrigerator or an air conditioner, the inertia J of the mechanical system can be assumed to some extent from the capacity and the rotational speed. In this case, the fluctuation range of the phase angle θ is defined to be within ± 5 degrees, for example, and the measurement frequency ω1t can be set from the equation (1) as shown in FIG.

  At this time, the necessary magnet magnetic flux Φ can also be set as a common-sense value if the PM motor capacity, rated voltage, and rotational speed are known.

  Furthermore, in the first embodiment, the direct current is increased in a ramp shape to align the rotor. However, if the load is highly viscous, such as a hydraulic pump at low temperatures, a stepped direct current can be applied. No problem. Furthermore, even if the direct current is not energized, it is not necessary to energize the direct current if the d-axis position is known. For example, if an initial position estimation method as shown in JP-A-2002-078392 is used, direct current energization is unnecessary.

  As described above, according to the first embodiment of the present invention, it is possible to accurately measure / adjust the inductance of the electric constant without fixing the rotor of the PM motor. In particular, it is not necessary to continue the direct current during the measurement, the burden on the inverter component device due to the continuous direct current can be reduced, and the failure of the inverter can be prevented.

  In the present embodiment (and the subsequent embodiments), a method for detecting the direct current of the inverter is used as a current detection method. However, a phase current detector for a PM motor is provided, and a direct phase current is detected. Is detected, the current reproducer 7 becomes unnecessary.

Embodiment 2:
Next, Embodiment 2 by this invention is demonstrated using FIGS.

  In the first embodiment, the measurement frequency in the automatic measurement mode is matched with the maximum drive frequency (maximum electrical angular frequency) of the PM motor. At this time, how much the rotor of the PM motor fluctuates and how much error occurs in the measurement result depends on the inertia J of the mechanical system. Therefore, in the second embodiment, a control device that takes this into consideration is provided.

  FIG. 5 is a partial view of a control block of the synchronous motor control device according to Embodiment 2 of the present invention. Only the relationship between the measurement frequency setter 14B and the motor constant automatic adjuster 15B in this embodiment is shown, and this is used in place of 14 and 15 in FIG. The operation will be described using the processing flow of FIG. 6 and the waveforms of FIG.

  FIG. 6 is an automatic measurement process flow of the synchronous motor control device according to the second embodiment of the present invention.

  FIG. 7 is a graph showing the change over time in the frequency of measurement AC, the current waveform of the motor, and the inductance measurement result in the synchronous motor control apparatus according to Embodiment 2 of the present invention.

  First, in FIG. 6, steps 201 to 205 are the same as the steps with the same reference numerals in FIG. 2, and the inductance Ld is measured.

  In step 601, the measurement frequency ω1t is set to an arbitrary frequency within the drive frequency range of the PM motor 5, the d-axis voltage command Vdc * is set to zero, and the q-axis voltage command Vqc * is set to a simple sine wave. Thus, a three-phase unbalanced AC voltage is applied to the PM motor 5. In step 602, the q-axis inductance Lq is calculated based on the current value detected at this time. This measurement result is stored in step 603 as the first Lq value Lq0 in FIG. Next, in step 604, the measurement frequency ω1t is slightly increased, and in step 605, the q-axis inductance Lq is calculated based on the detected current value. In step 606, the inductance Lq obtained this time is compared with the previously stored inductance Lq0. If the difference ΔLq is large, for example, if it exceeds 5%, the process returns to step 603 and this operation is repeated. By repeating this, the frequency for measurement is increased and the value of Lq is converged. During this time, the phase current of the PM motor 5 gradually increases in frequency as shown in FIG. If the difference between the value obtained last time and the value obtained this time is within a predetermined range of, for example, 5%, it may be considered that there is no frequency dependence, that is, the rotor fluctuation is not affected. it can. Accordingly, the process proceeds to step 607, where the value obtained this time is determined as the q-axis inductance Lq, and the measurement is terminated as shown in FIG. 7C.

  The frequency used for measurement is preferably as close to the actual operating conditions as possible. However, when the rotor vibrates, the measurement value of Lq includes the influence of Ld, and the error increases. This is greatly improved by adopting this embodiment.

Embodiment 3:
Next, Embodiment 3 by this invention is demonstrated using FIGS. 8-9.

  As shown in the first and second embodiments, in order to measure the electrical constant, it is necessary to flow alternating current through the electric motor. However, if an alternating current close to the rated current is flowed at once, it may become a trigger and the rotor may start to vibrate.

  FIG. 8 is a partial control block diagram of a synchronous motor control apparatus according to Embodiment 3 of the present invention that solves this problem. By using the motor constant automatic adjuster 15C shown in FIG. 8 instead of the motor constant automatic adjuster 15 shown in FIG.

  In FIG. 8, the motor constant automatic adjuster 15 </ b> C includes an Ld measuring device 18 and an Lq measuring device 19. Both internal configurations are the same, and only the Ld measuring device 18 shows a specific configuration. First, a measurement voltage amplitude command unit 20 that determines the amplitude of the applied voltage during constant measurement and a rate limiter 21 that limits the rate of change of the command are provided. Next, an AC function generator 22 that receives a measurement frequency ω1t and outputs a sinusoidal function, and a multiplier 23 that calculates the product of the output of the AC function generator 22 and the output of the rate limiter 21 are provided. I have. Furthermore, an Ld (or Lq) calculator 24 that calculates an inductance Ld (or Lq) based on the detected current Idc (or Iqc) is provided.

  FIG. 9 is a motor current waveform diagram in the electrical constant measurement mode in the synchronous motor control apparatus according to Embodiment 3 of the present invention.

  Next, the operation of FIG. 8 will be described with reference to the motor current waveform of FIG. The measurement voltage is applied in the order of d-axis → q-axis, and these operations are programmed in the measurement voltage amplitude command unit 20. A stepped voltage is output from the measurement voltage amplitude commander 20, but the rate of change is limited by the rate limiter 21. An amplitude command with a limited rate of change and a sine wave function output from the AC function generator 22 are multiplied by a multiplier 23 to obtain Vdc * (or Vqc *), and an unbalanced three for measurement corresponding thereto is obtained. Phase alternating current is applied to the PM motor. Since the applied voltage is an alternating current whose amplitude value gradually increases, the current value also gradually increases as shown in FIG. As a result, the influence as a disturbance is reduced and no vibration is generated. Also, at the end of measurement, the voltage amplitude is gradually reduced to prevent motor vibration.

  As described above, according to the third embodiment, it is possible to gradually increase / decrease the amplitude of alternating current for measuring constants such as inductance, thereby further effectively preventing the vibration of the rotor. Note that when measuring the d-axis inductance, the rate limiting effect is small, and therefore the rate limiting can be used only when measuring the q-axis inductance.

Embodiment 4:
Next, Embodiment 4 according to the present invention will be described with reference to FIG.

  As shown in the first to third embodiments, in order to measure the electrical constant, it is necessary to measure the motor constant from the application of the alternating voltage and the detected value of the alternating current associated therewith. In that case, it is necessary to analyze the relationship between the applied voltage and the detected current, including the phase information. Usually, it is common to introduce Fourier series expansion. However, accurately realizing the Fourier series expansion by numerical calculation processing increases the load of software processing and is problematic. In particular, when the measurement frequency ω1t is changed according to the motor rating, it is necessary to appropriately set the step period, the integration period, and the like. Therefore, the present embodiment provides a method for simultaneously obtaining an AC voltage waveform and analyzing a component of a detected current by a simple method to simultaneously obtain an inductance and a resistance component.

  FIG. 10 is a functional block diagram of the Ld measuring device of the synchronous motor control device according to the fourth embodiment of the present invention. By preparing an Lq measuring device having the same configuration as the Ld measuring device 18D and using it in the previous Embodiments 1 to 3, a highly accurate constant automatic adjusting device can be realized.

  The Ld measuring device 18D shown in FIG. 10 includes the same measurement voltage amplitude commander 20, a rate limiter 21 and a multiplier 23 as in FIG. 8, and an RL calculator 25 for calculating the resistance R and inductance L of the PM motor 5. . The RL calculator 25 integrates the measurement frequency ω1t and outputs a phase angle θ1t, and a sine wave and cosine wave generator that outputs sine and cosine waves based on the measurement frequency θ1t, respectively. 27 and 28 are provided. Also provided are multipliers 29 and 30 for multiplying the sine / cosine waveform and the current detection value Idc, and primary delay filters 31 and 32 for inputting these products and cutting the pulsation component. Furthermore, a constant calculator 33 for calculating the electric constant Ld (or Lq) of the electric motor based on the measurement voltage amplitude command Vt, the outputs of the first-order lag filters 31 and 32, and the measurement frequency ω1t is provided. .

  Next, the operation of the Ld measuring device 18D will be described.

  The measurement voltage amplitude command Vt is created in the same manner as the operation of the third embodiment. Based on the product of this Vt and the output of the cosine wave generator 28, Vdc * is expressed by equation (2), and an AC voltage is applied to the PM motor 5.

  On the other hand, the detection current Idc has a delay phase with respect to Vdc *, and therefore, the expression (3) is established.

  Equation (3) can be replaced as equation (4).

  When this Idc is multiplied by, for example, a sine wave function, the equation (5) is obtained, which is separated into a direct current component and a double pulsation component.

  If only a direct current component is output to this waveform via the first-order lag filter 31 in FIG. 10, Idsin (sine wave component) is obtained.

  Similarly, Idcos (cosine wave component) can be extracted by multiplying the cosine wave by the equation (4) and passing through the filter 32. At this time, if the time constant TLF of the first-order lag filters 31 and 32 is set to a certain level or more, there is no significant problem even if the frequency ω1t is variable. Therefore, it can be easily realized without using processing such as Fourier series expansion or FFT.

  Therefore, the outputs of the two first-order lag filters 31 and 32 are a sine wave component Idsin and a cosine wave component Idcos of Idc, respectively. Since a cosine wave is given as Vdc *, Ld and R are obtained by the constant calculator 33 from the relationship (6).

  In this embodiment, the d-axis electric constant is described as an example, but Lq and R can be calculated in the same manner with respect to the q-axis.

  As described above, according to the fourth embodiment of the present invention, it is possible to separate the components of the current flowing by the AC voltage with a simple configuration and obtain the electric constant of the electric motor.

Embodiment 5:
Next, Embodiment 5 according to the present invention will be described with reference to FIG.

  In the fourth embodiment described above, in order to measure the electrical constant, an alternating voltage is applied, and the detected value of the alternating current is separated into an in-phase component of the alternating voltage and a component different in phase by 90 degrees, and the electrical constant is calculated It was a thing.

  As an electric constant of an electric motor, it is known that an inductance is particularly susceptible to magnetic saturation, and the value varies depending on a measured current value. For this reason, it is better to measure the electrical constant by applying a current value of a predetermined value, for example, a motor rated current, instead of applying a voltage.

  According to this embodiment, it is possible to automatically measure the electrical constant using the current value as a parameter.

  FIG. 11 is a functional block diagram of the Ld measuring device 18E in the synchronous motor control apparatus according to Embodiment 5 of the present invention. By using this embodiment instead of the Ld measuring device 18D in the above-described embodiment, an automatic constant adjustment device using the current value as a parameter can be realized.

  In FIG. 11, the magnitude of the current command is set by the measurement current amplitude command unit 34. In this embodiment, the rate limiter is not added, but it can be added as in the previous examples. In the present embodiment, adder / subtractors 35 and 36 for adding or subtracting signals, integral compensators 37 and 38, a constant calculator 39, an adder 40, and multipliers 44 and 45 are used as new components. ing. In addition to these, the zero phase command generator 16, the integrator 26, the sine wave generator 27, the cosine wave generator 28, the multipliers 29 and 30, and the first-order lag filter 31 as described in the above embodiments. , 32.

  Next, the operation of the Ld measuring device 18E will be described.

  The current detection value Idc is multiplied by the cosine wave function and the sine wave function by the multipliers 29 and 30, and these multiplication values are input to the first-order lag filters 31 and 32, respectively, and only the DC component is output. The operation up to this point is exactly the same as the simplified Fourier series expansion described above. Thereafter, a difference value from the command value is calculated by adders / subtractors 35 and 36 for each component of Idsin and Idcos. For Idsin, a difference from It output from the measurement current amplitude command unit 34 is calculated, and the output voltage Vdsin is calculated by the integral compensator 37 so that this difference becomes zero. Similarly, Vdcos is calculated for Idcos by the zero phase command generator 16 so as to be controlled to zero. Vdsin and Vdcos are multiplied by a function of a sine wave / cosine wave in multipliers 44 and 45, respectively. These are voltage commands Vdc * to be applied to the electric motor after both are added by the adder 40. The present embodiment is characterized in that feedback control is performed so that each of the sine wave / cosine wave component of the current value matches the command value. The constant calculator 39 calculates the motor constant according to the equation (7) using the values of Vdsin, Vdcos, ω1t, and It.

  In this embodiment, the d-axis electric constant is described as an example, but Lq and R can be calculated in the same manner with the q-axis.

  Thus, according to the fifth embodiment of the present invention, the electric constant of the motor can be obtained with a simple configuration using the magnitude of the current as a parameter. As a result, it is possible to measure the magnetic saturation characteristic with respect to the current value, thereby further improving the control accuracy.

Embodiment 6:
Next, Embodiment 6 according to the present invention will be described with reference to FIG.

  In the fifth embodiment, the constant is automatically measured using the current amplitude as a parameter. Thus, by introducing feedback control, it is possible to perform measurement using the current value as a parameter, but there is a problem in the method of setting the feedback gain. That is, in order to quickly match the current value with the predetermined value, the control gain (the gain KiL of the integral compensator 34 in FIG. 11) must be set to an appropriate value according to the parameter to be controlled. However, if the electrical constant is completely unknown, it cannot be set to the best and the response must be sacrificed to some extent. In the worst case, divergent vibration does not stop, and measurement may be impossible. If the application is limited, such as a refrigerator motor, a certain degree of gain setting is possible, but there is a problem in performing automatic measurement of constants in a wider field.

  Therefore, the present embodiment provides a control device that further expands versatility and can measure a current value as a parameter.

  FIG. 12 is a functional block diagram of the Ld measuring device of the synchronous motor control device according to the sixth embodiment of the present invention. Although the basic configuration of the Ld measuring instrument 18F is the embodiment shown in FIG. 10, the way of giving the voltage amplitude command is greatly different. The Ld measuring device 18F of FIG. 12 adds a measurement voltage amplitude increase rate setting device 46, an integrator 47, a measurement current amplitude setting device 48, a current amplitude calculator 49, and a signal comparator 50 as new parts. Yes. The RL calculator 25 is exactly the same as that of the above-described fourth embodiment (FIG. 10), except that Idsin and Iqsin are output to the current amplitude calculator 49.

  Next, the operation of the Ld measuring device 18F will be described.

  The measurement voltage amplitude increase rate setting unit 46 sets an increase rate dV for gradually increasing the amplitude of the AC voltage applied to the PM motor. For example, dV is set such as 10% of the rated voltage per second. The switch 13 switches the switch to “1” or “0” according to the output of the comparator 50. In the case of “1”, dV is output as it is, and in the case of “0”, zero is output. . The integrator 47 performs integration according to the output of the switcher 13. On the other hand, the current amplitude calculator 49 calculates the current amplitude I0 by the equation (8) based on the values of Idcos and Idsin.

  This value is compared with the output It of the measured current amplitude setting device 48 by the comparator 50. If It> I0, the switch 13 is set to the “1” side, and if It <I0, “0”. Switch to the side. As a result, the constant measurement AC voltage continues to increase gradually from zero, and stops increasing when I0 = It. That is, the current value can be set to a predetermined value in a feedforward manner without introducing current feedback control. As a result, the electric constant range of the electric motor to be automatically measured is expanded, and the constant can be measured without adjustment regardless of the PM motor for any application. Although this method is also feedback control in a broad sense, it has a major advantage in that it has expanded versatility for unknown motors by losing linearity.

  In the present embodiment, the d-axis electric constant has been described as an example, but Lq and R can be calculated in the same manner with respect to the q-axis.

  As described above, according to the sixth embodiment of the present invention, it is possible to realize an automatic measurement of a highly versatile electrical constant using the magnitude of the current as a parameter.

Embodiment 7:
Next, Embodiment 7 of the present invention will be described with reference to FIGS.

  As described in the first embodiment, as the current detector in the present control device, one that detects the DC current of the inverter as shown in FIG. 1 is used. In practice, the current detector 6 is inexpensive and can be mounted in a small size, such as a shunt resistor.

  The seventh embodiment relates to a means for detecting a motor current from this direct current.

  FIG. 13 is a comparison diagram of applied voltage commands in the normal drive mode and the constant measurement mode in the synchronous motor control device according to Embodiment 7 of the present invention, and shows the difference in voltage waveforms in both modes. Here, it can be seen that a balanced three-phase AC voltage is applied during normal driving in FIG. 10A, while in the constant measurement mode including automatic adjustment (auto tuning) in FIG. A three-phase unbalanced voltage that is extremely out of balance is applied. In particular, in the section P near the zero cross in FIG. 13B, it can be seen that all the voltage commands become zero at the same time. A problem in current detection associated therewith will be described with reference to FIG.

  FIG. 14 is a current and voltage waveform diagram showing a problem in the synchronous motor control apparatus according to Embodiment 7 of the present invention. In the figure, the applied voltages Vu and Iu have a phase difference of almost 90 degrees. That is, since the measurement frequency ω1t is a high frequency, the load is almost inductive. As a result, the phase voltage Vu is close to zero in the section P near the peak of the current Iu. Under the condition that all the phase voltages are zero, no current flows through the current detector 6, and the DC current IDC does not flow in the section P in the vicinity of the current peak, which makes observation impossible. In this way, current detection is performed by omitting the vicinity of the current peak, and the accuracy is reduced.

  FIG. 15 is a partial functional block diagram of a synchronous motor control apparatus according to Embodiment 7 of the present invention, and shows a motor constant automatic adjuster 15G. FIG. 15 differs greatly from the previous embodiments (for example, FIG. 8 and the like) in that Vdc * and Vqc * output from the Ld measuring device 18 and the Lq measuring device 19 are lower limiters of the size. 51 and 52 are provided. As a result, a decrease in the voltage command near the zero cross is prevented, and current detection can be performed.

  FIG. 16 is a current and voltage waveform diagram in the synchronous motor control device according to the seventh embodiment of the present invention. As shown in the figure, the lower limit value of each phase voltage is limited in the section P in the vicinity of the zero cross, and there is a DC path current associated with the PWM control. On the other hand, correction will be added to the voltage command, but the error near the zero cross of the voltage has almost no effect globally, and it can enable highly accurate current detection. It becomes a very effective method.

  When the inverter circuit is actually configured using an IGBT, the output pulse width may be used as a guideline as the lower limit value of the voltage command absolute value. In consideration of the ringing phenomenon associated with the switching operation, it is possible to detect the current if a current supply width of at least 10 μs is secured.

Embodiment 8:
FIG. 17 is an overall external configuration diagram of a synchronous motor control apparatus according to Embodiment 8 of the present invention.

  In the figure, a microcomputer (hereinafter abbreviated as “microcomputer”) 53 includes the rotational speed command generator 1 and the mode switch 17 of FIG. Hereinafter, similarly, the same reference numerals as those in FIGS. 1, 2, 3, 5, 6, 41, 42, and 43 denote the same components. A command is transmitted from the microcomputer 53 to the power module 55 through the communication line 54. The power module 55 is obtained by integrating and miniaturizing the controller 2, the inverter 3, the current detector 6, and the diode bridge 42 of FIG. In addition, by connecting the AC power supply 41, the smoothing capacitor 43, and the PM motor 5 to the power module 55, a synchronous motor control device capable of automatically measuring / adjusting the electrical constant of the PM motor 5 can be realized.

  In the present embodiment, the controller 2 and the inverter 3 are modularized, so that the overall size of the device can be reduced, and any motor 5 can be easily driven as a control device that can be easily driven. The versatility can be expanded.

Embodiment 9:
Next, Embodiment 9 according to the present invention will be described with reference to FIG.

  FIG. 18 is an external configuration diagram of a control device of Embodiment 9 in which the present invention is applied to an air conditioner. In the figure, numerals 2, 3, 6, 42, and 43 represent the same parts as those in the first embodiment (FIG. 1) and the eighth embodiment (FIG. 17), respectively. In the present embodiment, an air conditioner outdoor unit 57 is configured in which a compressor 56 incorporating a motor is controlled by a power module 55 incorporating a controller 2, an inverter 3, a current detector 6, and a diode bridge 42. . The compressor 56 such as an air conditioner incorporates the PM motor 5 inside the hermetically sealed compressor 56, and it is difficult to detect the rotational speed of the PM motor 5, the position of the magnetic flux, and the like.

  However, by incorporating the control device according to the present invention as the power module 55, automatic measurement of the electrical constant of the motor 5 without detecting the rotation speed and position of the motor and with the motor 5 built in the compressor 56 is possible. / Adjustment is possible.

  Although the air conditioner has been described as an example of the embodiment, the same effect can be obtained in the case of other electric devices such as a packaged air conditioner and a refrigerator.

The control block diagram of the control apparatus of the synchronous motor by Embodiment 1 of this invention. The automatic measurement processing flowchart of the synchronous motor control apparatus by Embodiment 1 of this invention. The motor current waveform figure in the synchronous motor control apparatus by Embodiment 1 of this invention. The block diagram which shows the transfer function from the torque electric current to a rotor phase in the alternating current motor control apparatus by Embodiment 1 of this invention. The fragmentary view of the control block of the synchronous motor control device by Embodiment 2 of the present invention. The automatic measurement process flowchart of the synchronous motor control apparatus by Embodiment 2 of this invention. The graph which shows the time change of the frequency of the measurement alternating current in the synchronous motor control apparatus by Embodiment 2 of this invention, the electric current waveform of an electric motor, and an inductance measurement result. The fragmentary view of the control block of the synchronous motor control device by Embodiment 3 of the present invention. The motor current waveform figure in the synchronous motor control apparatus by Embodiment 3 of this invention. The functional block diagram of the Ld measuring device of the synchronous motor control apparatus by Embodiment 4 of this invention. The functional block diagram of the Ld measuring device of the synchronous motor control apparatus by Embodiment 5 of this invention. The functional block diagram of the Ld measuring device of the synchronous motor control apparatus by Embodiment 6 of this invention. The figure which shows the comparison of the applied voltage command at the time of the normal drive in the synchronous motor control apparatus by Embodiment 7 of this invention, and a constant measurement. The current and voltage waveform diagram which shows the subject in the synchronous motor control apparatus by Embodiment 7 of this invention. The partial functional block diagram of the synchronous motor control apparatus by Embodiment 7 of this invention. The current and voltage waveform diagram in the synchronous motor control device according to the seventh embodiment of the present invention. The external appearance block diagram which applied the synchronous motor control apparatus by Embodiment 8 of this invention. The external appearance block diagram of the control apparatus of Embodiment 9 which applied this invention to the air conditioner.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Speed command generator, 2 ... Controller, 3 ... Inverter, 4 ... DC power supply, 5 ... PM motor, 6 ... Current detector, 7 ... Current reproduction device, 8 ... Coordinate converter, 9 ... Voltage command calculation , 10 ... dq inverse converter, 11 ... PWM pulse generator, 12 ... Voltage command switcher, 13 ... Phase command switcher, 14 ... Measurement frequency setter, 15 ... Motor constant automatic adjuster, 16 ... Zero phase Command generator, 17 ... mode switch, 41 ... AC power supply, 42 ... diode bridge, 43 ... smoothing capacitor.

Claims (17)

  DC power supply, inverter that is fed from this DC power supply and outputs alternating current, PWM control unit that performs pulse width modulation control of this inverter, synchronous motor that is fed by the output alternating current of the inverter, and current that flows through this synchronous motor In the control apparatus for a synchronous motor, comprising: a current detection means for detecting; and a drive control apparatus that operates on the PWM control unit based on a speed command for the synchronous motor to drive the synchronous motor at a variable speed. Causes the inverter to output a three-phase unbalanced alternating current at a frequency of 25% or more of the maximum drive frequency of the synchronous motor, and outputs the three-phase unbalanced alternating current to the output of the current detection means. A control apparatus for a synchronous motor, comprising constant measurement control means for calculating a constant of the synchronous motor on the basis thereof.   2. The synchronous motor control device according to claim 1, wherein the three-phase unbalanced AC frequency is set to be equal to or higher than a maximum drive frequency of the synchronous motor.   2. The synchronous motor control device according to claim 1, further comprising means for outputting a direct current to the inverter prior to outputting the three-phase unbalanced alternating current.   4. The synchronous motor control device according to claim 3, further comprising means for gradually increasing the direct current.   2. The control apparatus for a synchronous motor according to claim 1, further comprising means for gradually increasing the three-phase unbalanced alternating current.   2. The synchronous motor control device according to claim 1, further comprising a lower limiter that regulates an upper limit value of the three-phase unbalanced AC voltage.   2. The control apparatus for a synchronous motor according to claim 1, wherein the current detection means is configured to detect a current on a DC side of the inverter, and is configured to ensure a current detection energization width of 10 μs or more. .   In Claim 1, the means for commanding the magnitude of the sine wave and cosine wave component of the three-phase unbalanced alternating current, the means for deriving the sine wave and cosine wave component of the output current of the current detection means, and these A control apparatus for a synchronous motor, comprising: means for controlling the sine wave and cosine wave components of the command so as to coincide with each component of the command.   The module according to claim 1, comprising a microprocessor constituting the drive control device including the constant measurement control means, and comprising a module in which the main circuit of the inverter, the current detection means, and the microprocessor are integrally formed. The control apparatus of the synchronous motor characterized by this.   2. The synchronous motor control device according to claim 1, wherein the synchronous motor driven by the drive control device is incorporated in an air conditioner or a compressor for freezing or refrigeration.   DC power supply, inverter that is fed from this DC power supply and outputs alternating current, PWM control unit that performs pulse width modulation control on this inverter, synchronous motor that is fed with output alternating current of the inverter, and current that flows through this synchronous motor In a control apparatus for a synchronous motor, comprising: current detection means for detecting; and a drive control apparatus that operates on the PWM control unit based on a speed command for the synchronous motor to drive the synchronous motor at a variable speed. Frequency setting means for setting a frequency for measuring electrical constants, means for causing the inverter to output a three-phase unbalanced alternating current of the frequency set by the frequency setting means, and at the time of outputting the three-phase unbalanced alternating current A control apparatus for a synchronous motor, comprising means for calculating an electric constant of the synchronous motor based on an output of the current detection means. According to claim 1 1, wherein the frequency setting means, when the output of the three-phase unbalanced AC, comprising means for continuously or stepwise changing the frequency, means for calculating the electrical constants, each of these frequencies A control device for a synchronous motor, comprising: means for calculating an electric constant of the synchronous motor based on the output of the current detection means. According to claim 1 1, wherein the frequency setting unit, the time to output the three-phase unbalanced AC, a means for continuously or stepwise changing the frequency, the output of the current detection means under each of these frequencies And a means for calculating an electric constant of the synchronous motor on the basis of the electric motor and a means for setting the electric constant when the calculated values converge within a predetermined range in the drive control device. apparatus.   DC power supply, inverter that is fed from this DC power supply and outputs alternating current, PWM control unit that performs pulse width modulation control on this inverter, synchronous motor that is fed with output alternating current of the inverter, and current that flows through this synchronous motor In the method for adjusting a synchronous motor control device, comprising: a current detection means for detecting; and a drive control device that operates on the PWM control unit based on a speed command for the synchronous motor to drive the synchronous motor at a variable speed. And supplying the three-phase unbalanced alternating current to the synchronous motor, and calculating the constant of the synchronous motor based on the output of the current detection means when the three-phase unbalanced alternating current is supplied. A method for adjusting a synchronous motor control device characterized by the above. According to claim 1 4, wherein the frequency of the three-phase unbalanced AC, adjusting method of the synchronous motor control device characterized by comprising the step of setting 25% or more of the frequency of the maximum driving frequency of the synchronous motor. According to claim 1 4, prior to the step of supplying an alternating current of the three-phase unbalanced, adjusting method of the synchronous motor control device characterized by comprising the step of supplying a direct current to the synchronous motor from the inverter. According to claim 1 4, the step of supplying an alternating current of the three-phase unbalanced includes the steps of continuously or stepwise changing the frequency of the three-phase unbalanced AC, the at three-phase unbalanced alternating supply of each frequency A step of calculating an electric constant of the synchronous motor based on the output of the current detection means, and a step of setting a constant when the change in the calculation result falls within a predetermined range in the drive control device. A method for adjusting a synchronous motor control device as a feature.

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