JP2006180605A - Controller for motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a controller for motor which reduces input power while suppressing cyclic disturbance in case that a load device, the target of rotation of an AC synchronous motor, causes cyclic disturbance. <P>SOLUTION: In torque control 22 that extracts 21 the components of pulsation of torque that the load device generates, and compensates it, this controller can achieve it by being provided with a limiter 227 which limits the components of currents to correct the components of pulsation. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a control device for an AC synchronous motor, and particularly to a technique for suppressing periodic torque disturbance generated by a load device or the motor itself.

  As a method for suppressing periodic torque disturbance generated by the load device of the electric motor, a technique described in Patent Document 1 is known. In Patent Document 1, a pulsation component included in a detected speed value of an electric motor is extracted, and the inverter output voltage is corrected so as to cancel it. When the fluctuation of the speed by the rotary compressor as a load connected to the motor to be controlled is smaller than the average speed fluctuation for the number of rotations of the compressor, the speed fluctuation compensation is not performed, and when the fluctuation is larger than the average speed fluctuation, Compensation corresponding to the primary component of the speed fluctuation is performed. Thus, it is described that the peak value of the motor current can be reduced as compared with the case where the control is always performed with the speed fluctuation being zero.

Japanese Patent Laid-Open No. 10-174488

  The so-called torque control described in Patent Document 1 can suppress periodic disturbance. However, no further electrical input reduction is mentioned.

  An object of the present invention is to provide a speed control device for an electric motor that reduces input while suppressing periodic disturbance.

  The object is to provide an electric motor that rotationally drives a load whose load torque fluctuates periodically, an inverter that performs pulse width modulation control on the electric motor, and a compensation signal that suppresses the periodically fluctuating load torque. And a torque fluctuation compensation means that performs torque fluctuation compensation by the torque fluctuation compensation means so that the torque fluctuation caused by the load that periodically varies the load torque remains. This is achieved by performing variation compensation.

  Another object of the present invention is to provide an electric motor that rotationally drives a load whose load torque fluctuates periodically, an inverter that performs pulse width modulation control on the electric motor, and a compensation signal that suppresses the load torque that fluctuates periodically. This is achieved by including a means for limiting a compensation amount of a compensation signal for suppressing a periodically varying load torque to be applied to the inverter.

  ADVANTAGE OF THE INVENTION According to this invention, the speed control apparatus of the electric motor which aimed at reduction of an input can be provided, suppressing a period disturbance.

  Hereinafter, with reference to FIGS. 1-8, embodiment of the air conditioner by this invention is described. In the following embodiments, a permanent magnet type synchronous motor (hereinafter abbreviated as PM motor) will be described as an electric motor, but the same applies to other synchronous motors (for example, a wound type synchronous motor, a reluctance motor, etc.). Is feasible.

  FIG. 1 is a block diagram showing a system configuration of Embodiment 1 of an AC motor control device according to the present invention. The control device according to the first embodiment calculates a rotational speed command generator 1 that gives a rotational speed command ωr * to the electric motor, and calculates an AC applied voltage of the electric motor, converts it into a pulse width modulation wave signal (PWM signal), and outputs it. A controller 2, an inverter 3 driven by the PWM signal, a DC power supply 4 for supplying power to the inverter 3, a PM motor 5 to be controlled, a compressor 6 as a load of the PM motor, and a DC power supply Comprises a current detector 7 for detecting a current I0 supplied to the inverter.

  The controller 2 detects the current I0 and reproduces the three-phase alternating currents Iu, Iv, Iw flowing through the PM motor 5 by calculation inside the controller, and the reproduced three-phase alternating current Iuc, A dq coordinate converter 9 for converting the coordinates Ivc and Iwc into components Idc and Iqc on the axes d and q according to the phase angle θdc (position of the magnet magnetic flux of the PM motor assumed inside the controller), q Iq * generator 10 that gives a command Iq * to the current component on the axis, and similarly, Id * generator 11 that gives a command Id * to the current component on the d-axis, Id *, Iq *, In addition, based on the electrical angular frequency command ω1 *, the voltage command calculator 12 that calculates the voltage commands Vdc * and Vqc *, and converts the Vdc * and Vqc * into the three-phase AC voltage commands Vu *, VV *, and Vw *. Based on dq inverse converter 13 and three-phase AC voltage command An error between the PWM pulse generator 14 that generates a pulse width modulation signal (PWM signal) for switching the inverter 3, the magnet magnetic flux position θd of the PM motor 5, and the position θdc assumed inside the controller 2. Δθ estimator 15 that estimates and calculates an angle (axis error) Δθ corresponding to the above, an adder / subtractor 16 that performs addition and subtraction, a zero generator 17 that gives a command to the axis error estimated value Δθdc, and Δθ equal to zero Conversion gain for converting the rotational speed command ωr * into the electrical angular frequency command ω1 * of the motor using the proportional compensator 18 for compensating the electrical angular frequency command ω1 * and the number of poles P of the PM motor. 19, an integrator 20 that integrates the electrical angular frequency and calculates the magnet magnetic flux position θdc, and a ΔTm estimator 21 that calculates a torque pulsation component from the axis error θdc.

  The DC power supply 4 that supplies power to the inverter 3 includes an AC power supply 41, a diode bridge 42 that rectifies the AC, and a smoothing capacitor 43 that suppresses a pulsating component included in the DC power supply.

  Next, the operation principle of this embodiment will be described with reference to FIG. The conversion gain 19 calculates and outputs the electrical angular frequency ω1 * of the PM motor based on the rotational speed command ωr * from the rotational speed command generator 1. Further, ω1 * is integrated using the integrator 20 to calculate the AC phase θdc. The current regenerator 8 reproduces the three-phase alternating current of the PM motor by calculation based on the power source current I0 detected by the current detector 7 by the method described in Japanese Patent Laid-Open No. Hei 8-19263. Next, the coordinate converter 9 converts the reproduced alternating currents Iuc, Ivc, and Iwc into current components Idc and Iqc on a rotating coordinate axis (dq axis) that rotates at an angular frequency ω1 * by θdc. Iqc is processed by the Iq * generator 10 to become a current command Iq * on the q axis. Further, the Id * generator 11 generates a current command Id * on the d axis (normally Id * = 0 in the case of a PM motor of a non-salient rotor). The voltage command calculator 12 calculates applied voltages Vdc * and Vqc * to the PM motor based on these commands (Id *, Iq *) and the angular frequency command ω1 *. Vdc * and Vqc * are again converted into an AC amount by the dq inverse converter 13, and further converted into a pulse width modulated wave signal by the PWM signal generator 14, and sent to the inverter 3. These basic operations are the same as those described in JP-A-2002-272194. Note that * is a sign meaning command.

  The Δθ estimator 15 performs an estimation calculation of the error Δθ between the position θd of the magnet magnetic flux in the PM motor and the position θdc in the controller. Δθ is defined by the vector diagram shown in FIG. The position of the actual magnet magnetic flux Φ inside the PM motor is defined as the d axis, and the axis orthogonal to the position is defined as the q axis. On the other hand, the dq axis assumed in the controller is defined as a dc-qc axis, and the deviation between the two corresponds to the axis error Δθ.

  If Δθ is obtained, it is possible to make the dq axis and the dc-qc axis coincide with each other by correcting this, and the magnetic pole position sensorless control of the PM motor can be realized. For example, as shown in FIG. 3, Δθ can be estimated by multiplying the difference between Iq * and Iqc by a proportional gain K0 to obtain an estimated value Δθdc of Δθ. Since Iqc fluctuates due to a shift in θd and θdc due to load fluctuation or the like, Δθ can be estimated conversely from the movement of Iqc. However, in the case of the configuration of FIG. 3, it is difficult to obtain Δθ with high accuracy. In order to improve the accuracy, for example, calculation may be performed according to Equation (3) in Japanese Patent Laid-Open No. 2002-272194.

  Based on the axis error estimated value Δθdc calculated by the Δθ estimator 15, feedback control is performed so that this becomes zero. The difference between the command (zero) of the zero generator 17 and Δθdc is obtained by the adder / subtractor 16, and the angular frequency ω 1 * is compensated via the proportional compensator 18. As shown in the vector diagram of FIG. 2, when Δθ is positive, the dc-qc axis is more advanced than the dq axis. Therefore, by reducing ω1 *, Δθ can be reduced. Conversely, when Δθ is negative, ω1 * is increased so that the dq axis coincides with the dc-qc axis. By controlling in this way, the phase angle θdc inside the controller can be made to coincide with the magnet magnetic flux position θd in the actual PM motor without using the position sensor of the magnetic pole axis of the PM motor. Can be realized.

Next, the ΔTm estimator will be described with reference to FIGS.
First, the principle of PM motor torque generation and the relationship between the shaft error Δθ will be briefly described with reference to FIGS. 4 and 5.

  FIG. 4 is a block diagram showing the principle from the voltage applied to the PM motor to the generation of the axis error Δθ. In each block of the figure, R is the winding resistance of the PM motor, L is the inductance of the PM motor, P is the number of poles of the PM motor, Ke is the power generation constant (magnet magnetic flux) of the PM motor, J is the PM motor and load device Total inertia, s, represents a differential operator used for Laplace transform.

  As shown in FIG. 4, the q-axis current Iq is generated from the relationship between the applied voltage Vq applied to the PM motor, a voltage disturbance VD described later, and the electric constants R and L of the motor. Iq is a component orthogonal to the magnet magnetic flux (d-axis) of the PM motor, and becomes the motor torque Tm by multiplying by the power generation constant Ke. The rotational speed ωr of the PM motor is obtained by integrating the difference between the motor torque Tm and the load torque TL. Here, the load torque TL has various characteristics depending on the type and application of the load device. The electric angular frequency ω1 of the motor is obtained by multiplying ωr by the number of pole pairs (P / 2), and its integrated value becomes the position θd of the PM motor. The axis error Δθ is obtained as a difference from the phase θdc in the controller.

  Here, it is considered that a periodic component is included in the voltage disturbance VD or the load torque TL.

  As the periodic voltage disturbance VD, for example, when the magnet magnetic flux of the PM motor is not uniform and there is a variation in magnetization, or when there is a variation in phase between windings, it is equivalently affected as a periodic voltage disturbance. Come on. Or the disturbance etc. by the influence of the arm short circuit prevention period (dead time) in an inverter generate | occur | produce as 6 times the drive frequency of an inverter.

  Further, as a periodic load torque disturbance, loads such as a reciprocating compressor, a single rotary compressor, a twin rotary compressor, and a two-stage compression rotary compressor can be considered. In the case of a reciprocating compressor and a rotary compressor, the load fluctuates violently with one rotation of the motor as one cycle.

  FIG. 5 shows a torque pulsation component (ΔTm), a rotational speed fluctuation (Δωr), and an axis error (Δθ) when the load torque TL includes a component that vibrates sinusoidally at an angular frequency ωd. Considering the steady state, the average values of Tm and TL coincide with each other, and ΔTm is only the vibration component (FIG. 5B). The vibration component Δωr included in the rotation speed is obtained by integrating ΔTm, and has a waveform whose phase is delayed by 90 degrees compared to ΔTm. The magnitude of the vibration itself varies depending on the inertia J, but the phase may be considered to be delayed by approximately 90 degrees. The axis error Δθ is obtained by further integrating Δωr and inverting the sign (inverting the sign according to the definition shown in FIG. 2), so that the phase is advanced by 90 degrees (90 degrees delayed by integration). In addition, since the sign is inverted, it is advanced 90 degrees). That is, the variation component of ΔTm is observed as a vibration waveform having the same phase at Δθ. This relationship is derived from the block diagram as follows.

  FIG. 6A shows a block diagram from ΔTm to Δθ. By inversely transforming this block diagram, a transfer function from Δθ to ΔTm can be obtained, as shown in FIG.

  If ΔTm is obtained according to FIG. 6C, the pulsation component of the torque can be directly estimated from Δθdc (estimated value of Δθ). However, it is practically impossible to differentially differentiate Δθdc. Since Δθdc is originally an estimated value and includes a large amount of noise in the detected value, using differential increases an estimation error and has a limit from the calculation cycle.

  Therefore, paying attention to the point that “disturbance component is a periodic function”, s = jωd is substituted into FIG. Then, as shown in FIG. 6D, a value obtained by multiplying Δθ by a constant can be estimated as ΔTm. This result agrees with the relationship between the waveforms (b) and (d) in FIG.

  A configuration embodying FIG. 6D is a ΔTm estimator 21 shown in FIG. The ΔTm estimator 21 includes a proportional gain 211 for multiplying Δθdc by 2J / P and two multipliers 212, and performs the calculation of FIG.

  From FIG. 7, it is possible to estimate the periodic disturbance component of the angular velocity ωd included in ΔTm from Δθdc.

  Next, the torque controller 22 provided with the limiter 227 will be described. The torque controller 22 in FIG. 1 includes a single phase-dq coordinate converter 223, a first-order lag filter 224, an integral controller 225, a dq-single phase inverse converter 226, an adder / subtractor 16, and a zero generator 17. Next, the operation of the torque controller 22 will be described.

  The output ΔTmc of the ΔTm estimator is decomposed into a SIN component and a COS component by the single phase-dq coordinate converter 223. The conversion formula of the single phase-dq coordinate converter 223 is as follows.

（数１）に従えば、ΔＴｍｃにωｄの周波数成分が含まれれば、その量に応じて、ΔＴｄｓ、ΔＴｑｓの平均値が零でない値になる。この平均値は、それぞれ、ΔＴｍｃに含まれるＣＯＳ成分、ならびにＳＩＮ成分に一致する。ただし、ΔＴｄｓ、ΔＴｑｓには、ωｄの２倍成分が多量に含まれるため、一次遅れフィルタ２２４にて、交流成分を削除する必要がある。この結果、ΔＴｄｓ、ΔＴｑｓは、ΔＴｍｃに含まれる脈動成分の、ＣＯＳ成分、ならびにＳＩＮ成分になる。次に、この各成分を零にするため、指令である「零」信号を、零発生器１７から与え、指令との偏差を加減算器１６で演算する。これらの偏差に基づき、積分制御器２２５が積分補償を行い、脈動成分を零に制御する。リミッタ部２２７では積分制御器２２５が算出した補正電流値に制限をかける。

According to (Equation 1), if the frequency component of ωd is included in ΔTmc, the average value of ΔTds and ΔTqs becomes a non-zero value according to the amount. This average value corresponds to the COS component and SIN component included in ΔTmc, respectively. However, since ΔTds and ΔTqs contain a large amount of the double component of ωd, it is necessary to delete the AC component by the first-order lag filter 224. As a result, ΔTds and ΔTqs become the COS component and SIN component of the pulsation component included in ΔTmc. Next, in order to make each component zero, a “zero” signal as a command is given from the zero generator 17, and a deviation from the command is calculated by the adder / subtractor 16. Based on these deviations, the integral controller 225 performs integral compensation and controls the pulsation component to zero. In the limiter unit 227, the correction current value calculated by the integration controller 225 is limited.

  Here, finally, the values of Ids and Iqs are inversely converted into single-phase signals, and IqSIN * is output. This inverse transformation is calculated according to the following equation.

脈動成分ΔＴｍｃは、（数１）にて座標変換された後は、直流量になるため、積分制御器２２５にて、偏差をなくすことができる。すなわち、このトルク制御器は、外部から見ると、角周波数ωｄにおいてゲインが無限大となる補償要素と、等価になる。

Since the pulsation component ΔTmc becomes a direct current amount after coordinate conversion in (Equation 1), the integration controller 225 can eliminate the deviation. That is, this torque controller is equivalent to a compensation element whose gain is infinite at the angular frequency ωd when viewed from the outside.

  Next, the vibration and input of the limiter and the compressor will be described.

  The limiter is based on the average value of Iq * in FIG. 1 being 100%. Since Iq * is the main component of torque current, it is based on this. The torque fluctuation component of the compressor is divided into a fluctuation (AC component) caused by torque pulsation in the compressor average load (DC component). By applying a torque current having an opposite phase to the pulsating component, the pulsating component can be canceled and brought close to zero. FIG. 8 shows the relationship among the limiter, vibration, and input power when a rotary compressor is used as the compressor. The horizontal axis limiter%, the left vertical axis shows the input power of the power supply, and the right vertical axis shows the vibration of the compressor. The horizontal axis limiter 100% is not defined by the current that cancels the pulsation component, and the torque component current command value Iq * is 100% (100% means that the limiter is not effective, opposite) 0% means no torque control at all). When performing the control of what percentage of torque control is applied, if the value that cancels the pulsation component is defined as 100%, the absolute value varies depending on the operating conditions. To determine the absolute amount of 100%. Since this is difficult in actual control, the torque current component command value Iq *, which is a value commensurate with the compressor average load (DC component), which is always larger than the pulsation component, is set to 100% for convenience.

  From FIG. 8, it can be seen that, by using torque control, the vibration falls to the lower right, but the input power of the power source increases. The vibration is a flat curve from about 75% to 100%, which indicates that the vibration is suppressed at about 80%. In this case, the vicinity of about 80% is the point where the torque control current is 100%. However, in this embodiment, since 100% is defined as the torque current component command value Iq *, the curve is flat between 75% and 100%.

  Now, it can be seen from FIG. 8 that there is a minimum value in the input power. For this reason, by setting the limiter optimally, the vibration of the compressor can be suppressed, and the input power can be made smaller than before performing this control.

  This is considered as follows. Inverter DC side current and U-phase motor current: (1) Without torque control (limiter = 0), (2) With torque control and limiter appropriately (3) Maximum torque control (limiter = 100) Compare the waveforms for. In the case of (1), the rotational speed fluctuation is the largest, the motor current waveform is uneven (the upper and lower peak values are not uniform in each cycle), and the pulsation of the direct current is also large. Further, in the case of (3), the fluctuation of the rotational speed is the smallest, but the motor current waveform is uneven and the direct current also pulsates.

  On the other hand, when the torque control was limited and applied, the fluctuations in the rotational speed were seen compared to (3), but the peak values of the motor current waveform were uniform and the pulsation of the DC current appeared small. If the crest values of the motor current are aligned, it means that the input is small. Also, the fact that the pulsation of the direct current is small means that when the pulsation is present, the effective value is 0 for direct current, whereas the effective value appears because there is an alternating current, and the input increases accordingly.

  If the torque control is limited based on the principle as described above, it is considered that although the speed (torque) fluctuation remains, the input can be reduced.

  Next, the compensation amount is limited by canceling the periodic disturbance. From FIG. 8, by limiting the torque control amount (maximum 10%) so that the input power is minimized, When used in an air conditioner, COP can be improved.

  Further, as described in detail in the explanation of the principle, the input power can be reduced if the limit of the correction amount for canceling the periodic disturbance is set so that the peak values of the alternating current flowing through the synchronous motor are substantially uniform.

  The above is an ideal embodiment, but when this embodiment is viewed as an input power reduction device without considering the aspect of torque control, the input power when the torque control is not performed is 490 W as shown in FIG. The amount decreases as the amount increases, and starts increasing from the point where the minimum value is exceeded. When the compensation amount reaches about 50%, the input power reaches 490 W again. That is, by making the compensation amount larger than 0% and 50% or less, an effect of reducing the input power can be obtained. Therefore, the limiter is set in a range larger than 0% and 50% or less.

  In particular, when the input power is significantly reduced, the limiter value may be set to 2% to 30%.

  According to the present embodiment, it is possible to realize an air conditioner that suppresses vibration and noise and enables highly efficient operation of the compressor. In addition, low-efficiency and high-efficiency motors are used for air conditioners that use compressors that could not be operated at low speed due to vibration and noise. And significant energy savings can be expected.

  In the above-described embodiment, the description has been made using various numerical values. However, these numerical values are merely examples, and other numerical values may be used as long as they match the control concept.

  The same effect can be obtained when a scroll compressor, rotary compressor, or reciprocating compressor is used as a load. In addition, as a current detection method, a method of reproducing the motor current from the current value I0 of the DC power supply 4 is used.

It is a block diagram which shows the system | strain structure of the Example of the synchronous motor control apparatus by this invention. It is a vector diagram which shows the definition of axial error (DELTA) (theta) in the Example of the synchronous motor control apparatus by this invention. It is a block diagram which shows the internal structure of the axis | shaft error estimator in the Example of the synchronous motor control apparatus by this invention. It is a block diagram explaining the principle from the applied voltage to an electric motor in the Example of the alternating current motor control apparatus by this invention to axial error generation | occurrence | production. It is a wave form diagram which shows the principle of the periodic disturbance torque in the Example of the alternating current motor control apparatus by this invention, the rotation pulsation resulting from it, and an axis | shaft error fluctuation | variation. It is a block diagram explaining the estimation principle of the pulsation torque component in the Example of the alternating current motor control apparatus by this invention. It is a block diagram which shows the internal structure of the (DELTA) Tm estimator in the Example of the synchronous motor control apparatus by this invention. It is a figure showing the relationship between the limiter in the Example of the synchronous motor control apparatus by this invention, an input voltage, and a vibration.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Speed command generator, 2 ... Controller, 3 ... Inverter, 4 ... DC power supply, 5 ... PM motor, 6 ... Compressor, 7 ... Current detector, 8 ... Current reproduction device, 9 ... dq coordinate converter DESCRIPTION OF SYMBOLS 10 ... Iq * generator, 11 ... Id * generator, 12 ... Voltage command calculator, 13 ... dq inverse converter, 14 ... PWM pulse generator, 15 ... Δ (theta) estimator, 16 ... Adder / subtractor, 17 ... Zero Generator: 18 ... Proportional compensator, 19 ... Conversion gain, 20 ... Integrator, 21 ... ΔTm estimator, 22 ... Torque controller.

Claims (5)

  An electric motor that rotationally drives a load whose load torque fluctuates periodically, an inverter that performs pulse width modulation control on the electric motor, and a torque fluctuation compensation that provides the inverter with a compensation signal that suppresses the periodically fluctuating load torque The torque fluctuation compensation so that the torque fluctuation caused by the load whose load torque fluctuates periodically remains in a state where the torque fluctuation compensation is performed by the torque fluctuation compensation means. Electric motor control device.   An electric motor that rotationally drives a load whose load torque fluctuates periodically, an inverter that performs pulse width modulation control on the electric motor, and a torque fluctuation compensation that provides the inverter with a compensation signal that suppresses the periodically fluctuating load torque An electric motor control apparatus comprising: means for limiting a compensation amount of a compensation signal for suppressing a periodically varying load torque applied to the inverter.   An electric motor that rotates the compressor, an inverter that performs pulse width modulation control on the electric motor, and a torque fluctuation compensation unit that provides the inverter with a compensation signal that suppresses a load torque that fluctuates periodically in the compressor. An electric motor control device comprising means for limiting a compensation amount of a compensation signal for suppressing a periodically varying load torque applied to the inverter.   A compressor for an air conditioner, a synchronous motor for driving the compressor, an inverter for applying a pulse-width-modulated voltage to the synchronous motor and driving it with a continuous alternating current, and a magnetic pole shaft in the synchronous motor The amount corresponding to the error between the assumed phase angle of the synchronous motor and the estimated phase angle of the synchronous motor's magnetic pole axis is calculated using the detected value of either the alternating current or direct current flowing through the synchronous motor. A compensator that calculates a periodic disturbance component from an amount corresponding to the axis error, cancels the periodic disturbance, and includes a controller that controls a voltage output from the inverter based on a detected value of the current. In the motor control apparatus, the controller includes a limiter that limits a correction amount for canceling the periodic disturbance.   The motor control device according to claim 1, wherein a limiting amount for limiting a correction amount for canceling the periodic disturbance is greater than 0% and equal to or less than 50% when an average value of torque current is defined as 100%.

Images