JP3806539B2 - Control method of permanent magnet type synchronous motor - Google Patents

Description

すなわち、トルクＴは、数式（２）で表わされる。
【００１９】
【数２】

ここで、
ω ：モータ軸の回転角速度で、ω＝ω₁／（ｐ／２）
ω₁：電気回転角周波数
ｐ ：モータの極数
Φ ：磁束
Ｌ_d，Ｌ_q：ｄ，ｑ軸のインダクタンス
であり、
【００２０】
【数３】
Ｅ₀＝ω₁・Φ …（３）
【００２１】
【数４】
Ｘ_d＝ω₁・Ｌ_d，Ｘ_q＝ω₁・Ｌ_q …（４）
である。このとき、同期モータ５１が円筒機の場合、Ｌ_d＝Ｌ_q＝Ｌであるから、トルクＴは、数式（５）のように表わされる。
【００２２】
【数５】
Ｔ＝ｋ・Φ・Ｉ_q （ｋは定数） …（５）
この結果、トルクＴは、電流のｑ軸成分Ｉ_q のみに比例する。したがって、トルクを制御するには、電流のｑ軸成分Ｉ_q だけを制御すればよいことがわかる。
次に同期モータの端子電圧Ｖ_t は、図２のベクトルから、数式（６）のように表わされる。
【００２３】
【数６】

数式（６）から、モータが一定速度で回転しているとき、端子電圧Ｖ_t は、ｄ軸電流成分を零としても、ｑ軸電流成分により変化する。すなわち、トルクを増加するために電流成分Ｉ_q を増加させると、端子電圧Ｖ_t は増加することが分かる。このとき、数式（６）から、ｄ軸電流成分Ｉ_d を負に制御すると、電圧のｑ軸成分の大きさが小さくなるので、ベクトル和としての端子電圧は抑制できることも分かる。
【００２４】
ここで、数式（６）のモータの端子電圧Ｖ_t にインバータの可出力電圧Ｖ_maxを代入し、さらに、Ｉ_d について解くと、数式（７）を得る。
【００２５】
【数７】

数式（７）から、モータ端子電圧抑制を行う弱め界磁制御の判断は、
数式（７）の結果がＩ_d≧０ならば弱め界磁制御不要（Ｉ_d ^*に０を設定）
数式（７）の結果がＩ_d＜０ならば弱め界磁制御必要（Ｉ_d ^*にＩ_dを設定）
で表すことができる。
【００２６】
ここで、Ｌ_q，Ｌ_d，Φはモータ定数であり一定として扱え、Ｉ_qとω₁はモータ運転状態で変わるので、それぞれ検出値もしくは制御装置内の指令値を用いればよい。一方、Ｖ_max はインバータに入力される直流電圧Ｖ_dcにより変動するので、次式で演算すればよい。
【００２７】
【数８】
Ｖ_max＝(Ａ_max・Ｖ_dc）／(２√２) …（８）
（８）式のＡ_max は、搬送波と変調波の振幅比で定義した変調度の最大値であり、通常、搬送波の振幅のピークを超えないようにＡ_max ＝１以下に設定する。このようにすると、モータの端子電圧波形に低時高調波成分が含まれない。すなわち、端子電圧波形が歪まないため、モータから発生するトルク脈動を抑えることができ、エレベーターなどの振動を抑制すべき用途には、好適であることが分かる。また、多少のトルク脈動を許す用途では、Ａ_max ＝１以下に設定する必要はなく、Ａ_max が１を超える値に設定すればよいことが分かる。
【００２８】
図３は上記の原理を応用したｄ軸電流指令装置６４の構成例である。図は円筒機の場合（Ｌ_d＝Ｌ_q＝Ｌ）であり、ｑ軸電流指令信号Ｉ_q ^*はｑ軸電流指令装置６３で速度制御装置６２からのトルク指令Ｔ^* を数式（５）の原理により変換したものである。ｄ軸電流演算器６４２は、数式（７）に基づいて、速度検出器５３からの信号とｑ軸電流指令装置６３からの信号Ｉ_q ^*及び可出力電圧設定器６４３からの信号Ｖ_max を入力し、仮のｄ軸電流指令Ｉ_drefを出力する。弱め界磁制御判定器６４５では、ｄ軸電流演算器６４２からの信号Ｉ_drefを入力し、上述した弱め界磁制御の判断に基づいて、ｄ軸電流指令Ｉ_d ^*を決定する。すなわち、Ｉ_dref≧０ならばＩ_d ^*＝０，Ｉ_dref＜０ならばＩ_d ^*＝Ｉ_drefとしている。なお、可出力電圧設定器６４３は、数式（８）に基づいて、直流電圧検出器７３の電圧検出信号Ｖ_dcからインバータの可出力電圧Ｖ_max を設定している。
【００２９】
このようにｄ軸電流指令を制御することにより、端子電圧の大きさを所定値に抑えることができるので、モータの耐圧や、インバータの制御電圧範囲を小さくできる。モータが最高回転付近でインバータの出力電圧に余裕がないときに効果的で、インバータの出力容量の低減ができる。さらに、インバータに入力される直流電圧の変動も考慮しているので、受電電圧が低下した場合においてもインバータが出力電圧不足になることはなく、安定な弱め界磁制御が実行される。
【００３０】
こうして得られた電流指令Ｉ_q ^*，Ｉ_d ^*は電流指令装置６５に入力される。図４は電流制御装置６５の具体的構成例を示す。本例の基本構成は周知であり、例えば、電気学会論文誌Ｄ，１１７巻，５号（１９９７年７月），５３９頁，図５に記載されている。図４の構成は、図２のベクトル図からｄ，ｑ軸の電圧成分を演算し、さらにｄ，ｑ軸の電流の指令と実際値との偏差に応じて働くＡＣＲ−ｄ，ＡＣＲ−ｑを備えている。また、図は電機子抵抗Ｒ_a をさらに考慮している。
【００３１】
Ｉ_d／Ｉ_q演算６５１は位置検出器５２からの界磁磁極位置（電気的回転角）θに応じた正弦または余弦信号を基準に、電流検出器７４からの３相の瞬時電流検出値ｉｕ，ｉｖ，ｉｗを用いて各電流の成分Ｉ_d，Ｉ_qが検出される。電流制御装置６５の出力はｄ，ｑ軸の電圧指令信号Ｖ_d ^*，Ｖ_q ^*であり、変調波発生装置７５では、このＩ_d／Ｉ_q演算６５１の逆演算を行って、ＰＷＭパルス発生装置７７でのＰＷＭパルス作成に用いる正弦波状の変調波信号ａ^* を作成している。この演算は周知なので省略する。
【００３２】
図５はこのような制御の有無による特性を示す図である。（ａ）は本発明の制御を行ったときの特性例、（ｂ）は本発明の制御を行わず、ｄ軸分の電流は零とし、ｑ軸分の電流制御のみをトルクに応じて行う従来の制御の特性例を示す。
【００３３】
（ａ）の例では、弱め界磁制御の条件になるトルク指令ａの点から、ｄ軸電流Ｉ_d をトルク指令の増加に応じて負の方向に増加させている。また、ｑ軸電流は数式（５）からわかるようにトルクに比例させる。ｄ軸電流をこのように制御すると、ｄ，ｑ軸電流成分のベクトル和である電機子電流Ｉは、Ｉ_q よりやや増加するものの、端子電圧Ｖ_t は無負荷時からほとんど増加しない。さらに、インバータの直流電圧が低下した場合のｂ〜ｃ点においても、ｄ軸電流が自動的に負の方向に増加され、インバータの可出力範囲内まで端子電圧を抑制できている。このため、インバータ出力電圧の可変範囲を必要最低限度の適切なものにすることができる。この結果、インバータの容量の増加を抑えられる。
【００３４】
一方、（ｂ）のようにｄ軸分を零にすると、端子電圧Ｖ_t はトルク指令の増加とともに増大する。また、インバータの直流電圧の低下時には、（ｂ）のようにインバータの可出力電圧範囲を超える場合がある。この結果、このような現象を防ぐために、インバータの出力電圧は大きなものが必要になり、その結果インバータの容量が増大する。
【００３５】
以上のように、本発明によれば、インバータの出力電圧の可変範囲が小さくできるので、インバータ容量を低減でき、また、モータの絶縁耐圧も増加させない。このため、小型かつ経済的なシステムを提供できる。
【００３６】
図６は図３に示したものと別の実施例を示す。図において、図３と同一番号のものは同一物を示す。図６の弱め界磁制御判定器６４１は、弱め界磁制御時の判定を変調波発生装置７５からの信号と搬送波発生装置７６からの信号に基づいて行い、切替え回路６４４に切替え信号を出力する。具体的には、変調波信号ａ^* の振幅が搬送波信号ｓ_c のピーク値（波高値）に達した時、ｄ軸電流指令Ｉ_d ^*にｄ軸電流演算器６４２の出力信号Ｉ_drefを選択する信号を切替え回路６４４に出力する。また、ｄ軸電流指令Ｉ_d ^*には、切替え回路６４４により、上述した状態以外において、０信号が入力される。
【００３７】
このようにすると、弱め界磁制御の判定（開始点）をインバータの出力電圧を決める搬送波と変調波の関係から直接決められるため、モータの端子電圧波形に低時高調波成分が含まれず、モータから発生するトルク脈動を確実に抑えることができ、端子電圧は所定値を超えることがないので、インバータの出力電圧の可変範囲の余裕を小さくでき、さらに、インバータの出力容量を低減できる。
【００３８】
なお、上記例では、変調波信号ａ^* の振幅が搬送波信号ｓ_c のピーク値（波高値）に達した時と説明したが、変調波信号ａ^* と搬送波信号ｓ_c の振幅比を定義し、この振幅比が予め設定した所定値に達した場合としてもよい。さらに、可出力電圧設定器６４３は、インバータの直流電圧検出器７３の電圧検出信号Ｖ_dcからインバータの可出力電圧Ｖ_max を設定しているが、上述したように弱め界磁制御の判定（開始点）を搬送波と変調波の関係から直接決めているため、直流電圧の検出信号を用いずに、可出力電圧Ｖ_max を予め設定した一定値としてもよい。図７は図３と図６に示したものとさらに別の実施例を示す。図において、図３及び図６と同一番号のものは同一物を示す。図７の弱め界磁制御判定器６４６は、弱め界磁制御時の判定をＰＷＭパルス発生装置７７からのＰＷＭパルス信号に基づいて行い、切替え回路６４４に切替え信号を出力する。具体的には、ＰＷＭパルス信号のオン時間が、搬送波信号の周期（インバータのキャリア周期）に達した時、ｄ軸電流指令Ｉ_d ^*にｄ軸電流演算器６４２の出力信号Ｉ_drefを選択する信号を切替え回路６４４に出力する。また、ｄ軸電流指令Ｉ_d ^*には、切替え回路６４４により、上述した状態以外において、０信号が入力される。
【００３９】
このようにすると、弱め界磁制御の判定（開始点）をインバータの出力電圧を決めるＰＷＭパルス信号から直接決められるため、図６の実施例と同様に、インバータの出力電圧の可変範囲の余裕を小さくでき、さらに、インバータの出力容量を低減できる。
【００４０】
なお、上記例では、ＰＷＭパルス信号のオン時間がインバータのキャリア周期に達した時と説明したが、ＰＷＭパルス信号のオン時間とキャリア周期の時間比を定義し、この時間比が予め設定した所定値に達した場合としてもよい。
【００４１】
【発明の効果】
以上説明したように、本発明によれば、永久磁石式同期モータの端子電圧の値が前記インバータの出力可能な最大出力電圧の範囲内となるように、前記モータの磁界と同方向の電流成分（ｄ軸電流成分）を制御するため、モータ端子電圧の増加の抑制ができるので、モータの出力電圧，電流の増加を小さくすることができ、この結果、インバータの容量及びモータの絶縁耐圧の増加を抑制することが可能になる。このため、制御システム全体を小型かつ経済的に構築することができる。
【図面の簡単な説明】
【図１】本発明の一実施形態による永久磁石同期モータの制御方法を示す図。
【図２】制御原理を説明するためのベクトル図。
【図３】図１におけるｄ軸電流指令装置の構成例を示す図。
【図４】電流制御装置の構成例を示す図。
【図５】本発明による特性を示す図。
【図６】ｄ軸電流指令装置の他の構成例を示す図。
【図７】ｄ軸電流指令装置の他の構成例を示す図。
【符号の説明】
５１…同期モータ、５２…位置検出器、５３…速度検出器、６１…速度指令装置、６２…速度制御装置、６３…ｑ軸電流指令装置、６４…ｄ軸電流指令装置、６５…電流制御装置、６６…インバータ、７０…交流電源、７１…コンバータ、７２…平滑コンデンサ、７３…直流電圧検出器、７４…電流検出器、７５…変調波発生装置、７６…搬送波発生装置、７７…ＰＷＭパルス発生装置、６４１，６４５，６４６…弱め界磁制御判定器、６４２…ｄ軸電流演算器、６４３…可出力電圧設定器、６４４…切替え回路、６５１…Ｉ_d／Ｉ_q演算。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a control method of a permanent magnet type synchronous motor using a small and strong permanent magnet as a field.
[0002]
[Prior art]
A synchronous motor using a small and strong permanent magnet as a field can be reduced in size, and the drive device including the motor can be reduced in size, and there is an advantage that efficiency is improved.
[0003]
The torque of the motor is controlled by controlling the current component (q-axis current component) perpendicular to the magnetic field direction (d-axis) corresponding to the magnetic pole position of the motor. This technique is described in Nakano, “Servo Technology and Power Electronics” (published in September 1994), pages 94-95. For the same purpose as a DC motor, there is also a method of controlling the d-axis current in order to weaken the field as the speed of the synchronous motor increases from a specified speed. This technique is described, for example, in JP-A-8-182398.
[0004]
However, the synchronous motor has an armature reaction. Therefore, the voltage increases as the load increases even when operating at a certain rotational speed. This phenomenon occurs in any of the above conventional control methods. In particular, when the load is larger than the rated torque, the voltage rise becomes significant. For this reason, when operating near the rated speed, the output voltage (maximum voltage that can be output) of the inverter that controls the motor must be increased to accommodate the increase in voltage due to an increase in load. As a result, there is a problem that the capacity of the inverter must be increased or the withstand voltage of the motor must be increased.
[0005]
Furthermore, since the inverter input DC voltage related to the output voltage of the inverter is normally generated from the AC power source by a rectifier composed of a diode and a capacitor, it is affected by the voltage fluctuation of the AC power source. In particular, when the AC power supply voltage decreases, the output voltage of the inverter also decreases with the decrease of the DC voltage. For this reason, in addition to the increase in voltage due to the increase in the motor load described above, a problem arises in that the apparatus becomes larger in order to increase the power reception voltage in advance by newly providing a transformer or the like so as to cope with the voltage fluctuation of the AC power supply. Also occurs. Moreover, even if the inverter input DC voltage is a battery or the like, the same countermeasure is necessary because it fluctuates by continuously applying the load.
[0006]
[Problems to be solved by the invention]
In view of the above problems, an object of the present invention is to provide a control method for a permanent magnet synchronous motor suitable for reducing an increase in output voltage and current of a motor and suppressing an increase in inverter capacity and motor dielectric strength. There is to do.
[0007]
[Means for Solving the Problems]
The above problem is to control the current component (d-axis current component) in the same direction as the magnetic field of the motor so that the value of the terminal voltage of the permanent magnet synchronous motor is within the range of the maximum output voltage that can be output by the inverter. This can be solved.
[0008]
Here, the d-axis current component is calculated by calculating a temporary command value of the d-axis current component based on the relationship between the DC voltage input to the inverter and the torque and rotational speed of the motor. When the value of the axial current command is zero or positive, the d-axis current component is zero, and when the temporary d-axis current command value obtained by the calculation is negative, the d-axis current component is the temporary d-axis current. Follow the command.
[0009]
Further, based on the relationship between the DC voltage input to the inverter and the torque and rotational speed of the motor, a temporary command value of a current component (d-axis current component) in the same direction as the magnetic field of the motor is calculated, Usually, the d-axis current component is controlled to zero in response to a predetermined relationship between a modulated wave for generating a PWM pulse input to the inverter and a carrier wave. Follow the provisional d-axis current command.
Further, based on the relationship between the DC voltage input to the inverter and the torque and rotational speed of the motor, a temporary command value of a current component (d-axis current component) in the same direction as the magnetic field of the motor is calculated, Normally, the d-axis current component controlled to zero is converted into the d-axis current component in response to the on-time of the PWM pulse input to the inverter and the carrier cycle of the inverter having a predetermined relationship. Follow the provisional d-axis current command.
[0010]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
[0011]
FIG. 1 shows a control method of a permanent magnet type synchronous motor according to an embodiment of the present invention.
In FIG. 1, the AC voltage of the AC power source 70 is converted into DC by a converter 71, this DC voltage is smoothed by a smoothing capacitor 72, and the smoothed DC is further converted by a inverter 66 into AC of variable voltage / variable frequency. The The output of the inverter 66 is supplied to the permanent magnet type synchronous motor 51, thereby driving the synchronous motor 51 at a variable speed. The rotating shaft of the permanent magnet type synchronous motor 51 is connected to a load (not shown), and further connected to a position detector 52 and a speed detector 53. The position detector 52 uses a resolver, an encoder, or the like, and detects the relative position of the armature of the synchronous motor 51 and the permanent magnet field, that is, the rotation angle. The speed detector 53 uses an encoder or the like and detects the rotational speed of the synchronous motor 51. In the illustrated example, the position detector 52 and the speed detector 53 are divided into functions and described separately. However, in practice, the position detector 52 and the speed detector 53 may be configured by the same device such as a resolver and an encoder.
[0012]
Now, when the speed command ω ^* is output from the speed command device 61, a deviation Δω from the output signal ω of the speed detector 53 is input to the speed control device 62. The speed control device 62 works according to this deviation, and its output signal becomes the torque command signal T ^* of the synchronous motor 51. The output signal T ^* of the speed control device 62 is input to the q-axis current command device 63, and the q-axis current command device 63 calculates a q-axis current command I _q ^* corresponding to the torque command signal T ^* . The q-axis current command I _q ^* is a command for a component orthogonal to the magnetic field direction of the armature current vector of the synchronous motor 51, and is input to the current control device 65. On the other hand, the d-axis current command device I _d ^* calculates the d-axis current command I _d ^* by a method as described later. The d-axis current command I _d ^* is a command having the same direction component as the magnetic field of the armature current vector of the synchronous motor 51. The main purpose of the command signal is to suppress not only the torque of the synchronous motor 51 but also an increase in output voltage. It is to issue a command signal for control. This d-axis current command signal I _d ^* is also input to the current controller 65.
[0013]
The current control device 65 is for controlling the actual current detected by the current detector 74 to flow as commanded based on the signal from the position detector 52, and outputs thereof are d-axis and q-axis. DC voltage commands V _d ^* and V _q ^* of The output signals V _d ^* and V _q ^* of the current control device 65 are input to the modulation wave generator 75, and the modulation wave generator 75 uses the DC voltage command V _d ^* , Modulation wave signal a ^* of AC amount corresponding to V _q ^* (In the case of three phases, there are three actual modulation wave signals of U, V, and W phases. Here, these are collectively a ^* . Output). The output signal a modulated wave generator 75 ^* is input to the PWM pulse generator 77 with the carrier signal s _c triangular, which is the output of a carrier generator 76.
[0014]
In the PWM pulse generator 77 compares the inputted modulated signal a ^* and carrier signal s _c, performs pulse width modulation to create a PWM pulse signal for driving the inverter 66, the PWM pulse signal to the inverter 66 Output.
[0015]
In the inverter 66, PWM control is executed by the PWM pulse signal from the PWM pulse generator 77, and the output voltage and output frequency of the inverter 66 are controlled.
[0016]
In this way, the current flowing through the synchronous motor 51 is controlled, and the torque and terminal voltage are controlled.
[0017]
FIG. 2 is a current and voltage vector diagram showing the control principle of FIG. In FIG.
I: armature currents I _d , I _q : d and q axis components E _{o of I} : no-load induced voltage V _t : terminal voltage X _d , X _q : reactance of d and q axes, X _d = X _ad + X ₁ , X _q = X _aq + X _l
X _ad , X _aq , X _l : d, q-axis armature reaction reactance, leakage reactance. From the figure, the output P of the synchronous motor is as shown in Equation (1).
[0018]
[Expression 1]

That is, the torque T is expressed by Equation (2).
[0019]
[Expression 2]

here,
ω: Rotational angular velocity of the motor shaft, ω = ω ₁ / (p / 2)
ω ₁ : electrical rotation angular frequency p: number of motor poles Φ: magnetic flux L _d , L _q : d, q-axis inductance,
[0020]
[Equation 3]
E ₀ = ω ₁ · Φ (3)
[0021]
[Expression 4]
X _d = ω ₁ · L _d , X _q = ω ₁ · L _q (4)
It is. At this time, when the synchronous motor 51 is a cylindrical machine, since L _d = L _q = L, the torque T is expressed as Equation (5).
[0022]
[Equation 5]
T = k ・ Φ ・ I _q (K is a constant) (5)
As a result, the torque T is proportional to only the q-axis component I _q of the current. Therefore, it can be seen that only the q-axis component I _q of the current needs to be controlled in order to control the torque.
Then the terminal voltage V _t of the synchronous motor, from the vector of Figure 2 is expressed as Equation (6).
[0023]
[Formula 6]

From equation (6), when the motor is rotating at a constant speed, the terminal voltage V _t can be zero the d-axis current component varies with the q-axis current component. That is, increasing the current component I _q to increase the torque, the terminal voltage V _t It can be seen to increase. At this time, it can be seen from Equation (6) that if the d-axis current component I _d is controlled to be negative, the magnitude of the q-axis component of the voltage is reduced, so that the terminal voltage as a vector sum can be suppressed.
[0024]
Here, when the inverter output voltage V _max is substituted for the motor terminal voltage V _t in equation (6) and further solved for I _d , equation (7) is obtained.
[0025]
[Expression 7]

From the formula (7), the determination of the field weakening control for suppressing the motor terminal voltage is as follows:
If the result of Equation (7) is I _d ≧ 0, field weakening control is not required (I _d ^* is set to 0)
If the result of Equation (7) is I _d <0, field weakening control is necessary (I _d ^* is set to I _d )
Can be expressed as
[0026]
Here, L _q , L _d , and Φ are motor constants that can be treated as being constant, and I _q and ω ₁ change depending on the motor operating state, and therefore, a detected value or a command value in the control device may be used. On the other hand, V _max varies depending on the DC voltage V _dc input to the inverter, and may be calculated by the following equation.
[0027]
[Equation 8]
V _max = (A _max · V _dc ) / (2√2) (8)
A _{max in} equation (8) is the maximum value of the modulation degree defined by the amplitude ratio of the carrier wave and the modulated wave, and is normally set to A _max = 1 or less so as not to exceed the peak of the carrier wave amplitude. If it does in this way, a low time harmonic component will not be included in the terminal voltage waveform of a motor. That is, since the terminal voltage waveform is not distorted, the torque pulsation generated from the motor can be suppressed, and it can be seen that the terminal voltage waveform is suitable for applications such as elevators where vibrations should be suppressed. Further, in applications that allow some of the torque pulsation is not necessary to set the following A _max = 1, A _max is found that may be set to a value greater than 1.
[0028]
FIG. 3 is a configuration example of the d-axis current command device 64 applying the above principle. The figure shows the case of a cylindrical machine (L _d = L _q = L). The q-axis current command signal I _q ^* is the q-axis current command device 63 and the torque command T ^* from the speed control device 62 is expressed by the equation (5). It is converted by the principle. The d-axis current calculator 642 receives the signal from the speed detector 53, the signal I _q ^* from the q-axis current command device 63, and the signal V _max from the output voltage setting unit 643 based on Expression (7). Then, a temporary d-axis current command I _dref is output. The field weakening control determiner 645 receives the signal I _dref from the d-axis current calculator 642 and determines the d-axis current command I _d ^* based on the above-described determination of field weakening control. That is, if I _dref ≧ 0, I _d ^* = 0, and if I _dref <0, I _d ^* = I _dref . The output voltage setting unit 643 sets the output voltage V _max of the inverter from the voltage detection signal V _dc of the DC voltage detector 73 based on the formula (8).
[0029]
By controlling the d-axis current command in this manner, the magnitude of the terminal voltage can be suppressed to a predetermined value, so that the withstand voltage of the motor and the control voltage range of the inverter can be reduced. This is effective when the output voltage of the inverter has no margin near the maximum rotation of the motor, and the output capacity of the inverter can be reduced. Furthermore, since the fluctuation of the DC voltage input to the inverter is also taken into consideration, even when the received voltage is lowered, the inverter does not run out of output voltage, and stable field-weakening control is executed.
[0030]
The current commands I _q ^* and I _d ^* thus obtained are input to the current command device 65. FIG. 4 shows a specific configuration example of the current control device 65. The basic configuration of this example is well known, and is described in, for example, IEEJ Transactions D, Vol. 117, No. 5 (July 1997), page 539, FIG. The configuration of FIG. 4 calculates the d and q axis voltage components from the vector diagram of FIG. 2, and further calculates ACR-d and ACR-q that work according to the deviation between the d and q axis current commands and the actual values. I have. Also, figure further consideration of the armature resistance R _a.
[0031]
The I _d / I _q calculation 651 is a three-phase instantaneous current detection value iu from the current detector 74 based on a sine or cosine signal corresponding to the field magnetic pole position (electrical rotation angle) θ from the position detector 52. , Iv, iw are used to detect the components I _d , I _{q of} each current. The output of the current controller 65 is d and q axis voltage command signals V _d ^* and V _q ^* , and the modulation wave generator 75 performs the reverse operation of the I _d / I _q operation 651 to generate a PWM pulse. A sinusoidal modulation wave signal a ^* used for PWM pulse generation in the device 77 is generated. Since this calculation is well known, it will be omitted.
[0032]
FIG. 5 is a diagram showing characteristics depending on the presence or absence of such control. (A) is a characteristic example when the control of the present invention is performed, (b) is the control of the present invention, the d-axis current is zero, and only the q-axis current control is performed according to the torque. The example of the characteristic of the conventional control is shown.
[0033]
In the example of (a), the d-axis current I _d is increased in the negative direction in accordance with the increase of the torque command from the point of the torque command a that becomes the condition of the field weakening control. Further, the q-axis current is proportional to the torque as can be seen from Equation (5). When the d-axis current controlling Thus, d, q-axis current armature current I is the vector sum of the components, although slightly increased than I _q, the terminal voltage V _t is hardly increased from no load. Furthermore, also at points b to c when the DC voltage of the inverter is lowered, the d-axis current is automatically increased in the negative direction, and the terminal voltage can be suppressed to within the output range of the inverter. For this reason, the variable range of the inverter output voltage can be made appropriate with the minimum necessary level. As a result, an increase in the capacity of the inverter can be suppressed.
[0034]
On the other hand, when the zero d-axis component so as in (b), however, the terminal voltage V _t increases with increasing torque command. Further, when the DC voltage of the inverter is reduced, the output voltage range of the inverter may be exceeded as shown in (b). As a result, in order to prevent such a phenomenon, the output voltage of the inverter needs to be large, and as a result, the capacity of the inverter increases.
[0035]
As described above, according to the present invention, the variable range of the output voltage of the inverter can be reduced, so that the inverter capacity can be reduced and the withstand voltage of the motor is not increased. For this reason, a small and economical system can be provided.
[0036]
FIG. 6 shows an alternative embodiment to that shown in FIG. In the figure, the same reference numerals as those in FIG. The field weakening control determination unit 641 in FIG. 6 performs the determination at the time of field weakening control based on the signal from the modulation wave generator 75 and the signal from the carrier wave generator 76 and outputs a switching signal to the switching circuit 644. Specifically, selecting the output signal I _dref of d-axis current calculation unit 642 when the amplitude of the modulated wave signal a ^* reaches a peak value of the carrier signal s _c (peak value), d-axis current command I _d ^* The signal to be output is output to the switching circuit 644. In addition, a 0 signal is input to the d-axis current command I _d ^* by the switching circuit 644 in a state other than the state described above.
[0037]
In this way, the field-weakening control decision (starting point) can be determined directly from the relationship between the carrier wave that determines the output voltage of the inverter and the modulated wave, so the motor terminal voltage waveform does not contain low-time harmonic components and is generated from the motor. Torque pulsation can be reliably suppressed, and the terminal voltage does not exceed a predetermined value. Therefore, the margin of the variable range of the output voltage of the inverter can be reduced, and further, the output capacity of the inverter can be reduced.
[0038]
In the above example, the amplitude of the modulated wave signal a ^* is explained that when the peak value of the carrier signal s _c (peak value), to define the amplitude ratio of the modulated signal a ^* and carrier signal s _c The amplitude ratio may reach a predetermined value set in advance. Further, the output voltage setting unit 643 sets the inverter output voltage V _max from the voltage detection signal V _dc of the inverter DC voltage detector 73. As described above, the field weakening control determination (starting point) is performed. Is directly determined from the relationship between the carrier wave and the modulated wave, the output voltage V _max may be set to a predetermined constant value without using the DC voltage detection signal. FIG. 7 shows a further embodiment different from that shown in FIGS. In the figure, the same reference numerals as those in FIG. 3 and FIG. The field weakening control determination unit 646 in FIG. 7 performs the determination at the time of field weakening control based on the PWM pulse signal from the PWM pulse generator 77 and outputs a switching signal to the switching circuit 644. Specifically, when the on-time of the PWM pulse signal reaches the period of the carrier wave signal (carrier period of the inverter), the output signal I _dref of the d-axis current calculator 642 is selected as the d-axis current command I _d ^*. The signal is output to the switching circuit 644. In addition, a 0 signal is input to the d-axis current command I _d ^* by the switching circuit 644 in a state other than the state described above.
[0039]
In this way, the field-weakening control determination (starting point) can be determined directly from the PWM pulse signal that determines the output voltage of the inverter, so that the margin of the variable range of the inverter output voltage can be reduced as in the embodiment of FIG. Furthermore, the output capacity of the inverter can be reduced.
[0040]
In the above example, the PWM pulse signal ON time has been described as reaching the carrier cycle of the inverter. However, the time ratio between the PWM pulse signal ON time and the carrier cycle is defined, and this time ratio is set to a predetermined value. It may be when the value is reached.
[0041]
【The invention's effect】
As described above, according to the present invention, the current component in the same direction as the magnetic field of the motor is set so that the value of the terminal voltage of the permanent magnet type synchronous motor is within the range of the maximum output voltage that can be output from the inverter. Since (d-axis current component) is controlled, an increase in motor terminal voltage can be suppressed, so that an increase in motor output voltage and current can be reduced, resulting in an increase in inverter capacity and motor dielectric strength. Can be suppressed. For this reason, the whole control system can be constructed in a small and economical manner.
[Brief description of the drawings]
FIG. 1 is a diagram illustrating a method for controlling a permanent magnet synchronous motor according to an embodiment of the present invention.
FIG. 2 is a vector diagram for explaining a control principle.
3 is a diagram showing a configuration example of a d-axis current command device in FIG. 1. FIG.
FIG. 4 is a diagram illustrating a configuration example of a current control device.
FIG. 5 is a graph showing characteristics according to the present invention.
FIG. 6 is a diagram showing another configuration example of the d-axis current command device.
FIG. 7 is a diagram showing another configuration example of the d-axis current command device.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 51 ... Synchronous motor 52 ... Position detector 53 ... Speed detector 61 ... Speed command device 62 ... Speed control device 63 ... q-axis current command device 64 ... d-axis current command device 65 ... Current control device , 66 ... Inverter, 70 ... AC power supply, 71 ... Converter, 72 ... Smoothing capacitor, 73 ... DC voltage detector, 74 ... Current detector, 75 ... Modulation wave generator, 76 ... Carrier wave generator, 77 ... PWM pulse generation device, 641,645,646 ... weakening control determiner, 642 ... d-axis current calculation unit, 643 ... variable output voltage setter 644 ... switching circuit, 651 ... I _d / I _q operation.

Claims (3)

A method for controlling a permanent magnet synchronous motor driven by a power converter composed of an inverter that inputs a DC voltage and converts it into an alternating current of variable voltage and variable frequency,
Based on the relationship between the torque and the rotational speed of the DC voltage input to the inverter motor, and calculates a command value of the temporary magnetic field in the same direction of the current component of the motor (d-axis current component), the temporary If the command value is zero or positive, the the d-axis current component to zero, if the command value of said provisional negative polarity, wherein the benzalkonium to follow the d-axis current component command value of the temporary A control method of a permanent magnet type synchronous motor. A method for controlling a permanent magnet synchronous motor driven by a power converter composed of an inverter that inputs a DC voltage and converts it into an alternating current of variable voltage and variable frequency,
A temporary command value of a current component (d-axis current component) in the same direction as the motor magnetic field based on the relationship between the set value or detected value of the DC voltage input to the inverter and the torque and rotation speed of the motor. calculating a, to follow the at the d-axis current component and the modulation wave and the carrier wave becomes a predetermined relationship to generate a PWM pulse input to the inverter command value of the temporary, it said in another state A control method of a permanent magnet type synchronous motor, wherein a d-axis current component is zero . A method for controlling a permanent magnet synchronous motor driven by a power converter composed of an inverter that inputs a DC voltage and converts it into an alternating current of variable voltage and variable frequency,
A temporary command value of a current component (d-axis current component) in the same direction as the motor magnetic field based on the relationship between the set value or detected value of the DC voltage input to the inverter and the torque and rotation speed of the motor. calculating a, to follow the at the d-axis current component and the carrier wave period becomes a predetermined relationship on time and the inverter PWM pulse input to the inverter command value of the temporary, it said in another state A control method of a permanent magnet type synchronous motor, wherein a d-axis current component is zero .

Images