WO2019073599A1 - Motor drive device, refrigeration cycle device equipped with same, and motor drive method - Google Patents

Abstract

Provided are a motor drive device and the like capable of both suppressing vibration and reducing loss. The motor drive device (50) is provided with a control unit (51) for changing, in accordance with the change in torque current on the q-axis of a rotating coordinate system of a motor (M), excitation current on the d-axis. By changing the excitation current in accordance with the change in the torque current like this, it is possible to suppress the vibration and noise of the motor (M) and the like and drastically reduce the loss of the motor (M) and the like in comparison with a conventional loss even if periodical variation of a load torque occurs during driving of the motor (M).

Description

Motor drive device, refrigeration cycle device including the same, and motor drive method

The present invention relates to a motor drive device and the like for driving a motor.

In rotary compressors and reciprocating compressors, it is known that load torque periodically fluctuates in the compression process of the refrigerant. As a technique of suppressing the vibration and noise accompanying such fluctuation of load torque, for example, the technique described in Patent Document 1 is known. That is, Patent Document 1 describes that the motor is controlled so as to make the difference between the output torque of the motor and the load torque (pulsating torque) of the compressor zero.

In the technology described in Patent Document 1 described above, although the vibration and the like of the compressor can be suppressed, the peak value of the motor current largely fluctuates with the periodic fluctuation of the load torque, leading to an increase in loss. Therefore, as a technique for reducing the loss of the motor, for example, Patent Document 2 describes that the peak value of the motor current is kept approximately constant.

Patent No. 4221307 Patent No. 4958431

As described above, although the technology described in Patent Document 1 can suppress the vibration of the compressor, it causes an increase in loss of a motor or the like (motor or circuit component). On the other hand, in the technique described in Patent Document 2, although loss of a motor or the like can be reduced, vibration of the compressor is relatively large. That is, the suppression of the vibration of the compressor and the reduction of the loss of the motor or the like are in a trade-off relationship.

Then, this invention makes it a subject to provide the motor drive device etc. which are compatible in suppression of a vibration, and reduction of a loss.

In order to solve the problems described above, the present invention is characterized in that the excitation current of the d-axis is changed according to the change of the torque current of the q-axis in the rotational coordinate system of the motor.

According to the present invention, it is possible to provide a motor drive device and the like that can achieve both suppression of vibration and reduction of loss.

It is explanatory drawing of the air conditioner provided with the motor drive device which concerns on 1st Embodiment of this invention. It is a block diagram of an air conditioner provided with a motor drive concerning a 1st embodiment of the present invention. FIG. 7 is an explanatory view showing a load torque of a compressor, an output torque of a motor, a rotational speed, and a motor current when the motor is rotated once at a mechanical angle in torque control. FIG. 6 is an explanatory view showing a load torque of a compressor, an output torque of a motor, a rotational speed, and a motor current when the motor is rotated once at a mechanical angle in constant current control. It is a block diagram of a control part with which a motor drive concerning a 1st embodiment of the present invention is provided. FIG. 5 is a Bode diagram in the case of using a predetermined transfer function in the motor drive device according to the first embodiment of the present invention. It is an experimental result which shows the waveform of the motor current of a dq coordinate system at the time of driving a motor on predetermined conditions based on "high efficiency torque control" of 1st Embodiment of this invention. It is an experimental result which shows the waveform of the three-phase motor current at the time of driving a motor on predetermined conditions based on "high efficiency torque control" of 1st Embodiment of this invention. It is a graph which shows the effective value and motor copper loss of a motor current in a comparative example, and the effective value and motor copper loss of a motor current in 1st Embodiment of this invention. It is a block diagram of the control part with which the motor drive concerning a 2nd embodiment of the present invention is provided. It is a block diagram containing the torque pulsation estimation part of the motor drive concerning a 2nd embodiment of the present invention. It is explanatory drawing of the speed fluctuation suppression control part with which the control part of the motor drive device which concerns on 2nd Embodiment of this invention is provided. It is explanatory drawing which shows the relationship between the torque current of a motor, and q axis | shaft inductance. It is explanatory drawing which shows the relationship between the excitation current of a motor, and d axis | shaft inductance. It is a comparative example which shows the waveform of the motor current of a dq coordinate system at the time of driving a motor on predetermined conditions based on "torque control." It is a comparative example which shows the waveform of the three-phase motor current at the time of driving a motor under predetermined conditions based on "torque control".

Below, the structure which drives the compressor 11 of the air conditioner 100 (refer FIG. 2) by the motor M is demonstrated as an example.

First Embodiment
<Configuration of air conditioner>
FIG. 1 is an explanatory view of an air conditioner 100 provided with a motor drive device according to the first embodiment.
The air conditioner 100 (refrigerating cycle device) is a device that performs air conditioning such as cooling operation and heating operation. As shown in FIG. 1, the air conditioner 100 includes an outdoor unit Go, an indoor unit Gi, and a remote control Re.

The outdoor unit Go houses the compressor 11 (see FIG. 2), the outdoor heat exchanger 13 and the like. The indoor unit Gi accommodates the indoor heat exchanger 14 (see FIG. 2), the indoor fan Fi, and the like. The outdoor unit Go and the indoor unit Gi are connected via a pipe k and connected via a communication line (not shown). The remote control Re transmits an operation signal such as an operation / stop command, a change of the set temperature, a change of the operation mode, and the like to the indoor unit Gi.

FIG. 2 is a block diagram of the air conditioner 100 provided with the motor drive device 50. As shown in FIG.
As shown in FIG. 2, the air conditioner 100 includes a refrigerant circuit 10, an outdoor fan Fo, and an indoor fan Fi. Moreover, the air conditioner 100 is provided with the motor M, the converter 20, the inverter 30, the electric current detector 40, and the motor drive device 50 other than the above-mentioned structure.

The refrigerant circuit 10 is a circuit through which the refrigerant circulates, and includes a compressor 11 (load), a four-way valve 12, an outdoor heat exchanger 13, an indoor heat exchanger 14, and an expansion valve 15. Ru.

The compressor 11 is a device that compresses a gaseous refrigerant, and is connected to the rotor of the motor M. The compressor 11 has a characteristic that the load torque (pulsating torque) periodically changes in the compression process of the refrigerant. As such a compressor 11, although a rotary compressor and a reciprocating compressor are mentioned, for example, it is not limited to this.
The motor M is, for example, a permanent magnet synchronous motor, and is connected to the compressor 11. As such a motor M, a salient pole type synchronous motor (salient pole machine) is mentioned.

The four-way valve 12 is a valve that switches the flow direction of the refrigerant. That is, at the time of heating operation (solid arrow in FIG. 2), the four-way valve 12 is controlled such that the indoor heat exchanger 14 functions as a condenser and the outdoor heat exchanger 13 functions as an evaporator. On the other hand, at the time of cooling operation (broken line arrow in FIG. 2), the four-way valve 12 is controlled so that the outdoor heat exchanger 13 functions as a condenser and the indoor heat exchanger 14 functions as an evaporator.

That is, the refrigerant circuit 10 includes the compressor 11, the condenser (one of the outdoor heat exchanger 13 and the indoor heat exchanger 14), the expansion valve 15, and the evaporator (the outdoor heat exchanger 13 and the indoor heat exchanger 14). And the other are sequentially connected in an annular fashion via the four-way valve 12. Then, the refrigerant circulates in a known refrigeration cycle (heat pump cycle) in the refrigerant circuit 10 based on an operation signal from the remote control Re (see FIG. 1) and detection values of various sensors (not shown).

The outdoor heat exchanger 13 is a heat exchanger in which heat exchange is performed between the outside air and the refrigerant.
The outdoor fan Fo is a fan that sends outside air to the outdoor heat exchanger 13 and is installed near the outdoor heat exchanger 13.

The indoor heat exchanger 14 is a heat exchanger in which heat exchange is performed between indoor air (air in a space to be air-conditioned) and a refrigerant.
The indoor fan Fi is a fan that sends indoor air to the indoor heat exchanger 14, and is installed near the indoor heat exchanger 14.

The expansion valve 15 is a valve that reduces the pressure of the refrigerant condensed by the above-described "condenser". The refrigerant decompressed by the expansion valve 15 is led to the aforementioned "evaporator".

Converter 20 is a power converter that converts an AC voltage applied from AC power supply E into a DC voltage.
The inverter 30 is a power converter that converts a DC voltage applied from the converter 20 into an AC voltage and applies the AC voltage to the winding of the motor M. As such an inverter 30, for example, a three-phase full bridge inverter can be used.

The current detector 40 is, for example, a shunt resistor, and detects a current supplied from the converter 20 to the inverter 30. The detection value of the current detector 40 is output to the control unit 51 of the motor drive device 50 described below.

The motor drive device 50 drives the motor M to drive the compressor 11 coupled to the motor M at variable speeds. As shown in FIG. 2, the motor drive device 50 includes a control unit 51. Although not shown, the control unit 51 includes electronic circuits such as a central processing unit (CPU), a read only memory (ROM), a random access memory (RAM), and various interfaces. Then, the program stored in the ROM is read and expanded in the RAM, and the CPU executes various processing.

<Torque control / constant current control / high efficiency torque control>
Next, after briefly describing “torque control” and “constant current control” which have been performed up to this point, “high efficiency torque control” of the present embodiment will be described.

FIG. 3 is an explanatory view showing a load torque of the compressor 11, an output torque of the motor M, a rotational speed, and a motor current when the motor M is rotated once at a mechanical angle in torque control (as appropriate). See).
The “torque control” is control for changing the output torque of the motor M so as to match the load torque of the compressor 11. As described above, when the motor M is rotated once at a mechanical angle, the load torque of the compressor 11 periodically pulsates. In the example shown in FIG. 3, the load torque shown by a broken line is pulsating once in the process of the motor M making one rotation at a mechanical angle.

In this torque control, the output torque (solid line) shown in FIG. 3 is made to coincide with the load torque (broken line), and the rotational speed of the motor M is made constant. Vibration and noise of the compressor 11 are thereby suppressed. On the other hand, since the crest value of the motor current fluctuates significantly with the fluctuation of the load torque, the loss of the motor M and the like (motor M and circuit components) becomes a relatively large value although not shown.

FIG. 4 is an explanatory view showing a load torque of the compressor 11, an output torque of the motor M, a rotational speed, and a motor current when the rotor of the motor M is rotated once at a mechanical angle in constant current control. See Figure 2 as appropriate).
The "constant current control" is control to make the output torque of the motor M constant regardless of the fluctuation of the load torque.

As shown in FIG. 4, in constant current control, the output torque of the motor M is maintained constant, so the peak value of the motor current also becomes constant. Thereby, the loss of the motor M or the like can be reduced. On the other hand, since the rotational speed of the motor M fluctuates greatly, the compressor 11 easily vibrates.

Thus, the suppression of the vibration of the compressor 11 and the reduction of the loss of the motor M etc. are in a trade-off relationship. So, in 1st Embodiment, in order to make suppression of the vibration of the compressor 11, and reduction of the loss of the motor M etc. make compatible, "high efficiency torque control" is performed.
“High-efficiency torque control” refers to the control unit 51 (see FIG. 2) according to the change in q-axis torque current I _q in the dq coordinate system (rotational coordinate system) of the motor in which cyclical fluctuations in load torque occur. There is a control to vary the excitation current I _d of d axis. The details of “high efficiency torque control” will be described later.

<Configuration of control unit>
FIG. 5 is a block diagram of the control unit 51 provided in the motor drive device 50. As shown in FIG.
As shown in FIG. 5, the control unit 51 includes a three-phase / two-axis conversion unit 51a, an axis error calculation unit 51b, a PLL circuit 51c, an integrator 51d, a subtractor 51e, and a speed control unit 51f. And a speed fluctuation suppression control unit 51g. In addition to the above-described configuration, the control unit 51 further includes an adder 51h, a subtractor 51i, an optimal phase control unit 51j, another subtractor 51k, a current control unit 51m, and a voltage command calculation unit 51n. , A 2-axis / 3-phase converter 51r, and a PWM signal generator 51s.

The control unit 51 reproduces the currents (I _u , I _v , I _w ) of the three-phase coordinate system based on the detection value of the current detector 40 (see FIG. 2). Then, the value of the reproduced current (I _u , I _v , I _w ) is input to the three-phase / two-axis conversion unit 51a as a current detection value.

The three-phase / two-axis conversion unit 51a is based on the phase θ _dc of the rotor of the motor M (see FIG. 2), and generates currents (I _u , I _v , I _w ) of the three-phase coordinate system It is converted into current detection values (I _dc , I _qc ). The direction of the actual magnet magnetic flux に お け る in the motor M is d axis, and the axis orthogonal to the d axis is q axis. Also, the subscript “c” in the current detection value (I _dc , I _qc ) means that it is based on the detection value of the current or the like.

The axis error calculation unit 51 b estimates an axis error Δθ between the real axis and the control axis regarding the magnet flux of the motor M. More specifically, the axis error computing unit 51b is an axis error between the phase of the actual magnet flux に お け る in the motor M and the phase θ _dc (control phase) which is the computation result of the integrator 51d described later. For example, Δθ is calculated using the following equation (1).
Here, R shown in the equation (1) is a winding resistance of the motor M, and ω _r is a calculation result of the rotational speed of the motor M. _{Further, L dc} is d-axis inductance of the motor _{M, L qc} is q-axis inductance of the motor M. Also, the superscript “*” attached to the d-axis voltage command V _d ^* or the like represents that it is a command value.

The momentary axis error Δθ that is the calculation result of the axis error calculator 51b is output to a PLL circuit 51c (Phase Locked Loop) shown in FIG. Thus, the control unit 51 estimates the momentary axis error Δθ based on the instantaneous value of the torque current I _{qc and} the instantaneous value of the excitation current I _dc . The control unit 51 controls the motor M without using a position sensor based on the axis error Δθ.

In addition, Formula (2) which is a principle formula (principle formula derived from the predetermined formula regarding the motor M) regarding calculation of axial error (DELTA) (theta) of the motor M is shown below for reference.

In Equation (2), the third and fourth terms of the denominator are differential terms including differential operations, and the third and fourth terms of the numerator are also differential terms. Conventionally, equation (3) shown below has been used in a form in which the differential terms whose processing is relatively complicated are omitted.

For example, in the stable state where there is no periodic torque fluctuation, the time differential value of the excitation current I _dc and the torque current I _qc becomes substantially zero, so that the axis error Δθ can be properly calculated also by the equation (3). On the other hand, in the present embodiment, as described later, the torque current I _qc is changed to resist the periodic load torque, and the excitation current I _dc is also changed according to the torque current I _qc. ing. In the case of such processing, in order to calculate the momentary axis error Δθ with high accuracy, in the present embodiment, equation (1) is used.

By the way, even in the case of the equation (1) in which the fourth term (differential term of the axis error Δθ) of the denominator and numerator of the equation (2) which is the principle equation is omitted, the axis error Δθ is sufficiently accurately calculated. it can.

The PLL circuit 51c shown in FIG. 5 calculates the rotational speed ω _r of the motor M based on PI control (Proportional Integral control) so that the above-mentioned axis error Δθ becomes zero. By this, since the dc axis and dq axis based on the current detection value and the like coincide with the d axis and q axis corresponding to the actual magnet flux モ ー タ of the motor M, the motor M can be controlled without position sensor.

The integrator 51 _d calculates the phase θ _dc of the rotor of the motor M by integrating the rotational speed ω _r .
Subtractor 51e, the upper system and the rotational speed command omega _r ^* inputted from the (not shown), the rotation speed omega _r of the motor M is inputted from the PLL circuit 51c, the difference (ω _r ^* -ω _r) the calculated as the rotational speed deviation [Delta] [omega _r.

Speed control unit 51f, based on the rotational speed deviation [Delta] [omega _r inputted from the subtractor 51e, for example, by the PI control, and calculates a torque current command I _q0 ^* corresponding to the average torque of the motor M.

The speed fluctuation suppression control unit 51 g calculates a pulsating torque current command I _qsin ^* that changes in a sinusoidal manner, in order to suppress the speed fluctuation associated with the periodic torque fluctuation of the motor M. Specifically, the speed variation suppression control unit 51g performs rotation based on the rotation speed command ω _r ^* and the rotation speed deviation Δω _r , for example, using a transfer function G (s) shown in the following equation (4). Pulsating torque current command I _qsin ^* is calculated such that speed deviation Δω _r becomes zero.

The subscript “sin” of the pulsating torque current command I _qsin ^* indicates that the waveform is sinusoidal (sin curve shape). Further, s shown in the equation (4) is a Laplace operator, K ₁ , K ₂ and K ₃ are control coefficients, and ω ₀ is a predetermined center frequency.

The transfer function G (s) shown in the equation (4) has a characteristic that it has sensitivity (gain) at a predetermined center frequency ω ₀ and almost no sensitivity to other frequencies. Therefore, by setting the value of the central frequency ω ₀ to the rotational speed command ω _r ^* , the speed fluctuation suppression control unit 51 g can be configured to react only to the angular frequency of the rotational speed command ω _r ^* . Thereby, it is possible to without raising almost the sensitivity of a frequency different from the rotational speed command omega _r ^*, the rotation speed command omega _r ^* of high sensitivity (high gain of). There is also an advantage that the rotational speed deviation Δω _r can be made substantially zero.

Note that another transfer function may be used as long as the transfer function is sensitive to a specific frequency. For example, instead of the equation (4), a transfer function G (s) shown in the following equation (5) may be used. Here, K ₄ and K ₅ shown in the equation (5) are control coefficients, and ω ₀ is a predetermined center frequency.

FIG. 6 is a Bode diagram when the transfer function G (s) of equation (5) is used.
As shown in FIG. 6, the gain and the phase are largely changed in the vicinity of the predetermined center frequency ω ₀ . As described above, by using the transfer function G (s) such as equation (5), it is possible to achieve high sensitivity (high gain) of the rotational speed command ω _r ^* .

The adder 51h shown in FIG. 5 is the sum of torque current command I _q0 ^* , which is the calculation result of speed control unit 51 _f, and pulsation torque current command I _qsin ^* , which is the calculation result of speed variation suppression control unit 51 g (I By taking _q0 ^* + I _qsin ^* ), a new torque current command I _q ^* is calculated.

As described above, the torque current command I _q0 ^* is a current command value corresponding to the average torque of the motor M. Further, the pulsating torque current command I _qsin ^* is a current command value for suppressing a periodic torque fluctuation. Adder torque current command is a calculation result of ^{_{51h I q * (= I q0}} * + I qsin *) , as against the load torque of the compressor 11 to vary periodically vary sinusoidally (pulsation) .

The subtractor 51i shown in FIG. 5 is a difference between the torque current command I _q ^* , which is the calculation result of the adder 51 _h , and the torque current I _qc (detection value), which is the calculation result of the three-phase / two-axis converter 51a. Calculate ΔI _q (= I _q ^* −I _qc ). The difference ΔI _q is input to a current control unit 51 m described later.

Optimum phase controller 51j on the basis of the adder torque current command is a calculation result of _{^{51h I q * (= I q0}} * + I qsin *), using the following equation (6), every moment of the exciting current command I Calculate _d ^* . Incidentally, Ke shown in equation (6) is the induced voltage constant, L _d is d-axis inductance of the motor M, L _q is q-axis inductance of the motor M.

So far, the excitation current command I _d ^* has been calculated based on the temporal average value of the torque current command I _q ^* in equation (6). On the other hand, in the present embodiment, the optimum phase control unit 51 j uses the equation (6) to calculate the exciting current command I _d ^* based on the instantaneous value (rather than the temporal average value) of the torque current command I _q ^{* .} Is calculated. Thus, in response to changes in momentary torque current I _q which pulsates sinusoidally, it is possible to vary the excitation current I _d.

More specifically, the control unit 51 periodically changes the torque current I _q in the positive region and periodically changes the excitation current I _d in the negative region (see FIG. 7A). Then, the control unit 51, as the absolute value of the torque current I _q is small, so that also decreases the absolute value of the exciting current I _d. It is to be noted that each of the above-mentioned "positive side area" and "negative side area" includes a value of zero.

In addition, from another viewpoint, the control unit 51 changes the excitation current _Id in a sine wave so as to be in reverse phase to the sine wave torque current I _q (see FIG. 7A).

When the load torque which fluctuates periodically during driving of the motor M is large (in FIG. 7A, the crank angle of the compressor 11: around 200 °, 560 °) by such “high efficiency torque control”, the control unit 51 increases the torque current I _q and increases the absolute value of the excitation current I _d (negative value). Thus, to take full advantage of the reluctance torque due to the excitation current I _d, while suppressing the vibration of the compressor 11, the loss can be reduced, such as a motor M.

Further, during driving of the motor M, when the periodically changing load torque is small (in FIG. 7A, the crank angle of the compressor 11: around 40 ° and 400 °), the control unit 51 reduces the torque current I _q . At the same time, the absolute value of the excitation current I _d (negative value) is reduced. As compared with the prior art (constant current control) in which a constant excitation current I _d flows regardless of the mechanical angle of the motor, this prevents unnecessary large current from flowing and, in turn, the loss of the motor M etc. It can be reduced.

Again, returning to FIG. 5, the explanation will be continued.
The subtractor 51k shown in FIG. 5 is the difference ΔI _d (= I _d ^* -) between the excitation current command I _d ^* and the excitation current I _dc (detection value) which is the calculation result of the 3-phase / 2-axis conversion unit 51a. Calculate I _dc ).

The current control unit 51m generates a second excitation current command I _d ^** such that the difference ΔI _d that is the calculation result of the subtractor 51k and the difference ΔI _q that is the calculation result of another subtractor 51i become zero. And a second torque current command I _q ^** .

Voltage command calculation unit 51 _n is based on second excitation current command I _d ^** and second torque current command I _q ^** , using a known voltage equation to obtain voltage commands (V _d ^* , V _q ^* ). Calculate

The two-axis / three-phase conversion unit 51r generates three-phase voltage commands (V _u ^* , V _d ^* , V _q ^* ) based on the phase θ _dc which is the calculation result of the integrator 51 _d ^. Convert to V _v ^* , V _w ^* ).

The PWM signal generation unit 51s generates a PWM signal based on PWM control (Pulse Width Modulation control) based on the three-phase voltage commands (V _u ^* , V _v ^* , V _w ^* ). The PWM signal switches on / off of each switching element (not shown) of the inverter 30 (see FIG. 2).

<Experimental result>
FIG. 13A is a comparative example showing the waveform of the motor current in the dq coordinate system when the motor M is driven under the conditions shown in Table 1 based on the “torque control”.
The horizontal axis in FIG. 13A is the crank angle of the compressor 11 (see FIG. 2), and the vertical axis is the motor current (excitation current I _d , torque current I _q ).

In the above-mentioned "torque control", the torque current I _q changes in a sine wave so as to resist the periodically changing load torque. On the other hand, regardless of the crank angle of the compressor 11 (mechanical angle of the motor M), the excitation current _Id is constant.

FIG. 13B is a comparative example showing waveforms of three-phase motor currents when the motor M is driven under the conditions shown in Table 1 based on “torque control”.
The horizontal axis in FIG. 13B is the crank angle of the compressor 11 (see FIG. 2), which corresponds to the crank angle in the horizontal axis of FIG. 13A. That is, the waveform diagram shown in FIG. 13B is obtained as a result of controlling the torque current I _q and the excitation current I _d of the motor M as shown in FIG. 13A. The vertical axis in FIG. 13B is three-phase motor current (current of U-phase, V-phase, W-phase).

FIG. 7A is an experimental result showing a waveform of a motor current in a dq coordinate system when the motor M is driven under the conditions shown in Table 1 based on the “high efficiency torque control” of the present embodiment.
As described above, the control unit 51 sinusoidally changes the torque current I _q so as to resist the periodically changing load torque. Further, the control unit 51 changes the excitation current _Id in a sine wave so that the phase is opposite to that of the torque current _Iq .

FIG. 7B is an experimental result showing a waveform of a three-phase motor current when the motor M is driven under the conditions shown in Table 1 based on the “high efficiency torque control” of this embodiment.
The crank angle on the horizontal axis in FIG. 7B corresponds to the crank angle on the horizontal axis in FIG. 7A.
For example, in a region where the three-phase motor current increases (a region requiring a large output torque), the control unit 51 increases the absolute value of the excitation current I _d (negative value) (see FIG. 7A). This makes it possible to reduce the peak of the peak value of the three-phase motor current because the reluctance torque is utilized to the maximum. Furthermore, compared with the comparative example of FIG. 13A, even if the peak value of the torque current I _q is relatively small (see FIG. 7A), a sufficient torque (pulsating torque) to resist the load torque can be obtained.

Further, in a region where the three-phase motor current decreases (a region where output torque is almost unnecessary), the control unit 51 reduces the absolute value of the excitation current I _d (negative value). As a result, it is possible to suppress the wasteful flow of the three-phase motor current. For example, in the experimental result shown in FIG. 7B, the crest value in the vicinity of the crank angle of 40 ° and 400 ° of the compressor 11 is smaller than that of the comparative example shown in FIG. 13B. As described above, according to the present embodiment, the peak value of the three-phase motor current in the region where output torque is almost unnecessary can be made close to zero. As a result, the loss of the motor M and the like can be significantly reduced as compared with the prior art.

Incidentally, the output torque T of the motor M is given by the following equation (7). Here, P _m shown in the equation (7) is a pole pair number.

As shown in the equation (7), in the output torque T of the motor M, the magnet torque (P _m · Ke · i _q ) by the magnet and the reluctance torque (−P _m ( L _d −L _q ) i _q • _id ) and are included. In the present embodiment, the exciting current i _d by optimally controlled according to the torque current i _q, it is possible to maximize the output torque T of the motor M (minimizing motor current).

FIG. 8 is a graph showing an effective value of motor current and motor copper loss in a comparative example, and an effective value of motor current and motor copper loss in the present embodiment.
A comparative example is an experimental result at the time of driving motor M on the conditions shown in Table 1 based on the above-mentioned "torque control." Moreover, the motor M is driven under the conditions shown in Table 1 also in the present embodiment in which "high efficiency torque control" is performed.

As shown in FIG. 8, in the present embodiment, the effective value of the motor current is smaller than that of the comparative example. Further, in the present embodiment, the motor copper loss is significantly reduced as compared with the comparative example. Although omitted in FIG. 8, it is possible to reduce, in addition to the motor copper loss, the conduction loss and the switching loss in each switching element (not shown) of the inverter 30 (see FIG. 2).

<Effect>
According to the first embodiment, the control unit 51, in accordance with the variation of the torque current I _q, it executes the "high-efficiency torque control" to vary the excitation current I _d. As a result, it is possible to reduce the loss of the motor M and the like (that is, to improve the efficiency) in a state where the torque control is applied to the maximum. As a result, the vibration and noise of the compressor 11 can be suppressed, and the loss of the motor M and the like can be significantly reduced as compared with the prior art.

Further, the control unit 51 estimates the axial error Δθ every moment based on the above equation (1) based on the instantaneous value of the torque current I _{qc and} the instantaneous value of the excitation current I _dc . As a result, the axis error Δθ can be calculated with high accuracy, and consequently, “high efficiency torque control” can be appropriately performed.

Second Embodiment
The second embodiment is different from the first embodiment in that a control unit 51A (see FIG. 9) of the motor drive device includes a torque pulsation estimation unit 51t (see FIG. 9). In the second embodiment, the configuration of the speed fluctuation suppression control unit 51Ag (see FIG. 9) is different from that of the first embodiment. The other aspects are the same as in the first embodiment. Therefore, only the parts different from the first embodiment will be described, and the descriptions of the overlapping parts will be omitted.

FIG. 9 is a block diagram of a control unit 51A provided in the motor drive device according to the second embodiment.
The control unit 51A includes a torque pulsation estimation unit 51t, a speed fluctuation suppression control unit 51Ag, and the like. The torque pulsation estimation unit 51t estimates a torque pulsation component (periodic disturbance) in the motor M.

FIG. 10 is a configuration diagram including a torque pulsation estimation unit 51t of the motor drive device.
As shown in FIG. 10, the torque pulsation estimation unit 51t includes a proportional gain calculation unit 511t and multipliers 512t and 513t.

The proportional gain calculation unit 511t multiplies the axis error Δθ, which is the calculation result of the axis error calculation unit 51b, by a predetermined proportional gain (2J / P). J included in the proportional gain (2J / P) is the inertia of the compressor 11 and the motor M, and P is the number of poles of the motor M.

The multiplier 512t calculates the square of the rotational speed ω _r of the motor M, which is the calculation result of the PLL circuit 51c. The other multiplier 513t calculates a torque pulsation component ΔT _m by multiplying the calculation result of the proportional gain calculation unit 511t and the calculation result of the multiplier 512t. The torque pulsation component ΔT _m is input to a speed fluctuation suppression control unit 51Ag (see FIG. 11) to be described next.

FIG. 11 is an explanatory diagram of the speed fluctuation suppression control unit 51Ag provided in the control unit 51A of the motor drive device.
As shown in FIG. 11, the speed variation suppression control unit 51Ag includes a signal generation unit g1, a Fourier transform unit g2, an integral compensator g3, and a Fourier inverse transform unit g4.

The signal generator g1 generates signals of sin component and cos component of the rotational speed command ω _r ^* .
The Fourier transform unit g2 receives the torque pulsation component ΔT _m and extracts the sin component and the cos component (first-order component) by Fourier transformation.
The integral compensator g3 is an integrator that calculates a predetermined sin component and cos component for zeroing the frequency component of the torque pulsation component ΔT _m extracted by the Fourier transform unit g2.

The inverse Fourier transform unit g4 converts the calculation result (sin component and cos component) of the integral compensator g3 into a pulsation torque current command I _qsin ^* by inverse Fourier transform. The pulsating torque current command I _qsin ^* is used to calculate the torque current command I _q ^* (= I _q0 ^* + I _{q sin} ^* ) by the adder _{51 h} (see FIG. 9).

Even in the configuration including the speed fluctuation suppression control unit 51Ag and the torque pulsation estimation unit 51t (see FIG. 10), the suppression of the vibration of the compressor 11 and the reduction of the loss of the motor M and the like are the same as in the first embodiment. Can be compatible.

<Effect>
According to the second embodiment, although the configuration is complicated compared to the first embodiment, the loss of the motor M and the like can be minimized while maximizing the effectiveness of the torque control.

«Modification»
As mentioned above, although motor drive device 50 grade | etc., Which concerns on this invention was demonstrated by each embodiment, this invention is not limited to these description, A various change can be made.
For example, when the inductance of the motor M changes with the motor current, at least one of the q-axis inductance L _q and the d-axis inductance L _d may be variable based on the motor current. Such processing will be described using FIGS. 12A and 12B.

Figure 12A is an explanatory diagram showing a relationship between a torque current _{i q} and the q-axis inductance _{L q} of the motor M.
The horizontal axis in FIG. 12A is the torque current _iq , and the vertical axis is the q-axis inductance _Lq . In the example shown in FIG. 12A, as the torque current i _q of the motor M increases, q-axis inductance L _q is small. A calculation formula or data table indicating such a relationship may be stored in advance in the control unit 51, and the control unit 51 may calculate the q-axis inductance L _q based on the momentary torque current i _q . By this, it is possible to calculate the axis error Δθ of the above-mentioned equation (1) with higher accuracy.
In particular, the q-axis inductance L _q has a large fluctuation with the motor current as compared with the d-axis inductance L _d, and therefore the influence on the accuracy of the axis error Δθ is large. Thus the control unit 51 based on the torque current i _q, estimated every moment of the q-axis inductance L _q of the motor M, based on the q-axis inductance L _q, estimates the axis error [Delta] [theta].

12B is an explanatory diagram showing the relationship between the excitation current _{i d} and the d-axis inductance _{L d} of the motor M.
The horizontal axis in FIG. 12B is the excitation current _{i d} is a negative value is obtained by multiplying the (-1) (-i _d), the vertical axis represents the d-axis inductance _{L d.} In the example shown in FIG. 12B, as the absolute value of the excitation current i _d of the motor M approaches zero, d-axis inductance L _d is gradually increased. Such a formula or data table showing the relationship previously was stored in the control unit 51, based on the excitation current i _d momentary, the control unit 51 may calculate the d-axis inductance L _d. By this, it is possible to calculate the axis error Δθ of the above-mentioned equation (1) with higher accuracy.
The control unit 51 may estimate both the q-axis inductance L _q and the d-axis inductance L _d , or may estimate one of them.

The control unit 51, N order component included in the torque current i _q (i.e., N order components in the Fourier analysis) in response to changes in, may be to vary the excitation current i _d. Here, N is a natural number. Thereby, the vibration of the compressor 11 can be effectively suppressed.

Further, for example, on the basis of the third order component and the fifth-order component included in the torque current i _q, the control unit 51 may be changed excitation current i _d. That is, in the torque current i _q, orders based on the respective extraction result of different frequency components, the control unit 51 may be changed excitation current i _d. By this, the vibration of the compressor 11 can be suppressed more effectively.

Further, in the first embodiment, the configuration in which the control unit 51 (see FIG. 5) controls the motor M based on the rotational speed deviation Δω _r or the like has been described, but the present invention is not limited thereto. That is, the same control can be performed by using a value related to the vibration of the compressor 11 or the motor M (for example, the vibration acceleration of the motor M) or the fluctuation range of the axis error Δθ. The same applies to the second embodiment.

Moreover, although each embodiment demonstrated the process which calculates axial difference | error (DELTA) (theta) using Formula (1) mentioned above, it does not restrict to this. For example, equation (2) may be used instead of equation (1). Although this makes the calculation somewhat complicated, the axis error Δθ can be calculated with higher accuracy. Moreover, you may use Formula (3) instead of Formula (1). Using the equation (3), although a slight error (error of the axis error Δθ itself) occurs, the same effect as each embodiment can be obtained.

Moreover, although the structure of FIG. 5 was illustrated as the control part 51 with which the motor drive device 50 is equipped in 1st Embodiment, it does not restrict to this. That is, as the configuration of the control unit 51, another known configuration regarding position sensorless vector control may be used.

Moreover, although each embodiment demonstrated the structure which controls the motor M by a position sensor-less, it does not restrict to this. For example, each embodiment can be applied to a configuration in which the rotational position of the motor M is detected by a sensor (not shown). When such a sensor is provided, it is not necessary to calculate the axis error Δθ.

Moreover, although each embodiment demonstrated the structure which drives the compressor 11 of the air conditioner 100 by the motor M, it does not restrict to this. For example, each embodiment can be applied to a configuration in which a motor (M) drives a compressor (load) that may cause periodic torque fluctuations, such as a refrigeration cycle apparatus such as a refrigerator.

Further, in each embodiment, the compressor 11 in which one torque fluctuation occurs in one mechanical angle rotation of the motor M has been described, but the present invention is not limited thereto. For example, each embodiment can be applied to a reciprocating compressor widely used in a refrigeration cycle apparatus such as a refrigerator as well as a twin rotary compressor.

Each embodiment is described in detail in order to explain the present invention in an easy-to-understand manner, and is not necessarily limited to one having all the configurations described. Moreover, it is possible to add, delete, and replace other configurations for part of the configurations of the respective embodiments.
In addition, each configuration, function, processing unit, processing means, etc. described above may be realized by hardware, for example, by designing part or all of them with an integrated circuit. Further, the mechanisms and configurations indicate what is considered to be necessary for the description, and not all the mechanisms and configurations are necessarily shown on the product.

100 Air Conditioner (Refrigeration Cycle Equipment)
10 Refrigerant circuit 11 Compressor 12 Four-way valve 13 Outdoor heat exchanger (condenser, evaporator)
14 Indoor heat exchanger (evaporator, condenser)
DESCRIPTION OF SYMBOLS 15 Expansion valve 20 Converter 30 Inverter 40 Current detector 50 Motor drive device 51, 51A Control part 51a 2 axis conversion part 51b Axis | shaft error calculating part 51c PLL circuit 51d Integrator 51e Subtractor 51f Speed control part 51g, 51Ag Speed fluctuation suppression control Unit 51h Adder 51i Subtractor 51j Optimal phase control unit 51k Subtractor 51m Current control unit 51n Voltage command calculation unit 51r 3-phase converter 51s PWM signal generator 51t Torque ripple estimation unit

Claims (9)

1. What is claimed is: 1. A motor driving apparatus comprising: a control unit that changes a d-axis excitation current according to a change in q-axis torque current in a rotational coordinate system of the motor.
2. The control unit
   The torque current is periodically changed in the positive side region, and the excitation current is periodically changed in the negative side region,
   The motor drive device according to claim 1, wherein the absolute value of the excitation current is made smaller as the absolute value of the torque current is smaller.
3. The motor drive device according to claim 1, wherein the control unit sinusoidally changes the excitation current so as to be in reverse phase to the sinusoidal torque current.
4. The control unit
   An axis error between a real axis and a control axis regarding a magnet flux of the motor is estimated, and the motor is controlled without a position sensor based on the axis error,
   The motor drive device according to claim 1, wherein the axis error is momentarily estimated based on an instantaneous value of the torque current and an instantaneous value of the excitation current.
5. The control unit
   The motor drive device according to claim 4, wherein an instantaneous q-axis inductance of the motor is estimated based on the torque current, and the axial error is estimated based on the q-axis inductance.
6. The motor drive device according to claim 1, wherein the control unit changes the excitation current according to a change in an N-order component included in the torque current.
   Here, N is a natural number.
7. The motor drive device according to any one of claims 1 to 6, wherein the motor is a salient pole machine.
8. A refrigerant circuit in which a compressor, a condenser, an expansion valve, and an evaporator are sequentially connected in an annular shape, and refrigerant is circulated in the refrigeration cycle;
   A motor driving device for driving a motor connected to the compressor;
   The refrigeration cycle apparatus according to claim 1, wherein the motor drive device changes the excitation current of the d axis in accordance with a change of a torque current of the q axis in a rotational coordinate system of the motor.
9. What is claimed is: 1. A motor driving method comprising: changing a d-axis excitation current according to a change in q-axis torque current in a rotational coordinate system of the motor.

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