JP3680618B2 - Control device for permanent magnet type synchronous motor - Google Patents

Description

【００２７】
ＩＰＭモータでリラクタンストルクを用いるためには、ｄ軸電流ｉ_dを常に負に制御する必要があるので、Ｌ_d＜Ｌ_qを考慮すると、数式６の復号は＋をとるときｉ_dが負となるので、以下の数式７となる。
【００２８】
【数７】

【００２９】
ＩＰＭモータにおける無効電力の関係を表す数式３の右辺にｉ_q ²＝Ｉ²−ｉ_d ²を代入して整理すると、以下の式となる。
【００３０】
【数式８】
Ｖｉ_Q＝ω（Ｌ_d−Ｌ_q）ｉ_d ²＋ωΨ_mｉ_d＋ωＬ_qＩ²
【００３１】
数式８に数式７を代入すると、数式９となる。
【００３２】
【数９】

【００３３】
上記数式９が本発明の基本原理を表している。すなわち、数式９の左辺が無効電力の実際値、数式９の右辺が最大トルク制御の条件を加えた無効電力の目標値であり、両辺が等しくなるように制御することで、ＩＰＭモータを高効率制御することができる。
【００３４】
次に、数式５の最大トルク制御の式を近似的に表し、制御構成を簡略化することを考える。数式５をＩ²について解くと、以下の数式１０となる。
【００３５】
【数１０】
Ｉ²＝２ｉ_d ²＋｛Ψ_m／（Ｌ_d−Ｌ_q）｝ｉ_d
【００３６】
一般的には、Ψ_m／（Ｌ_d−Ｌ_q）はｄ軸電流に比べて十分に大きい。このような条件の場合、以下の関係式が成り立つ。
【００３７】
【数１１】
２ｉ_d ²≪｛Ψ_m／（Ｌ_d−Ｌ_q）｝ｉ_d
【００３８】
この数式１１の関係から、数式１０を以下のように近似する。
【００３９】
【数１２】
Ｉ²≒ｋｉ_d
【００４０】
数式１２における係数ｋを書き換えて、数式１３を得る。
【００４１】
【数１３】
ｉ_d＝ＫＩ² （１／ｋ＝Ｋとおく。）
【００４２】
数式１３を数式８に代入すると、以下の式となる。
【００４３】
【数１４】
Ｖｉ_Q＝ωＩ²｛（Ｌ_d＋Ｌ_q＋Ψ_m・Ｋ）／２｝＝ωＩ²・Ｋ_M
（Ｋ_M＝（Ｌ_d＋Ｌ_q＋Ψ_m・Ｋ）／２）
【００４４】
よって、前述の数式１２のように近似すると、数式１４により、簡単な回路構成で近似的な最大トルク制御を実現することができる。
数式１４の電圧ベクトルの大きさＶに代えてＶの指令値Ｖ^*を用いても、その作用は同じである。
【００４５】
【発明の実施の形態】
以下、本発明の実施形態について、図面を参照しながら説明する。
図１は請求項１に係る発明の実施形態であり、永久磁石形同期電動機の高効率運転を示す制御ブロック図である。破線で囲んだ部分がこの実施形態において追加した機能である。図８の従来技術に示した安定化制御手段は図示を省略する。破線外は従来技術であるため、以下では破線内についてのみ説明する。
【００４６】
請求項１は、本発明の根本的な概念を表すものである。図１において、ＩＰＭモータのごとく突極性を有する永久磁石形同期電動機６の電圧、電流は無効電力演算手段８に取り込まれ、この演算手段８により、電動機６に入力されている無効電力の実際値が前述の数式１により演算される。また、無効電力目標値演算手段９により、高効率運転の条件式（前述の数式１４または数式９の右辺）を用いて電動機６に入力されるべき無効電力の目標値を演算する。なお、１１は周波数設定値ｆ^*を無効電力目標値の演算用に角周波数ωに変換する変換手段である。そして、調節手段１０により、電動機に入力されている無効電力が目標値に等しくなるように作用する値がｆ／Ｖ変換手段３の出力である電圧指令ｖ^*に帰還される。
【００４７】
これにより、Ｖ／ｆ制御されるＩＰＭモータにおいて、最大トルク制御を可能にして高効率運転を行うことができる。
【００４８】
図２は、請求項２に係る発明の実施形態を示す制御ブロック図である。
図２において、検出した２相の電流ｉ_u，ｉ_wは座標変換手段１２により電圧指令ベクトルに直交する電流成分ｉ_Qと電圧指令ベクトルに平行な電流成分ｉ_Pとに変換される。無効電力演算手段８では、電圧指令ベクトルと電流成分ｉ_Qとから無効電力の実際値を演算する。また、電流成分ｉ_Q，ｉ_Pと電動機６の等価回路定数（Ｌ_d，Ｌ_q，Ψ_m等）１３と角周波数ωとを用いて、無効電力目標値演算手段９が無効電力の目標値を演算する。調節手段１０により、電動機６に入力されている無効電力の実際値が目標値に等しくなるように作用する値が電圧指令ｖ^*に帰還される。
【００４９】
図３は、請求項３に係る発明の実施形態を示す制御ブロック図である。
図３において、Ｉ²演算手段１５はＩ²＝ｉ_P ²＋ｉ_Q ²の演算式に従い電流の大きさの２乗を求め、前記数式１４のカッコ内に示した係数Ｋ_M＝（Ｌ_d＋Ｌ_q＋Ψ_mＫ）／２を乗じ、更に角周波数ωを乗算器１７により乗算してω｛（Ｌ_d＋Ｌ_q＋Ψ_mＫ）／２｝Ｉ²を求め、これを無効電力の目標値とする。
その一方で、無効電力演算手段８により電圧指令ｖ^*とｉ_Qとを乗算してｖ^*ｉ_Qを求める。
【００５０】
これらのω｛（Ｌ_d＋Ｌ_q＋Ψ_mＫ）／２｝Ｉ²とｖ^*ｉ_Qとの差をとってその値を△とする。△は調節器１６に入力され、電圧指令ｖ^*を補正する△ｖ^*を生成する。
△が＋（プラス）の場合には実際に入力されている無効電力が目標とする無効電力よりも大きく、印加した電圧が過大であることを表し、−（マイナス）の場合には実際に入力されている無効電力が目標とする無効電力よりも小さく、電圧が過少であることを表すため、△ｖ^*は電圧に対し負帰還される。
なお、図３において１４は積算手段、１８は３相／２相変換手段である。
【００５１】
図３における調節器１６の具体的な構成について、図７を参照しながら説明する。
図７において、例▲１▼は１次遅れ要素を持たせた場合、例▲２▼は比例要素と積分要素を組合せたＰＩ調節器を構成した場合、例▲３▼は比例要素を持たせた場合、例▲４▼は積分要素を持たせた場合である。いずれの場合も、無効電力目標値と実際値との偏差である入力の△をゼロにするように電圧を調整する機能を持つ点では、目的は共通している。
【００５２】
また、図４は請求項４に係る発明の実施形態を示す制御ブロック図であり、破線で囲んだ部分が図３の実施形態と異なっている。
この実施形態では、数式１４を変形して求まる次式に基づいて制御回路を構成している。
【００５３】
【数１５】
Ｖｉ_Q／ω＝｛（Ｌ_d＋Ｌ_q＋Ψ_mＫ）／２｝Ｉ²＝Ｋ_M・Ｉ²
【００５４】
図４において、Ｉ²演算手段１５によりＩ²＝ｉ_P ²＋ｉ_Q ²の演算式に従い電流の大きさの２乗を求め、図３と同様の係数Ｋ_M＝（Ｌ_d＋Ｌ_q＋Ψ_mＫ）／２を乗算して｛（Ｌ_d＋Ｌ_q＋Ψ_mＫ）／２｝Ｉ²を求める。
一方で、電圧ｖ^*とｉ_Qとの積を除算器１９により角周波数ωで除して（ｖ^*ｉ_Q）／ωを求める。（ｖ^*ｉ_Q）／ωと｛（Ｌ_d＋Ｌ_q＋Ψ_mＫ）／２｝Ｉ²との差をとってその値を△とする。△は調節器１６に入力され、電圧指令ｖ^*を補正する△ｖ^*を生成する。調節器１６の構成は、図７と同じである。
【００５５】
このように、電動機６に入力されている無効電力の実際値に比例した値と電動機６の無効電力の目標値に比例した値とを用いて図３の実施形態と同様な調節器１６により制御回路を構成すれば、高効率な運転が可能である。また、電動機６に入力されている無効電力の実際値と無効電力の目標値とから同一値を加減した値を用いても、同一の効果が得られることは明らかである。
【００５６】
【発明の効果】
以上のように本発明によれば、埋込磁石構造等の突極性を有する永久磁石形同期電動機の制御装置であって、巻線に印加する電圧とその周波数とをほぼ比例させて駆動するＶ／ｆ制御において、簡単な制御によって高効率運転を実現することができる。すなわち、従来、磁極位置検出器付きの駆動装置によって実施されてきた最大トルク制御を検出器なしで実現可能とし、一方的に電圧を供給するだけのＶ／ｆ制御に負荷変動に応じて電圧を調整する機能が加わり、省エネルギー効果を発揮する等の効果がある。
【図面の簡単な説明】
【図１】請求項１に記載した発明の実施形態を示す制御ブロック図である。
【図２】請求項２に記載した発明の実施形態を示す制御ブロック図である。
【図３】請求項３に記載した発明の実施形態を示す制御ブロック図である。
【図４】請求項４に記載した発明の実施形態を示す制御ブロック図である。
【図５】本発明の原理を説明する図である。
【図６】本発明の原理を説明する図である。
【図７】図３、図４における調節器の構成図である。
【図８】従来技術を示す制御ブロック図である。
【符号の説明】
１ 周波数設定手段
２ 加減速演算手段
３ ｆ／Ｖ変換手段
４ ＰＷＭ手段
５ インバータ
６ 永久磁石形同期電動機
８ 無効電力演算手段
９ 無効電力目標値演算手段
１０ 調節手段
１１ 変換手段
１２ 座標変換手段
１３ 電動機の等価回路定数
１４ 積算手段
１５ Ｉ²演算手段
１６ 調節器
１７ 乗算手段
１８ ３相／２相変換手段
１９ 除算手段[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a control device for operating a permanent magnet synchronous motor having a saliency like an embedded magnet structure with high efficiency by using a semiconductor power converter such as an inverter.
[0002]
[Prior art]
In the control of a synchronous motor such as a permanent magnet type synchronous motor or a reluctance motor, generally, a position detection value for detecting the position of the rotor is required, and the phase of the current is controlled in synchronization with the detected position of the rotor. As the detector, a Hall element, an encoder, a resolver, or the like is used. Alternatively, so-called position sensorless control is also known in which the position of the rotor is electrically calculated from information on the voltage and current of the motor.
[0003]
High-efficiency operation in the case of detecting the rotor position or obtaining it by calculation can be realized relatively easily. In a permanent magnet type synchronous motor (hereinafter referred to as an SPM motor) having a surface magnet structure in which a permanent magnet is attached to the rotor surface, I _d = 0 which makes the current in the magnetic flux direction generated by the permanent magnet, that is, the d-axis current zero. Control is generally employed. This is because, in the case of an SPM motor, the d-axis current does not contribute to torque, so that copper loss can be suppressed to a minimum by I _d = 0 control.
[0004]
In a permanent magnet type synchronous motor (hereinafter referred to as an IPM motor) having an embedded magnet structure in which a permanent magnet is embedded in a rotor, a magnetic circuit is asymmetric and a saliency is generated. Maximum torque control using torque is adopted. In this maximum torque control, torque / current is maximized. That is, since the current is always minimized with respect to the desired torque, the copper loss can be minimized.
[0005]
On the other hand, in addition to the method of detecting or calculating the position of the rotor as described above, V / f control is also known in which the voltage applied to the synchronous motor is controlled in proportion to the frequency.
FIG. 8 shows a control block diagram of V / f control. The desired motor frequency is set by the frequency setting means 1 and the frequency is changed in a ramp function by the acceleration / deceleration calculation means 2. In the f / V conversion means 3, a voltage substantially proportional to the frequency is obtained by storage or calculation, and a voltage command v ^* is output according to the frequency command f ^* .
[0006]
Integrating means 7, the voltage by integrating the frequency command f ₁ ^* is a sum of the frequency correction component Delta] f ^* from stabilization control means 30 described later with the frequency instruction f ^*, is applied to the windings of the motor 6 phase Calculate θ. The PWM means 4 inputs the magnitude of the voltage command v ^* and the phase θ, performs pulse width modulation, and controls the switching elements constituting the inverter 5. From the inverter 5, an AC voltage whose pulse width is controlled is output and applied to the winding of the permanent magnet type synchronous motor 6.
[0007]
The V / f control has a problem in stability, such as the torque constantly oscillating or the operation stepping out when the load suddenly changes and the operation becomes impossible. Therefore, the stabilization control means 30 shown in the broken line in FIG. 8 converts the current component i _Q orthogonal to the voltage vector v and the parallel current component i _P from the primary current of the motor 6 to the three-phase / two-phase conversion means 31, coordinates. The correction component Δf ^* is calculated through the conversion means 32 and added / subtracted through the high-pass filters 33 and 34 and the proportional amplifiers 35 and 36, and this is fed back to the voltage frequency command f ^* to improve the stability. Yes.
[0008]
[Problems to be solved by the invention]
A control device with a position detector can easily realize high-efficiency operation by performing I _d = 0 control for an SPM motor and maximum torque control for an IPM motor, but it is difficult to reduce the size of the device. In addition, since a plurality of wires and receiving circuits for transmitting detector signals are required, there are problems in terms of reliability, workability, and price. Furthermore, in the position sensorless control, a position sensor is not necessary, but in order to obtain the rotor position, a high-speed and high-function calculation is necessary, and an expensive control device is essential.
[0009]
On the other hand, the conventional V / f control does not require a position detector, and the control is simple because the control is simple. However, since the position of the rotor is unknown, I _d = 0 control, IPM in the SPM motor In motors, it has been difficult to operate with high efficiency by performing maximum torque control and the like.
For example, since the V / f control applies a predetermined voltage with respect to the output frequency to the electric motor regardless of the state of the load, if the predetermined voltage is a value that is appropriate in the rated load state, When the load becomes lighter, an excessive voltage is supplied. As a result, an unnecessary current flows and loss increases, which causes a problem from the viewpoint of energy saving.
[0010]
In this regard, the present applicant has proposed, as Japanese Patent Application No. 11-33025, a control device that equivalently realizes i _d = 0 control by V / f control for the SPM motor, and solves the above problem. .
SUMMARY OF THE INVENTION The present invention solves the above problems in a permanent magnet synchronous motor having saliency, such as an IPM motor, and intends to provide a control apparatus for a synchronous motor capable of high-efficiency operation with a simple and inexpensive configuration. It is.
[0011]
[Means for Solving the Problems]
In order to solve the above problems, the present invention basically includes a means for controlling a voltage applied to a winding of a permanent magnet type motor having saliency, such as an embedded magnet structure, in a substantially proportional manner, and a motor. It should be input to the motor from the means for calculating the actual value of the input reactive power, the conditional expression for maximum torque control of the motor, and the relationship of the reactive power in the permanent magnet synchronous motor having saliency. Means for calculating a target value of the reactive power, and adjusting means for adjusting a voltage applied to the winding so that the actual value and the target value of the reactive power are equal to each other.
[0012]
Hereinafter, the principle of the present invention will be described.
The reactive power input to the motor can be calculated by taking the voltage and current as vectors. Various coordinate axes can be used as the vector coordinate axes. Here, the method of claim 2 will be described with reference to FIGS.
[0013]
As shown in FIG. 5, a PQ orthogonal coordinate in which the direction of the voltage vector v applied to the winding of the motor is taken as the P axis is considered. It is assumed that the PQ coordinate axis is dq orthogonal to the dq orthogonal coordinate in which the direction of the magnetic flux generated by the permanent magnet is taken as the d axis. This intersection angle substantially corresponds to an angle called a load angle or internal phase difference angle of the synchronous motor.
When the current vector is i, the vector i is the P-axis current i _P and Q-axis current i _Q observed on the PQ axis, or the d-axis current i _d and q-axis current observed on the dq axis. i _q can be decomposed.
Here, since the voltage vector v and the Q-axis current i _Q are orthogonal to each other, the reactive power Q _i input to the electric motor can be obtained from the product of both, and the mathematical expression can be obtained using the magnitude V of the voltage vector. 1 can be represented.
[0014]
[Expression 1]
Q _i = Vi _Q
[0015]
On the other hand, reactive power can also be calculated using an equivalent circuit constant of the motor, and this will be described with reference to FIG. Induced voltage flux linkage [psi _m of the permanent magnet makes occurs by rotating at an angular frequency ω, the size is present represented on the q axis .omega..psi _m. 6 represents the magnitude and e _m. The product of the current i _d to be perpendicular to the induced voltage, i.e. .omega..psi _m i _d is the reactive power. In addition, the magnitude of the reactance drop voltage due to the current i _d becomes ωL _d i _d using the d-axis inductance L _d . In FIG. 6, the magnitude is represented as e _Ld .
[0016]
Since the reactance drop voltage e _Ld (= ωL _d i _d ) and the d-axis current i _d are orthogonal to each other, the product of both, ie, ωL _d i _d ² is reactive power. Furthermore, the magnitude of the reactance voltage drop due to the q-axis current i _q is a .omega.L _q i _q using the inductance L _q of the q-axis. In FIG. 6, the magnitude is represented as e _Lq . Since the reactance drop voltage e _Lq (= ωL _q i _q ) and the q-axis current i _q are orthogonal, the product of both, that is, ωL _q i _q ² is reactive power. Therefore, the reactive power Q _m obtained from the above consideration can be expressed by Equation 2.
[0017]
[Expression 2]
Q _m = ωΨ _m i _d + ωL _d i _d ² + ωL _q i _q ²
[0018]
Since the reactive powers obtained by Equation 1 and Equation 2 are different from each other only in view, the relationship shown in Equation 3 is established.
[0019]
[Equation 3]
Vi _Q = ωΨ _m i _d + ωL _d i _d ² + ωL _q i _q ²
[0020]
Formula 3 is a formula representing the relationship of reactive power in the IPM motor, and is not directly related to the high-efficiency operation aimed by the present invention. Therefore, a conditional expression for high-efficiency operation of the IPM motor is substituted into the right side of Equation 3, the value is set as a target value of reactive power to be input to the motor, and the reactive power of the motor represented by the left side of Equation 3 is obtained. By controlling so that the actual value of becomes equal to the target value, it becomes possible to adjust the reactive power input to the motor and operate the IPM motor with high efficiency.
[0021]
Here, a description will be given using a conditional expression for maximum torque control in order to operate the IPM motor with high efficiency. The conditional expression for maximum torque control is shown below (refer to EE-Study Group materials “Characteristics of Ring Magnet Embedded PM Motor” RM-95-15).
[0022]
[Expression 4]
{Ψ _m + (L _d −L _q ) i _d } i _d − (L _d −L _q ) i _q ² = 0
[0023]
If there is a position detector that detects the position of the rotor, i _d and i _q can be easily obtained by coordinate conversion, and the control system is configured so as to satisfy Equation 4, so that highly efficient operation by maximum torque control can be achieved. Is possible. However, it is difficult to obtain i _d and i _q when there is no position detector.
Therefore, it is considered that maximum torque control is realized with an IPM motor using i _P , i _Q or i _P ² + i _Q ² = I ² (= _id ² + i _q ² ) which is relatively easily obtained.
First, when organized by substituting _{^{i q 2 = I 2 -i d}} 2 which is obtained from the _{^{I 2 = i d 2 + i}} q 2 in Equation 4, the Equation 5.
[0024]
[Equation 5]
2 (L _d −L _q ) i _d ² + Ψ _m i _d − (L _d −L _q ) I ² = 0
[0025]
When Formula 5 is solved for i _d , Formula 6 is obtained.
[0026]
[Formula 6]

[0027]
In order to use the reluctance torque in the IPM motor, it is necessary to always control the d-axis current i _d to be negative. Therefore, when L _d <L _q is taken into consideration, the decoding of Equation 6 is negative when i _d is negative. Therefore, the following Expression 7 is obtained.
[0028]
[Expression 7]

[0029]
Rearranging the right-hand side of Equation 3 represents the relationship between the reactive power in the IPM motor by substituting the ^{_{i q 2 = I 2 -i d}} 2, the following equation.
[0030]
[Formula 8]
Vi _Q = ω (L _d −L _q ) i _d ² + ωΨ _m i _d + ωL _q I ²
[0031]
Substituting Equation 7 into Equation 8 yields Equation 9.
[0032]
[Equation 9]

[0033]
The above formula 9 represents the basic principle of the present invention. That is, the left side of Equation 9 is the actual value of reactive power, the right side of Equation 9 is the target value of reactive power to which the maximum torque control condition is added, and the both sides are controlled to be equal. Can be controlled.
[0034]
Next, it is assumed that the maximum torque control formula of Formula 5 is approximately expressed to simplify the control configuration. Solving Equation 5 for I ² gives Equation 10 below.
[0035]
[Expression 10]
I ² = ² i _d ² + {Ψ _m / (L _d −L _q )} i _d
[0036]
In general, Ψ _m / (L _d −L _q ) is sufficiently larger than the d-axis current. In such a condition, the following relational expression holds.
[0037]
[Expression 11]
² i _d ² << {Ψ _m / (L _d −L _q )} i _d
[0038]
From the relationship of Equation 11, Equation 10 is approximated as follows.
[0039]
[Expression 12]
I ² ≒ ki _d
[0040]
The coefficient k in Expression 12 is rewritten to obtain Expression 13.
[0041]
[Formula 13]
i _d = KI ² (assuming 1 / k = K)
[0042]
When Expression 13 is substituted into Expression 8, the following expression is obtained.
[0043]
[Expression 14]
Vi _Q = ωI ² {(L _d + L _q + Ψ _m · K) / 2} = ωI ² · K _M
(K _M = (L _d + L _q + Ψ _m · K) / 2)
[0044]
Therefore, when approximated as the above-described Expression 12, approximate maximum torque control can be realized with Expression 14 by a simple circuit configuration.
Even if the command value V ^* of V is used in place of the magnitude V of the voltage vector in Expression 14, the operation is the same.
[0045]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is an embodiment of the invention according to claim 1 and is a control block diagram showing high-efficiency operation of a permanent magnet synchronous motor. The part enclosed by the broken line is a function added in this embodiment. The stabilization control means shown in the prior art in FIG. 8 is not shown. Since the portion outside the broken line is the prior art, only the inside of the broken line will be described below.
[0046]
Claim 1 represents the basic concept of the present invention. In FIG. 1, the voltage and current of a permanent magnet type synchronous motor 6 having a saliency like an IPM motor are taken into reactive power calculation means 8, and the actual value of reactive power input to the motor 6 by this calculation means 8. Is calculated by Equation 1 described above. Further, the reactive power target value calculation means 9 calculates the target value of reactive power to be input to the motor 6 using a conditional expression for high-efficiency operation (the right side of Expression 14 or Expression 9 described above). Reference numeral 11 denotes conversion means for converting the frequency set value f ^* into the angular frequency ω for the calculation of the reactive power target value. Then, a value that acts so that the reactive power input to the motor becomes equal to the target value is fed back to the voltage command v ^* that is the output of the f / V conversion unit 3 by the adjusting unit 10.
[0047]
Thereby, in the IPM motor controlled by V / f, it is possible to perform maximum efficiency control and perform high efficiency operation.
[0048]
FIG. 2 is a control block diagram showing an embodiment of the invention according to claim 2.
In FIG. 2, the detected two-phase currents i _u and i _w are converted by the coordinate conversion means 12 into a current component i _Q orthogonal to the voltage command vector and a current component i _P parallel to the voltage command vector. In reactive power computing means 8 calculates the actual value of the reactive power from the voltage command vector and the current component i _Q. Further, the reactive power target value calculating means 9 uses the current components i _Q and i _P , the equivalent circuit constants (L _d , L _q , Ψ _m, etc.) 13 of the motor 6 and the angular frequency ω, so that the reactive power target value calculating means 9 Is calculated. A value that acts so that the actual value of the reactive power input to the motor 6 becomes equal to the target value is fed back to the voltage command v ^* by the adjusting means 10.
[0049]
FIG. 3 is a control block diagram showing an embodiment of the invention according to claim 3.
In FIG. 3, the I ² calculation means 15 calculates the square of the magnitude of the current according to the calculation formula of I ² = i _P ² + i _Q ² , and the coefficient K _M = (L _d + L) shown in parentheses in the formula 14 above. _q + Ψ _m K) / 2 and further multiplied by the angular frequency ω by the multiplier 17 to obtain ω {(L _d + L _q + Ψ _m K) / 2} I ² , which is set as the target value of reactive power. .
On the other hand, obtaining the v ^* i _Q by multiplication of the voltage control v ^* and i _Q by the reactive power calculation means 8.
[0050]
The difference between these ω {(L _d + L _q + Ψ _m K) / 2} I ² and v ^* i _Q is taken, and the value is taken as Δ. △ is input to the controller 16 corrects the voltage command v ^* △ v ^* to generate.
If △ is + (plus), the reactive power actually input is larger than the target reactive power, indicating that the applied voltage is excessive, and if it is-(minus), it is actually input. Since ΔV ^* is smaller than the target reactive power and the voltage is too low, Δv ^* is negatively fed back with respect to the voltage.
In FIG. 3, 14 is an integrating means, and 18 is a three-phase / two-phase converting means.
[0051]
A specific configuration of the regulator 16 in FIG. 3 will be described with reference to FIG.
In FIG. 7, Example (1) has a first-order lag element, Example (2) has a PI controller that combines a proportional element and an integral element, and Example (3) has a proportional element. In this case, Example (4) is a case where an integral element is provided. In any case, the purpose is common in that it has a function of adjusting the voltage so that the Δ of the input, which is the deviation between the reactive power target value and the actual value, becomes zero.
[0052]
FIG. 4 is a control block diagram showing an embodiment of the invention according to claim 4, and a portion surrounded by a broken line is different from the embodiment of FIG.
In this embodiment, the control circuit is configured based on the following equation obtained by modifying Equation 14.
[0053]
[Expression 15]
Vi _Q / ω = {(L _d + L _q + Ψ _m K) / 2} I ² = K _M · I ²
[0054]
In FIG. 4, the square of the magnitude of the current is obtained by the I ² calculation means 15 according to the calculation formula of I ² = i _P ² + i _Q ² , and the coefficient K _M = (L _d + L _q + Ψ _m K as in FIG. 3). ) / 2 to find {(L _d + L _q + Ψ _m K) / 2} I ² .
On the other hand, the product of the voltages v ^* and i _Q is divided by the angular frequency ω by the divider 19 to obtain (v ^* i _Q ) / ω. The difference between (v ^* i _Q ) / ω and {(L _d + L _q + Ψ _m K) / 2} I ² is taken and the value is taken as Δ. △ is input to the controller 16 corrects the voltage command v ^* △ v ^* to generate. The configuration of the adjuster 16 is the same as that in FIG.
[0055]
As described above, control is performed by the regulator 16 similar to the embodiment of FIG. 3 using a value proportional to the actual value of the reactive power input to the motor 6 and a value proportional to the target value of the reactive power of the motor 6. If the circuit is configured, high-efficiency operation is possible. In addition, it is obvious that the same effect can be obtained even if a value obtained by adding or subtracting the same value from the actual value of reactive power input to the motor 6 and the target value of reactive power is used.
[0056]
【The invention's effect】
As described above, according to the present invention, there is provided a control device for a permanent magnet type synchronous motor having saliency such as an embedded magnet structure, which is driven with a voltage applied to a winding being approximately proportional to its frequency. In the / f control, high efficiency operation can be realized by simple control. In other words, the maximum torque control conventionally performed by a drive device with a magnetic pole position detector can be realized without a detector, and the voltage according to load fluctuation is applied to V / f control that only supplies voltage unilaterally. A function to adjust is added, and there is an effect such as exerting an energy saving effect.
[Brief description of the drawings]
FIG. 1 is a control block diagram showing an embodiment of the invention described in claim 1;
FIG. 2 is a control block diagram showing an embodiment of the invention as set forth in claim 2;
FIG. 3 is a control block diagram showing an embodiment of the invention as set forth in claim 3;
FIG. 4 is a control block diagram showing an embodiment of the invention as set forth in claim 4;
FIG. 5 is a diagram illustrating the principle of the present invention.
FIG. 6 is a diagram illustrating the principle of the present invention.
7 is a configuration diagram of a regulator in FIGS. 3 and 4. FIG.
FIG. 8 is a control block diagram showing a conventional technique.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 Frequency setting means 2 Acceleration / deceleration calculation means 3 f / V conversion means 4 PWM means 5 Inverter 6 Permanent magnet type synchronous motor 8 Reactive power calculation means 9 Reactive power target value calculation means 10 Adjustment means 11 Conversion means 12 Coordinate conversion means 13 Electric motor Equivalent circuit constant 14 Accumulating means 15 I ² computing means 16 Controller 17 Multiplicating means 18 Three-phase / two-phase converting means 19 Dividing means

Claims (4)

Means for controlling the voltage applied to the winding of the permanent magnet type synchronous motor having saliency and the frequency thereof approximately in proportion;
Means for calculating an actual value of reactive power input to the motor;
Means for calculating a target value of reactive power to be input to the motor from a conditional expression for controlling the maximum torque of the motor and a relationship of reactive power in the permanent magnet synchronous motor having saliency ;
Adjusting means for adjusting a voltage applied to the winding so that an actual value and a target value of the reactive power are equal;
A control device for a permanent magnet type synchronous motor.   2. The permanent magnet synchronous motor control apparatus according to claim 1, wherein an actual value of reactive power input to the motor from a product of a magnitude of a voltage vector applied to the winding and a current component orthogonal to the voltage vector. Means for calculating the target value of reactive power to be input to the motor from the equivalent circuit constant of the motor, the magnitude of current and the angular frequency of the voltage, and the actual value and target value of the reactive power. And a means for amplifying the difference and feeding back to the command value of the magnitude of the voltage vector. 3. The control apparatus for a permanent magnet synchronous motor according to claim 2, wherein the proportional constant determined from the equivalent circuit constant of the motor is K _M , the magnitude of the current is I, and the angular frequency of the voltage is input to the motor. A control device for a permanent magnet synchronous motor, characterized in that a target value of reactive power to be set is K _M ωI ² . Means for controlling the voltage applied to the winding of the permanent magnet synchronous motor having saliency and the frequency thereof in proportion to each other, and the actual value of the reactive power input to the motor is divided by the angular frequency of the voltage. means and said electric motor proportionality constant K _M which is determined from the equivalent circuit constant of the magnitude of the current when the I, means for calculating the value of K _M I ^2, the corners of the voltage actual value of the reactive power A control device for a permanent magnet type synchronous motor, comprising: adjusting means for adjusting a voltage applied to the winding so that a value divided by frequency and K _M I ² are equal to each other.

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