JP3123235B2 - Induction motor vector control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a vector control apparatus for an induction motor in a slip frequency control system for compensating for fluctuations in excitation inductance.

[0002]

2. Description of the Related Art Vector control of an induction motor for controlling a secondary magnetic flux and a secondary current orthogonal to the secondary magnetic flux without interference has been widely applied.

In this vector control, in the case of a three-phase induction motor, a current or a magnetic flux is treated as a vector of a dq coordinate system of two orthogonal axes rotating at the same speed as a rotating magnetic field generated by a power source.
This is a method in which the operation result is converted into a current command value of each phase of the three-phase power supply and controlled.

[0004] The specific method is as follows.
The voltage equation in the coordinate system is expressed by the following equation (1).

[0005]

(Equation 1)

Where ω _s = ω−ω _r , Lσ = (L ₁ L ₂ −M
²⁾ / L _2.

Here, v _1d and v _1q are the primary voltages d and q, respectively.
Axis components, i _1d and i _1q are the d and q axis components of the primary current, λ
_2d and λ _2q are the d and q axis components of the secondary magnetic flux, R ₁ and R ₂ are the primary and secondary resistances respectively, L ₁ , L ₂ and M are the primary and secondary, the excitation inductance, ω, ω _r and ω _s represent the primary power supply angular frequency, the rotor angular frequency, and the slip angular frequency, respectively, and P represents d / dt.

If the d-axis is set on the secondary magnetic flux in the dq coordinate system, λ _2q = 0. At this time, λ _2d = Φ ₂ = constant, i
_2d = 0, i _2q = i quadrature control similar torque and flux and ₂ next to the DC motor becomes possible.

On the other hand, the secondary magnetic flux has the following relationship.

[0010]

(Equation 2)

According to the vector control condition, i _2d = 0,
From equation (2), λ _2d = Mi _1d .

From λ _2q = 0, i _1q = −L ₂ / M ·
i _2q , where i _1q is proportional to the torque current.

Next, equation (3) is obtained from the fourth row of equation (1), and when the condition of the slip angular frequency is obtained from equation (3), ω _s is expressed by equation (4).

[0014]

(Equation 3)

The above is the vector control condition when the control is performed so that the secondary magnetic flux coincides with the d-axis. Therefore, in order to perform vector control, it is necessary to set i _1d to λ _2d / M and control ω _s so that equation (4) is satisfied.

Here, the resistance of the secondary resistor R ₂ used for calculating the slip angular frequency ω _s changes with the ambient temperature and temperature changes such as self-heating of the rotor, so that the resistance of the secondary resistor R ₂ is determined based on the output voltage of the motor. It is necessary to estimate the change and correct the target value of the slip angular frequency ω _{s with} the change to compensate for the generated torque change due to the change in the secondary resistance. If the change in the secondary resistance is neglected, the torque control accuracy and the torque response deteriorate. If such a change in the secondary resistance is estimated by using the output voltage of the inverter as it is, for example,
Since a change in the secondary resistance is taken in, it is desirable that the signal used for estimation is a signal that is not affected by the primary resistance.

For this reason, a control circuit shown in FIG. 5 has already been proposed. In the figure, reference numeral 1 denotes an excitation current command unit.
Until the angular frequency ω _r exceeds a certain value, λ _2d * / M * is i _1d
And the target value i _1d * of, ω _r is the i _1d * is reduced when it exceeds a certain value. Hereinafter, when the target value or the ideal value is indicated with *, the deviation between the speed commands ω _r * and ω _r is set to i _1q * through the speed amplifier 2, and the dq axis is determined based on i _1d * and i _1q *. The ideal values v _1d *, v _1q * of the above primary voltage are obtained by calculation, and the correction of the voltage fluctuation due to the change in the primary resistance and the secondary resistance is i _1d *.
= I _1d , i _1q * = i _1q , i _1d *
Δv _1d is obtained for the PI amplifier 3 ₁ that controls = i _1d ,
i _1q * = i is the PI amplifier 3 ₂ for controlling _1q Delta] v _1q is obtained. Since Δv _1d and Δv _1q include both the voltage change due to the change in the primary resistance and the secondary resistance, if the component that does not include the voltage change due to the change in the primary resistance is obtained, the compensation for the change in the secondary resistance is performed. Compensation that is not affected by the primary resistance change is possible. So taking the rotating coordinate gamma-[delta] axes spaced reference axis gamma on the vector of the primary current I _1, are determined by the primary voltage change Delta] v ₁ [delta] a slip correction calculation unit 3 ₃ of the [delta] axis. This Δv ₁ δ is represented by an expression not including the primary resistance R ₁ , and thus is not affected by the primary resistance R ₁ .

FIG. 6 is a vector diagram showing the relationship between the dq axis and the γ-δ axis and the voltage and current, and FIG . 7 is a vector diagram showing the primary voltage fluctuation . In FIG. 6, V and E represent the primary voltage, respectively . In FIG. 7, Δv ₁ is the primary voltage variation, Δv ₁ γ, Δv ₁ δ are the γ-axis components of the variation, δ
Axis component, φ is phase between γ axis and d axis, I ₀ is excitation current,
I ₂ is the torque current. Δv ₁ δ is represented by the following equation (5).

Δv ₁ δ = −Δv _1d · sin φ + Δv _1q cos φ (5) where cos φ = I ₀ / I ₁ = i _1d / i ₁ γ and sin φ = I ₂ / I
₁ = i _1q / i _{1 γ} and obtains a correction amount [Delta] [omega _s of the slip angular frequency corresponding to the secondary resistance variation based on the slip correction calculation unit 3, ₃ Delta] v ₁ [delta] in operation, the slip angular frequency calculation unit 3 ₄ The added value of ω _s * and Δω _s obtained in the above is set as the target value of the slip angular frequency, and the rotor angular frequency ω _r is added _thereto, and the angular frequency of the primary voltage
= Dθ / dt. 5 3 ₅ polar conversion unit, the 3 ₆ coordinate conversion unit, the 4 ₁ PWM circuit, 4 ₂ inverters, IM induction motor, PP pulse pickup unit, 4 ₃ is a speed detecting unit.

[0020]

(A) Since the primary voltage fluctuations Δv _1d and Δv _1q include both the fluctuation of the primary resistance and the fluctuation of the secondary resistance, the circuit shown in FIG. Delta] v _1d in part 3 _3, further calculates a Delta] v ₁ [delta] that is not affected by the primary resistance change from Delta] v _1q, calculates the [Delta] [omega _r further from the Delta] v ₁ [delta].

(B) When performing field control, λ _2d and i _1d
Is in the relationship of the following equation (6) from the third line of the equation (1). Also, since λ _2q = 0, equation (7) holds.

[0022]

(Equation 4)

From equation (7), it can be seen that during field control, i _1d is controlled in a first-order manner with respect to a change in λ _2d . That is, when the field command λ _2d * is changing, λ _2d = Mi _1d does not hold.

However, in the conventional circuit, since the field control is not taken into consideration, the theoretical development is performed with the excitation current i _1d constant, that is, i _1d = λ _2d / M, and the slip correction calculation is executed. Was. For this reason, in the field control region, the set value of the slip angular frequency cannot be accurately calculated, and is not an effective method. The equivalent circuit of the induction motor used for the vector control is configured as shown in FIG. 8, and the exciting inductance M 'changes depending on the frequency and the exciting current, and has characteristics as shown in FIG. Therefore, if the ratio between M ′ and I ₀ is controlled to a constant value, accurate torque control becomes impossible. In particular, the variation of M 'in the constant output region is large, and the torque accuracy at the constant output may be reduced.

[0025] As described above, the excitation command λ _2d * / M * is controlled to be constant in the constant torque range shown in FIG. 9, is in the constant output range is inversely proportional to the rotor angular frequency omega _r as shown in FIG. 9 Control. Therefore, in a constant output range in which the field control is performed by the magnetic saturation characteristics of the motor core, the excitation inductance M 'fluctuates greatly, and the torque control accuracy is reduced.

An object of the present invention has been made in view of the above circumstances, and a vector control device for an induction motor in which the accuracy of torque control is improved by compensating for fluctuations in excitation inductance over the entire operation range. To provide.

[0027]

SUMMARY OF THE INVENTION In order to achieve the above object, the present invention is directed to a first invention, which is a rotating coordinate system which rotates in synchronization with a power supply angular frequency of an induction motor and uses a secondary magnetic flux as a reference. Assuming that the coordinates used as axes are dq coordinates, target values i _1d *, i of the d-axis component and the q-axis component of the primary current of the induction motor
_A first means for calculating _1q *, respectively, and a coordinate having a phase φ different from tan ^-1 (i _1q * / i _1d *) with respect to the dq axes and having the primary current I ₁ as a reference axis is a γ-δ coordinate. Then, i _1d *, i _1q
* Based on the target value i ₁ γ * (=
I ₁ ) and a first coordinate converter for calculating the phase φ;
λ _2d * and the target value M * ratio of the λ _2d of magnetizing inductance *
/ M *, calculation result of first coordinate transformation unit and power supply angular frequency
Based on ω ₀ , the target values v ₁ γ of the γ and δ axis components of the primary voltage
*, V ₁ δ *, respectively, and the primary current detection value of the induction motor is converted to each axis component i ₁ γ, i _{1 of the} γ-δ coordinate.
a vector control device for an induction motor, comprising: a second coordinate conversion unit for converting into δ; and a slip angular frequency calculation unit for calculating a slip angular frequency based on an arithmetic expression including a set value of a secondary time constant. _The means for calculating _1d * is the target value of the d-axis component of the secondary magnetic flux according to the rotor angular frequency of the induction motor.
and means for outputting a λ _2d * / M *, this λ _2d * / M * the excitation
Means for output by dividing by the magnetic inductance change AMn
And the value output from this means to the first-order element operation unit
Means for calculating i _1d * by inputting the λ _2d * /
Divide M * by exciting inductance change AMn and output
Output from the means for changing the excitation inductance
AMn and the power supply angular frequency ω ₀ are calculated by
The target value i _1q * of the q-axis component is added to the obtained value, and the torque
Means for calculating the command value i _1q ′ * is provided .

(Equation 15)

According to a second aspect of the present invention, there is provided the primary advance element
The calculation unit has a differential term, before Symbol λ _2d to this differential term * / M *
And i _1d * is calculated by adding an output obtained by dividing the value by the exciting inductance change AMn .

[0029]

According to a third aspect of the present invention, there is provided an exciting inductance.
The amount of change AMn represents the δ-axis error voltage value △ v ₁ δ, the initial setting value M ² * / L ₂ * of the exciting inductance, the power supply angular frequency ω ₀
And the excitation command λ _2d * / M *,
Versus the exciting inductance variation AMn to the power source angular frequency ω ₀
It is characterized in that it is made into a data table.

According to a fourth aspect of the present invention, the data table
The excitation inductance change AMn to the excitation command λ _2d
* / M *.

According to a fifth aspect of the present invention, the excitation command λ _2d * / M * is divided using the excitation inductance change AMn in the form of a data table, and the obtained value is input to the first-order advance element calculation unit. The output of the primary current of the induction motor
It is characterized in that a target value i _1d * of the axis component is calculated.

According to a sixth aspect of the present invention, the excitation command λ _2d * / M * is divided using the excitation inductance change AMn in the form of a data table, and the obtained value is divided by the differential term of the first-order advanced element calculation unit. And the d-axis of the primary current of the induction motor
It is characterized by calculating a target value i _1d * of the component .

According to a seventh aspect of the present invention, an excitation command λ _2d * / M
* And the change AMn in the data table are input to the iron loss compensation circuit to calculate the iron loss compensation current. The obtained compensation current output and the primary current of the induction motor are calculated .
The torque command value i _1q ′ * is calculated by adding the target value i _1q * of the q-axis component .

[0035]

FIG. 1 is a circuit diagram showing a first embodiment of the present invention, in which the same reference numerals as in FIG. 5 denote the same parts. 1 ₁
Depending on the angular frequency ω _r of than the speed detection unit 4 ₃ λ _2d * / M
* Is a secondary magnetic flux command amplifier that outputs *, and when ω _r exceeds a certain value and enters the field control region, λ _2d * / M according to ω _r
* Becomes smaller. 1 ₂ is the formula (7), that is, λ _2d * / M *
This is a first-order advanced element operation unit that executes the operation of (1 + L ₂ * / R ₂ * · S).

5 ₁ is a first coordinate converter, i
Γ based on primary current I ₁ based on _1d * and i _1q *
It has a function of calculating i ₁ γ * on the −δ coordinate and the phase difference φ between the d axis and the γ axis, and specifically executes the calculation of the following equation.

[0037]

(Equation 5)

[0038] 5 ₂ is an ideal voltage calculation unit for calculating a target value of the primary voltage, the first coordinate conversion unit 5 ₁ than output the sin [phi, I _1, from cosφ and the secondary magnetic flux instruction amplifier 1 ₁ The following equation (8) is executed using λ _2d * / M * and the power supply angular frequency ω ₀ to calculate v ₁ γ * and v ₁ δ *.

[0039]

(Equation 6)

Reference numeral 6 denotes a second coordinate conversion unit, which converts the primary current detection values i _u , i _w into respective axis components i ₁ γ, i ₁ δ of the γ-δ coordinate. These i ₁ γ and i ₁ δ are the target values i ₁ γ *,
is compared with i ₁ δ * (= 0), and the deviation is input to PI amplifiers 7 and 8 which are current control amplifiers, respectively. Δv ₁ γ and Δv ₁ δ are output from the PI amplifiers 7 and 8, respectively. As described above, Δv ₁ γ is v ₁ γ * and Δv ₁ δ is v ₁ δ
* Is added to each.

[0041] 3 ₄ is a slip angular frequency calculating unit, λ _2d
* / M * and i _1q * are taken in and ω _s is obtained. By the way, when the computer executes the operation of each part of the circuit of FIG. 1, ω _s is calculated as follows.
That sequence of operations including operations and the slip angular frequency calculation of K is performed instantaneously by the clock signal, the slip angular frequency calculation unit 3 ₄ (n-1) th secondary resistance value obtained by the calculation n th of This is the set value in the calculation. n th K and R ₂ respectively K _n obtained by the calculation expressed as R _2n, assigning a value R ₂ * is set in advance in the initial value R ₂₀ of R _2n, 1
The calculations from the first to the nth time are as follows.

First time R ₂₁ = (1 + K ₁ ) · R ₂₀ = (1 + K ₁ ) · R ₂ * Second time R ₂₂ = (1 + K ₂ ) · R ₂₁ = (1 + K ₂ ) · (1 + K ₁ ) · R ₂ * : N-th time R _2n = (1 + K _n ) · R _{2 (n-1)} = (1 + K _n ) (1 + K _n-1 ) ... (1 + K ₁ ) · R ₂ * Therefore, ω _s obtained in the n-th calculation is ω _sn Expressed as
ω _sn is given by the following equation (9). ω _sn = (1 + K _n ) · ω _{s (n-1)} (9) The ω _{s (n-1)} obtained in the (n−1) th operation is keep in stores, omega by using K _n obtained by (9)
_sn is required.

In this case, the initial value ω _s1 is ω _s1 = (1 + K ₁ )
· R ₂ * · 1 / L ₂ * · i _1q * / (λ _2d * / M *). The obtained ω _s and the detected rotor angular frequency ω _{r of the} electric motor IM are added, and the added value ω ₀ is set as a target value of the power supply angular frequency.

[0044] 10 is a secondary resistance variation amplifier, the output K of the amplifier 10 adds the output omega _s * and the slip angular frequency calculation unit 3 ₄ via the multiplier 11. The multiplier 11 has ω _s *
Is given. If the secondary resistance change K is obtained as the secondary resistance compensation amplifier output as described above, the amplifier output K may be a constant value even when ω _s * changes. Therefore, even when the torque component current command i _1q * and the excitation component current command λ _2d * / M * change, and ω _s * changes suddenly, the response of the secondary resistance compensation becomes good.

Next, a compensation circuit 1 for the excitation inductance M '
2 will be described. This circuit 12 sets the speed of the motor to some point in the entire operation range during no-load operation (for example, 1/2).
The operation is performed at 20 points from 0 · Nmax to Nmax.
Nmax is the maximum rotation speed. ), In each of the measuring points [Delta] V ₁ [delta] _n or ΔV ₁ δ _In (n measures the data number) corresponding to each measurement point, obtains an exciting inductance variation A _Mn from the following equation (10).

[0046]

(Equation 7)

[0047] In the above equation (10), Δv ₁ δ _n is [delta] axis error voltage at a measuring point n, omega _on is primary angular frequency at the measuring point n. With the exciting inductance variation A _Mn Request exciting inductance M _'n * from the following equation (11).

[0048]

(Equation 8)

In the above equation (11), M ² * / L ₂ * is an initial setting value of the exciting inductance. M ′ in the above equation
The change data of the excitation inductance as shown in FIG. 10 is created using _n *, and linear interpolation is performed between the data. Using this M '*, an ideal voltage calculation is performed by the following equation (12).

[0050]

(Equation 9)

Further, by calculating M * of the excitation command λ _2do * / M * instead of M ′ *, the excitation command is also changed corresponding to the change of the excitation inductance. As a result, the secondary magnetic flux λ _2d is accurately controlled, and the torque control accuracy in the entire operation range is improved.

As described above, when the exciting inductance is compensated using the exciting inductance change A _Mn data measured during the no-load operation, the exciting inductance set value M ′ * = M ² * / L ₂ * and the exciting command λ M * of _2d * / M *
(Equivalently as M '*). Here, when calculating v ₁ δ * and v ₁ δ * in the ideal voltage calculation equation (8), there is a calculation of M * × λ _2d * / M *. But,
Here, M ′ * / M *, and the effect of M ′ fluctuation is equivalently canceled.

The slip frequency calculation is obtained from the following equation (13).

[0054]

(Equation 10)

Where R ' ₂ = (M / L ₂ ) ² · R ₂ , M'
= A M _² / L ^2. M ′ * / M in the calculation of ω _s *
Since the term “*” appears, the effect of the M ′ fluctuation is offset equivalently. As described above, the M 'fluctuation compensation only needs to be executed for the portion for calculating the excitation current command i _1d *. Therefore,
Figure 1 As shown in M 'if the input to the exciting inductance variation A _Mn and λ _2d * / M * and the division unit 13 calculating unit 1 ₂ division value primary proceeds at from the compensation circuit 12 M' The effects of fluctuations can be offset.

[0056] Figure 2 shows a second embodiment, in FIG. 2, the primary lead element calculation unit 1 ₂ in Figure 1 is M '
When no compensation is performed, i _1d * is calculated by the following equation.

[0057]

[Equation 11]

In the above equation (14), λ _2d * / M * × M * appears in the differential term with S. Therefore, the term of the differential time constant of the first-order advance element cancels out the influence of the M ′ variation. This may be configured as shown in FIG. That is, the first-order advanced differential term does not require M 'compensation. The following equation shows i _1d * in FIG.

[0059]

(Equation 12)

The third embodiment is M 'variation compensation in the case of performing iron loss compensation shown in FIG. The iron loss compensation current I _RM is I
_RM = R _m * / M * × λ _2d * / M * × 1 / ω _o, which is affected by M ′ fluctuation. Therefore, the excitation inductor
The inductance change A _Mn is calculated as R _m * / M of the iron loss compensation circuit 14.
Compensation for the variation of M 'is performed by giving to the item of *. this
The compensation current T _RM (in FIG. 1 and FIG. 2, indicated by iR _m)
When showing the operation expression of that) (16) becomes as expression. Also,
R _m is an equivalent iron loss resistance.

[0061]

(Equation 13)

Reference numeral 15 denotes a PW with a dead time compensation circuit.
It is an M circuit. In the above-described first to third embodiments, the change in M 'with respect to the power supply angular frequency (output frequency) [omega] ₀ is measured and compensated for in consideration of the fact that M' slightly changes depending on the frequency even in the constant torque range. It was a means. However, there are applications where it is desired to perform control similar to that of a DC machine in order to perform constant tension control in a winding machine of a steel line.
That is, the target value (current for torque) i _1q * of the primary current is set as the tension command, and the compensation due to the winding thickening is performed as the excitation command λ _2d * / M *.
Performed by At this time, the excitation command changes in proportion to the wound coil diameter. In such a case, the excitation command is determined not by the frequency but by the coil diameter.
If the fluctuation data (or A _Mn change data ) is made into a data table as change data for the excitation command,
Processing becomes easier.

[0063] Therefore, the speed of the motor during no-load operation as the fourth embodiment is set to several points of a constant output range, obtained from the respective measuring points, △ v by measuring the _{1 δ n} (10) formula A change in excitation inductance A _Mn is determined. this,
A data table is prepared for the excitation command (λ _2d * / M * ∝1 / ω _{r in the} constant output range). This was a data table as shown in FIG. 4 with the fourth embodiment excitation inductance compensation circuit 12 shown in the block diagram of a 'shown in FIG. 4 A _Mn data determined obtains the A _Mn by excitation command value. In FIG. 4, A _Mn is calculated by linear interpolation between data (measurement points). In the fourth embodiment, the same parts as those in the first embodiment are denoted by the same reference numerals, and description thereof is omitted.

[0064]

As described above, according to the present invention, the torque control accuracy can be improved by compensating for the variation in the excitation inductance in the entire operation range. In addition, since the constant tension control is performed, the excitation command changes in proportion to the wound coil diameter, and the torque control accuracy is improved.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a first embodiment and a third embodiment of the present invention;

FIG. 2 is a block diagram showing a main part of a second embodiment of the present invention;

FIG. 3 is a block diagram showing a fourth embodiment of the present invention;

FIG. 4 is a characteristic diagram showing excitation inductance change data with respect to an excitation command;

FIG. 5 is a block diagram showing a conventional example;

FIG. 6 is a vector diagram for explaining a conventional example;

FIG. 7 is a vector diagram for explaining a conventional example;

FIG. 8 is an equivalent circuit diagram of the induction motor,

FIG. 9 is a characteristic diagram showing a relationship between a primary angular frequency ω ₀ , excitation inductances M ′ and ω _0, and an excitation command;

FIG. 10 is a characteristic diagram showing change data of excitation inductance.

[Explanation of Signs] 1 ₁ ... Secondary magnetic flux command amplifier, 1 ₂ ... Primary lead element calculation unit,
2 ... speed amplifier, 3 ₄ ... slip angular frequency calculation unit, 5 ₁ ... first coordinate conversion unit, 5 ₂ ... ideal voltage calculation unit, 6 ... second coordinate conversion unit, 10 ... secondary resistance change calculation section , 12, 12 '
... M 'compensation circuit, 13 ... divider, 14 ... iron loss compensation circuit.

Claims (7)

(57) [Claims] 1. A d-axis coordinate of a primary current of an induction motor, assuming that d-q coordinates are rotation coordinates rotating in synchronization with a power supply angular frequency of the induction motor and having a secondary magnetic flux as a reference axis. a first means for calculating target values i _1d * and i _1q * of the q-axis component, respectively, and a primary current I _{1 having a} phase φ different from tan ^-1 (i _1q * / i _1d *) with respect to the d-q axes. Is the γ-δ coordinate, and the target value i ₁ γ * (= I ₁ ) of the γ-axis component of the primary current and the phase φ are calculated based on i _1d * and i _1q *. and 1 of the coordinate conversion unit, the ratio between the target value M * of lambda _2d * and the exciting inductance lambda _2d * / M *, the primary voltage on the basis of the calculation results and the power supply angular frequency omega ₀ of the first coordinate conversion unit γ , Δ
Second means for calculating the target values v ₁ γ * and v ₁ δ * of the axis components, respectively, and converting the primary current detection value of the induction motor into each axis component i ₁ γ and i ₁ δ of the γ-δ coordinate. a second coordinate transformation unit which, in the vector control apparatus for an induction motor having a slip angular frequency calculator for calculating the slip angular frequency based on the arithmetic expression including the setting value of the secondary time constant, the i _1d * The calculating means includes a target value λ _2d * / M * of the d-axis component of the secondary magnetic flux according to the rotor angular frequency of the induction motor .
Λ _2d * / M * and the exciting inductor
Means for dividing and outputting the result by the change in
_Is input to the primary-advanced element operation unit, and i _1d
* Means for calculating *, and the λ _2d * / M * is calculated by the exciting inductance change AMn.
The value output from the means for dividing and outputting
The following formula is used to calculate the change AMn and the power supply angular frequency ω ₀
And add the q-axis component target value i _1q * to the obtained value.
Means for calculating a torque command value i _1q ′ * . [Equation 14] 2. A differential term is provided in said first-order lead element computing section.
Then, the above-mentioned λ _2d * / M * is _added to this derivative term as an exciting inductor.
2. The vector control apparatus for an induction motor according to claim 1, wherein i _1d * is calculated by adding outputs divided by the change amount AMn . 3. The exciting inductance change AMn is δ
The axis error voltage value △ v ₁ δ is converted to the initial set value M ² * / L ₂ * of the excitation inductance, the power supply angular frequency ω _0, and the excitation command λ
_2d * / M * by dividing the value obtained by multiplying the exciting inductance variation AMn claim 1, characterized in that as the data table of the relative power supply angular frequency omega _0, 2 induction motor according Vector control device. 4. The vector control device for an induction motor according to claim 3, wherein the data table is created by applying an exciting inductance change AMn to an excitation command λ _2d * / M *. 5. An excitation command λ _2d * / M * is divided by using a change in the excitation inductance AMn stored in a data table, and the obtained value is input to a primary-advance element calculation unit, and the output is derived. The target value i _1d of the d-axis component of the primary current of the motor
5. The vector control device for an induction motor according to claim 3 , wherein * is calculated. 6. An excitation command λ _2d * / M * is divided by using an exciting inductance change AMn in a data table, and the obtained value is added to a derivative term of a primary-advance element operation unit to derive a guidance. Target value i _1d * of the d-axis component of the primary current of the motor
5. The vector control device for an induction motor according to claim 3, wherein 7. An iron loss compensating current is calculated by inputting an exciting command λ _2d * / M * and an exciting inductance change AMn tabulated in a data table to the iron loss compensating circuit, and calculating the obtained compensating current output and induction. Target value i of q-axis component of primary current of motor
_5. The vector control device for an induction motor according to claim 3, wherein a torque command value i _1q ′ * is calculated by adding _1q *.