JP2940167B2 - Controlling device for vector of induction motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a vector control device for an induction motor.

[0002]

2. Description of the Related Art Vector control of an induction motor for controlling a secondary magnetic flux and a secondary current orthogonal to the secondary magnetic flux without interference has been widely applied.

In this vector control, in the case of a three-phase induction motor, a current or a magnetic flux is treated as a vector of a dq coordinate system of two orthogonal axes rotating at the same speed as a rotating magnetic field generated by a power source.
This is a method in which the operation result is converted into a current command value of each phase of the three-phase power supply and controlled.

[0004] The specific method is as follows.
The voltage equation in the coordinate system is expressed by the following equation (1).

[0005]

(Equation 1)

Where ω _s = ω−ω _r , Lσ = (L ₁ L ₂ −M
²⁾ / L _2.

Here, v _1d and v _1q are the primary voltages d and q, respectively.
Axis components, i _1d and i _1q are primary current d and q axis components, λ _2d and λ _2q are secondary magnetic flux d and q axis components, respectively, and R ₁ and R ₂ are primary and secondary resistances, respectively. , L ₁ , L ₂ , and M represent primary, secondary, and exciting inductances, respectively, ω, ω _r , and ω _s represent primary power supply angular frequency, rotor angular frequency, slip angular frequency, and P represents d / dt. Things.

If the d-axis is set on the secondary magnetic flux in the dq coordinate system, λ _2q = 0. At this time, λ _2d = Φ ₂ = constant, i
_2d = 0, i _2q = i quadrature control similar torque and flux and ₂ next to the DC motor becomes possible.

On the other hand, the secondary magnetic flux has the following relationship.

[0010]

(Equation 2)

According to the vector control condition, i _2d = 0,
From equation (2), λ _2d = Mi _1d .

From λ _2q = 0, i _1q = −L ₂ / M ·
i _2q , where i _1q is proportional to the torque current.

Next, equation (3) is obtained from the fourth row of equation (1), and when the condition of the slip angular frequency is obtained from equation (3), ω _s is expressed by equation (4).

[0014]

(Equation 3)

The above is the vector control condition when the control is performed so that the secondary magnetic flux coincides with the d-axis. Therefore, in order to perform vector control, it is necessary to set i _1d to λ _2d / M and control ω _s so that equation (4) is satisfied.

Here, the resistance of the secondary resistor R ₂ used for calculating the slip angular frequency ω _s changes with the ambient temperature and temperature changes such as self-heating of the rotor, so that the resistance of the secondary resistor R ₂ is determined based on the output voltage of the motor. It is necessary to estimate the change and correct the target value of the slip angular frequency ω _{s with} the change to compensate for the generated torque change due to the change in the secondary resistance. If the change in the secondary resistance is neglected, the torque control accuracy and the torque response deteriorate. If such a change in the secondary resistance is estimated by using the output voltage of the inverter as it is, for example,
Since a change in the secondary resistance is taken in, it is desirable that the signal used for estimation is a signal that is not affected by the primary resistance.

For this reason, a control circuit shown in FIG. 8 has already been proposed. In the figure, reference numeral 1 denotes an excitation current command unit.
Until the angular frequency ω _r exceeds a certain value, λ _2d * / M * is i _1d
And the target value i _1d * of, ω _r is the i _1d * is reduced when it exceeds a certain value. Hereinafter, when the target value or the ideal value is indicated by *, the deviation between the speed commands ω _r * and ω _r is set to i _1q * through the speed amplifier 2, and the dq axis is determined based on i _1d * and i _1q *. The ideal values v _1d *, v _1q * of the above primary voltage are obtained by calculation, and the correction of the voltage fluctuation due to the change in the primary resistance and the secondary resistance is i _1d *.
= I _1d , i _1q * = i _1q , i _1d *
Δv _1d is obtained for the PI amplifier 3 ₁ that controls = i _1d ,
i _1q * = i is the PI amplifier 3 ₂ for controlling _1q Delta] v _1q is obtained. Since Δv _1d and Δv _1q include both the voltage change due to the change in the primary resistance and the secondary resistance, if the component that does not include the voltage change due to the change in the primary resistance is obtained, the compensation for the change in the secondary resistance is performed. Compensation that is not affected by the primary resistance change is possible. So taking the rotating coordinate gamma-[delta] axes spaced reference axis gamma on the vector of the primary current I _1, are determined by the primary voltage change Delta] v ₁ [delta] a slip correction calculation unit 3 ₃ of the [delta] axis. This Δv ₁ δ is represented by an expression not including the primary resistance R ₁ , and thus is not affected by the primary resistance R ₁ . Since Δv ₁ δ is also used in the present invention, it will be described in detail in the description of the present invention.

[0018] Figure 6 is a vector diagram illustrating d-q axis and the gamma-[delta] axis and voltage, the relationship between the current, a vector diagram illustrating the FIG. 7 primary voltage change, drawing V _1, E are each primary voltage , The secondary voltage, Δv ₁ is the primary voltage variation, Δv ₁ γ, Δv ₁ δ are the γ-axis component and δ-axis component of the variation, ψ is the phase between the γ-axis and the d-axis, and I ₀ is the excitation component. The current, I _2, is the torque current.
Δv ₁ δ is represented by the following equation (5).

Δv ₁ δ = −Δv _1d · sinψ + Δv _1q cosψ (5) where cosψ = I ₀ / I ₁ = i _1d / i ₁ γ, sinψ = I ₂ / I
₁ = i _1q / i _{1 γ} and obtains a correction amount [Delta] [omega _s of the slip angular frequency corresponding to the secondary resistance variation based on the slip correction calculation unit 3, ₃ Delta] v ₁ [delta] in operation, the slip angular frequency calculation unit 3 ₄ The added value of ω _s * and Δω _s obtained in the above is set as the target value of the slip angular frequency, and the rotor angular frequency ω _r is added _thereto, and the angular frequency of the primary voltage ω
= Dθ / dt. 8 3 ₅ polar conversion unit figure, 3 ₆ coordinate conversion unit, the 4 ₁ PWM circuit, 4 ₂ inverters, IM induction motor, PP pulse pickup unit, 4 ₃ is a speed detecting unit.

[0020]

(A) Since the primary voltage fluctuations Δv _1d and Δv _1q include both the fluctuation of the primary resistance and the fluctuation of the secondary resistance, the circuit shown in FIG. delta] v _1d in part 3 _3, further calculates a received not delta ₁ [delta] influences the primary resistance change from delta] v _1q, and further calculating the delta] v ₁ [delta] or al [delta] [omega _r.

(B) When performing field control, λ _2d and i _1d
Is in the relationship of the following equation (6) from the third line of the equation (1). Also, since λ _2q = 0, equation (7) holds.

[0022]

(Equation 4)

From equation (7), it can be seen that during field control, i _1d is controlled in a first-order manner with respect to a change in λ _2d . That is, when the field command λ _2d * is changing, λ _2d = Mi _1d does not hold.

However, in the conventional circuit, since the field control is not taken into consideration, the theoretical development is performed with the excitation current i _1d constant, that is, i _1d = λ _2d / M, and the slip correction calculation is executed. Was. For this reason, in the field control region, the set value of the slip angular frequency cannot be accurately calculated, and is not an effective method.

A first object of the present invention is to provide an ideal compensation which is not affected by a change in the primary resistance when compensating for a change in the secondary resistance in the equation for calculating the slip angular frequency. The purpose of the present invention is to propose a vector control device which is simpler in the calculation of the target value of the above than the method already proposed, and is effective even when performing the upper field control.

A second object of the present invention is to provide a vector control device capable of compensating for changes in primary resistance and exciting inductance.

A third object of the present invention is to provide a vector control device capable of improving the response of the secondary resistance fluctuation compensation and stabilizing the compensation.

A fourth object of the present invention is to provide a vector control device which improves the accuracy of secondary resistance fluctuation compensation and shortens the identification time.

[0029]

As described above, not only the secondary resistance but also the primary resistance changes with temperature, and therefore it is ideal to perform the secondary resistance compensation without being affected by the primary resistance change. is there. Here, in the present invention, the variation amount Δv ₁ δ of the δ-axis component of the primary voltage is used as in the circuit of FIG.
A further advanced control method was adopted.

That is, in the vector control shown in FIG. 8, the d-axis of the rotation coordinate dq-axis is made the same axis as the secondary magnetic flux, so that
The excitation current i _1d and the torque current i _1q are controlled so as to maintain orthogonality.

This time, a method of controlling by setting the γ-axis on the primary current I ₁ using the rotational coordinates as the γ-δ axis was studied.
However, since vector control requires control on the dq axis, it rotates at the same speed as the power supply angular frequency ω _0, and uses the dq axis and the γ-δ axis having different phases together. New control method.

When the primary voltage variation due to the secondary resistance change is detected as the variation Δv ₁ δ on the δ axis when considering on the γ-δ axis based on the primary current I ₁ , the voltage variation due to the primary resistance is included. Since there is no voltage component, robust secondary resistance compensation is possible.

Therefore, the ideal primary voltage v on the γ-δ axis
₁ γ * and v ₁ δ * are obtained by calculation, and the correction of the voltage fluctuation due to the change in the primary resistance and the secondary resistance is executed by controlling so that i ₁ γ = I ₁ and i ₁ δ = 0. By performing such control, Δv ₁ γ is obtained for the PI amplifier output controlling i ₁ γ = I _1, and Δv ₁ δ is obtained for the PI amplifier output controlling i ₁ δ = 0. Since Δv ₁ δ does not include a voltage component due to a primary resistance change, it can be used for compensating for a secondary resistance change. That is, Δv ₁ δ
, The ideal compensation independent of the primary resistance change can be performed.

As described above, using the γ-δ axis using the primary current I ₁ as a reference value has an advantage that primary voltage fluctuation data used for compensating for a change in the secondary resistance can be directly obtained on the δ axis.

In the present invention, when the field command λ _2d * is changed from the above equation (7), λ _2d = Mi _1d does not hold, so that λ _2d / M and i _1d are used separately. ing.

Hereinafter, the present invention will be described in detail.

(A) Vector control conditions when using γ-δ axes FIG. 3 is an asymmetric TI type equivalent circuit of an induction motor, and FIG. 4 is a vector diagram corresponding to this equivalent circuit.

Now, if the γ axis is set on the primary current I ₁ , i ₁ γ = I
₁ , i ₁ δ = 0. "Conventional technology" for γ-δ axis
Equation (1) holds true for the equation (1)
When d and q in the equation are changed to γ and δ, respectively,
Equations (8) and (9) hold from the fourth row.

[0039]

(Equation 5)

If the term including P is removed, ω always holds.
The condition of _s is required. The following equation is obtained from the equation (9).

R ₂ / L ₂ + P = −λ ₂ γ · ω _s / λ ₂ δ (10) By substituting equation (10) into equation (8), the following equation is obtained. Holds.

[0042]

(Equation 6)

Here, the relationship between λ _2d and λ ₂ γ, λ ₂ δ is as follows.

Λ ₂ γ = λ _2d cosψ, λ ₂ δ = −λ _2d sinψ (12) λ ₂ γ ² + λ ₂ δ ² = λ _2d2 (13) Equations (12) and (13) are _replaced by ( Substituting into equation (11) gives the following equation.

[0045]

(Equation 7)

As described above, the γ-axis is controlled on the primary current I ₁ so that i ₁ γ = I ₁ and i ₁ δ = 0, and d
If the vector control condition on the −q axis is satisfied, ω _s is expressed by the same formula as the formula (4) in the “Prior Art” section, and it has been found that the same condition can be obtained. However, if λ _2d ≠ Mi _1d is considered in consideration of the field control region,
Ω _s is expressed as in equation (15) from the first stage of equation (14).

[0047]

(Equation 8)

(B) Ideal voltage on the γ-δ axis On the γ-δ axis, i ₁ γ * = I ₁ and i ₁ δ * = 0 are controlled. The next (1
6) is obtained. If the term with P is omitted in equation (16), equation (17) is obtained.

[0049]

(Equation 9)

Here, when the vector control condition is satisfied, (18)
The equation holds. By satisfying the expression (18), (1
9) is obtained.

[0051]

(Equation 10)

When the torque current command i _1q * changes suddenly or when the exciting current command i _1d * changes due to the field control, the term of Pi ₁ γ applied to Lσ in equation (16) cannot be ignored. . The ideal voltage when this P term is considered is as follows.

[0053]

[Equation 11]

(C) Fluctuation of the secondary magnetic flux when the secondary resistance changes The fluctuation of the secondary magnetic flux when the secondary resistance changes will be examined. The following equation is obtained from the third and fourth lines of the equation (1).

[0055]

(Equation 12)

By multiplying the equations (21) and (22) by L ₂ / R ₂ , the following is obtained.

[0057]

(Equation 13)

When λ ₂ γ is obtained from equations (23) and (24), (23) × (1 + L ₂ P / R ₂ ) becomes equation (25), and (24) × L ₂ ω _s / R ₂ becomes Equation (26) is obtained. Also, from (25) + (26), λ ₂ γ is as shown in equation (27).

Next, when λ ₂ δ is obtained from equations (23) and (24), (23) × L ₂ ω _s / R ₂ becomes equation (28), and (24) × (1 + L ₂ P / R) ₂ ) becomes the equation (29). Further, when λ ₂ δ is obtained from (29)-(28), it is as shown in equation (30).

[0060]

[Equation 14]

Here, the following assumption is made.

(A) Assuming that the current is controlled to flow according to the command value, i ₁ γ * = i ₁ γ, i ₁ δ * = i ₁ δ =
Set to 0. The current on the dq axes is i _1d * = i _1d , i
_{Let 1q} * = i _1q .

(B) When the secondary resistance change is K, R ₂ =
Since (1 + K) R ₂ *, L ₂ ω _s / R ₂ in the equations (27) and (30) can be expressed as follows.

[0064]

(Equation 15)

(C) It is assumed that the exciting current is controlled as shown by the equation (7), so that the equation (32) is established.

[0066]

(Equation 16)

(D) Assuming that the secondary resistance compensation is performed, 1
+ L ₂ P / R ₂ transient term time constant L ₂ / R ₂ = L ₂ * / R ₂ *
Assume that (Short time and R ₂ does not change.) Therefore, the following equation is established.

[0068]

[Equation 17]

By substituting the above relational expressions into the expressions (27) and (30) and transforming them, the following expressions (34) and (35) are obtained.
Here, the ideal value of the secondary magnetic flux on the γ-δ axis is (36) to (3).
8) It is expressed by the equation.

[0070]

(Equation 18)

By multiplying the denominator and the numerator of the equation (34) by i ₁ γ * and using the equation (36), the secondary magnetic flux variation Δλ
₂ gamma is expressed as equation (39).

[0072]

[Equation 19]

By multiplying the denominator and the numerator of the equation (35) by i ₁ γ * and using the equation (37), the secondary magnetic flux variation Δ
λ ₂ δ is represented by equation (40).

[0074]

(Equation 20)

(D) Temporary Voltage Fluctuation When Secondary Magnetic Flux Fluctuates The primary voltage when the secondary magnetic flux fluctuates can be expressed as follows from equation (16).

[0076]

(Equation 21)

Since the ideal value of the primary voltage is expressed by the equation (19), the voltage fluctuations Δv ₁ γ and Δv ₁ δ are calculated from the equations (19) and (41) in consideration of the equation (18). become that way.
However, the expansion of Δλ ₂ δ and Δλ ₂ γ are (40) and (3
9) Equation is used.

[0078]

(Equation 22)

Here, since v ₁ γ includes the component of R ₁ i ₁ γ *, the voltage fluctuation due to the change in the primary resistance R ₁ is also v ₁ γ.
Will be included. Therefore, considering the change in the primary resistance R1, the equation (42) is as follows. However, K ₁ is the primary resistance variation.

[0080]

(Equation 23)

As described above, since Δv ₁ γ includes the variation of the primary resistance R ₁ , it is not suitable for use in compensating for the change in the secondary resistance R ₂ . On the other hand, since Δv ₁ δ does not include the component of R ₁ , it is considered to be a voltage fluctuation component due to a secondary resistance change. Therefore, by performing the detecting and secondary resistance compensation a variation Delta] v ₁ [delta] of the primary voltage v ₁ [delta] of [delta] axes, has the following advantages because it does not contain the influence of the primary resistance R _1.

[0082] (it) can be carried out secondary resistance compensation without being affected by the temperature variation of the primary resistance R _1.

(B) Although the influence of the voltage drop of R ₁ becomes large in the low-speed range, the primary voltage v ₁ δ on the δ axis does not include the voltage drop of R ₁ , so the secondary resistance is low even in the low-speed range. Compensation can be performed accurately.

[0084] (c) be performed Delta] v ₁ [delta] than the secondary resistance compensation, the Delta] v ₁ gamma only the voltage component due to R ₁ variation occurs. This allows estimation of R _1. R ₁ is considered to containing such V _CE partial voltage drop main circuit elements of the resistance component dead time of the primary resistance cable.

(E) Calculation of Secondary Resistance Change K By modifying equation (43), the following equation (45) is obtained.

[0086]

(Equation 24)

Accordingly, if the primary voltage variation Δv ₁ δ of δ can be detected, the secondary resistance variation K can be obtained from equation (45).

(F) Method of Identifying Primary Resistance and Exciting Inductance During No-Load Operation In the present invention, primary resistance and exciting inductance can be identified as described below by adding a change in secondary resistance. When the exciting inductance changes, the shunt ratio between the exciting current and the torque current changes, and the primary voltage also changes. Primary voltage in order to vary similarly be secondary resistance changes, it is impossible to distinguish changes in the excitation inductance M and the secondary resistance R _2. However, during the no-load operation, the torque current i _1q = 0, so that the primary voltage fluctuation does not have the effect of the secondary resistance change. Therefore, the excitation inductance can be compensated using the primary voltage fluctuation during the no-load operation.

A vector diagram at the time of no-load operation can be represented as shown in FIG. 5 by a TI type equivalent circuit. At the time of no-load operation, since the torque current i _1q = 0, the dq axis and the γ-δ axis match. Therefore, consider the dq axis. The primary voltage at the time of no-load operation can be expressed as follows from Expression (16). However, i _1q = 0, and the P term is ignored.

[0090]

(Equation 25)

Here, the following assumption is made.

(A) It is assumed that the current is controlled to flow as the command value, and i _1d * = i _1d .

(B) The change in the excitation inductance is expressed as A _M
far.

[0094] (c) put the A ₁ the amount of change in the primary resistance.

(D) Add * to the set value of the motor constant.

(E) Assuming that the leakage inductance Lσ is small, the change is ignored.

Now, the ideal voltage at the time of no-load operation can be expressed as follows from the equation (46).

[0098]

(Equation 26)

The primary voltage is expressed by using the primary resistance change A ₁ and the exciting inductance change A _M as follows.

[0100]

[Equation 27]

From equations (47) and (48), the primary voltage fluctuations Δv _1d and Δv _1q during no-load operation are as shown in equation (49). The (49) exciting inductance variation A _M and the primary resistance change in A ₁ from equation is as equation (50).

[0102]

[Equation 28]

In a steady state where the excitation command does not change, λ _2d *
= M * i _1d *, and A _M is as follows.

[0104]

(Equation 29)

As described above, it was found that the primary resistance and the exciting inductance can be identified by detecting the primary voltage fluctuation during the no-load operation. The summary is as follows.

(A) The primary resistance change A ₁ can be found from the primary voltage fluctuation Δv _1d on the d-axis.

[0107] (b) primary voltage variation Δv _1q than exciting inductance variation A _M of the q-axis is seen.

(G) Means of the Invention If the target value R ₂ * of the secondary resistance matches the actual secondary resistance, ω _s is obtained based on the equation (15), and this is calculated as ω _s * The secondary resistance changes with temperature. Therefore, in the present invention, K is calculated using Δv ₁ δ, and R ₂ * is corrected using this K to obtain ω _s *. To obtain ω _s *, the following equation (52) obtained from equation (15) is used.

[0109]

[Equation 30]

On the other hand, the primary resistance also changes depending on the temperature.
As can be seen from equation (43), v ₁ δ does not include the value of the primary resistance, and therefore does not depend on the change in the primary resistance when compensating for the secondary resistance. Although this point is common to the circuit shown in FIG. 8, the circuit shown in FIG. 8 performs current control in the dq coordinate system, whereas the present invention uses γ-current control.
The primary voltage is controlled on the basis of current control in the δ coordinate system, whereby the output of the current control amplifier is Δv ₁ γ, Δv ₁ δ
And the secondary resistance is compensated using Δv ₁ δ.

More specifically, a first coordinate converter for calculating a target value i ₁ γ * (= I ₁ ) of the γ-axis component of the primary current and the phase ψ based on i _1d * and i _1q * the ratio of the lambda _2d * and the excitation inductance M λ _2d / M, γ of the primary voltage based on the command value omega ₀ of the result and power supply angular frequency of the first coordinate conversion unit, a target value of δ-axis component v ₁ means for calculating γ * and v ₁ δ *, respectively, a second coordinate converter for converting the primary current detection value of the induction motor into each axis component i ₁ γ, i ₁ δ of γ-δ coordinates, Based on i ₁ γ *, the target value i ₁ δ * of the δ-axis component of the primary current, and i ₁ γ, i ₁ δ from the second coordinate conversion unit, v ₁ and variation Delta] v ₁ gamma from gamma *, means for calculating a variation Delta] v ₁ [delta] from v ₁ [delta] * in the [delta] -axis component of the current of the primary voltage, i _1d
*, I _1q *, i ₁ γ *, λ _2d *, primary power supply angular frequency ω ₀ ,
A secondary resistance change calculating unit for calculating a change with respect to the set value of the secondary resistance based on the set value M * of the exciting inductance and Δv ₁ δ, and calculating an added value of v ₁ γ * and Δv ₁ γ The target value v ₁ γ of the γ-axis component of the primary voltage is set as the target value v ₁ δ of the δ-axis component of the primary voltage, and the sum of v ₁ δ * and Δv ₁ δ is calculated.
The power supply voltage is controlled based on the target values v ₁ γ and v ₁ δ, and the set value of the secondary time constant by the slip angular frequency calculation unit and the calculation obtained by the secondary resistance change calculation unit. The secondary time constant at that time is obtained based on the result and the calculation is performed using the secondary time constant, i _1q * and λ _2d * / M *.

In the present invention, instead of using the secondary resistance change calculator, v ₁ in the δ-axis component of the current primary voltage is used.
A slip angular frequency control amplifier which inputs a deviation Δv ₁ δ from δ * and a deviation of the target value of Δ ₁ δ from zero and outputs a fluctuation Δω _s of the slip angular frequency from the target value ω _s *. The same operation and effect can be obtained even if the added value of Δω _s from the slip angular frequency control amplifier and ω _s * obtained by the slip angular frequency calculator is used as the target value of the slip angular frequency.

Further, in the present invention, Δv
_{Based on 1} γ, the set value of the primary resistance R ₁ * and i _1d *, a change in the set value of the primary resistance is calculated, and Δv
₁ δ, M *, secondary self inductance L ₂ *, ω ₀ and λ
An identification circuit unit for calculating a change in the excitation inductance with respect to the set value based on _2d * may be provided.

In addition, in the present invention, a secondary resistance change amplifier is provided in the slip angular frequency control amplifier, the output of this amplifier is multiplied by a slip frequency set value, and the obtained value is obtained as a change. ω _s * may be multiplied to obtain the slip angular frequency, or a limit may be applied to the slip angular frequency control amplifier and the secondary resistance change amplifier.

[0115]

FIG. 1 is a circuit diagram showing a first embodiment of the present invention, in which the same reference numerals as in FIG. 8 denote the same parts. 1 ₁
Depending on the angular frequency ω _r of than the speed detection unit 4 ₃ λ _2d * / M
* A secondary magnetic flux instruction amplifier for outputting, until beyond a certain value ω _r λ _2do * / M * outputs, omega _r depending on entering the omega _r a field control region beyond a certain value lambda _2d * / M *
Becomes smaller. 1 ₂ is the equation (7), that is, λ _2d * / M * (1
+ L ₂ * / R ₂ * · S).

5 ₁ is a first coordinate converter, i
Γ based on primary current I ₁ based on _1d * and i _1q *
It has a function of calculating i ₁ γ * on the −δ coordinate and the phase difference と between the d axis and the γ axis, and specifically executes the calculation of the following equation.

[0117]

(Equation 31)

[0118] 5 ₂ is an ideal voltage calculation unit for calculating a target value of the primary voltage, the first coordinate conversion unit 5 ₁ than output the Sinpusai, I _1, from cosψ and the secondary magnetic flux instruction amplifier 1 ₁ Λ _2d * / M * and the power supply angular frequency ω ₀ (1
9) Execute the calculation of the expression to calculate v ₁ γ * and v ₁ δ *.

Reference numeral 6 denotes a second coordinate conversion unit, which converts the primary current detection values i _u , i _w into respective axis components i ₁ γ, i ₁ δ of the γ-δ coordinate. These i ₁ γ and i ₁ δ are the target values i ₁ γ *,
is compared with i ₁ δ * (= 0), and the deviation is input to PI amplifiers 7 and 8 which are current control amplifiers, respectively. Δv ₁ γ and Δv ₁ δ are output from the PI amplifiers 7 and 8, respectively. As described above, Δv ₁ γ is v ₁ γ * and Δv ₁ δ is v ₁ δ
* Is added to each. Numeral 9 denotes a polar coordinate converter, which outputs the magnitude | V ₁ | of the primary voltage vector V ₁ and the phase angle φ with the γ-axis (see FIG. 3). This phase angle φ is added to ψ and θ (= ω ₀ t) described later, and the added value and | V ₁ |
: It is converted is inputted to the PWM circuit 4 ₁ U, V, the primary voltage command value corresponding to the W phase, thereby the inverter 4
The voltage of ₂ is controlled.

Numeral 10 denotes a secondary resistance change calculator, which is λ _2d
* / M *, i _1d *, i _1q *, ω ₀ , i ₁ γ * and Δv ₁ δ
And the calculation of the equation (45) is performed to obtain the secondary resistance change K. Numeral 11 denotes a slip angular frequency calculation unit which takes in K, λ _2d * / M * and i _1q (5
2) It has a function of obtaining ω _s by executing the equation. By the way, when the computer executes the operation of each part of the circuit of FIG. 1, ω _s is calculated as follows. That is, a series of calculations including the calculation of K and the slip angle frequency calculation are instantaneously performed by the clock signal, and the secondary resistance value obtained in the (n-1) th calculation in the slip angle frequency calculation unit 11 is calculated by the nth calculation. The set value in. K and R ₂ obtained in the n-th calculation are represented as K _n and R _2n , _respectively , and a predetermined value R ₂ * is assigned to an initial value R ₂₀ of R _2n. become that way.

First time R ₂₁ = (1 + K ₁ ) · R ₂₀ = (1 + K
₁ ) · R ₂ * 2nd time R ₂₂ = (1 + K ₂ ) · R ₂₁ = (1 + K ₂ ) · (1+
K ₁ ) · R ₂ *: n-th R _2n = (1 + K _n ) · R _{2 (n−1)} = (1 + K _n ) (1+
K _n−1 )... (1 + K ₁ ) · R ₂ * Therefore, when ω _s determined in the n-th operation is represented as ω _sn ,
ω _sn is given by the following equation (53). ω _sn = (1 + K _n ) · ω _{s (n-1)} (53) The ω _{s (n−1)} obtained in the (n−1) th calculation is keep in stores, omega _sn is obtained by using the K _n obtained by equation (53).

In this case, the initial value ω _s1 is ω _s1 = (1 + K ₁ )
· R ₂ * · 1 / L ₂ * · i _1q * / (λ _2d * / M *). The obtained ω _s and the detected rotor angular frequency ω _{r of the} electric motor IM are added, and the added value ω ₀ is set as a target value of the power supply angular frequency.

Numeral 12 denotes an identification circuit which takes in Δv ₁ γ and i _1d * during no-load operation and executes the upper equation of equation (50) to calculate the change A ₁ of the primary resistance. , The primary resistance is identified by Δv ₁ δ, ω ₀ and λ _2d *
/ M * is taken in, and the lower part of the equation (50) is executed to calculate the change A _M of the excitation inductance, thereby identifying the excitation inductance M ² / L ₂ . Here, the comparator 13 determines whether or not the operation is the no-load operation and the drive timing of the identification circuit unit 12. The comparator 13 sets the rated torque current to 100
%, For example, the value of 5% is set as a set value, and i _1q *
If i _1q * is lower than the set value, it is determined that no-load operation is to be performed, and the identification circuit 12 is driven. In this case, the influence of the change in the secondary resistance does not appear. The secondary resistance change calculator 10 is stopped.

[0124] In the above, in order to perform the calculation when calculating the v ₁ gamma * in the calculating portion 5 ₂ considering P term (20), a term relating to i ₁ gamma from R ₁ * R ₁ * ( 1 + Lσ
/ R ₁ * P), a more accurate ideal voltage can be given and the current response can be improved.

Next, a description will be given of a second embodiment in which the embodiment of FIG. 1 is improved. If the change in the secondary magnetic flux is ignored from the equation (16), the following equation (16a) is obtained. Where λ ₂ γ, λ ₂ δ
_Represents λ _2d using equation (18).

[0126]

(Equation 32)

[0127] by a value corresponding to the temporal change rate when the primary current as can be seen from this equation is suddenly changed v ₁ γ, v ₁ δ
Changes. That is, the change in v ₁ δ includes the time change rate of the primary current in addition to the change in the secondary resistance, and the change in v ₁ γ includes the change in the primary resistance and the exciting inductance. In addition to the minute, the temporal change rate of the primary current is also included. Therefore, in the first embodiment of FIG. 1, when the primary current suddenly changes, the change is regarded as a change in the secondary resistance, and also as a change in the primary resistance and a change in the excitation inductance. Become.

Therefore, in the second embodiment shown in FIG.
₁ γ, LσPi ₁ δ, the primary voltage fluctuations (which are termed Δv ₁ γ, Δv ₁ δ), and the primary voltage fluctuations that do not contain them (which are Δv ₁ γ _I , Δv ₁ δ _I ), The primary voltage is controlled using the former values Δv ₁ γ, Δv ₁ δ, and the secondary resistance change is compensated using the latter values Δv ₁ γ _I , Δv ₁ δ _I and The primary resistance and the like are to be identified.

More specifically, as shown in the second embodiment of FIG. 16, for the PI amplifier 7, the proportionality for calculating (i ₁ γ * -i ₁ γ) × Lσ / T _s corresponding to LσPi ₁ γ is calculated. Element 7 ₁
And and a integral element 7 ₂ integrating the _{(i 1 γ * -i 1 γ} ), outputs the sum of the proportional term output from the proportional element ₇₁ and the integral term output from the integral element 7 ₂ as Delta] v ₁ gamma while, and configured to output the integral term output as Δv ₁ γ _I. With respect to the PI amplifier 8, integrating corresponding to _{LσPi 1 δ (i 1 δ *} -i 1 δ) × Lσ / T s the calculating proportional element ₈₁ and the _{(i 1 γ * -i 1 γ} ) integral and a component _82, Delta] v and outputs the sum of the integral term output from the proportional term output and the integral element ₈₂ than the proportional element 8 ₁ as Delta] v ₁ [delta], the integral term output from the integral element ₈₂ It is configured to output a ₁ [delta] _I. Here, T _s indicates the operation cycle, and (i ₁ γ * −i ₁ γ) / T _s and (i ₁ δ * −i ₁ δ) /
T _s is calculated using a differential element.

According to such a configuration, even when the primary current changes abruptly, there is no effect on Δv ₁ γ _I and Δv ₁ δ _I , so that accurate secondary resistance compensation and identification of the primary voltage are performed. be able to.

FIG. 2 is a circuit diagram showing a third embodiment of the present invention. Instead of using the secondary resistance change calculator 10, a PI amplifier 14 which is a voltage fluctuation control amplifier is used.
Enter the deviation between the target value zero to the amplifier 14 Delta] v ₁ [delta] and Delta] v ₁ [delta] to obtain a variation [Delta] [omega _s from the target value omega _s * in the current slip angular frequency as the output signal. The slip angular frequency calculating unit 15 assumes that R ₂ does not fluctuate from an ideal value.

[0132]

[Equation 33]

Ω _s * is calculated based on the ω _s * and Δ
The value added to ω _s is set as the target value of the slip angular frequency.
According to such an embodiment, the target value of the slip angular frequency is automatically corrected according to the secondary resistance change. Reference numeral 14 denotes a switch unit for invalidating the output of the PI amplifier 14 by the output signal of the comparator 16.

[0134] Also in the embodiment shown in FIG. 2, using the PI amplifier 7 and 8 shown in each of FIG. 16 as the PI amplifier 7, and inputs a Delta] v ₁ gamma _I in the identification circuit 12, Delta] v
By entering the deviation between ₁ [delta] * and Delta] v ₁ [delta] _I to the PI amplifier 14, described above was as the secondary resistance change amount of the compensation or the like can be performed accurately.

Next, a fourth embodiment and a fifth embodiment are shown in FIGS. In the fourth and fifth embodiments, the slip frequency is shown in equation (15). Now, when the torque component current command i _1q changes suddenly or when the exciting current command λ _2d / M changes within the constant output range, the slip angular frequency ω
_s will change. Therefore, [Delta] [omega _s must also be changed when i _1q and lambda _2d / M changes the use of voltage change control amplifier for outputting a [Delta] [omega _s as a secondary resistance variation compensation amplifier. Therefore, the response of the secondary resistance fluctuation compensation when the torque current command i _1q or the excitation current command λ _2d / M changes is deteriorated. That is, if the secondary resistance change K is directly output as the output of the secondary resistance compensation amplifier, the above-described disadvantage can be solved.

Therefore, when the slip frequency is expressed by using the secondary resistance change K, the equation (53) is obtained.

[0137]

(Equation 34)

From equation (53), assuming that the change in the secondary resistance is a certain value K, the change in i _1q * and λ _2d * / M causes Δ Δ
It can be seen that ω _s changes. That is, it can be seen that the transient response deteriorates in the secondary resistance compensating amplifier (method for directly obtaining Δω _s ). (Δω due to the change of i _1q *, λ _2d * / M
_s amplifier output must also change)
And integral term output (error voltage) Delta] v ₁ [delta] _I in the present embodiment, provided with a compensation amplifier for outputting the Delta] v ₁ [delta] deviations from the secondary resistance variation K between the target value zero _I directly, the secondary resistance change From the equation (53), Δω _s = K × ω _s * using the minute K, and ω _s
* More to be added to obtain the ω _s.

FIGS. 9 and 10 show a fourth embodiment and a fifth embodiment. In FIG. 9, reference numeral 70 denotes a secondary resistance change amplifier. The output of the amplifier 70 is slid via a multiplier 71. It is added to the output ω _s * of the angular frequency calculation unit 15. The multiplier 71 is given ω _s *. FIG. 10 shows Δv ₁ δ _I *
= 0 and Delta] v ₁ [delta] Enter the deviation between _I to the secondary resistance variation amplifier 70, the output ω embodiments slip in the same manner as angular frequency arithmetic unit 11 of FIG. 9 through the multiplier 71 and the amplifier output K _s *
Is added.

If the secondary resistance change K is obtained as the secondary resistance compensation amplifier output as described above, the amplifier output K may be a constant value even when ω _s * changes. Therefore, even if the torque component current command i _1q * or the excitation component current command λ _2d * / M changes, and ω _s * changes suddenly, the response of the secondary resistance compensation is good.

Next, a sixth embodiment and a seventh embodiment will be described. Δv ₁ δ on the δ axis generated by the secondary resistance fluctuation can be expressed by equation (43). From equation (43), Δ
It can be seen that v ₁ δ changes in proportion to the primary angular frequency ω ₀ . For this reason, ω ₀ becomes very small in the low frequency region or when the motor is locked. As a result, Δv ₁ δ also becomes a very small value. If you are use a secondary resistance compensation amplifier ([Delta] [omega _s amplifier and the secondary resistance variation K amp), compensation response in the low frequency region becomes slow. (Input Δv ₁ δ or Δv ₁
δ _I will take the time to amp swings for a minute. )
In this embodiment, the response is improved by changing the gain of the PI amplifier according to the frequency. The gain of the PI amplifier is varied by the following equation. Kp = Kp * × ω _OTRQ
/ Ω ₀ , where Kp is the PI amplifier gain and Kp * is P
The gain setting value at the time of ω _{OTRQ of the} I amplifier, ω _OTRQ is the base angular frequency, and ω ₀ is the primary angular frequency. Also, considering the stability of the PI amplifier, the gain Kp ≦ limiter ≦ Kp _LIM (K
p _LIM is variable).
When ω ₀ is in the constant output range, ω ₀ = ω _OTRQ, and Kp =
Kp *. Here, the configuration of the PI amplifier gain is shown.

FIG. 11 shows a sixth embodiment, and FIG. 12 shows a seventh embodiment. In both figures, 72 is the PI amplifier gain, 7
3 is a limiter as far as it goes up and down. With the configuration shown in FIGS. 11 and 12, by changing the PI amplifier gain Kp in inverse proportion to the frequency, the compensation response in the low-speed range can be made faster. In addition, upper and lower limiters are provided, and the upper limiter Kp _LIM is determined in consideration of stability.
By setting the lower limiter to the set value Kp * at the time of the base angular frequency _ωOTRQ , stable compensation can be performed in the entire operation range.

Next, an eighth embodiment will be described. The Δv ₁ δ of the δ axis generated by the secondary resistance fluctuation includes a 1f ripple particularly in the signal at a low frequency. Therefore, it is necessary to insert a filter to perform stable secondary resistance fluctuation compensation. In this embodiment, a first-order lag filter is inserted into Δv ₁ δ, and the filter time constant is changed in inverse proportion to the first-order frequency. The following equation is the transfer function of the filter. G (S) = 1/1 + ST ₁ , T ₁ = 1 / f ₀ = 2π /
ω ₀ , where T ₁ is the time constant of the first-order lag filter, f ₀ is the output frequency of the inverter, and ω ₀ is the first-order angular frequency.
Also, upper and lower limiters for the filter time constant are provided to stabilize the secondary resistance compensation.

FIG. 13 shows an eighth embodiment having a first-order lag filter. In the figure, 74 is a first-order lag filter, 75 is an upper / lower limiter, and 76 is a time constant of the first-order lag filter. By inserting a first-order lag filter 74 as shown in FIG. 13 and making the filter time constant inversely proportional to the frequency, the compensation can be stabilized. The upper and lower limiters 7
If the filter effect 5 is provided so that the filter effect can be varied in the high-speed range and the extremely low-speed range, the compensation can be stabilized over a wide range.

Next, a ninth embodiment and a tenth embodiment will be described. As described in the sixth and seventh embodiments,
From equation (43), it can be seen that Δv ₁ δ changes in proportion to the primary angular frequency ω ₀ . For this reason, ω ₀ has a very small value in the low frequency region or when the motor is locked, so that Δv ₁ δ
Is also very small. When the secondary resistance compensation is performed in such a low frequency region, the calculation accuracy of the secondary resistance change calculation (45) becomes poor, or the secondary resistance change K amplifier or other PI amplifier is used. It takes time to identify the resistance compensation (it takes time for the amplifier to swing to the secondary resistance change K because the input Δv ₁ δ and Δv ₁ δ _{I of the} PI amplifier are very small).

Therefore, in this embodiment, the accuracy of the secondary resistance fluctuation compensation in the low frequency region is improved and the identification time is shortened. As shown in equation (43), Δv ₁ δ is proportional to ω ₀ . Since ω ₀ = ω _s when the motor is locked (ω _r = 0), Δv ₁ δ is proportional to the slip angular frequency ω _s . ω _s * is expressed by the following equation.

[0147]

(Equation 35)

The torque current command i _1q * is small (when lightly negative)
At times, ω _s * also decreases, and Δv ₁ δ also decreases. If Δv ₁ δ is small, the above-described problem occurs. Therefore, in applications such as a winder and an elevator of a steel line, a mechanical brake is attached to a motor, so that operation in a motor locked state is possible. In such a case, if the motor is braked to lock the motor and the torque current command i _1q * is made to flow at a large value (for example, a torque current command of about 50 to 100%), ω _s * also becomes large. Thus, a large value of Δv ₁ δ is obtained. Accordingly, by operating the secondary resistance change calculation (45) and the secondary resistance change K amplifier, it is possible to execute the secondary resistance change compensation with high accuracy and high identification speed. Performing this initial identification, holding the secondary resistance compensation data, and then entering normal operation with the hold data as the initial value eliminates the need for secondary resistance compensation identification time, and achieves highly accurate torque control. It becomes possible.

FIGS. 14 and 15 are flowcharts showing the ninth and tenth embodiments. FIG. 14 is an initial identification flowchart for calculating the secondary resistance change, and FIG. 15 is an initial identification flowchart for the secondary resistance change K amplifier. is there.

[0150]

According to the present invention, the δ-axis component v ₁ δ of the primary voltage on the rotating coordinate γ-δ axis using the primary current I ₁ as a reference axis.
Does not include the voltage drop of the primary resistance R ₁ , and as for the primary voltage fluctuation due to the secondary resistance change, the fluctuation component Δv ₁ δ does not show the influence of the primary resistance,
For example, Δv ₁ δ is obtained by a current control amplifier, and this is used to compensate for the secondary resistance change when obtaining the target value of the slip angular frequency, so that ideal compensation not affected by the primary resistance change can be performed. it can. Moreover, since the primary voltage ideal values v ₁ γ * and v ₁ δ * are calculated separately without _treating the excitation command λ _2d * / M * and the excitation current i _1d * as equal, the field voltage control is used. This is also effective vector control. Further, since voltage control is performed by obtaining Δv ₁ γ and Δv ₁ δ, voltage correction can be performed for changes in primary resistance and secondary resistance, and high torque control accuracy can be obtained and torque response is good. become.

In comparison with the circuit of FIG. 8, the voltage control is performed only on the dq coordinates in the circuit of FIG.
Since v _1d and Δv _1q include a change in both the primary resistance and the secondary resistance, the data affected by only the secondary resistance change from Δv _1d and Δv _1q and the influence of both changes are considered. Although it is necessary to separate the received data from the received data, in the present invention, it is possible to directly control by Δv ₁ γ and Δv ₁ δ without performing such separation.

If the secondary resistance is compensated by Δv ₁ δ, only the effect of the primary resistance change remains in Δv ₁ γ.
The primary resistance R ₁ can be estimated based on Δv ₁ γ.

Further, in the present invention, the primary voltage during no-load operation is analyzed, and attention is paid to the analysis result to obtain Δv ₁ γ (= Δ
v _1d ), a change in the primary resistance with respect to the set value is calculated, and the excitation inductance M ² / L ₂ is calculated based on Δv ₁ δ.
, The primary resistance and the excitation inductance can be compensated.

Also, the configuration of the current control amplifier is LσPi
_If the voltage deviation corresponding to 1γ, LσPi ₁ δ is obtained as a proportional term output, and the voltage variation generated by the secondary resistance change is obtained as an integral term output, LσPi ₁ γ, The term LσPi ₁ δ no longer appears in the integral term output. This improves the current response and
Using the integral term outputs Δv ₁ γ _I and Δv ₁ δ _I , the primary resistance, the secondary resistance, and the excitation inductance can be accurately compensated.

Further, in the present invention, there is an advantage that the secondary resistance compensation can be satisfactorily performed, the accuracy can be improved and the identification time can be shortened, and the compensation response in a low speed region can be shortened.

[Brief description of the drawings]

FIG. 1 is a block circuit diagram showing a first embodiment of the present invention.

FIG. 2 is a block circuit diagram showing a third embodiment of the present invention.

FIG. 3 is an equivalent circuit diagram of the induction motor.

FIG. 4 is a vector diagram.

FIG. 5 is a vector diagram.

FIG. 6 is a vector diagram.

FIG. 7 is a vector diagram.

FIG. 8 is a block circuit diagram showing a comparative example of the vector control device.

FIG. 9 is a main part block circuit diagram of a fourth embodiment.

FIG. 10 is a main part block circuit diagram of a fifth embodiment.

FIG. 11 is a main part block circuit diagram of a sixth embodiment.

FIG. 12 is a main part block circuit diagram of a seventh embodiment.

FIG. 13 is a main part block circuit diagram of an eighth embodiment.

FIG. 14 is a flowchart of the ninth embodiment.

FIG. 15 is a flowchart of the tenth embodiment.

FIG. 16 is a block circuit diagram of a second embodiment.

[Explanation of symbols]

1 ₁ ... Secondary magnetic flux command amplifier 1 ₂ ... Operation unit 2 ... Speed amplifier 5 ₁ ... First coordinate conversion unit 5 ₂ ... Ideal voltage calculation unit 6 ... Second coordinate conversion unit 7, 8 ... Current control amplifier PI amplifier 10 ... Secondary resistance change calculation unit 11, 15 ... Slip angular frequency calculation unit 12 ... Identification circuit unit 14 ... Voltage fluctuation control amplifier

Claims (13)

(57) [Claims] 1. A d-axis coordinate of a primary current of an induction motor, assuming that d-q coordinates are rotation coordinates rotating in synchronization with a power supply angular frequency of the induction motor and having a secondary magnetic flux as a reference axis. Induction motor having means for calculating target values i _1d * and i _1q * of the q-axis component, respectively, and a slip angular frequency calculator for calculating a slip angular frequency based on an arithmetic expression including a set value of a secondary time constant in the vector control device, means for calculating an i _1d * includes means for outputting a target value lambda _2d of d-axis component of the secondary flux * according to the rotor angular frequency of the induction motor, and the lambda _2d * derivative Means for calculating i _1d * based on the term, the phase ψ is different from tan ⁻¹ (i _1q * / i _1d *) with respect to the dq axis, and the primary current I ₁ is used as a reference axis. Assuming that the coordinates are γ-δ coordinates, the target value of the γ-axis component of the primary current based on i _1d *, i _1q * a first coordinate conversion unit for calculating i ₁ γ * (= I ₁ ) and the phase 、, λ _2d * and an exciting inductance M
Λ _2d * / M, the target value v ₁ γ *, v ₁ δ * of the γ- and δ-axis components of the primary current based on the calculation result of the first coordinate conversion unit and the command value ω ₀ of the power supply angular frequency. And a second coordinate conversion unit that converts the primary current detection value of the induction motor into each axis component i ₁ γ, i ₁ δ of the γ-δ coordinate. Γ-axis component of the current primary voltage based on i ₁ γ * and the target value i ₁ δ * of the δ-axis component of the primary current and i ₁ γ, i ₁ δ from the second coordinate conversion unit. , The variation Δv ₁ γ from v ₁ γ *, and v ₁ δ * in the δ-axis component of the current primary voltage.
Means for calculating a variation Delta] v ₁ [delta] _{from, i 1d *, i 1q *} , i 1 γ *, λ 2d *, the primary power source angular frequency omega _0, the set value of the excitation inductance M * and Delta] v ₁ [delta] based provided a secondary resistance variation calculator for calculating a change amount for the set value of the rotor resistance, v ₁ γ * and Delta] v ₁ target value of gamma-axis component of the primary voltage sum of the gamma v ₁ gamma Further, an addition value with v ₁ δ * Δ ₁ δ is set as a target value v ₁ δ of the δ-axis component of the primary voltage, and these target values v ₁ γ,
The power supply voltage is controlled based on v ₁ δ, and the slip angular frequency calculation unit calculates the secondary time constant based on the set value of the secondary time constant and the calculation result obtained by the secondary resistance change calculation unit. The next time constant is calculated, and this second time constant, i _1q *
And a calculation using λ _2d * / M *. (Equation 35) 2. The vector control device for an induction motor according to claim 1, wherein a variation Δv ₁ δ from v ₁ δ * in the δ-axis component of the current primary voltage is used instead of using the secondary resistance variation calculating section. And a deviation of the Δv ₁ δ from the target value of zero, and a voltage fluctuation control amplifier for outputting the fluctuation Δω _s of the slip angular frequency from the target value ω _s *. A vector control apparatus for an induction motor, characterized in that an addition value of Δω _{s of the above} and ω _s * obtained by the slip angular frequency calculation unit is set as a target value of the slip angular frequency. 3. The method of calculating a change in the primary resistance set value based on Δv ₁ γ, the primary resistance set value R ₁ * and i _1d * during the no-load operation, and calculating Δv ₁ δ, M *, secondary An identification circuit is provided for calculating a change in the set value of the excitation inductance based on the self-inductances L ₂ *, ω ₀ and λ _2d *, and the value calculated by the identification circuit is used to calculate the R of the ideal voltage calculation unit. ^{_{1 *, M * 2 / L}} 2 * vector control apparatus for an induction motor according to claim 1 or claim 2, wherein the identified and. 4. A d-axis component of a primary current of an induction motor, assuming that dq coordinates are rotation coordinates that rotate in synchronization with a power supply angular frequency of the induction motor and that use secondary magnetic flux as a reference axis. Induction motor having means for calculating target values i _1d * and i _1q * of the q-axis component, respectively, and a slip angular frequency calculator for calculating a slip angular frequency based on an arithmetic expression including a set value of a secondary time constant in the vector control device, means for calculating an i _1d * includes means for outputting a target value lambda _2d of d-axis component of the secondary flux * according to the rotor angular frequency of the induction motor, and the lambda _2d * derivative Means for calculating i _1d * based on the term, the phase ψ is different from tan ⁻¹ (i _1q * / i _1d *) with respect to the dq axis, and the primary current I ₁ is used as a reference axis. Assuming that the coordinates are γ-δ coordinates, the target value of the γ-axis component of the primary current based on i _1d *, i _1q * a first coordinate conversion unit for calculating i ₁ γ * (= I ₁ ) and the phase 、, λ _2d * and an exciting inductance M
Λ _2d * / M, the target value v ₁ γ *, v ₁ δ * of the γ- and δ-axis components of the primary voltage based on the calculation result of the first coordinate conversion unit and the command value ω ₀ of the power supply angular frequency. And a second coordinate conversion unit that converts the primary current detection value of the induction motor into each axis component i ₁ γ, i ₁ δ of the γ-δ coordinate. When, in proportion to the product of the target value i ₁ [delta] * and the second temporal change rate calculated by this and leakage inductance Lσ current deviation between i ₁ [delta] than the coordinate transformation unit of [delta] axis component of the primary current Term output, and an integral element that outputs the integrated value of the current deviation as an integral term output. The sum of the proportional term output and the integral term output is calculated as v in the δ-axis component of the current primary voltage. ₁ δ
* Outputs a voltage change Delta] v ₁ [delta] from, based on the integral term output Delta] v ₁ .DELTA..delta a current control amplifier to output as _I, i ₁ gamma * and i ₁ gamma than the second coordinate conversion unit Means for calculating the variation Δv ₁ γ from v ₁ γ * in the γ-axis component of the current primary voltage; i _1d *, i _1q *, i ₁ γ *, λ _2d *, and the primary power supply angular frequency ω _0, the set value M * and Delta] v ₁ [delta] and calculates the variation with respect to the set value of the rotor resistance based on the _I secondary resistance variation calculation part of the exciting inductance provided, v ₁ gamma * and Delta] v ₁ and gamma The added value is a target value v ₁ γ of the γ-axis component of the primary voltage, and the added value with v ₁ δ * Δ ₁ δ is a target value v ₁ δ of the δ-axis component of the primary voltage, and these target values v ₁ γ ,
While controlling the power supply voltage on the basis of v ₁ δ, the slip angular frequency calculation unit calculates the secondary time constant based on the set value of the secondary time constant and the calculation result obtained by the secondary resistance change calculation unit. The next time constant is calculated, and this second time constant, i _1q *
And a calculation using λ _2d * / M *. [Equation 36] 5. A vector control apparatus for an induction motor according to claim 4, instead of using the secondary resistance change calculation section, a target value of Delta] v ₁ [delta] _I integral term output Delta] v ₁ [delta] _I Toko current control amplifier A voltage fluctuation control amplifier for inputting a deviation from zero and outputting a fluctuation Δω _s from a slip angle frequency target value ω _s * is provided. Δω _s from this voltage fluctuation control amplifier and slip angular frequency calculation A vector control device for an induction motor, characterized in that an addition value to ω _s * obtained in a section is set as a target value of a slip angular frequency. 6. The means for calculating Δv ₁ γ calculates a temporal change rate of a current deviation between i ₁ γ * and i ₁ γ from the second coordinate conversion unit, and calculates the rate of change of the current deviation with the leakage inductance Lσ. Of the current primary voltage, and a proportional element that outputs the product of the proportional term and an integral element that outputs the integrated value of the current deviation as an integral term output. and outputs v as voltage change Delta] v ₁ gamma from ₁ gamma * in gamma-axis component, constituted by a current control amplifier outputting the integral term output as Δv ₁ γ _I, Δv ₁ during no-load operation
Based on γ _I , the set value of the primary resistance R ₁ * and i _1d *, a change in the set value of the primary resistance is calculated, and Δv ₁
δ _I , M *, secondary self inductance L ₂ *, ω ₀ and λ
An identification circuit unit for calculating a change in the set value of the excitation inductance based on _2d * is provided, and R ₁ *, M * ² /
The vector control device for an induction motor according to claim 4, wherein L ₂ * is identified. 7. A vector control apparatus for an induction motor according to claim 1, wherein, instead of using the secondary resistance change calculation section, Delta] v ₁ [delta] * variation from partial Delta] v ₁ [delta] in [delta] -axis component of the current of the primary voltage And a deviation of the Δv _{1q from} the target value zero is input, and a secondary resistance change amplifier for directly obtaining the secondary resistance change is provided as an output. The output of this amplifier and the target value ω _s * of the slip angular frequency are calculated. By multiplying, the variation Δω _s of the slip angular frequency from the target value ω _s is obtained, and the sum of Δω _s and ω _s * obtained by the slip angular frequency calculator is calculated as the target value of the slip angular frequency. A vector control device for an induction motor, comprising: 8. The vector control apparatus for an induction motor according to claim 4, instead of using the secondary resistance change calculation section, Delta] v ₁ of the integral term output Delta] v ₁ [delta] _I Toko current control amplifier
inputs the deviation between the target value zero [delta] _I, the secondary resistance variation amplifier to obtain a secondary resistance variation directly as the output provided, multiplying the target value omega _s * of the slip angular frequency The amplifier output Thus, the variation Δω _s of the slip angular frequency from the target value ω _s is determined, and the sum of Δω _s and ω _s * determined by the slip angular frequency calculator is set as the target value of the slip angular frequency. Characteristic vector control device for induction motor. 9. The vector control apparatus for an induction motor according to claim 2, wherein the gain of the voltage fluctuation control amplifier is changed in inverse proportion to the primary angular frequency, and the gain is subjected to upper and lower limiters. Vector control device for induction motor. 10. The vector control device for an induction motor according to claim 7, wherein the gain of the secondary resistance change amplifier is changed in inverse proportion to the primary angular frequency, and an upper and lower limiter is applied to the gain. Vector control device for induction motor. 11. The vector control device for an induction motor according to claim 4, wherein when outputting the integral term output of the current control amplifier to the secondary resistance change calculating section, the integral term output is given via a first-order lag filter. A vector control device for an induction motor, further comprising upper and lower limiters that are variable in a filter time constant. 12. The vector control device for an induction motor according to claim 1, wherein a change in the set value of the secondary resistance with respect to the set value of the secondary resistance is calculated by the secondary resistance change calculation unit based on the change Δv ₁ δ. The calculated secondary resistance change data is held as an initial value by the hold unit, and when the normal operation is started, the data of the hold unit is given to the output terminal of the secondary resistance change calculation unit as the initial data. Vector control device for induction motor. 13. The vector control device for an induction motor according to claim 7, wherein the variation Δv ₁ δ is input to a secondary resistance change amplifier, and an initial value of the secondary resistance change obtained at the amplifier output. A vector control device for an induction motor, characterized in that data is held in a holding unit and, when normal operation is started, the data in the holding unit is given as initial data to an output terminal of a secondary resistance change amplifier.