JP2943377B2 - Vector controller for induction motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a vector control device for an induction motor.

[0002]

2. Description of the Related Art Vector control of an induction motor for controlling a secondary magnetic flux and a secondary current orthogonal to the secondary magnetic flux without interference has been widely applied.

In this vector control, in the case of a three-phase induction motor, a current or a magnetic flux is treated as a vector of a dq coordinate system of two orthogonal axes rotating at the same speed as a rotating magnetic field generated by a power source.
This is a method in which the operation result is converted into a current command value of each phase of the three-phase power supply and controlled.

[0004] The specific method is as follows.
The voltage equation in the coordinate system is expressed by the following equation (1).

[0005]

(Equation 1)

Where ω _s = ω−ω _r , Lσ = (L ₁ L ₂ −M
²⁾ / L _2.

Here, v _1d and v _1q are the primary voltages d and q, respectively.
Axis components, i _1d and i _1q are the d and q axis components of the primary current, λ
_2d and λ _2q are the d and q axis components of the secondary magnetic flux, R ₁ and R ₂ are the primary and secondary resistances respectively, L ₁ , L ₂ and M are the primary and secondary, the excitation inductance, ω, ω _r and ω _s represent the primary power supply angular frequency, the rotor angular frequency, and the slip angular frequency, respectively, and P represents d / dt.

If the d-axis is set on the secondary magnetic flux in the dq coordinate system, λ _2q = 0. At this time, λ _2d = Φ ₂ = constant, i
_2d = 0, i _2q = i quadrature control similar torque and flux and ₂ next to the DC motor becomes possible.

On the other hand, the secondary magnetic flux has the following relationship.

[0010]

(Equation 2)

According to the vector control condition, i _2d = 0,
From equation (2), λ _2d = Mi _1d .

From λ _2q = 0, i _1q = −L ₂ / M ·
i _2q , where i _1q is proportional to the torque current.

Next, equation (3) is obtained from the fourth line of equation (1). When the condition of the slip angular frequency ω _s is obtained from equation (3), ω _s is expressed by equation (4).

[0014]

(Equation 3)

The above is the vector control condition when the control is performed so that the secondary magnetic flux coincides with the d-axis. Therefore, in order to perform vector control, it is necessary to set i _1d to λ _2d / M and control ω _s so that equation (4) is satisfied.

Here, the resistance of the secondary resistor R ₂ used for calculating the slip angular frequency ω _s changes with the ambient temperature and temperature changes such as self-heating of the rotor, so that the resistance of the secondary resistor R ₂ is determined based on the output voltage of the motor. It is necessary to estimate the change and correct the target value of the slip angular frequency ω _{s with} the change to compensate for the generated torque change due to the change in the secondary resistance. If the change in the secondary resistance is neglected, the torque control accuracy and the torque response deteriorate. If such a change in the secondary resistance is estimated by using the output voltage of the inverter as it is, for example,
Since a change in the secondary resistance is taken in, it is desirable that the signal used for estimation is a signal that is not affected by the primary resistance.

For this reason, a control circuit shown in FIG. 8 has already been proposed. In the figure, reference numeral 1 denotes an excitation current command unit.
Until the angular frequency ω _r exceeds a certain value, λ _2d * / M * is i _1d
And the target value i _1d * of, ω _r is the i _1d * is reduced when it exceeds a certain value. Hereinafter, when the target value or the ideal value is indicated with * , the deviation between the speed commands ω _r * and ω _r is set to i _1q * through the speed amplifier 2, and the dq axis is determined based on i _1d * and i _1q *. The ideal values v _1d *, v _1q * of the above primary voltage are obtained by calculation, and the correction of the voltage fluctuation due to the change in the primary resistance and the secondary resistance is i _1d *.
= I _1d , i _1q * = i _1q , i _1d *
Δv _1d is obtained for the PI amplifier 3 ₁ that controls = i _1d ,
i _1q * = i is the PI amplifier 3 ₂ for controlling _1q Delta] v _1q is obtained. Since Δv _1d and Δv _1q include both the voltage change due to the change in the primary resistance and the secondary resistance, if the component that does not include the voltage change due to the change in the primary resistance is obtained, the compensation for the change in the secondary resistance is performed. Compensation that is not affected by the primary resistance change is possible. So taking the rotating coordinate gamma-[delta] axes spaced reference axis gamma on the vector of the primary current I _1, are determined by the primary voltage change Delta] v ₁ [delta] a slip correction calculation unit 3 ₃ of the [delta] axis. This Δv ₁ δ is represented by an expression not including the primary resistance R ₁ , and thus is not affected by the primary resistance R ₁ . Since Δv ₁ δ is also used in the present invention, it will be described in detail in the description of the present invention.

FIG. 6 is a vector diagram showing the relationship between the dq axis and the γ-δ axis and the voltage and the current, and FIG. 7 is a vector diagram showing the primary voltage fluctuation . In FIG. In FIG. 7, Δv ₁ is the primary voltage variation, Δv ₁ γ, Δv ₁ δ are the γ-axis components of the variation, δ
Axis component, ψ is phase between γ axis and d axis, I ₀ is excitation current,
I ₂ is the torque current. Δv ₁ δ is represented by the following equation (5).

Δv ₁ δ = −Δv _1d · sinψ + Δv _1q cosψ (5) where cosψ = I ₀ / I ₁ = i _1d / i ₁ γ, sinψ = I ₂ / I
₁ = i _1q / i _{1 γ} and obtains a correction amount [Delta] [omega _s of the slip angular frequency corresponding to the secondary resistance variation based on the slip correction calculation unit 3, ₃ Delta] v ₁ [delta] in operation, the slip angular frequency calculation unit 3 ₄ The added value of ω _s * and Δω _s obtained in the above is set as the target value of the slip angular frequency, and the rotor angular frequency ω _r is added _thereto, and the angular frequency of the primary voltage ω
= Dθ / dt. 8 3 ₅ polar conversion unit figure, 3 ₆ coordinate conversion unit, the 4 ₁ PWM circuit, 4 ₂ inverters, IM induction motor, PP pulse pickup unit, 4 ₃ is a speed detecting unit.

[0020]

(A) Since the primary voltage fluctuations Δv _1d and Δv _1q include both the fluctuation of the primary resistance and the fluctuation of the secondary resistance, the circuit shown in FIG. Delta] v _1d in part 3 _3, further calculates a Delta] v ₁ [delta] that is not affected by the primary resistance change from Delta] v _1q, calculates the [Delta] [omega _r further from the Delta] v ₁ [delta].

(B) When performing field control, λ _2d and i _1d
Is in the relationship of the following equation (6) from the third line of the equation (1). Also, since λ _2q = 0, equation (7) holds.

[0022]

(Equation 4)

From equation (7), it can be seen that during field control, i _1d is controlled in a first-order manner with respect to a change in λ _2d . That is, when the field command λ _2d * is changing, λ _2d = Mi _1d does not hold.

However, in the conventional circuit, since the field control is not taken into consideration, the theoretical development is performed with the excitation current i _1d constant, that is, i _1d = λ _2d / M, and the slip correction calculation is executed. Was. For this reason, in the field control region, the set value of the slip angular frequency cannot be accurately calculated, and is not an effective method. The equivalent circuit of the induction motor used for the vector control is configured as shown in FIG. 3, and the exciting inductance M 'changes depending on the frequency and the exciting current, and has characteristics as shown in FIG. Therefore, M ′ and I ₀
If the ratio is controlled as a constant value, accurate torque control becomes impossible. In particular, the variation of M 'in the constant output region is large, and the torque accuracy at the constant output may be reduced.

It is an object of the present invention to provide a vector control device capable of improving the torque control accuracy by compensating for the fluctuation of the excitation inductance over the entire operation range.

[0026]

As described above, not only the secondary resistance but also the primary resistance changes with temperature, and therefore it is ideal to perform the secondary resistance compensation without being affected by the primary resistance change. is there. Here, in the present invention, the variation amount Δv ₁ δ of the δ-axis component of the primary voltage is used as in the circuit of FIG.
A further advanced control method was adopted.

That is, in the vector control shown in FIG. 8, the d-axis of the rotation coordinate dq axis is made the same axis as the secondary magnetic flux, so that
The excitation current i _1d and the torque current i _1q are controlled so as to maintain orthogonality.

This time, a method of controlling the γ-axis on the primary current I ₁ using the rotational coordinates as the γ-δ axis was studied.
However, since vector control requires control on the dq axis, it rotates at the same speed as the power supply angular frequency ω _0, and uses the dq axis and the γ-δ axis having different phases together. New control method.

Considering the primary current I ₁ on the γ-δ axis, if the primary voltage variation due to the secondary resistance change is detected as the variation Δv ₁ δ on the δ axis, the voltage variation due to the primary resistance is included. Since there is no voltage component, robust secondary resistance compensation is possible.

Therefore, the ideal primary voltage v on the γ-δ axis
₁ γ * and v ₁ δ * are obtained by calculation, and the correction of the voltage fluctuation due to the change in the primary resistance and the secondary resistance is executed by controlling so that i ₁ γ = I ₁ and i ₁ δ = 0. By performing such control, Δv ₁ γ is obtained for the PI amplifier output controlling i ₁ γ = I _1, and Δv ₁ δ is obtained for the PI amplifier output controlling i ₁ δ = 0. Since Δv ₁ δ does not include a voltage component due to a primary resistance change, it can be used for compensating for a secondary resistance change. That is, if the secondary resistance change is compensated for using Δv ₁ δ, ideal compensation independent of the primary resistance change can be performed.

As described above, using the γ-δ axis using the primary current I ₁ as a reference value has the advantage that primary voltage fluctuation data used for compensating for a change in the secondary resistance can be directly obtained on the δ axis.

In the present invention, when the field command λ _2d * is changed from the above equation (7), λ _2d = Mi _1d does not hold, so that λ _2d / M and i _1d are used separately. ing.

Hereinafter, the present invention will be described in detail.

(A) Vector Control Conditions When Using γ-δ Axis FIG. 3 is an asymmetric TI equivalent circuit of an induction motor, and FIG. 4 is a vector diagram corresponding to this equivalent circuit.

Now, if the γ axis is set on the primary current I ₁ , i ₁ γ = I
₁ , i ₁ δ = 0. "Conventional technology" for γ-δ axis
Equation (1) holds true for the equation (1)
When d and q in the equation are changed to γ and δ, respectively,
Equations (8) and (9) hold from the fourth row.

[0036]

(Equation 5)

If the term including P is removed, ω always holds.
The condition of _s is required. The following equation is obtained from the equation (9).

R ₂ / L ₂ + P = −λ ₂ γ · ω _s / λ ₂ δ (10) When the equation (10) is substituted into the equation (8), the following equation is obtained. Holds.

[0039]

(Equation 6)

Here, the relationship between λ _2d and λ ₂ γ, λ ₂ δ is as follows.

Λ ₂ γ = λ _2d cosψ, λ ₂ δ = −λ _2d sinψ (12) λ ₂ γ ² + λ ₂ δ ² = λ _2d ² (13) Equations (12) and (13) By substituting into equation (11), the following equation is obtained.

[0042]

(Equation 7)

As described above, the γ-axis is controlled on the primary current I ₁ so that i ₁ γ = I ₁ and i ₁ δ = 0, and d
If the vector control condition on the −q axis is satisfied, ω _s is expressed by the same formula as the formula (4) in the “Prior Art” section, and it has been found that the same condition can be obtained. However, if λ _2d ≠ Mi _1d is considered in consideration of the field control region,
From the equation at the first stage of equation (14) and the equivalent circuit of FIG. 3, R ₂ / L ₂
Becomes R ₂ '/ M', so that ω _s is expressed as in equation (15).

[0044]

(Equation 8)

(B) Ideal voltage on the γ-δ axis On the γ-δ axis, i ₁ γ * = I ₁ and i ₁ δ * = 0 are controlled. The next (1
6) is obtained. If the term with P is omitted in equation (16), equation (17) is obtained.

[0046]

(Equation 9)

Here, when the vector control condition is satisfied, (18)
The equation holds. By satisfying the expression (18), (1
9) is obtained.

[0048]

(Equation 10)

When the torque current command i _1q * changes suddenly or when the exciting current command i _1d * changes due to the field control, the term of Pi ₁ γ applied to Lσ in the equation (16) cannot be ignored. . The ideal voltage when this P term is considered is as follows.

[0050]

[Equation 11]

(C) Fluctuation of the secondary magnetic flux when the secondary resistance changes The fluctuation of the secondary magnetic flux when the secondary resistance changes will be examined. The following equation is obtained from the third and fourth lines of the equation (1).

[0052]

(Equation 12)

When L ₂ / R ₂ is multiplied by the equations (21) and (22), the following is obtained.

[0054]

(Equation 13)

When λ ₂ γ is obtained from equations (23) and (24), (23) × (1 + L ₂ P / R ₂ ) becomes equation (25), and (24) × L ₂ ω _s / R ₂ becomes Equation (26) is obtained. Also, from (25) + (26), λ ₂ γ is as shown in equation (27).

Next, when λ ₂ δ is obtained from equations (23) and (24), (23) × L ₂ ω _s / R ₂ becomes equation (28), and (24) × (1 + L ₂ P / R) ₂ ) becomes the equation (29). Further, when λ ₂ δ is obtained from (29)-(28), it is as shown in equation (30).

[0057]

[Equation 14]

Here, the following assumption is made.

(A) Assuming that the current is controlled to flow according to the command value, i ₁ γ * = i ₁ γ, i ₁ δ * = i ₁ δ =
Set to 0. The current on the dq axes is i _1d * = i _1d , i
_{Let 1q} * = i _1q .

(B) Assuming that the secondary resistance change is K, R ₂ =
Since (1 + K) R ₂ *, L ₂ ω _s / R ₂ in the equations (27) and (30) can be expressed as follows.

[0061]

(Equation 15)

(C) It is assumed that the exciting current is controlled as shown by the equation (7), so that the equation (32) holds.

[0063]

(Equation 16)

(D) Assuming that the secondary resistance compensation is performed, 1
+ L ₂ P / R ₂ transient term time constant L ₂ / R ₂ = L ₂ * / R ₂ *
Assume that (Short time and R ₂ does not change.) Therefore, the following equation is established.

[0065]

[Equation 17]

By substituting the above relational expressions into the expressions (27) and (30) and transforming them, the expressions (34) and (35) are obtained.
Here, the ideal value of the secondary magnetic flux on the γ-δ axis is (36) to (3).
8) It is expressed by the equation.

[0067]

(Equation 18)

By multiplying the denominator and the numerator of the equation (34) by i ₁ γ * and using the equation (36), the secondary magnetic flux variation Δλ
₂ gamma is expressed as equation (39).

[0069]

[Equation 19]

By multiplying the denominator and the numerator of equation (35) by i ₁ γ * and using equation (37), the secondary magnetic flux variation Δ
λ ₂ δ is represented by equation (40).

[0071]

(Equation 20)

(D) Temporary Voltage Fluctuation When Secondary Magnetic Flux Fluctuates The primary voltage when the secondary magnetic flux fluctuates can be expressed as follows from equation (16).

[0073]

(Equation 21)

Since the ideal value of the primary voltage is expressed by the equation (19), from the equations (19) and (41) in consideration of the equation (18), the voltage fluctuations Δv ₁ γ and Δv ₁ δ are become that way.
However, the expansion of Δλ ₂ δ and Δλ ₂ γ are (40) and (3
9) The formula is used.

[0075]

(Equation 22)

Here, since v ₁ γ includes the component of R ₁ i ₁ γ *, the voltage fluctuation due to the change in the primary resistance R ₁ is also v ₁ γ.
Will be included. Therefore, when also considering the change in the primary resistance R ₁ (42) equation is as follows. However, K ₁ is the primary resistance variation.

[0077]

(Equation 23)

As described above, since Δv ₁ γ includes the variation of the primary resistance R ₁ , it is not suitable for use in compensating for a change in the secondary resistance R ₂ . On the other hand, since Δv ₁ δ does not include the component of R ₁ , it is considered to be a voltage fluctuation component due to a secondary resistance change. Therefore, by performing the detecting and secondary resistance compensation a variation Delta] v ₁ [delta] of the primary voltage v ₁ [delta] of [delta] axes, has the following advantages because it does not contain the influence of the primary resistance R _1.

[0079] (it) can be carried out secondary resistance compensation without being affected by the temperature variation of the primary resistance R _1.

(B) Although the influence of the voltage drop of R ₁ becomes large in the low-speed region, the primary voltage v ₁ δ on the δ-axis does not include the voltage drop of R ₁ , so the secondary resistance is low even in the low-speed region. Compensation can be performed accurately.

[0081] (c) be performed Delta] v ₁ [delta] than the secondary resistance compensation, the Delta] v ₁ gamma only the voltage component due to R ₁ variation occurs. This allows estimation of R _1. R ₁ is considered to containing such V _CE partial voltage drop main circuit elements of the resistance component dead time of the primary resistance cable.

(E) Calculation of Secondary Resistance Change K By modifying equation (43), the following equation (45) is obtained.

[0083]

(Equation 24)

Therefore, if the primary voltage fluctuation Δv ₁ δ on the δ axis can be detected, the secondary resistance change K can be obtained from equation (45).

(F) Method of Identifying Primary Resistance and Exciting Inductance During No-Load Operation In the present invention, in addition to compensating for a change in secondary resistance, the primary resistance and exciting inductance can be identified as described below. When the exciting inductance changes, the shunt ratio between the exciting current and the torque current changes, and the primary voltage also changes. Primary voltage in order to vary similarly be secondary resistance changes, it is impossible to distinguish changes in the excitation inductance M and the secondary resistance R _2. However, during the no-load operation, the torque current i _1q = 0, so that the primary voltage fluctuation does not have the effect of the secondary resistance change. Therefore, the excitation inductance can be compensated using the primary voltage fluctuation during the no-load operation.

A vector diagram at the time of no-load operation can be represented as shown in FIG. 5 by a TI equivalent circuit. At the time of no-load operation, since the torque current i _1q = 0, the dq axis and the γ-δ axis match. Therefore, consider the dq axis. The primary voltage at the time of no-load operation can be expressed as follows from Expression (16). However, i _1q = 0, and the P term is ignored.

[0087]

(Equation 25)

Here, the following assumption is made.

(A) Assuming that the current is controlled so as to flow according to the command value, i _1d * = i _1d .

(B) The change in the excitation inductance is expressed as A _M
far.

(C) The change in the primary resistance is defined as A ₁ .

(D) Add * to the set value of the motor constant.

(E) Assuming that the leakage inductance Lσ is small, the change is ignored.

Now, the ideal voltage at the time of no-load operation can be expressed as follows from the equation (46).

[0095]

(Equation 26)

The primary voltage is expressed by using the primary resistance change A ₁ and the excitation inductance change A _M as follows.

[0097]

[Equation 27]

From equations (47) and (48), the primary voltage fluctuations Δv _1d and Δv _1q during no-load operation are as shown in equation (49). The (49) exciting inductance variation A _M and the primary resistance change in A ₁ from equation is as equation (50).

[0099]

[Equation 28]

In a steady state where the excitation command does not change, λ _2d *
= M * i _1d *, and A _M is as follows.

[0101]

(Equation 29)

As described above, it was found that the primary resistance and the excitation inductance can be identified by detecting the primary voltage fluctuation during the no-load operation. The summary is as follows.

[0103] (b) reveals primary resistance change in A ₁ than the primary voltage change Delta] v _1d of d-axis.

[0104] (b) primary voltage variation Δv _1q than exciting inductance variation A _M of the q-axis is seen.

(G) Means of the Invention If the target value R ₂ * of the secondary resistance matches the actual secondary resistance, ω _s is obtained based on the equation (15), and this is calculated as ω _s * The secondary resistance changes with temperature. Therefore, in the present invention, K is calculated using Δv ₁ δ, and R ₂ * is corrected using this K to obtain ω _s *. To obtain ω _s *, the following equation (52) obtained from equation (15) is used.

[0106]

[Equation 30]

On the other hand, the primary resistance also changes depending on the temperature.
As can be seen from equation (43), v ₁ δ does not include the value of the primary resistance, and therefore does not depend on the change in the primary resistance when compensating for the secondary resistance. Although this point is common to the circuit shown in FIG. 8, the circuit shown in FIG. 8 performs current control in the dq coordinate system, whereas the present invention uses γ-current control.
The primary voltage is controlled on the basis of current control in the δ coordinate system, whereby the output of the current control amplifier is Δv ₁ γ, Δv ₁ δ
And the secondary resistance is compensated using Δv ₁ δ.

More specifically, a first coordinate converter for calculating a target value i ₁ γ * (= I ₁ ) of the γ-axis component of the primary current and the phase ψ based on i _1d * and i _1q * , the ratio between the target value of lambda _2d * and the excitation inductance _{M * λ 2d * / M *} , γ of the primary voltage based on the command value omega ₀ of the result and power supply angular frequency of the first coordinate conversion unit, [delta] axis Component target values v ₁ γ *, v ₁ δ
And a second means for converting the detected value of the primary current of the induction motor into each axis component i ₁ γ, i ₁ δ of the γ-δ coordinate.
, The target value i ₁ δ * of i ₁ γ * and the δ-axis component of the primary current, and i ₁ γ, i ₁ from the second coordinate conversion unit.
Based on δ, v in the γ-axis component of the current primary voltage
₁ gamma * and variation Delta] v ₁ gamma from, means for calculating a variation Delta] v ₁ [delta] from v ₁ [delta] * in the [delta] -axis component of the current of the primary _{voltage, i 1d *, i 1q *} , i 1 γ *, Λ _2d *, primary power supply angular frequency ω ₀ , excitation inductance set value M * and Δv ₁
a secondary resistance change calculator for calculating a change with respect to the set value of the secondary resistance based on δ, and calculating a sum of v ₁ γ * and Δv ₁ γ as a target value v of the γ-axis component of the primary voltage. ₁ and gamma, also v ₁ [delta] * the sum of the a Delta] v ₁ [delta] and the target value v ₁ [delta] of [delta] axis component of the primary voltage, controls the power supply voltage on the basis of these target values v ₁ gamma, the v ₁ [delta] A secondary time constant at that time is obtained by the slip angular frequency calculator based on a set value of the secondary time constant and a calculation result obtained by the secondary resistance change calculator, and this secondary time constant is obtained. , I _1q * and λ _2d * / M *. In addition, during the no-load operation, a change with respect to the set value of the primary resistance is calculated based on Δv ₁ γ, the set value R ₁ * of the primary resistance and i _1d *, and Δv ₁ δ, M *, the secondary self inductance L ₂ *, ω
An identification circuit unit is provided for calculating a change in the exciting inductance with respect to a set value based on ₀ and λ _2d *. A data table of the change in the excitation inductance output from the identification circuit as a change in the excitation inductance at a point obtained by dividing the set value of the excitation inductance calculated by the identification circuit unit by an arbitrary number of divisions of the entire operation range. And an excitation inductance conversion data section obtained by linearly interpolating the data of the data table is provided. This data section outputs M 'change coefficient data KM'.

Further, in the present invention, instead of using the secondary resistance change calculator, v ₁ in the δ-axis component of the current primary voltage is used.
A slip angular frequency control amplifier which inputs a deviation Δv ₁ δ from δ * and a deviation of the target value of Δ ₁ δ from zero and outputs a fluctuation Δω _s of the slip angular frequency from the target value ω _s *. The same operation and effect can be obtained even if the added value of Δω _s from the slip angular frequency control amplifier and ω _s * obtained by the slip angular frequency calculator is used as the target value of the slip angular frequency. In addition, in the present invention, a secondary resistance change amplifier is provided in the slip angle frequency control amplifier, the output of the amplifier is multiplied by a slip frequency set value, and the obtained value is obtained as a variation.
The variation may be multiplied by ω _s * to obtain the slip angular frequency, or a limiter may be applied to the slip angular frequency control amplifier and the secondary resistance change amplifier.

[0110]

FIG. 1 is a circuit diagram showing a first embodiment of the present invention, in which the same reference numerals as in FIG. 8 denote the same parts. 1 ₁
Depending on the angular frequency ω _r of than the speed detection unit 4 ₃ λ _2d * / M
* A secondary magnetic flux instruction amplifier for outputting, omega _r to over a certain value and outputs a value obtained by dividing by the KM 'a lambda _2do * / M *, field control beyond a certain value omega _r region Λ _2d * / M * decreases according to ω _r . 1 ₂ (7), that is _{λ 2d * / M * (1} + L 2 * / R 2 * · S)
Is an operation unit that executes the operation of

5 ₁ is a first coordinate converter, i
Γ based on primary current I ₁ based on _1d * and i _1q *
It has a function of calculating i ₁ γ * on the −δ coordinate and the phase difference と between the d axis and the γ axis, and specifically executes the calculation of the following equation.

[0112]

(Equation 31)

[0113] 5 ₂ is an ideal voltage calculation unit for calculating a target value of the primary voltage, the first coordinate conversion unit 5 ₁ than output the Sinpusai, I _1, from cosψ and the secondary magnetic flux instruction amplifier 1 ₁ Λ _2d * / M * and the power supply angular frequency ω ₀ (1
9) Execute the calculation of the expression to calculate v ₁ γ * and v ₁ δ *.
Incidentally, the KM 'is supplied to the ideal voltage calculation unit 5 _2.

Reference numeral 6 denotes a second coordinate converter, which converts the primary current detection values i _u , i _w into respective axis components i ₁ γ, i ₁ δ of the γ-δ coordinate. These i ₁ γ and i ₁ δ are the target values i ₁ γ *,
is compared with i ₁ δ * (= 0), and the deviation is input to PI amplifiers 7 and 8 which are current control amplifiers, respectively. Δv ₁ γ and Δv ₁ δ are output from the PI amplifiers 7 and 8, respectively. As described above, Δv ₁ γ is v ₁ γ * and Δv ₁ δ is v ₁ δ
* Is added to each. Numeral 9 denotes a polar coordinate converter, which outputs the magnitude | V ₁ | of the primary voltage vector V ₁ and the phase angle φ with the γ-axis ( see FIG. 4 ). The phase angle phi, is summed with θ described later and ψ (= ω ₀ t), these sum values and | V ₁ | and is inputted to the PWM circuit 4 ₁ U, V, primary voltage corresponding to the W-phase It is converted into the command value, thereby the inverter 4 ₂
Is controlled.

Reference numeral 10 denotes a secondary resistance change calculating section, which is λ _2d
* / M *, i _1d *, i _1q *, ω ₀ , i ₁ γ * and Δv ₁ δ
And the calculation of the equation (45) is performed to obtain the secondary resistance change K. Reference numeral 11 denotes a slip angular frequency calculation unit which has a function of taking in K, λ _2d * / M * and i _1q * and executing the equation (52) to obtain ω _s . The slip angle frequency calculator 11 is supplied with the KM '. By the way, when the computer executes the operation of each part of the circuit of FIG. 1, ω _s is calculated as follows. That is, a series of calculations including the calculation of K and the slip angle frequency calculation are instantaneously performed by the clock signal, and the secondary resistance value obtained in the (n-1) th calculation in the slip angle frequency calculation unit 11 is calculated by the nth calculation. The set value in. represents K and R ₂ obtained in the n-th calculation each K _n, as R _2n, R
_{When a} preset value R ₂ * is assigned to the initial value R _{20 of 2n} , the first to n-th calculations are as follows.

First time R ₂₁ = (1 + K ₁ ) · R ₂₀ = (1 + K ₁ ) · R ₂ * Second time R ₂₂ = (1 + K ₂ ) · R ₂₁ = (1 + K ₂ ) · (1 + K ₁ ) · R ₂ * : N-th time R _2n = (1 + K _n ) · R _{2 (n-1)} = (1 + K _n ) (1 + K _n-1 ) ... (1 + K ₁ ) · R ₂ * Therefore, ω _s obtained in the n-th calculation is ω _sn Expressed as
ω _sn is given by the following equation (53). ω _sn = (1 + K _n ) · ω _{s (n-1)} (53) The ω _{s (n−1)} obtained in the (n−1) th calculation is keep in stores, omega _sn is obtained by using the K _n obtained by equation (53).

In this case, the initial value ω _s1 is ω _s1 = (1 + K ₁ )
· R ₂ * · 1 / L ₂ * · i _1q * / (λ _2d * / M *). The obtained ω _s and the detected rotor angular frequency ω _{r of the} electric motor IM are added, and the added value ω ₀ is set as a target value of the power supply angular frequency.

Numeral 12 denotes an identification circuit unit which takes in Δv ₁ γ and i _1d * during no-load operation and executes the upper equation of equation (50) to calculate a change A ₁ in the primary resistance. , The primary resistance is identified by Δv ₁ δ, ω ₀ and λ _2d *
/ M * is taken in, and the lower part of the equation (50) is executed to calculate the change A _M in the excitation inductance, thereby identifying the excitation inductance M ² / L ₂ . Here, it is determined whether or not the operation is the no-load operation.
Is driven by the comparator 13. Comparator 13, a set value, for example, the value of the 5% if the rated torque current is 100% as compared to i _1q * values, i _1q * is lower than the set value, determines that the no-load operation In this case, the identification circuit unit 12 is driven, and in this case, the influence of the secondary resistance change does not appear.

The change AM of the excitation inductance calculated by the identification circuit section 12 is at a measurement point at a certain point in the operating range. Thus, variations in the resulting AM M 'and R ₂ at one point with the operating range becomes impedance ratio, that distinction is Kashii very poorly as follows. That, R ₂ will vary with temperature, M 'is because changes due to excitation command frequency. For this reason, in this embodiment, no-load operation is performed, ΔV ₁ δ is measured at several measurement points in the operation range, the M ′ variation is calculated by the identification circuit unit 12, and the calculation result is referred to as the excitation inductance described later. It is stored in the conversion data section 77.

Then, during operation, M 'fluctuation compensation may be performed using the measured data. Specifically, the following two adjustments are performed.

(A) Change M * of the equivalent exciting current command λ ₂ d * / M * (make the magnetic flux λ ₂ d * constant). This approximates M ′ * = M *.

(B) M ′ * = M ₂ * / L ₂ * used for the γ, δ axis non-interference control voltage calculation is changed.

For the M ′, ΔV ₁ δ is measured in a no-load operation, and its variation is obtained. Considering the measurement time,
For example, n _max ≦ 16. Setting from the panel is a constant torque range measurement points n _T and a constant output range measurement points n _P. Each range is divided within the range of the number of measurement points to determine a measurement frequency. Here, FIG. 17B shows an example of measurement points when n _T = 2 and n _P = 9. And the variation of M ′ is (54)
Obtain from the formula.

[0124]

(Equation 32)

The M 'change coefficient data KM' obtained by using the above equation (54) is n _T points in the constant torque range and n _P points in the constant output range.
There are points. The obtained (measured) KM 'data is shown in FIG.
Is stored in the RAM area as image data and stored in the E ² PROM as setting data. K as a data table shown in FIG.
The M 'data has n _T points in the constant torque range and n _P points in the constant output range. The intermediate data is linearly interpolated based on the output frequency to obtain KM 'data.

The number of KM 'data is constant torque range n
There are _T and n _P constant output ranges. Ω _n data used for linear interpolation = constant torque range and constant output range of reference address pointer (n) are obtained as follows.

(A) Constant torque range

[0128]

[Equation 33]

Note that ω _PU is normalized data of ω at ω _TRQ .

(B) Constant output range

[0131]

(Equation 34)

Now, KM 'at the output angular frequency ω is linearly interpolated as follows.

Hereinafter, description will be made with reference to FIGS. The equation for linear interpolation is given by the following equation: (KM ′ _{n + 1} −KM ′ _n ) / (ω _{n + 1} −ω _n ) = (KM′−
KM ′ _n ) / (ω−ω _n ) By transforming this equation, KM′−KM ′ _n = (KM ′ _{n + 1} −KM ′ _n ) · (ω−
ω _n ) / (ω _{n + 1} −ω _n ). Here, if a = (ω−ω _n ) / (ω _{n + 1} −ω _n ), the above equation is expressed as KM ′ = a · KM ′ _{n + 1} + (1−a) · KM ′ _n Become.

Next, using the table reference address data n 'obtained above, a becomes as follows.

A = (n′−n) / (n + 1−n) = n′−n, 1−a = 1−n ′ + n = (n + 1) −n ′.

FIG. 21 shows the above relationship. That is, the integer part of the reference address data n 'in the KM' data table becomes address pointer data, and the decimal part becomes a data used for linear interpolation. Data obtained by subtracting a decimal part is (1-a) data.

The change (excitation) to the set value of the excitation inductance calculated by the identification circuit unit 12 as described above is determined as the change in the excitation inductance at a point obtained by dividing the entire operation range by an arbitrary number of divisions. And creates the data table described above. Then, an excitation inductance conversion data section 77 obtained by linearly interpolating the data of the created data table is provided. The output (KM ') of the data part 77 is supplied to the slip angular frequency calculation part 11, and means for dividing λ _2do * / M * and the target value V ₁ γ * of the γ and δ axis components of the primary voltage are provided. , V ₁ δ * are supplied to multiplying means provided in the means for calculating the respective values.

[0138] In the above, the arithmetic unit 5 ₂ v ₁ γ * in order to perform the operation of considering (20) a P term when calculating the in, the terms relating to i _{1 γ} * R ₁ * from R ₁ * (1 + L
σ / R ₁ * P)
A more accurate ideal voltage can be given, and the current response can be improved.

Next, a second embodiment in which the embodiment of FIG. 1 is improved will be described. If the change in the secondary magnetic flux is ignored from the equation (16), the following equation (16a) is obtained. Where λ ₂ γ, λ ₂ δ
_Represents λ _2d using equation (18).

[0140]

(Equation 35)

[0141] by a value corresponding to the temporal change rate when the primary current as can be seen from this equation is suddenly changed v ₁ γ, v ₁ δ
Changes. That is, the change in v ₁ δ includes the time change rate of the primary current in addition to the change in the secondary resistance, and the change in v ₁ γ includes the change in the primary resistance and the exciting inductance. In addition to the minute, the temporal change rate of the primary current is also included. Therefore, in the first embodiment of FIG. 1, when the primary current suddenly changes, the change is regarded as a change in the secondary resistance, and also as a change in the primary resistance and a change in the excitation inductance. Become.

Therefore, in the second embodiment shown in FIG.
₁ γ, LσPi ₁ δ, the primary voltage fluctuations (which are termed Δv ₁ γ, Δv ₁ δ), and the primary voltage fluctuations that do not contain them (which are Δv ₁ γ _I , Δv ₁ δ _I ), The primary voltage is controlled using the former values Δv ₁ γ, Δv ₁ δ, and the secondary resistance change is compensated using the latter values Δv ₁ γ _I , Δv ₁ δ _I and The primary resistance and the like are to be identified.

Specifically, as shown in the second embodiment of FIG. 16, for the PI amplifier 7, the proportionality for calculating (i ₁ γ * −i ₁ γ) × Lσ / T _s corresponding to LσPi ₁ γ is obtained. Element 7 ₁
And and a integral element 7 ₂ integrating the _{(i 1 γ * -i 1 γ} ), outputs the sum of the proportional term output from the proportional element ₇₁ and the integral term output from the integral element 7 ₂ as Delta] v ₁ gamma while, and configured to output the integral term output as Δv ₁ γ _I. With respect to the PI amplifier 8, integrating corresponding to _{LσPi 1 δ (i 1 δ *} -i 1 δ) × Lσ / T s the calculating proportional element ₈₁ and the _{(i 1 γ * -i 1 γ} ) integral and a component _82, and outputs the sum of the proportional term output from the proportional element ₈₁ and the integral term output from the integral element 8 ₂ as Delta] v ₁ [delta],
It is configured to output the integral term output from the integral element 8 ₂ as Δv ₁ δ _I. Where T _s indicates the operation cycle,
(I ₁ γ * −i ₁ γ) / T _s and (i ₁ δ * −i ₁ δ) / T _s are calculated by using a differential element.

According to such a configuration, even when the primary current changes abruptly, the effect does not appear on Δv ₁ γ _I and Δv ₁ δ _I , so that accurate secondary resistance compensation, identification of the primary voltage and the like are performed. be able to.

FIG. 2 is a circuit diagram showing a third embodiment of the present invention. Instead of using the secondary resistance change calculator 10, a PI amplifier 14 which is a voltage fluctuation control amplifier is used.
Enter the deviation between the target value zero to the amplifier 14 Delta] v ₁ [delta] and Delta] v ₁ [delta] to obtain a variation [Delta] [omega _s from the target value omega _s * in the current slip angular frequency as the output signal. The slip angular frequency calculating unit 15 assumes that R ₂ does not fluctuate from an ideal value.

[0146]

[Equation 36]

Ω _s * is calculated based on the ω _s * and Δ
The value added to ω _s is set as the target value of the slip angular frequency.
According to such an embodiment, the target value of the slip angular frequency is automatically corrected according to the secondary resistance change. Reference numeral 14 denotes a switch unit for invalidating the output of the PI amplifier 14 by the output signal of the comparator 16.

[0148] Also in the embodiment shown in FIG. 2, using the PI amplifier 7 and 8 shown in each of FIG. 16 as the PI amplifier 7, and inputs a Delta] v ₁ gamma _I in the identification circuit 12, Delta] v
By entering the deviation between ₁ [delta] * and Delta] v ₁ [delta] _I to the PI amplifier 14, it is possible to accurately perform such a secondary resistance variation compensation as previously described.

Next, a fourth embodiment and a fifth embodiment are shown in FIGS. In the fourth and fifth embodiments, the slip frequency is shown in equation (15). Now, when the torque component current command i _1q changes suddenly, or when the excitation current command λ _2d * / M * changes after entering the constant output range, the slip angular frequency ω _s changes. Therefore, [Delta] [omega _s must also be changed when i _1q and λ _2d * / M * is changed when using a voltage variation control amplifier for outputting a [Delta] [omega _s as a secondary resistance variation compensation amplifier. Therefore, the torque current command i _1q
And the response of the secondary resistance fluctuation compensation when the excitation current command λ _2d * / M * changes. That is, if the secondary resistance change K is directly output as the output of the secondary resistance compensation amplifier, the above-described disadvantage can be solved.

Therefore, when the slip frequency is expressed by using the secondary resistance change K, the equation (53) is obtained.

[0151]

(37)

From equation (53), assuming that the secondary resistance change is a constant value K, it can be seen that Δω _s changes due to changes in i _1q * and λ _2d * / M *. That is, it can be seen that the transient response deteriorates in the secondary resistance compensating amplifier (method for directly obtaining Δω _s ). And _{(i 1q *, λ 2d *} / M * of from must change [Delta] [omega _s amplifier output by a change) Therefore, the integral term output (error voltage) in the present embodiment Delta] v ₁ [delta] _I,
This Delta] v ₁ [delta] _I secondary resistance variation K from the deviation between the target value zero
Is provided, and the secondary resistance change K is used to obtain Δω _s = K × ω _s * from equation (53).
by adding the ω _s *, we seek ω _s.

FIGS. 9 and 10 show a fourth embodiment and a fifth embodiment. In FIG. 9, reference numeral 70 denotes an amplifier for changing the secondary resistance, and the output of the amplifier 70 is slid via a multiplier 71. It is added to the output ω _s * of the angular frequency calculation unit 15. The multiplier 71 is given ω _s *. FIG. 10 shows Δv ₁ δ _I *
= 0 and Delta] v ₁ [delta] Enter the deviation between _I to the secondary resistance variation amplifier 70, the output ω embodiments slip in the same manner as angular frequency arithmetic unit 11 of FIG. 9 through the multiplier 71 and the amplifier output K _s *
Is added.

If the secondary resistance change K is obtained as the secondary resistance compensation amplifier output as described above, the amplifier output K may be a constant value even when ω _s * changes. Therefore, even when the torque component current command i _1q * and the excitation component current command λ _2d * / M * change, and ω _s * changes suddenly, the response of the secondary resistance compensation becomes good.

Next, a sixth embodiment and a seventh embodiment will be described. Δv ₁ δ on the δ axis generated by the secondary resistance fluctuation can be expressed by equation (43). From equation (43), Δ
It can be seen that v ₁ δ changes in proportion to the primary angular frequency ω _O. For this reason, ω _O becomes extremely small in the low frequency region or when the motor is locked. As a result, Δv ₁ δ also has a very small value. When a secondary resistance compensation amplifier (Δω _s amplifier and secondary resistance change K amplifier) is used, the compensation response becomes slow in a low frequency region. (Input Δv ₁ δ or Δv ₁ δ of PI amplifier
_It takes time for the amplifier to swing because _I is very small. In this embodiment, the response is improved by changing the gain of the PI amplifier according to the frequency. The gain of the PI amplifier is varied by the following equation. Kp = Kp * × ω _OTRQ /
ω _O , where Kp is PI amplifier gain and Kp * is PI
The gain setting value at the time of ω _{OTRQ of the} amplifier, ω _OTRQ is the base angular frequency, and ω _O is the primary angular frequency. Further, taking into account the stability of the PI amplifier, the gain Kp ≦ limiter ≦ Kp _LIM (Kp
_LIM is variable). When ω _O is in the constant output range, ω _O = ω _OTRQ, and Kp = K
Let p *. Here, the configuration of the PI amplifier gain is shown.

FIG. 11 shows a sixth embodiment, and FIG. 12 shows a seventh embodiment. In both figures, 72 is the PI amplifier gain, 7
3 is an upper and lower limiter. With the configuration shown in FIGS. 11 and 12, by changing the PI amplifier gain Kp in inverse proportion to the frequency, the compensation response in the low-speed range can be made faster. Also, by providing upper and lower limiters,
By setting the upper limiter Kp _LIM in consideration of stability and setting the lower limiter to the set value Kp * at the base angular frequency _ωOTRQ , stable compensation can be performed in the entire operation range.

Next, an eighth embodiment will be described. The Δv ₁ δ of the δ axis generated by the secondary resistance fluctuation includes a 1f ripple particularly in the signal at a low frequency. Therefore, it is necessary to insert a filter to perform stable secondary resistance fluctuation compensation. In this embodiment, a first-order lag filter is inserted into Δv ₁ δ, and the filter time constant is changed in inverse proportion to the first-order frequency. The following equation is the transfer function of the filter. G (S) = 1/1 + ST ₁ , T ₁ = 1 / f ₀ = 2π /
ω _O , where T ₁ is the time constant of the first-order lag filter, f ₀ is the output frequency of the inverter, and ω _O is the first-order angular frequency.
In addition, the upper and lower limiters of the filter time constant are provided to stabilize the secondary resistance compensation.

FIG. 13 shows an eighth embodiment having a first-order lag filter. In the figure, 74 is a first-order lag filter, 75 is an upper / lower limiter, and 76 is a time constant of the first-order lag filter. By inserting a first-order lag filter 74 as shown in FIG. 13 and making the filter time constant inversely proportional to the frequency, the compensation can be stabilized. The upper and lower limiters 7
If the filter effect 5 is provided so that the filter effect can be varied in the high-speed range and the extremely low-speed range, the compensation can be stabilized over a wide range.

Next, a ninth embodiment and a tenth embodiment will be described. As described in the sixth and seventh embodiments,
From equation (43), it can be seen that Δv ₁ δ changes in proportion to the primary angular frequency ω _O. As a result, ω _O has a very small value in the low frequency region or when the motor is locked, so that Δv ₁ δ
Is also very small. When the secondary resistance compensation is performed in such a low frequency region, the calculation accuracy of the secondary resistance change calculation (45) becomes poor, or the secondary resistance change K amplifier or other PI amplifier is used. It takes time to identify the secondary resistance compensation (it takes time for the amplifier to swing to the secondary resistance change K because the input Δv ₁ δ and Δv ₁ δ _{I of the} PI amplifier are very small).

Therefore, in this embodiment, the accuracy of the secondary resistance fluctuation compensation in the low frequency region is improved and the identification time is shortened. As shown in equation (43), Δv ₁ δ is proportional to ω _O. When the motor is locked (ω _r = 0), ω _O = ω _s , so that Δv ₁ δ is proportional to the slip angular frequency ω _s . ω _s * is expressed by the following equation.

[0161]

(38)

When the torque current command i _1q * is small (lightly negative), ω _s * is also small, and Δv ₁ δ is also a small value. If Δv ₁ δ is small, the above-described problem occurs. Therefore, in applications such as a winder and an elevator of a steel line, a mechanical brake is attached to a motor, so that operation in a motor locked state is possible. In such a case, the motor is braked to lock the motor, and the torque current command i _1q * is set to a large value (for example, 50 to 100).
Do it to flow in the torque current command) of about%, omega _s
* Also increases, and a large value of Δv ₁ δ is obtained. Thus, by operating the secondary resistance change calculation (45) and the secondary resistance change K amplifier, it becomes possible to execute the secondary resistance change compensation with high accuracy and high identification speed. Performing this initial identification, holding the secondary resistance compensation data, and then entering normal operation with the hold data as the initial value eliminates the need for secondary resistance compensation identification time, and achieves highly accurate torque control. It becomes possible.

FIGS. 14 and 15 are flow charts showing the ninth and tenth embodiments. FIG. 14 is an initial identification flow chart for calculating the secondary resistance change, and FIG. 15 is an initial identification flow chart for the secondary resistance change K amplifier. is there.

[0164]

As described above, according to the present invention, the entire operation range is divided into arbitrary points, and ΔV ₁ δ or ΔV ₁ δI at each point during no-load operation is measured, and excitation is performed. An inductance change data table is created, an ideal voltage calculation is performed using the excitation inductance change data obtained by linearly interpolating between the measurement points, and an excitation command is calculated by substituting M * with the excitation inductance data. Therefore, the torque control accuracy can be improved by executing the compensation of the excitation inductance variation in the entire operation range.

[Brief description of the drawings]

FIG. 1 is a block circuit diagram showing a first embodiment of the present invention.

FIG. 2 is a block circuit diagram showing a third embodiment of the present invention.

FIG. 3 is an equivalent circuit diagram of the induction motor.

FIG. 4 is a vector diagram.

FIG. 5 is a vector diagram.

FIG. 6 is a vector diagram.

FIG. 7 is a vector diagram.

FIG. 8 is a block circuit diagram showing a comparative example of the vector control device.

FIG. 9 is a main part block circuit diagram of a fourth embodiment.

FIG. 10 is a main part block circuit diagram of a fifth embodiment.

FIG. 11 is a main part block circuit diagram of a sixth embodiment.

FIG. 12 is a main part block circuit diagram of a seventh embodiment.

FIG. 13 is a main part block circuit diagram of an eighth embodiment.

FIG. 14 is a flowchart of the ninth embodiment.

FIG. 15 is a flowchart of the tenth embodiment.

FIG. 16 is a block circuit diagram of a second embodiment.

17A is a characteristic diagram showing a relationship between a primary angular frequency ω ₀ and an excitation inductance M ′, and FIG. 17B is a characteristic diagram showing a relationship between a primary angular frequency ω ₀ and an excitation command.

FIG. 18 is an explanatory diagram of data storage in a RAM area.

FIG. 19 is an explanatory diagram of linear interpolation.

FIG. 20 is a detailed explanatory diagram of linear interpolation.

FIG. 21 is a detailed explanatory diagram of linear interpolation.

[Explanation of symbols]

1 ₁ ... Secondary magnetic flux command amplifier 1 ₂ ... Operation unit 2 ... Speed amplifier 5 ₁ ... First coordinate conversion unit 5 ₂ ... Ideal voltage calculation unit 6 ... Second coordinate conversion unit 7, 8 ... Current control amplifier PI amplifier 10 ... Secondary resistance change calculation unit 11, 15 ... Slip angular frequency calculation unit 12 ... Identification circuit unit 14 ... Voltage fluctuation control amplifier 77 ... Exciting inductance conversion data unit

Claims (11)

(57) [Claims] 1. A d-axis coordinate of a primary current of an induction motor, assuming that d-q coordinates are rotation coordinates rotating in synchronization with a power supply angular frequency of the induction motor and having a secondary magnetic flux as a reference axis. Induction motor having means for calculating target values i _1d * and i _1q * of the q-axis component, respectively, and a slip angular frequency calculator for calculating a slip angular frequency based on an arithmetic expression including a set value of a secondary time constant Means for calculating i _1d * comprises: means for outputting a target value λ _2d * / M * of the d-axis component of the secondary magnetic flux in accordance with the rotor angular frequency of the induction motor; and λ _2d * / M * and means for calculating i _1d * based on the differential term, wherein the phase ψ is different from tan ⁻¹ (i _1q * / i _1d *) with respect to the dq axis. And the coordinates with the primary current I ₁ as a reference axis are γ−δ coordinates, the primary current I _1d *, i _1q * a first coordinate conversion unit for calculating a target value _{i 1 γ * (= I 1} ) and the phase ψ of gamma-axis component, lambda _2d * and the target value M * ratio between lambda _2d of the excitation inductance * / M * , gamma command value omega ₀ based on the primary voltage of the operational results and power supply angular frequency of the first coordinate conversion unit, the target value v ₁ gamma * of [delta] -axis component, v ₁ [delta] * respectively below
Calculates the expression and converts the excitation inductance to the obtained value
And the ideal voltage calculation unit for calculating by multiplying the data, and the second coordinate conversion unit for converting the detection value of the primary current of the induction motor gamma-[delta] each axis component i ₁ gamma coordinates, to i ₁ [delta], i ₁ Based on γ *, the target value i ₁ δ * of the δ-axis component of the primary current, and i ₁ γ, i ₁ δ from the second coordinate conversion unit, v ₁ γ in the γ-axis component of the current primary voltage * Variation Δv ₁ γ and v ₁ δ * in the δ-axis component of the current primary voltage
And a means for calculating a variation Δv ₁ δ from the following _equation : i _1d *, i _1q *, i ₁ γ *, λ _2d *, primary power supply angular frequency ω ₀ , target value M * of excitation inductance, and Δv ₁ δ based provided a secondary resistance variation calculator for calculating a change in the target value of the secondary resistance, v ₁ γ * and Delta] v ₁ target value of gamma-axis component of the primary voltage sum of the gamma v ₁ gamma The sum of v ₁ δ * and Δ ₁ δ is set as a target value v ₁ δ of the δ-axis component of the primary voltage, and these target values v
_The power supply voltage is controlled based on 1γ, v ₁ δ, and the slip angular frequency calculation unit performs the control based on the set value of the secondary time constant and the calculation result obtained by the secondary resistance change calculation unit. The second order time constant is obtained, and this second order time constant, i
_1q * , _λ2d * / M * and excitation inductance conversion data
Is used to calculate the slip angular frequency to calculate the slip angular frequency.
The target value ω _s * is output , and Δv ₁ γ, the set value R ₁ * of the primary resistance and i
Calculate the change from the primary resistance set value based on _1d *
Thus, the primary resistance is identified , and Δv
₁ δ, M *, secondary self inductance L ₂ *, ω ₀ and λ
Calculate the change of the excitation inductance with respect to the set value based on _2d *, and identify the excitation inductance using this
An identification circuit unit to be provided, the change from the set value of the excitation inductance calculated by the identification circuit unit is output from the identification circuit unit as the excitation inductance change at a point divided by an arbitrary number of divisions of the entire operation range, A data table for the excitation inductance change is created, and an excitation inductance conversion data section obtained by linearly interpolating the data of the data table is provided.The output of the excitation inductance conversion data section is slipped.
Angular frequency calculator, ideal voltage calculator and excitation command λ _2d0 *
/ M *, a vector control device for an induction motor. [Equation 39] 2. The vector control apparatus for an induction motor according to claim 1, wherein a change Δv ₁ δ from v ₁ δ * in the δ-axis component of the current primary voltage is used instead of using the secondary resistance change calculation unit. inputs the deviation between the target value zero Delta] v ₁ [delta] of Toko, a voltage change control amplifier for outputting a variation [Delta] [omega _s from the target value omega _s * slip angular frequency provided from the voltage change control amplifier A vector control apparatus for an induction motor, characterized in that an addition value of Δω _{s of the above} and ω _s * obtained by the slip angular frequency calculation unit is set as a target value of the slip angular frequency. 3. A d-axis coordinate of a primary current of an induction motor, assuming that d-q coordinates are rotation coordinates that rotate in synchronization with a power supply angular frequency of the induction motor and have a secondary magnetic flux as a reference axis. Induction motor having means for calculating target values i _1d * and i _1q * of the q-axis component, respectively, and a slip angular frequency calculator for calculating a slip angular frequency based on an arithmetic expression including a set value of a secondary time constant Means for calculating i _1d * comprises: means for outputting a target value λ _2d * / M * of the d-axis component of the secondary magnetic flux in accordance with the rotor angular frequency of the induction motor; and λ _2d * / M * and means for calculating i _1d * based on the differential term, wherein the phase ψ is different from tan ⁻¹ (i _1q * / i _1d *) with respect to the dq axis. And the coordinates with the primary current I ₁ as a reference axis are γ−δ coordinates, the primary current I _1d *, i _1q * a first coordinate conversion unit for calculating a target value i ₁ γ * (= I ₁ ) of the γ-axis component and the phase ψ, a ratio λ _2d * / M * of λ _2d * and an exciting inductance target value M *, first gamma of the primary voltage based on the command value omega ₀ of the result and power supply angular frequency of the coordinate conversion unit, a target value of [delta] -axis component _{v 1 γ *, v 1 δ} * , respectively below Starring
Calculate the formula and convert the obtained value to the excitation inductance conversion data.
An ideal voltage calculation unit that calculates by multiplying the primary current by a motor; a second coordinate conversion unit that converts the detected value of the primary current of the induction motor into each axis component i ₁ γ, i ₁ δ of the γ-δ coordinate; a proportional term output the product of the target value i ₁ [delta] * and the second temporal change rate calculated by this and leakage inductance Lσ current deviation between i ₁ [delta] than the coordinate transformation unit of [delta]-axis component of the current And an integral element that outputs the integral value of the current deviation as an integral term output, and calculates the sum of the proportional term output and the integral term output as v ₁ δ in the δ-axis component of the current primary voltage.
* Outputs a voltage change Delta] v ₁ [delta] from, based the integral term output a current control amplifier to output as Delta] v ₁ [delta] _I, the i ₁ gamma * and i ₁ than the second coordinate conversion section gamma Means for calculating the variation Δv ₁ γ from v ₁ γ * in the γ-axis component of the current primary voltage; i _1d *, i _1q *, i ₁ γ *, λ _2d *, and the primary power supply angular frequency ω _0, the target value M * and Delta] v ₁ [delta] secondary resistance variation calculator for calculating a change in the target value of the secondary resistance based on the _I of the exciting inductance provided, v ₁ gamma * and Delta] v ₁ and gamma The added value is a target value v ₁ γ of the γ-axis component of the primary voltage, and the added value of v ₁ δ * and Δ ₁ δ is a target value v ₁ δ of the δ-axis component of the primary voltage.
_The power supply voltage is controlled based on 1γ, v ₁ δ, and the slip angular frequency calculation unit performs the control based on the set value of the secondary time constant and the calculation result obtained by the secondary resistance change calculation unit. The second order time constant is obtained, and this second order time constant, i
_1q * , _λ2d * / M * and excitation inductance conversion data
Is used to calculate the slip angular frequency to calculate the slip angular frequency.
The output target value omega _s *, means for calculating a Delta] v ₁ gamma is calculated temporal change rate of the current deviation of the i ₁ gamma than the i ₁ gamma * and the second coordinate conversion unit, which And a leakage inductance Lσ, including a proportional element that outputs a product of the proportional term, and an integral element that outputs a value obtained by integrating the current deviation as an integral term output. The sum of the proportional term output and the integral term output is represented by: A current control amplifier that outputs the voltage variation Δv ₁ γ from v ₁ γ * in the γ-axis component of the current primary voltage and outputs the integral term output as Δv ₁ γ _I , and outputs Δv ₁ during no-load operation. ₁ γ _I , a change in the primary resistance set value is calculated based on the primary resistance set value R ₁ * and i _1d * , whereby the primary resistance is identified and Δv ₁
δ _I , M *, secondary self inductance L ₂ *, ω ₀ and λ
Calculate the change of the excitation inductance with respect to the set value based on _2d *, and identify the excitation inductance using this
An identification circuit unit to be provided, the change from the set value of the excitation inductance calculated by the identification circuit unit is output from the identification circuit unit as the excitation inductance change at a point divided by an arbitrary number of divisions of the entire operation range, A data table for the excitation inductance change is created, and an excitation inductance conversion data section obtained by linearly interpolating the data of the data table is provided.The output of the excitation inductance conversion data section is slipped.
Angular frequency calculator, ideal voltage calculator and excitation command λ _2d0 *
/ M *, a vector control device for an induction motor. (Equation 40) 4. The vector control device for an induction motor according to claim 3, wherein the output of the integral term Δv ₁ δ _I of the current control amplifier and this Δv ₁
inputs the deviation between the target value zero [delta] _I, the voltage change control amplifier for outputting a variation [Delta] [omega _s from the target value omega _s * slip angular frequency provided, [Delta] [omega _s than the voltage change control amplifier A vector control device for an induction motor, characterized in that an added value of the slip angular frequency and a ω _s * obtained by a slip angular frequency calculation unit is set as a target value of the slip angular frequency. 5. A vector control apparatus for an induction motor according to claim 1, wherein, instead of using the secondary resistance change calculation section, Delta] v ₁ [delta] * variation from partial Delta] v ₁ [delta] in [delta] -axis component of the current of the primary voltage And a deviation of the Δv ₁ δ from the target value of zero is input, and a secondary resistance change amplifier for directly obtaining the secondary resistance change is provided as an output. The output of this amplifier and the target value ω _s * of the slip angular frequency and by multiplying the obtains the variation [Delta] [omega _s from the target value omega _s of the slip angular frequency, the Δ
vector control apparatus for an induction motor, characterized in that the target value of the sum of the slip angular frequency of omega _s * obtained in omega _s and slip angular frequency calculation unit. 6. The vector control apparatus for an induction motor according to claim 3, instead of using the secondary resistance change calculation section, Delta] v ₁ of the integral term output Delta] v ₁ [delta] _I Toko current control amplifier
inputs the deviation between the target value zero [delta] _I, the secondary resistance variation amplifier to obtain a secondary resistance variation directly as the output provided, multiplying the target value omega _s * of the slip angular frequency The amplifier output Thus, the variation Δω _s of the slip angular frequency from the target value ω _s is obtained, and the sum of Δω _s and ω _s * obtained by the slip angular frequency calculator is set as the target value of the slip angular frequency. Characteristic vector control device for induction motor. 7. The vector control device for an induction motor according to claim 2, wherein the gain of the voltage fluctuation control amplifier is changed in inverse proportion to the primary angular frequency, and an upper and lower limiter is applied to the gain. Vector control device for induction motor. 8. The vector control device for an induction motor according to claim 5, wherein the gain of the secondary resistance change amplifier is changed in inverse proportion to the primary angular frequency, and the gain is subjected to upper and lower limiters. Vector control device for induction motor. 9. The vector control device for an induction motor according to claim 3, wherein when the integral term output of the current control amplifier is given to the secondary resistance change calculating section, the integral term output is given via a first-order lag filter. A vector control device for an induction motor, further comprising upper and lower limiters that are variable in a filter time constant. 10. The vector control device for an induction motor according to claim 1, wherein a change with respect to a set value of the secondary resistance is calculated by a secondary resistance change calculator based on the change Δv ₁ δ. The calculated secondary resistance change data is held as an initial value by the hold unit, and when the normal operation is started, the data of the hold unit is given to the output terminal of the secondary resistance change calculation unit as the initial data. Vector control device for induction motor. 11. The vector control device for an induction motor according to claim 5, wherein the variation Δv ₁ δ is input to a secondary resistance change amplifier, and an initial value of the secondary resistance change obtained at the amplifier output. A vector control device for an induction motor, characterized in that data is held in a holding unit and, when normal operation is started, the data in the holding unit is given as initial data to an output terminal of a secondary resistance change amplifier.