JP2018019382A - Radio communication device and radio communication method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To utilize implicit feedback even in the case of constructing a radio system using FDD.SOLUTION: A radio communication device is configured to: communicate with another radio communication device using a first frequency used in an uplink and a second frequency used in a downlink; calculate correlation for each of combinations between an antenna element to be a reference and other antenna elements, using a digital signal corresponding to training signals converted by signal conversion units included in all antenna elements or some systems thereof; on the basis of a calculation result, calculate a complex phase rotation amount; predict the complex phase rotation amount on the basis of information on a ratio between the first frequency and second frequency and calculated information; manage the complex phase rotation amount on the basis of a predicted complex phase; and grant the managed complex phase rotation amount to a transmission signal of each antenna element to be used for transmission on an analog signal or digital signal, before performing transmission to the other radio communication device via the antenna element.SELECTED DRAWING: Figure 22

Description

  The present invention relates to a wireless communication apparatus and a wireless communication method.

[Background of 5th generation mobile communications]
Currently, high-functional mobile communication terminals such as smartphones are explosively spreading. With regard to mobile phones, the third generation mobile communication has shifted to the fourth generation mobile communication, and research and development related to the fifth generation mobile communication (commonly referred to as “5G”) is now underway. One of the studies that have been conducted on 5G is the use of macro cells and small cells.

  In conventional cellular phones, one service area is set to a radius of several kilometers, and this macro cell area is covered by one base station device. However, there are a very large number of users in such a macro cell. Since the limited system capacity as a whole is shared by each user, when accommodating a huge number of users, the throughput for each individual user is lowered.

  In order to avoid such a decrease in throughput, a technology for setting a small cell, which is a very small service area with a radius of several tens of meters, in a densely populated area where traffic is concentrated has been developed. With this technology, spot traffic is offloaded to the network without using a macro cell by utilizing a small cell. Here, it is assumed that the terminal device can use the communication capability in the small cell and the communication capability in the macro cell simultaneously. By using such a terminal device, user data is accommodated on the small cell side while exchanging information using the macro cell for control information. This makes it possible to make maximum use of the advantages of macro cells and small cells.

  In 5G mentioned above, the target value of transmission speed is set to 10 Gbit / s (gigabit per second) or more, and this small cell can perform efficient traffic offload by performing the same large-capacity communication. There is a need to. In a macro cell, it is assumed that a microwave band with a low frequency is used in order to allow long-distance propagation. However, considering the current state of the microwave band, where frequency resources are already being depleted, small cells that assume relatively short-distance communication are expected to use the quasi-millimeter wave band or the millimeter wave band with a relatively high frequency. Has been. The characteristic of this high frequency band is that propagation attenuation increases in inverse proportion to the square of the frequency. Therefore, it is preferable that the small cell base station is ideally installed near the user terminal. For example, in a place where installation is easy, such as on the roof of a building, the distance between the user terminal and the base station is too far away, which is not preferable in terms of circuit design.

  On the other hand, since the small cell is set in a place where traffic is concentrated, there is a case where it is strongly desired to install a base station apparatus even in a place where it is difficult to lay an optical fiber. For example, assuming that a small cell base station device is installed in a place where people are very crowded, such as in front of a station such as Shinjuku or Shibuya, propagation attenuation increases on the rooftop of a building adjacent to such a location. . For this reason, it may be required to be installed on a place lower in height than the rooftop of the building, such as a wall surface of the building. However, it may be difficult to install optical fiber on the wall of an existing building. In such a case, it may be necessary to provide a backhaul line to the base station apparatus using a radio line. is there.

  When providing such a backhaul line, it is necessary to use the millimeter wave band to perform a large capacity transmission of 10 Gbit / s or more in order to cope with a large capacity transmission of 10 Gbit / s or more required in a small cell. There is. In such an environment, since both opposing radio station apparatuses are fixedly installed in stable locations, it is natural that the line-of-sight is secured stably and the directional antennas are directed to each other. is there. In this case, reflected waves between buildings exist to some extent, but most of the received signals are line-of-sight components, and it is expected that it is difficult to say a multipath environment. This situation is the same for the access system if the small cell base station device is installed at a high place such as a building wall surface and looks down from the top to the bottom of the user and is generally used in a line-of-sight environment.

  Next, for large-capacity transmission of 10 Gbit / s or higher, which is the transmission rate required for 5G, it becomes possible to use frequency resources with a very wide bandwidth by utilizing the millimeter wave band, and this is feasible. Is growing. For example, if a backhaul line using the millimeter wave band is assumed, E band (71 to 76 GHz and 81 to 86 GHz) is used as an example, and if a bandwidth of 1 GHz is used, the frequency utilization efficiency is 10 bits. / S / Hz is sufficient. However, existing wireless equipment for achieving a frequency utilization efficiency of 10 bits / s / Hz generally employs spatial multiplexing transmission using a multiple-input multiple-output (MIMO) channel. Spatial multiplex transmission generally uses a multipath environment, and when the singular value decomposition of the channel matrix H expressing the transfer function of the MIMO channel in a matrix form is performed, the distribution of absolute values of the singular values obtained as a result thereof Represents the characteristics of the spatial multiplexing transmission. Specifically, the square value of the absolute value of the singular value is a value proportional to the signal-to-noise power ratio SNR (Signal to Noise Ratio), and not only the first singular value but also the first singular value for spatial multiplexing transmission. Communication is not possible if there is no sufficiently large value after the 2 singular value. The same is true for large-capacity transmission in small cells that are access systems, but it is indispensable to realize spatial multiplexing transmission in an environment where such line-of-sight waves are dominant. .

  As described above, in 5G, the use of the millimeter wave band is expected for both the access system and the backhaul line. As described above, the characteristic of the high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. For example, if the 2 GHz band (existing access system) and the 80 GHz band (future system using the millimeter wave band) are compared to 40 times the frequency, the propagation attenuation is 1600 times and the line gain of 32 dB is insufficient. It will be. Of course, it is not necessary to cover as much as 32 dB because it does not need to cover as wide a range as a macro cell. However, in the high frequency band, high power amplifiers in the transmission stage do not have very high output devices. In this case, it is considered that an additional line gain at a level of several tens of dB must be secured. Furthermore, since spatial multiplexing transmission is also expected in such an environment, both base stations and terminal stations are now considering wireless systems with significantly more antenna elements than conventional techniques. . Such a technique is called Massive MIMO. Below, the prior art regarding Massive MIMO is introduced.

  In Non-Patent Document 1, the base station side is equipped with 256 elements and the terminal station side is equipped with 16 elements, and aims at 16 streams of spatial multiplexing transmission utilizing a 256 × 16 channel matrix. In this non-patent document 1, in order to transmit a plurality of signal sequences (streams), the directivity is formed by analog beam forming using rotation of a complex phase amount in a wireless analog circuit and in a digital baseband circuit. Performed in combination with digital beam forming in the digital domain. As a result, avoiding heavy use of analog / digital (A / D) converters and digital / analog (D / A) converters, power consumption is reduced and channel gain shortage countermeasures are taken when channel information is fed back. Hereinafter, an outline of the technique disclosed in Non-Patent Document 1 will be described.

FIG. 27 is a functional block diagram showing a configuration example of a radio station apparatus in the prior art. The radio station apparatus shown in the figure includes modulators 901-1 to 901-N (MOD # 1 to MOD # N), a precoder 902, IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval). Giving circuits 903-1 to 903-N ₀ , D / A converters 904-1 to 904-N ₀ , up converters (UC) 905-1 to 905-N ₀ , and down converters (DC) 906-1 906-N ₀ , A / D converters 907-1 to 907-N ₀ , GI removal & FFT (Fast Fourier Transform) circuits 908-1 to 908-N ₀ , a post coder 909, Demodulators 910-1 to 910 -N (DEM # 1 to DEM # N), TDD switch (TDD-SW) 911, distribution coupler (HYB) 912-1 to 912 -N ₀ , and phase shifter 913 -1 Comprises a _{-1~913-N 0} -M 0, and distributor coupler (HYB) _{915-1~915-M 0,} and antenna element _{916-1~916-M 0.} Here, N corresponds to the number of multiplexed (number of streams) when performing spatial multiplexing, N ₀ represents the number of signal systems for performing signal processing for digital directivity formation, and M ₀ represents the number of antenna elements. (M ₀ ≧ N ₀ ≧ N). Further, the antenna elements 916-1 to 916 -M ₀ constitute an array antenna as a whole.

The distribution couplers 912-1 to 912 -N ₀ to the antenna elements 916-1 to 916 -M ₀ are common for transmission and reception. Further, the TDD switch 911 increases the connection from the distribution couplers 912-1 to 912-N ₀ to the antenna elements 916-1 to 916-M ₀ from the modulators 901-1 to 901-N corresponding to the transmission system. Switching is performed between converters 905-1 to 905-N ₀ and down converters 906-1 to 906-N ₀ corresponding to the reception system to demodulators 910-1 to 910-N. For example, the up converter 905-n and the distribution coupler 912-n are connected during transmission, and the down converter 906-n and the distribution coupler 912-n are connected during reception (n = 1,..., N ₀ ). An overall control circuit (not shown) manages the frame period and transmission / reception timing, and the TDD switch 911 is also switched by this control circuit.

The phase shifters 913-1-1 to 913 -N ₀ -M ₀ adjust the phase relationship of transmission and reception signals according to a predetermined beam pattern, and this phase rotation amount is controlled by a control circuit (not shown). Are also managed. The phase here is the same as the directivity control in the phased array antenna. For example, as directional gain in a predetermined direction across the antenna elements _{916-1~916-M 0} is the maximum, at each antenna element _{916-1~916-M 0} against incoming waves from that direction The complex phase corresponding to the value obtained by dividing the path length difference by the wavelength is adjusted. This enables each antenna element 916-1 to 916 -M ₀ to transmit and receive signals in the same phase.

The directivity here is obtained by dividing the horizontal azimuth angle θ and the vertical azimuth angle φ by a step size of a predetermined angle, and for each selectable (θ _i , φ _j ) menu, the corresponding complex The phase amount of the phase shifters 913-n-1 to 913-n-M ₀ (n = 1,..., N ₀ ) is adjusted with the set of phases as a set. As a result, for example, the antenna element 916-1 to 916 -M ₀ , the phase shifters 913-n-1 to 913 -n-M ₀ , and one hypothesis for the n-th signal sequence in the entire distribution coupler 912-n. Behaves as a directional antenna. These virtual directional antenna is physically via distribution couplers _{915-1~915-M 0,} will share the antenna element _{916-1~916-M 0.}

  Furthermore, in the following description, a case where communication is performed by forming a signal on the frequency axis as in an OFDM (Orthogonal Frequency Division Multiplexing) modulation method will be described as an example. Even in the case of single carrier transmission, when equalization processing on the frequency axis is performed, the signal is once converted to a frequency axis signal. Similar processing is possible regardless of whether OFDM or single carrier transmission is used.

The specific signal flow is as follows. First, signal transmission will be described. Modulators 901-1 to 901 -N each generate a transmission signal for each stream for spatial multiplexing. The precoder 902 appropriately combines signals between a plurality of virtual directional antennas so that signal separation can be efficiently performed on the receiving station side. This precoding processing corresponds to multiplication of a unitary transformation matrix when singular value decomposition is performed on, for example, a MIMO channel matrix between N ₀ virtual directional antennas and N signal systems actually transmitted and received. Thus, so-called eigenmode transmission is realized, and efficient transmission is realized. The IFFT & GI adding circuits 903-1 to 903 -N ₀ convert the transmission signal sequence formed in this way from a signal on the frequency axis to a signal on the time axis, and add a guard interval. If necessary, waveform shaping between symbols is also performed here. The D / A converters 904-1 to 904 -N ₀ convert the digital signals generated in this way into analog signals. Upconverter _{905-1~905-N 0} is the analog signal, converts the baseband signal to the radio frequency signals.

At the time of transmission, the TDD switch 911 connects the up converter 905-n and the distribution coupler 912-n (n = 1,..., N ₀ ). It should be noted, 1~N ₀ of subscript is all behave in the same way. Distribution coupler 912-n (n = 1,..., N ₀ ) distributes radio frequency band signals to signals of only the number of antenna systems M ₀ , and phase-shifters 913-n-1 to 913-n. input to -M _0. For example, the phase shifters 913-1-1 to 913-1 -M ₀ adjust the predetermined complex phase corresponding to the directivity direction (θ _i , φ _j ) corresponding to the first signal sequence on the analog signal. in is performed to transmit a first signal sequence after the adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} Similarly the phase shifter _{913-N 0 -1~913-N 0} -M 0 , the first _{N 0} directivity of the azimuth corresponding to the signal sequence of _{(θ i ', φ j'} ) predetermined complex phase corresponding to the the adjustments performed on the analog signal and transmits the signal sequence of the N after adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} The distribution couplers 915-m (m = 1,..., M ₀ ) receive the signals input from the corresponding phase shifters 913-1-m, 913-2-m, ..., 913-N ₀ -m. Combined and output to the antenna element 916-m.

Next, signal reception will be described. The signals received by the antenna elements 916-1 to 916 -M ₀ are distributed to the N ₀ system signals by the distribution couplers 915-1 to 915 -M ₀ , respectively, and the corresponding phase shifters 913-1-1. It is output to the _{~913-N 0} -M 0.
For example, the phase shifters 913-1-1 to 913-1 -M ₀ adjust the predetermined complex phase corresponding to the directivity direction (θ _i , φ _j ) corresponding to the first signal sequence on the analog signal. The first signal sequence after adjustment is input to the distribution coupler 912-1. The distribution coupler 912-1 combines these input signals, and inputs the combined signal to the down converter 906-1 via the TDD switch 911.
Similarly, the phase shifter _{913-N 0 -1~913-N 0} -M 0 , the directivity of the azimuth corresponding to the signal sequence of the _{N (θ i ', φ j} ') predetermined complex phase corresponding to the the adjustments performed on the analog signal, and inputs the N-th signal sequence after the adjustment to the distributor coupler 912-N _0. Distribution coupler 912-N ₀ synthesizes these inputted signals and inputs the synthesized signal to down converter 906-N ₀ via TDD switch 911.

The down converters 906-1 to 906-N ₀ down-convert radio frequency signals into baseband signals. The A / D converters 907-1 to 907 -N ₀ convert analog baseband signals obtained by down-conversion into digital baseband signals. Here, based on symbol timing managed by a timing detection circuit (not shown), the GI removal & FFT circuits 908-1 to 908 -N ₀ remove the guard interval from the digital baseband signal, Convert to frequency axis signal. Postcoder 909, the cross talk component between the signal sequence (stream) which has been processed by the GI removal & FFT circuit _{908-1~908-N 0} and signal separation in the frequency domain, corresponding signals after the crosstalk component separation Output to the demodulators 910-1 to 910 -N. Demodulators 910-1 to 910 -N reproduce and output data by a predetermined signal detection process.

Note that the power amplifier on the transmission side and the low noise amplifier on the reception side are not explicitly described here, but in general, the subsequent stage of the up converters 905-1 to 905-N ₀ (references “A ₁ ” to “A _N0 ”). ) And a low noise amplifier are installed in front of the down converters 906-1 to 906 -N ₀ (positions “B ₁ ” to “B _N0 ”). The power amplifier and the low noise amplifier have different complex phase rotation amounts, and there are cases where the transfer rotation amount differs for each frequency. However, a factor causing a difference in phase rotation amount between transmission and reception is eliminated between the TDD switch 911 and the antenna elements 916-1 to 916 -M ₀ , and the symmetry of the channel between transmission and reception is eliminated. Saved. For this reason, the same value can be used for setting the phase rotation amount of the phase shifters 913-1-1 to 913 -N ₀ -M ₀ at the time of transmission and at the time of reception.

The above is an outline of the Massive MIMO technology using hybrid beamforming. Here, the phase rotation amount set by the phase shifters 913-1-1 to 913 -N ₀ -M ₀ or the directionality of the directivity corresponding to each signal series (in the above example, the phase shifters 913-1-1-1). in _{913-1-M 0 (θ i,} φ j), the phase shifter _{913-N 0 -1~913-N 0} -M 0 (θ i ', φ j')) acquisition method, such as the present invention Although it is omitted because it is not directly related to the above feature, it can be obtained by conventional techniques such as Non-Patent Document 1.

[Expansion of Massive MIMO technology when line-of-sight wave is dominant]
In the above description of the Massive MIMO technology, since it is mainly assumed to be used in an access system, it is assumed that the environment is generally a multipath environment. However, even in the case of an access system, if a small cell base station is installed on the upper side and it is possible to secure a general outlook while looking down, operation in a non-multipath environment may be forced. In the case of a backhaul line in particular, this is conspicuous, and it is assumed that the so-called Rice coefficient K is 10 dB or more, and it is assumed to be used in an environment where only about 1/10 or less of the line-of-sight component is accompanied by multipath components. In this case, it is expected that the difference between the line gain corresponding to the first singular value and the line gain corresponding to the second singular value or more will be 20 dB or more. It is expected that

  In such an environment, as shown in Non-Patent Document 2, using the high efficiency of the line gain corresponding to the first singular value, all antenna elements are divided into a plurality of sets, and subarrays are set for each set. It is effective to take a configuration. Then, by installing the subarrays spatially separated, it is effective to reduce the correlation between the subarrays and to transmit the transmission corresponding to the first singular value in parallel with low correlation. Similarly, in Non-Patent Document 3, when the antenna aperture length of the subarray is narrow, the line-of-sight wave is dominant, and the correlation between the antenna elements in the subarray is sufficiently strong, the transmission / reception weight of each antenna element is the frequency. A “time-axis beamforming technique” has been proposed in which weights can be multiplied in units of sampling data on the time axis, which can be handled as constants having no dependency.

This is the relative channel information for each antenna element when the element spacing is narrow and the correlation between the antenna elements is strong (the relative value of the channel information with respect to the channel information at a certain antenna element, specifically the reference antenna When the complex phase of the channel information of the k-th frequency component of the element is ψ _ref ^(k) , the frequency dependence of the complex phase of Exp {−jψ _ref ^(k) } multiplied by each antenna element) The nature is due to being almost constant. For example, at the time of reception, the complex phase of the reception weight for synthesizing the reception signals of these antenna elements so that the complex phase is the same phase is generally constant in the entire frequency band. Thus, the same constant reception weight can be used. In general, when a function that is constant on the frequency axis is Fourier transformed to a δ function, the weight obtained by converting the reception weight on the frequency axis onto the time axis by IFFT only needs to consider the component at t = 0. Become. In other words, since signal processing in consideration of the delayed wave component is unnecessary, the time axis that is a predetermined coefficient for each antenna element is directly added to the sampling data obtained by sampling the analog baseband received signal by the A / D converter. By multiplying the reception weight, it is possible to realize directivity formation with only time-axis signal processing without converting the received signal into a signal on the frequency axis by FFT processing or the like.

  A coefficient for rotating the complex phase multiplied as a time axis weight is obtained by the following equations (1) to (3).

In the above equation, S _i (n) represents the sampling data of the n th sample of the i th antenna in the received training signal, and S _i (n) ^* represents the complex conjugate of S _i (n). Represent. N _FFT assumes a predetermined periodicity, and indicates a periodicity value that is meaningful in correlation detection, such as the number of FFT points of OFDM. ψ _j is the amount of rotation of the complex phase (on the receiving side) performed by time axis beamforming. The function angle (x) is a function representing the complex phase of the complex number x, and is a value determined by the ratio of the real part of x to the imaginary part of x and the sign of the real part and imaginary part. Although the in the correlation calculation in the formula (1) performs a correlation operation over N _FFT samples is the number of FFT points in the case of the OFDM signal as a predetermined periodicity, it is an integer multiple of example N _FFT Correlation calculation over other sample numbers may be performed so that the periodicity is maintained.

Here, as is clear from the equation (2), the sign of the complex phase of the complex coefficient c _j given by the above equation (1) and the complex phase of the time axis weight w _j given by the above equation (2) are inverted. It has become a thing. In this sense, in the embodiment of the present invention described later, the complex phase of the complex coefficient c _j given by the expression (1) corresponding to the relative channel information is obtained, and the complex phase of the time axis weight w _j is obtained. That is equivalent.

FIG. 28 is a functional block diagram showing a configuration example (subarray separation type) of a radio station apparatus using time-axis beamforming in the prior art described in Non-Patent Document 3. The radio station apparatus shown in the figure includes a baseband signal processing circuit 140 and transmission / reception signal processing circuits 929-1 to 929 -N. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 929-n (n = 1,..., N) includes a time-axis transmission weight multiplication circuit 921-n, D / A converters 922-n-1 to 922-n-M, and an up converter 923. -N-1 to 923-nM, down converter (DC) 924-n-1 to 924-nM, A / D converters 925-n-1 to 925-nM, time axis A reception weight multiplication circuit 926-n and a TDD switch (TDD-SW) 927-n are provided. The TDD switch 927-n is connected to the antenna elements 928-n-1 to 928-n-M. Here, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the transmission / reception signal processing circuits 929-1 to 929 -N are mounted for N systems as a whole. M represents the number of antenna elements of the subarray mounted on each of the transmission / reception signal processing circuits 929-1 to 929 -N. In the description of FIG. 27, the total number of antenna elements is M ₀ , but here, since the entire antenna has a subarray configuration, the number of antenna elements in each subarray is indicated as a different value M.

  Here, the transmission / reception signal processing circuits 929-1 to 929 -N (the transmission / reception signal processing circuit 929 -n is accompanied by subarray antenna elements 928 -n-1 to 928 -n-M) are described in Non-Patent Document 2. It is assumed that they will be installed spatially apart like The baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 929-1 to 929 -N by wire, and a digital baseband signal is transferred over this wire. Similarly to the case of FIG. 27, an entire control circuit (not shown) is mounted on the baseband signal processing circuit 140 to manage the frame period and transmission / reception timing. Here, the TDD switches 927-1 to 927-1 are used. The switching of 927-N is also managed here.

Furthermore, the time-axis beamforming technique basically assumes signal processing on the time axis, but even when signals on the frequency axis are formed as in the OFDM modulation method, signals on the frequency axis are processed by FFT processing and IFFT processing. Can be converted into a signal on the time axis, and by performing signal processing on this time axis signal, the time axis beam forming technique can be similarly applied to the OFDM modulation method as well as the single carrier transmission. However, in FIG. 27, assuming signal processing on the frequency axis, IFFT & GI adding circuits 903-1 to 903-N ₀ for IFFT processing and GI removal & FFT circuits 908-1 to 908-N ₀ for FFT processing Is shown separately from the modulators 901-1 to 901 -N and the demodulators 910-1 to 910 -N, but here, since signal processing on the frequency axis is not assumed, FIG. Even when the OFDM modulation method or the like is used, the functions of FFT processing and IFFT processing are included in the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N (or the signal separation circuit 141). These notations are omitted. Therefore, the input / output signals from the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are signals on the time axis regardless of the OFDM modulation scheme or single carrier transmission. . Further, the signal separation circuit 141 performs signal separation between each signal series, but here it is also possible to perform signal separation on the time axis, and once convert it to a frequency axis signal by FFT, Frequency-dependent signal separation may be performed. In this sense, the input to the demodulators 130-1 to 130-N is assumed to be a time-axis signal or a frequency-axis signal. Will be described as input.

  The specific signal flow is as follows. First, signal transmission will be described. Modulators 120-1 to 120 -N generate time base digital baseband transmission signals of the respective streams to be spatially multiplexed, and input them to time base transmission weight multiplication circuits 921-1 to 921 -N, respectively. To do. The time-axis transmission weight multiplication circuit 921-n (n = 1,..., N) each of the subarrays for forming the directivity of the digital signal input from the modulator 120-n by the transmission / reception signal processing circuit 929-n. The signal is converted into a digital signal multiplied by a transmission weight corresponding to the antenna elements 928-n-1 to 928-n-M. The D / A converters 922-n-1 to 922-n-M convert the digital signal multiplied by the transmission weight into an analog baseband signal, and the up-converters 923-n-1 to 923-n-M The analog baseband signal is converted into a radio frequency band signal. At the time of transmission, the TDD switch 927-n connects the up-converter 923-nm (m is an integer of 1 to M) to the antenna element 928-nm. Each antenna element 928-n-1 to 928-n-M transmits the respective signals input from the up-converters 923-n-1 to 923-n-M, and the transmission / reception signal processing circuits 929-1 to 929 are transmitted. A directional beam is formed every -N.

  Next, signal reception will be described. Signals received by the antenna elements 928-n-1 to 928-n-M (n = 1,..., N) are input to the down converters 924-n-1 to 924-n-M via the TDD switch 927-n. Is done. The down converters 924-n-1 to 924-n-M convert radio frequency signals into analog baseband signals. The A / D converters 925-n-1 to 925-n-M convert analog baseband signals into digital baseband signals. This digital baseband signal is input to the time axis reception weight multiplication circuit 926-n. The time axis reception weight multiplication circuit 926-n multiplies each input signal by a reception weight corresponding to each antenna element 928-n-1 to 928-n-M, and adds and combines the signals after reception weight multiplication. Thus, each is converted into one signal series. That is, a total of N signal series (streams) are converted by the time axis reception weight multiplication circuits 926-1 to 926 -N, and these signals are input to the signal separation circuit 141. The signal separation circuit 141 performs signal separation by suppressing crosstalk components between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. Demodulators 130-1 to 130-N reproduce and output data by a predetermined signal detection process.

  The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be performed once on the frequency axis after being converted into a frequency axis signal by FFT processing. Alternatively, only the signal processing performed by the transmission / reception signal processing circuits 929-1 to 929 -N is required, and the signal separation circuit 141 does not need to perform any processing. However, in any case, details of the signal separation method here are omitted because they are not directly related to the present application.

  The time axis transmission weights used in the time axis transmission weight multiplication circuits 921-1 to 921-N and the time axis reception weights used in the time axis reception weight multiplication circuits 926-1 to 926-N are shown here. No time axis transmission / reception weight acquisition means acquires. Similarly, a control circuit not shown here manages the value of the time axis transmission / reception weight used there. For example, a reference antenna element over a predetermined number of samples based on sampling data acquired by the A / D converters 925-n-1 to 925-n-M with respect to a training signal transmitted by a wireless station as a communication partner A correlation value between antenna elements (for example, 928-n-1) may be obtained and determined based on this complex phase. The values of the complex phase of the time axis reception weight and the time axis transmission weight generally do not match because the rotation amount of the complex phase such as a power amplifier and a low noise amplifier not shown here is different for each amplifier. By using the implicit feedback calibration method, conversion from the time axis reception weight to the time axis transmission weight is possible. The transmission / reception weight acquired in this way is stored in the memory for each corresponding radio station apparatus. At the time of transmission and reception, the time-axis transmission weight multiplication circuits 921-1 to 921-N and the time-axis reception weight multiplication circuits 926-1 to 926-N perform weight multiplication based on these transmission / reception weight values. It will be.

  FIG. 29 is a functional block diagram showing a configuration example (subarray shared type) of a radio station apparatus using time-axis beamforming in the prior art described in Non-Patent Document 3. In the figure, the same figure number is given to the function common to FIG. In the figure, a radio station apparatus 942 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 929-1 to 929-N, distribution couplers (HYB) 941-1 to 941-M, and an antenna element 928-. 1-928-M. In FIG. 28, the transmission / reception signal processing circuits 929-1 to 929 -N are accommodated in different housings assuming that they are installed in spatially separated locations for each sub-array, and baseband signal processing circuits in separate housings. A configuration in which a wired connection is made with the network 140 is shown. On the other hand, in FIG. 29, all the transmission / reception signal processing circuits 929-1 to 929 -N and the baseband signal processing circuit 140 are configured as a radio station apparatus 942 in the same casing, and the antenna elements 928-1 to 928 -M are entirely configured. Shared by. For this reason, for example, at the time of transmission, signals from the TDD switches 927-1 to 927 -N of the transmission / reception signal processing circuits 929-1 to 929 -N are synthesized by the distribution couplers 941-1 to 941 -M and synthesized. The transmitted signals are transmitted from the antenna elements 928-1 to 928-M. Similarly, at the time of reception, the signals received by the antenna elements 928-1 to 928-M are distributed by the distribution couplers 941-1 to 941-M. That is, the distribution coupler 941-m (m = 1,..., M) distributes and inputs the signal received by the antenna element 928-m to the TDD switches 927-1 to 927-N. All other signal processing is the same in FIGS.

  Similarly, FIG. 30 is a functional block diagram of another configuration example (subarray shared type) of a radio station apparatus using time-axis beamforming in the prior art described in Non-Patent Document 3. In this figure, the same parts as those in the radio station apparatus shown in FIG. 29 are denoted by the same reference numerals, and the description thereof is omitted. The radio station apparatus 945 shown in the figure includes a baseband signal processing circuit 140, time axis transmission weight multiplication circuits 921-1 to 921-N, addition synthesizers 943-1 to 943-M, and a D / A converter. 922-1 to 922-M, up-converters 923-1 to 923-M, down-converters 924-1 to 924-M, A / D converters 925-1 to 925-M, and duplicator 944-1 944-M, time axis reception weight multiplication circuits 926-1 to 926-N, a TDD switch 927, and antenna elements 928-1 to 928-M.

  In FIG. 29, the transmission / reception signal processing circuits 929-1 to 929 -N are individually mounted for each subarray, but the D / A converters 922-n-1 to 922 -n-M, the up-converters 923-n-1 to 923-nM, down converters 924-n-1 to 924-nM, A / D converters 925-n-1 to 925-nM, and TDD switches 927-1 to 927-N are shared. Possible (n = 1,..., N). Therefore, in FIG. 30, N digital baseband signals generated by the time-axis transmission weight multiplication circuits 921-1 to 921-N (where N planes are mounted as a whole) are added and synthesizer 943- 1 to 943-M are added and synthesized, and each of them is integrated into one system, and converted from a digital signal to an analog signal by a D / A converter 922-1 to 922-M. Similarly, on the receiving side, the duplicators 944-1 to 944-M duplicate the digital baseband signals generated by the A / D converters 925-1 to 925 -M into N signals in units of sampling data, The duplicated signal is input to time axis reception weight multiplication circuits 926-1 to 926 -N (where N planes are mounted as a whole). Each of the time axis reception weight multiplication circuits 926-1 to 926 -N multiplies the input signal by the reception weight, and adds and synthesizes the result to convert each signal into one system signal. That is, a total of N signals are converted by the time axis reception weight multiplication circuits 926-1 to 926 -N, and these signals are input to the signal separation circuit 941.

  As a result, the D / A converters 922-n-1 to 922-n-M are duplicated, the up converters 923-n-1 to 923-n-M are duplicated, and the down converters 924-n-1 to 924- n-M overlap mounting, A / D converters 925-n-1 to 925-n-M, TDD switches 927-1 to 927-N are avoided, circuit scale reduction, power consumption, etc. This has led to a reduction in energy consumption.

  Here, in actual operation, the radio station apparatus shown in FIG. 28 and the radio station apparatus shown in FIG. 29 or FIG. For example, the base station apparatus has a degree of freedom of installation like a building roof, and subarrays can be installed at a plurality of locations. On the other hand, the terminal station device side has a configuration in which FIG. 28 is a base station device and FIG. 29 or FIG. It is possible to increase the degree of freedom of installation by sharing the antenna. Or, for example, if the transmission capacity per terminal station device is such that spatial multiplexing is not required, FIG. 29 or FIG. 30 is a base station device, and FIG. 28 is one of transmission / reception signal processing circuits 929-1 to 929 -N. And a configuration for performing multi-user MIMO transmission with a plurality of terminal station devices and one base station device, assuming that the terminal station device is implemented only with (for example, only the transmission / reception signal processing circuit 929-1 in FIG. 28). It is also possible to do.

Incidentally, Figures 27 and 28, with regard to the correspondence between FIGS. 29 and 30, for example a phase shifter _{913-1-1~913-N} rotation amount of complex phase carried out at 0 -M ₀ in FIG. 27 [psi _alpha ( α is equivalent to an identification number for phase shifters 913-1-1 to 913 -N ₀ -M ₀ ), time axis transmission weight multiplication circuits 921-1 to 921 -N, time axis reception weight multiplication circuit 926 −1 to 926-N corresponds to multiplying the signal sequence of the corresponding antenna element by Exp {jψ _α }. In other words, in FIG. 27, signals transmitted and received from each antenna element are converted by analog circuits (that is, phase shifters 913-1-1 to 913 -N ₀ -M ₀ ), whereas FIGS. In FIG. 30, the signal transmitted / received from each antenna element is converted by a digital circuit (that is, time axis transmission weight multiplication circuits 921-1 to 921-N, time axis reception weight multiplication circuits 926-1 to 926-N). To do. Accordingly, in FIG. 27, the number of A / D converters 907-1 to 907-N ₀ and the number of D / A converters 904-1 to 904-N ₀ can be suppressed, while FIG. 29 and 30, as described in Non-Patent Document 3, there are advantages that the resolution of directivity formation is increased and that simple and efficient channel information feedback is possible.

Note that the phase rotation by the phase shifter is usually performed by selectively passing a delay line corresponding to the amount of phase rotation on the device. For this reason, when phase rotation of x is given as an absolute value, the phase rotation (delay) of the complex phase rotation amount is negative with respect to the delay corresponding to the phase x as a signal, and the sign consistency cannot be obtained. . However, in the following description, for convenience, when the phase shift corresponding to the multiplication of the coefficient Exp {jψ _α } is given as a signal by the phase shifter, the amount of rotation of the complex phase performed by the phase shifter is referred to as ψ _α. To do.

[Calibration technology in channel information feedback]
In general, when performing directivity formation using a plurality of antenna elements on the transmission side, it is necessary to feed back channel information of the MIMO channel, including the techniques from Non-Patent Document 1 to Non-Patent Document 3 described above. At this time, if the number of antenna elements becomes enormous, the amount of channel information to be fed back becomes enormous, and various ideas are required. In the Massive MIMO system as described above, in order to obtain the forward link channel information in the transmission direction, the reverse link channel information in the reception direction is used, and the complex phase of the received signal generated by a circuit such as a low noise amplifier used during reception Is converted to the amount of rotation of the complex phase of the transmission signal generated by a circuit such as a high power amplifier used at the time of transmission, and the channel information of the reverse link is multiplied by a predetermined calibration coefficient to convert the forward link Channel information can be acquired. In general, these techniques are known as implicit feedback techniques (see, for example, Non-Patent Document 4). Details of the calibration process are shown below.

  In an actual wireless communication apparatus, in signal processing on the transmission side, signal amplification is often performed by a high power amplifier immediately before transmission. In this case, there is an error in the amplification factor due to the individual difference of the high power amplifier, and the complex phase may rotate at a different value for each high power amplifier in the high power amplifier. Similarly, in signal processing on the reception side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, there is an error in the amplification factor due to individual differences of the low noise amplifiers, and the complex phase may rotate with a different value for each low noise amplifier in the low noise amplifier. In some cases, the amplification factor and the amount of phase rotation may have frequency dependency. When the individual difference between the amplification factor and the amount of rotation of the complex phase is so large that it cannot be ignored, it is necessary to perform a calibration process when estimating the downlink channel information from the uplink channel information. Since the error of the amplification factor and the amount of phase rotation is almost stable in time, measure the error of the amplification factor and the amount of phase rotation in advance, and use a coefficient to cancel the influence of the error. Converts link channel information to downlink channel information. In the following description, the description will focus on calibration processing performed on the base station apparatus side having a plurality of antenna elements, but the same can be done on the terminal station apparatus side, and is common to general radio station apparatuses. It is an explanation.

  In the above description, the amplitude and the complex phase may be changed by a high power amplifier or a low noise amplifier (strictly speaking, a transmission system and a reception system circuit including other circuits such as a filter). In this case, it has been described that a calibration coefficient for correcting in accordance with a change in amplitude or complex phase is acquired in advance and used for correction. Although a known technique may be used for the calibration process, an example of the calibration process will be described below.

  FIG. 31 is a diagram illustrating asymmetry of channel information between the uplink and the downlink. In the figure, reference numerals 955-1 to 955-3 denote wireless modules, reference numerals 951-1 to 951-3 denote high power amplifiers (HPA), and reference numerals 952-1 to 952-3 denote low noise amplifiers (LNA). Reference numerals 953-1 to 953-3 denote TDD switches, and reference numerals 954-1 to 954-3 denote antenna elements.

Here, in order to explain the calibration technique, only the functions that affect the channel information in the radio station apparatus are extracted, and thus configurations other than those illustrated are omitted. Therefore, for the convenience of the configuration of the wireless modules 955-1 to 955-3, the high power amplifiers 951-1 to 951-3, the low noise amplifiers 952-1 to 952-3, the TDD switches 953-1 to 953-3, the antenna Although only the elements 954-1 to 954-3 are shown, functions such as an up-converter and a down-converter are mounted in the subsequent stage (previous stage). Further, for example, assuming that a radio station apparatus having a plurality of antennas has a plurality of radio modules 955-1 to 955-2 mounted in one housing, the radio module 955-3 counters this. This corresponds to a diagram in which the radio module 955-3 corresponding to one antenna element of the radio station apparatus that communicates is extracted and described. Further, when the signal passes through each of the high power amplifiers 951-1 to 951-3, the amplitude and the complex phase are Z _{HPA # 1} (f _k ), Z _{HPA # 2} (f _k ), Z _{HPA # 3} (f _k ) It shall change. Further, when the signal passes through the low noise amplifiers 952-1 to 952-3, the amplitude and the complex phase are Z _{LNA # 1} (f _k ), Z _{LNA # 2} (f _k ), and Z _{LNA # 3} (f _k). ) Change. Here, it is assumed that there is a frequency dependency as a general condition, and the frequency “(f _k )” for the k-th frequency component is described.

Here, for example, channel information when signals are transmitted from the wireless module 955-1 and the wireless module 955-2 to the test wireless module 955-3 will be described. Here, the channel information on the space between the antenna element 954-1 of the wireless module 955-1 and the antenna element 954-3 of the wireless module 955-3 is represented by h ₁ (f _k ), and the wireless module 955 -2 channel element information in the space between the antenna element 954-2 of -2 and the antenna element 954-3 of the wireless module 955-3 is represented by h ₂ (f _k ).

At this time, the channel information when the signal is actually transmitted from the wireless module 955-1 to the wireless module 955-3 indicates a change associated with the passage of the high power amplifier 951-1 in h ₁ (f _k ) in space. A coefficient Z _{HPA # 1} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying the passage of the low noise amplifier 952-3 are observed as multiplied values. Similarly, the channel information when transmitting a signal from the wireless module 955-2 to the wireless module 955-3 is a coefficient Z indicating a change associated with the passage of the high power amplifier 951-2 at h ₂ (f _k ) in space. _{HPA # 2} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying the passage through the low noise amplifier 952-3 are observed as multiplied values. Therefore, the channel from the wireless module 955-1 to the wireless module 955-3 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 3} (f _k ). A channel from the wireless module 955-2 to the wireless module 955-3 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 3} (f _k ). For this reason, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 955-1 and the wireless module 955-2 have a relative Z _{HPA # 2} (f _k ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same on the receiving side, and when the signal transmitted from the wireless module 955-3 is received by the wireless module 955-1, the channel information is stored in the high power amplifier 951 at h ₁ (f _k ) in space. Observed as a value obtained by multiplying a coefficient Z _{HPA # 3} (f _k ) indicating a change accompanying the passage of −3 and a coefficient Z _{LNA # 1} (f _k ) indicating a change accompanying the passing of the low noise amplifier 952-1 Is done. Similarly, when the signal transmitted from the wireless module 955-3 is received by the wireless module 955-2, the channel information changes in the space h ₂ (f _k ) with the passage of the high power amplifier 951-3. The coefficient Z _{HPA # 3} (f _k ) shown is multiplied by the coefficient Z _{LNA # 2} (f _k ) showing the change accompanying the passage of the low noise amplifier 952-2. Therefore, the channel from the wireless module 955-3 to the wireless module 955-1 is represented by Z _{HPA # 3} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). A channel from the wireless module 955-3 to the wireless module 955-2 is represented by Z _{HPA # 3} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). For this reason, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 955-1 and the wireless module 955-2 have relatively Z _{LNA # 2} (f _k ) / Z _{LNA # 1} (f _k ) difference occurs.

Here again, the channel information when the radio module 955-1 receives a signal transmitted from the radio module 955-3 corresponding to the reverse link in the left radio station apparatus is h ₁ (f _k ) · Z. _{HPA # 3} (f _k ) · Z _{LNA # 1} (f _k ). The channel information when the signal transmitted from the wireless module 955-1 corresponding to the forward link is received by the wireless module 955-3 is h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA. # 3} (f _k ). The calibration coefficient of the wireless module 955-1 is given by the following equation (4).

Similarly, channel information when the radio module 955-2 receives a signal transmitted from the radio module 955-3 corresponding to the reverse link in the left radio station apparatus is h ₁ (f _k ) · Z _{HPA # 3.} (F _k ) · Z _{LNA # 2} (f _k ). The channel information when the signal transmitted from the wireless module 955-2 corresponding to the forward link is received by the wireless module 955-3 is h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA. # 3} (f _k ). The calibration coefficient of the wireless module 955-2 is given by the following equation (5).

Here, for example, the channel information in the reverse link acquired by the wireless modules 955-1 to 955-2 is h ₁ (f _k ) · Z _{HPA # 3} (f _k ) · Z _{LNA # 1} (f _k ) and h, respectively. ₁ (f _k ) · Z _{HPA # 3} (f _k ) · Z _{LNA # 2} (f _k ), which is multiplied by calibration coefficients C ₁ (f _k ) and C ₂ (f _k ), h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA # 3} (f _k ) and h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA # 3} (f _k ) .

  In actual operation, when the wireless module of the actual communication partner is different from the wireless module 955-3, strictly speaking, the estimated value of the forward link channel information obtained by multiplying the calibration coefficient is the forward link. The channel information itself indicates a different value. However, even in that case, the value obtained by multiplying the true forward link channel information regarding the wireless module 955-1 and the wireless module 955-2 by the common coefficient coincides with the above estimated value, and the directivity formation is performed. Considering that there is no effect even if a common constant is multiplied for all antenna elements, there is no problem as feedback of channel information.

  In the above description, the transmitting side of the focused radio station apparatus is described as the forward link, and the receiving side is defined as the reverse link. Is called the downlink and the reverse link is called the uplink. Similarly, when the focused radio station apparatus is a terminal station apparatus or a relay station apparatus, the forward link is usually called an uplink and the reverse link is called a downlink.

Atsushi Suyama, Atsunori Ohara, Shin Yoon, Yukihiko Okumura, “Effects of Analog Beamforming Configuration in Massive MIMO Using High Frequency Band Hybrid Beamforming”, IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, March 2015 , Vol.114, no.490, RCS2014-337, p.213-218 Atsushi Ohta, Takuto Arai, Hiroshi Shirato, Kazuki Maruta, Yasuhiko Iwakuni, Masataka Iizuka, “Line-of-sight Millimeter-Wave Spatial Multiplexing Technology Using Parallel Transmission in the First Eigenmode: System Proposal and Basic Characteristics Evaluation Results”, IEICE Technical Report, IEICE, August 2015, vol.115, no.181, RCS2015-144, p.73-78 Atsushi Ohta, Hiroshi Shirato, Kazuki Maruta, Takuto Arai, Yasuhiko Iwakuni, Masataka Iizuka, “Efficient use of first eigenmode transmission in Massive MIMO for line-of-sight”, IEICE Technical Report, IEICE, 2015 November, vol.115, no.288, RCS2015-239, p.293-298 Hayato Fukuzono, Tomonori Murakami, Riichi Kudo, Yasushi Takatori, Hayato Mizoguchi, “Experimental evaluation of implicit feedback in downlink multi-user MIMO-OFDM system”, IEICE Technical Report, IEICE, November 2013 , Vol.113, no.301, RCS2013-187, p.79-84

In the above description, regardless of time axis beamforming and other methods, the implicit feedback technique is applied in channel information feedback, and the forward link channel information is acquired from the reverse link channel information by the calibration process. It was a premise. This premise is that channel information h ₁ (f _k ) and h ₂ (f _k ) between antenna elements 204-1 and 204-2 and antenna element 204-3 in FIG. It was assumed that the values would be equal. However, even in the millimeter wave band, which is said to be an abundant frequency resource, it is difficult to secure a bandwidth at the 1 GHz level in a relatively low frequency band, and currently 71 to 76 GHz / 81 to 86 GHz called E band are used in FDD, for example. It has become one of the promising options. This band is assumed to be used in FDD, but if the frequency is different from f _k ≠ f ′ _{k on} the upstream and downstream, h ₁ (f _k ) ≠ h ₁ (f ′ _k ) and h ₂ (f _k) ) ≠ h ₂ (f ′ _k ) and symmetry is not expected to be maintained. Therefore, when the above-described wireless system is constructed using FDD, channel symmetry is broken by using different frequencies, so that there is a problem that implicit feedback cannot be used.

  In view of the above circumstances, an object of the present invention is to provide a technique that can use implicit feedback even when a wireless system is constructed using FDD.

  One embodiment of the present invention is a wireless communication apparatus that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with other wireless communication apparatuses, and includes an uplink. A communication unit that communicates with the other wireless communication device using a first frequency used in the second frequency and a second frequency different from the first frequency used in the downlink, and a plurality of antenna elements or some of the plurality of antenna elements. A signal conversion unit provided in the system for converting an analog signal of a radio frequency received by the antenna element into a baseband digital signal, and a training signal transmitted by the other wireless communication device converted by the signal conversion unit Using a digital signal corresponding to the antenna element serving as a reference selected from a plurality of antenna elements provided with the signal conversion unit, and other antenna elements A correlation calculation unit that calculates a correlation for each combination; a rotation amount calculation unit that calculates a rotation amount of a complex phase to be given to a received signal of an individual antenna element based on a correlation calculation result by the correlation calculation unit; Based on the information on the ratio between the first frequency and the second frequency and the information calculated by the correlation calculation unit or the rotation amount calculation unit, the complex to be given to the transmission signal of the individual antenna element A rotation amount prediction unit that predicts a rotation amount of the phase, and a phase rotation that manages the rotation amount of the complex phase to be given to the transmission signal for each antenna element based on the complex phase predicted by the rotation amount prediction unit An amount management unit, a transmission signal generation unit that generates a signal addressed to the other wireless communication device, and a complex phase rotation amount managed by the phase rotation amount management unit for each antenna element used for transmission. A phase rotator for applying an analog signal or a digital signal to a signal; and a signal transmitter for transmitting a signal output via the phase rotator to the other wireless communication device via the antenna element; , A wireless communication device.

  One aspect of the present invention is the above-described wireless communication apparatus, further comprising a first signal distribution unit that branches the transmission signal generated by the transmission signal generation unit on an analog signal for each antenna element, and the phase The rotation unit sets the rotation amount of the complex phase managed by the phase rotation amount management unit in a phase shifter provided for each antenna element.

  One aspect of the present invention is the above-described wireless communication apparatus, further comprising a second signal distribution unit that branches the transmission signal generated by the transmission signal generation unit for each antenna element on a digital signal, and the phase The rotation unit multiplies the complex phase rotation amount ψ for each antenna element managed by the phase rotation amount management unit by a coefficient Exp (jψ) for each antenna element on the digital signal.

  One embodiment of the present invention is a wireless communication method performed by a wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device. A communication step of communicating with the other wireless communication device using a first frequency used in the uplink and a second frequency different from the first frequency used in the downlink; and all antennas A training signal transmitted by the other wireless communication device converted by a signal conversion unit that converts an analog signal of a radio frequency received by the antenna element into a baseband digital signal, which is provided in the element or a part of a plurality of systems. Using a digital signal corresponding to the antenna element serving as a reference selected from a plurality of antenna elements provided with the signal converter, and other antennas. A correlation calculation step for calculating a correlation for each combination of tenor elements, and a rotation amount for calculating a rotation amount of a complex phase to be given to a received signal of an individual antenna element based on a correlation calculation result in the correlation calculation step Based on the calculation step, the information about the ratio between the first frequency and the second frequency, and the calculation result of the correlation calculation step or the rotation amount calculation step, it is given to the transmission signal of the individual antenna element A rotation amount prediction step for predicting a rotation amount of the power complex phase, and a rotation amount of the complex phase to be given to the transmission signal for each antenna element are managed based on the complex phase predicted in the rotation amount prediction step. A phase rotation amount management step, a transmission signal generation step for generating a signal addressed to the other wireless communication device, and the phase rotation amount management step. The amount of rotation of the complex phase managed in the group is set in a phase rotation unit that gives an analog signal or a digital signal to the transmission signal for each antenna element used for transmission, and is output via the phase rotation unit. A signal transmission step of transmitting the received signal to the other wireless communication device via the antenna element.

  According to the present invention, it is possible to use implicit feedback even when a wireless system is constructed using FDD.

It is a functional block diagram which shows the structural example of the radio station apparatus in embodiment of this invention. It is a functional block diagram which shows the structural example of the radio station apparatus in the embodiment. It is a figure which shows the structural example of the communication system in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the radio station apparatus in 2nd Embodiment. It is a functional block diagram which shows the structural example of the radio station apparatus in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a figure which shows the outline | summary of the channel information prediction in the linear array by which the antenna element was arrange | positioned linearly in 3rd Embodiment. It is a figure which shows the specific example of the array antenna comprised in planar shape in the same embodiment. It is a figure which shows the specific example of the array antenna comprised in planar shape in the same embodiment. It is a figure which shows the specific example of the square array antenna comprised in planar shape in the same embodiment. It is a figure which shows the specific example of the square array antenna comprised in planar shape in the same embodiment. It is a figure which shows the specific example of the square array antenna comprised in planar shape in the same embodiment. It is a figure which shows the specific example of the array antenna by which transmission / reception separated by the planar shape in the same embodiment was isolate | separated. It is a figure which shows the specific example of the array antenna by which transmission / reception separated by the planar shape in the same embodiment was isolate | separated. It is a figure which shows the example of the antenna pattern used for prediction of the rotation amount of the complex phase in the same embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows another structural example of the transmission / reception signal processing circuit using the digital signal processing in the embodiment. It is a figure which shows the outline | summary of the conversion method of the complex phase in the 4th Embodiment of this invention. It is a flowchart which shows the flow of a process of the implicit feedback method at the time of FDD application in the 4th Embodiment of this invention. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a figure which shows the calibration process in a prior art.

  Embodiments of the present invention will be described in detail with reference to the drawings. First, the basic principle of the embodiment of the present invention will be described. The terms “time axis” and “frequency axis” used in this specification are sometimes expressed as “time domain” and “frequency domain”, but here they are unified as “time domain” and “frequency domain”. Give an explanation.

[Overview of basic principles]
In the time-axis beamforming technique, for example, at the time of reception, a main path of an incoming wave between transmitting and receiving stations is extracted, and signal processing for directing the directivity gain of the antenna element group in that direction is performed. The transmission / reception weight used at that time is a constant having no frequency dependence, and as a result, signal processing is reduced in various respects. However, since it is based on digital signal processing, each antenna element is individually multiplied by the transmission / reception weight in digital signal processing, and accompanying this, individual A / D (analog / digital) conversion is performed for each antenna system. And a D / A (digital / analog) converter.

  However, if the multiplication processing of the coefficient having no frequency dependency is limited to, for example, the complex phase rotation processing without amplitude change, digital signal processing is not necessarily required. Specifically, by using a phase shifter that is an analog circuit and passing the analog signal through this phase shifter set to a predetermined complex phase rotation, signal processing substantially equivalent to weight multiplication processing is performed. Can be realized. In Non-Patent Document 1, directivity control is realized using a phase shifter. For example, the phase rotation amount of each antenna element is set for each predetermined direction set in increments of 5 degrees in the horizontal / vertical direction. A combination set was determined in advance, and the phase rotation amount of each phase shifter was set according to the direction in which the beam obtained by some control procedure should be directed. However, this phase rotation amount combination set is selected from a pre-set menu, and it is necessary to search for a direction with the highest reception level while transmitting a training signal individually for each direction. . However, in time-axis beamforming, the phase rotation amount of each antenna element is optimized based on the training signal transmitted from the terminal station apparatus side, so the calculation of the phase rotation amount used for directivity formation is simply fed back. In addition, the combination set of complex phase rotation amounts can be set with a much higher degree of freedom without requiring a prior menu or the like.

  Therefore, the wireless station device (wireless communication device) in the embodiment of the present invention employs digital assist type analog beam forming. That is, the radio station apparatus performs the calculation process of the complex phase rotation amount performed by each phase shifter by digital signal processing, and controls the phase shifter using the complex phase value obtained by the digital signal processing. By doing so, the desired complex phase is rotated to form directivity on the analog signal.

  Here, the process of calculating the complex phase of the transmission / reception weight of the time-axis beamforming performed by digital signal processing does not necessarily have to be performed simultaneously by each antenna element. Or, even when digital signal processing is performed when calculating the complex phase of the transmission / reception weight of time-axis beamforming, the time rate for performing signal processing for this is the same as that of normal communication for transmitting / receiving user data. It is expected to be overwhelmingly less than the time rate. Therefore, the radio station apparatus according to the present embodiment performs the operations of these digital circuits in a limited time.

  Further, time axis beam forming corresponds to control for directing directivity to a path having the maximum intensity of an incoming wave. Therefore, in the plane wave approximation of the incoming wave from the direction of the path with the maximum intensity, the length of the perpendicular drawn from the planarly arranged antenna element to the plane having the same phase as the wavefront of the plane wave is the path length difference, The path length difference has a geometric periodicity and is given as a function of coordinates. The transmission / reception weight is a coefficient corresponding to the rotation of the complex phase that cancels the path length difference.

  Of course, in actuality, since it includes a multipath component and also includes a measurement error, the amount of rotation of the complex phase for each antenna element slightly deviates from clean periodicity. However, approximately, the x-axis and y-axis are set on the plane on which the antenna is arranged, and the three-dimensional notation is performed with respect to the coordinate point of the antenna element using the z-axis as the complex phase rotation amount. When the rotation amount of the complex phase is plotted against the coordinates, all plot points are on one plane in the plane wave approximation. Note that the complex phase rotation amount ψ is equivalent to the complex number Exp (−j {ψ ± 2π × integer}) even if ± 2π × integer is added. Depending on the coordinates of the element, there may be cases where it should be regarded as a value obtained by adding ± 2π × integer on the plane, but considering such a complex phase offset, the amount of rotation of the complex phase can be obtained with a small number of antenna elements. The remaining antenna elements can approximately acquire the rotation amount of the complex phase by linear interpolation processing based on the path length difference.

  Even if the uplink and downlink frequencies are different as in FDD, or if there is an asymmetry of the uplink / downlink channel that separates the transmission antenna and the reception antenna, this path length difference should be considered. It is possible to approximately obtain the rotation amount of the complex phase. For example, if the path length difference is ΔL and the wavelength is λ, the complex phase rotation amount for canceling the path length difference is given by (2πΔL / λ).

For example, the uplink frequency is F ₁ and the wavelength is λ ₁ , and the downlink frequency is F ₂ and the wavelength is λ ₂ . If the complex phase rotation amount Δθ _UL at the time of uplink reception is (2πΔL / λ ₁ ), the complex phase rotation amount Δθ _DL at the time of downlink reception is (2πΔL / λ ₂ ). The relational expression of Expression (6) holds.

Therefore, even in the case of FDD, if the amount of rotation of the complex phase is obtained based on the channel estimation of the frequency F _{1 in} the uplink, the value is multiplied by (λ ₁ / λ ₂ ) to approximate the frequency F _{2 in the} downlink. The amount of rotation of the complex phase is obtained. Since the uplink complex phase rotation amount does not need to be obtained for all antenna elements as described above, when the complex phase rotation amount of the frequency F ₁ is obtained for a part of the uplink antenna elements, the remaining amount is determined. It is also possible to determine the amount of rotation of the complex phase of the frequency F ₁ of the antenna element. If this is obtained, it is also possible to calculate the amount of rotation of the complex phase of the frequency F _{2 in} the downlink.

  At this time, if the rotation amount of the complex phase in at least three antenna elements is obtained, the rotation amount of the complex phase of the remaining antenna elements is calculated from the rotation amount of the complex phase of each antenna element coordinate on the plane including the three points. It can be calculated. If the amount of rotation of the complex phase can be calculated with four or more antenna elements, the amount of rotation of the complex phase in the three-dimensional space given by the coordinates of each antenna element and the amount of rotation of the complex phase, and the amount of rotation of the calculated complex phase Calculate the plane with the smallest sum of squared errors with the least-squares approach to the error, and calculate the rotation amount of the complex phase for each antenna coordinate on that plane, and set this as the phase shifter May correspond.

It should be noted here that the complex phase rotation amount obtained by the correlation calculation includes the uncertainty of the complex phase of 2π period, but the complex to be multiplied by the coefficient (λ ₁ / λ ₂ ). The amount of phase rotation must exclude the uncertainty of the complex phase of 2π period. For example, with respect to the first to sixth antenna elements arranged linearly, the rotation amount of the complex phase with reference to the first antenna element is “0”, “π / 2”, “ It is assumed that “π”, “3π / 2”, “0”, “π / 2”. Here, up to “3π / 2” of the fourth antenna element increases monotonously and has continuity, but the complex phase rotation amount “0” of the fifth antenna element should actually be regarded as “2π”. Similarly, the rotation amount “π / 2” of the complex phase of the sixth antenna element should actually be regarded as “5π / 2”. This is because, for example, if the assumption that the difference in the amount of rotation of the complex phase between adjacent antenna elements does not change by more than ± π due to the design condition of the antenna element spacing and the arrival direction, the complex phase of 2π period can be easily obtained. Can be removed, and the rotation amount of the complex phase may be multiplied by a coefficient (λ ₁ / λ ₂ ). As an example, if the coefficient (λ ₁ / λ ₂ ) is 0.9, the complex phase rotation amounts “0”, “π / 2”, “π” of the first to sixth antenna elements obtained from the uplink ”,“ 3π / 2 ”,“ 0 ”,“ π / 2 ”, the downlink complex phase rotation amount before the calibration is“ 0 ”,“ 0.9 × π / 2 ”, “0.9 × π”, “0.9 × 3π / 2”, “0.9 × 2π”, and “0.9 × 5π / 2”.
Even if the uplink and downlink antenna elements have different configurations, if the uplink antenna elements and the downlink antenna elements are mixed to take the entire antenna configuration, the uplink antenna elements When the amount of rotation of the complex phase of the frequency F ₁ of the antenna element is obtained by the receiving antenna, the amount of rotation of the complex phase of the frequency F ₂ when it is assumed that other downlink antenna elements are also used in the uplink is calculated. It is possible to perform a calibration process based on the phase rotation amount of the antenna element, and calculate the rotation amount of the complex phase of the frequency F ₂ of the antenna element for downlink.

Or, even when the uplink antenna element group and the downlink antenna element group are physically separated, the uplink antenna element group and the downlink antenna element group are simply parallel. When in a moving relationship (that is, on the same plane and the relative positional relationship of each antenna element does not change between the transmitting and receiving antennas), transmit the amount of downlink complex phase rotation obtained from the receiving antenna as it is Even if it is used as the complex phase of the corresponding antenna element on the side, if the distance between the transmitting and receiving stations is sufficiently far away and a common plane wave approximation is expected to be possible, a sufficiently effective directivity can be obtained. Formation is expected to be possible.
Hereinafter, a detailed embodiment to which the basic principle is applied will be described with reference to the drawings.

[First Embodiment]
FIG. 1 is a functional block diagram showing a configuration example (subarray separation type) of a radio station apparatus 450 in the present embodiment. In this figure, the same parts as those of the conventional radio station apparatus shown in FIGS. In the present embodiment, as corresponding to FIG. 28 and FIG. 29 of the prior art, a “sub-array separation type” configuration in which a directional beam is formed by being separated into a plurality of sub-arrays, and a plurality of directional beams in one array. There is a variation of the configuration by “sub-array shared type” (which may be understood as “integrated array” because it is not separated into sub-arrays strictly), but here “sub-array separated type” Give an explanation.

  The radio station apparatus 450 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N (N is an integer of 1 or more), and a control circuit 460. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 451-n (n = 1,..., N) includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, an A / D converter 125-n, TDD (Time Division Duplex) switch (TDD-SW) 127-n, phase shifters 402-n-1 to 402-n-M (M is 2 or more) Integer), switches 403-n-1 to 403-n-M, distribution coupler (HYB) 404-n, correlation calculation circuit 405-n, phase shift control circuit 406-n, down converter 424- n-1 to 424-n-M and A / D converters 425-n-1 to 425-n-M. The phase shifters 402-n-1 to 402-n-M are connected to the antenna elements 401-n-1 to 401-n-M, respectively. The D / A converters 122-1 to 122-N, the up converters 123-1 to 123-N, the down converters 124-1 to 124-N, and the A / D converters 125-1 to 125-N are respectively shown in FIG. 28 and the D / A converters 922-1-1 to 922-NM, the up converters 923-1-1 to 923-NM, and the down converters 924-1-1 to 924-NM. The same A / D converters 925-1-1 to 925 -N-M can be used.

  Here, in the down converters 424-1-1 to 424 -N-M, in order to perform frequency conversion between a radio frequency signal and a baseband signal, it is necessary to input a signal from a local oscillator. That is, a common local signal is used for the combination of the down converters 424-n-1 to 424-n-M in the same transmission / reception signal processing circuit 451-n (n = 1,..., N), and each down converter 424- It is necessary to suppress temporal changes in the relative relationship of the complex phases in n-1 to 424-n-M. In this sense, a configuration in which a local oscillator exists substantially outside the down converters 424-1-1 to 424-NM is taken. However, since the description is complicated, an external local is used here as a simple description. Oscillator specification is omitted. It is necessary to share the local oscillator used between the same n down-converters 424-n-1 to 424-n-M, but down-converters 424-n'-1 to 424-having different values of n. In combination with n′-M, the local oscillator to be used need not be shared. Further, these local oscillators do not need to be shared between the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. The only important thing is to make the local signal common between the signal sequences that implement the directivity formation in a coordinated manner.

Here, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the radio station apparatus 450 has a total of N transmission / reception signal processing circuits 451-1 to 451-N mounted therein. M represents the number of antenna elements of the subarray mounted on each of the transmission / reception signal processing circuits 451-1 to 451-N. In the description of FIG. 27 it has a total number of antenna elements and M _0, but because in this case as in the case of FIG. 28 is a subarray constitute the entire antenna number of antenna elements of each subarray were labeled as different values M. Each of the transmission / reception signal processing circuits 451-n is associated with sub-array antenna elements 401-n-1 to 401-n-M, and the transmission / reception signal processing circuits 451-1 to 451-N are described in Non-Patent Document 2. It is assumed that they are installed spatially apart. The baseband signal processing circuit 140 is connected to the transmission / reception signal processing circuits 451-1 to 451-N by wire, and a digital baseband signal is transferred over the wire. Although not shown in FIG. 27, the radio station device 450 includes an overall control circuit 460. In the figure, an example is shown in which the radio station device 450 mounts the control circuit 460 on the baseband signal processing circuit 140. The control circuit 460 manages the frame period and transmission / reception timing, and also manages the switching of the TDD switches 127-1 to 127 -N. The control circuit 460 determines whether to connect the distribution coupler 404-n to the up-converter 123-n or to the down-converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by dividing.

  Furthermore, in this embodiment, the estimation process of the complex phase rotation amount basically performed by the variable phase shifter corresponding to the transmission / reception weight is performed by digital signal processing, and the actual complex phase rotation processing is realized by analog signal processing. . For this reason, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are premised on signal processing on the frequency axis like the OFDM (Orthogonal Frequency Division Multiplexing) modulation method. Even in such a case, even when signal processing on the time axis is assumed as in SC-FDE (Single-Carrier Frequency Domain Equalization), it is possible to cope with either method. However, the signal separation circuit 141 performs signal separation between each signal series, but here it is also possible to perform signal separation on the time axis, and once convert it to a frequency axis signal by FFT, Frequency-dependent signal separation may be performed. In this sense, the input to the demodulators 130-1 to 130-N is assumed to be a time-axis signal or a frequency-axis signal. Will be described as input. In this way, the concept regarding variations of communication systems such as the OFDM modulation system and SC-FDE is the same in the following description.

The specific signal flow is as follows.
First, signal transmission will be described. In the wireless station device 450, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmits the signal with the up-converter 123-n and the distribution coupler 404-n connected (n = 1,..., N).

  Each of the modulators 120-1 to 120-N generates a time-axis digital baseband transmission signal of each stream to be spatially multiplexed and inputs the transmission signals to the transmission / reception signal processing circuits 451-1 to 451-N. The transmission / reception signal processing circuit 451-n (n = 1,..., N) receives the time base digital baseband transmission signal generated by the modulator 120-n. The D / A converter 122-n of the transmission / reception signal processing circuit 451-n (n = 1,..., N) converts the transmission signal input from the modulator 120-n into an analog baseband signal, and improves it. Input to converter 123-n. The up-converter 123-n converts the signal input from the D / A converter 122-n from a baseband signal to a radio frequency band signal and inputs the signal to the TDD switch 127-n. The TDD switch 127-n inputs the signal input from the up-converter 123-n to the distribution coupler 404-n.

  The distribution coupler 404-n branches the analog signal input from the TDD switch 127-n into M analog signals. The branched analog signals are input to the phase shifters 402-n-1 to 402-n-M via the switches 403-n-1 to 403-n-M. The phase shifters 402-n-1 to 402-n-M apply a predetermined complex phase rotation on the analog signal. For example, the phase shifters 402-n-1 to 402 -n-M apply a predetermined amount of complex phase rotation to a signal in the radio frequency band of the analog signal format. The analog signals to which complex phase rotation has been applied by the phase shifters 402-n-1 to 402-n-M are transmitted via the antenna elements 401-n-1 to 401-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 402-n-1 to 402-n -M. For example, the antenna elements 401-1-1 to 401-1 -M and the antenna elements 401-N-1 to 401 -N-M are individually formed with directivity, and the radio station apparatus in the directivity direction Communicate.

  Next, signal reception will be described. In the wireless station device 450, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receives a signal in a state where the distribution coupler 404-n and the down converter 124-n are connected (n = 1,..., N).

  Signals received by the antenna elements 401-n-1 to 401-n-M (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and passes through the switches 403-n-1 to 403-n-M. To the distribution coupler 404-n. The distribution coupler 404-n synthesizes the signals of the respective antenna systems input via the switches 403-n-1 to 403-n-M on analog signals, and the synthesized signal via the TDD switch 127-n. To the down converter 124-n. The down converter 124-n converts a radio frequency signal input via the TDD switch 127-n into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the signal input from the down converter 124-n from an analog baseband signal to a digital baseband signal and inputs the signal to the signal separation circuit 141.

  The signal separation circuit 141 performs signal separation by suppressing crosstalk components between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted into a frequency axis signal by FFT processing and performed on the frequency axis. When the crosstalk component is suppressed on the time axis, first, the correlation between signal sequences input to the signal separation circuit 141 is calculated based on the digital baseband signal corresponding to the received training signal. . Then, based on a MIMO (Multiple Input Multiple Output) channel matrix given by the calculated correlation, a general MIMO signal separation process such as ZF (Zero Forcing) type or MMSE (Maximum Mean Square Error) type signal separation is performed. A matrix similar to the above may be calculated, and this matrix may be multiplied by the sampling data unit by regarding the signal sequence input to the signal separation circuit 141 as a vector. In general MIMO signal separation processing such as ZF type and MMSE type signal separation on the general frequency axis, a different matrix is used for each frequency component, whereas almost the same matrix on the frequency axis. Is used, signal separation processing can be performed on the time axis in units of sampling data. Alternatively, only signal processing performed by the transmission / reception signal processing circuits 451-1 to 451-N is sufficient, and the signal separation circuit 141 does not need to perform any processing. However, in any case, details of the signal separation method here are omitted because they are not directly related to the features of the present embodiment. Demodulators 130-1 to 130-N demodulate the signal in which the crosstalk component is suppressed in signal separation circuit 141.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402 -N-M will be described. In this signal processing, the switches 403-n-1 to 403-n-M (n = 1,..., N) are shifted by the phase shifters 402-n-1 to 402-n-M and the down converters 424-n-1 to 424-n-1. 424-n-M is connected. These switch switchings are managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. The phase shifters 402-n-1 to 402-n -M (n = 1,..., N) are connected to the distribution coupler 404-n except when calculating the rotation amount of the complex phase. Further, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402 -NM are set to a predetermined value. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference with respect to the initial predetermined value. For example, in the most obvious example, the phase shifters 402-1-1 to 402-N-M may be set to all zero (or all the same value), and in this case, the rotation amount of the obtained complex phase May be used as the phase rotation amount of the phase shifters 402-1-1 to 402 -NM at the time of subsequent communication. Or, the initial predetermined values of the phase shifters 402-1-1-1 to 402-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the complex phase rotation amount is + α degrees. If it is + β degrees, + γ degrees,..., The phase rotation amounts of the phase shifters 402-1-1-402-NM during the subsequent communication are + (α + 10) degrees, + (β + 20) degrees. , + (Γ + 30) degrees, and so on.

  As an actual process, first, a radio station apparatus of a communication partner that should acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and the radio station apparatus 450 receives this training signal. Signals received by the antenna elements 401-n-1 to 401-n-M (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and passes through the switches 403-n-1 to 403-n-M. To the down converters 424-n-1 to 424-n-M. Each of the down converters 424-n-1 to 424-n-M converts the input radio frequency signal into an analog baseband signal and supplies the analog baseband signal to the A / D converters 425-n-1 to 425-n-M. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal, and outputs it to the correlation calculation circuit 405-n.

  Correlation calculation circuits 405-1 to 405-N calculate the rotation amount of the complex phase using equations (1) to (3), respectively. In addition, when the calibration calculation circuits 405-1 to 405-N require calibration processing as necessary, the complex phase on the transmission side is set as a value in consideration of the calibration coefficient in the equations (1) to (3). Determine the amount of rotation. The amount of rotation of the complex phase obtained by the correlation calculation circuit 405-n (n = 1,..., N) is the identification number of the wireless station device with which it communicates (when communicating with a plurality of wireless station devices. Together with a single wireless station apparatus in the P-P (point-to-point) type, the identification number is not necessary.) And input to the phase shift control circuit 406-n (n = 1,..., N) Is done. The phase shift control circuit 406-n stores the complex phase rotation amount to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the radio station apparatus of the communication partner. Manage it by memorizing it.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 460 grasps a wireless station device as a communication partner, and the phase shift control circuit 406-n (n = 1,... , N) to instruct the phase shifters 402-n-1 to 402-n-M to set the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication. The phase shift control circuit 406-n acquires the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication by reading from the memory or the like, and acquires the rotation amount of the complex phase by the phase shifters 402-n-1 to 402. Set to -n-M to realize beamforming on analog.

Although not clearly shown in the figure, for example, if a high-power amplifier (HPA) on the transmission side is arranged, it is arranged at a point described as “A” in the figure, and a low-noise amplifier on the reception side ( LNA) and the like are arranged at points described as “B” and “C _1-1 ” to “C _N-M ” in the drawing. Since the points described as “A” and “B” are common within the same transmission / reception signal processing circuits 451-1 to 451-N, the complex phases of the individual high power amplifier and low noise amplifier are uncertain. The calibration process for removing the property is unnecessary.

On the other hand, regarding the low-noise amplifiers described as “C _1-1 ” to “C _N-M ”, when the amount of rotation of the complex phase can fluctuate with time, the same transmission / reception signal processing circuit 451-1. 1 to 451 -N can cause complex phase uncertainty between the antenna elements 401-1-1 to 401 -N-M, and therefore, as in the case of the conventional implicit feedback calibration method. Therefore, it is necessary to remove the uncertainty of the complex phase of the low-noise amplifier of each system. Note that this embodiment can be applied to an arbitrary method, and a specific method of calibration processing is not limited. Considering this calibration result, for example, if the amount of rotation of the complex phase in each of “C _1-1 ”, “C _1-2 ”, and “C _1-3 ” is +10 degrees, +20 degrees, and +30 degrees, The phase rotation amount is adjusted by performing corrections of −10 degrees, −20 degrees, and −30 degrees for the complex phase rotation amounts obtained by the equations (1) to (3), respectively. The calibration result information is collected by a calibration circuit (not shown), and the phase shift control circuits 406-1 to 406-N or correlation calculation circuits 405-1 to 405-N store this information. To make corrections.

  In addition, since the transmission / reception signal processing circuits 451-1 to 451-N having a subarray configuration can be physically separated to reduce the correlation of directional beams formed on analog, generally, Install at a distance greater than a predetermined distance.

  Further, the same applies to all the following descriptions (including other embodiments). In FIG. 1, the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N are connected by wire. In this configuration, digital baseband signals are transmitted and received over the wire. The D / A converters 122-1 to 122-N and the A / D converters 125-1 to 125-N are connected to the baseband signal processing circuit 140. In the case of mounting, a signal flowing on a wired connection between the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N can be an analog baseband signal.

  Next, similarly to FIG. 29 with respect to FIG. 28 in the prior art, the radio station apparatus of this embodiment can also be configured as a “subarray shared type” that realizes a plurality of directional beam formations with a single array antenna. is there. This configuration is shown in FIG.

  FIG. 2 is a functional block diagram showing a configuration example (subarray shared type) of the radio station apparatus 452 in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The radio station apparatus 452 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N, distribution couplers (HYB) 407-1 to 407-M, and an antenna element 401-. 1-401-M and a control circuit 460. The distribution coupler (HYB) 407-m (m = 1,..., M) is a transmission / reception signal processing circuit 451-1, 451-2,. 2-m,..., 402-Nm, and antenna elements 401-m (m = 1,..., M).

  In radio station apparatus 452, as with radio station apparatus 450 shown in FIG. 1, down converters 424-1-1 to 424-NM perform frequency conversion between radio frequency signals and baseband signals. Therefore, input of a signal from the local oscillator is required. A common local signal is used for each combination of down converters 424-n-1 to 424-n-M (n = 1,..., N) in each of the transmission / reception signal processing circuits 451-1 to 451-N. It is necessary to suppress the time change of the relative relationship of the complex phase in the down converter. In this sense, a configuration in which a local oscillator exists substantially outside the transmission / reception signal processing circuits 451-1 to 451-N is taken. However, since the description is complicated, the description of the external local oscillator is used here as a simple description. The description is omitted. It is not necessary to share the local oscillator used in each of the transmission / reception signal processing circuits 451-1 to 451-N. Further, it is not necessary to share these local oscillators with the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. The only important thing is to make the local signal common between the signal sequences that implement the directivity formation in a coordinated manner. However, unlike the case of FIG. 1, the down converters 424-1-1 to 424-NM are all housed in the same housing, so that all local signals can be shared in the case of FIG. Is possible.

  Similar to the above description, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and M represents the number of antenna elements of a common array antenna. In radio station apparatus 450 shown in FIG. 1, transmission / reception signal processing circuits 451-1 to 451-N are mounted in physically different housings for each system. On the other hand, in the radio station apparatus 452 shown in FIG. 2, all the N systems of the transmission / reception signal processing circuits 451-1 to 451-N are mounted in one casing. Further, in the radio station apparatus 452, the transmission / reception signal processing circuits 451-1 to 451-N share the antenna elements 401-1 to 401-M via the distribution couplers 407-1 to 407-M. Here, in the radio station apparatus 450 shown in FIG. 1 and the radio station apparatus 452 shown in FIG. 2, the internal processing in the transmission / reception signal processing circuits 451-1 to 451-N is the same. Further, FIG. 2 shows a case where the control circuit 460 is mounted on the baseband signal processing circuit 140, but the control circuit 460 may be mounted at an arbitrary location in the wireless station device 452. The control circuit 460 manages the frame period, transmission / reception timing, and switching of the TDD switches 127-1 to 127 -N.

The following shows a specific signal flow in the radio station apparatus 452, focusing on the difference from the radio station apparatus 450 shown in FIG.
First, signal transmission will be described. In the wireless station device 452, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmits the signal with the up-converter 123-n and the distribution coupler 404-n connected (n = 1,..., N). This point is common to the radio station apparatus 450 shown in FIG.

  Each of the modulators 120-1 to 120-N generates a time-base digital baseband transmission signal of each stream for spatial multiplexing, and inputs the generated signals to the transmission / reception signal processing circuits 451-1 to 451-N. Radio frequency analog signals that have undergone processing for directivity formation are output from phase shifters 402-1-1 to 402 -N-M by processing similar to that of radio station apparatus 450 shown in FIG. 1. The phase shifters 402-1-1 through 402 -N-M send the signals of these systems to the distribution couplers 407-1 through 407 -M connected to the corresponding antenna elements 401-1 through 401 -M. input. That is, the phase shifters 402-1-m, 402-2-m,..., 402-Nm (m = 1,..., M) input signals to the distribution coupler 407-m. Distribution couplers 407-1 to 407 -M synthesize the input signals, and the combined signals are transmitted via antenna elements 401-1 to 401 -M.

  Next, signal reception will be described. In the radio station apparatus 452, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receives a signal in a state where the distribution coupler 404-n and the down converter 124-n are connected (n = 1,..., N). This point is common to the radio station apparatus 450 shown in FIG.

  Signals received by the antenna elements 401-1 to 401-M are input to the distribution couplers 407-1 to 407-M, respectively. Distribution coupler 407-m (m = 1,..., M) distributes the input signal to N systems, and phase shifters 402-1-m, 402-2-m,..., 402-N-m. To enter. The transmission / reception signal processing circuits 451-1 to 451-N perform signal processing in the radio station device 450 shown in FIG. 1 on the signals input to the phase shifters 402-1-1-1 to 402-NM as described above. Similar signal processing is performed, and a digital baseband signal is input to the signal separation circuit 141. The signal separation circuit 141 performs signal separation by suppressing the crosstalk component, and the demodulators 130-1 to 130-N demodulate the separated signals.

  For signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402 -N-M, the transmission / reception signal processing circuits 451-1 to 451-N are implemented in FIGS. The signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. 1 is the same as the signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. The description is omitted here.

  FIG. 3 is a diagram illustrating a configuration example of a communication system in the present embodiment. The communication system shown in the figure includes a radio station apparatus 452 shown in FIG. 2 and a radio station apparatus 450 shown in FIG. In radio station apparatus 450, each of a plurality of transmission / reception signal processing circuits 451-1 to 451-N forms one beam as a subarray. On the other hand, in the radio station apparatus 452, the (sub) array is shared, and one array antenna forms a plurality of beams. In actual operation, as shown in FIG. 3, the wireless station device 450 faces the wireless station device 452, so that N systems of signals can be spatially multiplexed.

  Next, another configuration example of the transmission / reception signal processing circuit according to the present embodiment will be described with reference to FIGS. The transmission / reception signal processing circuits shown in FIGS. 4 to 6 can be replaced with the transmission / reception signal processing circuits 451-1 to 451-N included in the wireless station device 450 shown in FIG. 1 or the wireless station device 452 shown in FIG. . Below, the case where it replaces with the transmission-and-reception signal processing circuits 451-1 to 451-N with which the radio station apparatus 450 is provided is demonstrated to an example. Note that when the transmission / reception signal processing circuits 451-1 to 451-N included in the wireless station device 452 are replaced, the antenna elements 401-n-1 to 401-n-M in FIGS. 401-M.

  FIG. 4 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 453-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The transmission / reception signal processing circuit 453-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a distribution coupler 414-n, and a phase shifter 409-n-1. 409-n-M, TDD switches 408-n-1 to 408-n-M, phase shifters 402-n-1 to 402-n-M, and switches 403-n-1 to 403-n-M Distribution coupler 415-n, down converter (DC) 124-n, A / D converter 125-n, down converter (DC) 424-n-1 to 424-n-M, A / D D converters 425-n-1 to 425-n-M, a correlation calculation circuit 405-n, and a phase shift control circuit 406-n are provided. The TDD switches 408-n-1 to 408-n-M are connected to the antenna elements 401-n-1 to 401-n-M, respectively. Here, as in FIG. 1, the specification of the external local oscillator is omitted.

  The difference between the transmission / reception signal processing circuit 453-n and the transmission / reception signal processing circuits 451-1 to 451-N shown in FIG. 1 or 2 is that the TDD switches 408-n-1 to 408-n-M are the antenna elements 401. -N-1 to 401-n-M are arranged in the immediate vicinity, and as a result, the path from the up-converter 123-n to the antenna elements 401-n-1 to 401-n-M in the transmission system and the antenna in the reception system The path from the elements 401-n-1 to 401-n-M to the down converter 124-n is physically separated.

In the case of the transmission / reception signal processing circuit 451-n shown in FIG. 1 or FIG. 2, for example, the high-power amplifier (or power amplifier) of the transmission system is arranged at the location described as “A” after the up-converter 123-n. The low-noise amplifier of the reception system is a place described as “B” in the previous stage of the down converter 124-n (in addition, “C _n-1 ” in the previous stage of the down converters 424-n-1 to 424-n-M, "C _n-2", ..., which is arranged in "C _n-M" and described location), thereby enabling shared by transmission and reception circuits from the TDD switch 127-n to the antenna terminal. This point is greatly different from the transmission / reception signal processing circuit 453-n. The merit of adopting the configuration as shown in FIG. 4 is that high power amplifiers and low noise amplifiers can be mounted as many as the number of antenna elements, and as a result, the total transmission power is improved, and the TDD switch in FIGS. It is possible to suppress the effects of insertion loss and distribution and coupling loss of various circuits between 127-n and antenna elements 401-n-1 to 401-n-M. Therefore, although not shown here, in the transmission and reception signal processing circuit 453-n, "D _1", "D _2", ..., high-power amplifier location as "D _M" is "E It is preferable that a low noise amplifier is disposed at a place where “ ₁ ”, “E ₂ ”,..., “E _M ” is described. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is placed at a location marked “A” and a low-noise amplifier is placed at a location marked “B”, then “D ₁ ”, “D ₂ ”,. _M "location and" E ₁ described as "," E ₂ ", ..., high-power amplifier in each location described as" E _M ", not necessarily lead to a low-noise amplifier is disposed.

The following shows a specific signal flow in the transmission / reception signal processing circuit 453-n focusing on the difference.
First, signal transmission will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M include antenna elements 401-n-1 to 401-n-M and phase shifters 409-n-1 to 409-n-. Signals are transmitted with M connected to each other.

  The transmission / reception signal processing circuit 453-n receives a time base digital baseband transmission signal of one stream from a modulator 120-n not shown here. The D / A converter 122-n converts the input transmission signal into an analog baseband signal and inputs the analog baseband signal to the upconverter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a radio frequency band signal and inputs the signal to the distribution coupler 414-n.

  Distribution coupler 414-n branches the analog signal in the radio frequency band input from up-converter 123-n into M-system analog signals and inputs them to phase shifters 409-n-1 to 409-n-M. . Each of the phase shifters 409-n-1 to 409-n-M applies a predetermined complex phase rotation on the analog signal to the signal input from the distributor / coupler 414-n, to thereby generate a TDD switch 408-n-1. Are input to the antenna elements 401-n-1 to 401-n-M through .about.408-n-M. The antenna elements 401-n-1 to 401-n-M transmit the input transmission signal. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 409-n-1 to 409-n-M.

  Next, signal reception will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M are connected to the antenna elements 401-n-1 to 401-n-M and the phase shifters 402-n-1 to 402-n-. M is connected, and the switches 403-n-1 to 403-n-M receive signals in a state where the phase shifters 402-n-1 to 402-n-M and the distribution coupler 415-n are connected. Do.

  The signals received by the antenna elements 401-n-1 to 401-n-M are respectively phase shifters 402-n-1 to 402-n-M via the TDD switches 408-n-1 to 408-n-M. Is input. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on the analog signal to the input signal, and switches 403-n-1 to 403-n-M To the distribution coupler 415-n. Distribution coupler 415-n synthesizes the signals of each antenna system on an analog signal and inputs the resultant signal to down converter 124-n. The down converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal. The A / D converter 125-n inputs the converted digital baseband signal to the signal separation circuit 141 in the baseband signal processing circuit 140 not shown here. The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402 -N-M will be described. The transmission / reception signal processing circuit 453-n performs this signal processing with the switches 403-n-1 to 403-n-M and the phase shifters 402-n-1 to 402-nM and the down converters 424-n-1 to 403-n-1. 424-n-M is connected. These switch switchings are managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. The phase shifters 402-n-1 to 402-n -M are connected to the distribution coupler 415-n except when calculating the rotation amount of the complex phase. Further, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402 -NM are set to a predetermined value. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference with respect to the initial predetermined value, as described above. The transmission / reception signal processing circuits 453-1 to 453 -N perform the same processing all at once under the management of the control circuit 460.

  As an actual process, first, a radio station apparatus of a communication partner that should acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and includes radio signal processing circuits 453-1 to 453-N. Receives this training signal. The signals received by the antenna elements 401-n-1 to 401-n-M are respectively phase shifters 402-n-1 to 402-n-M via the TDD switches 408-n-1 to 408-n-M. Is input. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and passes through the switches 403-n-1 to 403-n-M. To the down converters 424-n-1 to 424-n-M. Each of the down converters 424-n-1 to 424-n-M converts the input radio frequency signal into an analog baseband signal and supplies the analog baseband signal to the A / D converters 425-n-1 to 425-n-M. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal, and outputs it to the correlation calculation circuit 405-n.

  The correlation calculation circuit 405-n calculates the amount of rotation of the complex phase as described in FIG. The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the phase shift control circuit 406-n together with the identification number of the radio station apparatus with which communication is performed. The phase shift control circuit 406-n stores the complex phase rotation amount to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the radio station apparatus of the communication partner. Manage it by memorizing it.

The amount of rotation of the complex phase described above relates to the amount of phase rotation of the phase shifters 402-n-1 to 402-n-M in the reception system, but the amount of complex phase rotation in a low noise amplifier, a high power amplifier, and the like. In order to cancel the individual difference, the correlation calculation circuit 405-n performs calibration processing in the implicit feedback of the prior art, and performs correction in the transmission system by correction corresponding to the calibration processing based on the amount of complex phase rotation in the reception system. The amount of rotation of the complex phase is converted to a value set in the phase shifters 409-n-1 to 409-n-M. However, when a low noise amplifier is arranged at a location described as “E ₁ ”, “E ₂ ”,..., “E _M ”, a sufficient reception level can be obtained, so “C ₁ ”, “C ₂ ”. ,..., “C _M ” does not require a low-noise amplifier. Therefore, the rotation of the complex phase in each low-noise amplifier of the reception system does not need to be distinguished from the rotation of the complex phase in space. The complex phase rotation amount calculated by the correlation calculation circuit 405-n can be used as it is as the complex phase rotation amount set in the phase shifters 402-n-1 to 402-n-M of the system. The phase shift control circuit 406-n is a complex in the transmission system to be set in each of the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M. The amount of phase rotation is managed by associating it with the identification number of the radio station apparatus of the communication partner and storing it in the memory in the same manner.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 460 grasps a wireless station device that is a communication partner and communicates with the phase shift control circuit 406-n. Instructing the phase shifters 402-n-1 to 402-n -M and the phase shifters 409-n-1 to 409-n-M to set the rotation amount of the complex phase corresponding to the radio station apparatus to be performed. The phase shift control circuit 406-n acquires the rotation amount of the complex phase in each of the reception system and the transmission system corresponding to the wireless station apparatus that performs communication by reading from the memory. The phase shift control circuit 406-n sets the amount of rotation of the complex phase of the reception system to the phase shifters 402-n-1 to 402-n-M, and sets the amount of rotation of the complex phase of the transmission system to the phase shifter 409. -N-1 to 409-n-M is set to realize analog beamforming.

  FIG. 5 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 454-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  Differences between the transmission / reception signal processing circuit 454-n shown in the figure and the transmission / reception signal processing circuit 453-n shown in FIG. 4 are as follows. That is, the transmission / reception signal processing circuit 453-n shown in FIG. 4 shares the antenna elements 401-n-1 to 401-n-M for transmission / reception. On the other hand, the transmission / reception signal processing circuit 454-n shown in FIG. 5 has a configuration in which these are separated by transmission and reception, a pair is formed by transmission / reception, and a transmission / reception antenna is disposed as a set in close proximity. Therefore, when the transmission / reception signal processing circuit 451-n of the wireless station device 450 is replaced with the transmission / reception signal processing circuit 454-n, antenna elements 441-n-1 to 441-n-M are added and the TDD switch 408-n- 1-408-n-M is omitted. When the transmission / reception signal processing circuit 451-n of the wireless station device 452 is replaced with the transmission / reception signal processing circuit 454-n, the transmission / reception signal processing circuit 454- is replaced with the antenna elements 401-n-1 to 401-n-M. 1 to 454-N, corresponding to antenna elements 401-1 to 401-M and antenna elements 441-1 to 441-M (and antenna elements 401-1 to 401-M and antenna elements 441-1 to 441-M). Distributed coupler).

  In the description of FIG. 4, for the sake of convenience, description has been made assuming that the antenna element up to the immediate vicinity is regarded as the transmission / reception signal processing circuit 453-n, but the TDD switches 408-n-1 to 408-n-M If considered, FIG. 4 and FIG. 5 are completely equivalent diagrams. Also in the details of the signal processing, in the transmission / reception signal processing circuit 453-n shown in FIG. 4 in the reception system, the antenna elements 401-n-1 to 401-n-M receive signals, whereas FIG. In the transmission / reception signal processing circuit 454-n shown in FIG. 4, the antenna elements 441-n-1 to 441-n-M receive signals, and transmission / reception does not pass through the TDD switches 408-n-1 to 408-n-M. Except for this point, all signal processing in the transmission / reception signal processing circuit 454-n shown in FIG. 5 and the transmission / reception signal processing circuit 453-n shown in FIG. 4 is common.

  However, in consideration of the physically different transmission / reception antennas, in addition to simple calibration processing, it is also possible to perform correction in consideration of physically different antenna element coordinates. In the above description, the transmission / reception antennas are arranged as a set in close proximity. However, as long as correction is performed in consideration of the physical differences in the coordinates of the antenna elements, the transmission / reception antennas are not necessarily set. There is no need to place them.

  FIG. 6 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 455-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

Also in the transmission / reception signal processing circuit 455-n shown in FIG. 6, the TDD switches 408-n-1 to 408-n-M are regarded as functions on the antenna element side, similarly to the transmission / reception signal processing circuit 454-n shown in FIG. Otherwise, FIG. 6 is also completely equivalent to FIG. In this sense, the transmission / reception signal processing circuit 455-n illustrated in FIG. 6 also receives the transmission antenna elements 401-n-1 to 401-n-M and the reception, similarly to the transmission / reception signal processing circuit 454-n illustrated in FIG. The antenna elements 441-n-1 to 441-n-M are separated from each other. However, in the transmission / reception signal processing circuit 454-n shown in FIG. 5, a pair of individual transmission / reception antenna elements (for example, a pair of the antenna element 401-n-1 and the antenna element 441-n-1) is integrally disposed in the vicinity. On the other hand, in the transmission / reception signal processing circuit 455-n shown in FIG. 6, the transmitting antenna elements 401-n-1 to 401-n-M constitute one transmitting antenna array and receive antenna elements. 441-n-1 to 441-n-M are assumed to constitute a single receiving antenna array. Accordingly, there is no difference between FIG. 5 and FIG. 6 except for the physical arrangement of the antenna elements (or geometrical differences on the wiring connecting the antenna elements).
Note that when the transmission / reception antennas are separated and operated as shown in FIG. 6, a wall-like obstacle may be arranged in order to avoid leakage of signals from the transmission antenna to the reception antenna.

  As described above, according to the present embodiment, the radio station apparatus uses the A / D converter for each antenna element only when calculating the rotation amount of the complex phase for performing directivity control. The radio station apparatus substitutes analog beam forming using a phase shifter instead of digital beam forming at the time of actual signal transmission. On the other hand, when receiving a signal, the radio station device uses a switch to set the output destination of the signal from the receiving antenna to the A / D converter when receiving the training signal, and to the receiving circuit when performing signal synthesis and demodulation processing. Switch. As a result, in Massive MIMO, the number of A / D converters and D / A converters, which are conventionally required constantly when performing digital beam forming, can be greatly reduced. Therefore, a situation in which a large number of A / D converters and D / A converters continue to operate constantly during communication is avoided, and a huge amount of A / D converters and D / A conversions that have been required particularly during broadband communication are avoided. It is possible to reduce the power consumption and reduce the amount of heat generated. Furthermore, since the A / D converter and the D / A converter do not require a large heat sink that is necessary for heat generation, not only low power consumption but also miniaturization of the radio station apparatus is achieved. It is possible to reduce the cost.

[Second Embodiment]
In the first embodiment, separate down converters and A / D converters are mounted on all (transmitting) and receiving antennas, but these circuits are used only for channel estimation. This works effectively to reduce overall power consumption and suppress the amount of heat generated, but from the viewpoint of miniaturization of the device, a circuit that is only temporarily used is mounted by a huge number of antenna elements. Is inefficient. In particular, A / D converters that operate at a very high speed due to the wide bandwidth tend to be expensive in price per unit, which may lead to an increase in the price of the entire apparatus. In the second embodiment, a configuration for consolidating these down converters, A / D converters, and the like into one system is shown.

  FIG. 7 is a functional block diagram showing a configuration example (subarray separation type) of the radio station apparatus 550 in the present embodiment. Similarly to the first embodiment, the second embodiment also forms a directional beam into a plurality of subarrays so as to correspond to FIGS. 1 and 2 in the first embodiment. ”And a“ sub-array shared type ”that realizes multiple directional beams in one array (strictly speaking, it is not separated into sub-arrays and may be understood as“ integrated array ”). There are variations. In FIG. 7, the “sub-array separation type” will be described. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The radio station apparatus 550 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N (N is an integer of 1 or more), and a control circuit 560. The transmission / reception signal processing circuit 551-n (n = 1,..., N) includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, and an A / D converter 125-n. A TDD switch 127-n, phase shifters 502-n-1 to 502-nM, switches 503-n-1 to 503-nM, a distribution coupler 504-n, and a correlation calculation circuit. 505-n and a phase shift control circuit 506-n. Phase shifters 502-n-1 to 502-n-M are connected to antenna elements 501-n-1 to 501-n-M, respectively.

  As with the radio station apparatus 450 shown in FIG. 1, N corresponds to the number of multiplexed streams (number of streams) when performing spatial multiplexing, and the radio station apparatus 550 includes transmission / reception signal processing circuits 551-1 to 551 -N as a whole. Only N systems are installed. Similarly to the radio station apparatus 450, M represents the number of antenna elements of the subarray mounted on each of the transmission / reception signal processing circuits 551-1 to 551 -N. Each of the transmission / reception signal processing circuits 551-n is accompanied by sub-array antenna elements 501-n-1 to 501-n-M, and the transmission / reception signal processing circuits 551-1 to 551-N are as shown in Non-Patent Document 2. It is assumed that they will be installed spatially apart from each other. The baseband signal processing circuit 140 is connected to the transmission / reception signal processing circuits 551-1 to 551 -N by wire, and a digital baseband signal is transferred over the wire. Although not shown in FIG. 27, an example is shown in which the radio station device 550 mounts the entire control circuit 560 on the baseband signal processing circuit 140. The control circuit 560 manages the frame period and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N. The control circuit 560 determines whether to connect the distribution coupler 504-n to the up-converter 123-n or to the down-converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by dividing. In addition, the control circuit 560 instructs the correlation calculation circuit 505-n (n = 1,..., N) to control the switches 503-n-1 to 503-n-M.

  Furthermore, in the present embodiment, the complex phase rotation amount calculation processing basically performed by the phase shifter corresponding to the transmission / reception weight is performed in the form of digital signal processing, and the actual complex phase rotation processing is realized by analog signal processing. For this reason, even if the modulators 120-1 to 120-N and the demodulator 130-1 to 130-N assume signal processing on the frequency axis as in the OFDM modulation scheme, the time axis as in SC-FDE. Even when the above signal processing is premised, it is possible to cope with either system, and the concept regarding variations of the communication system such as the OFDM modulation system and SC-FDE is explained in the first embodiment. It is the same.

  In addition, in the up converters 123-1 to 123-N and the down converters 124-1 to 124-N, in order to perform frequency conversion between a radio frequency signal and a baseband signal, a signal input from a local oscillator is input. Although necessary, coordinated signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551 -N, and therefore it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description. Although omitted in the following description, it is of course possible to perform cooperative signal processing between the transmission / reception signal processing circuits 551-1 to 551 -N as an additional function. You may do.

A specific signal flow in radio station apparatus 550 is as follows.
First, signal transmission will be described. Radio station apparatus 550 sets all switches 503-n-1 to 503-n-M to ON (connected state to distribution coupler 504-n), and TDD switch 127-n is up-converter 123-n-1 to 123. Signal transmission is performed with n-N and distribution coupler 504-n connected (n = 1,..., N). These conditions are the same in all the transmission / reception signal processing circuits 551-1 to 551 -N.

  Each of the modulators 120-1 to 120-N generates a time base digital baseband transmission signal of each stream to be spatially multiplexed, and inputs the transmission signal to the transmission / reception signal processing circuits 551-1 to 551-N, and then transmits and receives Processing until the up-converter 123-n of the signal processing circuit 551-n (n = 1,..., N) inputs a radio frequency band signal to the subsequent stage is the same as that of the radio station device 450 shown in FIG. . The up-converter 123-n inputs a signal to the distribution coupler 504-n via the TDD switch 127-n.

  The distribution coupler 504-n branches the analog signal input from the TDD switch 127-n into M-system analog signals, and the phase shifter 502-n via the switches 503-n-1 to 503-n-M. Input to -1 to 502-n-M. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation on the analog signal to the input signal. Analog signals to which complex phase rotation has been applied by the phase shifters 502-n-1 to 502-n-M are transmitted via the antenna elements 501-n-1 to 501-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 502-n-1 to 502-n-M, and communicates with the radio station apparatus ahead of the directivity.

  The above description is signal processing common to the transmission / reception signal processing circuits 551-1 to 551 -N mounted in N systems, and the same processing is performed in parallel while maintaining a substantially synchronous relationship such as operating with a common clock. carry out.

  Next, the flow of signals related to reception will be described. In the radio station apparatus 550, all the switches 503-n-1 to 503-n-M are turned ON (connected state with the distribution coupler 504-n), and the TDD switch 127-n is connected to the distribution coupler 504-n and the down converter. A signal is received in a state in which 124-n is connected (n = 1,..., N). These conditions are the same in all the transmission / reception signal processing circuits 551-1 to 551 -N.

  Signals received by the antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to the phase shifters 502-n-1 to 502-n-M, respectively. The phase shifters 502-n-1 to 502-n-M respectively apply a predetermined complex phase rotation on the analog signals to the input signals, and pass through the switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. The distribution coupler 504-n combines the signals of the respective antenna systems input via the switches 503-n-1 to 503-n-M on the analog signal, and combines the combined signals via the TDD switch 127-n. To the down converter 124-n. The subsequent processing is the same as that of the radio station apparatus 450 shown in FIG.

  Each of the phase shifters 502-n-1 to 502-n-M performs a predetermined complex phase rotation on the analog signal with respect to the signals received via the antenna elements 501-n-1 to 501-n-M. Are added to form a predetermined directivity, and communication is performed with a radio station apparatus ahead of the directivity. The above description is signal processing common to the transmission / reception signal processing circuits 551-1 to 551 -N mounted in N systems, and the same processing is performed in parallel while maintaining a substantially synchronous relationship such as operating with a common clock. carry out.

  The signal separation circuit 141 performs the same processing as the radio station device 450 shown in FIG. 1 on the signals input from the transmission / reception signal processing circuits 551-1 to 551 -N, and removes them in the above-described directivity formation. A process of suppressing crosstalk components between radio station apparatuses that cannot be fully processed is performed. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted into a frequency axis signal by FFT processing and performed on the frequency axis. Alternatively, only signal processing for directivity formation performed by the transmission / reception signal processing circuits 551-1 to 551 -N is required, and the signal separation circuit 141 does not need to perform any particular processing (in this case, the signal separation circuit 141 Is optional). However, in any case, the details of the signal separation method here are not directly related to the present embodiment, and can be implemented using the conventional MIMO signal processing technique, and thus the description thereof is omitted here. .

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1-1 to 502-NM will be described. In this signal processing, any one of the switches 503-n-1 to 503-n-M (n = 1,..., N) is operated by the phase shifters 502-n-1 to 502-n-M and the down converter 124-. n is connected (ON) while the remaining switches are in a state (OFF) in which the connection with the down converter 124-n is disconnected. Of the switches 503-n-1 to 503-n-M (n = 1,..., N), the targets to be connected (turned on) to the down converter 124-n are sequentially switched. These switch switchings are managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. Note that during normal operation other than when calculating the amount of rotation of the complex phase described here, the phase shifters 502-n-1 to 502-n-M (n = 1,..., N) as described above. ) Are all connected to distribution coupler 504-n. Further, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amount of the phase shifters 502-1-1-1 to 502-NM is set to a predetermined value. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference with respect to the initial predetermined value. For example, in the most obvious example, the phase shifters 502-1-1-1 to 502-NM may be set to all zero (or all the same value), and in this case, the rotation amount of the obtained complex phase May be used as the phase rotation amount of the phase shifters 502-1-1-1 to 502-NM during subsequent communication. Or, the initial predetermined values of the phase shifters 502-1-1-1 to 502-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the rotation amount of the complex phase is + α degrees. If it is + β degrees, + γ degrees,..., The phase rotation amounts of the phase shifters 502-1-1-1 to 502-NM during the subsequent communication are + (α + 10) degrees and + (β + 20) degrees. , + (Γ + 30) degrees,.

  As an actual process, first, a radio station apparatus as a communication partner to acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and the radio station apparatus 550 receives this training signal. Signals received by the antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to the phase shifters 502-n-1 to 502-n-M, respectively. The phase shifters 502-n-1 to 502-n-M respectively apply a predetermined complex phase rotation on the analog signals to the input signals, and pass through the switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. Here, since all of the switches 503-n-1 to 503-n-M are OFF, the signal synthesized in the distribution coupler 504-n is effectively converted to the switch 503-n. Only the signals received by the antennas of the systems to which the switches are connected (ON) among n-1 to 503-n-M are output. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 504-n, as a whole, one antenna is selected from the antenna group of the antenna elements 501-n-1 to 501-n-M. A process of extracting the element 501 -n-k (k is an integer from 1 to M) is performed. In addition, it switches with a predetermined period so that k may take any value from 1 to M. The reception signal of the antenna element 501-nk selected in this manner is input to the down converter 124-n via the TDD switch 127-n. The down converter 124-n converts the input radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  Each of the correlation calculation circuits 505-1 to 505-N continuously records the digital baseband signal until the signals from all the switches are received while switching. That is, the correlation calculation circuit 505-n receives digital baseband signals from all the switches 503-n-1 to 503-n-M while switching the switches 503-n-1 to 503-n-M. And record. The correlation calculation circuit 505-n considers the periodicity of the training signal (for example, the periodicity such that the training signal having the same content is repeated in a 2048 sample period) with respect to the recorded digital baseband signal, and Sampling data of each antenna element 501-n-k is extracted so that the sampling timing in the period corresponds, and in the same manner as the correlation calculation circuits 405-1 to 405-N of the first embodiment, Using equation (3), the amount of rotation of the complex phase to be set for each of the phase shifters 502-n-1 to 502-n-M is calculated. Note that this utilizes the fact that the channel fluctuation can be ignored if the radio station apparatus is fixedly installed at a high place and is in a line-of-sight environment. Further, when calibration processing is necessary as necessary, the correlation calculation circuits 505-1 to 505-N are similar to the correlation calculation circuits 405-1 to 405-N of the first embodiment. The amount of rotation of the complex phase on the transmission side is determined as a value in consideration of the calibration coefficient in 1) to Equation (3).

In the first embodiment, the sampling data of each antenna element 401-n-k can be acquired at the same time. However, in the second embodiment, since sampling is performed at different timings, the periodicity of the training signal Therefore, it has been devised so that it can be regarded as equivalently sampled at the same time. At this time, if the frequency error cannot be ignored on the transmission side and the reception side, sampling may not be considered equivalently at the same time just by the periodicity of the training signal. In such a case, the frequency error is corrected. You may do. For example, assuming that the training signal has a periodicity of N _FFT samples with respect to the sampling data S _k (n) of the training signal while one switch 503-nk is continuously ON, the N _Test cycle Suppose that sampling data can be secured. If the frequency error is Δf, Δf can be estimated by obtaining Δf ′ that maximizes the Q value in the following equation (7) for various Δf ′.

Here, attention is focused only on the information of the switches 503-nk, but Δf may be obtained for the sampling data of each of the switches 503-nk and averaged. If Δf is estimated in this way, the influence of the frequency error can be eliminated by performing the correction shown in the following equation (8) on the sampling data S _k (n ′).

  Here, n 'means a serial number that is continuous between switching from the switch 503-n-1 to the switch 503-n-1, regardless of switch switching. Since the phase of 2πjΔfn ′ rotates during the time of sampling period × n ′, the rotation is reversely corrected.

  Thus, the rotation amount of the complex phase obtained by the correlation calculation circuit 505-n (n = 1,..., N) is the identification number of the radio station apparatus with which it communicates (communication with a plurality of radio station apparatuses). In the same manner as in the first embodiment, an identification number is not necessary when communication is performed with a single radio station apparatus in the P-P type.) And is input to the phase shift control circuit 506-n. The phase shift control circuit 506-n associates the amount of rotation of the complex phase to be set to each of the phase shifters 502-n-1 to 502-nM with the identification number of the communication partner radio station apparatus in the memory. Manage it by memorizing it.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 560 grasps the wireless station device that is a communication partner, and the phase shift control circuit 506-n (n = 1,... , N), the phase shifters 502-n-1 to 502-n-M are instructed to set the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication. The phase shift control circuit 506-n acquires the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication by reading it from the memory, and acquires the rotation amount of the complex phase by the phase shifters 502-n-1 to 502. Set to -n-M to realize beamforming on analog.

  Although not clearly shown in the figure, for example, if a high-power amplifier (HPA) on the transmission side is arranged, it is arranged at a point described as “A” in the figure, and a low-noise amplifier on the reception side ( LNA) or the like is arranged at a point described as “B” in the drawing. Regarding the points described as “A” and “B”, in the same transmission / reception signal processing circuits 551-1 to 551 -N, the antenna elements 501 -n- 1 to 501 -n-M with respect to the same n. In this embodiment, which basically does not assume cooperative transmission between the transmission / reception signal processing circuits 551-1 to 551 -N, the individual high power amplifiers and low noise amplifiers are used. A calibration process for removing the uncertainty of the complex phase is unnecessary.

  Further, since the transmission / reception signal processing circuits 551-1 to 551 -N having a subarray configuration can be physically separated to reduce the correlation of directional beams formed on analog, generally, Install at a distance greater than a predetermined distance.

  Next, as in FIG. 2 with respect to FIG. 1 in the first embodiment, the radio station apparatus of this embodiment is configured by a “sub-array shared type” that realizes a plurality of directional beam formations with one array antenna. Is also possible. This configuration is shown in FIG.

  FIG. 8 is a functional block diagram showing a configuration example (subarray shared type) of the wireless station device 552 in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The radio station apparatus 552 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N, distribution couplers (HYB) 507-1 to 507-M, and an antenna element 501-. 1-501-M and a control circuit 560. Distribution couplers (HYB) 507-m (m = 1,..., M) are the phase shifters 502-1-m, 502- of the transmission / reception signal processing circuits 551-1, 551-2,. 2-m,..., 502-Nm, and the antenna element 501-m.

  Also in the radio station apparatus 552, as in the radio station apparatus 550 shown in FIG. 7, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the transmission / reception signal processing circuits 551-1 to 551 -N This is implemented for N systems. M represents the number of antenna elements of the subarray mounted on the wireless station device 552. In the radio station apparatus 550 shown in FIG. 7, since the number of antenna elements of each subarray is M, the total number of antennas is N × M. However, in this drawing shared as in FIG. Of the value M. In actual operation, the radio station apparatus 550 shown in FIG. 7 and the radio station apparatus 552 shown in FIG. 8 do not necessarily have the same number of antenna elements.

  The baseband signal processing circuit 140 is connected by wire to each of the transmission / reception signal processing circuits 551-1 to 551 -N, and a digital baseband signal is transferred over this wire. Similarly to the case of FIG. 1, FIG. 8 shows an example in which the entire control circuit 560 is mounted on the baseband signal processing circuit 140.

  Similar to the wireless station device 550, the wireless station device 552 basically performs digital phase processing for the complex phase rotation amount performed by the phase shifter corresponding to the transmission / reception weight, and the actual complex phase rotation processing is analog signal processing. To achieve. Therefore, similarly to the radio station device 550, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N of the radio station device 552 have a time axis when assuming signal processing on the frequency axis. Both methods can be used when the above signal processing is assumed.

  Further, in the up converters 123-1 to 123-N and the down converters 124-1 to 124-N, in order to perform frequency conversion between a radio frequency signal and a baseband signal, a signal input from a local oscillator is input. Necessary. However, cooperative signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551 -N, and it is not always necessary to use a common local oscillator. Unlike the case of the radio station apparatus 550 shown in FIG. 7, in the radio station apparatus 552, the up converters 123-1 to 123-N and the down converters 124-1 to 124-N are all contained in the same casing. For this reason, it is expensive to prepare individual local oscillators for the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. Therefore, there is substantially a local oscillator shared externally. The configuration is common. However, since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  As described above, in the wireless station device 552, the transmission / reception signal processing circuits 551-1 to 551-N for N systems are mounted in one housing, and are connected via the distribution couplers 507-1 to 507-M. The antenna elements 501-1 to 501-M are shared. 8 illustrates an example in which the control circuit 560 is mounted on the baseband signal processing circuit 140, but the control circuit 560 may be mounted at any location in the wireless station device 552. The control circuit 560 manages the frame period and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N.

  The difference between the radio station device 550 shown in FIG. 7 and the radio station device 552 shown in FIG. 8 is that the transmission / reception signal processing circuits 551-1 to 551 -N are aggregated in the housing of one radio station device 552 and shared. Each of the sub-array antenna elements 501-1 to 501-M is distributed by distribution couplers 507-1 to 507-M and connected to transmission / reception signal processing circuits 551-1 to 551-N. is there. Therefore, the processing in the transmission / reception signal processing circuits 551-1 to 551 -N is the same for both the radio station device 550 shown in FIG. 7 and the radio station device 552 shown in FIG. 8.

Hereinafter, focusing on the difference from the radio station apparatus 550 illustrated in FIG. 7, a specific signal flow that is the difference in the radio station apparatus 552 is shown.
First, signal transmission will be described. In radio station apparatus 552, processing for directivity formation is performed from each of phase shifters 502-n-1 to 502-n-M (n = 1,..., N) of transmission / reception signal processing circuit 551-n. A radio frequency analog signal is output. Each of these M signals is input to distribution couplers 507-1 to 507-M connected to the corresponding antenna elements 501-1 to 501-M. That is, the phase shifters 502-1-m, 502-2-m,..., 502-Nm (m = 1,..., M) input signals to the distribution coupler 507-m. Distribution couplers 507-1 to 507 -M synthesize signals input from N transmission / reception signal processing circuits 551-1 to 551 -N, and the combined signals are antenna elements 501-1 to 501 -M. Sent through.

  Next, signal reception will be described. Signals received by antenna elements 501-1 to 501-M of radio station apparatus 552 are distributed to N systems in distribution couplers 507-1 to 507-M, respectively, and transmission / reception signal processing circuits 551-1 to 551- are respectively received. N is input. For example, a signal received by the antenna element 501-m (m = 1,..., M) is distributed to N systems by a distribution coupler 507-m, and phase shifters 502-1-m, 502-2-m, ..., input to 502-Nm. The radio station device 552 performs signal processing similar to the signal reception in the radio station device 550 shown in FIG. 7 on the signals input to the phase shifters 502-1-1 to 502 -N-M in this way.

  In the signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1-1 to 502-NM, the signals received by the antenna elements 501-1 to 501-M are respectively similar to the reception processing. In the distribution couplers 507-1 to 507 -M, the signals are distributed to the N systems and are respectively input to the transmission / reception signal processing circuits 551-1 to 551 -N. The radio station device 552 uses the same signal as the calculation of the amount of rotation of the complex phase in the radio station device 550 shown in FIG. 7 as the signal input to the phase shifters 502-1-1-1 to 502-NM. Process.

  Other operations are basically performed in the same manner as the radio station apparatus 550 shown in FIG. Further, the explanation regarding the HPA on the transmission side and the LNA on the reception side is the same as that of the radio station apparatus 550 shown in FIG.

  Note that the communication system according to the present embodiment includes a wireless station device 550 instead of the wireless station device 450 illustrated in FIG. 3 and a wireless station device 552 instead of the wireless station device 452. Note that the radio station device 450 and the radio station device 552 may face each other, or the radio station device 550 and the radio station device 452 may face each other.

  Next, another configuration example of the transmission / reception signal processing circuit according to the present embodiment will be described with reference to FIGS. The transmission / reception signal processing circuits shown in FIGS. 9 to 12 can be replaced with the transmission / reception signal processing circuits 551-1 to 551 -N included in the wireless station device 550 shown in FIG. 7 or the wireless station device 552 shown in FIG. 8. . Below, the case where it replaces with the transmission-and-reception signal processing circuits 551-1 to 551-N with which the radio station apparatus 550 is provided is demonstrated to an example. Note that when the transmission / reception signal processing circuits 551-1 to 551-N included in the wireless station device 552 are replaced, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 in FIGS. ˜501-n-M correspond to the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M, the antenna elements 501-1 to 501-M, and the antenna elements 541-1 to 541-M. Distributed coupler (HYB)).

  FIG. 9 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 553-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The transmission / reception signal processing circuit 553-n shown in the figure includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, an A / D converter 125-n, and a circulator 521-. n, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, distribution couplers 504-n, correlation calculation circuit 505-n, phase Shift control circuit 506-n. Phase shifters 502-n-1 to 502-n-M are connected to antenna elements 501-n-1 to 501-n-M, respectively.

  In this figure, the antenna elements 501-n-1 to 501-n-M are shown in the figure for easy understanding of the comparison with FIG. 7, but the transmission / reception signal processing circuit 553-n corresponds to the inside of the solid line. In this frame, it corresponds to the transmission / reception signal processing circuit 551-n in FIG.

  The difference between the transmission / reception signal processing circuit 551-n shown in FIG. 7 and FIG. 8 and the transmission / reception signal processing circuit 553-n shown in FIG. 7 is the same as that of the transmission / reception signal processing circuit 551-n shown in FIG. Although the input / output flow of the transmission signal and the reception signal is controlled by n, the transmission / reception signal processing circuit 553-n shown in FIG. 9 controls this input / output flow using the circulator 521-n. That is, circulator 521-n passes the input signal from up-converter 123-n to distribution coupler 504-n, and passes the input signal from distribution coupler 504-n to down-converter 124-n. The circulator 521-n suppresses other signal flows. Therefore, similar signal processing can be performed without using the TDD switches 127-1 to 127 -N as in the transmission / reception signal processing circuit 551-n shown in FIGS. 7 and 8. Except for this point, all other signal processing is equivalent to the signal processing of the transmission / reception signal processing circuit 551-n shown in FIGS. The explanation of the HPA on the transmission side and the LNA on the reception side is the same as that of the transmission / reception signal processing circuit 551-n shown in FIGS.

  FIG. 10 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 554-n in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The transmission / reception signal processing circuit 554-n shown in the figure includes a D / A converter 122-n, an up converter 123-n, a distribution coupler 514-n, and phase shifters 509-n-1 to 509-n. -M, TDD switches 508-n-1 to 508-nM, phase shifters 502-n-1 to 502-nM, switches 503-n-1 to 503-nM, and distribution A coupler 515-n, a down converter 124-n, an A / D converter 125-n, a correlation calculation circuit 505-n, and a phase shift control circuit 506-n are provided. TDD switches 508-n-1 to 508-n-M are connected to antenna elements 501-n-1 to 501-n-M, respectively.

  Here, similarly to the transmission / reception signal processing circuits 551-1 to 551 -N shown in FIG. 7, the up converter 123-n and the down converter 124-n perform frequency conversion between a radio frequency signal and a baseband signal. Therefore, input of a signal from the local oscillator is required. However, since no cooperative signal processing is assumed between the transmission / reception signal processing circuits 554-1 to 554 -N, it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  Further, the entire control circuit 560 is mounted in the baseband signal processing circuit 140 of the radio station apparatus 550 or the radio station apparatus 552 or in the radio station apparatus 552. The control circuit 560 manages transmission / reception timing. Further, since the plurality of transmission / reception signal processing circuits 554-1 to 554 -N operate in conjunction with each other, when the control circuit 560 performs control, the plurality of transmission / reception signal processing circuits 554-1 to 554 -N are timed. Works together.

  The difference between the transmission / reception signal processing circuit 554-n and the transmission / reception signal processing circuits 551-1 to 551-N shown in FIG. 7 or 8 is that the TDD switches 508-n-1 to 508-n-M are the antenna elements 501. -N-1 to 501-n-M (or antenna elements 501-1 to 501-M) are arranged in the immediate vicinity, and as a result, from the D / A converter 122-n in the transmission system to the antenna element 501-n-1 The path from .about.501-n-M and the path from the antenna elements 501-n-1 to 501-n-M to the A / D converter 125-n in the receiving system are physically separated. .

In the case of the transmission / reception signal processing circuit 551-n shown in FIG. 7 or FIG. 8, for example, the high power amplifier (or power amplifier) of the transmission system is arranged at the place described as “A” after the up converter 123-n. The low-noise amplifier of the receiving system is arranged at a location described as “B” in the previous stage of the down converter 124-n, and the circuit from the TDD switch 127-n to the antenna end can be shared by transmission and reception. This point is greatly different from the transmission / reception signal processing circuit 554-n. The merit of adopting the configuration as shown in FIG. 10 is that high power amplifiers and low noise amplifiers can be mounted as many as the number of antenna elements. As a result, the total transmission power is improved, and the TDD switch in FIGS. It is a point that it becomes possible to suppress the influence of insertion loss and distribution and coupling loss of various circuits between 127-n and antenna elements 501-1 to 501-M. For this reason, although not shown here, high power amplifiers are provided at locations where “D ₁ ”, “D ₂ ”,..., “D _M ” are described, “E ₁ ”, “E ₂ ”,. It is preferable that a low-noise amplifier is arranged at a place described as “E _M ”. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is disposed at a location described as “A” and a low-noise amplifier is disposed at a location described as “B”, “D ₁ ”, “D ₂ ” _,. There is no necessity that a high power amplifier and a low noise amplifier are arranged at each of the point locations described as “E ₁ ”, “E ₂ ”,..., “E _M ”.

  The above changes correspond to the transmission / reception signal processing circuit 451-n in FIG. 1 and FIG. 2 in the first embodiment being changed to the transmission / reception signal processing circuit 453-n shown in FIG. 7 and 8 in the second embodiment is changed to the transmission / reception signal processing circuit 554-n shown in FIG. This corresponds to the change to 453-n.

The following shows a specific signal flow in the transmission / reception signal processing circuit 554-n, which is a difference from the transmission / reception signal processing circuit 551-n, focusing on the characteristic part of the present embodiment.
First, signal transmission will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M have antenna elements 501-n-1 to 501-n-M and phase shifters 509-n-1 to 509-n-. Signals are transmitted with M connected to each other.

  The transmission / reception signal processing circuit 554-n receives a time-base digital baseband transmission signal of one stream from a modulator 120-n not shown here. The D / A converter 122-n converts the input transmission signal into an analog baseband signal and outputs the analog baseband signal to the upconverter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a radio frequency band signal and inputs the signal to the distribution coupler 514-n.

  Distribution coupler 514-n branches the analog signal in the radio frequency band input from up-converter 123-n into M-system analog signals and inputs them to phase shifters 509-n-1 to 509-n-M. . Each of the phase shifters 509-n-1 to 509-n-M applies a predetermined complex phase rotation on the analog signal to the signal input from the distribution / combining device 514-n, so that the TDD switch 508-n-1 is applied. Are input to the antenna elements 501-n-1 to 501-nM through .about.508-nM. The antenna elements 501-n-1 to 501-n-M transmit the input transmission signals. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 509-n-1 to 509-n-M.

  Next, signal reception will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M are connected to the antenna elements 501-n-1 to 501-nM and the phase shifters 502-n-1 to 502-n-. M is connected, and the switches 503-n-1 to 503-n-M receive signals in a state where the distribution couplers 515-n and the phase shifters 502-n-1 to 502-n-M are respectively connected. I do.

  Signals received by the antenna elements 501-n-1 to 501-n-M are respectively phase shifters 502-n-1 to 502-n-M via TDD switches 508-n-1 to 508-n-M. Is input. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and switches 503-n-1 to 503-n-M. To the distribution coupler 515-n. Distribution coupler 515-n combines the signals of the respective antenna systems input via switches 503-n-1 to 503-n-M on an analog signal and inputs the combined signal to down converter 124-n. The down converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal input from the down converter 124-n into a digital baseband signal, and a signal in the baseband signal processing circuit 140 not shown here. This is input to the separation circuit 141. The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1-1 to 502-NM will be described. In this signal processing, any one of the switches 503-n-1 to 503-n-M connects (ON) the distribution coupler 515-n and the phase shifters 502-n-1 to 502-n-M. On the other hand, the remaining switches are performed in a state where the connection with the distribution coupler 515-n is disconnected (OFF). These switch switchings are managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. During normal operation other than when calculating the amount of rotation of the complex phase described here, all of the phase shifters 502-n-1 to 502-n-M are distributed couplers 515-n as described above. In addition, the phase shifters 509-n-1 to 509-n-M are all connected to the distribution coupler 514-n. Similarly to the transmission / reception signal processing circuit 551-n shown in FIG. 7 or FIG. 8, the phase rotation of the phase shifters 502-1-1-502-1-M is performed when the complex phase rotation amount calculation processing is performed. The amount is set to a predetermined value, and the rotation amount of the complex phase obtained by the subsequent processing is set as a difference with respect to the initial predetermined value.

  As an actual process, first, a radio station apparatus of a communication partner that should acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and has radio signal processing circuits 554-1 to 554 -N. Receives this signal. Signals received by antenna elements 501-n-1 to 501-n-M are input to phase shifters 502-n-1 to 502-n-M, respectively. The phase shifters 502-n-1 to 502-n-M respectively apply a predetermined complex phase rotation on the analog signals to the input signals, and pass through the switches 503-n-1 to 503-n-M. To the distribution coupler 515-n. Here, since all of the switches 503-n-1 to 503-n-M are OFF, the signal synthesized in the distribution coupler 515-n is effectively converted to the switch 503-n. Only the signal received by the antenna of the system to which the switch is connected (ON) among n-1 to 503-n-M is output. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 515-n as a whole, one antenna element 501-n from the antenna elements 501-n-1 to 501-n-M is obtained. The process of extracting -k (k is an integer from 1 to M) is performed. Note that k is changed to take any value from 1 to M. The reception signal of the antenna element 501-nk selected in this way is input to the down converter 124-n. The down converter 124-n converts the input radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal into a digital baseband signal and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  The correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 554-n are the correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 551-n shown in FIG. The same processing is performed. That is, the correlation calculation circuit 505-n calculates the amount of rotation of the complex phase on the transmission side to be set in each of the phase shifters 502-n-1 to 502-n-M. The phase shift control circuit 506-n associates the amount of rotation of the complex phase to be set to each of the phase shifters 502-n-1 to 502-nM with the identification number of the communication partner radio station apparatus in the memory. Manage it by memorizing it.

In addition, the correlation calculation circuit 505-n has a high power amplifier in the place described as “D ₁ ”, “D ₂ ”,..., “D _M ”, and “E ₁ ”, “E ₂ ”,. In the case where the low noise amplifier is arranged at a place described as “E _M ”, similarly to the correlation calculation circuit 405-n of the transmission / reception signal processing circuit 453-n in FIG. The rotation amount may be subjected to a calibration process in the implicit feedback of the prior art, and converted to the complex phase rotation amount in the transmission system by correction corresponding to the calibration process based on the rotation amount of the complex phase in the reception system. . The phase shift control circuit 506-n associates the rotation amount of the complex phase in the transmission system to be set to each of the phase shifters 509-n-1 to 509-n-M with the identification number of the radio station apparatus of the communication counterpart. Similarly, it is managed by storing it in a memory.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 560 grasps a wireless station device that is a communication partner and communicates with the phase shift control circuit 506-n. It instructs the phase shifters 502-n-1 to 502-n-M and the phase shifters 509-n-1 to 509-n-M to set the complex phase rotation amount corresponding to the radio station apparatus to be performed. The phase shift control circuit 506-n acquires the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication by reading it from the memory, and acquires the rotation amount of the complex phase by the phase shifters 502-n-1 to 502. -N-M and phase shifters 509-n-1 to 509-n-M are set to realize analog beamforming. These processes are similarly performed in the transmission / reception signal processing circuits 554-1 to 554-N.

  FIG. 11 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 555-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  Differences between the transmission / reception signal processing circuit 555-n shown in the figure and the transmission / reception signal processing circuit 554-n shown in FIG. 10 are as follows. That is, the transmission / reception signal processing circuit 554-n shown in FIG. 10 shares the antenna elements 501-n-1 to 501-n-M for transmission / reception. On the other hand, the transmission / reception signal processing circuit 555-n shown in FIG. 11 has a configuration in which these are separated by transmission and reception, a pair is formed by transmission / reception, and a transmission / reception antenna is set as a set at a close location. Therefore, when the transmission / reception signal processing circuit 551-n of the wireless station device 550 is replaced with the transmission / reception signal processing circuit 555-n, antenna elements 541-n-1 to 541-n-M are added, and the TDD switch 508-n- 1-508-n-M is omitted. When the transmission / reception signal processing circuit 551-n of the wireless station device 552 is replaced with the transmission / reception signal processing circuit 555-n, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 to 541 are used. In place of −n−M, the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M (and the antenna elements 501-1 to 501) shared by the transmission / reception signal processing circuits 555-1 to 555-N are used. -M, a distribution coupler corresponding to the antenna elements 541-1 to 541-M).

  In the description of FIG. 10, for the sake of convenience, the description up to the immediate vicinity of the antenna element is described as the transmission / reception signal processing circuit 554-n, but the TDD switches 508-n-1 to 508-n-M are assumed to be functions on the antenna element side. If considered, FIG. 10 and FIG. 11 are completely equivalent diagrams. Also in the details of the signal processing, in the receiving system, the transmission / reception signal processing circuit 554-n shown in FIG. 10 receives signals from the antenna elements 501-n-1 to 501-n-M, whereas FIG. The transmission / reception signal processing circuit 555-n shown in FIG. 5 receives data using the antenna elements 541-n-1 to 541-n-M, and does not pass through the TDD switches 508-n-1 to 508-n-M in transmission / reception. Except for this point, all signal processing in the transmission / reception signal processing circuit 554-n shown in FIG. 10 and the transmission / reception signal processing circuit 555-n shown in FIG. 11 is common.

  However, in consideration of the physically different transmission / reception antennas, in addition to simple calibration processing, it is also possible to perform correction in consideration of physically different antenna element coordinates.

  In the above description, the transmission / reception antennas are arranged as a set in close proximity. However, as long as correction is performed in consideration of the physical differences in the coordinates of the antenna elements, the transmission / reception antennas are not necessarily set. There is no need to place them. FIG. 12 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 556-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  Also in the transmission / reception signal processing circuit 556-n shown in FIG. 12, the TDD switches 508-n-1 to 508-n-M have the functions on the antenna element side as in the transmission / reception signal processing circuit 555-n shown in FIG. Considering this, FIG. 10 is also completely equivalent to FIG. In this sense, also in the transmission / reception signal processing circuit 556-n shown in FIG. 12, similarly to the transmission / reception signal processing circuit 555-n shown in FIG. 11, the transmission antenna elements 501-n-1 to 501-n-M and the reception are received. The antenna elements 541-n-1 to 541-n-M are separated. However, in the transmission / reception signal processing circuit 555-n shown in FIG. 11, a pair of individual transmission / reception antenna elements (for example, a pair of the antenna element 501-n-1 and the antenna element 541-n-1) is integrally disposed in the vicinity. On the other hand, in the transmission / reception signal processing circuit 556-n shown in FIG. 12, the transmitting antenna elements 501-n-1 to 501-n-M constitute one transmitting antenna array and receive antennas. The elements 541-n-1 to 541-n-M are assumed to constitute a single receiving antenna array. Therefore, there is no difference in signal processing between FIG. 11 and FIG. 12 except for the physical arrangement of antenna elements.

  When performing digital beam forming in Massive MIMO, the conventional radio station apparatus requires expensive A / D converters and D / A converters corresponding to the number of signal sequences, so the apparatus is expensive. In addition, there is a problem that power consumption increases. Therefore, the radio station apparatus according to the present embodiment switches the switches so that only the target signal sequence is connected to the A / D converter only when weight information for performing directivity control is acquired. In addition, when transmitting data, the radio station apparatus converts a signal before separation for each antenna element into an analog signal by a D / A converter, and uses the phase shifter after separating the converted analog signal for each antenna element. To perform analog beam forming. As a result, it is possible to reduce the number of A / D converters and D / A converters necessary for acquiring weight information during data transmission / reception, and to reduce power consumption.

[Third Embodiment]
In general, when acquiring channel information of a plurality of antenna elements, in a multipath environment, the correlation of channel information is greatly reduced in the case of antenna elements separated by about a half wavelength. However, in a case where the line of sight environment is generally obtained or the reflection point is limited, the plane wave approximation for the incoming wave having the strongest intensity tends to be applicable. In this case, if the wavefronts arriving at the same complex phase can be extracted, the path length difference for each antenna element between the antenna plane and the wavefront of the arriving wave has geometric regularity, and some antennas The complex phase of the channel information of the remaining antenna elements can be estimated from the complex phase of the channel information of the elements.

  FIG. 13 shows an outline of channel information prediction in a linear array in which antenna elements are linearly arranged in the third embodiment of the present invention. In the figure, a state in which the antenna elements 401-1 to 401-5 are arranged in a straight line is shown. First, consider a plane wave coming from an angle θ direction with respect to the front direction of the linear array. Also, let d be the spacing between the elements of each antenna. Here, the received signal at the antenna element 401-1 is Φ1 (t), the received signal at the antenna element 401-2 is Φ2 (t),..., And the received signal at the antenna element 401-5 is Φ5 (t), Let ΔLs be the path length difference of the s-th antenna element (s is an integer of 2 or more) with reference to the antenna element 401-1. For convenience, ΔL1 is 0.

  In general, when the wavelength is λ, the complex phase rotates by 2πΔL / λ through the path length ΔL. Assuming plane wave approximation, the path length difference between the antenna element 401-1 and the antenna element 401-2 is ΔL2 = d · Sinθ. Similarly, the path length difference between the antenna element 401-1 and the antenna element 401-3 is ΔL3 = 2 × d · Sinθ, and the path length difference between the antenna element 401-1 and the antenna element 401-4 is ΔL4 = 3. × d · Sinθ, the path length difference between the antenna element 401-1 and the antenna element 401-5 is ΔL5 = 4 × d · Sinθ.

  Therefore, if attention is paid to the frequency component of a certain wavelength λ, the plane is a wave that is approximated, and Φ2 (t) ≈Exp {−2πjΔL2 / λ} Φ1 (t), Φ3 (t) ≈Exp {−2πjΔL3 / λ } Φ1 (t), Φ4 (t) ≈Exp {−2πjΔL4 / λ} Φ1 (t), Φ5 (t) ≈Exp {−2πjΔL5 / λ} Φ1 (t) holds. Using the relationship of ΔLs = (s−1) d · Sinθ, Φs (t) ≈Exp {−2πj (s−1) d · Sinθ / λ} Φ1 (t), The complex phase is rotated by Exp {−2πjd · Sinθ / λ}. Therefore, for example, when channel estimation is performed by the antenna element 401-1 and the antenna element 401-5, and 1/4 of the difference of the complex phase is rotated for each antenna element based on the amount of rotation of the complex phase between them. It becomes possible to predict.

  Similar prediction is possible when the antenna elements are arranged two-dimensionally in addition to the case where the antenna elements are arranged one-dimensionally. Next, a specific example of channel information prediction of a planar array antenna according to this embodiment will be described with reference to FIGS. 14 and 15. In FIG. 14, alphabets a to z and A to K indicated by “◯” represent antenna elements. Here, an example of a shape in which equilateral triangles are spread in close-packed form is shown, but the shape may be any other configuration.

  For example, it is assumed that the radio station apparatus performs channel estimation of three points of antenna elements i, m, and q indicated by double black circles, and obtains a complex phase of the channel information of the three points. Here, for example, the antenna element i is a reference antenna, the rotation amount of the complex phase between the antenna element i and the antenna element m is ζ, and the rotation amount of the complex phase between the antenna element i and the antenna element q is η. . In this case, if attention is paid to the antenna element on the straight line connecting the antenna element i and the antenna element m, the antenna element c is ζ × 1/3, the antenna element d is ζ × 2/3, and the antenna element B is ζ × 4 /. 3. The antenna element u can be approximated to −ζ × 1/3. Similarly, when attention is paid to the antenna element on a straight line connecting the antenna element i and the antenna element q, the antenna element b is η × 1/3, the antenna element g is η × 2/3, and the antenna element G is η × 4 /. 3. The antenna element v can be approximated to −η × 1/3. If this is extended, two-dimensional prediction is also possible. For example, in the case of the antenna element a, ζ × 1/3 + η × 1/3, and in the case of the antenna element H, −ζ × 1/3 + η × 4/3, In the case of the antenna element E, it can be predicted as ζ × 2/3 + η × 3/3.

  As a condition for enabling the above prediction, the amount of rotation of the complex phase needs to be π or less between the antenna element that acquires the channel information. For example, taking the case of FIG. 13 as an example, whether the complex phase difference between the antenna element 401-1 and the antenna element 401-5 is π / 2 or -3π / 2, Exp { It cannot be distinguished from j · π / 2} = Exp {−j · 3π / 2}. If the complex phase difference is π / 2, the complex phase difference between the antenna element 401-1 and the antenna element 401-2 should be π / 2 × (¼). Is −3π / 2, the complex phase difference between the antenna element 401-1 and the antenna element 401-2 should be −3π / 2 × (1/4).

  Since channel information cannot be correctly predicted in the situation where there is uncertainty as described above, the complex phase difference between antenna elements that acquire channel information needs to be π or less. In actual measurement, complex phase fluctuations are also expected due to measurement errors due to noise and the effects of multipath. Therefore, in practice, the complex phase difference needs to be a small value with a margin more than π. The reference value is different depending on the magnitude of the influence of the reflected wave and cannot be generally stated. However, when the antenna element interval d is reduced or the arrival angle θ is sufficiently small, the path length difference ΔL is reduced. Therefore, the amount of change in the complex phase accompanying the path length difference can be made sufficiently small. In the example of FIG. 14, the antenna elements i, m, and q are positioned relatively apart from each other, but the mutual element spacing is small so that the complex phase difference between the antenna elements is less than or equal to π. Adjacent antenna elements (for example, antenna elements a, e, and f) can be used. Although FIG. 14 shows an example in which channels are estimated with three antenna elements and the complex phases of the remaining antenna elements are approximated in a two-dimensional plane, channel information is acquired with more antenna elements. Therefore, it becomes possible to predict channel information a little more finely.

  FIG. 15 shows another example of channel information prediction of an array antenna configured in a planar shape according to the third embodiment of the present invention. The difference from FIG. 14 is that the number of antenna elements for acquiring channel information is changed from seven antenna elements a, w, z, C, F, I, and t indicated by double black circles. That is. In this case, the hexagonal array antenna has six triangles {a, w, z}, {a, z, C}, {a, C, F}, {a, F, I}, {a, I, t}, {a, t, w}. The points in each triangle may be predicted based on the complex phase of the vertices of each triangle. For example, paying attention to the triangle {a, w, z}, the antenna element a is the reference antenna as before, the rotation amount of the complex phase between the antenna element a and the antenna element w is ζ, and the antenna element a and the antenna element Let η be the amount of rotation of the complex phase between z.

  In this case, for example, when attention is paid to the antenna element on a straight line connecting the antenna element a and the antenna element w, the antenna element c can be approximated to ζ × 1/3, and the antenna element j can be approximated to ζ × 2/3. Similarly, if attention is paid to the antenna element on a straight line connecting the antenna element a and the antenna element z, it can be approximated to η × 1/3 for the antenna element d and η × 2/3 for the antenna element l. For other antenna elements k, ζ × 1/3 + η × 1/3, for antenna element y, ζ × 1/3 + η × 2/3, for antenna element x, ζ × 2/3 + η × 1/3, etc. Is predictable. Other triangles can be similarly predicted. The difference from FIG. 14 is that, in an array antenna having a spatial extension, the influence of multipaths in antenna elements at remote locations weakens the correlation, so that the triangles {a, w, z}, {a, z, C }, {A, C, F}, {a, F, I}, {a, I, t}, and {a, t, w} are individually approximated to arrive at the entire array antenna. Even if the arrival directions of are slightly different, there is a feature that the accuracy of approximation can be improved. Needless to say, there is no essential restriction on the arrangement of the antenna elements, so there is no necessity for the close-packed structure as shown in FIGS. 14 and 15. For example, a square lattice array can be used. .

  Next, a specific example of channel information prediction of a square array antenna configured in a planar shape in the present embodiment will be described with reference to FIGS. In FIG. 16, alphabets a to z and A to J indicated by “◯” or “●” represent antenna elements. Here, an example of a shape in which equilateral triangles are spread in a close-packed form is shown, but any other configuration may be used. For example, it is assumed that the radio station apparatus performs channel estimation of three points of antenna elements a, l, and p indicated by double black circles, and finds the complex phase of the channel information of the three points.

  For example, the antenna element a is a reference antenna, the complex phase rotation amount between the antenna element a and the antenna element l is ζ, and the complex phase rotation amount between the antenna element a and the antenna element p is η. In this case, linear prediction is a little more complicated than in FIGS. 14 and 15, but the basic idea is the same. For example, a square lattice is considered as a lattice point on the xy plane, and the antenna element a is regarded as the origin. At this time, for example, if the coordinates are defined such that the antenna s is (1, 0) and the antenna v is (0, 1) (here, for the sake of convenience, the y-axis has a positive downward direction). In this case, the antenna l corresponds to (5, 3) and the antenna p corresponds to (1, 5).

When the amount of phase rotation is represented by the z axis and expressed in three dimensions, (0, 0, 0) for antenna a, (5, 3, ζ) for antenna l, and (1, 5, η) for antenna p. Become. The equation representing the two-dimensional plane uses a (b−c ₀ ) + b (y−y ₀ ) + c (z−) when coefficients a, b, and c (this coefficient is not related to the antenna element identifier) are used. z ₀ ) = 0. For example, since (0, 0, 0) is a point on the plane, if (x ₀ , y ₀ , z ₀ ) = ( ₀ , ₀ , ₀ ), a relational expression of ax + by + cz = 0 is obtained. Here, (a, b, c) is a normal vector of this plane, and the absolute value of the vector itself has no meaning. Therefore, if a ′ = a / c and b ′ = b / c, the unconstant is There are two relational expressions, a′x + b′y + z = 0. On the other hand, since the coordinates (5, 3, ζ) and (1, 5, η) are on the plane, a binary linear equation for a ′ and b ′ can be established and solved. Thus, the values of a ′ and b ′ can be easily obtained.

If a ′ and b ′ obtained as described above are used, z = − (a′x + b′y). Therefore, if the coordinates of each antenna element are substituted into x and y, the complex phase of each antenna element is obtained. Will be required. For example, since the coordinate for the antenna e is (3, 1), z = − (3a ′ + b ′) is a desired value, and for the antenna r, the coordinate is (5, 5), so z = − ( 5a ′ + 5b ′) is the desired value. In this way, the channel information of the remaining antenna elements can be estimated from the channel information regarding a small number of antenna elements by the same plane wave approximation without depending on the configuration of the antenna arrangement.
It should be noted that a slight supplement is added to the method of using the equation indicating the relationship between the coordinates of each antenna element and the complex phase of each antenna element. In the above description regarding FIG. 14 and FIG. 15, the case where the complex phases of the three antenna elements are obtained and the complex phases of the other antenna elements are obtained therefrom by linear approximation has been described. As a method of handling complex phase information of a, w, z, C, F, I, and t, all antenna elements a to z and A to K may be approximated by one plane wave using the least square method. Is possible. For example, in the two-dimensional plane where the antenna elements a to z and A to K exist, an arbitrary orthogonal x axis and y axis are defined, and the coordinates of the k th antenna element are (x _k , y _k ). The complex phase φ _k of the k antenna element is given by the following equation (9) using the coefficients α, β, and γ.

On the other hand, the complex phase of the k-th antenna element actually estimated with the first antenna element (for example, antenna element a in the figure) as a reference antenna is displayed as ~ φ _k (tilde φ indicates “˜” above φ. (Hereinafter, the same shall be described.), The combination of (α, β, γ) that minimizes the following evaluation function W (α, β, γ) can be obtained by the method of least squares. (Formula (10)).

Based on the combination of (α, β, γ) obtained by the above least square method, the necessary complex phase is calculated from the coordinates (x _k , y _k ) of the k-th antenna element using Equation (9). It becomes possible. In this way, when the least square method is used, an arbitrary number of antenna elements for obtaining the complex phase can be selected. Originally, the present invention approximates an incoming wave with a plane wave. However, since it includes components other than a line-of-sight wave (plane wave) due to the influence of a reflected wave, it has a clean relationship as shown in Equation (9). Must not. The error from this plane affects the estimation accuracy of the complex phase. By increasing the number of antenna elements used for the least-squares method, the effect of this reflected wave can be averaged and estimated. The accuracy can be improved. On the other hand, if the number of antenna elements used for the least squares method is increased, the circuit scale increases. Therefore, the number of antenna elements is set in each trade-off.

  In the above description, the procedure for predicting the complex phase of the channel information is shown. However, after obtaining the complex phase of the channel information, a value obtained by multiplying the complex phase by -1 is the complex phase to be executed by the phase shifter. Since it corresponds to the rotation amount, the rotation amount of the complex phase may be set as the z-axis value, and the calculation process for directly obtaining the rotation amount of the complex phase may be performed.

  In FIG. 16, antenna elements indicated by “◯” corresponding to a to r and antenna elements indicated by “●” corresponding to s to z and A to J are arranged in a nested manner. For example, as described with reference to FIGS. 5 and 11, when the transmission antenna and the reception antenna are separated and arranged in pairs, the antenna elements a to r represented by “◯” are the reception antenna, “●”. If the antenna elements s to z and A to J represented by the above are used as transmitting antennas and are arranged in the vicinity, channel information is acquired by some receiving antennas by the above-described method, and the information is based on the information. If the information of other antenna elements is predicted, the antenna element to be predicted may be a transmission antenna or a reception antenna. Therefore, even in a transmission antenna that cannot actually receive a signal, channel information Can be predicted (acquired). Note that the transmitting antenna and the receiving antenna do not need to be arranged differently at re-adjacent lattice points, and may be other general arrangements.

  FIG. 17 shows another example of channel information prediction of a square array antenna configured in a planar shape according to the third embodiment of the present invention. The difference from FIG. 16 is that in FIG. 16, the transmitting antenna and the receiving antenna are alternately arranged like an Othello cell, whereas in FIG. 17, the transmitting antenna or the receiving antenna is aligned in a vertical column. It is a point that is lined up. In this case, for example, one transmission / reception signal processing circuit 454-n, or one pair of transmission / reception signal processing circuits 555-n, antenna elements 401-n-1 and 441-n-1, or antenna element 501-n- 1 and 541-n-1 may be arranged as adjacent antennas such as antenna elements a and s and antenna elements d and v in FIG.

  FIG. 18 shows another example of channel information prediction of a square array antenna configured in a planar shape according to the third embodiment of the present invention. The difference from FIG. 17 is that the transmitting antenna and the receiving antenna are arranged in a vertical column in FIG. 17, but the transmitting antenna and the receiving antenna are alternately arranged in adjacent columns. In FIG. 18, all the receiving antennas a to r are gathered in one place, and similarly, all the transmitting antennas s to z and A to J are gathered in one place, and each is arranged in a different area. ing. This corresponds to FIGS. 6 and 12 described above.

  The merit of separating the transmission system and the reception system is, for example, to avoid leaking the transmission system noise to the reception system when operating without turning off the power of the transmission high-power amplifier even during signal reception. Since the entire transmission system and the entire reception system are separated, it is easy to ensure mutual isolation. On the other hand, as shown in FIGS. 16 and 17, it is not preferable in terms of configuration to estimate channel information in an antenna element that has not actually acquired channel information by signal reception by the above-described method. However, if the antenna arrangements of the transmitting and receiving antennas of the radio station apparatus # 1 and the radio station apparatus # 2 that oppose each other have a certain symmetry, the channel information acquired on the receiving side is appropriately calibrated. On the assumption that the transmission is performed, it can be handled as it is as channel information on the transmission side. As an example, the receiving antenna element a and the transmitting antenna element s, the receiving antenna element b and the transmitting antenna element t, the receiving antenna element c and the transmitting antenna element u, the receiving antenna element d and the transmitting antenna element v. A range in which plane wave approximation is possible even if the amount of rotation of the complex phase of the receiving antennas a to r is applied to the transmitting antennas s to J as they are in consideration of such symmetry. So it can be considered that there is no big difference.

  Next, a specific example of channel information prediction of an array antenna with separated transmission / reception configured in a planar shape according to the present embodiment will be described with reference to FIGS. 19 and 20. In FIG. 19, the array antenna mounted by the radio station apparatus # 1 is 561, the array antenna mounted by the radio station apparatus # 2 is 562, and the transmission antennas mounted by the radio station apparatus # 1 are 563-1 to 563-2. The reception antennas mounted by the wireless station device # 1 are 564-1 to 564-2, the transmission antennas mounted by the wireless station device # 2 are 565-1 to 565-2, and the wireless station device # 2 is mounted. Assume that the receiving antennas are 566-1 to 566-2.

  The feature of the geometric positional relationship between the array antenna 561 implemented by the radio station apparatus # 1 and the array antenna 562 implemented by the radio station apparatus # 2 shown in FIG. 19 is that each antenna element of the array antenna 561 exists. If the plane and the plane on which each antenna element of the array antenna 562 exists are substantially parallel and face each other, and as a result, the line-of-sight wave is generally dominant, the antenna element 563 of the radio station apparatus # 1 -1 and channel information when received by the antenna element 566-1 of the radio station apparatus # 2, and the antenna element of the radio station apparatus # 2 transmitted from the antenna element 563-2 of the radio station apparatus # 1 The relationship of the relative channel information with the channel information received at 566-2 is transmitted from the antenna element 565-1 of the wireless station device # 2, and the wireless station device Channel information when received by the first antenna element 564-1, and channel when transmitted from the antenna element 565-2 of the radio station apparatus # 2 and received by the antenna element 564-2 of the radio station apparatus # 1 It is considered that it is generally consistent with the relationship of the channel information with the information.

  This is because the relationship between the transmission antenna mounted on the radio station apparatus # 1 and the reception antenna mounted on the radio station apparatus # 2 is symmetrically folded between transmission and reception on a mirror surface, and further translated in parallel. This is due to the similarity between the transmission antenna mounted by 2 and the reception antenna mounted by the radio station apparatus # 1. Therefore, the geometrical correspondence of each antenna element can be translated on the transmitting antenna group to correspond to the receiving antenna group, so that channel information obtained on the receiving side can be utilized on the transmitting side. Become.

  If the transmitting and receiving antennas are not shared and the transmitting antenna group and the receiving antenna group are physically separated as shown in FIG. 18 (and FIG. 19), the geometry as described in FIG. The approximation using the relation of the parallel translation relationship may be less accurate. In this case, it is also possible to cope with it by applying some contrivances.

  FIG. 20 shows another example of channel information prediction of an array antenna having a transmission and reception separated in a planar shape according to the third embodiment of the present invention. 20, only the configuration different from the configuration of FIG. 19 will be described. In FIG. 20, one transmission antenna 567 is included in reception antennas 564-1 to 564-2 in radio station apparatus # 1. In FIG. 20, in the radio station apparatus # 2, three reception antennas 568-1 to 568-3 are included in the transmission antennas 565-1 to 565-2. Note that the calculation of the reception weight of the radio station apparatus # 2 is the same as that in FIG. 19, but slight changes are made to the calculation of the transmission weight of the radio station apparatus # 2.

  Here, the antenna element 567 of the radio station apparatus # 1 is an antenna element in the vicinity of the center of gravity in the reception antenna group, and a training signal can be transmitted from the antenna element 567. The training signal transmitted from the antenna element is received by the receiving antennas of the antenna elements 568-1 to 568-3 of the radio station apparatus # 2, and the above-described means is based on the channel information obtained by these antenna elements. Thus, channel information of each antenna element of the transmission antenna group of the radio station apparatus # 2 can be predicted, and the amount of rotation of the complex phase in the transmission directivity control can be calculated based on the information. These additional functions are unique functions in the overall configuration, but are effective when the antenna scale is enormous or the distance between the transmitting antenna group and the receiving antenna group is increased.

  In FIG. 20, when the amount of rotation of the complex phase related to each antenna element of the transmission antenna group of radio station apparatus # 2 is predicted, the rotation of the complex phase obtained by the receiving antenna elements 568-1 to 568-3 is performed. Although the amount of rotation of other complex phases is predicted based on the amount, it is of course possible to use other patterns of antenna elements. FIG. 21 shows an example of an antenna pattern used for predicting the complex phase rotation amount in the third embodiment of the present invention. FIG. 21 shows an example of four patterns (a) to (d) in a 5 × 5 square array. ● and ◎ indicated by symbols a to y in the figure represent antenna elements, and other complex phases of the antenna elements indicated by ● based on the amount of rotation of the complex phase obtained using the antenna elements indicated by ◎. The amount of rotation is predicted. Here, a case where five antenna elements are used for obtaining the amount of rotation of the complex phase is shown as an example, but it is of course possible to implement with other numbers of elements of 3 or more.

For example, taking FIG. 21A as an example, the amount of rotation of the complex phase is obtained using the antenna elements h, l, m, n, and r, and the correlation between the antenna element m near the center of gravity and other antenna elements is obtained. The complex phase of the correlation value is calculated for the antenna elements h, l, n, and r. The complex phase obtained in this way is set on the z-axis, and the least square method similar to that described in equations (9) and (10) is used, and the relational expression shown in equation (9) is used for each antenna element. The amount of rotation of the complex phase may be estimated. The same applies to FIG.
In addition, in order to expand the spatial extent of the antenna elements used for the prediction of the amount of rotation of the complex phase, elements other than the first proximity element and the second proximity element of the reference antenna are used for channel information prediction. It is also possible to use the pattern of (d). This case is the same as in FIGS. 21A and 21B. For example, in the case of FIG. 21C, the antenna element r is set as a reference antenna, and the antenna element r, the antenna element l, and the antenna element n After calculating the complex phase of the correlation value, for the antenna element f and the antenna element j, instead of directly performing the correlation calculation with the reference antenna element r, the correlation between the antenna element l and the antenna element f is performed. You may perform a calculation and the correlation calculation of the antenna element n and the antenna element j. The sum of the relative value of the complex phase of the antenna element l with respect to the antenna element r and the relative value of the complex phase of the antenna element f with respect to the antenna element l is approximately the relative value of the complex phase of the antenna element f with respect to the antenna element r. Can be considered. Similarly, the sum of the relative value of the complex phase of the antenna element n with respect to the antenna element r and the relative value of the complex phase of the antenna element j with respect to the antenna element n is approximately the complex value of the antenna element j with respect to the antenna element r. It can be regarded as a relative value of the phase.

Furthermore, in order to directly calculate the correlation value between the antenna element r and the antenna elements f and j, the relative value of the complex phase of the antenna element l with respect to the antenna element r in order to remove the uncertainty of the complex phase of 2π period. And the addition value of the relative value of the complex phase of the antenna element f with respect to the antenna element l, and the addition value of the relative value of the complex phase of the antenna element n with respect to the antenna element r and the relative value of the complex phase of the antenna element j with respect to the antenna element n. It may be used in another form. In this case, a value obtained by directly adding a correlation value between the antenna element r and the antenna elements f and j to an integer multiple of 2π, a relative value of the complex phase of the antenna element l with respect to the antenna element r, and the antenna element l The addition value of the relative value of the complex phase of the antenna element f with respect to the antenna element f and the addition value of the relative value of the complex phase of the antenna element n with respect to the antenna element r and the relative value of the complex phase of the antenna element j with respect to the antenna element n are closest. In addition, the correlation value between the antenna element r and the antenna elements f and j may be corrected.
The reason for including the process of calculating and adding the complex phase difference divided into a plurality of stages in this way is that when the complex phase difference between the desired antenna elements is ± π or more, the phase of the complex phase varies due to the periodicity of the complex phase. Since the uncertainty of the 2π period cannot be ignored, the complex phase difference between the adjacent antenna elements is prevented from becoming more than π by adding the complex phases of the correlation values between the adjacent antenna elements. As a result, it becomes possible to avoid the uncertainty of the complex phase of 2π period.
Note that such an antenna pattern is similar to the estimation of the amount of rotation of the complex phase of the receiving antenna, as shown in FIG. 20, in the estimation of the amount of rotation of the complex phase of the receiving antenna as in FIGS. Can be used.

  Hereinafter, the configuration of the transmission / reception signal processing circuit for realizing the third embodiment of the present invention will be described with reference to FIGS. The transmission / reception signal processing circuits in FIG. 22 are the transmission / reception signal processing circuits 451-1 to 451-N in the first embodiment in FIG. 1 and the transmission / reception signal processing circuits 551-1 to 551- in the second embodiment in FIG. This corresponds to another embodiment in which both of the transmission / reception signal processing circuits corresponding to N are combined.

  FIG. 22 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 651-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The transmission / reception signal processing circuit 651-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A / D converter 125-. n, TDD switch (TDD-SW) 127-n, phase shifters 402-n-1 to 402-n-M, switches 403-n-1 to 403-n-M, and a down converter (DC) 424-n-1 to 424-n-2, A / D converters 425-n-1 to 425-n-2, correlation calculation circuit 405-n, phase shift control circuit 406-n, complex phase A rotation amount prediction circuit 410-n. In the figure, the down converter 424-n and the A / D converter 425-n show two configurations. However, the down converter 424-n and the A / D converter 425-n are less than M. Any value may be used. In the following description, there are two down converters 424-n and A / D converters 425-n (down converters 424-n-1 to 424-n-2 and A / D converters 425-n-1 to 425- The case of n-2) will be described as an example.

  The difference from the transmission / reception signal processing circuits 451-1 to 451-N in the first embodiment in FIG. 1 or FIG. 2 is that the down converter 424-n and the A / D converter 425-n are not arranged in all antenna systems. Is a point. For example, the down converter 424-n and the A / D converter 425-n are not arranged in the antenna element 401-2 to the antenna element 401-3. In addition, for example, down converter 424-n, A / D converter 425-n, etc. are provided in three systems (generally, three systems or more and M-1 systems or less) among antenna elements 401-1 to 401-M. If the channel information acquisition procedure described with reference to FIGS. 14 to 17 is performed, the down converter 424-n, the A / D converter 425-n, and the like are not necessary in the system of the remaining antenna elements. is there.

  Hereinafter, a specific signal flow that becomes the difference will be described by paying attention to the difference. The signal processing related to transmission and reception is basically the same as in FIG. 1 and FIG. 2, and the difference is signal processing when calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n -M. Exists only in. The details are shown below.

  In the signal processing when calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M, the switch 403-n-1 includes the down converter 424-n-1 and the phase shifter 402-. n-1 and a switch 403-n-4 connects a down converter 424-n-2 and a phase shifter 402-n-4 (in practice, a down converter not shown here) The same applies to the switch of the antenna system provided with the A / D converter), and the remaining switches are left unconnected. In this state, first, a training signal for channel estimation is transmitted from the radio station apparatus of the communication partner whose complex phase rotation amount is to be acquired, and the radio station apparatus receives this signal.

  Signals received by the antenna elements 401-n-1 to M are input to the phase shifters 402-n-1 to 402-n-M and are analogized by the phase shifters 402-n-1 to 402-n-M. A predetermined complex phase rotation is applied on the signal, and each is input to the switches 403-n-1 to 403-n -M. Switches 403-n-1 and 403-n-4 (actually, the switches of the antenna system having a down converter and an A / D converter not shown here are the same, but the explanation is simplified. Therefore, the signals input to these two antenna systems will be input to the down converters 424-n-1 to 424-n-2. The down converters 424-n-1 to 424-n-2 convert the input radio frequency signals into analog baseband signals. The A / D converters 425-n-1 to 425-n-2 convert analog signals into digital baseband signals.

  Information converted into digital baseband signals by the A / D converters 425-n-1 to 425-n-2 is input to the correlation calculation circuit 405-n. The correlation calculation circuit 405-n calculates the amount of rotation of the complex phase using the expressions (1) to (3) based on the input information. When calibration processing is necessary as necessary, the amount of rotation of the complex phase on the transmission side is determined as a value in consideration of the calibration coefficient in Equations (1) to (3). The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the complex phase rotation amount prediction circuit 410-n together with the identification number of the radio station apparatus that is the communication partner.

  The complex phase rotation amount prediction circuit 410-n predicts channel information of the remaining antenna elements based on channel information of a limited antenna system. In the case of FIG. 22, the complex phase rotation amount prediction circuit 410-n includes antenna elements 401-1 and 401-4 (actually, a down converter and an A / D converter not shown here are provided). The channel information of the remaining antenna elements is predicted based on the channel information received in the system antenna elements). The complex phase rotation amount prediction circuit 410-n calculates the amount of rotation of the complex phase to be performed by the phase shifters 402-n-1 to 402-n-M, along with the identification number of the wireless station apparatus with which it communicates, Input to the phase shift control circuit 406-n. In the phase shift control circuit 406-n, the value input here is managed by being stored in a memory.

  When actual data communication is performed, that is, when transmission or reception processing is performed, a control circuit (not shown in FIG. 22) grasps a wireless station apparatus that is a communication partner, and a phase shift control circuit 406-n. In response to this, the amount of rotation of the complex phase corresponding to the wireless station apparatus that performs communication is instructed to the phase shifters 402-n-1 to 402-n-M, and the amount of rotation of the complex phase is indicated. In 1-402-n-M, the beam forming on analog is realized by setting the rotation amount of this complex phase.

Although not clearly shown in FIG. 22, for example, if a high-power amplifier (HPA) on the transmission side is arranged, it is arranged at a point with a description “A” in the figure, and a low-noise amplifier on the reception side ( LNA) or the like is arranged at a point where “B” and “C ₁ ” to “C ₂ ” are described in the figure. With regard to “A” and “B”, the transmission / reception signal processing circuit 651-n does not assume mutual cooperation. Therefore, calibration processing for removing the uncertainty of the complex phases of the individual high power amplifier and low noise amplifier Is not necessary, but for the low-noise amplifiers described in “C ₁ ” to “C ₂ ”, the antenna elements 401-1 and 401-4 in the same transmission / reception signal processing circuit 651-n are connected to each other. Therefore, it is necessary to remove the uncertainty of complex phase of low-noise amplifiers of each system by using the implicit feedback calibration method of the prior art. .

The present invention can be applied to any method, and a specific method of calibration processing is not limited. Considering this calibration result, for example, if the amount of rotation of the complex phase at “C ₁ ” and “C ₂ ” is +10 degrees and +20 degrees, the complex phase obtained by Expression (1) to Expression (3) The phase rotation amount is adjusted by correcting −10 degrees and −20 degrees with respect to the rotation amount. The information of the calibration result is collected by a calibration circuit (not shown here), and this information is collected by the phase shift control circuit 406-n, the complex phase rotation amount prediction circuit 410-n or the correlation calculation circuit 405-n. Perform correction using the information. In the above description, FIG. 2 and FIG. 4 to FIG. 11 have been described as various variations with respect to the description of FIG. 1 and FIG. 7. It is possible to utilize it.
In this figure, regarding the switches 403-n-2, 403-n-3, 403-n-5 to 403-n-M of the system not provided with the down converter and the A / D converter, the phase shifter 402 -N-1 to 402-n-M In the signal processing when calculating the rotation amount of the complex phase, the connection state is established, or these switches 403-n-2, 403-n-3, 403-n- Even if 5 to 403-n-M is omitted, there is no practical problem.

  FIG. 23 is a functional block diagram showing another configuration example of the transmission / reception signal processing circuit 652-n in the present embodiment. FIG. 23 is a combination of the transmission / reception signal processing circuit 455-n in the first embodiment of FIG. 6 and the transmission / reception signal processing circuit corresponding to the transmission / reception signal processing circuit 556-n in the second embodiment of FIG. This corresponds to another implementation. The same functional blocks as those in the previous figure are given the same numbers.

  The transmission / reception signal processing circuit 652-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A / D converter 125-. n, distribution coupler 414-n, distribution coupler 415-n, phase shifters 402-n-1 to 402-n-M, switches 403-n-1 to 403-n-M, Phasers 409-n-1 to 409-n-M, down converters (DC) 424-n-1 to 424-n-2, A / D converters 425-n-1 to 425-n-2, A correlation calculation circuit 405-n, a phase shift control circuit 406-n, and a complex phase rotation amount prediction circuit 410-n.

  The difference from the transmission / reception signal processing circuit 455-n in the first embodiment in FIG. 6 is that the down converter 424-n and the A / D converter 425-n are not arranged in all antenna systems. For example, the down converter 424-n and the A / D converter 425-n are not arranged in the antenna elements 441-2 to 441-3. Further, for example, down converter 424-n, A / D converter 425-n, etc. are provided in three systems (generally, three systems or more and M-1 systems or less) among antenna elements 441-1 to 441-M. If the channel information acquisition procedure described with reference to FIGS. 14 to 17 is performed, the down converter 424-n, the A / D converter 425-n, and the like are not necessary in the system of the remaining antenna elements. is there.

  The following shows a specific signal flow that becomes the difference by focusing on the difference. Signal processing related to transmission and reception is basically the same as in the description of FIG. 6 (and the description of FIG. 4 related to FIG. 6), and the difference is a complex performed by the phase shifters 402-n-1 to 402-n-M. It exists only in the signal processing when calculating the amount of phase rotation. The details are shown below. In the signal processing when calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M, the switch 403-n-1 includes the down converter 424-n-1 and the phase shifter 402-. n-1 and a switch 403-n-4 connects a down converter 424-n-2 and a phase shifter 402-n-4 (in practice, a down converter not shown here) The same applies to the switch of the antenna system provided with the A / D converter), and the remaining switches are left unconnected.

  The switching of these switches is managed by the correlation calculation circuit 405-n under the instruction of a control circuit not shown here, and the phase shifter 402-n-1 is used except when calculating the rotation amount of the complex phase. ˜402-n-M are connected to the distribution coupler 415-n. Further, when this processing is performed, the phase rotation amounts of the phase shifters 402-n-1 to 402-n-M are set to predetermined values. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference with respect to the initial predetermined value, as described above. Further, in the other transmission / reception signal processing circuits 652-n not shown here, the same processing is performed all together under the control of the control circuit not shown here.

  In this state, first, a training signal for channel estimation is transmitted from the radio station apparatus of the communication partner whose complex phase rotation amount is to be acquired, and the radio station apparatus receives this signal. Signals received by the antenna elements 441-n-1 to 441-n-M are input to the phase shifters 402-n-1 to 402-n-M, and the phase shifters 402-n-1 to 402-n-. At M, a predetermined complex phase rotation is applied on the analog signal, and each is input to the switches 403-n-1 to 403-n-M. Switches 403-n-1 and 403-n-4 (actually, the switches of the antenna system having a down converter and an A / D converter not shown here are the same, but the explanation is simplified. Therefore, the signals input to these two antenna systems will be input to the down converters 424-n-1 to 424-n-2. The down converters 424-n-1 to 424-n-2 convert the input radio frequency signals into analog baseband signals, and the A / D converters 425-n-1 to 425-n-2 provide analog signals. Convert signal to digital baseband signal.

  Information converted into digital baseband signals by the A / D converters 425-n-1 to 425-n-2 is input to the correlation calculation circuit 405-n. The correlation calculation circuit 405-n calculates the complex phase rotation amount using Expressions (1) to (3). When calibration processing is necessary as necessary, the amount of rotation of the complex phase on the transmission side is determined as a value in consideration of the calibration coefficient in Equations (1) to (3). The complex phase rotation amount obtained by the correlation calculation circuit 405-n is input to the complex phase rotation amount prediction circuit 410-n together with the identification number of the radio station apparatus with which it communicates.

  The complex phase rotation amount prediction circuit 410-n predicts channel information of the remaining antenna elements based on channel information of a limited antenna system. In the case of FIG. 22, the complex phase rotation amount prediction circuit 410-n calculates the rotation amount of the complex phase to be performed by the phase shifters 402-n-1 to 402-n-M, and is a wireless device with which to communicate. The information is input to the phase shift control circuit 406-n together with the identification number of the station device. In the phase shift control circuit 406-n, the value input here is managed by being stored in a memory.

  The amount of rotation of the complex phase described above relates to the amount of phase rotation of the phase shifters 402-n-1 to 402-n-M in the reception system, but the amount of complex phase rotation in a low noise amplifier, a high power amplifier, and the like. In order to cancel the individual difference, calibration processing in the implicit feedback of the prior art is performed, the amount of rotation of the complex phase in the transmission system is converted, and values set in the phase shifters 409-n-1 to 409-n-M Similarly, it is managed by being stored in the memory.

  When actual data communication is performed, that is, when transmission or reception processing is performed, a control circuit (not shown in FIG. 23) grasps a wireless station apparatus that is a communication partner, and a phase shift control circuit 406-n. On the other hand, the amount of rotation of the complex phase corresponding to the wireless station apparatus that performs communication is complexed to the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M. The phase rotation amount is instructed, and the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M are set to the complex phase rotation amounts and are Realizes beamforming.

Although not clearly shown in the figure, for example, if a high-power amplifier (HPA) on the transmission side is arranged, the points “A” and “D ₁ ” to “D _M ” in the figure are described. If a low-noise amplifier (LNA) or the like on the receiving side is arranged, it is arranged at a point where “B” and “E ₁ ” to “E _M ” in the figure are described. With regard to “A” and “B”, the transmission / reception signal processing circuit 652-n does not assume mutual cooperation. Therefore, calibration processing for removing the uncertainty of the complex phases of the individual high-power amplifier and low-noise amplifier Is not necessary, but the same transmission / reception signal processing circuit is used for the high-power amplifier having the description of “D ₁ ” to “D _M ” and the low-noise amplifier having the description of “E ₁ ” to “E _M ”. 652-n can cause complex phase uncertainty between each antenna element 401-n-1 to M and antenna element 441-n-1 to 441-n-M. By using the implicit feedback calibration method, it is necessary to remove the uncertainty of the complex phase of the low-noise amplifier of each system.

The present invention can be applied to any method, and a specific method of calibration processing is not limited. Considering this calibration result, for example, assuming that the amount of rotation of the complex phase at “C ₁ ”, “C ₂ ”, etc. is +10 degrees, +20 degrees, etc., it is obtained by Expressions (1) to (3) The amount of phase rotation is adjusted by correcting the amount of rotation of the complex phase by -10 degrees, -20 degrees, or the like. The information of the calibration result is collected by a calibration circuit (not shown here), and this information is collected by the phase shift control circuit 406-n, the complex phase rotation amount prediction circuit 410-n or the correlation calculation circuit 405-n. Perform correction using the information.
Similarly to FIG. 22, phase shifts are made for the switches 403-n-2, 403-n-3, and 403-n-5 to 403-n-M of the system not provided with the down converter and the A / D converter. In the signal processing when calculating the amount of rotation of the complex phase performed by the devices 402-n-1 to 402-n-M, or these switches 403-n-2, 403-n-3, 403- Even if n-5 to 403-n-M are omitted, there is no practical problem.

  Although the above description has focused on the case where the rotation of the complex phase of each antenna element is performed on an analog signal, it can also be used in digital signal processing. In this case, since the A / D converter is mounted on all the antenna elements in the received signal processing, the other antenna elements are evaluated based on the complex phase rotation amount evaluation results performed using a small number of antenna elements. Although there is no direct merit in obtaining the amount of rotation of the complex phase, the signal processing on the transmission side can be effectively used when the transmission antenna and the reception antenna are physically different.

  FIG. 24 is a functional block diagram showing another configuration example of the transmission / reception signal processing circuit 653-n using digital signal processing in the third embodiment of the present invention. FIG. 24 is a diagram for applying the present invention to the transmission / reception signal processing circuits 929-1 to 929 -N in the configuration example of the radio station apparatus using the time-axis beam forming in the prior art shown in FIGS. 28 and 29. It is an improvement. The difference from the transmission / reception signal processing circuit 929-n shown in FIGS. 28 and 29 is that a correlation calculation circuit 405-n, a complex phase rotation amount prediction circuit 410-n, and a transmission weight calculation circuit 411-n are newly added. The transmission weight calculation circuit 411-n provides information related to the transmission weight input to the axis transmission weight multiplication circuit.

  For example, when the transmission antenna and the reception antenna are different as shown in FIGS. 16 and 17, all or some of the A / D converters 925-n-1 to 925-n-M used for reception are used. Sampling data relating to several systems is input to the correlation calculation circuit 405-n, and correlation calculation between the reference antenna element and other antenna elements is performed based on this information. The correlation value is input to the complex phase rotation amount prediction circuit 410-n. Based on the input complex phase and the coordinate information of the antenna element, the complex value of all the antenna elements is calculated by the least square method or the linear estimation method described above. The phase rotation amount is estimated, and the estimation result is input to the transmission weight calculation circuit 411-n. In the transmission weight calculation circuit 411-n, in order to cancel the relative complex phase difference of each antenna element with respect to the reference antenna, a complex number corresponding to Exp (−jψ) is transmitted to the time axis of each antenna element with respect to the complex phase ψ. The weight is input to the time-axis transmission weight multiplication circuit 921-n, and the time-axis transmission weight multiplication circuit 921-n multiplies this value for each antenna element, and then D / A converters 922-n-1 to 922. Input to -n-M. In this way, transmission directivity control using a transmission antenna different from the reception antenna can be performed based on information obtained by some reception antennas even in digital signal processing.

  In FIG. 24, the number of systems input from the A / D converters 925-n-1 to 925-n-M to the correlation calculation circuit 405-n is not clearly shown, but as in FIGS. In addition, a configuration may be adopted in which signals from A / D converters (a part of 925-n-1 to 925-n-M) related to a part of the entire antenna system are input, or a configuration in which all of them are input. I do not care. Furthermore, as shown in FIG. 20, when a part of the antenna elements 568-1 to 568-3 of the antenna element group constituting the transmitting antenna 565 is a receiving antenna, only the output of the corresponding receiving antenna is output. May be input to the correlation calculation circuit 405-n (in this case, the reception antenna may not be used for the reception signal processing of user data).

  As described above, according to this embodiment, the radio station apparatus acquires channel information of some antenna elements without acquiring channel information of all antenna elements. The radio station apparatus calculates weight information from the acquired channel information, and approximates the weight information of other antenna elements based on the calculated weight information. Thereby, it is not necessary to newly provide a switching mechanism such as a switch, and the number of A / D converters and D / A converters necessary for acquiring channel information can be suppressed. Further, as described above, by calculating the weight information of other antenna elements by approximation, it is possible to use implicit feedback even when the transmitting antenna and the receiving antenna are different. Therefore, it is possible to reduce the size and cost, and it is possible to use implicit feedback even when the transmission antenna and the reception antenna are different.

[Fourth Embodiment]
In the conventional implicit feedback technique, a technique for estimating downlink channel information from uplink channel information by utilizing channel symmetry has been common. Therefore, for example, when the transmission antenna and the reception antenna are different, or when the uplink frequency and the downlink frequency are different, it is considered that the implicit feedback utilizing the symmetry of the channel cannot be used. However, as shown in the third embodiment, it is possible to predict a channel even for an antenna element different from the antenna element used for channel feedback, and implicit feedback can be achieved by using such predicted channel information. It has been shown in the third embodiment that this is also possible.

This uses the characteristics of the environment where the sight wave is dominant that the incoming wave can be approximated by a plane wave, and in all cases, when the arrival direction of the plane wave is determined, the path length difference of each antenna element is determined, and depending on the path length difference Thus, the complex phase of the channel information of each antenna element (or the complex phase rotated by the phase shifter, that is, the transmission / reception weight) can be predicted according to the geometric rule. When such a condition is satisfied, considering that the complex phase of 2πΔL / λ rotates using the path length difference ΔL and the wavelength λ, the receiving side (assuming the base station apparatus, here the uplink a rotation amount [Delta] [theta] _UL of complex phase obtained in that), assuming a transmission side (base station apparatus, wherein between the rotational amount [Delta] [theta] _UL of complex phase be determined by the downlink), a predetermined Rules exist.

FIG. 25 is a diagram showing an outline of a complex phase conversion method according to the fourth embodiment of the present invention. In FIG. 25, FIG. 25A shows an incoming wave of frequency F ₁ (wavelength λ ₁ ), and FIG. 25B shows an incoming wave of frequency F ₂ (wavelength λ ₂ ). Here, F ₁ <F ₂ is assumed, and the wavelength λ ₂ is shorter than the wavelength λ ₁ . Even for the same path length difference ΔL, the amount of rotation of the complex phase differs depending on the difference in wavelength or frequency, and the difference can be derived using Equation (6).

  FIG. 26 is a flowchart showing the process flow of the implicit feedback method when FDD is applied in the fourth embodiment of the present invention. First, when the process is started, the training signal transmitted from the transmitting-side radio station apparatus is received by the receiving-side radio station apparatus, and the receiving-side radio station apparatus receives the digital sampling data of the received signal received in a predetermined cycle. (Step S101). The radio station apparatus on the receiving side uses the sampling data for the acquired training signal, performs a correlation operation with the reference antenna according to the equation (1), and calculates the rotation amount of the complex phase from the complex phase of the correlation value (step S102). ). When this is complex phase rotation amount information regarding some antenna elements, the radio station apparatus on the receiving side predicts complex phase rotation amount information of other antenna elements as in the third embodiment (step S103). After that, the receiving-side radio station apparatus predicts downlink complex phase rotation amount information of different frequencies from the uplink complex phase rotation amount information using Equation (6) (step S104). Since the correspondence between the high-power amplifier and the low-noise amplifier differs between the uplink and the downlink, the radio station device on the receiving side acquires the rotation amount information of the downlink complex phase by the implicit feedback technology using the calibration process. (Step S105). Based on the predicted complex phase rotation amount information, the reception-side radio station apparatus acquires the complex phase rotation amount in transmission directivity formation by inverting this complex phase (step S106). The above is the extension method of the implicit feedback method to FDD in the fourth embodiment of the present invention. The rotation amount of the complex phase obtained here is set in the phase shifter when the directivity is formed by analog signal processing, and the complex phase for each antenna element obtained when the directivity is formed by digital signal processing. For ψ, Exp (jψ) is set as the transmission weight of the corresponding antenna element.

In the above description, the rotation amount of the complex phase corresponding to the complex coefficient c _j of the correlation calculation result of the above formula (1) is first obtained in steps S101 to S105 as the application target of the third embodiment. Although the case where the amount of rotation of the complex phase used in the transmission directivity control corresponding to the above equation (2) with the complex phase inverted is obtained later in step S106 has been described as an example, the rotation of the complex phase acquired in the processing of step S102 First, the rotation amount of the complex phase in the reception directivity formation in which the sign is inverted using the amount information is acquired, and the third embodiment is applied to the rotation amount of the complex phase to receive the remaining antenna elements. The rotation amount of the complex phase on the side is calculated first, and the conversion process for different frequencies using the equation (6) is performed as the processing of step S104, and the complex phase rotation amount at the time of forming the transmission directivity is calculated. You may calculate.

  Further, the processing in step S104 described above is performed first if it is performed in the complex phase rotation amount prediction circuit 410-n (n = 1,..., N) in FIGS. 22 and 23 of the third embodiment. Phase shifters 402-n-1 to 402-n-M (in the case of FIG. 22) or phase shifters 409-n-1 to 409-n-M (in the case of FIG. 23) during transmission directivity formation in the above description. It is possible to cope with this by changing the amount of rotation of the complex phase to be set according to the frequency. In addition, if the complex phase rotation amount prediction circuit 410-n or the transmission weight calculation circuit 411-n (n = 1,..., N) in FIG. This can be dealt with by changing the transmission weight set in the time axis transmission weight multiplication circuit at the time of forming the transmission directivity in the description given above according to the frequency. Also in the first embodiment and the second embodiment, the complex phase rotation amount prediction circuit 410 is provided between the correlation calculation circuit 405-n or 505-n and the phase shift control circuit 406-n or 506-n. If -n is arranged and the same processing is performed here, the fourth embodiment can be applied even in combination with the first embodiment and the second embodiment.

  At this time, in FIGS. 1 and 2, the downlink phase is the phase shifters 402-1-1 to 402 -N-M, and in FIGS. 4 to 6, the downlink phase is the phase shifter 409-. n-1 to 409-n-M, and in FIGS. 7 to 9, the downlink phase is phase shifters 502-1-1-1 to 502-NM, and in FIGS. The phase shifters 509-n-1 to 509-n-M perform rotation processing of the converted complex phase. Therefore, in FIG. 1, FIG. 2 and FIG. 7 to FIG. 9, even in the same communication partner station, the amount of rotation of the complex phase of the uplink and downlink should be managed so as to have different values. A series of management is managed by the control circuit 460.

  As described above, according to this embodiment, the radio station apparatus calculates channel information and weight information at a certain frequency in each antenna element and a frequency different from the certain frequency based on a path length difference between certain frequencies. Accordingly, it is possible to use implicit feedback even when a radio system such as FDD having different uplink and downlink frequencies is constructed.

  In each of the embodiments described above, the complex phase is rotated on the radio frequency analog signal. However, the position of the up-converter is changed, and the complex phase is rotated with respect to the baseband or intermediate frequency analog signal. It is good also as a structure which performs frequency conversion with a radio frequency in the latter stage or the front | former stage.

[Additional matters regarding the embodiment]
Supplementary matters regarding the embodiment of the present invention described above will be described below.
You may make it implement | achieve the radio station apparatus in embodiment mentioned above with a computer. In that case, a program for realizing this function may be recorded on a computer-readable recording medium, and the program recorded on this recording medium may be read into a computer system and executed. Here, the “computer system” includes an OS and hardware such as peripheral devices. The “computer-readable recording medium” refers to a storage device such as a flexible medium, a magneto-optical disk, a portable medium such as a ROM and a CD-ROM, and a hard disk incorporated in a computer system. Furthermore, a “computer-readable recording medium” dynamically holds a program for a short time, like a communication line when transmitting a program via a network such as the Internet or a communication line such as a telephone line. In this case, a volatile memory in a computer system serving as a server or a client in that case may be included and a program that holds a program for a certain period of time. Further, the program may be for realizing a part of the functions described above, and may be a program capable of realizing the functions described above in combination with a program already recorded in the computer system. It may be realized using hardware such as PLD (Programmable Logic Device) or FPGA (Field Programmable Gate Array).

  As mentioned above, although embodiment of this invention has been described with reference to drawings, the said embodiment is only the illustration of this invention, and it is clear that this invention is not limited to the said embodiment. is there. Therefore, additions, omissions, substitutions, and other modifications of the components may be made without departing from the technical idea and scope of the present invention.

  The present invention can be applied to a wireless communication device that transmits and receives wireless signals using a plurality of antennas.

120-1 to 120-N ... modulators 122-1 to 122-N, 122-n ... D / A converters 123-1 to 123-N, 123-n ... up-converters 124-1 to 124-N, 124 -N ... down converters 125-1 to 125-N, 125-n ... A / D converters 127-1 to 127-N ... TDD switches 130-1 to 130-N ... demodulator 140 ... baseband signal processing circuit 141 ... Signal separation circuits 401-1 to 401 -M ... antenna elements 401-1-1 to 401 -NM, 401 -n-1 to 401 -nM ... antenna elements 402-1-1 to 402 -N- M, 402-n-1 to 402-nM ... Phase shifters 403-1-1 to 403-NM, 403-n-1 to 403-nM ... Switches 404-1 to 404-N ... Distribution couplers 405-1 to 405-N, 40 -N ... correlation calculation circuits 406-1 to 406-N, 406-n ... phase shift control circuits 407-1 to 407-M ... distribution couplers 408-n-1 to 408-n-M ... TDD switch 409-n -1 to 409-n-M ... phase shifter 410-n ... complex phase rotation amount prediction circuit 411-n ... transmission weight calculation circuit 414-n ... distribution coupler 415-n ... distribution coupler 424-1-1. 424-NM, 424-n-1 to 424-nM, down converters 425-1-1 to 425-NM, 425-n-1 to 425-nM, A / D converter 441 -1 to 441-M ... antenna elements 441-n-1 to 441-nM ... antenna elements 450 ... radio station devices 451-1 to 451-N ... transmission / reception signal processing circuit 452 ... radio station devices 453-n ... transmission / reception Signal processing circuit 454-n ... transmission / reception signal Processing circuits 455-n ... reception signal processing circuit 460 ... control circuit 901-1~901-N ... modulator 902 ... precoder _903-1~903-N 0 ... IFFT & GI imparting circuit _904-1~904-N 0 ... D / A converters 905-1 to 905-N ₀ ... up converters 906-1 to 906-N ₀ ... down converters 907-1 to 907-N ₀ ... A / D converters 908-1 to 908-N ₀ ... GI removal & FFT circuit 909... Post coder 910-1 to 910 -N... Demodulator 911... TDD switch 912-1 to 912 -N ₀ ... distribution coupler 913-1-1 to 913 -N ₀ -M ₀ ... phase shifter 915 -1~915-M 0 _... distributor coupler 916-1~916-M 0 _... antenna elements 921-1~921-N ... time base transmission weight multiplying circuit 922-1～922- ... D / A converters 922-1-1 to 922-NM ... D / A converters 923-1 to 923-M ... Up converters 923-1-1 to 923-NM ... Up converters 924-1 924-M... Down converters 924-1-1 to 924-NM. Down converters 925-1 to 925-M... A / D converters 925-1-1 to 925-N. Units 926-1 to 926 -N: Time axis reception weight multiplication circuits 927-1 to 927 -N: TDD switches 928-1 to 928 -M: Antenna elements 928-1-1 to 928 -NM: Antenna elements 929 -1 to 929-N ... transmission / reception signal processing circuits 941-1 to 941-M ... distribution coupler 942 ... radio station apparatuses 943-1 to 943-M ... addition synthesizers 944-1 to 944-M ... duplicator 945 ... Radio station apparatus 951- 1-951-3 High power amplifiers 952-1 to 952-3 Low noise amplifiers 953-1 to 953-3 TDD switches 954-1 to 954-3 Antenna elements 955-1 to 955-3 Wireless modules

Claims (4)

A wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with other wireless communication devices,
A communication unit that communicates with the other wireless communication device using a first frequency used in the uplink and a second frequency different from the first frequency used in the downlink;
A signal conversion unit that is provided in all antenna elements or a part of a plurality of systems, and converts a radio frequency analog signal received by the antenna element into a baseband digital signal;
Using a digital signal corresponding to a training signal transmitted by the other radio communication device converted by the signal conversion unit, a reference antenna selected from a plurality of antenna elements provided with the signal conversion unit A correlation calculation unit for calculating a correlation for each combination of the element and another antenna element;
A rotation amount calculation unit that calculates a rotation amount of a complex phase to be given to a reception signal of an individual antenna element based on a correlation calculation result by the correlation calculation unit;
A complex phase to be given to a transmission signal of an individual antenna element based on information on a ratio between the first frequency and the second frequency and information calculated by the correlation calculation unit or the rotation amount calculation unit A rotation amount prediction unit for predicting the rotation amount of
Based on the complex phase predicted by the rotation amount prediction unit, a phase rotation amount management unit that manages the rotation amount of the complex phase to be given to the transmission signal for each antenna element;
A transmission signal generator for generating a signal addressed to the other wireless communication device;
A phase rotation unit that rotates the amount of rotation of the complex phase managed by the phase rotation amount management unit on an analog signal or a digital signal with respect to a transmission signal for each antenna element used for transmission;
A signal transmission unit that transmits the signal output via the phase rotation unit to the other wireless communication device via the antenna element;
A wireless communication device comprising: A first signal distribution unit that branches the transmission signal generated by the transmission signal generation unit for each antenna element on an analog signal;
The phase rotation unit is
The radio communication apparatus according to claim 1, wherein the rotation amount of the complex phase managed by the phase rotation amount management unit is set in a phase shifter provided for each antenna element. A second signal distribution unit that branches the transmission signal generated by the transmission signal generation unit for each antenna element on a digital signal;
The phase rotation unit is
The wireless communication apparatus according to claim 1, wherein a coefficient Exp (jψ) is multiplied for each antenna element on a digital signal by a complex phase rotation amount ψ for each antenna element managed by the phase rotation amount management unit. A wireless communication method executed by a wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with other wireless communication devices,
A communication step of communicating with the other wireless communication device using a first frequency used in the uplink and a second frequency different from the first frequency used in the downlink;
Transmitted by the other radio communication device provided in all antenna elements or some of a plurality of systems and converted by a signal conversion unit that converts a radio frequency analog signal received by the antenna element into a baseband digital signal Correlation for calculating a correlation for each combination of an antenna element serving as a reference selected from a plurality of antenna elements provided with the signal conversion unit and other antenna elements using a digital signal corresponding to a training signal A calculation step;
A rotation amount calculation step for calculating a rotation amount of a complex phase to be given to a reception signal of an individual antenna element based on a correlation calculation result in the correlation calculation step;
Based on the information on the ratio between the first frequency and the second frequency and the calculation result of the correlation calculation step or the rotation amount calculation step, the complex phase to be given to the transmission signal of the individual antenna element A rotation amount prediction step for predicting the rotation amount;
Based on the complex phase predicted in the rotation amount prediction step, a phase rotation amount management step for managing the rotation amount of the complex phase to be given to the transmission signal for each antenna element;
A transmission signal generation step of generating a signal addressed to the other wireless communication device;
The rotation amount of the complex phase managed in the phase rotation amount management step is set in the phase rotation unit that rotates the phase on an analog signal or a digital signal with respect to the transmission signal for each antenna element used for transmission, and the phase rotation A signal transmission step of transmitting a signal output via the unit to the other wireless communication device via the antenna element;
A wireless communication method.

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