JP6712240B2 - Wireless communication device and wireless communication method - Google Patents

Description

The present invention relates to a wireless communication device and a wireless communication method.

[Background surrounding 5th generation mobile communication]
Currently, high-performance mobile communication terminals such as smartphones have been explosively popularized. Regarding mobile phones, the third-generation mobile communication has shifted to the fourth-generation mobile communication, and at present, further research and development on the fifth-generation mobile communication (commonly referred to as “5G”) are underway. One of the studies conducted on this 5G is the use of macro cells and small cells.

In the conventional mobile phones, one service area is set to a radius of several kilometers, and one base station device covers this macrocell area. However, there is an extremely large number of users in such a macro cell. Since the entire limited system capacity is shared by each user, when accommodating a huge number of users, the throughput for each individual user decreases.

In order to avoid such a decrease in throughput, a technology has been developed for setting a small cell, which is a very small service area with a radius of several tens of meters, in a densely populated area where traffic is concentrated. This technology uses small cells to offload spot-like traffic to the network without going through macro cells. Here, it is assumed that the terminal device is capable of simultaneously using the communication capacity of the small cell and the communication capacity of the macro cell. By using such a terminal device, the user data is accommodated on the small cell side while exchanging information on the control information by utilizing the macro cell. This makes it possible to maximize the advantages of macro cells and small cells.

In 5G mentioned above, the target value of the transmission rate is set to 10 Gbit/s (gigabit per second) or higher, and even in this small cell, the same large capacity communication is performed to realize efficient offload of traffic. There is a need to. In the macro cell, it is premised that the microwave band of low frequency is used to allow long-distance propagation. However, considering the current state of the microwave band, where frequency resources are already exhausted, small cells that are supposed to communicate at relatively short distances are expected to use the quasi-millimeter wave band or millimeter wave band with relatively high frequencies. Has been done. The characteristic of this high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. Therefore, it is preferable that the small cell base station is ideally installed near the user terminal. For example, in a place such as a rooftop of a building where installation is easy, the distance between the user terminal and the base station becomes too large, which is not preferable in terms of circuit design.

On the other hand, since the small cells are set in a place where traffic is concentrated, there are cases where installation of a base station device is strongly desired even in a place where it is difficult to install optical fibers. For example, assuming a small cell base station device is installed in a place where many people are crowded, such as in front of a station such as Shinjuku or Shibuya, propagation attenuation will be large on the roof of a building adjacent to such a place. .. Therefore, it may be required to be installed on a place lower than the roof of the building, for example, on the wall of the building. However, it may be difficult to lay an optical fiber on the wall of an existing building, and in such a case, it may be necessary to provide a backhaul line to the base station device using a wireless line. is there.

When providing such a backhaul line, in order to support the large capacity transmission of 10 Gbit/s or more required in a small cell, it is necessary to similarly utilize the millimeter wave band to perform large capacity transmission of 10 Gbit/s or more. There is. In such an environment, since the opposing wireless station devices are fixedly installed in stable places, it is natural that the line-of-sight is naturally ensured and the directional antennas face each other. is there. In this case, although reflected waves between buildings and the like exist to some extent, most of the received signals are line-of-sight components, and it is expected that this is unlikely to be a multipath environment. This situation is the same for the access system as long as the base station device for the small cell is installed at a high place such as a wall surface of a building and the user looks down from above and is used in a generally line-of-sight environment.

Next, for large-capacity transmission at 10 Gbit/s or higher, which is the transmission speed required for 5G, it is possible to use frequency resources with an extremely wide bandwidth by utilizing the millimeter wave band, which makes it possible to achieve this. Is rising. For example, assuming a backhaul line using the millimeter wave band, if the E band (71 to 76 GHz and 81 to 86 GHz) is used as an example and the bandwidth of 1 GHz is used, the frequency utilization efficiency is 10 bits. /S/Hz will suffice. However, existing wireless equipment for achieving a frequency utilization efficiency of 10 bits/s/Hz generally employs spatial multiplexing transmission using MIMO (Multiple-Input Multiple-Output) channels. Spatial multiplex transmission generally utilizes a multipath environment, and when performing singular value decomposition of a channel matrix H that represents the transfer function of a MIMO channel in a matrix format, the distribution of the absolute values of the singular values obtained as a result. Represents the characteristics of the spatial multiplexing transmission. Specifically, the square value of the absolute value of the singular value is a value proportional to the signal-to-noise power ratio SNR (Signal to Noise Ratio), and not only the first singular value but also the first singular value for spatial multiplexing transmission. Communication cannot be established unless it has a sufficiently large value even after the two singular values. The same applies to large-capacity transmission in small cells that are access systems, but realization of spatial multiplex transmission in such an environment in which line-of-sight waves are dominant is indispensable for realizing the target wireless system. ..

As described above, in 5G, it is expected to use the millimeter wave band for both the access system and the backhaul line. As described above, the characteristic of the high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. For example, if the frequency of the 2 GHz band (existing access system) and the 80 GHz band (future system using the millimeter wave band) are compared with each other, the frequency is 40 times, so that the propagation attenuation is 1600 times and the line gain of 32 dB is insufficient. It will be. Of course, it is not necessary to cover as much as 32 dB because it does not need to cover a wider area than a macro cell, but on the other hand, in the high frequency band, there is no device with high output in the high power amplifier in the transmission stage, so this point is also taken into consideration. Then, it is considered that additional line gain must be secured at the level of several tens of dB. Furthermore, since spatial multiplexing transmission is also expected in such an environment, a radio system including much more antenna elements than in the conventional technique has been studied in both the base station and the terminal station. .. Such a technique is called Massive MIMO. Below, the related art regarding Massive MIMO will be introduced.

In Non-Patent Document 1, a base station side is provided with an antenna having 256 elements and a terminal station side is provided with an antenna having 16 elements, and a channel matrix having a size of 256×16 is utilized to aim at spatial multiplexing transmission of 16 streams. In this Non-Patent Document 1, in order to transmit a plurality of signal sequences (streams), directivity formation is performed by analog beamforming using rotation of a complex phase amount in a wireless analog circuit and by digital baseband circuit. Performed in combination with digital beamforming in the digital domain. As a result, the analog/digital (A/D) converter and the digital/analog (D/A) converter are avoided from being heavily used, and power consumption is reduced and a channel gain shortage measure at the time of channel information feedback is taken. However, for the analog beam formed here, for example, a plurality of fixed patterns are set in both the horizontal direction and the vertical direction with a predetermined step width (for example, at intervals of 5 degrees), and an analog beam having a relatively large received power is set in the predetermined pattern. Select and use only the number. As a matter of course, since the analog beam cannot be accurately directed to the direction of arrival of the line-of-sight wave, selecting a redundant number of analog beams can sufficiently compress the required number of A/D converters and D/A converters. I can't.

[Expansion of Massive MIMO technology when line of sight is dominant]
In the above description of the Massive MIMO technology, it is assumed that the environment is generally a multipath environment because the usage in the access system is mainly assumed. However, even in the case of an access system, if the small cell base station is installed above, and it is possible to secure a general view by looking down, it may be forced to operate in a non-multipath environment. Especially in the case of a backhaul line, this is remarkable, and it is expected to be used in an environment where the so-called Rice coefficient K is 10 dB or more and the multipath component is accompanied by only about 1/10 or less of the line-of-sight component. In this case, the difference between the line gain corresponding to the first singular value and the line gain corresponding to the second singular value or more is expected to be 20 dB or more, and spatial multiplexing transmission of two or more streams is substantially inefficient. Is expected.

In such an environment, as shown in Non-Patent Document 2, all antenna elements are divided into a plurality of sets by utilizing the high efficiency of the line gain corresponding to the first singular value, and a sub array is set for each set. The configuration is effective. Then, by arranging the subarrays spatially separated from each other, it is effective to reduce the correlation between the subarrays and perform parallel transmission of the transmission corresponding to the first singular value with low correlation. Similarly, in Non-Patent Document 3, when the antenna aperture length of the sub array is narrow, the line-of-sight wave is dominant, and the correlation between the antenna elements in the sub array is sufficiently strong, the transmission/reception weight of each antenna element is A "time-axis beamforming technique" has been proposed in which it can be treated as a constant having no dependency, and in this case, weight multiplication can be performed in units of sampling data on the time-axis.

This is a relative channel information for each antenna element (a relative value of the channel information with respect to the channel information of a certain reference antenna element, specifically, the reference antenna when the element spacing is narrow and the correlation between the antenna elements is strong. When the complex phase of the channel information of the kth frequency component of the element is ψ _ref ^(k) , the frequency dependence of the complex phase of the information obtained by multiplying each antenna element by Exp{-jψ _ref ^(k) } The sex is a method resulting from the fact that it is almost constant. For example, at the time of reception, the complex phase of the reception weight for combining the received signals of these antenna elements so that the complex phase becomes the same phase is almost constant over the entire frequency band. It is possible to use reception weights with the same constant in. In general, when a function that is a constant on the frequency axis is Fourier-transformed into a δ function, the weight obtained by converting the reception weight on the frequency axis on the time axis by IFFT only needs to consider the component at t=0. Become. In other words, since signal processing considering the delayed wave component is not required, the sampling data obtained by sampling the analog baseband received signal with the A/D converter is directly added to the time axis which is a predetermined coefficient for each antenna element. By multiplying the reception weights, it becomes possible to realize directivity formation only by signal processing on the time axis without converting the received signal into a signal on the frequency axis even once by FFT processing or the like.

The coefficient for rotating the complex phase to be multiplied as the time axis weight is obtained by the following equations (1) to (3).

In the above equation, S _i (n) represents the sampling data of the n-th sample of the i-th antenna in the received training signal, and S _i (n) ^* is the complex conjugate of S _i (n). Represent N _FFT assumes a predetermined periodicity, and indicates a periodicity value that has meaning in correlation detection such as the number of FFT points in OFDM. ψ _j is the amount of rotation of the complex phase (on the receiving side) performed by time-axis beamforming. The function angle(x) is a function representing the complex phase of the complex number x, and is a value determined by the ratio of the real number part of x and the imaginary number part of x, and the sign of the real part and imaginary part. Further, in the correlation calculation in Expression (1), the correlation calculation is performed over N _FFT samples, which is the number of FFT points in the case of an OFDM signal, as the predetermined periodicity, but even if it is an integer multiple of N _FFT , for example. Correlation calculation may be performed over other numbers of samples so that the periodicity is maintained.

Here, as is apparent from the equation (2), the complex phase of the complex coefficient c _j given by the above equation (1) and the complex phase of the time axis weight w _j given by the above equation (2) are inverted in sign. It has become a thing. In this sense, in the background art and embodiments of the present invention described later, the complex phase of the complex coefficient c _j given by the equation (1) corresponding to the relative channel information is obtained, and the complex of the time axis weight w _j is calculated. Determining the phase is equivalent.

FIG. 6 is a functional block diagram showing a configuration example (sub-array separation type) of a wireless station device using time-axis beamforming in the conventional technique described in Non-Patent Document 3. The wireless station device shown in the figure includes a baseband signal processing circuit 140 and transmission/reception signal processing circuits 929-1 to 929-N. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD#1 to MOD#N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM#1 to DEM#). N) and a control circuit 460. The transmission/reception signal processing circuit 929-n (n=1,..., N) includes a time axis transmission weight multiplication circuit 921-n, D/A converters 922-n-1 to 922-n-M, and an up converter 923. -N-1 to 923-n-M, down converter (DC) 924-n-1 to 924-n-M, A/D converters 925-n-1 to 925-n-M, time axis A reception weight multiplication circuit 926-n and a TDD switch (TDD-SW) 927-n are provided. The TDD switch 927-n is connected to the antenna elements 928-n-1 to 928-n-M. Here, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the transmission/reception signal processing circuits 929-1 to 929-N are mounted for N systems as a whole. Further, M represents the number of antenna elements of the sub-array mounted on each of the transmission/reception signal processing circuits 929-1 to 929-N.

Here, the transmission/reception signal processing circuits 929-1 to 929-N (antenna elements 928-n-1 to 928-n-M of the sub array are attached to the transmission/reception signal processing circuit 929-n) are described in Non-Patent Document 2 It is assumed that they will be installed spatially apart from each other. The baseband signal processing circuit 140 is wire-connected to each of the transmission/reception signal processing circuits 929-1 to 929-N, and the digital baseband signal is transferred on the wire. The control circuit 460 manages the frame cycle and the transmission/reception timing, and the switching of the TDD switches 927-1 to 927-N is also managed here.

Furthermore, the time-axis beamforming technology basically presupposes signal processing on the time axis, but even when signals on the frequency axis are formed as in the OFDM modulation method, signals on the frequency axis are processed by FFT processing and IFFT processing. Can be converted into a signal on the time axis, and by performing the signal processing on the time axis signal, the time axis beamforming technique can be similarly applied to the OFDM modulation method as well as the single carrier transmission. However, since the signal processing on the frequency axis is not presupposed here, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-in FIG. 6 even if the OFDM modulation method or the like is used. It is assumed that the N (or the signal separation circuit 141) has the functions of the FFT processing and the IFFT processing inside, and the description thereof is omitted. Therefore, it is assumed that the input/output signals from the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are signals on the time axis regardless of the OFDM modulation method or single carrier transmission. .. Further, the signal separation circuit 141 performs signal separation between the respective signal sequences, but here it is also possible to perform signal separation on the time axis, and once it is converted to a frequency axis signal by FFT and then on the frequency axis. Frequency-dependent signal separation may be performed. In this sense, the input to the demodulators 130-1 to 130-N may be a time axis signal or a frequency axis signal, but here, for simplicity, the time axis signal is used. Will be described as inputting.

The specific signal flow is as follows. First, signal transmission will be described. The modulators 120-1 to 120-N generate time-axis digital baseband transmission signals of the respective streams that are spatially multiplexed, and input the respective signals to the time-axis transmission weight multiplication circuits 921-1 to 921-N. To do. The time axis transmission weight multiplication circuits 921-n (n=1,..., N) are subarrays for directivity formation of the digital signal input from the modulator 120-n by the transmission/reception signal processing circuit 929-n. The transmission weight corresponding to the antenna elements 928-n-1 to 928-n-M is converted into a digital signal multiplied by the transmission weight. The D/A converters 922-n-1 to 922-n-M convert the digital signals multiplied by the transmission weights into analog baseband signals, and the up converters 923-n-1 to 923-n-M are , Converts analog baseband signals to radio frequency band signals. At the time of transmission, the TDD switch 927-n connects the up converter 923-n-m (m is an integer of 1 or more and M or less) to the antenna element 928-n-m. The respective signals input from the up converters 923-n-1 to 923-n-M are transmitted from the respective antenna elements 928-n-1 to 928-n-M, and the transmission/reception signal processing circuits 929-1 to 929 are provided. A directional beam is formed for each N.

Next, signal reception will be described. The signals received by the antenna elements 928-n-1 to 928-n-M (n=1,..., N) are input to the down converters 924-n-1 to 924-n-M via the TDD switch 927-n. To be done. The down converters 924-n-1 to 924-n-M convert radio frequency signals into analog baseband signals. The A/D converters 925-n-1 to 925-n-M convert the analog baseband signal into a digital baseband signal. This digital baseband signal is input to the time axis reception weight multiplication circuit 926-n. The time-axis reception weight multiplication circuit 926-n multiplies each input signal by the reception weight corresponding to each antenna element 928-n-1 to 928-n-M, and adds and synthesizes the signals after the reception weight multiplication. Then, each is converted into one signal sequence. In other words, the time axis reception weight multiplication circuits 926-1 to 926-N convert the signals into a total of N signal sequences (streams), and these signals are input to the signal separation circuit 141. The signal separation circuit 141 suppresses crosstalk components between the streams to perform signal separation, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The demodulators 130-1 to 130-N reproduce and output data by a predetermined signal detection process.

The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be performed on the frequency axis by once converting into a frequency axis signal by FFT processing. Alternatively, only the signal processing performed by the transmission/reception signal processing circuits 929-1 to 929-N is sufficient, and the signal separation circuit 141 does not need to perform any processing. However, in any case, the details of the signal separation method here are not directly related to the present application, and therefore omitted.

The time axis transmission weights used in the time axis transmission weight multiplication circuits 921-1 to 921-N and the time axis reception weights used in the time axis reception weight multiplication circuits 926-1 to 926-N are shown here. It is acquired by the time axis transmission/reception weight acquisition means. Then, similarly, a control circuit not shown here manages the value of the time axis transmission/reception weight used there. For example, based on the sampling data acquired by the A/D converters 925-n-1 to 925-n-M for the training signal transmitted by the wireless station that is the communication partner, the reference antenna element is used for a predetermined number of samples. The correlation value between the antenna element and (for example, 928-n-1) may be obtained, and the correlation value may be determined based on this complex phase. The values of the complex phase of the time axis reception weight and the time axis transmission weight do not generally match because the amount of rotation of the complex phase of the power amplifier and the low noise amplifier, which are not shown here, are different in each amplifier. By using the implicit feedback calibration method, it is possible to convert the time axis reception weight to the time axis transmission weight. The transmission/reception weights thus obtained are stored in the memory for each corresponding wireless station device. Then, at the time of transmission and reception, the time axis transmission weight multiplication circuits 921-1 to 921-N and the time axis reception weight multiplication circuits 926-1 to 926-N perform weight multiplication based on the values of these transmission and reception weights. It will be.

FIG. 7 is a functional block diagram showing a configuration example (sub-array shared type) of a wireless station device using time-axis beamforming in the conventional technique described in Non-Patent Document 3. In the same figure, the same figure number is given to the function common to FIG. In the figure, the wireless station device 942 includes a baseband signal processing circuit 140, transmission/reception signal processing circuits 929-1 to 929-N, distribution couplers (HYB) 941-1 to 941-M, and an antenna element 928-. 1-928-M. In FIG. 6, the transmission/reception signal processing circuits 929-1 to 929-N are housed in different housings on the assumption that they are installed in spatially separated locations for each subarray, and the baseband signal processing circuit in a different housing. The configuration shown in FIG. On the other hand, in FIG. 7, all the transmission/reception signal processing circuits 929-1 to 929-N and the baseband signal processing circuit 140 are configured as the wireless station device 942 in the same housing, and the antenna elements 928-1 to 928-M are entirely configured. It is shared with. Therefore, for example, at the time of transmission, the signals from the TDD switches 927-1 to 927-N of the transmission/reception signal processing circuits 929-1 to 929-N are combined by the distribution couplers 941-1 to 941-M and combined. The transmitted signal is transmitted from the antenna elements 928-1 to 928-M. Similarly, at the time of reception, the signals received by the antenna elements 928-1 to 928-M are distributed by the distribution couplers 941-1 to 941-M. That is, the distribution coupler 941-m (m=1,..., M) distributes the signal received by the antenna element 928-m to the TDD switches 927-1 to 927-N and inputs it. All other signal processing is common to FIGS. 6 and 7.

Here, in actual operation, the wireless station device shown in FIG. 6 and the wireless station device shown in FIG. 7 face each other and perform communication. For example, the base station device has a degree of freedom in installation like a rooftop of a building, and subarrays can be installed in a plurality of places. On the other hand, on the terminal station device side, when there are large restrictions on the installation of building wall surfaces, etc., the terminal station device shares the sub-array with one array antenna by configuring the base station device in FIG. 6 and the terminal station device in FIG. It is possible to increase the degree of freedom of installation in the form of. Alternatively, for example, if the transmission capacity per terminal station device does not require spatial multiplexing, only one system of the base station device in FIG. 7 and the transmission/reception signal processing circuits 929-1 to 929-N in FIG. , A transmission/reception signal processing circuit 929-1 of FIG. 6 is assumed to be mounted, and a configuration may be adopted in which multi-user MIMO transmission is performed by a plurality of terminal station devices and one base station device. It is possible.

[Calibration technology for channel information feedback]
Generally, when directivity is formed using a plurality of antenna elements on the transmitting side, it is necessary to feed back channel information of MIMO channels, including the techniques of Non-Patent Documents 1 to 3 described above. At this time, if the number of antenna elements becomes enormous, the amount of channel information to be fed back becomes enormous, so that various measures are required. In the Massive MIMO system as described above, in order to obtain the channel information of the forward link in the transmission direction, the channel information of the reverse link in the reception direction is used, and the complex phase of the received signal generated by the circuit such as the low noise amplifier used at the time of reception is used. And the amount of rotation of the complex phase of the transmission signal generated by a circuit such as a high-power amplifier used at the time of transmission are converted, and the channel information of the reverse link is multiplied by a predetermined calibration coefficient It is possible to obtain channel information. Generally, these techniques are known as implicit feedback techniques (see Non-Patent Document 4, for example). Although details of the calibration process are omitted here, the amount of rotation of the complex phase of the reception signal generated by a circuit such as a low noise amplifier used at the time of reception using any conventional technique and the amount of rotation caused by a circuit such as a high power amplifier used at the time of transmission It is assumed that the individual difference and uncertainty of the rotation amount of the complex phase of the transmission signal for each antenna system can be eliminated.

In the wireless transmission system using the high frequency band such as the millimeter wave band described above, there are problems regarding the A/D converter, the D/A converter, and the up converter and the down converter as described below. To do.

In the time-axis beamforming technology described in the above “Expansion of Massive MIMO technology when line-of-sight waves dominate”, A/D converters, D/A converters, up-converters, and down-converters are provided by the number of antenna elements. You will need it. Further, in the configuration example shown in FIG. 7, these are required for the number of antenna elements×the number of streams. Generally, the power consumption of the A/D converter and the D/A converter used in an environment with a very high clock speed due to the wide band tends to increase, and as a result, the amount of heat generated is also at a level that cannot be ignored. In the case of a device that generates a large amount of heat, a heat radiating plate or the like for radiating the heat is required, and these are physically required to have a size proportional to the amount of heat generated. In addition to the design requirements of the circuit itself, it is also difficult to miniaturize the A/D converter and the D/A converter themselves that constantly consume power because of the necessity of heat radiation processing for the heat generation amount expected during operation. And increase in size.

Originally, in a system assuming use of the millimeter wave band, the wavelength of the antenna element itself becomes shorter as the frequency increases, and it is possible to reduce the size of the antenna element alone in proportion to this wavelength. For example, the size of the antenna element can be designed to be sufficiently small with respect to the wavelength, and when the antenna elements are arranged at about 1/2 wavelength, the size of the entire array antenna should be designed to be small even if the number of antenna elements is large. Will be possible. For example, assuming the E band and assuming that the center frequency is 80 GHz, one wavelength is 3.75 mm. If an array antenna of 16×16=256 elements having a square array structure with a wavelength interval of 1×16 is configured, the size of the entire array antenna is about 6 cm×6 cm. However, in order to realize the mounting in this size, the A/D converter and the D/A converter themselves are configured to be within one wavelength of 3.75 mm square or less, and the amount of heat generated during operation is expected. A heat sink for heat dissipation must also be placed in this. If this is not possible, the actual size of the array antenna depends on the sizes of the A/D converter and the D/A converter, and the size of the heat sink for radiating the heat. For example, if a wide band signal having a bandwidth of 1 GHz is to be handled, it is necessary to perform the sampling process in the base band at a super speed of at least 1 GHz. The higher the speed, the higher the power consumption and the amount of heat generated, and eventually the size cannot be reduced. In this way, despite the Massive MIMO technology that can be miniaturized because of the high frequency band, the situation is that miniaturization cannot be achieved due to the magnitude of power consumption. For these reasons, reduction of the power consumed by the A/D converter and the D/A converter is a big issue.

Furthermore, in addition to power consumption, ultra-high-speed A/D converters and D/A converters are also expensive in price. Therefore, if the number of antenna elements is 100, the cost will be further reduced. It is required to reduce the total number of A/D converters and D/A converters. This also applies to the up-converter and the down-converter, and mounting the up-converter and the down-converter for the number of antenna elements leads to an increase in cost. Further, since the local oscillators input to the up-converter and the down-converter need to eliminate the uncertainty of the complex phase in order to form the directivity, sharing of the local signal is required. The effects of loss due to enormous distribution of these high-frequency signals for each antenna element and mutual pre-interference between systems (leakage of signals on board wiring) cannot be ignored. In particular, in order to avoid mutual pre-interference between each system, it is ideal to separate the signals of each system from the physical distance, but for that purpose, the physical size of the array antenna is large. Lead to change.

Therefore, in order to make the array antenna of each radio station apparatus smaller, even when the time-axis beamforming is used, it is consumed by the A/D converter and the D/A converter mounted as much as possible. Power needs to be reduced.

Further, as described above, in the Massive MIMO technique using the high frequency band, the antenna size can be physically reduced due to the short wavelength, and the overall size of the device can be reduced. At this time, for example, when using the TDD switch or the like to switch between the transmission system and the reception system while sharing the transmission/reception antenna, the TDD switch avoids the transmission signal from leaking into the reception system and ensures sufficient isolation. There is a need. However, as the scale of the device becomes smaller, the TDD switch also needs to have a simple structure. In order to secure the isolation between the transmitting and receiving systems, there is also a method of increasing the isolation by making the switch multistage, but this is not suitable for miniaturization because the physical size of the multistage switch is required at the time of mounting. .. In order to avoid such a problem, it is assumed that the transmitting antenna and the receiving antenna are physically separated, but when the transmitting antenna and the receiving antenna are different, the channel of the implicit feedback Since the symmetry is broken, there is a problem that implicit feedback cannot be used.

In Non-Patent Document 5 and Non-Patent Document 6, as a technique for solving such a problem, based on the result of channel estimation using some antenna elements, the complex phase of all other antenna elements is calculated. A method of acquiring the rotation amount information or the information about the transmission/reception weight has been proposed (see Non-Patent Documents 5 and 6).

In the time-axis beamforming technique, for example, at the time of reception, the main path of the incoming wave between the transmitting and receiving stations is extracted, and signal processing for directing the directional gain of the antenna element group in that direction is performed. The transmission/reception weight used at that time is a constant having no frequency dependence, and as a result, signal processing is reduced at various points. However, since it is based on digital signal processing, each antenna element is individually multiplied by the transmission/reception weight in digital signal processing, and accompanying this, individual A/D (analog/digital) conversion is performed for each antenna system. Device and D/A (digital/analog) converter.

However, the multiplication processing of the coefficient having no frequency dependency does not necessarily require digital signal processing, if it is limited to, for example, the rotation processing of the complex phase without the change of the amplitude. Specifically, by using a phase shifter which is an analog circuit and passing an analog signal through this phase shifter set to a predetermined complex phase rotation, signal processing substantially equivalent to weight multiplication processing is performed. Can be realized. In Non-Patent Document 1, directivity control is realized by using a phase shifter. This is, for example, for the phase rotation amount of each antenna element for each predetermined direction set in increments of 5 degrees in the horizontal/vertical direction. The combination set is determined in advance, and the phase rotation amount of each phase shifter is set according to the direction in which the beam obtained by some control procedure should be directed. However, this combination set of phase rotation amounts was selected from a preset menu, and it was necessary to search for the direction with the highest reception level while transmitting the training signal individually for each direction. .. However, in time-axis beamforming, the phase rotation amount of each antenna element is optimized based on the training signal transmitted from the terminal station device side, and therefore the calculation of the phase rotation amount used for directivity formation is simply fed back. In addition, the combination set of the rotation amount of the complex phase can be set with a remarkably high degree of freedom without requiring a menu in advance.

Therefore, a wireless station device (wireless communication device) in the related art adopts digital assist type analog beamforming. That is, the wireless station device performs the calculation processing of the rotation amount of the complex phase performed by each phase shifter by digital signal processing, and controls the phase shifter using the value of the complex phase obtained by the digital signal processing. By doing so, the desired complex phase is rotated and directivity is formed on the analog signal.

First, the time-axis beamforming corresponds to the control of directing the directivity to the path having the maximum intensity of the incoming wave. Therefore, in the plane wave approximation of the incoming wave from the direction of the path with the maximum intensity, the length of the perpendicular line drawn from the planarly arranged antenna element to the wave front of the plane wave (the plane where the complex phases of the signals are in phase) is It becomes a path length difference, and the path length difference has a geometric periodicity and is given by a function of coordinates. Further, the transmission/reception weight becomes a coefficient corresponding to the rotation of the complex phase that cancels the path length difference.

Of course, in reality, since the multipath component is involved and a measurement error is also included, the rotation amount of the complex phase for each antenna element is slightly deviated from the clean periodicity. However, approximately, the x-axis and the y-axis are set on the plane where the antenna is arranged, and three-dimensional notation is performed with the z-axis as the amount of rotation of the complex phase with respect to the coordinate points of the antenna element. When plotting the amount of rotation of the complex phase with respect to the coordinates, all the plot points are on one plane in the plane wave approximation. Note that the rotation amount ψ of the complex phase is completely equivalent to the complex number Exp (−j{ψ±2π×integer}) even if ±2π×integer is added. Depending on the coordinate of the element, it may be considered that it exists on the plane as a value obtained by adding ±2π×integer, but if such a complex phase offset is taken into account, the rotation amount of the complex phase is calculated with a small number of antenna elements. The remaining antenna elements can approximately obtain the rotation amount of the complex phase by linear interpolation processing based on the path length difference.

In addition, due to the design of the device, even if the uplink/downlink channel asymmetry that separates the transmitting antenna and the receiving antenna is involved, the rotation amount of the complex phase can be approximately obtained by considering this path length difference. It is possible to do so. In a system using a plurality of antenna elements, it is common to install the antenna elements at a distance of half a wavelength or more in order to avoid coupling between the antenna elements. Conversely speaking, it is possible to operate with an element interval of about half a wavelength at the shortest, but in a high frequency band system with a short wavelength, the physical distance between antenna elements is extremely narrowed and mounted. Expected to do. For example, assuming the 80 GHz band, the wavelength is about 3.75 mm, and the antenna elements may be arranged at intervals of several millimeters. Here, when transmitting and receiving are switched in time division and the transmitting and receiving antennas are shared, a high power amplifier for transmission and a low noise amplifier for reception are mounted in this interval. When is close to, the signal from the high power amplifier may leak to the low noise amplifier side.

Also, the switch for switching in time division is small enough to fit in this size, but it is necessary to ensure sufficient isolation between the transmitting side and the receiving side, which is a severe condition. Furthermore, when introducing a time division switch, the signal amplified by the high power amplifier is input to the antenna element via the time division switch, so the transmission output of the high power amplifier passes through the time division switch. You will lose only the passing loss by doing. Similarly, since the signal received by the antenna passes through the time-division switch before being input to the low-noise amplifier, this also causes a loss corresponding to the passage loss due to passing through the time-division switch. The SNR characteristic deteriorates. In particular, when the time divisional switch is composed of multiple stages of switches in order to secure transmission/reception isolation, this passage loss becomes large, and this effect cannot be ignored. From such a background, a device may be designed by separating a transmitting antenna and a receiving antenna, and a countermeasure corresponding to the asymmetry of the uplink/downlink channel is required.

However, even in such a case, if the path length difference is ΔL and the wavelength is λ, the amount of rotation of the complex phase for canceling the path length difference is common (2πΔL/λ) for both transmission and reception. Given in. At this time, if the rotation amount of the complex phase in at least three antenna elements is obtained, the rotation amount of the complex phase of the remaining antenna elements can be calculated from the rotation amount of the complex phase of each antenna element coordinate on the plane including the three points. Can be calculated. If the complex phase rotation amount can be calculated with four or more antenna elements, the z-axis coordinate of the plane in the three-dimensional space (complex phase rotation amount) given by the coordinates of each antenna element and the complex phase rotation amount can be calculated. Calculate the plane with the smallest sum of squared errors by the method of least squares that minimizes the error with the rotation amount of the complex phase, and find the rotation amount of the complex phase for each antenna coordinate on that plane. Alternatively, this may be set as a phase shifter to deal with it.

Even if the uplink and downlink (that is, transmission/reception) antenna elements have different configurations as described above, the uplink antenna element and the downlink antenna element are mixed to form the entire antenna configuration. When taking, the amount of rotation of the complex phase of that antenna element is calculated in the uplink receiving antenna, and the amount of rotation of the complex phase of the uplink when it is assumed that other downlink antenna elements are also used in the uplink. It is also possible to calculate, and it is possible to calculate the amount of complex phase rotation of the downlink antenna element by performing calibration processing based on the amount of phase rotation of the antenna element in the uplink.

Or, even if the uplink antenna element group and the downlink antenna element group are physically separated, the uplink antenna element group and the downlink antenna element group are simply parallel to each other. When there is a moving relationship (that is, when the relative positional relationship of each antenna element does not change between the transmitting and receiving antennas on the same plane), the amount of rotation of the downlink complex phase obtained by the receiving antenna is transmitted as is. Even if it is used as the complex phase of the corresponding antenna element on the side, if the distance between the transmitting and receiving stations is sufficiently large and a common plane wave approximation is expected, it is possible to obtain a sufficiently approximate directivity. It is expected that formation is possible.

As shown in FIG. 6 and FIG. 7, a "sub array separation type" configuration in which a directional beam is formed by being divided into a plurality of sub arrays, and a "sub array shared use" that realizes a plurality of directional beams in one array Although there are variations in the configuration based on the "type" (strictly, it may be understood as an "integrated array" because it is not divided into subarrays), the "subarray separation type" will be described here.

FIG. 8 shows an outline of complex phase rotation amount prediction in a linear array in which antenna elements are linearly arranged in the related art. In the figure, a state is shown in which the antenna elements 401-1 to 401-5 are linearly arranged. First, let us consider a plane wave coming from the direction of the angle θ with respect to the front direction of the linear array. The distance between the elements of each antenna is d. Here, the received signal at the antenna element 401-1 is Φ1(t), the received signal at the antenna element 401-2 is Φ2(t),..., The received signal at the antenna element 401-5 is Φ5(t), The path length difference of the sth antenna element (s is an integer of 2 or more) based on the antenna element 401-1 is ΔLs. For convenience, ΔL1 is 0.

Generally, when the wavelength is λ and the path length ΔL is passed, the complex phase is rotated by 2πΔL/λ. Assuming the plane wave approximation, the path length difference between the antenna element 401-1 and the antenna element 401-2 is ΔL2=d·Sinθ. Similarly, the path length difference between the antenna element 401-1 and the antenna element 401-3 is ΔL3=2×d·Sinθ, and the path length difference between the antenna element 401-1 and the antenna element 401-4 is ΔL4=3. ×d·Sinθ, the path length difference between the antenna element 401-1 and the antenna element 401-5 is ΔL5=4×d·Sinθ.

Therefore, if attention is paid to the frequency component of a certain wavelength λ, Φ2(t)≈Exp{-2πjΔL2/λ}Φ1(t), Φ3(t)≈Exp{-2πjΔL3/λ} in a wave that satisfies the plane wave approximation. Φ1(t), Φ4(t)≈Exp{-2πjΔL4/λ}Φ1(t), Φ5(t)≈Exp{-2πjΔL5/λ}Φ1(t). If the above-mentioned relation of ΔLs=(s−1)d·Sinθ is used, Φs(t)≈Exp{−2πj(s−1)d·Sinθ/λ}Φ1(t), and each antenna element has This means that the complex phase is rotated by Exp{−2πjd·Sin θ/λ}. Therefore, for example, if channel estimation is performed by the antenna element 401-1 and the antenna element 401-5, and 1/4 of the difference in the complex phase is rotated for each antenna element based on the rotation amount of the complex phase during this period. It becomes possible to predict.

Similar prediction is possible not only when the antenna elements are arranged one-dimensionally but also when they are arranged two-dimensionally. Next, a specific example of the prediction of the rotation amount of the complex phase of the array antenna configured in a plane according to the conventional technique will be described with reference to FIG. In FIG. 9, alphabets a to z and A to K indicated by “◯” represent antenna elements. Here, an example of a shape in which equilateral triangles are spread in a closest packing shape is shown, but the shape may be any other configuration.

For example, it is assumed that the wireless station device performs channel estimation of three points of antenna elements i, m, and q indicated by double black circles, and obtains the complex phase of the channel information of the three points. Here, for example, the antenna element i is used as a reference antenna, the complex phase rotation amount between the antenna element i and the antenna element m is ζ, and the complex phase rotation amount between the antenna element i and the antenna element q is η. .. In this case, paying attention to an antenna element on a straight line connecting the antenna element i and the antenna element m, the antenna element c is ζ×1/3, the antenna element d is ζ×2/3, and the antenna element B is ζ×4/. 3, the antenna element u can be approximated to −ζ×1/3. Similarly, focusing on the antenna element on the straight line connecting the antenna element i and the antenna element q, η×1/3 for the antenna element b, η×2/3 for the antenna element g, and η×4/ for the antenna element G. 3 and the antenna element v can be approximated as −η×⅓. If this is expanded, two-dimensional prediction is also possible. For example, if the antenna element a is ζ×1/3+η×1/3, and if the antenna element H is −ζ×1/3+η×4/3, The antenna element E can be predicted as ζ×2/3+η×3/3.

As a condition for enabling the above prediction, the rotation amount of the complex phase needs to be π or less between the two antenna elements among the antenna elements that acquire the rotation amount information of the complex phase. For example, taking the case of FIG. 8 as an example, even if the complex phase difference between the antenna element 401-1 and the antenna element 401-5 is π/2 or −3π/2, Exp{ Since j·π/2}=Exp{−j·3π/2}, it cannot be distinguished. If the complex phase difference is π/2, the complex phase difference between the antenna element 401-1 and the antenna element 401-2 should be π/2×(1/4). Is −3π/2, the complex phase difference between the antenna element 401-1 and the antenna element 401-2 should be −3π/2×(1/4).

The complex phase difference between the antenna elements that obtain the complex phase rotation amount information must be π or less because the complex phase rotation amount information cannot be predicted correctly under the above-mentioned uncertainty. There is. In the actual measurement, fluctuations in the complex phase are expected due to measurement errors due to noise and the effects of multipath. Therefore, in reality, the complex phase difference needs to be a small value with a margin larger than π. The reference value cannot be generally stated because it depends on the magnitude of the reflected wave, but if the antenna element spacing d is small or the arrival angle θ is sufficiently small, the path length difference ΔL becomes small. Therefore, the amount of change in the complex phase due to the path length difference can be made sufficiently small. In the example of FIG. 9, the three points of the antenna elements i, m and q are in a relatively distant positional relationship, but the mutual element spacing is small in order to make the complex phase difference between the antenna elements π or less. It is also possible to use adjacent antenna elements (for example, antenna elements a, e, f, etc.). Note that FIG. 9 shows an example in which the channel is estimated by three antenna elements and the complex phases of the remaining antenna elements are approximated in a two-dimensional plane. In the case of acquiring, it becomes possible to predict the rotation amount information of a slightly more complex phase. Of course, since there is no essential restriction on the arrangement of the antenna elements, it is not necessary to have the close-packed structure as shown in FIG. 9, and for example, a square lattice array can be used.

Next, a specific example of the complex phase rotation amount information prediction of the planar array square array antenna according to the related art will be described with reference to FIGS. Although details will be described later, here, description will be made assuming a configuration in which a transmitting antenna and a receiving antenna are separated. In FIG. 10, a to r represented by “◯” represent reception antenna elements, and s to z and A to J represented by “●” represent transmission antenna elements. Here, an example is shown in which the antenna elements are arranged in a square lattice shape instead of the shape in which the equilateral triangles are spread in the closest packing shape. For example, it is assumed that the wireless station device performs channel estimation of three points of antenna elements a, l, and p indicated by double circles, and obtains the complex phase of the channel information of the three points.

For example, the antenna element a is used as a reference antenna, the complex phase rotation amount between the antenna element a and the antenna element l is ζ, and the complex phase rotation amount between the antenna element a and the antenna element p is η. In this case, linear prediction is slightly more complicated than that in FIG. 9, but the basic idea is the same. For example, the square lattice is considered as a lattice point on the xy plane, and the antenna element a is regarded as the origin. At this time, for example, if the coordinates of the antenna s are defined as (1, 0) and the antenna v is defined as (0, 1) (here, the y-axis is a positive direction in which the downward direction is a positive direction). In this case, the antenna 1 corresponds to (5,3) and the antenna p corresponds to (1,5).

If the amount of phase rotation is represented by the z-axis and expressed three-dimensionally, the antenna a is (0,0,0), the antenna l is (5,3,ζ), and the antenna p is (1,5,η). Become. The equation expressing the two-dimensional plane is a(x−x ₀ )+b(y−y ₀ )+c(z− if the coefficients of a, b, and c (this coefficient is not related to the identifier of the antenna element) are used. It is represented by z ₀ )=0. For example, since (0,0,0) is a point on the plane, if (x ₀ , y ₀ , z ₀ )=( ₀ , ₀ , ₀ ), the relational expression of ax+by+cz=0 is obtained. Here, (a, b, c) is a normal vector of this plane, and the absolute value of the vector itself has no meaning. Therefore, if a′=a/c and b′=b/c, the unconstant is It becomes two, and the relational expression of a′x+b′y+z=0 is obtained. On the other hand, since the coordinates (5, 3, ζ) and (1, 5, η) are on the plane, it is possible to set up a binary linear simultaneous equation for a′ and b′, and solve this. The values of a'and b'can be easily obtained with.

Using a'and b'determined as described above, z=-(a'x+b'y), so if the coordinates of each antenna element are substituted for x and y, the complex phase of each antenna element Will be obtained. For example, for antenna e, the coordinates are (3,1), so z=-(3a'+b') is the desired value, and for antenna r, the coordinates are (5,5), so z=-( 5a'+5b') is the desired value. By doing this, it is possible to estimate the rotation amount information of the complex phase of the remaining antenna elements from the rotation amount information of the complex phase for a small number of antenna elements by the same plane wave approximation without depending on the configuration of the antenna arrangement. It is possible.

A little supplement will be added to the method of utilizing the equation showing the relationship between the coordinates of each antenna element and the complex phase of each antenna element. In the above description with reference to FIG. 9, the case where the complex phases of the antenna elements at three points are obtained and the complex phases of the other antenna elements are obtained by linear approximation therefrom is described. It is also possible to approximate all the antenna elements a to z and A to K with one plane wave by using the multiplication method. For example, when an arbitrary orthogonal x-axis and y-axis are defined in the two-dimensional plane where the antenna elements a to z and AK exist, and the coordinates of the k-th antenna element are (x _k , y _k ), It is assumed that the complex phase φ _k of the k antenna element is given by the following equation (4) using the coefficients α, β and γ.

On the other hand, the complex phase of the k-th antenna element actually estimated using the first antenna element (for example, the antenna element a in the figure) as a reference antenna is represented by ~φ _k (tilde φ represents "~" above φ). In the following, the same description will be made.) The following combination of (α, β, γ) that minimizes the evaluation function W(α, β, γ) can be obtained by the least squares method. (Formula (5)).

Based on the combination of (α, β, γ) obtained by the above least squares method, the required complex phase is calculated from the coordinates (x _k , y _k ) of the k-th antenna element using equation (4). It will be possible. When the least squares method is used as described above, the number of antenna elements for obtaining the complex phase can be selected arbitrarily. Originally, the present invention approximates an incoming wave with a plane wave, but in reality, due to the influence of the reflected wave, the components other than the line-of-sight wave (plane wave) are included, so that a clear relationship such as equation (4) is obtained. Don't The error from this plane affects the estimation accuracy of the complex phase, but by increasing the number of antenna elements used in the least squares method, it becomes possible to average the effect of this reflected wave. The accuracy can be improved. On the other hand, since increasing the number of antenna elements used in the least squares method increases the circuit scale, the number of antenna elements must be set at each trade-off.

In the above description, the procedure for predicting the complex phase of the channel information is shown. However, after obtaining the complex phase of the channel information, the value obtained by multiplying the complex phase by -1 is the complex phase of the phase shifter. Since it corresponds to the amount of rotation, the amount of rotation of the complex phase may be set as the value of the z axis and the arithmetic processing for directly obtaining the amount of rotation of the complex phase may be performed.

In FIG. 10, receiving antenna elements indicated by “◯” corresponding to a to r and transmitting antenna elements indicated by “●” corresponding to s to z and A to J are arranged side by side in a nested manner. For example, if the transmitting antenna and the receiving antenna are separated and they are arranged relatively close to each other, the rotation amount information of the complex phase is acquired by some receiving antennas by the above-mentioned method, and based on the information. If the information of other antenna elements is predicted, the antenna element to be predicted may be a transmitting antenna or a receiving antenna, so even if the transmitting antenna that cannot actually receive a signal has a complex phase It becomes possible to predict the rotation amount information. The transmitting antenna and the receiving antenna do not have to be arranged differently between re-adjacent lattice points, and other general arrangements may be used.

FIG. 11 shows another example of a transmission/reception antenna arrangement of a square array antenna configured in a plane according to the related art. The difference from FIG. 10 is that the transmission antennas and the reception antennas are alternately arranged like the grid of Othello in FIG. 10, whereas the transmission antennas or the reception antennas are arranged in a vertical column in FIG. That is the point. In this case, for example, the antenna elements forming a pair of one transmission signal processing circuit and one reception signal processing circuit may be arranged as adjacent antennas such as the antenna elements a and s and the antenna elements d and v in FIG. It will be.

FIG. 12 shows another example of the arrangement of transmitting and receiving antennas of a square array antenna configured in a plane according to the related art. The difference from FIG. 11 is that in FIG. 11, the transmitting antennas and the receiving antennas are aligned in a vertical row, but the transmitting antennas and the receiving antennas are alternately arranged in the adjacent rows. 12, in FIG. 12, all the receiving antennas a to r are arranged in one place, and similarly, all the transmitting antennas s to z and A to J are also arranged in one place, and each of them is arranged in a different area. ing.

The merit of separating the transmission system and the reception system is to prevent the noise of the transmission system from leaking to the reception system when operating without turning off the power of the high power amplifier of the transmission system even when receiving a signal, for example. By separating the entire transmission system and the entire reception system, it is easy to ensure mutual isolation. On the other hand, as shown in FIGS. 10 and 11, in order to estimate the rotation amount information of the complex phase in the antenna element which is not actually acquiring the rotation amount information of the complex phase by the signal reception, It is not structurally preferable. However, if the antenna arrangements of the transmitting antenna and the receiving antenna of the opposing wireless station device #1 and wireless station device #2 have some symmetry, the rotation amount of the complex phase acquired on the receiving side (channel information ) Can be treated as it is as the rotation amount information of the complex phase on the transmission side on the assumption that an appropriate calibration process is performed. As an example, in FIG. 12, the receiving antenna element a and the transmitting antenna element s, the receiving antenna element b and the transmitting antenna element t, the receiving antenna element c and the transmitting antenna element u, the receiving antenna element d and the transmitting antenna element v,... Is in a geometrically parallel positional relationship, and even if the rotation amounts of the complex phases of the receiving antennas a to r are directly applied to the transmitting antennas s to J in consideration of such symmetry, a plane wave approximation is obtained. It can be considered that there is no great difference within the possible range.

It should be noted that the antenna element used for predicting the rotation amount of the complex phase can be selected not only by using the antenna elements at relatively distant positions as shown in FIG. 9 and FIG. It is also possible to use. FIG. 13 shows an example of an antenna pattern used for predicting the rotation amount of a complex phase in the related art. FIG. 13 shows an example of four patterns (a) to (d) in a 5×5 square array. In the figure, the symbols a to y indicate ● and ◎, which represent antenna elements, and the complex phase of the other antenna elements indicated by ● based on the amount of rotation of the complex phase obtained using the antenna element indicated by ◎. Predict the amount of rotation of. Here, the case where five antenna elements are used to obtain the amount of rotation of the complex phase is shown as an example, but it is of course possible to use other number of elements of 3 or more.

For example, taking FIG. 13A as an example, the rotation amount of the complex phase is acquired using the antenna elements h, 1, m, n, and r, and the antenna element m near the center of gravity and the other antenna elements h and l are acquired. , N, r are calculated, and the complex phase of the correlation value is obtained for the antenna elements h, 1, n, r. The complex phase thus obtained is set on the z-axis, the least squares method similar to that described in equations (4) and (5) is used, and each antenna element of the relational expression shown in equation (4) is used. The amount of rotation of the complex phase may be estimated. The same applies to FIG. 13(b).

In addition, in order to extend the spatial spread of the antenna element used for predicting the rotation amount of the complex phase, elements other than the first proximity element and the second proximity element of the reference antenna are used for predicting the rotation amount information of the complex phase. It is also possible to use patterns 13(c) and 13(d). This case is similar to the case of FIGS. 13A and 13B, but in the case of FIG. 13C, for example, the antenna element r is set as the reference antenna, and the antenna element r, the antenna element l, and the antenna element n are set. After obtaining the complex phase of the correlation value with respect to the antenna element f and the antenna element j, instead of directly performing the correlation calculation with the reference antenna element r, the correlation between the antenna element l and the antenna element f is calculated. The calculation and the correlation calculation of the antenna element n and the antenna element j may be performed. The sum of the relative value of the complex phase of the antenna element l with respect to the antenna element r and the relative value of the complex phase of the antenna element f with respect to the antenna element l is approximately the relative value of the complex phase of the antenna element f with respect to the antenna element r. It is possible to see. Similarly, the sum of the relative value of the complex phase of the antenna element n with respect to the antenna element r and the relative value of the complex phase of the antenna element j with respect to the antenna element n is approximately the complex value of the antenna element j with respect to the antenna element r. It can be regarded as a relative value of the phase.

Furthermore, the correlation value between the antenna element r and the antenna elements f and j is directly calculated, while the relative value of the complex phase of the antenna element l with respect to the antenna element r is removed in order to remove the uncertainty of the complex phase of 2π period. And an added value of the relative value of the complex phase of the antenna element f to the antenna element l, and an added value of the relative value of the complex phase of the antenna element n to the antenna element r and the relative value of the complex phase of the antenna element j to the antenna element n. You may use it in another form. In this case, a value obtained by adding an integral multiple of 2π to the value obtained by directly calculating the correlation value between the antenna element r and the antenna elements f and j, the relative value of the complex phase of the antenna element l with respect to the antenna element r, and the antenna element l. So that the sum of the relative values of the complex phase of the antenna element f with respect to, and the sum of the relative value of the complex phase of the antenna element n with respect to the antenna element r and the relative value of the complex phase of the antenna element j with respect to the antenna element n are closest to each other. Alternatively, the correlation value between the antenna element r and the antenna elements f and j may be corrected.

In this way, the reason for including the process of calculating and adding the complex phase difference in multiple stages is that if the complex phase difference between the desired antenna elements is ±π or more, the phase is due to the periodicity of the complex phase. Since the uncertainty of the 2π period cannot be ignored, the complex phase difference between the antenna elements used for estimating the rotation amount of the complex phase is π by adding and using the complex phase of the correlation value between the adjacent antenna elements. It is possible to avoid the above, and as a result, it is possible to avoid the uncertainty of the complex phase of 2π period.

FIG. 14 is a diagram showing a configuration example of a communication system in the related art. In FIG. 14, 450 and 452 are wireless station devices. Here, for example, a configuration is shown in which the wireless station device 450 corresponding to the base station device communicates with the wireless station device 452 corresponding to the terminal station, but there are a plurality of wireless station devices 452 corresponding to the terminal stations, and at the same time. It is also possible to perform spatial multiplexing transmission on the same frequency.

Here, the wireless station device 450 shown in the figure includes a baseband signal processing circuit 140 and transmission/reception signal processing circuits 651-1 to 651-N. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD#1 to MOD#N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM#1 to DEM#). N) and a control circuit 460. In the wireless station device 450, each of the plurality of transmission/reception signal processing circuits 651-1 to 651-N forms one beam as a sub array. On the other hand, in the wireless station device 452, the (sub)array is shared and one array antenna forms a plurality of beams. In actual operation, as shown in FIG. 14, the wireless station device 450 faces the wireless station device 452, so that signals of N systems can be spatially multiplexed. The transmission/reception signal processing circuits 651-1 to 651-N have a common configuration, a signal from the modulator 120-n is input, and an output is output to the signal separation circuit 141.
Hereinafter, the configurations of the transmission/reception signal processing circuits 651-1 to 651-N according to the related art will be described with reference to FIGS. 15 and 16.

FIG. 15 is a functional block diagram showing a configuration example of a transmission/reception signal processing circuit 651-n (n=1,..., N) in the conventional technique shown in Non-Patent Document 5.
The transmission/reception signal processing circuit 651-n shown in the figure includes a D/A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A/D converter 125-n. n, a TDD switch (TDD-SW) 127-n, a phase shifter 402-n-1 to 402-n-M, a switch 403-n-1 to 403-n-M, and a distribution coupler 404-. n, down converters (DC) 424-n-1 to 424-n-2, A/D converters 425-n-1 to 425-n-2, correlation calculation circuit 405-n, and phase shift control The circuit 406-n and the complex phase rotation amount prediction circuit 410-n are provided.

The antenna elements 401-1 to 401-M and the baseband signal processing circuit 140 are connected to the transmission/reception signal processing circuit 651-n. Although the down converter 424-n and the A/D converter 425-n have two configurations in the figure, the down converter 424-n and the A/D converter 425-n have a value smaller than M. Any value will do as long as it is available. In the following description, there are two down converters 424-n and A/D converters 425-n (down converters 424-n-1 to 424-n-2 and A/D converters 425-n-1 to 425-n. The case of n-2) will be described as an example.

A down converter 424-n, an A/D converter 425-n, and the like are arranged in three systems (generally three systems or more and M-1 system or less) of the antenna elements 401-1 to 401-M. If the procedure of acquiring the rotation amount information of the complex phase described in FIGS. 9 to 11 is executed, the down converter 424-n and the A/D converter 425-n are effectively unnecessary in the system of the remaining antenna elements. is there.

The specific signal flow is shown below. The signal processing is roughly divided into three types, that is, signal processing, signal reception processing, and signal transmission processing when calculating the rotation amount of the complex phase performed by the phase shifters 402-n-1 to 402-n-M. And explain. The control circuit 460 shown in FIG. 14 is a circuit that manages the entire communication, and although not described in detail, manages the entire timing management and various processes, for example, switching of the TDD switch 127-n and the like, and phase shifting. It manages a series of management such as management of setting timing of the devices 402-n-1 to 402-n-M. Therefore, strictly speaking, signals may be input/output to/from each functional block, but they are omitted here for simplification.

First, the signal processing when calculating the rotation amount of the complex phase performed by the phase shifters 402-n-1 to 402-n-M will be described. In the signal processing for calculating the rotation amount of the complex phase, the switch 403-n-1 connects the down converter 424-n-1 and the phase shifter 402-n-1, and the switch 403-n-4 is down. The converter 424-n-2 and the phase shifter 402-n-4 are connected (actually, a down converter and an antenna system switch equipped with an A/D converter which are not shown here are also the same), The remaining switches are not connected. In this state, first, a training signal for channel estimation is transmitted from the wireless station device of the communication partner that should acquire the rotation amount of the complex phase, and the wireless station device receives this signal.

The signals received by the antenna elements 401-n-1 to 401-M are input to the phase shifters 402-n-1 to 402-n-M, and are analogized by the phase shifters 402-n-1 to 402-n-M. A predetermined complex phase rotation (which may be 0 degree, for example) is added to the signal, and each is input to the switches 403-n-1 to 403-n-M. Switches 403-n-1 and 403-n-4 (actually, a switch of an antenna system including a down converter and an A/D converter, which are not shown here, are also the same, but the explanation is simplified. Therefore, the signals input to the two antenna systems will be described) are input to the down converters 424-n-1 to 424-n-2. The down converters 424-n-1 to 424-n-2 convert the input radio frequency signal into an analog baseband signal. The A/D converters 425-n-1 to 425-n-2 convert analog signals into digital baseband signals.

The information converted into the digital baseband signal by the A/D converters 425-n-1 to 425-n-2 is input to the correlation calculation circuit 405-n. The correlation calculation circuit 405-n calculates the rotation amount of the complex phase using the equations (1) to (3) based on the input information. Further, when the calibration process is required as necessary, the rotation amount of the complex phase on the transmission side is determined as a value in which the calibration coefficient is considered in the equations (1) to (3). The rotation amount of the complex phase calculated by the correlation calculation circuit 405-n is input to the complex phase rotation amount prediction circuit 410-n together with the identification number of the wireless station device that is a communication partner.

The complex phase rotation amount prediction circuit 410-n predicts the complex phase rotation amount information of the remaining antenna elements based on the complex phase rotation amount information (or channel information) of the limited antenna system. In the case of FIG. 15, the complex phase rotation amount prediction circuit 410-n includes antenna elements 401-1 and 401-4 (actually, a down converter and an A/D converter which are not shown here are provided). The same applies to the antenna elements of the system), and the complex phase rotation amount information of the remaining antenna elements is predicted based on the complex phase rotation amount information acquired. The complex phase rotation amount prediction circuit 410-n calculates the rotation amount of the complex phase to be performed by the phase shifters 402-n-1 to 402-n-M, and together with the identification number of the wireless station device that is a communication partner, It is input to the phase shift control circuit 406-n. The phase shift control circuit 406-n manages the value input here by storing it in a memory. It should be noted that the control circuit 460 performs overall adjustment, timing control, and the like in a series of these processes.

Next, signal processing when actual data communication is performed, that is, when transmission or reception processing is performed will be described. First, when performing data communication, all of the switches 403-n-1 to 403-n-M are connected to the distribution coupler 404-n. In addition, the control circuit 460 (not shown) grasps the wireless station device as the communication partner, and notifies the phase shift control circuit 406-n of the rotation amount of the complex phase corresponding to the wireless station device that performs communication. Instruct the phase shifters 402-n-1 to 402-n-M.

First, the signal processing when transmitting a signal will be described. At the time of transmission, the TDD switch 127-n transmits a signal with the up converter 123-n and the distribution coupler 404-n connected (n=1,..., N). The modulator 120-n (not shown) generates a time-axis digital baseband transmission signal of each stream to be spatially multiplexed and inputs the transmission signal to the transmission/reception signal processing circuit 651-n. The transmission/reception signal processing circuit 651-n (n=1,..., N) receives the transmission signal of the time base digital baseband generated by the modulator 120-n (not shown).

The D/A converter 122-n of the transmission/reception signal processing circuit 651-n (n=1,..., N) converts the transmission signal input from the modulator 120-n into an analog baseband signal and up. Input to the converter 123-n. The up converter 123-n converts the signal input from the D/A converter 122-n from a baseband signal into a radio frequency band signal, and inputs the signal to the TDD switch 127-n. The TDD switch 127-n inputs the signal input from the up converter 123-n to the distribution coupler 404-n.

The distribution coupler 404-n branches the analog signal input from the TDD switch 127-n into an analog signal of M system. The branched analog signal is input to the phase shifters 402-n-1 to 402-n-M via the switches 403-n-1 to 403-n-M. The phase shifters 402-n-1 to 402-n-M apply a predetermined complex phase rotation on the analog signal. For example, the phase shifters 402-n-1 to 402-n-M add a predetermined amount of complex phase rotation to the signals in the radio frequency band in the analog signal format. The analog signals to which the complex phase rotation is applied by the phase shifters 402-n-1 to 402-n-M are transmitted via the antenna elements 401-n-1 to 401-n-M, respectively. The transmission signal has a predetermined directivity formed by the complex phase rotation in the phase shifters 402-n-1 to 402-n-M. Although one transmission/reception signal processing circuit 651-n has been described here, for example, the antenna elements 401-1 to 401-1-M and the antenna elements 401-N-1 to 401-N-M are used. In, directivity is formed individually, and communication is performed with the wireless station device in the directivity direction.

Next, signal reception will be described. At the time of reception, the TDD switch 127-n receives a signal with the distribution coupler 404-n and the down converter 124-n connected (n=1,..., N). The signals received by the antenna elements 401-n-1 to 401-n-M (n=1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M adds a predetermined complex phase rotation on the analog signal to the input signal, and outputs the added signal via switches 403-n-1 to 403-n-M. Input to the distribution coupler 404-n.

The distribution coupler 404-n synthesizes the signals of the respective antenna systems input via the switches 403-n-1 to 403-n-M on an analog signal, and synthesizes the synthesized signal via the TDD switch 127-n. Input to the down converter 124-n. The down converter 124-n converts the radio frequency signal input via the TDD switch 127-n into an analog baseband signal, and inputs the analog baseband signal to the A/D converter 125-n. The A/D converter 125-n converts the signal input from the down converter 124-n from an analog baseband signal into a digital baseband signal, and outputs it to a signal separation circuit not shown here. ..

The signal separation circuit 141 in FIG. 14 suppresses crosstalk components between streams to perform signal separation, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be performed on the frequency axis by once converting into a frequency axis signal by FFT processing. When suppressing the crosstalk component on the time axis, first, the correlation between the signal sequences input to the signal separation circuit 141 is calculated based on the digital baseband signal corresponding to the received training signal. .. Then, based on the MIMO channel matrix given by the calculated correlation, a matrix similar to a general MIMO signal separation process such as ZF (Zero Forcing) type or MMSE (Maximum Mean Square Error) type signal separation is calculated. Then, this matrix may be multiplied by the sampling data unit with respect to the signal vector input to the signal separation circuit 141.

This is because while a general MIMO signal separation process such as a ZF-type or MMSE-type signal separation on a general frequency axis uses different matrices for each frequency component, a matrix that is almost the same on the frequency axis is used. In the case of using, the signal separation processing can be performed on the time axis in units of sampling data. Alternatively, only the signal processing performed by the transmission/reception signal processing circuits 651-1 to 651-N is sufficient, and the signal separation circuit 141 does not need to perform any processing. However, in any case, the details of the signal separation method here are not directly related to the features of the present embodiment, and therefore omitted. The demodulators 130-1 to 130-N demodulate the signals in which the crosstalk component is suppressed in the signal separation circuit 141.

Although not shown in FIG. 15, if a high power amplifier (HPA) or the like on the transmission side is arranged, for example, it is arranged at a point where “A” is described in the figure, and a low noise amplifier on the reception side ( if placing LNA) or the like, will be placed at a point with a description of "B" and "C _1" - "C _2" in FIG. Regarding "A" and "B", since the transmission/reception signal processing circuits 651-n are not supposed to cooperate with each other, the calibration processing for removing the uncertainty of the complex phase of the individual high power amplifier and low noise amplifier. Although it is not necessary, the low noise amplifier at the point where “C ₁ ”to “C ₂ ” is described is the same as the antenna element 401-1 and the antenna element 401-4 in the same transmission/reception signal processing circuit 651-n. It is necessary to remove the uncertainty of the complex phase of the low noise amplifier of each system by using the calibration method of the implicit feedback of the prior art because it may cause the uncertainty of the complex phase between ..

The present invention can be applied to any method, and a specific method of calibration processing does not matter. Considering this calibration result, for example, assuming that the rotation amounts of the complex phase at “C ₁ ”and “C ₂ ” are +10 degrees and +20 degrees, the complex phase obtained by the equations (1) to (3) is obtained. The phase rotation amount is adjusted by correcting -10 degrees and -20 degrees with respect to the rotation amount. The calibration result information is collected by a calibration circuit (not shown), and the phase shift control circuit 406-n, the complex phase rotation amount prediction circuit 410-n, or the correlation calculation circuit 405-n collects the information. Perform the correction using the information.

In addition, in the figure, regarding the switches 403-n-2, 403-n-3, 403-n-5 to 403-n-M of the system not including the down converter and the A/D converter, the phase shifter 402 -N-1 to 402-n-M, in the signal processing when calculating the amount of rotation of the complex phase, or in the connection state, or these switches 403-n-2, 403-n-3, 403-n- Even if 5 to 403-n-M are omitted, there is no practical problem.
Note that the phase rotation by the phase shifter is normally given by selectively passing a delay line corresponding to the amount of phase rotation on the device. Therefore, if a phase rotation of x is given as an absolute value, a negative phase rotation (delay) of the complex phase rotation amount is performed as a signal with a delay corresponding to the phase x, and the sign consistency cannot be obtained. .. However, in the following description for convenience, will be referred to as "the rotation amount of complex phase carried out in the phase shifter [psi _alpha" when giving a phase rotation equivalent to multiplication of the coefficient Exp {jψ _α} as a signal with the phase shifter To do.

FIG. 16 is a functional block diagram showing another configuration example of the transmission/reception signal processing circuit 652-n in the conventional technique shown in Non-Patent Document 5. The same numbers are assigned to the same functional blocks as those in the above figures.

The transmission/reception signal processing circuit 652-n shown in the figure includes a D/A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A/D converter 125-n. n, distribution coupler 414-n, distribution coupler 415-n, phase shifters 402-n-1 to 402-n-M, switches 403-n-1 to 403-n-M, and Phase converters 409-n-1 to 409-n-M, down converters (DC) 424-n-1 to 424-n-2, and A/D converters 425-n-1 to 425-n-2. , A correlation calculation circuit 405-n, a phase shift control circuit 406-n, and a complex phase rotation amount prediction circuit 410-n. Further, antenna elements 401-n-1 to 401-n-M and antenna elements 441-n-1 to 441-n-M are connected to the transmission/reception signal processing circuit 652-n.

The difference from FIG. 15 is that the antenna elements 401-n-1 to 401-n-M for transmission and the antenna elements 441-n-1 to 441-n-M for reception are separated, and as a result, TDD. While the switch 127-n becomes unnecessary, the distribution coupler 404-n shared by transmission and reception is separated into a transmission distribution coupler 414-n and a reception distribution coupler 415-n. The shared phase shifters 402-n-1 to 402-n-M are the phase shifters 409-n-1 to 409-n-M for transmission and the phase shifters 402-n-1 to 402 for reception. It is a point separated into -n-M. Other configurations are the same.

Similar to FIG. 15, regarding the signal processing, roughly, the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M perform the rotation amount of the complex phase. The signal processing, the signal receiving processing, and the signal transmitting processing when calculating is separately described. Similarly, the control circuit 460 shown in FIG. 14 is a circuit that manages the entire communication. Although not described in detail, the control circuit 460 manages the overall timing management and various processes, and for example, the phase shifters 402-n-1 to 402-n. -M and a phase shifter 409-n-1 to 409-n-M manage a series of management such as management of setting timing. Therefore, strictly speaking, signals may be input/output to/from each functional block, but they are omitted here for simplification.

First, the signal processing for calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M will be described. .. In the signal processing for calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M, the switch 403-n- 1 connects the down converter 424-n-1 and the phase shifter 402-n-1, and the switch 403-n-4 connects the down converter 424-n-2 and the phase shifter 402-n-4. (Actually, the same applies to the switches of the antenna system including the down converter and the A/D converter, which are not shown here), but the remaining switches are not connected.

These switches are controlled by the correlation calculation circuit 405-n under the instruction of the control circuit 460 (not shown), and the phase shifter 402-n- is used except when calculating the rotation amount of the complex phase. 1 to 402-n-M are connected to the distribution coupler 415-n. When performing this process, the phase rotation amount of the phase shifters 402-n-1 to 402-n-M is set to a predetermined value. The rotation amount of the complex phase obtained in the subsequent process is set as a difference with respect to the initial predetermined value, as in the above description. Further, the other transmission/reception signal processing circuits 652-n not shown here also perform the same processing all at once under the control of the control circuit 460 not shown here.

In this state, first, a training signal for channel estimation is transmitted from the wireless station device of the communication partner that should acquire the rotation amount of the complex phase, and the wireless station device receives this signal. The signals received by the antenna elements 441-n-1 to 441-n-M are input to the phase shifters 402-n-1 to 402-n-M, and the phase shifters 402-n-1 to 402-n- are input. A predetermined complex phase rotation is added on the analog signal at M, and each is input to the switches 403-n-1 to 403-n-M. Switches 403-n-1 and 403-n-4 (actually, a switch of an antenna system including a down converter and an A/D converter, which are not shown here, are also the same, but the explanation is simplified. Therefore, the signals input to the two antenna systems will be described) are input to the down converters 424-n-1 to 424-n-2. The down converters 424-n-1 to 424-n-2 convert the input radio frequency signals into analog baseband signals, and the A/D converters 425-n-1 to 425-n-2 perform analog conversion. Convert the signal to a digital baseband signal.

The information converted into the digital baseband signal by the A/D converters 425-n-1 to 425-n-2 is input to the correlation calculation circuit 405-n. The correlation calculation circuit 405-n calculates the rotation amount of the complex phase using the equations (1) to (3). Further, when the calibration process is required as necessary, the rotation amount of the complex phase on the transmission side is determined as a value in which the calibration coefficient is considered in the equations (1) to (3). The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the complex phase rotation amount prediction circuit 410-n together with the identification number of the wireless station device that is a communication partner.

The complex phase rotation amount prediction circuit 410-n predicts the complex phase rotation amount information of the remaining antenna elements based on the limited complex phase rotation amount information of the antenna system. In the case of FIG. 16, the complex phase rotation amount prediction circuit 410-n calculates the rotation amount of the complex phase to be performed by the phase shifters 402-n-1 to 402-n-M, and the wireless communication partner. It is input to the phase shift control circuit 406-n together with the identification number of the station device. The phase shift control circuit 406-n manages the value input here by storing it in a memory.

Further, the rotation amount of the complex phase described above relates to the phase rotation amount of the phase shifters 402-n-1 to 402-n-M in the reception system, but the complex phase rotation amount of the low noise amplifier and the high power amplifier In order to cancel the individual difference, a calibration process in the implicit feedback of the prior art is performed, the amount of rotation of the complex phase in the transmission system is converted, and the value is set in the phase shifters 409-n-1 to 409-n-M. Are similarly stored in the memory and managed.

When performing actual data communication, that is, when performing transmission or reception processing, the control circuit 460 (not shown in FIG. 16) grasps the wireless station device as the communication partner, and the phase shift control circuit 406- For n, the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M receive the rotation amount of the complex phase corresponding to the wireless station device that performs communication. The phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M are set to this complex phase rotation amount to indicate the complex phase rotation amount. Achieve the above beamforming.

Although not shown in the figure, if a high-power amplifier (HPA) on the transmission side is arranged, for example, there is a description of “A” and/or “D ₁ ”to “D _M ” in the figure. place the point, if placing such reception side low-noise amplifier (LNA), it will be located in a point on the description of "B" and or "E _1" - "E _M" in FIG. Regarding "A" and "B", since the transmission/reception signal processing circuit 652-n is not supposed to cooperate with each other, the calibration processing for removing the uncertainty of the complex phase of the individual high power amplifier and low noise amplifier. However, the same transmission/reception signal processing circuit is used for the high-power amplifier at the points described as “D ₁ ”to “D _M ”and the low-noise amplifier at the points described as “E ₁ ”-“E _M ”. Since it may cause the uncertainty of the complex phase between the antenna elements 401-n-1 to M and the antenna elements 441-n-1 to 441-n-M in 652-n, By using the implicit feedback calibration method, it is necessary to remove the uncertainty of the complex phase of the low noise amplifier of each system.

The present invention can be applied to any method, and a specific method of calibration processing does not matter. Considering this calibration result, assuming that the rotation amounts of the complex phase at “E ₁ ”, “E ₄ ”, etc. are +10 degrees, +20 degrees, etc., they are obtained by Equations (1) to (3). The phase rotation amount is adjusted by correcting the rotation amount of the complex phase by -10 degrees, -20 degrees, or the like. The calibration result information is collected by a calibration circuit (not shown), and the phase shift control circuit 406-n, the complex phase rotation amount prediction circuit 410-n, or the correlation calculation circuit 405-n collects the information. Perform the correction using the information.

Further, similarly to FIG. 15, the switches 403-n-2, 403-n-3, 403-n-5 to 403-n-M of the system not including the down converter and the A/D converter are moved. In the signal processing at the time of calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M, the connected state is established, or these switches 403-n-2, 403-n-3, 403 are used. Even if -n-5 to 403-n-M are omitted, there is no practical problem.

Next, signal processing when actual data communication is performed, that is, when transmission or reception processing is performed will be described. First, when performing data communication, all the switches 403-n-1 to 403-n-M are connected to the distribution coupler 415-n. In addition, the control circuit 460 (not shown) grasps the wireless station device as the communication partner, and notifies the phase shift control circuit 406-n of the rotation amount of the complex phase corresponding to the wireless station device that performs communication. Instructing the phase shifters 402-n-1 to 402-n-M and/or the phase shifters 409-n-1 to 409-n-M.

First, the signal processing when transmitting a signal will be described. At the time of transmission, as in the case of FIG. 15, the transmission/reception signal processing circuits 652-n (n=1,..., N) have the time-axis digital baseband generated by the modulator 120-n (not shown). Is input, the D/A converter 122-n converts the transmission signal input from the modulator 120-n into an analog baseband signal, and inputs the analog baseband signal to the up-converter 123-n. The up converter 123-n converts the signal input from the D/A converter 122-n from a baseband signal into a radio frequency band signal, and inputs the signal to the distribution coupler 414-n.

The distribution coupler 414-n branches the analog signal input from the up converter 123-n into an M-system analog signal. The branched analog signal is input to the phase shifters 409-n-1 to 409-n-M. The phase shifters 409-n-1 to 409-n-M add a predetermined complex phase rotation on the analog signal. The analog signals to which the complex phase rotation is applied by the phase shifters 409-n-1 to 409-n-M are transmitted via the antenna elements 401-n-1 to 401-n-M, respectively. The transmission signal has a predetermined directivity formed by the complex phase rotation in the phase shifters 409-n-1 to 409-n-M. For example, the antenna elements 401-1 to 401-1-M and the antenna elements 401-N-1 to 401-N-M are formed with individual directivities, and the wireless station device in the directivity direction is used. Communicate.

Next, signal reception will be described. At the time of reception, the switches 403-n-1 to 403-n-M receive signals with the distribution coupler 415-n and the phase shifters 402-n-1 to 402-n-M connected (n = 1,...,N). The signals received by the antenna elements 441-n-1 to 441-n-M (n=1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M adds a predetermined complex phase rotation on the analog signal to the input signal, and outputs the added signal via switches 403-n-1 to 403-n-M. Input to the distribution coupler 415-n.

The distributor/combiner 415-n combines the signals of the respective antenna systems input via the switches 403-1 to 403-n-M on an analog signal and inputs the combined signal to the down converter 124-n. To do. The down converter 124-n converts the input radio frequency signal into an analog baseband signal, and inputs the analog baseband signal to the A/D converter 125-n. The A/D converter 125-n converts the signal input from the down converter 124-n from an analog baseband signal into a digital baseband signal, and outputs it to a signal separation circuit not shown here. .. Other processes are the same as those in FIG.

In the wireless station device 450 shown in FIG. 14, the antennas included in the transmission/reception signal processing circuits 651-1 to 651-N are independent of each other, but each of the transmission/reception signal processing circuits 651-1 to 651-N is shown. It is also possible to share the antenna elements 401-1 to 401-M. For example, when the transmission/reception signal processing circuits 651-1 to 651-N illustrated in FIG. 14 have the configuration illustrated in FIG. 15 (that is, when the transmission antenna and the reception antenna are shared), the transmission/reception signal processing circuit 651-n. The signal line from (1≦n≦N) to the antenna element 401-nk (1≦k≦M) is combined by the distribution coupler 407-k, and the antenna element 401-k is connected to the end. It may be configured. This configuration is the one shown in FIG.

In the case where the transmission/reception signal processing circuits 651-1 to 651-N shown in FIG. 14 are replaced with the transmission/reception signal processing circuits 652-1 to 652-N shown in FIG. 15 (that is, the transmission antenna and the reception antenna are If separated), the signal line from the transmission/reception signal processing circuit 652-n (1≦n≦N) to the antenna element 401-n-k (1≦k≦M) is combined by the distribution coupler 407-k. Then, the antenna element 401-k is connected to the end thereof, and the signal line from the transmission/reception signal processing circuit 652-n (1≦n≦N) to the antenna element 441-n-k (1≦k≦M) is connected. A configuration may be used in which the elements are combined by the distribution coupler 447-k and the antenna element 441-k is connected to the end. This structure is the structure shown in FIG.

Although the wireless station device 452 has not been described in the description of FIG. 14, the configuration of the wireless station device 452 shown in FIG. 14 is the same as that of the wireless station device 456 shown in FIG. 17 or the wireless station device 457 shown in FIG. It is also possible to configure. In this way, it is possible to form a plurality of directional beams with one (sub)array antenna and perform spatial multiplex transmission on the same frequency at the same time.

As described above, in the conventional technology, basically, the estimation process of the complex phase rotation amount performed by the variable phase shifter corresponding to the transmission/reception weight is performed by the digital signal processing, and the actual complex phase rotation process is performed by the analog signal processing. Will be realized. Therefore, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are premised on signal processing on the frequency axis like the OFDM (Orthogonal Frequency Division Multiplexing) modulation method. In either case, both systems can be supported even if signal processing on the time axis is premised such as single carrier transmission and SC-FDE (Single-Carrier Frequency Domain Equalization).

However, the signal separation circuit 141 performs signal separation between each signal series, but here it is also possible to perform signal separation on the time axis, and once it is converted to a frequency axis signal by FFT and then on the frequency axis. Frequency-dependent signal separation may be performed. In this sense, the input to the demodulators 130-1 to 130-N may be a time axis signal or a frequency axis signal, but here, for simplicity, the time axis signal is used. Will be described as inputting. In this way, the concept regarding variations of communication systems such as the OFDM modulation system and SC-FDE is the same in the following description.

In addition, since the transmission/reception signal processing circuits 651-1 to 651-N having the sub-array configuration can be physically separated from each other, the correlation of the directional beams formed on the analog can be reduced. Are installed at a distance more than a predetermined distance.
Further, the same applies to all the following descriptions (including the embodiments of the present invention), but the baseband signal processing circuit 140 and the transmission/reception signal processing circuits 651-1 to 651-N are connected by wire, and Although the digital baseband signal is transmitted and received on the line, the baseband signal processing circuit 140 mounts the D/A converters 122-1 to 122-N and the A/D converters 125-1 to 125-N. In this case, the signal flowing on the wired connection between the baseband signal processing circuit 140 and the transmission/reception signal processing circuits 651-1 to 651-N can be the analog baseband signal.

Satoshi Suyama, Tatsunori Ohara, Shinkiyun, Yukihiko Okumura, "Influence of analog beamforming configuration in Massive MIMO with high frequency hybrid beamforming", IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, March 2015 ，RCS2014-337，p.213-218 Atsushi Ota, Takuto Arai, Hiroshi Shirato, Kazuki Maruta, Tatsuhiko Iwakuni, Masataka Iizuka, “Analytical Environment Millimeter-Wave Band Spatial Multiplexing Technique Using Parallel Transmission of 1st Eigenmode-Proposal and Basic Characteristic Evaluation Results-”, IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, August 2015, RCS2015-144, p.73-78 Atsushi Ota, Hiroshi Shirato, Kazuki Maruta, Takuto Arai, Tatsuhiko Iwakuni, Masataka Iizuka, "Effective Utilization of First Eigenmode Transmission in Massive MIMO for Viewing Environment", IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, 2015 November, RCS2015-239, p.293-298 Fukuzono Hayato, Murakami Tomoki, Kudo Riichi, Takatori Yasushi, Mizoguchi Masato, “Experimental evaluation of implicit feedback in downlink multi-user MIMO-OFDM system”, IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, November 2013 , RCS2013-187, p.79-84 Hiroshi Shirato, Kazuki Maruta, Ken Tanaka, Takuto Arai, Tatsuhiko Iwakuni, Satoshi Kurosaki, Atsushi Ota, “Proposal of Digital Assisted Analog Beamforming (DAABF) Technology-Expansion Technology for Wireless Entrance with Channel Time Fluctuation”, 2016 IEICE Communications Society Conference, B-5-87, The Institute of Electronics, Information and Communication Engineers, September 2016 Ken Tanaka, Kazuki Maruta, Hiroshi Shirato, Takuto Arai, Tatsuhiko Iwakuni, Satoshi Kurosaki, Atsushi Ota, "Proposal of Digital Assisted Analog Beamforming (DAABF) Technology-Expansion to FDD System", 2016 IEICE Communication Society Conference, B-5-89, The Institute of Electronics, Information and Communication Engineers, September 2016

In FIG. 8, if the difference in the amount of rotation of the complex phase due to the difference in path length between adjacent antenna elements becomes π or more, the amount of rotation of the complex phase cannot be accurately estimated due to the uncertainty of ±2nπ of the complex phase. There are cases. For example, consider a case where the antenna element spacing in FIG. 8 is one wavelength and the angle of arrival θ is 30 degrees. In this case, the path length difference between the closest antenna elements is λSin30=λ/2, and π rotation occurs when converted into a complex phase. The communication environment assumed here is an environment in which the line-of-sight wave is dominant, but the measured value is slightly different because it is slightly affected by the reflected wave.

For example, in the antenna elements 401-1 401-2, and 401-3, it is assumed that the rotation amount of the complex phase of the line-of-sight wave changes as 0→π→2π→. At that time, for positive small errors δ ₁ and δ ₂ , the antenna element 401-2 measures the rotation amount of π−δ ₁ by the reflected wave, and the antenna element 401-3 measures the rotation amount of the complex phase of 2π+δ ₂ by the reflected wave. Suppose At this time, the rotation amount of the complex phase between 401-2 and the antenna element 401-3 is (2π+δ ₂ )−(π−δ ₁ )=π+(δ ₁ +δ ₂ ), and the absolute value exceeds π. On the other hand, if the complex phase rotation amount of the antenna element 401-3 is corrected by 2π+δ ₂ −2π and 2π, the complex phase rotation amount in that case is (2π+δ ₂ −2π)−(π−δ ₁ )= It becomes −π+(δ ₁ +δ ₂ ), and the absolute value of the rotation amount of the complex phase becomes smaller than π. In this case, since it cannot be determined which is correct, assuming that the complex phase rotation amount is within ±π, the complex phase rotation amount of the latter antenna element 401-3 is erroneously (2π+δ ₂ − 2π). In this way, due to the uncertainty of the complex phase of 2π period, if the least squares estimation processing is performed using the incorrect complex phase rotation amount, the obtained complex phase rotation amount is an inappropriate value. Therefore, proper directivity control cannot be performed.

On the other hand, when the size of the entire antenna becomes large, the beam width of the directional beam becomes narrow as the antenna aperture increases. When forming a narrow directional beam with a large-scale antenna, use more antenna elements or use distant antenna elements as shown in FIGS. 9 and 10 instead of adjacent or adjacent antenna elements. Is preferred. However, increasing the number of antenna elements used for estimating the direction of arrival and estimating the amount of rotation of the complex phase leads to the use of many A/D converters that are relatively expensive, and from the viewpoint of cost reduction. Not preferable. Further, if the antenna elements that are separated from each other are used, the rotation amount of the complex phase between the antenna elements has an uncertainty of 2π period as described above, and thus the rotation amount of the complex phase cannot be calculated correctly. Therefore, it is possible to accurately estimate the rotation amount of the complex phase while avoiding the uncertainty of the 2π period of the rotation amount of the complex phase while adopting a configuration in which the A/D converter is mounted with a smaller number of antenna elements. A technique for estimating the amount of complex phase rotation is required.

In view of the above circumstances, it is an object of the present invention to provide a technique capable of reducing the cost and accurately estimating the rotation amount of a complex phase.

One embodiment of the present invention is a wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device. Image information acquisition means for acquiring image information in the front direction of the antenna plane including the antenna element, and input means for specifying the other wireless communication device to be a communication partner from the image information acquired by the image information acquisition means. An azimuth information acquisition unit that analyzes the video information and acquires azimuth information in which the other wireless communication device specified by the input unit exists; and a signal transmitted by the other wireless communication device, A signal receiving unit for receiving via each antenna element, and a part of the plurality of antenna elements, each of which is provided in a plurality of systems, and a radio frequency analog signal received by the antenna element is a baseband digital signal. Based on an output signal of the signal conversion unit when a training signal transmitted by the other wireless communication device is received by the plurality of antenna elements of the plurality of systems, The reference antenna element selected from among the antenna elements of the system, and obtains the correlation information for each combination of the other antenna elements of the antenna elements of the plurality of systems, based on the correlation information a correlation calculation unit for calculating a rotation amount of the complex phase of the antenna elements of said part of a plurality of systems, the based on the acquired azimuth information by azimuth information acquisition means, the part calculated by the correlation calculating section Of a plurality of systems of antenna elements, a rotation amount correction unit that performs correction for eliminating the uncertainty of the complex phase with respect to the rotation amount of a complex phase, and the partial plurality of systems of antenna elements corrected by the rotation amount correction unit A first rotation amount prediction unit that predicts a rotation amount of the complex phase to be given to the reception signals of the plurality of antenna elements based on the rotation amount of the complex phase of , and a prediction by the first rotation amount prediction unit. first phase rotation amount management unit that stores the rotation amount of the complex phase that is, the amount of rotation of said first of said complex phase stored in the phase rotation amount management unit, for each of the antenna elements for receiving signals a first phase rotation unit that Ru is phase rotation on an analog signal or on a digital signal on the received signal, the first of the received signal phase-rotated by the phase rotation section, for each of the array antennas, use the reception Based on the received signal synthesized by the signal synthesizing unit and the signal synthesizing unit that synthesizes an analog signal or a digital signal over all or some of the antenna elements. And a signal reproducing unit that reproduces a signal transmitted by the other wireless communication device.

One aspect of the present invention is the above-described wireless communication apparatus, the other of said antenna elements for each of the transmission signal to Jo Tokoro value on the digital signal on or analog signals used to transmit the transmission signal to the wireless communication device based only the second phase rotation unit that Ru is phase rotation, the rotation amount before Symbol first rotation amount prediction unit in predicted the complex phase rotation of said second complex phase should be given by the phase rotation unit a second rotation amount prediction unit for calculating an amount, and stores the rotation amount of the double prime phase calculated by the second rotation amount prediction unit, a second phase rotation to be set in the second phase rotation unit and quantity management unit, a transmission signal generator for generating a digital signal of the other radio communication device addressed, the antenna element using a signal which the transmission signal generating unit has generated to the transmission on a digital signal on or analog signal is branched for each, and the signal distribution unit for outputting the second phase rotation unit, a signal after phase rotation in front Stories second phase rotation unit, via the antenna element as an analog signal of the radio frequency And a signal transmitting unit for transmitting the signal.

One aspect of the present invention is the above-described wireless communication apparatus, when the radio frequency of the received signal and the transmission signal is different from the received signal to the amount of rotation of the complex phase should be given by the second phase rotation unit and further comprising a third rotation amount prediction unit intends line correction of multiplying the ratio of the radio frequency of the transmission signal.

One embodiment of the present invention is a wireless communication method performed by a wireless communication device which forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device. And a video information acquisition step of acquiring video information in a front direction of an antenna plane including the plurality of antenna elements, and the other wireless communication device serving as a communication partner from the video information acquired in the video information acquisition step. An input step of accepting the designation of:, an azimuth information acquisition step of analyzing the video information, and acquiring azimuth information of the existence of the other wireless communication device specified by the input step, and the other wireless communication device transmitting the signal, a signal receiving step of receiving via each of the plurality of antenna elements, the antenna elements of a portion of a plurality of systems in said another wireless communication device transmitted training signal pre SL plurality of antenna elements Based on a signal obtained by converting an analog signal of a radio frequency obtained by reception into a digital signal of a base band, a reference antenna element selected from among the antenna elements of the plurality of systems, and the part of the antenna element. Correlation calculation step of acquiring correlation information for each combination with other antenna elements of the antenna elements of the plurality of systems, and calculating the amount of rotation of the complex phase of the antenna elements of the plurality of systems based on the correlation information If, on the basis of the said orientation information acquired by the azimuth information acquisition step, uncertainty of the complex phase with respect to the rotation amount of the complex phase of the antenna elements of a plurality of systems of said portion calculated by the correlation calculating step Based on the rotation amount correction step of performing a correction to eliminate the , the rotation amount of the complex phase of the antenna elements of the plurality of systems corrected in the rotation amount correction step, with respect to the received signal of the plurality of antenna elements first and rotational amount prediction step, a first phase rotation amount managing step of storing a rotation amount of predicted the complex phase in the first rotation amount prediction steps for predicting the amount of rotation of the complex phase to give Te When the amount of rotation of said stored complex phase in the first phase rotation amount management step, is used to receive the relative received signal of each antenna element and the first to Ru by phase rotation on an analog signal or on a digital signal a phase rotation step, the first of the received signal phase-rotated by the phase rotation step for each of the array antenna, on all used for reception or analog signals over a part of the antenna element or on a digital signal The signal combining step to combine and the signal combining step And a signal reproducing step of reproducing a signal transmitted by the other wireless communication device based on the more combined received signal.

According to the present invention, the cost can be reduced and the complex phase rotation amount can be estimated with high accuracy.

It is a figure which shows an example of the antenna element in 1st Embodiment. It is a functional block diagram which shows the structural example of the transmission/reception signal processing circuit in the same embodiment. It is a figure which shows the structural example of the complex phase rotation amount prediction circuit in the same embodiment. It is a functional block diagram which shows the structure of the transmission/reception signal processing circuit in 2nd Embodiment. It is a functional block diagram which shows the structure of the transmission/reception signal processing circuit in 4th Embodiment. It is a functional block diagram which shows the structural example of the wireless station apparatus in a prior art. It is a functional block diagram which shows the structural example of the wireless station apparatus in a prior art. FIG. 10 is a diagram showing an outline of complex phase rotation amount prediction in a linear array in which antenna elements are linearly arranged in the related art. It is a figure which shows the outline|summary of rotation amount prediction of the complex phase of the array antenna comprised in the planar form in the prior art. It is a figure which shows the outline|summary of the rotation amount prediction of the complex phase of the square array antenna comprised by the planar shape in a prior art. It is a figure which shows the outline|summary of the rotation amount prediction of the complex phase of the square array antenna comprised by the planar shape in a prior art. It is a figure which shows the outline|summary of the rotation amount prediction of the complex phase of the square array antenna comprised by the planar shape in a prior art. It is a figure which shows the example of the antenna pattern used for prediction of the rotation amount of a complex phase in a prior art. It is a figure which shows the structural example of the communication system in a prior art. It is a functional block diagram which shows the structural example of the transmission/reception signal processing circuit in a prior art. It is a functional block diagram which shows the structural example of the transmission/reception signal processing circuit in a prior art. It is a functional block diagram which shows the structural example of the wireless station apparatus in a prior art. It is a functional block diagram which shows the structural example of the wireless station apparatus in a prior art.

Embodiments of the present invention will be described in detail with reference to the drawings. Note that the terms “time axis” and “frequency axis” used in this specification may be expressed as “time domain” and “frequency domain”, but here, the terms “time axis” and “frequency axis” are unified. And explain.

[First Embodiment]
The present invention is directed to a radio station device under such conditions in the state where the radio station device is fixedly installed and in which the time variation of the channel between the radio station devices can be ignored, as shown in the background art. Is a technology to be used when applying. Therefore, it is premised that the following processing is performed in advance in each wireless station device before the actual service operation starts. Further, in the first embodiment, the antenna elements used when calculating the rotation amount of the complex phase are composed of antenna elements linearly arranged in the horizontal direction and antenna elements linearly arranged in the vertical direction. It is assumed that

The basic principle of the present invention will be described below with reference to the drawings. FIG. 1 shows an example of an antenna element according to the first embodiment of the present invention. 1 has a similar arrangement to that of FIG. 10, but the difference is that the antenna elements used for estimating the amount of rotation of the complex phase are antenna elements f, l, p, q, and r (indicated by ⊚ in FIG. 1). Antenna element). Here, it is assumed that the antenna elements f, l, and p are aligned in the vertical direction, and the antenna elements p, q, and r are aligned in the horizontal direction. Similar to FIG. 10, in FIG. 1, a to r represented by “◯” are reception antenna elements used for data reception, and s to z and A to J represented by “●” are transmission antenna elements used in data transmission. Represents.

As an example, it is assumed that one side of each square lattice of the square lattice composed of a to r represented by “◯”, s to z represented by “●”, and A to J has one wavelength interval. Therefore, the distance between the antenna elements f, l, p, q, and r indicated by “⊚” is two wavelengths. Focusing on the antenna elements p, q, and r arranged in the horizontal direction, for example, the path length difference for each antenna element of the received signal coming from the horizontal angle difference θ direction is 2λ×Sinθ. Considering the case of θ=14.5 degrees, for example, 2λ×Sin θ=λ/2, which results in a change of π in phase. In reality, this is affected by the reflected wave and causes a slight error around π, so that it may not be within ±π between the antenna elements in some cases. However, if a small camera is mounted and the angle difference φ _{H in} the horizontal direction from the front direction (and the angle difference φ _E also in the vertical direction) can be obtained as an approximate value from the result of image analysis, it is approximate. It is possible to predict the amount of rotation of the complex phase, and by selecting a value that is a complex phase difference of ±π with respect to the predicted value, it is possible to eliminate the uncertainty of the complex phase of 2π period. Is.

For example, the image of a small camera is displayed on the monitor, the installer of the wireless station device specifies the position of the wireless station device to be the communication partner, and the specified place is horizontally analyzed from the front by image analysis. If it can be obtained as the angle difference in the vertical direction and the angle difference in the vertical direction, this value can be used as the angle difference φ _H and φ _E. At this time, if the distance between the antenna elements (for example, antenna elements p and q, antenna elements q and r) used for estimating the rotation amount of the complex phase in the horizontal direction is d _H , the rotation amount of the complex phase between the elements is approximately d. It is expected to be _{H 2} /λ×Sin(φ _H ). Accordingly, the rotation amount delta _H of complex phase is expected to be in the range of formula 6 below.

Similarly, if the distance between the antenna elements (for example, antenna elements f and l, antenna elements l and r) used for estimating the amount of complex phase rotation in the vertical direction is d _E , the amount of complex phase rotation between the elements is approximately d. _E /λ×Sin(φ _E ). Therefore, it is expected that the rotation amount Δ _E of the complex phase is in the range shown by the following Expression 7.

In this way, the uncertainty of the complex phase is eliminated, and the rotation amount of the complex phase of the antenna elements f, l, p, q, and r is calculated. Multiplication should be applied. As a result, for example, assuming an antenna element with a two-wavelength interval as shown in FIG. 1, when the horizontal and vertical angles are not within the range of ±14.5 degrees, when applying the background art of the present invention. Although it should be impossible to eliminate the uncertainty of the complex phase of 2π period, the application of the embodiment of the present invention makes it possible to eliminate the uncertainty of the complex phase of 2π period for a wider range of wireless station devices. In other words, the spatial spread of the antenna element used for estimating the rotation amount of the complex phase is expanded without being restricted by the angular range in which the wireless station device exists, and the antenna element with a wide aperture can equivalently perform high-precision estimation. It will be possible.

FIG. 2 is a functional block diagram showing a configuration example of a transmission/reception signal processing circuit for realizing the first embodiment. 2 corresponds to the transmission/reception signal processing circuit 651-n shown in FIG. 14, and the description will focus on the differences from the transmission/reception signal processing circuit 651-n shown in FIG.

A transmission/reception signal processing circuit 655-n shown in the figure includes a D/A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A/D converter 125-n. n, a TDD switch (TDD-SW) 127-n, a phase shifter 402-n-1 to 402-n-M, a switch 403-n-1 to 403-n-M, and a distribution coupler 404-. n, down converters (DC) 424-n-1 to 424-n-2, A/D converters 425-n-1 to 425-n-2, correlation calculation circuit 405-n, and phase shift control The circuit 406-n, the complex phase rotation amount prediction circuit 417-n, the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, and the information input unit 664 are provided. Further, antenna elements 401-1 to 401-M and a baseband signal processing circuit 140 are connected to the transmission/reception signal processing circuit 655-n. Although the down converter 424-n and the A/D converter 425-n each have two configurations in the figure, the down converter 424-n and the A/D converter 425-n have values smaller than M. Any value may be used as long as it is. In the following description, the case where there are two down converters 424-n and A/D converters 425-n will be described as an example. In the figure, the same numbers are assigned to the same functional blocks as those in the above figures, and the detailed description thereof will be omitted as necessary.

The difference from FIG. 15 is that a small camera 661, an image processing circuit 662, an arrival angle estimation circuit 663 and an information input means 664 are added to the transmission/reception signal processing circuit, and the complex phase rotation amount prediction circuit 410-n. It is replaced with the complex phase rotation amount prediction circuit 417-n.
Except for the above differences, the region in which signals flow during transmission and reception in data communication is the same as in FIG. 15, and the signal processing relating to data transmission/reception via antenna elements 401-1 to 401-M is the same as in FIG. Is equivalent to the signal processing of the background art of the present invention. The difference in the signal processing exists only in the signal processing when calculating the rotation amount of the complex phase performed by the phase shifters 402-n-1 to 402-n-M. The signal processing when calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M will be described below.

First, when installing each wireless station device, the installation operator installs the wireless station device at the installation location and confirms the image captured through the small camera 661 with the information input unit 664. The small camera 661 may be composed of a plurality of cameras. For example, if two small cameras 661 with different horizontal positions and two cameras with different vertical positions are used, stereoscopic image information can be obtained, and the accuracy of orientation estimation in image processing can be improved. It is also possible to improve. Here, the information input unit 664 is, for example, a monitor screen and a unit for the installation operator to input (for example, a touch panel of the monitor screen). Even when a plurality of small cameras 661 are used as described above, only the images of one small camera 661 are displayed on the monitor screen, and the images of the other cameras are displayed at locations corresponding to each other in image processing. Be searchable. If a touch panel is used here, the wireless station device to be the communication partner is designated by touching the screen. Alternatively, the location on the screen may be designated with a mouse or the like. The information input from the information input unit 664 in this manner is compared with reference points in the horizontal direction and the vertical direction in the image processing circuit 662, and various existing image processing techniques are used to determine the positional relationship. The arrival angle estimation circuit 663 calculates an angle difference φ _{H in the} horizontal direction and an angle difference φ _E in the vertical direction with respect to the front direction. In this sense, the image processing circuit 662 and the arrival angle estimation circuit 663 may function integrally. The result calculated in this way is input to the complex phase rotation amount prediction circuit 417-n. The above processing is performed in advance. Then, the process of calculating the rotation amount of the complex phase is performed.

In the signal processing for calculating the rotation amount of the complex phase, the switch 403-n-1 connects the down converter 424-n-1 and the phase shifter 402-n-1, and the switch 403-n-4 is down. The converter 424-n-2 and the phase shifter 402-n-4 are connected (actually, a down converter and an antenna system switch equipped with an A/D converter which are not shown here are also the same), The remaining switches are not connected. In this state, first, a training signal for channel estimation is transmitted from the wireless station device of the communication partner that should acquire the rotation amount of the complex phase, and the wireless station device receives this signal.

The signal received by the antenna element 401-n-1 is input to the phase shifter 402-n-1, and the phase shifter 402-n-1 rotates a predetermined complex phase (for example, 0 degree) on the analog signal. ) Is added and input to the switch 403-n-1. The signal input to the switch 403-n-1 is input to the down converter 424-n-1. The signal received by the antenna element 401-n-4 is input to the phase shifter 402-n-4, and the phase shifter 402-n-4 rotates a predetermined complex phase (for example, 0 degree) on the analog signal. However, it may be input) to the switches 403-n-4. The switch 403-n-4 (actually, a switch of an antenna system including a down converter and an A/D converter not shown here is also the same, but for simplification of description, this one antenna is used. Signal will be input to the down converter 424-n-4.

The down converters 424-n-1 to 424-n-2 convert the input radio frequency signal into an analog baseband signal. The A/D converters 425-n-1 to 425-n-2 convert analog signals into digital baseband signals. The information converted into the digital baseband signal by the A/D converters 425-n-1 to 425-n-2 is input to the correlation calculation circuit 405-n. The correlation calculation circuit 405-n calculates the rotation amount of the complex phase using the equations (1) to (3) based on the input information. Further, when the calibration process is required as necessary, the rotation amount of the complex phase on the transmission side is determined as a value in which the calibration coefficient is considered in the equations (1) to (3). The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the complex phase rotation amount prediction circuit 417-n together with the identification number of the wireless station device that is a communication partner.

In the complex phase rotation amount prediction circuit 417-n, first, in the example of FIG. 1, since the complex phase regarding the antenna elements f, l, p, q, and r involves the uncertainty of the complex phase of 2π period, the arrival angle estimation is performed. The horizontal angle difference φ _H and the vertical angle difference φ _E with respect to the front direction input from the circuit 663 are applied to the equations (7) and (8) to eliminate the uncertainty of the complex phase. For example, taking the antenna element r and the antenna element q as an example, the rotation amount of the complex phase between these two elements is within the range of Expression (6). Therefore, when the antenna element r is used as the reference antenna, the complex phase rotation amount of the antenna element q with respect to the reference antenna element r is added to the complex phase rotation amount obtained by the correlation calculation circuit 405-n by ±2πn, and the expression ( Find the complex phase within the range of 6). Rotational amount of the complex phase of the antenna element q in this way (here delta _q and rear) (to remove the uncertainty of the complex phase of 2π cycles) to determine the. After that, the complex phase rotation amount of the antenna element p with respect to the antenna element q is calculated, the uncertainty of the complex phase rotation amount is similarly removed by the formula (6), and Δ _q is added to this to add the reference antenna element r. The amount of rotation of the complex phase of the antenna element p with respect to (corresponding to _Δp ) is calculated. In this case, Δ _q and Δ _q satisfy the following conditional expression 8.

Alternatively, the fact that the element spacing between the reference antenna element r and the antenna element p is 2d _H may be directly used to obtain the value directly by the following Expression 9.

The above process is performed on the antenna elements (antenna elements p, q, and r in this figure) for estimating the rotation amount of the complex phase arranged in the horizontal direction. After that, the same processing is performed on the antenna elements (antenna elements f, l, and r in this figure) for estimating the rotation amount of the complex phase arranged in the vertical direction. In this way, the uncertainty of the complex phase of the 2π period with respect to the antenna elements f, l, p, q, and r is eliminated, and based on this result, the complex phase rotation amount prediction circuit 417-n uses the complex phase in FIG. Similar to the rotation amount prediction circuit 410-n, the least squares method is applied to calculate the final estimation formula corresponding to the formula (4). The result is applied to the coordinates of each antenna element to estimate the rotation amount of the complex phase of each antenna element.

In this way, the spatial spread of the antenna element used for estimating the amount of rotation of the complex phase is expanded while eliminating the uncertainty of the 2π period of the amount of rotation of the complex phase of the antenna element, and the equivalent wide aperture is used. The antenna element enables highly accurate estimation. In this way, the complex phase rotation amount prediction circuit 417-n calculates the rotation amount of the complex phase to be performed by the phase shifters 402-n-1 to 402-n-M, and the wireless station device of the other party to communicate with. It is input to the phase shift control circuit 406-n together with the identification number. The phase shift control circuit 406-n manages the value input here by storing it in a memory. It should be noted that the control circuit 460 performs overall adjustment, timing control, and the like in a series of these processes.

FIG. 3 is a diagram showing a configuration example of the complex phase rotation amount prediction circuit 417-n in the first embodiment. In FIG. 3, the complex phase rotation amount prediction circuit 417-n includes a provisional phase rotation amount calculation circuit 418-n and a complex phase rotation amount correction circuit 419-n. Taking the description of FIG. 2 as an example, the angle difference φ _{H in the} horizontal direction and the angle difference φ _{E in the} vertical direction with respect to the front direction are input to the provisional phase rotation amount calculation circuit 418-n from the arrival angle estimation circuit 663. .. The provisional phase rotation amount calculation circuit 418-n is predicted from the azimuth information by using the input horizontal angle difference φ _H and vertical direction angle difference φ _E and Expressions (6) to (9). The amount of rotation of the complex phase is estimated. The complex phase rotation amount correction circuit 419-n limits the rotation amount of the complex phase within the range of ±π with respect to the estimated rotation amount of the complex phase, and rotates the complex phase rotation input from the correlation calculation circuit 405-n. The uncertainty of the complex phase of the quantity is eliminated and the amount of rotation of the complex phase with respect to the antenna elements f, l, p, q, r is determined. Further, the complex phase rotation amount correction circuit 419-n calculates information about the rotation amount of the complex phase of all the antenna elements corresponding to the equation (4) by using the least squares method using these.

Substituting the coordinates of each antenna element into the equation corresponding to equation (4) obtained in this way, the amount of rotation of the complex phase to be set in the phase shifter of each antenna element is calculated, and this is used for phase shift control. Input to the circuit. In this way, the processing basically equivalent to that of the complex phase rotation amount prediction circuit 410-n of FIG. 15 is performed except the means for eliminating the uncertainty of the complex phase.

As described above, according to the present embodiment, the wireless station device uses the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, and the information input means 664 to determine the arrival angle in the horizontal and vertical directions. Based on the estimated arrival angle, the uncertainty of 2π period regarding the rotation amount of the complex phase is eliminated. Thereby, the rotation amount of the complex phase can be estimated with high accuracy. Therefore, it is possible to set the directivity with high accuracy while suppressing the number of A/D converters and D/A converters required to acquire channel information or complex phase rotation amount information. Further, the wireless station device can reduce the power consumption and the cost of configuring the device.

Although the transmission/reception signal processing circuit 655-n corresponding to the transmission/reception signal processing circuit 651-n of FIG. 15 in the background art of the present invention has been described with reference to FIG. 2, it may also be realized by the configuration corresponding to FIG. It is possible. That is, it can be similarly used in a configuration in which the transmitting antenna and the receiving antenna are separated.

[Second Embodiment]
FIG. 4 is a functional block diagram showing the configuration of a transmission/reception signal processing circuit for realizing the second embodiment. The difference from FIG. 16 is that a small camera 661, an image processing circuit 662, an arrival angle estimation circuit 663, and an information input unit 664 are added to the transmission/reception signal processing circuit, as in the case of FIG. The complex phase rotation amount prediction circuit 410-n is replaced with the complex phase rotation amount prediction circuit 417-n.

Compared with FIG. 16, except that the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, the information input means 664 are added, and the processing of the complex phase rotation amount prediction circuit 417-n is excluded, The signal processing relating to the reception of data via the antenna elements 441-1 to 442-M is equivalent to the signal processing of the background art of the present invention in FIG. Similarly, the signal processing relating to the transmission of data via the antenna elements 401-1 to 401-M is equivalent to the signal processing of the background art of the present invention in FIG. The difference of the signal processing is only the signal processing when calculating the rotation amount of the complex phase performed by the phase shifters 402-n-1 to 402-n-M and the phase shifters 442-n-2 to 442-n-M. Exists in. Further, the signal processing when calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M and the phase shifters 442-n-2 to 442-n-M is the first. This is equivalent to the signal processing of FIG. 3 in the embodiment. Therefore, the complex phase rotation amount prediction circuit 417-n is also realized by the configuration of FIG.

[Third Embodiment]
As described in Non-Patent Document 6, the background art of the present invention describes a method of acquiring the rotation amount of a complex phase set in a phase shifter in the case of FDD in which frequencies used for transmission and reception are different. Here, for example, with respect to the values of α, β, and γ regarding the rotation amount of the complex phase given by Expression (4), and the frequency f _{BW in the} receiving direction (backward direction) and the frequency f _{FW in the} transmitting direction (forward direction), A conversion formula for the rotation amount of the complex phase in the transmission direction is given by the following formula 10 with respect to the formula (4) obtained in the reception direction by the following formula.

Therefore, if the function of performing this conversion is implemented in the complex phase rotation amount prediction circuit 417-n (for example, the complex phase rotation amount correction circuit 419-n), the present invention can be used in FDD. ..

[Fourth Embodiment]
In the configurations of FIGS. 2 and 4 described in the first to third embodiments of the present invention, the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, and the information input unit 664 are the installers of the wireless station device. It is used only when the radio station equipment is installed by the. Therefore, the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, and the information input unit 664 are not necessarily essential to the wireless station device, and can be realized by a removable external device.

FIG. 5 is a functional block diagram showing the configuration of a transmission/reception signal processing circuit for realizing the fourth embodiment. The configuration shown in FIG. 5 is a configuration in which the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, and the information input means 664 in the configuration shown in FIG. Interface circuits 665 and 666 for connection are mounted on each of the 656-n and the external device 657. Except that information is exchanged via the interface circuits 665 and 666, the processing content is completely equivalent to that of the transmission/reception signal processing circuit 655-n shown in FIG. The external device 657 can be removed and connected to another transmission/reception signal processing circuit 655-n.

Further, this configuration is a configuration in which the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, and the information input means 664 in the configuration shown in FIG. It is also possible to realize a configuration in which the small camera 661, the image processing circuit 662, the arrival angle estimation circuit 663, and the information input means 664 are detachable external devices in the configuration shown.

As described above, according to each of the above embodiments, the wireless station device does not acquire the complex phase rotation amount information of all the antenna elements, but acquires the complex phase rotation amount information of some of the antenna elements. To do. The wireless station device calculates weight information from the acquired complex phase rotation amount information, and approximately calculates weight information of other antenna elements based on the calculated weight information. At this time, if the angles of arrival in the horizontal and vertical directions can be estimated using the small camera and the image analysis means, the rotation amount of the complex phase of a part of the antenna elements used when estimating the rotation amount of the complex phase is 2π cycles. Complex phase uncertainties can be eliminated. By eliminating this uncertainty of the complex phase, it is possible to use the antenna elements relatively distant, which could not be set in the background art of the present invention, for estimating the rotation amount of the complex phase. As a result, the antenna element used for estimating the rotation amount of the complex phase while suppressing the number of A/D converters and D/A converters required to acquire the channel information or the rotation amount information of the complex phase. It is possible to widen the spatial spread (antenna aperture), and as a result, the directivity to be formed can be set with high accuracy. Further, as described above, by calculating the weight information of other antenna elements by approximation, even if the transmitting antenna and the receiving antenna are different, implicit feedback can be used. Therefore, it is possible to reduce the size and cost and to use the implicit feedback even when the transmitting antenna and the receiving antenna are different. Further, even when the frequency is different between the transmission and the reception, the extended implicit feedback is used to enable the directivity control with high accuracy.

In each of the above-described embodiments, the rotation of the complex phase is performed on the analog signal of the radio frequency, but the position of the up converter is changed to rotate the complex phase of the analog signal of the baseband or the intermediate frequency. It may be configured such that the frequency conversion is performed and the frequency conversion with the radio frequency is performed in the subsequent stage or the previous stage.

[Supplementary information regarding the embodiment]
The supplementary items regarding the embodiment of the present invention described above will be shown below.
The wireless station device in the above-described embodiment may be realized by a computer. In that case, the program for realizing this function may be recorded in a computer-readable recording medium, and the program recorded in this recording medium may be read by a computer system and executed. The “computer system” mentioned here includes an OS and hardware such as peripheral devices. Further, the “computer-readable recording medium” refers to a portable medium such as a flexible disk, a magneto-optical disk, a ROM, a CD-ROM, or a storage device such as a hard disk built in a computer system. Further, the "computer-readable recording medium" means that a program is dynamically held for a short time like a communication line when transmitting the program through a network such as the Internet or a communication line such as a telephone line. In this case, a volatile memory inside the computer system that serves as a server or a client in that case may hold a program for a certain period of time. Further, the program may be for realizing a part of the functions described above, or may be a program that can realize the functions described above in combination with a program already recorded in a computer system, It may be realized using hardware such as PLD (Programmable Logic Device) or FPGA (Field Programmable Gate Array).

Although the embodiments of the present invention have been described with reference to the drawings, the above embodiments are merely examples of the present invention, and it is obvious that the present invention is not limited to the above embodiments. is there. Therefore, additions, omissions, replacements, and other changes of the constituent elements may be made without departing from the technical idea and scope of the present invention.

The present invention can be applied to a wireless communication device that transmits and receives wireless signals by using a plurality of antennas.

120-1 to 120-N... Modulator 122-1 to 122-N, 122-n... D/A converter 123-1 to 123-N, 123-n... Up converter 124-1 to 124-N, 124 -N... Down converters 125-1 to 125-N, 125-n... A/D converters 127-1 to 127-N... TDD switches 130-1 to 130-N... Demodulator 140... Baseband signal processing circuit 141 ... Signal separation circuits 401-1 to 401-M... Antenna elements 401-1 to 401-NM, 401-n-1 to 401-n-M... Antenna elements 402-1-1 to 402-N- M, 402-n-1 to 402-n-M... Phase shifter 403-1-1 to 403-NM, 403-n-1 to 403-n-M... Switch 404-1 to 404-N... Distribution couplers 405-1 to 405-N, 405-n... Correlation calculation circuits 406-1 to 406-N, 406-n... Phase shift control circuits 407-1 to 407-M... Distribution coupler 408-n-1 ~ 408-n-M... TDD switch 409-n-1 to 409-n-M... Phase shifter 410-n... Complex phase rotation amount prediction circuit 411-n... Transmission weight calculation circuit 414-n... Distribution coupler 415 -N... Distribution coupler 417-n... Complex phase rotation amount prediction circuit 418-n... Temporary phase rotation amount calculation circuit 419-n... Complex phase rotation amount correction circuit 424-1-1 to 424-NM, 424- n-1 to 424-n-M... Down converter 425-1-1 to 425-NM, 425-n-1 to 425-n-M... A/D converter 441-1 to 441-M... Antenna Element 441-n-1 to 441-n-M... Antenna element 450... Radio station apparatus 451-1 to 451-N... Transmission/reception signal processing circuit 452... Radio station apparatus 453-n... Transmission/reception signal processing circuit 454-n... Transmission/reception Signal processing circuit 455-n... Transmission/reception signal processing circuit 460... Control circuit 655-n... Transmission/reception signal processing circuit 656-n... Transmission/reception signal processing circuit 657... External device 661... Small camera 662... Image processing circuit 663... Arrival angle estimation Circuit 664... Information input means 665... I/F (interface circuit)
666... I/F (interface circuit)
901-1 to 901-N... Modulator 902... Precoder 903-1 to 903-N ₀ ... IFFT & GI assigning circuit 904-1 to 904-N ₀ ... D/A converter 905-1 to 905-N ₀ ... Up converter 906-1~906-N 0 _... downconverter _907-1~907-N 0 ... A / D converter _908-1~908-N 0 ... GI removing & FFT circuit 909 ... postcoders 910-1~910-N ... demodulator 911 ... TDD switch 912-1~912-N 0 _... distributor coupler _{913-1-1~913-N} 0 -M 0 _... phase shifter 915-1~915-M 0 _... distributor coupler 916-1 ... 916-M ₀ ... Antenna elements 921-1 to 921-N... Time axis transmission weight multiplication circuits 922-1 to 922-M... D/A converters 922-1-1 to 922-NM... D/A Converters 923-1 to 923-M... Up converters 923-1-1 to 923-NM... Up converters 924-1 to 924-M... Down converters 924-1-1 to 924-NM... Down converters 925-1 to 925-M... A/D converters 925-1 to 925-NM... A/D converters 926-1 to 926-N... Time axis reception weight multiplication circuits 927-1 to 927- N... TDD switch 928-1 to 928-M... Antenna element 928-1-1 to 928-NM... Antenna element 929-1 to 929-N... Transmit/receive signal processing circuit 941-1 to 941-M... Distribution coupling Unit 942... Wireless station devices 943-1 to 943-M... Adder/combiner 944-1 to 944-M... Duplicator 945... Wireless station devices 951-1 to 951-3... High power amplifiers 952-1 to 952-3 ...Low noise amplifier 953-1 to 953-3... TDD switch 954-1 to 954-3... Antenna element 955-1 to 955-3... Wireless module

Claims (4)

A wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements, and wirelessly communicates with another wireless communication device,
Video information acquisition means for acquiring video information in the front direction of the antenna plane including the plurality of antenna elements,
Input means for designating the other wireless communication device to be a communication partner from the video information acquired by the video information acquisition means;
Azimuth information acquisition means for analyzing the video information and acquiring azimuth information in which the other wireless communication device specified by the input means exists.
A signal reception unit that receives a signal transmitted by the other wireless communication device via each of the plurality of antenna elements,
A signal converter provided in each of a plurality of systems of a part of the plurality of antenna elements, for converting an analog signal of a radio frequency received by the antenna element into a baseband digital signal,
Based on an output signal of the signal conversion unit when the training signal transmitted by the other wireless communication device is received by the antenna elements of the plurality of systems, a selection is made from among the antenna elements of the plurality of systems. To obtain correlation information for each combination of an antenna element serving as a reference and an antenna element of the plurality of antenna elements of the plurality of systems, and based on the correlation information, the antennas of the plurality of systems. A correlation calculation unit that calculates the rotation amount of the complex phase of the element ,
Based on the orientation information acquired by the azimuth information acquisition means, the rotation amount of the complex phase of the antenna element of the plurality of lines of said portion calculated by the correlation calculating unit to eliminate the uncertainty of complex phase A rotation amount correction unit that performs correction ,
Predicting the rotation amount of the complex phase to be given to the reception signals of the plurality of antenna elements based on the rotation amount of the complex phase of the antenna elements of the plurality of systems corrected by the rotation amount correction unit 1 rotation amount prediction unit,
First phase rotation amount management unit that stores the rotation amount of the predicted the complex phase in the first rotation amount estimation section,
The amount of rotation of said first of said complex phase stored in the phase rotation amount management unit, a first phase rotation Ru by phase rotation in the upper analog signal on the received signal of each antenna element or on a digital signal for use in reception Department,
The received signal phase-rotated by said first phase rotation unit, the array for each antenna, the signal synthesizing unit for synthesizing on an analog signal or on a digital signal over all or a portion of the antenna element used for reception When,
A signal reproducing unit that reproduces a signal transmitted by the other wireless communication device based on the received signal combined by the signal combining unit,
A wireless communication device. A second phase rotation unit that Ru is only phase rotation Jo Tokoro value on the digital signal on or analog signal to the transmission signal for each of the antenna elements used for transmission of the transmission signal to the other wireless communication device,
Before SL based on the rotation amount of predicted the complex phase at a first rotational amount prediction unit, a second rotation amount prediction unit for calculating a rotation amount of the complex phase should be given by the second phase rotation unit,
Stores the rotation amount of the double prime phase calculated by the second rotation amount prediction unit, a second phase rotation amount management unit to be set in the second phase rotation unit,
A transmission signal generation unit that generates a digital signal addressed to the other wireless communication device;
A signal distributor for said signal to the transmission signal generating unit has generated is branched for each of the antenna elements used for transmission on a digital signal on or analog signal, and outputs the second phase rotation unit,
The signal after phase rotation in front Stories second phase rotation unit, a signal transmitting unit for transmitting via the antenna element as an analog signal of the radio frequency,
The wireless communication device according to claim 1, further comprising: When the radio frequency of the received signal and the transmitted signal is different, intends row correction for multiplying the ratio of the radio frequency of the received signal and the transmitted signal to the amount of rotation of the complex phase should be given by the second phase rotation unit The wireless communication device according to claim 2 , further comprising a third rotation amount prediction unit. A radio communication method executed by a radio communication device that forms a directivity using one or a plurality of array antennas configured to include a plurality of antenna elements, and that performs radio communication with another radio communication device,
A video information acquisition step of acquiring video information in the front direction of the antenna plane including the plurality of antenna elements;
An input step of accepting designation of the other wireless communication device to be a communication partner from the video information acquired in the video information acquisition step;
An azimuth information acquisition step of acquiring the azimuth information in which the other wireless communication device specified by the input step is analyzed by analyzing the video information,
A signal receiving step of receiving a signal transmitted by the other wireless communication device via each of the plurality of antenna elements,
The part of the radio frequency analog signal obtained by receiving at the antenna elements of a plurality of systems into a digital signal of baseband signals in said other wireless communication apparatus before Symbol plurality of antenna elements of the training signal transmitted On the basis of the above , the correlation information for each combination of the reference antenna element selected from the antenna elements of the plurality of systems and the other antenna element of the antenna elements of the plurality of systems A correlation calculation step of acquiring, and calculating the amount of rotation of the complex phase of the antenna elements of the plurality of systems based on the correlation information ,
Based on the orientation information acquired by the azimuth information acquisition step, eliminating the uncertainty of complex phase with respect to the rotation amount of the complex phase of the antenna element of the plurality of lines of said portion calculated by the correlation calculating step a rotation amount correction step of performing correction for,
Predicting a rotation amount of a complex phase to be given to a reception signal of the plurality of antenna elements based on a rotation amount of a complex phase of the antenna elements of the plurality of systems corrected in the rotation amount correction step 1 rotation amount prediction step,
A first phase rotation amount managing step of storing a rotation amount of predicted the complex phase in the first rotation amount prediction steps,
Wherein the amount of rotation of said stored complex phase in the first phase rotation amount management step, a first phase rotation Ru by phase rotation on an analog signal or on a digital signal on the received signal for each of said antenna elements for receiving signals Steps,
Signal combining step of combining the received signal phase-rotated by said first phase rotation step for each of the array antenna, on all used for reception or analog signal or on a digital signal over a portion of the antenna element When,
A signal reproducing step of reproducing a signal transmitted by the other wireless communication device based on the received signal combined by the signal combining step,
And a wireless communication method.

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