JP6673775B2 - Wireless communication device and wireless communication method - Google Patents

Description

  The present invention relates to a wireless communication device and a wireless communication method.

[Background surrounding 5th generation mobile communication]
At present, high-performance mobile communication terminals such as smartphones are exploding. With regard to mobile phones, the third generation mobile communication has shifted to the fourth generation mobile communication, and research and development on the further fifth generation mobile communication (commonly known as "5G") are currently underway. One of the studies performed on 5G is the use of macro cells and small cells.

  In conventional mobile phones, one service area is set to a radius of several kilometers, and one macro station area is covered by one base station device. However, there is a very large number of users in such a macrocell. Since the entire system capacity is shared by each user, when accommodating an enormous number of users, the throughput for each individual user decreases.

  In order to avoid such a decrease in throughput, a technology for setting a small cell, which is a very small service area with a radius of about several tens of meters, in a densely populated area where traffic is concentrated has been developed. In this technology, small traffic is utilized to offload spot-like traffic to a network without going through a macro cell. Here, it is assumed that the terminal device can simultaneously and concurrently use the communication capability in the small cell and the communication capability in the macro cell. By using such a terminal device, user data is accommodated on the small cell side while exchanging control information using a macro cell. This makes it possible to maximize the advantages of the macro cell and the small cell.

  In 5G described above, the target value of the transmission rate is set to 10 Gbit / s (gigabits per second) or more, and even in this small cell, efficient offload of traffic is realized by performing similar large-capacity communication. There is a need to. In a macro cell, it is assumed that a microwave band having a low frequency is used to allow long-distance propagation. However, considering the current situation in the microwave band where frequency resources are already being depleted, small cells that assume communication at relatively short distances are expected to use quasi-millimeter wave bands or millimeter wave bands with relatively high frequencies. Have been. The feature of this high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. Therefore, it is preferable that the small cell base station is ideally installed at a location close to the user terminal. For example, in a place where installation is easy, such as the rooftop of a building, the distance between the user terminal and the base station is too large, which is not preferable in line design.

  On the other hand, since the small cell is set in a place where traffic concentrates, even in a place where it is difficult to lay an optical fiber, there is a case where the installation of the base station device is strongly desired. For example, assuming that small cell base station equipment is installed in a place where there are very many people, such as in front of a station such as Shinjuku or Shibuya, propagation attenuation increases on the rooftop of a building adjacent to such a place. . Therefore, installation on a place lower than the roof of a building, for example, on a wall surface of the building, may be required. However, it may be difficult to lay optical fiber on the wall of an existing building, and in such cases, it may be necessary to provide a backhaul line to the base station device using a wireless line. is there.

  In order to provide such a backhaul line, in order to cope with a large capacity transmission of 10 Gbit / s or more required in a small cell, it is necessary to similarly perform a large capacity transmission of 10 Gbit / s or more utilizing a millimeter wave band. There is. In such an environment, since both opposing radio station devices are fixedly installed in a stable place, it is natural that the line of sight is generally secured naturally and the directional antennas face each other. is there. In this case, although reflected waves between buildings and the like exist to some extent, most of the received signals are line-of-sight components, and are expected to be in a state that cannot be said to be a multipath environment. This situation is the same for the access system if the small cell base station device is installed in a high place such as a building wall surface and is used in a line-of-sight environment in a manner that looks down on the user from above and below.

  Next, for large-capacity transmission of 10 Gbit / s or more, which is the transmission rate required for 5G, frequency resources with a very wide bandwidth can be used by utilizing the millimeter-wave band, which makes it feasible. Is growing. For example, assuming a backhaul line using a millimeter wave band, the E-band (71 to 76 GHz and 81 to 86 GHz) is used as an example, and if a 1 GHz bandwidth is used, the frequency utilization efficiency is 10 bits. / S / Hz. However, existing radio equipment for achieving a frequency utilization efficiency of 10 bit / s / Hz generally employs spatial multiplexing transmission using a multiple-input multiple-output (MIMO) channel. Spatial multiplexing transmission generally uses a multipath environment, and when a singular value decomposition of a channel matrix H expressing a transfer function of a MIMO channel in a matrix format is performed, a distribution of an absolute value of a singular value obtained as a result. Represents the characteristics of the spatial multiplexing transmission. Specifically, the square of the absolute value of the singular value is a value proportional to the signal-to-noise ratio (SNR). For spatial multiplexing transmission, not only the first singular value but also the second singular value is used. Communication is not established unless it has a sufficiently large value even after the second singular value. The same is true for large-capacity transmission in small cells that are access systems, but realizing spatial multiplexing transmission in such an environment where line-of-sight waves are dominant is indispensable for realizing the target wireless system. .

  As described above, in 5G, use of the millimeter wave band is expected for both the access system and the backhaul line. As described above, a characteristic of the high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. For example, comparing the 2 GHz band (existing access system) with the 80 GHz band (future system using the millimeter wave band), since the frequency is 40 times, the propagation attenuation is 1600 times and the line gain of 32 dB is insufficient. Will be. Of course, it is not necessary to cover all of 32 dB because it is not necessary to cover a wider area than a macro cell. On the other hand, in a high frequency band, a high power amplifier in a transmission stage does not have a device with a very high output. In this case, it is considered that additional line gain must be secured at a level of several tens of dB. Furthermore, since spatial multiplexing transmission is also expected in such an environment, a radio system equipped with a much larger number of antenna elements than in the prior art has been studied in both the base station and the terminal station. . Such a technique is called Massive MIMO. The following describes a conventional technique related to Massive MIMO.

  In Non-Patent Document 1, a base station side has an antenna of 256 elements, and a terminal station side has an antenna of 16 elements, and aims at spatial multiplexing transmission of 16 streams using a channel matrix of 256 × 16 size. In Non-Patent Document 1, in order to transmit a plurality of signal sequences (streams), the directivity is formed by analog beam forming using rotation of a complex phase amount in a wireless analog circuit and digital baseband circuit. Performed in conjunction with digital beamforming in the digital domain. This avoids excessive use of analog / digital (A / D) converters and digital / analog (D / A) converters, reduces power consumption, and takes measures against insufficient line gain when feeding back channel information. Hereinafter, an outline of the technique disclosed in Non-Patent Document 1 will be described.

FIG. 27 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. The wireless station apparatus shown in FIG. 1 includes modulators 901-1 to 901-N (MOD # 1 to MOD # N), a precoder 902, an IFFT (Inverse Fast Fourier Transform) and a GI (Guard Interval). and applying circuit _{903-1~903-N 0,} a D / a converter _{904-1~904-N 0,} the up-converter (UC) _{905-1~905-N 0,} downconverter (DC) 906-1 a _{~906-N 0,} and a / D converter _{907-1~907-N 0,} GI removal & FFT: a (fast Fourier transform) circuit _{908-1~908-N 0,} the postcoder 909, a demodulator 910-1~910-N (DEM # 1~DEM # N), a TDD switch (TDD-SW) 911, a distributor coupler (HYB) _{912-1~912-N 0,} the phase shifter 913 -1 Comprises a _{-1~913-N 0} -M 0, and distributor coupler (HYB) _{915-1~915-M 0,} and antenna element _{916-1~916-M 0.} Where N corresponds to the number of multiplexed time of performing spatial multiplexing (the number of streams), N ₀ is the number of signal lines for performing signal processing for digitally beamforming, M ₀ is representative of the number of antenna elements (M ₀ ≧ N ₀ ≧ N). Furthermore the antenna elements _{916-1~916-M 0} constitute an array antenna as a whole.

Antenna element _{916-1~916-M 0} from the distributor coupler _{912-1~912-N 0} is common to transmission and reception. Also, TDD switches 911, up from the distributor coupler _{912-1~912-N 0} a connection to the antenna element _{916-1~916-M 0,} from the modulator 901-1~901-N corresponding to the transmission system converter and _{905-1~905-N 0,} switching between the demodulator 910-1~910-N from the down-converter _{906-1~906-N 0} corresponding to the receiving system. For example, the up converter 905-n and the distribution coupler 912-n are connected during transmission, and the down converter 906-n and the distribution coupler 912-n are connected during reception (n = 1,..., N ₀ ). An entire control circuit (not shown) manages the frame period and the transmission / reception timing, and switching of the TDD switch 911 is also performed by this control circuit.

Further, the phase shifter _{913-1-1~913-N 0} -M 0 adjusts the phase relationship between the transmit and receive signals in accordance with the beam pattern defined in advance, the phase rotation amount by the control circuit (not shown) Is also managed. The phase here is the same as the directivity control in the phased array antenna. For example, as directional gain in a predetermined direction across the antenna elements _{916-1~916-M 0} is the maximum, at each antenna element _{916-1~916-M 0} against incoming waves from that direction A complex phase corresponding to a value obtained by dividing the path length difference by the wavelength is adjusted. Thus, each antenna element _{916-1~916-M 0} to be able to transmit and receive signals in phase.

Here, the directivity is obtained by dividing a horizontal azimuth angle θ and a vertical azimuth angle φ by a predetermined angle step size, and selecting a corresponding complex (θ _i , φ _j ) for each menu. phase shifter sets of phase as a set _{913-n-1~913-n-} M 0 (n = 1, ..., n 0) to adjust the amount of phase. As a result, for example, one virtual element for the n-th signal sequence in the whole antenna elements 916-1 to 916 -M ₀ , the phase shifters 913 -n-1 to 913 -n-M ₀ , and the distribution coupler 912-n Behave as a directional antenna. These virtual directional antenna is physically via distribution couplers _{915-1~915-M 0,} will share the antenna element _{916-1~916-M 0.}

  Further, in the following description, a case will be described as an example in which communication is performed by forming a signal on the frequency axis as in an OFDM (Orthogonal Frequency Division Multiplexing) modulation method. Note that even in the case of single carrier transmission, when performing equalization processing on the frequency axis, the signal is once converted into a signal on the frequency axis, so that precoding processing and postcoding processing are performed on such a frequency axis. The same discussion is possible without regard to OFDM or single-carrier transmission, assuming that the processing is performed after the signal is converted into a signal of the same type.

The specific signal flow is as follows. First, signal transmission will be described. Each of the modulators 901-1 to 901-N generates a transmission signal of each stream to be spatially multiplexed. The precoder 902 appropriately performs signal synthesis between a plurality of virtual directional antennas, and enables efficient signal separation on the receiving station side. The pre-coding processing is equivalent to multiplication example N ₀ systematic virtual directional antenna actually unitary transformation matrix at the time of singular value decomposition of the MIMO channel matrix between the signal lines of the N lines for transmitting and receiving. Thereby, so-called eigenmode transmission is realized, and efficient transmission is realized. IFFT & GI imparting circuit _{903-1~903-N 0} is a transmission signal sequence formed in this way, into a signal on the time axis from the signal on the frequency axis, imparting guard intervals. If necessary, waveform shaping between symbols is also performed here. D / A converter _{904-1~904-N 0} is a digital signal generated in this manner, into an analog signal. Upconverter _{905-1~905-N 0} is the analog signal, converts the baseband signal to the radio frequency signals.

At the time of transmission, the TDD switch 911 connects the up-converter 905-n and the distribution coupler 912-n (n = 1,..., N ₀ ). It should be noted, 1~N ₀ of subscript is all behave in the same way. The distributing coupler 912-n (n = 1,..., N ₀ ) distributes the signal in the radio frequency band to signals of the number of antenna systems M ₀ , and disperses the signals to the phase shifters 913-n-1 to 913-n. input to -M _0. For example the phase shifter _{913-1-1~913-1-M 0} is first directional orientation corresponding to the signal sequence (θ _i, φ _j) on an analog signal to adjust the predetermined complex phase corresponding in is performed to transmit a first signal sequence after the adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} Similarly, the phase shifters 913 -N _{0-1 to} 913 -N ₀ -M ₀ are provided with a predetermined complex phase corresponding to the directivity direction (θ _{i ′} , φ _{j ′} ) corresponding to the N _{0 th} signal sequence. the adjustments performed on the analog signal and transmits the signal sequence of the N after adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} Incidentally, the distribution coupler 915-m (m = 1, ..., M 0) , the corresponding phase shifters 913-1-m, 913-2-m, ..., a signal input from _913-N 0 -m The signals are combined and output to the antenna element 916-m.

Next, reception of a signal will be described. The signals received by the antenna elements 916-1 to 916 -M ₀ are respectively distributed to the N ₀ signals by the distribution couplers 915-1 to 915 -M ₀ , and the respective phase shifters 913-1-1 correspond to the respective N ₀ systems. It is output to the _{~913-N 0} -M 0.
For example the phase shifter _{913-1-1~913-1-M 0} is first directional orientation corresponding to the signal sequence (θ _i, φ _j) on an analog signal to adjust the predetermined complex phase corresponding The adjusted first signal sequence is input to the distribution coupler 912-1. The distribution coupler 912-1 combines these input signals, and inputs the combined signal to the down converter 906-1 via the TDD switch 911.
Similarly, the phase shifters 913 -N _{0-1 to} 913 -N ₀ -M ₀ are provided with a predetermined complex phase corresponding to the direction (θ _{i ′} , φ _{j ′} ) of the directivity corresponding to the Nth signal sequence. the adjustments performed on the analog signal, and inputs the N-th signal sequence after the adjustment to the distributor coupler 912-N _0. Distributor coupler _{912-N 0} synthesizes these input signals, and inputs the combined signal, the down-converter _{906-N 0} via the TDD switch 911.

Downconverter _{906-1~906-N 0} is down-converts the signal of the radio frequency into a baseband signal. The A / D converters 907-1 to 907 -N ₀ convert an analog baseband signal obtained by down-conversion into a digital baseband signal. Here, based on the symbol timing managed by the timing detection circuit (not shown), GI removal & FFT circuit 908-1~908-N ₀ removes a guard interval from the digital baseband signal, the signal in the time axis Convert to frequency axis signal. Postcoder 909, the cross talk component between the signal sequence (stream) which has been processed by the GI removal & FFT circuit _{908-1~908-N 0} and signal separation in the frequency domain, corresponding signals after the crosstalk component separation To demodulators 910-1 to 910-N. The demodulators 910-1 to 910-N reproduce and output data by a predetermined signal detection process.

Here, although low-noise amplifier of the power amplifier and the receiver of the transmission side is not explicitly described, generally downstream of the up-converter 905-1~905-N ₀ (code "A _1" - "A _N0" the power amplifier installed in the position), front (code _{"B 1"} in the down converter _{906-1~906-N 0} ~ installing low-noise amplifier in a position) of the _{"B N0".} The power amplifier and the low-noise amplifier have different amounts of complex phase rotation, and the amount of transfer rotation may be different for each frequency. However, factors to get a difference in phase rotation amount on both transmit and receive between the TDD switch 911 and antenna element 916-1~916-M ₀ is excluded, the symmetry of the channel at the time of reception and time of transmission Will be saved. Therefore, setting the phase rotation quantity of the phase shifter _{913-1-1~913-N 0} -M 0, it is possible to use the same value when it transmits and receives.

The above is the outline of the Massive MIMO technology using hybrid beamforming. Here in the example directional orientation (above corresponding to the phase rotation amount or the signal sequence for setting in phase shifters _{913-1-1~913-N 0} -M 0, the phase shifter 913-1-1～ in _{913-1-M 0 (θ i,} φ j), the phase shifter _{913-N 0 -1~913-N 0} -M 0 (θ i ', φ j')) acquisition method, such as the present invention Although not described because it is not directly related to the features described above, it can be obtained by conventional techniques such as Non-Patent Document 1.

[Expansion of Massive MIMO technology when line of sight is dominant]
In the above description of the Massive MIMO technology, since it is mainly assumed to be used in an access system, it is generally assumed that the environment is a multipath environment. However, even in the access system, if the small cell base station is installed at the top, and it is possible to look down and look roughly, it may be necessary to operate in a non-multipath environment. This is particularly remarkable in the case of a backhaul line, and is assumed to be used in an environment in which a so-called Rice coefficient K is 10 dB or more and a multipath component is only about 1/10 or less of a line-of-sight component. In this case, the difference between the line gain corresponding to the first singular value and the line gain corresponding to the second singular value or more is expected to be 20 dB or more, and spatial multiplex transmission of two or more streams is inefficient. It is expected that

  In such an environment, as shown in Non-Patent Document 2, all antenna elements are divided into a plurality of sets by utilizing the high line gain efficiency corresponding to the first singular value, and a sub-array is set for each set. The configuration is effective. By arranging the sub-arrays spatially separated, it is effective to reduce the correlation between the sub-arrays and to transmit the transmission corresponding to the first singular value in parallel with low correlation. Similarly, in Non-Patent Document 3, if the antenna aperture length of the subarray here is narrow, the line-of-sight wave is dominant, and the correlation between the antenna elements in the subarray is sufficiently strong, the transmission / reception weight of each antenna element is the frequency. A "time-axis beamforming technique" has been proposed, which can be treated as a constant having no dependence, and in this case, weight multiplication can be performed in units of sampling data on the time axis.

This is a relative value of channel information for each antenna element (a relative value of channel information with respect to channel information of an antenna element serving as a reference, when the element spacing is narrow and the correlation of antenna elements is strong, If the complex phase of the channel information of the k-th frequency component of the element is ψ _ref ^(k) , the frequency dependence of the complex phase of information obtained by multiplying each antenna element by Exp {−j _ref ^(k) } The gender is a method based on the fact that it is almost constant. For example, at the time of reception, the complex phase of the reception weight for synthesizing the received signals of these antenna elements such that the complex phases are the same is substantially constant in the entire frequency band, and Thus, it is possible to use reception weights having the same constant. Generally, a Fourier transform of a function that becomes a constant on the frequency axis becomes a δ function. Therefore, the weight obtained by transforming the reception weight on the frequency axis on the time axis by IFFT only needs to consider the component at t = 0. Become. In other words, since it is not necessary to perform signal processing in consideration of the delay wave component, the time base, which is a predetermined coefficient for each antenna element, is directly added to the sampling data obtained by sampling the analog / baseband received signal by the A / D converter. If the reception weight is multiplied, the directivity can be completely formed only by signal processing on the time axis without converting the reception signal into a signal on the frequency axis by FFT processing or the like.

  The coefficient for rotation of the complex phase to be multiplied as the time axis weight is obtained by the following equations (1) to (3).

In the above equation, S _i (n) represents the sampling data of the n-th sample of the i-th antenna in the received training signal, and S _i (n) ^* represents the complex conjugate of S _i (n). Represent. N _FFT assumes a predetermined periodicity, and indicates a periodicity value that is significant in correlation detection, such as the number of FFT points in OFDM. ψ _j is the amount of rotation of the complex phase (on the receiving side) performed by time axis beamforming. The function angle (x) is a function representing the complex phase of the complex number x, and is a value determined by the ratio of the real part of x to the imaginary part of x and the sign of the real part and the imaginary part. Although the in the correlation calculation in the formula (1) performs a correlation operation over N _FFT samples is the number of FFT points in the case of the OFDM signal as a predetermined periodicity, be an integer multiple of example N _FFT Correlation calculations over other sample numbers may be performed so that periodicity is maintained.

Here, as is clear from the equation (2), the complex phase of the complex coefficient c _j given by the above equation (1) and the complex phase of the time axis weight w _j given by the above equation (2) have inverted signs. It has become something. In this sense, in an embodiment of the present invention to be described later, the complex phase of the complex coefficient c _j given by the equation (1) corresponding to the relative channel information and the complex phase of the time axis weight w _j are found. Are equivalent.

FIG. 28 is a functional block diagram showing a configuration example (separated sub-array type) of a wireless station device using time-axis beamforming in the conventional technology described in Non-Patent Document 3. The wireless station device shown in FIG. 1 includes a baseband signal processing circuit 140 and transmission / reception signal processing circuits 929-1 to 929 -N. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 929-n (n = 1,..., N) includes a time axis transmission weight multiplication circuit 921-n, D / A converters 922-n-1 to 922-n-M, and an up converter 923. -N-1 to 923-n-M, down converter (DC) 924-n-1 to 924-n-M, A / D converters 925-n-1 to 925-n-M, and time axis A reception weight multiplication circuit 926-n and a TDD switch (TDD-SW) 927-n are provided. TDD switch 927-n is connected to antenna elements 928-n-1 to 928-n-M. Here, N corresponds to the multiplexing number (the number of streams) when performing spatial multiplexing, and the transmission / reception signal processing circuits 929-1 to 929 -N are mounted in N systems in total. M represents the number of antenna elements of the sub-array mounted on each of the transmission / reception signal processing circuits 929-1 to 929-N. In the description of FIG. 27 it has a total number of antenna elements and M _0, but here since the subarray constitute the entire antenna number of antenna elements of each subarray were labeled as different values M.

  Here, the transmission / reception signal processing circuits 929-1 to 929-N (the transmission / reception signal processing circuit 929-n has sub-array antenna elements 928-n-1 to 928-n-M) are described in Non-Patent Document 2. It is assumed that they are installed spatially apart from each other. Further, the baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 929-1 to 929 -N by wire, and the digital baseband signal is transferred over this wire. As in the case of FIG. 27, the entire control circuit, not shown, is mounted on the baseband signal processing circuit 140 and manages the frame period and the transmission / reception timing. Switching of 927-N is also managed here.

Furthermore, the time-axis beamforming technology basically presupposes signal processing on the time axis. However, even when a signal on the frequency axis is formed as in the OFDM modulation method, the signal on the frequency axis is formed by FFT processing and IFFT processing. Can be converted into a signal on the time axis, and by performing signal processing on the time axis signal, the time axis beamforming technique can be similarly applied to the OFDM modulation method together with the single carrier transmission. However, assuming the signal processing on the frequency axis in FIG. 27, a GI removal & FFT circuit _{908-1~908-N 0} for IFFT & GI imparting circuit _{903-1~903-N 0} and FFT processing for IFFT processing Have been described separately from the modulators 901-1 to 901-N and the demodulators 910-1 to 910-N. However, since signal processing on the frequency axis is not assumed here, FIG. Even when the OFDM modulation method is used, the functions of the FFT processing and the IFFT processing are included in the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N (or the signal separation circuit 141). Therefore, these notations are omitted. Therefore, irrespective of the OFDM modulation method or single carrier transmission, the input / output signals from the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are signals on the time axis. . Further, the signal separation circuit 141 performs signal separation between the respective signal sequences. Here, it is also possible to perform signal separation on the time axis. Frequency-dependent signal separation may be performed. In this sense, it is assumed that the input to the demodulators 130-1 to 130-N is a signal on the time axis and a signal on the frequency axis. The description will be made assuming that the user inputs “.

  The specific signal flow is as follows. First, signal transmission will be described. Each of the modulators 120-1 to 120-N generates a transmission signal of a time base digital baseband of each stream to be spatially multiplexed, and inputs each of the signals to a time base transmission weight multiplying circuit 921-1 to 921-N. I do. The time axis transmission weight multiplying circuits 921-n (n = 1,..., N) are each a sub-array for forming the directivity of the digital signal input from the modulator 120-n by the transmission / reception signal processing circuit 929-n. The signal is converted into a digital signal obtained by multiplying the transmission weights corresponding to the antenna elements 928-n-1 to 928-n-M. The D / A converters 922-n-1 to 922-n-M convert the digital signal multiplied by the transmission weight into an analog baseband signal, and the up-converters 923-n-1 to 923-n-M , Convert the analog baseband signal into a signal in a radio frequency band. At the time of transmission, the TDD switch 927-n connects the upconverter 923-nm (m is an integer of 1 to M) to the antenna element 928-nm. The signals input from the up-converters 923-n-1 to 923-n-M are transmitted from the antenna elements 928-n-1 to 928-n-M, and the transmission / reception signal processing circuits 929-1 to 929 are transmitted. A directional beam is formed every −N.

  Next, reception of a signal will be described. Signals received by the antenna elements 928-n-1 to 928-n-M (n = 1,..., N) are input to the down-converters 924-n-1 to 924-n-M via the TDD switch 927-n. Is done. The down converters 924-n-1 to 924-n-M convert radio frequency signals into analog baseband signals. The A / D converters 925-n-1 to 925-n-M convert an analog baseband signal into a digital baseband signal. This digital baseband signal is input to the time axis reception weight multiplying circuit 926-n. The time axis reception weight multiplying circuit 926-n multiplies each of the input signals by the reception weight corresponding to each of the antenna elements 928-n-1 to 928-n-M, and adds and synthesizes the signals after the reception weight multiplication. And converts them into a single signal sequence. That is, the signals are converted into a total of N signal sequences (streams) by the time axis reception weight multiplying circuits 926-1 to 926 -N, and these signals are input to the signal separation circuit 141. The signal separation circuit 141 performs signal separation by suppressing the crosstalk component between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The demodulators 130-1 to 130-N reproduce and output data by a predetermined signal detection process.

  The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted to a frequency axis signal by FFT processing and then performed on the frequency axis. Alternatively, only the signal processing performed by the transmission / reception signal processing circuits 929-1 to 929 -N may be sufficient, and the signal separation circuit 141 may not perform any processing. In any case, however, the details of the signal separation method here are omitted because they are not directly related to the present application.

  The time axis transmission weights used in the time axis transmission weight multiplication circuits 921-1 to 921-N and the time axis reception weights used in the time axis reception weight multiplication circuits 926-1 to 926-N are shown in FIG. No time axis transmission / reception weight acquisition means acquires. Similarly, a control circuit (not shown) manages the value of the time axis transmission / reception weight used there. For example, based on the sampling data acquired by the A / D converters 925-n-1 to 925-n-M for a training signal transmitted by a wireless station as a communication partner, the reference antenna element is transmitted over a predetermined number of samples. A correlation value between the antenna elements (for example, 928-n-1) may be determined, and the correlation value may be determined based on the complex phase. The complex phase values of the time-axis reception weight and the time-axis transmission weight generally do not match because the amount of rotation of the complex phase of a power amplifier (not shown) and a low-noise amplifier differs between individual amplifiers. By using the calibration method of the implicit feedback, it is possible to convert the time axis reception weight to the time axis transmission weight. The transmission and reception weights thus obtained are stored in the memory for each corresponding wireless station device. Then, at the time of transmission and reception, weight multiplication is performed by the time axis transmission weight multiplication circuits 921-1 to 921-N and the time axis reception weight multiplication circuits 926-1 to 926-N based on the values of these transmission and reception weights. Will be.

  FIG. 29 is a functional block diagram showing a configuration example (shared sub-array type) of a wireless station device using time-axis beam forming in the conventional technology described in Non-Patent Document 3. 28, the same functions as those in FIG. 28 are denoted by the same reference numerals. In the figure, a radio station apparatus 942 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 929-1 to 929-N, distribution couplers (HYB) 941-1 to 941-M, and an antenna element 928- 1 to 928-M. In FIG. 28, the transmission / reception signal processing circuits 929-1 to 929-N are housed in different housings on the assumption that they are installed at locations spatially separated for each sub-array, and a baseband signal processing circuit of another housing is provided. A configuration in which a wired connection is made between the wireless LAN and the wireless LAN 140 is shown. On the other hand, in FIG. 29, all the transmission / reception signal processing circuits 929-1 to 929 -N and the baseband signal processing circuit 140 are configured as a wireless station device 942 in the same housing, and the entire antenna elements 928-1 to 928 -M are used. Is shared by For this reason, for example, at the time of transmission, the signals from the TDD switches 927-1 to 927-N of the transmission / reception signal processing circuits 929-1 to 929-N are combined by the distribution couplers 941-1 to 941-M and combined. The transmitted signal is transmitted from the antenna elements 928-1 to 928-M. Similarly, at the time of reception, the signals received by the antenna elements 928-1 to 928-M are distributed by the distribution couplers 941-1 to 941-M. That is, the distribution coupler 941-m (m = 1,..., M) distributes the signal received by the antenna element 928-m to the TDD switches 927-1 to 927-N and inputs the signal. All other signal processing is common to FIGS. 28 and 29.

  Similarly, FIG. 30 is a functional block diagram of another configuration example (shared sub-array type) of a wireless station device using time-axis beamforming in the conventional technology described in Non-Patent Document 3. 29, the same parts as those of the radio station apparatus shown in FIG. 29 are denoted by the same reference numerals, and description thereof will be omitted. The wireless station device 945 shown in FIG. 14 includes a baseband signal processing circuit 140, time-base transmission weight multiplying circuits 921-1 to 921-N, addition and synthesis units 943-1 to 943-M, and a D / A converter. 922-1 to 922-M, up converters 923-1 to 923-M, down converters 924-1 to 924-M, A / D converters 925-1 to 925-M, and a duplicator 944-1 944-M, a time axis reception weight multiplying circuit 926-1 through 926-N, a TDD switch 927, and antenna elements 928-1 through 928-M.

  In FIG. 29, the transmission / reception signal processing circuits 929-1 to 929-N are individually mounted for each sub-array, but the D / A converters 922-n-1 to 922-n-M and the up-converters 923-n-1 to 923-n-1 923-n-M, down converters 924-n-1 to 924-n-M, A / D converters 925-n-1 to 925-n-M, and TDD switches 927-1 to 927-N are shared. It is possible (n = 1,..., N). Therefore, in FIG. 30, an N-system digital baseband signal generated by the time-base transmission weight multiplying circuits 921-1 to 921-N (N planes are mounted as a whole) is added to the adder / synthesizer 943 in sampling data units. 1-943-M, and combine them into one system, and convert them from digital signals to analog signals by D / A converters 922-1 to 922-M. Similarly, on the receiving side, the duplicators 944-1 to 944-M duplicate the digital baseband signals generated by the A / D converters 925-1 to 925-M into N-system signals in sampling data units, The duplicated signal is input to a time axis reception weight multiplying circuit 926-1 to 926-N (N planes are mounted as a whole). Each of the time axis reception weight multiplying circuits 926-1 to 926 -N multiplies an input signal by a reception weight, and adds and combines the results to convert them into one system signal. That is, the signals are converted into a total of N signals by the time axis reception weight multiplying circuits 926-1 to 926 -N, and these signals are input to the signal separating circuit 941.

  Thereby, the D / A converters 922-n-1 to 922-n-M are redundantly mounted, the up-converters 923-n-1 to 923-n-M are redundantly mounted, and the down-converters 924-n-1 to 924-924. Avoid duplicate implementation of nM, duplicate implementation of A / D converters 925-n-1 to 925-n-M, duplicate implementation of TDD switches 927-1 to 927-N, reduce circuit scale, reduce power consumption, etc. Has led to the reduction of

  Here, in actual operation, the wireless station device shown in FIG. 28 and the wireless station device shown in FIG. 29 or 30 communicate with each other. For example, a base station apparatus has a degree of freedom in installation like a building rooftop, and a sub-array can be installed at a plurality of locations. On the other hand, when the restriction on the installation of the wall surface of the building or the like is large, the terminal station apparatus has a configuration in which FIG. 28 is a base station apparatus and FIG. 29 or 30 is a terminal station apparatus. It is possible to increase the degree of freedom in installation by sharing the antenna. Alternatively, for example, if the transmission capacity per terminal station apparatus does not require spatial multiplexing, FIG. 29 or FIG. 30 is a base station apparatus, and FIG. 28 is one of transmission / reception signal processing circuits 929-1 to 929-N. It is assumed that the terminal station apparatus is mounted with only the terminal station apparatus (for example, only the transmission / reception signal processing circuit 929-1 in FIG. 28), and a plurality of terminal station apparatuses and one base station apparatus perform multi-user MIMO transmission. It is also possible.

Incidentally, Figures 27 and 28, with regard to the correspondence between FIGS. 29 and 30, for example a phase shifter _{913-1-1~913-N} rotation amount of complex phase carried out at 0 -M ₀ in FIG. 27 [psi _alpha ( α is a phase shifter _{913-1-1~913-N} 0 if the corresponding) to the identification number for the -M _0, the time axis transmission weight multiplying circuit 921-1~921-N, the time axis reception weight multiplication circuits 926 -1 to 926-N correspond to multiplying the signal sequence of the corresponding antenna element by Exp {j} _α }. In other words, while was conversion of an analog circuit the signals transmitted and received from each antenna element in Figure 27 (i.e. the phase shifter _{913-1-1~913-N 0} -M 0), 28, 29 and In FIG. 30, it is equivalent to converting a signal transmitted / received from each antenna element by a digital circuit (that is, a time axis transmission weight multiplication circuit 921-1 to 921-N, a time axis reception weight multiplication circuit 926-1 to 926-N). I do. This has the advantage that the number of A / D converters 907-1 to 907-N ₀ and D / A converters 904-1 to 904-N ₀ can be reduced in FIG. 27, while FIG. 29 and FIG. 30, as described in Non-Patent Document 3, has the advantage that the resolution of directivity formation is increased and that simple and efficient feedback of channel information is possible.

It should be noted that the phase rotation by the phase shifter normally gives phase rotation by selectively passing through a delay line corresponding to the amount of phase rotation on the device. Therefore, when a phase rotation of x is given as an absolute value, a negative phase rotation (delay) of the complex phase rotation amount is performed with a delay corresponding to the phase x as a signal, and code consistency cannot be obtained. . However, in the following description for convenience, will be referred to as "the rotation amount of complex phase carried out in the phase shifter [psi _alpha" when giving a phase rotation equivalent to multiplication of the coefficient Exp {jψ _α} as a signal with the phase shifter I do.

[Calibration technology in channel information feedback]
In general, when directivity is formed using a plurality of antenna elements on the transmission side, it is necessary to feed back MIMO channel information including the techniques described in Non-Patent Documents 1 to 3. At this time, if the number of antenna elements becomes enormous, the amount of channel information to be fed back becomes enormous, so various measures are required. In the Massive MIMO system as described above, in order to acquire channel information of a forward link in a transmission direction, channel information of a reverse link in a reception direction is used, and a complex phase of a reception signal generated by a circuit such as a low noise amplifier used at the time of reception is used. The relationship between the amount of rotation and the amount of rotation of the complex phase of the transmission signal generated by a circuit such as a high power amplifier used at the time of transmission is converted, and the channel information of the reverse link is multiplied by a predetermined calibration coefficient to convert the It is possible to obtain channel information. Generally, these techniques are known as implicit feedback techniques (for example, see Non-Patent Document 4). The details of the calibration process will be described below.

  In an actual wireless communication device, in signal processing on the transmission side, a signal is often amplified by a high power amplifier immediately before transmission. In this case, there is an error in the amplification factor due to the individual difference of the high power amplifier, and the complex phase may rotate with a different value for each high power amplifier in the high power amplifier. Similarly, in signal processing on the receiving side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, the amplification factor may have an error due to the individual difference of the low noise amplifier, and the complex phase may rotate with a different value for each low noise amplifier in the low noise amplifier. In some cases, the amplification factor and the amount of phase rotation may have frequency dependency. If the individual difference between the amplification factor and the rotation amount of the complex phase is so large that it cannot be ignored, it is necessary to perform a calibration process when estimating downlink channel information from uplink channel information. Since the errors in the amplification factor and the amount of phase rotation are substantially stable in time, the errors in the amplification factor and the amount of phase rotation are measured in advance, and are increased using a coefficient for canceling the influence of the error. The link channel information is converted into downlink channel information. In the following description, a description will be given mainly of the calibration processing performed on the base station apparatus side having a plurality of antenna elements. However, the same is also possible on the terminal station apparatus side, and the common processing is common to general wireless station apparatuses. It is an explanation.

  In the above description, the amplitude and the complex phase may be changed by a high-power amplifier or a low-noise amplifier (strictly, a circuit of a transmission system and a reception system including circuits of other filters and the like). In this case, it has been described that the calibration coefficient for performing the correction according to the change in the amplitude or the complex phase is acquired in advance, and this is used for the correction. A known technique may be used for the calibration process, but an example of the calibration process will be described below.

  FIG. 31 is a diagram illustrating asymmetry of channel information between the uplink and the downlink. In the figure, reference numerals 955-1 to 955-3 indicate wireless modules, reference numerals 951-1 to 951-3 indicate high power amplifiers (HPAs), and reference numerals 952-1 to 952-3 indicate low noise amplifiers (LNA). Reference numerals 953-1 to 953-3 indicate TDD switches, and reference numerals 954-1 to 954-3 indicate antenna elements.

Here, for the explanation of the calibration technique, only the functions that affect the channel information in the wireless station device are extracted, so that the configurations other than those illustrated are omitted. Therefore, regarding the configuration of the wireless modules 955-1 to 955-3, for convenience, the high power amplifiers 951-1 to 951-3, the low noise amplifiers 952-1 to 952-3, the TDD switches 953-1 to 953-3, and the antenna Although only the elements 954-1 to 954-3 are shown, functions such as an up-converter and a down-converter are mounted in the latter stage (the former stage). Further, for example, it is assumed that a wireless station device having a plurality of antennas has a plurality of wireless modules 955-1 to 955-2 mounted in one housing, and a wireless module 955-3 opposes this. This corresponds to a diagram that extracts and describes a wireless module 955-3 corresponding to one antenna element of a wireless station device that communicates. When the signal passes through each of the high power amplifiers 951-1 to 951-3, the amplitude and the complex phase are Z _{HPA # 1} (f _k ), Z _{HPA # 2} (f _k ), and Z _{HPA # 3} (f _k ) shall change. When the signal passes through each of the low-noise amplifiers 952-1 to 952-3, the amplitude and the complex phase are Z _{LNA # 1} (f _k ), Z _{LNA # 2} (f _k ), and Z _{LNA # 3} (f _k ) Shall change. Here, it is assumed that there is a frequency dependency as a general condition, and the frequency “(f _k )” is described for the k-th frequency component.

Here, for example, channel information when a signal is transmitted from the wireless module 955-1 and the wireless module 955-2 to the test wireless module 955-3 will be described. Here, the antenna element 954-1 of the wireless module 955-1, the channel information in the space between the antenna elements 954-3 of the wireless module 955-3 is represented by _h 1 _{(f k),} the wireless module 955 channel information on the space between the antenna elements 954-2 and antenna element 954-3 of the wireless module 955-3 -2 is represented by _h 2 _{(f k).}

At this time, the channel information when the signal is actually transmitted from the wireless module 955-1 to the wireless module 955-3 indicates a change in the space h ₁ (f _k ) accompanying the passage of the high power amplifier 951-1. It is observed as a value multiplied by a coefficient Z _{HPA # 1} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying the passage through the low noise amplifier 952-3. Similarly, channel information when a signal is transmitted from the wireless module 955-2 to the wireless module 955-3 includes a coefficient Z indicating a change associated with the passage of the high power amplifier 951-2 in h ₂ (f _k ) in space. It is observed as a value multiplied by _{HPA # 2} ( _fk ) and a coefficient ZLNA _{# 3} ( _fk ) indicating a change accompanying passage through the low noise amplifier 952-3. Therefore, a channel from the wireless module 955-1 to the wireless module 955-3 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 3} (f _k ). The channel from the wireless module 955-2 to the wireless module 955-3 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 3} (f _k ). Therefore, between the wireless module 955-1 and the wireless module 955-2, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), Z _{HPA # 2} (f _k ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same on the receiving side. When the signal transmitted from the wireless module 955-3 is received by the wireless module 955-1, the channel information is stored in the high power amplifier 951 in h ₁ (f _k ) in space. Observed as a value obtained by multiplying a coefficient Z _{HPA # 3} (f _k ) indicating a change accompanying passage of -3 and a coefficient Z _{LNA # 1} (f _k ) indicating a change accompanying passage of the low noise amplifier 952-1. Is done. Similarly, when a signal transmitted from the wireless module 955-3 is received by the wireless module 955-2, the channel information indicates a change in h ₂ (f _k ) in space due to the passage of the high power amplifier 951-3. The coefficient Z _{HPA # 3} (f _k ) is multiplied by a coefficient Z _{LNA # 2} (f _k ) indicating a change accompanying the passage through the low noise amplifier 952-2. Therefore, a channel from the wireless module 955-3 to the wireless module 955-1 is represented by Z _{HPA # 3} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). A channel from the wireless module 955-3 to the wireless module 955-2 is represented by Z _{HPA # 3} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). Therefore, between the wireless module 955-1 and the wireless module 955-2, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), Z _{LNA # 2} (f _k) ) / Z _{LNA # 1} (f _k ).

To summarize Here again, the channel information when receiving the signal transmitted from the wireless module 955-3 corresponding to the reverse link on the left side of the radio station apparatus by the wireless module _{955-1 h 1 (f k) ·} Z _{HPA # 3} (f _k ) · Z _{LNA # 1} (f _k ). When the signal transmitted from the wireless module 955-1 corresponding to the forward link is received by the wireless module 955-3, the channel information is h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA # 3} (f _k ). The calibration coefficient of the wireless module 955-1 is given by the following equation (4).

Similarly, the channel information when receiving the signal transmitted from the wireless module 955-3 corresponding to the reverse link on the left side of the radio station apparatus by the wireless module _{955-2 h 1 (f k) ·} Z HPA # 3 (F _k ) · Z _{LNA # 2} (f _k ). When the signal transmitted from the wireless module 955-2 corresponding to the forward link is received by the wireless module 955-3, the channel information is h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA # 3} (f _k ). The calibration coefficient of the wireless module 955-2 is given by the following equation (5).

Here, for example, each channel information in the reverse link to be acquired by the wireless module _{955-1~955-2 h 1 (f k) ·} Z HPA # 3 (f k) · Z LNA # 1 (f k) and h ₁ (f _k ) · Z _{HPA # 3} (f _k ) · Z _{LNA # 2} (f _k ). When this is multiplied by the calibration coefficients C ₁ (f _k ) and C ₂ (f _k ), h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA # 3} (f _k ) and h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA # 3} (f _k ) .

  At the time of actual operation, when the wireless module of the actual communication partner is different from the wireless module 955-3, strictly speaking, the estimated value of the channel information of the forward link obtained by multiplying this calibration coefficient is Of the channel information itself. However, even in that case, the value obtained by multiplying the channel information of the true forward link regarding the radio modules 955-1 and 955-2 by the common coefficient and the above-mentioned estimated value match, and the directivity formation is performed. Considering that there is no effect even if a common constant is multiplied to all antenna elements, there is no problem as feedback of channel information.

  Also, in the above description, the transmitting side of the target wireless station apparatus is described as a forward link, and the receiving side is described as a reverse link. However, when the target wireless station apparatus is a base station apparatus, the forward link is generally used. Is called a downlink, and the reverse link is called an uplink. Similarly, when the radio station apparatus of interest is a terminal station apparatus or a relay station apparatus, the forward link is usually called an uplink and the reverse link is called a downlink.

Satoshi Suyama, Tatsunori Ohara, Kiyun Shin, Yukihiko Okumura, "Effect of Analog Beamforming Configuration in Massive MIMO Using High Frequency Band Hybrid Beamforming", IEICE Technical Report, IEICE, March 2015 , Vol.114, no.490, RCS2014-337, p.213-218 Atsushi Ohta, Takuto Arai, Hirofumi Shirato, Kazuki Maruta, Tatsuhiko Iwakuni, Masataka Iizuka, "Line-of-sight environment millimeter-wave spatial multiplexing transmission technology using parallel transmission of the first eigenmode: method proposal and basic performance evaluation results," IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, August 2015, vol.115, no.181, RCS2015-144, p.73-78 Atsushi Ota, Hiroshi Shirato, Kazuki Maruta, Takuto Arai, Tatsuhiko Iwakuni, Masataka Iizuka, "Efficient Use of First Eigenmode Transmission in Massive MIMO with Line-of-Sight Environment", IEICE Technical Report, IEICE, 2015 November, vol.115, no.288, RCS2015-239, p.293-298 Hayato Fukuzono, Tomonori Murakami, Riichi Kudo, Yasushi Takatori, Masato Mizoguchi, "Experimental evaluation of implicit feedback in downlink multi-user MIMO-OFDM system", IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, November 2013 , Vol.113, no.301, RCS2013-187, p.79-84

In the above description, the implicit feedback technique is applied to the feedback of the channel information, and the channel information of the forward link is obtained from the channel information of the reverse link by the calibration process, regardless of the time axis beamforming and other methods. It was assumed that. This is based on the premise that channel information h ₁ (f _k ) and h ₂ (f _k ) between antenna elements 204-1 and 204-2 and antenna element 204-3 in FIG. It was assumed that the values would be the same between the two. However, even in the millimeter wave band, which is said to be abundant frequency resource, it is difficult to secure a bandwidth at the 1 GHz level in a relatively low frequency band, and currently, for example, 71 to 76 GHz / 81 to 86 GHz called E band is used for FDD. Is one of the promising options. This band is assumed to be used in FDD, but if the frequency is different from f _k ≠ f ′ _{k in} the uplink and downlink, h ₁ (f _k ) ≠ h ₁ (f ′ _k ) and h ₂ (f _k It is expected that symmetry with ≠ h ₂ (f ′ _k ) will not be maintained. Therefore, when the above-described wireless system is constructed using FDD, the symmetry of the channel is broken by using different frequencies, so that there has been a problem that implicit feedback cannot be used.

  In view of the above circumstances, an object of the present invention is to provide a technique that can use implicit feedback even when a wireless system is constructed using FDD.

  One embodiment of the present invention is a wireless communication device that forms directivity using one or more array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device, A communication unit that communicates with the other wireless communication device by using a first frequency used in the first communication unit and a second frequency different from the first frequency used in the downlink; A signal conversion unit provided in the system for converting a radio frequency analog signal received by the antenna element into a baseband digital signal; and a training signal transmitted by the other wireless communication device converted by the signal conversion unit. Using a digital signal corresponding to the, an antenna element serving as a reference selected from a plurality of antenna elements provided with the signal conversion unit, and other antenna elements A correlation calculation unit that calculates a correlation for each combination, and a rotation amount calculation unit that calculates a rotation amount of a complex phase to be given to a received signal of an individual antenna element based on a calculation result of the correlation by the correlation calculation unit. A complex to be given to a transmission signal of an individual antenna element based on information on a ratio between the first frequency and the second frequency and information calculated by the correlation calculator or the rotation amount calculator. A rotation amount prediction unit that predicts a rotation amount of a phase, and a phase rotation that manages a rotation amount of a complex phase to be given to a transmission signal for each antenna element based on the complex phase predicted by the rotation amount prediction unit. An amount management unit, a transmission signal generation unit that generates a signal addressed to the another wireless communication device, and a transmission amount for each antenna element used for transmission, the rotation amount of the complex phase managed by the phase rotation amount management unit. A phase rotation unit that applies a signal on an analog signal or a digital signal, and a signal transmission unit that transmits a signal output via the phase rotation unit to the other wireless communication device via the antenna element. And a wireless communication device comprising:

  One aspect of the present invention is the wireless communication device described above, further comprising a first signal distribution unit that branches a transmission signal generated by the transmission signal generation unit on an analog signal for each of the antenna elements, The rotation unit sets the rotation amount of the complex phase managed by the phase rotation amount management unit in a phase shifter provided for each antenna element.

  One embodiment of the present invention is the wireless communication device described above, further comprising a second signal distribution unit that branches the transmission signal generated by the transmission signal generation unit on a digital signal for each of the antenna elements, The rotation unit multiplies the rotation amount の of the complex phase for each antenna element managed by the phase rotation amount management unit by a coefficient Exp (jψ) on a digital signal for each antenna element.

  One aspect of the present invention is a wireless communication method performed by a wireless communication device that forms directivity using one or more array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device. A communication step of communicating with the other wireless communication device by using a first frequency used in an uplink and a second frequency different from the first frequency used in a downlink; A training signal transmitted by the other wireless communication device, which is provided in an element or a plurality of systems and converted by a signal conversion unit that converts a radio frequency analog signal received by the antenna element into a baseband digital signal. An antenna element serving as a reference selected from a plurality of antenna elements provided with the signal conversion unit using a digital signal corresponding to A correlation calculation step of calculating a correlation for each combination of tena elements, and a rotation amount for calculating a rotation amount of a complex phase to be given to a received signal of an individual antenna element based on a calculation result of the correlation in the correlation calculation step Based on a calculation step, information on a ratio between the first frequency and the second frequency, and a calculation result of the correlation calculation step or the rotation amount calculation step, the information is given to a transmission signal of an individual antenna element. A rotation amount prediction step for predicting a rotation amount of a power phase, and a rotation amount of a complex phase to be given to a transmission signal for each antenna element based on the complex phase predicted in the rotation amount prediction step. A phase rotation amount management step, a transmission signal generation step of generating a signal addressed to the other wireless communication device, and a phase rotation amount management step. The rotation amount of the complex phase managed in the loop is set to a phase rotation unit that gives an analog signal or a digital signal to a transmission signal for each antenna element used for transmission, and is output via the phase rotation unit. Transmitting the transmitted signal to the other wireless communication device via the antenna element.

  According to the present invention, it is possible to use implicit feedback even when a wireless system is constructed using FDD.

FIG. 3 is a functional block diagram illustrating a configuration example of a wireless station device according to the embodiment of the present invention. FIG. 3 is a functional block diagram illustrating a configuration example of a wireless station device according to the embodiment. FIG. 2 is a diagram illustrating a configuration example of a communication system according to the first embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. It is a functional block diagram showing the example of composition of the radio station device in a 2nd embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a wireless station device according to the embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. It is a figure showing an outline of channel information prediction in a linear array in which antenna elements are arranged in a straight line in a 3rd embodiment. It is a figure showing the example of the array antenna constituted in plane in the embodiment. It is a figure showing the example of the array antenna constituted in plane in the embodiment. FIG. 3 is a diagram illustrating a specific example of a square array antenna configured in a planar shape in the same embodiment. FIG. 3 is a diagram illustrating a specific example of a square array antenna configured in a planar shape in the same embodiment. FIG. 3 is a diagram illustrating a specific example of a square array antenna configured in a planar shape in the same embodiment. It is a figure which shows the specific example of the array antenna by which transmission and reception were isolate | separated in the planar form in the same embodiment. It is a figure which shows the specific example of the array antenna by which transmission and reception were isolate | separated in the planar form in the same embodiment. It is a figure showing an example of an antenna pattern used for prediction of the amount of rotation of a complex phase in the embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating another configuration example of the transmission / reception signal processing circuit using digital signal processing in the same embodiment. It is a figure showing an outline of a conversion method of a complex phase in a 4th embodiment of the present invention. It is a flow chart which shows a flow of processing of an implicit feedback method at the time of application of FDD in a 4th embodiment of the present invention. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 11 is a diagram illustrating a calibration process in the related art.

  An embodiment of the present invention will be described in detail with reference to the drawings. First, the basic principle of the embodiment of the present invention will be described. The terms “time axis” and “frequency axis” used in this specification may be expressed as “time domain” and “frequency domain”, but here, they are unified as “time axis” and “frequency axis”. Give an explanation.

[Overview of basic principles]
In the time axis beamforming technique, for example, at the time of reception, a main path of an incoming wave between a transmitting and receiving station is extracted, and signal processing is performed to direct the directional gain of the antenna element group in the direction. The transmission and reception weights used at this time are constants having no frequency dependence, and as a result, signal processing is reduced at various points. However, since it is based on digital signal processing, transmission / reception weights are multiplied individually for each antenna element in digital signal processing, and an A / D (analog / digital) conversion is performed separately for each antenna system. And a D / A (digital / analog) converter.

  However, multiplication processing of coefficients having no frequency dependency does not necessarily require digital signal processing if it is limited to, for example, only rotation processing of a complex phase without a change in amplitude. Specifically, by using a phase shifter that is an analog circuit and passing the analog signal through the phase shifter set to a predetermined complex phase rotation, signal processing substantially equivalent to weight multiplication processing is performed. Can be realized. In Non-Patent Document 1, directivity control is realized using a phase shifter. This is performed, for example, in a predetermined direction set in horizontal / vertical directions at intervals of 5 degrees. A combination set is determined in advance, and the phase rotation amount of each phase shifter is set according to the direction in which the beam obtained by some control procedure should be directed. However, this combination set of phase rotation amounts is selected from a preset menu, and it is necessary to search for the direction in which the reception level is the highest while individually transmitting a training signal for each direction. . However, in the time-axis beamforming, the amount of phase rotation of each antenna element is optimized based on the training signal transmitted from the terminal station apparatus side, so that the calculation of the amount of phase rotation used for forming directivity is simply fed back. In addition to this, the combination set of the amount of rotation of the complex phase can be set with a remarkably high degree of freedom without requiring a menu or the like in advance.

  Therefore, the wireless station device (wireless communication device) according to the embodiment of the present invention employs digital assist type analog beamforming. That is, the wireless station device performs the calculation of the amount of rotation of the complex phase performed by each phase shifter by digital signal processing, and controls the phase shifter using the complex phase value obtained by the digital signal processing. Thus, a desired complex phase is rotated to form directivity on an analog signal.

  Here, the process of calculating the complex phase of the transmission / reception weight of time axis beam forming performed in digital signal processing need not necessarily be performed simultaneously by each antenna element. Alternatively, even when digital signal processing is performed when calculating the complex phase of the transmission / reception weight of time axis beamforming, the time rate at which signal processing for this is performed is the same as that of normal communication for transmitting and receiving user data. It is expected to be overwhelmingly less than the hourly rate. Therefore, the wireless station device of the present embodiment performs the operations of these digital circuits in a limited time.

  In addition, time-axis beam forming corresponds to control for directing a directivity to a path where the intensity of an incoming wave is maximum. Therefore, in the plane wave approximation of the arriving wave from the direction of the path having the maximum intensity, the length of the perpendicular drawn from the planarly arranged antenna element to the plane having the same phase as the wavefront of the plane wave becomes a path length difference, The path length difference has a geometric periodicity and is given as a function of coordinates. The transmission / reception weight is a coefficient corresponding to the rotation of the complex phase for canceling the path length difference.

  Of course, the rotation amount of the complex phase for each antenna element slightly deviates from the beautiful periodicity because it actually involves a multipath component and also includes a measurement error. However, approximately, an x-axis and a y-axis are set on a plane on which the antenna is arranged, and three-dimensional notation is performed with respect to a coordinate point of the antenna element by using the z-axis as a rotation amount of a complex phase. When plotting the amount of rotation of the complex phase with respect to the coordinates, all plot points are present on one plane in the plane wave approximation. In addition, since the complex number Exp (−jψ ± 2π × integer ψ) is completely equivalent even when ± 2π × integer is added to the rotation amount 複 素 of the complex phase, each antenna is large when the entire antenna aperture is large. Depending on the coordinates of the elements, there may be cases where it is considered that they exist on a plane as a value obtained by adding ± 2π × integer. However, considering such a complex phase offset, the rotation amount of the complex phase can be obtained with a small number of antenna elements. The remaining antenna elements can approximately acquire the rotation amount of the complex phase by linear interpolation based on the path length difference.

  Even when the uplink and downlink frequencies are different, such as FDD, or when there is an asymmetry in the uplink / downlink channel that separates the transmission antenna and the reception antenna, the path length difference can be considered. It is possible to approximately acquire the rotation amount of the complex phase. For example, if the path length difference is ΔL and the wavelength is λ, the amount of rotation of the complex phase for canceling the path length difference is given by (2πΔL / λ).

For example, the frequency of the uplink and ₁ wavelength lambda in F _1, the frequency of the downlink wavelength F ₂ is the lambda _2. Assuming that the rotation amount Δθ _UL of the complex phase at the time of receiving the _uplink is (2πΔL / λ ₁ ), the rotation amount Δθ _DL of the complex phase at the time of receiving the downlink is (2πΔL / λ ₂ ). Equation (6) holds.

Therefore, even in the case of FDD, if the amount of rotation of the complex phase is obtained based on the channel estimation of the frequency F _{1 on} the uplink, and the value is multiplied by (λ ₁ / λ ₂ ), the frequency F ₂ on the downlink is approximately obtained. Of the complex phase is obtained. Rotation of the complex phase of the uplink, there is no need to seek in all of the antenna elements as described above, After determining the amount of rotation of the complex phase of the frequencies F ₁ part of the antenna elements of the uplink, the remaining it is also possible to determine the amount of rotation of the complex phase of the frequencies F ₁ of the antenna element, which likewise makes it possible also to calculate the amount of rotation of the complex phase of the frequency F ₂ in the downlink if Motomare.

  At this time, if the amount of rotation of the complex phase of at least three antenna elements is obtained, the amount of rotation of the complex phase of the remaining antenna elements is obtained from the amount of rotation of the complex phase of each antenna element coordinate on the plane including the three points. Can be calculated. If the amount of rotation of the complex phase can be calculated by four or more antenna elements, the amount of rotation of the complex phase in a three-dimensional space given by the coordinates of each antenna element and the amount of rotation of the complex phase, and the amount of rotation of the calculated complex phase With the least-squares approach to the error, the plane with the smallest sum of the square errors is calculated, the amount of rotation of the complex phase with respect to each antenna coordinate on the plane is calculated, and this is set in the phase shifter. You may respond.

It should be noted here that the rotation amount of the complex phase obtained by the correlation operation has an uncertainty of the complex phase of 2π period, but the complex amount to be multiplied by the coefficient (λ ₁ / λ ₂ ). The amount of phase rotation must exclude the uncertainty of the complex phase of the 2π period. For example, for the first to sixth antenna elements arranged linearly, the rotation amount of the complex phase with respect to the first antenna element is “0”, “π / 2”, “ π ”,“ 3π / 2 ”,“ 0 ”, and“ π / 2 ”. Here, up to “3π / 2” of the fourth antenna element monotonically increases and has continuity, but the rotation amount “0” of the complex phase of the fifth antenna element should actually be regarded as “2π”. Similarly, the rotation amount “π / 2” of the complex phase of the sixth antenna element should be regarded as “5π / 2”. This can be easily achieved by introducing the assumption that the difference in the amount of rotation of the complex phase between adjacent antenna elements does not change by more than ± π due to the design conditions of the antenna element spacing and the direction of arrival, for example. Can be removed, and the rotation amount of the complex phase may be multiplied by a coefficient (λ ₁ / λ ₂ ). As an example, if the coefficient (λ _{1 /} λ ₂₎ is 0.9, the amount of rotation of the complex phase of the first to sixth antenna element determined from the uplink "0", "[pi / 2", "[pi , “3π / 2”, “0”, and “π / 2”, the rotation amount of the downlink complex phase before performing the calibration is “0”, “0.9 × π / 2”, It is given by “0.9 × π”, “0.9 × 3π / 2”, “0.9 × 2π”, and “0.9 × 5π / 2”.
Even if the uplink and downlink antenna elements have different configurations, if the uplink antenna element and the downlink antenna element are mixed and the overall antenna configuration is adopted, the uplink After determining the amount of rotation of the receiving antenna in the complex phase of the frequencies F ₁ of the antenna elements also calculates the rotation amount of the complex phase of the frequency F ₂ assuming that used in other antenna elements uplink for the downlink it is possible, it is possible that the calibration process is performed based on the phase rotation amount of the antenna elements, it calculates a rotation amount of the complex phase of the frequency F ₂ of the antenna element for the downlink.

Alternatively, even when the uplink antenna element group and the downlink antenna element group are physically separated, the uplink antenna element group and the downlink antenna element group are simply arranged in parallel. In the case of a moving relationship (that is, when the relative positional relationship between the antenna elements does not change between the transmitting and receiving antennas on the same plane), the amount of rotation of the downlink complex phase obtained by the receiving antenna is transmitted as it is. Even if it is used as the complex phase of the corresponding antenna element on the side, if the distance between the transmitting and receiving stations is sufficiently large and it is expected that a common plane wave approximation is possible, the directivity that is sufficiently approximately effective It is expected that formation is possible.
Hereinafter, a detailed embodiment to which the basic principle is applied will be described with reference to the drawings.

[First Embodiment]
FIG. 1 is a functional block diagram illustrating a configuration example (sub-array separated type) of a wireless station device 450 according to the present embodiment. In the figure, the same parts as those of the radio station apparatus according to the prior art shown in FIGS. 28 to 30 are denoted by the same reference numerals. In the present embodiment, as shown in FIGS. 28 and 29 of the related art, a configuration using a “sub-array separation type” in which a directional beam is divided into a plurality of sub-arrays and a plurality of directional beams are formed in one array There is a variation of the configuration by the "sub-array shared type" (strictly speaking, it is not divided into sub-arrays, so it may be understood as "integrated array"). Give an explanation.

  The wireless station device 450 shown in FIG. 7 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N (N is an integer of 1 or more), and a control circuit 460. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 451-n (n = 1,..., N) includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, D converter 125-n, TDD (Time Division Duplex) switch (TDD-SW) 127-n, and phase shifters 402-n-1 to 402-n-M (M is 2 or more) Integer), switches 403-n-1 to 403-n-M, distribution coupler (HYB) 404-n, correlation calculation circuit 405-n, phase shift control circuit 406-n, and down converter 424-n. n-1 to 424-nM and A / D converters 425-n-1 to 425-nM. The phase shifters 402-n-1 to 402-nM are connected to the antenna elements 401-n-1 to 401-nM, respectively. The D / A converters 122-1 to 122-N, the up converters 123-1 to 123-N, the down converters 124-1 to 124-N, and the A / D converters 125-1 to 125-N are shown in FIG. 28 and D / A converters 922-1-1 to 922-NM shown in FIG. 29, up-converters 923-1-1 to 923-NM, and down-converters 924-1-1 to 924-NM. , A / D converters 925-1-1 to 925-NM can be used.

  Here, in the down converters 424-1-1 to 424-NM, a signal from a local oscillator is required to perform frequency conversion between a radio frequency signal and a baseband signal. That is, a common local signal is used for the combination of the down converters 424-n-1 to 424-n-M in the same transmission / reception signal processing circuit 451-n (n = 1,..., N), and each down converter 424-n-M is used. It is necessary to suppress a temporal change in the relative relationship between the complex phases in n-1 to 424-n-M. In this sense, a configuration in which a local oscillator exists outside the down converters 424-1-1 to 424-NM is practically used. The description of the oscillator is omitted. The local oscillator used must be shared among the down converters 424-n-1 to 424-n-M of the same n, but the down converters 424-n'-1 to 424-n having different values of n. The local oscillator used in combination with n'-M does not need to be shared. Further, these local oscillators do not need to be shared between the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. To the last, it is only important to share the local signal between the signal sequences that cooperate in forming the directivity.

Here, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the wireless station device 450 includes a total of N transmission / reception signal processing circuits 451-1 to 451-N for N systems. M represents the number of antenna elements of the sub-array mounted on each of the transmission / reception signal processing circuits 451-1 to 451-N. In the description of FIG. 27 it has a total number of antenna elements and M _0, but because in this case as in the case of FIG. 28 is a subarray constitute the entire antenna number of antenna elements of each subarray were labeled as different values M. Each of the transmission / reception signal processing circuits 451-n is accompanied by sub-array antenna elements 401-n-1 to 401-n-M, and the transmission / reception signal processing circuits 451-1 to 451-N are described in Non-Patent Document 2. It is assumed that they are installed spatially separated as described above. Further, the baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 451-1 to 451-N by wire, and the digital baseband signal is transferred over this wire. Although not shown in FIG. 27, the wireless station device 450 includes an overall control circuit 460. FIG. 3 shows an example in which the wireless station device 450 mounts the control circuit 460 on the baseband signal processing circuit 140. The control circuit 460 manages a frame cycle and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N. The control circuit 460 determines whether to connect the distribution coupler 404-n to the up converter 123-n or the down converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by splitting.

  Further, in the present embodiment, the process of estimating the amount of complex phase rotation basically performed by the variable phase shifter corresponding to the transmission / reception weight is performed by digital signal processing, and the actual complex phase rotation process is realized by analog signal processing. . For this reason, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are premised on signal processing on the frequency axis like the OFDM (Orthogonal Frequency Division Multiplexing) modulation method. In both cases, even if signal processing on the time axis is premised, such as SC-FDE (Single-Carrier Frequency Domain Equalization), both methods can be supported. However, the signal separation circuit 141 performs signal separation between each signal sequence. Here, it is also possible to perform signal separation on the time axis. Frequency-dependent signal separation may be performed. In this sense, it is assumed that the input to the demodulators 130-1 to 130-N is a signal on the time axis and a signal on the frequency axis. The description will be made assuming that the user inputs “. As described above, the concept regarding the variation of the communication scheme such as the OFDM modulation scheme and the SC-FDE is the same in the following description.

The specific signal flow is as follows.
First, signal transmission will be described. In the wireless station device 450, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmit signals while the up-converter 123-n and the distribution coupler 404-n are connected (n = 1,..., N).

  Each of the modulators 120-1 to 120-N generates a transmission signal of a time base digital baseband of each stream to be spatially multiplexed, and inputs the transmission signal to the transmission / reception signal processing circuits 451-1 to 451-N. The transmission signal of the time base digital baseband generated by the modulator 120-n is input to the transmission / reception signal processing circuit 451-n (n = 1,..., N). The D / A converter 122-n of the transmission / reception signal processing circuit 451-n (n = 1,..., N) converts the transmission signal input from the modulator 120-n into an analog baseband signal, and Input to converter 123-n. The up-converter 123-n converts the signal input from the D / A converter 122-n from a baseband signal to a signal in a radio frequency band, and inputs the signal to the TDD switch 127-n. The TDD switch 127-n inputs the signal input from the up-converter 123-n to the distribution coupler 404-n.

  The distribution coupler 404-n branches the analog signal input from the TDD switch 127-n into M-system analog signals. The branched analog signals are input to phase shifters 402-n-1 to 402-n-M via switches 403-n-1 to 403-n-M. The phase shifters 402-n-1 to 402-n-M apply a predetermined complex phase rotation on the analog signal. For example, the phase shifters 402-n-1 to 402-n-M apply a predetermined amount of complex phase rotation to a signal in a radio frequency band in an analog signal format. The analog signals subjected to the complex phase rotation by the phase shifters 402-n-1 to 402-n-M are transmitted via the antenna elements 401-n-1 to 401-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 402-n-1 to 402-n-M. For example, individual directivities are formed in the antenna elements 401-1-1 to 401-1-M and the antenna elements 401-N-1 to 401-NM, and a radio station device in the directivity direction is formed. Perform communication.

  Next, reception of a signal will be described. In the wireless station device 450, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receive signals while the distribution coupler 404-n and the down-converter 124-n are connected (n = 1,..., N).

  The signals received by the antenna elements 401-n-1 to 401-n-M (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on an analog signal to the input signal, and outputs the result through switches 403-n-1 to 403-n-M. And input to the distribution coupler 404-n. The distribution coupler 404-n combines signals of the respective antenna systems input via the switches 403-n-1 to 403-n-M on an analog signal, and combines the combined signals via the TDD switch 127-n. Input to the down-converter 124-n. The down-converter 124-n converts a radio frequency signal input via the TDD switch 127-n into an analog baseband signal, and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the signal input from the downconverter 124-n from an analog baseband signal to a digital baseband signal, and inputs the digital baseband signal to the signal separation circuit 141.

  The signal separation circuit 141 performs signal separation by suppressing the crosstalk component between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted to a frequency axis signal by FFT processing and then performed on the frequency axis. When suppressing the crosstalk component on the time axis, first, the correlation between the signal sequences input to the signal separation circuit 141 is calculated based on the digital baseband signal corresponding to the received training signal. . Then, based on a MIMO (Multiple Input Multiple Output) channel matrix given by the calculated correlation, general MIMO signal separation processing such as ZF (Zero Forcing) type or MMSE (Maximum Mean Square Error) type signal separation. A matrix similar to the above may be calculated, and this matrix may be multiplied in units of sampling data by regarding the signal sequence input to the signal separation circuit 141 as a vector. This is because, in general MIMO signal separation processing such as ZF-type or MMSE-type signal separation on the frequency axis, different matrices are used for each frequency component, but substantially the same matrix on the frequency axis. In the case where is used, it corresponds to the fact that signal separation processing can be performed on the time axis in units of sampling data. Alternatively, only the signal processing performed by the transmission / reception signal processing circuits 451-1 to 451-N may be performed, and the signal separation circuit 141 may not perform any processing. In any case, however, the details of the signal separation method here are omitted because they do not directly relate to the features of the present embodiment. The demodulators 130-1 to 130-N demodulate the signal in which the crosstalk component has been suppressed in the signal separation circuit 141.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402-NM will be described. In this signal processing, the switches 403-n-1 to 403-n-M (n = 1,..., N) are controlled by the phase shifters 402-n-1 to 402-n-M and the down converters 424-n-1 to 424-n-1. 424-nM. The switching of these switches is managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. Except when calculating the rotation amount of the complex phase, the phase shifters 402-n-1 to 402-n-M (n = 1,..., N) are connected to the distribution coupler 404-n. In addition, when performing the calculation processing of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402-NM are set to predetermined values. The rotation amount of the complex phase obtained in the subsequent processing is set as a difference from an initial predetermined value. For example, in the simplest example, all of the phase shifters 402-1-1 to 402-NM may be set to zero (or the same value), and in this case, the rotation amount of the obtained complex phase May be used as it is as the phase rotation amount of the phase shifters 402-1-1 to 402-NM during the subsequent communication. Alternatively, the initial predetermined values of the phase shifters 402-1-1 to 402-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the amount of rotation of the complex phase is + α degrees, . +-. Degree., + .Gamma.,..., The phase rotation amounts of the phase shifters 402-1-1 to 402-NM at the time of the subsequent communication are + (. Alpha. + 10) degrees and + (. Beta. + 20) degrees. , + (Γ + 30) degrees,...

  As the actual processing, first, the wireless station device of the communication partner from which the rotation amount of the complex phase is to be acquired transmits a training signal for channel estimation, and the wireless station device 450 receives this training signal. The signals received by the antenna elements 401-n-1 to 401-n-M (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on an analog signal to the input signal, and outputs the result through switches 403-n-1 to 403-n-M. To down converters 424-n-1 to 424-n-M. The down-converters 424-n-1 to 424-n-M respectively convert the input radio frequency signals into analog baseband signals, and convert the signals into A / D converters 425-n-1 to 425-n-M. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal, and outputs the signal to the correlation calculation circuit 405-n.

  The correlation calculation circuits 405-1 to 405-N calculate the amount of rotation of the complex phase using Expressions (1) to (3), respectively. When the calibration processing is necessary as needed, the correlation calculation circuits 405-1 to 405-N determine the complex phase on the transmission side as a value in consideration of the calibration coefficient in the equations (1) to (3). Determine the amount of rotation. The rotation amount of this complex phase obtained by the correlation calculation circuit 405-n (n = 1,..., N) is the identification number of the wireless station device with which to communicate (in the case of performing communication with a plurality of wireless station devices. An identification number is not required when performing communication with one wireless station device in a PP (point-to-point) type.) And input to a phase shift control circuit 406-n (n = 1,..., N). Is done. The phase shift control circuit 406-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

  Note that when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 460 grasps the wireless station apparatus to be the communication partner, and the phase shift control circuit 406-n (n = 1,...). , N) to instruct the phase shifters 402-n-1 to 402-n-M to set the amount of rotation of the complex phase corresponding to the wireless station device that performs communication. The phase shift control circuit 406-n obtains the amount of rotation of the complex phase corresponding to the wireless station device with which the communication is performed, for example, by reading out from the memory, and acquires the amount of rotation of the complex phase from the phase shifters 402-n-1 to 402. -N-M is set to realize beamforming on analog.

Although not explicitly shown in the figure, if a transmitting-side high-power amplifier (HPA) is disposed, for example, the transmitting-side high-power amplifier (HPA) is disposed at a point described as “A” in the drawing, and the receiving-side low-noise amplifier (HPA) is disposed. if placing LNA) or the like, arranged at a point which is described as "B" and "C _1-1" - "C _N-M" in FIG. Regarding the points described as “A” and “B”, since the same transmission / reception signal processing circuits 451-1 to 451-N are common, the complex phases of the individual high power amplifiers and low noise amplifiers are uncertain. Calibration processing for removing the characteristic is unnecessary.

On the other hand, regarding the low-noise amplifiers at the points described as “C _1-1 ” to “C _N-M ”, when the rotation amount of the complex phase can vary with time, the same transmission / reception signal processing circuit 451 is used. 1 to 451-N, which can cause the uncertainty of the complex phase among the respective antenna elements 401-1-1 to 401-N-M, similarly to the prior art implicit feedback calibration method. It is necessary to remove the uncertainty of the complex phase of the low-noise amplifier of each system. Note that the present embodiment can be applied to any method, and the specific method of the calibration process is not limited. Considering this calibration result, for example, assume that the rotation amount of the complex phase at each of “C _1-1 ”, “C _1-2 ”, and “C _1-3 ” is +10 degrees, +20 degrees, and +30 degrees. -10, -20, and -30 degrees are respectively corrected for the complex phase rotation amounts obtained by Equations (1) to (3) to adjust the phase rotation amounts. The information on the calibration result is collected by a calibration circuit (not shown), and the information is collected by the phase shift control circuits 406-1 to 406-N or the correlation calculation circuits 405-1 to 405-N. And perform correction.

  In addition, since the transmission / reception signal processing circuits 451-1 to 451-N having a sub-array configuration can be installed physically apart from each other to reduce the correlation between directional beams formed on analog, generally, Are set apart by a predetermined distance or more.

  Further, the same applies to all the following descriptions (including other embodiments), but in FIG. 1, the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N are connected by wire. In this configuration, digital baseband signals are transmitted and received over the wire, but the D / A converters 122-1 to 122-N and the A / D converters 125-1 to 125-N are connected to the baseband signal processing circuit 140. Is implemented, a signal flowing on a wired connection between the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N may be an analog baseband signal.

  Next, similarly to FIG. 29 with respect to FIG. 28 in the related art, the wireless station apparatus of the present embodiment can be configured by a “shared sub-array type” in which a plurality of directional beam formations are realized by one array antenna. is there. This configuration is shown in FIG.

  FIG. 2 is a functional block diagram illustrating a configuration example (shared sub-array) of the wireless station device 452 according to the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The wireless station device 452 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N, distribution couplers (HYB) 407-1 to 407-M, and an antenna element 401-. 1 to 401 -M, and a control circuit 460. The distribution coupler (HYB) 407-m (m = 1,..., M) includes phase shifters 402-1-m, 402- of the transmission / reception signal processing circuits 451-1, 451-2,. , 402-N-m, and the antenna element 401-m (m = 1,..., M).

  Also in the wireless station device 452, similarly to the wireless station device 450 shown in FIG. 1, the down converters 424-1-1 to 424-NM perform frequency conversion between a radio frequency signal and a baseband signal. Therefore, it is necessary to input a signal from a local oscillator. A common local signal is used for each combination of the down converters 424-n-1 to 424-n-M (n = 1,..., N) in each of the transmission / reception signal processing circuits 451-1 to 451-N. It is necessary to suppress a temporal change in the relative relationship between the complex phases in the down converter. In this sense, a configuration in which a local oscillator exists substantially outside the transmission / reception signal processing circuits 451-1 to 451-N is used. However, since the description is complicated, a simple description of the external local oscillator is used here. The description is omitted. It is not necessary to share the local oscillator used in each of the transmission / reception signal processing circuits 451-1 to 451-N. Further, it is not necessary to share these local oscillators with the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. To the last, it is only important to share the local signal between the signal sequences that cooperate in forming the directivity. However, unlike the case of FIG. 1, all the down converters 424-1-1 to 424-NM are housed in the same housing, so that all the local signals can be shared in the case of FIG. Is possible.

  As described above, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and M represents the number of antenna elements of the array antenna that is shared. In the wireless station device 450 shown in FIG. 1, the transmission / reception signal processing circuits 451-1 to 451-N are mounted in physically different housings for each system. On the other hand, in the wireless station device 452 shown in FIG. 2, all N systems of the transmission / reception signal processing circuits 451-1 to 451-N are mounted in one housing. Further, in the wireless station device 452, the transmission / reception signal processing circuits 451-1 to 451-N share the antenna elements 401-1 to 401-M via the distribution couplers 407-1 to 407-M. Here, in the wireless station device 450 shown in FIG. 1 and the wireless station device 452 shown in FIG. 2, the internal processing in the transmission / reception signal processing circuits 451-1 to 451-N is the same. FIG. 2 shows a case where the control circuit 460 is mounted on the baseband signal processing circuit 140, but the control circuit 460 can be mounted at an arbitrary place in the wireless station device 452. The control circuit 460 manages the frame period, transmission / reception timing, and switching of the TDD switches 127-1 to 127 -N.

The following shows a specific signal flow in the wireless station device 452, focusing on the difference from the wireless station device 450 shown in FIG.
First, signal transmission will be described. In the wireless station device 452, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmit signals while the up-converter 123-n and the distribution coupler 404-n are connected (n = 1,..., N). This point is common to the wireless station device 450 shown in FIG.

  Each of the modulators 120-1 to 120-N generates a transmission signal of a time base digital baseband of each stream to be spatially multiplexed, and inputs each of the signals to the transmission / reception signal processing circuits 451-1 to 451-N. By the same processing as that of the radio station apparatus 450 shown in FIG. 1, analog signals of radio frequencies subjected to processing for forming directivity are output from the phase shifters 402-1-1 to 402-NM. The phase shifters 402-1-1 to 402-NM transmit these signals of each system to the distribution couplers 407-1 to 407-M connected to the corresponding antenna elements 401-1 to 401-M. input. That is, the phase shifters 402-1-m, 402-2-m,..., 402-Nm (m = 1,..., M) input signals to the distribution coupler 407-m. Distribution couplers 407-1 to 407-M respectively combine the input signals, and the combined signals are transmitted via antenna elements 401-1 to 401-M.

  Next, reception of a signal will be described. In the wireless station device 452, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receive signals while the distribution coupler 404-n and the down-converter 124-n are connected (n = 1,..., N). This point is common to the wireless station device 450 shown in FIG.

  Signals received by antenna elements 401-1 to 401-M are input to distribution couplers 407-1 to 407-M, respectively. The distribution coupler 407-m (m = 1,..., M) distributes the input signal to N systems, and the phase shifters 402-1-m, 402-2-m,. To enter. The transmission / reception signal processing circuits 451-1 to 451-N process the signals input to the phase shifters 402-1-1 to 402-NM in the wireless station device 450 shown in FIG. The same signal processing is performed, and a digital baseband signal is input to the signal separation circuit 141. The signal separation circuit 141 suppresses crosstalk components to perform signal separation, and the demodulators 130-1 to 130-N demodulate the separated signals.

  Signal processing for calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402-NM is implemented by the transmission / reception signal processing circuits 451-1 to 451-N in FIGS. Although there is a difference in the units performed, the signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. 1 and the signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. Here, the description is omitted.

  FIG. 3 is a diagram illustrating a configuration example of a communication system according to the present embodiment. The communication system shown in the figure includes the wireless station device 452 shown in FIG. 2 and the wireless station device 450 shown in FIG. In the wireless station device 450, each of the plurality of transmission / reception signal processing circuits 451-1 to 451-N forms one beam as a sub-array. On the other hand, the wireless station device 452 has a common (sub) array, and one array antenna forms a plurality of beams. In an actual operation, as shown in FIG. 3, the wireless station apparatus 450 faces the wireless station apparatus 452, so that it is possible to spatially multiplex N-system signals.

  Next, another configuration example of the transmission / reception signal processing circuit of the present embodiment will be described with reference to FIGS. 4 to 6 can be replaced with the transmission and reception signal processing circuits 451-1 to 451-N included in the wireless station device 450 shown in FIG. 1 or the wireless station device 452 shown in FIG. . Hereinafter, a case will be described as an example where the transmission and reception signal processing circuits 451-1 to 451-N included in the wireless station device 450 are replaced. Note that when replacing with the transmission / reception signal processing circuits 451-1 to 451-N provided in the wireless station device 452, the antenna elements 401-n-1 to 401-n-M in FIGS. 401-M.

  FIG. 4 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 453-n (n = 1,..., N) in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The transmission / reception signal processing circuit 453-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a distribution coupler 414-n, and a phase shifter 409-n-1 to 409-n-1. 409-n-M, TDD switches 408-n-1 to 408-n-M, phase shifters 402-n-1 to 402-n-M, and switches 403-n-1 to 403-n-M. , A distribution coupler 415-n, a down converter (DC) 124-n, an A / D converter 125-n, down converters (DC) 424-n-1 to 424-n-M, It includes D converters 425-n-1 to 425-n-M, a correlation calculation circuit 405-n, and a phase shift control circuit 406-n. The TDD switches 408-n-1 to 408-n-M are connected to the antenna elements 401-n-1 to 401-n-M, respectively. Here, as in FIG. 1, the description of the external local oscillator is omitted.

  The difference between the transmission / reception signal processing circuit 453-n and the transmission / reception signal processing circuits 451-1 to 451-N shown in FIG. 1 or FIG. 2 is that the TDD switches 408-n-1 to 408-n-M are the antenna elements 401. -N-1 to 401-n-M, and as a result, the path from the up-converter 123-n in the transmission system to the antenna elements 401-n-1 to 401-n-M and the antenna in the reception system The point is that the paths from the elements 401-n-1 to 401-nM to the downconverter 124-n are physically separated.

In the case of the transmission / reception signal processing circuit 451-n shown in FIG. 1 or FIG. 2, for example, a high power amplifier (or power amplifier) of a transmission system is arranged at a place described as “A” in the subsequent stage of the up converter 123-n. The low-noise amplifier of the receiving system is provided at the place described as “B” before the down-converter 124-n (further, “C _n−1 ” before the down-converters 424-n-1 to 424-n-M, "C _n-2", ..., which is arranged in "C _n-M" and described location), thereby enabling shared by transmission and reception circuits from the TDD switch 127-n to the antenna terminal. This point is significantly different from the transmission / reception signal processing circuit 453-n. The advantage of adopting the configuration shown in FIG. 4 is that high power amplifiers and low noise amplifiers can be mounted by the number of antenna elements, resulting in an increase in total transmission power, and the TDD switch shown in FIG. 1 or FIG. The point is that it is possible to suppress the effects of insertion loss and distribution and coupling loss of various circuits from 127-n to antenna elements 401-n-1 to 401-n-M. Therefore, although not shown here, in the transmission and reception signal processing circuit 453-n, "D _1", "D _2", ..., high-power amplifier location as "D _M" is "E It is preferable that a low-noise amplifier be disposed at a location described as “ ₁ ”, “E ₂ ”,..., “E _M ”. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is placed at a place described as “A” and a low-noise amplifier is placed at a place described at “B”, “D ₁ ”, “D ₂ ”,. _It is not necessary that a high-power amplifier and a low-noise amplifier be arranged in each of the places described as “ _M ” and the places described as “E ₁ ”, “E ₂ ”,..., “E _M ”.

The following shows a specific signal flow in the transmission / reception signal processing circuit 453-n focusing on the above difference.
First, signal transmission will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M have the antenna elements 401-n-1 to 401-n-M and the phase shifters 409-n-1 to 409-n-. A signal is transmitted in a state where M and M are connected.

  The transmission / reception signal processing circuit 453-n receives a time-base digital baseband transmission signal of one stream from a modulator 120-n (not shown). The D / A converter 122-n converts the input transmission signal into an analog baseband signal and inputs the signal to the up-converter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a signal in a radio frequency band, and inputs the signal to the distribution coupler 414-n.

  The distribution coupler 414-n branches the radio frequency band analog signal input from the up converter 123-n into an M-system analog signal and inputs the signal to the phase shifters 409-n-1 to 409-n-M. . Each of the phase shifters 409-n-1 to 409-n-M applies a predetermined complex phase rotation on an analog signal to the signal input from the distribution coupler 414-n, and a TDD switch 408-n-1. 408-n-M through the antenna elements 401-n-1 to 401-n-M. The antenna elements 401-n-1 to 401-n-M transmit the input transmission signal. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 409-n-1 to 409-n-M.

  Next, reception of a signal will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M have the antenna elements 401-n-1 to 401-n-M and the phase shifters 402-n-1 to 402-n-. M and the switches 403-n-1 to 403-n-M receive signals while the phase shifters 402-n-1 to 402-n-M and the distribution coupler 415-n are connected. Do.

  The signals received by the antenna elements 401-n-1 to 401-n-M are respectively transmitted through the TDD switches 408-n-1 to 408-n-M and the phase shifters 402-n-1 to 402-n-M. Is input to Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on the analog signal to the input signal, and switches 403-n-1 to 403-n-M. And input to the distribution coupler 415-n. The distribution coupler 415-n combines the signals of the respective antenna systems on an analog signal and inputs the combined signal to the down converter 124-n. The down-converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal. The A / D converter 125-n inputs the converted digital baseband signal to a signal separation circuit 141 in a baseband signal processing circuit 140 (not shown). The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402-NM will be described. The transmission / reception signal processing circuit 453-n performs this signal processing by using the switches 403-n-1 to 403-n-M and the phase shifters 402-n-1 to 402-n-M and the down converter 424-n-1 to 403-n-M. 424-nM. The switching of these switches is managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. Note that the phase shifters 402-n-1 to 402-n-M are connected to the distribution coupler 415-n except when calculating the rotation amount of the complex phase. In addition, when performing the calculation processing of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402-NM are set to predetermined values. The rotation amount of the complex phase obtained in the subsequent processing is set as a difference from an initial predetermined value, as in the above description. The transmission / reception signal processing circuits 453-1 to 453-N simultaneously perform the same processing under the control of the control circuit 460.

  As the actual processing, first, the wireless station device of the communication partner from which the rotation amount of the complex phase is to be acquired transmits a training signal for channel estimation, and the wireless station device including the transmission / reception signal processing circuits 453-1 to 453-N Receives this training signal. The signals received by the antenna elements 401-n-1 to 401-n-M are respectively transmitted through the TDD switches 408-n-1 to 408-n-M and the phase shifters 402-n-1 to 402-n-M. Is input to Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on an analog signal to the input signal, and outputs the result through switches 403-n-1 to 403-n-M. To down converters 424-n-1 to 424-n-M. The down-converters 424-n-1 to 424-n-M respectively convert the input radio frequency signals into analog baseband signals, and convert the signals into A / D converters 425-n-1 to 425-n-M. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal, and outputs the signal to the correlation calculation circuit 405-n.

  The correlation calculation circuit 405-n calculates the amount of rotation of the complex phase as in the description of FIG. The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the phase shift control circuit 406-n together with the identification number of the wireless station device with which communication is performed. The phase shift control circuit 406-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

The above-described complex phase rotation amount relates to the phase rotation amount of the phase shifters 402-n-1 to 402-n-M in the receiving system, but the complex phase rotation amount in a low noise amplifier, a high power amplifier, or the like. In order to cancel the individual difference, the correlation calculation circuit 405-n performs a calibration process in the implicit feedback of the related art, and performs a correction corresponding to the calibration process on the basis of the rotation amount of the complex phase in the reception system in the transmission system. The rotation amount of the complex phase is converted to a value set in the phase shifters 409-n-1 to 409-n-M. However, if a low-noise amplifier is placed in a place described as “E ₁ ”, “E ₂ ”,..., “E _M ”, a sufficient reception level can be obtained, so “C ₁ ”, “C ₂ ” ,..., The low-noise amplifier is unnecessary at the place where “C _M ” is described. Therefore, the rotation of the complex phase in each low-noise amplifier of the receiving system does not need to be distinguished from the rotation of the complex phase in space. As the rotation amount of the complex phase set in the phase shifters 402-n-1 to 402-n-M of the system, the rotation amount of the complex phase calculated by the correlation calculation circuit 405-n can be used as it is. The phase shift control circuit 406-n includes a complex in a transmission system to be set in each of the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M. The amount of phase rotation is managed by, for example, storing it in a memory in association with the identification number of the wireless station device of the communication partner.

  Note that when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 460 grasps the wireless station apparatus to be a communication partner and communicates with the phase shift control circuit 406-n. The phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M are instructed to set the amount of rotation of the complex phase corresponding to the wireless station device to be performed. The phase shift control circuit 406-n obtains the amount of rotation of the complex phase in each of the reception system and the transmission system corresponding to the wireless station device that performs communication by reading from the memory. The phase shift control circuit 406-n sets the rotation amount of the complex phase of the reception system to the phase shifters 402-n-1 to 402-n-M, and sets the rotation amount of the complex phase of the transmission system to the phase shifter 409. −n−1 to 409−n−M to realize beamforming on analog.

  FIG. 5 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 454-n (n = 1,..., N) in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The difference between the transmission / reception signal processing circuit 454-n shown in FIG. 4 and the transmission / reception signal processing circuit 453-n shown in FIG. 4 is as follows. That is, the transmission / reception signal processing circuit 453-n illustrated in FIG. 4 shares the antenna elements 401-n-1 to 401-n-M for transmission and reception. On the other hand, the transmission / reception signal processing circuit 454-n illustrated in FIG. 5 is configured to separate them for transmission and reception, form a pair for transmission / reception, and arrange a set of transmission / reception antennas in a close place. Therefore, when replacing the transmission / reception signal processing circuit 451-n of the wireless station device 450 with the transmission / reception signal processing circuit 454-n, the antenna elements 441-n-1 to 441-n-M are added, and the TDD switch 408-n-M is added. 1 to 408-nM are omitted. When replacing the transmission / reception signal processing circuit 451-n of the wireless station device 452 with the transmission / reception signal processing circuit 454-n, the transmission / reception signal processing circuit 454-n is used instead of the antenna elements 401-n-1 to 401-n-M. The antenna elements 401-1 to 401-M and the antenna elements 441-1 to 441-M shared by the antenna elements 1 to 454-N (and correspond to the antenna elements 401-1 to 401-M and the antenna elements 441-1 to 441-M) Distribution coupler).

  In the description of FIG. 4, for convenience, the description has been made assuming that the portion immediately near the antenna element is the transmission / reception signal processing circuit 453-n. However, the TDD switches 408-n-1 to 408-n-M are provided with the function of the antenna element side. If considered, FIGS. 4 and 5 are completely equivalent diagrams. In the details of the signal processing, in the transmission / reception signal processing circuit 453-n shown in FIG. 4 in the reception system, the antenna elements 401-n-1 to 401-n-M receive signals, whereas in FIG. In the transmission / reception signal processing circuit 454-n shown in (1), reception is performed using the antenna elements 441-n-1 to 441-n-M, and transmission / reception does not pass through the TDD switches 408-n-1 to 408-n-M. Except for this point, all the signal processing in the transmission / reception signal processing circuit 454-n shown in FIG. 5 and the transmission / reception signal processing circuit 453-n shown in FIG. 4 are common.

  However, in consideration of the fact that the transmitting and receiving antennas are physically different, it is also possible to perform correction in consideration of the fact that the coordinates of the antenna elements are physically different, in addition to the simple calibration processing. In the above description, the transmitting and receiving antennas are arranged as a set in a close place. However, as long as the correction is performed in consideration of the fact that the coordinates of the antenna elements are physically different, the transmitting and receiving antennas are not necessarily set. No need to place.

  FIG. 6 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 455-n (n = 1,..., N) in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

In the transmission / reception signal processing circuit 455-n shown in FIG. 6, similarly to the transmission / reception signal processing circuit 454-n shown in FIG. 5, the TDD switches 408-n-1 to 408-n-M are regarded as functions on the antenna element side. 6 is a diagram completely equivalent to FIG. In this sense, also in the transmission / reception signal processing circuit 455-n shown in FIG. 6, similarly to the transmission / reception signal processing circuit 454-n shown in FIG. 5, the transmission antenna elements 401-n-1 to 401-n-M and the reception antenna Antenna elements 441-n-1 to 441-n-M are separated from each other. However, in the transmission / reception signal processing circuit 454-n illustrated in FIG. 5, a pair of individual transmission / reception antenna elements (for example, a pair of the antenna element 401-n-1 and the antenna element 441-n-1) is integrally disposed in the vicinity. In the transmission / reception signal processing circuit 455-n shown in FIG. 6, the transmission antenna elements 401-n-1 to 401-n-M constitute one transmission antenna array, and the reception antenna element 441-n-1 to 441-n-M assume a configuration in which they constitute one receiving antenna array. Therefore, there is no difference between FIGS. 5 and 6 except for the physical arrangement of the antenna elements (or the geometrical difference in the wiring connecting the antenna elements).
In the case where the transmitting and receiving antennas are operated separately as shown in FIG. 6, a wall-shaped obstacle may be arranged to avoid signal leakage from the transmitting antenna to the receiving antenna.

  As described above, according to the present embodiment, the wireless station device uses the A / D converter for each antenna element only when calculating the amount of rotation of the complex phase for performing directivity control. Then, at the time of actual signal transmission, the wireless station apparatus substitutes analog beamforming using a phase shifter instead of digital beamforming. On the other hand, when receiving a signal, the wireless station device uses a switch to output a signal from a receiving antenna to an A / D converter when receiving a training signal, and to a receiving circuit when performing signal demodulation by performing signal synthesis. Switch. This makes it possible to greatly reduce the number of A / D converters and D / A converters that have conventionally been required steadily when performing digital beamforming in Massive MIMO. Therefore, it is possible to avoid a situation in which a large number of A / D converters and D / A converters constantly operate during communication, and to avoid a huge amount of A / D converters and D / A conversion particularly during wideband communication. It is possible to reduce the power consumption of the vessel and the amount of heat generated. Furthermore, since a large heat sink required for heat generation is not required for the A / D converter and the D / A converter, not only low power consumption but also miniaturization of the wireless station device can be achieved. And cost can be reduced.

[Second embodiment]
In the first embodiment, individual down converters and A / D converters are mounted for all (transmitting) receiving antennas, but these circuits are used only for channel estimation. This works effectively to reduce the overall power consumption and the amount of heat generated.However, from the standpoint of reducing the size and cost of the device, it is necessary to mount a circuit that is only used temporarily for a huge number of antenna elements. Is inefficient. In particular, an A / D converter that operates at a very high speed because of a wide band tends to be expensive per unit, which may lead to an increase in the price of the entire device. In the second embodiment, a configuration for integrating these down converters, A / D converters, and the like into one system will be described.

  FIG. 7 is a functional block diagram illustrating a configuration example (sub-array separation type) of the wireless station device 550 according to the present embodiment. Similarly to the first embodiment, in the second embodiment as well, as shown in FIGS. 1 and 2 in the first embodiment, a “sub-array separation type” in which a directional beam is divided and formed into a plurality of sub-arrays. And a "sub-array shared type" that realizes a plurality of directional beams with one array (strictly speaking, since it is not separated into sub-arrays, it may be understood as an "integrated array") Variations exist. FIG. 7 illustrates the “sub-array separation type”. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The wireless station device 550 shown in FIG. 7 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N (N is an integer of 1 or more), and a control circuit 560. The transmission / reception signal processing circuit 551-n (n = 1,..., N) includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, and an A / D converter 125-n. , TDD switch 127-n, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, distribution coupler 504-n, and correlation calculation circuit 505-n and a phase shift control circuit 506-n. The phase shifters 502-n-1 to 502-n-M are connected to the antenna elements 501-n-1 to 501-n-M, respectively.

  Similarly to the wireless station device 450 shown in FIG. 1, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the wireless station device 550 includes the transmission / reception signal processing circuits 551-1 to 551-N as a whole. Only N systems are implemented. Similarly to the wireless station device 450, M represents the number of antenna elements of the sub-array mounted in each of the transmission / reception signal processing circuits 551-1 to 551-N. Each of the transmission / reception signal processing circuits 551-n has sub-array antenna elements 501-n-1 to 501-n-M attached thereto. It is assumed that they will be installed spatially apart from each other. Further, the baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 551-1 to 551-N by wire, and the digital baseband signal is transferred over this wire. Although not shown in FIG. 27, an example is shown in which the wireless station device 550 mounts the entire control circuit 560 on the baseband signal processing circuit 140. The control circuit 560 manages a frame cycle and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N. The control circuit 560 determines whether to connect the distribution coupler 504-n to the up converter 123-n or the down converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by splitting. The control circuit 560 instructs the correlation calculation circuits 505-n (n = 1,..., N) to control the switches 503-n-1 to 503-n-M.

  Further, in the present embodiment, the complex phase rotation amount calculation processing basically performed by the phase shifter corresponding to the transmission / reception weight is performed by digital signal processing, and the actual rotation processing of the complex phase is realized by analog signal processing. For this reason, even when the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N assume signal processing on the frequency axis like the OFDM modulation method, the time axis like the SC-FDE is used. Even if the above signal processing is assumed, it is possible to cope with either method, and the concept regarding the variation of the communication method such as the OFDM modulation method and the SC-FDE is described in the first embodiment described above. Is the same as

  In the up-converters 123-1 to 123-N and the down-converters 124-1 to 124-N, a signal from a local oscillator is input in order to perform frequency conversion between a radio frequency signal and a baseband signal. Although necessary, cooperative signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551-N, and thus it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description. Although omitted in the following description, it is of course possible to perform cooperative signal processing among the transmission / reception signal processing circuits 551-1 to 551-N as an additional function. In this case, a common local oscillator is used. You can go.

A specific signal flow in the wireless station device 550 is as follows.
First, signal transmission will be described. The wireless station device 550 turns on all the switches 503-n-1 to 503-n-M (the connection state with the distribution coupler 504-n) and sets the TDD switch 127-n to the up converters 123-n-1 to 123. Signal transmission is performed in a state where -n-N and distribution coupler 504-n are connected (n = 1,..., N). These conditions are the same for all the transmission / reception signal processing circuits 551-1 to 551-N.

  Each of the modulators 120-1 to 120-N generates a transmission signal of a time base digital baseband of each stream to be spatially multiplexed, and inputs the generated signal to the transmission / reception signal processing circuits 551-1 to 551-N. Processing until the up-converter 123-n of the signal processing circuit 551-n (n = 1,..., N) inputs a signal in the radio frequency band to the subsequent stage is the same as that of the radio station device 450 shown in FIG. . The up-converter 123-n inputs a signal to the distribution coupler 504-n via the TDD switch 127-n.

  The distribution coupler 504-n branches the analog signal input from the TDD switch 127-n into an M-system analog signal, and outputs the phase shifter 502-n via the switches 503-n-1 to 503-n-M. -1 to 502-n-M. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation on the analog signal to the input signal. The analog signals subjected to the complex phase rotation by the phase shifters 502-n-1 to 502-n-M are transmitted via the antenna elements 501-n-1 to 501-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 502-n-1 to 502-n-M, and communicates with the wireless station device having the directivity.

  The above description is the signal processing common to the transmission / reception signal processing circuits 551-1 to 551-N mounted on the N systems, and the similar processing is performed in parallel while maintaining a substantially synchronous relationship such as operating with a common clock. carry out.

  Next, the flow of signals related to reception will be described. The radio station apparatus 550 turns on all the switches 503-n-1 to 503-n-M (the connection state with the distribution coupler 504-n), and sets the TDD switch 127-n to the distribution coupler 504-n and the down converter. 124-n is connected, and a signal is received (n = 1,..., N). These conditions are the same for all the transmission / reception signal processing circuits 551-1 to 551-N.

  Signals received by antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to phase shifters 502-n-1 to 502-n-M, respectively. Each of the phase shifters 502-n-1 to 502-n-M performs a predetermined complex phase rotation on an analog signal with respect to the input signal, and outputs the result through switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. The distribution coupler 504-n combines signals of the respective antenna systems input via the switches 503-n-1 to 503-n-M on an analog signal, and combines the combined signals via the TDD switch 127-n. Input to the down-converter 124-n. Subsequent processing is the same as that of the wireless station device 450 shown in FIG.

  Each of the phase shifters 502-n-1 to 502-n-M performs a predetermined complex phase rotation on an analog signal with respect to the signal received via the antenna element 501-n-1 to 501-n-M. Are added to each other to form a predetermined directivity, and perform communication with the wireless station device of the directivity. The above description is the signal processing common to the transmission / reception signal processing circuits 551-1 to 551-N mounted on the N systems, and the similar processing is performed in parallel while maintaining a substantially synchronous relationship such as operating with a common clock. carry out.

  The signal separation circuit 141 performs the same processing as that of the wireless station device 450 shown in FIG. 1 on the signals input from the transmission / reception signal processing circuits 551-1 to 551-N, and removes the signals in the above-described directivity formation. A process of suppressing crosstalk components between wireless station devices that cannot be completed is performed. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted to a frequency axis signal by FFT processing and then performed on the frequency axis. Alternatively, only the signal processing for forming the directivity performed by the transmission / reception signal processing circuits 551-1 to 551-N may be performed, and the signal separation circuit 141 may not perform any processing (in this case, the signal separation circuit 141 Can be omitted). However, in any case, the details of the signal separation method here are not directly related to the present embodiment, and can be implemented using the conventional MIMO signal processing technology, and thus the description thereof is omitted here. .

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1 to 502-NM will be described. In this signal processing, one of the switches 503-n-1 to 503-n-M (n = 1,..., N) includes one of the phase shifters 502-n-1 to 502-n-M and the down-converter 124-n-M. n, while the remaining switches are disconnected (OFF) from the downconverter 124-n. Among the switches 503-n-1 to 503-n-M (n = 1,..., N), the objects to be connected (turned on) to the down-converter 124-n are sequentially switched. The switching of these switches is managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. During normal operation other than when calculating the amount of rotation of the complex phase described here, the phase shifters 502-n-1 to 502-n-M (n = 1,..., N ) Are all connected to distribution couplers 504-n. In addition, when performing the calculation processing of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 502-1-1 to 502-NM are set to predetermined values. The rotation amount of the complex phase obtained in the subsequent processing is set as a difference from an initial predetermined value. For example, in the simplest example, all of the phase shifters 502-1-1 to 502-NM may be set to zero (or the same value), and in this case, the rotation amount of the obtained complex phase May be used as it is as the phase rotation amount of the phase shifters 502-1-1 to 502-NM during the subsequent communication. Alternatively, the initial predetermined values of the phase shifters 502-1-1 to 502-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the amount of rotation of the complex phase is + α degrees, . +-. Degree., + .Gamma.,..., The phase rotation amounts of the phase shifters 502-1-1 to 502-NM during the subsequent communication are changed by + (. Alpha. + 10) degrees and + (. Beta. + 20) degrees. , + (Γ + 30) degrees,...

  As an actual process, first, the wireless station device of the communication partner to acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and the wireless station device 550 receives this training signal. Signals received by antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to phase shifters 502-n-1 to 502-n-M, respectively. Each of the phase shifters 502-n-1 to 502-n-M performs a predetermined complex phase rotation on an analog signal with respect to the input signal, and outputs the result through switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. Here, since all of the switches 503-n-1 to 503-n-M are OFF except one, the signal synthesized by the distribution coupler 504-n is effectively the switch 503-n. Only the signal received by the antenna of the system to which the switch is connected (ON) is output only among n-1 to 503-n-M. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 504-n, one antenna is selected from the antenna group of the antenna elements 501-n-1 to 501-n-M as a whole. Processing for extracting the element 501-nk (k is any integer from 1 to M) is performed. Note that k is switched at a predetermined cycle so that k takes any value from 1 to M. The reception signal of the antenna element 501-nk selected in this way is input to the down-converter 124-n via the TDD switch 127-n. The down-converter 124-n converts the input radio frequency signal into an analog baseband signal, and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal, and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  Each of the correlation calculation circuits 505-1 to 505-N continuously records the digital baseband signal while switching, until the signals from all the switches are completely received. That is, the correlation calculation circuit 505-n receives the digital baseband signals from all the switches 503-n-1 to 503-n-M while switching the switches 503-n-1 to 503-n-M. And record. The correlation calculation circuit 505-n considers the periodicity of the training signal (for example, a periodicity such that a training signal having the same content is repeated at a cycle of 2048 samples) with respect to the recorded digital baseband signal, and The sampling data of each antenna element 501-nk is extracted so that the sampling timings in the cycle correspond to each other, and the equations (1) to (N) are calculated similarly to the correlation calculation circuits 405-1 to 405-N of the first embodiment. The amount of rotation of the complex phase to be set for each of the phase shifters 502-n-1 to 502-n-M is calculated using Expression (3). This utilizes the fact that the time variation of the channel can be neglected if the wireless station device is fixedly installed at a high place and has a line-of-sight environment. Further, when the calibration processing is necessary as needed, the correlation calculation circuits 505-1 to 505-N use the formula (5) similarly to the correlation calculation circuits 405-1 to 405-N of the first embodiment. The rotation amount of the complex phase on the transmitting side is determined as a value in consideration of the calibration coefficients in 1) to (3).

In the first embodiment, the sampling data of each of the antenna elements 401-nk can be obtained at the same time. However, in the second embodiment, the sampling is performed at different timings. Therefore, it is devised that it can be regarded as equivalently sampled at the same time. At this time, if the frequency error cannot be neglected between the transmitting side and the receiving side, sampling may not be equivalently regarded as sampling at the same time only with the periodicity of the training signal. You can go. For example, assuming that the training signal has a periodicity of N _FFT samples for the sampling data S _k (n) of the training signal while one switch 503 -nk is continuously ON, the N _Test period Assume that enough sampling data has been secured. Assuming that the frequency error is Δf, it is possible to estimate Δf by obtaining Δf ′ that maximizes the Q value of the following equation (7) for various Δf ′.

Here, attention is paid only to the information of the switches 503-nk, but Δf may be obtained for the sampled data of each switch 503-nk and averaged. After Δf is estimated in this manner, the influence of the frequency error can be removed by performing the correction represented by the following equation (8) on the sampling data S _k (n ′).

  Here, n 'means a serial number that is continuous during switching from the switch 503-n-1 to the switch 503-n-1 regardless of the switch switching. Since the phase is rotated by 2πjΔfn ′ during the sampling period × n ′, the rotation is inversely corrected.

  The rotation amount of the complex phase obtained by the correlation calculation circuit 505-n (n = 1,..., N) in this manner is based on the identification number of the wireless station device with which communication is performed (communication with a plurality of wireless station devices). In the case similar to the first embodiment, an identification number is not required when performing communication with a single wireless station device in the PP type.) And input to the phase shift control circuit 506-n. The phase shift control circuit 506-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 502-n-1 to 502-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

  Note that when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 560 grasps the wireless station apparatus to be the communication partner, and the phase shift control circuit 506-n (n = 1,...). , N) to instruct the phase shifters 502-n-1 to 502-n-M to set the amount of rotation of the complex phase corresponding to the wireless station device that performs communication. The phase shift control circuit 506-n obtains the rotation amount of the complex phase corresponding to the wireless station device with which the communication is performed, for example, by reading out from the memory, and acquires the rotation amount of the complex phase. -N-M is set to realize beamforming on analog.

  Although not explicitly shown in the figure, if a transmitting-side high-power amplifier (HPA) is disposed, for example, the transmitting-side high-power amplifier (HPA) is disposed at a point described as “A” in the drawing, and the receiving-side low-noise amplifier (HPA) is disposed. If LNA) are to be arranged, they are arranged at points described as "B" in the figure. Regarding the points described as “A” and “B”, in the same transmission / reception signal processing circuits 551-1 to 551-N, for the same n, the antenna elements 501-n-1 to 501-n-M In the present embodiment which does not basically assume cooperative transmission among the transmission / reception signal processing circuits 551-1 to 551-N, the individual high power amplifier and low noise amplifier The calibration process for removing the uncertainty of the complex phase is unnecessary.

  In addition, the transmission / reception signal processing circuits 551-1 to 551-N having a sub-array configuration can be installed physically apart from each other to reduce the correlation between directional beams formed on analog. Are set apart by a predetermined distance or more.

  Next, similarly to FIG. 2 with respect to FIG. 1 in the first embodiment, the wireless station apparatus of the present embodiment is configured by a “shared sub-array type” that realizes a plurality of directional beam formations by one array antenna. Is also possible. This configuration is shown in FIG.

  FIG. 8 is a functional block diagram illustrating a configuration example (shared sub-array) of the wireless station device 552 in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The wireless station device 552 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N, distribution couplers (HYB) 507-1 to 507-M, and an antenna element 501-M. 1 to 501-M and a control circuit 560. The distribution coupler (HYB) 507-m (m = 1,..., M) includes phase shifters 502-1-m, 502-502 of the transmission / reception signal processing circuits 551-1, 551-2,. 2-m, ..., 502-N-m, and the antenna element 501-m.

  In the wireless station device 552, similarly to the wireless station device 550 shown in FIG. 7, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the transmission / reception signal processing circuits 551-1 to 551-N have And only N systems are implemented. M represents the number of antenna elements of the sub-array mounted on the wireless station device 552. In the wireless station apparatus 550 shown in FIG. 7, the number of antenna elements of each sub-array is M, so the total number of antennas is N × M. However, in this drawing shared as in the case of FIG. And the value of M. In actual operation, it is not necessary to use the same number of antenna elements for the wireless station device 550 shown in FIG. 7 and the wireless station device 552 shown in FIG.

  The baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 551-1 to 551-N by wire, and the digital baseband signal is transferred over the wire. Also, as in the case of FIG. 1, FIG. 8 shows an example in which the entire control circuit 560 is mounted on the baseband signal processing circuit 140.

  Like the wireless station device 550, the wireless station device 552 basically performs digital signal processing on the amount of complex phase rotation performed by the phase shifter corresponding to the transmission / reception weight, and performs actual complex phase rotation processing on analog signal processing. It is realized by. Therefore, similarly to the wireless station device 550, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N of the wireless station device 552 assume a signal processing on the frequency axis, It is possible to cope with any of the above-mentioned methods based on the signal processing.

  In the up-converters 123-1 to 123-N and the down-converters 124-1 to 124-N, a signal from a local oscillator is input in order to perform frequency conversion between a radio frequency signal and a baseband signal. Required. However, since cooperative signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551-N, it is not always necessary to use a common local oscillator. Unlike the wireless station device 550 shown in FIG. 7, in the wireless station device 552, the up converters 123-1 to 123-N and the down converters 124-1 to 124-N are all contained in the same housing. Therefore, if individual local oscillators are prepared for the up-converters 123-1 to 123-N and the down-converters 124-1 to 124-N, the cost increases, and a local oscillator that is shared externally substantially exists. The configuration is common. However, since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  As described above, in the wireless station device 552, the transmission / reception signal processing circuits 551-1 to 551-N for N systems are mounted in one housing, and are connected via the distribution couplers 507-1 to 507-M. The antenna elements 501-1 to 501-M are shared. FIG. 8 shows an example in which control circuit 560 is implemented in baseband signal processing circuit 140, but may be implemented in any location in wireless station apparatus 552. The control circuit 560 manages a frame cycle and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N.

  The difference between the wireless station device 550 shown in FIG. 7 and the wireless station device 552 shown in FIG. 8 is that the transmission / reception signal processing circuits 551-1 to 551-N are aggregated in the housing of one wireless station device 552 and shared. Antenna elements 501-1 to 501 -M of the sub-array thus divided by distribution couplers 507-1 to 507 -M and connected to transmission / reception signal processing circuits 551-1 to 551-N. is there. Therefore, the processing inside the transmission / reception signal processing circuits 551-1 to 551-N is the same for the wireless station device 550 shown in FIG. 7 and the wireless station device 552 shown in FIG.

The following description focuses on the difference from the wireless station device 550 shown in FIG.
First, signal transmission will be described. In the wireless station device 552, processing for forming directivity is performed from each of the phase shifters 502-n-1 to 502-n-M (n = 1,..., N) of the transmission / reception signal processing circuit 551-n. An analog signal of a radio frequency is output. These M-system signals are input to distribution couplers 507-1 to 507-M connected to the corresponding antenna elements 501-1 to 501-M, respectively. That is, the phase shifters 502-1-m, 502-2-m,..., 502-Nm (m = 1,..., M) input signals to the distribution coupler 507-m. Distribution couplers 507-1 to 507-M respectively combine signals input from N transmission / reception signal processing circuits 551-1 to 551-N, and the combined signals are used as antenna elements 501-1 to 501-M. Sent via

  Next, reception of a signal will be described. The signals received by the antenna elements 501-1 to 501-M of the wireless station device 552 are distributed to N systems in the distribution couplers 507-1 to 507-M, respectively, and the transmission and reception signal processing circuits 551-1 to 551-M are respectively provided. N. For example, a signal received by the antenna element 501-m (m = 1,..., M) is distributed to N systems by the distribution coupler 507-m, and the phase shifters 502-1-m, 502-2-m, .., 502-Nm. The wireless station device 552 performs the same signal processing as the signal reception in the wireless station device 550 shown in FIG. 7 on the signals input to the phase shifters 502-1-1 to 502-NM in this manner.

  In the signal processing for calculating the amount of rotation of the complex phase in the phase shifters 502-1-1 to 502-NM, similarly to the reception processing, the signals received by the antenna elements 501-1 to 501-M are respectively , And are distributed to N systems in the distribution couplers 507-1 to 507-M, and are respectively input to the transmission / reception signal processing circuits 551-1 to 551-N. The wireless station apparatus 552 adds the same signals as those calculated by the wireless station apparatus 550 shown in FIG. 7 in calculating the amount of rotation of the complex phase to the signals input to the phase shifters 502-1-1 to 502-NM. Perform processing.

  Other operations are basically performed in the same manner as the wireless station device 550 shown in FIG. The description regarding the HPA on the transmitting side and the LNA on the receiving side is the same as that of the wireless station device 550 shown in FIG.

  The communication system according to the present embodiment has a configuration in which a wireless station device 550 is provided instead of the wireless station device 450 shown in FIG. 3 and a wireless station device 552 is provided instead of the wireless station device 452. Note that the wireless station device 450 and the wireless station device 552 may face each other, or the wireless station device 550 and the wireless station device 452 may face each other.

  Next, another configuration example of the transmission / reception signal processing circuit of the present embodiment will be described with reference to FIGS. 9 to 12 can be replaced with the transmission / reception signal processing circuits 551-1 to 551-N included in the wireless station device 550 shown in FIG. 7 or the wireless station device 552 shown in FIG. . Hereinafter, a case will be described as an example where the transmission and reception signal processing circuits 551-1 to 551-N included in the wireless station device 550 are replaced. Note that, when replacing with the transmission / reception signal processing circuits 551-1 to 551-N included in the wireless station device 552, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 in FIGS. To 501-n-M correspond to the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M, and the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M. Distribution coupler (HYB)).

  FIG. 9 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 553-n (n = 1,..., N) in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The transmission / reception signal processing circuit 553-n shown in the figure includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, an A / D converter 125-n, and a circulator 521-n. n, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, distribution coupler 504-n, correlation calculation circuit 505-n, and phase And a shift control circuit 506-n. The phase shifters 502-n-1 to 502-n-M are connected to the antenna elements 501-n-1 to 501-n-M, respectively.

  In FIG. 7, the antenna elements 501-n-1 to 501-n-M are shown in the figure for easy understanding in comparison with FIG. 7, but the transmission / reception signal processing circuit 553-n corresponds to a frame indicated by a solid line. This frame corresponds to the transmission / reception signal processing circuit 551-n in FIG.

  The difference between the transmission / reception signal processing circuit 551-n shown in FIGS. 7 and 8 and the transmission / reception signal processing circuit 553-n shown in FIG. 7 is the TDD switch 127-n in the transmission / reception signal processing circuit 551-n shown in FIGS. Although the input / output flow of the transmission signal and the reception signal is controlled by n, the transmission / reception signal processing circuit 553-n shown in FIG. 9 controls the input / output flow using the circulator 521-n. That is, the circulator 521-n passes the input signal from the up-converter 123-n to the distribution coupler 504-n, and passes the input signal from the distribution coupler 504-n to the down converter 124-n. The circulator 521-n suppresses other signal flows. Therefore, similar signal processing can be performed without using the TDD switches 127-1 to 127-N as in the transmission / reception signal processing circuit 551-n illustrated in FIGS. Except for this point, all other signal processing is the same as the signal processing of the transmission / reception signal processing circuit 551-n shown in FIGS. The description regarding the HPA on the transmission side and the LNA on the reception side is the same as that of the transmission / reception signal processing circuit 551-n illustrated in FIGS.

  FIG. 10 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 554-n according to the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The transmission / reception signal processing circuit 554-n shown in the figure includes a D / A converter 122-n, an up-converter 123-n, a distribution coupler 514-n, and phase shifters 509-n-1 to 509-n. -M, TDD switches 508-n-1 to 508-n-M, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, and distribution It includes a coupler 515-n, a down converter 124-n, an A / D converter 125-n, a correlation calculation circuit 505-n, and a phase shift control circuit 506-n. The TDD switches 508-n-1 to 508-n-M are connected to the antenna elements 501-n-1 to 501-n-M, respectively.

  Here, similarly to the transmission / reception signal processing circuits 551-1 to 551-N shown in FIG. 7, the up-converter 123-n and the down-converter 124-n perform frequency conversion between a radio frequency signal and a baseband signal. Therefore, it is necessary to input a signal from a local oscillator. However, since cooperative signal processing is not assumed between the transmission / reception signal processing circuits 554-1 to 554-N, it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  Further, the entire control circuit 560 is implemented in the base station signal processing circuit 140 of the wireless station device 550 or the wireless station device 552 or in the wireless station device 552. The control circuit 560 manages transmission / reception timing. Further, since the plurality of transmission / reception signal processing circuits 554-1 to 554-N operate in conjunction with each other, when the control circuit 560 performs control, the plurality of transmission / reception signal processing circuits 554-1 to 554-N Work together.

  The difference between the transmission / reception signal processing circuit 554-n and the transmission / reception signal processing circuits 551-1 to 551-N shown in FIG. 7 or FIG. 8 is that the TDD switches 508-n-1 to 508-n-M are the antenna elements 501. -N-1 to 501-n-M (or antenna elements 501-1 to 501-M), and as a result, the D / A converter 122-n to the antenna element 501-n-1 in the transmission system. The path from the antenna element 501-n-M to the A / D converter 125-n is physically separated from the path from the antenna element 501-n-1 to 501-n-M in the receiving system. .

In the case of the transmission / reception signal processing circuit 551-n shown in FIG. 7 or FIG. 8, for example, a high-power amplifier (or power amplifier) of the transmission system is arranged at a location described as “A” in the subsequent stage of the up-converter 123-n. The low-noise amplifier of the receiving system is arranged at the location described as "B" in the preceding stage of the down-converter 124-n, and the circuit from the TDD switch 127-n to the antenna end can be shared for transmission and reception. This point is significantly different from the transmission / reception signal processing circuit 554-n. The advantage of adopting the configuration as shown in FIG. 10 is that high power amplifiers and low noise amplifiers can be mounted by the number of antenna elements, resulting in an increase in total transmission power, and the TDD switch shown in FIGS. 7 and 8. The point is that it becomes possible to suppress the effects of insertion loss and distribution and coupling loss of various circuits from 127-n to the antenna elements 501-1 to 501-M. For this reason, although not shown here, high-power amplifiers are placed at the locations described as “D ₁ ”, “D ₂ ”,..., “D _M ”, and “E ₁ ”, “E ₂ ”,. It is preferable that a low-noise amplifier is arranged at a location described as “E _M ”. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is placed at a place described as “A” and a low-noise amplifier is placed at a place described as “B”, “D ₁ ”, “D ₂ ” _,. There is no necessity to arrange a high power amplifier and a low noise amplifier at each of the locations of the points described as “E ₁ ”, “E ₂ ”,..., “E _M ”.

  The above changes correspond to the change of the transmission / reception signal processing circuit 451-n in FIGS. 1 and 2 in the first embodiment to the transmission / reception signal processing circuit 453-n shown in FIG. The transmission / reception signal processing circuit 551-n in FIGS. 7 and 8 in the second embodiment is changed to the transmission / reception signal processing circuit 554-n shown in FIG. 453-n.

The following is a description of a specific signal flow in the transmission / reception signal processing circuit 554-n, which is a difference from the transmission / reception signal processing circuit 551-n, focusing on a characteristic part of the present embodiment.
First, signal transmission will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M have the antenna elements 501-n-1 to 501-n-M and the phase shifters 509-n-1 to 509-n-M. A signal is transmitted in a state where M and M are connected.

  A transmission / reception signal processing circuit 554-n receives a time-base digital baseband transmission signal of one stream from a modulator 120-n (not shown). The D / A converter 122-n converts the input transmission signal into an analog baseband signal and outputs the signal to the up-converter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a signal in a radio frequency band, and inputs the signal to the distribution coupler 514-n.

  The distribution coupler 514-n branches the analog signal of the radio frequency band input from the up-converter 123-n into M-system analog signals and inputs the signals to the phase shifters 509-n-1 to 509-n-M. . Each of the phase shifters 509-n-1 to 509-n-M applies a predetermined complex phase rotation on the analog signal to the signal input from the distribution coupler 514-n, and a TDD switch 508-n-1. 508-nM to the antenna elements 501-n-1 to 501-nM. The antenna elements 501-n-1 to 501-n-M transmit the input transmission signal. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 509-n-1 to 509-n-M.

  Next, reception of a signal will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M have the antenna elements 501-n-1 to 501-n-M and the phase shifters 502-n-1 to 502-n-. M and the switches 503-n-1 to 503-n-M connect the distribution coupler 515-n and the phase shifters 502-n-1 to 502-n-M, respectively, to receive signals. I do.

  Signals received by the antenna elements 501-n-1 to 501-n-M are respectively transmitted through the TDD switches 508-n-1 to 508-n-M via the phase shifters 502-n-1 to 502-n-M. Is input to Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation on the analog signal to the input signal, and switches 503-n-1 to 503-n-M. And input to the distribution coupler 515-n. The distribution coupler 515-n combines signals of the respective antenna systems input via the switches 503-n-1 to 503-n-M on an analog signal, and inputs the combined signal to the down converter 124-n. The down-converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal input from the downconverter 124-n into a digital baseband signal, and outputs a signal in a baseband signal processing circuit 140 (not shown). Input to the separation circuit 141. The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1 to 502-NM will be described. In this signal processing, one of the switches 503-n-1 to 503-n-M connects (ON) the distribution coupler 515-n and the phase shifters 502-n-1 to 502-n-M. On the other hand, the remaining switches are performed in a state where the connection with the distribution coupler 515-n is cut off (OFF). The switching of these switches is managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. Note that during normal operation other than when calculating the amount of rotation of the complex phase described here, all of the phase shifters 502-n-1 to 502-n-M are distributed couplers 515-n as described above. The phase shifters 509-n-1 to 509-n-M are all connected to the distribution coupler 514-n. Also, similarly to the transmission / reception signal processing circuit 551-n shown in FIG. 7 or FIG. 8, when calculating the amount of rotation of the complex phase, the phase shifters 502-1-1 to 502-1-M perform the phase rotation. The amount is set to a predetermined value, and the rotation amount of the complex phase obtained in the subsequent processing is set as a difference from the initial predetermined value.

  As the actual processing, first, the wireless station device of the communication partner from which the rotation amount of the complex phase should be acquired transmits a training signal for channel estimation, and the wireless station device having the transmission / reception signal processing circuits 554-1 to 554-N Receives this signal. The signals received by antenna elements 501-n-1 to 501-n-M are input to phase shifters 502-n-1 to 502-n-M, respectively. Each of the phase shifters 502-n-1 to 502-n-M performs a predetermined complex phase rotation on an analog signal with respect to the input signal, and outputs the result through switches 503-n-1 to 503-n-M. And input to the distribution coupler 515-n. Here, all of the switches 503-n-1 to 503-n-M are OFF except one, so that the signal combined in the distribution coupler 515-n is effectively the switch 503-n-M. Only the signal received by the antenna of the system to which the switch is connected (ON) is output from only n-1 to 503-n-M. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 515-n, one antenna element 501-n is selected from the antenna elements 501-n-1 to 501-n-M as a whole. A process for extracting −k (k is any integer from 1 to M) is performed. Note that the switching is performed so that k takes any value from 1 to M. The reception signal of the antenna element 501-nk selected in this way is input to the down converter 124-n. The down-converter 124-n converts the input radio frequency signal into an analog baseband signal, and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal into a digital baseband signal and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  The correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 554-n correspond to the correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 551-n shown in FIG. The same processing is performed. That is, the correlation calculation circuit 505-n calculates the rotation amount of the complex phase on the transmission side to be set in each of the phase shifters 502-n-1 to 502-n-M. The phase shift control circuit 506-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 502-n-1 to 502-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

Further, the correlation calculation circuit 505-n outputs the high-power amplifiers at the locations described as “D ₁ ”, “D ₂ ”,..., “D _M ” and “E ₁ ”, “E ₂ ”,. When the low-noise amplifier is arranged at the location described as “E _M ”, similarly to the correlation calculation circuit 405-n of the transmission / reception signal processing circuit 453-n in FIG. 4, the complex phase of the reception system calculated above is calculated. The rotation amount may be subjected to a calibration process in the implicit feedback of the related art, and may be converted to a rotation amount of the complex phase in the transmission system by a correction corresponding to the calibration process based on the rotation amount of the complex phase in the reception system. . The phase shift control circuit 506-n associates the rotation amount of the complex phase in the transmission system to be set in each of the phase shifters 509-n-1 to 509-n-M with the identification number of the wireless station device of the communication partner. Similarly, it is managed by storing it in a memory.

  Note that, when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 560 grasps the wireless station apparatus to be a communication partner and communicates with the phase shift control circuit 506-n. The phase shifters 502-n-1 to 502-n-M and the phase shifters 509-n-1 to 509-n-M are instructed to set the amount of rotation of the complex phase corresponding to the wireless station device to be performed. The phase shift control circuit 506-n obtains the amount of rotation of the complex phase corresponding to the wireless station device with which the communication is performed, for example, by reading out from the memory, and acquires the amount of rotation of the complex phase from the phase shifters 502-n-1 to 502. -N-M and phase shifters 509-n-1 to 509-n-M are set to implement analog beamforming. These processes are similarly performed in the transmission / reception signal processing circuits 554-1 to 554-N.

  FIG. 11 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 555-n (n = 1,..., N) in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The difference between the transmission / reception signal processing circuit 555-n shown in FIG. 10 and the transmission / reception signal processing circuit 554-n shown in FIG. 10 is as follows. That is, the transmission / reception signal processing circuit 554-n shown in FIG. 10 shares the antenna elements 501-n-1 to 501-n-M for transmission and reception. On the other hand, the transmission / reception signal processing circuit 555-n shown in FIG. 11 has a configuration in which these are separated for transmission and reception, pairs are formed for transmission / reception, and a set of transmission / reception antennas is arranged in a close place. Therefore, when replacing the transmission / reception signal processing circuit 551-n of the wireless station device 550 with the transmission / reception signal processing circuit 555-n, the antenna elements 541-n-1 to 541-n-M are added and the TDD switch 508-n- 1 to 508-nM are omitted. When replacing the transmission / reception signal processing circuit 551-n of the wireless station device 552 with the transmission / reception signal processing circuit 555-n, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 to 541 are used. In place of −nM, the antenna elements 501-1 to 501-M and 541-1 to 541-M (and the antenna elements 501-1 to 501-501) shared by the transmission / reception signal processing circuits 555-1 to 555-N are used. -M, a distribution coupler corresponding to the antenna elements 541-1 to 541-M).

  In the description of FIG. 10, for the sake of convenience, the description has been made assuming that the portion immediately near the antenna element is the transmission / reception signal processing circuit 554-n. However, the TDD switches 508-n-1 to 508-n-M are provided with the functions of the antenna element side. If considered, FIGS. 10 and 11 are completely equivalent diagrams. Regarding the details of the signal processing, in the receiving system, the transmission / reception signal processing circuit 554-n shown in FIG. 10 is different from the antenna element 501-n-1 to 501-n-M in that the signal is received in FIG. In the transmission and reception signal processing circuit 555-n shown in FIG. Except for this point, all the signal processing in the transmission / reception signal processing circuit 554-n shown in FIG. 10 and the transmission / reception signal processing circuit 555-n shown in FIG. 11 are common.

  However, in consideration of the fact that the transmitting and receiving antennas are physically different, it is also possible to perform correction in consideration of the fact that the coordinates of the antenna elements are physically different, in addition to the simple calibration processing.

  In the above description, the transmitting and receiving antennas are arranged as a set in a close place. However, as long as the correction is performed in consideration of the fact that the coordinates of the antenna elements are physically different, the transmitting and receiving antennas are not necessarily set. No need to place. FIG. 12 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 556-n (n = 1,..., N) in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  In the transmission / reception signal processing circuit 556-n shown in FIG. 12, similarly to the transmission / reception signal processing circuit 555-n shown in FIG. 11, the TDD switches 508-n-1 to 508-n-M have the functions on the antenna element side. If considered, FIG. 10 is a diagram completely equivalent to FIG. In this sense, also in the transmission / reception signal processing circuit 556-n shown in FIG. 12, similarly to the transmission / reception signal processing circuit 555-n shown in FIG. 11, the transmitting antenna elements 501-n-1 to 501-n-M and the reception antenna Antenna elements 541-n-1 to 541-n-M are separated from each other. However, in the transmission / reception signal processing circuit 555-n illustrated in FIG. 11, a pair of individual transmission / reception antenna elements (for example, a pair of the antenna element 501-n-1 and the antenna element 541-n-1) is integrally arranged in the vicinity. In the transmission / reception signal processing circuit 556-n shown in FIG. 12, the transmission antenna elements 501-n-1 to 501-n-M constitute a single transmission antenna array, and the reception antenna The elements 541-n-1 to 541-n-M are assumed to constitute a single receiving antenna array. Therefore, there is no difference in signal processing between FIGS. 11 and 12 except for the physical arrangement of antenna elements.

  When digital beamforming is performed in Massive MIMO, a conventional radio station requires expensive A / D converters and D / A converters in a number corresponding to the number of signal sequences, so that the apparatus is expensive. At the same time, power consumption increases. Therefore, the wireless station device of the present embodiment switches the switch so that only the signal sequence targeted only when obtaining weight information for performing directivity control is connected to the A / D converter. Also, at the time of data transmission, the wireless station device converts a signal before being separated for each antenna element into an analog signal by a D / A converter, separates the converted analog signal for each antenna element, and then uses a phase shifter. To perform analog beamforming. As a result, it is possible to reduce the number of A / D converters and D / A converters required to acquire weight information at the time of data transmission and reception, and to reduce power consumption.

[Third Embodiment]
Generally, when acquiring channel information of a plurality of antenna elements, in a multipath environment, in the case of antenna elements separated by a distance of about １ ／ wavelength, the correlation of the channel information is greatly reduced. However, when the environment is generally in the line-of-sight environment or when the number of reflection points is limited, the plane wave approximation for the arriving wave having the highest intensity tends to be applicable. In this case, if a wavefront arriving with the same complex phase can be extracted, the path length difference between each antenna element between the antenna plane and the wavefront of the arriving wave has a geometric regularity, and some antennas The complex phase of the channel information of the remaining antenna element can be estimated from the complex phase of the channel information of the element.

  FIG. 13 shows an outline of channel information prediction in a linear array in which antenna elements are linearly arranged according to the third embodiment of the present invention. FIG. 1 shows a state in which the antenna elements 401-1 to 401-5 are linearly arranged. First, consider a plane wave arriving from the angle θ direction with respect to the front direction of the linear array. The distance between the elements of each antenna is d. Here, the received signal at the antenna element 401-1 is Φ1 (t), the received signal at the antenna element 401-2 is Φ2 (t),..., The received signal at the antenna element 401-5 is Φ5 (t), Let ΔLs be the path length difference of the s-th antenna element (s is an integer of 2 or more) with respect to antenna element 401-1. ΔL1 is set to 0 for convenience.

  In general, when the wavelength is λ and passes through the path length ΔL, the complex phase rotates by 2πΔL / λ. Assuming plane wave approximation, the path length difference between antenna element 401-1 and antenna element 401-2 is ΔL2 = d · Sin θ. Similarly, the path length difference between antenna element 401-1 and antenna element 401-3 is ΔL3 = 2 × d · Sin θ, and the path length difference between antenna element 401-1 and antenna element 401-4 is ΔL4 = 3 × d · Sin θ, and the path length difference between the antenna element 401-1 and the antenna element 401-5 is ΔL5 = 4 × d · Sin θ.

  Therefore, focusing on the frequency component of a certain wavelength λ, the plane is a wave for which approximation is established, and Φ2 (t) ≒ Exp ｛-2πjΔL2 / λ｝ Φ1 (t), Φ3 (t) ≒ Exp ｛-2πjΔL3 / λ The relationship of｝ Φ1 (t), Φ4 (t) ≒ Exp ｛-2πjΔL4 / λ｝ Φ1 (t), Φ5 (t) ≒ Exp ｛-2πjπL5 / λ｝ Φ1 (t) holds. Using the above-described relationship of ΔLs = (s−1) d · Sinθ, Φs (t) ≒ Exp ｛−2πj (s−1) d · Sinθ / λ｝ Φ1 (t). This means that the complex phase is rotated by Exp {−2πjd · Sinθ / λ}. Therefore, for example, channel estimation is performed by the antenna elements 401-1 and 401-5, and based on the rotation amount of the complex phase between them, す る と of the difference of the complex phase rotates for each antenna element. It becomes possible to predict.

  Similar prediction is possible not only when the antenna elements are arranged one-dimensionally but also when the antenna elements are arranged two-dimensionally. Next, a specific example of channel information prediction of an array antenna configured in a planar shape in the present embodiment will be described with reference to FIGS. 14 and 15. In FIG. 14, a to z and A to K in the alphabets shown by “O” represent antenna elements. Here, an example of a shape in which equilateral triangles are spread in a close-packed manner is shown, but the shape may be any other configuration.

  For example, it is assumed that the wireless station device performs channel estimation at three points of antenna elements i, m, and q indicated by double black circles, and determines the complex phase of the channel information at the three points. Here, for example, the antenna element i is used as a reference antenna, the rotation amount of the complex phase between the antenna element i and the antenna element m is ζ, and the rotation amount of the complex phase between the antenna element i and the antenna element q is η. . In this case, focusing on the antenna element on the straight line connecting the antenna element i and the antenna element m, ζ × １ ／ for the antenna element c, ζ × ２ for the antenna element d, and ζ × 4 / for the antenna element B. 3. With the antenna element u, it can be approximated to -ζ × １ ／. Similarly, focusing on the antenna element on the straight line connecting the antenna element i and the antenna element q, η × １ ／ for the antenna element b, η × ２ for the antenna element g, and η × 4 / for the antenna element G 3. With the antenna element v, it can be approximated to -η × １ ／. If this is extended, two-dimensional prediction is also possible. For example, for the antenna element a, ζ × １ ／ + η × １ ／, for the antenna element H, −ζ × １ ／ + η × 4/3, For the antenna element E, it can be predicted as ζ × 2/3 + η × 3/3.

  As a condition for enabling the above prediction, it is necessary that the rotation amount of the complex phase between the antenna elements for acquiring the channel information is not more than π. For example, taking the case of FIG. 13 as an example, even if the complex phase difference between the antenna element 401-1 and the antenna element 401-5 is π / 2 or -3π / 2, Exp ｛ Since jππ / 2｝ = Exp ｛-j ・ 3π / 2｝, it cannot be distinguished. If the complex phase difference is π / 2, the complex phase difference between the antenna element 401-1 and the antenna element 401-2 should be π / 2 × (1/4). Is −3π / 2, the complex phase difference between antenna element 401-1 and antenna element 401-2 should be −3π / 2 × (１ ／).

  Since channel information cannot be predicted correctly in the situation where there is uncertainty as described above, the complex phase difference between antenna elements for acquiring channel information needs to be π or less. In actual measurement, fluctuation of the complex phase is also expected due to a measurement error due to noise and the influence of multipath. Therefore, the complex phase difference actually needs to be a small value with a margin larger than π. The standard value cannot be unconditionally determined because the influence of the reflected wave differs depending on the magnitude of the reflected wave. However, when the antenna element distance d is reduced or the arrival angle θ is sufficiently small, the path length difference ΔL decreases. Therefore, the amount of change in the complex phase due to the path length difference can be made sufficiently small. In the example of FIG. 14, the three points of the antenna elements i, m, and q are relatively separated from each other. However, in order to reduce the complex phase difference between the antenna elements to π or less, the mutual element spacing is small. It is also possible to use adjacent antenna elements (for example, antenna elements a, e, f, etc.). Although FIG. 14 shows an example in which the channels are estimated by three antenna elements and the complex phases of the remaining antenna elements are approximated in a two-dimensional plane, a case where channel information is acquired by more antenna elements , It is possible to make a more detailed prediction of channel information.

  FIG. 15 shows another example of channel information prediction of an array antenna configured in a plane according to the third embodiment of the present invention. The difference from FIG. 14 is that the number of antenna elements for acquiring channel information is changed from three to seven antenna elements a, w, z, C, F, I and t indicated by double black circles. That is. In this case, the hexagonal array antenna has six triangles {a, w, z}, {a, z, C}, {a, C, F}, {a, F, I}, {a, I, t}, {a, t, w}. The points in each triangle may be predicted based on the complex phase of the vertices of each triangle. For example, focusing on the triangle {a, w, z}, the antenna element a is used as a reference antenna, and the rotation amount of the complex phase between the antenna element a and the antenna element w is ζ, and the antenna element a and the antenna element The rotation amount of the complex phase during z is η.

  In this case, for example, when attention is paid to an antenna element on a straight line connecting the antenna element a and the antenna element w, it can be approximated to ζ × １ ／ for the antenna element c and ζ × ２ for the antenna element j. Similarly, if attention is paid to an antenna element on a straight line connecting the antenna element a and the antenna element z, it can be approximated as η × １ ／ for the antenna element d and η × ２ for the antenna element l. For other antenna element k, ζ × １ ／ + η × １ ／, for antenna element y, ζ × １ ／ + η × ２, for antenna element x, ζ × ／ + η × １ ／. As predictable. Other triangles can be similarly predicted. The difference from FIG. 14 is that in the case of an array antenna having a spatial spread, the influence of multipath on antenna elements at distant locations has a weaker correlation, so that triangles {a, w, z}, {a, z, C近似, {a, C, F}, {a, F, I}, {a, I, t}, {a, t, w}, by performing individual approximation to arrive at the entire array antenna. Has a feature that it is possible to improve the approximation accuracy even when the arrival directions are slightly different. Note that, of course, there is no essential restriction on the arrangement of the antenna elements. Therefore, there is no necessity of a densely packed structure as shown in FIGS. 14 and 15, and for example, a square lattice array can be used. .

  Next, a specific example of channel information prediction of a square array antenna formed in a planar shape in the present embodiment will be described with reference to FIGS. In FIG. 16, a to z and A to J of alphabets indicated by “O” or “●” represent antenna elements. Here, an example of a shape in which equilateral triangles are spread in a close-packed manner is shown, but any other configuration may be used. For example, it is assumed that the wireless station device performs channel estimation at three points of the antenna elements a, l, and p indicated by double black circles, and determines the complex phase of the channel information at the three points.

  For example, let the antenna element a be a reference antenna, the rotation amount of the complex phase between the antenna element a and the antenna element 1 be ζ, and the rotation amount of the complex phase between the antenna element a and the antenna element p be η. In this case, the linear prediction is slightly more complicated than in FIGS. 14 and 15, but the basic concept is the same. For example, the square lattice is considered as a lattice point on the xy plane, and the antenna element a is regarded as the origin. At this time, for example, if the coordinates are defined as (1, 0) for the antenna s and (0, 1) for the antenna v (here, for convenience, the downward direction of the y axis is the positive direction). In this case, the antenna 1 corresponds to (5, 3) and the antenna p corresponds to (1, 5).

When the amount of phase rotation is represented by the z-axis and expressed in three dimensions, (0,0,0) for the antenna a, (5,3, は) for the antenna 1, and (1,5, η) for the antenna p. Become. Equation expressing a two-dimensional plane, a, b, the use of coefficients c (this factor not related to the identifier of the antenna _{elements), a (x-x 0} ) + b (y-y 0) + c (z- z ₀ ) = 0. For example, since (0, 0, 0) is a point on the plane, if (x ₀ , y ₀ , z ₀ ) = ( ₀ , ₀ , ₀ ), the relational expression of ax + by + cz = 0 is obtained. Here, (a, b, c) is a normal vector of this plane, and since the absolute value of the vector has no meaning, if a ′ = a / c and b ′ = b / c, the unconstant is In other words, a relational expression of a'x + b'y + z = 0 is obtained. On the other hand, since the coordinates (5, 3, ζ) and (1, 5, η) are on a plane, a two-dimensional linear simultaneous equation for a ′ and b ′ can be established and solved. Then, the values of a 'and b' can be easily obtained.

When a 'and b' obtained as described above are used, z =-(a'x + b'y). Therefore, by substituting the coordinates of each antenna element into x and y, the complex phase of each antenna element is obtained. Is required. For example, z =-(3a '+ b') is a desired value for the antenna e because the coordinates are (3,1), and z =-(for the antenna r since the coordinates are (5,5). 5a ′ + 5b ′) is a desired value. By doing so, it is possible to estimate channel information on the remaining antenna elements from channel information on a small number of antenna elements by similar plane wave approximation without depending on the configuration of the antenna arrangement.
It should be noted that the method of utilizing such an equation indicating the relationship between the coordinates of each antenna element and the complex phase of each antenna element will be supplemented slightly. In the above description with reference to FIGS. 14 and 15, the case where the complex phases of the three antenna elements are obtained and the complex phases of the other antenna elements are obtained by linear approximation therefrom has been described. As a method of handling the information of the complex phases of a, w, z, C, F, I, and t, all the antenna elements a to z and A to K may be approximated by one plane wave using the least squares method. It is possible. For example, in a two-dimensional plane where the antenna elements a to z and A to K exist, arbitrary orthogonal x-axis and y-axis are determined, and the coordinates of the k-th antenna element are (x _k , y _k ). It is assumed that the complex phase φ _k of the k antenna elements is given by the following equation (9) using the coefficients α, β, and γ.

On the other hand, the complex phase of the k-th antenna element actually estimated using the first antenna element (for example, antenna element a in the figure) as a reference antenna is represented by ~ φ _k (tilde φ indicates “~” above φ). In the following, a combination of (α, β, γ) that minimizes the following evaluation function W (α, β, γ) can be obtained by the least squares method. (Equation (10)).

Based on the combination of (α, β, γ) obtained by the above least squares method, a required complex phase is calculated from the coordinates (x _k , y _k ) of the k-th antenna element using Expression (9). It becomes possible. When the least squares method is used, an arbitrary number of antenna elements for obtaining a complex phase can be selected. Originally, the present invention approximates an incoming wave with a plane wave, but actually includes components other than the line-of-sight wave (plane wave) under the influence of the reflected wave, so that a clean relationship such as equation (9) is obtained. Not be. The error from this plane affects the estimation accuracy of the complex phase.By increasing the number of antenna elements used for the least squares method, it is possible to average the effect of this reflected wave, The accuracy can be improved. On the other hand, when the number of antenna elements used for the least squares method is increased, the circuit scale is increased. Therefore, the number of antenna elements is set in each trade-off.

  In the above description, the procedure for predicting the complex phase of the channel information has been described. However, after the complex phase of the channel information is obtained, a value obtained by multiplying the complex phase by −1 is the complex phase of the complex phase implemented by the phase shifter. Since the amount of rotation corresponds to the amount of rotation, the amount of rotation of the complex phase may be set as a value on the z-axis, and a calculation process for directly calculating the amount of rotation of the complex phase may be performed.

  In FIG. 16, the antenna elements indicated by “O” corresponding to a to r and the antenna elements indicated by “●” corresponding to s to z and A to J are nested. For example, as described with reference to FIGS. 5 and 11, when the transmitting antenna and the receiving antenna are separated and arranged in pairs, the antenna elements a to r represented by “「 ”are the receiving antennas, and“ ● ” If the antenna elements s to z and A to J represented by are designated as transmitting antennas and arranged in the vicinity, channel information is acquired by some of the receiving antennas by the above-described method, and the information is used as a basis. If information on other antenna elements is predicted, the antenna element to be predicted may be a transmission antenna or a reception antenna. Can be predicted (acquired). Note that the transmitting antenna and the receiving antenna do not need to be arranged differently at the re-adjacent grid points, and may be arranged in another general manner.

  FIG. 17 shows another example of channel information prediction of a square array antenna configured in a plane according to the third embodiment of the present invention. The difference from FIG. 16 is that, in FIG. 16, the transmitting antennas and the receiving antennas are alternately arranged like Othello cells, whereas in FIG. 17, the transmitting antennas or the receiving antennas are arranged in a vertical column. It is a point that is aligned. In this case, for example, one transmission / reception signal processing circuit 454-n, or one of the antenna elements 401-n-1 and 441-n-1 as a pair of the transmission / reception signal processing circuit 555-n, or the antenna element 501-n- 1 and 541-n-1 and the like may be arranged as adjacent antennas such as the antenna elements a and s and the antenna elements d and v in FIG.

  FIG. 18 shows another example of channel information prediction of a square array antenna configured in a plane according to the third embodiment of the present invention. The difference from FIG. 17 is that, in FIG. 17, the transmitting antennas and the receiving antennas are arranged in a vertical column and the transmitting antennas and the receiving antennas are alternately arranged between adjacent columns. In FIG. 18, all the receiving antennas a to r are put together in one place, and similarly, all the transmitting antennas s to z and A to J are put together in one place, and each is arranged in a different area. ing. This corresponds to FIGS. 6 and 12 described above.

  The merit of separating the transmission system and the reception system is that, for example, when operating without turning off the power of the high power amplifier of the transmission system even at the time of signal reception, it is possible to prevent the noise of the transmission system from leaking to the reception system. In addition, since the entire transmission system and the entire reception system are separated from each other, mutual isolation can be easily ensured. On the other hand, as shown in FIG. 16 and FIG. 17, it is not preferable in terms of configuration to estimate the channel information of the antenna element for which the channel information is not actually obtained by signal reception by the above-described method. However, if the antenna arrangement of the transmitting and receiving antennas of the opposing radio station apparatus # 1 and radio station apparatus # 2 has a certain kind of symmetry, the channel information acquired on the receiving side is subjected to an appropriate calibration process. On the premise of performing the above, it is possible to treat it as it is as channel information on the transmission side. As an example, the receiving antenna element a and the transmitting antenna element s, the receiving antenna element b and the transmitting antenna element t, the receiving antenna element c and the transmitting antenna element u, the receiving antenna element d and the transmitting antenna element v. Considering such symmetry, even if the rotation amounts of the complex phases of the receiving antennas a to r are applied to the transmitting antennas s to J as they are, a plane wave approximation is possible. Then it can be assumed that there is no big difference.

  Next, a specific example of channel information prediction of an array antenna in which transmission and reception are separated in a planar shape according to the present embodiment will be described with reference to FIGS. In FIG. 19, the array antenna mounted on the wireless station device # 1 is 561, the array antenna mounted on the wireless station device # 2 is 562, and the transmitting antennas mounted on the wireless station device # 1 are 563-1 to 563-2. The receiving antennas mounted on the wireless station device # 1 are 564-1 to 564-2, the transmitting antennas mounted on the wireless station device # 2 are 565-1 to 565-2, and the wireless station device # 2 is mounted. The receiving antennas are 566-1 to 566-2.

  The feature of the geometric positional relationship between the array antenna 561 mounted on the wireless station device # 1 and the array antenna 562 mounted on the wireless station device # 2 shown in FIG. 19 is that each antenna element of the array antenna 561 exists. If the plane and the plane on which each antenna element of the array antenna 562 is present are substantially parallel and face each other, and as a result, if the line of sight wave is dominant, as a result, the antenna element 563 of the radio station apparatus # 1 will be described. -1 and the channel information when received by the antenna element 566-1 of the radio station apparatus # 2, and the antenna element transmitted from the antenna element 563-2 of the radio station apparatus # 1 and the antenna element of the radio station apparatus # 2 The relative channel information relationship with the channel information received at 566-2 is transmitted from antenna element 565-1 of radio station apparatus # 2 and transmitted to radio station apparatus Channel information when received by the first antenna element 564-1 and a channel when transmitted by the antenna element 565-2 of the wireless station apparatus # 2 and received by the antenna element 564-2 of the wireless station apparatus # 1 It is considered that the relationship between the information and the relative channel information is substantially the same.

  This is based on the relationship between the transmitting antenna mounted on the wireless station device # 1 and the receiving antenna mounted on the wireless station device # 2. 2 is similar to the relationship between the transmitting antenna mounted on the wireless antenna device 2 and the receiving antenna mounted on the wireless station device # 1. Therefore, the geometrical correspondence of each antenna element can be translated to the receiving antenna group by translating the transmitting antenna group, so that the channel information obtained on the receiving side can be generally utilized on the transmitting side. Become.

  In the case where the transmitting and receiving antennas are not shared and the transmitting and receiving antenna groups are physically separated and installed as shown in FIG. 18 (and FIG. 19), the geometry as described in FIG. In the approximation utilizing the relation of the parallel translation, accuracy may be reduced. In this case, it is also possible to cope with the situation by applying some measures.

  FIG. 20 shows another example of channel information prediction of an array antenna in which transmission and reception are separated from each other and arranged in a plane in the third embodiment of the present invention. 20, only the configuration different from the configuration in FIG. 19 will be described. In FIG. 20, one transmitting antenna 567 is included in receiving antennas 564-1 to 564-2 in radio station apparatus # 1. In FIG. 20, three receiving antennas 568-1 to 568-3 are included in transmitting antennas 565-1 to 565-2 in radio station apparatus # 2. The calculation of the reception weight of the wireless station device # 2 is the same as that of FIG. 19, but the calculation of the transmission weight of the wireless station device # 2 is slightly modified.

  Here, antenna element 567 of wireless station apparatus # 1 is an antenna element near the center of gravity in the receiving antenna group, and can transmit a training signal from antenna element 567. The training signal transmitted from this antenna element is received by the receiving antennas of the antenna elements 568-1 to 568-3 of the radio station apparatus # 2, and based on the channel information obtained by these antenna elements, Thus, it is possible to predict channel information of each antenna element of the transmission antenna group of the wireless station apparatus # 2, and calculate the amount of rotation of the complex phase in the transmission directivity control based on the information. These additional functions are unique functions in the overall configuration, but are effective when the antenna scale is enormous or when the distance between the transmitting antenna group and the receiving antenna group is large.

  In FIG. 20, when estimating the amount of rotation of the complex phase for each antenna element of the transmitting antenna group of wireless station apparatus # 2, the rotation of the complex phase obtained by reception antenna elements 568-1 to 568-3 is estimated. Although the amount of rotation of other complex phases is predicted based on the amount, it is naturally possible to use antenna elements of other patterns. FIG. 21 shows an example of an antenna pattern used for predicting a rotation amount of a complex phase according to the third embodiment of the present invention. FIG. 21 shows examples of four patterns (a) to (d) in a 5 × 5 square array. In the drawing, ● and ◎ indicated by symbols a to y represent antenna elements, and based on the amount of rotation of the complex phase obtained using the antenna element indicated by ◎, the complex phases of other antenna elements indicated by ● Predict the amount of rotation. Here, a case where five antenna elements are used to determine the amount of rotation of the complex phase is shown as an example, but it is needless to say that the present invention can be implemented with another number of elements equal to or more than three.

For example, taking FIG. 21A as an example, the rotation amount of the complex phase is obtained using the antenna elements h, l, m, n, and r, and the correlation between the antenna element m near the center of gravity and the other antenna elements is obtained. Then, the complex phase of the correlation value is calculated for the antenna elements h, l, n, and r. The complex phase obtained in this manner is set on the z-axis, and the least squares method similar to that described in Expressions (9) and (10) is used. The rotation amount of the complex phase may be estimated. FIG. 21B is the same.
In addition, in order to expand the spatial extent of the antenna element used for the prediction of the amount of rotation of the complex phase, elements other than the first and second adjacent elements of the reference antenna are used for the prediction of the channel information in FIG. It is also possible to use the patterns (d) and (d). This case is similar to the case of FIGS. 21A and 21B, but in the case of FIG. 21C, for example, the antenna element r is set as the reference antenna, and the antenna element r, the antenna element 1, and the antenna element n are After calculating the complex phase of the correlation value, for the antenna elements f and j, instead of directly performing the correlation calculation with the reference antenna element r, the correlation between the antenna element l and the antenna element f is calculated. The calculation and the correlation calculation between antenna element n and antenna element j may be performed. The sum of the relative value of the complex phase of the antenna element l with respect to the antenna element r and the relative value of the complex phase of the antenna element f with respect to the antenna element l is approximately equal to the relative value of the complex phase of the antenna element f with respect to the antenna element r. It can be considered. Similarly, the sum of the relative value of the complex phase of antenna element n with respect to antenna element r and the relative value of the complex phase of antenna element j with respect to antenna element n is approximately equal to the complex value of antenna element j with respect to antenna element r. It can be considered as a relative value of the phase.

More specifically, the correlation value between the antenna element r and the antenna elements f and j is directly calculated, while the relative value of the complex phase of the antenna element l with respect to the antenna element r is removed in order to remove the uncertainty of the complex phase of the 2π period. The sum of the relative value of the complex phase of antenna element f with respect to antenna element l and the sum of the relative value of the complex phase of antenna element n with antenna element r and the relative value of the complex phase of antenna element j with antenna element n You may use it in another form. In this case, a value obtained by adding an integral multiple of 2π to a value obtained by directly calculating the correlation value between the antenna element r and the antenna elements f and j, the relative value of the complex phase of the antenna element l with respect to the antenna element r, and the antenna element l , And the sum of the relative value of the complex phase of the antenna element n with respect to the antenna element r and the relative value of the complex phase of the antenna element j with respect to the antenna element n is the closest. Alternatively, the correlation value between the antenna element r and the antenna elements f and j may be corrected.
The reason for including the process of calculating and adding the complex phase difference in a plurality of stages as described above is that when the complex phase difference between the desired antenna elements becomes ± π or more, the phase of the complex phase is reduced due to the periodicity of the complex phase. Since the uncertainty of the 2π period cannot be ignored, by adding and using the complex phase of the correlation value between adjacent antenna elements, the complex phase difference between adjacent antenna elements should not be more than π. As a result, it is possible to avoid the uncertainty of the 2π period complex phase.
Such an antenna pattern is used in the estimation of the rotation amount of the complex phase of the reception antenna in the same manner as in FIGS. 14, 15, and 16, as in the estimation of the rotation amount of the complex phase of the transmission antenna as shown in FIG. Can be used.

  Hereinafter, the configuration of the transmission / reception signal processing circuit for realizing the third embodiment of the present invention will be described with reference to FIGS. The transmission / reception signal processing circuits in FIG. 22 are the transmission / reception signal processing circuits 451-1 to 451-N in the first embodiment in FIG. 1 and the transmission / reception signal processing circuits 551-1 to 551-551 in the second embodiment in FIG. This corresponds to another embodiment in which both of the transmission / reception signal processing circuits corresponding to N are combined.

  FIG. 22 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 651-n (n = 1,..., N) in the present embodiment. In this figure, the same functional blocks as those in the above-mentioned figures are given the same numbers, and their explanations are omitted.

  The transmission / reception signal processing circuit 651-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A / D converter 125-n. n, a TDD switch (TDD-SW) 127-n, phase shifters 402-n-1 to 402-n-M, switches 403-n-1 to 403-n-M, and a down converter (DC) 424-n-1 to 424-n-2, A / D converters 425-n-1 to 425-n-2, correlation calculation circuit 405-n, phase shift control circuit 406-n, and complex phase And a rotation amount prediction circuit 410-n. Although FIG. 2 shows two configurations of the down converter 424-n and the A / D converter 425-n, the down converter 424-n and the A / D converter 425-n have values smaller than M. Any value may be used. In the following description, there are two down converters 424-n and A / D converters 425-n (down converters 424-n-1 to 424-n-2 and A / D converters 425-n-1 to 425-n). The case of n-2) will be described as an example.

  The difference from the transmission / reception signal processing circuits 451-1 to 451-N in the first embodiment in FIG. 1 or FIG. 2 is that the down converter 424-n and the A / D converter 425-n are not arranged in all antenna systems. Is a point. For example, the antenna elements 401-2 to 401-3 do not include the down converter 424-n and the A / D converter 425-n. Further, for example, a down converter 424-n and an A / D converter 425-n are provided in three systems (generally three systems or more and M-1 systems or less) among the antenna elements 401-1 to 401-M. By arranging and performing the procedure for acquiring channel information described with reference to FIGS. 14 to 17, the down converter 424-n and the A / D converter 425-n are not required in the system of the remaining antenna elements. is there.

  Hereinafter, the flow of a specific signal that is a difference will be described focusing on the difference. The signal processing for transmission and reception is basically the same as in FIGS. 1 and 2, and the difference is the signal processing for calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M. Exists only in The details are shown below.

  In the signal processing for calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M, the switch 403-n-1 includes the down converter 424-n-1 and the phase shifter 402-n. n-1 and a switch 403-n-4 connects the downconverter 424-n-2 and the phase shifter 402-n-4 (actually, a downconverter, not shown here, The same applies to an antenna system switch provided with an A / D converter), and the remaining switches are left unconnected. In this state, first, a training signal for channel estimation is transmitted from the wireless station device of the communication partner from which the rotation amount of the complex phase is to be obtained, and the wireless station device receives this signal.

  The signals received by the antenna elements 401-n-1 to 401-M are input to the phase shifters 402-n-1 to 402-n-M, and are analogized by the phase shifters 402-n-1 to 402-n-M. A predetermined complex phase rotation is applied on the signal, and each is input to the switches 403-n-1 to 403-n-M. Switches 403-n-1 and 403-n-4 (actually, the same applies to an antenna system switch including a down converter and an A / D converter not shown here, Therefore, the signals input to the two antenna systems will be input to the down converters 424-n-1 to 424-n-2. The down converters 424-n-1 to 424-n-2 convert the input radio frequency signals into analog baseband signals. The A / D converters 425-n-1 to 425-n-2 convert analog signals into digital baseband signals.

  The information converted into digital baseband signals by the A / D converters 425-n-1 to 425-n-2 is input to the correlation calculation circuit 405-n. The correlation calculation circuit 405-n calculates the amount of rotation of the complex phase using the equations (1) to (3) based on the input information. When the calibration process is necessary as needed, the rotation amount of the complex phase on the transmission side is determined as a value in consideration of the calibration coefficient in Expressions (1) to (3). The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the complex phase rotation amount prediction circuit 410-n together with the identification number of the wireless station device to communicate with.

  The complex phase rotation amount prediction circuit 410-n predicts channel information of the remaining antenna elements based on channel information of a limited antenna system. In the case of FIG. 22, the complex phase rotation amount prediction circuit 410-n includes the antenna elements 401-1 and 401-4 (actually, a down converter and an A / D converter not shown are provided. Based on the channel information received by the system antenna elements, channel information of the remaining antenna elements is predicted. The complex phase rotation amount prediction circuit 410-n calculates the amount of rotation of the complex phase to be performed by the phase shifters 402-n-1 to 402-n-M, and together with the identification number of the wireless station device to communicate with, Input to the phase shift control circuit 406-n. In the phase shift control circuit 406-n, the value input here is managed by being stored in a memory or the like.

  Note that when performing actual data communication, that is, when performing transmission or reception processing, a control circuit (not shown in FIG. 22) grasps a wireless station device to be a communication partner and performs phase shift control circuit 406-n , The phase shifters 402-n-1 to 402-n-M instruct the phase shifters 402-n-1 to 402-n-M for the amount of rotation of the complex phase corresponding to the wireless station device that performs communication. In the case of 1 to 402-nM, the amount of rotation of the complex phase is set to realize analog beamforming.

Although not explicitly shown in FIG. 22, for example, if a high power amplifier (HPA) on the transmission side is arranged, it is arranged at a point indicated by “A” in the figure, and a low noise amplifier (HP if placing LNA) or the like, will be placed at a point with a description of "B" and "C _1" - "C _2" in FIG. Regarding “A” and “B”, since the transmission / reception signal processing circuit 651-n is not assumed to cooperate with each other, a calibration process for removing uncertainty of the complex phase of the individual high power amplifier and low noise amplifier Is unnecessary, but regarding the low-noise amplifier at the point where “C ₁ ” to “C ₂ ” are described, the antennas 401-1, 401-4, etc. in the same transmission / reception signal processing circuit 651-n are not connected. It is necessary to remove the uncertainty of the complex phase of the low-noise amplifier of each system by using, for example, a calibration technique of the implicit feedback of the related art, because the uncertainty of the complex phase may be caused between the two. .

The present invention can be applied to any method, and the specific method of the calibration process is not limited. Considering this calibration result, for example, assuming that the rotation amounts of the complex phases at “C ₁ ” and “C ₂ ” are +10 degrees and +20 degrees, the complex phases obtained by the equations (1) to (3) are obtained. Is corrected by -10 degrees and -20 degrees to adjust the phase rotation amount. The information on the calibration result is collected by a calibration circuit (not shown), and is collected by the phase shift control circuit 406-n, the complex phase rotation amount prediction circuit 410-n, or the correlation calculation circuit 405-n. The correction is performed using the information. In the above description, FIGS. 2 and 4 to 11 are described as various variations with respect to the description of FIGS. 1 and 7, but if the same extension is performed, it is the same as other operation methods. It can be used for
In the figure, the switches 403-n-2, 403-n-3, 403-n-5 to 403-n-M of the system without the down converter and the A / D converter have the phase shifter 402. −n−1 to 402−n−M, the connection state is set in the signal processing when calculating the amount of rotation of the complex phase, or these switches 403-n−2, 403−n−3, and 403−n− Omitting 5 to 403-nM does not cause any problem in practice.

  FIG. 23 is a functional block diagram illustrating another configuration example of the transmission / reception signal processing circuit 652-n according to the present embodiment. FIG. 23 illustrates a combination of the transmission / reception signal processing circuit 455-n in the first embodiment of FIG. 6 and the transmission / reception signal processing circuit 556-n corresponding to the transmission / reception signal processing circuit 556-n in the second embodiment of FIG. It corresponds to another implementation. The same numbers are assigned to the same functional blocks as those in the above-described drawings.

  The transmission / reception signal processing circuit 652-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A / D converter 125-n. n, distribution couplers 414-n, distribution couplers 415-n, phase shifters 402-n-1 to 402-n-M, switches 403-n-1 to 403-n-M, Phasers 409-n-1 to 409-n-M, downconverters (DC) 424-n-1 to 424-n-2, A / D converters 425-n-1 to 425-n-2, , A correlation calculation circuit 405-n, a phase shift control circuit 406-n, and a complex phase rotation amount prediction circuit 410-n.

  The difference from the transmission / reception signal processing circuit 455-n in the first embodiment in FIG. 6 is that the down converter 424-n and the A / D converter 425-n are not arranged in all antenna systems. For example, the antenna elements 441-2 to 441-3 do not include the down converter 424-n and the A / D converter 425-n. Further, for example, a down converter 424-n, an A / D converter 425-n, and the like are provided for three systems (generally, three systems or more and M-1 systems or less) among the antenna elements 441-1 to 441-M. By arranging and performing the procedure for acquiring channel information described with reference to FIGS. 14 to 17, the down converter 424-n and the A / D converter 425-n are not required in the system of the remaining antenna elements. is there.

  The following shows a specific signal flow that becomes a difference by focusing on the difference. Basically, the signal processing related to transmission and reception is the same as the description of FIG. 6 (and the description of FIG. 4 related to FIG. 6), and the difference is calculated by the phase shifters 402-n-1 to 402-n-M. It exists only in signal processing when calculating the amount of phase rotation. The details are shown below. In the signal processing for calculating the amount of rotation of the complex phase performed by the phase shifters 402-n-1 to 402-n-M, the switch 403-n-1 includes the down converter 424-n-1 and the phase shifter 402-n. n-1 and a switch 403-n-4 connects the downconverter 424-n-2 and the phase shifter 402-n-4 (actually, a downconverter, not shown here, The same applies to an antenna system switch provided with an A / D converter), and the remaining switches are left unconnected.

  The switching of these switches is managed by the correlation calculating circuit 405-n under the instruction of a control circuit (not shown), and the phase shifter 402-n-1 is used except when calculating the amount of rotation of the complex phase. 402-nM are connected to the distribution coupler 415-n. When performing this processing, the phase rotation amounts of the phase shifters 402-n-1 to 402-n-M are set to predetermined values. The rotation amount of the complex phase obtained in the subsequent processing is set as a difference from an initial predetermined value, as in the above description. Also, other transmission / reception signal processing circuits 652-n (not shown) perform the same processing all at once under the control of a control circuit (not shown).

  In this state, first, a training signal for channel estimation is transmitted from the wireless station device of the communication partner from which the rotation amount of the complex phase is to be obtained, and the wireless station device receives this signal. The signals received by the antenna elements 441-n-1 to 441-n-M are input to the phase shifters 402-n-1 to 402-n-M, and the phase shifters 402-n-1 to 402-n-M are input. At M, a predetermined complex phase rotation is applied on the analog signal, and each is input to the switches 403-n-1 to 403-n-M. Switches 403-n-1 and 403-n-4 (actually, the same applies to an antenna system switch including a down converter and an A / D converter not shown here, Therefore, signals input to these two antenna systems will be input to down converters 424-n-1 to 424-n-2. The down-converters 424-n-1 to 424-n-2 convert the input radio frequency signals into analog baseband signals, and the A / D converters 425-n-1 to 425-n-2 convert the signals into analog baseband signals. Converts a signal to a digital baseband signal.

  The information converted into digital baseband signals by the A / D converters 425-n-1 to 425-n-2 is input to the correlation calculation circuit 405-n. The correlation calculation circuit 405-n calculates the rotation amount of the complex phase by using Expressions (1) to (3). When the calibration process is necessary as needed, the rotation amount of the complex phase on the transmission side is determined as a value in consideration of the calibration coefficient in Expressions (1) to (3). The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the complex phase rotation amount prediction circuit 410-n together with the identification number of the wireless station device with which to communicate.

  The complex phase rotation amount prediction circuit 410-n predicts channel information of the remaining antenna elements based on channel information of a limited antenna system. In the case of FIG. 22, the complex phase rotation amount prediction circuit 410-n calculates the amount of rotation of the complex phase to be performed by the phase shifters 402-n-1 to 402-n-M, and the wireless communication partner It is input to the phase shift control circuit 406-n together with the identification number of the station device. In the phase shift control circuit 406-n, the value input here is managed by being stored in a memory or the like.

  Further, the rotation amount of the complex phase described above relates to the phase rotation amount of the phase shifters 402-n-1 to 402-n-M in the receiving system, but the rotation amount of the complex phase rotation amount in a low noise amplifier, a high power amplifier, and the like. In order to cancel the individual difference, a calibration process in the implicit feedback of the related art is performed, the rotation amount of the complex phase in the transmission system is converted, and a value set in the phase shifters 409-n-1 to 409-n-M Is similarly managed by being stored in a memory.

  Note that when performing actual data communication, that is, when performing transmission or reception processing, a control circuit (not shown in FIG. 23) grasps a wireless station apparatus to be a communication partner and performs phase shift control circuit 406-n , The amount of rotation of the complex phase corresponding to the wireless station device performing communication is transmitted to the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M. The amount of phase rotation is specified, and the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M set the amount of rotation of the complex phase to be analog Beam forming is realized.

Although not explicitly shown in the drawing, if a high-power amplifier (HPA) or the like on the transmission side is to be disposed, for example, points “A” and “D ₁ ” to “D _M ” in the drawing will be described. arranged, if placing a recipient of the low-noise amplifier (LNA) or the like, will be placed at a point with a description of "B" and "E _1" - "E _M" in FIG. Regarding "A" and "B", since the transmission / reception signal processing circuit 652-n does not assume that they cooperate with each other, calibration processing for removing the uncertainty of the complex phases of the individual high-power amplifiers and low-noise amplifiers Are unnecessary, but the same transmission / reception signal processing circuit is used for the high power amplifier with the description of “D ₁ ” to “D _M ” and the low noise amplifier with the description of “E ₁ ” to “E _M ” 652-n may cause uncertainty in the complex phase between each of the antenna elements 401-n-1 to M and the antenna elements 441-n-1 to 441-n-M. It is necessary to remove the uncertainty of the complex phase of the low-noise amplifier of each system by using the calibration method of the implicit feedback.

The present invention can be applied to any method, and the specific method of the calibration process is not limited. Considering this calibration result, for example, assuming that the rotation amount of the complex phase at “C ₁ ”, “C ₂ ”, etc. is +10 degrees, +20 degrees, etc., the equations (1) to (3) are obtained. The phase rotation amount is adjusted by correcting the rotation amount of the complex phase by, for example, -10 degrees or -20 degrees. The information on the calibration result is collected by a calibration circuit (not shown), and is collected by the phase shift control circuit 406-n, the complex phase rotation amount prediction circuit 410-n, or the correlation calculation circuit 405-n. The correction is performed using the information.
Similarly to FIG. 22, the phase shift is performed for the switches 403-n-2, 403-n-3, 403-n-5 to 403-n-M of the system without the down converter and the A / D converter. Switches 402-n-1 to 402-n-M in the signal processing when calculating the amount of rotation of the complex phase, or switch 403-n-2, 403-n-3, 403- Even if n-5 to 403-nM is omitted, there is no practical problem.

  Although the above description has been made mainly on the case where the rotation of the complex phase of each antenna element is performed on an analog signal, it can also be used in digital signal processing. In this case, since the A / D converters are mounted on all the antenna elements in the reception signal processing, the rotation amounts of the other antenna elements are evaluated based on the complex phase rotation amount evaluation result performed using a small number of antenna elements. Although there is no direct merit in determining the amount of rotation of the complex phase, the signal processing on the transmitting side can be effectively used when the transmitting antenna and the receiving antenna are physically different.

  FIG. 24 is a functional block diagram showing another configuration example of the transmission / reception signal processing circuit 653-n using digital signal processing according to the third embodiment of the present invention. FIG. 24 is a diagram for explaining the present invention applied to the transmission / reception signal processing circuits 929-1 to 929 -N in the configuration example of the wireless station device using time-axis beamforming in the conventional technique shown in FIGS. 28 and 29. It is an improvement. The difference from the transmission / reception signal processing circuit 929-n shown in FIGS. 28 and 29 is obtained by adding a correlation calculation circuit 405-n, a complex phase rotation amount prediction circuit 410-n, and a transmission weight calculation circuit 411-n. The transmission weight calculation circuit 411-n provides information on the transmission weight to be input to the axis transmission weight multiplication circuit.

  For example, when the transmitting antenna and the receiving antenna are different as shown in FIGS. 16 and 17, all or some of the A / D converters 925-n-1 to 925-n-M used for reception are used. Sampling data for some systems is input to a correlation calculation circuit 405-n, and a correlation operation between a reference antenna element and another antenna element is performed based on this information. The correlation value is input to the complex phase rotation amount prediction circuit 410-n. Based on the input complex phase and the coordinate information of the antenna element, the complex value of all the antenna elements is calculated by the least square method or the linear estimation method described above. The amount of phase rotation is estimated, and the estimation result is input to the transmission weight calculation circuit 411-n. In order to cancel the relative complex phase difference of each antenna element with respect to the reference antenna, the transmission weight calculation circuit 411-n transmits a complex number corresponding to Exp (−j) to the complex phase ψ on the time axis of each antenna element. The weight is input to a time axis transmission weight multiplying circuit 921-n. The time axis transmission weight multiplying circuit 921-n multiplies this value for each antenna element and then D / A converters 922-n-1 to 922. -Input to n-M. In this way, even in digital signal processing, transmission directivity control using a transmission antenna different from the reception antenna can be performed based on information obtained from some reception antennas.

  In FIG. 24, the number of input systems from the A / D converters 925-n-1 to 925-n-M to the correlation calculation circuit 405-n is not explicitly shown. , A signal from A / D converters (parts of 925-n-1 to 925-n-M) related to a part of the entire antenna system, or a configuration in which all of the signals are input. I do not care. Further, as shown in FIG. 20, when some of the antenna elements 568-1 to 568-3 of the antenna element group forming the transmitting antenna 565 are used as receiving antennas, the output of only the corresponding receiving antenna is used. May be input to the correlation calculation circuit 405-n (in this case, the reception antenna may not be used for the reception signal processing of the user data).

  As described above, according to the present embodiment, the wireless station device acquires channel information of some antenna elements without acquiring channel information of all antenna elements. The wireless station device calculates weight information from the acquired channel information, and calculates the weight information of other antenna elements by approximation based on the calculated weight information. Accordingly, there is no need to newly provide a switching mechanism such as a switch, and the number of A / D converters and D / A converters required to acquire channel information can be reduced. Further, as described above, by calculating the weight information of the other antenna elements by approximation, even if the transmitting antenna and the receiving antenna are different, the implicit feedback can be used. Therefore, it is possible to reduce the size and cost and to use the implicit feedback even when the transmitting antenna and the receiving antenna are different.

[Fourth embodiment]
In the conventional implicit feedback technique, a technique of estimating downlink channel information from uplink channel information by using channel symmetry has been general. For this reason, for example, when the transmitting antenna and the receiving antenna are different, or when the frequency of the uplink is different from the frequency of the downlink, it has been considered that implicit feedback utilizing the symmetry of the channel cannot be used. However, as shown in the third embodiment, it is possible to predict a channel for an antenna element different from the antenna element used for channel feedback, and to use implicit feedback by using such predicted channel information. The third embodiment has shown that this is also possible.

This utilizes the characteristic of the environment where the incoming wave can be approximated by a plane wave, and the line of sight wave is dominant.In all cases, when the arrival direction of the plane wave is determined, the path length difference of each antenna element is determined, and according to the path length difference Thus, the complex phase of the channel information of each antenna element (or the complex phase rotated by the phase shifter, that is, the transmission / reception weight) could be predicted according to the geometric rules. When such a condition is satisfied, considering that the complex phase of 2πΔL / λ is rotated using the path length difference ΔL and the wavelength λ, the receiving side (assuming a base station apparatus, here an uplink a rotation amount [Delta] [theta] _UL of complex phase obtained in that), assuming a transmission side (base station apparatus, wherein between the rotational amount [Delta] [theta] _UL of complex phase be determined by the downlink), a predetermined Rules exist.

FIG. 25 is a diagram illustrating an outline of a method of converting a complex phase according to the fourth embodiment of the present invention. 25A shows an incoming wave of frequency F ₁ (wavelength λ ₁ ), and FIG. 25B shows an incoming wave of frequency F ₂ (wavelength λ ₂ ). Here, it is assumed that F ₁ <F ₂ , and the wavelength λ ₂ is shorter than the wavelength λ ₁ . Even for the same path length difference ΔL, the amount of rotation of the complex phase differs according to the difference in wavelength or frequency, and the difference can be derived using equation (6).

  FIG. 26 is a flowchart showing the flow of processing of the implicit feedback method when FDD is applied in the fourth embodiment of the present invention. First, when the process is started, the training signal transmitted from the transmitting-side wireless station device is received by the receiving-side wireless station device, and the receiving-side wireless station device transmits digital sampling data of the received signal at a predetermined period. (Step S101). The wireless station device on the receiving side performs a correlation operation with the reference antenna according to Expression (1) using the sampling data for the acquired training signal, and calculates the rotation amount of the complex phase from the complex phase of the correlation value (step S102). ). If this is the complex phase rotation amount information on some of the antenna elements, the receiving-side radio station device predicts the complex phase rotation amount information on the other antenna elements as in the third embodiment (step S103). Thereafter, the wireless station device on the receiving side predicts the rotation amount information of the complex phase of the downlink having a different frequency from the rotation amount information of the complex phase of the uplink using Expression (6) (step S104). Since the correspondence between the high-power amplifier and the low-noise amplifier differs between the uplink and the downlink, the wireless station device on the receiving side acquires the rotation amount information of the complex phase of the downlink by an implicit feedback technique using a calibration process. (Step S105). The wireless station device on the receiving side acquires the rotation amount of the complex phase in the transmission directivity formation in which the complex phase is inverted, based on the predicted rotation amount information of the complex phase (step S106). The above is the extension of the implicit feedback method to FDD in the fourth embodiment of the present invention. The amount of rotation of the complex phase obtained here is set to the phase shifter when the directivity is formed by analog signal processing, and the obtained complex phase for each antenna element is formed when the directivity is formed by digital signal processing. For ψ, Exp (jψ) is set as the transmission weight of the corresponding antenna element.

In the above description, the rotation amount of the complex phase corresponding to the complex coefficient c _{j of the} result of the correlation operation of the above equation (1) is determined first in steps S101 to S105 for the target to which the third embodiment is applied. The case where the rotation amount of the complex phase used in the transmission directivity control corresponding to the above equation (2) with the complex phase inverted is obtained later in step S106 has been described as an example, but the rotation of the complex phase acquired in the processing of step S102 has been described. Using the amount information, the amount of rotation of the complex phase in forming the reception directivity whose sign is inverted is acquired first, and the third embodiment is applied to the amount of rotation of the complex phase to receive the remaining antenna elements. The amount of rotation of the complex phase on the side is calculated first, and a conversion process for different frequencies using equation (6) is performed as the process of step S104, and the amount of complex phase rotation at the time of forming the transmission directivity is calculated. It may be calculated.

  Further, the processing in step S104 described above is performed first if it is performed by the complex phase rotation amount prediction circuit 410-n (n = 1,..., N) in FIGS. 22 and 23 of the third embodiment. Phase shifters 402-n-1 to 402-n-M (in the case of FIG. 22) or phase shifters 409-n-1 to 409-n-M (in the case of FIG. 23) when the transmission directivity is formed in the above description. Can be handled by changing the amount of rotation of the complex phase set in accordance with the frequency. Further, if it is implemented by the complex phase rotation amount prediction circuit 410-n or the transmission weight calculation circuit 411-n (n = 1,..., N) in FIG. 24 of the third embodiment, similarly in digital signal processing. This can be dealt with by changing the transmission weight set in the time axis transmission weight multiplication circuit at the time of forming the transmission directivity in the description given above in accordance with the frequency and operating. In the first and second embodiments, the complex phase rotation amount prediction circuit 410 is provided between the correlation calculation circuit 405-n or 505-n and the phase shift control circuit 406-n or 506-n. If -n is arranged and the same processing is performed here, it is possible to apply the fourth embodiment even in combination with the first embodiment and the second embodiment.

  In this case, the phase of the downlink is phase shifters 402-1-1 to 402-NM in FIGS. 1 and 2, and the phase of the downlink is phase shifters 409- in FIGS. 7 to 9, the phase of the downlink is phase shifters 502-1-1 to 502-NM, and in FIGS. 10 to 12, the phase of the downlink is Are phase shifters 509-n-1 to 509-n-M, which perform rotation processing of the converted complex phase. Therefore, in FIGS. 1 and 2 and FIGS. 7 to 9, even in the same communication partner station, the rotation amounts of the uplink and downlink complex phases should be managed so as to be different values. A series of management is managed by the control circuit 460.

  As described above, according to the present embodiment, the wireless station device calculates channel information and weight information at a certain frequency in each antenna element and a different frequency from each other based on a path length difference at a certain frequency. This makes it possible to use implicit feedback even when constructing a wireless system in which the uplink frequency and the downlink frequency are different, such as FDD.

  In each of the embodiments described above, the complex phase rotation is performed on the radio frequency analog signal. However, the position of the up converter is changed, and the complex phase rotation is performed on the baseband or intermediate frequency analog signal. It is also possible to adopt a configuration in which a frequency conversion with a radio frequency is performed in a subsequent stage or a preceding stage.

[Supplementary information on the embodiment]
The supplementary items relating to the embodiment of the present invention described above are shown below.
The wireless station device in the above-described embodiment may be realized by a computer. In that case, a program for realizing this function may be recorded on a computer-readable recording medium, and the program recorded on this recording medium may be read and executed by a computer system. Here, the “computer system” includes an OS and hardware such as peripheral devices. The “computer-readable recording medium” refers to a portable medium such as a flexible disk, a magneto-optical disk, a ROM, and a CD-ROM, and a storage device such as a hard disk built in a computer system. Further, a “computer-readable recording medium” refers to a communication line for transmitting a program via a network such as the Internet or a communication line such as a telephone line. Such a program may include a program that holds a program for a certain period of time, such as a volatile memory in a computer system serving as a server or a client in that case. The program may be for realizing a part of the functions described above, or may be a program that can realize the functions described above in combination with a program already recorded in a computer system, It may be realized using hardware such as a PLD (Programmable Logic Device) or an FPGA (Field Programmable Gate Array).

  As described above, the embodiments of the present invention have been described with reference to the drawings. However, it is apparent that the above embodiments are merely examples of the present invention, and the present invention is not limited to the above embodiments. is there. Therefore, additions, omissions, replacements, and other changes may be made to the components without departing from the technical spirit and scope of the present invention.

  The present invention is applicable to a wireless communication device that transmits and receives a wireless signal using a plurality of antennas.

120-1 to 120-N ... Modulators 122-1 to 122-N, 122-n ... D / A converters 123-1 to 123-N, 123-n ... Up-converters 124-1 to 124-N, 124 -N: Down converters 125-1 to 125-N, 125-n: A / D converters 127-1 to 127-N: TDD switches 130-1 to 130-N: Demodulator 140: Baseband signal processing circuit 141 ... signal separation circuits 401-1 to 401-M ... antenna elements 401-1-1 to 401-NM, 401-n-1 to 401-nM ... antenna elements 402-1-1 to 402-N- M, 402-n-1 to 402-nM ... phase shifters 403-1-1 to 403-NM, 403-n-1 to 403-nM ... switches 404-1 to 404-N ... Distribution couplers 405-1 to 405-N, 40 -N correlation calculation circuits 406-1 to 406-N, 406-n phase shift control circuits 407-1 to 407-M distribution couplers 408-n-1 to 408-n-M TDD switches 409-n -1 to 409-nM phase shifter 410-n complex phase rotation amount prediction circuit 411-n transmission weight calculation circuit 414-n distribution coupler 415-n distribution coupler 424-1-1 424-NM, 424-n-1 to 424-n-M... Downconverters 425-1-1 to 425-NM, 425-n-1 to 425-nm,... A / D converter 441 -1 to 441-M ... antenna elements 441-n-1 to 441-n-M ... antenna elements 450 ... radio station devices 451-1 to 451-N ... transmission / reception signal processing circuit 452 ... radio station devices 453-n ... transmission and reception Signal processing circuit 454-n: transmission / reception signal Processing circuits 455-n ... reception signal processing circuit 460 ... control circuit 901-1~901-N ... modulator 902 ... precoder _903-1~903-N 0 ... IFFT & GI imparting circuit _904-1~904-N 0 ... D / A converter 905-1~905-N 0 _... upconverter 906-1~906-N 0 _... downconverter _907-1~907-N 0 ... A / D converter _908-1~908-N 0 ... GI removing & FFT circuit 909 ... postcoders 910-1~910-N ... demodulator 911 ... TDD switch 912-1~912-N 0 _... distributor coupler _{913-1-1~913-N} 0 -M 0 _... phase shifter 915 -1~915-M 0 _... distributor coupler 916-1~916-M 0 _... antenna elements 921-1~921-N ... time base transmission weight multiplying circuit 922-1～922- ... D / A converters 922-1-1 to 922-NM ... D / A converters 923-1 to 923-M ... up-converters 923-1-1 to 923-NM ... up-converters 924-1 ... 924-M... Down-converters 924-1-1 through 924-NM... Down-converters 925-1 through 925-M... A / D converters 925-1-1 through 925-NM. 926-1 to 926-N time reception weight multiplying circuits 927-1 to 927-N TDD switches 928-1 to 928-M antenna elements 928-1-1 to 928-N-M antenna elements 929 −1 to 929-N transmission / reception signal processing circuits 941-1 to 941-M distribution coupler 942 wireless station devices 943-1 to 943-M addition and combining units 944-1 to 944-M replication device 945 Radio station device 951- 1-951-3 High-power amplifiers 952-1 to 952-3 Low-noise amplifiers 953-1 to 953-3 TDD switches 954-1 to 954-3 Antenna elements 955-1 to 955-3 Wireless module

Claims (4)

A wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device,
A communication unit configured to communicate with the other wireless communication device using a first frequency used in an uplink and a second frequency different from the first frequency used in a downlink;
A signal conversion unit that is provided in all antenna elements or a plurality of partial systems and converts a radio frequency analog signal received by the antenna element into a baseband digital signal,
Using a digital signal corresponding to the training signal transmitted by the other wireless communication device converted by the signal conversion unit, a reference antenna selected from a plurality of antenna elements provided with the signal conversion unit Element, a correlation calculation unit that calculates a correlation for each combination of other antenna elements,
A rotation amount calculation unit that calculates a rotation amount of a complex phase to be given to a reception signal of an individual antenna element based on a calculation result of the correlation by the correlation calculation unit,
A complex phase to be given to a transmission signal of an individual antenna element based on information on a ratio between the first frequency and the second frequency and information calculated by the correlation calculator or the rotation amount calculator. A rotation amount prediction unit for predicting the rotation amount of
Based on the complex phase predicted by the rotation amount prediction unit, a phase rotation amount management unit that manages the rotation amount of the complex phase to be given to the transmission signal for each antenna element,
A transmission signal generation unit that generates a signal addressed to the other wireless communication device,
A rotation amount of the complex phase managed by the phase rotation amount management unit, a phase rotation unit that performs a phase rotation on an analog signal or a digital signal for a transmission signal for each antenna element used for transmission,
A signal transmitting unit that transmits a signal output via the phase rotation unit to the other wireless communication device via the antenna element,
A wireless communication device comprising: A first signal distributor that branches the transmission signal generated by the transmission signal generator on an analog signal for each antenna element;
The phase rotation unit,
The wireless communication device according to claim 1, wherein the amount of rotation of the complex phase managed by the phase rotation amount management unit is set in a phase shifter provided for each antenna element. A second signal distribution unit that branches the transmission signal generated by the transmission signal generation unit on a digital signal for each of the antenna elements,
The phase rotation unit,
The wireless communication apparatus according to claim 1, wherein the rotation amount ψ of the complex phase for each antenna element managed by the phase rotation amount management unit is multiplied by a coefficient Exp (jψ) on a digital signal for each antenna element. A wireless communication method executed by a wireless communication device that wirelessly communicates with another wireless communication device by forming directivity using one or more array antennas including a plurality of antenna elements,
A communication step of communicating with the other wireless communication device using a first frequency used in an uplink and a second frequency different from the first frequency used in a downlink;
Transmitted by the other wireless communication device which is provided in all antenna elements or a plurality of partial systems, and which is converted by a signal conversion unit which converts a radio frequency analog signal received by the antenna element into a baseband digital signal. Using a digital signal corresponding to a training signal, a correlation for calculating a correlation for each combination of a reference antenna element selected from a plurality of antenna elements provided with the signal conversion unit and other antenna elements. A calculating step;
A rotation amount calculating step of calculating a rotation amount of a complex phase to be given to a received signal of an individual antenna element based on a calculation result of the correlation in the correlation calculating step,
Based on information on the ratio between the first frequency and the second frequency and the calculation result of the correlation calculation step or the rotation amount calculation step, a complex phase to be given to a transmission signal of an individual antenna element A rotation amount prediction step of predicting the rotation amount;
Based on the complex phase predicted in the rotation amount prediction step, a phase rotation amount management step of managing a rotation amount of a complex phase to be given to the transmission signal for each antenna element,
A transmission signal generation step of generating a signal addressed to the other wireless communication device,
The rotation amount of the complex phase managed in the phase rotation amount management step is set in a phase rotation unit that performs a phase rotation on an analog signal or a digital signal with respect to a transmission signal for each antenna element used for transmission, and A signal transmission step of transmitting a signal output through the unit to the other wireless communication device via the antenna element,
A wireless communication method comprising:

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