JP6646544B2 - Wireless communication device and wireless communication method - Google Patents

Description

  The present invention relates to a wireless communication device and a wireless communication method.

[Background surrounding 5th generation mobile communication]
At present, high-performance mobile communication terminals such as smartphones are exploding. With regard to mobile phones, the third generation mobile communication has shifted to the fourth generation mobile communication, and research and development on the further fifth generation mobile communication (commonly known as "5G") are currently underway. One of the studies performed on 5G is the use of macro cells and small cells.

  In conventional mobile phones, one service area is set to a radius of several kilometers, and one macro station area is covered by one base station device. However, there is a very large number of users in such a macrocell. Since the entire system capacity is shared by each user, when accommodating an enormous number of users, the throughput for each user decreases.

  In order to avoid such a decrease in throughput, a technology for setting a small cell, which is a very small service area with a radius of about several tens of meters, in a densely populated area where traffic is concentrated has been developed. In this technology, small traffic is used to offload spot-like traffic to a network without going through a macro cell. Here, it is assumed that the terminal device can simultaneously and concurrently use the communication capability in the small cell and the communication capability in the macro cell. By using such a terminal device, user data is accommodated on the small cell side while exchanging control information using a macro cell. This makes it possible to maximize the advantages of the macro cell and the small cell.

  In 5G described above, the target value of the transmission rate is set to 10 Gbit / s (gigabits per second) or more, and even in this small cell, the same large-capacity communication is performed to achieve efficient offload of traffic. There is a need to. In a macro cell, it is assumed that a microwave band having a low frequency is used to allow long-distance propagation. However, in consideration of the current situation of the microwave band where frequency resources are already depleted, small cells that are expected to communicate at relatively short distances are expected to use quasi-millimeter wave bands or millimeter wave bands with relatively high frequencies. Have been. The feature of this high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. Therefore, it is preferable that the small cell base station is ideally installed at a location close to the user terminal. For example, in a place where installation is easy, such as on the roof of a building, the distance between the user terminal and the base station is too large, which is not preferable in line design.

  On the other hand, since the small cell is set in a place where traffic is concentrated, even in a place where it is difficult to lay an optical fiber, there is a case where the installation of the base station device is strongly desired. For example, assuming that small cell base station equipment is installed in a place where there are very many people, such as in front of a station such as Shinjuku or Shibuya, propagation attenuation increases on the rooftop of a building adjacent to such a place. . Therefore, installation on a place lower than the roof of the building, for example, on the wall surface of the building, may be required. However, it may be difficult to lay optical fiber on the wall of an existing building, and in such cases, it may be necessary to provide a backhaul line to the base station device using a wireless line. is there.

  When such a backhaul line is provided, it is necessary to perform a large-capacity transmission of 10 Gbit / s or more by utilizing a millimeter wave band in order to cope with a large-capacity transmission of 10 Gbit / s or more required in a small cell. There is. In such an environment, since both opposing radio station devices are fixedly installed in a stable place, it is natural that the line of sight is generally secured naturally and the directional antennas are directed toward each other. is there. In this case, although reflected waves between buildings and the like exist to some extent, most of the received signals are line-of-sight components and are expected to be in a state that is difficult to say in a multipath environment. This situation is the same for the access system if the small cell base station device is installed in a high place such as a building wall surface and is generally used in a line-of-sight environment in a manner of looking down on the user from above.

  Next, for large-capacity transmission of 10 Gbit / s or more, which is the transmission rate required for 5G, frequency resources with a very wide bandwidth can be used by utilizing the millimeter wave band. Is growing. For example, if a backhaul line using a millimeter wave band is assumed, the E-band (71 to 76 GHz and 81 to 86 GHz) is used as an example, and if a 1 GHz bandwidth is used, the frequency utilization efficiency is 10 bits. / S / Hz. However, existing radio equipment for achieving a frequency utilization efficiency of 10 bit / s / Hz generally employs spatial multiplexing transmission using a multiple-input multiple-output (MIMO) channel. Spatial multiplexing transmission generally uses a multipath environment, and when a singular value decomposition of a channel matrix H expressing a transfer function of a MIMO channel in a matrix format is performed, a distribution of an absolute value of a singular value obtained as a result is obtained. Represents the characteristics of the spatial multiplexing transmission. Specifically, the square of the absolute value of the singular value is a value proportional to the signal-to-noise ratio (SNR). Communication is not established unless it has a sufficiently large value after the second singular value. The same applies to large-capacity transmission in small cells that are access systems, but realizing spatial multiplexing transmission in an environment where such line-of-sight waves are dominant is indispensable for realizing a target wireless system. .

  As described above, in 5G, use of the millimeter wave band is expected for both the access system and the backhaul line. As described above, the characteristic of the high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. For example, comparing the 2 GHz band (existing access system) with the 80 GHz band (future system using the millimeter wave band), since the frequency is 40 times, the propagation attenuation is 1600 times and the line gain of 32 dB is insufficient. Will be. Of course, it is not necessary to cover all of 32 dB because it is not necessary to cover a wider area than a macro cell. However, in a high frequency band, a high power amplifier in a transmission stage does not have a high output device. In this case, it is considered that additional line gain must be secured at a level of several tens of dB. Furthermore, since spatial multiplexing transmission is also expected in such an environment, a radio system equipped with a much larger number of antenna elements than in the prior art has been studied in both the base station and the terminal station. . Such a technique is called Massive MIMO. In the following, conventional techniques related to Massive MIMO are introduced.

  In Non-Patent Document 1, the base station side has 256 elements and the terminal station side has 16 elements, and aims at spatial multiplexing transmission of 16 streams using a channel matrix of 256 × 16 size. In this Non-Patent Document 1, in order to transmit a plurality of signal sequences (streams), the directivity is formed by analog beamforming using rotation of a complex phase amount in a wireless analog circuit and digital baseband circuit. Performed in conjunction with digital beamforming in the digital domain. This avoids heavy use of analog / digital (A / D) converters and digital / analog (D / A) converters, and reduces power consumption and takes measures against insufficient line gain when feeding back channel information. Hereinafter, an outline of the technology disclosed in Non-Patent Document 1 will be described.

FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. The wireless station apparatus shown in FIG. 1 includes modulators 901-1 to 901-N (MOD # 1 to MOD # N), a precoder 902, an IFFT (Inverse Fast Fourier Transform) and a GI (Guard Interval). and applying circuit _{903-1~903-N 0,} a D / a converter _{904-1~904-N 0,} the up-converter (UC) _{905-1~905-N 0,} downconverter (DC) 906-1 a _{~906-N 0,} and a / D converter _{907-1~907-N 0,} GI removal & FFT: a (fast Fourier transform) circuit _{908-1~908-N 0,} the postcoder 909, a demodulator 910-1~910-N (DEM # 1~DEM # N), a TDD switch (TDD-SW) 911, a distributor coupler (HYB) _{912-1~912-N 0,} the phase shifter 913 -1 Comprises a _{-1~913-N 0} -M 0, and distributor coupler (HYB) _{915-1~915-M 0,} and antenna element _{916-1~916-M 0.} Where N corresponds to the number of multiplexed time of performing spatial multiplexing (the number of streams), N ₀ is the number of signal lines for performing signal processing for digitally beamforming, M ₀ is representative of the number of antenna elements (M ₀ ≧ N ₀ ≧ N). Furthermore the antenna elements _{916-1~916-M 0} constitute an array antenna as a whole.

Antenna element _{916-1~916-M 0} from the distributor coupler _{912-1~912-N 0} is common to transmission and reception. Also, TDD switches 911, up from the distributor coupler _{912-1~912-N 0} a connection to the antenna element _{916-1~916-M 0,} from the modulator 901-1~901-N corresponding to the transmission system converter and _{905-1~905-N 0,} switching between the demodulator 910-1~910-N from the down-converter _{906-1~906-N 0} corresponding to the receiving system. For example, the up converter 905-n and the distribution coupler 912-n are connected during transmission, and the down converter 906-n and the distribution coupler 912-n are connected during reception (n = 1,..., N ₀ ). An entire control circuit (not shown) manages the frame period and the transmission / reception timing, and switching of the TDD switch 911 is also performed by this control circuit.

Further, the phase shifter _{913-1-1~913-N 0} -M 0 adjusts the phase relationship between the transmit and receive signals in accordance with the beam pattern defined in advance, the phase rotation amount by the control circuit (not shown) Is also managed. The phase here is the same as the directivity control in the phased array antenna. For example, as directional gain in a predetermined direction across the antenna elements _{916-1~916-M 0} is the maximum, at each antenna element _{916-1~916-M 0} against incoming waves from that direction A complex phase corresponding to a value obtained by dividing the path length difference by the wavelength is adjusted. Thus, each antenna element _{916-1~916-M 0} to be able to transmit and receive signals in phase.

Here, the directivity is obtained by dividing a horizontal azimuth angle θ and a vertical azimuth angle φ by a predetermined angle step size, and selecting a corresponding complex (θ _i , φ _j ) for each menu. phase shifter sets of phase as a set _{913-n-1~913-n-} M 0 (n = 1, ..., n 0) to adjust the amount of phase. As a result, for example, one virtual element for the n-th signal sequence in the whole antenna elements 916-1 to 916 -M ₀ , the phase shifters 913 -n-1 to 913 -n-M ₀ , and the distribution coupler 912-n Behave as a directional antenna. These virtual directional antenna is physically via distribution couplers _{915-1~915-M 0,} will share the antenna element _{916-1~916-M 0.}

  Further, in the following description, as an example, a case is described in which communication is performed by forming a signal on the frequency axis as in an OFDM (Orthogonal Frequency Division Multiplexing) modulation method. Note that even in the case of single carrier transmission, when performing equalization processing on the frequency axis, the signal is temporarily converted to a signal on the frequency axis. The same discussion is possible without regard to OFDM or single-carrier transmission, assuming that the processing is performed after the signal has been converted into a signal of the same type.

The specific signal flow is as follows. First, signal transmission will be described. Each of the modulators 901-1 to 901-N generates a transmission signal of each stream to be spatially multiplexed. The precoder 902 appropriately performs signal synthesis between a plurality of virtual directional antennas, and enables efficient signal separation on the receiving station side. The pre-coding processing is equivalent to multiplication example N ₀ systematic virtual directional antenna actually unitary transformation matrix at the time of singular value decomposition of the MIMO channel matrix between the signal lines of the N lines for transmitting and receiving. Thereby, so-called eigenmode transmission is realized, and efficient transmission is realized. IFFT & GI imparting circuit _{903-1~903-N 0} is a transmission signal sequence formed in this way, into a signal on the time axis from the signal on the frequency axis, imparting guard intervals. If necessary, waveform shaping between symbols is also performed here. D / A converter _{904-1~904-N 0} is a digital signal generated in this manner, into an analog signal. Upconverter _{905-1~905-N 0} is the analog signal, converts the baseband signal to the radio frequency signals.

At the time of transmission, the TDD switch 911 connects the up-converter 905-n and the distribution coupler 912-n (n = 1,..., N ₀ ). It should be noted, 1~N ₀ of subscript is all behave in the same way. The distributing coupler 912-n (n = 1,..., N ₀ ) distributes the signal in the radio frequency band to signals of the number of antenna systems M ₀ , and disperses the signals to the phase shifters 913-n-1 to 913-n. input to -M _0. For example the phase shifter _{913-1-1~913-1-M 0} is first directional orientation corresponding to the signal sequence (θ _i, φ _j) on an analog signal to adjust the predetermined complex phase corresponding in is performed to transmit a first signal sequence after the adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} Similarly, the phase shifters 913 -N _{0-1 to} 913 -N ₀ -M ₀ are provided with a predetermined complex phase corresponding to the directivity direction (θ _{i ′} , φ _{j ′} ) corresponding to the N _{0 th} signal sequence. the adjustments performed on the analog signal and transmits the signal sequence of the N after adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} Incidentally, the distribution coupler 915-m (m = 1, ..., M 0) , the corresponding phase shifters 913-1-m, 913-2-m, ..., a signal input from _913-N 0 -m The signals are combined and output to the antenna element 916-m.

Next, reception of a signal will be described. The signals received by the antenna elements 916-1 to 916 -M ₀ are respectively distributed to the N ₀ signals by the distribution couplers 915-1 to 915 -M ₀ , and the respective phase shifters 913-1-1 correspond to the respective N ₀ systems. It is output to the _{~913-N 0} -M 0.
For example the phase shifter _{913-1-1~913-1-M 0} is first directional orientation corresponding to the signal sequence (θ _i, φ _j) on an analog signal to adjust the predetermined complex phase corresponding The adjusted first signal sequence is input to the distribution coupler 912-1. The distribution coupler 912-1 combines these input signals, and inputs the combined signal to the down converter 906-1 via the TDD switch 911.
Similarly, the phase shifters 913 -N _{0-1 to} 913 -N ₀ -M ₀ are provided with a predetermined complex phase corresponding to the direction (θ _{i ′} , φ _{j ′} ) of the directivity corresponding to the Nth signal sequence. the adjustments performed on the analog signal, and inputs the N-th signal sequence after the adjustment to the distributor coupler 912-N _0. Distributor coupler _{912-N 0} synthesizes these input signals, and inputs the combined signal, the down-converter _{906-N 0} via the TDD switch 911.

Downconverter _{906-1~906-N 0} is down-converts the signal of the radio frequency into a baseband signal. The A / D converters 907-1 to 907 -N ₀ convert an analog baseband signal obtained by down-conversion into a digital baseband signal. Here, based on the symbol timing managed by the timing detection circuit (not shown), GI removal & FFT circuit 908-1~908-N ₀ removes a guard interval from the digital baseband signal, the signal in the time axis Convert to frequency axis signal. Postcoder 909, the cross talk component between the signal sequence (stream) which has been processed by the GI removal & FFT circuit _{908-1~908-N 0} and signal separation in the frequency domain, corresponding signals after the crosstalk component separation To demodulators 910-1 to 910-N. The demodulators 910-1 to 910-N reproduce and output data by a predetermined signal detection process.

Here, although low-noise amplifier of the power amplifier and the receiver of the transmission side is not explicitly described, generally downstream of the up-converter 905-1~905-N ₀ (code "A _1" - "A _N0" the power amplifier installed in the position), front (code _{"B 1"} in the down converter _{906-1~906-N 0} ~ installing low-noise amplifier in a position) of the _{"B N0".} The power amplifier and the low-noise amplifier have different amounts of complex phase rotation, and the amount of transfer rotation may be different for each frequency. However, factors to get a difference in phase rotation amount on both transmit and receive between the TDD switch 911 and antenna element 916-1~916-M ₀ is excluded, the symmetry of the channel at the time of reception and time of transmission Will be saved. Therefore, setting the phase rotation quantity of the phase shifter _{913-1-1~913-N 0} -M 0, it is possible to use the same value when it transmits and receives.

The above is the outline of the Massive MIMO technology using hybrid beamforming. Here in the example directional orientation (above corresponding to the phase rotation amount or the signal sequence for setting in phase shifters _{913-1-1~913-N 0} -M 0, the phase shifter 913-1-1～ in _{913-1-M 0 (θ i,} φ j), the phase shifter _{913-N 0 -1~913-N 0} -M 0 (θ i ', φ j')) acquisition method, such as the present invention Although not described because it is not directly related to the features described above, it can be obtained by a conventional technique such as Non-Patent Document 1.

[Expansion of Massive MIMO technology when line of sight is dominant]
In the above description of the Massive MIMO technology, since it is mainly assumed to be used in an access system, it is generally assumed that the environment is a multipath environment. However, even in the access system, if the small cell base station is installed at the top, and it is possible to look down and look roughly, it may be necessary to operate in a non-multipath environment. This is particularly remarkable in the case of a backhaul line, and is assumed to be used in an environment in which a so-called Rice coefficient K is 10 dB or more and a multipath component is only about 1/10 or less of a line-of-sight component. In this case, the difference between the line gain corresponding to the first singular value and the line gain corresponding to the second singular value or more is expected to be 20 dB or more, and spatial multiplex transmission of two or more streams is inefficient. It is expected that

  In such an environment, as shown in Non-Patent Document 2, all antenna elements are divided into a plurality of sets by utilizing the high line gain efficiency corresponding to the first singular value, and a sub-array is set for each set. The configuration is effective. By arranging the sub-arrays spatially separated, it is effective to reduce the correlation between the sub-arrays and to transmit the transmission corresponding to the first singular value in parallel with low correlation. Similarly, in Non-Patent Document 3, when the antenna aperture length of the subarray here is narrow, the line of sight wave is dominant, and the correlation between the antenna elements in the subarray is sufficiently strong, the transmission / reception weight of each antenna element is the frequency. A "time-axis beamforming technique" has been proposed, which can be treated as a constant having no dependence, and in this case, weight multiplication can be performed in units of sampling data on the time axis.

This is a relative value of channel information for each antenna element (a relative value of channel information with respect to channel information of an antenna element serving as a reference, when the element spacing is narrow and the correlation of antenna elements is strong, If the complex phase of the channel information of the k-th frequency component of the element is ψ _ref ^(k) , the frequency dependence of the complex phase of information obtained by multiplying each antenna element by Exp {−j _ref ^(k) } The gender is a method based on the fact that it is almost constant. For example, at the time of reception, the complex phase of the reception weight for synthesizing the received signals of these antenna elements such that the complex phases are the same is substantially constant in the entire frequency band, and Thus, it is possible to use reception weights having the same constant. Generally, a Fourier transform of a function that becomes a constant on the frequency axis becomes a δ function. Therefore, the weight obtained by transforming the reception weight on the frequency axis on the time axis by IFFT only needs to consider the component at t = 0. Become. In other words, since it is not necessary to perform signal processing in consideration of the delay wave component, the time base, which is a predetermined coefficient for each antenna element, is directly added to the sampling data obtained by sampling the analog / baseband received signal by the A / D converter. If the reception weight is multiplied, the directivity can be completely formed only by signal processing on the time axis without converting the reception signal into a signal on the frequency axis by FFT processing or the like.

  The coefficient for rotation of the complex phase to be multiplied as the time axis weight is obtained by the following equations (1) to (3).

In the above equation, S _i (n) represents the sampling data of the n-th sample of the i-th antenna in the received training signal, and S _i (n) ^* represents the complex conjugate of S _i (n). Represent. N _FFT assumes a predetermined periodicity, and indicates a periodicity value that is significant in correlation detection, such as the number of FFT points in OFDM. ψ _j is the amount of rotation of the complex phase (on the receiving side) performed by time axis beamforming. The function angle (x) is a function representing the complex phase of the complex number x, and is a value determined by the ratio of the real part of x to the imaginary part of x and the sign of the real part and the imaginary part. Although the in the correlation calculation in the formula (1) performs a correlation operation over N _FFT samples is the number of FFT points in the case of the OFDM signal as a predetermined periodicity, be an integer multiple of example N _FFT Correlation calculations over other sample numbers may be performed so that periodicity is maintained.

Here, as is clear from the equation (2), the complex phase of the complex coefficient c _j given by the above equation (1) and the complex phase of the time axis weight w _j given by the above equation (2) have inverted signs. It has become something. In this sense, in an embodiment of the present invention to be described later, the complex phase of the complex coefficient c _j given by the equation (1) corresponding to the relative channel information and the complex phase of the time axis weight w _j are found. Are equivalent.

FIG. 12 is a functional block diagram illustrating a configuration example (sub-array separation type) of a wireless station device using time-axis beamforming in the related art described in Non-Patent Document 3. The wireless station device shown in FIG. 1 includes a baseband signal processing circuit 140 and transmission / reception signal processing circuits 929-1 to 929 -N. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 929-n (n = 1,..., N) includes a time axis transmission weight multiplication circuit 921-n, D / A converters 922-n-1 to 922-n-M, and an up converter 923. -N-1 to 923-n-M, down converter (DC) 924-n-1 to 924-n-M, A / D converters 925-n-1 to 925-n-M, and time axis A reception weight multiplication circuit 926-n and a TDD switch (TDD-SW) 927-n are provided. TDD switch 927-n is connected to antenna elements 928-n-1 to 928-n-M. Here, N corresponds to the multiplexing number (the number of streams) when performing spatial multiplexing, and the transmission / reception signal processing circuits 929-1 to 929 -N are mounted in N systems in total. M represents the number of antenna elements of the sub-array mounted on each of the transmission / reception signal processing circuits 929-1 to 929-N. In the description of Figure 11 it had a total number of antenna elements and M _0, but here since the subarray constitute the entire antenna number of antenna elements of each subarray were labeled as different values M.

  Here, the transmission / reception signal processing circuits 929-1 to 929-N (the transmission / reception signal processing circuit 929-n has sub-array antenna elements 928-n-1 to 928-n-M) are described in Non-Patent Document 2. It is assumed that they are installed spatially apart from each other. In addition, the baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 929-1 to 929 -N by wire, and the digital baseband signal is transferred over this wire. As in the case of FIG. 11, an entire control circuit (not shown) is mounted on the baseband signal processing circuit 140 to manage a frame cycle and transmission / reception timing. Switching of 927-N is also managed here.

Furthermore, the time-axis beamforming technology basically presupposes signal processing on the time axis. However, even when a signal on the frequency axis is formed as in the OFDM modulation method, the signal on the frequency axis is formed by FFT processing and IFFT processing. Can be converted into a signal on the time axis, and by performing signal processing on the time axis signal, the time axis beamforming technique can be similarly applied to the OFDM modulation method together with the single carrier transmission. However, assuming the signal processing on the frequency axis in FIG. 11, a GI removal & FFT circuit _{908-1~908-N 0} for IFFT & GI imparting circuit _{903-1~903-N 0} and FFT processing for IFFT processing Has been described separately from the modulators 901-1 to 901-N and the demodulators 910-1 to 910-N. However, since signal processing on the frequency axis is not assumed here, FIG. Even when the OFDM modulation method is used, the functions of the FFT processing and the IFFT processing are included in the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N (or the signal separation circuit 141). Therefore, these notations are omitted. Therefore, irrespective of the OFDM modulation method or single carrier transmission, the input / output signals from the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are signals on the time axis. . Further, the signal separation circuit 141 performs signal separation between the respective signal sequences. Here, it is also possible to perform signal separation on the time axis. Frequency-dependent signal separation may be performed. In this sense, it is assumed that the input to the demodulators 130-1 to 130-N is a signal on the time axis and a signal on the frequency axis. The description will be made assuming that the user inputs "."

  The specific signal flow is as follows. First, signal transmission will be described. The modulators 120-1 to 120-N generate transmission signals of the time base digital baseband of the respective streams to be spatially multiplexed, and input these to the time base transmission weight multiplying circuits 921-1 to 921-N. I do. The time-axis transmission weight multiplying circuits 921-n (n = 1,..., N) are each a sub-array for forming the directivity of the digital signal input from the modulator 120-n by the transmission / reception signal processing circuit 929-n. The signal is converted into a digital signal obtained by multiplying the transmission weights corresponding to the antenna elements 928-n-1 to 928-n-M. The D / A converters 922-n-1 to 922-n-M convert the digital signal multiplied by the transmission weight into an analog baseband signal, and the up-converters 923-n-1 to 923-n-M , Convert the analog baseband signal into a signal in a radio frequency band. At the time of transmission, the TDD switch 927-n connects the upconverter 923-nm (m is an integer of 1 or more and M or less) to the antenna element 928-nm. The signals input from the up-converters 923-n-1 to 923-n-M are transmitted from the antenna elements 928-n-1 to 928-n-M, and the transmission / reception signal processing circuits 929-1 to 929 are transmitted. A directional beam is formed every −N.

  Next, reception of a signal will be described. The signals received by the antenna elements 928-n-1 to 928-n-M (n = 1,..., N) are input to the down converters 924-n-1 to 924-n-M via the TDD switch 927-n. Is done. The down converters 924-n-1 to 924-n-M convert radio frequency signals into analog baseband signals. The A / D converters 925-n-1 to 925-n-M convert an analog baseband signal into a digital baseband signal. This digital baseband signal is input to the time axis reception weight multiplying circuit 926-n. The time axis reception weight multiplying circuit 926-n multiplies each of the input signals by the reception weight corresponding to each of the antenna elements 928-n-1 to 928-n-M, and adds and synthesizes the signals after the reception weight multiplication. And converts them into a single signal sequence. That is, the signals are converted into a total of N signal sequences (streams) by the time axis reception weight multiplying circuits 926-1 to 926 -N, and these signals are input to the signal separating circuit 141. The signal separation circuit 141 performs signal separation by suppressing a crosstalk component between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The demodulators 130-1 to 130-N reproduce and output data by a predetermined signal detection process.

  The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted to a frequency axis signal by FFT processing and then performed on the frequency axis. Alternatively, only the signal processing performed by the transmission / reception signal processing circuits 929-1 to 929 -N may be performed, and the signal separation circuit 141 may not perform any processing. However, in any case, the details of the signal separation method here are omitted because they are not directly related to the present application.

  The time axis transmission weights used in the time axis transmission weight multiplying circuits 921-1 to 921-N and the time axis reception weights used in the time axis reception weight multiplication circuits 926-1 to 926-N are shown here. The time axis transmission / reception weight acquisition means obtains it. Similarly, a control circuit (not shown) manages the value of the time axis transmission / reception weight used there. For example, based on the sampling data obtained by the A / D converters 925-n-1 to 925-n-M for a training signal transmitted by a wireless station as a communication partner, the reference antenna element is transmitted over a predetermined number of samples. A correlation value between the antenna elements (for example, 928-n-1) may be determined, and the correlation value may be determined based on the complex phase. The complex phase values of the time axis reception weight and the time axis transmission weight generally do not match because the amount of rotation of the complex phase of a power amplifier and a low noise amplifier (not shown) is different between individual amplifiers. By using the calibration method of the implicit feedback, it is possible to convert the time axis reception weight into the time axis transmission weight. The transmission / reception weights thus obtained are stored in the memory for each corresponding wireless station device. Then, at the time of transmission and reception, the time axis transmission weight multiplication circuits 921-1 to 921-N and the time axis reception weight multiplication circuits 926-1 to 926-N perform weight multiplication based on the values of these transmission and reception weights. Will be.

  FIG. 13 is a functional block diagram showing a configuration example (shared sub-array type) of a wireless station device using time-axis beam forming in the conventional technology described in Non-Patent Document 3. In the figure, functions common to those in FIG. 12 are given the same figure numbers. In the figure, a radio station apparatus 942 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 929-1 to 929-N, distribution couplers (HYB) 941-1 to 941-M, and an antenna element 928- 1 to 928-M. In FIG. 12, the transmission / reception signal processing circuits 929-1 to 929-N are housed in different housings on the assumption that they are installed in places spatially separated for each sub-array, and the baseband signal processing circuit of another housing is provided. 140 shows a configuration in which a wired connection is made between the wireless LAN and the wireless LAN. On the other hand, in FIG. 13, all of the transmission / reception signal processing circuits 929-1 to 929 -N and the baseband signal processing circuit 140 are configured as a wireless station device 942 in the same housing, and the antenna elements 928-1 to 928 -M are formed as a whole. Is shared by For this reason, for example, at the time of transmission, signals from the TDD switches 927-1 to 927-N of the respective transmission / reception signal processing circuits 929-1 to 929-N are combined by the distribution couplers 941-1 to 941-M and combined. The transmitted signal is transmitted from the antenna elements 928-1 to 928-M. Similarly, at the time of reception, the signals received by the antenna elements 928-1 to 928-M are distributed by the distribution couplers 941-1 to 941-M. That is, the distribution coupler 941-m (m = 1,..., M) distributes the signal received by the antenna element 928-m to the TDD switches 927-1 to 927-N and inputs the signal. All other signal processing is common to FIG. 12 and FIG.

  Similarly, FIG. 14 is a functional block diagram of another configuration example (sub-array shared type) of a wireless station device using time-axis beamforming in the related art described in Non-Patent Document 3. In the figure, the same parts as those of the radio station apparatus shown in FIG. The wireless station device 945 shown in the figure includes a baseband signal processing circuit 140, time-base transmission weight multiplying circuits 921-1 to 921-N, summing and combining units 943-1 to 943-M, and a D / A converter. 922-1 to 922-M, up converters 923-1 to 923-M, down converters 924-1 to 924-M, A / D converters 925-1 to 925-M, and a duplicator 944-1. To 944-M, a time axis reception weight multiplying circuit 926-1 to 926-N, a TDD switch 927, and antenna elements 928-1 to 928-M.

  In FIG. 13, the transmission / reception signal processing circuits 929-1 to 929-N are individually mounted for each sub-array, but the D / A converters 922-n-1 to 922-n-M and the up-converters 923-n-1 to 923-n-1 923-n-M, down converters 924-n-1 to 924-n-M, A / D converters 925-n-1 to 925-n-M, and TDD switches 927-1 to 927-N are shared. It is possible (n = 1,..., N). Therefore, in FIG. 14, an N-system digital baseband signal generated by the time-base transmission weight multiplying circuits 921-1 to 921-N (N planes are mounted as a whole) is added to the adder / synthesizer 943 in sampling data units. 1-943-M, the signals are combined into one system, and D / A converters 922-1 to 922-M convert the digital signals into analog signals. Similarly, on the receiving side, the duplicators 944-1 to 944-M duplicate the digital baseband signals generated by the A / D converters 925-1 to 925-M into N-system signals in sampling data units, The duplicated signal is input to the time axis reception weight multiplying circuits 926-1 to 926-N (N planes are mounted as a whole). Each of the time axis reception weight multiplication circuits 926-1 to 926 -N multiplies the input signal by the reception weight, and adds and combines the results to convert them into one system signal. That is, the signals are converted into a total of N signals by the time axis reception weight multiplying circuits 926-1 to 926 -N, and these signals are input to the signal separating circuit 941.

  Thereby, the D / A converters 922-n-1 to 922-n-M are redundantly mounted, the up-converters 923-n-1 to 923-n-M are redundantly mounted, and the down-converters 924-n-1 to 924-924 are mounted. Avoid duplicate implementation of nM, duplicate implementation of A / D converters 925-n-1 to 925-n-M, duplicate implementation of TDD switches 927-1 to 927-N, reduce circuit scale, reduce power consumption, etc. Has led to the reduction of

  Here, in actual operation, the wireless station device shown in FIG. 12 and the wireless station device shown in FIG. 13 or FIG. For example, the base station apparatus has a degree of freedom in installation as on the roof of a building, and a sub-array can be installed at a plurality of locations. On the other hand, when there is a large restriction on the installation of a building wall or the like, the terminal station apparatus has a configuration in which FIG. 12 is a base station apparatus and FIG. 13 or FIG. 14 is a terminal station apparatus. It is possible to increase the degree of freedom in installation by sharing the antenna. Alternatively, for example, if the transmission capacity per terminal station device does not require spatial multiplexing, FIG. 13 or FIG. 14 shows the base station device, and FIG. 12 shows one of the transmission / reception signal processing circuits 929-1 to 929-N. Assuming that the terminal station apparatus is equipped with only the terminal station apparatus (for example, only the transmission / reception signal processing circuit 929-1 in FIG. 12), multi-user MIMO transmission is performed by a plurality of terminal station apparatuses and one base station apparatus. It is also possible.

Incidentally, 11 and 12, with regard to the correspondence between FIGS. 13 and 14, for example a phase shifter _{913-1-1~913-N} 0 rotation amount of complex phase carried out by -M ₀ in FIG. 11 [psi _alpha ( α is a phase shifter _{913-1-1~913-N} 0 if the corresponding) to the identification number for the -M _0, the time axis transmission weight multiplying circuit 921-1~921-N, the time axis reception weight multiplication circuits 926 -1 to 926-N correspond to multiplying the signal sequence of the corresponding antenna element by Exp {j} _α }. In other words, while was conversion of an analog circuit the signals transmitted and received from each antenna element in Fig. 11 (i.e. the phase shifter _{913-1-1~913-N 0} -M 0), 12, 13 and In FIG. 14, it is equivalent to converting a signal transmitted / received from each antenna element by a digital circuit (that is, a time axis transmission weight multiplication circuit 921-1 to 921-N, a time axis reception weight multiplication circuit 926-1 to 926-N). I do. This has the advantage that the number of A / D converters 907-1 to 907-N ₀ and the number of D / A converters 904-1 to 904-N ₀ can be reduced in FIG. 11, while FIG. 13 and FIG. 14, as described in Non-Patent Document 3, has the advantage that the resolution of directivity formation is increased and that simple and efficient feedback of channel information is possible.

It should be noted that the phase rotation by the phase shifter normally gives phase rotation by selectively passing through a delay line corresponding to the amount of phase rotation on the device. Therefore, when a phase rotation of x is given as an absolute value, a negative phase rotation (delay) of the complex phase rotation amount is performed with a delay corresponding to the phase x as a signal, and code consistency cannot be obtained. . However, in the following description for convenience, will be referred to as "the rotation amount of complex phase carried out in the phase shifter [psi _alpha" when giving a phase rotation equivalent to multiplication of the coefficient Exp {jψ _α} as a signal with the phase shifter I do.

[Calibration technology in channel information feedback]
In general, when directivity is formed using a plurality of antenna elements on the transmission side, it is necessary to feed back MIMO channel information including the techniques described in Non-Patent Documents 1 to 3. At this time, if the number of antenna elements becomes enormous, the amount of channel information to be fed back becomes enormous, so various measures are required. In the Massive MIMO system as described above, in order to acquire channel information of a forward link in a transmission direction, channel information of a reverse link in a reception direction is used, and a complex phase of a reception signal generated by a circuit such as a low noise amplifier used at the time of reception is used. The relationship between the amount of rotation and the amount of rotation of the complex phase of the transmission signal generated by a circuit such as a high power amplifier used at the time of transmission is converted, and the channel information of the reverse link is multiplied by a predetermined calibration coefficient to convert the It is possible to obtain channel information. Generally, these techniques are known as implicit feedback techniques (for example, see Non-Patent Document 4). The details of the calibration process will be described below.

  In an actual wireless communication apparatus, in signal processing on the transmission side, signal amplification is often performed by a high power amplifier immediately before transmission. In this case, the amplification factor may have an error due to the individual difference of the high power amplifier, and the complex phase may rotate in the high power amplifier with a different value for each high power amplifier. Similarly, in signal processing on the receiving side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, the amplification factor has an error due to the individual difference of the low-noise amplifier, and the complex phase in the low-noise amplifier may rotate with a different value for each low-noise amplifier. In some cases, the amplification factor and the amount of phase rotation may have frequency dependency. When the individual difference between the amplification factor and the rotation amount of the complex phase is so large that it cannot be ignored, it is necessary to perform a calibration process when estimating the downlink channel information from the uplink channel information. Since the errors in the amplification factor and the amount of phase rotation are substantially stable in time, the errors in the amplification factor and the amount of phase rotation are measured in advance, and are increased using a coefficient for canceling the influence of the error. The link channel information is converted into downlink channel information. In the following description, the calibration process performed on the base station device side having a plurality of antenna elements will be mainly described. It is an explanation.

  In the above description, the amplitude and the complex phase may be changed by a high-power amplifier or a low-noise amplifier (strictly, a circuit of a transmission system and a reception system including circuits of other filters and the like). In this case, it has been described that the calibration coefficient for performing the correction according to the change in the amplitude or the complex phase is acquired in advance, and this is used for the correction. A known technique may be used for the calibration processing, but an example of the calibration processing will be described below.

  FIG. 15 is a diagram illustrating asymmetry of channel information between the uplink and the downlink. In the figure, reference numerals 955-1 to 955-3 indicate wireless modules, reference numerals 951-1 to 951-3 indicate high power amplifiers (HPAs), and reference numerals 952-1 to 952-3 indicate low noise amplifiers (LNA). Reference numerals 953-1 to 953-3 indicate TDD switches, and reference numerals 954-1 to 954-3 indicate antenna elements.

Here, for the explanation of the calibration technique, only the functions that affect the channel information in the wireless station device are extracted, so that the configurations other than those illustrated are omitted. Therefore, regarding the configuration of the wireless modules 955-1 to 955-3, for convenience, the high power amplifiers 951-1 to 951-3, the low noise amplifiers 952-1 to 952-3, the TDD switches 953-1 to 953-3, and the antenna Although only the elements 954-1 to 954-3 are shown, functions such as an up-converter and a down-converter are mounted in the latter stage (the former stage). Further, for example, it is assumed that a wireless station device having a plurality of antennas has a plurality of wireless modules 955-1 to 955-2 mounted in one housing, and a wireless module 955-3 opposes this. This corresponds to a diagram that extracts and describes a wireless module 955-3 corresponding to one antenna element of a wireless station device that communicates. When the signal passes through each of the high power amplifiers 951-1 to 951-3, the amplitude and the complex phase are Z _{HPA # 1} (f _k ), Z _{HPA # 2} (f _k ), and Z _{HPA # 3} (f _k ) shall change. When the signal passes through each of the low-noise amplifiers 952-1 to 952-3, the amplitude and the complex phase are Z _{LNA # 1} (f _k ), Z _{LNA # 2} (f _k ), and Z _{LNA # 3} (f _k ) Shall change. Here, it is assumed that there is a frequency dependency as a general condition, and the frequency “(f _k )” is described for the k-th frequency component.

Here, for example, channel information when a signal is transmitted from the wireless module 955-1 and the wireless module 955-2 to the test wireless module 955-3 will be described. Here, the antenna element 954-1 of the wireless module 955-1, the channel information in the space between the antenna elements 954-3 of the wireless module 955-3 is represented by _h 1 _{(f k),} the wireless module 955 channel information on the space between the antenna elements 954-2 and antenna element 954-3 of the wireless module 955-3 -2 is represented by _h 2 _{(f k).}

At this time, the channel information when the signal is actually transmitted from the wireless module 955-1 to the wireless module 955-3 indicates a change in the space h ₁ (f _k ) accompanying the passage of the high power amplifier 951-1. It is observed as a value multiplied by a coefficient Z _{HPA # 1} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying the passage through the low noise amplifier 952-3. Similarly, channel information when a signal is transmitted from the wireless module 955-2 to the wireless module 955-3 includes a coefficient Z indicating a change associated with the passage of the high power amplifier 951-2 in h ₂ (f _k ) in space. It is observed as a value multiplied by _{HPA # 2} ( _fk ) and a coefficient ZLNA _{# 3} ( _fk ) indicating a change accompanying passage through the low noise amplifier 952-3. Therefore, a channel from the wireless module 955-1 to the wireless module 955-3 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 3} (f _k ). The channel from the wireless module 955-2 to the wireless module 955-3 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 3} (f _k ). Therefore, between the wireless module 955-1 and the wireless module 955-2, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), Z _{HPA # 2} (f _k ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same on the receiving side. When the signal transmitted from the wireless module 955-3 is received by the wireless module 955-1, the channel information is stored in the high power amplifier 951 in h ₁ (f _k ) in space. Observed as a value obtained by multiplying a coefficient Z _{HPA # 3} (f _k ) indicating a change accompanying passage of -3 and a coefficient Z _{LNA # 1} (f _k ) indicating a change accompanying passage of the low noise amplifier 952-1. Is done. Similarly, when a signal transmitted from the wireless module 955-3 is received by the wireless module 955-2, the channel information indicates a change in h ₂ (f _k ) in space due to the passage of the high power amplifier 951-3. The coefficient Z _{HPA # 3} (f _k ) is multiplied by a coefficient Z _{LNA # 2} (f _k ) indicating a change accompanying the passage through the low noise amplifier 952-2. Therefore, a channel from the wireless module 955-3 to the wireless module 955-1 is represented by Z _{HPA # 3} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). A channel from the wireless module 955-3 to the wireless module 955-2 is represented by Z _{HPA # 3} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). Therefore, between the wireless module 955-1 and the wireless module 955-2, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), Z _{LNA # 2} (f _k) ) / Z _{LNA # 1} (f _k ).

To summarize Here again, the channel information when receiving the signal transmitted from the wireless module 955-3 corresponding to the reverse link on the left side of the radio station apparatus by the wireless module _{955-1 h 1 (f k) ·} Z _{HPA # 3} (f _k ) · Z _{LNA # 1} (f _k ). When the signal transmitted from the wireless module 955-1 corresponding to the forward link is received by the wireless module 955-3, the channel information is h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA # 3} (f _k ). The calibration coefficient of the wireless module 955-1 is given by the following equation (4).

Similarly, the channel information when receiving the signal transmitted from the wireless module 955-3 corresponding to the reverse link on the left side of the radio station apparatus by the wireless module _{955-2 h 1 (f k) ·} Z HPA # 3 (F _k ) · Z _{LNA # 2} (f _k ). When the signal transmitted from the wireless module 955-2 corresponding to the forward link is received by the wireless module 955-3, the channel information is h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA # 3} (f _k ). The calibration coefficient of the wireless module 955-2 is given by the following equation (5).

Here, for example, each channel information in the reverse link to be acquired by the wireless module _{955-1~955-2 h 1 (f k) ·} Z HPA # 3 (f k) · Z LNA # 1 (f k) and h ₁ (f _k ) · Z _{HPA # 3} (f _k ) · Z _{LNA # 2} (f _k ). When this is multiplied by the calibration coefficients C ₁ (f _k ) and C ₂ (f _k ), h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA # 3} (f _k ) and h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA # 3} (f _k ) .

  At the time of actual operation, when the wireless module of the actual communication partner is different from the wireless module 955-3, strictly speaking, the estimated value of the channel information of the forward link obtained by multiplying this calibration coefficient is Of the channel information itself. However, even in this case, the value obtained by multiplying the channel information of the true forward link regarding the wireless modules 955-1 and 955-2 by the common coefficient and the above-described estimated value match, and the directivity formation is performed. Considering that there is no effect even if all antenna elements are multiplied by a common constant, there is no problem as feedback of channel information.

  Further, in the above description, the transmitting side of the target wireless station apparatus is described as the forward link, and the receiving side is described as the reverse link. However, when the target wireless station apparatus is the base station apparatus, the forward link is usually Is called a downlink, and the reverse link is called an uplink. Similarly, when the radio station apparatus of interest is a terminal station apparatus or a relay station apparatus, usually, the forward link is called an uplink, and the reverse link is called a downlink.

Satoshi Suyama, Tatsunori Ohara, Kiyun Shin, Yukihiko Okumura, "Effect of Analog Beamforming Configuration in Massive MIMO Using High Frequency Band Hybrid Beamforming", IEICE Technical Report, IEICE, March 2015 , Vol.114, no.490, RCS2014-337, p.213-218 Atsushi Ota, Takuto Arai, Hiroshi Shirato, Kazuki Maruta, Tatsuhiko Iwakuni, Masataka Iizuka, "Spatial multiplex transmission technology for millimeter-wave band in the forward-looking environment using parallel transmission of the first eigenmode-scheme proposal and basic performance evaluation results," IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, August 2015, vol.115, no.181, RCS2015-144, p.73-78 Atsushi Ota, Hiroshi Shirato, Kazuki Maruta, Takuto Arai, Tatsuhiko Iwakuni, Masataka Iizuka, "Efficient Use of First Eigenmode Transmission in Massive MIMO with Line-of-Sight Environment", IEICE Technical Report, IEICE, 2015 November, vol.115, no.288, RCS2015-239, p.293-298 Hayato Fukuzono, Tomonori Murakami, Riichi Kudo, Yasushi Takatori, Masato Mizoguchi, "Experimental evaluation of implicit feedback in downlink multi-user MIMO-OFDM system", IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, November 2013 , Vol.113, no.301, RCS2013-187, p.79-84

  In the wireless transmission system using the high frequency band such as the millimeter wave band described above, there are problems regarding the A / D converter and the D / A converter, and the up-converter and the down-converter as described below. I do.

  In the time-axis beamforming technology described in the above “Extension of Massive MIMO technology when line-of-sight waves are dominant”, the number of A / D converters, D / A converters, upconverters, and downconverters is equal to the number of antenna elements. Will be needed. Further, in the configuration example shown in FIG. 13, these are required by the number of antenna elements × the number of streams. In general, the power consumption of A / D converters and D / A converters used in an environment with a very high clock speed due to the wide band tends to increase, and as a result, the heat generation also becomes a non-negligible level. In the case of a device that generates a large amount of heat, a heat radiating plate or the like for radiating the heat is required, and these devices physically need to have a size proportional to the amount of the generated heat. Also, it is difficult to reduce the size of the A / D converter and the D / A converter, which constantly consume power, due to the necessity of heat radiation processing for the heat generation expected during operation in addition to the design requirements of the circuit itself. And become larger.

  Originally, in a system that assumes use of the millimeter wave band, the wavelength of the antenna element itself becomes shorter as the frequency increases, and the size of the antenna element itself can be reduced in a form proportional to this wavelength. For example, if the size of the antenna element can be designed to be sufficiently small with respect to the wavelength, and the antenna elements are arranged at about 1/2 wavelength, the size of the entire array antenna should be designed to be small even if the number of antenna elements is large. Becomes possible. For example, if the center frequency is 80 GHz assuming the E band, one wavelength is 3.75 mm. If an array antenna of 16 × 16 = 256 elements is formed with a wavelength array and a square array structure, the size of the entire array antenna will fall within a size of about 6 cm × 6 cm. However, in order to realize mounting in this size, the A / D converter and the D / A converter themselves are also configured to fit within 3.75 mm square, which is one wavelength, and the amount of heat generation expected during operation is reduced. A heat sink for heat dissipation must also be accommodated in this. If this is not possible, the actual size of the array antenna will depend on the size of the A / D converter and the D / A converter, and the size of the heat sink for heat dissipation. For example, if a wideband signal having a bandwidth of 1 GHz is handled, it is necessary to perform the sampling process performed in the baseband at a super speed of at least 1 GHz. The higher the speed, the higher the power consumption and the larger the amount of heat generated, and ultimately cannot achieve miniaturization. As described above, the size cannot be reduced due to the large power consumption, despite the use of the Massive MIMO technology which can be reduced in size due to the high frequency band. For these reasons, reducing the power consumed by the A / D converter and the D / A converter is a major issue.

  Further, in addition to the problem of power consumption, A / D converters and D / A converters that operate at high speed are expensive. It is required to reduce the total number of / D converters and D / A converters. The same applies to the up converter and the down converter, and mounting the up converter and the down converter for the number of antenna elements leads to an increase in cost. In addition, local oscillators input to the up-converter and the down-converter need to eliminate the uncertainty of the complex phase in order to form directivity. Therefore, it is required to share local signals. The loss due to distributing an enormous number of these high-frequency signals for each antenna element and the influence of mutual interference from these many signal lines to the signal lines of each system cannot be ignored. In particular, to avoid mutual interference between the systems, it is ideal to separate the signals of the respective systems from each other at a physical distance. It leads to conversion.

  Therefore, in order to make the array antenna of each wireless station device smaller, even when utilizing time-axis beamforming, it is consumed by the A / D converter and the D / A converter mounted as much as possible. Power needs to be reduced.

  In view of the above circumstances, an object of the present invention is to provide a wireless communication device and a wireless communication method that can be reduced in size and cost.

  One embodiment of the present invention is a wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device. The signal transmitted by the wireless communication device, a signal receiving unit that receives via each of the antenna elements used for signal reception, provided at least for each array antenna used for signal reception, of the signal received by the signal receiving unit Among them, for each of the reception signals received via the antenna elements constituting the corresponding array antenna, a first phase rotation unit for rotating and outputting a complex phase on an analog signal, and at least a signal used for signal reception. The antenna element is provided for each array antenna and determines whether or not to pass the reception signal output from the corresponding first phase rotation unit. A switching unit that can be switched for each unit, and when the reception signal is a reproduction target, passes all or a part of the reception signal output from the corresponding first phase rotation unit, and the reception signal When the training signal is for calculating the rotation amount of the complex phase, the switching unit switches the reception signal output from the corresponding first phase rotation unit one by one for each antenna element system. A switching control unit for controlling the array antenna, at least for each of the array antennas used for signal reception, a signal combining unit for combining the received signals passed by the corresponding switching unit, and at least for each of the array antennas used for signal reception A signal conversion unit that converts the received signal synthesized by the corresponding signal synthesis unit from an analog signal to a baseband digital signal. A signal reproducing unit that reproduces a signal transmitted by the other wireless communication device based on the received signal to be reproduced converted by the signal converting unit; and the switching unit passes one by one for each system of the antenna element. A correlation calculating unit that calculates a correlation for each combination of a reference antenna element among the antenna elements and another antenna element of the training signal using the training signal, based on a calculation result of the correlation by the correlation calculating unit. A first rotation amount calculation unit for calculating a rotation amount of a complex phase to be given in the first phase rotation unit.

  One aspect of the present invention is the above-described wireless communication device, wherein the transmission signal generation unit that generates an analog transmission signal to be transmitted to another wireless communication device using the array antenna, A signal distribution unit that is provided for each array antenna and branches the analog transmission signal transmitted using the corresponding array antenna in accordance with each of the antenna elements configuring the array antenna, and is used at least for signal transmission. A second phase rotation unit provided for each of the array antennas and rotating a complex phase on an analog signal with respect to each of the transmission signals branched by the corresponding signal distribution unit; and the second phase rotation unit The antenna element that constitutes the array antenna corresponding to the second phase rotation unit by transmitting the transmission signal whose phase has been rotated. A signal transmitting unit that transmits the signal through the second unit, a calculation result of the correlation calculated by the correlation calculation unit, a calculation result of the rotation amount of the complex phase calculated by the first rotation amount calculation unit, or a calibration process on the calculation result. And a second rotation amount calculation unit that calculates the rotation amount of the complex phase to be given in the second phase rotation unit based on any of the information on the rotation amount of the complex phase obtained by performing .

  One embodiment of the present invention is the above wireless communication device, wherein the processing of the first phase rotation unit and the second phase rotation are performed in place of the first phase rotation unit and the second phase rotation unit. A phase rotation unit for performing the processing of the rotation unit is provided for each of the array antennas.

  One embodiment of the present invention is the above-described wireless communication device, wherein, instead of the signal combining unit and the signal distributing unit, a signal combining and distributing unit that performs processing of the signal combining unit and processing of the signal distributing unit is provided. A switching unit is provided for each of the array antennas, and a switching unit that temporally switches between processing of the signal combining unit and processing of the signal distributing unit in the signal combining and distributing unit is further provided for each of the array antennas.

  One embodiment of the present invention is the above-described wireless communication device, wherein the signal synthesis unit and the signal distribution unit are individually independent, and the first phase rotation unit and the second phase rotation unit Are independent.

  One aspect of the present invention is a wireless communication method executed by a wireless communication device that forms directivity using one or more array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device A signal receiving unit is provided for each of the array antennas used for signal reception, and a signal receiving step of receiving a signal transmitted by the other wireless communication device via each of the antenna elements used for signal reception. The phase rotator is configured to perform a complex phase on an analog signal for each of the antenna element systems with respect to each of the received signals received through the antenna elements constituting the corresponding array antenna among the signals received in the signal receiving process. And a switching unit provided for each of the array antennas used for signal reception at least. If it is a reproduction target, all or a part of the reception signal whose complex phase has been rotated in the phase rotation process by the corresponding phase rotation unit is passed, and the reception signal is used for calculating the rotation amount of the complex phase. When the training signal, the switching process of passing the reception signal rotated the complex phase in the phase rotation process by the corresponding phase rotation unit one by one for each antenna element system, and at least the signal used for signal reception A signal combining section provided for each array antenna, a signal combining step of combining the received signals passed in the switching step by the corresponding switching section, and a signal converting section provided at least for each array antenna used for signal reception Converts the received signal synthesized in the signal synthesis process by the corresponding signal synthesis unit into an analog signal. A signal conversion step of converting a signal into a baseband digital signal, and a signal transmitted by the other wireless communication device based on the received signal to be reproduced converted to a digital signal in the signal conversion step. A signal reproducing step of reproducing the reference signal and a reference antenna element among the antenna elements of the training signal using the training signal passed one by one for each antenna element in the switching step. A correlation calculation step of calculating a correlation for each combination with the antenna element, and a rotation amount calculation unit calculates a rotation amount of a complex phase to be given in the phase rotation unit based on a calculation result of the correlation in the correlation calculation step. And calculating the amount of rotation.

  According to the present invention, it is possible to reduce the size and cost of a wireless communication device.

FIG. 2 is a functional block diagram illustrating a configuration example of a wireless station device according to the first embodiment of the present invention. FIG. 3 is a functional block diagram illustrating a configuration example of a wireless station device according to the embodiment. FIG. 2 is a diagram illustrating a configuration example of a communication system according to the first embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 9 is a functional block diagram illustrating a configuration example of a wireless station device according to a second embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a wireless station device according to the embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 3 is a functional block diagram illustrating a configuration example of a transmission / reception signal processing circuit in the same embodiment. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 11 is a functional block diagram illustrating a configuration example of a wireless station device according to the related art. FIG. 7 is a diagram illustrating a calibration process in the related art.

  An embodiment of the present invention will be described in detail with reference to the drawings. First, the basic principle of the embodiment of the present invention will be described. The terms “time axis” and “frequency axis” used in this specification may be expressed as “time domain” and “frequency domain”. Give an explanation.

[Overview of basic principles]
In the time axis beamforming technique, for example, at the time of reception, a main path of an incoming wave between a transmitting and receiving station is extracted, and signal processing for directing the directional gain of the antenna element group in the direction is performed. The transmission and reception weights used at this time are constants having no frequency dependence, and as a result, signal processing is reduced at various points. However, since it is based on digital signal processing, transmission / reception weights are multiplied on individual digital signal processing for each antenna element, and associated A / D (analog / digital) conversion is performed for each antenna system. And a D / A (digital / analog) converter.

  However, multiplication processing of coefficients having no frequency dependence does not necessarily require digital signal processing if it is limited to, for example, only rotation processing of a complex phase without a change in amplitude. Specifically, by using a phase shifter that is an analog circuit and passing the analog signal through this phase shifter set to a predetermined complex phase rotation, signal processing substantially equivalent to weight multiplication processing is performed. Can be realized. In Non-Patent Document 1, directivity control is realized using a phase shifter. This is performed, for example, in a predetermined direction set in horizontal / vertical directions at intervals of 5 degrees. A combination set is determined in advance, and the phase rotation amount of each phase shifter is set according to the direction in which the beam obtained by some control procedure should be directed. However, this combination set of phase rotation amounts is selected from a preset menu, and it is necessary to search for the direction in which the reception level is the highest while individually transmitting a training signal for each direction. . However, in time-axis beamforming, the amount of phase rotation of each antenna element is optimized based on the training signal transmitted from the terminal station apparatus side, so that the calculation of the amount of phase rotation used for forming directivity is simply fed back. In addition to this, the combination set of the amount of rotation of the complex phase can be set with a remarkably high degree of freedom without requiring a menu or the like in advance.

  Therefore, the wireless station device (wireless communication device) according to the embodiment of the present invention employs digital assist type analog beamforming. That is, the wireless station device performs the calculation of the amount of rotation of the complex phase performed by each phase shifter by digital signal processing, and controls the phase shifter using the complex phase value obtained by the digital signal processing. By doing so, the desired complex phase is rotated to form directivity on the analog signal.

Here, the process of calculating the complex phase of the transmission / reception weight of time axis beam forming performed in digital signal processing need not necessarily be performed simultaneously by each antenna element. Alternatively, even when digital signal processing is performed when calculating the complex phase of the transmission / reception weight of time axis beamforming, the time rate at which signal processing for this is performed is the same as that of normal communication for transmitting and receiving user data. It is expected to be overwhelmingly less than the hourly rate. Therefore, the wireless station device of the present embodiment performs the operations of these digital circuits in a limited time.
Hereinafter, a detailed embodiment to which the basic principle is applied will be described with reference to the drawings.

[First Embodiment]
FIG. 1 is a functional block diagram illustrating a configuration example (sub-array separated type) of a wireless station device 450 according to the present embodiment. In the figure, the same parts as those of the radio station apparatus according to the prior art shown in FIGS. 12 to 14 are denoted by the same reference numerals. In this embodiment, as shown in FIGS. 12 and 13 of the related art, a “sub-array separation type” configuration in which a directional beam is divided into a plurality of sub-arrays and a plurality of directional beams are formed in one array There is a variation of the configuration by the "sub-array shared type" (strictly speaking, it is not divided into sub-arrays, so it may be understood as "integrated array"). Give an explanation.

  The wireless station device 450 shown in FIG. 7 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N (N is an integer of 1 or more), and a control circuit 460. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 451-n (n = 1,..., N) includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, and an A / A converter. D converter 125-n, TDD (Time Division Duplex) switch (TDD-SW) 127-n, and phase shifters 402-n-1 to 402-n-M (M is 2 or more) Integer), switches 403-n-1 to 403-n-M, distribution coupler (HYB) 404-n, correlation calculation circuit 405-n, phase shift control circuit 406-n, and down converter 424-n. n-1 to 424-nM and A / D converters 425-n-1 to 425-nM. Phase shifters 402-n-1 to 402-nM are connected to antenna elements 401-n-1 to 401-nM, respectively. The D / A converters 122-1 to 122-N, the up converters 123-1 to 123-N, the down converters 124-1 to 124-N, and the A / D converters 125-1 to 125-N are shown in FIG. 12 and D / A converters 922-1-1 to 922-NM shown in FIG. 13, up-converters 923-1-1 to 923-NM, and down-converters 924-1-1 to 924-NM. , A / D converters 925-1-1 to 925-NM can be used.

  Here, in the down converters 424-1-1 to 424-NM, a signal from a local oscillator is required to perform frequency conversion between a radio frequency signal and a baseband signal. That is, a common local signal is used for the combination of the down converters 424-n-1 to 424-n-M in the same transmission / reception signal processing circuit 451-n (n = 1,..., N), and each down converter 424-n-M is used. It is necessary to suppress a temporal change in the relative relationship between the complex phases in n-1 to 424-n-M. In this sense, a configuration in which a local oscillator exists outside the down converters 424-1-1 to 424-NM is practically used. The description of the oscillator is omitted. The local oscillator used must be shared among the down converters 424-n-1 to 424-n-M of the same n, but the down converters 424-n'-1 to 424-n having different values of n. The local oscillator used in combination with n'-M does not need to be shared. Further, these local oscillators do not need to be shared between the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. To the last, it is only important to share a local signal between signal sequences that cooperate in forming directivity.

Here, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the wireless station device 450 includes a total of N transmission / reception signal processing circuits 451-1 to 451-N for N systems. M represents the number of antenna elements of the sub-array mounted on each of the transmission / reception signal processing circuits 451-1 to 451-N. In the description of Figure 11 it had a number of all the antenna elements and M _0, but because here as in the case of FIG. 12 is a subarray constitute the entire antenna number of antenna elements of each subarray were labeled as different values M. Each of the transmission / reception signal processing circuits 451-n is accompanied by sub-array antenna elements 401-n-1 to 401-n-M, and the transmission / reception signal processing circuits 451-1 to 451-N are described in Non-Patent Document 2. It is assumed that they are installed spatially separated as described above. Further, the baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 451-1 to 451-N by wire, and the digital baseband signal is transferred over this wire. Although not shown in FIG. 11, the wireless station device 450 includes an overall control circuit 460. FIG. 3 shows an example in which the wireless station device 450 mounts the control circuit 460 on the baseband signal processing circuit 140. The control circuit 460 manages a frame cycle and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N. The control circuit 460 determines whether to connect the distribution coupler 404-n to the up converter 123-n or the down converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by splitting.

  Furthermore, in the present embodiment, the process of estimating the amount of complex phase rotation performed by the variable phase shifter basically corresponding to the transmission / reception weight is performed by digital signal processing, and the actual complex phase rotation process is realized by analog signal processing. . For this reason, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are premised on signal processing on the frequency axis like the OFDM (Orthogonal Frequency Division Multiplexing) modulation method. In either case, even if signal processing on the time axis is assumed, such as SC-FDE (Single-Carrier Frequency Domain Equalization), it is possible to cope with either method. However, the signal separation circuit 141 performs signal separation between each signal sequence. Here, it is also possible to perform signal separation on the time axis. Frequency-dependent signal separation may be performed. In this sense, the input to the demodulators 130-1 to 130-N is assumed to be a signal on the time axis or a signal on the frequency axis. The description will be made assuming that the user inputs "." As described above, the concept regarding the variation of the communication scheme such as the OFDM modulation scheme and the SC-FDE is the same in the following description.

The specific signal flow is as follows.
First, signal transmission will be described. In the wireless station device 450, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmit signals while the up-converter 123-n and the distribution coupler 404-n are connected (n = 1,..., N).

  Each of the modulators 120-1 to 120-N generates a transmission signal of a time base digital baseband of each stream to be subjected to spatial multiplexing, and inputs the signal to the transmission / reception signal processing circuits 451-1 to 451-N. The transmission / reception signal processing circuit 451-n (n = 1,..., N) receives the transmission signal of the time base digital baseband generated by the modulator 120-n. The D / A converter 122-n of the transmission / reception signal processing circuit 451-n (n = 1,..., N) converts the transmission signal input from the modulator 120-n into an analog baseband signal, and Input to converter 123-n. The up-converter 123-n converts the signal input from the D / A converter 122-n from a baseband signal to a signal in a radio frequency band, and inputs the signal to the TDD switch 127-n. The TDD switch 127-n inputs the signal input from the up converter 123-n to the distribution coupler 404-n.

  The distribution coupler 404-n branches the analog signal input from the TDD switch 127-n into M-system analog signals. The branched analog signals are input to phase shifters 402-n-1 to 402-n-M via switches 403-n-1 to 403-n-M. The phase shifters 402-n-1 to 402-n-M apply a predetermined complex phase rotation on the analog signal. The analog signals subjected to the complex phase rotation by the phase shifters 402-n-1 to 402-n-M are transmitted via the antenna elements 401-n-1 to 401-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 402-n-1 to 402-n-M. For example, the antenna elements 401-1-1 to 401-1-M and the antenna elements 401-N-1 to 401-NM have individual directivities formed, and a radio station device in the directivity direction is provided. Perform communication.

  Next, reception of a signal will be described. In the wireless station apparatus 450, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receive signals while the distribution coupler 404-n and the down-converter 124-n are connected (n = 1,..., N).

  The signals received by the antenna elements 401-n-1 to 401-nM (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-nM, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on an analog signal to the input signal, and outputs the result through switches 403-n-1 to 403-n-M. And input to the distribution coupler 404-n. The distribution coupler 404-n combines signals of the respective antenna systems input via the switches 403-n-1 to 403-n-M on an analog signal, and combines the combined signals via the TDD switch 127-n. And input to the down-converter 124-n. The down-converter 124-n converts a radio frequency signal input via the TDD switch 127-n into an analog baseband signal, and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the signal input from the downconverter 124-n from an analog baseband signal to a digital baseband signal, and inputs the digital baseband signal to the signal separation circuit 141.

  The signal separation circuit 141 performs signal separation by suppressing a crosstalk component between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted to a frequency axis signal by FFT processing and then performed on the frequency axis. When suppressing the crosstalk component on the time axis, first, the correlation between the signal sequences input to the signal separation circuit 141 is calculated based on the digital baseband signal corresponding to the received training signal. . Then, based on a MIMO (Multiple Input Multiple Output) channel matrix given by the calculated correlation, general MIMO signal separation processing such as ZF (Zero Forcing) type or MMSE (Maximum Mean Square Error) type signal separation. A matrix similar to the above may be calculated, and the matrix may be multiplied in units of sampling data by regarding the signal sequence input to the signal separation circuit 141 as a vector. This is because, in general MIMO signal separation processing such as ZF-type or MMSE-type signal separation on the frequency axis, different matrices are used for each frequency component, but almost the same matrix on the frequency axis In the case where is used, it corresponds to the fact that signal separation processing can be performed on the time axis in units of sampling data. Alternatively, only the signal processing performed by the transmission / reception signal processing circuits 451-1 to 451-N may be performed, and the signal separation circuit 141 may not perform any processing. However, in any case, the details of the signal separation method here are omitted because they are not directly related to the features of the present embodiment. The demodulators 130-1 to 130-N demodulate the signal in which the crosstalk component has been suppressed in the signal separation circuit 141.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402-NM will be described. In this signal processing, the switches 403-n-1 to 403-n-M (n = 1,..., N) are connected to the phase shifters 402-n-1 to 402-n-M and the down converters 424-n-1 to 424-n-1. 424-nM. The switching of these switches is managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. Note that the phase shifters 402-n-1 to 402-n-M (n = 1,..., N) are connected to the distribution coupler 404-n except when calculating the rotation amount of the complex phase. In addition, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402-NM are set to predetermined values. The rotation amount of the complex phase obtained in the subsequent processing is set as a difference from an initial predetermined value. For example, in the simplest example, all of the phase shifters 402-1-1 to 402-NM may be set to zero (or the same value), and in this case, the rotation amount of the obtained complex phase May be used as it is as the phase rotation amount of the phase shifters 402-1-1 to 402-NM during subsequent communication. Alternatively, the initial predetermined values of the phase shifters 402-1-1 to 402-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the rotation amount of the complex phase is + α degrees, If + β degrees, + γ degrees,. , + (Γ + 30) degrees,...

  As the actual processing, first, the wireless station device of the communication partner from which the amount of rotation of the complex phase is to be acquired transmits a training signal for channel estimation, and the wireless station device 450 receives this training signal. The signals received by the antenna elements 401-n-1 to 401-nM (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-nM, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on an analog signal to the input signal, and outputs the result through switches 403-n-1 to 403-n-M. Input to the down converters 424-n-1 to 424-n-M. The down-converters 424-n-1 to 424-n-M convert the input radio frequency signals into analog baseband signals, respectively. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal and outputs the signal to the correlation calculation circuit 405-n.

  The correlation calculation circuits 405-1 to 405-N calculate the amount of rotation of the complex phase using equations (1) to (3), respectively. When the calibration processing is necessary as needed, the correlation calculation circuits 405-1 to 405-N determine the complex phase on the transmission side as a value in consideration of the calibration coefficient in the equations (1) to (3). Determine the amount of rotation. The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n (n = 1,..., N) is the identification number of the wireless station device with which to communicate (in the case of performing communication with a plurality of wireless station devices. An identification number is not required when communicating with one wireless station device in a PP (point-to-point) type.) And input to a phase shift control circuit 406-n (n = 1,..., N). Is done. The phase shift control circuit 406-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

  Note that when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 460 grasps the wireless station apparatus to be the communication partner, and the phase shift control circuit 406-n (n = 1,...). , N) to instruct the phase shifters 402-n-1 to 402-n-M to set the rotation amount of the complex phase corresponding to the wireless station device that performs communication. The phase shift control circuit 406-n obtains the amount of rotation of the complex phase corresponding to the wireless station device that performs communication, for example, by reading out from the memory, and acquires the amount of rotation of the complex phase from the phase shifters 402-n-1 to 402. -N-M is set to realize beamforming on analog.

Although not explicitly shown in the figure, if a transmitting-side high-power amplifier (HPA) is disposed, for example, the transmitting-side high-power amplifier (HPA) is disposed at a point described as “A” in the drawing, and the receiving-side low-noise amplifier (HPA) is disposed. if placing LNA) or the like, arranged at a point which is described as "B" and "C _1-1" - "C _N-M" in FIG. Regarding the points described as “A” and “B”, since the same transmission / reception signal processing circuits 451-1 to 451-N are common, the complex phases of the individual high power amplifiers and low noise amplifiers are uncertain. Calibration processing for removing the characteristic is unnecessary.

On the other hand, regarding the low-noise amplifiers at the points described as “C _1-1 ” to “C _N-M ”, when the rotation amount of the complex phase can vary with time, the same transmission / reception signal processing circuit 451 is used. 1 to 451-N, which can cause the uncertainty of the complex phase among the respective antenna elements 401-1-1 to 401-N-M, similarly to the prior art implicit feedback calibration method. It is necessary to remove the uncertainty of the complex phase of the low-noise amplifier of each system. Note that the present embodiment can be applied to any method, and the specific method of the calibration process is not limited. Considering this calibration result, for example, assume that the rotation amount of the complex phase at each of “C _1-1 ”, “C _1-2 ”, and “C _1-3 ” is +10 degrees, +20 degrees, and +30 degrees. -10, -20, and -30 degrees are respectively corrected for the complex phase rotation amounts obtained by Equations (1) to (3) to adjust the phase rotation amounts. The information on the calibration result is collected by a calibration circuit (not shown), and the information is collected by the phase shift control circuits 406-1 to 406-N or the correlation calculation circuits 405-1 to 405-N. And perform correction.

  In addition, since the transmission / reception signal processing circuits 451-1 to 451-N having a sub-array configuration can be installed physically apart from each other to reduce the correlation between directional beams formed on analog, generally, Are set apart from each other by a predetermined distance or more.

  Further, the same applies to all the following descriptions (including other embodiments), but in FIG. 1, the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N are connected by wire. In this configuration, digital baseband signals are transmitted and received over this wire. Is implemented, a signal flowing on a wired connection between the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N can be an analog baseband signal.

  Next, similarly to FIG. 13 with respect to FIG. 12 in the related art, the wireless station apparatus of the present embodiment can be configured by a “shared sub-array type” in which a plurality of directional beam formations are realized by one array antenna. is there. This configuration is shown in FIG.

  FIG. 2 is a functional block diagram illustrating a configuration example (shared sub-array type) of the wireless station device 452 in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

  The wireless station device 452 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N, distribution couplers (HYB) 407-1 to 407-M, and an antenna element 401-. 1 to 401 -M, and a control circuit 460. The distribution coupler (HYB) 407-m (m = 1,..., M) includes phase shifters 402-1-m, 402-N of the transmission / reception signal processing circuits 451-1, 451-2,. , 402-N-m, and the antenna element 401-m (m = 1,..., M).

  Also in the wireless station device 452, similarly to the wireless station device 450 shown in FIG. 1, the down converters 424-1-1 to 424-NM perform frequency conversion between a radio frequency signal and a baseband signal. Therefore, it is necessary to input a signal from a local oscillator. A common local signal is used for each combination of the down converters 424-n-1 to 424-n-M (n = 1,..., N) in each of the transmission / reception signal processing circuits 451-1 to 451-N. It is necessary to suppress a temporal change in the relative relationship between the complex phases in the down converter. In this sense, a configuration in which a local oscillator exists substantially outside the transmission / reception signal processing circuits 451-1 to 451-N is adopted. However, since the description is complicated, a simple description of the external local oscillator is used here. The description is omitted. It is not necessary to share the local oscillator used in each of the transmission / reception signal processing circuits 451-1 to 451-N. Further, it is not necessary to share these local oscillators with the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. To the last, it is only important to share a local signal between signal sequences that cooperate in forming directivity. However, unlike the case of FIG. 1, the down converters 424-1-1 to 424-NM are all housed in the same housing, so that all the local signals can be shared in the case of FIG. Is possible.

  Similarly to the above description, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and M represents the number of antenna elements of the array antenna that is shared. In the wireless station device 450 shown in FIG. 1, the transmission / reception signal processing circuits 451-1 to 451-N are mounted in physically different housings for each system. On the other hand, in the wireless station device 452 shown in FIG. 2, all the N systems of the transmission / reception signal processing circuits 451-1 to 451-N are mounted in one housing. Further, in the wireless station device 452, the transmission / reception signal processing circuits 451-1 to 451-N share the antenna elements 401-1 to 401-M via the distribution couplers 407-1 to 407-M. Here, the internal processing in the transmission / reception signal processing circuits 451-1 to 451-N is the same between the wireless station device 450 shown in FIG. 1 and the wireless station device 452 shown in FIG. FIG. 2 shows a case where the control circuit 460 is mounted on the baseband signal processing circuit 140; however, the control circuit 460 can be mounted at an arbitrary place in the wireless station device 452. The control circuit 460 manages a frame cycle, transmission / reception timing, and switching of the TDD switches 127-1 to 127 -N.

The following shows a specific signal flow in the wireless station device 452, focusing on the difference from the wireless station device 450 shown in FIG.
First, signal transmission will be described. In the wireless station device 452, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmit signals while the up-converter 123-n and the distribution coupler 404-n are connected (n = 1,..., N). This point is common to the wireless station device 450 shown in FIG.

  Each of the modulators 120-1 to 120-N generates a transmission signal of a time base digital baseband of each stream to be spatially multiplexed, and inputs each of the signals to the transmission / reception signal processing circuits 451-1 to 451-N. By a process similar to that of the radio station device 450 shown in FIG. 1, radio frequency analog signals subjected to a process for forming directivity are output from the phase shifters 402-1-1 to 402-NM. The phase shifters 402-1-1 to 402-NM transmit these signals of each system to distribution couplers 407-1 to 407-M connected to the corresponding antenna elements 401-1 to 401-M. input. That is, the phase shifters 402-1-m, 402-2-m,..., 402-Nm (m = 1,..., M) input signals to the distribution coupler 407-m. Distribution couplers 407-1 to 407-M respectively combine the input signals, and the combined signals are transmitted via antenna elements 401-1 to 401-M.

  Next, reception of a signal will be described. In the wireless station device 452, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receive signals while the distribution coupler 404-n and the down-converter 124-n are connected (n = 1,..., N). This point is common to the wireless station device 450 shown in FIG.

  Signals received by antenna elements 401-1 to 401-M are input to distribution couplers 407-1 to 407-M, respectively. The distribution coupler 407-m (m = 1,..., M) distributes the input signal to N systems, and outputs the phase shifters 402-1-m, 402-2-m,. To enter. The transmission / reception signal processing circuits 451-1 to 451-N process the signals input to the phase shifters 402-1-1 to 402-NM in the radio station device 450 shown in FIG. The same signal processing is performed, and a digital baseband signal is input to the signal separation circuit 141. The signal separation circuit 141 suppresses crosstalk components to perform signal separation, and the demodulators 130-1 to 130-N demodulate the separated signals.

  The signal processing for calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402-NM is implemented by the transmission / reception signal processing circuits 451-1 to 451-N in FIGS. Although the units performed are different, the signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. 1 is the same as the signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. Here, the description is omitted.

  FIG. 3 is a diagram illustrating a configuration example of a communication system according to the present embodiment. The communication system shown in the figure includes the wireless station device 452 shown in FIG. 2 and the wireless station device 450 shown in FIG. In the wireless station device 450, each of the plurality of transmission / reception signal processing circuits 451-1 to 451-N forms one beam as a sub-array. On the other hand, the wireless station device 452 has a common (sub) array, and one array antenna forms a plurality of beams. In an actual operation, as shown in FIG. 3, the radio station apparatus 450 faces the radio station apparatus 452, so that N-system signals can be spatially multiplexed.

  Next, another configuration example of the transmission / reception signal processing circuit of the present embodiment will be described with reference to FIG. The transmission / reception signal processing circuit illustrated in FIG. 4 can be replaced with the transmission / reception signal processing circuits 451-1 to 451-N included in the wireless station device 450 illustrated in FIG. 1 or the wireless station device 452 illustrated in FIG. Hereinafter, a case will be described as an example in which the transmission and reception signal processing circuits 451-1 to 451-N included in the wireless station device 450 are replaced. Note that when replacing with the transmission / reception signal processing circuits 451-1 to 451-N provided in the wireless station device 452, the antenna elements 401-n-1 to 401-n-M in FIG. Becomes

  FIG. 4 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 453-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

  The transmission / reception signal processing circuit 453-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a distribution coupler 414-n, and a phase shifter 409-n-1 to 409-n-1. 409-n-M, TDD switches 408-n-1 to 408-n-M, phase shifters 402-n-1 to 402-n-M, and switches 403-n-1 to 403-n-M. , A distribution coupler 415-n, a down converter (DC) 124-n, an A / D converter 125-n, down converters (DC) 424-n-1 to 424-n-M, It includes D converters 425-n-1 to 425-n-M, a correlation calculation circuit 405-n, and a phase shift control circuit 406-n. The TDD switches 408-n-1 to 408-n-M are connected to the antenna elements 401-n-1 to 401-n-M, respectively. Here, as in FIG. 1, the description of the external local oscillator is omitted.

  The difference between the transmission / reception signal processing circuit 453-n and the transmission / reception signal processing circuits 451-1 to 451-N shown in FIG. 1 or FIG. -N-1 to 401-n-M, and as a result, the path from the up-converter 123-n in the transmission system to the antenna elements 401-n-1 to 401-n-M and the antenna in the reception system The point is that the paths from the elements 401-n-1 to 401-nM to the downconverter 124-n are physically separated.

In the case of the transmission / reception signal processing circuit 451-n shown in FIG. 1 or FIG. 2, for example, a high power amplifier (or power amplifier) of a transmission system is arranged at a location described as “A” in the subsequent stage of the up-converter 123-n. The low-noise amplifier of the receiving system is provided at the place described as “B” before the down-converter 124-n (further, “C _n−1 ” before the down-converters 424-n-1 to 424-n-M, "C _n-2", ..., which is arranged in "C _n-M" and described location), thereby enabling shared by transmission and reception circuits from the TDD switch 127-n to the antenna terminal. This point is significantly different from the transmission / reception signal processing circuit 453-n. The advantage of adopting the configuration shown in FIG. 4 is that high power amplifiers and low noise amplifiers can be mounted by the number of antenna elements, resulting in an increase in total transmission power, and the TDD switch shown in FIG. 1 or FIG. The point is that it is possible to suppress the effects of insertion loss and distribution and coupling loss of various circuits from 127-n to antenna elements 401-n-1 to 401-n-M. Therefore, although not shown here, in the transmission and reception signal processing circuit 453-n, "D _1", "D _2", ..., high-power amplifier location as "D _M" is "E It is preferable that a low-noise amplifier be disposed at a location described as “ ₁ ”, “E ₂ ”,..., “E _M ”. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is placed at a place described as “A” and a low-noise amplifier is placed at a place described at “B”, “D ₁ ”, “D ₂ ”,. _It is not necessary that a high-power amplifier and a low-noise amplifier be arranged in each of the places described as “ _M ” and the places described as “E ₁ ”, “E ₂ ”,..., “E _M ”.

The following shows a specific signal flow in the transmission / reception signal processing circuit 453-n focusing on the above difference.
First, signal transmission will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M have the antenna elements 401-n-1 to 401-n-M and the phase shifters 409-n-1 to 409-n-. A signal is transmitted in a state where M and M are connected.

  The transmission / reception signal processing circuit 453-n receives a time-base digital baseband transmission signal of one stream from a modulator 120-n (not shown). The D / A converter 122-n converts the input transmission signal into an analog baseband signal and inputs the signal to the up-converter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a signal in a radio frequency band, and inputs the signal to the distribution coupler 414-n.

  The distribution coupler 414-n branches the radio frequency band analog signal input from the up-converter 123-n into an M-system analog signal, and inputs the signal to the phase shifters 409-n-1 to 409-n-M. . Each of the phase shifters 409-n-1 to 409-n-M applies a predetermined complex phase rotation on an analog signal to the signal input from the distribution coupler 414-n, and a TDD switch 408-n-1. To 408-nM to antenna elements 401-n-1 to 401-nM. The antenna elements 401-n-1 to 401-n-M transmit the input transmission signal. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 409-n-1 to 409-n-M.

  Next, reception of a signal will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M have the antenna elements 401-n-1 to 401-n-M and the phase shifters 402-n-1 to 402-n-. M, and switches 403-n-1 to 403-n-M receive signals while the phase shifters 402-n-1 to 402-n-M are connected to the distribution coupler 415-n. Do.

  Signals received by the antenna elements 401-n-1 to 401-n-M are respectively transmitted through the TDD switches 408-n-1 to 408-n-M and the phase shifters 402-n-1 to 402-n-M. Is entered. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on the analog signal to the input signal, and switches 403-n-1 to 403-n-M. Is input to the distribution coupler 415-n. The distribution coupler 415-n combines the signals of the respective antenna systems on an analog signal and inputs the combined signal to the down converter 124-n. The down-converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal. The A / D converter 125-n inputs the converted digital baseband signal to a signal separation circuit 141 in a baseband signal processing circuit 140 (not shown). The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402-NM will be described. The transmission / reception signal processing circuit 453-n performs this signal processing by using the switches 403-n-1 to 403-n-M and the phase shifters 402-n-1 to 402-n-M and the down converter 424-n-1 to 403-n-M. 424-nM. The switching of these switches is managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. Note that the phase shifters 402-n-1 to 402-n-M are connected to the distribution coupler 415-n except when calculating the rotation amount of the complex phase. In addition, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402-NM are set to predetermined values. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference from an initial predetermined value, as in the above description. The transmission / reception signal processing circuits 453-1 to 453-N perform similar processing all at once under the control of the control circuit 460.

  As the actual processing, first, the wireless station device of the communication partner from which the rotation amount of the complex phase should be acquired transmits a training signal for channel estimation, and the wireless station device including the transmission / reception signal processing circuits 453-1 to 453-N Receives this training signal. Signals received by the antenna elements 401-n-1 to 401-n-M are respectively transmitted through the TDD switches 408-n-1 to 408-n-M and the phase shifters 402-n-1 to 402-n-M. Is entered. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on an analog signal to the input signal, and outputs the result through switches 403-n-1 to 403-n-M. Input to the down converters 424-n-1 to 424-n-M. The down-converters 424-n-1 to 424-n-M convert the input radio frequency signals into analog baseband signals, respectively. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal and outputs the signal to the correlation calculation circuit 405-n.

  The correlation calculation circuit 405-n calculates the amount of rotation of the complex phase as in the description of FIG. The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the phase shift control circuit 406-n together with the identification number of the wireless station device with which communication is performed. The phase shift control circuit 406-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

The above-described complex phase rotation amount relates to the phase rotation amount of the phase shifters 402-n-1 to 402-n-M in the receiving system, but the complex phase rotation amount in a low noise amplifier, a high power amplifier, or the like. In order to cancel the individual difference, the correlation calculation circuit 405-n performs a calibration process in the implicit feedback of the related art, and performs a correction corresponding to the calibration process on the basis of the rotation amount of the complex phase in the reception system in the transmission system. The rotation amount of the complex phase is converted to a value set in the phase shifters 409-n-1 to 409-n-M. However, if a low-noise amplifier is placed in a place described as “E ₁ ”, “E ₂ ”,..., “E _M ”, a sufficient reception level can be obtained, so “C ₁ ”, “C ₂ ” ,..., The low-noise amplifier is unnecessary at the place where “C _M ” is described. Therefore, the rotation of the complex phase in each low-noise amplifier of the receiving system does not need to be distinguished from the rotation of the complex phase in space. As the rotation amount of the complex phase set in the phase shifters 402-n-1 to 402-n-M of the system, the rotation amount of the complex phase calculated by the correlation calculation circuit 405-n can be used as it is. The phase shift control circuit 406-n includes a complex in a transmission system to be set in each of the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M. The amount of phase rotation is managed by, for example, storing it in a memory in association with the identification number of the wireless station device of the communication partner.

  Note that when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 460 grasps the wireless station apparatus to be a communication partner and communicates with the phase shift control circuit 406-n. The phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M are instructed to set the amount of rotation of the complex phase corresponding to the wireless station device to be performed. The phase shift control circuit 406-n obtains the amount of rotation of the complex phase in each of the reception system and the transmission system corresponding to the wireless station device that performs communication, for example, by reading from a memory. The phase shift control circuit 406-n sets the rotation amount of the complex phase of the reception system to the phase shifters 402-n-1 to 402-nM, and sets the rotation amount of the complex phase of the transmission system to the phase shifter 409. −n−1 to 409−n−M to realize beamforming on analog.

  When performing digital beamforming in Massive MIMO, the conventional radio station apparatus requires a large number of A / D converters and D / A converters corresponding to the number of signal sequences, and thus has the problem of increasing power consumption. Had. On the other hand, the wireless station apparatus according to the above-described embodiment uses the A / D converter for each antenna element corresponding to the signal sequence only when calculating the amount of rotation of the complex phase for performing the directivity control, and At the time of transmission, analog beamforming using a phase shifter is performed. Therefore, the wireless station apparatus of the present embodiment uses a switch to output the received signal to the A / D converter when receiving the training signal used for calculating the amount of rotation of the complex phase, and to output the received signal when receiving data. To the receiving circuit. As a result, at the time of data transmission and reception, it becomes possible to stop the operation of the A / D converter and the D / A converter for each antenna element required for calculating the amount of rotation of the complex phase, thereby reducing power consumption. It becomes possible.

  As described above, according to this embodiment, the wireless station device uses the A / D converter for each antenna element only when calculating the amount of rotation of the complex phase for performing directivity control. Then, at the time of actual signal transmission, the radio station apparatus substitutes analog beamforming using a phase shifter instead of digital beamforming. On the other hand, at the time of signal reception, the wireless station apparatus uses a switch to output the signal output from the reception antenna to the A / D converter when receiving the training signal and to the reception circuit when performing signal synthesis and demodulation processing. Switch. This makes it possible to greatly reduce the number of A / D converters and D / A converters that have conventionally been required steadily when performing digital beamforming in Massive MIMO. Therefore, it is possible to avoid a situation in which a large number of A / D converters and D / A converters constantly operate during communication, and to avoid a huge number of A / D converters and D / A conversions that have been performed particularly during wideband communication. It is possible to reduce the power consumption of the vessel and the amount of heat generated. Furthermore, since a large heat sink required for heat generation is not required for the A / D converter and the D / A converter, not only low power consumption but also miniaturization of the wireless station device can be achieved. And cost can be reduced.

[Second embodiment]
In the first embodiment, individual down converters and A / D converters are mounted for all (transmitting) receiving antennas, but these circuits are used only for channel estimation. This works effectively to reduce the overall power consumption and the amount of heat generated.However, from the standpoint of reducing the size and cost of the device, it is necessary to mount a circuit that is only used temporarily for a huge number of antenna elements. Is inefficient. In particular, an A / D converter that operates at a very high speed because of a wide band tends to be expensive per unit, which may lead to an increase in the price of the entire device. In the second embodiment, a configuration for integrating these down converters, A / D converters, and the like into one system will be described.

  FIG. 5 is a functional block diagram illustrating a configuration example (sub-array separated type) of the wireless station device 550 according to the present embodiment. Similarly to the first embodiment, in the second embodiment, as shown in FIGS. 1 and 2 in the first embodiment, a “sub-array separation type” in which a directional beam is divided into a plurality of sub-arrays and formed. And a "sub-array shared type" that realizes a plurality of directional beams with one array (strictly speaking, since it is not separated into sub-arrays, it may be understood as an "integrated array") Variations exist. FIG. 5 illustrates the “sub-array separation type”. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

  The wireless station device 550 shown in FIG. 7 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N (N is an integer of 1 or more), and a control circuit 560. The transmission / reception signal processing circuit 551-n (n = 1,..., N) includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, and an A / D converter 125-n. , TDD switch 127-n, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, distribution coupler 504-n, and correlation calculation circuit 505-n and a phase shift control circuit 506-n. The phase shifters 502-n-1 to 502-n-M are connected to the antenna elements 501-n-1 to 501-n-M, respectively.

  As in the wireless station device 450 shown in FIG. 1, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the wireless station device 550 includes the transmission / reception signal processing circuits 551-1 to 551-N as a whole. Only N systems are implemented. Similarly to the wireless station device 450, M represents the number of antenna elements of the sub-array mounted on each of the transmission / reception signal processing circuits 551-1 to 551-N. Each of the transmission / reception signal processing circuits 551-n has sub-array antenna elements 501-n-1 to 501-n-M attached thereto. It is assumed that they will be installed spatially separated from each other. Further, the baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 551-1 to 551-N by wire, and the digital baseband signal is transferred over this wire. Although not shown in FIG. 11, an example is shown in which the wireless station device 550 mounts the entire control circuit 560 on the baseband signal processing circuit 140. The control circuit 560 manages a frame cycle and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N. The control circuit 560 determines whether the distribution coupler 504-n is connected to the up converter 123-n or the down converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by splitting. The control circuit 560 instructs the correlation calculation circuits 505-n (n = 1,..., N) to control the switches 503-n-1 to 503-n-M.

  Further, in the present embodiment, the complex phase rotation amount calculation processing basically performed by the phase shifter corresponding to the transmission / reception weight is performed by digital signal processing, and the actual complex phase rotation processing is realized by analog signal processing. For this reason, even if the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N assume signal processing on the frequency axis like the OFDM modulation method, the time axis like SC-FDE is used. Even when the above signal processing is premised, it is possible to cope with either system, and the concept regarding the variation of the communication system such as the OFDM modulation system and the SC-FDE is described in the first embodiment described above. Is the same as

  In the up-converters 123-1 to 123-N and the down-converters 124-1 to 124-N, a signal from a local oscillator is input in order to perform frequency conversion between a radio frequency signal and a baseband signal. Although required, cooperative signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551-N, and thus it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description. Although omitted in the following description, it is of course possible to perform cooperative signal processing among the transmission / reception signal processing circuits 551-1 to 551-N as an additional function. You may go.

A specific signal flow in the wireless station device 550 is as follows.
First, signal transmission will be described. The wireless station device 550 turns on all the switches 503-n-1 to 503-n-M (the connection state with the distribution coupler 504-n) and sets the TDD switch 127-n to the up converters 123-n-1 to 123. Signal transmission is performed in a state where -n-N and distribution coupler 504-n are connected (n = 1,..., N). These conditions are the same for all the transmission / reception signal processing circuits 551-1 to 551-N.

  Each of the modulators 120-1 to 120-N generates a transmission signal of a time base digital baseband of each stream to be subjected to spatial multiplexing, inputs the signal to the transmission / reception signal processing circuits 551-1 to 551-N, and then transmits / receives the signals. Processing until the up-converter 123-n of the signal processing circuit 551-n (n = 1,..., N) inputs a signal in the radio frequency band to the subsequent stage is the same as that of the radio station device 450 shown in FIG. . The up-converter 123-n inputs a signal to the distribution coupler 504-n via the TDD switch 127-n.

  The distribution coupler 504-n splits the analog signal input from the TDD switch 127-n into M-system analog signals, and outputs the phase shifter 502-n via the switches 503-n-1 to 503-n-M. -1 to 502-n-M. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation on the analog signal to the input signal. The analog signals subjected to the complex phase rotation by the phase shifters 502-n-1 to 502-n-M are transmitted via the antenna elements 501-n-1 to 501-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 502-n-1 to 502-n-M, and communicates with the wireless station device having the directivity.

  The above description is the signal processing common to the transmission / reception signal processing circuits 551-1 to 551-N mounted on the N systems. carry out.

  Next, the flow of signals related to reception will be described. The wireless station device 550 turns on all the switches 503-n-1 to 503-n -M (the connection state with the distribution coupler 504-n) and sets the TDD switch 127-n with the distribution coupler 504-n and the down converter. 124-n is connected, and a signal is received (n = 1,..., N). These conditions are the same for all the transmission / reception signal processing circuits 551-1 to 551-N.

  The signals received by the antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to the phase shifters 502-n-1 to 502-n-M, respectively. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation to the input signal on an analog signal, and outputs the result through switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. The distribution coupler 504-n combines signals of the respective antenna systems input via the switches 503-n-1 to 503-n-M on an analog signal, and combines the combined signals via the TDD switch 127-n. And input to the down-converter 124-n. Subsequent processing is the same as that of the wireless station device 450 shown in FIG.

  Each of phase shifters 502-n-1 to 502-n-M performs a predetermined complex phase rotation on an analog signal with respect to a signal received via antenna elements 501-n-1 to 501-n-M. Are added to each other to form a predetermined directivity, and communicate with the wireless station device of the directivity. The above description is the signal processing common to the transmission / reception signal processing circuits 551-1 to 551-N mounted on the N systems. carry out.

  Note that the signal separation circuit 141 performs the same processing as that of the radio station device 450 shown in FIG. A process of suppressing crosstalk components between wireless station devices that cannot be completed is performed. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted to a frequency axis signal by FFT processing and then performed on the frequency axis. Alternatively, only the directivity forming signal processing performed by the transmission / reception signal processing circuits 551-1 to 551-N may be performed, and the signal separation circuit 141 may not perform any processing (in this case, the signal separation circuit 141 Can be omitted). However, in any case, the details of the signal separation method here are not directly related to the present embodiment, and can be implemented using the conventional MIMO signal processing technology, and thus the description thereof is omitted here. .

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1 to 502-NM will be described. In this signal processing, one of the switches 503-n-1 to 503-n-M (n = 1,..., N) includes one of the phase shifters 502-n-1 to 502-n-M and the down converter 124-n. n, while the remaining switches are disconnected (OFF) from the downconverter 124-n. Among the switches 503-n-1 to 503-n-M (n = 1,..., N), the objects to be connected (turned on) to the down-converter 124-n are sequentially switched. The switching of these switches is managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. During normal operation other than when calculating the amount of rotation of the complex phase described here, the phase shifters 502-n-1 to 502-n-M (n = 1,..., N ) Are all connected to distribution couplers 504-n. In addition, when performing the calculation processing of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 502-1-1 to 502-NM are set to predetermined values. The rotation amount of the complex phase obtained in the subsequent processing is set as a difference from an initial predetermined value. For example, in the simplest example, all of the phase shifters 502-1-1 to 502-NM may be set to zero (or the same value), and in this case, the rotation amount of the obtained complex phase May be used as it is as the phase rotation amount of the phase shifters 502-1-1 to 502-NM during the subsequent communication. Alternatively, the initial predetermined values of the phase shifters 502-1-1 to 502-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the rotation amount of the complex phase is + α degrees, . +-. Degree., + .Gamma.,..., The phase rotation amounts of the phase shifters 502-1-1 to 502-NM during the subsequent communication are changed by + (. Alpha. + 10) degrees and + (. Beta. , + (Γ + 30) degrees,...

  As the actual processing, first, the wireless station device of the communication partner from which the rotation amount of the complex phase is to be acquired transmits a training signal for channel estimation, and the wireless station device 550 receives this training signal. The signals received by the antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to the phase shifters 502-n-1 to 502-n-M, respectively. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation to the input signal on an analog signal, and outputs the result through switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. Here, all of the switches 503-n-1 to 503-n-M are OFF except one, so that the signal combined in the distribution coupler 504-n is effectively the switch 503-n. Only the signal received by the antenna of the system to which the switch is connected (ON) is output from n-1 to 503-n-M. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 504-n, one antenna is selected from the antenna group of the antenna elements 501-n-1 to 501-n-M as a whole. The processing for extracting the element 501-nk (k is any integer from 1 to M) is performed. Note that k is switched at a predetermined cycle so that k takes any value from 1 to M. The reception signal of the antenna element 501-nk selected in this way is input to the down converter 124-n via the TDD switch 127-n. The down-converter 124-n converts the input radio frequency signal into an analog baseband signal, and inputs the signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal, and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  Each of the correlation calculation circuits 505-1 to 505-N continuously records the digital baseband signal while switching, until the signals from all the switches are completely received. That is, the correlation calculation circuit 505-n receives the digital baseband signals from all the switches 503-n-1 to 503-n-M while switching the switches 503-n-1 to 503-n-M. And record. The correlation calculation circuit 505-n considers the periodicity of the training signal with respect to the recorded digital baseband signal (for example, periodicity such that a training signal having the same content is repeated at a cycle of 2048 samples), and The sampling data of each antenna element 501-nk is extracted so that the sampling timing in the cycle corresponds, and similarly to the correlation calculation circuits 405-1 to 405-N of the first embodiment, the equations (1) to Using the equation (3), the amount of rotation of the complex phase to be set for each of the phase shifters 502-n-1 to 502-n-M is calculated. This utilizes the fact that the time variation of the channel can be ignored if the wireless station device is fixedly installed at a high place and has a line-of-sight environment. Further, when the calibration processing is necessary as needed, the correlation calculation circuits 505-1 to 505-N use the formula (5) similarly to the correlation calculation circuits 405-1 to 405-N of the first embodiment. The rotation amount of the complex phase on the transmission side is determined as a value in consideration of the calibration coefficients in 1) to (3).

In the first embodiment, the sampling data of each of the antenna elements 401-nk can be obtained at the same time. However, in the second embodiment, the sampling is performed at different timings. Therefore, it is devised that it can be regarded as equivalently sampled at the same time. At this time, if the frequency error cannot be neglected between the transmitting side and the receiving side, sampling may not be equivalently regarded as sampling at the same time only with the periodicity of the training signal. You can go. For example, assuming that the training signal has a periodicity of N _FFT samples for the sampling data S _k (n) of the training signal while one switch 503 -nk is continuously ON, the N _Test period Assume that enough sampling data has been secured. Assuming that the frequency error is Δf, it is possible to estimate Δf by obtaining Δf ′ that maximizes the Q value of the following equation (6) for various Δf ′.

Here, attention is paid only to the information of the switches 503-nk, but Δf may be obtained for the sampled data of each switch 503-nk and averaged. After Δf is estimated in this manner, the influence of the frequency error can be removed by performing the correction represented by the following equation (7) on the sampling data S _k (n ′).

  Here, n 'means a serial number which is continuous during switching from the switch 503-n-1 to the switch 503-n-1 irrespective of switch switching. Since the phase is rotated by 2πjΔfn ′ during the sampling period × n ′, the rotation is inversely corrected.

  The rotation amount of the complex phase obtained by the correlation calculation circuit 505-n (n = 1,..., N) in this manner is based on the identification number of the wireless station device with which the communication is performed (communication with a plurality of wireless station devices). In the case similar to the first embodiment, an identification number is not required when performing communication with a single wireless station apparatus in the PP type) and input to the phase shift control circuit 506-n. The phase shift control circuit 506-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 502-n-1 to 502-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

  Note that, when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 560 grasps the wireless station apparatus to be a communication partner, and the phase shift control circuit 506-n (n = 1,...). , N) to instruct the phase shifters 502-n-1 to 502-n-M to set the amount of rotation of the complex phase corresponding to the wireless station device performing the communication. The phase shift control circuit 506-n acquires the amount of rotation of the complex phase corresponding to the wireless station device with which the communication is performed, for example, by reading out from the memory, and acquires the amount of rotation of the complex phase. -N-M is set to realize beamforming on analog.

  Although not explicitly shown in the figure, if a transmitting-side high-power amplifier (HPA) is disposed, for example, the transmitting-side high-power amplifier (HPA) is disposed at a point described as “A” in the drawing, and the receiving-side low-noise amplifier (HPA) is disposed. If LNA) and the like are to be arranged, they are arranged at points described as “B” in the figure. Regarding the points described as “A” and “B”, in the same transmission / reception signal processing circuits 551-1 to 551-N, for the same n, the antenna elements 501-n-1 to 501-n-M In the present embodiment which does not basically assume cooperative transmission among the transmission / reception signal processing circuits 551-1 to 551-N, individual high power amplifiers and low noise amplifiers The calibration process for removing the uncertainty of the complex phase is unnecessary.

  In addition, the transmission / reception signal processing circuits 551-1 to 551-N having a sub-array configuration can be installed physically apart from each other, so that the correlation between directional beams formed on analog can be reduced. Are set apart from each other by a predetermined distance or more.

  Next, similarly to FIG. 2 with respect to FIG. 1 in the first embodiment, the radio station apparatus of the present embodiment is configured by a “sub-array shared type” in which a plurality of directional beam formations are realized by one array antenna. Is also possible. This configuration is shown in FIG.

  FIG. 6 is a functional block diagram illustrating a configuration example (shared sub-array) of the wireless station device 552 in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

  The wireless station device 552 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N, distribution couplers (HYB) 507-1 to 507-M, and an antenna element 501- 1 to 501-M and a control circuit 560. The distribution coupler (HYB) 507-m (m = 1,..., M) includes phase shifters 502-1-m, 502-502 of the transmission / reception signal processing circuits 551-1, 551-2,. 2-m, ..., 502-N-m, and the antenna element 501-m.

  In the wireless station device 552, similarly to the wireless station device 550 shown in FIG. 5, N corresponds to the multiplexing number (stream number) when performing spatial multiplexing, and the transmission / reception signal processing circuits 551-1 to 551-N are entirely And only N systems are implemented. M represents the number of antenna elements of the sub-array mounted on the wireless station device 552. In the wireless station apparatus 550 shown in FIG. 5, the number of antenna elements in each sub-array is M, so that the total number of antennas is N × M. However, in this drawing shared as in FIG. And the value of M. In actual operation, it is not necessary to use the same number of antenna elements for the wireless station device 550 shown in FIG. 5 and the wireless station device 552 shown in FIG.

  The baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 551-1 to 551-N by wire, and a digital baseband signal is transferred over the wire. Also, as in the case of FIG. 1, FIG. 6 shows an example in which the entire control circuit 560 is mounted on the baseband signal processing circuit 140.

  Similarly to the wireless station apparatus 550, the wireless station apparatus 552 basically performs digital signal processing for the amount of complex phase rotation performed by the phase shifter corresponding to the transmission / reception weight, and performs actual complex phase rotation processing in analog signal processing. It is realized by. Therefore, similarly to the wireless station device 550, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N of the wireless station device 552 assume a signal processing on the frequency axis, It is possible to cope with any of the above-described methods based on the signal processing.

  Further, in the up converters 123-1 to 123-N and the down converters 124-1 to 124-N, the input of a signal from a local oscillator is performed in order to perform frequency conversion between a radio frequency signal and a baseband signal. Required. However, since cooperative signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551-N, it is not always necessary to use a common local oscillator. Unlike the case of the wireless station device 550 shown in FIG. 5, in the wireless station device 552, the upconverters 123-1 to 123-N and the downconverters 124-1 to 124-N are all contained in the same housing. Therefore, if individual local oscillators are prepared for the up-converters 123-1 to 123-N and the down-converters 124-1 to 124-N, the cost increases, so that there is a local oscillator that is substantially shared outside. The configuration is common. However, since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  As described above, in the wireless station device 552, the transmission / reception signal processing circuits 551-1 to 551-N for N systems are mounted in one housing, and are provided via the distribution couplers 507-1 to 507-M. The antenna elements 501-1 to 501-M are shared. FIG. 6 illustrates an example in which the control circuit 560 is mounted on the baseband signal processing circuit 140, but may be mounted at an arbitrary location in the wireless station device 552. The control circuit 560 manages a frame cycle and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N.

  The difference between the wireless station device 550 shown in FIG. 5 and the wireless station device 552 shown in FIG. 6 is that the transmission / reception signal processing circuits 551-1 to 551-N are aggregated in the housing of one wireless station device 552 and shared. Antenna elements 501-1 to 501 -M of the sub-array thus divided by distribution couplers 507-1 to 507 -M and connected to transmission / reception signal processing circuits 551-1 to 551-N. is there. Therefore, the processing inside the transmission / reception signal processing circuits 551-1 to 551-N is the same for the wireless station device 550 shown in FIG. 5 and the wireless station device 552 shown in FIG.

In the following, focusing on the difference from the wireless station device 550 shown in FIG. 5, a specific signal flow as the difference in the wireless station device 552 will be described.
First, signal transmission will be described. In the wireless station device 552, processing for forming directivity is performed from each of the phase shifters 502-n-1 to 502-n-M (n = 1,..., N) of the transmission / reception signal processing circuit 551-n. An analog signal of a radio frequency is output. These M-system signals are input to distribution couplers 507-1 to 507-M connected to the corresponding antenna elements 501-1 to 501-M, respectively. That is, the phase shifters 502-1-m, 502-2-m,..., 502-Nm (m = 1,..., M) input signals to the distribution coupler 507-m. Distribution couplers 507-1 to 507-M respectively combine signals input from N transmission / reception signal processing circuits 551-1 to 551-N, and the combined signals are used as antenna elements 501-1 to 501-M. Sent via

  Next, reception of a signal will be described. The signals received by the antenna elements 501-1 to 501-M of the wireless station device 552 are distributed to N systems in the distribution couplers 507-1 to 507-M, respectively, and the transmission and reception signal processing circuits 551-1 to 551-M are respectively provided. N. For example, a signal received by the antenna element 501-m (m = 1,..., M) is distributed to N systems by the distribution coupler 507-m, and the phase shifters 502-1-m, 502-2-m, .., 502-Nm. The wireless station device 552 performs the same signal processing as the signal reception in the wireless station device 550 illustrated in FIG. 5 on the signals input to the phase shifters 502-1-1 to 502-NM in this manner.

  In the signal processing for calculating the amount of rotation of the complex phase in the phase shifters 502-1-1 to 502-NM, similarly to the reception processing, the signals received by the antenna elements 501-1 to 501-M are respectively , And are distributed to N systems in the distribution couplers 507-1 to 507-M, and are respectively input to the transmission / reception signal processing circuits 551-1 to 551-N. The wireless station device 552 adds, to the signals input to the phase shifters 502-1-1 to 502-NM in this manner, a signal similar to the calculation of the amount of rotation of the complex phase in the wireless station device 550 shown in FIG. Perform processing.

  Other operations are basically performed in the same manner as the wireless station device 550 shown in FIG. The description regarding the HPA on the transmitting side and the LNA on the receiving side is the same as that of the wireless station device 550 shown in FIG.

  The communication system according to the present embodiment has a configuration in which a wireless station device 550 is provided instead of the wireless station device 450 shown in FIG. 3 and a wireless station device 552 is provided instead of the wireless station device 452. Note that the wireless station device 450 and the wireless station device 552 may face each other, or the wireless station device 550 and the wireless station device 452 may face each other.

  Next, another configuration example of the transmission / reception signal processing circuit of the present embodiment will be described with reference to FIGS. The transmission / reception signal processing circuits shown in FIGS. 7 to 10 can be replaced with the transmission / reception signal processing circuits 551-1 to 551-N included in the wireless station device 550 shown in FIG. 5 or the wireless station device 552 shown in FIG. . Hereinafter, a case will be described as an example where the transmission and reception signal processing circuits 551-1 to 551-N included in the wireless station device 550 are replaced. Note that when replacing with the transmission / reception signal processing circuits 551-1 to 551-N provided in the wireless station device 552, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 in FIGS. To 501-n-M correspond to the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M, and the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M. And a distributed coupler (HYB).

  FIG. 7 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 553-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

  The transmission / reception signal processing circuit 553-n shown in the figure includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, an A / D converter 125-n, and a circulator 521-n. n, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, distribution coupler 504-n, correlation calculation circuit 505-n, and phase And a shift control circuit 506-n. The phase shifters 502-n-1 to 502-n-M are connected to the antenna elements 501-n-1 to 501-n-M, respectively.

  In FIG. 5, the antenna elements 501-n-1 to 501-n-M are shown in the figure for easy understanding of comparison with FIG. 5, but the transmission / reception signal processing circuit 553-n corresponds to a frame indicated by a solid line. This frame corresponds to the transmission / reception signal processing circuit 551-n in FIG.

  The difference between the transmission / reception signal processing circuit 551-n shown in FIGS. 5 and 6 and the transmission / reception signal processing circuit 553-n shown in FIG. 5 is the difference between the TDD switch 127-n in the transmission / reception signal processing circuit 551-n shown in FIGS. Although the input / output flow of the transmission signal and the reception signal is controlled by n, the transmission / reception signal processing circuit 553-n shown in FIG. 7 controls the input / output flow using the circulator 521-n. That is, the circulator 521-n passes the input signal from the up-converter 123-n to the distribution coupler 504-n, and passes the input signal from the distribution coupler 504-n to the down converter 124-n. The circulator 521-n suppresses other signal flows. Therefore, similar signal processing can be performed without using the TDD switches 127-1 to 127-N as in the transmission / reception signal processing circuit 551-n illustrated in FIGS. Except for this point, all other signal processing is the same as the signal processing of the transmission / reception signal processing circuit 551-n shown in FIGS. The description regarding the HPA on the transmission side and the LNA on the reception side is the same as that of the transmission / reception signal processing circuit 551-n illustrated in FIGS.

  FIG. 8 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 554-n according to the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

  The transmission / reception signal processing circuit 554-n shown in the figure includes a D / A converter 122-n, an up-converter 123-n, a distribution coupler 514-n, and phase shifters 509-n-1 to 509-n. -M, TDD switches 508-n-1 to 508-n-M, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, and distribution It includes a coupler 515-n, a down converter 124-n, an A / D converter 125-n, a correlation calculation circuit 505-n, and a phase shift control circuit 506-n. The TDD switches 508-n-1 to 508-n-M are connected to the antenna elements 501-n-1 to 501-n-M, respectively.

  Here, similarly to the transmission / reception signal processing circuits 551-1 to 551-N shown in FIG. 5, the up-converter 123-n and the down-converter 124-n perform frequency conversion between a radio frequency signal and a baseband signal. Therefore, it is necessary to input a signal from a local oscillator. However, since cooperative signal processing is not assumed between the transmission / reception signal processing circuits 554-1 to 554-N, it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  Further, the entire control circuit 560 is implemented in the base station signal processing circuit 140 of the wireless station device 550 or the wireless station device 552 or in the wireless station device 552. The control circuit 560 manages transmission / reception timing. Further, since the plurality of transmission / reception signal processing circuits 554-1 to 554-N operate in conjunction with each other, when the control circuit 560 performs control, the plurality of transmission / reception signal processing circuits 554-1 to 554-N Work together.

  The difference between the transmission / reception signal processing circuit 554-n and the transmission / reception signal processing circuits 551-1 to 551-N shown in FIG. 5 or FIG. 6 is that the TDD switches 508-n-1 to 508-n-M are the antenna elements 501. -N-1 to 501-n-M (or antenna elements 501-1 to 501-M), and as a result, the D / A converter 122-n to the antenna element 501-n-1 in the transmission system. The path from the antenna element 501-n-M to the A / D converter 125-n is physically separated from the path from the antenna element 501-n-1 to 501-n-M in the receiving system. .

In the case of the transmission / reception signal processing circuit 551-n shown in FIG. 5 or FIG. 6, for example, a high-power amplifier (or power amplifier) of the transmission system is arranged at a location described as “A” in the subsequent stage of the up-converter 123-n. The low-noise amplifier of the receiving system is arranged at the location described as "B" in the preceding stage of the down-converter 124-n, and the circuit from the TDD switch 127-n to the antenna end can be shared for transmission and reception. This point is significantly different from the transmission / reception signal processing circuit 554-n. The merit of adopting the configuration as shown in FIG. 8 is that high power amplifiers and low noise amplifiers can be mounted by the number of antenna elements, and as a result, the total transmission power is improved, and the TDD switch shown in FIGS. The point is that it becomes possible to suppress the effects of insertion loss and distribution and coupling loss of various circuits from 127-n to the antenna elements 501-1 to 501-M. For this reason, although not shown here, high-power amplifiers are placed at the locations described as “D ₁ ”, “D ₂ ”,..., “D _M ”, and “E ₁ ”, “E ₂ ”,. It is preferable that a low-noise amplifier is arranged at a location described as “E _M ”. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is placed at a place described as “A” and a low-noise amplifier is placed at a place described as “B”, “D ₁ ”, “D ₂ ” _,. There is no necessity to arrange a high power amplifier and a low noise amplifier at each of the locations of the points described as “E ₁ ”, “E ₂ ”,..., “E _M ”.

  The above changes correspond to the change of the transmission / reception signal processing circuit 451-n in FIGS. 1 and 2 in the first embodiment to the transmission / reception signal processing circuit 453-n shown in FIG. The transmission / reception signal processing circuit 551-n shown in FIGS. 5 and 6 in the second embodiment is changed to the transmission / reception signal processing circuit 554-n shown in FIG. 453-n.

The following is a description of a specific signal flow in the transmission / reception signal processing circuit 554-n, which is a difference from the transmission / reception signal processing circuit 551-n, focusing on a characteristic part of the present embodiment.
First, signal transmission will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M have the antenna elements 501-n-1 to 501-n-M and the phase shifters 509-n-1 to 509-n-M. A signal is transmitted in a state where M and M are connected.

  A transmission / reception signal processing circuit 554-n receives a time-base digital baseband transmission signal of one stream from a modulator 120-n (not shown). The D / A converter 122-n converts the input transmission signal into an analog baseband signal and outputs the signal to the up-converter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a signal in a radio frequency band, and inputs the signal to the distribution coupler 514-n.

  The distribution coupler 514-n branches the analog signal of the radio frequency band input from the up-converter 123-n into M-system analog signals and inputs the signals to the phase shifters 509-n-1 to 509-n-M. . Each of the phase shifters 509-n-1 to 509-n-M applies a predetermined complex phase rotation on the analog signal to the signal input from the distribution coupler 514-n, and a TDD switch 508-n-1. 508-nM to the antenna elements 501-n-1 to 501-nM. The antenna elements 501-n-1 to 501-n-M transmit the input transmission signal. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 509-n-1 to 509-n-M.

  Next, reception of a signal will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M have the antenna elements 501-n-1 to 501-n-M and the phase shifters 502-n-1 to 502-n-. M and the switches 503-n-1 to 503-n-M connect the distribution coupler 515-n and the phase shifters 502-n-1 to 502-n-M, respectively, to receive signals. I do.

  Signals received by the antenna elements 501-n-1 to 501-n-M are respectively transmitted through the TDD switches 508-n-1 to 508-n-M via the phase shifters 502-n-1 to 502-n-M. Is input to Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation on the analog signal to the input signal, and switches 503-n-1 to 503-n-M. And input to the distribution coupler 515-n. The distribution coupler 515-n combines signals of the respective antenna systems input via the switches 503-n-1 to 503-n-M on an analog signal, and inputs the combined signal to the down converter 124-n. The down-converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal input from the downconverter 124-n into a digital baseband signal, and outputs a signal in a baseband signal processing circuit 140 (not shown). Input to the separation circuit 141. The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1 to 502-NM will be described. In this signal processing, one of the switches 503-n-1 to 503-n-M connects (ON) the distribution coupler 515-n and the phase shifters 502-n-1 to 502-n-M. On the other hand, the remaining switches are performed in a state where the connection with the distribution coupler 515-n is cut off (OFF). The switching of these switches is managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. Note that during normal operation other than when calculating the amount of rotation of the complex phase described here, all of the phase shifters 502-n-1 to 502-n-M are distributed couplers 515-n as described above. The phase shifters 509-n-1 to 509-n-M are all connected to the distribution coupler 514-n. Also, as in the case of the transmission / reception signal processing circuit 551-n shown in FIG. 5 or FIG. 6, when calculating the rotation amount of the complex phase, the phase shifters 502-1-1 to 502-1-M perform the phase rotation. The amount is set to a predetermined value, and the rotation amount of the complex phase obtained in the subsequent processing is set as a difference from the initial predetermined value.

  As the actual processing, first, the wireless station device of the communication partner from which the rotation amount of the complex phase is to be acquired transmits a training signal for channel estimation, and the wireless station device having the transmission / reception signal processing circuits 554-1 to 554-N Receives this signal. Signals received by antenna elements 501-n-1 to 501-n-M are input to phase shifters 502-n-1 to 502-n-M, respectively. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation to the input signal on an analog signal, and outputs the result through switches 503-n-1 to 503-n-M. And input to the distribution coupler 515-n. Here, all of the switches 503-n-1 to 503-n-M are OFF except one, so that the signal combined in the distribution coupler 515-n is effectively the switch 503-n-M. Only the signal received by the antenna of the system to which the switch is connected (ON) is output from n-1 to 503-n-M. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 515-n, one antenna element 501-n is selected from the antenna elements 501-n-1 to 501-n-M as a whole. A process for extracting −k (k is any integer from 1 to M) is performed. Note that the switching is performed so that k takes any value from 1 to M. The reception signal of the antenna element 501-nk selected in this way is input to the down converter 124-n. The down-converter 124-n converts the input radio frequency signal into an analog baseband signal, and inputs the signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal into a digital baseband signal and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  The correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 554-n correspond to the correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 551-n shown in FIG. The same processing is performed. That is, the correlation calculation circuit 505-n calculates the rotation amount of the complex phase on the transmission side to be set in each of the phase shifters 502-n-1 to 502-n-M. The phase shift control circuit 506-n stores the amount of rotation of the complex phase to be set in each of the phase shifters 502-n-1 to 502-n-M in the memory in association with the identification number of the wireless station device of the communication partner. Manage by memorizing.

Further, the correlation calculation circuit 505-n outputs the high-power amplifiers at the locations described as “D ₁ ”, “D ₂ ”,..., “D _M ” and “E ₁ ”, “E ₂ ”,. When the low-noise amplifier is arranged at the location described as “E _M ”, similarly to the correlation calculation circuit 405-n of the transmission / reception signal processing circuit 453-n in FIG. 4, the complex phase of the reception system calculated above is calculated. The rotation amount may be subjected to a calibration process in the implicit feedback of the related art, and may be converted to a rotation amount of the complex phase in the transmission system by a correction corresponding to the calibration process based on the rotation amount of the complex phase in the reception system. . The phase shift control circuit 506-n associates the rotation amount of the complex phase in the transmission system to be set in each of the phase shifters 509-n-1 to 509-n-M with the identification number of the wireless station device of the communication partner. Similarly, it is managed by storing it in a memory.

  Note that, when performing actual data communication, that is, when performing transmission processing or reception processing, the control circuit 560 grasps the wireless station apparatus to be a communication partner and communicates with the phase shift control circuit 506-n. The phase shifters 502-n-1 to 502-n-M and the phase shifters 509-n-1 to 509-n-M are instructed to set the rotation amount of the complex phase corresponding to the wireless station device to be performed. The phase shift control circuit 506-n obtains the amount of rotation of the complex phase corresponding to the wireless station device with which the communication is performed, for example, by reading out from the memory, and acquires the amount of rotation of the complex phase from the phase shifters 502-n-1 to 502. -N-M and phase shifters 509-n-1 to 509-n-M are set to implement analog beamforming. These processes are similarly performed in the transmission / reception signal processing circuits 554-1 to 554-N.

  FIG. 9 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 555-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

  The difference between the transmission / reception signal processing circuit 555-n shown in FIG. 8 and the transmission / reception signal processing circuit 554-n shown in FIG. 8 is as follows. That is, the transmission / reception signal processing circuit 554-n illustrated in FIG. 8 shares the antenna elements 501-n-1 to 501-n-M for transmission and reception. On the other hand, the transmission / reception signal processing circuit 555-n shown in FIG. 9 is configured to separate them for transmission and reception, form a pair for transmission / reception, and arrange a set of transmission / reception antennas in a close place. Therefore, when replacing the transmission / reception signal processing circuit 554-n of the wireless station device 550 with the transmission / reception signal processing circuit 555-n, the antenna elements 541-n-1 to 541-n-M are added, and the TDD switch 508-n-M is added. 1 to 508-nM are omitted. When replacing the transmission / reception signal processing circuit 551-n of the wireless station device 552 with the transmission / reception signal processing circuit 555-n, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 to 541 are used. In place of −nM, the antenna elements 501-1 to 501-M and 541-1 to 541-M (and the antenna elements 501-1 to 501-501) shared by the transmission / reception signal processing circuits 555-1 to 555-N are used. -M, a distribution coupler corresponding to the antenna elements 541-1 to 541-M).

  In the description of FIG. 8, for the sake of convenience, the description has been made assuming that the portion immediately adjacent to the antenna element is the transmission / reception signal processing circuit 554-n. If considered, FIGS. 8 and 9 are completely equivalent diagrams. Regarding the details of the signal processing, in the receiving system, the transmission / reception signal processing circuit 554-n shown in FIG. 8 has a configuration in which the antenna elements 501-n-1 to 501-n-M receive signals, In the transmission / reception signal processing circuit 555-n shown in (1), reception is performed using the antenna elements 541-n-1 to 541-n-M, and transmission / reception does not pass through the TDD switches 508-n-1 to 508-n-M. Except for this point, all the signal processing in the transmission / reception signal processing circuit 554-n shown in FIG. 8 and the transmission / reception signal processing circuit 555-n shown in FIG. 9 are common.

  However, in consideration of the fact that the transmitting and receiving antennas are physically different, it is also possible to perform correction in consideration of the fact that the coordinates of the antenna elements are physically different, in addition to the simple calibration processing.

  In the above description, the transmitting and receiving antennas are arranged as a set in a close place. However, as long as the correction is performed in consideration of the fact that the coordinates of the antenna elements are physically different, the transmitting and receiving antennas are not necessarily set. No need to place. FIG. 10 is a functional block diagram illustrating a configuration example of the transmission / reception signal processing circuit 556-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figures are assigned the same numbers, and their explanations are omitted.

In the transmission / reception signal processing circuit 556-n shown in FIG. 10, similarly to the transmission / reception signal processing circuit 555-n shown in FIG. 9, the TDD switches 508-n-1 to 508-n-M have the functions of the antenna element side. If considered, FIG. 8 is a diagram completely equivalent to FIG. In this sense, in the transmission / reception signal processing circuit 556-n shown in FIG. 10, similarly to the transmission / reception signal processing circuit 555-n shown in FIG. 9, the transmission antenna elements 501-n-1 to 501-n-M and the reception antenna Antenna elements 541-n-1 to 541-n-M are separated from each other. However, in the transmission / reception signal processing circuit 555-n illustrated in FIG. 9, a pair of individual transmission / reception antenna elements (for example, a pair of the antenna element 501-n-1 and the antenna element 541-n-1) is integrally disposed in the vicinity. In the transmission / reception signal processing circuit 556-n shown in FIG. 10, the transmission antenna elements 501-n-1 to 501-n-M constitute a single transmission antenna array, and the reception antenna The elements 541-n-1 to 541-n-M are assumed to constitute a single receiving antenna array. Therefore, there is no difference in signal processing between FIG. 9 and FIG. 10 except for the physical arrangement of the antenna elements.
When the transmitting and receiving antennas are operated separately as shown in FIG. 10, a wall-shaped obstacle may be arranged in order to avoid signal leakage from the transmitting antenna to the receiving antenna.

  When digital beamforming is performed in Massive MIMO, a conventional radio station requires expensive A / D converters and D / A converters in a number corresponding to the number of signal sequences, so that the apparatus is expensive. At the same time, power consumption increases. Therefore, the wireless station device of the present embodiment switches the switch so that only the signal sequence targeted only when obtaining weight information for performing directivity control is connected to the A / D converter. Also, at the time of data transmission, the wireless station device converts a signal before being separated for each antenna element into an analog signal by a D / A converter, separates the converted analog signal for each antenna element, and then uses a phase shifter. To perform analog beamforming. As a result, it is possible to reduce the number of A / D converters and D / A converters required to acquire weight information at the time of data transmission and reception, and to reduce power consumption.

  As described above, according to the wireless station device of the present embodiment, a switch is provided for each antenna element, and when the amount of rotation of the complex phase serving as weight information is calculated from the state of the received signal for each receiving antenna element, Only one of the receiving antenna elements is turned on to connect to the A / D converter, and a switch is devised so that the antenna element connected to this A / D converter is periodically switched. As a result, the radio station apparatus can individually grasp the states of the reception signals of all antenna elements by one RF system (including from the down converter to the A / D converter), and based on the information, the rotation amount of the complex phase. Is calculated. In Massive MIMO, it is necessary to calculate weight information from at least the state of a received signal for each antenna element. Conventionally, A / D converters and D / A converters have become very expensive, especially in a wide band, and it has been difficult to reduce the size of the devices, including the measures of heat generation. According to the present embodiment, the number of A / Ds and D / As to be mounted can be minimized, and the size, the power consumption, and the economy of the wireless station device can be reduced.

  As described above, according to the present embodiment, when calculating the rotation amount of the complex phase for performing the directivity control, only one of all the receiving antenna elements is turned ON. The down converter and the A / D converter are used while switching the switches as described above. Then, at the time of actual signal transmission, the radio station apparatus substitutes analog beamforming using a phase shifter instead of digital beamforming. On the other hand, when receiving a signal, the wireless station device turns on all switches, synthesizes a signal that has been subjected to analog beamforming using a phase shifter, and performs demodulation processing. As a result, the number of A / D converters and D / A converters conventionally required when performing digital beamforming in Massive MIMO can be significantly reduced. Therefore, it is possible to avoid a situation in which a large number of A / D converters and D / A converters constantly operate during communication, and to avoid a huge number of A / D converters and D / A conversions that have been performed particularly during wideband communication. It is possible to reduce the power consumption of the vessel and the amount of heat generated. Further, the number of necessary A / D converters and D / A converters is greatly reduced, and a large heat radiating plate required for heat generation accompanying enormous power consumption is not required. It is possible to reduce the size of the wireless station device as well as the power.

  In the above-described embodiment, the rotation of the complex phase is performed on the analog signal of the radio frequency. However, the positions of the up converter and the down converter are changed, and the complex A configuration may be adopted in which rotation is given, and frequency conversion with a radio frequency is performed in a subsequent or preceding stage.

  According to the present embodiment, a wireless communication device such as the wireless station devices 550 and 552 forms directivity using one or a plurality of sub-arrays including a plurality of antenna elements, and communicates with other wireless communication devices. Perform wireless communication. However, although the expression “sub-array” is used in this specification, the signal processing feature of the present invention does not necessarily have to be configured to implement a plurality of “sub-arrays”, and the wireless communication device includes A single array antenna composed of all the antenna elements may be used, or an array antenna composed of a part of all the antenna elements included in the wireless device may be used. The wireless communication device includes a plurality of antenna elements, a signal receiving unit, a first phase rotation unit, a switching unit, a switching control unit, a signal combining unit, a signal converting unit, a signal reproducing unit, and a correlation calculating unit. Unit, a first rotation amount calculation unit, and a first phase rotation amount control unit. Note that the wireless communication device includes a first phase rotation unit, a switching unit, a signal synthesis unit, and a signal conversion unit at least for each sub-array used for signal reception. Further, the wireless communication device includes a transmission signal generation unit, a signal distribution unit, a second phase rotation unit, a signal transmission unit, a second rotation amount calculation unit, and a second phase rotation amount control unit. Prepare. Note that the wireless communication device includes a signal distribution unit and a second phase rotation unit for at least each sub-array used for signal transmission. Here, if the phase rotation amount control unit is a PP (point-to-point) type, it can be dealt with by directly setting the rotation amount of the complex phase calculated by the rotation amount calculation unit to the phase rotation unit. However, in this case, it can be understood that one of the rotation amount calculation unit and the phase rotation unit implicitly includes a function of a phase rotation amount control unit.

  When receiving a signal to be reproduced or receiving a training signal, the signal receiving unit transmits a signal transmitted by another wireless communication device of a communication partner by wireless communication through each of a plurality of antenna elements forming a sub-array used for signal reception. To receive. The first phase rotator is configured to output, from among the signals received by the signal receiving unit, a received signal received via an antenna element forming a sub-array corresponding to the function unit, on an analog signal basis for each antenna element system. Output the complex phase rotated by a predetermined value. It should be noted that if the signal simply passes through the first phase rotation unit at the time of receiving a training signal or the like, it is understood that a zero-degree complex phase is rotated.

  The switching unit switches whether to pass the reception signal output from the first phase rotation unit corresponding to the same sub-array as the self-function unit for each system of the antenna element. When the received signal is a reproduction target, the switching control unit allows all or a part of the received signal output from the first phase rotation unit corresponding to the switching unit to pass when the received signal is a reproduction target. Is a training signal, control is performed such that the reception signal output from the first phase rotation unit corresponding to the switching unit is passed one by one for each antenna element system. The signal combining unit combines the received signals passed by the switching unit corresponding to the same sub-array as the own function unit for each of the antenna element systems. The signal converting unit converts the received signal synthesized by the signal synthesizing unit corresponding to the same sub-array as the function unit from a radio frequency analog signal to a baseband digital signal.

  At the time of receiving the training signal, the correlation calculating unit continuously records the signal sequence of the input digital signal, and considers the periodicity of the training signal from the recorded digital signal sequence, and the sampling timing in the cycle corresponds. Thus, a digital signal sequence of each antenna element is extracted. Further, the correlation calculator calculates a correlation of the digital signal sequence with the training signal between the antenna elements corresponding to the function unit. Specifically, the correlation calculation unit determines a reference antenna element from among the plurality of antenna elements, and uses the training signal passed by the switching unit one by one to use the reference antenna element and other antenna elements. The correlation of the digital signal sequence with the training signal having the same sampling timing is calculated for each combination of The first rotation amount calculation unit calculates a rotation amount of a complex phase to be given in the first phase rotation unit based on a calculation result of the correlation by the correlation calculation unit. The first phase rotation amount control unit controls a rotation amount of the complex phase given by the first phase rotation unit based on a calculation result of the rotation amount of the complex phase by the first rotation amount calculation unit.

  Further, the second rotation amount calculation unit may calculate the correlation result calculated by the correlation calculation unit, the calculation result of the rotation amount of the complex phase calculated by the first rotation amount calculation unit, or perform a calibration process on the calculation result. , The amount of rotation of the complex phase to be given in the second phase rotation unit is calculated based on any of the information on the amount of rotation of the complex phase obtained by performing the above. The second phase rotation amount control unit controls the amount of rotation of the complex phase given by the second phase rotation unit based on the result of calculation of the amount of rotation of the complex phase by the second rotation amount calculation unit.

  Upon receiving the signal to be reproduced, the signal reproducing unit reproduces a signal transmitted by another wireless communication device of the communication partner based on the received signal synthesized by the signal synthesizing unit. On the other hand, at the time of signal transmission, the transmission signal generation unit generates an analog transmission signal to be transmitted to another wireless communication device of the communication partner using the sub-array. The signal distribution unit branches a transmission signal to be transmitted using a sub-array corresponding to the function unit of the analog transmission signal generated by the transmission signal generation unit, in correspondence with each of the antenna elements constituting the sub-array. The second phase rotation unit rotates the complex phase by a predetermined value on the analog signal for each of the transmission signals branched by the signal distribution unit corresponding to the same sub-array as the self-function unit for each antenna element. The signal transmission unit transmits the transmission signal whose phase has been rotated by the second phase rotation unit via an antenna element forming a sub-array corresponding to the second phase rotation unit.

Note that the wireless communication apparatus uses a sub-array such as a correlation calculation circuit 505-1 to 505-N that integrates a correlation calculation unit, a first rotation amount calculation unit, and a second rotation amount calculation unit. A correlation calculation unit, a first rotation amount calculation unit, and a second rotation amount calculation unit may be separately provided.
Further, the radio communication device includes a functional unit in which a first phase rotation amount control unit and a second phase rotation amount control unit are integrated for each sub-array, such as phase shift control circuits 506-1 to 506-N. Alternatively, the first phase rotation amount control unit and the second phase rotation amount control unit may be separately provided.

  In addition, the wireless communication device includes phase shifters 502-n-1 to 502-n-M (n = 1,..., N) included in the transmission / reception signal processing circuits 551-1 to 551-N or the transmission / reception signal processing circuits 553-n. ), A phase rotator integrating the first phase rotator and the second phase rotator may be provided for each sub-array, and the transmission / reception signal processing circuits 554-n, 555-n, and 556-n The first phase rotation unit and the second phase rotation unit are individually provided like phase shifters 502-n-1 to 502-n-M and 509-n-1 to 509-n-M. Is also good.

  In addition, the wireless communication device is similar to the distribution couplers 504-1 to 504-N included in the transmission / reception signal processing circuits 551-1 to 551-N and the distribution coupler 504-n included in the transmission / reception signal processing circuits 553-n. Alternatively, a functional unit obtained by integrating a signal combining unit and a signal distributing unit may be provided for each sub-array. In this case, the wireless communication device is a switching unit that temporally switches between processing of the signal combining unit and processing of the signal distributing unit in the signal combining and distributing unit, such as the TDD switches 127-1 to 127 -N, or the circulator 521-n. As described above, an output port may be different for each input port, and a switching unit for switching the direction of a flowing signal may be further provided for each sub-array. In addition, the wireless communication device independently controls the signal combining unit and the signal distribution unit like the distribution couplers 514-n and 515-n included in the transmission / reception signal processing circuits 554-n, 555-n, and 556-n. May be provided.

[Supplementary information on the embodiment]
The supplementary items relating to the embodiment of the present invention described above are shown below.
The wireless station device in the above-described embodiment may be realized by a computer. In that case, a program for realizing this function may be recorded on a computer-readable recording medium, and the program recorded on this recording medium may be read and executed by a computer system. Here, the “computer system” includes an OS and hardware such as peripheral devices. The “computer-readable recording medium” refers to a portable medium such as a flexible disk, a magneto-optical disk, a ROM, and a CD-ROM, and a storage device such as a hard disk built in a computer system. Further, a “computer-readable recording medium” refers to a communication line for transmitting a program via a network such as the Internet or a communication line such as a telephone line. Such a program may include a program that holds a program for a certain period of time, such as a volatile memory in a computer system serving as a server or a client in that case. The program may be for realizing a part of the functions described above, or may be a program that can realize the functions described above in combination with a program already recorded in a computer system, It may be realized using hardware such as a PLD (Programmable Logic Device) or an FPGA (Field Programmable Gate Array).

  As described above, the embodiments of the present invention have been described with reference to the drawings. However, it is apparent that the above embodiments are merely examples of the present invention, and the present invention is not limited to the above embodiments. is there. Therefore, additions, omissions, replacements, and other changes may be made to the components without departing from the technical spirit and scope of the present invention.

  The present invention is applicable to a wireless communication device that transmits and receives a wireless signal using a plurality of antennas.

120-1 to 120-N ... Modulators 122-1 to 122-N, 122-n ... D / A converters 123-1 to 123-N, 123-n ... Up-converters 124-1 to 124-N, 124 -N: Down converters 125-1 to 125-N, 125-n: A / D converters 127-1 to 127-N: TDD switches 130-1 to 130-N: Demodulator 140: Baseband signal processing circuit 141 ... signal separation circuits 401-1 to 401-M ... antenna elements 401-1-1 to 401-NM, 401-n-1 to 401-nM ... antenna elements 402-1-1 to 402-N- M, 402-n-1 to 402-nM ... phase shifters 403-1-1 to 403-NM, 403-n-1 to 403-nM ... switches 404-1 to 404-N ... Distribution couplers 405-1 to 405-N, 40 -N correlation calculation circuits 406-1 to 406-N, 406-n phase shift control circuits 407-1 to 407-M distribution couplers 408-n-1 to 408-n-M TDD switches 409-n -1 to 409-nM phase shifter 414-n distribution coupler 415-n distribution couplers 424-1-1 to 424-NM, 424-n-1 to 424-n-M Down-converters 425-1-1 to 425-NM, 425-n-1 to 425-nm, A / D converter 450, wireless station devices 451-1 to 451-N, transmission / reception signal processing circuit 452 Radio station apparatus 453-n transmission / reception signal processing circuit 460 control circuits 501-1 to 501-M antenna elements 501-1-1 to 501-NM, 501-n-1 to 501-n-N antenna Elements 502-1-1 to 502-NM, 502 n-1 to 502-n-N phase shifters 503-1-1 to 503-NM, 503-n-1 to 503-n-N switches 504-1 to 504-N, 504-n ... Distribution couplers 505-1 to 505-N, 505-n Correlation calculation circuits 506-1 to 506-N, 506-n Phase shift control circuits 507-1 to 507-M Distribution coupler 508-n-1 508-nM TDD switches 509-n-1 to 509-nM, 509-n-1 to 509-nN ... phase shifters 514-n ... distribution couplers 515-n ... distribution couplers 521-n circulators 541-1 to 541-M antenna elements 541-n-1 to 541-n-M antenna elements 550 radio station devices 551-1 to 551-N transmission / reception signal processing circuit 552 radio station Device 553-n... Transmission / reception signal processing circuit 554 n ... reception signal processing circuit 555-n ... reception signal processing circuit 556-n ... reception signal processing circuit 560 ... control circuit 901-1~901-N ... modulator 902 ... precoder _903-1~903-N 0 ... IFFT & GI imparting circuit _904-1~904-N 0 ... D / A converter 905-1~905-N 0 _... upconverter 906-1~906-N 0 _... downconverter _907-1~907-N 0 ... A / D converter 908-1 to 908-N ₀ GI removal & FFT circuit 909 Post coder 910-1 to 910-N Demodulator 911 TDD switch 912-1 to 912-N ₀ Distribution coupler 913-1-1 _913-N 0 -M 0 _... phase shifter 915-1~915-M 0 _... distributor coupler 916-1~916-M 0 _... antenna elements 921-1~921-N ... time Axis transmission weight multiplying circuits 922-1 to 922 -M D / A converters 922-1 to 922 -N -M D / A converters 923-1 to 923 -M ... up-converters 923-1-1 923-NM ... Up converters 924-1 to 924-M ... Down converters 924-1-1 to 924-NM ... Down converters 925-1 to 925-M ... A / D converters 925-1- 1-925-NM: A / D converters 926-1 to 926-N: time-axis reception weight multiplying circuits 927-1 to 927-N: TDD switches 928-1 to 928-M: antenna elements 928-1 -1 to 928-NM Antenna elements 929-1 to 929-N Transmission / reception signal processing circuits 941-1 to 941-M Distribution coupler 942 Radio station devices 943-1 to 943-M Addition combiner 944-1 44-M duplicater 945 wireless station devices 951-1 to 951-3 high power amplifiers 952-1 to 952-3 low noise amplifiers 953-1 to 953-3 TDD switches 954-1 to 954-3 Antenna elements 955-1 to 955-3 ... wireless module

Claims (6)

A wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device,
A signal receiving unit that receives the signal transmitted by the other wireless communication device via each of the antenna elements used for signal reception,
At least for each of the array antennas used for signal reception, of the signals received by the signal receiving unit, for each of the received signals received via the antenna elements constituting the corresponding array antenna, a system of antenna elements A first phase rotation unit for rotating and outputting a complex phase on an analog signal every time;
A switching unit provided at least for each of the array antennas used for signal reception, and capable of switching whether or not to pass the reception signal output from the corresponding first phase rotation unit for each system of the antenna element,
If the received signal is to be reproduced, all or a part of the received signal output from the corresponding first phase rotation unit is passed, and the received signal is subjected to training for calculating a complex phase rotation amount. A switching control unit that controls switching in the switching unit so that the reception signal output from the corresponding first phase rotation unit is passed one by one for each system of the antenna element when the signal is a signal;
A signal combining unit that is provided at least for each of the array antennas used for signal reception, and combines the received signals passed by the corresponding switching unit,
A signal converter provided at least for each of the array antennas used for signal reception and converting the received signal synthesized by the corresponding signal synthesizer from an analog signal to a baseband digital signal,
A signal reproducing unit that reproduces a signal transmitted by the other wireless communication device based on the received signal to be reproduced converted by the signal conversion unit,
A correlation calculating unit that calculates a correlation for each combination of a reference antenna element among the antenna elements and another antenna element of the training signal using the training signal that is passed one by one for each system of the antenna element; A calculating unit;
A first rotation amount calculation unit that calculates a rotation amount of a complex phase to be given in the first phase rotation unit based on a calculation result of the correlation by the correlation calculation unit;
With
The correlation calculation unit acquires the training signal having the periodicity of the N _FFT samples passed by the switching unit for a plurality of cycles for the same system of the antenna element, and obtains n-th (n is 1) in a plurality of cycles . To the N _FFT ) , add the sampling data of the training signal and the result of performing the frequency error correction to the sampling data of the n-th training signal in a cycle subsequent to the training signal. Performing an addition process, estimating a frequency error so as to maximize the sum of the results of the addition process, and the switching unit passes one or more of the training signals passed one by one for each antenna element system. Performing a process of removing the effect of the frequency error estimated for the system, the frequency error Calculating a correlation of each combination of the reference antenna element and the other antenna elements in said antenna elements of said training signal using the training signal to remove effects,
A wireless communication device characterized by the above-mentioned. A transmission signal generation unit that generates an analog transmission signal to be transmitted to another wireless communication device using the array antenna,
A signal distribution unit that is provided at least for each of the array antennas used for signal transmission and branches the analog transmission signal transmitted using the corresponding array antenna in accordance with each of the antenna elements constituting the array antenna. ,
A second phase rotation unit that is provided at least for each of the array antennas used for signal transmission and that rotates the complex phase on an analog signal for each of the transmission signals branched by the corresponding signal distribution unit.
A signal transmission unit that transmits the transmission signal whose phase is rotated by the second phase rotation unit, via the antenna element that configures the array antenna corresponding to the second phase rotation unit;
The calculation result of the correlation calculated by the correlation calculation unit, the calculation result of the rotation amount of the complex phase calculated by the first rotation amount calculation unit, or the complex phase obtained by performing a calibration process on the calculation result A second rotation amount calculation unit that calculates a rotation amount of a complex phase to be given in the second phase rotation unit based on any of the information on the rotation amount of
The wireless communication device according to claim 1, wherein: A phase rotator that performs processing of the first phase rotator and processing of the second phase rotator instead of the first phase rotator and the second phase rotator is provided for each of the array antennas. Prepare,
The wireless communication device according to claim 2, wherein: In place of the signal synthesizing unit and the signal distributing unit, a signal synthesizing and distributing unit that performs a process of the signal synthesizing unit and a process of the signal distributing unit is provided for each of the array antennas,
A switching unit that switches between processing of the signal combining unit and processing of the signal distributing unit in the signal combining and distributing unit is further provided for each array antenna.
The wireless communication device according to claim 2 or 3, wherein: The signal synthesizing unit and the signal distributing unit are independent and independent, and the first phase rotating unit and the second phase rotating unit are independent and independent.
The wireless communication device according to claim 2 or 3, wherein: A wireless communication method executed by a wireless communication device that wirelessly communicates with another wireless communication device by forming directivity using one or more array antennas including a plurality of antenna elements,
A signal receiving unit, the signal transmitted by the other wireless communication device, a signal receiving step of receiving via each of the antenna elements used for signal reception,
At least a phase rotator provided for each of the array antennas used for signal reception is configured such that, among signals received in a signal receiving process, a reception signal received via each of the antenna elements constituting the corresponding array antenna is an antenna. A phase rotation process of rotating a complex phase on an analog signal for each element system,
At least a switching unit provided for each of the array antennas used for signal reception, when the reception signal is to be reproduced, all of the reception signals obtained by rotating a complex phase in the phase rotation process by the corresponding phase rotation unit. Or, when the received signal is a training signal for calculating the amount of rotation of a complex phase, the received signal obtained by rotating the complex phase in the phase rotation process by the corresponding phase rotator is used as an antenna. A switching process of passing one by one for each element system;
A signal synthesizing unit provided at least for each of the array antennas used for signal reception, a signal synthesizing step of synthesizing the received signal passed in the switching step by the corresponding switching unit,
A signal conversion unit provided at least for each of the array antennas used for signal reception converts the received signal synthesized by the corresponding signal synthesis unit in the signal synthesis process from an analog signal to a baseband digital signal. Process
A signal reproducing unit that reproduces a signal transmitted by the other wireless communication device based on the received signal to be reproduced converted into a digital signal in the signal converting step;
The correlation calculation unit uses the training signal that is passed one by one for each antenna element system in the switching process, for each combination of a reference antenna element and another antenna element among the antenna elements of the training signal. A correlation calculation process for calculating the correlation,
A rotation amount calculation unit calculates a rotation amount of a complex phase to be given in the phase rotation process, based on a calculation result of the correlation in the correlation calculation process,
Has,
The correlation calculation unit acquires the training signal having a periodicity of N _FFT samples passed in the switching process for the same antenna element system for a plurality of cycles, and obtains n-th (n is 1) in a plurality of cycles . To the N _FFT ) , add the sampling data of the training signal and the result of performing the frequency error correction to the sampling data of the n-th training signal in a cycle subsequent to the training signal. Further comprising an estimation step of estimating a frequency error so as to maximize the sum of the results of the addition processing ,
In the correlation calculation step, a process of removing the influence of the frequency error estimated for one or more of the systems with respect to the training signals passed one by one for each of the antenna element systems in the switching step. Perform, calculate the correlation for each combination of a reference antenna element and other antenna elements in the antenna element of the training signal using the training signal to remove the effect of the frequency error,
A wireless communication method, comprising:

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