JP2018019380A - Radio communication device and radio communication method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To implement downsizing and cost reduction of a radio communication device.SOLUTION: Each sub array includes a first phase rotation unit for rotating a complex phase on an analog signal with respect to a reception signal by each antenna element constituting an array antenna. Each array antenna includes a changeover unit that allows all or part of reception signals output from the first phase rotation unit to pass when the reception signals are a reproduction target and allows a reception signal to pass one by one when the reception signal is a training signal. Each array antenna includes a signal synthesis unit for synthesizing reception signals allowed by and passing through a corresponding changeover unit, and a corresponding signal conversion unit converts a synthesized reception signal to a baseband digital signal. A signal reproduction unit reproduces a reception signal that is a reproduction target and is converted by a conversion unit. A correlation calculation unit, using training signals allowed by and passing through the changeover unit, calculates training signal correlation among antenna elements. A first phase rotation amount control unit, on the basis of a complex phase rotation amount calculated on the basis of the correlation, controls a rotation amount to be applied by the first phase rotation unit.SELECTED DRAWING: Figure 5

Description

  The present invention relates to a wireless communication apparatus and a wireless communication method.

[Background of 5th generation mobile communications]
Currently, high-functional mobile communication terminals such as smartphones are explosively spreading. With regard to mobile phones, the third generation mobile communication has shifted to the fourth generation mobile communication, and research and development related to the fifth generation mobile communication (commonly referred to as “5G”) is now underway. One of the studies that have been conducted on 5G is the use of macro cells and small cells.

  In conventional cellular phones, one service area is set to a radius of several kilometers, and this macro cell area is covered by one base station device. However, there are a very large number of users in such a macro cell. Since the limited system capacity as a whole is shared by each user, when accommodating a huge number of users, the throughput for each individual user is lowered.

  In order to avoid such a decrease in throughput, a technology for setting a small cell, which is a very small service area with a radius of several tens of meters, in a densely populated area where traffic is concentrated has been developed. With this technology, spot traffic is offloaded to the network without using a macro cell by utilizing a small cell. Here, it is assumed that the terminal device can use the communication capability in the small cell and the communication capability in the macro cell simultaneously. By using such a terminal device, user data is accommodated on the small cell side while exchanging information using the macro cell for control information. This makes it possible to make maximum use of the advantages of macro cells and small cells.

  In 5G mentioned above, the target value of transmission speed is set to 10 Gbit / s (gigabit per second) or more, and this small cell can perform efficient traffic offload by performing the same large-capacity communication. There is a need to. In a macro cell, it is assumed that a microwave band with a low frequency is used in order to allow long-distance propagation. However, considering the current state of the microwave band, where frequency resources are already being depleted, small cells that assume relatively short-distance communication are expected to use the quasi-millimeter wave band or the millimeter wave band with a relatively high frequency. Has been. The characteristic of this high frequency band is that propagation attenuation increases in inverse proportion to the square of the frequency. Therefore, it is preferable that the small cell base station is ideally installed near the user terminal. For example, in a place where installation is easy, such as on the roof of a building, the distance between the user terminal and the base station is too far away, which is not preferable in terms of circuit design.

  On the other hand, since the small cell is set in a place where traffic is concentrated, there is a case where it is strongly desired to install a base station apparatus even in a place where it is difficult to lay an optical fiber. For example, assuming that a small cell base station device is installed in a place where people are very crowded, such as in front of a station such as Shinjuku or Shibuya, propagation attenuation increases on the rooftop of a building adjacent to such a location. . For this reason, it may be required to be installed on a place lower in height than the rooftop of the building, such as a wall surface of the building. However, it may be difficult to install optical fiber on the wall of an existing building. In such a case, it may be necessary to provide a backhaul line to the base station apparatus using a radio line. is there.

  When providing such a backhaul line, it is necessary to use the millimeter wave band to perform a large capacity transmission of 10 Gbit / s or more in order to cope with a large capacity transmission of 10 Gbit / s or more required in a small cell. There is. In such an environment, since both opposing radio station apparatuses are fixedly installed in stable locations, it is natural that the line-of-sight is secured stably and the directional antennas are directed to each other. is there. In this case, reflected waves between buildings exist to some extent, but most of the received signals are line-of-sight components, and it is expected that it is difficult to say a multipath environment. This situation is the same for the access system if the small cell base station device is installed at a high place such as a building wall surface and looks down from the top to the bottom of the user and is generally used in a line-of-sight environment.

  Next, for large-capacity transmission of 10 Gbit / s or higher, which is the transmission rate required for 5G, it becomes possible to use frequency resources with a very wide bandwidth by utilizing the millimeter wave band, and this is feasible. Is growing. For example, if a backhaul line using the millimeter wave band is assumed, E band (71 to 76 GHz and 81 to 86 GHz) is used as an example, and if a bandwidth of 1 GHz is used, the frequency utilization efficiency is 10 bits. / S / Hz is sufficient. However, existing wireless equipment for achieving a frequency utilization efficiency of 10 bits / s / Hz generally employs spatial multiplexing transmission using a multiple-input multiple-output (MIMO) channel. Spatial multiplex transmission generally uses a multipath environment, and when the singular value decomposition of the channel matrix H expressing the transfer function of the MIMO channel in a matrix form is performed, the distribution of absolute values of the singular values obtained as a result thereof Represents the characteristics of the spatial multiplexing transmission. Specifically, the square value of the absolute value of the singular value is a value proportional to the signal-to-noise power ratio SNR (Signal to Noise Ratio), and not only the first singular value but also the first singular value for spatial multiplexing transmission. Communication is not possible if there is no sufficiently large value after the 2 singular value. The same is true for large-capacity transmission in small cells that are access systems, but it is indispensable to realize spatial multiplexing transmission in an environment where such line-of-sight waves are dominant. .

  As described above, in 5G, the use of the millimeter wave band is expected for both the access system and the backhaul line. As described above, the characteristic of the high frequency band is that the propagation attenuation increases in inverse proportion to the square of the frequency. For example, if the 2 GHz band (existing access system) and the 80 GHz band (future system using the millimeter wave band) are compared to 40 times the frequency, the propagation attenuation is 1600 times and the line gain of 32 dB is insufficient. It will be. Of course, it is not necessary to cover as much as 32 dB because it does not need to cover as wide a range as a macro cell. However, in the high frequency band, high power amplifiers in the transmission stage do not have very high output devices. In this case, it is considered that an additional line gain at a level of several tens of dB must be secured. Furthermore, since spatial multiplexing transmission is also expected in such an environment, both base stations and terminal stations are now considering wireless systems with significantly more antenna elements than conventional techniques. . Such a technique is called Massive MIMO. Below, the prior art regarding Massive MIMO is introduced.

  In Non-Patent Document 1, the base station side is equipped with 256 elements and the terminal station side is equipped with 16 elements, and aims at 16 streams of spatial multiplexing transmission utilizing a 256 × 16 channel matrix. In this non-patent document 1, in order to transmit a plurality of signal sequences (streams), the directivity is formed by analog beam forming using rotation of a complex phase amount in a wireless analog circuit and in a digital baseband circuit. Performed in combination with digital beam forming in the digital domain. As a result, avoiding heavy use of analog / digital (A / D) converters and digital / analog (D / A) converters, power consumption is reduced and channel gain shortage countermeasures are taken when channel information is fed back. Hereinafter, an outline of the technique disclosed in Non-Patent Document 1 will be described.

FIG. 11 is a functional block diagram showing a configuration example of a radio station apparatus in the prior art. The radio station apparatus shown in the figure includes modulators 901-1 to 901-N (MOD # 1 to MOD # N), a precoder 902, IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval). Giving circuits 903-1 to 903-N ₀ , D / A converters 904-1 to 904-N ₀ , up converters (UC) 905-1 to 905-N ₀ , and down converters (DC) 906-1 906-N ₀ , A / D converters 907-1 to 907-N ₀ , GI removal & FFT (Fast Fourier Transform) circuits 908-1 to 908-N ₀ , a post coder 909, Demodulators 910-1 to 910 -N (DEM # 1 to DEM # N), TDD switch (TDD-SW) 911, distribution coupler (HYB) 912-1 to 912 -N ₀ , and phase shifter 913 -1 Comprises a _{-1~913-N 0} -M 0, and distributor coupler (HYB) _{915-1~915-M 0,} and antenna element _{916-1~916-M 0.} Here, N corresponds to the number of multiplexed (number of streams) when performing spatial multiplexing, N ₀ represents the number of signal systems for performing signal processing for digital directivity formation, and M ₀ represents the number of antenna elements. (M ₀ ≧ N ₀ ≧ N). Further, the antenna elements 916-1 to 916 -M ₀ constitute an array antenna as a whole.

The distribution couplers 912-1 to 912 -N ₀ to the antenna elements 916-1 to 916 -M ₀ are common for transmission and reception. Further, the TDD switch 911 increases the connection from the distribution couplers 912-1 to 912-N ₀ to the antenna elements 916-1 to 916-M ₀ from the modulators 901-1 to 901-N corresponding to the transmission system. Switching is performed between converters 905-1 to 905-N ₀ and down converters 906-1 to 906-N ₀ corresponding to the reception system to demodulators 910-1 to 910-N. For example, the up converter 905-n and the distribution coupler 912-n are connected during transmission, and the down converter 906-n and the distribution coupler 912-n are connected during reception (n = 1,..., N ₀ ). An overall control circuit (not shown) manages the frame period and transmission / reception timing, and the TDD switch 911 is also switched by this control circuit.

The phase shifters 913-1-1 to 913 -N ₀ -M ₀ adjust the phase relationship of transmission and reception signals according to a predetermined beam pattern, and this phase rotation amount is controlled by a control circuit (not shown). Are also managed. The phase here is the same as the directivity control in the phased array antenna. For example, as directional gain in a predetermined direction across the antenna elements _{916-1~916-M 0} is the maximum, at each antenna element _{916-1~916-M 0} against incoming waves from that direction The complex phase corresponding to the value obtained by dividing the path length difference by the wavelength is adjusted. This enables each antenna element 916-1 to 916 -M ₀ to transmit and receive signals in the same phase.

The directivity here is obtained by dividing the horizontal azimuth angle θ and the vertical azimuth angle φ by a step size of a predetermined angle, and for each selectable (θ _i , φ _j ) menu, the corresponding complex The phase amount of the phase shifters 913-n-1 to 913-n-M ₀ (n = 1,..., N ₀ ) is adjusted with the set of phases as a set. As a result, for example, the antenna element 916-1 to 916 -M ₀ , the phase shifters 913-n-1 to 913 -n-M ₀ , and one hypothesis for the n-th signal sequence in the entire distribution coupler 912-n. Behaves as a directional antenna. These virtual directional antenna is physically via distribution couplers _{915-1~915-M 0,} will share the antenna element _{916-1~916-M 0.}

  Furthermore, in the following description, a case where communication is performed by forming a signal on the frequency axis as in an OFDM (Orthogonal Frequency Division Multiplexing) modulation method will be described as an example. Even in the case of single carrier transmission, when equalization processing on the frequency axis is performed, the signal is once converted to a frequency axis signal. Similar processing is possible regardless of whether OFDM or single carrier transmission is used.

The specific signal flow is as follows. First, signal transmission will be described. Modulators 901-1 to 901 -N each generate a transmission signal for each stream for spatial multiplexing. The precoder 902 appropriately combines signals between a plurality of virtual directional antennas so that signal separation can be efficiently performed on the receiving station side. This precoding processing corresponds to multiplication of a unitary transformation matrix when singular value decomposition is performed on, for example, a MIMO channel matrix between N ₀ virtual directional antennas and N signal systems actually transmitted and received. Thus, so-called eigenmode transmission is realized, and efficient transmission is realized. The IFFT & GI adding circuits 903-1 to 903 -N ₀ convert the transmission signal sequence formed in this way from a signal on the frequency axis to a signal on the time axis, and add a guard interval. If necessary, waveform shaping between symbols is also performed here. The D / A converters 904-1 to 904 -N ₀ convert the digital signals generated in this way into analog signals. Upconverter _{905-1~905-N 0} is the analog signal, converts the baseband signal to the radio frequency signals.

At the time of transmission, the TDD switch 911 connects the up converter 905-n and the distribution coupler 912-n (n = 1,..., N ₀ ). It should be noted, 1~N ₀ of subscript is all behave in the same way. Distribution coupler 912-n (n = 1,..., N ₀ ) distributes radio frequency band signals to signals of only the number of antenna systems M ₀ , and phase-shifters 913-n-1 to 913-n. input to -M _0. For example, the phase shifters 913-1-1 to 913-1 -M ₀ adjust the predetermined complex phase corresponding to the directivity direction (θ _i , φ _j ) corresponding to the first signal sequence on the analog signal. in is performed to transmit a first signal sequence after the adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} Similarly the phase shifter _{913-N 0 -1~913-N 0} -M 0 , the first _{N 0} directivity of the azimuth corresponding to the signal sequence of _{(θ i ', φ j'} ) predetermined complex phase corresponding to the the adjustments performed on the analog signal and transmits the signal sequence of the N after adjustment from the antenna elements _{916-1~916-M 0} through the distribution coupler _{915-1~915-M 0.} The distribution couplers 915-m (m = 1,..., M ₀ ) receive the signals input from the corresponding phase shifters 913-1-m, 913-2-m, ..., 913-N ₀ -m. Combined and output to the antenna element 916-m.

Next, signal reception will be described. The signals received by the antenna elements 916-1 to 916 -M ₀ are distributed to the N ₀ system signals by the distribution couplers 915-1 to 915 -M ₀ , respectively, and the corresponding phase shifters 913-1-1. It is output to the _{~913-N 0} -M 0.
For example, the phase shifters 913-1-1 to 913-1 -M ₀ adjust the predetermined complex phase corresponding to the directivity direction (θ _i , φ _j ) corresponding to the first signal sequence on the analog signal. The first signal sequence after adjustment is input to the distribution coupler 912-1. The distribution coupler 912-1 combines these input signals, and inputs the combined signal to the down converter 906-1 via the TDD switch 911.
Similarly, the phase shifter _{913-N 0 -1~913-N 0} -M 0 , the directivity of the azimuth corresponding to the signal sequence of the _{N (θ i ', φ j} ') predetermined complex phase corresponding to the the adjustments performed on the analog signal, and inputs the N-th signal sequence after the adjustment to the distributor coupler 912-N _0. Distribution coupler 912-N ₀ synthesizes these inputted signals and inputs the synthesized signal to down converter 906-N ₀ via TDD switch 911.

The down converters 906-1 to 906-N ₀ down-convert radio frequency signals into baseband signals. The A / D converters 907-1 to 907 -N ₀ convert analog baseband signals obtained by down-conversion into digital baseband signals. Here, based on symbol timing managed by a timing detection circuit (not shown), the GI removal & FFT circuits 908-1 to 908 -N ₀ remove the guard interval from the digital baseband signal, Convert to frequency axis signal. Postcoder 909, the cross talk component between the signal sequence (stream) which has been processed by the GI removal & FFT circuit _{908-1~908-N 0} and signal separation in the frequency domain, corresponding signals after the crosstalk component separation Output to the demodulators 910-1 to 910 -N. Demodulators 910-1 to 910 -N reproduce and output data by a predetermined signal detection process.

Here, the power amplifier on the transmission side and the low noise amplifier on the reception side are not explicitly described, but in general, the latter stage of the up converters 905-1 to 905-N ₀ (references “A ₁ ” to “A _N0 ”). ) And a low noise amplifier are installed in front of the down converters 906-1 to 906 -N ₀ (positions “B ₁ ” to “B _N0 ”). The power amplifier and the low noise amplifier have different complex phase rotation amounts, and there are cases where the transfer rotation amount differs for each frequency. However, a factor causing a difference in phase rotation amount between transmission and reception is eliminated between the TDD switch 911 and the antenna elements 916-1 to 916 -M ₀ , and the symmetry of the channel between transmission and reception is eliminated. Saved. For this reason, the same value can be used for setting the phase rotation amount of the phase shifters 913-1-1 to 913 -N ₀ -M ₀ at the time of transmission and at the time of reception.

The above is an outline of the Massive MIMO technology using hybrid beamforming. Here, the phase rotation amount set by the phase shifters 913-1-1 to 913 -N ₀ -M ₀ or the directionality of the directivity corresponding to each signal series (in the above example, the phase shifters 913-1-1-1). in _{913-1-M 0 (θ i,} φ j), the phase shifter _{913-N 0 -1~913-N 0} -M 0 (θ i ', φ j')) acquisition method, such as the present invention Although it is omitted because it is not directly related to the above feature, it can be obtained by conventional techniques such as Non-Patent Document 1.

[Expansion of Massive MIMO technology when line-of-sight wave is dominant]
In the above description of the Massive MIMO technology, since it is mainly assumed to be used in an access system, it is assumed that the environment is generally a multipath environment. However, even in the case of an access system, if a small cell base station is installed on the upper side and it is possible to secure a general outlook while looking down, operation in a non-multipath environment may be forced. In the case of a backhaul line in particular, this is conspicuous, and it is assumed that the so-called Rice coefficient K is 10 dB or more, and it is assumed to be used in an environment where only about 1/10 or less of the line-of-sight component is accompanied by multipath components. In this case, it is expected that the difference between the line gain corresponding to the first singular value and the line gain corresponding to the second singular value or more will be 20 dB or more. It is expected that

  In such an environment, as shown in Non-Patent Document 2, using the high efficiency of the line gain corresponding to the first singular value, all antenna elements are divided into a plurality of sets, and subarrays are set for each set. It is effective to take a configuration. Then, by installing the subarrays spatially separated, it is effective to reduce the correlation between the subarrays and to transmit the transmission corresponding to the first singular value in parallel with low correlation. Similarly, in Non-Patent Document 3, when the antenna aperture length of the subarray is narrow, the line-of-sight wave is dominant, and the correlation between the antenna elements in the subarray is sufficiently strong, the transmission / reception weight of each antenna element is the frequency. A “time-axis beamforming technique” has been proposed in which weights can be multiplied in units of sampling data on the time axis, which can be handled as constants having no dependency.

This is the relative channel information for each antenna element when the element spacing is narrow and the correlation between the antenna elements is strong (the relative value of the channel information with respect to the channel information at a certain antenna element, specifically the reference antenna When the complex phase of the channel information of the k-th frequency component of the element is ψ _ref ^(k) , the frequency dependence of the complex phase of Exp {−jψ _ref ^(k) } multiplied by each antenna element) The nature is due to being almost constant. For example, at the time of reception, the complex phase of the reception weight for synthesizing the reception signals of these antenna elements so that the complex phase is the same phase is generally constant in the entire frequency band. Thus, the same constant reception weight can be used. In general, when a function that is constant on the frequency axis is Fourier transformed to a δ function, the weight obtained by converting the reception weight on the frequency axis onto the time axis by IFFT only needs to consider the component at t = 0. Become. In other words, since signal processing in consideration of the delayed wave component is unnecessary, the time axis that is a predetermined coefficient for each antenna element is directly added to the sampling data obtained by sampling the analog baseband received signal by the A / D converter. By multiplying the reception weight, it is possible to realize directivity formation with only time-axis signal processing without converting the received signal into a signal on the frequency axis by FFT processing or the like.

  A coefficient for rotating the complex phase multiplied as a time axis weight is obtained by the following equations (1) to (3).

In the above equation, S _i (n) represents the sampling data of the n th sample of the i th antenna in the received training signal, and S _i (n) ^* represents the complex conjugate of S _i (n). Represent. N _FFT assumes a predetermined periodicity, and indicates a periodicity value that is meaningful in correlation detection, such as the number of FFT points of OFDM. ψ _j is the amount of rotation of the complex phase (on the receiving side) performed by time axis beamforming. The function angle (x) is a function representing the complex phase of the complex number x, and is a value determined by the ratio of the real part of x to the imaginary part of x and the sign of the real part and imaginary part. Although the in the correlation calculation in the formula (1) performs a correlation operation over N _FFT samples is the number of FFT points in the case of the OFDM signal as a predetermined periodicity, it is an integer multiple of example N _FFT Correlation calculation over other sample numbers may be performed so that the periodicity is maintained.

Here, as is clear from the equation (2), the sign of the complex phase of the complex coefficient c _j given by the above equation (1) and the complex phase of the time axis weight w _j given by the above equation (2) are inverted. It has become a thing. In this sense, in the embodiment of the present invention described later, the complex phase of the complex coefficient c _j given by the expression (1) corresponding to the relative channel information is obtained, and the complex phase of the time axis weight w _j is obtained. That is equivalent.

FIG. 12 is a functional block diagram showing a configuration example (subarray separation type) of a radio station apparatus using time-axis beamforming in the prior art described in Non-Patent Document 3. The radio station apparatus shown in the figure includes a baseband signal processing circuit 140 and transmission / reception signal processing circuits 929-1 to 929 -N. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 929-n (n = 1,..., N) includes a time-axis transmission weight multiplication circuit 921-n, D / A converters 922-n-1 to 922-n-M, and an up converter 923. -N-1 to 923-nM, down converter (DC) 924-n-1 to 924-nM, A / D converters 925-n-1 to 925-nM, time axis A reception weight multiplication circuit 926-n and a TDD switch (TDD-SW) 927-n are provided. The TDD switch 927-n is connected to the antenna elements 928-n-1 to 928-n-M. Here, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the transmission / reception signal processing circuits 929-1 to 929 -N are mounted for N systems as a whole. M represents the number of antenna elements of the subarray mounted on each of the transmission / reception signal processing circuits 929-1 to 929 -N. In the description of FIG. 11, the total number of antenna elements is M ₀ , but here the whole antenna has a subarray configuration, so the number of antenna elements in each subarray is indicated as a different value M.

  Here, the transmission / reception signal processing circuits 929-1 to 929 -N (the transmission / reception signal processing circuit 929 -n is accompanied by subarray antenna elements 928 -n-1 to 928 -n-M) are described in Non-Patent Document 2. It is assumed that they will be installed spatially apart like The baseband signal processing circuit 140 is connected to each of the transmission / reception signal processing circuits 929-1 to 929 -N by wire, and a digital baseband signal is transferred over this wire. Similarly to the case of FIG. 11, an overall control circuit (not shown) is mounted on the baseband signal processing circuit 140 to manage the frame period and transmission / reception timing. Here, the TDD switches 927-1 to 927-1 are used. The switching of 927-N is also managed here.

Furthermore, the time-axis beamforming technique basically assumes signal processing on the time axis, but even when signals on the frequency axis are formed as in the OFDM modulation method, signals on the frequency axis are processed by FFT processing and IFFT processing. Can be converted into a signal on the time axis, and by performing signal processing on this time axis signal, the time axis beam forming technique can be similarly applied to the OFDM modulation method as well as the single carrier transmission. However, in FIG. 11, signal processing on the frequency axis is assumed, and IFFT & GI adding circuits 903-1 to 903-N ₀ for IFFT processing and GI removal & FFT circuits 908-1 to 908-N ₀ for FFT processing Is shown separately from the modulators 901-1 to 901 -N and the demodulators 910-1 to 910 -N, but here, since signal processing on the frequency axis is not assumed, FIG. Even when the OFDM modulation method or the like is used, the functions of FFT processing and IFFT processing are included in the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N (or the signal separation circuit 141). These notations are omitted. Therefore, the input / output signals from the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are signals on the time axis regardless of the OFDM modulation scheme or single carrier transmission. . Further, the signal separation circuit 141 performs signal separation between each signal series, but here it is also possible to perform signal separation on the time axis, and once convert it to a frequency axis signal by FFT, Frequency-dependent signal separation may be performed. In this sense, the input to the demodulators 130-1 to 130-N is assumed to be a time-axis signal or a frequency-axis signal. Will be described as input.

  The specific signal flow is as follows. First, signal transmission will be described. Modulators 120-1 to 120 -N generate time base digital baseband transmission signals of the respective streams to be spatially multiplexed, and input them to time base transmission weight multiplication circuits 921-1 to 921 -N, respectively. To do. The time-axis transmission weight multiplication circuit 921-n (n = 1,..., N) each of the subarrays for forming the directivity of the digital signal input from the modulator 120-n by the transmission / reception signal processing circuit 929-n. The signal is converted into a digital signal multiplied by a transmission weight corresponding to the antenna elements 928-n-1 to 928-n-M. The D / A converters 922-n-1 to 922-n-M convert the digital signal multiplied by the transmission weight into an analog baseband signal, and the up-converters 923-n-1 to 923-n-M The analog baseband signal is converted into a radio frequency band signal. At the time of transmission, the TDD switch 927-n connects the up-converter 923-nm (m is an integer of 1 to M) to the antenna element 928-nm. Each antenna element 928-n-1 to 928-n-M transmits the respective signals input from the up-converters 923-n-1 to 923-n-M, and the transmission / reception signal processing circuits 929-1 to 929 are transmitted. A directional beam is formed every -N.

  Next, signal reception will be described. Signals received by the antenna elements 928-n-1 to 928-n-M (n = 1,..., N) are input to the down converters 924-n-1 to 924-n-M via the TDD switch 927-n. Is done. The down converters 924-n-1 to 924-n-M convert radio frequency signals into analog baseband signals. The A / D converters 925-n-1 to 925-n-M convert analog baseband signals into digital baseband signals. This digital baseband signal is input to the time axis reception weight multiplication circuit 926-n. The time axis reception weight multiplication circuit 926-n multiplies each input signal by a reception weight corresponding to each antenna element 928-n-1 to 928-n-M, and adds and combines the signals after reception weight multiplication. Thus, each is converted into one signal series. That is, a total of N signal series (streams) are converted by the time axis reception weight multiplication circuits 926-1 to 926 -N, and these signals are input to the signal separation circuit 141. The signal separation circuit 141 performs signal separation by suppressing crosstalk components between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. Demodulators 130-1 to 130-N reproduce and output data by a predetermined signal detection process.

  The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be performed once on the frequency axis after being converted into a frequency axis signal by FFT processing. Alternatively, only the signal processing performed by the transmission / reception signal processing circuits 929-1 to 929 -N is required, and the signal separation circuit 141 does not need to perform any processing. However, in any case, details of the signal separation method here are omitted because they are not directly related to the present application.

  The time axis transmission weights used in the time axis transmission weight multiplication circuits 921-1 to 921-N and the time axis reception weights used in the time axis reception weight multiplication circuits 926-1 to 926-N are shown here. No time axis transmission / reception weight acquisition means acquires. Similarly, a control circuit not shown here manages the value of the time axis transmission / reception weight used there. For example, a reference antenna element over a predetermined number of samples based on sampling data acquired by the A / D converters 925-n-1 to 925-n-M with respect to a training signal transmitted by a wireless station as a communication partner A correlation value between antenna elements (for example, 928-n-1) may be obtained and determined based on this complex phase. The values of the complex phase of the time axis reception weight and the time axis transmission weight generally do not match because the rotation amount of the complex phase such as a power amplifier and a low noise amplifier not shown here is different for each amplifier. By using the implicit feedback calibration method, conversion from the time axis reception weight to the time axis transmission weight is possible. The transmission / reception weight acquired in this way is stored in the memory for each corresponding radio station apparatus. At the time of transmission and reception, the time-axis transmission weight multiplication circuits 921-1 to 921-N and the time-axis reception weight multiplication circuits 926-1 to 926-N perform weight multiplication based on these transmission / reception weight values. It will be.

  FIG. 13 is a functional block diagram showing a configuration example (subarray shared type) of a radio station apparatus using time-axis beamforming in the conventional technique described in Non-Patent Document 3. In the figure, the same figure number is given to the function common to FIG. In the figure, a radio station apparatus 942 includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 929-1 to 929-N, distribution couplers (HYB) 941-1 to 941-M, and an antenna element 928-. 1-928-M. In FIG. 12, the transmission / reception signal processing circuits 929-1 to 929 -N are accommodated in different housings assuming that they are installed in spatially separated locations for each sub-array, and baseband signal processing circuits in separate housings. A configuration in which a wired connection is made with the network 140 is shown. On the other hand, in FIG. 13, all the transmission / reception signal processing circuits 929-1 to 929 -N and the baseband signal processing circuit 140 are configured as a radio station device 942 in the same casing, and the antenna elements 928-1 to 928 -M are entirely configured. Shared by. For this reason, for example, at the time of transmission, signals from the TDD switches 927-1 to 927 -N of the transmission / reception signal processing circuits 929-1 to 929 -N are synthesized by the distribution couplers 941-1 to 941 -M and synthesized. The transmitted signals are transmitted from the antenna elements 928-1 to 928-M. Similarly, at the time of reception, the signals received by the antenna elements 928-1 to 928-M are distributed by the distribution couplers 941-1 to 941-M. That is, the distribution coupler 941-m (m = 1,..., M) distributes and inputs the signal received by the antenna element 928-m to the TDD switches 927-1 to 927-N. All other signal processing is the same in FIGS.

  Similarly, FIG. 14 is a functional block diagram of another configuration example (subarray shared type) of a radio station apparatus using time-axis beamforming in the prior art described in Non-Patent Document 3. In this figure, the same parts as those in the radio station apparatus shown in FIG. The radio station apparatus 945 shown in the figure includes a baseband signal processing circuit 140, time axis transmission weight multiplication circuits 921-1 to 921-N, addition synthesizers 943-1 to 943-M, and a D / A converter. 922-1 to 922-M, up-converters 923-1 to 923-M, down-converters 924-1 to 924-M, A / D converters 925-1 to 925-M, and duplicator 944-1 944-M, time axis reception weight multiplication circuits 926-1 to 926-N, a TDD switch 927, and antenna elements 928-1 to 928-M.

  In FIG. 13, the transmission / reception signal processing circuits 929-1 to 929 -N are individually mounted for each subarray, but the D / A converters 922-n-1 to 922 -n-M, the up-converters 923-n-1 to 923-nM, down converters 924-n-1 to 924-nM, A / D converters 925-n-1 to 925-nM, and TDD switches 927-1 to 927-N are shared. Possible (n = 1,..., N). Therefore, in FIG. 14, N digital baseband signals generated by the time-axis transmission weight multiplication circuits 921-1 to 921-N (where N planes are mounted as a whole) are added and synthesizer 943- 1 to 943-M are added and synthesized, and each of them is integrated into one system, and converted from a digital signal to an analog signal by a D / A converter 922-1 to 922-M. Similarly, on the receiving side, the duplicators 944-1 to 944-M duplicate the digital baseband signals generated by the A / D converters 925-1 to 925 -M into N signals in units of sampling data, The duplicated signal is input to time axis reception weight multiplication circuits 926-1 to 926 -N (where N planes are mounted as a whole). Each of the time axis reception weight multiplication circuits 926-1 to 926 -N multiplies the input signal by the reception weight, and adds and synthesizes the result to convert each signal into one system signal. That is, a total of N signals are converted by the time axis reception weight multiplication circuits 926-1 to 926 -N, and these signals are input to the signal separation circuit 941.

  As a result, the D / A converters 922-n-1 to 922-n-M are duplicated, the up converters 923-n-1 to 923-n-M are duplicated, and the down converters 924-n-1 to 924- n-M overlap mounting, A / D converters 925-n-1 to 925-n-M, TDD switches 927-1 to 927-N are avoided, circuit scale reduction, power consumption, etc. This has led to a reduction in energy consumption.

  Here, in actual operation, the radio station apparatus shown in FIG. 12 and the radio station apparatus shown in FIG. 13 or FIG. For example, the base station apparatus has a degree of freedom of installation like a building roof, and subarrays can be installed at a plurality of locations. On the other hand, the terminal station device side has a configuration in which the base station device is shown in FIG. 12 and the terminal station device is shown in FIG. 13 or FIG. It is possible to increase the degree of freedom of installation by sharing the antenna. Alternatively, for example, if the transmission capacity per terminal station apparatus is such that spatial multiplexing is not required, FIG. 13 or FIG. 14 shows the base station apparatus, and FIG. 12 shows one system among the transmission / reception signal processing circuits 929-1 to 929 -N. And a configuration in which multi-user MIMO transmission is performed by a plurality of terminal station apparatuses and one base station apparatus, assuming that the terminal station apparatus is implemented only with (for example, only the transmission / reception signal processing circuit 929-1 in FIG. 12). It is also possible to do.

Incidentally, 11 and 12, with regard to the correspondence between FIGS. 13 and 14, for example a phase shifter _{913-1-1~913-N} 0 rotation amount of complex phase carried out by -M ₀ in FIG. 11 [psi _alpha ( α is equivalent to an identification number for phase shifters 913-1-1 to 913 -N ₀ -M ₀ ), time axis transmission weight multiplication circuits 921-1 to 921 -N, time axis reception weight multiplication circuit 926 −1 to 926-N corresponds to multiplying the signal sequence of the corresponding antenna element by Exp {jψ _α }. In other words, in FIG. 11, signals transmitted and received from each antenna element are converted by analog circuits (that is, phase shifters 913-1-1 to 913 -N ₀ -M ₀ ), whereas FIGS. In FIG. 14, the signal transmitted / received from each antenna element is converted by a digital circuit (that is, time-axis transmission weight multiplication circuits 921-1 to 921-N, time-axis reception weight multiplication circuits 926-1 to 926-N). To do. Accordingly, FIG. 11 has an advantage that the number of A / D converters 907-1 to 907 -N ₀ and D / A converters 904-1 to 904 -N ₀ can be suppressed, while FIG. 13 and 14, as described in Non-Patent Document 3, there are advantages that the resolution of directivity formation is increased and that simple and efficient channel information feedback is possible.

Note that the phase rotation by the phase shifter is usually performed by selectively passing a delay line corresponding to the amount of phase rotation on the device. For this reason, when phase rotation of x is given as an absolute value, the phase rotation (delay) of the complex phase rotation amount is negative with respect to the delay corresponding to the phase x as a signal, and the sign consistency cannot be obtained. . However, in the following description, for convenience, when the phase shift corresponding to the multiplication of the coefficient Exp {jψ _α } is given as a signal by the phase shifter, the amount of rotation of the complex phase performed by the phase shifter is referred to as ψ _α. To do.

[Calibration technology in channel information feedback]
In general, when performing directivity formation using a plurality of antenna elements on the transmission side, it is necessary to feed back channel information of the MIMO channel, including the techniques from Non-Patent Document 1 to Non-Patent Document 3 described above. At this time, if the number of antenna elements becomes enormous, the amount of channel information to be fed back becomes enormous, and various ideas are required. In the Massive MIMO system as described above, in order to obtain the forward link channel information in the transmission direction, the reverse link channel information in the reception direction is used, and the complex phase of the received signal generated by a circuit such as a low noise amplifier used during reception Is converted to the amount of rotation of the complex phase of the transmission signal generated by a circuit such as a high power amplifier used at the time of transmission, and the channel information of the reverse link is multiplied by a predetermined calibration coefficient to convert the forward link Channel information can be acquired. In general, these techniques are known as implicit feedback techniques (see, for example, Non-Patent Document 4). Details of the calibration process are shown below.

  In an actual wireless communication apparatus, in signal processing on the transmission side, signal amplification is often performed by a high power amplifier immediately before transmission. In this case, there is an error in the amplification factor due to the individual difference of the high power amplifier, and the complex phase may rotate at a different value for each high power amplifier in the high power amplifier. Similarly, in signal processing on the reception side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, there is an error in the amplification factor due to individual differences of the low noise amplifiers, and the complex phase may rotate with a different value for each low noise amplifier in the low noise amplifier. In some cases, the amplification factor and the amount of phase rotation may have frequency dependency. When the individual difference between the amplification factor and the amount of rotation of the complex phase is so large that it cannot be ignored, it is necessary to perform a calibration process when estimating the downlink channel information from the uplink channel information. Since the error of the amplification factor and the amount of phase rotation is almost stable in time, measure the error of the amplification factor and the amount of phase rotation in advance, and use a coefficient to cancel the influence of the error. Converts link channel information to downlink channel information. In the following description, the description will focus on calibration processing performed on the base station apparatus side having a plurality of antenna elements, but the same can be done on the terminal station apparatus side, and is common to general radio station apparatuses. It is an explanation.

  In the above description, the amplitude and the complex phase may be changed by a high power amplifier or a low noise amplifier (strictly speaking, a transmission system and a reception system circuit including other circuits such as a filter). In this case, it has been described that a calibration coefficient for correcting in accordance with a change in amplitude or complex phase is acquired in advance and used for correction. Although a known technique may be used for the calibration process, an example of the calibration process will be described below.

  FIG. 15 is a diagram illustrating asymmetry of channel information between the uplink and the downlink. In the figure, reference numerals 955-1 to 955-3 denote wireless modules, reference numerals 951-1 to 951-3 denote high power amplifiers (HPA), and reference numerals 952-1 to 952-3 denote low noise amplifiers (LNA). Reference numerals 953-1 to 953-3 denote TDD switches, and reference numerals 954-1 to 954-3 denote antenna elements.

Here, in order to explain the calibration technique, only the functions that affect the channel information in the radio station apparatus are extracted, and thus configurations other than those illustrated are omitted. Therefore, for the convenience of the configuration of the wireless modules 955-1 to 955-3, the high power amplifiers 951-1 to 951-3, the low noise amplifiers 952-1 to 952-3, the TDD switches 953-1 to 953-3, the antenna Although only the elements 954-1 to 954-3 are shown, functions such as an up-converter and a down-converter are mounted in the subsequent stage (previous stage). Further, for example, assuming that a radio station apparatus having a plurality of antennas has a plurality of radio modules 955-1 to 955-2 mounted in one housing, the radio module 955-3 counters this. This corresponds to a diagram in which the radio module 955-3 corresponding to one antenna element of the radio station apparatus that communicates is extracted and described. Further, when the signal passes through each of the high power amplifiers 951-1 to 951-3, the amplitude and the complex phase are Z _{HPA # 1} (f _k ), Z _{HPA # 2} (f _k ), Z _{HPA # 3} (f _k ) It shall change. Further, when the signal passes through the low noise amplifiers 952-1 to 952-3, the amplitude and the complex phase are Z _{LNA # 1} (f _k ), Z _{LNA # 2} (f _k ), and Z _{LNA # 3} (f _k). ) Change. Here, it is assumed that there is a frequency dependency as a general condition, and the frequency “(f _k )” for the k-th frequency component is described.

Here, for example, channel information when signals are transmitted from the wireless module 955-1 and the wireless module 955-2 to the test wireless module 955-3 will be described. Here, the channel information on the space between the antenna element 954-1 of the wireless module 955-1 and the antenna element 954-3 of the wireless module 955-3 is represented by h ₁ (f _k ), and the wireless module 955 -2 channel element information in the space between the antenna element 954-2 of -2 and the antenna element 954-3 of the wireless module 955-3 is represented by h ₂ (f _k ).

At this time, the channel information when the signal is actually transmitted from the wireless module 955-1 to the wireless module 955-3 indicates a change associated with the passage of the high power amplifier 951-1 in h ₁ (f _k ) in space. A coefficient Z _{HPA # 1} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying the passage of the low noise amplifier 952-3 are observed as multiplied values. Similarly, the channel information when transmitting a signal from the wireless module 955-2 to the wireless module 955-3 is a coefficient Z indicating a change associated with the passage of the high power amplifier 951-2 at h ₂ (f _k ) in space. _{HPA # 2} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying the passage through the low noise amplifier 952-3 are observed as multiplied values. Therefore, the channel from the wireless module 955-1 to the wireless module 955-3 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 3} (f _k ). A channel from the wireless module 955-2 to the wireless module 955-3 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 3} (f _k ). For this reason, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 955-1 and the wireless module 955-2 have a relative Z _{HPA # 2} (f _k ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same on the receiving side, and when the signal transmitted from the wireless module 955-3 is received by the wireless module 955-1, the channel information is stored in the high power amplifier 951 at h ₁ (f _k ) in space. Observed as a value obtained by multiplying a coefficient Z _{HPA # 3} (f _k ) indicating a change accompanying the passage of −3 and a coefficient Z _{LNA # 1} (f _k ) indicating a change accompanying the passing of the low noise amplifier 952-1 Is done. Similarly, when the signal transmitted from the wireless module 955-3 is received by the wireless module 955-2, the channel information changes in the space h ₂ (f _k ) with the passage of the high power amplifier 951-3. The coefficient Z _{HPA # 3} (f _k ) shown is multiplied by the coefficient Z _{LNA # 2} (f _k ) showing the change accompanying the passage of the low noise amplifier 952-2. Therefore, the channel from the wireless module 955-3 to the wireless module 955-1 is represented by Z _{HPA # 3} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). A channel from the wireless module 955-3 to the wireless module 955-2 is represented by Z _{HPA # 3} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). For this reason, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 955-1 and the wireless module 955-2 have relatively Z _{LNA # 2} (f _k ) / Z _{LNA # 1} (f _k ) difference occurs.

Here again, the channel information when the radio module 955-1 receives a signal transmitted from the radio module 955-3 corresponding to the reverse link in the left radio station apparatus is h ₁ (f _k ) · Z. _{HPA # 3} (f _k ) · Z _{LNA # 1} (f _k ). The channel information when the signal transmitted from the wireless module 955-1 corresponding to the forward link is received by the wireless module 955-3 is h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA. # 3} (f _k ). The calibration coefficient of the wireless module 955-1 is given by the following equation (4).

Similarly, channel information when the radio module 955-2 receives a signal transmitted from the radio module 955-3 corresponding to the reverse link in the left radio station apparatus is h ₁ (f _k ) · Z _{HPA # 3.} (F _k ) · Z _{LNA # 2} (f _k ). The channel information when the signal transmitted from the wireless module 955-2 corresponding to the forward link is received by the wireless module 955-3 is h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA. # 3} (f _k ). The calibration coefficient of the wireless module 955-2 is given by the following equation (5).

Here, for example, the channel information in the reverse link acquired by the wireless modules 955-1 to 955-2 is h ₁ (f _k ) · Z _{HPA # 3} (f _k ) · Z _{LNA # 1} (f _k ) and h, respectively. ₁ (f _k ) · Z _{HPA # 3} (f _k ) · Z _{LNA # 2} (f _k ), which is multiplied by calibration coefficients C ₁ (f _k ) and C ₂ (f _k ), h ₁ (f _k ) · Z _{HPA # 1} (f _k ) · Z _{LNA # 3} (f _k ) and h ₁ (f _k ) · Z _{HPA # 2} (f _k ) · Z _{LNA # 3} (f _k ) .

  In actual operation, when the wireless module of the actual communication partner is different from the wireless module 955-3, strictly speaking, the estimated value of the forward link channel information obtained by multiplying the calibration coefficient is the forward link. The channel information itself indicates a different value. However, even in that case, the value obtained by multiplying the true forward link channel information regarding the wireless module 955-1 and the wireless module 955-2 by the common coefficient coincides with the above estimated value, and the directivity formation is performed. Considering that there is no effect even if a common constant is multiplied for all antenna elements, there is no problem as feedback of channel information.

  In the above description, the transmitting side of the focused radio station apparatus is described as the forward link, and the receiving side is defined as the reverse link. Is called the downlink and the reverse link is called the uplink. Similarly, when the focused radio station apparatus is a terminal station apparatus or a relay station apparatus, the forward link is usually called an uplink and the reverse link is called a downlink.

Atsushi Suyama, Atsunori Ohara, Shin Yoon, Yukihiko Okumura, “Effects of Analog Beamforming Configuration in Massive MIMO Using High Frequency Band Hybrid Beamforming”, IEICE Technical Report, The Institute of Electronics, Information and Communication Engineers, March 2015 , Vol.114, no.490, RCS2014-337, p.213-218 Atsushi Ohta, Takuto Arai, Hiroshi Shirato, Kazuki Maruta, Yasuhiko Iwakuni, Masataka Iizuka, “Line-of-sight Millimeter-Wave Spatial Multiplexing Technology Using Parallel Transmission in the First Eigenmode: System Proposal and Basic Characteristics Evaluation Results”, IEICE Technical Report, IEICE, August 2015, vol.115, no.181, RCS2015-144, p.73-78 Atsushi Ohta, Hiroshi Shirato, Kazuki Maruta, Takuto Arai, Yasuhiko Iwakuni, Masataka Iizuka, “Efficient use of first eigenmode transmission in Massive MIMO for line-of-sight”, IEICE Technical Report, IEICE, 2015 November, vol.115, no.288, RCS2015-239, p.293-298 Hayato Fukuzono, Tomonori Murakami, Riichi Kudo, Yasushi Takatori, Hayato Mizoguchi, “Experimental evaluation of implicit feedback in downlink multi-user MIMO-OFDM system”, IEICE Technical Report, IEICE, November 2013 , Vol.113, no.301, RCS2013-187, p.79-84

  In the wireless transmission system using the high frequency band such as the millimeter wave band described above, there are problems related to the A / D converter and the D / A converter, as well as the up converter and the down converter, as shown below. To do.

  In the time-axis beamforming technology described in “Expansion of Massive MIMO Technology when Line-of-Sight Wave is Dominant”, the A / D converter, D / A converter, up-converter, and down-converter are the same as the number of antenna elements. I need it. Further, in the configuration example shown in FIG. 13, these are required by the number of antenna elements × the number of streams. In general, A / D converters and D / A converters that are used in an environment where the clock speed is very high due to the wide band tend to increase power consumption. In the case of a device having a large calorific value, a heat radiating plate or the like for radiating the heat is required, and these are physically required to be proportional to the calorific value. In addition to the design requirements of the circuit itself, it is difficult to reduce the size of the A / D converter and the D / A converter itself that regularly consume power because of the necessity of heat dissipation processing for the amount of heat generated during operation. Become larger.

  Originally, in a system that assumes use of the millimeter wave band, the wavelength of the antenna element itself decreases as the frequency increases, and the size of the antenna element itself can be reduced in proportion to the wavelength. For example, the size of the antenna element can be designed to be sufficiently small with respect to the wavelength, and when the antenna elements are arranged at about ½ wavelength, the size of the entire array antenna should be designed to be small even if the number of antenna elements is large. Is possible. For example, assuming the E band and assuming that the center frequency is 80 GHz, one wavelength is 3.75 mm. If a 16 × 16 = 256 element array antenna is formed with a one-wavelength interval and a square array structure, the size of the entire array antenna will be about 6 cm × 6 cm. However, in order to realize mounting at this size, the A / D converter and the D / A converter itself are configured to be within a wavelength of 3.75 mm square or less, and the amount of heat generated during operation is expected. A heat sink for heat dissipation must also be stored in this space. If this is not possible, the actual array antenna size will depend on the size of the A / D converter and D / A converter, the size of the heat sink for heat dissipation, and the like. For example, if a wideband signal having a bandwidth of 1 GHz is handled, the sampling process performed in the baseband must be performed at a super speed of at least 1 GHz. The higher the speed, the higher the power consumption and the greater the amount of heat generated. In the end, miniaturization cannot be realized. As described above, despite the Massive MIMO technology that can be downsized due to the high frequency band, the size cannot be reduced due to the power consumption. For these reasons, reducing the power consumed by the A / D converter and the D / A converter is a major issue.

  Furthermore, in addition to the problem of power consumption, A / D converters and D / A converters that operate at high speed are expensive. Therefore, if the number of antenna elements is in units of 100, it is possible to reduce the cost even further. It is required to reduce the total number of / D converters and D / A converters. The same applies to up-converters and down-converters, and mounting up-converters and down-converters for the number of antenna elements leads to an increase in cost. In addition, the local oscillator input to the up-converter and the down-converter needs to eliminate the uncertainty of the complex phase in order to form directivity, so that it is required to share the local signal. Effects such as loss due to distribution of a huge number of these high-frequency signals for each antenna element and mutual pre-interference from these many signal lines to the signal lines of each system cannot be ignored. In particular, in order to avoid mutual pre-interference between systems, it is ideal to isolate the physical distance of the signals of each system. Leads to

  Therefore, in order to make the array antenna of each radio station apparatus more compact, even when using time-axis beamforming, it is consumed by the A / D converter and D / A converter to be mounted as much as possible. There is a need to reduce power.

  In view of the above circumstances, an object of the present invention is to provide a wireless communication apparatus and a wireless communication method that can be reduced in size and cost.

  One aspect of the present invention is a wireless communication apparatus that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication apparatus. A signal receiving unit that receives a signal transmitted by the wireless communication device via each of the antenna elements used for signal reception, and at least for each of the array antennas used for signal reception, and the signal received by the signal receiving unit Among these, for each received signal received via the antenna elements constituting the corresponding array antenna, a first phase rotation unit that outputs a rotated complex phase on an analog signal, and at least used for signal reception It is provided for each array antenna, and determines whether or not to pass the received signal output from the corresponding first phase rotation unit. When the switching unit that can be switched for each series and the received signal is a reproduction target, all or part of the received signal output from the corresponding first phase rotation unit is passed, and the received signal is Switching in the switching unit so as to pass the received signal output from the corresponding first phase rotation unit one by one for each system of antenna elements when it is a training signal for calculating the rotation amount of the complex phase A switching control unit that controls at least each of the array antennas used for signal reception, a signal synthesis unit that synthesizes the received signals passed by the corresponding switching unit, and at least each of the array antennas used for signal reception A signal converter that converts the received signal synthesized by the corresponding signal synthesizer from an analog signal to a baseband digital signal; A signal reproduction unit that reproduces a signal transmitted by the other wireless communication device based on the reception signal to be reproduced, which is converted by the signal conversion unit, and the switching unit that passes one for each antenna element system. A correlation calculation unit that calculates a correlation for each combination of a reference antenna element in the antenna elements of the training signal and another antenna element using the training signal, and a correlation calculation result by the correlation calculation unit And a first rotation amount calculation unit for calculating a rotation amount of the complex phase to be given in the first phase rotation unit.

  One aspect of the present invention is the above-described wireless communication apparatus, wherein the array antenna is used to generate an analog transmission signal to be transmitted to another wireless communication apparatus, and at least used for signal transmission. A signal distribution unit provided for each array antenna and for branching the analog transmission signal transmitted using the corresponding array antenna in correspondence with each of the antenna elements constituting the array antenna, and at least used for signal transmission A second phase rotation unit provided for each array antenna and rotating a complex phase on an analog signal for each of the transmission signals branched by the corresponding signal distribution unit; and the second phase rotation unit The antenna element that constitutes the array antenna corresponding to the second phase rotation unit, the transmission signal whose phase is rotated by A calculation result of the correlation calculated by the signal calculation unit and the correlation calculation unit, a calculation result of the rotation amount of the complex phase calculated by the first rotation amount calculation unit, or a calibration process on the calculation result And a second rotation amount calculation unit for calculating a rotation amount of the complex phase to be given in the second phase rotation unit based on any of the information on the rotation amount of the complex phase obtained by performing .

  One aspect of the present invention is the above-described wireless communication device, in which the first phase rotation unit and the second phase are replaced with the first phase rotation unit and the second phase rotation unit. A phase rotation unit that performs processing of the rotation unit is provided for each array antenna.

  One aspect of the present invention is the above-described wireless communication device, in which a signal combining / distributing unit that performs processing of the signal combining unit and processing of the signal distributing unit is provided instead of the signal combining unit and the signal distributing unit. A switching unit that is provided for each array antenna and that temporally switches between processing of the signal combining unit and processing of the signal distributing unit in the signal combining and distributing unit is provided for each array antenna.

  One aspect of the present invention is the above-described wireless communication device, wherein the signal synthesis unit and the signal distribution unit are individually independent, and the first phase rotation unit and the second phase rotation unit Are independent.

  One embodiment of the present invention is a wireless communication method performed by a wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with another wireless communication device. A signal receiving unit is provided for each of the array antennas used for signal reception, and at least for each of the array antennas used for signal reception, for receiving signals transmitted by the other wireless communication devices via the antenna elements used for signal reception. For each received signal received through the antenna elements constituting the corresponding array antenna among the signals received in the signal reception process, the phase rotation unit is configured to have a complex phase on the analog signal for each antenna element system. A phase rotation process for rotating the signal, and at least a switching unit provided for each array antenna used for signal reception includes the received signal. When it is a reproduction target, all or part of the reception signal whose complex phase is rotated in the phase rotation process by the corresponding phase rotation unit is passed, and the reception signal is used for calculating the rotation amount of the complex phase. When it is a training signal, a switching process for passing the received signal, which is obtained by rotating the complex phase in the phase rotation process by the corresponding phase rotation unit one by one for each antenna element system, and at least used for signal reception A signal combining unit provided for each array antenna, a signal combining unit configured to combine the received signals passed in the switching step by the corresponding switching unit, and a signal conversion unit provided for each of the array antennas used at least for signal reception The received signal synthesized in the signal synthesis process by the corresponding signal synthesis unit A signal conversion process for converting a signal to a baseband digital signal, and a signal transmitted by the other wireless communication device based on the reception signal to be reproduced, which is converted into a digital signal by the signal reproduction unit in the signal conversion process A signal reproduction process for reproducing the reference signal, and a correlation calculation unit using the training signal that has been passed one by one for each antenna element system in the switching process, and a reference antenna element in the antenna element of the training signal and the like. The correlation calculation process for calculating the correlation for each combination with the antenna element and the rotation amount calculation unit calculate the rotation amount of the complex phase to be given by the phase rotation unit based on the correlation calculation result in the correlation calculation process A rotation amount calculating process.

  According to the present invention, it is possible to reduce the size and cost of a wireless communication device.

It is a functional block diagram which shows the structural example of the radio station apparatus in the 1st Embodiment of this invention. It is a functional block diagram which shows the structural example of the radio station apparatus in the embodiment. It is a figure which shows the structural example of the communication system in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the radio station apparatus in 2nd Embodiment. It is a functional block diagram which shows the structural example of the radio station apparatus in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the transmission / reception signal processing circuit in the embodiment. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a functional block diagram which shows the structural example of the radio station apparatus in a prior art. It is a figure which shows the calibration process in a prior art.

  Embodiments of the present invention will be described in detail with reference to the drawings. First, the basic principle of the embodiment of the present invention will be described. The terms “time axis” and “frequency axis” used in this specification are sometimes expressed as “time domain” and “frequency domain”, but here they are unified as “time domain” and “frequency domain”. Give an explanation.

[Overview of basic principles]
In the time-axis beamforming technique, for example, at the time of reception, a main path of an incoming wave between transmitting and receiving stations is extracted, and signal processing for directing the directivity gain of the antenna element group in that direction is performed. The transmission / reception weight used at that time is a constant having no frequency dependence, and as a result, signal processing is reduced in various respects. However, since it is based on digital signal processing, each antenna element is multiplied by a transmission / reception weight on the individual digital signal processing, and accompanying this, individual A / D (analog / digital) conversion is performed for each antenna system. And a D / A (digital / analog) converter.

  However, if the multiplication processing of the coefficient having no frequency dependency is limited to, for example, the complex phase rotation processing without amplitude change, digital signal processing is not necessarily required. Specifically, by using a phase shifter that is an analog circuit and passing the analog signal through this phase shifter set to a predetermined complex phase rotation, signal processing substantially equivalent to weight multiplication processing is performed. Can be realized. In Non-Patent Document 1, directivity control is realized using a phase shifter. For example, the phase rotation amount of each antenna element is set for each predetermined direction set in increments of 5 degrees in the horizontal / vertical direction. A combination set was determined in advance, and the phase rotation amount of each phase shifter was set according to the direction in which the beam obtained by some control procedure should be directed. However, this phase rotation amount combination set is selected from a pre-set menu, and it is necessary to search for a direction with the highest reception level while transmitting a training signal individually for each direction. . However, in time-axis beamforming, the phase rotation amount of each antenna element is optimized based on the training signal transmitted from the terminal station apparatus side, so the calculation of the phase rotation amount used for directivity formation is simply fed back. In addition, the combination set of complex phase rotation amounts can be set with a much higher degree of freedom without requiring a prior menu or the like.

  Therefore, the wireless station device (wireless communication device) in the embodiment of the present invention employs digital assist type analog beam forming. That is, the radio station apparatus performs the calculation process of the complex phase rotation amount performed by each phase shifter by digital signal processing, and controls the phase shifter using the complex phase value obtained by the digital signal processing. By doing so, the desired complex phase is rotated to form directivity on the analog signal.

Here, the process of calculating the complex phase of the transmission / reception weight of the time-axis beamforming performed by digital signal processing does not necessarily have to be performed simultaneously by each antenna element. Or, even when digital signal processing is performed when calculating the complex phase of the transmission / reception weight of time-axis beamforming, the time rate for performing signal processing for this is the same as that of normal communication for transmitting / receiving user data. It is expected to be overwhelmingly less than the time rate. Therefore, the radio station apparatus according to the present embodiment performs the operations of these digital circuits in a limited time.
Hereinafter, a detailed embodiment to which the basic principle is applied will be described with reference to the drawings.

[First Embodiment]
FIG. 1 is a functional block diagram showing a configuration example (subarray separation type) of a radio station apparatus 450 in the present embodiment. In this figure, the same parts as those of the conventional radio station apparatus shown in FIGS. 12 to 14 are denoted by the same reference numerals. In the present embodiment, as corresponding to FIGS. 12 and 13 of the prior art, a “sub-array separation type” configuration in which a directional beam is formed by being separated into a plurality of sub-arrays, and a plurality of directional beams in one array. There is a variation of the configuration by “sub-array shared type” (which may be understood as “integrated array” because it is not separated into sub-arrays strictly), but here “sub-array separated type” Give an explanation.

  The radio station apparatus 450 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N (N is an integer of 1 or more), and a control circuit 460. The baseband signal processing circuit 140 includes modulators 120-1 to 120-N (MOD # 1 to MOD # N), a signal separation circuit 141, and demodulators 130-1 to 130-N (DEM # 1 to DEM #). N). The transmission / reception signal processing circuit 451-n (n = 1,..., N) includes a D / A converter 122-n, an up converter (UC) 123-n, a down converter (DC) 124-n, an A / D converter 125-n, TDD (Time Division Duplex) switch (TDD-SW) 127-n, phase shifters 402-n-1 to 402-n-M (M is 2 or more) Integer), switches 403-n-1 to 403-n-M, distribution coupler (HYB) 404-n, correlation calculation circuit 405-n, phase shift control circuit 406-n, down converter 424- n-1 to 424-n-M and A / D converters 425-n-1 to 425-n-M. The phase shifters 402-n-1 to 402-n-M are connected to the antenna elements 401-n-1 to 401-n-M, respectively. The D / A converters 122-1 to 122-N, the up converters 123-1 to 123-N, the down converters 124-1 to 124-N, and the A / D converters 125-1 to 125-N are respectively shown in FIG. 12 and the D / A converters 922-1-1 to 922-NM, the up converters 923-1-1 to 923-NM, and the down converters 924-1-1 to 924-NM. The same A / D converters 925-1-1 to 925 -N-M can be used.

  Here, in the down converters 424-1-1 to 424 -N-M, in order to perform frequency conversion between a radio frequency signal and a baseband signal, it is necessary to input a signal from a local oscillator. That is, a common local signal is used for the combination of the down converters 424-n-1 to 424-n-M in the same transmission / reception signal processing circuit 451-n (n = 1,..., N), and each down converter 424- It is necessary to suppress temporal changes in the relative relationship of the complex phases in n-1 to 424-n-M. In this sense, a configuration in which a local oscillator exists substantially outside the down converters 424-1-1 to 424-NM is taken. However, since the description is complicated, an external local is used here as a simple description. Oscillator specification is omitted. It is necessary to share the local oscillator used between the same n down-converters 424-n-1 to 424-n-M, but down-converters 424-n'-1 to 424-having different values of n. In combination with n′-M, the local oscillator to be used need not be shared. Further, these local oscillators do not need to be shared between the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. The only important thing is to make the local signal common between the signal sequences that implement the directivity formation in a coordinated manner.

Here, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and the radio station apparatus 450 has a total of N transmission / reception signal processing circuits 451-1 to 451-N mounted therein. M represents the number of antenna elements of the subarray mounted on each of the transmission / reception signal processing circuits 451-1 to 451-N. In the description of FIG. 11, the total number of antenna elements is M ₀ , but here the whole antenna has a subarray configuration as in FIG. 12, and therefore the number of antenna elements in each subarray is indicated as a different value M. Each of the transmission / reception signal processing circuits 451-n is associated with sub-array antenna elements 401-n-1 to 401-n-M, and the transmission / reception signal processing circuits 451-1 to 451-N are described in Non-Patent Document 2. It is assumed that they are installed spatially apart. The baseband signal processing circuit 140 is connected to the transmission / reception signal processing circuits 451-1 to 451-N by wire, and a digital baseband signal is transferred over the wire. Although not shown in FIG. 11, the wireless station device 450 includes an overall control circuit 460. In the figure, an example is shown in which the radio station device 450 mounts the control circuit 460 on the baseband signal processing circuit 140. The control circuit 460 manages the frame period and transmission / reception timing, and also manages the switching of the TDD switches 127-1 to 127 -N. The control circuit 460 determines whether to connect the distribution coupler 404-n to the up-converter 123-n or to the down-converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by dividing.

  Furthermore, in this embodiment, the estimation process of the complex phase rotation amount basically performed by the variable phase shifter corresponding to the transmission / reception weight is performed by digital signal processing, and the actual complex phase rotation processing is realized by analog signal processing. . For this reason, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N are premised on signal processing on the frequency axis like the OFDM (Orthogonal Frequency Division Multiplexing) modulation method. Even in such a case, even when signal processing on the time axis is assumed as in SC-FDE (Single-Carrier Frequency Domain Equalization), it is possible to cope with either method. However, the signal separation circuit 141 performs signal separation between each signal series, but here it is also possible to perform signal separation on the time axis, and once convert it to a frequency axis signal by FFT, Frequency-dependent signal separation may be performed. In this sense, the input to the demodulators 130-1 to 130-N is assumed to be a time-axis signal or a frequency-axis signal. Will be described as input. In this way, the concept regarding variations of communication systems such as the OFDM modulation system and SC-FDE is the same in the following description.

The specific signal flow is as follows.
First, signal transmission will be described. In the wireless station device 450, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmits the signal with the up-converter 123-n and the distribution coupler 404-n connected (n = 1,..., N).

  Each of the modulators 120-1 to 120-N generates a time-axis digital baseband transmission signal of each stream to be spatially multiplexed and inputs the transmission signals to the transmission / reception signal processing circuits 451-1 to 451-N. The transmission / reception signal processing circuit 451-n (n = 1,..., N) receives the time base digital baseband transmission signal generated by the modulator 120-n. The D / A converter 122-n of the transmission / reception signal processing circuit 451-n (n = 1,..., N) converts the transmission signal input from the modulator 120-n into an analog baseband signal, and improves it. Input to converter 123-n. The up-converter 123-n converts the signal input from the D / A converter 122-n from a baseband signal to a radio frequency band signal and inputs the signal to the TDD switch 127-n. The TDD switch 127-n inputs the signal input from the up-converter 123-n to the distribution coupler 404-n.

  The distribution coupler 404-n branches the analog signal input from the TDD switch 127-n into M analog signals. The branched analog signals are input to the phase shifters 402-n-1 to 402-n-M via the switches 403-n-1 to 403-n-M. The phase shifters 402-n-1 to 402-n-M apply a predetermined complex phase rotation on the analog signal. The analog signals to which complex phase rotation has been applied by the phase shifters 402-n-1 to 402-n-M are transmitted via the antenna elements 401-n-1 to 401-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 402-n-1 to 402-n -M. For example, the antenna elements 401-1-1 to 401-1 -M and the antenna elements 401-N-1 to 401 -N-M are individually formed with directivity, and the radio station apparatus in the directivity direction Communicate.

  Next, signal reception will be described. In the wireless station device 450, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receives a signal in a state where the distribution coupler 404-n and the down converter 124-n are connected (n = 1,..., N).

  Signals received by the antenna elements 401-n-1 to 401-n-M (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and passes through the switches 403-n-1 to 403-n-M. To the distribution coupler 404-n. The distribution coupler 404-n synthesizes the signals of the respective antenna systems input via the switches 403-n-1 to 403-n-M on analog signals, and the synthesized signal via the TDD switch 127-n. To the down converter 124-n. The down converter 124-n converts a radio frequency signal input via the TDD switch 127-n into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the signal input from the down converter 124-n from an analog baseband signal to a digital baseband signal and inputs the signal to the signal separation circuit 141.

  The signal separation circuit 141 performs signal separation by suppressing crosstalk components between the streams, and inputs the separated signals to the corresponding demodulators 130-1 to 130-N. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted into a frequency axis signal by FFT processing and performed on the frequency axis. When the crosstalk component is suppressed on the time axis, first, the correlation between signal sequences input to the signal separation circuit 141 is calculated based on the digital baseband signal corresponding to the received training signal. . Then, based on a MIMO (Multiple Input Multiple Output) channel matrix given by the calculated correlation, a general MIMO signal separation process such as ZF (Zero Forcing) type or MMSE (Maximum Mean Square Error) type signal separation is performed. A matrix similar to the above may be calculated, and this matrix may be multiplied by the sampling data unit by regarding the signal sequence input to the signal separation circuit 141 as a vector. In general MIMO signal separation processing such as ZF type and MMSE type signal separation on the general frequency axis, a different matrix is used for each frequency component, whereas almost the same matrix on the frequency axis. Is used, signal separation processing can be performed on the time axis in units of sampling data. Alternatively, only signal processing performed by the transmission / reception signal processing circuits 451-1 to 451-N is sufficient, and the signal separation circuit 141 does not need to perform any processing. However, in any case, details of the signal separation method here are omitted because they are not directly related to the features of the present embodiment. Demodulators 130-1 to 130-N demodulate the signal in which the crosstalk component is suppressed in signal separation circuit 141.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402 -N-M will be described. In this signal processing, the switches 403-n-1 to 403-n-M (n = 1,..., N) are shifted by the phase shifters 402-n-1 to 402-n-M and the down converters 424-n-1 to 424-n-1. 424-n-M is connected. These switch switchings are managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. The phase shifters 402-n-1 to 402-n -M (n = 1,..., N) are connected to the distribution coupler 404-n except when calculating the rotation amount of the complex phase. Further, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402 -NM are set to a predetermined value. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference with respect to the initial predetermined value. For example, in the most obvious example, the phase shifters 402-1-1 to 402-N-M may be set to all zero (or all the same value), and in this case, the rotation amount of the obtained complex phase May be used as the phase rotation amount of the phase shifters 402-1-1 to 402 -NM at the time of subsequent communication. Or, the initial predetermined values of the phase shifters 402-1-1-1 to 402-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the complex phase rotation amount is + α degrees. If it is + β degrees, + γ degrees,..., The phase rotation amounts of the phase shifters 402-1-1-402-NM during the subsequent communication are + (α + 10) degrees, + (β + 20) degrees. , + (Γ + 30) degrees, and so on.

  As an actual process, first, a radio station apparatus of a communication partner that should acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and the radio station apparatus 450 receives this training signal. Signals received by the antenna elements 401-n-1 to 401-n-M (n = 1,..., N) are input to the phase shifters 402-n-1 to 402-n-M, respectively. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and passes through the switches 403-n-1 to 403-n-M. To the down converters 424-n-1 to 424-n-M. Each of the down converters 424-n-1 to 424-n-M converts the input radio frequency signal into an analog baseband signal and supplies the analog baseband signal to the A / D converters 425-n-1 to 425-n-M. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal, and outputs it to the correlation calculation circuit 405-n.

  Correlation calculation circuits 405-1 to 405-N calculate the rotation amount of the complex phase using equations (1) to (3), respectively. In addition, when the calibration calculation circuits 405-1 to 405-N require calibration processing as necessary, the complex phase on the transmission side is set as a value in consideration of the calibration coefficient in the equations (1) to (3). Determine the amount of rotation. The amount of rotation of the complex phase obtained by the correlation calculation circuit 405-n (n = 1,..., N) is the identification number of the wireless station device with which it communicates (when communicating with a plurality of wireless station devices. Together with a single wireless station apparatus in the P-P (point-to-point) type, the identification number is not necessary.) And input to the phase shift control circuit 406-n (n = 1,..., N) Is done. The phase shift control circuit 406-n stores the complex phase rotation amount to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the radio station apparatus of the communication partner. Manage it by memorizing it.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 460 grasps a wireless station device as a communication partner, and the phase shift control circuit 406-n (n = 1,... , N) to instruct the phase shifters 402-n-1 to 402-n-M to set the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication. The phase shift control circuit 406-n acquires the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication by reading from the memory or the like, and acquires the rotation amount of the complex phase by the phase shifters 402-n-1 to 402. Set to -n-M to realize beamforming on analog.

Although not clearly shown in the figure, for example, if a high-power amplifier (HPA) on the transmission side is arranged, it is arranged at a point described as “A” in the figure, and a low-noise amplifier on the reception side ( LNA) and the like are arranged at points described as “B” and “C _1-1 ” to “C _N-M ” in the drawing. Since the points described as “A” and “B” are common within the same transmission / reception signal processing circuits 451-1 to 451-N, the complex phases of the individual high power amplifier and low noise amplifier are uncertain. The calibration process for removing the property is unnecessary.

On the other hand, regarding the low-noise amplifiers described as “C _1-1 ” to “C _N-M ”, when the amount of rotation of the complex phase can fluctuate with time, the same transmission / reception signal processing circuit 451-1. 1 to 451 -N can cause complex phase uncertainty between the antenna elements 401-1-1 to 401 -N-M, and therefore, as in the case of the conventional implicit feedback calibration method. Therefore, it is necessary to remove the uncertainty of the complex phase of the low-noise amplifier of each system. Note that this embodiment can be applied to an arbitrary method, and a specific method of calibration processing is not limited. Considering this calibration result, for example, if the amount of rotation of the complex phase in each of “C _1-1 ”, “C _1-2 ”, and “C _1-3 ” is +10 degrees, +20 degrees, and +30 degrees, The phase rotation amount is adjusted by performing corrections of −10 degrees, −20 degrees, and −30 degrees for the complex phase rotation amounts obtained by the equations (1) to (3), respectively. The calibration result information is collected by a calibration circuit (not shown), and the phase shift control circuits 406-1 to 406-N or correlation calculation circuits 405-1 to 405-N store this information. To make corrections.

  In addition, since the transmission / reception signal processing circuits 451-1 to 451-N having a subarray configuration can be physically separated to reduce the correlation of directional beams formed on analog, generally, Install at a distance greater than a predetermined distance.

  Further, the same applies to all the following descriptions (including other embodiments). In FIG. 1, the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N are connected by wire. In this configuration, digital baseband signals are transmitted and received over the wire. The D / A converters 122-1 to 122-N and the A / D converters 125-1 to 125-N are connected to the baseband signal processing circuit 140. In the case of mounting, a signal flowing on a wired connection between the baseband signal processing circuit 140 and the transmission / reception signal processing circuits 451-1 to 451-N can be an analog baseband signal.

  Next, similarly to FIG. 13 for FIG. 12 in the prior art, the radio station apparatus of this embodiment can also be configured by a “subarray shared type” that realizes a plurality of directional beam formations with a single array antenna. is there. This configuration is shown in FIG.

  FIG. 2 is a functional block diagram showing a configuration example (subarray shared type) of the radio station apparatus 452 in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The radio station apparatus 452 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 451-1 to 451-N, distribution couplers (HYB) 407-1 to 407-M, and an antenna element 401-. 1-401-M and a control circuit 460. The distribution coupler (HYB) 407-m (m = 1,..., M) is a transmission / reception signal processing circuit 451-1, 451-2,. 2-m,..., 402-Nm, and antenna elements 401-m (m = 1,..., M).

  In radio station apparatus 452, as with radio station apparatus 450 shown in FIG. 1, down converters 424-1-1 to 424-NM perform frequency conversion between radio frequency signals and baseband signals. Therefore, input of a signal from the local oscillator is required. A common local signal is used for each combination of down converters 424-n-1 to 424-n-M (n = 1,..., N) in each of the transmission / reception signal processing circuits 451-1 to 451-N. It is necessary to suppress the time change of the relative relationship of the complex phase in the down converter. In this sense, a configuration in which a local oscillator exists substantially outside the transmission / reception signal processing circuits 451-1 to 451-N is taken. However, since the description is complicated, the description of the external local oscillator is used here as a simple description. The description is omitted. It is not necessary to share the local oscillator used in each of the transmission / reception signal processing circuits 451-1 to 451-N. Further, it is not necessary to share these local oscillators with the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. The only important thing is to make the local signal common between the signal sequences that implement the directivity formation in a coordinated manner. However, unlike the case of FIG. 1, the down converters 424-1-1 to 424-NM are all housed in the same housing, so that all local signals can be shared in the case of FIG. Is possible.

  Similar to the above description, N corresponds to the number of multiplexing (the number of streams) when performing spatial multiplexing, and M represents the number of antenna elements of a common array antenna. In radio station apparatus 450 shown in FIG. 1, transmission / reception signal processing circuits 451-1 to 451-N are mounted in physically different housings for each system. On the other hand, in the radio station apparatus 452 shown in FIG. 2, all the N systems of the transmission / reception signal processing circuits 451-1 to 451-N are mounted in one casing. Further, in the radio station apparatus 452, the transmission / reception signal processing circuits 451-1 to 451-N share the antenna elements 401-1 to 401-M via the distribution couplers 407-1 to 407-M. Here, in the radio station apparatus 450 shown in FIG. 1 and the radio station apparatus 452 shown in FIG. 2, the internal processing in the transmission / reception signal processing circuits 451-1 to 451-N is the same. Further, FIG. 2 shows a case where the control circuit 460 is mounted on the baseband signal processing circuit 140, but the control circuit 460 may be mounted at an arbitrary location in the wireless station device 452. The control circuit 460 manages the frame period, transmission / reception timing, and switching of the TDD switches 127-1 to 127 -N.

The following shows a specific signal flow in the radio station apparatus 452, focusing on the difference from the radio station apparatus 450 shown in FIG.
First, signal transmission will be described. In the wireless station device 452, the switches 403-n-1 to 403-n-M connect the distribution coupler 404-n and the phase shifters 402-n-1 to 402-n-M, and the TDD switch 127-n. Transmits the signal with the up-converter 123-n and the distribution coupler 404-n connected (n = 1,..., N). This point is common to the radio station apparatus 450 shown in FIG.

  Each of the modulators 120-1 to 120-N generates a time-base digital baseband transmission signal of each stream for spatial multiplexing, and inputs the generated signals to the transmission / reception signal processing circuits 451-1 to 451-N. Radio frequency analog signals that have undergone processing for directivity formation are output from phase shifters 402-1-1 to 402 -N-M by processing similar to that of radio station apparatus 450 shown in FIG. 1. The phase shifters 402-1-1 through 402 -N-M send the signals of these systems to the distribution couplers 407-1 through 407 -M connected to the corresponding antenna elements 401-1 through 401 -M. input. That is, the phase shifters 402-1-m, 402-2-m,..., 402-Nm (m = 1,..., M) input signals to the distribution coupler 407-m. Distribution couplers 407-1 to 407 -M synthesize the input signals, and the combined signals are transmitted via antenna elements 401-1 to 401 -M.

  Next, signal reception will be described. In the radio station apparatus 452, the switches 403-n-1 to 403-n-M connect the phase shifters 402-n-1 to 402-n-M and the distribution coupler 404-n, and the TDD switch 127-n. Receives a signal in a state where the distribution coupler 404-n and the down converter 124-n are connected (n = 1,..., N). This point is common to the radio station apparatus 450 shown in FIG.

  Signals received by the antenna elements 401-1 to 401-M are input to the distribution couplers 407-1 to 407-M, respectively. Distribution coupler 407-m (m = 1,..., M) distributes the input signal to N systems, and phase shifters 402-1-m, 402-2-m,..., 402-N-m. To enter. The transmission / reception signal processing circuits 451-1 to 451-N perform signal processing in the radio station device 450 shown in FIG. 1 on the signals input to the phase shifters 402-1-1-1 to 402-NM as described above. Similar signal processing is performed, and a digital baseband signal is input to the signal separation circuit 141. The signal separation circuit 141 performs signal separation by suppressing the crosstalk component, and the demodulators 130-1 to 130-N demodulate the separated signals.

  For signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402 -N-M, the transmission / reception signal processing circuits 451-1 to 451-N are implemented in FIGS. The signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. 1 is the same as the signal processing in the transmission / reception signal processing circuits 451-1 to 451-N in FIG. The description is omitted here.

  FIG. 3 is a diagram illustrating a configuration example of a communication system in the present embodiment. The communication system shown in the figure includes a radio station apparatus 452 shown in FIG. 2 and a radio station apparatus 450 shown in FIG. In radio station apparatus 450, each of a plurality of transmission / reception signal processing circuits 451-1 to 451-N forms one beam as a subarray. On the other hand, in the radio station apparatus 452, the (sub) array is shared, and one array antenna forms a plurality of beams. In actual operation, as shown in FIG. 3, the wireless station device 450 faces the wireless station device 452, so that N systems of signals can be spatially multiplexed.

  Next, another configuration example of the transmission / reception signal processing circuit of this embodiment will be described with reference to FIG. The transmission / reception signal processing circuit illustrated in FIG. 4 can be replaced with the transmission / reception signal processing circuits 451-1 to 451-N included in the wireless station device 450 illustrated in FIG. 1 or the wireless station device 452 illustrated in FIG. Below, the case where it replaces with the transmission-and-reception signal processing circuits 451-1 to 451-N with which the radio station apparatus 450 is provided is demonstrated to an example. Note that when the transmission / reception signal processing circuits 451-1 to 451-N included in the wireless station device 452 are replaced, the antenna elements 401-n-1 to 401-n-M in FIG. It becomes.

  FIG. 4 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 453-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The transmission / reception signal processing circuit 453-n shown in the figure includes a D / A converter 122-n, an up converter (UC) 123-n, a distribution coupler 414-n, and a phase shifter 409-n-1. 409-n-M, TDD switches 408-n-1 to 408-n-M, phase shifters 402-n-1 to 402-n-M, and switches 403-n-1 to 403-n-M Distribution coupler 415-n, down converter (DC) 124-n, A / D converter 125-n, down converter (DC) 424-n-1 to 424-n-M, A / D D converters 425-n-1 to 425-n-M, a correlation calculation circuit 405-n, and a phase shift control circuit 406-n are provided. The TDD switches 408-n-1 to 408-n-M are connected to the antenna elements 401-n-1 to 401-n-M, respectively. Here, as in FIG. 1, the specification of the external local oscillator is omitted.

  The difference between the transmission / reception signal processing circuit 453-n and the transmission / reception signal processing circuits 451-1 to 451-N shown in FIG. 1 or 2 is that the TDD switches 408-n-1 to 408-n-M are the antenna elements 401. -N-1 to 401-n-M are arranged in the immediate vicinity, and as a result, the path from the up-converter 123-n to the antenna elements 401-n-1 to 401-n-M in the transmission system and the antenna in the reception system The path from the elements 401-n-1 to 401-n-M to the down converter 124-n is physically separated.

In the case of the transmission / reception signal processing circuit 451-n shown in FIG. 1 or FIG. 2, for example, the high-power amplifier (or power amplifier) of the transmission system is arranged at the location described as “A” after the up-converter 123-n. The low-noise amplifier of the reception system is a place described as “B” in the previous stage of the down converter 124-n (in addition, “C _n-1 ” in the previous stage of the down converters 424-n-1 to 424-n-M, "C _n-2", ..., which is arranged in "C _n-M" and described location), thereby enabling shared by transmission and reception circuits from the TDD switch 127-n to the antenna terminal. This point is greatly different from the transmission / reception signal processing circuit 453-n. The merit of adopting the configuration as shown in FIG. 4 is that high power amplifiers and low noise amplifiers can be mounted as many as the number of antenna elements, and as a result, the total transmission power is improved, and the TDD switch in FIGS. It is possible to suppress the effects of insertion loss and distribution and coupling loss of various circuits between 127-n and antenna elements 401-n-1 to 401-n-M. Therefore, although not shown here, in the transmission and reception signal processing circuit 453-n, "D _1", "D _2", ..., high-power amplifier location as "D _M" is "E It is preferable that a low noise amplifier is disposed at a place where “ ₁ ”, “E ₂ ”,..., “E _M ” is described. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is placed at a location marked “A” and a low-noise amplifier is placed at a location marked “B”, then “D ₁ ”, “D ₂ ”,. _M "location and" E ₁ described as "," E ₂ ", ..., high-power amplifier in each location described as" E _M ", not necessarily lead to a low-noise amplifier is disposed.

The following shows a specific signal flow in the transmission / reception signal processing circuit 453-n focusing on the difference.
First, signal transmission will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M include antenna elements 401-n-1 to 401-n-M and phase shifters 409-n-1 to 409-n-. Signals are transmitted with M connected to each other.

  The transmission / reception signal processing circuit 453-n receives a time base digital baseband transmission signal of one stream from a modulator 120-n not shown here. The D / A converter 122-n converts the input transmission signal into an analog baseband signal and inputs the analog baseband signal to the upconverter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a radio frequency band signal and inputs the signal to the distribution coupler 414-n.

  Distribution coupler 414-n branches the analog signal in the radio frequency band input from up-converter 123-n into M-system analog signals and inputs them to phase shifters 409-n-1 to 409-n-M. . Each of the phase shifters 409-n-1 to 409-n-M applies a predetermined complex phase rotation on the analog signal to the signal input from the distributor / coupler 414-n, to thereby generate a TDD switch 408-n-1. Are input to the antenna elements 401-n-1 to 401-n-M through .about.408-n-M. The antenna elements 401-n-1 to 401-n-M transmit the input transmission signal. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 409-n-1 to 409-n-M.

  Next, signal reception will be described. In the transmission / reception signal processing circuit 453-n, the TDD switches 408-n-1 to 408-n-M are connected to the antenna elements 401-n-1 to 401-n-M and the phase shifters 402-n-1 to 402-n-. M is connected, and the switches 403-n-1 to 403-n-M receive signals in a state where the phase shifters 402-n-1 to 402-n-M and the distribution coupler 415-n are connected. Do.

  The signals received by the antenna elements 401-n-1 to 401-n-M are respectively phase shifters 402-n-1 to 402-n-M via the TDD switches 408-n-1 to 408-n-M. Is input. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation on the analog signal to the input signal, and switches 403-n-1 to 403-n-M To the distribution coupler 415-n. Distribution coupler 415-n synthesizes the signals of each antenna system on an analog signal and inputs the resultant signal to down converter 124-n. The down converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal. The A / D converter 125-n inputs the converted digital baseband signal to the signal separation circuit 141 in the baseband signal processing circuit 140 not shown here. The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 402-1-1 to 402 -N-M will be described. The transmission / reception signal processing circuit 453-n performs this signal processing with the switches 403-n-1 to 403-n-M and the phase shifters 402-n-1 to 402-nM and the down converters 424-n-1 to 403-n-1. 424-n-M is connected. These switch switchings are managed by the correlation calculation circuits 405-1 to 405-N under the instruction of the control circuit 460. The phase shifters 402-n-1 to 402-n -M are connected to the distribution coupler 415-n except when calculating the rotation amount of the complex phase. Further, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amounts of the phase shifters 402-1-1 to 402 -NM are set to a predetermined value. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference with respect to the initial predetermined value, as described above. The transmission / reception signal processing circuits 453-1 to 453 -N perform the same processing all at once under the management of the control circuit 460.

  As an actual process, first, a radio station apparatus of a communication partner that should acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and includes radio signal processing circuits 453-1 to 453-N. Receives this training signal. The signals received by the antenna elements 401-n-1 to 401-n-M are respectively phase shifters 402-n-1 to 402-n-M via the TDD switches 408-n-1 to 408-n-M. Is input. Each of the phase shifters 402-n-1 to 402-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and passes through the switches 403-n-1 to 403-n-M. To the down converters 424-n-1 to 424-n-M. Each of the down converters 424-n-1 to 424-n-M converts the input radio frequency signal into an analog baseband signal and supplies the analog baseband signal to the A / D converters 425-n-1 to 425-n-M. input. Each of the A / D converters 425-n-1 to 425-n-M converts the input signal from an analog signal to a digital baseband signal, and outputs it to the correlation calculation circuit 405-n.

  The correlation calculation circuit 405-n calculates the amount of rotation of the complex phase as described in FIG. The rotation amount of the complex phase obtained by the correlation calculation circuit 405-n is input to the phase shift control circuit 406-n together with the identification number of the radio station apparatus with which communication is performed. The phase shift control circuit 406-n stores the complex phase rotation amount to be set in each of the phase shifters 402-n-1 to 402-n-M in the memory in association with the identification number of the radio station apparatus of the communication partner. Manage it by memorizing it.

The amount of rotation of the complex phase described above relates to the amount of phase rotation of the phase shifters 402-n-1 to 402-n-M in the reception system, but the amount of complex phase rotation in a low noise amplifier, a high power amplifier, and the like. In order to cancel the individual difference, the correlation calculation circuit 405-n performs calibration processing in the implicit feedback of the prior art, and performs correction in the transmission system by correction corresponding to the calibration processing based on the amount of complex phase rotation in the reception system. The amount of rotation of the complex phase is converted to a value set in the phase shifters 409-n-1 to 409-n-M. However, when a low noise amplifier is arranged at a location described as “E ₁ ”, “E ₂ ”,..., “E _M ”, a sufficient reception level can be obtained, so “C ₁ ”, “C ₂ ”. ,..., “C _M ” does not require a low-noise amplifier. Therefore, the rotation of the complex phase in each low-noise amplifier of the reception system does not need to be distinguished from the rotation of the complex phase in space. The complex phase rotation amount calculated by the correlation calculation circuit 405-n can be used as it is as the complex phase rotation amount set in the phase shifters 402-n-1 to 402-n-M of the system. The phase shift control circuit 406-n is a complex in the transmission system to be set in each of the phase shifters 402-n-1 to 402-n-M and the phase shifters 409-n-1 to 409-n-M. The amount of phase rotation is managed by associating it with the identification number of the radio station apparatus of the communication partner and storing it in the memory in the same manner.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 460 grasps a wireless station device that is a communication partner and communicates with the phase shift control circuit 406-n. Instructing the phase shifters 402-n-1 to 402-n -M and the phase shifters 409-n-1 to 409-n-M to set the rotation amount of the complex phase corresponding to the radio station apparatus to be performed. The phase shift control circuit 406-n acquires the rotation amount of the complex phase in each of the reception system and the transmission system corresponding to the wireless station apparatus that performs communication by reading from the memory. The phase shift control circuit 406-n sets the amount of rotation of the complex phase of the reception system to the phase shifters 402-n-1 to 402-n-M, and sets the amount of rotation of the complex phase of the transmission system to the phase shifter 409. -N-1 to 409-n-M is set to realize analog beamforming.

  When performing digital beam forming in Massive MIMO, the conventional radio station apparatus requires a large number of A / D converters and D / A converters corresponding to the number of signal sequences, which increases power consumption. Had. On the other hand, the above-described radio station apparatus of the present embodiment uses an A / D converter for each antenna element corresponding to a signal sequence only when calculating the amount of rotation of a complex phase for performing directivity control, and data At the time of transmission, analog beam forming using a phase shifter is performed. Therefore, the radio station apparatus according to the present embodiment uses a switch, when receiving a training signal used for calculating the amount of rotation of the complex phase, the output destination of the received signal to the A / D converter, and when receiving data, the output destination of the received signal To the receiving circuit. As a result, at the time of data transmission / reception, it becomes possible to stop the operation of the A / D converter and the D / A converter for each antenna element required for calculating the amount of rotation of the complex phase, thereby reducing power consumption. It becomes possible.

  As described above, according to the present embodiment, the radio station apparatus uses the A / D converter for each antenna element only when calculating the rotation amount of the complex phase for performing directivity control. The radio station apparatus substitutes analog beam forming using a phase shifter instead of digital beam forming at the time of actual signal transmission. On the other hand, when receiving a signal, the radio station device uses a switch to set the output destination of the signal from the receiving antenna to the A / D converter when receiving the training signal, and to the receiving circuit when performing signal synthesis and demodulation processing. Switch. As a result, in Massive MIMO, the number of A / D converters and D / A converters, which are conventionally required constantly when performing digital beam forming, can be greatly reduced. Therefore, a situation in which a large number of A / D converters and D / A converters continue to operate constantly during communication is avoided, and a huge amount of A / D converters and D / A conversions that have been required particularly during broadband communication are avoided. It is possible to reduce the power consumption and reduce the amount of heat generated. Furthermore, since the A / D converter and the D / A converter do not require a large heat sink that is necessary for heat generation, not only low power consumption but also miniaturization of the radio station apparatus is achieved. It is possible to reduce the cost.

[Second Embodiment]
In the first embodiment, separate down converters and A / D converters are mounted on all (transmitting) and receiving antennas, but these circuits are used only for channel estimation. This works effectively to reduce overall power consumption and suppress the amount of heat generated, but from the viewpoint of miniaturization of the device, a circuit that is only temporarily used is mounted by a huge number of antenna elements. Is inefficient. In particular, A / D converters that operate at a very high speed due to the wide bandwidth tend to be expensive in price per unit, which may lead to an increase in the price of the entire apparatus. In the second embodiment, a configuration for consolidating these down converters, A / D converters, and the like into one system is shown.

  FIG. 5 is a functional block diagram showing a configuration example (subarray separation type) of the wireless station device 550 in the present embodiment. Similarly to the first embodiment, the second embodiment also forms a directional beam into a plurality of subarrays so as to correspond to FIGS. 1 and 2 in the first embodiment. ”And a“ sub-array shared type ”that realizes multiple directional beams in one array (strictly speaking, it is not separated into sub-arrays and may be understood as“ integrated array ”). There are variations. In FIG. 5, the “sub-array separation type” will be described. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The radio station apparatus 550 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N (N is an integer of 1 or more), and a control circuit 560. The transmission / reception signal processing circuit 551-n (n = 1,..., N) includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, and an A / D converter 125-n. A TDD switch 127-n, phase shifters 502-n-1 to 502-nM, switches 503-n-1 to 503-nM, a distribution coupler 504-n, and a correlation calculation circuit. 505-n and a phase shift control circuit 506-n. Phase shifters 502-n-1 to 502-n-M are connected to antenna elements 501-n-1 to 501-n-M, respectively.

  As with the radio station apparatus 450 shown in FIG. 1, N corresponds to the number of multiplexed streams (number of streams) when performing spatial multiplexing, and the radio station apparatus 550 includes transmission / reception signal processing circuits 551-1 to 551 -N as a whole. Only N systems are installed. Similarly to the radio station apparatus 450, M represents the number of antenna elements of the subarray mounted on each of the transmission / reception signal processing circuits 551-1 to 551 -N. Each of the transmission / reception signal processing circuits 551-n is accompanied by sub-array antenna elements 501-n-1 to 501-n-M, and the transmission / reception signal processing circuits 551-1 to 551-N are as shown in Non-Patent Document 2. It is assumed that they will be installed spatially apart from each other. The baseband signal processing circuit 140 is connected to the transmission / reception signal processing circuits 551-1 to 551 -N by wire, and a digital baseband signal is transferred over the wire. Although not shown in FIG. 11, an example is shown in which the radio station device 550 mounts the entire control circuit 560 on the baseband signal processing circuit 140. The control circuit 560 manages the frame period and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N. The control circuit 560 determines whether to connect the distribution coupler 504-n to the up-converter 123-n or to the down-converter 124-n by the TDD switch 127-n (n = 1,..., N). Switch by dividing. In addition, the control circuit 560 instructs the correlation calculation circuit 505-n (n = 1,..., N) to control the switches 503-n-1 to 503-n-M.

  Furthermore, in the present embodiment, the complex phase rotation amount calculation processing basically performed by the phase shifter corresponding to the transmission / reception weight is performed in the form of digital signal processing, and the actual complex phase rotation processing is realized by analog signal processing. For this reason, even if the modulators 120-1 to 120-N and the demodulator 130-1 to 130-N assume signal processing on the frequency axis as in the OFDM modulation scheme, the time axis as in SC-FDE. Even when the above signal processing is premised, it is possible to cope with either system, and the concept regarding variations of the communication system such as the OFDM modulation system and SC-FDE is explained in the first embodiment. It is the same.

  In addition, in the up converters 123-1 to 123-N and the down converters 124-1 to 124-N, in order to perform frequency conversion between a radio frequency signal and a baseband signal, a signal input from a local oscillator is input. Although necessary, coordinated signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551 -N, and therefore it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description. Although omitted in the following description, it is of course possible to perform cooperative signal processing between the transmission / reception signal processing circuits 551-1 to 551 -N as an additional function. You may do.

A specific signal flow in radio station apparatus 550 is as follows.
First, signal transmission will be described. Radio station apparatus 550 sets all switches 503-n-1 to 503-n-M to ON (connected state to distribution coupler 504-n), and TDD switch 127-n is up-converter 123-n-1 to 123. Signal transmission is performed with n-N and distribution coupler 504-n connected (n = 1,..., N). These conditions are the same in all the transmission / reception signal processing circuits 551-1 to 551 -N.

  Each of the modulators 120-1 to 120-N generates a time base digital baseband transmission signal of each stream to be spatially multiplexed, and inputs the transmission signal to the transmission / reception signal processing circuits 551-1 to 551-N, and then transmits and receives Processing until the up-converter 123-n of the signal processing circuit 551-n (n = 1,..., N) inputs a radio frequency band signal to the subsequent stage is the same as that of the radio station device 450 shown in FIG. . The up-converter 123-n inputs a signal to the distribution coupler 504-n via the TDD switch 127-n.

  The distribution coupler 504-n branches the analog signal input from the TDD switch 127-n into M-system analog signals, and the phase shifter 502-n via the switches 503-n-1 to 503-n-M. Input to -1 to 502-n-M. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation on the analog signal to the input signal. Analog signals to which complex phase rotation has been applied by the phase shifters 502-n-1 to 502-n-M are transmitted via the antenna elements 501-n-1 to 501-n-M, respectively. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 502-n-1 to 502-n-M, and communicates with the radio station apparatus ahead of the directivity.

  The above description is signal processing common to the transmission / reception signal processing circuits 551-1 to 551 -N mounted in N systems, and the same processing is performed in parallel while maintaining a substantially synchronous relationship such as operating with a common clock. carry out.

  Next, the flow of signals related to reception will be described. In the radio station apparatus 550, all the switches 503-n-1 to 503-n-M are turned ON (connected state with the distribution coupler 504-n), and the TDD switch 127-n is connected to the distribution coupler 504-n and the down converter. A signal is received in a state in which 124-n is connected (n = 1,..., N). These conditions are the same in all the transmission / reception signal processing circuits 551-1 to 551 -N.

  Signals received by the antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to the phase shifters 502-n-1 to 502-n-M, respectively. The phase shifters 502-n-1 to 502-n-M respectively apply a predetermined complex phase rotation on the analog signals to the input signals, and pass through the switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. The distribution coupler 504-n combines the signals of the respective antenna systems input via the switches 503-n-1 to 503-n-M on the analog signal, and combines the combined signals via the TDD switch 127-n. To the down converter 124-n. The subsequent processing is the same as that of the radio station apparatus 450 shown in FIG.

  Each of the phase shifters 502-n-1 to 502-n-M performs a predetermined complex phase rotation on the analog signal with respect to the signals received via the antenna elements 501-n-1 to 501-n-M. Are added to form a predetermined directivity, and communication is performed with a radio station apparatus ahead of the directivity. The above description is signal processing common to the transmission / reception signal processing circuits 551-1 to 551 -N mounted in N systems, and the same processing is performed in parallel while maintaining a substantially synchronous relationship such as operating with a common clock. carry out.

  The signal separation circuit 141 performs the same processing as the radio station device 450 shown in FIG. 1 on the signals input from the transmission / reception signal processing circuits 551-1 to 551 -N, and removes them in the above-described directivity formation. A process of suppressing crosstalk components between radio station apparatuses that cannot be fully processed is performed. The suppression of the crosstalk component performed by the signal separation circuit 141 can be performed on the time axis, or can be temporarily converted into a frequency axis signal by FFT processing and performed on the frequency axis. Alternatively, only signal processing for directivity formation performed by the transmission / reception signal processing circuits 551-1 to 551 -N is required, and the signal separation circuit 141 does not need to perform any particular processing (in this case, the signal separation circuit 141 Is optional). However, in any case, the details of the signal separation method here are not directly related to the present embodiment, and can be implemented using the conventional MIMO signal processing technique, and thus the description thereof is omitted here. .

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1-1 to 502-NM will be described. In this signal processing, any one of the switches 503-n-1 to 503-n-M (n = 1,..., N) is operated by the phase shifters 502-n-1 to 502-n-M and the down converter 124-. n is connected (ON) while the remaining switches are in a state (OFF) in which the connection with the down converter 124-n is disconnected. Of the switches 503-n-1 to 503-n-M (n = 1,..., N), the targets to be connected (turned on) to the down converter 124-n are sequentially switched. These switch switchings are managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. Note that during normal operation other than when calculating the amount of rotation of the complex phase described here, the phase shifters 502-n-1 to 502-n-M (n = 1,..., N) as described above. ) Are all connected to distribution coupler 504-n. Further, when performing the calculation process of the rotation amount of the complex phase, the phase rotation amount of the phase shifters 502-1-1-1 to 502-NM is set to a predetermined value. The amount of rotation of the complex phase obtained in the subsequent processing is set as a difference with respect to the initial predetermined value. For example, in the most obvious example, the phase shifters 502-1-1-1 to 502-NM may be set to all zero (or all the same value), and in this case, the rotation amount of the obtained complex phase May be used as the phase rotation amount of the phase shifters 502-1-1-1 to 502-NM during subsequent communication. Or, the initial predetermined values of the phase shifters 502-1-1-1 to 502-NM are +10 degrees, +20 degrees, +30 degrees,..., And the calculated value of the rotation amount of the complex phase is + α degrees. If it is + β degrees, + γ degrees,..., The phase rotation amounts of the phase shifters 502-1-1-1 to 502-NM during the subsequent communication are + (α + 10) degrees and + (β + 20) degrees. , + (Γ + 30) degrees,.

  As an actual process, first, a radio station apparatus as a communication partner to acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and the radio station apparatus 550 receives this training signal. Signals received by the antenna elements 501-n-1 to 501-n-M (n = 1,..., N) are input to the phase shifters 502-n-1 to 502-n-M, respectively. The phase shifters 502-n-1 to 502-n-M respectively apply a predetermined complex phase rotation on the analog signals to the input signals, and pass through the switches 503-n-1 to 503-n-M. And input to the distribution coupler 504-n. Here, since all of the switches 503-n-1 to 503-n-M are OFF, the signal synthesized in the distribution coupler 504-n is effectively converted to the switch 503-n. Only the signals received by the antennas of the systems to which the switches are connected (ON) among n-1 to 503-n-M are output. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 504-n, as a whole, one antenna is selected from the antenna group of the antenna elements 501-n-1 to 501-n-M. A process of extracting the element 501 -n-k (k is an integer from 1 to M) is performed. In addition, it switches with a predetermined period so that k may take any value from 1 to M. The reception signal of the antenna element 501-nk selected in this manner is input to the down converter 124-n via the TDD switch 127-n. The down converter 124-n converts the input radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the input analog baseband signal into a digital baseband signal and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  Each of the correlation calculation circuits 505-1 to 505-N continuously records the digital baseband signal until the signals from all the switches are received while switching. That is, the correlation calculation circuit 505-n receives digital baseband signals from all the switches 503-n-1 to 503-n-M while switching the switches 503-n-1 to 503-n-M. And record. The correlation calculation circuit 505-n considers the periodicity of the training signal (for example, the periodicity such that the training signal having the same content is repeated in a 2048 sample period) with respect to the recorded digital baseband signal, and Sampling data of each antenna element 501-n-k is extracted so that the sampling timing in the period corresponds, and in the same manner as the correlation calculation circuits 405-1 to 405-N of the first embodiment, Using equation (3), the amount of rotation of the complex phase to be set for each of the phase shifters 502-n-1 to 502-n-M is calculated. Note that this utilizes the fact that the channel fluctuation can be ignored if the radio station apparatus is fixedly installed at a high place and is in a line-of-sight environment. Further, when calibration processing is necessary as necessary, the correlation calculation circuits 505-1 to 505-N are similar to the correlation calculation circuits 405-1 to 405-N of the first embodiment. The amount of rotation of the complex phase on the transmission side is determined as a value in consideration of the calibration coefficient in 1) to Equation (3).

In the first embodiment, the sampling data of each antenna element 401-n-k can be acquired at the same time. However, in the second embodiment, since sampling is performed at different timings, the periodicity of the training signal Therefore, it has been devised so that it can be regarded as equivalently sampled at the same time. At this time, if the frequency error cannot be ignored on the transmission side and the reception side, sampling may not be considered equivalently at the same time just by the periodicity of the training signal. In such a case, the frequency error is corrected. You may do. For example, assuming that the training signal has a periodicity of N _FFT samples with respect to the sampling data S _k (n) of the training signal while one switch 503-nk is continuously ON, the N _Test cycle Suppose that sampling data can be secured. If the frequency error is Δf, Δf can be estimated by obtaining Δf ′ that maximizes the Q value of the following equation (6) for various Δf ′.

Here, attention is focused only on the information of the switches 503-nk, but Δf may be obtained for the sampling data of each of the switches 503-nk and averaged. If Δf is estimated in this way, the influence of the frequency error can be eliminated by performing the correction shown in the following equation (7) on the sampling data S _k (n ′).

  Here, n 'means a serial number that is continuous between switching from the switch 503-n-1 to the switch 503-n-1, regardless of switch switching. Since the phase of 2πjΔfn ′ rotates during the time of sampling period × n ′, the rotation is reversely corrected.

  Thus, the rotation amount of the complex phase obtained by the correlation calculation circuit 505-n (n = 1,..., N) is the identification number of the radio station apparatus with which it communicates (communication with a plurality of radio station apparatuses). In the same manner as in the first embodiment, an identification number is not necessary when communication is performed with a single radio station apparatus in the P-P type.) And is input to the phase shift control circuit 506-n. The phase shift control circuit 506-n associates the amount of rotation of the complex phase to be set to each of the phase shifters 502-n-1 to 502-nM with the identification number of the communication partner radio station apparatus in the memory. Manage it by memorizing it.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 560 grasps the wireless station device that is a communication partner, and the phase shift control circuit 506-n (n = 1,... , N), the phase shifters 502-n-1 to 502-n-M are instructed to set the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication. The phase shift control circuit 506-n acquires the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication by reading it from the memory, and acquires the rotation amount of the complex phase by the phase shifters 502-n-1 to 502. Set to -n-M to realize beamforming on analog.

  Although not clearly shown in the figure, for example, if a high-power amplifier (HPA) on the transmission side is arranged, it is arranged at a point described as “A” in the figure, and a low-noise amplifier on the reception side ( LNA) or the like is arranged at a point described as “B” in the drawing. Regarding the points described as “A” and “B”, in the same transmission / reception signal processing circuits 551-1 to 551 -N, the antenna elements 501 -n- 1 to 501 -n-M with respect to the same n. In this embodiment, which basically does not assume cooperative transmission between the transmission / reception signal processing circuits 551-1 to 551 -N, the individual high power amplifiers and low noise amplifiers are used. A calibration process for removing the uncertainty of the complex phase is unnecessary.

  Further, since the transmission / reception signal processing circuits 551-1 to 551 -N having a subarray configuration can be physically separated to reduce the correlation of directional beams formed on analog, generally, Install at a distance greater than a predetermined distance.

  Next, as in FIG. 2 with respect to FIG. 1 in the first embodiment, the radio station apparatus of this embodiment is configured by a “sub-array shared type” that realizes a plurality of directional beam formations with one array antenna. Is also possible. This configuration is shown in FIG.

  FIG. 6 is a functional block diagram showing a configuration example (subarray shared type) of the wireless station device 552 in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The radio station apparatus 552 shown in the figure includes a baseband signal processing circuit 140, transmission / reception signal processing circuits 551-1 to 551-N, distribution couplers (HYB) 507-1 to 507-M, and an antenna element 501-. 1-501-M and a control circuit 560. Distribution couplers (HYB) 507-m (m = 1,..., M) are the phase shifters 502-1-m, 502- of the transmission / reception signal processing circuits 551-1, 551-2,. 2-m,..., 502-Nm, and the antenna element 501-m.

  In the radio station apparatus 552 as well, as in the radio station apparatus 550 shown in FIG. This is implemented for N systems. M represents the number of antenna elements of the subarray mounted on the wireless station device 552. In the radio station apparatus 550 shown in FIG. 5, since the number of antenna elements of each sub-array is M, the total number of antennas is N × M. Of the value M. In actual operation, the radio station device 550 shown in FIG. 5 and the radio station device 552 shown in FIG. 6 do not necessarily have the same number of antenna elements.

  The baseband signal processing circuit 140 is connected by wire to each of the transmission / reception signal processing circuits 551-1 to 551 -N, and a digital baseband signal is transferred over this wire. Similarly to the case of FIG. 1, FIG. 6 shows an example in which the entire control circuit 560 is mounted on the baseband signal processing circuit 140.

  Similar to the wireless station device 550, the wireless station device 552 basically performs digital phase processing for the complex phase rotation amount performed by the phase shifter corresponding to the transmission / reception weight, and the actual complex phase rotation processing is analog signal processing. To achieve. Therefore, similarly to the radio station device 550, the modulators 120-1 to 120-N and the demodulators 130-1 to 130-N of the radio station device 552 have a time axis when assuming signal processing on the frequency axis. Both methods can be used when the above signal processing is assumed.

  Further, in the up converters 123-1 to 123-N and the down converters 124-1 to 124-N, in order to perform frequency conversion between a radio frequency signal and a baseband signal, a signal input from a local oscillator is input. Necessary. However, cooperative signal processing is not assumed between the transmission / reception signal processing circuits 551-1 to 551 -N, and it is not always necessary to use a common local oscillator. Unlike the case of the radio station apparatus 550 shown in FIG. 5, in the radio station apparatus 552, the up converters 123-1 to 123-N and the down converters 124-1 to 124-N are all contained in the same casing. For this reason, it is expensive to prepare individual local oscillators for the up converters 123-1 to 123-N and the down converters 124-1 to 124-N. Therefore, there is substantially a local oscillator shared externally. The configuration is common. However, since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  As described above, in the wireless station device 552, the transmission / reception signal processing circuits 551-1 to 551-N for N systems are mounted in one housing, and are connected via the distribution couplers 507-1 to 507-M. The antenna elements 501-1 to 501-M are shared. 6 shows an example in which the control circuit 560 is mounted on the baseband signal processing circuit 140, but the control circuit 560 may be mounted at any location in the wireless station device 552. The control circuit 560 manages the frame period and transmission / reception timing, and also manages switching of the TDD switches 127-1 to 127 -N.

  The difference between the radio station apparatus 550 shown in FIG. 5 and the radio station apparatus 552 shown in FIG. 6 is that the transmission / reception signal processing circuits 551-1 to 551 -N are aggregated in the housing of one radio station apparatus 552 and shared. Each of the sub-array antenna elements 501-1 to 501-M is distributed by distribution couplers 507-1 to 507-M and connected to transmission / reception signal processing circuits 551-1 to 551-N. is there. Therefore, the processing in the transmission / reception signal processing circuits 551-1 to 551 -N is the same for both the radio station apparatus 550 shown in FIG. 5 and the radio station apparatus 552 shown in FIG. 6.

In the following, focusing on the difference from radio station apparatus 550 shown in FIG. 5, a specific signal flow that is the difference in radio station apparatus 552 is shown.
First, signal transmission will be described. In radio station apparatus 552, processing for directivity formation is performed from each of phase shifters 502-n-1 to 502-n-M (n = 1,..., N) of transmission / reception signal processing circuit 551-n. A radio frequency analog signal is output. Each of these M signals is input to distribution couplers 507-1 to 507-M connected to the corresponding antenna elements 501-1 to 501-M. That is, the phase shifters 502-1-m, 502-2-m,..., 502-Nm (m = 1,..., M) input signals to the distribution coupler 507-m. Distribution couplers 507-1 to 507 -M synthesize signals input from N transmission / reception signal processing circuits 551-1 to 551 -N, and the combined signals are antenna elements 501-1 to 501 -M. Sent through.

  Next, signal reception will be described. Signals received by antenna elements 501-1 to 501-M of radio station apparatus 552 are distributed to N systems in distribution couplers 507-1 to 507-M, respectively, and transmission / reception signal processing circuits 551-1 to 551- are respectively received. N is input. For example, a signal received by the antenna element 501-m (m = 1,..., M) is distributed to N systems by a distribution coupler 507-m, and phase shifters 502-1-m, 502-2-m, ..., input to 502-Nm. The radio station apparatus 552 performs signal processing similar to reception of signals in the radio station apparatus 550 shown in FIG. 5 on the signals input to the phase shifters 502-1-1 to 502 -N-M in this way.

  In the signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1-1 to 502-NM, the signals received by the antenna elements 501-1 to 501-M are respectively similar to the reception processing. In the distribution couplers 507-1 to 507 -M, the signals are distributed to the N systems and are respectively input to the transmission / reception signal processing circuits 551-1 to 551 -N. The radio station device 552 uses the same signal as the calculation of the amount of rotation of the complex phase in the radio station device 550 shown in FIG. 5 as the signal input to the phase shifters 502-1-1-1 to 502-NM in this way. Process.

  Other operations are basically performed in the same manner as the radio station apparatus 550 shown in FIG. Further, the explanation regarding the HPA on the transmitting side and the LNA on the receiving side is the same as that of the radio station apparatus 550 shown in FIG.

  Note that the communication system according to the present embodiment includes a wireless station device 550 instead of the wireless station device 450 illustrated in FIG. 3 and a wireless station device 552 instead of the wireless station device 452. Note that the radio station device 450 and the radio station device 552 may face each other, or the radio station device 550 and the radio station device 452 may face each other.

  Next, another configuration example of the transmission / reception signal processing circuit according to the present embodiment will be described with reference to FIGS. The transmission / reception signal processing circuits shown in FIGS. 7 to 10 can be replaced with the transmission / reception signal processing circuits 551-1 to 551 -N included in the wireless station device 550 shown in FIG. 5 or the wireless station device 552 shown in FIG. 6. . Below, the case where it replaces with the transmission-and-reception signal processing circuits 551-1 to 551-N with which the radio station apparatus 550 is provided is demonstrated to an example. Note that when the transmission / reception signal processing circuits 551-1 to 551-N included in the wireless station device 552 are replaced, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 in FIGS. ˜501-n-M correspond to the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M, the antenna elements 501-1 to 501-M, and the antenna elements 541-1 to 541-M. The distribution coupler (HYB).

  FIG. 7 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 553-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The transmission / reception signal processing circuit 553-n shown in the figure includes a D / A converter 122-n, an up converter 123-n, a down converter 124-n, an A / D converter 125-n, and a circulator 521-. n, phase shifters 502-n-1 to 502-n-M, switches 503-n-1 to 503-n-M, distribution couplers 504-n, correlation calculation circuit 505-n, phase Shift control circuit 506-n. Phase shifters 502-n-1 to 502-n-M are connected to antenna elements 501-n-1 to 501-n-M, respectively.

  In this figure, the antenna elements 501-n-1 to 501-n-M are shown in the figure for easy understanding of the comparison with FIG. 5, but the transmission / reception signal processing circuit 553-n corresponds to the inside of the solid line. In this frame, it corresponds to the transmission / reception signal processing circuit 551-n in FIG.

  The difference between the transmission / reception signal processing circuit 551-n shown in FIGS. 5 and 6 and the transmission / reception signal processing circuit 553-n shown in FIG. 5 is the same as that of the transmission / reception signal processing circuit 551-n shown in FIGS. Although the input / output flow of the transmission signal and the reception signal is controlled by n, the transmission / reception signal processing circuit 553-n shown in FIG. 7 controls the input / output flow using the circulator 521-n. That is, circulator 521-n passes the input signal from up-converter 123-n to distribution coupler 504-n, and passes the input signal from distribution coupler 504-n to down-converter 124-n. The circulator 521-n suppresses other signal flows. Therefore, similar signal processing can be performed without using the TDD switches 127-1 to 127 -N as in the transmission / reception signal processing circuit 551-n shown in FIGS. 5 and 6. Except for this point, all other signal processing is equivalent to the signal processing of the transmission / reception signal processing circuit 551-n shown in FIGS. The explanation regarding the HPA on the transmission side and the LNA on the reception side is the same as that of the transmission / reception signal processing circuit 551-n shown in FIGS.

  FIG. 8 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 554-n in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  The transmission / reception signal processing circuit 554-n shown in the figure includes a D / A converter 122-n, an up converter 123-n, a distribution coupler 514-n, and phase shifters 509-n-1 to 509-n. -M, TDD switches 508-n-1 to 508-nM, phase shifters 502-n-1 to 502-nM, switches 503-n-1 to 503-nM, and distribution A coupler 515-n, a down converter 124-n, an A / D converter 125-n, a correlation calculation circuit 505-n, and a phase shift control circuit 506-n are provided. TDD switches 508-n-1 to 508-n-M are connected to antenna elements 501-n-1 to 501-n-M, respectively.

  Here, similarly to the transmission / reception signal processing circuits 551-1 to 551 -N shown in FIG. 5, the up converter 123-n and the down converter 124-n perform frequency conversion between a radio frequency signal and a baseband signal. Therefore, input of a signal from the local oscillator is required. However, since no cooperative signal processing is assumed between the transmission / reception signal processing circuits 554-1 to 554 -N, it is not always necessary to use a common local oscillator. Since the description is complicated, the description of the external local oscillator is omitted here as a simple description.

  Further, the entire control circuit 560 is mounted in the baseband signal processing circuit 140 of the radio station apparatus 550 or the radio station apparatus 552 or in the radio station apparatus 552. The control circuit 560 manages transmission / reception timing. Further, since the plurality of transmission / reception signal processing circuits 554-1 to 554 -N operate in conjunction with each other, when the control circuit 560 performs control, the plurality of transmission / reception signal processing circuits 554-1 to 554 -N are timed. Works together.

  The difference between the transmission / reception signal processing circuit 554-n and the transmission / reception signal processing circuits 551-1 to 551-N shown in FIG. 5 or 6 is that the TDD switches 508-n-1 to 508-n-M are the antenna elements 501. -N-1 to 501-n-M (or antenna elements 501-1 to 501-M) are arranged in the immediate vicinity, and as a result, from the D / A converter 122-n in the transmission system to the antenna element 501-n-1 The path from .about.501-n-M and the path from the antenna elements 501-n-1 to 501-n-M to the A / D converter 125-n in the receiving system are physically separated. .

In the case of the transmission / reception signal processing circuit 551-n shown in FIG. 5 or FIG. 6, for example, the high power amplifier (or power amplifier) of the transmission system is arranged at the place described as “A” after the up converter 123-n. The low-noise amplifier of the receiving system is arranged at a location described as “B” in the previous stage of the down converter 124-n, and the circuit from the TDD switch 127-n to the antenna end can be shared by transmission and reception. This point is greatly different from the transmission / reception signal processing circuit 554-n. The advantage of adopting the configuration as shown in FIG. 8 is that high power amplifiers and low noise amplifiers can be mounted as many as the number of antenna elements. As a result, the total transmission power is improved, and the TDD switch in FIGS. It is a point that it becomes possible to suppress the influence of insertion loss and distribution and coupling loss of various circuits between 127-n and antenna elements 501-1 to 501-M. For this reason, although not shown here, high power amplifiers are provided at locations where “D ₁ ”, “D ₂ ”,..., “D _M ” are described, “E ₁ ”, “E ₂ ”,. It is preferable that a low-noise amplifier is arranged at a place described as “E _M ”. In this case, a calibration process for removing the uncertainty of the complex phase of each high power amplifier and low noise amplifier is required. However, if a high-power amplifier is disposed at a location described as “A” and a low-noise amplifier is disposed at a location described as “B”, “D ₁ ”, “D ₂ ” _,. There is no necessity that a high power amplifier and a low noise amplifier are arranged at each of the point locations described as “E ₁ ”, “E ₂ ”,..., “E _M ”.

  The above changes correspond to the transmission / reception signal processing circuit 451-n in FIG. 1 and FIG. 2 in the first embodiment being changed to the transmission / reception signal processing circuit 453-n shown in FIG. The transmission / reception signal processing circuit 551-n in FIG. 5 and FIG. 6 in the second embodiment is changed to the transmission / reception signal processing circuit 554-n shown in FIG. This corresponds to the change to 453-n.

The following shows a specific signal flow in the transmission / reception signal processing circuit 554-n, which is a difference from the transmission / reception signal processing circuit 551-n, focusing on the characteristic part of the present embodiment.
First, signal transmission will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M have antenna elements 501-n-1 to 501-n-M and phase shifters 509-n-1 to 509-n-. Signals are transmitted with M connected to each other.

  The transmission / reception signal processing circuit 554-n receives a time-base digital baseband transmission signal of one stream from a modulator 120-n not shown here. The D / A converter 122-n converts the input transmission signal into an analog baseband signal and outputs the analog baseband signal to the upconverter 123-n. The up-converter 123-n converts the analog baseband signal input from the D / A converter 122-n into a radio frequency band signal and inputs the signal to the distribution coupler 514-n.

  Distribution coupler 514-n branches the analog signal in the radio frequency band input from up-converter 123-n into M-system analog signals and inputs them to phase shifters 509-n-1 to 509-n-M. . Each of the phase shifters 509-n-1 to 509-n-M applies a predetermined complex phase rotation on the analog signal to the signal input from the distribution / combining device 514-n, so that the TDD switch 508-n-1 is applied. Are input to the antenna elements 501-n-1 to 501-nM through .about.508-nM. The antenna elements 501-n-1 to 501-n-M transmit the input transmission signals. The transmission signal has a predetermined directivity formed by complex phase rotation in the phase shifters 509-n-1 to 509-n-M.

  Next, signal reception will be described. In the transmission / reception signal processing circuit 554-n, the TDD switches 508-n-1 to 508-n-M are connected to the antenna elements 501-n-1 to 501-nM and the phase shifters 502-n-1 to 502-n-. M is connected, and the switches 503-n-1 to 503-n-M receive signals in a state where the distribution couplers 515-n and the phase shifters 502-n-1 to 502-n-M are respectively connected. I do.

  Signals received by the antenna elements 501-n-1 to 501-n-M are respectively phase shifters 502-n-1 to 502-n-M via TDD switches 508-n-1 to 508-n-M. Is input. Each of the phase shifters 502-n-1 to 502-n-M applies a predetermined complex phase rotation to the input signal on the analog signal, and switches 503-n-1 to 503-n-M. To the distribution coupler 515-n. Distribution coupler 515-n combines the signals of the respective antenna systems input via switches 503-n-1 to 503-n-M on an analog signal and inputs the combined signal to down converter 124-n. The down converter 124-n converts a radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal input from the down converter 124-n into a digital baseband signal, and a signal in the baseband signal processing circuit 140 not shown here. This is input to the separation circuit 141. The baseband signal processing circuit 140 performs subsequent signal processing.

  Next, signal processing when calculating the amount of rotation of the complex phase in the phase shifters 502-1-1-1 to 502-NM will be described. In this signal processing, any one of the switches 503-n-1 to 503-n-M connects (ON) the distribution coupler 515-n and the phase shifters 502-n-1 to 502-n-M. On the other hand, the remaining switches are performed in a state where the connection with the distribution coupler 515-n is disconnected (OFF). These switch switchings are managed by the correlation calculation circuits 505-1 to 505-N under the instruction of the control circuit 560. During normal operation other than when calculating the amount of rotation of the complex phase described here, all of the phase shifters 502-n-1 to 502-n-M are distributed couplers 515-n as described above. In addition, the phase shifters 509-n-1 to 509-n-M are all connected to the distribution coupler 514-n. Similarly to the transmission / reception signal processing circuit 551-n shown in FIG. 5 or FIG. 6, the phase rotation of the phase shifters 502-1-1 to 502-1-M is performed when the complex phase rotation amount calculation processing is performed. The amount is set to a predetermined value, and the rotation amount of the complex phase obtained by the subsequent processing is set as a difference with respect to the initial predetermined value.

  As an actual process, first, a radio station apparatus of a communication partner that should acquire the rotation amount of the complex phase transmits a training signal for channel estimation, and has radio signal processing circuits 554-1 to 554 -N. Receives this signal. Signals received by antenna elements 501-n-1 to 501-n-M are input to phase shifters 502-n-1 to 502-n-M, respectively. The phase shifters 502-n-1 to 502-n-M respectively apply a predetermined complex phase rotation on the analog signals to the input signals, and pass through the switches 503-n-1 to 503-n-M. To the distribution coupler 515-n. Here, since all of the switches 503-n-1 to 503-n-M are OFF, the signal synthesized in the distribution coupler 515-n is effectively converted to the switch 503-n. Only the signal received by the antenna of the system to which the switch is connected (ON) among n-1 to 503-n-M is output. That is, in the switches 503-n-1 to 503-n-M and the distribution coupler 515-n as a whole, one antenna element 501-n from the antenna elements 501-n-1 to 501-n-M is obtained. The process of extracting -k (k is an integer from 1 to M) is performed. Note that k is changed to take any value from 1 to M. The reception signal of the antenna element 501-nk selected in this way is input to the down converter 124-n. The down converter 124-n converts the input radio frequency signal into an analog baseband signal and inputs the analog baseband signal to the A / D converter 125-n. The A / D converter 125-n converts the analog baseband signal into a digital baseband signal and inputs the digital baseband signal to the correlation calculation circuit 505-n.

  The correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 554-n are the correlation calculation circuit 505-n and the phase shift control circuit 506-n of the transmission / reception signal processing circuit 551-n shown in FIG. The same processing is performed. That is, the correlation calculation circuit 505-n calculates the amount of rotation of the complex phase on the transmission side to be set in each of the phase shifters 502-n-1 to 502-n-M. The phase shift control circuit 506-n associates the amount of rotation of the complex phase to be set to each of the phase shifters 502-n-1 to 502-nM with the identification number of the communication partner radio station apparatus in the memory. Manage it by memorizing it.

In addition, the correlation calculation circuit 505-n has a high power amplifier in the place described as “D ₁ ”, “D ₂ ”,..., “D _M ”, and “E ₁ ”, “E ₂ ”,. In the case where the low noise amplifier is arranged at a place described as “E _M ”, similarly to the correlation calculation circuit 405-n of the transmission / reception signal processing circuit 453-n in FIG. The rotation amount may be subjected to a calibration process in the implicit feedback of the prior art, and converted to the complex phase rotation amount in the transmission system by correction corresponding to the calibration process based on the rotation amount of the complex phase in the reception system. . The phase shift control circuit 506-n associates the rotation amount of the complex phase in the transmission system to be set to each of the phase shifters 509-n-1 to 509-n-M with the identification number of the radio station apparatus of the communication counterpart. Similarly, it is managed by storing it in a memory.

  When actual data communication is performed, that is, when transmission processing or reception processing is performed, the control circuit 560 grasps a wireless station device that is a communication partner and communicates with the phase shift control circuit 506-n. It instructs the phase shifters 502-n-1 to 502-n-M and the phase shifters 509-n-1 to 509-n-M to set the complex phase rotation amount corresponding to the radio station apparatus to be performed. The phase shift control circuit 506-n acquires the rotation amount of the complex phase corresponding to the wireless station apparatus that performs communication by reading it from the memory, and acquires the rotation amount of the complex phase by the phase shifters 502-n-1 to 502. -N-M and phase shifters 509-n-1 to 509-n-M are set to realize analog beamforming. These processes are similarly performed in the transmission / reception signal processing circuits 554-1 to 554-N.

  FIG. 9 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 555-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

  Differences between the transmission / reception signal processing circuit 555-n shown in the figure and the transmission / reception signal processing circuit 554-n shown in FIG. 8 are as follows. That is, the transmission / reception signal processing circuit 554-n shown in FIG. 8 shares the antenna elements 501-n-1 to 501-n-M for transmission / reception. On the other hand, the transmission / reception signal processing circuit 555-n shown in FIG. 9 has a configuration in which these are separated by transmission and reception, a pair is formed by transmission / reception, and a transmission / reception antenna is placed as a set in close proximity. Therefore, when the transmission / reception signal processing circuit 554-n of the wireless station device 550 is replaced with the transmission / reception signal processing circuit 555-n, antenna elements 541-n-1 to 541-n-M are added, and the TDD switch 508-n- 1-508-n-M is omitted. When the transmission / reception signal processing circuit 551-n of the wireless station device 552 is replaced with the transmission / reception signal processing circuit 555-n, the antenna elements 501-n-1 to 501-n-M and the antenna elements 541-n-1 to 541 are used. In place of −n−M, the antenna elements 501-1 to 501-M and the antenna elements 541-1 to 541-M (and the antenna elements 501-1 to 501) shared by the transmission / reception signal processing circuits 555-1 to 555-N are used. -M, a distribution coupler corresponding to the antenna elements 541-1 to 541-M).

  In the description of FIG. 8, for the sake of convenience, the description up to the immediate vicinity of the antenna element has been described as the transmission / reception signal processing circuit 554-n, but the TDD switches 508-n-1 to 508-n-M are assumed to be functions on the antenna element side. If considered, FIG. 8 and FIG. 9 are completely equivalent diagrams. Also in the details of the signal processing, in the receiving system, the transmission / reception signal processing circuit 554-n shown in FIG. 8 receives signals from the antenna elements 501-n-1 to 501-n-M, whereas FIG. The transmission / reception signal processing circuit 555-n shown in FIG. 5 receives data using the antenna elements 541-n-1 to 541-n-M, and does not pass through the TDD switches 508-n-1 to 508-n-M in transmission / reception. Except for this point, all signal processing in the transmission / reception signal processing circuit 554-n shown in FIG. 8 and the transmission / reception signal processing circuit 555-n shown in FIG. 9 is common.

  However, in consideration of the physically different transmission / reception antennas, in addition to simple calibration processing, it is also possible to perform correction in consideration of physically different antenna element coordinates.

  In the above description, the transmission / reception antennas are arranged as a set in close proximity. However, as long as correction is performed in consideration of the physical differences in the coordinates of the antenna elements, the transmission / reception antennas are not necessarily set. There is no need to place them. FIG. 10 is a functional block diagram showing a configuration example of the transmission / reception signal processing circuit 556-n (n = 1,..., N) in the present embodiment. In the figure, the same functional blocks as those in the above-mentioned figure are given the same numbers, and the description thereof is omitted.

Also in the transmission / reception signal processing circuit 556-n shown in FIG. 10, the TDD switches 508-n-1 to 508-n-M have the functions on the antenna element side as in the transmission / reception signal processing circuit 555-n shown in FIG. Considering this, FIG. 8 is also completely equivalent to FIG. In this sense, also in the transmission / reception signal processing circuit 556-n shown in FIG. 10, the transmitting antenna elements 501-n-1 to 501-n-M and the reception are received in the same manner as the transmission / reception signal processing circuit 555-n shown in FIG. The antenna elements 541-n-1 to 541-n-M are separated. However, in the transmission / reception signal processing circuit 555-n shown in FIG. 9, a pair of individual transmission / reception antenna elements (for example, a pair of the antenna element 501-n-1 and the antenna element 541-n-1) is integrally disposed in the vicinity. On the other hand, in the transmission / reception signal processing circuit 556-n shown in FIG. 10, the transmitting antenna elements 501-n-1 to 501-n-M constitute one transmitting antenna array, and receive antennas. The elements 541-n-1 to 541-n-M are assumed to constitute a single receiving antenna array. Therefore, there is no difference in signal processing between FIG. 9 and FIG. 10 except for the physical antenna element arrangement.
In the case where the transmission / reception antennas are separated and operated as shown in FIG. 10, a wall-like obstacle may be arranged in order to avoid signal leakage from the transmission antenna to the reception antenna.

  When performing digital beam forming in Massive MIMO, the conventional radio station apparatus requires expensive A / D converters and D / A converters corresponding to the number of signal sequences, so the apparatus is expensive. In addition, there is a problem that power consumption increases. Therefore, the radio station apparatus according to the present embodiment switches the switches so that only the target signal sequence is connected to the A / D converter only when weight information for performing directivity control is acquired. In addition, when transmitting data, the radio station apparatus converts a signal before separation for each antenna element into an analog signal by a D / A converter, and uses the phase shifter after separating the converted analog signal for each antenna element. To perform analog beam forming. As a result, it is possible to reduce the number of A / D converters and D / A converters necessary for acquiring weight information during data transmission / reception, and to reduce power consumption.

  Thus, according to the radio station apparatus of the present embodiment, each antenna element has a switch, and when calculating the amount of rotation of the complex phase that becomes weight information from the state of the reception signal for each reception antenna element, Only one of all receiving antenna elements is turned on and connected to the A / D converter, and switching of the switch is devised so as to periodically switch the antenna element connected to the A / D converter. As a result, the radio station apparatus can individually grasp the received signal states of all the antenna elements by one RF system (including from the down converter to the A / D converter), and the amount of rotation of the complex phase based on the information. Is calculated. In Massive MIMO, it is necessary to calculate weight information from at least the state of the received signal for each antenna element. Conventionally, the A / D converter and the device of the D / A converter have become very expensive especially in a wide band, and it has been difficult to reduce the size of the device including the countermeasures against heat generation. According to this embodiment, it is possible to minimize the number of A / Ds and D / As to be mounted, and to reduce the size, power consumption, and economy of the radio station device.

  As described above, according to the present embodiment, when the radio station apparatus calculates the rotation amount of the complex phase for performing directivity control, only one of all the receiving antenna elements is turned on. In this way, a down converter and an A / D converter are used while switching the switches. The radio station apparatus substitutes analog beam forming using a phase shifter instead of digital beam forming at the time of actual signal transmission. On the other hand, at the time of signal reception, the radio station apparatus turns on all the switches, synthesizes a signal subjected to analog beam forming using a phase shifter, and performs demodulation processing. Thereby, in Massive MIMO, the number of A / D converters and D / A converters conventionally required when performing digital beam forming can be significantly reduced. Therefore, a situation in which a large number of A / D converters and D / A converters continue to operate constantly during communication is avoided, and a huge amount of A / D converters and D / A conversions that have been required particularly during broadband communication are avoided. It is possible to reduce the power consumption and reduce the amount of heat generated. In addition, the number of A / D converters and D / A converters required is greatly reduced, eliminating the need for large heat sinks required for heat generation associated with enormous power consumption. It is possible to reduce not only the power but also the size of the radio station device.

  In the above-described embodiment, the complex phase is rotated on the radio frequency analog signal. However, the position of the up-converter and the down-converter is changed, and the complex phase of the base-band or intermediate frequency analog signal is changed. A configuration may be adopted in which rotation is applied and frequency conversion with a radio frequency is performed at the subsequent stage or the preceding stage.

  According to the present embodiment, wireless communication devices such as the wireless station devices 550 and 552 form directivity using one or a plurality of subarrays configured to include a plurality of antenna elements, and with other wireless communication devices. Wireless communication. However, although the expression “sub-array” is used in this specification, the signal processing feature of the present invention does not necessarily have to be configured to mount a plurality of “sub-arrays”, and the wireless communication apparatus is provided. A single array antenna constituted by all antenna elements may be used, or an array antenna constituted by a part of all antenna elements provided in the radio apparatus may be used. The wireless communication device includes a plurality of antenna elements, a signal receiving unit, a first phase rotating unit, a switching unit, a switching control unit, a signal combining unit, a signal converting unit, a signal reproducing unit, and a correlation calculation. A first rotation amount calculation unit, and a first phase rotation amount control unit. Note that the wireless communication device includes a first phase rotation unit, a switching unit, a signal synthesis unit, and a signal conversion unit at least for each subarray used for signal reception. Further, the wireless communication device includes a transmission signal generation unit, a signal distribution unit, a second phase rotation unit, a signal transmission unit, a second rotation amount calculation unit, and a second phase rotation amount control unit. Prepare. The wireless communication apparatus includes a signal distribution unit and a second phase rotation unit at least for each subarray used for signal transmission. Here, if the phase rotation amount control unit is a P-P (point-to-point) type, it can be dealt with by directly setting the rotation amount of the complex phase calculated by the rotation amount calculation unit in the phase rotation unit. However, in this case, it can be understood that either the rotation amount calculation unit or the phase rotation unit implicitly includes the function of the phase rotation amount control unit.

  When receiving a signal to be played back or receiving a training signal, the signal receiving unit transmits a signal transmitted by another wireless communication device with a wireless communication partner via each of a plurality of antenna elements constituting a subarray used for signal reception. Receive. The first phase rotation unit outputs an analog signal for each antenna element system for each received signal received via an antenna element constituting a sub-array corresponding to the self-function part among signals received by the signal receiving unit. The complex phase is rotated by a predetermined value and output. In the case of simply passing through the first phase rotation unit at the time of receiving a training signal, it is understood that the complex phase of zero degrees has been rotated.

  The switching unit switches for each antenna element system whether or not to pass the reception signal output from the first phase rotation unit corresponding to the same sub-array as the self-function unit. The switching control unit allows the switching in each switching unit to pass all or part of the reception signal output from the first phase rotation unit corresponding to the switching unit when the reception signal is a reproduction target, and receives the received signal. Is a training signal, the reception signal output from the first phase rotation unit corresponding to the switching unit is controlled to pass one by one for each antenna element system. The signal synthesizer synthesizes the received signal that has been passed for each system of antenna elements by the switching unit corresponding to the same sub-array as the self-function unit. The signal conversion unit converts the reception signal synthesized by the signal synthesis unit corresponding to the same sub-array as the self-function unit from a radio frequency analog signal to a baseband digital signal.

  When receiving the training signal, the correlation calculation unit continuously records the signal sequence of the input digital signal, and considers the periodicity of the training signal from the recorded digital signal sequence, and the sampling timing in the cycle corresponds. Thus, the digital signal sequence of each antenna element is extracted. Further, the correlation calculation unit calculates the correlation of the digital signal sequence with respect to the training signal between the antenna elements corresponding to the self-function unit. Specifically, the correlation calculation unit determines a reference antenna element from among a plurality of antenna elements, and uses the training signal passed by the switching unit one by one, and the reference antenna element and other antenna elements The correlation of the digital signal sequence with respect to the training signal having the same sampling timing is calculated for each combination. The first rotation amount calculation unit calculates the rotation amount of the complex phase to be given by the first phase rotation unit, based on the correlation calculation result by the correlation calculation unit. The first phase rotation amount control unit controls the rotation amount of the complex phase provided by the first phase rotation unit based on the calculation result of the rotation amount of the complex phase by the first rotation amount calculation unit.

  Further, the second rotation amount calculation unit may perform a calibration process on the calculation result of the correlation calculated by the correlation calculation unit, the calculation result of the rotation amount of the complex phase calculated by the first rotation amount calculation unit, or the calculation result. The amount of rotation of the complex phase to be given by the second phase rotation unit is calculated based on any of the information on the amount of rotation of the complex phase obtained by performing the above. The second phase rotation amount control unit controls the rotation amount of the complex phase provided by the second phase rotation unit based on the calculation result of the rotation amount of the complex phase by the second rotation amount calculation unit.

  When receiving the signal to be reproduced, the signal reproduction unit reproduces the signal transmitted by the other wireless communication device of the communication partner based on the reception signal synthesized by the signal synthesis unit. On the other hand, at the time of signal transmission, the transmission signal generation unit generates an analog transmission signal to be transmitted to another wireless communication device of the communication partner using the subarray. A signal distribution part branches the transmission signal transmitted using the subarray corresponding to a self-function part among the analog transmission signals which the transmission signal generation part produced | generated corresponding to each antenna element which comprises the subarray. The second phase rotation unit rotates the complex phase by a predetermined value on the analog signal for each transmission signal branched for each antenna element by the signal distribution unit corresponding to the same sub-array as the self-function unit. The signal transmission unit transmits a transmission signal whose phase has been rotated by the second phase rotation unit via an antenna element that constitutes a subarray corresponding to the second phase rotation unit.

Note that the wireless communication apparatus includes a functional unit in which a correlation calculation unit, a first rotation amount calculation unit, and a second rotation amount calculation unit are integrated, such as correlation calculation circuits 505-1 to 505-N. You may provide for every, and you may provide a correlation calculation part, a 1st rotation amount calculation part, and a 2nd rotation amount calculation part separately.
In addition, the wireless communication apparatus includes a functional unit that integrates the first phase rotation amount control unit and the second phase rotation amount control unit for each subarray, such as phase shift control circuits 506-1 to 506-N. Alternatively, the first phase rotation amount control unit and the second phase rotation amount control unit may be provided separately.

  Further, the wireless communication device includes phase shifters 502-n-1 to 502-nM (n = 1,..., N) included in the transmission / reception signal processing circuits 551-1 to 551-N or the transmission / reception signal processing circuit 553-n. ), A phase rotation unit that integrates the first phase rotation unit and the second phase rotation unit may be provided for each sub-array, and transmission / reception signal processing circuits 554-n, 555-n, and 556-n are provided. The phase shifters 502-n-1 to 502-nM and 509-n-1 to 509-n-M provided with the first phase rotation unit and the second phase rotation unit individually Also good.

  In addition, the wireless communication device includes distribution couplers 504-1 to 504-N included in the transmission / reception signal processing circuits 551-1 to 551-N and a distribution coupler 504-n included in the transmission / reception signal processing circuit 553-n. A function unit in which the signal synthesis unit and the signal distribution unit are integrated may be provided for each subarray. In this case, the wireless communication device is a switching unit or a circulator 521-n that temporally switches between the processing of the signal combining unit and the processing of the signal distributing unit in the signal combining / distributing unit such as the TDD switches 127-1 to 127-N. As described above, the output port may be different for each input port, and a switching unit that switches the direction of the flowing signal may be further provided for each subarray. In addition, the wireless communication apparatus separately sets the signal synthesis unit and the signal distribution unit, like the distribution couplers 514-n and 515-n included in the transmission / reception signal processing circuits 554-n, 555-n, and 556-n. You may prepare.

[Additional matters regarding the embodiment]
Supplementary matters regarding the embodiment of the present invention described above will be described below.
You may make it implement | achieve the radio station apparatus in embodiment mentioned above with a computer. In that case, a program for realizing this function may be recorded on a computer-readable recording medium, and the program recorded on this recording medium may be read into a computer system and executed. Here, the “computer system” includes an OS and hardware such as peripheral devices. The “computer-readable recording medium” refers to a storage device such as a flexible medium, a magneto-optical disk, a portable medium such as a ROM and a CD-ROM, and a hard disk incorporated in a computer system. Furthermore, a “computer-readable recording medium” dynamically holds a program for a short time, like a communication line when transmitting a program via a network such as the Internet or a communication line such as a telephone line. In this case, a volatile memory in a computer system serving as a server or a client in that case may be included and a program that holds a program for a certain period of time. Further, the program may be for realizing a part of the functions described above, and may be a program capable of realizing the functions described above in combination with a program already recorded in the computer system. It may be realized using hardware such as PLD (Programmable Logic Device) or FPGA (Field Programmable Gate Array).

  As mentioned above, although embodiment of this invention has been described with reference to drawings, the said embodiment is only the illustration of this invention, and it is clear that this invention is not limited to the said embodiment. is there. Therefore, additions, omissions, substitutions, and other modifications of the components may be made without departing from the technical idea and scope of the present invention.

  The present invention can be applied to a wireless communication device that transmits and receives wireless signals using a plurality of antennas.

120-1 to 120-N ... modulators 122-1 to 122-N, 122-n ... D / A converters 123-1 to 123-N, 123-n ... up-converters 124-1 to 124-N, 124 -N ... down converters 125-1 to 125-N, 125-n ... A / D converters 127-1 to 127-N ... TDD switches 130-1 to 130-N ... demodulator 140 ... baseband signal processing circuit 141 ... Signal separation circuits 401-1 to 401 -M ... antenna elements 401-1-1 to 401 -NM, 401 -n-1 to 401 -nM ... antenna elements 402-1-1 to 402 -N- M, 402-n-1 to 402-nM ... Phase shifters 403-1-1 to 403-NM, 403-n-1 to 403-nM ... Switches 404-1 to 404-N ... Distribution couplers 405-1 to 405-N, 40 -N ... correlation calculation circuits 406-1 to 406-N, 406-n ... phase shift control circuits 407-1 to 407-M ... distribution couplers 408-n-1 to 408-n-M ... TDD switch 409-n -1 to 409-nM ... phase shifter 414-n ... distribution coupler 415-n ... distribution coupler 424-1-1 to 424-NM, 424-n-1 to 424-nM ... Down converters 425-1-1 to 425-NM, 425-n-1 to 425-nM ... A / D converter 450 ... wireless station devices 451-1 to 451-N ... transmission / reception signal processing circuit 452 ... Radio station apparatus 453-n ... transmission / reception signal processing circuit 460 ... control circuits 501-1 to 501-M ... antenna elements 501-1-1 to 501-NM, 501-n-1 to 501-nN ... antennas Elements 502-1-1-1 to 502-NM, 502 n-1 to 502-nN phase shifters 503-1-1 to 503-NM, 503-n-1 to 503-nN, switches 504-1 to 504-N, 504-n,. Distribution couplers 505-1 to 505-N, 505-n ... correlation calculation circuits 506-1 to 506-N, 506-n ... phase shift control circuits 507-1 to 507-M ... distribution couplers 508-n-1 508-n-M TDD switches 509-n-1 to 509-nM, 509-n-1 to 509-n-N phase shifter 514-n, distribution coupler 515-n, distribution coupler. 521-n ... circulators 541-1 to 541-M ... antenna element 541-n-1 to 541-n-M ... antenna element 550 ... radio station apparatus 551-1 to 551-N ... transmission / reception signal processing circuit 552 ... radio station Device 553-n. Transmission / reception signal processing circuit 554- n ... transmission / reception signal processing circuit 555-n ... transmission / reception signal processing circuit 556-n ... transmission / reception signal processing circuit 560 ... control circuit 901-1 to 901-N ... modulator 902 ... precoder 903-1 to 903-N ₀ ... IFFT & GI assignment Circuits 904-1 to 904-N ₀ ... D / A converters 905-1 to 905-N ₀ ... Up converters 906-1 to 906-N ₀ ... Down converters 907-1 to 907-N ₀ ... A / D conversion 908-1 to 908-N ₀ GI removal & FFT circuit 909 Post coder 910-1 to 910-N Demodulator 911 TDD switch 912-1 to 912-N ₀ Distribution coupler 913-1-1 913-N ₀ -M ₀ ... phase shifters 915-1 to 915-M ₀ ... distribution couplers 916-1 to 916-M ₀ ... antenna elements 921-1 to 921-N ... time Axis transmission weight multiplication circuit 922-1 to 922-M... D / A converters 922-1-1 to 922-NM ... D / A converters 923-1 to 923-M ... Upconverter 923-1-1 923-NM ... Up converters 924-1 to 924-M ... Down converters 924-1-1 to 924-NM ... Down converters 925-1 to 925-M ... A / D converter 925-1 1 to 925-NM ... A / D converters 926-1 to 926-N ... time-axis reception weight multiplication circuits 927-1 to 927-N ... TDD switches 928-1 to 928-M ... antenna elements 928-1 -1 to 928-NM ... Antenna elements 929-1 to 929-N ... Transmission / reception signal processing circuits 941-1 to 941-M ... Distribution coupler 942 ... Radio station apparatuses 943-1 to 943-M ... Adder / synthesizer 944-1 44-M, replicator 945, radio station apparatuses 951-1 to 951-3, high power amplifiers 952-1 to 952-3, low noise amplifiers 953-1 to 953-3, and TDD switches 954-1 to 954-3. Antenna elements 955-1 to 955-3 ... wireless module

Claims (6)

A wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with other wireless communication devices,
A signal receiving unit that receives a signal transmitted by the other wireless communication device via each of the antenna elements used for signal reception;
At least for each array antenna used for signal reception, among the signals received by the signal receiving unit, a system of antenna elements for each received signal received via the antenna element constituting the corresponding array antenna A first phase rotation unit that rotates and outputs a complex phase on an analog signal each time;
A switching unit that is provided at least for each of the array antennas used for signal reception and can switch for each system of antenna elements whether or not to pass the received signal output from the corresponding first phase rotation unit;
When the received signal is to be reproduced, all or part of the received signal output from the corresponding first phase rotation unit is passed, and the received signal is subjected to training for calculating the rotation amount of the complex phase. A switching control unit that controls switching in the switching unit so that the received signal output from the corresponding first phase rotation unit is passed one by one for each system of antenna elements when it is a signal,
A signal synthesizing unit that is provided for at least each of the array antennas used for signal reception and synthesizes the received signal that has been passed by the corresponding switching unit;
A signal conversion unit that is provided for at least each of the array antennas used for signal reception and converts the reception signal synthesized by the corresponding signal synthesis unit from an analog signal to a baseband digital signal;
A signal reproduction unit that reproduces a signal transmitted by the other wireless communication device based on the reception signal to be reproduced converted by the signal conversion unit;
Correlation for calculating a correlation for each combination of a reference antenna element and another antenna element in the antenna element of the training signal using the training signal that the switching unit passes one for each antenna element system A calculation unit;
A first rotation amount calculation unit that calculates a rotation amount of a complex phase to be given in the first phase rotation unit based on a correlation calculation result by the correlation calculation unit;
A wireless communication apparatus comprising: A transmission signal generating unit that generates an analog transmission signal to be transmitted to another wireless communication device using the array antenna;
A signal distribution unit provided at least for each of the array antennas used for signal transmission and branching the analog transmission signal transmitted using the corresponding array antenna in correspondence with each of the antenna elements constituting the array antenna; ,
A second phase rotation unit that is provided for at least each of the array antennas used for signal transmission and rotates a complex phase on an analog signal for each of the transmission signals branched by the corresponding signal distribution unit;
A signal transmission unit configured to transmit the transmission signal whose phase is rotated by the second phase rotation unit via the antenna element constituting the array antenna corresponding to the second phase rotation unit;
The calculation result of the correlation calculated by the correlation calculation unit, the calculation result of the rotation amount of the complex phase calculated by the first rotation amount calculation unit, or the complex phase obtained by performing calibration processing on the calculation result A second rotation amount calculation unit that calculates a rotation amount of a complex phase to be given in the second phase rotation unit based on any of the information on the rotation amount of
The wireless communication apparatus according to claim 1. Instead of the first phase rotation unit and the second phase rotation unit, a phase rotation unit that performs processing of the first phase rotation unit and processing of the second phase rotation unit is provided for each array antenna. Prepare
The wireless communication apparatus according to claim 2. Instead of the signal synthesis unit and the signal distribution unit, each array antenna includes a signal synthesis distribution unit that performs the processing of the signal synthesis unit and the processing of the signal distribution unit,
A switching unit that switches between the processing of the signal combining unit and the processing of the signal distributing unit in the signal combining / distributing unit is further provided for each array antenna.
The wireless communication apparatus according to claim 2 or claim 3, wherein The signal synthesis unit and the signal distribution unit are individually independent, and the first phase rotation unit and the second phase rotation unit are individually independent.
The wireless communication apparatus according to claim 2 or claim 3, wherein A wireless communication method executed by a wireless communication device that forms directivity using one or a plurality of array antennas including a plurality of antenna elements and wirelessly communicates with other wireless communication devices,
A signal receiving process in which the signal receiving unit receives the signal transmitted by the other wireless communication device via each of the antenna elements used for signal reception;
At least a phase rotation unit provided for each array antenna used for signal reception receives an antenna for each received signal received through the antenna element constituting the corresponding array antenna among signals received in the signal receiving process. A phase rotation process for rotating a complex phase on an analog signal for each element system;
At least a switching unit provided for each array antenna used for signal reception, when the received signal is a reproduction target, all of the received signals whose complex phases have been rotated in the phase rotation process by the corresponding phase rotation unit. Alternatively, when the received signal is a training signal for calculating the amount of rotation of the complex phase through a part of the received signal, the received signal obtained by rotating the complex phase in the phase rotation process by the corresponding phase rotating unit is an antenna. A switching process of passing one by one for each element system;
A signal combining unit that combines at least a signal combining unit provided for each array antenna used for signal reception to combine the received signals passed in the switching step by the corresponding switching unit;
A signal conversion unit provided at least for each array antenna used for signal reception converts the received signal synthesized in the signal synthesis process by the corresponding signal synthesis unit from an analog signal to a baseband digital signal. Process,
A signal reproduction process in which a signal reproduction unit reproduces a signal transmitted by the other wireless communication device based on the reception signal to be reproduced, which is converted into a digital signal in the signal conversion process;
The correlation calculation unit uses the training signal that has been passed one by one for each antenna element system in the switching process, for each combination of the reference antenna element in the antenna element of the training signal and another antenna element. A correlation calculation process for calculating the correlation;
A rotation amount calculation step in which a rotation amount calculation unit calculates a rotation amount of a complex phase to be given in the phase rotation process, based on a correlation calculation result in the correlation calculation process;
A wireless communication method comprising:

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