JP6397383B2 - Wireless communication apparatus and wireless communication method - Google Patents

Description

  The present invention relates to a wireless communication apparatus and a wireless communication method.

[Preface]
Currently, with the explosive spread of smartphones, convenient frequency resources in the microwave band are depleted. As countermeasures, a shift from a third-generation mobile phone to a fourth-generation mobile phone and the allocation of a new frequency band are being carried out. However, since there are many businesses that want to provide services, the frequency resources allocated to each business are limited.

  In mobile phone services, studies are underway to improve frequency utilization efficiency with a multi-antenna system that uses multiple antenna elements. The already popular wireless standard IEEE (Institute of Electrical and Electronics Engineers, Inc.) 802.11n uses MIMO (Multiple Input Multiple Output) transmission technology that uses a plurality of antenna elements for both transmission and reception. To perform spatial multiplexing transmission. Thereby, in IEEE 802.11n, the transmission capacity is increased to improve the frequency utilization efficiency. The term MIMO is generally used on the assumption that both the transmitting station and the receiving station are provided with a plurality of antenna elements. When the receiving side is a single antenna element, the term MISO (Multiple Input Single Output) is used instead of MIMO. However, in the following, the term MIMO is used to encompass all of these.

  Moreover, as a recent communication technology, as in OFDM (Orthogonal Frequency Division Multiplexing) modulation scheme and SC-FDE (Single Carrier Frequency Domain Equalization) scheme, it is divided into a plurality of frequency components (subcarriers) on the frequency axis. A method of performing signal processing is common. The following description will be made using the term “subcarrier” on the premise of a general scheme common to them without particularly distinguishing between OFDM and SC-FDE.

In the MIMO transmission technique, it is possible to perform more efficient transmission by knowing transmission path information between a transmitting station and a receiving station. As the simplest example, the case of a plurality of antennas on the receiving side has been shown. However, when N antenna elements are provided on the transmitting side and only one antenna element is provided on the receiving side, Directivity control is performed on the transmission side so that the signal to be transmitted is synthesized in phase at the antenna element on the reception side. Thereby, the line gain can be increased. Specifically, when channel information between the j-th antenna element of the transmitting station and the antenna element of the receiving station in the k-th subcarrier is h _j ^(k) , the following equation ( The transmission weight w _j ^(k) of ¹⁾ is calculated, and the product obtained by multiplying this by the transmission signal is transmitted from each antenna element (equal gain synthesis). Note that the channel information in this specification is based on circuit configurations and signal processing and control procedures in actual operation, and strictly speaking, amplifiers, filters, etc. in transmission and reception RF (Radio Frequency) circuits. It is assumed that it includes complex phase rotation and amplitude variation information.

A vector (h ₁ ^(k) ,..., H _j ^(k) ,..., H _N ^(k) ) having channel information corresponding to each of the first to N-th antenna elements on the transmission side as a channel vector h ^(K) . Also, vectors (w ₁ ^(k) ,..., W _j ^(k) ,..., W _N ^(k) ) ^T (T) having transmission weights corresponding to the first to Nth antenna elements on the transmission side as components. Represents a transposition.) Is called a transmission weight vector w ^(k) . Strictly speaking, the channel vector in the downlink → h ^(k) (the symbol “→” before “h ^(k) ” is a symbol given to h to represent the vector) is a row. The vector, transmission weight vector → w ^(k) should be expressed as a column vector. However, in the following, for simplicity, the symbol “→” is omitted and the row vector and the column vector are not distinguished. In the following description, the notation regarding the reception signal Rx, the transmission signal Tx, and the noise n should be clearly indicated by adding “→” to be a vector, but “→” because there is no other confusing notation. The description is omitted. The reception signal Rx is given by the following equation (2) with respect to the transmission signal Tx and the noise n.

When Expression (1) is substituted into Expression (2), a value obtained by adding the absolute values of the components h _j ^(k) of the channel vector h ^(k) over all antenna components is obtained as the channel gain. If the transmission power from each antenna element is kept the same as when transmitting with a single antenna, with N antenna elements, the amplitude of the received signal is N times that when transmitting with one antenna element. Expected. The received signal power is improved to N ² times because it is proportional to the square of the amplitude. This value is a gain when a plurality of antenna elements are used as an array antenna. Strictly speaking, the gain of the array antenna itself is generally interpreted as N times the reception power (which is the result of evaluation under a constant total transmission power). Assuming the environment, the case where the transmission power per element is constant will be described as a reference.

  In general, it is known from Shannon's theorem that the increase in transmission capacity with respect to the improvement in SNR (Signal-Noise Ratio) is larger in the low SNR region and smaller in the high SNR region. For this reason, it is often the case that a plurality of antenna elements are provided on the receiving side and the transmission capacity is improved by spatial multiplexing rather than aiming to improve the transmission capacity by improving the line gain. MIMO transmission technology aims to increase transmission capacity by spatial multiplexing.

FIG. 1 shows an overview of MIMO transmission. Here, as an explanation focusing on a certain frequency, the subscript “k” representing a subcarrier or frequency is omitted. In FIG. 1, reference numeral 11 represents a transmitting station, and reference numeral 12 represents a receiving station. In this example, both the transmitting station 11 and the receiving station 12 are provided with two antenna elements, and channel information (amplitude and complex phase) between the transmitting antenna # 1 of the transmitting station 11 and the receiving antenna # 1 of the receiving station 12 is provided. information) indicating the amount of rotation h _11, the transmission antenna # 1 and channel information between the receiving antenna # 2 of the receiving station 12 (amplitude, information indicating the amount of rotation of the complex phase) to h ₂₁ of the transmission station _11, transmission Channel information between the transmitting antenna # 2 of the station 11 and the receiving antenna # 1 of the receiving station 12 (information indicating the amount of rotation of the amplitude and complex phase) is h ₁₂ , and the transmitting antenna # 2 of the transmitting station 11 and the receiving station 12 If channel information (information indicating amplitude and complex phase rotation amount) between the receiving antenna # 2 and h ₂₂ is represented as h ₂₂ , signals t ₁ and t ₂ transmitted from the two transmitting antennas of the transmitting station 11 and , Two reception of the receiving station 12 Between the signals r ₁ and r ₂ received by the antenna, the noise signals n ₁ and n ₂ are used and expressed by the following formula (3).

  Basically, in MIMO transmission, the signal on the transmission side is estimated based on the reception signal on the reception side and the channel matrix. If the noise term in Equation (3) is sufficiently small, the transmission signal can be estimated from the reception signal by multiplying both sides by the inverse matrix of the channel matrix. By multiplying a predetermined transmission weight matrix on the transmission side and further multiplying a predetermined reception weight on the reception side, the transmission characteristics can be improved, and more efficient transmission becomes possible. For example, when channel information between a plurality of antenna elements on the transmitting side and antenna elements on the receiving side is known, the channel matrix is subjected to singular value decomposition (SVD) and transmission is performed in the eigenmode. This maximizes the transmission capacity.

  Specifically, as shown in the following equation (4), the channel matrix H is decomposed into a diagonal matrix D having unitary matrices U and V and a singular value λ as diagonal components.

  At this time, if the unitary matrix V is used as the transmission weight matrix, the reception signal vector Rx is given by the following equation (5) with respect to the transmission signal vector Tx and the noise vector n.

On the receiving side, the following equation (6) is obtained by multiplying Hermitian conjugate matrix U ^H of unitary matrix U.

In equation (6), since the non-diagonal component of the diagonal matrix D is zero, the cross term of the transmission signal is already canceled and the signal is separated. In addition, the noise vector is multiplied by the reception weight matrix U ^H and converted into a noise vector n ′ expressed by rotating the coordinate axis, but the statistical characteristics of the vector remain equivalent to the original noise vector. FIG. 2 shows a conceptual diagram of eigenmode transmission. 2A shows a basic MIMO channel, FIG. 2B shows a situation where multiplication of transmission / reception weight matrices is performed, and FIG. 2C shows a virtual transmission path formed by eigenmode transmission. Yes. The channel matrix H between the transmitting and receiving antennas of the transmitting station 11 and the receiving station 12 includes a matrix D having a singular value λ in each diagonal component by singular value decomposition and two unitary as shown in the equation (4). matrix U, represented by the product of ^{V H.} Here, when the transmission weight matrix V is used on the transmission side and the reception weight matrix U ^H is used on the reception side as shown in FIG. 2B and Equation (5), the unitary matrix portion of Equation (4) is multiplied and unit. As a result, as shown in the equation (6), the non-diagonal component is zero and only the diagonal component can be represented by a non-zero matrix D. As shown in FIG. 2C, this corresponds to a situation in which transmission paths of virtual channels represented by singular values λ ₁ , λ ₂ . At this time, the square value of the absolute value of each singular value λ corresponds to the line gain of the individual signal series. Each singular value λ is different for each signal system. By optimizing the transmission mode in accordance with the singular value of this eigenmode, the transmission capacity can be maximized. The transmission mode is a specific mode of signal transmission determined by a combination of the modulation multi-level number and the error correction coding rate.

  Here, if each component of the channel matrix of the MIMO channel is independent and uncorrelated, the absolute value of each singular value is a relatively large value. For example, when there are many reflected waves and the received power of the line-of-sight wave is relatively low, each singular value has a relatively large value as described above. On the other hand, when the transmission / reception antenna is in a line-of-sight environment and there are not many reflected waves, only the absolute value of the first singular value is extremely large, and the absolute value of the singular value after the second singular value is extremely large. There is a tendency to become smaller. For this reason, it is generally said that MIMO transmission is suitable for a multipath environment, and is not so suitable when a line-of-sight wave is dominant.

  The above is a description of the single user MIMO transmission technique assuming one base station apparatus and one terminal station apparatus. The same description can be extended to multi-user MIMO that performs communication on the same frequency axis at the same time between one base station apparatus and a plurality of terminal station apparatuses. In multi-user MIMO, each terminal station apparatus generally performs communication with a smaller number of antenna elements than the total number of signal systems to be spatially multiplexed. Therefore, on the downlink, directivity control for suppressing inter-user interference is performed in advance on the transmission side. Although the specific expressions are slightly different, basically, as in the above eigenmode transmission, the transmission weight corresponding to the channel matrix is used after grasping the channel matrix.

  Further, in the above description, the description has been focused on the downlink. However, in the uplink as well, communication using the channel information can be performed after grasping the channel information in advance. For example, in the processing as the array antenna described first, the weight of in-phase synthesis given by equation (1) is used as a reception weight, and the weight of maximum ratio synthesis is given by equation (7) below. It is also possible to use one.

The constant C in Equation (7) is a coefficient that is determined as appropriate. Among the components of the vector, C having a large absolute value of h _j ^(k) is added with a large weight, and C is determined so that a small signal is added with a small weight. As a result, a signal with a large SNR is emphasized, and adjustment is made so that noise of a signal with a small SNR does not excessively affect the signal.

  Note that downlink channel information is required to calculate the transmission weight, which can be realized by various forms of channel feedback. In the simplest example, it is possible for the terminal station apparatus to receive the training signal transmitted from the base station apparatus in the downlink, and to notify the reception result accommodated in the uplink control information. In general, a channel estimation method for performing such a loopback is called explicit feedback. In addition to this, for example, a calibration coefficient necessary for conversion between uplink channel information and downlink channel information inside the base station apparatus is acquired in advance, and channel estimation is performed on the uplink. Then, it is also possible to estimate downlink channel information by multiplying the calibration coefficient. This method is called implicit feedback. In the case of a large-scale antenna with a large number of antennas, implicit feedback is generally advantageous. However, the channel feedback method is not particularly limited in the present invention, and a general channel estimation method can be used.

[Direction of future mobile networks]
As described above, with the explosive spread of smartphones, further increase in transmission capacity is required. In the current wireless communication research, the technical study for the 5th generation mobile phone following the 4th generation mobile phone is being studied. Here, the transmission capacity is more than 10 times that of the 4th generation. It is demanded. Here, not only an increase in transmission capacity of one base station apparatus and a radio system under the base station apparatus but also an increase in transmission capacity per unit area is required. Specifically, in areas where many people gather, such as Shinjuku, Shibuya, Ginza, and Otemachi, it is not possible to deal with simply by increasing the transmission capacity of the wireless system, and the area area covered by one base station device is reduced (hereinafter, It is called “small cell”), and the same transmission capacity is realized in a smaller area, and a transmission capacity proportional to the number of small cells is realized by setting a large number of small cells. However, since this small cell is required to be installed in a place where people gather and further transmission capacity is required, it is not possible to sufficiently design the station placement like a macro cell having a large area. Originally, when multiple frequency channels are available to introduce a small cell while the frequency resources are depleted, it is not used as frequency repetition (frequency reuse), but multiple resources are used at the same place. It is preferable to increase the total transmission capacity by using the frequency channels. Therefore, there is a need for a technique that enables repeated installation of small cells at a relatively short distance without designing a station even for the same frequency channel.

  Furthermore, in order to realize a transmission capacity of 10 Gbit / s or more in the depletion of frequency resources in the microwave band, it is necessary to secure a certain frequency bandwidth. For this purpose, utilization of a higher frequency band is expected. The However, in terms of line design, the free space propagation loss increases by 20 dB when the frequency is increased by 10 times. Therefore, if the transmission power is the same, the propagation distance is reduced to 1/10 in the line-of-sight environment. Furthermore, increasing the output of a high-power amplifier on the transmission side becomes more difficult as the frequency becomes higher, and the transmission power per antenna is limited, but a transmission capacity of 10 Gbit / s or more is sufficient for the line design. A technology that can be realized is required.

  From this point of view, large-scale MIMO (Massive MIMO) transmission technology is currently attracting attention. In Massive MIMO transmission technology, the number of antennas on the base station apparatus side is increased by at least an order of magnitude from the conventional MIMO transmission, which is about several at the maximum, and a large number of antenna elements of tens to hundreds are used. Thus, the line gain to the terminal station device as the destination is improved, and the interference to the terminal station devices other than the destination is reduced. There are various variations in the implementation method and application method of Massive MIMO. For so-called small cells, the reduction of interference to terminal station apparatuses other than the destination is utilized for “inter-cell interference suppression”. As a matter of course, Massive MIMO transmission technology is not limited to single-user MIMO in which a large-scale MIMO matrix is simply used in a single terminal station device, as well as in the conventional technology, and simultaneous communication is performed in a plurality of terminal station devices. Also used as multi-user MIMO for performing Here, in the use as multi-user MIMO, unlike the case of a small cell, it is also possible to utilize a very redundant number of antenna elements for “suppression of inter-user interference” between terminal station devices in the same area. It is. In the following, a large-scale antenna system (see, for example, Non-Patent Document 1 to Non-Patent Document 4) will be briefly described as an example of a technique related to these large-scale antennas.

[Outline of large-scale antenna system]
FIG. 3 is a diagram showing an outline of a large-scale antenna system. In FIG. 3, the base station apparatus 1, the radio station apparatus 2, the line-of-sight wave 3, the stable reflected wave 4 by the structure, the multiple reflected waves 5-6 near the ground, and the structure 7 are shown. In the large-scale antenna system of FIG. 3, the base station apparatus 1 includes a large number (for example, 100 or more) of antenna elements, and is installed at a high place such as a rooftop of a building or a high steel tower. Similarly, the radio station apparatus 2 is installed at a high place such as on the roof of a building, on the roof of a house, on a telephone pole or a steel tower. Therefore, the base station apparatus 1 and the radio station apparatus 2 are generally in a line-of-sight environment, and in the meantime, in addition to the path of the line-of-sight wave 3, the path of the stable reflected wave 4 by the large stable structure 7, etc. There are mixed paths of multiple reflected waves 5 and 6 due to nearby vehicles such as cars and people. Note that the reception level of the multiple reflected waves 5 and 6 near the ground is lower than that of the line-of-sight wave 3 and the stable reflected wave 4, particularly when a directional antenna is used.

  FIG. 4 is a diagram illustrating impulse responses in the line-of-sight environment and the non-line-of-sight environment. 4A shows an impulse response in a non-line-of-sight environment, and FIG. 4B shows an impulse response in a line-of-sight environment. 4A and 4B, the horizontal axis represents the delay time, and the vertical axis represents the reception level of each delayed wave. In the case of the non-line-of-sight environment shown in FIG. 4 (a), there are no direct wave components in the line-of-sight section, many multiple reflected waves of various paths exist as components, and each amplitude and complex phase becomes intense with time. fluctuate.

  In contrast, when a line-of-sight environment such as the large-scale antenna system shown in FIG. 3 is assumed, the level of the stable path of the line-of-sight 3 and the stable reflected wave 4 by the structure 7 is high. When the delay amount is generally larger than the stable reflected wave 4 by the line-of-sight wave 3 and the structure 7, the multiple reflected wave of the variable path is relative as shown in FIG. The level becomes smaller. When such channel information is acquired a plurality of times and averaged, the components of the stable path are stable each time in both amplitude and complex phase, and the same value is obtained. However, the components of the time-varying path are synthesized and averaged randomly in the complex space and approach the average value 0. Therefore, only stable components can be extracted effectively by averaging. The absolute channel information depends on the symbol timing, and if the symbol timing is different, channel information cannot be properly averaged. In order to solve such a problem, Non-Patent Document 2 considers that relative channel information (or each channel information is divided by the channel information of the reference antenna) based on the complex phase of the reference antenna element. The technology that utilizes (good) is introduced. When such averaging is not performed, it is possible to discuss using absolute channel information without using relative channel information. There is no problem even if it is used. In the following description, the channel information is basically handled as relative channel information based on the complex phase of the reference antenna in order to comprehensively handle it including the averaging process.

  The base station apparatus 1 (see FIG. 3) calculates transmission / reception weights based on the channel information of the stable path that does not vary with time obtained in this way. The base station apparatus 1 performs directivity control for performing in-phase synthesis with a large number of antenna elements using the calculated transmission / reception weights. By using the transmission / reception weight described above, the base station apparatus 1 can increase the directivity gain to the radio station apparatus of the communication partner that is the target of directivity control in proportion to the square of the number N of antennas.

In addition, since the directivity gain of interference to radio station apparatuses other than the destination remains N times, a gap of N times is generated between the desired signal and the interference signal by simple calculation. As a result, the expected value of SIR (Signal to Interference Ratio) is ₁₀ Log ₁₀ (N) [dB]. This expected value is 20 dB when N is 100. Furthermore, when a radio station apparatus with a small correlation is selectively spatially multiplexed, further improvement in SIR characteristics is expected, and higher spatial multiplexing can be realized.

In Non-Patent Document 3 and Non-Patent Document 4, a technique for further suppressing interference that cannot be suppressed by the transmission / reception weights described above and a combination of radio station apparatuses having lower correlation of channel information (channel correlation) are selected. Technology is introduced. In order to realize super-high-order spatial multiplexing, it is important to combine radio station apparatuses having a small correlation of channel information. A channel vector h _j ^(k) (“h _j ^(k) ” ⁾ is a vector including channel information on the k-th subcarrier between a number of antenna elements of the base station apparatus and the antenna elements of the j-th radio station apparatus. Yes, the symbol “→” should be added to h to clearly indicate that it is a vector, but this is omitted. The channel correlation with the channel vector h _i ^(k) in the device is given by the following equation (8).

The system assumed a sight environment, assuming a virtual channel model consisting only of sight waves, when the correlation of the channel vector h _i between the antenna and the base station apparatus of the radio station apparatus ^_(k) is small Is suitable for spatial multiplexing, and conversely, when the correlation is large, it is not suitable for spatial multiplexing.

[About large-scale MIMO in small cells]
In the description of the above equation (8), assuming that the radio station apparatus side is a single antenna, the correlation between channel vectors is generally low due to the spatial spread of different radio station apparatuses. It was supposed. On the other hand, in the fifth generation mobile phone, for the purpose of improving throughput per user, a large number of antennas are mounted on the radio station apparatus side, and similarly, a large number of antennas are mounted on the base station apparatus side. In recent research reports, the number of antenna elements on the base station apparatus side is 256 elements, and the number of antenna elements on the radio station apparatus side is 16 elements. Yes. Here, it is assumed that the radio station device carried by the user is a small-sized device that is portable in size. Furthermore, for example, base station devices are also installed on the walls of existing buildings (for example, about 20 meters above ground) in order to reduce mutual interference between adjacent small cells and to install in places where people concentrate. A form in which directivity is given to each antenna element and a limited area is irradiated by looking down from above is expected. In this case, it is not preferable to install a large antenna on the wall surface of a building from the viewpoint of safety and ease of installation. When using high frequency bands such as millimeter waves and quasi-millimeter waves, antenna elements can be made smaller and antenna directivity can be easily formed as the wavelength becomes shorter. It is expected to use a small antenna set packed with many antennas.

  In this case, for example, if the channel correlation between the j-th antenna and the i-th antenna among a plurality of antennas in a single radio station apparatus is obtained by the above equation (8), the antenna interval of the user-side radio station apparatus is Under conditions that are very short and the line of sight between the base station apparatus and the radio station apparatus can be secured, it can be regarded as a MIMO channel with a very large channel correlation between antenna elements. As described above, when the base station device is installed at a high place such as a wall surface of a building and looks down, and the user uses a smartphone or the like in his / her hand, it is between each antenna element of the base station device and the radio station device. Is generally expected to be an outlook environment, and such a situation is expected to be a general use environment.

  In this case, when the MIMO matrix is subjected to singular value decomposition, the absolute value of the first singular value becomes a very high value using the line-of-sight component, but the higher-order singular values after the second singular value are It tends to be relatively small compared to the singular value. That is, in spatial multiplexing transmission using the MIMO channel, the first path corresponding to the first singular value is in a situation where the line gain is very large, but the higher-order path that is equal to or higher than the second singular value. Is a relatively inefficient situation.

  In order to avoid this problem, for example, it is ideal to secure a spatial spread without consolidating the antennas of the base station apparatus in a small casing. For example, if the base station apparatus antennas are assembled in a linear array with a linear arrangement of about 10 m, even if the line-of-sight wave is dominant, the spatial expansion of the antennas of the base station apparatus will result in a MIMO channel. Can be expected to increase the absolute value of higher-order singular values greater than or equal to the second singular value and contribute to an increase in capacity. However, such a large-scale structure is not preferable in terms of the antenna installation structure. For example, when the number of antenna elements on the base station apparatus side is 256 elements, installing 256 antennas individually on the wall surface of the building at intervals of about 4 cm increases the burden of installation work. On the other hand, when a structure already assembled in a linear array is installed on the wall surface of a building, it is difficult to install it on the wall surface of the building because the structure becomes large. Further, in order to transmit cooperatively with all the antenna elements, it is necessary to distribute a radio frequency signal from the casing of one base station apparatus to the antenna elements having a spatial extent of about 10 m with a cable. The cable loss of signal transmission in the high frequency band cannot be ignored. For example, assuming 20 GHz as a radio frequency, a cable loss of 10 dB or more at 10 m may occur, and the line gain earned by increasing the number of antenna elements may be impaired.

  For this reason, in large-scale MIMO operations where dominated by line-of-sight waves is required, at least when the antenna on the radio station side is required to be downsized, the capacity is increased by spatial multiplexing transmission using single-user MIMO. It becomes difficult to plan.

[For wireless entrance]
Although the above situation has been explained mainly in the small cell environment, the same problem remains in the radio entrance line as long as the line-of-sight wave is dominant. For example, when a small cell base station device for a fifth generation mobile phone is installed on the wall of a building, a large-capacity entrance line is required there. In general, the entrance line to the base station apparatus is generally provided using an optical fiber, but it is not easy to install the optical fiber on the wall of an existing building. For example, if an optical fiber is pulled to somewhere on the rooftop or the ground, and a cable is installed along the wall of the building, the appearance of the already built building may be damaged, and the building owner's consent is obtained. Hateful. Regarding the construction to penetrate the hole in the wall of the building in order to provide the optical fiber from the inside of the building to the wall of the building, it may be more difficult to obtain the consent of the building owner. In this case, for example, an optical fiber may be drawn to the rooftop of the building facing the building where the small cell base station device is installed, and an entrance line may be provided from the rooftop of the building to the small cell base station device on the wall of the building via a wireless line. Conceivable.

  The problem in this case is that the base station equipment on the wall of the building is assumed to be sufficiently downsized as before, so in the line-of-sight environment, the higher-order path above the second singular value of the MIMO channel is spatially multiplexed. It is assumed that the contribution to is limited, and positively utilizing the first singular value is effective for stable transmission.

[Wireless entrance to moving bodies such as trains]
Next, another major problem will be described in consideration of the usage form of the fifth generation mobile phone. As described above, when there are a large number of users in a very small area, it is important to be able to provide a large-capacity wireless line to the large number of users. Since the small cell, as the name suggests, constitutes a relatively small service area, the target for providing a large-capacity wireless link is a user who moves at a low walking speed at most. Therefore, it is not assumed that the majority of users move from one small cell to another (or between a small cell and a macro cell), and the load of exchange of control information for the switching does not matter so much. However, assuming a user who moves on a train or the like, for example, the timing of handover of each user is generally the same, and instantaneous mass processing is required on the network side. The load of this signal processing and the compression of transmission capacity due to the generation of a large amount of control signals are inefficient. Normally, each user in a train aggregates traffic once in the train, and this aggregated traffic is aggregated. It is preferable to secure an entrance line to an external network by bundling it as viewed from the train. This avoids a load due to an unnecessary huge amount of handover and a compression of transmission capacity due to an enormous amount of control information. This idea is grasped by the concept of a moving cell in the sense that a cell, which is a service area of a fifth generation mobile phone, moves with a train.

  The entrance line to the small cell base station device of the 5th generation mobile phone on the wall of the building just before could not be provided by optical fiber, but this train moving cell (hereinafter, Assuming a “train moving cell”), it is impossible to use an optical fiber for the entrance line, and establishing an efficient method of constructing a wireless entrance to this train moving cell is extremely This is important for mobile phones.

  Here, for example, taking an urban train as an example, the required line capacity is estimated. For example, on the Yamanote Line, it is assumed that there are 11 trains and about 100 passengers per vehicle are on board. If it is a commuting time, the majority of businessmen and university students who carry smartphones etc. occupy, and it is assumed that as many as 550 users of all the vehicles are trying to access the Internet by wireless communication. In the age of the fifth generation, if one user is communicating at an average of 10 Mbit / s due to an increase in content capacity, a capacity of about 5.5 Gbit / s is required. Considering temporal fluctuations, if a line capacity of about twice that is secured, a capacity of about 10 Gbit / s is required as a radio entrance of a train moving cell.

  In order to provide such a large-capacity line, of course, it is necessary to provide a line in a high frequency band such as millimeter wave or quasi-millimeter wave, and unlike the small cell described above, a high speed of 100 km / h or more is required. Broadband large-capacity transmission to a moving body assuming movement is unavoidable. In order to avoid frequent handovers as described above, in the radio entrance of a train moving cell, it is preferable that a section of about several hundred meters can be provided by the same base station device, and free space propagation loss in a high frequency band In view of the fact that the large is large, it is necessary to secure the line gain by the large-scale MIMO.

  Here, when a transmission capacity of about 10 Gbit / s is realized as described above, for example, assuming a frequency use efficiency of about 3 bits / Hz · s, a bandwidth of about 3 to 4 GHz is obtained without spatial multiplexing. I need. For example, when an e-band band of 70 to 80 GHz is used, when one channel occupies a bandwidth of 3 to 4 GHz, only one channel can be secured. However, on the other hand, considering that the frequency channels are segregated for each of a plurality of routes running in parallel, the bandwidth of one channel is suppressed to about 1 GHz, and operation by a plurality of channels is required. In this case, it is necessary to secure the insufficient capacity by spatial multiplexing. In the above example, about four spatial multiplexing is required.

  As explained earlier, it is essential to apply large-scale MIMO even in train moving cells. In that case, channel information necessary for directivity formation is highly accurate for gain expansion by large-scale MIMO. It is necessary to know. However, the wavelength is short in the high frequency band, and the time variation of the channel appears to be relatively large with respect to the movement of the radio station apparatus. Furthermore, when the opposing trains pass each other, the external environment fluctuates more severely than the fluctuation of the channel due to the normal linear movement, and the reflected wave component is greatly disturbed. When transmitting to the train from the base station device side, channel information is essential for directivity formation. However, if channel information requires a certain degree of estimation accuracy, channel information is very frequently used. Need to exchange each other. However, in spite of aiming to increase the transmission capacity, the overhead for control information called channel information feedback becomes a tip over at the end, avoiding a decrease in transmission capacity due to channel information feedback, Therefore, it is required that large-capacity spatial multiplexing can be realized stably even in an environment where the time fluctuations are large.

[Device configuration example of MIMO transmission]
(Overall circuit configuration)
FIG. 5 is a schematic block diagram illustrating an example of the configuration of the base station apparatus 80 in the multiuser MIMO system. Here, a multi-user MIMO system will be described. Is the same as a single user MIMO system.

  As shown in FIG. 5, the base station device 80 includes a transmission unit 81, a reception unit 85, an interface circuit 87, a MAC (Medium Access Control) layer processing circuit 88, and a communication control circuit 820. The MAC layer processing circuit 88 has a scheduling processing circuit 881.

  The base station apparatus 80 inputs / outputs data to / from an external device or network via the interface circuit 87. The interface circuit 87 detects data to be transferred on the wireless line among the input data, and outputs the detected data to the MAC layer processing circuit 88. The MAC layer processing circuit 88 performs processing related to the MAC layer in accordance with an instruction from the communication control circuit 820 that performs management control of the operation of the entire base station apparatus 80. Here, the processing related to the MAC layer includes conversion of data input / output by the interface circuit 87 and data transmitted / received on the wireless line, addition of header information of the MAC layer, and the like. In this process, the scheduling processing circuit 881 performs various scheduling processes including a combination of terminal station apparatuses that simultaneously perform spatial multiplexing in multiuser MIMO transmission. The scheduling processing circuit 881 outputs the scheduling result to the communication control circuit 820. In multi-user MIMO, signals are transmitted to a plurality of terminal station devices at a time, so that a plurality of signal sequences are output from the MAC layer processing circuit 88 to the transmission unit 81.

(Circuit configuration of the transmitter 81)
FIG. 6 is a schematic block diagram illustrating an example of the configuration of the transmission unit 81 in the base station apparatus 80 in the multiuser MIMO system. As shown in FIG. 6, the transmission unit 81 includes transmission signal processing circuits 811-1 to 811-N _SDM (N _SDM is an integer of 2 or more) and addition / synthesis circuits 812-1 to 812-N _BS-Ant (N _BS-Ant is an integer of 2 or more), IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval) giving circuits 813-1 to 813-N _BS-Ant , and D / A (digital) / Analog) converters 814-1 to 814-N _BS-Ant , local oscillator 815, mixers 816-1 to 816-N _BS-Ant , filters 817-1 to 817-N _BS-Ant and high power includes amplifier (HPA) and _{818-1~818-N BS-Ant,} an antenna element _{819-1~819-N BS-Ant,} and a transmission weight processing unit 830 That. Transmission signal processing circuits 811-1 to 811-N _SDM and transmission weight processing unit 830 are connected to communication control circuit 820 shown in FIG.

The transmission weight processing unit 830 includes a channel information acquisition circuit 831, a channel information storage circuit 832, and a multiuser MIMO (MU-MIMO) transmission weight calculation circuit 833. Here, _{N SDM} subscript of the transmission signal processing circuit _{811-1~811-N SDM} in FIG. 6 represents a multiplex number for performing spatial multiplexing simultaneously. In addition, N _BS-Ant of the subscript of the circuit from the adder / synthesizer circuit 812-1 to 812-N _BS-Ant to the antenna elements 819-1 to 819-N _BS-Ant is the number of antenna elements provided in the base station apparatus 80 Represents. N _BS-Ant is, for example, 100.

In multi-user MIMO, in order to transmit signals to a plurality of terminal station devices at once, a plurality of signal sequences are input from the MAC layer processing circuit 88 to the transmission unit 81, and the input plurality of signal sequences are transmitted signals. Input to the processing circuits 811-1 to 811-N _SDM . When the transmission signal processing circuits 811-1 to 811-N _SDM receive data (data input # 1 to #N _SDM ) to be transmitted to each destination terminal station device from the MAC layer processing circuit 88, the transmission signal processing circuits 811-1 to 811-N _SDM A radio packet to be transmitted is generated and modulated. Here, for example, if the OFDM modulation method is used, the signal of each signal sequence is modulated for each subcarrier. Further, the baseband signal subjected to modulation processing is multiplied by a transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819 -N _BS-Ant is subjected to the remaining signal processing as necessary, and is added and synthesized as a sampling signal of the transmission signal in the baseband. −1 to 812-N Input to _BS-Ant .

The signals input to the adder / synthesizers 812-1 to 812-N _BS-Ant are synthesized for each subcarrier. The synthesized signal is converted from a signal on the frequency axis to a signal on the time axis by IFFT & GI adding circuits 83-1 to 813-N _BS-Ant , and further, a guard interval is inserted or between OFDM symbols (SC-FDE ( In the case of single-carrier frequency domain equalization), processing such as waveform shaping between blocks is performed, and D / A converters 814-1 to 814-1 are provided for each of the antenna elements 819-1 to 819 -N _BS-Ant. 814-N _BS-Ant converts the digital sampling data into a baseband analog signal. Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816-N _BS-Ant and up-converted to a radio frequency signal. Here, since the upconverted signal includes a signal in a region outside the band of the channel to be transmitted, the out _- of-band component is removed by the filters 817-1 to 817-N _BS-Ant and the electric signal to be transmitted is transmitted. A typical signal. The generated signal is amplified by the high-power amplifier _{818-1~818-N BS-Ant,} and transmitted from the antenna elements _{819-1~819-N BS-Ant.}

In FIG. 6, after adding and synthesizing the signals of the subcarriers by the adding and synthesizing circuits 812-1 to 812-N _BS-Ant , processing such as IFFT processing, insertion of guard intervals, waveform shaping, and the like is performed. However, the transmission signal processing circuits 811-1 to 811-N _SDM perform these processes, and the IFFT & GI adding circuits 83-1 to 813-N _BS-Ant may be omitted at this position in terms of function distribution. Good. In this case, the remaining signal processing as necessary after transmission weight multiplication in the transmission signal processing circuits 811-1 to 811-N _SDM refers to processing such as IFFT processing, insertion of guard intervals, waveform shaping, and the like.

Also, the transmission weights multiplied by the transmission signal processing circuits 811-1 to 811-N _SDM are acquired from the multi-user MIMO transmission weight calculation circuit 833 provided in the transmission weight processing unit 830 at the time of signal transmission processing. In the transmission weight processing unit 830, the channel information acquisition circuit 831 separately acquires the channel information acquired by the reception unit 85 via the communication control circuit 820, and sequentially updates the channel information in the channel information storage circuit 832. Remember. At the time of signal transmission, in accordance with an instruction from the communication control circuit 820, the multi-user MIMO transmission weight calculation circuit 833 reads channel information corresponding to the destination terminal station device from the channel information storage circuit 832 and based on the read channel information A transmission weight corresponding to a combination of terminal station apparatuses as destinations is calculated. The multiuser MIMO transmission weight calculation circuit 833 outputs the calculated transmission weight to the transmission signal processing circuits 811-1 to 811-N _SDM .

  In addition, the communication control circuit 820 manages control related to overall communication such as management of the destination terminal station apparatus and overall timing control. The communication control circuit 820 outputs information indicating a destination terminal station apparatus and the like to the transmission weight processing unit 830 that performs signal processing related to the above-described transmission weight calculation.

(Circuit configuration of the receiver 85)
FIG. 7 is a schematic block diagram illustrating an example of the configuration of the reception unit 85 in the base station apparatus 80 in the multiuser MIMO system. As shown in FIG. 7, the receiving unit 85 includes antenna elements 851-1 to 851-N _BS-Ant , low noise amplifiers (LNA) 852-1 to 852-N _BS-Ant , a local oscillator 853, and a mixer 854. -1 to 854-N _BS-Ant , filters 855-1 to 855-N _BS-Ant , A / D (analog / digital) converters 856-1 to 856-N _BS-Ant , and FFT (Fast Fourier) Transform (fast Fourier transform) circuits 857-1 to 857 -N _BS-Ant , reception signal processing circuits 858-1 to 858 -N _SDM, and a reception weight processing unit 860. Reception signal processing circuits 858-1 to 858 -N _SDM and reception weight processing unit 860 are connected to communication control circuit 820 shown in FIG. 5. Reception weight processing section 860 includes channel information estimation circuit 861 and multiuser MIMO (MU-MIMO) reception weight calculation circuit 862.

The signals received by the antenna elements 851-1 to 851-N _BS-Ant are amplified by the low noise amplifiers 852-1 to 852-N _BS-Ant . The amplified signal and the local oscillation signal output from the local oscillator 853 are multiplied by the mixers 854-1 to 854-N _BS-Ant , and the amplified signal is down-converted from a radio frequency signal to a baseband signal. The Since the downconverted signal includes a signal even in a region outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855-NBS _-Ant . The signal from which the out-of-band component is removed is converted into a digital baseband signal by the A / D converters 856-1 to 856-N _BS-Ant . All digital baseband signals are input to the FFT circuits 857-1 to 857 -NBS- _Ant , and signals on the time axis are frequency axes at a predetermined symbol timing determined by a timing detection circuit omitted here. Convert to the above signal (separate into each subcarrier signal). The signals separated into the subcarriers are input to reception signal processing circuits 858-1 to 858 -N _SDM and also input to channel information estimation circuit 861.

The channel information estimation circuit 861 uses the antenna element of each terminal station apparatus and the base station apparatus 80 based on a known signal for channel estimation (a preamble signal or the like given to the head of a radio packet) separated into each subcarrier. Channel information between each of the antenna elements 851-1 to 851-NBS- _Ant is estimated for each subcarrier, and the estimation result is output to the multiuser MIMO reception weight calculation circuit 862. Multiuser MIMO reception weight calculation circuit 862 calculates reception weights to be multiplied for each subcarrier based on the input channel information. At this time, reception weights for synthesizing signals received by the respective antenna elements 851-1 to 851-NBS- _ant differ for each signal series, and the received signal processing circuits 858-1 to 858-1 corresponding to the signal series to be extracted. 858-N input to each _SDM .

In the reception signal processing circuits 858-1 to 858 -N _SDM , the signals for each subcarrier input from the FFT circuits 857-1 to 857 -N _{BS-Ant are} input from the multiuser MIMO reception weight calculation circuit 862. The reception weights are multiplied, and the signals received by the antenna elements 851-1 to 851-N _BS-Ant are added and combined for each subcarrier. The received signal processing circuits 858-1 to 858 -N _SDM perform demodulation processing on the added and combined signals and output the reproduced data to the MAC layer processing circuit 88. In this demodulation processing, for example, a configuration may be adopted in which final signal detection is performed by, for example, performing a soft decision on a received signal once, performing a deinterleaving process as necessary, and then performing an error correction process.

Here, different received signal processing circuits 858-1 to 858 -N _SDM perform signal processing of different signal sequences. The MAC layer processing circuit 88 performs processing related to the MAC layer (for example, conversion between data input / output to / from the interface circuit 87 and data transmitted / received on the wireless line, termination of header information of the MAC layer, etc.). Do. In this process, the scheduling processing circuit 881 performs various scheduling processes including a combination of terminal station apparatuses that simultaneously perform spatial multiplexing in multiuser MIMO transmission, and outputs a scheduling result to the communication control circuit 820. The received data processed by the MAC layer processing circuit 88 is output to an external device or network via the interface circuit 87.

  In addition, the communication control circuit 820 manages control related to overall communication such as management of a transmission source terminal station and overall timing control. Also, information indicating the transmission source terminal station apparatus and the like is input from the communication control circuit 820 to the reception weight processing unit 860 that performs signal processing related to the calculation of the reception weight described above.

As for signal reception, as in the case of transmission, in the wideband system using the OFDM modulation scheme or SC-FDE scheme, the above-described reception weight multiplication is performed for each subcarrier. That is, the signals output from the A / D converters 856-1 to 856-N _BS-Ant are subjected to FFT in the FFT circuits 857-1 to 857-N _BS-Ant to be separated into subcarriers, and the separated subcarriers are separated. For each carrier, signal processing in the channel information estimation circuit 861 and reception signal processing in the reception signal processing circuits 858-1 to 858 -N _SDM are performed.

The above is description of the structure of the base station apparatus 80, the transmission part 81, and the receiving part 85 in a multiuser MIMO system. As described above, for example considers the transmission signal processing circuit 811-1～N _SDM and the reception signal processing circuit 858-1～N _SDM respectively a signal processing circuit for the signal sequence of _{N SDM} systems of single terminal station device, further If the multi-user MIMO transmission weight calculation circuit 833 and the multi-user MIMO reception weight calculation circuit 862 are regarded as transmission / reception weight calculation circuits related to single-user MIMO, basically the base station apparatus 80 in the single-user MIMO system, transmission in the above description The structure of the part 81 and the receiving part 85 is represented.

What is important here is that the local oscillator 815 in the transmission unit 81 is shared by the mixers 816-1 to 816-N _BS-Ant in each antenna system of the transmission unit 81, and the local oscillator 853 in the reception unit 85 is the reception unit. It is a point shared by mixers 854-1 to 854-NBS _-Ant in each of the 85 antenna systems. The phase of the transmission / reception signal is adjusted by each antenna. By making the phase relationship of the signal input from each local oscillator 815 or local oscillator 853 constant, transmission / reception is performed in any phase relationship. It is possible to determine whether to multiply the weight. In the case where a plurality of asynchronous oscillators are used in the transmission unit 81 or the reception unit 85, directivity control that multiplies transmission weights at least in the transmission unit 81 does not function effectively. Care should be taken in this regard when designing the device.

[Characteristics of MIMO channel dominated by line-of-sight]
First, the spatial multiplexing characteristics in the propagation path where the line-of-sight wave is dominant will be summarized. H _LOS is a channel matrix when the transmitting station and the receiving station are in a line-of-sight environment, and H _{i. i. d} . For simplicity, the _HLOS is replaced with a matrix in which each component is all “1”, and the following channel matrix is used for 16 transmission antennas and 16 reception antennas, and for two transmission antennas: 256 transmission antennas and 16 reception antennas. For the two cases, the absolute value distribution of the 16 singular values of the channel matrix given by the following equation (9) is evaluated.

FIG. 8 shows distribution characteristics of absolute values of singular values for each channel matrix. FIG. 8 (a) shows Hi _{i. i. In} the case of _d. In this case, the right side is a case where the transmission side has 256 lines. When the number of transmission / reception antennas is the same as 16, the distribution of absolute values of singular values tends to widen, and the gap between the absolute values of the first singular value and the 16th singular value tends to widen. However, when the number of transmitting or receiving antennas becomes redundant, for example, when the number of transmitting antennas is 256, the absolute value of the singular value is H _{i. i. It} is not affected by the random number value for each evaluation of _d. , and the difference between the distribution probability 0% and the 100% value is almost eliminated. This is common in FIG. 8A and FIG. 8B, but _{Hi. i. In} the case of _d. only, the gap from the first singular value to the 16th singular value is very small, whereas in the case of the rice channel of FIG. The gap between the absolute values is 20 dB, which is 10 dB larger than the Rice coefficient, while the gap between the second singular value and the sixteenth singular value is small. That is, as can be seen from FIGS. 8A and 8B, when the line-of-sight wave is dominant, the top of FIG. 2C corresponding to the first singular value is increased even if the number of antenna elements is increased. The line gain is excessively concentrated on the transmission line corresponding to the pipe of (λ ₁ ), and the second (λ ₂ ) and the third (λ ₃ ) pipe from the top in FIG. This means that the corresponding transmission line can hardly be used.

  On the other hand, when a millimeter wave or the like is used, the free space propagation loss increases depending on the frequency. For example, a gain of about 24 dB must be gained somewhere at 80 GHz with respect to 5 GHz. For this reason, it is effective to increase the size of the antenna, but it is necessary to implement the increase in the size of the antenna for spatial multiplexing and the increase in the size of the antenna for increasing the line gain from different viewpoints. .

[About channel information estimation and feedback]
In large-scale MIMO, the improvement of the line gain by utilizing a large number of antenna elements is an effect obtained by performing transmission / reception directivity control (beamforming). Information is needed. However, since the signal between one antenna element pair and one antenna element is a signal at a stage where the line gain is not improved before the directivity is formed, the estimation accuracy of the acquired channel information is generally low. . Although it is possible to improve the SNR by repeatedly receiving and averaging the same signal, this increases the overhead of channel feedback. For example, odd or even subcarriers of all subcarriers are used, and only either odd or even subcarriers are intermittently assigned to the two antenna elements and used. If channel estimation is performed based on channel information obtained from subcarriers, channel estimation for two antennas can be performed simultaneously, and the SNR of subcarriers for channel feedback can be improved by about 3 dB. It is. In principle, it is possible to improve the SNR of subcarriers for which channel feedback is performed if the period of intermittent subcarriers is changed from two subcarrier intervals to a larger subcarrier interval as described above. In the case of general channel information, there is a problem that the frequency dependency is very large, and the accuracy of channel interpolation between discontinuous subcarriers is greatly deteriorated. Conversely, in order to improve the channel interpolation accuracy between the sub-carrier, to limit the number of subcarriers N _SC to decimation, where improvement amount of the resulting SNR is to be limited. Therefore, there is a need for a technique that realizes highly accurate channel estimation and highly efficient channel feedback in a high SNR environment.

  In response to such a problem, for example, in Non-Patent Document 5, the base station apparatus combines a plurality of antennas with a predetermined transmission / reception weight, and a high-gain superdirective beam with a very narrow beam width is horizontally and vertically oriented. Proposals have been made to deal with the problem by forming a large number of steps at a predetermined angle. For example, when the terminal station apparatus exists in an area of ± 60 degrees (total of 120 degrees) in the horizontal direction, if the beam is formed in increments of 5 degrees, the horizontal direction can be covered with 25 directional beams. For example, if the user exists only in an area of about 30 degrees, for example, if the beam is formed in increments of 5 degrees, the vertical direction can be covered with seven directional beams. In order to simultaneously cover the vertical direction and the horizontal direction, a total of 175 patterns of fixed directional beams may be used. If this step size is set to 10 degrees, the entire pattern can be similarly covered with 52 patterns of fixed directional beams. If a plurality of such fixed beams are selected and considered as MIMO channels between the number of virtual antenna elements and the antenna elements of the terminal station apparatus, this virtual antenna element can be used. Thus, channel information can be acquired in a situation where a sufficient line gain can be secured. However, if the training signal is not transmitted for the number of virtual antenna elements, the terminal station side does not know which virtual antenna element to select to secure a good communication environment. The overhead required for transmission of a typical antenna element is required. In this way, it is possible to acquire MIMO channel information for realizing high-order spatial multiplexing while ensuring a sufficient line gain, but still to efficiently acquire it with a small overhead. Further ingenuity is required.

[Calibration and implicit feedback]
In an actual wireless communication apparatus, in signal processing on the transmission side, signal amplification is often performed by a high power amplifier immediately before transmission. In this case, there is an error in the amplification factor due to the individual difference of the high power amplifier, and the complex phase may rotate at a different value for each high power amplifier in the high power amplifier. Similarly, in signal processing on the reception side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, there is an error in the amplification factor due to individual differences of the low noise amplifiers, and the complex phase may rotate with a different value for each low noise amplifier in the low noise amplifier. Strictly speaking, there are individual differences in transmission and reception circuits including other RF circuits such as filters.

  In general, the amplification factor and the amount of phase rotation of the high power amplifier and the low noise amplifier have frequency dependency. If the individual difference in the amplification factor with frequency dependency and the amount of rotation of the complex phase is so large that it cannot be ignored, it is necessary to perform calibration processing when estimating the downlink channel information from the uplink channel information. is there. If the gain and phase rotation error are almost stable in time, measure the gain and phase rotation error in advance and use a coefficient to cancel the effect of the error. Converts link channel information to downlink channel information. In general, a channel estimation method using such characteristics is called implicit feedback, and is distinguished from explicit feedback in which channel information estimated in the downlink is stored in control information as digital data and notified in the uplink. ing.

  An example of the calibration process will be described below. FIG. 9 is a diagram illustrating asymmetry of channel information between the uplink and the downlink. In FIG. 9, reference numerals 25-0 to 25-2 denote wireless modules, reference numerals 21-0 to 21-2 denote high power amplifiers (HPAs), and reference numerals 22-0 to 22-2 denote low noise amplifiers (LNA). Reference numerals 23-0 to 23-2 denote time division switches (TDD-SW), and reference numerals 24-0 to 24-2 denote antenna elements.

Here, since only the functions that affect the channel information are extracted in the base station apparatus, configurations other than those illustrated are omitted, but the wireless modules 25-0 to 25-2 include other functions. Further, when the signal passes through the high power amplifiers 21-0 to 21-2, the amplitude and the complex phase are Z _{HPA # 0} (f _k ), Z _{HPA # 1} (f _k ), Z _{HPA # 2} (f _k ). Further, when the amplitude passes through various amplifiers and the complex phase rotates by θ, the influence can be expressed by a coefficient of Z = A · e ^jθ . When the signal passes through the low noise amplifiers 22-0 to 22-2, the coefficient of influence due to the amplitude and the complex phase is Z _{LNA # 0} (f _k ), Z _{LNA # 1} (f _k ), Z _{LNA # 2} ( f _k ). Here, it is assumed that there is frequency dependency as a general condition, and the frequency “(f _k )” for the k-th subcarrier is described.

Here, for example, channel information when signals are transmitted from the wireless module 25-1 and the wireless module 25-2 to the test wireless module 25-0 will be described. Here, spatial channel information between the antenna element 24-1 of the wireless module 25-1 and the antenna element 24-3 of the wireless module 25-0 is represented by h ₁ (f _k ), and the wireless module 25 -2 antenna element 24-2 and the channel information on the space between the antenna element 24-0 of the wireless module 25-0 are represented by h ₂ (f _k ).

At this time, the channel information when the signal is actually transmitted from the wireless module 25-1 to the wireless module 25-0 indicates a change accompanying the passage of the high power amplifier 21-1 at h ₁ (f _k ) in space. A coefficient Z _{HPA # 1} (f _k ) and a coefficient Z _{LNA # 0} (f _k ) indicating a change accompanying the passage of the low noise amplifier 22-0 are observed as multiplied values.

Similarly, channel information when transmitting a signal from the wireless module 25-2 to the wireless module 25-0 is a coefficient Z indicating a change associated with the passage of the high power amplifier 21-2 at h ₂ (f _k ) in space. _{HPA # 2} (f _k ) and a coefficient Z _{LNA # 0} (f _k ) indicating a change accompanying the passage through the low noise amplifier 22-0 are observed as multiplied values.

Therefore, the channel from the wireless module 25-1 to the wireless module 25-0 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 0} (f _k ). A channel from the wireless module 25-2 to the wireless module 25-0 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 0} (f _k ). Therefore, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 25-1 and the wireless module 25-2 have relatively Z _{HPA # 2} (f _k ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same in the reverse direction communication. When the signal transmitted from the wireless module 25-0 is received by the wireless module 25-1, the channel information is high power in h ₁ (f _k ) in space. As a value obtained by multiplying a coefficient Z _{HPA # 0} (f _k ) indicating a change accompanying the passage of the amplifier 21-0 and a coefficient Z _{LNA # 1} (f _k ) indicating a change accompanying the passage of the low noise amplifier 22-1 Observed.

Similarly, when the signal transmitted from the wireless module 25-0 is received by the wireless module 25-2, the channel information changes in the space h ₂ (f _k ) with the passage of the high power amplifier 21-0. It is observed as a value obtained by multiplying the indicated coefficient Z _{HPA # 0} (f _k ) by the coefficient Z _{LNA # 2} (f _k ) indicating the change accompanying the passage of the low noise amplifier 22-2.

Therefore, the channel from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). A channel from the wireless module 25-0 to the wireless module 25-2 is represented by Z _{HPA # 0} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). Therefore, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 25-1 and the wireless module 25-2 have relatively Z _{LNA # 2} (f _k ) / Z _{LNA # 1} (f _k ) difference occurs.

  As described above, the base station apparatus can acquire channel information including changes by the low-noise amplifiers 22-1 to 22-2 connected to each antenna element on the uplink, but the base station apparatus is on the downlink. Channel information cannot be obtained directly. Therefore, downlink channel information is obtained by conversion from uplink channel information. For this conversion, the influence of individual differences between the low noise amplifiers 22-0 to 22-2 and the high power amplifiers 21-0 to 21-2 connected to the antenna elements 24-0 to 24-2 is canceled. There is a need.

  Therefore, in the manufacturing stage of the base station apparatus, a test radio module 25-0 serving as a reference is prepared, and the antenna terminal of the test radio module 25-0 and the antenna terminals of the radio modules 25-1 and 25-2 are prepared. And connect directly with a cable. In an environment where channel information on the propagation path has a common value, channel information including changes by the high power amplifiers 21-0 to 21-2 and the low noise amplifiers 22-0 to 22-2 is measured, and the measured channel information is used. To correct.

  FIG. 10 is a diagram showing an outline of calibration. In FIG. 10, reference numerals 26-0 to 26-2 indicate antenna terminals, and reference numeral 27 indicates a coaxial cable. In addition, the same code | symbol is attached | subjected to the function part same as the function part shown in FIG.

  FIG. 10A shows a configuration in which the wireless module 25-0 and the wireless module 25-1 are connected by a coaxial cable. FIG. 10B shows a configuration in which the wireless module 25-0 and the wireless module 25-2 are connected by a coaxial cable. FIG. 9 shows a state in which a signal propagates in an actual space, whereas FIG. 10 shows a state in which a signal propagates on a coaxial cable without passing through an antenna element.

The channel information of the coaxial cable 27 as a propagation path connecting the wireless modules 25-1 and 25-2 and the wireless module 25-0 is h ₀ (f _k ).
At this time, channel information from the wireless module 25-1 to the wireless module 25-0 is expressed by Z _{HPA # 1} (f _k ) · h ₀ (f _k ) · Z _{LNA # 0} (f _k ). The channel information from the wireless module 25-2 to the wireless module 25-0 is represented by Z _{HPA # 2} (f _k ) · h ₀ (f _k ) · Z _{LNA # 0} (f _k ).

Further, channel information from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₀ (f _k ) · Z _{LNA # 1} (f _k ), and the wireless module 25 The channel information from −0 to the wireless module 25-2 is expressed by Z _{HPA # 0} (f _k ) · h ₀ (f _k ) · Z _{LNA # 2} (f _k ).
Therefore, after measuring these channel information, calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) represented by the following equations (10) and (11) are calculated.

The channel information from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ), and the wireless module 25- It has been described that the channel information from 0 to the wireless module 25-2 is represented by Z _{HPA # 0} · (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). When these are multiplied by the calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) of the equations (10) and (11), the following equations (12) and (13) are obtained.

  The right sides of Expression (12) and Expression (13) are the channel information from the wireless module 25-1 to the wireless module 25-0 and the channel information from the wireless module 25-2 to the wireless module 25-0, as described above. It matches.

  In this way, calibration coefficients corresponding to the equations (10) and (11) are acquired in the manufacturing stage of the base station device, and stored in the base station device, thereby calibrating them. The downlink channel information can be calculated from the uplink channel information using the coefficient.

  In the following description, these calibration coefficients are obtained in advance and the values are used mainly for digital signal processing. Of course, on the analog circuit, these calibration coefficients are all substantially constant (if the complex phase is constant, the absolute value itself may be different) and in the base station apparatus. If adjustment is performed in the terminal station apparatus, it can be read as processing in which all the calibration coefficients are regarded as 1. In particular, since the frequency dependency of the amplification factor of the amplitude is directly related to the waveform distortion, it is generally designed industrially so as to have a constant (flat waveform) on the frequency axis as a characteristic of the amplifier.

  Similarly, when the complex phase of the uplink and downlink is adjusted to be a constant value, it is collectively adjusted in the latter stage of the digital signal processing (at the time of reception) and the former stage of the IFFT (at the time of transmission). As a result, since the complex phase of the calibration coefficient expressed by the equations (10) and (11) becomes a substantially constant value for all the antenna elements, the same effect can be obtained.

  The above is a description regarding calibration of one-to-multiple SIMO channels for transmission / reception antenna elements. However, in the case of calibration of a channel matrix of a plurality of antennas to a plurality of antennas, a description in a slightly corrected form is necessary. It is. For example, as shown in FIG. 11, a case where there are two transmission / reception antenna elements is considered. In order to simplify the explanation, the left side is regarded as the base station apparatus 41 and the right side is the terminal station apparatus 51. Each of the downlink that is communication from the left to the right and the uplink that is communication from the right to the left Organize channel matrix relationships.

  First, when considering the downlink, the channel information observed at the antenna element of the receiving-side terminal station apparatus 51 includes the base station apparatus in addition to each component of the channel matrix on the space between the transmitting antenna end and the receiving antenna end. The amplification factor and complex phase rotation amount in the transmission high power amplifier 41 are multiplied by the amplification factor and complex phase rotation amount in the low noise amplifier of the terminal station apparatus. The amount of rotation differs for each high power amplifier corresponding to the plurality of antenna elements of the base station apparatus, and also varies for each low noise amplifier corresponding to the plurality of antenna elements of the terminal station apparatus. On the other hand, in the uplink, the channel information observed at the antenna element of the base station device on the receiving side is not only the components of the channel matrix on the space between the transmitting antenna end and the receiving antenna end, The amplification factor and complex phase rotation amount in the transmission high power amplifier are multiplied by the amplification factor and complex phase rotation amount in the low noise amplifier of the base station apparatus. These rotation amounts differ for each high power amplifier corresponding to the plurality of antenna elements of the terminal station apparatus, and differ for each low noise amplifier corresponding to the plurality of antenna elements of the base station apparatus. Here, regarding the channel matrix in the space between the transmitting antenna end and the receiving antenna end, the value of each component itself is not changed due to the symmetry of the space, but the transmitting antenna and the receiving antenna are not connected in the uplink and the downlink. Since the relationship between the base station apparatus and the terminal station apparatus is reversed, the channel matrix in space can be converted into an uplink and a downlink by transposing the matrix. In addition to this, if the same processing as the calibration in the above-mentioned SIMO channel is performed, calibration is performed to convert the channel state between the uplink and the downlink with an arbitrary test transmission / reception apparatus. It is possible to acquire a downlink channel matrix from an uplink channel matrix by using the distribution coefficient.

Here, θ _{BS-HPA, m} ^(k) is the phase rotation amount of the high power amplifier corresponding to the m-th antenna element of the base station apparatus, and A _{BS-HPA, m} ^(k) is the m-th antenna of the base station apparatus. The amplitude amplification amount of the high power amplifier corresponding to the element, θ _{BS-LNA, m} ^(k) , the phase rotation amount of the low noise amplifier corresponding to the m-th antenna element of the base station apparatus, and A _{BS-LNA, m} ^{( k)} is an amplification amount of the amplitude of the low noise amplifier corresponding to the m-th antenna element of the base station apparatus. Furthermore, θ _{MT-HPA, m} ^(k) is the amount of phase rotation of the high power amplifier corresponding to the m-th antenna element of the terminal station apparatus, and A _{MT-HPA, m} ^(k) is the m-th antenna element of the terminal station apparatus. The amount of amplification of the amplitude of the corresponding high power amplifier, θ _{MT-LNA, m} ^(k) is the amount of phase rotation of the low noise amplifier corresponding to the m-th antenna element of the terminal station device, and A _{MT-LNA, m} ^(k) Is the amount of amplification of the amplitude of the low-noise amplifier corresponding to the m-th antenna element of the terminal station apparatus. Also, θ _Test-HPA ^(k) is the phase rotation amount of the high power amplifier of the test station, A _Test-HPA ^(k) is the amplitude amplification amount of the high power amplifier of the test station, and θ _Test-LNA ^{( k)} is the phase rotation amount of the test station, and A _Test-LNA ^(k) is the amplification amount of the amplitude of the low noise amplifier of the test station. If the BS (base station apparatus) and MT (terminal station apparatus) subscripts of each coefficient are generalized and shown as STA (base station apparatus or terminal station apparatus), the calibration coefficient related to the mth antenna element of the STA is expressed by an equation ( It is possible to obtain the downlink channel matrix from the uplink channel matrix as shown in equation (15) using the definition as shown in 14).

In general, a coefficient that is equally multiplied by all elements of a channel matrix has no effect both when evaluating the characteristics of the propagation path indicated by the channel matrix and when calculating a transmission / reception weight matrix (vector). It is possible to treat it as a coefficient without any problems. In this sense, the equation (15) includes the coefficients θ _Test-HPA ^(k) , A _Test-HPA ^(k) , θ _Test-LNA ^(k) , and A _Test-LNA ^{(k )} for the test station. ⁾ Although a contains, for all of which are coefficients equal multiplied to all the elements of the channel matrix, this value is not particularly affect especially when obtaining reception weight matrix (vector). Furthermore, a local oscillator signal input to the mixer when converting the baseband signal to the radio frequency on the transmitting station side, and a local oscillator input to the mixer when converting the radio frequency signal to the baseband signal on the receiving station side The state of the complex phase of each signal of the signal in fact also affects the complex phase of each component of the observed channel matrix, but this is also because all the coefficients are multiplied equally by all antennas. In addition, all the components of the channel matrix are equally multiplied as coefficients. Since this coefficient also does not affect the transmission / reception weight matrix (vector), it is possible to perform evaluation while ignoring these effects. As described above, if the calibration coefficient is known in advance, the channel matrix between the uplink and the downlink can be converted based on the calibration coefficient. Similarly, if a reception weight vector can be acquired, a transmission weight vector can be calculated by applying a calibration coefficient to the reception weight vector.

[Wireless entrance at mobile fronthaul]
The same applies to the fourth-generation mobile phones, but most of the base station devices are assumed to be improved in maintainability of base station devices that need to be installed in large numbers and coordinated transmission from a plurality of base station devices. Functions (BBU: Base Band Unit) in one place, (Function 1) Generation of analog baseband signal by D / A conversion from digital sampling data, (Function 2) Radio frequency from analog baseband signal Up-conversion to band signals, (Function 3) Power amplification and transmission from antenna, (Function 4) Reception and power amplification at antenna, (Function 5) Down-conversion from radio frequency band signal to analog baseband signal (Function 6) Remote radio head having only functions such as generation of digital sampling data from analog baseband signals by A / D conversion ( A cloud-type radio access network (C-RAN: Centralized Radio Access Network or Cloud Radio Access Network) in which only RRH (Remote Radio Head) is distributed is effective. This is a technique called “fronthaul” or “mobile fronthaul” in contrast to the so-called “backhaul line”, and CPRI (Interface as an interface standard for transmitting digital sampling data over an optical fiber) Common Public Radio Interface) has been standardized.

Here, in order to perform coordinated transmission (JT: Joint Transmit) or coordinated reception (JR: Joint Reception) with a plurality of RRHs, the delay jitter can be suppressed with an accuracy of the order of 10 ⁻⁹ , and on the optical fiber. The bit error rate is required to have a high quality of the order of 10 ^{−12 and} a low delay of 100 μs or less. This mobile fronthaul can be provided not only by an optical fiber but also by a wireless line. In this case, although the wireless front line is a wireless line, the delay jitter is 10 ⁻⁹ , the bit error rate is 10 ⁻¹² , and the transmission delay is 100 μs or less. High quality must be realized.

Among existing wireless entrance devices, a large parabolic antenna forms an ultra-high gain and directivity of 40 dBi to 50 dBi, and this antenna is opposed so that multipath components are almost eliminated and stable without nonlinear distortion. Some transmission lines can realize a bit error rate 10 ⁻¹² equivalent to that of an optical fiber in one transmission without retransmission control. However, in the case where the directivity gain is secured by utilizing large-scale MIMO as described above, the redundancy of the antenna is used, and the multi-path component is used similarly to the normal spatial multiplexing, so that a plurality of streams can be obtained. Multiplexing was also possible. However, when such multipath components are taken in, the influence of nonlinear distortion cannot be ignored. In general, the error rate characteristic draws an error floor, and a sufficiently low error rate cannot be achieved even in a high SNR environment. It was difficult. In this case, it is necessary to improve error rate characteristics by performing retransmission control, but generally retransmission control is accompanied by a large transmission processing delay, and it is difficult to complete retransmission control with a low delay of 100 μs or less. Further, when performing highly efficient retransmission control in ultra-wideband large-capacity transmission, the process of selective retransmission control becomes complicated, and in addition, due to the restriction on delay of 100 μs or less, it is 2-3 times regardless of the success / failure of transmission. It is also necessary to discontinue retransmission and discard at the upper limit of the number of times of transmission. In the retransmission abortion, a mismatch may occur between the transmission side and the reception side, but this mismatch between the transmission side and the reception side leads to a communication deadlock state. However, if negotiation processing for avoiding such a state is performed, the processing becomes further complicated, which is not preferable. There is a need to avoid such complication of signal processing and to realize highly efficient retransmission control at ultra-high speed.

  In this mobile fronthaul, the digital sampling data of the signal to be transmitted from the RRH antenna element is transmitted between the BBU and the RRH as described above, and the above (Function 1) to (Function 6) are transmitted by the RRH. ) Function to transmit and receive actual radio frequency signals. However, in the case of using a large-scale antenna in the fifth generation small cell base station apparatus, if following the conventional method of transmitting digital sampling data over an optical fiber, each antenna element It was necessary to transmit digital sampling data. In general, it is said that the transmission capacity of digital sampling data is about 16 times the amount of user data to be actually transmitted in the wireless section, and this is a case where 1 Gbit / s data transmission is performed in the wireless section. However, digital sampling data equivalent to 16 Gbit / s must be transmitted in the mobile fronthaul section. However, if a large-scale MIMO technology using 256 antenna elements is adopted in the fifth generation small cell base station apparatus, the transmission capacity required for the mobile fronthaul is 4 Tbit / s, and the transmission medium Even if it is an optical fiber, it is an unrealistic area at present. In general, a system that can be used at a relatively low cost is a system that has a low transmission rate and is widespread, and is currently a communication system of about 10 Gbit / s. Considering future technological innovations, the data capacity to be transmitted on the mobile fronthaul is not allowed to diverge as the number of antenna elements increases, and at least digital sampling for the number of spatially multiplexed signal systems・ It is required to fit in the data capacity.

  From this point of view, the review of the function distribution between BBU and RRH is currently under consideration. In addition to the functions (Function 1) to (Function 6) described above, the function is originally implemented on the BBU side. In addition to the physical layer modulation function and demodulation function in power signal processing, studies are underway to implement complex MIMO signal processing, FFT / IFFT signal processing, and the like on the RRH side. As a result, the information capacity to be transmitted by the mobile fronthaul is greatly compressed, which is far from the original idea of the mobile fronthaul. In other words, since (Function 1) to (Function 6) do not include functions specialized for the modulation / demodulation method peculiar to the wireless system to be adopted, only a change on the BBU side at the time of future system upgrade If the RRH was able to deal with RRH without any changes, but the function allocation was revised as described above, it would be possible to upgrade even if the basic design parameters were changed, for example, the number of subcarriers was changed. For this purpose, the RRH itself must be remodeled.

  Therefore, there is a need for a device that can cope with large-scale MIMO but does not increase the transmission capacity of digital sampling data to be transmitted on the mobile fronthaul.

  In the following, the mobile fronthaul will be described with reference to the drawings. FIG. 12 is a diagram showing an outline of the function sharing between the mobile fronthaul and the mobile backhaul in the prior art. In particular, FIG. 12A shows a mobile fronthaul, and FIG. 12B shows a mobile backhaul. Here, only functions related to signal transmission in the direction from the network side to the user (in the case of a mobile fronthaul, from BBU to RRH) are extracted. In the figure, 401 is a MAC layer processing circuit, 402 is a transmission signal processing circuit, 403 is a time axis signal generation circuit, 404 is an optical interface circuit, 405 is an optical fiber, 406 is an optical interface circuit, 407 is a D / A converter, Reference numeral 408 denotes an RF processing circuit, 409 denotes an antenna element, 411-1 denotes a BBU, 412-1 denotes an RRH, 414 denotes an optical interface circuit, 415 denotes an optical fiber, and 416 denotes an optical interface. The MAC layer processing circuit 401, the transmission signal processing circuit 402, and the time axis signal generation circuit 403 constitute regions 400 and 410 that perform baseband signal processing related to radio as a whole.

  In FIG. 12A, when a signal to be transmitted is input from the network side to the BBU 441-1, the MAC layer processing circuit 401 performs the MAC layer signal processing, the frame format used for transmission / reception in the wireless section, and the network The frame format of the data flowing on the side is converted and terminated, and a radio packet format signal is input to the transmission signal processing circuit 402. The transmission signal processing circuit 402 performs transmission signal processing of a radio signal. Here, the transmission scheme in the radio section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. The signal generated in this manner is converted into a time axis signal by the time axis signal generation circuit 403. For example, taking the case of OFDM as an example, IFFT is performed to convert a frequency axis signal into a time axis signal, and a guard interval is inserted to perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a continuous signal in time series. Data of these sampling values is converted into a predetermined frame format by the optical interface circuit 404, converted from an electrical signal to an optical signal, and output to the optical fiber 405. The signal output to the optical fiber 405 is transmitted to the RRH 442-1 side. In the RRH 442-1, the optical interface circuit 406 converts the optical signal into an electrical signal, terminates the signal in a predetermined format, and serves as a digital baseband signal. An information string of sampling values is generated. The D / A converter 407 converts this signal to an analog baseband signal at a predetermined clock rate, and the RF processing circuit 408 converts it to a radio frequency signal using an up-converter. The signal is amplified by a power amplifier and transmitted from an antenna 409 to the space. As described above, the functions corresponding to the radio signal base station apparatus as a whole are accommodated by dividing the functions into BBU 441-1 and RRH 442-1 mediated by optical fibers. The feature here is that the wireless digital baseband signal processing is integrated into the BBU provided in the network-side office, so that even if there is a change in the wireless communication system, all changes are made on the BBU side. There is a merit that it is finished.

  On the other hand, the mobile backhaul has the configuration shown in FIG. 12B as the configuration corresponding to FIG. For example, user data is transmitted as it is to a base station apparatus installed on the rooftop of a building, a telephone pole, or the like through an optical fiber, and all of the user data is closed in the base station apparatus to perform radio signal processing. Specifically, on the network side, user data to be transmitted by a wireless line is transmitted from the optical interface circuit 414 to the optical interface circuit 416 in the base station apparatus via the optical fiber 415. The optical interface circuit 416 converts the optical signal into an electrical signal, terminates the signal in a predetermined format, and outputs user data to the MAC layer processing circuit 401. The MAC layer processing circuit 401 performs MAC layer signal processing, converts and terminates the frame format used for transmission / reception in the wireless section and the frame format of data flowing on the network side, and performs transmission signal processing on the signal in the wireless packet format. Input to the circuit 402. The transmission signal processing circuit 402 performs transmission signal processing of a radio signal. Here, the transmission scheme in the radio section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. The signal generated in this manner is converted into a time axis signal by the time axis signal generation circuit 403. For example, taking the case of OFDM as an example, IFFT is performed to convert a frequency axis signal into a time axis signal, and a guard interval is inserted to perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a continuous signal in time series. The data of these sampling values is input to the D / A converter 407, which converts it into an analog baseband signal at a predetermined clock rate, and the RF processing circuit 408 uses the up-converter to convert the radio frequency signal. After the out-of-band radiation signal is removed by the filter, the signal is amplified by the high power amplifier and transmitted to the space from the antenna 409. The description of the signal processing and circuit configuration of the base station apparatus described so far is based on this mobile backhaul.

  In the description of FIG. 12 described above, the RRH 442-1 has one antenna element. However, in future mobile networks, it is assumed that the Massive MIMO technology using a multi-element antenna is applied. The amount of information to be transmitted increases. FIG. 13 is a diagram showing an outline of the function sharing between the mobile fronthaul and the mobile backhaul in the prior art when a multi-element antenna is used. In particular, FIG. 13A shows a mobile fronthaul, and FIG. 13B shows a mobile backhaul. Here, as in FIG. 12, only functions related to signal transmission in the direction from the network side toward the user (in the case of a mobile fronthaul, BBU to RRH) are extracted. In FIG. 13, the same components as those in FIG. In FIG. 13, 412 is a transmission signal processing circuit, 424 is an optical interface circuit, 425 is an optical fiber, 426 is an optical interface circuit, 427 is a D / A converter, 428 is an RF processing circuit, 429 is an antenna element, and 430 is a transmission A weight multiplication circuit, 431 is a transmission weight processing circuit, 433 is a time axis signal generation circuit, 441-2 is BBU, and 442-2 is RRH. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 constitute an area 420 for performing baseband signal processing related to radio as a whole. Similarly, the MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 constitute an area 420 or 440 that performs baseband signal processing related to radio as a whole. To do. In FIG. 13 (a), when a signal to be transmitted is input from the network side to the BBU 441-2, the MAC layer processing circuit 401 performs the MAC layer signal processing, the frame format used for transmission / reception in the wireless section, and the network The frame format of the data flowing on the side is converted and terminated, and a radio packet format signal is input to the transmission signal processing circuit 412. The transmission signal processing circuit 412 performs transmission signal processing of radio signals. Here, the transmission scheme in the radio section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. In addition, for example, reception weight information collected on the BBU reception side (not shown) is provided to the transmission weight processing circuit 431 via the transmission signal processing circuit 412. The transmission weight processing circuit 431 performs a calibration process on the reception weight information and the like, calculates a transmission weight, and outputs this to the transmission weight multiplication circuit 430. The transmission weight multiplication circuit 430 multiplies the transmission signal input from the transmission signal processing circuit 412 by the transmission weight. This processing is performed on the frequency axis when OFDM is used, for example. That is, a different transmission weight is used for each subcarrier calculated based on a different reception weight for each subcarrier, and a transmission signal that is different for each subcarrier is multiplied for each subcarrier. The signal generated in this way is converted into a time-axis signal for each antenna system by the time-axis signal generation circuit 433. For example, taking the case of OFDM as an example, IFFT is performed to convert a frequency axis signal into a time axis signal, insert a guard interval, and perform waveform shaping processing between symbols, etc. It is carried out every time. As a result, each sampling value of the digital baseband signal is converted into a signal for each antenna element that is continuous in time series. Data of these sampling values is converted into a predetermined frame format by the optical interface circuit 424, converted from an electrical signal to an optical signal, and output to the optical fiber 425. Here, unlike the case of FIG. 12, it is necessary to simultaneously accommodate signals corresponding to the number of antenna elements on the optical fiber, and the capacity can be increased by any method such as wavelength multiplexing, TDM, or a plurality of optical fibers. Is transmitted on the transmission line of the optical fiber that has been made, and is input to the RRH 442-2 side. In the RRH 442-2, the optical interface circuit 426 converts an optical signal into an electrical signal, terminates a signal in a predetermined format, and generates an information string of sampling values as a digital baseband signal. As described above, here, sampling signals corresponding to the number of antenna elements are reproduced. The D / A converter 427 converts the signal into an analog baseband signal for each antenna system at a predetermined clock rate, and the RF processing circuit 428 uses a common local oscillator for each antenna system to generate a radio frequency signal for each antenna system. After the out-of-band radiation signal is individually removed by a filter, the signal is amplified by a high power amplifier individually, and is transmitted to the space from each antenna element 429. As described above, the functions corresponding to the radio signal base station apparatus as a whole are accommodated by dividing the functions into BBU 441-2 and RRH 442-2 mediated by optical fibers.

  The above explanation is based on the mobile fronthaul policy, but considering the transmission capacity of the signal actually flowing on the optical fiber, the number of antenna elements is limited to a very small number, such as a sector antenna of several sectors. It is thought that. With 100 or more antenna elements assumed in Massive MIMO or the like, a general idea is that the cost becomes enormous considering the number of installed optical fibers, the capacity of data to be transmitted, and the like.

  On the other hand, the mobile backhaul has a configuration shown in FIG. 13B as a configuration corresponding to FIG. Compared to FIG. 12B, the area 410 for performing wireless signal processing is replaced with an area 440 for performing wireless signal processing having the same function as the area 420 for performing wireless signal processing in FIG. The processing after the A converter 427 is exactly the same as in FIG. 13A except that the processing is performed for each antenna system. The feature here is that the number of antenna elements in the base station apparatus has increased, but the information capacity on the transmission path from the optical interface circuit 414 to the optical interface circuit 416 in the base station apparatus via the optical fiber 415 has increased. It is not. Therefore, the problem as shown in FIG. 13A does not occur. However, the problem that the mobile fronthaul as shown in FIG. 12 (a) is originally useful is not solved at all. In general, the mobile fronthaul is suppressed while suppressing the transmission capacity on the optical fiber. There has been a demand for a configuration that can achieve both advantages.

  In such a study, a review of the functional distribution between BBU and RRH is under consideration as a mobile midhole between the fronthaul and the backhaul. FIG. 14 shows a configuration example of a mobile midhole in the prior art. Here, as in FIG. 12, only functions related to signal transmission (BBU to RRH direction) in the direction from the network side toward the user are extracted. The same components as those in FIGS. 12 and 13 are denoted by the same reference numerals. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time-axis signal generation circuit 433 have regions 445-1 and 445-2 that perform baseband signal processing related to radio as a whole. Although different from FIG. 12A and FIG. 13A, the MAC layer processing circuit 401 has a transmission signal processing circuit 412, a transmission weight multiplication circuit 430, and transmission weight processing in an area 445-1 on the BBU side. The circuit 431 and the time axis signal generation circuit 433 are arranged separately in the region 445-2 on the RRH side, and the overall function distribution is reviewed.

  By reviewing the above function allocation, the transmission capacity on the optical fiber in the case of the mobile midhole of FIG. 14 is almost the same as that of FIG. 12A, and the amount of information is greatly increased as shown in FIG. There is no need to increase it. In this sense, the effect of suppressing the transmission capacity on the optical fiber is sufficiently obtained. However, from the perspective of eliminating the processing dependent on the radio transmission system from the RRH 442-3 side, which is the purpose of the mobile fronthaul in the first place, and consolidating all the radio system dependent processing on the BBU 441-3 side, The reason why the processing of the region 445-2 is arranged on the RRH 442-3 side is not sufficiently suitable for the purpose of the front hole as compared with the configuration of the mobile front hole shown in FIG. In this sense, it is also called a midhole as an intermediate between the fronthaul and the backhaul, but ideally it has many antenna elements on the RRH side but depends on the radio transmission system on the BBU side. It is preferable that the information capacity flowing on the optical fiber can be reduced to the same extent as the backhaul.

[Other issues]
In the case of a multipath environment, it is known that techniques such as OFDM and SC-FDE are generally effective for removing multipath components. However, in existing systems such as OFDM and SC-FDE, generally, the base station apparatus and the radio station apparatus use their own asynchronous clocks and local oscillators, respectively, and perform transmission processing in a burst mode. In the burst mode, there is no problem because the processing is reset for each burst even if there is a clock error. Further, the clock error and the local oscillator frequency error are not required to be synchronized with higher accuracy if the error is not to break the orthogonality of each subcarrier when performing FFT. Therefore, if a certain amount of frequency error is detected, the frequency error is corrected by AFC processing on the transmitting side and the receiving side, and the remaining residual frequency error is estimated by the pilot subcarrier in the case of OFDM, for example. The correction process could be performed.

However, as described above, in the case of mobile fronthaul, the delay jitter is required to be 10 ⁻⁹ or less, and the block transmission technology such as OFDM and SC-FDE does not satisfy the required delay jitter. could not.

  Furthermore, assuming use of an e-band band of 70 to 80 GHz, the bandwidth in which phase noise exists becomes wider than subcarrier intervals such as OFDM and SC-FDE, and within a time shorter than the OFDM symbol period. Due to phase noise that is an unstable fluctuation of the complex phase, a component that breaks orthogonality between subcarriers such as OFDM cannot be ignored. This is a problem because FFT processing is performed solely. For single carrier transmission, single carrier signal processing is performed in a time period shorter than the block length as a block transmission unit such as OFDM or SC-FDE. It was possible to correct the phase fluctuation component before the accumulation of phase noise increased. However, if spatial multiplexing transmission is performed using the MIMO channel as described above, the transmission and reception weights for separating the signals of each stream have different values for each subcarrier, and the frequency dependence of this weight. In order to perform signal processing reflecting the above, it is necessary to convert the received signal into a signal on the frequency axis by FFT once. Thus, FFT processing is essential for spatially multiplexed signal separation, but if FFT is performed, phase noise cannot be removed, which is a problem. However, in a state before performing FFT processing and performing spatial separation on the frequency axis, single-carrier signals of a plurality of streams interfere with each other, and thus single-carrier signal detection is difficult. Thus, if signal separation can be performed on the time axis without performing FFT processing, then phase noise compensation can be performed by signal processing of a single carrier signal, and such a technique is required.

Atsushi Ota, Kaoru Kurosaki, Kazuki Maruta, Takuto Arai, Masataka Iizuka, “B-5-175 Proposal of Large-scale Antenna Wireless Entrance System”, Proceedings of the IEICE General Conference 2013 (Communication_1), 2013 March 5, p. 585 Takuto Arai, Atsushi Ota, Atsushi Kurosaki, Kazuki Maruta, Masataka Iizuka, “B-5-176 Calculation Method of Transmit / Receive Weights in Large Antenna Radio Entrance System”, IEICE General Conference Proceedings 2013 (Communication_1) March 5, 2013, p. 586 Kazuteru Maruta, Atsushi Ota, Atsushi Kurosaki, Takuto Arai, Masataka Iizuka, “B-5-177 Inter-User Interference Suppression Method in Large Antenna Wireless Entrance System”, IEICE General Conference Proceedings 2013 (Communication_1) ), March 5, 2013, p. 587 Satoshi Kurosaki, Atsushi Ota, Kazuki Maruta, Takuto Arai, Masataka Iizuka, "B-5-178 Low Correlation Scheduling Method in Large Antenna Radio Entrance System", IEICE General Conference Proceedings 2013 (Communication_1) March 5, 2013, p. 588 Ohara Yasunori, Suyama Kaoru, Shinki Yun, Okumura Yukihiko, “Combined processing of fixed beamforming and eigenmode precoding in ultra-high-speed Massive MIMO transmission using RCS 2013-349”, IEICE technical report, Wireless communication System (RCS), Vol. 113, No. 456, February 24, 2014, p. 259-p. H.264

  In the future, the environment in which a large transmission path is required in a mobile network is as follows: (Environment 1) Large-capacity transmission from a base station device to a mobile user's terminal station device in an access system for a mobile user; For example, when a cell radio base station device is installed on a wall surface of a building or the like, large-capacity transmission at a radio entrance that makes the entrance line to the base station device wireless for construction reasons, (Environment 3) It is conceivable to transmit a large number of mobile users once in a train, and to transmit a large amount of wireless entrance as an entrance line for collectively transferring the collected data to a ground network. In the case of (Environment 1), for the purpose of efficient transmission to users existing in a relatively small area, a service area is formed in a direction looking down from a high place such as a wall surface of a building or a rooftop, and the terminal The station environment generally has a line-of-sight environment due to the physical characteristics that radio waves arrive from above, and the antenna elements mounted on the terminal stations also have some directivity in consideration of the arrival of radio waves from above. If an antenna with a gain is used, the line-of-sight wave becomes a very dominant environment as a result. Similarly, in the case of (Environment 2), since the installation location is fixedly installed, a line-of-sight environment is secured, and it is expected that a high-gain directional antenna will be used as the antenna element. 1) The prospect wave is more dominant than above. In the case of (Environment 3), for example, a train or the like is a moving body, so a directional antenna as high as (Environment 2) cannot be used, but considering the one-dimensional movement of the train, It is unlikely that a directional antenna suitable for the one-dimensional route is used and an obstacle is interrupted on the line, and the line-of-sight wave is expected to be dominant more than (Environment 1).

  Here, as explained earlier, in general, multi-path environments with many reflected waves are effective in spatial multiplexing transmission using a MIMO channel matrix, and it is not suitable when the direct line-of-sight component is dominant. Not known to be. In order to avoid this problem and perform high-order spatial multiplexing transmission in an environment where the line-of-sight wave is dominant, it is necessary to make the antenna element of the base station device large in scale (widen the antenna aperture length). Is one choice. That is, it is possible to approximate a multipath environment by forming a situation in which the path lengths between the antenna elements on the transmission path are randomly different.

  On the other hand, as described above, in the future mobile networks, there is a need to increase the capacity of transmission lines, and considering operation with a wide frequency bandwidth, operation in high frequency bands such as quasi-millimeter waves and millimeter waves Will be forced. It should be noted that when using a high frequency band, each device or device and antenna are connected by a coaxial cable, etc., and when transmitting / receiving signals etc. are transmitted over this cable, transmission over the cable is required. There is a problem that the loss becomes very large. Specifically, a 60 GHz band signal or the like depends on the cable, but a loss of about 15 dB per meter is expected.

  Even using a millimeter wave band, which is disadvantageous as a gain in line design, even if many antennas are installed to secure the line gain, it is enormous on the cable connecting the spatial spread of the antennas. It is meaningless if cable loss occurs. Furthermore, when the number of antenna elements is increased in order to ensure line gain, a huge number of cables are stretched between the spatial extents of the antennas, and transmission loss between them is reduced. If a low-loss cable is used for the reduction, the cable routing during that time also has a low degree of freedom in installation such as bending of the cable, and the restrictions on the installation of the base station apparatus become very large and impractical.

In other words, in the wireless entrance or access system in the future mobile network, in order to realize large-capacity transmission by high-order spatial multiplexing using a high frequency band such as millimeter waves, even though the line-of-sight wave is the dominant environment. Needs to solve these problems and build a wireless system.
In addition to the above discussions specific to wireless lines, the above-mentioned wireless trend cannot be ignored in the provision of mobile fronthaul lines using optical fibers to base station devices equipped with a large number of antenna elements. Is the situation. As described above, the transmission of sampling data on the mobile fronthaul line originally requires an optical line having a capacity of about 16 times the amount of information of actual user data. In 5G mobile, the target is to increase the capacity of 10 Gbps as user data by utilizing the (quasi) millimeter wave band, etc., but if it becomes 16 times that, a capacity of 160 Gbps is required for the mobile fronthaul line, and further As the number of antenna elements increases, it is theoretically required to increase the capacity up to 160 Gbps × the number of antenna elements. Even if the speed of increasing the capacity of a radio line and the speed of technological development for increasing the capacity of an optical line are equal, this increase in capacity for the number of antenna elements far exceeds the allowable limit of an optical line. Yes, not very selectable in terms of cost.
In this way, even in the use of mobile fronthaul lines using optical fibers in future mobile networks, these problems will be solved in order to economically lay radio base station devices equipped with a large number of antenna elements. There is a need to build a future mobile network that combines light and radio.

  In view of the above circumstances, the present invention is a radio communication apparatus capable of realizing efficient channel feedback by performing channel estimation with a small number of subcarriers and estimating channel information of subcarriers that have not been acquired with high accuracy. And it aims at providing the radio | wireless communication method.

  One aspect of the present invention relates to the sixth embodiment, the distance d [m] between the nearest antenna elements, the assumed maximum signal arrival direction θ [rad], the bandwidth W [Hz], the speed of light c [m / s], a wireless communication apparatus that operates under the condition that α = dW sin θ / c has a value of 1 or less, and is limited to all subcarriers within a communication band and some subcarriers. Channel estimation means for receiving the transmitted training signal, performing channel estimation of the subcarrier based on the received training signal, and acquiring channel information, and using the complex phase of the channel information or the complex phase of the reference antenna as a reference A regression line is obtained by applying a 2π period complex phase loopback to the complex phase of the relative channel information and the frequency of the subcarrier, and the channel of each subcarrier is based on the regression line. Complex phase calculation means for calculating the complex phase of the channel information or relative channel information, and reception weight calculation means for calculating the reception weight based on the channel information calculated by the complex phase calculation means or the complex phase of the relative channel information. A wireless communication device.

  One aspect of the present invention is the wireless communication device, in which the complex phase calculation unit obtains the regression line by a least square method, and the least square method uses the channel estimation in the subcarrier directly performed. With respect to the complex phase of the channel information or the relative channel information, values of 0 and ± 2π are assumed as offsets of the complex phase, and the accumulated value of the square error of the least squares method is minimized by adding the offset. The complex phase of the channel information or the relative channel information of each subcarrier obtained based on the information of the regression line obtained as a result of searching for the state is calculated.

  One aspect of the present invention is the wireless communication device, wherein the complex phase calculation unit uses the least squares method of the regression line when the antenna elements are configured and used in a linear array. A constraint condition is determined so that the inclination is in a proportional relationship, and the least square method over all antenna elements is collectively applied under the constraint condition.

  One aspect of the present invention relates to the sixth embodiment, and the element spacing d [m] of the nearest antenna, the assumed maximum signal arrival direction θ [rad], the bandwidth W [Hz], the speed of light c [m / s], a wireless communication method performed by a wireless communication device that operates under the condition that α = dWsin θ / c is 1 or less, and includes a part of subcarriers from all subcarriers in the communication band. A channel estimation step of receiving a training signal transmitted only to a carrier, performing channel estimation of the subcarrier based on the received training signal, and acquiring channel information; and a complex phase of the channel information or a complex of a reference antenna A regression line is obtained by applying a 2π period complex phase loopback to the complex phase of the relative channel information based on the phase and the frequency of the subcarrier, and each subline is based on the regression line. A complex phase calculating step for calculating a complex phase of the channel information or relative channel information of a subcarrier, and a reception weight for calculating a reception weight based on the complex phase of the channel information or the relative channel information calculated in the complex phase calculating step. And a calculation step.

  According to the present invention, while channel estimation is performed with a small number of subcarriers, channel information of subcarriers that have not been acquired can be estimated with high accuracy, so that while maintaining efficient channel feedback, accuracy is maintained. The effect is that channel estimation can be performed.

It is a figure which shows the outline | summary of the example of MIMO transmission. It is a conceptual diagram of the example of eigenmode transmission. It is a figure which shows the outline | summary of the example of a large-scale antenna system. It is a figure showing the example of the impulse response in a line-of-sight environment and a non-line-of-sight environment. It is a schematic block diagram which shows an example of a structure of the base station apparatus in a multiuser MIMO system. It is a schematic block diagram which shows an example of a structure of the transmission part in the base station apparatus in a multiuser MIMO system. It is a schematic block diagram which shows an example of a structure of the receiving part in the base station apparatus in a multiuser MIMO system. It is a figure which shows the example of the distribution characteristic of the absolute value of the singular value for every channel matrix. It is a figure which shows the example of the asymmetry of the channel information of an uplink and a downlink. It is a figure which shows the outline | summary of the example of a calibration. It is a figure which shows the example of the calibration of a multiple-to-multiple channel matrix. It is a figure which shows the outline | summary of the function sharing of a mobile fronthaul and a mobile backhaul. It is a figure which shows the outline | summary of the function sharing of the mobile fronthaul and mobile backhaul in a prior art in the case of using a multi-element antenna. It is a figure which shows the structural example of a mobile midhole. It is a figure which shows the example of the linear array in which 100 antenna elements of the base station apparatus are arrange | positioned at equal intervals in 1st Embodiment. It is a figure which shows the example of the linear array in which 100 antenna elements of the base station apparatus in 1st Embodiment were divided | segmented and arrange | positioned for every 4 groups. It is a schematic block diagram which shows an example of a structure of the base station apparatus in 1st Embodiment. It is a schematic block diagram which shows an example of a structure of the 1st transmission signal process part of the base station apparatus in 1st Embodiment. It is a schematic block diagram which shows an example of a structure of the 1st received signal processing part of the base station apparatus in 1st Embodiment. It is a schematic block diagram which shows an example of a structure of the terminal station apparatus in 1st Embodiment. It is a schematic block diagram which shows the 1st example of a structure of the transmission part of the terminal station apparatus in 1st Embodiment. It is a schematic block diagram which shows the 1st example of a structure of the receiving part of the terminal station apparatus in 1st Embodiment. It is a schematic block diagram which shows the 2nd example of a structure of the receiving part of the terminal station apparatus in 1st Embodiment. It is a schematic block diagram which shows the example of the apparatus structure of the 2nd received signal processing part of the base station apparatus in the 1st Embodiment, or the 2nd received signal processing circuit of the terminal station apparatus. It is a schematic block diagram which shows the 2nd example of a structure of the transmission part of the terminal station apparatus in 1st Embodiment. It is a flowchart which shows the outline | summary of the signal processing flow at the time of signal transmission in 1st Embodiment. It is a flowchart which shows the outline | summary of the 1st example of the signal processing flow at the time of signal reception in 1st Embodiment. It is a flowchart which shows the outline | summary of the 2nd example of the signal processing flow at the time of signal reception in 1st Embodiment. It is a figure which shows the 1st example of the case where four parabolic antennas were mounted in the base station apparatus, and 16 antenna elements were mounted in the terminal station apparatus in the linear array form in 2nd Embodiment. It is a figure which shows the example of the path | route difference for every antenna element by the side of the terminal station apparatus from the parabolic antenna by the side of the base station apparatus in 2nd Embodiment. It is a figure which shows the 2nd example of the case in which 4 parabolic antennas were mounted in the base station apparatus, and 16 antenna elements were mounted in the terminal station apparatus in the linear array form in 2nd Embodiment. It is a figure which shows the 1st example which implement | achieves transmission / reception of the signal from a base station apparatus to a satellite base station apparatus in a 3rd Embodiment by a radio link. It is a figure which shows the 2nd example which implement | achieves transmission / reception of the signal from a control base station apparatus to a satellite base station apparatus in a 3rd Embodiment by a radio link. It is a figure which shows the outline | summary of the signal processing between an integrated base station apparatus and a satellite base station apparatus, and the signal processing between a satellite base station apparatus and a terminal station apparatus in 3rd Embodiment. It is a figure which shows the circuit structure of the satellite base station apparatus in 3rd Embodiment. The outline | summary of a structure of the radio | wireless communications system in 4th Embodiment is shown. It is a figure which shows the outline | summary of the signal processing in the train moving cell which used the virtual transmission line corresponding to the 1st singular value in 4th Embodiment in multiple systems. It is a figure which shows the outline | summary of the offline signal processing performed after acquiring the coordinate information and channel information in 4th Embodiment. It is a figure which shows the outline | summary of another signal processing in the train moving cell which used the virtual transmission line corresponding to the 1st singular value in 4th Embodiment in multiple systems. It is a figure which shows the outline | summary of another signal processing performed offline after acquiring the coordinate information and channel information in 4th Embodiment. It is a figure which shows the example of the signal processing in the case of acquiring a transmission / reception weight vector from the coordinate information in service operation in the radio | wireless communications system of 4th Embodiment. It is a block diagram which shows the structural example of the base station apparatus in 4th Embodiment. It is a block diagram which shows the structural example of the terminal station apparatus in 4th Embodiment. It is a graph which shows the function F ((alpha)) in 5th Embodiment. It is a figure explaining the removal method of the interference signal in a 5th embodiment. It is a flowchart which shows the transmission / reception weight acquisition process of the time-axis beam forming in the radio | wireless communications system of 5th Embodiment. It is a figure which shows the structural example of the 1st transmission signal processing part with which the base station apparatus in 5th Embodiment is provided. It is a block diagram which shows the structural example of the 1st received signal processing part of the base station apparatus in 5th Embodiment. It is a block diagram which shows the example of the circuit structure of the transmission part with which the terminal station apparatus in 5th light embodiment is provided. It is a block diagram which shows the structural example of the receiving part of the terminal station apparatus in 5th Embodiment. It is a figure which shows the outline | summary of the received signal received with the several antenna element. It is a figure which shows the outline | summary of the channel estimation by a limited subcarrier. It is a flowchart which shows the processing operation of the approximate solution of channel matrix acquisition in 6th Embodiment. It is a flowchart which shows the processing operation of the approximate solution of a channel matrix using a plurality of antennas. It is a flowchart which shows the processing operation | movement which calculates | requires the approximate solution of a channel matrix from a bidirectional | two-way channel estimation result. It is a figure which shows the result of having evaluated the distribution characteristic by simulation. It is a figure which shows the gain characteristic in the transmission / reception weight vector approximate solution using the antenna away from the gravity center. It is a figure which shows the example of the antenna element used for transmission of the training signal in the case of using a linear array. It is a figure which shows the example of the antenna element used for transmission of the training signal in the case of using the two-dimensional antenna arrangement of the close-packed form. It is a figure which shows the example in the case of adding the antenna element for training signal transmission to the gravity center vicinity of a linear array. It is a figure which shows the specific example of the linear interpolation of an approximate weight value. It is a flowchart which shows the operation | movement of the determination process of the complex phase offset in the linear interpolation of an approximate weight value. It is a flowchart which shows the processing operation | movement of the approximate solution of the weight vector corresponding to a some 1st singular value. It is a figure which shows the apparatus structure which performs simultaneous channel estimation between the some small cells which exist in the same area. It is a figure which shows the example of the frame structure in 8th Embodiment. It is a flowchart which shows the processing operation for the communication start between the terminal station apparatus which newly entered in the area, and a base station apparatus. It is a flowchart which shows the transmission / reception signal processing operation | movement of a base station apparatus. It is a flowchart which shows the transmission / reception signal processing operation | movement of a terminal station apparatus. It is a figure which shows the flame | frame structure used as a precondition. It is a figure which shows the example which abbreviate | omits a training signal and performs communication only with a data payload. It is a flowchart which shows the processing operation of other 2nd reception weight matrix calculation. It is a flowchart which shows the other signal processing operation | movement of time-axis beam forming. It is a figure which shows the structure of the base station apparatus and terminal station apparatus at the time of multiuser MIMO application in 10th Embodiment. It is a figure showing the circuit structure of the 1st transmission signal processing part of the base station apparatus at the time of multiuser MIMO application in 10th Embodiment. It is a figure which shows the circuit structure of the 1st received signal processing part of the base station apparatus at the time of multiuser MIMO application in 10th Embodiment. It is a figure which shows the outline | summary of the function sharing of the mobile fronthaul using the virtual transmission line corresponding to the 1st singular value in 11th Embodiment. It is a figure which shows the outline | summary (downlink) of the function sharing of the mobile fronthaul using the virtual transmission line corresponding to the some 1st singular value in 11th Embodiment. It is a figure which shows the outline | summary (uplink) of the function sharing of the mobile fronthaul using the virtual transmission line corresponding to the some 1st singular value in 11th Embodiment. It is a figure which shows the outline | summary of the correlation detection circuit in 11th Embodiment. It is a figure which shows the outline | summary (uplink) of the function sharing of another mobile fronthaul using the virtual transmission line corresponding to the some 1st singular value in 11th Embodiment. It is a figure which shows the outline | summary (downlink) of the function sharing at the time of the multiuser MIMO application of the mobile fronthaul using the virtual transmission line corresponding to the some 1st singular value in 11th Embodiment. It is a figure which shows the outline | summary (uplink) of the function sharing at the time of the multiuser MIMO application of the mobile fronthaul using the virtual transmission line corresponding to the some 1st singular value in 11th Embodiment. It is a figure which shows the example of a structure in the case of applying a 1st signal processing part and a 2nd signal processing part to access system in embodiment.

Embodiments of the present invention will be described in detail with reference to the drawings.
Symbols in the following embodiments will be described.
N _SDM : Spatial multiplexing number.
N, M, N ′, M ′: natural numbers.
i, j, m, n: serial numbers of antenna elements or the like (general integers).
k: subcarrier number (frequency component number).
N _BS-Ant : Total number of antenna elements of the base station apparatus.
N ′ _BS-Ant : Number of antenna elements included in the first transmission signal processing unit or the first reception signal processing unit of the base station apparatus.
N _MT-Ant : The number of antenna elements of the terminal station apparatus.
N _Ant : Number of antenna elements of the base station apparatus or terminal station apparatus.
N _SC : Number of subcarriers.
N _FFT : Number of FFT points.
L: Distance.
K: Rice coefficient.
λ _k : wavelength of the k-th subcarrier.
r _ji : Distance between the i-th antenna element on the transmission side and the j-th antenna element on the reception side.
h _ji , h ′ _ji : Channel information between the i-th antenna element on the transmission side and the j-th antenna element on the reception side (because it has frequency dependence, it is the k-th frequency component if necessary for explanation) May explicitly indicate this).
d: Distance between antenna elements.
Δd _mn : Distance between the n-th antenna element and the m-th antenna element.
ΔL _m : path length difference of the m-th antenna element with respect to the first antenna element.
c: Speed of light (3 × 10 ⁸ m / s).
f _c : Center frequency [Hz] of the radio signal.
f _k : frequency [Hz] of the k-th subcarrier of the baseband signal.
t: Time.
W: Bandwidth [Hz].
Δt: Sampling cycle (Δt = 1 / W).
ψ _j (t), Φ _j (t): received signal (sampling value) at the j-th antenna element at time t.
φ _j ^(k) (t): the received signal of the k-th subcarrier at the j-th antenna element at time t (value focusing on a predetermined subcarrier in the sampling value).
η _k : an offset value of the k-th subcarrier considering a complex phase of 2π period when the least square method is used.
u _m : m-th left singular vector.
v _m : m-th right singular vector.

[First Embodiment]
[Spatial multiplex transmission using virtual transmission lines corresponding to a plurality of first singular values]
(Outline of the basic principle according to the first embodiment)
As described in FIG. 8, in the case where the line-of-sight wave is dominant as shown in FIG. 8B, the gap between the absolute values of the first singular value and the second singular value is increased, and the second singular value is obtained. In the case of using a transmission line corresponding to a singular value greater than or equal to the value, the Hi _{. i.} Communication is performed with a slight line gain earned by using the component _d . However, for example, when irradiating the inside of a limited small cell area below a base station apparatus installed on the wall of a building, the base station apparatus side is equipped with an antenna with high directivity gain. Furthermore, in situations where it is expected that a small antenna element capable of obtaining a directional gain using features such as millimeter waves with a short wavelength will be mounted on the terminal station side, the transmitting side and the receiving side Compared to a microwave band system or the like in which both antennas are omnidirectional, the multipath component is expected to be very limited. Therefore, the characteristics of MIMO transmission when only the line-of-sight wave is taken into consideration are organized.

FIG. 15 is a diagram illustrating an example of a linear array in which 100 antenna elements of the base station apparatus are arranged at equal intervals. In FIG. 15, reference numeral 40 denotes a wireless communication system, reference numeral 301 denotes a base station apparatus, and reference numeral 302 denotes a terminal station apparatus. In FIG. 15, 100 antenna elements of the base station apparatus 301 are mounted in a linear array. 100 antenna elements of the base station apparatus 301 is arranged at regular intervals over the length D _1. Further, 16 of the antenna element of the terminal station device 302 is equally spaced linear array shape over the length D _2.

FIG. 16 is a diagram illustrating an example of a linear array in which 100 antenna elements of the base station apparatus are arranged in groups of 4 every 25 antennas. In FIG. 16, reference numeral 50 is a wireless communication system, reference numeral 303 is a base station apparatus, reference numeral 302 is a terminal station apparatus, reference numerals 304-1 to 304-4 are first signal processing units, and reference numeral 305 is a second signal. A processing unit (strictly including other base station device functions such as an interface circuit, a MAC layer processing circuit, and a communication control circuit). In FIG. 16, the 100 antenna elements of the base station apparatus 303 are divided into four groups for every 25 antenna elements. 25 antenna elements in the same group, in a very narrow interval as compared with the case of FIG. 15, over a short length D ₃ than the length D _1, are arranged in a linear array form.

In FIG. 16, 100 antenna elements of the base station apparatus 303 are mounted in a linear array for each group (25 antenna elements). That is, 100 antenna elements of the base station apparatus 303 are mounted in a linear array for each first signal processing unit 304. The first signal processing units 304-1 to 304-4 form one directional beam for each group (25 antenna elements) by signal processing. Further, 16 of the antenna element of the terminal station device 302 is equally spaced linear array shape over the length D _2.

  Here, in each of the two cases shown in FIGS. 15 and 16, the transmission characteristics when the four signal systems are transmitted by being spatially multiplexed (four multiplexed) will be compared. The characteristics of transmission can be grasped by the gain of each transmission path shown in FIG. 2C, which corresponds to the square value of the absolute value of the singular value obtained by performing the singular value decomposition of the channel matrix. In the case illustrated in FIG. 15, for example, assuming that the base station apparatus 301 is the transmitting station 11 and the terminal station apparatus 302 is the receiving station 12 assuming a downlink, the size of the channel matrix is 16 × 100. Singular value decomposition is performed on this matrix.

On the other hand, in the case shown in FIG. 16, the following procedure is assumed and its characteristics are grasped. First, base station apparatus 303 is based on a 16 × 25 channel matrix (in the case of the downlink) formed by 25 antenna elements of base station apparatus 303 and 16 antenna elements of terminal station apparatus 302. Suppose that singular value decomposition is performed on the first right singular vector and transmitted. Specifically, the base station device 303 includes a portion between each of the 25 antenna elements connected to the first signal processing units 304-1 to 304-4 and the 16 antenna elements of the terminal station device 302. The singular value decomposition is performed on the channel matrices H _{1 to} H ₄ . The partial channel matrices H _{1 to} H ₄ are shown in Equation (16).

Here the parts of the channel matrix _H 1 to H ₄ is a matrix of 16 × 25. Therefore, v _ij forming each right singular vector is a 25-dimensional vector, and represents the right singular vector corresponding to the j-th singular value in the i-th group of the four groups of antenna groups. Similarly, u _ij forming each left singular vector is a 16-dimensional vector, which represents the left singular vector corresponding to the j th singular value in the i th group of the four groups of antennas. The overall channel matrix between all antenna elements of the base station apparatus 303 and the terminal station apparatus 302 is shown in Expression (17).

The transmission weight matrix W _Tx here is shown in Equation (18).

Here for convenience of notation, is used a Hermitian conjugate representation of the transmission weight matrix W _Tx, the size of the transmit weight matrix W _Tx itself is 100 × 4. As a result, the product of the entire channel matrix and the transmission weight matrix is expressed by Equation (19).

Here, H _i v _i1 is a 16 × 1 matrix (column vector), which coincides with λ _i u _{i1 according} to equation (16). As a result, the total size of the product of the entire channel matrix and the transmission weight matrix is 16 × 4. In general, since the first left singular vectors of the partial channel matrices H _{1 to} H ₄ are not orthogonal to each other, a reception weight for signal separation is formed and multiplied at the time of reception. However, when the first left singular vectors of the partial channel matrices H _{1 to} H ₄ are substantially orthogonal to each other, the matrix represented by the product of the entire channel matrix and the transmission weight matrix is subjected to singular value decomposition. The square values of the absolute values of the four singular values substantially coincide with the line gain of the transmission line in FIG. In this evaluation, it is assumed that each element of the channel matrix is expressed by the following equation (20) by a free space propagation model considering only the line-of-sight wave.

Here, r _ij represents the distance between the i-th antenna on the transmitting side and the j-th antenna on the receiving side, and λ represents the wavelength. In order to grasp the overall characteristics, the coefficients multiplied as coefficients as a whole are omitted here for simplicity.

Therefore, in FIG. 16, the characteristics of FIG. 15 installed with the same antenna aperture length are compared for L = 100 m, D ₁ = 12 m, D ₂ = 10 cm, D ₃ = 30 cm, and frequency 80 GHz. Here, when the absolute value of the singular value is X as the line gain, the line gain is evaluated as 20 Log (X) [dB]. At this time, the gains of the four lines in FIG. 15 are -56.5 dB, -83.4 dB, -118.3 dB, and -157.2 dB, respectively, while the above processing is applied to FIG. Respectively becomes -62.5 dB. In the case of FIG. 15, only the line with the maximum gain corresponding to the first singular value in FIG. 2 (c) has a large value, and the gain of the line corresponding to the remaining singular value is relatively small. Although it depends on the values of parameters such as antenna gain, only a line corresponding to the first singular value can be used. On the other hand, in the case of FIG. 16, it can be seen that the four transmission lines can be used almost equally. Here, the gain difference with respect to the first singular value in FIG. 15 and the gain difference with respect to the singular value in FIG. 16 is 6 dB. In FIG. 16, the number of antenna elements used for directional beam formation is reduced from 100 to 25. 4, corresponding to 10 Log (1/4) = − 6 dB. In other words, the efficiency is reduced to 1/4 by dividing the antenna element group into four. However, for the channel capacity due to the Shannon limit, the transmission capacity is increased by multiplexing four signal sequences rather than improving the SNR by 6 dB. In terms of increase, it is overwhelmingly efficient.

  The higher the received signal power of the reflected wave component is, the more the line gain of the line corresponding to the singular value after the second singular value can be effectively used by setting parameter values such as transmission power and antenna gain. In general, if the number of antenna elements is increased to compensate for the line gain that decreases with the use of a high frequency band such as millimeter waves, the line gain of the line corresponding to the singular value after the second singular value is sufficient. In general, it is difficult to think of the situation, and as data transmission, the transmission capacity is increased in FIG. 16 in which transmission for four lines is performed in parallel rather than in the case of FIG. 15 in which transmission for substantially one line is performed. Can be seen as suitable. It is effective to group antennas in this way and efficiently use the virtual transmission line corresponding to the first singular value in each group.

(Singular value decomposition and calibration)
Here, consider the case where the amplification factor of the low noise amplifier or the high power amplifier is not different for each antenna element (or can be approximated to be a constant value) in the above-described equation (14). In order to simplify the equation, the uplink channel matrix is H _UL , the downlink channel matrix is H _DL , the base station apparatus side calibration matrix is C _BS , and the terminal station apparatus side calibration matrix is C _MT. Furthermore, the singular value decomposition results of the uplink channel matrix H _UL are shown in Equation (21) and Equation (22).

Here, it is assumed that the absolute values of the diagonal terms in Equation (21) are all equal in each matrix. In this case, both equations (21) are unitary matrices, and this term behaves as a rotation of the coordinate axis.
Here, the downlink channel matrix _HDL is expressed by using the calibration matrices C _BS and C _MT and the uplink channel matrix H _UL , and the equation (22) is substituted into the equation (23).

Here, expressions on both sides of the _{DUL on} the right side are shown in Expression (24) and Expression (25).

  When the absolute value of each component of the calibration matrix is approximately equal, the uplink channel matrix is calibrated and the singular value decomposition is performed to obtain the transmission / reception weight, and the uplink channel matrix is the singular value. Expression (24) and Expression (24) are the same as the result of performing the calibration process by multiplying each component of the first right singular vector and the first left singular vector obtained by the decomposition by the calibration coefficient. It can be seen from (25).

  In other words, if singular value decomposition is performed on the uplink channel matrix, calibration processing is performed on each component of the obtained reception weight vector (or matrix) without performing singular value decomposition on the downlink. For example, it can be seen that a desired transmission weight vector (or matrix) can be acquired. For this reason, the singular value decomposition needs to be performed only once.

(About the circuit configuration of the base station apparatus in the present invention)
Hereinafter, the circuit configuration of the base station apparatus 303 in the first embodiment of the present invention will be described with reference to the drawings.
FIG. 17 is a schematic block diagram showing an example of the configuration of the base station apparatus 70 in the MIMO system of the present invention. Although FIG. 16 illustrates the case of point-to-point type one-to-one communication with one base station device 303 and one terminal station device 302, of course, there are a plurality of terminal station devices 302. It does not matter. The transmission / reception of the signal in FIG. 16 communicates with only one terminal station apparatus 302 at the same time when viewed from the subcarrier of interest, and is in the form of single-user MIMO transmission. Is selected. If OFDMA is used for access control, a different terminal station apparatus 302 may be allocated for each subcarrier. However, if attention is paid to each subcarrier, the allocation is limited to one terminal station apparatus 302. Further, when the terminal is fixed by Point-to-Point type communication, the process of selecting the communication partner terminal station device 302 in the scheduling becomes unnecessary. In the first embodiment of the present invention, the point-to-multipoint type one-to-many communication multi-user MIMO transmission mode can be used. However, in the following description, regardless of such a variation. Next, general single-user MIMO transmission will be described. Further, in the following description, in order to simplify the description, a case where a broadband system is assumed and signal processing for each subcarrier on the frequency axis is performed as in OFDM or SC-FDE will be described. (For example, a narrow band single carrier system) can be extended.

  As illustrated in FIG. 17, the base station device 70 corresponding to the base station device 303 includes a first transmission signal processing unit 181-1 to 181-4, a second transmission signal processing unit 71, and a first reception. A signal processing unit 185-1 to 185-4, a second reception signal processing unit 75, an interface circuit 77, a MAC (Medium Access Control) layer processing circuit 78, and a communication control circuit 120 are provided. The MAC layer processing circuit 78 has a scheduling processing circuit 781.

  The base station device 70 inputs and outputs data with an external device or a network via the interface circuit 77. The interface circuit 77 detects data to be transferred on the wireless line among the input data, and outputs the detected data to the MAC layer processing circuit 78. The MAC layer processing circuit 78 performs processing related to the MAC layer in accordance with an instruction of the communication control circuit 120 that performs management control of the operation of the entire base station device 70. Here, the processing related to the MAC layer includes conversion between data input / output by the interface circuit 77 and data transmitted / received on the wireless line, addition of header information of the MAC layer, and the like. In this process, the scheduling processing circuit 781 performs various scheduling processes of the terminal station apparatus 302 that performs spatial multiplexing. The scheduling processing circuit 781 outputs the scheduling result to the communication control circuit 120. In MIMO transmission, signals of a plurality of signal sequences are spatially multiplexed at a time and transmitted, so that a plurality of signal sequences are output from the MAC layer processing circuit 78 to the second transmission signal processing unit 71.

  Although the operation of the second transmission signal processing unit 71 will be described later, basically, a predetermined modulation process is performed on a plurality of series of signals from the MAC layer processing circuit 78, and some precoding process (on the transmission side) is performed as necessary. And the like, and the like are processed and output to the first transmission signal processing units 181-1 to 181-4. At this time, regardless of whether OFDM or SC-FDE is used, when the first transmission signal processing units 181-1 to 181-4 perform signal processing on the frequency axis, the second transmission signal processing unit 71 generates a signal on the frequency axis and outputs it to the first transmission signal processing units 181-1 to 181-4. When signal processing on the time axis is performed by the first transmission signal processing unit as in a fifth embodiment to be described later, a time axis signal may be output. As shown in FIG. 16, each of the first transmission signal processing units 181-1 to 181-4 is connected to a plurality of antennas and outputs a transmission signal to each antenna. At this time, among the antenna element groups grouped for each of the first transmission signal processing units 181-1 to 181-4, a signal obtained by multiplying the transmission weight vector corresponding to the first singular value (strictly speaking, For example, in the case of OFDM, a signal obtained by synthesizing the signals of the subcarriers is converted into a time axis component, and the signal is upconverted to a radio frequency) and transmitted from each antenna.

  Next, at the time of reception, signals received by a plurality of antennas connected to the respective first reception signal processing units 185-1 to 185-4 (more precisely, the received radio frequency signals are down-converted into baseband signals). For example, in the case of OFDM, the time axis signal is converted into a frequency axis signal by FFT) and multiplied by a predetermined reception weight vector to convert it into one complex scalar quantity for each subcarrier, To the received signal processing unit 75. In this example, the second received signal processing unit 75 refers to four received signal sequences, and first performs channel estimation for each subcarrier using a known training signal provided at the head of the received signal. A × 4 MIMO channel matrix is acquired for each subcarrier. A reception weight matrix is calculated based on this channel matrix, and a detection process of the transmitted signal is performed based on the acquired reception weight matrix. For example, a ZF (Zero Forcing) type inverse matrix or a MMSE (Maximum Mean Square Error) type reception weight matrix is used. If there is a margin in signal processing, MLD (Maximum Likelihood Detection), simple MLD (QR-MLD) using QR decomposition, or the like may be used. A signal detected by the reception signal processing is output to the MAC layer processing circuit 78, performs predetermined MAC layer processing, and is output to the network side via the interface circuit 77.

FIG. 18 is a schematic block diagram illustrating an example of the configuration of the first transmission signal processing unit 181 in the base station device 70 according to the first embodiment of the present invention. As shown in FIG. 18, the first transmission signal processing unit 181 includes a first transmission signal processing circuit 111, an IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval) giving circuit 813. −1 to 813-N ′ _BS-Ant , D / A (digital / analog) converters 814-1 to 814-(N ′ _BS-Ant ), local oscillator 815, mixers 816-1 to 816-( N'BS _-Ant ), filters 817-1 to 817- (N'BS _-Ant ), high power amplifiers (HPA) 818-1 to 818- (N'BS _-Ant ), and antenna element 819-1 ˜819- (N ′ _BS-Ant ) and a first transmission weight processing unit 130. N ′ _BS-Ant is a value obtained by dividing the total number of antenna elements of the base station apparatus 70 by the spatial multiplexing number (= N _BS-Ant / N _SDM ). In short, N ′ _BS-Ant represents the number of a plurality of antenna elements included in one first transmission signal processing unit 181. The first transmission signal processing circuit 111 is connected to the second transmission signal processing unit 71 shown in FIG. Further, the first transmission signal processing circuit 111 and the first transmission weight processing unit 130 are connected to the communication control circuit 120 via the second transmission signal processing unit 71 shown in FIG. In the example of FIG. 17, four first transmission signal processing units (181-1 to 181-4) are connected to the base station device 70, and an explanation focusing on one of them will be given below.

The first transmission weight processing unit 130 includes a first channel information acquisition circuit 131, a first channel information storage circuit 132, and a first transmission weight calculation circuit 133. Here, the subscript (N ′ _BS-Ant ) of the circuit from the IFFT & GI giving circuits 813-1 to 813-(N ′ _BS-Ant ) to the antenna elements 819-1 to 819-(N ′ _BS-Ant ) is , Represents the number of antenna elements included in the first transmission signal processing unit 181 of the base station device 70.

In the present invention according to the first embodiment, since a signal of a plurality of systems N _SDM (= 4) is spatially multiplexed and transmitted to one terminal station apparatus 302, a plurality of signal sequences are transmitted from the MAC layer processing circuit 78. A transmission signal is input to each of the first transmission signal processing units 181-1 to 181-4 via the second transmission signal processing unit 71. When the data to be transmitted to the destination terminal station 302 is input from the MAC layer processing circuit 78, the second transmission signal processing unit 71 generates a wireless packet to be transmitted through a wireless line and performs modulation processing. Here, for example, if the OFDM modulation method is used, the signal of each signal sequence is modulated for each subcarrier. The signal subjected to modulation processing is subjected to precoding processing as necessary. Here, the precoding processing may be multiplication of a transmission weight matrix for suppressing signal leakage between virtual transmission lines corresponding to a plurality of first singular values. Alternatively, such precoding processing need not be performed.
The N _SDM system signals generated in this way are input to the first transmission signal processing units 181-1 to 181-4. In each of the first transmission signal processing units 181-1 to 181-4, the input digital baseband signal input #i (i is 1 to N _SDM ) is input to the first transmission signal processing circuit 111-i. The The first transmission signal processing circuit 111 basically performs transmission weight multiplication and signal processing of the remaining physical layers. For example, if the OFDM modulation method is used, the input baseband signal subjected to the modulation process is multiplied by a transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819-(N ′ _BS-Ant ) is subjected to the remaining signal processing as necessary, and IFFT & GI adding circuits 813-1 to 813-( N ′ _BS−Ant ) is converted from a signal on the frequency axis to a signal on the time axis. The converted signal is further subjected to processing such as insertion of a guard interval and waveform shaping between OFDM symbols (between block transmission blocks in the case of SC-FDE (Single-Carrier Frequency Domain Equalization)), and the like. Every 1 to 819- (N ′ _BS-Ant ) is converted from digital sampling data to a baseband analog signal by D / A converters 814-1 to 814-(N ′ _BS-Ant ). Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816- (N ′ _BS-Ant ), and is up-converted to a radio frequency signal. Here, since the upconverted signal includes a signal outside the band of the channel to be transmitted, the signal outside the band is removed by the filters 817-1 to 817-(N ′ _BS-Ant ) and transmitted. Generate an electrical signal to power. The generated signal is 'is amplified by _(BS-Ant, antenna elements 819-1~819- (N high-power amplifier 818-1~818- N)' is transmitted from the _BS-Ant).

  Note that the transmission weight vector multiplied by the first transmission signal processing circuit 111 is acquired from the first transmission weight calculation circuit 133 provided in the first transmission weight processing unit 130 during the signal transmission processing. In the first transmission weight processing unit 130, the channel information acquired by the first reception signal processing units 185-1 to 185-4 in the first channel information acquisition circuit 131 is transmitted via the communication control circuit 120 (strictly Are obtained separately by the second reception signal processing unit 75 and the second transmission signal processing unit 71 together, and are stored in the first channel information storage circuit 132 while being sequentially updated. At the time of signal transmission, according to an instruction from the communication control circuit 120, the first transmission weight calculation circuit 133 reads channel information corresponding to the destination terminal station device from the first channel information storage circuit 132, and the read channel information The transmission weight vector is calculated based on

  There are several variations in the channel estimation method and the transmission / reception weight calculation method when the virtual transmission path corresponding to the first singular value is utilized, and details of a method for efficiently acquiring this will be described later. As an example, for the transmission weight vector, for example, a singular value decomposition may be performed on the acquired channel matrix, and a first right singular vector obtained as a result may be used.

Or, paying attention to one central part of the antenna on the terminal station device 302 side, one of the antenna and one received signal processing unit (any one of 185-1 to 185-4) of the base station device 70 Based on the channel vector between the antenna elements 819-1 to 819-4 provided and N ′ _BS-Ant , each component of the transmission weight vector may be obtained by Expression (7) (maximum ratio combining weight). Alternatively, all the absolute values may be made constant with respect to the value given by the equation (7) (weight of equal gain synthesis).

  Or, when the transmitting side is transmitting a signal by multiplying a plurality of antenna elements by a predetermined transmission weight vector, it is effectively transmitted from a plurality of transmitting antennas even though it is actually transmitted. Is equivalent to that transmitted from the virtual antenna element, and a training signal is transmitted from the one virtual antenna element by multiplying by a predetermined transmission weight vector. A vector of channel information to and from the receiving antenna is obtained, and each component of the received weight vector may be obtained from Equation (7) based on this vector (maximum ratio combining weight) or Equation (7). All absolute values may be made constant with respect to a given value (equal gain combining weight).

  If the channel vector at the time of reception is known, it is possible to acquire uplink channel information by the method of implicit feedback. Based on the uplink channel vector obtained in this way, the transmission weight vector May be calculated similarly. Similarly, the transmission weight vector may be calculated by an implicit feedback method in which a calibration process is directly performed on the uplink reception weight vector.

  The first transmission weight calculation circuit 133 outputs the transmission weight vector calculated in this way to the first transmission signal processing circuit 111. Further, the communication control circuit 120 manages the control related to the entire communication such as the management of the destination terminal station apparatus and the entire timing control. The communication control circuit 120 outputs information indicating the destination terminal station apparatus and the like to the first transmission weight processing unit 130 that performs signal processing related to the above-described transmission weight calculation.

  In the above description, the first channel information acquisition circuit 131 acquires the channel information acquired by the first reception signal processing units 185-1 to 185-4 via the communication control circuit 120, and this channel information. However, it is not necessary to frequently update the channel information in a high-level sighting environment in which channel time fluctuation can be ignored. For example, the first channel information acquisition circuit 131 acquires channel information in advance before starting service operation, and further stores a transmission weight vector calculated from the value of the channel information (not shown in the figure). In this case, the “transmission weight storage circuit” is mounted and recorded), and it may be used repeatedly. Further, as an intermediate between them, the transmission weight vector is basically read from the first transmission weight storage circuit, but the transmission weight vector is updated based on the sequentially acquired channel information, and the updated channel information is updated. It is also possible to adopt a configuration in which the transmission weight vector is updated based on a predetermined time interval.

FIG. 19 is a schematic block diagram illustrating an example of the configuration of the first reception signal processing unit 185 in the base station apparatus 70 according to the first embodiment of the present invention. As shown in FIG. 19, the first received signal processing unit 185 includes antenna elements 851-1 to 851- (N ′ _BS-Ant ) and low noise amplifiers (LNA) 852-1 to 852- (N ′ _{BS− Ant} ), local oscillator 853, mixers 854-1 to 854- (N′BS _-Ant ), filters 855-1 to 855- (N′BS _-Ant ), and A / D (analog / digital) conversion Units 856-1 to 856- (N'BS _-Ant ), FFT (Fast Fourier Transform) circuits 857-1 to 857- (N'BS _-Ant ), and a first reception weight processing unit 160. And a first received signal processing circuit 158. The first reception signal processing circuits 158-1 to 158-N _SDM (= 4) are connected to the second reception signal processing unit 75 shown in FIG. Further, the first reception signal processing circuits 158-1 to 158-N _SDM (= 4) and the first reception weight processing unit 160 are connected via the second reception signal processing unit 75 shown in FIG. The communication control circuit 120 is connected. The first reception weight processing unit 160 includes a first channel information estimation circuit 161 and a first reception weight calculation circuit 162. Similar to the description of the first transmission signal processing unit 181, four first reception signal processing units (185-1 to 185-4) are connected to the base station apparatus 70 in the example of FIG. 17. However, an explanation focusing on one of them will be given below.

First, the signals received by the antenna elements 851-1 to 851- (N ′ _BS-Ant ) are amplified by the low noise amplifiers 852-1 to 852- (N ′ _BS-Ant ). The amplified signal and the local oscillation signal output from the local oscillator 853 are multiplied by mixers 854-1 to 854- (N ′ _BS-Ant ), and the amplified signal is changed from a radio frequency signal to a baseband signal. Down converted. Since the down-converted signal includes a signal outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855- (N ′ _BS-Ant ). The signal from which the out-of-band component is removed is converted into a digital baseband signal by the A / D converters 856-1 to 856- (N ′ _BS-Ant ). In the case of OFDM, for example, all the digital baseband signals are input to FFT circuits 857-1 to 857-(N ′ _BS-Ant ), and a predetermined symbol timing determined by a timing detection circuit omitted here. Then, the signal on the time axis is converted into a signal on the frequency axis (separated into signals of each subcarrier). The signal separated into the subcarriers is input to the first received signal processing circuit 158 and also input to the first channel information estimation circuit 161. In FIG. 19, the circuit for detecting the symbol timing of OFDM is omitted, but the symbol timing can be grasped by some existing method.

In the first channel information estimation circuit 161, an antenna element of each terminal station device 302 based on a known signal for channel estimation separated into each subcarrier (a preamble signal added to the head of a radio packet), The channel information between each antenna element 851 of the base station apparatus 70 is estimated for each subcarrier, and the estimation result is output to the first reception weight calculation circuit 162. The first reception weight calculation circuit 162 calculates a reception weight vector to be multiplied for each subcarrier based on the input channel information. At this time, the reception weights for combining the signals received by the respective antenna elements 851-1 to 851- (N ′ _BS-Ant ) are the first reception signal processing units 185-1 to 185-N _SDM (= 4). The first received signal processing units 185-1 to 185-N _SDM are calculated separately.

In the first reception signal processing circuit 158, signals for each subcarrier input from the FFT circuits 857-1 to 857- (N ′ _BS-Ant ) (more precisely, signals from a plurality of antenna elements are used as elements. A result of multiplying the received signal vector) by the received weight input from the first received weight calculation circuit 162 (more precisely, a received weight vector having received weights corresponding to a plurality of antenna elements). The signals received by the antenna elements 851-1 to 851- (N ′ _BS-Ant ) are added and combined for each subcarrier. The first received signal processing circuit 158 outputs the added and synthesized signal to the second received signal processing unit 75. Here, the addition synthesis means addition after multiplication of each component of the vector in the vector product for each subcarrier, and the multiplication of the reception signal and the reception weight and the total addition synthesis of the result are mathematically vectors. Corresponds to product processing.

  The reception weight vector multiplied by the first reception signal processing circuit 158 is acquired from the first reception weight calculation circuit 162 provided in the first reception weight processing unit 160 at the time of signal reception processing. The first reception weight processing unit 160 uses the channel information acquired by the first channel information estimation circuit 161 to calculate a reception weight vector by the first reception weight calculation circuit 162. For example, if the terminal station device 302 is performing signal transmission on the virtual transmission path corresponding to the first singular value toward the plurality of antenna elements 851 of the first received signal processing unit 185, the terminal station device 302 Since the transmission weight for the virtual transmission line corresponding to the first singular value is used to transmit to each first received signal processing unit 185 using one virtual antenna element. Then, channel information on the reception side between the one antenna and the plurality of antenna elements 851 of the first reception signal processing unit 185 is obtained, and each component of the reception weight vector with respect to this channel vector is expressed by Equation (7). It may be obtained (weight for maximum ratio combining), or all absolute values may be fixed with respect to the value given by equation (7) (weight for equal gain combining). Alternatively, the singular value decomposition is performed on the channel matrix between the antenna element included in the terminal station device 302 and the plurality of antenna elements 851 of the first reception signal processing unit 185 included in the base station device 70. Each of the left singular vectors may be used as a reception weight vector.

  The first reception weight calculation circuit 162 outputs the reception weight vector calculated in this way to the first reception signal processing circuit 158. Further, the communication control circuit 120 manages control related to overall communication such as management of a transmission source station and overall timing control.

  In the above description, the channel information acquired by the first channel information estimation circuit 161 has been described as being used for sequential reception weight calculation. It is not necessary to update the channel information. The first channel information estimation circuit 161 acquires channel information in advance before starting service operation, for example, and stores a reception weight vector calculated from the value of the channel information (not shown in the figure). However, in this case, the “first reception weight storage circuit” is implemented in a configuration in which the first reception weight calculation circuit 162 is mounted and recorded in the subsequent stage), and it may be used repeatedly. In this case, the communication control circuit 120 outputs information indicating the terminal station device or the like of the transmission source to the first reception weight storage circuit that outputs the reception weight. Further, as an intermediate between them, the reception weight vector is basically read from the first reception weight storage circuit, but the reception weight vector is updated based on the sequentially acquired channel information, and the updated channel information is updated. It is also possible to adopt a configuration in which the reception weight vector is updated based on a predetermined time interval.

  Note that in the second received signal processing unit 75 in FIG. 17, received signals that have been multiplied by the reception weight vectors from the first received signal processing units 185-1 to 185-4 described above and collected in one system are received. Although these signals are input, these are substantially equivalent to 4 × 4 MIMO channel reception signals, and it is possible to perform processing of a spatially multiplexed signal sequence by reception signal detection processing of the prior art. Specifically, 4 × 4 when four signal sequences transmitted on the transmission side are received by four sets of virtual antennas configured by a plurality of antennas on the reception side (base station device 70 side). For the MIMO channel, a 4 × 4 channel matrix is obtained for each subcarrier using a known training signal for channel estimation given to the head of the received signal. A reception weight matrix is calculated based on this channel matrix, and a detection process of the transmitted signal is performed based on the acquired reception weight matrix. For example, a ZF (Zero Forcing) type inverse matrix or a MMSE (Maximum Mean Square Error) type reception weight matrix is used. If there is a margin in signal processing, MLD (Maximum Likelihood Detection), simple MLD (QR-MLD) using QR decomposition, or the like may be used. In addition, in the signal detection processing here, for example, a soft decision of the received signal is performed once, deinterleaving processing is performed as necessary, and then error correction processing is performed to perform final signal detection. good. A signal detected by the reception signal processing is output to the MAC layer processing circuit 78, performs predetermined MAC layer processing, and is output to the network side via the interface circuit 77.

  The MAC layer processing circuit 78 also performs processing related to the MAC layer (for example, conversion between data input / output to / from the interface circuit 77 and data transmitted / received on the wireless line, that is, a wireless packet, termination of header information of the MAC layer) Etc.). In the case of Point-to-Point type communication, the scheduling processing circuit 781 is substantially unnecessary. However, when Point-to-MultiPoint type communication is performed with a plurality of terminal station devices 302, Various scheduling processes for selecting the terminal station device 302 to perform communication are performed, and the scheduling result is output to the communication control circuit 120. The received data processed by the MAC layer processing circuit 78 is output to an external device or network via the interface circuit 77.

  In addition, the communication control circuit 120 manages control related to overall communication such as management of the terminal station device 302 as a transmission source and overall timing control. In addition, information indicating the transmission source terminal station apparatus and the like is input from the communication control circuit 120 to the first reception weight processing unit 160 that performs signal processing related to the calculation of the reception weight described above.

As for signal reception, as in the case of transmission, in the wideband system using the OFDM modulation scheme or SC-FDE scheme, the above-described reception weight multiplication is performed for each subcarrier. That is, FFT is performed on the signals output from the A / D converters 856-1 to 856- (N′BS _-Ant ) by the FFT circuits 857-1 to 857- (N′BS _-Ant ) to each subcarrier. The signal processing in the first channel information estimation circuit 161 and the reception signal processing in the first reception signal processing circuit 158 are performed for each separated subcarrier.

The above is the description of the base station apparatus 70 in the first embodiment of the present invention. What is important here is that the local oscillator 815 in the first transmission signal processing unit 181 is common to the mixers 816-1 to 816- (N ′ _BS-Ant ) in each antenna system in the same first transmission signal processing unit 181. On the other hand, the local oscillator 815 is not shared between different first transmission signal processing units 181. Similarly, the local oscillator 853 in the first reception signal processing unit 185 is shared by the mixers 854-1 to 854- (N ′ _BS-Ant ) in each antenna system in the same first reception signal processing unit 185. On the other hand, it is also important that the local oscillator 853 is not shared between different first received signal processing units 185. As shown in FIG. 16, in general, the first transmission signal processing units 181-1 to 181-4 (in FIG. 16, the first signal processing units 304-1 to 304-4) are physically separated on the order of several meters. In a high frequency band such as millimeter waves, the loss when the cable is routed is as large as 10 dB or more per meter. Since there are a large number of antenna elements connected to the first transmission signal processing unit 181 and the first reception signal processing unit 185, the mixers 816 and 854 need to branch and input signals to a plurality of systems. Considering the drop in level due to this branching, the cable length loss of several meters is not negligible, so it is closed within the individual first transmission signal processing unit 181 and the individual first reception signal processing unit 185 and is locally A configuration in which the oscillators 815 and 853 are respectively shared and not shared between the different first transmission signal processing units 181 and the first reception signal processing units 185 is effective.

  Here, in each antenna, the phase of the transmission / reception signal is adjusted for directivity control, but the same first transmission signal processing units 181-1 to 181-4 (the first signal processing unit in FIG. 16). 304-1 to 304-4), it is easy to make the phase relationship of the signals inputted from the respective local oscillators 815 or 853 always constant, and the local oscillator 815 or local oscillator 853 It is possible to determine what kind of phase relationship should be multiplied by the transmission / reception weights in the non-dependent part. However, when the local oscillator 815 uses a plurality of asynchronous ones in the first transmission signal processing unit 181 (or the local oscillator 853 in the first reception signal processing unit 185), at least the first transmission signal is used. When multiplying the transmission weight in the processing unit 181, it is necessary to adjust in consideration of the complex phase relationship between the plurality of local oscillators 815 (or 853). If this adjustment is omitted, directivity control functions effectively. No longer. Care should be taken in this regard when designing the device. In the first embodiment of the present invention, the same first transmission signal processing unit 181 shares the local oscillator 815 for the reason described above, and the same first reception signal processing unit 185 uses the local oscillator 853 for the reason described above. However, since it is possible to perform signal separation between four signal sequences that are spatially multiplexed on the reception side, it is not necessary to perform complete signal separation on the transmission side as in multi-user MIMO. .

  In this way, a plurality of signal sequences can be basically separated by signal processing on the receiving side, but the phase between the first transmission signal processing units 181-1 to 181-4 by channel feedback or the like, for example. If the relationship is known, it is also possible to multiply (ie, transmit precoding) a signal separation transmission weight matrix in advance on the transmission side. In this case, the second transmission signal processing unit 71 of the base station device 70 multiplies this transmission weight matrix. In the calculation of the transmission weight matrix, the transmission weight matrix is acquired using the calibration process based on the uplink channel information regarding the four signal sequences after being multiplied by the reception weight vector acquired by the second reception signal processing unit 75. It doesn't matter. However, as described above, since the channel matrix can be obtained from the training signal added to the transmission signal in the terminal station device 302 on the reception side even in the downlink, the second processing is not necessarily performed if the signal processing on the reception side is utilized. The transmission signal matrix 71 need not multiply the transmission weight matrix.

(About the circuit configuration of the terminal station apparatus 60 corresponding to the terminal station apparatus 302 in the present invention)
FIG. 20 is a schematic block diagram illustrating an example of the configuration of the terminal station device 60 corresponding to the terminal station device 302 in the first embodiment of the present invention. As illustrated in FIG. 20, the terminal station device 60 includes a transmission unit 61, a reception unit 65, an interface circuit 67, a MAC (Medium Access Control) layer processing circuit 68, and a communication control circuit 121.

  The terminal station device 60 inputs and outputs data with an external device or a network via the interface circuit 67. The interface circuit 67 detects data to be transferred on the wireless line among the input data, and outputs the detected data to the MAC layer processing circuit 68. The MAC layer processing circuit 68 performs processing related to the MAC layer in accordance with an instruction from the communication control circuit 121 that performs management control of the operation of the entire terminal station device 60. Here, the processing related to the MAC layer includes conversion between data inputted / outputted by the interface circuit 67 and data transmitted / received on the wireless line, that is, a wireless packet, addition of MAC layer header information, and the like. In the MIMO transmission, a signal is spatially multiplexed and transmitted to one terminal station device 60, so that a plurality of signal sequences are output from the MAC layer processing circuit 68 to the transmission unit 61.

FIG. 21 is a schematic block diagram illustrating an example of the configuration of the transmission unit 61 in the terminal station device 60 according to the first embodiment of the present invention. As illustrated in FIG. 21, the transmission unit 61 includes transmission signal processing circuits 811-1 to 811-N _SDM (N _SDM is an integer equal to or greater than 2) and addition / synthesis circuits 812-1 to 812-N _MT-Ant (N _MT-Ant is an integer of 2 or more), IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval) giving circuits 813-1 to 813-N _MT-Ant , and D / A (digital) / Analog) converters 814-1 to 814-N _MT-Ant , local oscillator 815, mixers 816-1 to 816-N _MT-Ant , filters 817-1 to 817-N _MT-Ant , and high power amplifier (HPA) and _{818-1~818-N MT-Ant,} an antenna element _{819-1~819-N MT-Ant,} a first transmission weight processing unit 140 It is provided. The transmission signal processing circuits 811-1 to 811-N _SDM and the first transmission weight processing unit 140 are connected to the communication control circuit 121 shown in FIG.

The first transmission weight processing unit 140 includes a channel information acquisition circuit 141, a channel information storage circuit 142, and a first transmission weight calculation circuit 143. Here, _{N SDM} subscripts of the transmission signal processing circuit _{811-1~811-N SDM} in FIG 21 represents a multiplex number for performing spatial multiplexing simultaneously. In addition, N _MT-Ant of the subscript of the circuit from the adder / synthesizer circuit 812-1 to 812-N _MT-Ant to the antenna elements 819-1 to 819-N _MT-Ant is the number of antenna elements provided in the terminal station device 60 Represents. N _MT-Ant is 16, for example.

In the first embodiment, since one terminal station device 60 spatially multiplexes and transmits a signal to the base station device 70, a plurality of signal sequences are input from the MAC layer processing circuit 68 to the transmission unit 61 and input. The plurality of signal series are input to the transmission signal processing circuits 811-1 to 811-N _SDM . The transmission signal processing circuits 811-1 to 811-N _SDM transmit data (wireless data) transmitted from the MAC layer processing circuit 68 to the data (data input # 1 to #N _SDM ) to be transmitted to the destination base station apparatus 70 via a wireless line. When a packet) is input, modulation processing is performed on this. Here, for example, if the OFDM modulation method is used, the signal of each signal sequence is modulated for each subcarrier. Further, the baseband signal subjected to modulation processing is multiplied by a transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819 -N _MT-Ant is subjected to the remaining signal processing as necessary, and each transmission signal processing is performed as sampling data of the transmission signal in the baseband. It is input from the circuit _{811-1~811-N SDM} to additive synthesis circuit _{812-1~812-N MT-Ant.}

The signals input to the adder / synthesizers 812-1 to 812-N _MT-Ant are synthesized for each subcarrier. The combined signal is converted from a signal on the frequency axis to a signal on the time axis by IFFT & GI adding circuits 813-1 to 813 -N _MT-Ant , and further, insertion of a guard interval or between OFDM symbols (SC-FDE ( In the case of single-carrier frequency domain equalization), processing such as waveform shaping between blocks is performed, and D / A converters 814-1 to 814-1 are provided for each of the antenna elements 819-1 to 819-N _MT-Ant. 814-N _MT-Ant converts digital sampling data to baseband analog signal. Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816-N _MT-Ant and up-converted to a radio frequency signal. Here, since the upconverted signal includes a signal in an area outside the band of the channel to be transmitted, the signal outside the band should be removed by the filters 817-1 to 817-N _MT-Ant and transmitted. Generate a signal. The generated signal is amplified by the high-power amplifier _{818-1~818-N MT-Ant,} and transmitted from the antenna elements _{819-1~819-N MT-Ant.}

In FIG. 21, after adding and synthesizing the signals of the subcarriers by the adding and synthesizing circuits 812-1 to 812-N _MT-Ant , processing such as IFFT processing, insertion of guard intervals, and waveform shaping is performed. However, the transmission signal processing circuits 811-1 to 811-N _SDM perform these processes, and the IFFT sampling signals on the time axis are combined by the addition combining circuits 812-1 to 812-N _MT-Ant. The IFFT & GI adding circuits 813-1 to 813 -N _MT-Ant may be omitted (strictly, these are included in the transmission signal processing circuits 811-1 to 811 -N _SDM ). In this case, the remaining signal processing as necessary after transmission weight multiplication in the transmission signal processing circuits 811-1 to 811-N _SDM refers to processing such as IFFT processing, insertion of guard intervals, waveform shaping, and the like.

The transmission weights multiplied by the transmission signal processing circuits 811-1 to 811-N _SDM are acquired from the first transmission weight calculation circuit 143 provided in the first transmission weight processing unit 140 at the time of signal transmission processing. To do. In the first transmission weight processing unit 140, the channel information acquisition circuit 141 separately acquires the channel information acquired by the reception unit 65 via the communication control circuit 121, and sequentially updates the channel information while storing the channel information. Store in circuit 142. When transmitting a signal, in accordance with an instruction from the communication control circuit 121, the first transmission weight calculation circuit 143 reads channel information corresponding to the destination station from the channel information storage circuit 142, and calculates a transmission weight based on the read channel information. To do. The first transmission weight calculation circuit 143 outputs the calculated transmission weight to the transmission signal processing circuits 811-1 to 811-N _SDM . In normal communication, the terminal station device with which the terminal station device communicates is limited to a specific base station device. Therefore, in the above description, management related to the terminal station device as the destination is explicitly shown. Of course, it is also possible to carry out the processing as one.

  The feature of the present invention according to the first embodiment is that the transmission weight is calculated between the terminal station device 60 and the first transmission signal processing units 181-1 to 181-4 of the base station device 70. The virtual transmission path corresponding to the first singular value is used. There are several variations in the channel estimation method and the transmission / reception weight calculation method when the virtual transmission path corresponding to the first singular value is utilized, and a method for efficiently acquiring this will be described in detail later. To do. For example, the first right singularity when the singular value decomposition is performed on each channel matrix in the uplink from the terminal station device 60 to the first transmission signal processing units 181-1 to 181-4 of the base station device 70. A vector may be used as the transmission weight vector. In this case, the first transmission weight calculation circuit 143 has a function of calculating the first right singular vector.

  Or, when the base station device 70 side transmits a signal by multiplying each of the plurality of antenna elements of the first transmission signal processing units 181-1 to 181-4 by a predetermined transmission weight vector, actually, Despite being transmitted from a plurality of transmission antennas, each of the first transmission signal processing units 181-1 to 181-4 is effectively equivalent to that transmitted from one virtual antenna element. It is. For this reason, a channel information vector between the one virtual antenna element and each receiving antenna of the terminal station apparatus 60 is acquired, and the uplink is performed by an implicit feedback technique in which calibration processing is performed on the channel vector. It is also possible to acquire the channel information. Based on the uplink channel vector thus determined, the first transmission weight calculation circuit 143 obtains a vector obtained by taking the complex conjugate of this channel vector as shown in Equation (7), or each component of the vector. Any of the vectors whose absolute values are all constant may be used as the transmission weight vector.

  As an alternative to this, the first transmission weight calculation circuit 143 includes the terminal station device 60 and one antenna element in the first transmission signal processing units 181-1 to 181-4 of the base station device 70. The channel vector of the channel information on the receiving side is obtained between the two, and either the vector obtained by taking the complex conjugate of this channel vector as shown in the equation (7), or the vector in which the absolute values of each component of the vector are all constant May be used as a transmission weight vector. Further, the communication control circuit 121 manages control related to overall communication such as overall timing control.

FIG. 22 is a schematic block diagram illustrating an example of the configuration of the reception unit 65 in the terminal station device 60 according to the first embodiment of the present invention. As shown in FIG. 22, the receiving unit 65 includes antenna elements 851-1 to 851-N _MT-Ant , low noise amplifiers (LNA) 852-1 to 852-N _MT-Ant , a local oscillator 853, and a mixer 854. -1 to 854-N _MT-Ant , filters 855-1 to 855-N _MT-Ant , A / D (analog / digital) converters 856-1 to 856-N _MT-Ant , and FFT (Fast Fourier) Transform (fast Fourier transform) circuits 857-1 to 857-N _MT-Ant , reception signal processing circuits 145-1 to 145-N _SDM, and a first reception weight processing unit 144 are provided. Reception signal processing circuits 145-1 to 145-N _SDM and first reception weight processing unit 144 are connected to communication control circuit 121 shown in FIG. The first reception weight processing unit 144 includes a first channel information estimation circuit 146 and a first reception weight calculation circuit 147.

First, signals received by antenna elements _{851-1~851-N MT-Ant} is amplified by the low noise amplifier _{852-1~852-N MT-Ant.} The amplified signal and the local oscillation signal output from the local oscillator 853 are multiplied by the mixers 854-1 to 854-N _MT-Ant , and the amplified signal is down-converted from a radio frequency signal to a baseband signal. The Since the down-converted signal includes a signal outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855-N _MT-Ant . The signal from which the out-of-band component is removed is converted into a digital baseband signal by the A / D converters 856-1 to 856-N _MT-Ant . For example, in the case of using OFDM, the digital baseband signal is input to the FFT circuits 857-1 to 857 -N _MT-Ant , and the time axis at a predetermined symbol timing determined by the timing detection circuit omitted here. The upper signal is converted into a signal on the frequency axis (separated into signals of each subcarrier). The signals separated into the subcarriers are input to the reception signal processing circuits 145-1 to 145-N _SDM and also input to the first channel information estimation circuit 146.

In the first channel information estimation circuit 146, the first transmission signal of the base station apparatus 70 is based on a known signal for channel estimation separated into each subcarrier (such as a preamble signal added to the head of the radio packet). Channel information channel between the virtual antenna elements formed by the transmission weight vectors of the processing units 181-1 to 181-4 and the antenna elements 851-1 to 851-N _MT-Ant of the terminal station device 60 The vector is estimated for each subcarrier, and the estimation result is output to the first reception weight calculation circuit 147. The first reception weight calculation circuit 147 calculates the reception weight to be multiplied for each subcarrier based on the input channel information. For this reception weight, for example, as described above, a ZF type pseudo inverse matrix is used, or an MMSE type reception weight matrix is used. At this time, the reception weight vector for synthesizing the signals received by the antenna elements 851-1 to 851-N _MT-Ant differs for each signal sequence, and the above-described ZF type pseudo inverse matrix or MMSE type reception is performed. It corresponds to a row vector such as a weight matrix and is input to received signal processing circuits 145-1 to 145-N _SDM corresponding to the signal sequence to be extracted.

In the reception signal processing circuits 145-1 to 145-N _SDM , the signal for each subcarrier input from the FFT circuits 857-1 to 857-N _{MT-Ant is} input from the first reception weight calculation circuit 147. The reception weight is multiplied, and the signals received by the antenna elements 851-1 to 851-N _MT-Ant are added and synthesized for each subcarrier. The received signal processing circuits 145-1 to 145-N _SDM demodulate the added and combined signals and output the reproduced data to the MAC layer processing circuit 68.

Here, different received signal processing circuits 145-1 to 145-N _SDM perform signal processing of different signal sequences. Further, as received signal processing across a plurality of received signal processing circuits 145-1 to 145-N _SDM , MLD, simplified MLD using QR decomposition, or the like may be used. The MAC layer processing circuit 68 also performs processing related to the MAC layer (for example, conversion between data input / output to / from the interface circuit 67 and data transmitted / received on the wireless line, that is, a wireless packet, termination of header information of the MAC layer) Etc.). The reception data processed by the MAC layer processing circuit 68 is output to an external device or a network via the interface circuit 67. Further, the communication control circuit 121 manages control related to overall communication such as overall timing control.

  Here, similarly to the transmission side, the terminal station device 60 side at the time of reception can also perform signal processing that consciously uses the virtual transmission path corresponding to the first singular value. FIG. 23 illustrates an example of another configuration of the receiving unit 65 in the terminal station device 60 according to the first embodiment.

23, reference numeral 154 is a first reception weight processing unit, reference numeral 155 is a first reception signal processing circuit, reference numeral 156 is a first channel information estimation circuit, reference numeral 157 is a first reception weight calculation circuit, reference numeral 159 Indicates a second received signal processing circuit, and the others are the same as in FIG. In the above description, the first reception weight calculation circuit 157 has been described as calculating the reception weight for directly separating the signal sequence of the N _SDM system. However, the virtual circuit corresponding to the first singular value is temporarily used. It is also possible to remove residual interference between the remaining signal sequences in two stages while performing signal separation on the transmission path.

  In this case, in the first reception weight calculation circuit 157, for example, each antenna element of the first transmission signal processing units 181-1 to 181-4 of the base station apparatus 70 is directed to each antenna element of the terminal station apparatus 60. When the channel information can be acquired, the first left singular vector when the singular value decomposition is performed on the channel matrix having the channel information as a component is calculated as a reception weight vector. Or, when the first transmission signal processing units 181-1 to 181-4 use one virtual antenna element formed by multiplying the transmission weight vector, A channel vector between the corresponding virtual antenna element and each antenna element of the terminal station device 60 may be obtained, and each component of the reception weight vector may be obtained by Expression (7) based on this vector (maximum ratio combining). Or the absolute value of the value given by Equation (7) may be given constant (weight for equal gain synthesis).

The reception weight vector obtained in this way is input to each of the first reception signal processing circuits 155-1 to 155-N _SDM . However, since signal processing for separating residual interference is performed in the second reception signal processing circuit 159 in this subsequent stage, there is no need to frequently update the reception weight vector in real time. It is also possible to reuse a common reception weight vector in a time domain where channel information is expected not to fluctuate rapidly.

The first channel information estimation circuit 156 and the first reception weight calculation circuit 157 do not sequentially update the reception weight from such a viewpoint, but for example, the first channel information estimation circuit 156 sends a certain amount of channel estimation results. The channel estimation accuracy is improved by averaging, and the first reception weight calculation circuit 157 calculates a first reception weight vector at a predetermined period based on the averaged channel information. The reception signal processing circuits 155-1 to 155-N may be configured to be input to the _SDM . In this case, when averaging, the channel information may be considered as relative channel information based on the complex phase of the reference antenna (for example, the first antenna) (or each channel information divided by the channel information of the reference antenna). It is preferable to utilize (good).

In the first reception signal processing circuits 155-1 to 155-N _SDM , signals from the virtual transmission line corresponding to these first singular values are input to the second reception signal processing circuit 159. The function sharing between the first received signal processing circuits 155-1 to 155-N _SDM and the second received signal processing circuit 159 is the same as that of the first received signal processing unit 185 of the base station apparatus shown in FIG. This is similar to the relationship of the received signal processing unit 75. That is, a signal obtained by reducing a huge number of received signal signals corresponding to the number of antenna elements N _MT-Ant to the number of signal sequences N _SDM spatially multiplexed by the first received signal processing circuits 155-1 to 155-N _SDM . And is input to the second received signal processing circuit 159, and the second received signal processing circuit 159 performs general MIMO signal processing in a space with reduced dimensions.

Specifically, the second received signal processing circuit 159 obtains an N _SDM × N _SDM channel matrix for the signal sequence of the N _SDM system with reference to a known training signal given to the head of the received signal. The received signal detection process is performed based on the channel matrix. As described above, the second received signal processing circuit 159 multiplies a ZF-type inverse matrix or an MMSE-type linear reception weight matrix, or nonlinear such as MLD or simple MLD (QR-MLD, etc.). It is also possible to perform the signal processing. The second received signal processing circuit 159 performs demodulation processing on the signal of the N _SDM system that is signal-separated in this way, and outputs the reproduced data to the MAC layer processing circuit 68. This is equivalent to the signal processing of the second received signal processing unit 75 of the base station apparatus.

Here, an example of a device configuration in the second received signal processing unit 75 of the base station device 70 or the second received signal processing circuit 159 of the terminal station device 60 (basically, the processing is common to the base station device and the terminal station device). Is shown in FIG. The basic operation is as described above, and the N _SDM sequence signal sequence is converted into the second received signal processing circuit 190 (the first received signal processing circuit 190 in FIG. 23) as the received signal of the virtual transmission line corresponding to the N _SDM first singular values. 2 (corresponding to the reception signal processing circuit 159 of No. 2) (not shown here, the signal of each subcarrier is input and the same signal processing is performed), and then the channel matrix acquisition circuit 191 Then, a known training signal given to the head of the received signal is referred to, and an N _SDM × N _SDM channel matrix is obtained for the signal sequence of the N _SDM system. The reception weight matrix calculation circuit 192 calculates a reception weight matrix as a ZF-type inverse matrix or MMSE-type linear reception weight matrix based on the channel matrix, and inputs this to the reception weight matrix multiplication circuit 193. The reception weight matrix multiplication circuit 193 multiplies subsequent data by a reception weight matrix, and suppresses an interference signal that is a crosstalk component between different virtual transmission lines. The signal detection circuit 194 performs signal detection on each signal with improved SINR characteristics. The signal detection here is intended for general demodulation processing. For example, the received signal is softly determined, error correction is performed after deinterleaving, and final signal detection is performed. The data expanded into a plurality of signal series and transmitted in parallel is converted into one series data by parallel / serial conversion, and these are output to the MAC layer processing circuit. Although a linear signal processing example is shown here as a typical example, the signal detection circuit 194 can also perform nonlinear signal processing such as MLD or QR-MLD.

In addition, as the second transmission signal processing unit 71 of FIG. 17 is explicitly shown in the description of the configuration of the base station device 70, the terminal station device 60 side also shows the second transmission signal in FIG. Add the transmission signal processing circuit 148, the transmission signal processing circuit _{811-1~811-N SDM} replaces the first transmission signal processing circuit _{821-1~821-N SDM,} further first transmission weight calculating circuit 143 Can be replaced by the transmission weight calculation circuit 149. In this case, the second transmission signal processing circuit 148 shown in FIG. 21 has the function of the second transmission signal processing unit 71 in FIG. 17, and the second transmission signal processing circuit 148 generates a wireless packet to be transmitted through a wireless line. Then, a modulation weighting process is performed, and a transmission weight matrix for compensating for leakage of signals between virtual transmission lines corresponding to each first singular value is acquired from the transmission weight calculation circuit 149, and the N _SDM system The transmission signal vector is multiplied by a transmission weight matrix (transmission precoding) and output to the first transmission signal processing circuit 821. On the other hand, in the first transmission signal processing circuits 821-1 to 821-N _SDM , each of the transmission signals of the N _SDM system input from the second transmission signal processing circuit 148 has a virtual corresponding to the first singular value. The transmission weight for forming the transmission path is multiplied. Here, the function of the transmission weight calculation circuit 149 is to calculate a transmission weight for forming a virtual transmission line corresponding to the first singular value in the first transmission signal processing circuits 821-1 to 821-N _SDM . And a function for calculating a transmission weight matrix for compensating for leakage of signals between virtual transmission lines corresponding to each first singular value. In this case, the transmission weight matrix is obtained on the basis of channel information acquired by, for example, a method using implicit feedback with calibration processing or a method using direct explicit feedback.

  The feature of the present invention according to the first embodiment is that a virtual corresponding to the first singular value is between the terminal station device 60 and the first transmission signal processing units 181-1 to 181-4 of the base station device 70. Using a common transmission line. Therefore, for each channel matrix from the terminal station device 60 toward the first transmission signal processing units 181-1 to 181-4 of the base station device 70, the transmission weight is transmitted to the first right singular vector when the singular value decomposition is performed. A vector is used, and the first transmission weight calculation circuit 143 has a function of calculating the first right singular vector. As an approximate solution of the first right singular vector, details will be described later, but between the first antenna element in the first transmission signal processing units 181-1 to 181-4 of the base station device 70, Either a vector obtained by obtaining a channel vector and taking the complex conjugate of the channel vector or a vector in which the absolute values of the components of the vector are all constant may be used as the transmission weight vector. Originally, the antenna element group of the first transmission signal processing units 181-1 to 181-4 of the base station apparatus 70 forms a virtual directional antenna as a whole. This is equivalent to considering the case represented by one antenna physically present near the center in the antenna element group as an approximate solution. In this case, although the weight of the approximate solution is different from the weight of the exact solution, the gain obtained as a result of evaluation by simulation does not have an extremely large deterioration as will be described later.

The above is description of the structure of the terminal station apparatus 60, the transmission part 61, and the receiving part 65 in 1st Embodiment. What is important here is that the local oscillator 815 in the transmission unit 61 is shared by the mixers 816-1 to 816-N _MT-Ant in each antenna system of the transmission unit 61, and the local oscillator 853 in the reception unit 65 is the reception unit. It is a point shared by mixers 854-1 to 854-N _MT-Ant in each of the 65 antenna systems. In directivity control, the phase of a transmission / reception signal is adjusted for each antenna element. By making the phase relationship of signals input from each local oscillator 815 or local oscillator 853 always constant, It is possible to determine whether the transmission / reception weights should be multiplied in such a phase relationship. On the other hand, when the local oscillator 815 or the local oscillator 853 uses a plurality of asynchronous ones in the transmission unit 61 or the reception unit 65, directivity control for multiplying at least the transmission weight in the transmission unit 61 functions effectively. Disappear. Care should be taken in this regard when designing the device. Note that the local oscillator 815 and the local oscillator 853 can be shared.

(About extension to Point-to-Multipoint)
In the above description, the description has been focused on the point-to-point type wireless entrance line. Therefore, if the arrangement of the first signal processing unit 304 is operated in a state optimized by some technique, the first signal As many spatial multiplexing transmissions as the number of processing units 304 are possible. The optimization method for this may be the antenna arrangement method described later, or may be set by searching for an arrangement in which the virtual transmission lines are substantially orthogonal while setting the antenna at several locations. However, when one base station apparatus 70 and a plurality of terminal station apparatuses 60 perform point-to-multipoint communication at the same time, an ideal antenna arrangement on the base station apparatus 70 side is common to all terminal station apparatuses. Since it is difficult to make a random arrangement, a redundant number of first signal processing units 304 are installed, and communication is performed by selecting different combinations of first signal processing units 304 for each terminal station device 60 to communicate with. It does not matter. In this case, since the optimal combination of the first signal processing units 304 is determined by the communication control circuit, the optimal combination of the first signal processing units 304 for each terminal station device 60 is included in the communication control circuit. It is necessary to provide functions such as a database for information.

(Processing flow of signal processing in the present invention)
Hereinafter, a processing flow of signal processing in the first embodiment of the present invention will be described. Basically, the signal processing flows of the base station apparatus and the terminal station apparatus are the same, but the names and code numbers of the circuits to be cited are different. Further, as described above, the signal processing in the second transmission signal processing unit 71 (corresponding to the second transmission signal processing circuit 148 in FIG. 25 having an explicit description in the case of a terminal station apparatus) It is also possible not to perform weight matrix multiplication (transmission precoding). In this case, the signal processing portion can be omitted.

  FIG. 26 shows an outline of a signal processing flow at the time of signal transmission in the first embodiment. When the transmission process is started, the communication control circuit 120 instructs the first signal processing unit 304 and the second signal processing unit 305 to read out the first transmission weight vector and the second transmission weight matrix (step S1). S2601) (Step S2602).

  The reading of the transmission weight vector and the transmission weight matrix here is, for example, in the case of Point-to-Multipoint type communication and in the case of the base station apparatus 70, the terminal station apparatus 60 which is a communication partner station is different every time communication is performed. Therefore, a read process is performed every time transmission is performed. On the other hand, in the case of fixed point-to-point communication, or when the communication partner station is always fixed to the base station device 70 as in the terminal station device 60, the processing is omitted without reading each time. Is also possible. However, when the transmission variation of the channel cannot be ignored and the transmission weight is changed every time, the configuration may be such that the communication partner is read out every time even when the communication partner is fixed. Or it is good also as a structure which reads the transmission weight updated by the predetermined period. Further, when the matrix multiplication in the second transmission weight multiplication circuit in the second signal processing unit 305 is omitted as described above, it is not necessary to read out the transmission weight matrix in the second signal processing unit 305. .

The second signal processing unit 305 generates transmission signals for N _SDM systems that perform spatial multiplexing (step S2603), and multiplies the transmission signals by a second transmission weight matrix (step S2604). The transmission data for the N _SDM systems after multiplication is transferred to the corresponding first signal processing unit 304 (step S2605). The second signal processing unit 305 determines whether transmission of transmission data has been completed (step S2606). When transmission of transmission data has not ended (step S2606: NO), the second signal processing unit 305 returns the process to step S2603. When the transmission of the transmission data has ended (step S2606: YES), the second signal processing unit 305 ends the signal processing flow at the time of signal transmission.

On the other hand, the transmission signal (one-dimensional signal) is multiplied by the transmission weight vector using the transmission weight vector for each first signal processing unit 304 read in advance by the first signal processing unit 304 (step S2607). ), The number of transmission antennas is converted into a transmission signal vector of N _Ant dimension, and transmission signal processing is performed for each antenna system (step S2608).

  For example, when an OFDM modulation scheme is assumed, transmission signal generation is performed for each subcarrier and for each OFDM symbol, and transmission weight matrix multiplication and transmission weight vector multiplication are all individually performed for each subcarrier. Done. As the transmission signal processing here, IFFT processing is performed on the transmission signal for each subcarrier on the frequency axis, a guard interval is added, and waveform shaping between symbols is added as necessary. Sample data is D / A converted to generate an analog baseband signal, which is upconverted by a mixer, then out-of-band components are removed by a filter, and a signal amplified by a high power amplifier is transmitted from an antenna. .

  The above is an example in the case of using the OFDM modulation method, but the same applies to other methods such as SC-FDE, and basically various conventional communication methods can be applied. In addition, it is not always necessary to perform signal processing on the frequency axis. When performing transmission weight vector multiplication on the time axis as described later, sampling is performed using a single transmission weight vector (and matrix). A configuration in which transmission weight vectors (and matrices) are multiplied for each data may be used. Further, processing such as error correction coding and interleaving is included in the transmission signal generation processing, and is omitted in the description here.

  Next, FIG. 27 shows an outline of a first example of a signal processing flow at the time of signal reception in the first embodiment. When receiving the signal, the communication control circuit 120 instructs the first signal processing units 304 to read out the respective first reception weight vectors (step S2701).

  In the case of the point-to-multipoint communication, for example, in the case of the base station device 70, the reading of the reception weight vector here is the same as the case of the transmission side. Since they are different, the reading process is performed every time transmission is performed. On the other hand, in the case of fixed point-to-point communication, or when the communication partner station is always fixed to the base station device 70 as in the terminal station device 60, the processing is omitted without reading each time. It is also possible. However, when the time variation of the channel cannot be ignored and the reception weight is changed every time, the configuration may be such that the communication partner is read out every time even if the communication partner is fixed. Or it is good also as a structure which reads the receiving weight updated by the predetermined period.

  Thereafter, when an actual signal is received, signal reception processing is performed for each antenna system (step S2702). The signal reception processing here is, for example, amplifying the reception signal with a low noise amplifier, down-converting with a mixer, removing out-of-band frequency components with a filter, and then digital baseband with an A / D converter Convert to signal sample value. In these received signal processes, for example, in the case of the OFDM modulation method, the signal is cut out for each OFDM symbol, the guard interval is removed, and converted to a signal on the frequency axis for each subcarrier component by FFT.

  For the signal subjected to the above reception signal processing, the reception signal vector having the signal for each antenna element as a vector component is multiplied by the reception weight vector (step S2703), and this result is used as the second signal processing. The data is transferred to the unit 305 (step S2704). The first signal processing unit 304 determines whether or not reception of received data has ended (step S2705). If the received data continues further (step S2705: YES), the process returns to the received signal processing shown in step S2702, and the process continues. If the received data ends (step S2705: NO), The processing of the first signal processing unit 304 ends.

On the other hand, the second signal processing unit 305 determines whether or not the signal transferred from the first signal processing unit 304 is, for example, a training signal for channel estimation added to the head of the wireless packet (step S2706). If it is a training signal (step S2706: YES), channel estimation is performed (step S2707), and a second reception weight matrix is generated based on the obtained channel matrix (step S2708). On the other hand, if the data portion is not a training signal (step S2706: NO), the spatially multiplexed N _SDM- dimensional received signal vector is multiplied by the second reception weight matrix (step S2709), and after signal separation A signal detection process is performed (step S2710). Signal detection processing here means processing for estimating a transmission signal through, for example, soft decision processing, deinterleaving processing, error correction processing, etc. for the transmission signal, and after completion of signal processing of these physical layers, the MAC layer It is output to the processing circuit.

  In the received signal processing, general conventional techniques such as the SC-FDE method as well as the OFDM modulation method can be similarly applied. Further, in the description, the signal processing for each subcarrier has been described. However, the signal processing is not necessarily performed on the frequency axis, and the reception weight vector and the reception weight matrix are multiplied on the time axis as will be described later. In some cases, for example, a single reception weight vector and reception weight matrix may be used to multiply the reception weight vector and reception weight matrix for each sampling data. For example, the MIMO signal processing corresponding to step S2709 for multiplying the second reception weight matrix (and step S2710 for performing signal detection processing after signal separation) is not necessarily multiplied by the second reception weight matrix as linear signal processing. There is no need to do this, and it is also possible to apply a general MIMO signal detection process such as QR-MLD or MLD.

Note that the above description assumes that the local oscillator 853 used in the first signal processing unit 304 is asynchronous for each first signal processing unit 304 as in the base station apparatus 70, and the second reception weight matrix is a signal. In the case of the terminal station device 60, for example, the local oscillator 853 used for the up-conversion processing and / or down-conversion processing of all antenna elements is shared, so that reception is not necessarily performed. The second reception weight matrix does not change every time. As described above, when the local oscillator 853 is shared as a whole and the time variation of the channel can be ignored, the second reception weight matrix acquired in the past can be used. FIG. 28 shows an outline of a second example of the signal processing flow at the time of signal reception in the first embodiment of the present invention. Steps S2801 to S2805 shown in FIG. 28 are the same as steps S2701 to S2705 shown in FIG. The difference from FIG. 27 is that in the signal processing of the second signal processing unit 305, the second signal processing unit 305 receives the second reception weight matrix from the communication control circuit 120 when the signal reception processing is started. In response to the read instruction, the second reception weight matrix is read (step S2806). After that, when the received signal is transferred from the first signal processing unit 304, the second received weight matrix is multiplied by the spatially multiplexed N _SDM- dimensional received signal vector (step S2807), and then signal detection is performed. Processing is performed (step S2808). Other signal processing is the same as in FIG.

  As described above, the wireless communication system 50 according to the first embodiment includes the base station device 70 (first wireless station device) and the terminal station device 60 (second wireless station device). The base station device 70 includes a plurality of first transmission signal processing units 181, a plurality of first reception signal processing units 185, a second transmission signal processing unit 71, and a second reception signal processing unit 75. . The plurality of first transmission signal processing units 181 includes a plurality of antenna elements 819. The plurality of first reception signal processing units 185 includes a plurality of antenna elements 851. The second transmission signal processing unit 71 executes radio communication signal processing associated with the plurality of first transmission signal processing units 181. The second received signal processing unit 75 executes signal processing for wireless communication associated with the plurality of first received signal processing units 185. The terminal station device 60 includes a plurality of antenna elements 819, a plurality of antenna elements 851, a transmission unit 61, and a reception unit 65. The transmission unit 61 performs wireless communication with the plurality of first reception signal processing units 185 via the plurality of antenna elements 819. The transmission unit 61 wirelessly communicates with a plurality of first reception signal processing units 185 and a second reception signal processing unit 75 associated with the first reception signal processing unit 185 via a plurality of antenna elements 819. Communication may be performed. The receiving unit 65 performs wireless communication with the plurality of first transmission signal processing units 181 via the plurality of antenna elements 851. The receiving unit 65 wirelessly communicates with the plurality of first transmission signal processing units 181 and the second transmission signal processing unit 71 associated with the first transmission signal processing unit 181 via the plurality of antenna elements 851. Communication may be performed. The first transmission signal processing unit 181 and the first reception signal processing unit 185 are a transmission weight vector and a reception weight for a MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group. At least one of the vectors is at least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix, or an approximate solution of the first right singular vector and an approximate solution of the first left singular vector. It calculates based on at least one. The plurality of first transmission signal processing units 181 and first reception signal processing units 185 include the transmission weight vector and the reception weight vector corresponding to the first transmission signal processing unit 181 or the first reception signal processing unit 185, respectively. At least one of them is used to spatially transmit independent signal sequences.

  Accordingly, the base station device 70, the terminal station device 60, the wireless communication system 50, and the wireless communication method according to the first embodiment can increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. For example, the base station device 70, the terminal station device 60, the radio communication system 50, and the radio communication method use high-order spatial multiplexing and a high frequency band while the line-of-sight environment is dominant in the radio communication system in the future mobile network. As a result, a large capacity can be realized.

  The base station device 70 as the base station device 303 of the first embodiment includes a first signal processing unit 304 in which antenna elements are bundled in a narrow area, and a second signal processing unit 305 that aggregates them. The first signal processing unit 304 performs closed beam forming in each signal processing unit, and does not perform beam forming across different first signal processing units 304. The individual first signal processing unit 304 generates transmission / reception weights for the “virtual transmission path corresponding to the first singular value” that makes the best use of the “line-of-sight wave”, and performs a plurality of first signal processing Spatial multiplexing transmission is performed using individual transmission / reception weights in the unit 304.

  Here, in the above description, it has been described as a typical embodiment that the sight environment is the dominant environment. However, for example, even when the sight environment is not perfect, a very strong reflected wave arrives from a specific direction. When a diffracted wave that is not inferior to a line-of-sight wave arrives, a virtual transmission line corresponding to the first singular value is formed in the arrival direction. In this sense, the virtual transmission line corresponding to the first singular value is not necessarily a transmission line constituted by line-of-sight waves, and in the first embodiment of the present invention, the virtual transmission line corresponding to the first singular value is used. The first embodiment of the present invention can be applied even in a non-line-of-sight environment because it is a spatial multiplexing transmission that actively utilizes a plurality of transmission paths. The circuit configuration and processing flow in that case can be applied as they are without changing from the above description.

[Second Embodiment]
[Antenna placement conditions for orthogonalization of line-of-sight MIMO transmission]
(Outline of the basic principle according to the second embodiment)
Here, the conditions for a plurality of virtual transmission paths to be approximately orthogonal are arranged. In FIG. 16, since one directional beam is effectively formed for every 25 antenna elements, this is approximately one antenna having a very high directivity gain (for example, a parabolic antenna). Is equivalent to using. Therefore, FIG. 29 shows an example of a case where four parabolic antennas are mounted on the base station apparatus and 16 antenna elements are mounted on the terminal station apparatus in a linear array shape. 29, reference numeral 306 indicates a base station apparatus, reference numeral 302 indicates a terminal station apparatus, and reference numerals 307-1 to 307-4 indicate parabolic antennas. The terminal station device 302 corresponds to the terminal station device 60. Base station apparatus 306 is equivalent to base station apparatus 70 equivalently. Basically, transmission equivalent to the parabolic antennas 307-1 to 307-4 shown in FIG. 29 is realized by grouping a large number of antenna elements of the base station apparatus 70.

  Here, the conditions under which the four singular values of the MIMO channel in FIG. 29 in which the line-of-sight wave is dominant are substantially equally large as in the case described above are arranged. For example, a patent document (Japanese Patent No. 5488894) stipulates a condition for establishing MIMO transmission in a line-of-sight environment. The basic idea of these prior arts is an antenna element provided in a base station device and a terminal station device. As a result, the distance between individual transmitting antennas and receiving antennas varies, and the relationship between the distances becomes a pseudo-random relationship (or a predetermined relationship). Attempt to adjust to.

  On the other hand, for example, in the phased array antenna technology, if the distance between the antenna elements of a one-dimensional linear array is d, the value of d is set to be small, so that it can be viewed from an antenna far away in the direction in which directivity should be directed. The radio wave that arrives (or is sent out) at each antenna element of the linear array can be approximated to a plane wave, and if the direction of the incoming wave is expressed by an angle θ, the path difference for each antenna element is high at d · sinθ. It becomes possible to approximate the accuracy. However, in the above-described basic concept of the prior art, in order to set the distance between the antenna elements as wide as possible, such a plane wave approximation cannot be performed, and the approximation needs to be performed under different conditions. It was. As a result, the accuracy of approximation in the prior art is reduced. In fact, when performing MIMO communication in a line-of-sight environment, a large parabolic antenna was assumed at both the transmitting and receiving stations. Usually, the size of the parabolic antenna is much larger than the wavelength, so that the path length difference as described above cannot be approximated to a plane wave with a negligible accuracy with respect to the wavelength.

However, while increasing the effective aperture length of the antenna only on the base station device 70 side, the distance between the antenna elements on the terminal station device 60 side is set to about several wavelengths (the distance between the base station device 70 and the terminal station device 60). If it is set to the order of 3 wavelengths or less, for example, the path difference for each antenna element on the terminal station apparatus 302 side from the parabolic antenna on the base station apparatus 306 side in FIG. It can be approximated. Therefore, as shown in FIG. 30, base station apparatus antenna #j (310-j) (1 ≦ j ≦ M) of base station apparatus 70 corresponding to base station apparatus 306 and terminal station corresponding to terminal station apparatus 302 If the distance between the device 60 and the terminal station antenna # 1 (320-1) is L _j and the angle difference from the front direction of the terminal station antenna # 1 (320-1) is θ _j , A channel matrix (channel information between the j-th antenna on the base station device 70 side and the i-th (1 ≦ i ≦ N _MT-Ant ) antenna of the terminal station device 60 using the antenna interval d of the terminal station device 60 is expressed as h _ij. Can be approximately expressed by the following equation (26). In Expression (26), N _MT-Ant is expressed as N in order to avoid complication. In FIG. 29, the number of parabolic antennas 307-1 to 307-4 included in the base station apparatus 306 is four, but here the number of antenna elements is M as a general number.

Here, although displayed in the form of a matrix product, the (j, j) component of the second item indicates channel information between the j-th antenna of the base station device 70 and the first antenna of the terminal station device 60. The difference information of each component of the N _MT-Ant antennas of the terminal station device 60 corresponds to the column vector of the j-th column of the first item. Since the (j, j) component of the second item is a common term applied to all components of the column vector, each singular value is stable and large if each column vector of the first item is orthogonal. Become. The condition that the i-th column vector and the j-th column vector in the first term are orthogonal is expressed by the following equation (27) that the inner product of the complex vectors is zero.

In the above formula modification, the formula of the sum of geometric series is used. Since the last condition is that the denominator is not zero and the numerator is zero, from the condition that the term in parentheses of exp of the numerator is an integral multiple of 2πi, Is guided. In the following equations, N is described as the original number of antennas N _MT-Ant .

The constant K _{1 in the} equation (28) is an integer that is not a multiple of the number of antennas N _MT-Ant of the terminal station device 60. Here, as shown in FIG. 30, it is assumed that the j-th antenna on the base station device 70 side has a displacement of _{dj in} the horizontal direction from the front of the antenna on the terminal station device 60 side, and the difference between the i-th and j-th antennas is Δd _ij And The distance between the base station apparatus 70 and the terminal station apparatus 60 is L, and both sides of the first expression of Expression (28) are multiplied by L / d. Although the distance between each antenna element of the base station apparatus 70 and the first antenna of the terminal station apparatus 60 is slightly different, if the distance L is sufficiently larger than the displacement d _j , it can be approximated as L≈L _j .

That is, with _respect to the number of antennas N _MT-Ant of the terminal station device 60, the antenna element interval d of the terminal station device 60, the terminal station device 60 and the base for an integer K ₂ that is neither an integer multiple of N _MT-Ant nor zero. What is necessary is just to set so that the space | interval of arbitrary two antenna elements may become the space | interval which satisfy | fills Formula (29) with respect to the distance L of the station apparatus 70, and the wavelength (lambda) of the signal wave of radio | wireless communication. If this condition is interpreted geometrically, this condition can be easily realized, for example, under the following conditions. For example, the simplest condition is that an arbitrary offset is allowed in the linear direction on a straight line whose distance is L from the linear array on the terminal station device 60 side, and N _{MT− at an} interval of λL / N _MT-Ant d. _The point of the _Ant point is set linearly, the M point is selected from the N _MT-Ant points, and the M antenna elements of the base station device 70 are arranged. This is because, when two arbitrary integers are selected from consecutive N _MT-Ant integers and the difference between them is obtained, the absolute value is always less than (N _MT-Ant −1), and K ₂ is N _{MT -Utilizes} the fact that it is possible to avoid an integral multiple of _Ant . It is also possible to select the antenna installation location of the base station apparatus from other broader conditions by searching and setting an arrangement in which the difference between any two individual integers is not an integer multiple of N _MT-Ant. It is.

As shown in FIG. 29, when the antenna element on the terminal station device 60 side is downsized, the number of antenna elements N _MT-Ant on the terminal station device 60 side is increased correspondingly, while generally the antenna of the base station device 70 is increased. Since the number of elements M is set to the upper limit of the assumed number of spatial multiplexing, it is assumed that the value of M is smaller than the number of antenna elements N _MT-Ant on the terminal station device 60 side. That is, when N _MT-Ant > M, the M antennas on the base station device 70 side do not have to be evenly spaced even under the simplest conditions described above, and are intermittent from the above N _MT-Ant points. The arrangement location of the antenna element may be set as follows. In the case of FIG. 16, the actual number of antenna elements is 100. However, M in FIG. Should be considered. Although the approximation with very high accuracy is performed until the derivation of the equation (28), the approximation used in the equation (29) has a slightly low accuracy. However, depending on the parameter setting, if the distance between the antenna elements of the base station apparatus 70 in the equation (29) is about 1 m, even if there is an error of several percent of about ± several centimeters, it is almost orthogonal. There is no difference. Although the same applies to the first embodiment of the present invention, basically, even when there is an interference component between a plurality of virtual transmission paths on the receiving side, it is possible to suppress the interference by signal processing on the receiving side. If such signal processing is assumed, loss can be minimized by making the state almost orthogonal to the last. Therefore, the approximation accuracy of Expression (29) does not have a great influence on the system operation.

  Considering such characteristics, if each grouped antenna element group shown in FIG. 16 is regarded as one virtual antenna, the center point of a multi-element antenna having a physical extension is assumed to be virtual. If a plurality of antenna element groups are installed in the antenna arrangement described above, a weight vector represented by the first singular vector is used, and a plurality of signal sequences are obtained with desired characteristics. It becomes possible to perform spatial multiplexing and transmission.

In the above description, the antenna element on the base station apparatus side and the antenna element on the terminal apparatus side have been described taking an example where they face each other and face each other as shown in FIG. 29. As shown in FIG. 31, the case where there is no direct relationship is assumed. This is due to restrictions on installation locations of the base station device 306 and the terminal station device 302. In such a case, a desired effect can be derived by slightly converting each parameter. In FIG. 31, when the terminal station device 302 is viewed from the base station device 306, the terminal station device 302 exists in a direction deviated from the front by an angle θ, and when the base station device 306 is viewed from the terminal station device 302. The base station apparatus 306 exists in a direction that is shifted from the front by an angle θ ′. In this case, the distance D ₁ between the parabolic antennas 307-1 and 307-4 of the base station device 306 appears equivalently narrowed to D ₁ ′ (= D ₁ cos θ). Similarly, the width D ₂ of the linear array 312 of the terminal station apparatus 302 appears equivalently narrowed to D ₂ ′ (= D ₂ cos θ ′). In other words, it can be considered that the antenna element interval d is converted to dcosθ × cosθ ′ times. If the distance between the approximate center of gravity of the parabolic antennas 307-1 to 307-4 of the base station device 306 and the terminal station device 302 is L, these antennas are converted into an optimal antenna arrangement using the above conditional expressions. Can be calculated. That is, the base station device 70 virtually projects each linear array on an axis orthogonal to the straight line in the line-of-sight direction. Similarly, the terminal station device 60 virtually projects each linear array on an axis orthogonal to the straight line in the line-of-sight direction. Based on the distance between the virtually projected antenna elements on the axis, d in Expression (29) is converted to d cos θ × cos θ ′. Based on the distance between the virtually projected antenna elements on the axis, d in Expression (29) is converted by a conversion device or a person.

  Incidentally, in FIG. 29 and FIG. 31, the distance between the center of gravity of the parabolic antennas 307-1 to 307-4 of the base station apparatus 306 and the distance between the terminal station apparatus 302 is L, but the exact distance between the individual antenna elements is the same. Not the distance L but each has an error. However, as can be seen from the fact that approximation is used in the middle of the calculation of equation (29), the distance L here can be allowed to have a certain degree of error, and the antenna directly facing as shown in FIG. The distance between the elements (the distance between two straight lines of the antenna elements arranged on a parallel straight line) may be used, or as shown in FIG. As described above, the distance L may be handled as an approximate value or an approximate value.

  In the description in FIGS. 29 and 30 in this description, a case where the terminal station apparatus is configured by a linear array is shown as a typical example. This feature is that the parabolic antennas 307-1 to 307-4 or 310-1 to 310-M (corresponding to the antenna element groups 304-1 to 304-4 in FIG. 16) on the base station apparatus side are arranged on a straight line. Similarly, the antenna elements on the terminal station side are also arranged in a linear array on a straight line. In the above calculation of the orthogonalization condition, the two straight lines are described as being in a parallel relationship. Specifically, for example, a horizontal axis parallel to the wall surface of the building is assumed on the wall surface of the building or the roof of the building, and a plurality of first signal processing units and antenna elements (groups) 307 are arranged along the horizontal axis. When installing -1 to 307-4, it is preferable that the linear array of the terminal station apparatus is also arranged on a horizontal axis parallel to the wall surface of the opposite building across the road. Here, a vertical axis orthogonal to the horizontal axis on the terminal station apparatus side is set, a grid parallel to the horizontal axis and the vertical axis is assumed, and the linear array of the terminal station apparatus described above is set to N in the vertical direction. 'Consider the case of using a stacked square lattice array. At this time, the total number of antenna elements of the terminal station apparatus is N × N ′ elements. However, when the antenna elements (groups) 307-1 to 307-4 of the plurality of first signal processing units of the base station apparatus are viewed from this terminal station apparatus, there is an angle difference with respect to the horizontal direction. Since there is no difference between the antenna elements (groups) 307-1 to 307-4 with respect to the elevation angle in the direction, each of the N ′ stages of the above-described square lattice array is not regarded as an independent antenna element, and is substantially vertical. It is understood that the N ′ elements arranged in the direction behave equivalently as one virtual antenna element, and this virtual antenna element is arranged in a linear array on the horizontal axis. In fact, this effect has been confirmed also in the simulation evaluation. The antenna element of the terminal station apparatus is parallel to this axis with respect to the axis connecting the antenna elements (groups) 307-1 to 307-4 on the base station apparatus side. In the case where N × N ′ elements are arranged in a lattice shape composed of an axis and an axis orthogonal to this axis, the base station apparatus side antenna elements (groups) 307-1 to 307-4 are parallel to the axis connecting them. If it is regarded as a linear array of N elements on the axis and the element spacing of the N elements, the element spacing of the antenna elements (groups) 307-1 to 307-4 on the base station apparatus side is optimized by applying Equation (18). good. Of course, when the antenna elements (groups) 307-1 to 307-4 on the base station apparatus side are vertically aligned with the wall surface of the building, the antenna elements on the terminal station apparatus side are also N elements in the vertical direction. The N ′ elements are arranged in a grid pattern in the horizontal direction, and this is regarded as an equivalent N element linear array arranged in the vertical direction, and the equation (18) may be applied. Although the antenna arrangement on the terminal station apparatus side has been described as a square lattice array here, it is not necessarily a square lattice, and the element spacing in the horizontal direction and the vertical direction may be different in a rectangular shape. In this case, the interval between the antenna elements arranged on the axis on the terminal station apparatus side parallel to the axis on which the antenna elements (groups) 307-1 to 307-4 on the base station apparatus side are aligned is the antenna element of formula (18) What is necessary is just to calculate as the space | interval d.

As described above, the wireless communication system 53 of the second embodiment includes the base station device 306 (first wireless station device) and the terminal station device 302 (second wireless station device). The base station apparatus 306 includes antenna elements (parabolic antennas) 307-1 to 307-4, and the terminal station apparatus 302 includes a terminal station apparatus antenna element group 312. The plurality of parabolic antennas (antenna elements) 307-1 to 307-4 of the base station apparatus 306, as shown in the equation (29), the distance Δd _mn between the m-th element and the n-th element, and the terminal station apparatus 302 based on the distance L between the base station apparatus 306, the wavelength λ of the radio communication signal wave, the number of antennas N _MT-Ant of the terminal station apparatus 302, and the distance d between the antenna elements of the terminal station apparatus 302. The Each of the parabolic antennas 307-1 to 307-4 is, for example, a single high gain antenna element (single antenna element). The high gain antenna element is a parabolic antenna 307, for example. Parabolic antennas 307-1 to 307-4 may be a group of antenna elements instead of a single high gain antenna element. When the antenna element group is used in this way, the base station device 306 may be the base station device 303 shown in FIG. 16 and the base station device 17 shown in FIG. 17 includes a plurality of first transmission signal processing units 181, a plurality of first reception signal processing units 185, a second transmission signal processing unit 71, and a second reception signal processing unit 75. And have. The plurality of first transmission signal processing units 181 includes a plurality of antenna elements 819. The plurality of first reception signal processing units 185 includes a plurality of antenna elements 851. The second transmission signal processing unit 71 executes radio communication signal processing associated with the plurality of first transmission signal processing units 181. The second received signal processing unit 75 executes signal processing for wireless communication associated with the plurality of first received signal processing units 185. The terminal station device 60 includes a plurality of antenna elements 819, a plurality of antenna elements 851, a transmission unit 61, and a reception unit 65. The transmission unit 61 performs wireless communication with the plurality of first reception signal processing units 185 via the plurality of antenna elements 819. The receiving unit 65 performs wireless communication with the plurality of first transmission signal processing units 181 via the plurality of antenna elements 851.

  That is, the interval between the antenna elements or the interval between the antenna element groups depends on the distance L between the base station apparatus 306 (first radio station apparatus) and the terminal station apparatus 302 (second radio station apparatus), and the signal wave of radio communication. , A number N of linear array antenna elements constituting the second antenna element group, or a number N of longitudinal or lateral antenna elements arranged in a grid, and the second antenna elements Each single antenna element constituting the plurality of first antenna groups or an integer multiple of a value calculated based on the distance d between the antenna elements in the longitudinal direction or the transverse direction constituting the group Antenna element groups are arranged.

  Accordingly, the base station device 306 or 70, the terminal station device 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method according to the second embodiment have a transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. It can be increased. The base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the radio communication system 53 or 50, and the antenna element arrangement method according to the second embodiment transmit a plurality of signal sequences in a spatially multiplexed manner with desired characteristics. Is possible. The base station device 306 or 70, the terminal station device 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method according to the second embodiment can improve the SIR characteristics. The base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the radio communication system 53 or 50, and the antenna element arrangement method according to the second embodiment can realize higher spatial multiplexing.

  The base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method according to the second embodiment can In the MIMO channels of the plurality of parabolic antennas 307 and the plurality of element antennas on the terminal station device 302 or 60 side, the antenna installation conditions for orthogonalizing each channel are defined. However, here, in order to simplify the discussion, the conditions for mounting a parabolic antenna on the base station side are shown. Even when the first signal processing unit as shown in FIG. 16 is used, orthogonalization between the transmission paths can be generally achieved under the same conditions. As a result, it is possible to improve the communication quality on the transmission path and increase the transmission capacity.

[Third Embodiment]
[Technology applied to the access system of a plurality of virtual transmission lines corresponding to the first singular value]
(Basic principle according to the third embodiment)
In the above description, the environment in which the line-of-sight wave is dominant in the transmission path between the base station apparatus and the terminal station apparatus has been taken up. As an example of such use, a case where the terminal station device 60 is regarded as a relay station and the above-described technology is used in an entrance line from the base station device 70 to the relay station can be considered. However, the virtual transmission line corresponding to the first singular value is not limited to use in the wireless entrance line. Also in the above description, the terminal station device 60 corresponding to the relay station is assumed to be small, and it is assumed that many small antennas are arranged in a very narrow area. Therefore, for example, if a large number of antennas are mounted on a smartphone or the like, and especially reception in a small cell of the fifth generation mobile communication system (5G) is assumed, use of millimeter waves or quasi-millimeter waves is assumed. Except for the user moving, it can be regarded as an environment where the antennas are condensed and arranged in a narrow space just like the case of the above-described wireless entrance. That is, as an example, when a small cell base station is arranged at a relatively high place such as a wall of a building, the base station device 70 and the terminal station device 60 are generally in a line-of-sight environment from directly above, and slightly in the surface direction of the smartphone. Considering the situation of mounting an antenna having a directivity gain, it is expected that communication will be performed in an environment where the line-of-sight wave is dominant. In such a case, it is effective to group the antennas of the base station apparatus into a plurality as in FIG. 16, and to actively use the virtual transmission path corresponding to the first singular value in each group as well. Become.

  Here, considering the difference between the case of FIG. 16 and the case of the access system, in FIG. 16, the straight line where the antenna elements on the base station apparatus 70 are aligned and the straight line where the antennas on the terminal station apparatus 60 are aligned are almost parallel. However, in general, the three-dimensional positional relationship of each antenna is determined by the combination of the antenna arrangement of the smartphone, the angle at which the user holds the smartphone, the direction in which the user is facing, etc. It is expected that the relationship is not parallel but twisted. In this case, there is generally no optimum value as shown in the equation (29), but for example, a grouped antenna element group is provided with a little redundancy, and an optimum antenna of a combination having a low correlation in selective diversity. What is necessary is just to select and use an element group. However, in this case, it is necessary to set the interval between the antenna element groups slightly larger so that the correlation of the channel can be reduced as much as possible. In that case, it may be effective not to arrange all antenna element groups only on the wall surface of the same structure (building or the like) but to install the antennas across a plurality of structures. In such a case, in FIG. 16, it is difficult to connect the second signal processing unit 305 and the first signal processing units 304-1 to 304-4 with a wired cable (including an optical fiber).

  Therefore, the antenna element group (first signal processing unit 304) of the base station device 303 is installed as a satellite base station, and the general base station device corresponding to the second signal processing unit 305 of the base station device 303 It is effective to realize transmission / reception of signals to / from the satellite base station apparatus through a wireless line. FIG. 32 shows a third embodiment of the present invention. In FIG. 32, reference numeral 31 represents a general base station apparatus, reference numerals 32-1 to 32-3 represent satellite base station apparatuses, and reference numeral 33 represents a terminal station apparatus. The terminal station device 33 corresponds to the terminal station device 60. Reference numeral 37 represents a structure. Reference numeral 38 represents a structure. The general base station apparatus 31 is disposed on the upper surface (the roof of a building or the like) of the structure 37, for example. The satellite base station device 32 is disposed on, for example, a side surface (wall surface of a building or the like) of the structure 38. The terminal station device 33 is arranged in a cell 39 between the structure 37 and the structure 38, for example.

  The overall base station apparatus 31 and satellite base station apparatuses 32-1 to 32-3 realize the functions of the base station apparatus 70 described above as a whole. The overall base station apparatus 31 includes a large number of antenna elements, and constitutes a small antenna as a whole so that the distance between the antenna elements is approximately the same as the wavelength (for example, between several wavelengths to ½ wavelength). On the other hand, each of the satellite base station apparatuses 32 is a small size in which a large number of antenna elements are arranged at intervals of about the same wavelength (for example, between several wavelengths to ½ wavelength), like the first signal processing unit 304 of FIG. Equipped with an antenna. Communication is performed between the overall base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 using a first singular vector corresponding to the first singular value of each antenna. Further, the terminal station device 33 includes a plurality of antennas. For example, assuming a smartphone or the like, a plurality of antennas are mounted in a relatively narrow area. Here, communication is also performed between the terminal station device 33 and the satellite base station devices 32-1 to 32-3 using the first singular vector corresponding to the first singular value of each antenna as a transmission / reception weight vector. Furthermore, communication is also performed between the satellite base station apparatuses 32-1 to 32-3 and the central base station apparatus 31 by using the first singular vector corresponding to the first singular value of each antenna as a transmission / reception weight vector. This makes it possible to achieve stable MIMO transmission with a small channel correlation by utilizing a plurality of different satellite base station apparatuses 32-1 to 32-3 while utilizing a transmission / reception weight vector that maximizes a line-of-sight gain. It becomes possible.

  Next, FIG. 33 shows another example of the third embodiment of the present invention. In FIG. 33, reference numerals 34-1 to 34-2 represent central base station apparatuses, reference numerals 35-1 to 35-6 represent satellite base station apparatuses, and reference numerals 36-1 to 36-2 represent terminal station apparatuses. FIG. 33 is the same as FIG. 32 described above, but a number of satellite base station devices 35-1 to 35-6 arranged on the side surface (wall surface of the building) of the structure 38 and the upper surface (the roof of the building) of the structure 37. A difference from FIG. 32 is that the correspondence between the integrated base station devices 34-1 to 34-2 arranged in FIG. The satellite base station devices 35-1 to 35-6 can provide an access link in a flexible combination with a plurality of central base station devices 34-1 to 34-2. For example, when viewed from the general base station apparatus 34-1 close to the terminal station apparatus 36-1, the nearest satellite base station apparatuses are the satellite base station apparatuses 35-1 to 35-3. However, if the satellite base station devices having a small correlation from the terminal station device 36-1 are the satellite base station devices 35-1, 35-2, and 35-4, the general base station device 34-1 is the next general base station. It is also possible to provide communication using the virtual transmission path corresponding to the first singular value to the terminal station device 36-1 by utilizing the satellite base station device 35-4 which is close to the station device 34-2. become.

  Incidentally, one of the reasons for utilizing the virtual transmission path corresponding to the first singular value in such an access system is that the channel information of the virtual transmission path corresponding to the first singular value is the other higher order. Compared to the channel information of the virtual transmission path corresponding to the singular value, the time variation is relatively small in comparison. That is, the first right (left) singular vector when the singular value decomposition is performed represents a path corresponding to the line-of-sight wave, and there is no large variation in transmission / reception weight to be applied in a slight movement. However, since the higher-order right (left) singular vector corresponds to a path that contains a large amount of reflected wave components, the channel state after synthesis changes significantly due to slight movement of the terminal, resulting in the right (left) singular vector. Will vary greatly. In other words, if the channel information changes greatly, the transmission / reception weight vector for securing the gain also fluctuates. Unavoidable. In particular, if the terminal station device 36 is a small portable terminal such as a smartphone, the channel correlation between antennas is large, and it may be difficult to increase the capacity with single user MIMO. In such a case, as shown in an embodiment described later, it is effective to increase the overall total capacity by multi-user MIMO that simultaneously accommodates a large number of terminals having a small transmission capacity. Mutual interference, which deteriorates the SINR characteristic, leading to a reduction in transmission capacity and instability of communication. However, by actively using the virtual transmission line corresponding to the first singular value with relatively small channel time fluctuation, even when multi-user MIMO is used, the mutual interference between terminals is greatly increased. A secondary effect that it can be reduced can also be obtained.

 The multi-user MIMO generally means that one base station apparatus performs spatial multiplexing transmission with a plurality of terminal station apparatuses, but here, as shown in FIG. 32, a plurality of general base station apparatuses. It can be understood including cases where 34-1 to 34-2 provide services to a plurality of terminal station devices 36-1 to 36-2 simultaneously. In this case, each central base station apparatus and terminal station apparatus may actually be one-to-one communication, but satellite base station apparatuses 35-1 to 35-35 that use the same frequency channel for communication and further utilize them. 6 is also operated while mutually using the resources of the satellite base station devices 35-1 to 35-6 so as to allow some overlap, a certain satellite base station device 32 is simultaneously connected to a plurality of terminal station devices 36. In this case, it is possible to consider such an operation form as a multi-user MIMO in a broad sense, and even in such a case, the effect of reducing the influence of the above-mentioned channel time variation works effectively. Become.

(Outline of non-regenerative re-beamforming relay in satellite base station equipment)
A basic operation when relay transmission is performed via a plurality of satellite base station apparatuses 32 using the virtual transmission path corresponding to the above-described first singular value will be described with reference to the drawings. In the configuration using the satellite base station apparatus 32 shown in FIG. 32, the second signal processing unit 305 and the first signal processing units 304-1 to 304- of the base station apparatus 303 shown in FIG. 4 corresponds to the function of the base station apparatus as a whole configured by the central base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 in FIG. 32. In FIG. And the first signal processing units 304-1 to 304-4 are connected by wire, whereas in FIG. 32, the central base station device 31 and the satellite base station devices 32-1 to 32-3 This corresponds to the configuration in which the connection is made wirelessly. In FIG. 16, the signal transmission between the second signal processing unit 305 and the first signal processing units 304-1 to 304-4 ensures that there is no mutual signal interference because the transmission medium is wired. Therefore, in FIG. 32, when the wired part is completely rewired, each satellite base station device 32 is also used in the wireless signal between the central base station device 31 and the satellite base station devices 32-1 to 32-3. The situation where there is no signal interference between -1 and 32-3 will be ensured. However, in the third embodiment, signal processing for signal separation is performed in the terminal station device 33 (in the case of the downlink) or the overall base station device 31 (in the case of the uplink). The point in the present embodiment is that it is not necessary to completely separate the signals in the portions. However, if signal transmission / reception is performed between the general base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 with omnidirectionality, a line gain corresponding to the first singular value cannot be obtained. Between the base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3, transmission / reception signal processing is performed that uses the first singular vector for utilizing the virtual transmission path corresponding to the first singular value as the transmission / reception weight. .

FIG. 34 shows signal processing between the central base station device 31 and the satellite base station device 32, and signal processing between the satellite base station device 32 and the terminal station device 33 in the third embodiment of the present invention. The outline of is shown. 34, reference numeral 31 denotes a general base station apparatus, reference numeral 32 denotes a satellite base station apparatus, reference numeral 33 denotes a terminal station apparatus, and reference numeral 52 denotes a wireless communication system, each corresponding to the same number in FIG. There are a plurality of satellite base station apparatuses 32, and a plurality of satellite base station apparatuses 32 are shown in the center of FIG. In Figure 34, the general base station apparatus 31 and the satellite base station apparatus 32 performs communication using the frequency f _1, the satellite base station apparatus 32 and the terminal station is communicating at the frequency f _2. However, assuming communication by FDD (Frequency Division Duplex), the general base station apparatus 31 and the satellite base station apparatus 32 communicate at the frequencies f _1-1 and f _1-2 , and the satellite base The station apparatus 32 and the terminal station apparatus may be communicating with the frequencies _f2-1 and _f2-2 . Here, for the sake of simplicity, it is assumed that TDD (Time Division Duplex) is assumed and the uplink and downlink of each section use the same frequency.

Also, as can be seen from FIG. 34, except for the difference between the frequencies f ₁ and f ₂ , the antenna base station apparatuses 32-1 to 32-3 are substantially symmetrically folded. 1, the overall base station apparatus 31 is similar to the terminal station apparatus 302 of FIG. 16, and has a circuit configuration and signal processing. Has a configuration corresponding to the terminal station apparatus 302. Therefore, in the following description citing the circuit configuration, the description will be made with reference to the reference numerals shown in FIG.

  Furthermore, for example, when OFDM is used, it is necessary to perform individual signal processing in each subcarrier. However, for the sake of simplicity of explanation, the description for identifying the subcarrier in each signal and transmission / reception weight is omitted here. To do.

FIG. 34 particularly shows signal processing at the time of signal transmission from the overall base station apparatus 31 toward the terminal station apparatus 33. Although not specifically shown here, it should be noted that these signal processes are all performed for each subcarrier as in the above-described process. However, when the same transmission / reception weight can be used within the entire frequency bandwidth as will be described later, it is possible to extend the signal processing on the time axis for each sampling. First, in the case where the central base station apparatus 31 transmits N _SDM system signal sequences S _{1 to} S _N (here, N = N _SDM for convenience of subscript notation), In the second transmission signal processing circuit 148, in addition to the generation of the signal sequences S _{1 to} S _N as general transmission signal processing, the signal processing shown in (i) in FIG. _Assuming that the transmission weight matrix for signal separation is W _tx , signal conversion is performed by the following equation (30).

Here, as described in FIG. 25, in the signal processing corresponding to the second transmission signal processing circuit in FIG. 25, it is not always necessary to introduce the transmission weight matrix W _tx , and W _tx is regarded as a unit matrix. The transmission signal sequences t _{1 to} t _N may be left as the signal sequences S _{1 to} S _N. Next, directivity formation corresponding to the first singular value is performed on each satellite base station apparatus 32-1 to 32-N _SDM . Specifically, the channel matrix for each antenna of the k-th _S satellite base station apparatus 32 of interest is given by the first right singular vector when the matrix is subjected to singular value decomposition from each antenna of the overall base station apparatus 31. Transmission weight vectors w _{tx, kS, 1 to} w _{tx, kS, Nc} (Nc is the number of antenna elements provided in the overall base station apparatus for convenience of description here) is expressed by equation (31) (( Multiply the transmission signal t _kS as shown in ii)).

As shown in Expression (32) ((iii) in FIG. 34), _NSDM system signals addressed to each satellite base station apparatus 32 are added and combined for each antenna element to generate a time-axis signal.

Further, the IFFT & GI adding circuit 813 of the overall base station apparatus 31 performs IFFT conversion on the signal on the frequency axis shown in Expression (32) and converts it to a signal on the time axis (strictly, for example, OFDM) This includes signal processing such as insertion of a guard interval and a wave forming system, but the description is omitted here). The D / A converter 814 of the overall base station apparatus 31 D / A converts the IFFT conversion result to generate an analog baseband signal. The mixer 816 of the general base station apparatus 31, an analog baseband signal, multiplies the signal from the local oscillator 815 of the frequency of f _1, upconverts the analog baseband signal to a frequency band of the signal f _1. Filter 817 of general base station apparatus 31 removes out-of-band signals from the signal in the frequency band of f _1. The HPA 818 of the overall base station apparatus 31 amplifies the signal in the frequency band of f ₁ and transmits the amplified signal from the antenna element 819.

Antenna elements of the satellite base station apparatus 32 receives a signal in the frequency band of f _1. Low-noise amplifier of the satellite base station apparatus 32 (LNA) amplifies the signal in the frequency band of _{f 1.} The mixer of the satellite base station apparatus 32 down-converts the signal in the frequency band of f ₁ into an analog baseband signal. The filter of the satellite base station apparatus 32 removes out-of-band signals from the analog baseband signal. The A / D converter of the satellite base station apparatus 32 converts the analog baseband signal into a digital baseband signal. The FFT circuit of the satellite base station apparatus 32 converts the digital baseband signal into a subcarrier reception signal based on the symbol timing of the sampled signal.

Here, in the satellite base station apparatus 32, the first left singular vector obtained when the matrix of each antenna of the satellite base station apparatus 32 of interest is subjected to singular value decomposition from the respective antennas of the satellite base station apparatus 32 of interest. The received weight vectors w ′ _{rx, kS, 1 to} w ′ _{rx, kS, M given} by the equation (33) ((iv) in FIG. 34) are received signal vectors r ′ _{kS, 1 to} r ′. Multiply _{kS, M.} Here, M is the number of antenna elements of the satellite base station apparatus 32, and represents the number of antenna elements that communicate with the overall base station apparatus 31.

The scalar signal R ′ _kS obtained here is not an estimate of the transmission signal of the overall base station apparatus 31, but is an intermediate one-dimensional signal for extracting the virtual transmission path of the first singular value. This is different from regenerative relay. However, unlike normal non-regenerative relaying, this corresponds to generating the above-mentioned intermediate signal R ′ _kS and separately performing reception directional beam forming and transmission directional beam forming. In the transmission from the satellite base station apparatus 32 corresponding to the last access system to the terminal station apparatus 33, the last one-hop channel of the intermediate signal R ′ _kS obtained in this way is used. Signal transmission is performed using a virtual transmission line corresponding to the first singular value. Specifically, the transmission weight vector given by the first right singular vector obtained by performing singular value decomposition on the channel matrix for each antenna of the terminal station device 33 from each antenna of the satellite base station device 32 of interest. The received signal vector R ′ _kS is multiplied by w ′ _{tx, kS, 1 to} w ′ _{tx, kS, M} as shown in Expression (34) ((v) in FIG. 34).

Here, in the satellite base station apparatus 32, the number of antenna elements for the terminal station apparatus 33 has been described as M, but this does not necessarily need to match the number of antenna elements on the overall base station apparatus 31 side. Therefore, the number M of antenna elements may be understood as the number m of antenna elements. The transmission signal vector of each subcarrier thus obtained is converted into a time axis signal by IFFT (strictly speaking, for example, in the case of OFDM, signal processing such as insertion of a guard interval and wave forming system) The analog baseband signal is generated by D / A conversion and is multiplied by a signal from a local oscillator having a frequency of f ₂ by a mixer, and the frequency band of f ₂ is obtained. The signal is up-converted to the above signal, and after the out-of-band signal is removed by the filter, the signal is amplified by the high power amplifier and transmitted.

Here, the expressions (33) and (34) are obtained by multiplying the reception signal vector by the reception weight vector, and then multiplying the obtained R ′ _kS by the transmission weight vector. The transmission weight vector and the reception weight vector are multiplied in advance to generate a transmission weight matrix, and the received signal vectors r ′ _{kS, 1 to} r ′ _{kS, M} are directly multiplied by this. I do not care. However, the amount of calculation at this time is 2 × M complex multiplications when performing the equations (33) and (34) separately, whereas when performing them all together, the generation of the transmission weight matrix and the matrix A total of 2 × M ² complex multiplications is required for both multiplications of × vectors, and it is more efficient to divide them into individual operations.

Next, signal processing for the signal received by the terminal station device 33 will be described. Antenna elements of the terminal station 33 851, receives a signal in the frequency band of _{f 2.} LNA852 amplifies the signal in the frequency band of _{f 2.} The mixer 854 down-converts the signal in the frequency band of f ₂ to the analog baseband signal. Filter 855 removes out-of-band signals from the analog baseband signal. The A / D converter 856 converts the analog baseband signal into a digital baseband signal. The FFT circuit 857 converts the digital baseband signal into a subcarrier reception signal based on the symbol timing of the sampled signal.

Here, in the terminal station apparatus 33, first, the first channel information estimation circuit 146 or the first channel information estimation circuit 156 includes a virtual antenna element formed by each antenna element of each satellite base station apparatus 32 using a transmission weight vector. The channel information between each antenna of the terminal station apparatus 33 is acquired from the training signal given to the received signal. The first reception weight calculation circuit 147 or the first reception weight calculation circuit 157 generates a reception weight from the acquired channel information. The reception weight here may be a ZF-type or MMSE-type reception weight matrix for directly separating the signal sequence of the N _SDM system in one step (corresponding to FIG. 22), and corresponds to the first singular value. It is also possible to use a two-stage reception weight multiplication process (corresponding to FIG. 23) for once separating the signals into virtual transmission lines and then removing the mutual interference between the virtual transmission lines. In the latter case, this vector is based on the channel vector between the virtual antenna element formed by the transmission weight vector of each antenna element of each satellite base station apparatus 32 and each antenna of the terminal station apparatus 33. In addition, each component of the reception weight vector may be obtained by Expression (7) (maximum ratio combining weight), or all absolute values may be given constant with respect to the value given by Expression (7). (Weight of equal gain synthesis). The reception weight is set to w _{rx, kS, 1} to w _{rx, kS, Nm} (N _m is the number of antenna elements N _MT-Ant provided in the terminal station device for convenience of subscript here), and the terminal station Assuming that the signal vector received by each antenna element of the device 33 is Rx, the signal is once separated into a virtual transmission path by the equation given by Equation (35) ((vi) in FIG. 34).

Here, r _kS is a received signal of a virtual transmission path via the k-th _S satellite base station apparatus 32. At this time, since it is a signal of N _SDM × N _SDM MIMO transmission, the second received signal processing circuit 159 of the terminal station device 33 thereafter performs general MIMO signal processing. Specifically, referring to a known training signal given to the head of the received signal, an N _SDM × N _SDM channel matrix is obtained for the signal sequence of the N _SDM system, and the received signal is detected based on the channel matrix. Process. As shown above, it is also possible to multiply a ZF type inverse matrix or MMSE type linear reception weight matrix, or to perform nonlinear signal processing such as MLD. The N-system signals separated in this way are demodulated, and the reproduced data is output to the MAC layer processing circuit 68. As an example, when a ZF type or MMSE type linear reception weight matrix W _rx is used, the calculation shown in Expression (36) ((vii) in FIG. 34) is performed. Here, in order to avoid complications, N _SDM is represented as N in the equations (36) and (37).

In addition, as described above, instead of the two-stage signal processing of Expression (35) and Expression (36), direct signal separation may be performed using the received signal of each antenna element of the terminal station device 33 in one stage. Is possible. This is expressed by Equation (37) using a ZF-type or MMSE-type linear reception weight matrix ~ W _rx (~ is represented on the matrix W _rx ) that directly separates the received signal vector Rx. This corresponds to performing signal processing.

  The feature of the embodiment of the present invention described above is that the satellite base station apparatus 32 does not have the regenerative relay function in the prior art, but is not limited to the function of only the normal non-regenerative relay, An intermediate process of multiplying the reception weight vector and further multiplying the signal by the transmission weight vector is performed, and the transmission / reception weight vector basically effectively uses a virtual transmission line corresponding to a plurality of first singular values. This is a directivity control for this purpose.

  FIG. 35 shows a circuit configuration of the satellite base station apparatus according to the embodiment of the present invention. In FIG. 35, reference numeral 700 denotes an antenna element. Reference numeral 701 denotes TDD-SW. Reference numeral 702 denotes a low noise amplifier (LNA). Reference numeral 703 denotes a mixer. Reference numeral 704 denotes a filter. Reference numeral 705 denotes an A / D converter. Reference numeral 706 denotes an FFT circuit. Reference numeral 707 denotes a reception weight multiplication circuit. Reference numeral 708 denotes a transmission weight multiplication circuit. Reference numeral 709 denotes an IFFT & GI adding circuit. Reference numeral 710 denotes a D / A converter. Reference numeral 711 denotes a mixer. Reference numeral 712 denotes a filter. Reference numeral 713 denotes a high power amplifier. Reference numeral 714 denotes TDD-SW. Reference numeral 721 denotes an antenna element. Reference numeral 722 denotes a low noise amplifier (LNA). Reference numeral 723 denotes a mixer. Reference numeral 724 denotes a filter. Reference numeral 725 denotes an A / D converter. Reference numeral 726 denotes an FFT circuit. Reference numeral 727 denotes a reception weight multiplication circuit. Reference numeral 728 denotes a transmission weight multiplication circuit. Reference numeral 729 denotes an IFFT & GI adding circuit. Reference numeral 730 denotes a D / A converter. Reference numeral 731 denotes a mixer. Reference numeral 732 denotes a filter. Reference numeral 733 denotes a high power amplifier. Reference numeral 740 denotes a transmission weight processing unit. Reference numeral 741 denotes a local oscillator. Reference numeral 742 denotes a local oscillator. Reference numeral 743 denotes a communication control circuit. Hereinafter, for convenience of explanation, the left side is described as the side that communicates with the overall base station apparatus 31, and the right side is described as the side that communicates with the terminal station apparatus 33.

  First, the downlink direction (general base station apparatus 31 → satellite base station apparatus 32 → terminal station apparatus 33 direction) will be described. A signal received by each antenna element 700 is input to the low noise amplifier 702 via the TDD-SW 701. Here, the TDD-SW 701 is switched by the communication control circuit 743 in accordance with the transmission / reception timing. The weak signal input to the low noise amplifier 702 is amplified, and the amplified signal and the local oscillation signal output from the local oscillator 741 are multiplied by the mixer 703. The amplified signal is converted from a radio frequency signal to a baseband signal. Down-converted to Since the down-converted signal includes a frequency component outside the frequency band to be received, the out-of-band component is removed by the filter 704. The signal from which the out-of-band component is removed is converted into a digital baseband signal by the A / D converter 705. All digital baseband signals are input to the FFT circuit 706, and a signal on the time axis is converted to a signal on the frequency axis at a predetermined symbol timing determined by a timing detection circuit (not shown here). Separated into signals). The signal separated into each frequency component is input to the reception weight multiplication circuit 707 and also input to the transmission / reception weight processing unit 740-1.

  The reception weight multiplication circuit 707 multiplies the reception weight vector of the first singular value in correspondence with the virtual transmission path carried by the satellite base station apparatus 32, and from one vector-like signal by a plurality of antenna elements, one scalar-type Converted to a signal. This signal is input to the transmission weight multiplication circuit 708, and the transmission weight multiplication circuit 708 transmits the transmission weight vector corresponding to the virtual transmission line corresponding to the first singular value from the satellite base station apparatus 32 of interest to the terminal station apparatus 33. Is multiplied to generate a signal corresponding to each antenna element 721, which is input to the IFFT & GI giving circuit 709 of each antenna element system. Furthermore, processing such as guard interval insertion and waveform shaping between OFDM symbols (between block transmission blocks in the case of SC-FDE (Single-Carrier Frequency Domain Equalization)) is performed for each antenna element 721. D / A converter 710 converts digital sampling data to baseband analog It is converted to issue. Further, each analog signal is multiplied by a local oscillation signal input from the local oscillator 742 by a mixer 711 and up-converted to a radio frequency signal. Here, since the upconverted signal includes a signal in a frequency component outside the band of the channel to be transmitted, the filter 712 removes the frequency component outside the band and generates an electrical signal to be transmitted. . The generated signal is amplified by the high power amplifier 713 and transmitted from the antenna element 721 via the TDD-SW 714.

  Next, the uplink direction (terminal station apparatus 33 → satellite base station apparatus 32 → overall base station apparatus 31 direction) will be described. A signal received by each antenna element 721 is input to the low noise amplifier 722 via the TDD-SW 714. Here, the switching of the TDD-SW 714 is performed by the communication control circuit 743 in accordance with the transmission / reception timing. The weak signal input to the low noise amplifier 722 is amplified, and the amplified signal and the local oscillation signal output from the local oscillator 742 are multiplied by the mixer 723. The amplified signal is converted from the radio frequency signal to the baseband signal. Down-converted to Since the down-converted signal includes a frequency component outside the frequency band to be received, the out-of-band component is removed by the filter 724. The signal from which the out-of-band component is removed is converted into a digital baseband signal by the A / D converter 725. All the digital baseband signals are input to the FFT circuit 726, and a signal on the time axis is converted into a signal on the frequency axis at a predetermined symbol timing determined by a timing detection circuit (not shown here) (for each frequency component). Separated into signals). The signal separated into each frequency component is input to the reception weight multiplication circuit 727 and also input to the transmission / reception weight processing unit 740-2.

  In the reception weight multiplication circuit 727, the reception weight vector of the first singular value is multiplied corresponding to the virtual transmission line carried by the satellite base station apparatus 32, and a single scalar-type signal is obtained from the vector-like signal by the plurality of antenna elements. Converted to a signal. This signal is input to the transmission weight multiplication circuit 728, and the transmission weight multiplication circuit 728 transmits the transmission weight corresponding to the virtual transmission path corresponding to the first singular value from the satellite base station apparatus 32 of interest to the general base station apparatus 31. The vector is multiplied to generate a signal corresponding to each antenna element 700, and these signals are input to the IFFT & GI adding circuit 729 of each antenna element system. The IFFT & GI adding circuit 729 generates a signal on the time axis from the signal on the frequency axis. After being converted into a signal, processing such as insertion of a guard interval and waveform shaping between OFDM symbols (between block transmission blocks in the case of SC-FDE (Single-Carrier Frequency Domain Equalization)) is performed for each antenna element 700. D / A converter 730 converts the digital sampling data into baseband analog It is converted into grayed signal. Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 741 by the mixer 731 and up-converted to a radio frequency signal. Here, since the signal is included in the frequency component outside the band of the channel to be transmitted in the upconverted signal, the frequency component outside the band is removed by the filter 732 to generate an electrical signal to be transmitted. . The generated signal is amplified by the high power amplifier 733 and transmitted from the antenna element 700 via the TDD-SW 701.

  The transmission / reception weight vector multiplied by the reception weight multiplication circuit 707 and the transmission weight multiplication circuit 728 is calculated by the transmission / reception weight processing unit 740-1. The transmission / reception weight vector multiplied by the reception weight multiplication circuit 727 and the transmission weight multiplication circuit 708 is calculated by the transmission / reception weight processing unit 740-2. A reception weight vector is calculated based on channel information acquired by using the downlink and uplink FFT circuits, and a transmission weight vector in the opposite direction is calculated by performing a calibration process. In transmission / reception with the central base station apparatus 31, once the channel information is acquired because the propagation path is fixed and the line-of-sight environment is high, the value may be continuously used thereafter. Is possible.

  Therefore, it is possible to repeatedly perform channel estimation before the start of service operation, and to use a highly accurate channel estimation result (in this case, using relative channel information) and a transmission / reception weight vector calculated therefrom. Is possible. On the other hand, channel information between the satellite base station device 32 and the terminal station device 33 changes due to movement of the terminal station device 33 and its surroundings and environmental fluctuations, so channel information acquisition and weight vector updating are necessary in real time. It becomes. Any means may be used as the channel information acquisition (channel feedback) method. For the channel information acquired in this way, the transmission / reception weight processing units 740-1 and 740-2 set the transmission / reception weight vector of the virtual transmission path corresponding to the first singular value by a technique such as singular value decomposition. This is calculated and input to reception weight multiplication circuits 707 and 727 and transmission weight multiplication circuits 708 and 728. Note that the communication control circuit 743 grasps transmission / reception timing and symbol timing, and gives various instructions to the TDD-SWs 701 and 714 and various circuits. The reason why the communication control circuit 743 grasps such timing can be grasped by synchronizing with the central base station apparatus 31 in communication with the central base station apparatus 31, and can be understood by GPS or other wireless systems (for example, It may be performed using information acquired from a signal from a macro cell base station).

  As described above, the wireless communication system 52 according to the third embodiment includes the overall base station device 31 (first wireless station device), the plurality of satellite base station devices 32 (second wireless station device), and a terminal. A station device 33 (third wireless station device). The overall base station apparatus 31 includes a first radio station signal processing unit that performs radio communication with the plurality of satellite base station apparatuses 32 via the first antenna element group. The satellite base station apparatus 32 performs wireless communication with the overall base station apparatus 31 via the second antenna element group, and executes wireless communication with the terminal station apparatus 33 via the third antenna element group. 2 radio station signal processing unit.

  The second radio station signal processing unit of the satellite base station apparatus 32 (second radio station apparatus) of the third embodiment is used for radio communication between the first antenna element group and the second antenna element group. The received weight vector for the MIMO channel matrix is set to at least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix, or an approximate solution of the first right singular vector and the first left singular vector. Calculation is based on at least one of the approximate solutions of singular vectors. The second radio station signal processing unit generates a transmission signal sequence by multiplying a signal received via the second antenna element group by a reception weight vector. The second radio station signal processing unit transmits a signal based on the generated signal sequence to the terminal station device 33 (third radio station device) via the third antenna element group.

  The second radio station signal processing unit of the satellite base station device 32 (second radio station device) of the third embodiment includes the fourth antenna element group and the satellite of the terminal station device 33 (third radio station device). A transmission weight vector for the MIMO channel matrix used for wireless communication with the third antenna element group of the base station device 32 (second wireless station device) is a first right corresponding to the first singular value of the MIMO channel matrix. Calculated based on at least one of the singular vector and the first left singular vector, or at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector, and via the third antenna element group A plurality of signal sequences are generated by multiplying the received signal by a transmission weight vector, and signals based on the plurality of signal sequences are transmitted via the second antenna element group. And it transmits to end station device 33 (third radio station).

  The first radio station signal processing unit of the radio communication system 52 of the third embodiment calculates a transmission weight vector for a MIMO channel matrix used for radio communication between the first antenna element group and the second antenna element group, At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix, or at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. To generate a signal sequence for each second radio station device, multiply the generated signal sequence by a transmission weight vector corresponding to each signal sequence, and add and synthesize the results for all signal systems Then, a signal based on the added and synthesized signal sequence is transmitted to the satellite base station apparatus 32 (second radio station apparatus) via the first antenna element group.

  The first wireless station signal processing unit of the wireless communication system 52 of the third embodiment is configured to calculate a reception weight vector for a MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group, At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix, or at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. Is calculated for each satellite base station apparatus 32 (second radio station apparatus) based on the signal, and the signal received via the first antenna element group is multiplied by the reception weight vector for each second radio station apparatus. A plurality of signal series are generated, and a residual interference component is removed from a signal based on the plurality of signal series.

  As a result, the central base station device 31, the satellite base station device 32, the terminal station device 33, the wireless communication system 52, and the wireless communication method of the third embodiment increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. It becomes possible to make it. The overall base station apparatus 31, satellite base station apparatus 32, terminal station apparatus 33, wireless communication system 52, and wireless communication method of the third embodiment can increase the transmission capacity by non-regenerative re-beamforming relay. It becomes. The central base station device 31, the satellite base station device 32, the terminal station device 33, the wireless communication system 52, and the wireless communication method according to the third embodiment may transmit a plurality of signal sequences in a spatially multiplexed manner with desired characteristics. It becomes possible.

  In the first embodiment, the base station device 70, the terminal station device 60, the radio communication system 50, and the radio communication method use a plurality of virtual transmission paths corresponding to the first singular value for the access system, and perform spatial multiplexing. It is also possible to perform. However, since the coordinates of the terminal station device 60 cannot be designed fixedly, it is preferable to arrange a plurality of first signal processing units 304 on the base station device 70 side in a spatially multiplexed manner. For this reason, when a function corresponding to the first signal processing unit 304 is arranged as the satellite base station apparatus 32, it is preferable to perform relaying wirelessly rather than being wired around. When realizing this mode of use, signal processing is effective in that only beamforming is performed at the relay station while it is a non-regenerative relay, and an implementation method for that is defined.

[Fourth Embodiment]
[Train moving cell using multiple virtual transmission lines corresponding to the first singular value]
(Outline of train moving cell and basic principles)
As described in the previous third embodiment, in the future wireless access for 5G (fifth generation), in particular, instead of accommodating the traffic of a large number of users in the train in the macrocell, It is effective to collect and deallocate user data to the wired network side at the wireless entrance and offload all at once in order to effectively utilize the macrocell frequency resources and improve the total transmission capacity of the entire system. is there. Here, no distinction is made between trains and trains, and all vehicles such as conventional lines and bullet trains are labeled as trains.

  However, in the case of trains, the movement speed is generally high, and if it is a Shinkansen, it will exceed 300 km / h, and the conventional line will be constructed assuming a transmission speed of 100 km / h or more. Must. In particular, in order to provide a large-capacity line, it is necessary to use a high frequency band such as a millimeter wave, and in terms of line design, a gain loss proportional to the square of the wavelength and a high-power amplifier of the millimeter wave band Considering that the output is very small compared to the microwave band and that the high output is expensive, it is necessary to secure a line gain by making a large-scale antenna using a large number of low-output antenna elements. In particular, considering the train moving speed, in order to reduce the frequency of handover and reduce the control load, it is necessary to cover an area of about 500 [m] intervals with the base station apparatus on the entrance side. In securing the gain of a large-scale antenna, it is necessary to perform feedback of channel information with high accuracy and to perform high-precision directivity control, but the time variation of the channel is generally large due to the fast movement speed of the train, At this time, it is necessary to follow the fluctuation with high accuracy and to form high-speed directivity.

  As mentioned above, the time variation of the MIMO channel generally tends to be relatively small in the virtual transmission line corresponding to the one having the larger singular value obtained when the singular value decomposition of the channel matrix is performed. On the contrary, in a virtual transmission line corresponding to a small singular value, channel time variation tends to become very large. Here, since it is assumed that the transmitting and receiving stations are in the line-of-sight environment, the virtual transmission line corresponding to the first singular value is a transmission line corresponding to the line-of-sight wave. In this case, taking into consideration that the train track is almost straight, a highly directional antenna is installed near the top of the track (such as an overhead pole) with directivity in the direction parallel to the track. If the wireless signals are transmitted and received in a face-to-face manner, the line-of-sight environment can be secured stably for a long time. The channel information between the antenna elements changes very rapidly because the moving direction of the train and the arrival direction of the radio waves are substantially parallel. On the other hand, the relative relationship (complex phase difference) of the channel information of each path between the antenna elements is in a state that can be approximated to a plane wave, and therefore hardly changes with respect to time variation. In other words, although the channel itself fluctuates at a high speed, the transmission / reception weight of the virtual transmission line corresponding to the first singular value changes very slowly, and even if a constant transmission / reception weight value is kept for a certain period, it is almost ideal. It is possible to maintain the directivity formation.

  On the other hand, the virtual transmission line corresponding to the second singular value or later actively uses many reflected waves, and the incident waves from various directions that are not parallel to the moving direction are dominated. It becomes the target. At this time, when waves from various angular directions are combined, the situation is clearly different from that of a plane wave, and the relative relationship of channel information corresponding to each path between antenna elements varies greatly with time variation. To do. Therefore, it is necessary to update the transmission / reception weight at high speed in order to follow fluctuations in the relative relationship of channel information. If high-speed updating cannot be performed, appropriate directivity cannot be formed. Here, even if implicit feedback is introduced to reduce the overhead of feedback of channel information, the training signal for channel estimation must be exchanged very frequently in order to follow this rapid fluctuation of channel information. It is difficult to achieve the accuracy that is actually required.

  Here, various circuits for transmission / reception (high power amplifier, low noise amplifier, filter, mixer, etc.) provided in the terminal station device and the base station device of the wireless entrance installed in the train are complex that change due to temperature characteristics and the like. It is assumed that the rotation amount of the phase can be handled as being almost constant without time variation by a highly accurate calibration process. In this case, if attention is paid to a certain train, the movement path of the train (strictly, each antenna element of the terminal station device mounted on the train) will show a one-dimensional displacement with extremely high accuracy if a passing track is determined. If the coordinates on the one-dimensional moving route can be grasped, the channel information when there is a train at the same place every time becomes almost the same channel.

  Of course, for example, assuming 80 GHz, the wavelength is 3.75 mm, and a deviation of coordinates of only 1 mm results in an error of coordinates of 25% or more of the wavelength, and the absolute channel information changes greatly. This is the relative channel information necessary for calculating the transmission / reception weight. In the case of a line-of-sight wave that arrives in a plane wave, as long as the plane and the plane in which the antenna elements are two-dimensionally arranged are substantially parallel, the relative channel information is not greatly affected by the coordinate error. In general, with regard to fluctuations in transmission / reception weights, the transition is in the vertical direction compared to the case where the transition is perpendicular to the line-of-sight direction (the direction parallel to the straight line connecting the transmission / reception antennas) and the case where the transition is parallel to the line-of-sight direction. In this case, the fluctuation amount of the transmission / reception weight becomes larger. Of course, the play between the track and the width of the train wheel may be several millimeters or more, so it does not pass exactly the same coordinates, and it also involves some errors in the direction perpendicular to this line-of-sight direction. Become. However, in the case of the virtual transmission line corresponding to the first singular value as described above, it can be approximated by a plane wave, and if the error is such a level, the change in the transmission / reception weight is within an allowable range. On the other hand, in the case of transition in a direction parallel to the line-of-sight direction, the accuracy of the determination of the train position is lowered due to the high moving speed. However, in the case of a plane wave, the transmission / reception weight is quite insensitive compared to the case where the train transits in the vertical direction, and can be sufficiently handled.

  Note that although the transmission / reception weight itself may have a small time fluctuation, the complex phase of the signal synthesized using this transmission / reception weight fluctuates with the coordinates, which is observed as a Doppler shift. However, since the Doppler shift itself is not essentially different from the frequency error, even if OFDM is used, for example, if it is smaller than the subcarrier interval, it does not break the orthogonality of OFDM. Therefore, it is possible to cope with the residual frequency error by performing tracking processing or the like.

  First, a mechanism for determining the position (coordinates) of a train in a one-dimensional train movement with high accuracy is introduced. For example, marks that specify some coordinates are arranged at predetermined intervals on the side of the track, the marks are photographed with a camera or the like, and the image is subjected to image processing to measure the coordinates. For example, barcode-like geometric coordinate information marks are arranged at intervals of 10 [m], and an image photographed by a camera is analyzed to identify the coordinates of the mark from the barcode. Although a time lag occurs between the time when the mark actually passes through and the time when the coordinates of the mark can be specified by image analysis, it is possible to obtain in advance how long this time lag will be. On the other hand, since this time lag is estimated to be substantially the same for all the landmarks, if the landmarks are installed at intervals of 10 [m], the difference in time at which the landmarks are recognized by image analysis From the relationship of the moving distance of 10 [m], it is possible to estimate the moving speed at that moment with high accuracy. Furthermore, from the moving speed and the time lag described above, it is possible to accurately estimate where the train is actually at the moment when the coordinates of the landmark are determined, and the coordinates when a certain minute time has passed from there are also the same. Can be estimated.

  In this way, it is possible to accurately obtain instantaneous coordinates in a one-dimensional movement. Here, a barcode is shown as an example, but an optical signal such as an LED may be used, or coordinates may be specified using an electromagnetic signal such as a wireless tag. The current GPS coordinate estimation accuracy is not so high, but when GPS correction value information is acquired via a radio line (different frequency band or via a different system) separately set as Assisted GPS It is known that the accuracy of GPS positioning is suppressed to about 5 to 10 [m]. Furthermore, in the case of trains, the travel route is one-dimensional in order to travel on the track, and even when using GPS or the like, if the location is limited to that one-dimensional route (on the track), the accuracy is further increased. . In the future, it is said that coordinate estimation using a quasi-zenith satellite can be performed with an accuracy of several centimeters, and such a future GPS function can naturally be used. Furthermore, as with car speedometers and distance meters, it is also possible to obtain the distance traveled from the number of wheel rotations. When stopping at a station platform, etc., high-precision coordinate correction is performed. If a combination is used, such as a combination of the movement distance, GPS information, and other coordinate acquisition means, the measurement accuracy is greatly improved. In this way, highly accurate coordinate estimation is possible without using a camera and an image recognition function as described above. Even if the coordinate information is acquired by any means, the present embodiment can be similarly applied.

  In the above description, the case where the coordinates are estimated on the train side has been described. Similarly, it is necessary to specify the coordinates of the train as a communication partner on the base station apparatus side as well. Basically, the base station apparatus acquires the coordinates of the train using the coordinate specifying means of the traveling train connected via the base station apparatus and the wired network. In this way, when specifying the coordinates of the train on the wired network side, for example, a configuration may be used in which a light emitter and its light receiver are arranged along the track and light is blocked when passing through the train. Alternatively, a configuration may be adopted in which a light emitter is installed on the train side, and a light receiver along the line detects the light of the light emitter. It is unrealistic in terms of cost to install a large number of cameras beside the track, but it is possible to obtain the coordinates with high accuracy by other methods. Or, if the coordinate information acquired by the train can be notified to the base station device side via a wireless line (which may be in a different frequency band or via a different system) that is separately set with low delay, It is also possible to specify the coordinates of the train on the base station apparatus side without using the coordinate information acquisition means connected in this way.

  Next, in such a state that the coordinates of such a train can be specified, the channel matrix at that time is calculated while driving an actual vehicle in a time zone outside normal service such as midnight or before service operation starts. It is acquired in advance in association with the coordinate information, and stored in the base station apparatus or terminal station apparatus. If the average value is obtained by measuring several times, the correspondence relationship between the channel matrix and the train coordinates can be obtained. When obtaining this average value, instead of using absolute channel information as measured, relative channel information based on the complex phase of the reference antenna is used, and transmission and reception weights are calculated for this relative channel information. To do. Channel matrix measurement is performed not only on the train side but also on the base station device on the wired network side. Alternatively, uplink channel information may be acquired from downlink channel information by a technique such as implicit feedback.

  As described above, a database of the relationship between train coordinates and uplink and downlink channel matrices is constructed in advance before actual service operation starts. More precisely, based on the channel matrix, the transmission / reception weight of the virtual transmission line corresponding to the first singular value is calculated in advance, and the transmission / reception weight is stored in the database. For example, on the train side, transmission weight vectors used in the uplink of each virtual transmission path and reception weight vectors used in the downlink are stored in a database in association with discrete coordinate information. Similarly, on the base station apparatus side, the reception weight vector used in the uplink of each virtual transmission path and the transmission weight vector in the downlink are stored in the database in association with the discrete coordinate information.

  FIG. 36 shows an outline of the configuration of the wireless communication system in the fourth embodiment. In the figure, reference numeral 600 is a base station apparatus, reference numerals 601-1 to 601-8 are first signal processing units of the base station apparatus, reference numerals 602-1 to 602-8 are geometric coordinate information, and reference numeral 603 is a train. Reference numeral 604 denotes a terminal station apparatus, reference numeral 605 denotes a camera, reference numeral 606 denotes a train side weight matrix / coordinate database, and reference numeral 607 denotes a base station apparatus side weight vector / coordinate database. In addition, when showing either of the 1st signal processing parts 601-1 to 601-8, it describes as the 1st signal processing part 601.

  The train 603 is provided with a terminal station device 604 provided with a number of antenna elements above the top of the leading vehicle in the traveling direction or above the last portion of the last vehicle. The installation location here may be a configuration in which signals are transmitted and received through the window from inside the train 603, or may be installed outside the train 603 and covered with a cover (radome) for further reducing the influence of rain and wind. good. Further, the base station apparatus 600 includes a plurality of first signal processing units 601-1 to 601-8 such as the first signal processing units 304-1 to 304-4 illustrated in FIG. 16 in the first embodiment. The first signal processing units 601-1 to 601-8 are installed with a physical distance. For example, a large number of first signal processing units 601-1 to 601-8 are installed at intervals of 1 m or 2 m. In contrast to the first signal processing units 601-1 to 601-8, the base station apparatus 600 also includes the second signal processing unit 305 illustrated in FIG. The first signal processing units 601-1 to 601-8 included in the base station device 600 are installed on an overhead pole provided to cross the track in a direction perpendicular to the track of the train 603, for example. . The first signal processing units 601-1 to 601-8 include a large number of antenna elements and a circuit for signal processing, like the first signal processing units 304-1 to 304-4 illustrated in FIG. 16.

  Similarly, the terminal station apparatus 604 includes a large number of antenna elements. The antenna element of the terminal station device 604 and the antenna elements of the first signal processing units 601-1 to 601-8 each have a configuration in which directivity is increased in the forward direction as viewed from the antenna element in order to ensure the line gain. Since the movement of the train 603 may involve a curve rather than a complete straight line, it is not necessary that the directivity gain is excessively high. The base station device 600 is not viewed from the terminal station device 604 installed in the train 603. It is ideal that the antenna element has a high directivity gain within a range in which the directivity gain does not decrease with respect to the existing orientation and the orientation in which the terminal station device 604 of the train 603 is present when viewed from the base station device 600. is there.

  In the example shown in FIG. 36, a camera 605 is mounted on the train 603, and a device that captures the geometric coordinate information 602 on the side of the track and grasps the coordinates of the train 603 using an image analysis function that is not clearly shown in the drawing is used. 603 includes. The base station apparatus 600 includes a weight vector / coordinate database 607, and the terminal station apparatus 604 of the train 603 includes a weight matrix / coordinate database 606. The geometric coordinate information 602-1 to 602-8 is installed at the side of the track at approximately equal intervals, and is arranged at a position where the camera 605 mounted on the train 603 can be photographed. The geometric coordinate information 602-1 to 602-8 includes coordinate information indicating the coordinates where the geometric coordinate information 602-1 to 602-8 is installed or geometric coordinate information 602-1 to 602-8. Is recorded with a barcode or the like.

  For example, when the train 603 travels while shooting the geometric coordinate information 602-1 to 602-8 with the camera 605 and recognizes the image of the geometric coordinate information 602-7, the geometric coordinate information The coordinates of the geometric coordinate information 602-7 are specified from the information of 602-7. If there is a time lag in image recognition, the time lag time ΔT is known at the time of factory shipment or product design. Since the geometric coordinate information 602-1 to 602-8 is sequentially recognized by the camera 605, the geometric coordinate information 602-8 is recognized at a certain time t1 [s], and another time t2 [s]. If the geometric coordinate information 602-7 is recognized, the distance x [m] between the geometric coordinate information 602-8 and the geometric coordinate information 602-7 is v = x / At (t2−t1), the second speed [m / s] is grasped, and at time t2 + Δt when a minute time Δt has elapsed from time t2, v × ΔT [m] is ahead of the coordinates of the geometric coordinate information 602-7. It is judged that it is located in.

  On the other hand, before the start of service operation, the train 603 travels on a predetermined track, and the training signal transmitted from each of the first signal processing units 601-1 to 601-8 of the base station device 600 is transmitted to the terminal station. The information is received by each antenna element of the device 604, and channel information of the MIMO channel is acquired. Based on the acquired channel information, the reception weight vector and the transmission weight vector used in the communication using the virtual transmission path corresponding to the first singular value are used as the first signal processing units 601-1 to 601-8. And is recorded in the weight matrix / coordinate database 606. When the reception weight vector and the transmission weight vector are acquired a plurality of times, it is possible to improve accuracy by obtaining an average value thereof. In this averaging, for example, the transmission weight vector of the antenna element included in the terminal station device 604 is used in a form based on the complex phase of the transmission / reception weight vector of a predetermined antenna element among the antenna elements included in the terminal station device 604. It is preferable to perform processing capable of averaging even information acquired at different times, such as making each component of the above information a relative transmission / reception weight.

  A similar database is created on the base station apparatus 600 side and stored in the weight vector / coordinate database 607. In the above description, the case where the training signal is transmitted from the base station apparatus 600 side has been described, but the coordinates and time information of the train 603 are recorded on the train 603 (or terminal station apparatus 604) side by some method described later, A training signal is transmitted from the terminal station device 604 side, received by each antenna element of the first signal processing units 601-1 to 601-8 of the base station device 600, and the channel information of the MIMO channel together with the reception time information Record. Based on the recorded time information and train coordinate information, and the relationship between the time information and channel information, reception for predetermined train coordinates used in communication using the virtual transmission line corresponding to the first singular value described above The weight vector and the transmission weight vector are acquired for each of the first signal processing units 601-1 to 601-8 and recorded in the weight vector / coordinate database 607. Alternatively, as long as the downlink channel information can be acquired on the train 603 side, it is also possible to acquire the uplink channel information using the calibration technique. Therefore, the training signal transmitted from the base station apparatus 600 side can be obtained. After acquiring a set of channel information on the train 603 side, uplink channel information is acquired by calibration processing offline, and the first signal processing units 601-1 to 601-601 are based on this channel information. The transmission weight vector and the reception weight vector may be calculated every 8 and may be stored in the weight vector / coordinate database 607 as a database.

  In the above processing, information acquired on the base station device 600 side and information acquired on the terminal station device 604 side should be comprehensively processed and recorded in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607. In order to generate information, necessary information is once collected in an environment such as another computer, and information processed by the information is separately stored in the weight vector / coordinate database 607 of the base station device 600 and the terminal station device 604. The weight matrix / coordinate database 606 may be recorded.

  Here, the coordinate interval corresponding to the transmission / reception weight vector in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607 matches the interval between the positions where the geometric coordinate information 602-1 to 602-8 are installed. There is absolutely no necessity. As described above, the coordinate information can be acquired from the moving speed and elapsed time of the train at any position between the geometric coordinate information 602-1 to 602-8, and the channel information acquired at the arbitrary position. Therefore, it is possible to calculate the transmission / reception weight vector at the arbitrary position, so that the interval between the positions (coordinates) corresponding to the transmission / reception weight vector in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607 is arbitrary. Can be set.

  In the weight matrix / coordinate database 606 of the terminal station device 604 and the weight vector / coordinate database 607 on the base station device 600 side, transmission / reception weight vectors for discrete coordinates are recorded, and for continuous data transmission / reception. Needs to obtain a transmission / reception weight vector between coordinates corresponding to the discrete transmission / reception weight vector. However, if the step size of coordinates corresponding to the transmission / reception weight vector / matrix recorded in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607 is sufficiently short, the transmission / reception weight vector / matrix corresponding to each of the consecutive coordinates. If the difference is sufficiently small, the same transmission / reception weight vector / matrix is continuously used until the coordinates where the next transmission / reception weight vector / matrix is recorded, so that there is no problem. If there are no restrictions on the storage capacity of the weight vector / coordinate database 606 and the weight vector / coordinate database 607, the same transmission / reception weight vector / matrix can continue to be used up to the newly recorded coordinate point. The transmission / reception weight vector / matrix of the coordinate information is preferably recorded in the weight vector / coordinate database 606 and the weight vector / coordinate database 607. However, when the storage capacity is limited, for example, by using a transmission / reception weight vector / matrix corresponding to each of the coordinates of two consecutive points, interpolation is performed in an interval between the coordinates of the two points, and approximate The transmission / reception weight vector / matrix in the section may be calculated.

  Note that when the train 603 is moving at a high speed, a Doppler shift occurs in the frequency of the received signal observed by the base station apparatus 600 and the terminal station apparatus 604. For this reason, in the above description, the channel information measured in advance before the start of service operation naturally includes the influence of the Doppler shift depending on the moving speed of the train 603. In this Doppler shift, since the wavelength shifts with the frequency offset, the value of the channel information itself varies depending on the speed of the train 603. In normal operation of train 603, it is expected to run at the same speed in the same place, but in reality, the possibility of slowing down due to some trouble in operation cannot be denied, and in such a situation the train It is necessary to avoid deterioration of the communication quality of the entrance line, which causes a problem when a user in 603 accesses the Internet.

  However, each antenna element of the first signal processing units 601-1 to 601-8 of the base station apparatus 600 and each antenna element of the terminal station apparatus 604 correspond to the first singular values. When using a virtual transmission line, it is assumed that the antenna elements are concentrated in a very narrow area. In this case, the path length difference between the first signal processing units 601-1 to 601-8 of the base station device 600 and the antenna elements of the terminal station device 604 is small. Furthermore, the direction in which each antenna element is visible is in the same direction with very high accuracy. In this case, the amount of Doppler shift generated in each antenna element is almost the same value, and the complex phase difference of the relative channel state between each antenna element is a value that does not depend on the moving speed of the train 603 with high accuracy. . That is, the channel information (and the channel vector having the channel information as a component) has different values due to the moving speed of the train 603, but the transmission / reception weight vector corresponding to the channel vector itself does not depend on the moving speed of the train 603. It will be. Therefore, it is possible to calculate the moving speed of the train 603 by the above-described method, and it is also possible to estimate the Doppler shift amount from the relationship between the moving speed and coordinates and to correct the channel vector for an arbitrary moving speed. In the wireless communication system according to the fourth embodiment that actively uses the virtual transmission path corresponding to the first singular value, the Doppler shift is possible. It is possible to use the transmission / reception weight vector stored in the database without any correction.

(Overview of signal processing in the wireless communication system of the fourth embodiment)
The outline of signal processing performed in the wireless communication system of the fourth embodiment will be described below with reference to the drawings. The training signal used here may be, for example, a training signal of an OFDM signal that can acquire channel information of each frequency component, or may be a training signal with a small autocorrelation for calculating a time correlation. Also, even when using an OFDM signal as a training signal, it is not always necessary to provide a guard interval like a general OFDM signal, and it is possible to use a training signal in which each frequency component is continuous without a guard interval. is there. This is because absolute channel information (or channel matrix, channel vector) is not always necessary in the calculation of transmission / reception weight vectors, and it is sufficient if only the relative relationship of complex phases between antenna elements can be obtained. This is because there is no necessity to detect the symbol timing of the OFDM signal with high accuracy if the phase relationship with respect to any one reference antenna element can be obtained. Moreover, even in the case of an OFDM signal, there is no need for a training signal using all subcarriers to be used, as described later, and the number of subcarriers allocated as training signals is limited to increase the transmission power per subcarrier. Thus, it may be a training signal used in a state where the SNR on the receiving side is increased.

<When transmitting a training signal from the base station device>
FIG. 37 is a diagram illustrating an outline of signal processing in a train moving cell using a plurality of virtual transmission lines corresponding to the first singular value in the fourth embodiment. Here, an example is shown in which a training signal is transmitted from the base station apparatus to the terminal station apparatus side, and coordinate information and channel information are recorded on the terminal station apparatus side. Although not explicitly described in the following description, for example, in the case of a broadband system such as when using OFDM, it is necessary to acquire channel information for each subcarrier and calculate transmission / reception weights. However, in order to simplify the explanation, descriptions such as subscripts related to subcarriers and the like and detailed explanations are omitted.

  FIG. 37A is a flowchart showing processing performed by the base station apparatus 600. When starting the process, the base station device 600 continuously transmits training signals (step S3701). The base station apparatus 600 ends the process after continuously transmitting training signals over a predetermined period. For example, the base station device 600 detects that the train 603 exists in the area of the base station device 600 and continuously transmits a training signal over a period in which the train 603 exists in the area. The terminal station device 604 provided in the train 603 receives the training signal transmitted from the base station device 600. The detection of the train 603 by the base station apparatus 600 is performed based on the operation information of the train 603, or is performed by some other method.

  FIG. 37B is a flowchart showing processing performed by the terminal station device 604 in the train 603. When the terminal station device 604 starts processing, the terminal station device 604 obtains coordinate information by an arbitrary method such as analysis of a captured image captured by the GPS or the camera 605 during traveling (step S3711), and the terminal station device 604 performs arbitrary processing according to the acquisition of the coordinate information. Time information is also acquired by the technique (step S3712). The terminal station device 604 associates and stores the coordinate information acquired in step S3711 and the time information corresponding to each coordinate information acquired in step S3712 (step S3713), and ends the process. The time acquisition method in this process may use, for example, information from other wireless communication systems such as GPS or a macro cell. A clock that has been set in advance is mounted, and the time recorded by the clock is set. You may record as it is. Here, when the coordinate information and the time information are recorded in association with each other in step S3713, not the time information after the time lag for analyzing the coordinate information but the raw information including the coordinate information before the time lag occurs (for example, the camera When the time information and the coordinate information after the analysis are recorded together, when processing information in the offline processing described later, this time lag is related to the time lag. There is no need to make corrections. The correspondence between the time information and the coordinate information is not necessarily recorded by the terminal station device 604 of the wireless communication system, and even the time information and the coordinate information can be recorded in another completely independent system. Any device may be used.

  FIG. 37 (c) is also a flowchart showing processing performed by the terminal station device 604 in the train 603. The terminal station device 604 performs the process shown in FIG. 37C in addition to the process shown in FIG. When starting the process, the terminal station device 604 receives a training signal (step S3721), and acquires time information in accordance with the reception of the training signal (step S3722). The terminal station device 604 performs reception signal processing on the received training signal and performs channel estimation (step S3723). As the received signal processing here, for example, the received signal is amplified by a low noise amplifier, converted from a radio frequency to a baseband signal through a mixer, a sampling signal on the time axis is generated by A / D conversion, and OFDM If the modulation method is used, FFT processing is performed at a predetermined symbol timing, and channel estimation is performed by comparison with a known training signal. If the training signal is a continuous signal that does not include a guard interval, the FFT processing may be performed at an arbitrary timing. Further, the channel estimation result may be divided by the channel estimation result of the reference antenna element or its complex phase, and converted into relative channel information that is actually required for calculation of the transmission / reception weight vector.

  The terminal station device 604 stores the channel information acquired in step S3723 in association with the time information acquired in step S3722 (step S3724), and ends the process. Here, the timing at which the channel information can be acquired in step S3723 has a time lag unlike the time when the training signal is received in step S3721, but the time information acquired in step S3722 is the time when the training signal is received in step S3721. If managed in this manner, even if a time lag occurs until the timing at which channel information can be acquired in step S3723, it is possible to eliminate the time lag from the time information recorded in association with the channel information in step S3724. In the description, it is assumed that the time lag is not included in the acquisition time of the channel information. Note that the above-described signal processing and channel estimation can be separately performed as offline processing at different times, so that raw sampling data may be recorded as it is. However, in the following description, for the sake of convenience, the description will be made assuming that the channel estimation result (relative channel information) is recorded.

  FIG. 38 is a diagram illustrating an outline of offline signal processing performed after acquiring coordinate information and channel information according to the fourth embodiment. Here, as in FIG. 37, an example of offline processing when a training signal is transmitted from the base station apparatus 600 to the terminal station apparatus 604 side and coordinate information and channel information are recorded on the terminal station apparatus 604 side is shown. Yes. This off-line signal processing is performed by a signal processing device. The signal processing apparatus may be provided in the terminal station apparatus 604, or may be provided as an independent apparatus that can acquire the coordinate information, time information, and channel information acquired in the terminal station apparatus 604.

  When the offline signal processing is started, the signal processing device reads the data of the combination of the recorded time information and coordinate information (step S3801), and the coordinate difference from the coordinate information corresponding to the previous time in time series (Movement distance) is calculated (step S3802). In addition, the signal processing device calculates the elapsed time as a difference from the time information corresponding to the previous time also for the time information (step S3803). The signal processing device calculates the moving speed by dividing these differences (step S3804), and calculates the distance traveled by the train 603 during the time lag from the time lag required when actually acquiring coordinate information separately measured. The coordinate information obtained by calculating and adding only the moving distance is calculated (step S3805). Here, the time lag means, for example, the time required for image processing or the like when the image captured by the camera is subjected to image processing to acquire coordinate information, and the actual image acquisition and the coordinate information are recorded. It corresponds to the time until. Since the acquisition time of the channel information and the acquisition time of the coordinate information are not necessarily the same, the signal processing device converts the correspondence between the time and the coordinate from the relationship between the coordinate information, the time information, and the moving speed here. Expressions are extracted and managed (step S3806). However, when the time information and the coordinate information are recorded in association with each other in the form in which the time lag is eliminated in the above-described prior process, the correction process related to the time lag in step S3805 can be omitted.

  On the other hand, when the signal processing apparatus reads the combination of the channel information and the time information in parallel with the processing from step S3801 to step S3806 (step S3807), the signal processing apparatus estimates the coordinate information from the time information by the above conversion formula (step S3807). S3808). The signal processing apparatus calculates a channel matrix from the channel information (step S3809), and performs singular value decomposition on the channel matrix (step S3810). The signal processing apparatus records the first left singular vector obtained by the singular value decomposition in step S3810 in association with the coordinate information as the reception weight vector of the terminal station apparatus 604 (step S3811), and the first left obtained by the singular value decomposition. The right singular vector is recorded in association with the coordinate information as the transmission weight vector of the base station apparatus 600 (step S3812).

  Further, the signal processing apparatus performs calibration processing on the first left singular vector (reception weight vector) and the first right singular vector (transmission weight vector) calculated in step S3810 (step S3813). The signal processing apparatus records the vector obtained by the calibration process for the reception weight vector in association with the coordinate information as the transmission weight vector of the terminal station apparatus 604 (step S3814), and obtains the vector by the calibration process for the transmission weight vector. The vector is recorded as the reception weight vector of the base station apparatus 600 in association with the coordinate information (step S3815). The above processing is performed for each virtual transmission line corresponding to each first singular value related to the first signal processing units 601-1 to 601-8 on the base station device 600 side, and information recording is performed for each of them. Is done.

  In the processing so far, the processing for each virtual transmission path corresponding to each first singular value related to all the first signal processing units 601-1 to 601-8 on the base station device 600 side has been shown. In the situation where the distance between the base station apparatus 600 and the terminal station apparatus 604 changes every moment due to movement of the first signal processing unit 601-according to a specific distance as shown in the second embodiment. The interval from 1 to 601-8 cannot be optimized, and as a result, not all virtual transmission paths are generally orthogonal. Therefore, the signal processing apparatus selects an efficient part of the virtual transmission path among the slightly redundant number of first signal processing units 601-1 to 601-8 on the base station apparatus 600 side (step S3816). ). Data transmission between the base station apparatus 600 and the terminal station apparatus 604 is performed using a transmission / reception weight vector corresponding to the virtual transmission path selected in step S3816.

For example, when the target spatial multiplexing number is _NSDM , among the reception weight vectors on the terminal station device 604 side corresponding to the virtual transmission line related to the first signal processing unit 601 on the base station device 600 side, the correlation There a combination of small N _SDM number of receiving weight vector is selected from a plurality of receiving weight vectors recorded. This selection is performed, for example, by searching for a combination of N _SDM received weight vectors having a minimum correlation value from a plurality of recorded received weight vectors. As an example of this search method, for example, when N _SDM received weight vectors are selected, among the correlation values calculated for each combination of two received weight vectors included in the N _SDM received weight vectors The maximum correlation value may be a correlation value in a combination of N _SDM reception weight vectors, and a combination of reception weight vectors that minimizes the correlation value may be selected. The correlation value calculated for the combination of the two reception weight vectors here corresponds to the inner product value of the normalized vectors. Note Besides, for example, the correlation values in N _SDM number of receiving weight vectors, Shun power sum of the absolute value of the correlation value calculated for each combination of the two receiving weight vectors included in N _SDM number of receiving weight vector The combination that minimizes the power sum may be selected.

  In this way, a combination of reception weight vectors having a small correlation according to some criterion is selected, and the combination information (that is, this is included in all the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side). (Which corresponds to a combination of which first signal processing unit is selected and communicated) is also associated with the above-described transmission / reception weight vector, and the transmission / reception weight vector for the terminal station device 604 on the train 603 side and the base station device 600 side The transmission / reception weight vector and the combination information in the first signal processing units 601-1 to 601-8 to be used are recorded and managed. This record management interval is a constant value between minute coordinate differences in which the time variation of the transmission / reception weight vector associated with the channel time variation is negligible, and coordinate information and transmission / reception weight vector information each time a significant difference appears. May be recorded and managed.

In the above description, the number of utilization of the virtual transmission path, that is, the number of spatial multiplexing is fixed as N _SDM . However, this number is variable and the correlation value (standard) among all combinations of reception weight vectors. The maximum number of virtual transmission paths may be used in a range in which the maximum value of the inner product value of the converted vectors) falls below a predetermined threshold. Similarly, the maximum number of virtual transmission paths may be used in a range where the sum of powers of absolute values of correlation values falls below a predetermined threshold. In the case of a wideband system, the number of multiplexing for each subcarrier is not necessarily constant, so that it is possible to manage the number of multiplexing different for each subcarrier. In these cases, together with the information on the transmission / reception weight vectors that are made into a database, the number of multiplexed signals and which of the first signal processing units 601 the virtual transmission path to use is managed.

  In this case, the communication control circuit provided in the base station apparatus 600 refers to the weight vector / coordinate database 607 and instructs the first signal processing unit used for communication at each coordinate to transmit and receive signals for each coordinate. It is good also as a structure to perform. Further, the coordinate information used in the database here does not need to represent the two-dimensional or three-dimensional geographical coordinates, but may be one-dimensional like a distance from an arbitrary starting point on the track. However, any information may be used as long as it can be associated with the channel information.

  Further, although the previous coordinate information and the previous time information are required here, the moving speed information cannot be acquired for the data at the head of the database because there is no previous coordinate and time information. However, there is an overlap between the area covered by one base station apparatus 600 and the area covered by the adjacent base station apparatus 600, and if data acquisition is actually performed before the service area is switched, There is no problem if the information at the beginning of the recorded data is ignored. For example, in the case of the base station apparatus 600 including the station platform in the service area, there is no problem even if the leading data is discarded if the data is acquired from the station platform while it is stopped. Or, the moving speed can be obtained directly by measuring the number of rotations of the train wheel. In that case, the moving speed can be obtained directly without calculating the moving speed from the difference in coordinates and the difference in time It is also possible to record the obtained value together with the moving speed when measuring the channel information and use this value. Thus, by analyzing the acquired data offline, it becomes possible to process and prepare necessary information.

<When transmitting a training signal from the terminal station side>
FIG. 39 is a diagram showing an outline of another signal processing in a train moving cell using a plurality of virtual transmission lines corresponding to the first singular value in the fourth embodiment. The signal processing shown in FIG. 39 differs from the signal processing shown in FIG. 37 in that a training signal is transmitted from the terminal station apparatus 604 to the base station apparatus 600, the terminal station apparatus 604 acquires coordinate information, and the base station apparatus 600 The signal processing for recording channel information is shown.

  FIG. 39A is a flowchart of processing performed by the terminal station device 604. When the terminal station device 604 starts processing along with the operation of the train 603, the terminal station device 604 continuously transmits a training signal (step S3901). The terminal station device 604 ends the processing after continuously transmitting training signals while performing the channel information collection processing between the terminal station device 604 and the base station device 600 before starting the service operation.

  The terminal station device 604 performs the process of FIG. 39B in parallel with the process of FIG. When the terminal station device 604 starts the channel information collection process before starting the service operation, the terminal station device 604 acquires coordinate information and time information by some means (steps S3911 and S3912). The terminal station device 604 records data in which the acquired coordinate information and time information are associated with each other (step S3913), and ends the process.

  On the other hand, for example, when the base station apparatus 600 detects that the train 603 is present in the area of the base station apparatus 600, the base station apparatus 600 starts the process of FIG. 39C and receives the training signal transmitted from the terminal station apparatus 604 (Step S3921). The base station apparatus 600 also acquires time information together with the reception of the training signal (step S3922). Base station apparatus 600 performs channel estimation based on the training signal after reception signal processing similar to the reception signal processing in terminal station apparatus 604 described above (step S3923). The base station apparatus 600 records the obtained channel information and time information in association with each other (step S3924), and ends the process. Note that the presence of the train 603 in the area of the base station device 600 is detected using some method such as a method using operation information.

  These processes are simultaneously performed in parallel by all the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side. The time information may be acquired by each of the first signal processing units 601-1 to 601-8, or the entire base station apparatus 600 including a plurality of first signal processing units 601-1 to 601-8. It is also possible to provide a function for acquiring time information in common, and notify the time information to all the first signal processing units connected by wire. Or the channel information acquired by the plurality of first signal processing units 601-1 to 601-8 is aggregated to the base station device 600 via a wire, and the time information acquired by the base station device 600 and the plurality of first signal processings. The channel information from the sections 601-1 to 601-8 may be recorded and managed in association with each other.

  FIG. 40 is a diagram illustrating an outline of another offline signal processing performed after acquiring coordinate information and channel information in the fourth embodiment. Here, as in FIG. 39, a training signal is transmitted from the terminal station device 604 to the base station device 600 side, coordinate information is acquired on the terminal station device 604 side, and channel information is recorded on the base station device 600 side. An example of offline signal processing is shown. The basic processing shown in FIG. 40 is the same as the signal processing shown in FIG. 38. The processing from step S4001 to step S4010, the processing in step S4013, and the processing in step S4016 are each shown in FIG. This is the same process as the process from step S3801 to step S3810, the process of step S3813, and the process of step S3816. Here, overlapping description is omitted, and steps S4011, S4012, S4014, and S4015 will be described.

  The process shown in FIG. 40 is a process based on an uplink training signal, and the process shown in FIG. 38 is a process based on a downlink training signal. Because of this difference, the first left singular vector is the reception weight vector of the terminal station device 604 on the train 603 side in step S3811 in FIG. 38, whereas in step S4011 in the process shown in FIG. The vector is the reception weight vector of the base station apparatus 600. In step S3812, the first right singular vector is used as the transmission weight vector of base station apparatus 600, whereas in step S4012 of FIG. 40, the first right singular vector is used as the transmission weight vector of terminal station apparatus 604 on the train 603 side. It is said.

  Also, the handling of the vectors obtained by the calibration process for the first right singular vector and the first left singular vector obtained by the singular value decomposition in step S4010 of FIG. 40 is similarly different. In step S4014 of FIG. 40, the first right singular vector, that is, the vector obtained by the calibration process for the reception weight vector of base station apparatus 600 is used as the transmission weight vector of base station apparatus 600. Also, in step S4015 of FIG. 40, the first left singular vector, that is, the vector obtained by the calibration process for the transmission weight vector of the terminal station device 604 is used as the reception weight vector of the terminal station device 604.

  In the above description of FIGS. 38 and 40, the transmission weight vector and the reception weight vector on the base station apparatus 600 side and the transmission weight vector and the reception weight vector on the terminal station apparatus 604 side are shown in each figure by calibration processing. However, since the calculation is naturally performed in duplicate, for example, only FIG. 38 or FIG. 40 can be substituted, and the processing after step S3813 in FIG. 38, and FIG. The processing for obtaining the remaining transmission / reception weight vectors by calibration in the processing after step S4013 may be omitted.

  In the above description, since the first signal processing units 601-1 to 601-8 are physically independent on the base station apparatus 600 side, the transmission / reception weight vector is described. However, the terminal station apparatus 604 is described. Since the independent first signal processing units 601-1 to 601-8 are not present on the side like the base station apparatus 600, in the description, they are expressed as transmission / reception weight vectors. It is correctly expressed as a transmission / reception weight matrix composed of vectors, and is actually expressed as a weight matrix / coordinate database 606 of the terminal station device 604. However, since this is not essential in understanding the present embodiment, this point is expressed in accordance with the description before and after.

<Outline of signal processing during train operation>
FIG. 41 is a diagram illustrating an example of signal processing when a transmission / reception weight vector is acquired from coordinate information during service operation in the wireless communication system according to the fourth embodiment. The transmission / reception weight vector here requires both a transmission weight vector and a reception weight vector, and both the base station apparatus 600 side and the terminal station apparatus 604 side need to acquire transmission / reception weight vectors. Both processes are represented in common. Of course, the weight vector / coordinate database 607 of the base station apparatus 600 and the weight matrix / coordinate database 606 of the terminal station apparatus 604 used at this time are different, and the method for acquiring the coordinate information of the train 603 is the train 603 side. And the base station apparatus 600 are different. For example, when acquiring the coordinate information of the train 603 on the base station device 600 side, the position may be detected by providing a plurality of sensors beside the track as described above, or the coordinate information acquired by the train 603 side may be used. A configuration may be adopted in which transmission is performed by a wireless line (a different frequency band such as a macro cell or via a different system) set separately. Even if the coordinate information of the train 603 is acquired by any method, for example, if there is a difference in the time lag of information acquisition, the coordinate information at the time of information acquisition can be accurately estimated by adjusting the time lag. Even if a method is used, the present embodiment can essentially be used in the same manner.

  First, the current coordinate information of the train 603 is acquired by any method in either the terminal station device 604 or the base station device 600 on the train side 603 (step S4101). Also, time information at that time is acquired (step S4102). The difference between the previous coordinate information and the previous time information is calculated (steps S4103 and S4104), and the movement speed is calculated by dividing it (step S4105), and the current coordinates are corrected by taking into account the processing time lag. (Step S4106), and the corresponding transmission / reception weight vector is read from the weight matrix / coordinate database 606 or the weight vector / coordinate database 607 using the coordinate information as an argument (step S4107). The processes in steps S4101 to S4107 are repeated every time the coordinate information of the train is acquired or at a predetermined cycle, and the transmission / reception weight vector / matrix at that moment is always acquired.

  In parallel with the above processing, when signals are received, virtual transmission lines corresponding to a plurality of first singular values (a plurality of first signal processing units 601-1 to 601-8 of the base station apparatus 600). In step S4107, the reception weight vector of the uplink or downlink transmission path corresponding to any of the above is sequentially acquired (step S4111), and reception signal processing is performed on a symbol basis when receiving a signal (step S4112). For example, in the case of an OFDM signal, the signal on the frequency axis separated into each frequency component is multiplied by the reception weight vector for each frequency component (step S4113), and the reception signal on each virtual transmission path is extracted individually. . Here, signals from a plurality of virtual transmission lines are extracted. If these signal separations are necessary, a received signal on each virtual transmission line is used by using a training signal given to the head of a series of received signals. Channel estimation is performed, signal detection is performed by arbitrary signal detection processing using the channel matrix of the MIMO channel obtained by this channel estimation (step S4114), and if there is subsequent data, the above processing is repeated ( If the data is finished (step S4115: YES), the reception process is finished.

  Similarly, when transmitting a signal in parallel with the above processing, a virtual transmission path corresponding to a plurality of first singular values (corresponding to a plurality of first signal processing units of base station apparatus 600). The transmission weight vector read in step S4107 is sequentially acquired (step S4121), and the transmission weight vector is generated for each virtual transmission path along with the generation of the transmission signal to be transmitted on each virtual transmission path (step S4122). Multiplication is performed (step S4123), and transmission signal processing of the signal is further performed (step S4124). If transmission data remains, the processing is continued (step S4125: NO), and the processing ends when the data ends (step S4125). Step S4125: YES).

  The above description has been described collectively with the processing performed by the base station apparatus 600 and the terminal station apparatus 604, but the details are slightly different. For example, in the base station apparatus 600, signal processing related to one virtual transmission path is performed for each of the plurality of first signal processing units 601-1 to 601-8, but the terminal station apparatus 604 includes all the virtual transmission paths. Signal processing for signals is performed in parallel in the terminal station apparatus. For example, at the time of reception signal processing, the reception signal received by each antenna element is multiplied by a different reception weight vector, and a signal detection process of MIMO transmission is performed on a plurality of signal sequences obtained as a result. At the time of transmission signal processing, an individual transmission weight vector is multiplied for each signal sequence of each virtual transmission path. However, in the terminal station apparatus 604, a signal obtained by multiplying the transmission weight vector in the actual transmission signal processing is added and combined for each antenna element with signal components related to all virtual transmission paths, and the transmission signal processing is performed on the combined result. Will be given. In the base station apparatus 600, the first signal used for actual signal transmission / reception among the first signal processing units 601-1 to 601-8 installed redundantly compared to the actual spatial multiplexing number. The first signal processing unit which manages whether the processing unit is the first signal processing unit 601-1 to 601-8, and the communication control circuit or the like refers to the database in transmission or reception Is instructed to perform signal processing in each of the first signal processing units. Except for such slight differences, the base station apparatus 600 and the terminal station apparatus 604 can explain the signal processing flow with this processing flow.

(Selection of the first signal processing unit of a train moving cell using a plurality of virtual transmission lines corresponding to the first singular value)
Here, in the configuration of the wireless communication system of the present embodiment shown in FIG. 36, eight first signal processing units 601-1 to 601-8 are shown, but this is a spatial multiplexing of eight signal sequences simultaneously. Does not necessarily mean to do. For example, when the transmission weight vectors corresponding to the respective first signal processing units 601-1 to 601-8 recorded in the weight vector / coordinate database 606 on the train 603 side and the reception weight vectors show a high correlation. Means that communication between a certain first signal processing unit 601 and a terminal station device 604 interferes with communication between another first signal processing unit 601 and the terminal station device 604. In the example shown in FIG. 29 in the second embodiment, the optimum condition of the distance between the antenna elements (corresponding to the first signal processing unit) on the base station apparatus side is shown by design. A distance L between the station apparatus and the terminal station apparatus is included. That is, in the wireless communication system of the present embodiment in which the distance between the base station device and the terminal station device changes with the movement of the train 603, the first signal processing unit is always satisfied under the condition that satisfies the relationship of Expression (29). It is impossible to install 601.

Therefore, a situation may occur where the correlation between the transmission / reception weight vectors corresponding to the individual first signal processing units 601 becomes high. In such a case, since efficient spatial multiplexing cannot be performed, it is used when simultaneously performing spatial multiplexing transmission using a combination of transmission weight vectors or reception weight vectors having a low correlation among the first signal processing units 601. A predetermined number of first signal processing units 601 are selected. This selection can also be performed in advance before starting the service operation. For example, if the target spatial multiplexing number is 4, each transmission / reception weight in 70 patterns of ₈ C ₄ combinations A combination pattern that minimizes the maximum value of the inner product of vectors normalized by vectors (absolute values of six inner products are calculated with ₄ C ₂ ), or the sum of powers of the absolute value of the inner product is minimum Spatial multiplex transmission is performed using a transmission / reception weight vector having a combination pattern or a combination having a relatively low correlation. The combination pattern of the first signal processing unit 601 used in the combination is also recorded on the weight vector / coordinate database 607 side on the base station device 600 side, and the recorded combination pattern of the first signal processing unit 601 is used. Communication may be performed.

  Note that, since broadband communication is normally performed, the optimum combination pattern of the first signal processing unit 601 is different for each frequency component to be used. Therefore, this combination pattern of the first signal processing unit 601 is different for each subcarrier. It is good also as a structure using. Alternatively, the same combination pattern of the first signal processing units 601 may be used for all subcarriers. In this case, for example, the absolute value of the inner product in each subcarrier may be equal to the inner product of all subcarriers. A combination pattern of transmission / reception weight vectors that minimizes the maximum absolute value may be selected, or a combination pattern of transmission / reception weight vectors that minimizes the sum of all frequency components of the power value of the absolute value of each inner product is selected. May be.

(Configuration example of a base station device of a train moving cell using a plurality of virtual transmission lines corresponding to the first singular value)
FIG. 42 is a block diagram illustrating a configuration example of the base station apparatus 600 according to the fourth embodiment. The base station apparatus 600 includes an interface circuit 77, a MAC layer processing circuit 78, a second transmission signal processing unit 71, first transmission signal processing units 181-1 to 181-4, and a first reception signal processing. Units 185-1 to 185-4, a second received signal processing unit 75, a communication control circuit 614, a weight vector / coordinate database 607, a coordinate information acquisition circuit 612, and a time information acquisition circuit 613.

  Also in the wireless communication system of the fourth embodiment, in order to use spatial multiplexing using virtual transmission paths corresponding to a plurality of first singular values, the base station apparatus 600 is shown in FIG. The configuration is the same as that of the base station device 70 shown, but the following configuration is different. The base station apparatus 600 according to the fourth embodiment includes a weight vector / coordinate database 607, a coordinate information acquisition circuit 612, and a time information acquisition circuit 613, and a communication control circuit 614 instead of the communication control circuit 120. Is different from the base station apparatus 70.

  The weight vector / coordinate database 607 stores transmission / reception weight vectors associated with the coordinate information acquired as described above. When some of the plurality of first signal processing units 601 provided in the base station apparatus 600 are selected and used, information indicating the first signal processing unit 601 to be used in addition to the transmission / reception weight vector. Are associated with each other and stored in the weight vector / coordinate database 607. Each of the plurality of first signal processing units 601 includes one first transmission signal processing unit 181 and one first reception signal processing unit 185, respectively.

  The coordinate information acquisition circuit 612 acquires coordinate information indicating the position of the traveling train 603 by the above-described means. The time information acquisition circuit 613 acquires time information indicating the current time. The communication control circuit 614 reads a transmission / reception weight vector corresponding to the coordinate information acquired by the coordinate information acquisition circuit 612 from the weight vector / coordinate database 607. The communication control circuit 614 outputs the transmission weight vector among the read transmission / reception weight vectors to the first transmission signal processing units 181-1 to 181-4 via the second transmission signal processing unit 71. The communication control circuit 614 instructs the first transmission signal processing units 181-1 to 181-4 to transmit using the transmission weight vector. Note that the communication control circuit 614 may output the transmission weight vector directly to the first transmission signal processing units 181-1 to 181-4 without going through the second transmission signal processing unit 71. Further, the communication control circuit 614 outputs the reception weight vector among the read transmission / reception weight vectors to the first reception signal processing units 185-1 to 185-4 via the second reception signal processing unit 75. The communication control circuit 614 instructs the first reception signal processing units 185-1 to 185-4 to receive using the reception weight vector. Note that the communication control circuit 614 may output the reception weight vector directly to the first reception signal processing units 185-1 to 185-4 without going through the second reception signal processing unit 75.

  Therefore, the first transmission weight processing unit 130 included in the first transmission signal processing units 181-1 to 181-4 in the fourth embodiment performs the first transmission shown in FIG. 18 in the first embodiment. Like the first transmission weight processing unit 130 included in the signal processing unit 181, it is necessary to include the first channel information acquisition circuit 131, the first channel information storage circuit 132, and the first transmission weight calculation circuit 133. Instead, the transmission weight vector instructed from the communication control circuit 614 is output to the first transmission signal processing circuit 111. Similarly, the first reception weight processing unit 160 included in the first reception signal processing units 185-1 to 185-4 in the fourth embodiment is the same as the first reception weight processing unit 160 illustrated in FIG. 19 in the first embodiment. Unlike the first reception weight processing unit 160 provided in the reception signal processing unit 185, it is not necessary to include the first channel information estimation circuit 161 and the first reception weight calculation circuit 162, and the communication control circuit 614 instructs The received weight vector is output to the first received signal processing circuit 158.

  However, when channel information is acquired on the base station apparatus 600 side before the service operation is started (corresponding to FIG. 39), it is necessary to perform channel estimation on the received signal, so the first reception weight processing Unit 160 may be configured to include first channel information estimation circuit 161. Similarly, the first reception weight processing unit 160 may be configured to include a first reception weight calculation circuit 162 for calculating the reception weight vector. However, since these channel estimation and reception weight vector calculation can be performed by offline processing as pre-processing before the start of service operation, it is not always necessary to implement these. As described above, if the first reception weight processing unit 160 is mounted with only the first channel information estimation circuit 161, the first reception weight processing unit 160 is controlled by the first channel information estimation circuit 161. The estimated channel information is output to the communication control circuit 614. The communication control circuit 614 records the channel information acquired from the first reception weight processing unit 160 in association with the time information acquired in the process of step S3922 shown in FIG. In FIG. 42, it is assumed that the functions to be recorded are mounted in the communication control circuit 614 and are not explicitly shown.

  When the first reception weight processing unit 160 does not include the first channel information estimation circuit 161, the first reception weight processing unit 160 uses the acquired sampling value of the training signal for channel estimation. Output to the communication control circuit 614. The communication control circuit 614 may record the sampling information acquired from the first reception weight processing unit 160 in association with the time information acquired in the process of step S3922 shown in FIG. . In this case, for example, instead of the process of step S4007 in FIG. 40, the sampling data may be read and the channel estimation process may be performed.

  Note that the time information acquisition circuit 613 is necessary when acquiring channel information in advance before the start of service operation, and thus this function can be omitted during actual service operation. Further, since the train moving cell is basically Point-to-Point communication, the scheduling processing circuit 781 can be omitted. However, since the terminal station device 604 is mounted on a plurality of trains with respect to the base station device 600 that is fixedly installed, the information recorded in the weight vector / coordinate database 607 may differ slightly for each train. For this reason, in that case, the weight vector / coordinate database 607 records a database on all the trains that may pass through the location, and the scheduling processing circuit 781 refers to the operation information etc. to determine which train. It may be provided with a function of managing whether it is the timing to pass.

  In acquiring channel information before starting service operation, various information acquired in the communication control circuit 614 is recorded. Of course, such information is recorded in other circuits. The information may be read via the communication control circuit 614.

(Configuration example of a terminal unit of a train moving cell using a plurality of virtual transmission lines corresponding to the first singular value)
FIG. 43 is a block diagram illustrating a configuration example of the terminal station apparatus 604 according to the fourth embodiment. The terminal station device 604 includes an interface circuit 67, a MAC layer processing circuit 68, a transmission unit 61, a reception unit 65, a communication control circuit 624, a weight matrix / coordinate database 606, a coordinate information acquisition circuit 622, a time And an information acquisition circuit 623.

  Also in the wireless communication system of the fourth embodiment, the terminal station apparatus 604 is shown in FIG. 20 in the first embodiment in order to use spatial multiplexing using a virtual transmission path corresponding to a plurality of first singular values. However, the following configuration is different. The terminal station device 604 according to the fourth embodiment includes a weight matrix / coordinate database 606, a coordinate information acquisition circuit 622, and a time information acquisition circuit 623, and includes a communication control circuit 624 instead of the communication control circuit 121. Is different from the terminal station device 60.

  The weight matrix / coordinate database 606 stores the transmission / reception weight matrix (vector) associated with the coordinate information acquired as described above. In addition, regarding the base station apparatus 600, when some of the plurality of first signal processing units 601 provided in the base station apparatus 600 are selected and used, in addition to the transmission / reception weight vector, the first used Information indicating the signal processing unit 601 is associated with the coordinate information and stored in the weight matrix / coordinate database 606, but the terminal station device 604 selects the physically different first signal processing unit 601. Therefore, the vector information constituting the transmission / reception weight matrix includes which first signal processing unit 601 in the base station apparatus 600 is used, and which first signal processing is explicitly specified. It is not necessary to record whether the part 601 is used. However, the combination of the first signal processing unit 601 assumed to be used by the base station device 600 and the combination of the first signal processing unit 601 actually used in the transmission / reception weight matrix assumed to be used by the terminal station device 604 Must be consistent with each other.

  The coordinate information acquisition circuit 622 acquires coordinate information indicating the position of the running train 603 by the above-described means. The time information acquisition circuit 623 acquires time information indicating the current time. The communication control circuit 624 reads a transmission / reception weight matrix associated with the coordinate information acquired by the coordinate information acquisition circuit 622 from the weight matrix / coordinate database 606. The communication control circuit 624 outputs the read transmission weight matrix to the transmission unit 61 and causes the transmission unit 61 to perform transmission using the transmission weight matrix. The communication control circuit 624 outputs the read reception weight matrix to the reception unit 65 at the time of reception, and causes the reception unit 65 to perform reception using the reception weight matrix.

Therefore, the first transmission weight processing unit 140 included in the transmission unit 61 in the fourth embodiment is the same as the first transmission weight processing unit 140 included in the transmission unit 61 illustrated in FIG. 21 in the first embodiment. Similarly, it is not always necessary to provide the channel information acquisition circuit 141, the channel information storage circuit 142, and the first transmission weight calculation circuit 143 (if implemented, the channel information acquisition circuit 141, the channel information storage circuit 142, and the first transmission weight calculation circuit 143 are used only in the pre-processing before starting the service. The transmission weight vector (column vector constituting the transmission weight matrix) instructed from the communication control circuit 624 is output to the transmission signal processing circuits 811-1 to 811-N _SDM . Similarly, the first reception weight processing unit 44 provided in the reception unit 65 in the fourth embodiment is the first reception weight processing unit 144 provided in the reception unit 65 illustrated in FIG. 22 in the first embodiment. As described above, during service operation, the reception weight vector instructed from the communication control circuit 624 is output to the reception signal processing circuit 145.

  However, when the terminal station device 604 on the train 603 side acquires the channel information before starting the service operation (when the process shown in FIG. 37 is performed), the terminal station device 604 determines whether the channel information is based on the received training signal. Since it is necessary to perform estimation, the first reception weight processing unit 144 may be configured to include the first channel information estimation circuit 146. However, since these channel estimation and reception weight vector (matrix) calculation can be performed by offline processing, it is not always necessary to implement these. As described above, when the first reception weight processing unit 144 includes only the first channel information estimation circuit 146, the first reception weight processing unit 144 is estimated by the first channel information estimation circuit 146. The channel information is output to the communication control circuit 624. The communication control circuit 624 records the channel information acquired from the first reception weight processing unit 144 in association with the time information acquired in the process of step S3722 shown in FIG. In FIG. 43, it is assumed that the functions to be recorded are mounted in the communication control circuit 624, and are not explicitly shown. In the first embodiment, the above-described configuration example of the reception unit 65 of the terminal station device 60 illustrated in FIG. 23 and the configuration example of the transmission unit 61 of the terminal station device 60 illustrated in FIG. Except for the difference, the same configuration can be used.

  As described above, when traffic such as information terminals used by users who ride on the train 603 is aggregated to try to offload from the macro cell at the wireless entrance, the terminal station device 604 provided in the train 603 moving at high speed. And channel time fluctuation between the base station apparatus 600 and the base station apparatus 600 becomes a problem. Therefore, in the wireless communication system according to the fourth embodiment, in an environment in which the arrival direction of radio waves and the movement direction of the train 603 are substantially aligned, the time variation of the transmission path corresponding to the first singular value tends to be small. To use. Specifically, a plurality of first signal processing units 601 are installed on an overhead pole, and on the other hand, a relationship between coordinate information and channel information is acquired in advance and converted into a database, and a weight matrix / coordinate database 606 and weight vector / It is recorded in the coordinate database 607 and used. The terminal station device 604 provided in the train 603 sequentially acquires its own coordinates, and reads out the transmission / reception weight matrix from the weight matrix / coordinate database 606 as an argument, thereby enabling stable communication and the communication line Large capacity can be made possible. Similarly, the base station apparatus 600 acquires the coordinate information of the train 603 sequentially, and reads out the transmission / reception weight vector used by the first signal processing unit 601 from the weight vector / coordinate database 607 using this as an argument. Communication can be performed and the capacity of the communication line can be increased.

  In the wireless communication system of the present embodiment, each of the first transmission signal processing units 181 multiplies a baseband signal obtained by performing modulation processing in the second transmission signal processing unit 71 by a transmission weight vector. To generate signal vector components. A signal vector composed of signal components generated in each of the first transmission signal processing units 181 is transmitted from an antenna element provided in the first transmission signal processing unit 181. A signal vector component transmitted from each antenna element is a signal vector component obtained by multiplication with a transmission weight vector corresponding to the antenna element.

  In the wireless communication system of the present embodiment, each of the first reception signal processing units 185 acquires a baseband signal vector from reception signals received by a plurality of antenna elements provided, and acquires the acquired baseband signal vector and reception weight. One signal sequence is obtained by multiplying the vector. The second received signal processing unit 75 reproduces the data transmitted from the terminal station apparatus from each of the one signal series acquired by each first received signal processing unit 185.

  In the wireless communication system of the fourth embodiment, the base station device 600 spatially multiplex-transmits a plurality of data sequences # 1 to #L with the terminal station device 604. May be transmitted to and received from the terminal station device 604.

  Further, when the antenna element provided in the first transmission signal processing unit 181 is a single polarization antenna element, the first transmission signal processing unit 181 may transmit one signal sequence. . Further, when the antenna element provided in the first transmission signal processing unit 181 is a plurality of types of polarization antenna elements and a plurality of types of polarization antenna elements are used at the same time, the first transmission signal processing unit 181 includes one system. Alternatively, a signal sequence of up to two systems may be transmitted. Further, when the antenna element provided in the first reception signal processing unit 185 is a single polarization antenna element, the first reception signal processing unit 185 acquires only one system of baseband signals from the reception signal. You may do it. In addition, when the antenna element provided in the first reception signal processing unit 185 is a plurality of types of polarization antenna elements and a plurality of types of polarization antenna elements are used at the same time, the first reception signal processing unit 185 includes one system. You may make it acquire the baseband signal only to 2 systems.

[Fifth Embodiment]
[About time-axis beamforming]
(Overview of basic principles)
First, consider two antenna elements (a first antenna element and a second antenna element) with an interval of d [m]. If the transmission signal from the communication partner station has a very strong directivity and is a line-of-sight environment, the signals arriving at the two first and second antenna elements can be approximated by plane waves, The received signal at time t of the second antenna element is represented by φ ₁ (t) and φ ₂ (t). If the signal component of the k-th frequency component at time t is S _k (t), the distance between the transmitting station and the first antenna element is L, the center frequency is f _c , the frequency bandwidth is W, and f _c is the center − If the frequency of the k-th frequency component between W / 2 and + W / 2 is f _k , and the wavelength of the k-th frequency component in the radio frequency is λ _k , the received signal φ ₁ (t) is expressed by Expression (38). be able to.

Here, the direction of arrival and an angle theta, (the c here the speed of light) the received signal phi _{2 (t)} is the received signal phi _{1 (t)} is (dsin) / c in the case where it can approximate plane wave only delayed Therefore, the received signal φ ₂ (t) can be approximated as shown in Equation (39).

  Here, Δt represents a sampling interval Δt [s] (= 1 / W). The coefficient α is expressed by Expression (40) using the signal bandwidth W [Hz], the arrival direction angle θ, the antenna element interval d [m], and the speed of light c [m / s].

When this equation (38) is compared with equation (39), each frequency component k in equation (39 ⁾ is multiplied by a weight (coefficient) w _k ^{(freq−domain)} on the frequency axis shown in equation (41). Yes.

Since this coefficient is a coefficient on the frequency axis, when converted to a coefficient on the time axis by IFFT, the weight (coefficient) of the nth sample (0 ≦ n ≦ N _FFT −1, where N _FFT is the number of FFT points) on the time axis w _n ^{(time-domain)} is expressed by Expression (42).

Here, the ratio of the sum of the square of the absolute value of the first n = 0 time axis weight and the square of the absolute value of the weight from n = 1 to N _FFT −1 is expressed in [dB]. Is expressed by the following equation (43) as a function F (α) of the coefficient α.

  FIG. 44 is a graph showing a function F (α) in the fifth embodiment. In the figure, the vertical axis indicates the value of the function F (α) and the relative power ratio, and the horizontal axis indicates the value of α. The function F (α) corresponds to the ratio of the received power of the preceding wave and the received power of all the delayed waves other than the preceding wave. From the figure, it can be seen that when α is sufficiently small, one component of the preceding wave occupies most of the entire power, and conversely, when α increases, the delayed wave component increases.

As can be seen from the equation (40), although depending on the bandwidth W, generally, if the antenna element interval d is sufficiently narrow, α becomes a small value. Therefore, the reception signal can be reduced by sufficiently narrowing the antenna element interval. An approximate value of φ ₂ (t) can be obtained by multiplying the value of the received signal φ ₁ (t) by the coefficient of the preceding wave component. In other words, since the weight w _k ^{(freq-domain)} on the frequency axis is different for each frequency component, conventionally, the multiplication of the weight is performed after converting the signal on the time axis into the signal on the frequency axis by FFT. It was. However, in FIG. 44, if the design is made such that the coefficient α is reduced by narrowing the antenna element interval so that the function F (α) becomes a predetermined value (for example, 30 [dB]) or more, the predetermined coefficient is applied to the time-axis signal. it is possible to calculate as only the receiving signal phi _{1 (t)} from the received signal phi _{2 (t)} the equation multiplied (44).

Here, the relationship between the received signal φ ₁ (t) and the received signal φ ₂ (t) is shown. However, if N antenna elements are installed at intervals d in a linear array, the m-th antenna element The received signal φ _m (t) can be approximated by Expression (45) using the received signal φ _m-1 (t) of the adjacent (m−1) th antenna element.

That is, the reception signal φ _m (t) of the _m- th antenna element can be approximated by Expression (46) using the reception signal φ ₁ (t) of the first antenna element.

However, _cm in Expression (46) is given by Expression (47).

  Here, the mathematical meaning of the above description is organized using Equation (41) and Equation (43). First, the situation where energy concentrates on one time axis component when IFFT, that is, inverse Fourier transform is performed, means that a delta function behavior is seen on the time axis. It is known that when the delta function is Fourier transformed, it becomes a constant on the frequency axis. On the other hand, if a component that appears constant on the frequency axis is IFFT, mathematical description can be made with a single time axis component with high accuracy on the time axis.

When the first half of the right side of equation (41) is a constant that does not depend on the frequency components, focusing on exponent of the natural logarithm of the second half, and has a -2πjf _k αΔt. Since the frequency f _k of the k-th frequency component takes a value between −W / 2 and + W / 2 and the sampling interval Δt is 1 / W, f _k Δt takes a value between −1/2 and 1/2. take. The complex phase of Exp (−2πjf _k αΔt) is from −απ to + απ. If α is sufficiently small, the complex phase is approximately 0, and Exp (−2πjf _k αΔt) can be approximated by a constant of 1. That is, if α is sufficiently small, w _k ^{(freq-domain)} has negligible frequency dependence. If α is large to some extent, the frequency dependence cannot be ignored, and the frequency component obtained after the inverse Fourier transform deviates from the behavior of the delta function type, and the delayed wave component cannot be ignored.

Here, in order to discuss the value of the coefficient α quantitatively, a frequency bandwidth of 1 GHz is assumed in the 80 GHz band. The wavelength is 3.75 mm, antenna elements are arranged at intervals of 5 mm, and d = 0.005. Assuming that the angle range of the incoming wave is ± 10 degrees, the value of α is 0.005 × sin (10 °) × 10 ⁹ / (3 × 10 ⁸ ) = 0.002894 from Equation (40). This is 1.04 degrees in angle, and it can be seen that this is a constant having almost no frequency dependence. That is, it can be seen that if the frequency dependence is small, the value of the function F (α) in Equation (43) becomes a very large value and is more like a delta function.

Thus, in a situation where the coefficient α is sufficiently small and Exp (−2πjf _k αΔt) can be approximated by 1, Equation (42) becomes a non-zero value only when n = 0, Can be approximated in the same way.

  When this is substituted into equation (47), the coefficient of the m-th antenna element is represented by equation (49).

  In this way, if the complex phase relationship in the received signals of each of the N antenna elements can be acquired, the reception weight for combining the received signals of the respective antenna elements can be determined as in Expression (50).

  Note that the reception weight vector of this synthesis is the weight of the maximum ratio synthesis, but since it is expected that the reception level is almost the same level in the line-of-sight environment, it is the weight of the equal gain synthesis equation (51). May be.

That is, the signal from the target transmission antenna element is extracted by multiplying the reception signal vector of each antenna element by the reception weight vector of (~ w ₁ , ~ w ₀ , ..., ~ w _N ). Is possible.

  Here, α in Expression (40) is a coefficient depending on the path length difference, and if it is a linear array, it can be expressed in the form of Expression (49). If the geometric feature is utilized, such a reception weight can be described in the same procedure.

The coefficient _cm can be directly obtained in addition to the expression (47). In particular, in the case of other than linear array, even if the characteristics of the geometric arrangement are not reflected in the mathematical formula, the periodic training signal transmitted from the transmitting station side is acquired for one period, and the training signal Based on the cross-correlation, it is also possible to calculate as in Expression (52). However, j = m (j and m are values indicating antenna element numbers, and are expressed as j instead of m).

Here, S _j (n) represents the nth sampling value of the jth antenna element. In this case, since the signal processing is performed on the signal from one antenna element to one antenna element, the value for one symbol is not used as it is for the line gain, but the average value of many times is used. It is desirable to increase the SNR value. For example, if 100 symbols are averaged, an improvement in the SNR characteristic of 20 dB can be expected. In this averaging, it is also preferable to use relative channel information based on the complex phase of the reference antenna element. By the way, the influence of phase noise becomes a problem in high frequency bands such as millimeter waves, but since the degree of phase fluctuation can be expected to have some degree of randomness, the averaging process here also suppressed the influence of phase noise. A correlation value in the form can be obtained.

  Here, the above-described training signal may be a signal with an OFDM guard interval or a continuous signal without a guard interval. In order to acquire absolute channel information, it is necessary to accurately acquire the symbol timing and measure the channel. However, in order to calculate the transmission / reception weight, the complex phase is different from the reference antenna element. This is because it is sufficient to know relatively what kind of relationship it is. In addition, other training signals can be used in the same manner as long as the signals have low autocorrelation in the time direction.

  Based on the coefficient obtained in this way, the value of the reception weight may be obtained by Expression (50) or Expression (51). For the reception weight obtained in this way, for example, as a reception signal process, a time series signal sequence ^ S (n) of one system may be calculated by Expression (53).

Here, in formula (53), ˜w ₁ is 1, and in formula (53), ˜w _j S _j (n) from j = 2 to N ′ _BS-Ant is added to S ₁ (n). It is carried out. The signal sequence ^ S (n) obtained in this way is regarded as a signal received by a single virtual receiving antenna for receiving a virtual transmission line signal corresponding to the first singular value. The received signal processing may be performed.

  In general, if a narrow band signal is synthesized by adding a complex phase difference of 2π (d · sin θ / c) / λ to the direction of angle θ, directivity can be directed in the direction of angle θ. However, for a broadband signal having a bandwidth of 1 GHz, it is necessary to change the complex phase difference for each frequency component. Further, in order to achieve signal separation with a sharp directivity with a narrow beam width, it is preferable to widen the antenna element interval. In this case, however, the complex phase difference for each frequency component is increased (that is, the frequency dependence is increased). It showed up as an effect. For this reason, in general, communication is performed in such a way that the antenna element interval is widened to widen the antenna aperture length, and the directivity is stably formed over a wide band with a different weight for each frequency component while narrowing the directional beam. I was aiming for that. However, if the directivity can be reduced by making the number of antenna elements enormous, the antenna aperture length is rather narrowed, and the delay other than the preceding wave is determined by the evaluation function F (α) shown in Expression (43). If the antenna element interval and the range of the direction θ of the incoming wave are limited so that the contribution of the wave component can be determined to be negligible (that is, the value of α is limited), different antenna elements can be used for wideband signals. By multiplying the received signal by a predetermined coefficient, a signal coming from a predetermined direction is efficiently extracted, and signal separation at the time of spatial multiplexing can be realized on the time axis using the coefficient.

  For example, when spatial multiplexing is performed using a plurality of virtual transmission paths corresponding to a plurality of first singular values as shown in FIG. 16 in the first embodiment, the equations (49) to (52) are used. When the coefficient corresponding to is calculated in correspondence with each individual virtual transmission line and the weight vector is calculated using the equations (50) to (51), the multiple signal sequences are spatially multiplexed. It becomes possible to calculate the reception weight vector on the time axis for the first stage signal separation.

In addition, the calibration process similar to the calibration process on the frequency axis can be performed on the time axis, and if each received weight vector is multiplied by the calibration coefficient on the time axis, the time axis for transmission is similarly transmitted. It is also possible to obtain a transmission weight vector.

(Outline of residual interference component removal method)
As described above, it is possible to extract a signal of a virtual transmission line corresponding to the target first singular value. However, between virtual transmission lines that are not completely orthogonal to each other, Although it is weak, an interference signal leaks. Hereinafter, a method for removing the interference signal will be described. The following interference signal removal method corresponds to signal processing in the second received signal processing unit 75 in the base station apparatus. Also in the description of the first embodiment described above, the interference signal that cannot be completely removed by the first received signal processing unit 185 is removed in two stages using the second received signal processing unit 75. Is equivalent to the signal processing.

  In the following description, for example, as shown in FIG. 16 in the first embodiment, a plurality of first signal processing units including a plurality of antenna element groups are mounted on the base station apparatus side, and the terminal station apparatus side Is assumed to have a single antenna element group composed of a plurality of antenna elements. FIG. 45 is a diagram for explaining an interference signal removal method according to the fifth embodiment. As shown in the figure, the wireless communication system in the fifth embodiment includes a base station device 204 and a terminal station device 206. The base station apparatus 204 includes a plurality of first signal processing units, and the first signal processing unit includes a plurality of antenna elements. The plurality of antenna elements provided in each first signal processing unit form virtual antenna elements 201, 202, and 203, respectively. The terminal station device 206 includes an antenna element group 205 formed by a plurality of antenna elements. In the wireless communication system according to the fifth embodiment, the communication path between the base station apparatus 204 and the terminal station apparatus 206 communicates between the virtual antenna elements 201, 202, 203 and the antenna element group 205. Can be seen as a system.

FIG. 45A is a diagram illustrating interference components generated in the downlink in the wireless communication system according to the fifth embodiment. A directional beam (transmission beam) directed to the terminal station device 206 formed by the virtual antenna element 201 of the base station device 204 is directed to the virtual antenna element 201 by the antenna element group 205 of the terminal station device 206. Consider the case of receiving with a directional beam (receive beam). In this case, the signal ψ transmitted from the virtual antenna element 202 in addition to the signal ψ ₁ transmitted from the virtual antenna element 201 to the reception beam directed to the virtual antenna element 201 of the antenna element group 205. Consider that when a part of ₂ leaks, a part of the signal ψ ₂ is removed as an interference component. This is equivalent to removing the interference component in the uplink shown in FIG. 45B, and can be processed in the same way in terms of signal processing. In the uplink, when the signal ψ ₁ is transmitted with a transmission beam formed in the antenna element group 205 of the terminal station device 206 and directed to the virtual antenna element 201, the virtual antenna element 201 is A part of the signal ψ ₂ transmitted from the antenna element group 205 toward the virtual antenna element 202 leaks into the reception beam toward the antenna element group 205 of the terminal station device 206, and this leaked signal ψ ₂ Is equivalent to removing a part of the signal as an interference component.

First, taking the downlink as an example, for the incoming wave from the angle θ direction toward the virtual antenna element 201 in the antenna element group 205, the signal is expressed by the reception weight vector represented by the equations (48) to (52). Wait. Here, the signal ψ ₁ from the virtual antenna element 201 is awaited, but here the signal ψ ₂ from the virtual antenna element 202 in the direction of the angle θ ′ leaks. At this time, if the coefficient β is obtained by substituting θ ′ into θ in the equation (40), the coefficient α in the equation (49) is replaced with the coefficient β for each antenna element of the antenna element group 205 of the terminal station device 206. The signal ψ ₂ is received with the given coefficient. However, each element of the reception weight vector to be multiplied when the antenna element group 205 of the terminal station device 206 is waiting with a directional beam directed to the virtual antenna element 201 is an expression in which the coefficient α is not replaced with the coefficient β. This is a reception weight represented by Expression (50) taking the complex conjugate of (49). For this reason, the complex phase is not originally canceled to 0 by multiplication of Expression (49) and Expression (50) by multiplication of the reception weight. When the sum is obtained by multiplying the coefficient Exp {2πjf _c (α−β) × (m−1) Δt} in the m-th antenna element constituting the antenna element group 205, the respective complex phases are synthesized with different complex phases. Will be. Note that m = 1, 2,..., K, and K is the number of antenna elements N _MT-Ant constituting the antenna element group 205 provided in the terminal station device 206.

However, although it has been described here as a combination of different complex phases, since it is actually a regular combination, it satisfies the “antenna arrangement condition for orthogonalization of line-of-sight MIMO transmission” described in the second embodiment. In the situation, the summed value converges to approximately zero. Incidentally, when the reception weight represented by the equation (50) or the equation (51) is applied to the signal ψ ₁ from the virtual antenna element 201, the signals received by all the antenna elements constituting the antenna element group 205 are represented. The complex phase has a constant value. As a result, signals can be added and synthesized in the same phase, the amplitude of the signal is N _MT-Ant (the number of antenna elements), and a large gain is obtained.

By such a procedure, when directivity is directed to the jth virtual antenna element of the base station apparatus 204 on the receiving side, how the signal from the ith virtual antenna element of the base station apparatus 204 is received. You just have to figure out. In the above-described example, a signal indicating how the signal ψ ₂ transmitted from the virtual antenna element 202 leaks in a state where it waits with a directional beam directed to the virtual antenna element 201 is expressed by Equation (49). Is replaced by (α−β) and the sum is obtained for all m. Since this sum is given as the sum of the geometric series as in the left side of the formula (54), it is expressed by the following formula by the formula of the sum of the geometric series.

Here, when f _c (α−β) N _MT-Ant Δt is an integer, the sum of the geometric series becomes zero and no residual interference exists, but in general, residual interference remains. Therefore, suppression of the interference component is necessary.

On the other hand, ψ ₁ (t) is transmitted with a directional beam directed from the virtual antenna element 201 to the antenna element group 205, and is waited with a directional beam directed from the antenna element group 205 to the virtual antenna element 201. The signal ^ ψ ₁ (t) received in the state indicates the relationship between the signal ψ ₁ (t) transmitted from the antenna element group 205 toward the first virtual antenna element and the first virtual antenna element. The desired signal component is included as ^ h ₁₁ × ψ ₁ (t) using the indicated coefficient ^ h ₁₁ . Similarly, a signal ψ _j (t) transmitted from the j-th virtual antenna element and a signal ^ ψ _i (t) received by the reception beam from the i-th virtual antenna element toward the antenna element group 205. It can be expressed that the component of the signal ψ _i (t) includes only ^ h _ij × ψ ₁ (t) using the coefficient ^ h _ij indicating the relationship between the two.

However, the point to be noted here is that when the weight on the frequency axis in Expression (41) has no frequency dependence, only the time axis weight in Expression (42) is n = 0 and all others are non-zero. Since it becomes zero, the above-described description can be made. However, when the frequency dependence remains in the weight of the equation (41), n = 1 of the time axis weight of the equation (42) and the subsequent terms are also included. It has a value and, as a result, cannot be expressed in such a simple expression. Therefore, if this signal is converted into a signal on the frequency axis and understood, generally, the coefficient ^ h _{ij, the} signal ψ ₁ (t) and the signal ψ ψ _i (t) are decomposed into frequency components, With respect to the k-th subcarrier, the signal ^ ψ _i ^(k) (t) is converted into the signal ^ ψ _i ^(k) (t) using the coefficient ^ h _ij ^(k) , the signal ψ ₁ ^(k) (t), and the signal ψψ _i ^(k) (t). Needs to be treated as if the component of the signal ψ ₁ ^(k) (t) is included in ^ _ij ^(k) × ψ ₁ ^(k) (t).

However, if the weight on the frequency axis of equation (41) has almost no frequency dependence, n = 1 of the time axis weight of equation (42) and the subsequent terms approximate to 0 with high accuracy. In this case, the signal processing on the time axis can be performed for the subsequent processing. As described above, whether the signal processing on the frequency axis is necessary or can be dealt with by simple signal processing on the time axis depends on the frequency dependence of the weight on the frequency axis in the equation (41). Yes, depending on various parameters of the target system. Therefore, when the frequency dependence of all coefficients ^ h _ij can be ignored in terms of system design and each signal can be separated by signal processing between sampling values on the time axis, FIG. The communication state shown can be expressed by equation (55). Here, M represents the number of spatial multiplexing (the number of virtual transmission lines) for convenience of description.

Since this is the same format as the conventional MIMO signal processing on the frequency axis, for example, a signal vector (^ ψ ₁₎ that is observed with some interference signals leaked by an inverse matrix of this channel matrix or linear processing by MMSE. _{(t), ^ ψ 2 (} t), ···, ^ ψ M (t)) from the cancellation signal vector interference components _{(ψ 1 (t), ψ} 2 (t), ···, ψ M (T)) can be obtained. In this manner, signals between a plurality of virtual transmission lines can be separated.

  What is important here is the intention of this equation, and the time t in equation (55) can be applied to any time, and if this relational expression is understood at any time as an instantaneous value, This means that this relationship is established between the sampling values at the same time with respect to the sampling values. That is, if sampling is performed, it is possible to perform signal separation with the sampling value without performing FFT or the like.

  For this reason, in the above description, the description has been made assuming that the OFDM modulation method is mainly applied. However, a training signal is added to the head of the transmission signal, and Equation (55) is obtained by some method using this training signal. If the corresponding channel matrix can be acquired in advance, it is possible to perform signal separation on the time axis without performing FFT, and in the case of performing single carrier transmission, directly to the signal separated by Equation (55), It is possible to perform signal processing for single carrier transmission.

  In general, in the case of a high frequency band such as millimeter wave, phase noise cannot be ignored. When a wideband OFDM signal is used, the phase noise component breaks the orthogonality between OFDM subcarriers by FFT processing, It is expected that the frequency component is dispersed as noise and crosstalk between subcarriers, making it difficult to correct phase noise. With single carrier transmission, it is possible to estimate and compensate for the phase noise component from the relationship between the signal detected for each symbol and the received signal, and it is possible to cope with relatively large phase noise. However, in the prior art, when multiple series of single-carrier transmission signals are combined in a spatial MIMO channel, the signals cannot be separated unless they are first converted to a frequency signal. The implementation of FFT was indispensable. However, since the phase noise is converted into a state that cannot be compensated by performing the FFT, a signal separation technique that can be realized without performing the FFT is necessary for the phase noise compensation.

  For the MIMO channel on the time axis of Equation (55), in addition to the theoretical analysis calculation as shown in Equation (54), for example, using a training signal for one period equivalent to one OFDM symbol, It is also possible to directly obtain each component of the channel matrix. In the signal processing when multiplying the reception signal vector received by a plurality of antenna elements by the reception weight vector, the SNR of the original signal is low, and thus a device such as averaging to some extent is required. However, since the signal obtained by multiplying the reception weight vector for extracting the signal of the virtual transmission line corresponding to the first singular value has already been improved in SNR due to the directivity gain, the reception weight vector is obtained therefrom. When performing the operation of calculating the value by correlation calculation, it is not always necessary to perform a plurality of averaging processes. For example, there is no problem even if it is acquired by a single process using a training signal for one period corresponding to one OFDM symbol. .

In this equation, N _{FFT FFT} sample points are used. However, since FFT is not necessarily used, N _{FFT in} equation (56) does not necessarily have to be the number of OFDM FFT points, and is an arbitrary value. It is also possible.

  The approximation condition here is that the virtual transmission line corresponding to the first singular value is assumed that the interference component is largely suppressed by directivity control of both transmission and reception, in other words, the pair of MIMO channel matrices. When the absolute value of the corner component is relatively large compared to the absolute value of the off-diagonal component, approximation can be performed with high accuracy.

Using the coefficient ^ h _ij obtained in this way, the signal separation of the time axis signal is performed by performing an operation such as multiplying both sides by a linear weight by the ZF method or the MMSE method on the channel matrix of Equation (55). If, for example, single carrier signal processing is applied to the resulting time axis signal, phase noise is a problem, using existing single carrier phase noise countermeasures with high accuracy. Signal detection processing can be performed.

  In the above description, when the frequency dependence of the weight on the frequency axis in Expression (41) is almost negligible, that is, n = 1 of the time axis weight in Expression (42) and the subsequent terms are 0 with high accuracy. However, when such a condition is not satisfied, the FFT processing is straightforwardly performed, and signal separation processing is individually performed on the frequency axis.

(Time axis beam forming process flow)
FIG. 46 is a flowchart illustrating processing for acquiring transmission / reception weights for time-axis beamforming in the wireless communication system according to the fifth embodiment. The flowchart shown in FIG. 46A is a flowchart of processing for acquiring the transmission / reception weight of the first stage of time-axis beamforming. Here, a case will be described in which a correlation coefficient as shown in Expression (52) is used instead of a method for obtaining a coefficient on the time axis from various parameters as shown in Expression (49). However, these values differ only in their weight calculation methods, and analytical methods can be similarly applied by replacing the calculation methods. Also, the acquisition of the transmission / reception weight vector is necessary on the base station apparatus side and the terminal station apparatus side as well, and the processing described here is performed in both directions. Hereinafter, processing performed in the downlink will be described.

  First, the base station apparatus transmits a training signal for channel estimation (step S4601). The training signal here is a signal for channel estimation, and does not depend on the transmission method used in data communication after channel estimation. For example, a continuous OFDM signal without a guard interval may be used, and there is little autocorrelation. A training signal having a predetermined periodicity may be used. Of course, it is also possible to use the training signal used in the transmission method used in data communication as it is.

When the terminal station apparatus receives the training signal transmitted from the base station apparatus by each antenna element (step S4611), the terminal station apparatus individually performs down-conversion processing from the radio frequency to the baseband, and performs A / D conversion. Record the sampling value after. This training signal is acquired at a predetermined period (for example, 100 periods), and the signals of each period are added and synthesized in accordance with the periodicity and averaged (step S4612). If the averaged sampling value of the n-th sample of the j-th antenna element is S _j (n), the correlation coefficient c _j of the j-th antenna element with respect to the first antenna element is obtained by Expression (52) (step S4613). ).

The reception weight of the j-th antenna element is calculated by taking the complex conjugate of the correlation coefficient c _j obtained in step S4613 (or further dividing that value by the absolute value), and combining these to obtain the reception weight vector Is obtained (step S4614). In particular, when spatial multiplexing is not performed, the reception weight vector is simply calculated in this manner on the time axis for directivity formation. However, when a plurality of systems of signals are spatially multiplexed as in FIG. 16 shown in the first embodiment, the above-described reception weight vectors for the first signal processing units 304-1 to 304-4 in FIG. Is calculated.

  If the calibration coefficient when performing implicit feedback does not have frequency dependence, use a common calibration coefficient in the frequency direction, perform calibration processing on the received weight vector on the time axis, A transmission weight vector is acquired (step S4615). The acquired transmission / reception weight vector is recorded and managed (step S4616), and the transmission / reception weight vector is used for subsequent signal transmission / reception. If the training signal can be received with a high SNR by any method, the averaging process can be omitted. Even if the calibration coefficient is frequency-dependent, if the deviation for each frequency is relatively small, the average value of the calibration coefficients for all subcarriers is regarded as the calibration coefficient common to all subcarriers. Processing may be performed. In this manner, the first-stage reception weight vector and transmission weight vector for extracting the virtual transmission line corresponding to each first singular value can be obtained.

  FIG. 46B is a diagram for obtaining a second-stage (transmission) reception weight matrix for removing residual interference with respect to a signal of each virtual transmission path received with increased directivity gain as described above. It is a flowchart which shows the process of. First, the base station apparatus transmits a training signal using the transmission weight vector obtained by the process shown in FIG. 46A (step S4621). The training signal here is a signal for performing channel estimation and may not depend on a transmission method used in data communication after channel estimation.

  When the terminal station apparatus receives the training signal transmitted from the base station apparatus by each antenna element (step S4631), the terminal station apparatus individually performs down-conversion processing from the radio frequency to the baseband, and performs A / D conversion. The sampling value is acquired by The reception signal vector at each antenna element is multiplied by the reception weight vector (step S4632). The cross-correlation of the training signal for each received weight vector (that is, for each virtual transmission path corresponding to the first singular value) acquired in this way is acquired by Expression (56) (step S4633), and a channel matrix is acquired. (Step S4634). The terminal station apparatus calculates a reception weight matrix based on the acquired channel matrix (step S4635), and records and manages the calculated reception weight matrix as a second-stage time axis weight matrix (step S4636).

  In the case of a wireless entrance where both the base station device and the terminal station device are fixedly installed, the weights of the first stage and the second stage hardly change with time. If you obtain it in advance, you can continue to use it. On the other hand, for example, in a train moving cell, the channel is time-varying at high speed. However, the first-stage received weight vector is acquired in advance and stored in a database, so that the first-stage There is no need to calculate the eye transmit / receive weight vector. However, the second-stage received weight matrix (vector) is for removing interference signals remaining in different states depending on the incompleteness of the first-stage received weight vector. These need to be updated sequentially. In this case, the processing from step S4631 to step S4636 may be performed every time a wireless packet signal is received.

  In addition, when a plurality of virtual transmission lines corresponding to the first singular value are used in the access system, not only the second-stage reception weight matrix (vector) but also the first-stage transmission / reception weight vector is updated sequentially. There is a need to. In the frame configuration shown in FIG. 65 described in the eighth embodiment described later, in the example shown in the eighth embodiment described later as a slot for accommodating a training signal for acquiring a transmission / reception weight vector therefor, The first slot of the frame and four slots from the end of the frame are used. However, the training signal for calculating the second-stage reception weight matrix can be arranged in the above-mentioned slots, but the sub-slots of subdivided sub-slots in the middle of the slots other than these slots. It is also possible to arrange at the head (that is, the head of the radio packet) and calculate the second reception weight matrix based on this.

(Time-axis beamforming circuit configuration)
FIG. 47 is a diagram illustrating a configuration example of the first transmission signal processing unit 381-j provided in the base station apparatus in the fifth embodiment. Note that j indicates a serial number in the plurality of first transmission signal processing units 381 provided in the base station apparatus. As shown in the figure, the first transmission signal processing unit 381-j includes an IFFT & GI adding circuit 813-j, a first transmission signal processing circuit 311-j, and D / A converters 84-1 to 814-. N ′ _BS-Ant , local oscillator 815, mixers 816-1 to 816-N ′ _BS-Ant , filters 817-1 to 817-N ′ _BS-Ant , and high power amplifier (HPA) 818-1 818-N ′ _BS-Ant , antenna elements 819-1 to 819 -N ′ _BS-Ant, and a first transmission weight processing unit 330. The first transmission weight processing unit 330 includes a first channel information acquisition circuit 332, a first channel information storage circuit 333, and a first transmission weight calculation circuit 334. The configuration of the base station apparatus in the fifth embodiment is the same as that of the base station apparatus 70 (FIG. 17) in the first embodiment, but is shown in FIG. 47 instead of the first transmission signal processing unit 181. The first transmission signal processing unit 381 is different.

  When the first transmission signal processing unit 381-j according to the fifth embodiment applies time-axis beamforming to the first transmission signal processing unit 181-j illustrated in FIG. 18 in the first embodiment. This is the circuit configuration. The difference between the first transmission signal processing unit 381-j and the first transmission signal processing unit 181-j shown in FIG. 18 is that a large number of IFFT & GI adding circuits 813 are integrated into one. The transmission signal processing circuit 111-j and the first transmission weight processing unit 130 are replaced with a first transmission signal processing circuit 311 and a first transmission weight processing unit 330, and the IFFT & GI giving circuit 813 is a first This is the point arranged in the previous stage of the transmission signal processing circuit 311.

  In addition, in the first transmission signal processing unit 181-j shown in FIG. 18, multiplication of the transmission weight vector on the frequency axis is indispensable. Therefore, whether it is OFDM or SC-FDE, IFFT processing is essential in any arrangement as transmission signal processing. On the other hand, since the first transmission signal processing unit 381-j in the fifth embodiment can perform directional beam forming signal processing in units of sampling data on the time axis, the IFFT & GI adding circuit 813 is provided. Can also be omitted, and pure single carrier signal processing can also be used. In this sense, the IFFT & GI giving circuit 813 shown in FIG. 47 is expressed by a dotted square.

An operation of the first transmission signal processing unit 381-j provided in the base station apparatus of the fifth embodiment will be described. When the digital baseband signal is input from the second transmission signal processing unit 71 to the first transmission signal processing unit 381-j, if this is a signal on the frequency axis, the IFFT & GI adding circuit 813-j The signal on the axis is converted into a signal on the time axis. In the case of a system in which sampling data on the time axis is input like a pure single carrier signal, the digital signal is not passed through the IFFT & GI adding circuit 813-j (the IFFT & GI adding circuit 813-j is omitted). The baseband signal is input to the first transmission signal processing circuit 311-j. The first transmission signal processing circuit 311-j multiplies the signal series of one system by the transmission weight vector on the time axis for each sampling value, and each component of the resultant transmission signal vector is D / D for each antenna element system. Output to A converters 814-1 to 814 -N ′ _BS-Ant . The subsequent processing is equivalent to the processing performed in the first transmission signal processing unit 181-j shown in FIG.

  Note that the transmission weight vector multiplied by the signal sequence in the first transmission signal processing circuit 311-j is the same as that in the first transmission signal processing unit 181-j shown in FIG. Obtained from the first transmission weight calculation circuit 334 provided in the transmission weight processing unit 330. In the first transmission weight processing unit 330, the first channel information acquisition circuit 332 acquires the channel information acquired by the reception unit via the communication control circuit 120, and stores it in the first channel information storage circuit 333. Store and update sequentially. At the time of transmission, in accordance with an instruction from the communication control circuit 120, the first transmission weight calculation circuit 334 reads channel information corresponding to the destination terminal station device from the first channel information storage circuit 333, and based on the read channel information. The transmission weight vector is calculated. As a method for calculating the transmission weight vector here, an arbitrary method as described above can be used.

  Note that the communication control circuit 120 shown in FIG. 17 and the communication control circuit 120 shown in FIG. 47 are slightly different in details such as channel information transferred through the communication control circuit 120. For example, since FIG. 17 is intended for a general communication method, it is necessary to transfer different channel information for each subcarrier, but in the case of FIG. Only the channel information common to all subcarriers is transferred. However, since the other functions are equivalent except for such a specific difference in information, description will be made using the same reference numerals.

  Further, in the first transmission signal processing unit 181-j shown in FIG. 18, individual transmission weight vectors for each frequency component are obtained, whereas in the first transmission signal processing unit 381-j in the fifth embodiment, a single transmission weight vector is obtained. The difference from the first transmission signal processing unit 181-j in FIG. 18 is that only the transmission weight vector for one time axis component is obtained. In addition, when the communication partner radio station apparatus is fixedly installed, the transmission weight vector itself is calculated and recorded in advance, and is simply read out to be transmitted to the first transmission signal processing circuit 311. The configuration may be such that the weight vector is notified, or if the point-to-point type one-to-one communication is used, the first transmission signal processing circuit 311 may store the transmission weight vector directly. good.

FIG. 48 is a block diagram illustrating a configuration example of the first reception signal processing unit 385-j of the base station apparatus according to the fifth embodiment. As shown in the figure, the first received signal processing unit 385-j includes antenna elements 851-1 to 851-N ′ _BS-Ant and low noise amplifiers (LNA) 852-1 to 852-N ′ _BS-Ant. A local oscillator 853, a mixer 854-1 to 854-N ′ _BS-Ant , a filter 855-1 to 855-N ′ _BS-Ant, and an A / D converter 856-1 to 856-N ′ _{BS−. Ant} , a first reception signal processing circuit 358-j, an FFT circuit 857-j, and a first reception weight processing unit 360 are provided. The first reception weight processing unit 360 includes a first channel information estimation circuit 362 and a first reception weight calculation circuit 363.

  When the first received signal processing unit 385-j in the fifth embodiment applies time-axis beamforming to the first received signal processing unit 185-j shown in FIG. 19 in the first embodiment. This is the circuit configuration. The difference from the first reception signal processing unit 185-j shown in FIG. 19 is that a large number of FFT circuits 857 are integrated into one, the reception signal processing circuit 858 and the first reception weight processing unit 160 are different. Instead, a first reception signal processing circuit 358 and a first reception weight processing unit 360 are provided, and an FFT circuit 857 is further provided at a subsequent stage of the first reception signal processing circuit 358.

  Further, in the first reception signal processing unit 185-j shown in FIG. 19, since multiplication of the reception weight vector on the frequency axis is indispensable, whether it is OFDM or SC-FDE, In any arrangement, FFT processing is essential as received signal processing. On the other hand, since the first received signal processing unit 385-j in the fifth embodiment can perform directional beam forming signal processing in units of sampling data on the time axis, the first received signal It is also possible to omit the FFT circuit 857-j from the processing unit 385-j and perform pure single carrier signal processing. Further, as shown in FIG. 46 (b), the FFT circuit 857-j here is omitted even when the reception weight matrix of the time axis is used in the weight after the two-stage weight multiplication. Furthermore, it is possible to adopt a configuration in which signal processing on the time axis is continued. In this sense, the FFT circuit 857-j shown in FIG. 48 is represented by a dotted square.

The operation of the first received signal processing unit 385-j provided in the base station apparatus of the fifth embodiment will be described. Among the processes for the signals received by the antenna elements 851-1 to 851-N ′ _BS-Ant , the low noise amplifiers 852-1 to 852-N ′ from the _BS-Ant to the A / D converters 856-1 to 856-N ′ _The processing performed up to _BS-Ant is the same as the processing in the first reception signal processing unit 185-j of the first embodiment. By this processing, the received signal is converted into a digital baseband signal (sampling data). The A / D converters 856-1 to 856 -N ′ _BS-Ant input a received signal vector composed of digital baseband signals in each sampling to the first received signal processing circuit 358-j. In accordance with an instruction from the communication control circuit 120, the first reception signal processing circuit 358-j multiplies the reception signal vector on the time axis input from the first reception weight processing unit 360 by the sampling value by the reception signal vector. And converted into a single signal sequence.

  When signal processing on the frequency axis is necessary after this, as in OFDM or SC-FDE, and when processing using a reception weight matrix on the frequency axis is performed later, the first received signal processing The circuit 358-j outputs one signal series obtained by the conversion to the FFT circuit 857-j, and the FFT circuit 857-j converts the signal on the time axis into a signal on the frequency axis. In the case of a system that performs all signal processing on the time axis like a pure single carrier signal, or when continuously multiplying the reception weight matrix on the time axis in the subsequent stage, it is necessary to pass through the FFT circuit 857-j. (The FFT circuit 857-j is omitted) and this signal is output. The FFT circuit 857-j outputs the signal input from the first reception signal processing circuit 358-j at a predetermined symbol timing determined by the timing detection circuit, which is not shown here, by FFT on the time axis. The signal is converted into a signal on the frequency axis (digital baseband signal for each subcarrier). The FFT circuit 857-j outputs the digital baseband signal for each subcarrier to the second received signal processing unit 75. As described above, when the first reception signal processing unit 385-j does not include the FFT circuit 857-j, the first reception signal processing circuit 358-j converts the reception weight vector on the time axis for each sampling value. A signal for one system obtained by multiplying the received signal vector with the received signal vector is output to the second received signal processing unit 75.

In addition, when each sampled received signal vector acquired by each A / D converter 856-1 to 856-N ′ _BS-Ant is based on a training signal for channel estimation, this received signal vector Is input to the first channel information estimation circuit 362. The first channel information estimation circuit 362 uses the antenna elements 819-1 to 819 -N _MT-Ant of each terminal station device and the antenna elements 851-1 to 851 -N of the base station device 70 by some method as described above. 'Channel information with the virtual antenna element constituted by _BS-Ant is estimated on the time axis (for example, equivalent to the correlation calculation shown in Equation (52)), and the estimation result is used as the first reception weight. The result is output to the calculation circuit 363. Based on the input channel information on the time axis, the first reception weight calculation circuit 363 is on the time axis to be multiplied given by the complex conjugate or the complex conjugate value divided by the absolute value for each component. A reception weight vector is calculated. In the first embodiment, the first received signal processing unit 185 shown in FIG. 19 obtains an individual received weight vector of each frequency component, whereas the first received signal of the base station apparatus in the fifth embodiment. The difference between the processing unit 385 and the first reception signal processing unit 185 is that only the reception weight vector of a single time axis component is obtained. The first reception weight calculation circuit 363 outputs the reception weight vector calculated in this way to the first reception signal processing circuit 358-j. The calculation of the reception weight vector by the first reception weight calculation circuit 363 corresponds to the processing described in FIG.

  Further, the communication control circuit 120 manages control related to overall communication such as management of a transmission source station and overall timing control. Here, when the terminal station apparatus of the communication partner is fixedly installed, the reception weight vector itself is calculated and recorded in advance, and is simply read out to the first reception signal processing circuit 358-j. Alternatively, the reception weight vector may be notified, or the point-to-point type one-to-one communication may store the reception weight vector directly in the first reception signal processing circuit 358. May be.

  Although the above description is signal processing only for the virtual transmission line corresponding to the first singular value, the virtual transmission line corresponding to the first singular value mainly conscious of the line-of-sight wave is effectively used. However, when a relatively clean reflected wave component such as a large structure is also used as an auxiliary, for example, several samples other than addition synthesis by multiplying coefficients between sampling values at the same time between multiple antenna elements. A configuration including a delay component may be used. For example, when the reference antenna is the first antenna and a coefficient for leaking a delayed wave of one sampling clock into the j-th antenna is obtained, the following equation (57) obtained by expanding equation (52) may be used.

Here, as in the equation (52), S _j (n) represents the n-th sampling value of the j-th antenna element. Here, since the signal processing is performed on the signal of one antenna element pair to one antenna element, the value of one symbol is not used as it is for the line gain, but the SNR is obtained by using the average value of many times. It is desirable to increase the value. Based on this coefficient, the value of the reception weight may be obtained in the same manner as in equation (50) or equation (51). For the reception weight obtained in this way, for example, as a reception signal process, a signal sequence on one time axis may be calculated by the following equation (58).

Here, ˜w ′ _j is a reception weight calculated using the formula (50) or the formula (51) with respect to the coefficient c ′ _j . In this way, it is possible to perform processing in consideration of a delayed wave for one sampling clock. Similarly, when considering a delayed wave of two sampling clocks, the following equation (59) may be applied.

Here, ˜w ″ _j is a reception weight calculated using the equation (50) or the equation (51) with respect to the coefficient c ″ _j . If the same extension is performed, it is possible to consider further delayed wave components. However, both Equation (58) and Equation (60) are the same for each antenna element at the same time shown in Equation (53). It is a common feature that it includes a component that adds and synthesizes a signal obtained by multiplying the sampling value signal by the reception weight to w _j , and the essence of this embodiment is here.

  In the present embodiment, in the synthesis of transmission / reception signals of a large number of antenna elements, frequency-dependent transmission / reception weights are used, and conventional directivity control is performed by multiplying different transmission / reception weights for each individual subcarrier on the frequency axis. Instead, by using a transmission / reception weight common to all subcarriers, directivity control on the time axis can be performed for each sampling value, which can be used in both the base station apparatus and the terminal station apparatus in the same manner. It is also possible to implement only in the base station apparatus or the terminal station apparatus. Although the above description was about the base station apparatus, the terminal station apparatus will be described below. Here, as in the case of the base station apparatus, it is possible to apply time-axis beamforming on both the transmission side and the reception side.

FIG. 49 is a block diagram illustrating an example of a circuit configuration of the transmission unit 361 included in the terminal station apparatus according to the fifth embodiment. As shown in the figure, the transmission unit 361 includes a second transmission signal processing circuit 348, IFFT & GI adding circuits 83-1 to 813-N _SDM, and first transmission signal processing circuits 331-1 to 331-N _SDM. , Adder / synthesizer circuit 812-1 to 812-N _MT-Ant , D / A converters 814-1 to 814-N _MT-Ant , local oscillator 815, mixers 816-1 to 816-N _MT-Ant And filters 817-1 to 817-N _MT-Ant , high power amplifiers (HPA) 818-1 to 818-N _MT-Ant , antenna elements 819-1 to 819-N _MT-Ant , A transmission weight processing unit 340. The first transmission weight processing unit 340 includes a first channel information acquisition circuit 341, a first channel information storage circuit 342, and a first transmission weight calculation circuit 343. The configuration of the terminal station apparatus in the fifth embodiment has the same configuration as that of the terminal station apparatus 60 (FIG. 20) in the first embodiment, but differs in that a transmission unit 361 is provided instead of the transmission unit 61.

  The transmission unit 361 in the fifth embodiment has a circuit configuration in the case where time-axis beamforming is applied to the transmission unit 61 illustrated in FIG. 25 in the first embodiment. The difference between the transmission unit 361 and the transmission unit 61 shown in FIG. 25 is that a large number of IFFT & GI assignment circuits 813 are integrated into one, and the first transmission weight processing unit 140 is replaced with the first transmission weight. The processing unit 340 is provided, and further, an IFFT & GI adding circuit 813 is arranged in the previous stage of the first transmission signal processing circuit 331.

  In addition, since the transmission unit 61 shown in FIG. 25 requires multiplication of the transmission weight vector on the frequency axis, IFFT processing is used as transmission signal processing on the frequency axis regardless of OFDM or SC-FDE. It was essential in either arrangement. On the other hand, the transmission unit 361 according to the fifth embodiment can perform directional beam forming signal processing in units of sampling data on the time axis. Therefore, the IFFT & GI adding circuit 813 is also omitted, and a pure single It is also possible to perform carrier signal processing. In this sense, the IFFT & GI giving circuit 813 in the drawing is represented by a dotted square.

Operation | movement of the transmission part 361 with which the terminal station apparatus of 5th Embodiment is provided is demonstrated. When data corresponding to the number of spatial multiplexing N _SDM systems (data input # 1 to #N _SDM ) is input from the MAC layer processing circuit 68 to the second transmission signal processing circuit 348, the second transmission signal processing circuit 348 Then, transmission signal processing is performed on each input data system, and a digital baseband signal is generated for each data system. This signal processing is general, and any type of radio signal may be used, such as OFDM, single carrier, or SC-FDE. When arbitrary precoding is performed on the transmission signal, the second transmission signal processing circuit 348 performs precoding signal processing. When the signal output from the second transmission signal processing circuit 348 is a signal on the frequency axis, the IFFT & GI giving circuit 813 converts the signal on the frequency axis into a signal on the time axis for each signal system. In the case of a system in which sampling data on the time axis is input like a pure single carrier signal, the second transmission is performed without the IFFT & GI adding circuit 813 (that is, the IFFT & GI adding circuit 813 is omitted). The signal processing circuit 348 inputs the signals of each data system obtained by the signal processing to the first transmission signal processing circuits 331-1 to 331 -N _SDM .

The first transmission signal processing unit 381-j provided in the base station apparatus shown in FIG. 47 performs signal processing on one signal sequence. On the other hand, since the terminal station apparatus performs processing of a plurality of signal systems, the terminal station apparatus includes a plurality of first transmission signal processing circuits 331-1 to 331 -N _SDM . Each of the first transmission signal processing circuits 331-1 to 331 -N _SDM multiplies the signal series of the corresponding data system by a transmission weight vector on the time axis for each sampling value, and each of the resultant transmission signal vectors. The components are output to the adder / synthesizer circuit 812-1 to 812-N _MT-Ant . The adder / synthesizer circuits 812-1 to 812-N _MT-Ant add and synthesize all the signal series signals input from the first transmission signal processing circuits 331-1 to 331-N _SDM , respectively. The D / A converters 814-1 to 814 -N output to the _SDM . The subsequent processing is equivalent to the processing performed in the transmission unit 61 shown in FIG. Note that the transmission weight vector on the time axis multiplied by the signal sequence of each data system in the first transmission signal processing circuits 331-1 to 331 -N _SDM is similar to the transmission unit 61 during signal transmission processing. It is acquired from the first transmission weight calculation circuit 343 provided in the first transmission weight processing unit 340. Alternatively, the transmission weight vector on the time axis acquired in advance as described above is stored in the first transmission weight processing unit 340, and then the first transmission signal processing circuit 331-1 is used. 331-N It may be configured to instruct the _SDM .

In the first transmission weight processing unit 340, the first channel information acquisition circuit 341 separately acquires the channel information acquired by the reception unit 365 via the communication control circuit 121, and sequentially updates this, The data is stored in the first channel information storage circuit 342. At the time of signal transmission, in accordance with an instruction from the communication control circuit 120, the first transmission weight calculation circuit 343 reads channel information corresponding to the destination terminal station device from the first channel information storage circuit 342, and the read channel information Based on, the transmission weight vector on the time axis is calculated. Here, the calculation method of the transmission weight vector can use an arbitrary method similar to that of the above-described base station apparatus. Furthermore, the transmission unit 61 shown in FIG. 25 obtains individual transmission weight vectors for each frequency component, whereas the first transmission weight calculation circuit 343 in the fifth embodiment transmits a transmission weight with a single time axis component. A point for obtaining only the vector is a difference from the transmission unit 61. Further, when the terminal station apparatus is fixedly installed, the transmission weight vector itself is calculated and recorded in advance, and the first transmission signal processing circuits 331-1 to 331 -N _SDM are simply read out. The transmission weight vector may be notified to the first transmission signal processing circuits 331-1 to 331 -N in the _{SDM for} point-to-point type one-to-one communication. The direct transmission weight vector may be stored.

FIG. 50 is a block diagram illustrating a configuration example of the receiving unit 365 of the terminal station apparatus according to the fifth embodiment. As shown in the figure, the receiving unit 365 includes antenna elements 851-1 to 851-N _MT-Ant , low noise amplifiers 852-1 to 852-N _MT-Ant , a local oscillator 853, and mixers 854-1 to 854-1. 854-N, filters 855-1 to 855-N, A / D converters 856-1 to 856-N, first received signal processing circuits 355-1 to 355-N _SDM , and FFT circuit 857- 1-857-N _SDM , the 2nd received signal processing circuit 359, and the 1st received weight processing part 354 are provided. The first reception weight processing unit 354 includes a first channel information estimation circuit 356 and a first reception weight calculation circuit 357.

The receiving unit 365 in the fifth embodiment has a circuit configuration when time-axis beamforming is applied to the receiving unit 65 shown in FIG. 23 in the first embodiment. The difference from the receiving unit 65 shown in FIG. 23 is that a large number of FFT circuits 857 are integrated into one, the first received signal processing circuit 155, the second received signal processing circuit 159, and the first received weight processing. A first reception signal processing circuit 355, a second reception signal processing circuit 359, and a first reception weight processing unit 354 instead of the unit 154, and further, FFT circuits 857-1 to 857-N _SDM , Is arranged at the subsequent stage of the first reception signal processing circuit 355.

Further, in the receiving unit 65 shown in FIG. 23, since multiplication of the reception weight vector on the frequency axis is indispensable, FFT processing is performed as reception signal processing on the frequency axis regardless of whether it is OFDM or SC-FDE. It was essential in either arrangement. On the other hand, the receiving unit 365 according to the fifth embodiment can perform directional beam forming signal processing in units of sampling data on the time axis, and thus includes FFT circuits 857-1 to 857 -N _SDM . It may be omitted and pure single carrier signal processing may be used. In this sense, the FFT circuits 857-1 to 857 -N _SDM shown in FIG. 50 are represented by dotted-line squares.

The operation of the receiving unit 365 provided in the terminal station apparatus according to the fifth embodiment will be described. Of the processes for the signals received by the antenna elements 851-1 to 851-N _MT-Ant , the low noise amplifiers 852-1 to 852-N _MT-Ant to A / D converters 856-1 to 856-N _MT-Ant The processing performed up to is the same processing as the processing in the receiving unit 65 of the first embodiment. The A / D converters 856-1 to 856-N _MT-Ant input a received signal vector composed of digital baseband signals in each sampling to the first received signal processing circuits 355-1 to 355-N _SDM . . In the base station apparatus, as shown in FIG. 48, the first received signal processing unit 385-j performs processing of one signal series. On the other hand, in order to perform processing of a plurality of signal systems in the terminal station apparatus, the receiving unit 365 is configured to include a plurality of first received signal processing circuits 355-1 to 355 -N _SDM . The first reception signal processing circuits 355-1 to 355 -N _SDM multiply the reception signal vector for each sampling value by the reception weight vector on the time axis corresponding to each signal series in accordance with an instruction from the communication control circuit 121. , Converted into signals of each signal series. The reception weight vector corresponding to each signal series is input from the first reception weight processing unit 354. The first reception signal processing circuits 355-1 to 355 -N _SDM output the obtained signals of each signal series to the FFT circuits 857-1 to 857 -N _SDM corresponding to the signal series.

  Strictly speaking, the communication control circuit 121 shown in FIG. 20 and the communication control circuit 121 shown in FIG. 50 are slightly different in details such as channel information transferred via the communication control circuit 121. For example, in FIG. 20, since a general communication method is targeted, it is necessary to transfer different channel information for each subcarrier. In the case of FIG. 50, single channel information on the time axis (that is, frequency dependence) Only the channel information common to all subcarriers is transferred. However, since the other functions are equivalent except for such a specific difference in information, description will be made using the same reference numerals.

When signal processing on the frequency axis is necessary after this, as in OFDM and SC-FDE, the signal of each signal sequence on the time axis is converted to the frequency axis by the FFT circuits 857-1 to 857-N _SDM . Convert to the above signal. In the case of a system that performs all signal processing on the time axis like a pure single carrier signal, or in the case where signal processing is performed using the reception weight matrix on the time axis even in the subsequent stage, the FFT circuit 857-1 The signal of each signal series is output to the second received signal processing circuit 359 without passing through the 857-N _SDM (that is, the FFT circuit 857-1 to 857 -N _SDM is omitted).

For example, when the second received signal processing circuit 359 performs signal processing on the time axis, the first received signal processing circuits 355-1 to 355-N _SDM to the spatial multiplexing number N _SDM on the time axis. The signal of each signal sequence is input, multiplied by the reception weight matrix on the time axis, and the signal separation of the spatially multiplexed signal sequence is performed on the time axis, and the separated signals are required as necessary. The signal detection process is performed, and the reproduced data series is output to the MAC layer processing circuit 68.

Here, the reception weight matrix by which the second reception signal processing circuit 359 multiplies each signal series signal is a reception weight matrix obtained by the processing shown in FIG. Here, a reception weight matrix acquired in advance may be used, or the head of a signal (wireless packet) input from the first reception signal processing circuits 355-1 to 355 -N _SDM each time a wireless packet is received. A channel matrix related to channel information is obtained by the calculation shown in Expression (56) using the training signal for channel estimation given to, and, as a general MIMO signal processing, ZF method, MMSE method, etc. A linear weight may be calculated and used as a reception weight matrix. The signal of the N _SDM system that is signal-separated in this way may be processed by single carrier if it is a single carrier signal, or timing detection that is omitted here if it is OFDM or SC-FDE. The data on the time axis is converted into a signal on the frequency axis (digital baseband signal for each subcarrier) by FFT at a predetermined symbol timing determined by the circuit for the data, and the data is reproduced by performing normal demodulation processing The series is output to the MAC layer processing circuit 68.

On the other hand, when the second received signal processing circuit 359 performs signal processing on the frequency axis, signals on the frequency axis from the FFT circuits 857-1 to 857 -N _{SDM are} sent to the second received signal processing circuit 359. A reception weight matrix that is input and acquired in advance may be used, or the head of a signal (wireless packet) input from each of the first reception signal processing circuits 355-1 to 355 -N _SDM each time a wireless packet is received. The channel matrix is obtained by performing channel estimation based on the relationship between the information of the training signal part for channel estimation and the known training signal given to ZF and MMSE as general MIMO signal processing. It is possible to calculate a linear weight such as a reception weight matrix, or to perform signal detection processing by nonlinear processing such as MLD or simplified QR-MLD. The data series reproduced by performing such signal detection processing is output to the MAC layer processing circuit 68. Note that the signal detection process or the demodulation process in the above description may include a deinterleaving process, an error correction process, or the like as necessary.

In addition, when the received signal vector which uses each sampling value acquired in each A / D converter 856-1 to 856-N _MT-Ant as a component corresponds to the training signal for channel estimation, this The received signal vector is input to first channel information estimation circuit 356. The first channel information estimation circuit 356 uses the antenna elements 851-1 to 851-N _MT-Ant of each terminal station device and the virtual antennas of the base station device 70 by some method similar to that of the base station device 70 described above. The channel information between the elements is estimated on the time axis, and the estimation result is output to the first reception weight calculation circuit 357. The first reception weight calculation circuit 357 calculates a reception weight vector to be multiplied based on the input channel information on the time axis. In the first embodiment, the receiving unit 65 shown in FIG. 23 obtains an individual transmission weight vector of each frequency component, whereas the receiving unit 365 of the terminal station apparatus in the fifth embodiment has a single time axis. The difference from the receiving unit 65 is that only the transmission weight vector for the component is obtained. The first reception weight calculation circuit 357 outputs the reception weight vector calculated in this way to the first reception signal processing circuits 355-1 to 355-N _SDM . Alternatively, the reception weight vector on the time axis acquired in advance as described above is stored in the first reception weight processing unit 354, and from there, the first reception signal processing circuit 355-1 is stored. ~ 355-N _SDM may be instructed.

In addition, the communication control circuit 121 manages control related to overall communication such as management of a transmission source station and overall timing control. Here, when the terminal station apparatus is fixedly installed, the reception weight vector itself is calculated and recorded in advance, and is simply read out to the first reception signal processing circuit 355 to receive the reception weight vector. In addition, in the case of point-to-point type one-to-one communication, the reception weight vector directly in the first reception signal processing circuits 355-1 to 355 -N _SDM is used. You may remember. The above is the description of the configuration example of the reception unit 365 of the terminal station apparatus in the fifth embodiment.

  The point of the fifth embodiment is that when a plurality of antenna elements are provided, the difference between the path lengths of the line-of-sight wave is sufficiently reduced by narrowing the distance between the antenna elements, and the reception weight in the frequency domain is reduced. There is a tendency for frequency dependence to be reduced. At this time, when the reception weight is converted into the time domain by IFFT or the like, almost the first wave component (line-of-sight wave component) occupies most of the entire received power. In such a situation, MIMO signal separation performed in the frequency domain can be performed in the time domain. Also, MIMO signal separation that can be performed in the time domain using only the first wave component can be performed for each sampling value.

  Therefore, in the wireless communication system according to the fifth embodiment, for example, at the time of signal reception, the first reception signal processing unit 385 included in the base station device and the first reception signal processing in the reception unit 365 that the terminal station device can include. The circuit 355 performs first-stage signal separation for each sampling value on the time axis, which separates the signal of the virtual transmission line corresponding to the first singular value obtained by singular value decomposition of the channel matrix from the received signal. To do. Thereafter, the second received signal processing unit 75 provided in the subsequent stage of the first received signal processing unit 385 in the base station device and the second received signal processing circuit 359 in the receiving unit 365 that the terminal station device may include are virtually Second-stage signal separation for separating signals between transmission lines is performed. In the apparatus on the receiving side of the wireless communication system according to the fifth embodiment, when the second stage signal separation is performed on the time axis, the reception weight matrix used in the signal separation is the signal received for each antenna element. It is possible to calculate based on the channel matrix obtained by obtaining the correlation value.

  By performing the MIMO signal separation in the time domain, it is possible to omit the FFT required for the signal separation, and to reduce the computation load and the computation circuit. In addition, when performing spatially multiplexed single carrier transmission, it is not necessary to convert the signal on the frequency axis, so that the effects of phase noise, which is a problem in high frequency bands such as millimeter waves, can be reduced with existing pure single The phase noise compensation technique in carrier transmission can be used. Further, while reducing the calculation load for MIMO signal separation in the base station apparatus and the terminal station apparatus, as in the first to fourth embodiments, the band can be expanded in the situation where the direct wave in the line-of-sight environment is dominant. be able to.

  In the wireless communication system according to the fifth embodiment, the configuration for obtaining the reception weight / transmission weight from the correlation coefficient has been described. However, channel information may be fed back to some subcarriers, and the reception weight / transmission weight of all frequency bands may be calculated using the average value.

[Sixth Embodiment]
[Approximate method for obtaining channel matrix when line-of-sight wave is dominant]
(Outline of the basic principle according to the sixth embodiment)
In the above description, spatial multiplexing transmission that actively uses virtual transmission paths corresponding to a plurality of first singular values has been described. In this spatial multiplexing transmission, it is necessary to decompose the channel matrix by singular value decomposition and calculate the first right singular vector or the first left singular vector. Moreover, in order to acquire the original channel matrix used for this singular value decomposition, channel estimation and feedback of channel information are necessary. Here, in order to improve the gain by using a large number of antenna elements, it is necessary to obtain a channel matrix component corresponding to the product of the respective numbers of a large number of transmitting antennas and a large number of receiving antennas. In general, the overhead associated with transmission of a training signal for channel estimation increases in proportion to the number of components of the channel matrix, and this amount cannot be ignored.

  In the implicit feedback, for example, when an uplink channel matrix can be acquired, a downlink channel matrix is estimated by a calibration process, and overhead associated with an increase in the number of antenna elements of the base station apparatus can be reduced. it can. On the other hand, if there are multiple antenna elements on the terminal side, an uplink channel matrix cannot be obtained unless a training signal is individually transmitted from each antenna element, and the overhead associated with the increase in the number of antenna elements on the terminal side is It cannot be reduced.

  When the number of antenna elements on the terminal side is relatively small, for example, if two antennas are used, each antenna transmits a training signal on even-numbered subcarriers and odd-numbered subcarriers. Channel estimation is possible. In this case, channel information of subcarriers for which channel estimation has not been performed is estimated by interpolation. In addition, it is possible to efficiently perform channel estimation by using a plurality of orthogonalized preamble signals. However, for example, in the case of including 16 or 32 multi-element antenna elements, there is a problem that the accuracy of channel estimation by interpolation is extremely reduced because the subcarrier interval is too wide, which is efficient. There are limitations to channel estimation and its feedback.

  Furthermore, even when a large-scale antenna is applied to improve the line gain, a single element antenna cannot be expected to have a high SNR (signal-to-noise ratio) that allows effective channel estimation in line design. For this reason, for example, long-time measurement and averaging processing on the premise that there is no channel time variation are required. However, long-time averaging cannot be adopted in a case where time variation occurs, as in the case of a train moving cell, for example, and it is necessary to acquire channel information of all necessary paths instantaneously. In a general MIMO system, for example, when N antenna elements are provided on the transmission side, channel estimation is performed using N training symbols of 1 OFDM symbol. However, when both the transmitting and receiving stations are provided with a large number of antenna elements as described above, there is a possibility that the channels may fluctuate from time to time during the estimation of N channels. For this reason, a channel estimation method that finishes obtaining necessary channel information in as short a time as possible is required.

  Here, when the virtual transmission path for the first singular value is actively used, the antenna group of the terminal device and the antenna group of each first signal processing unit of the base station device are in a very narrow area. It is concentrated. For this reason, the channel information slightly changes due to the subtle position shift. However, if attention is paid only to the line-of-sight component, the change is expected to change in a manner that maintains regularity between the components of the channel vector. For example, when a signal transmitted from an antenna element on the transmitting side is received by an antenna group on the receiving side, the channel information changes depending on the signal path length difference. For this reason, what is the regularity is acquired on the propagation model considering only “line-of-sight”, and channel estimation is efficiently performed by actively using the regularity. Can do. Hereinafter, a method for efficiently performing this channel estimation will be described.

FIG. 51 is a diagram showing an outline of received signals received by a plurality of antenna elements. In this figure, reference numerals 221-1 to 221-5 are antenna elements, and the antenna elements 221-1 to 221-5 are other antenna elements (in FIG. It shows the situation when receiving a signal from (not shown). Here, the received signal at time t at the m-th antenna element 221-m is represented as Φ _m (t). When plane wave approximation is performed with the distance from the transmission antenna of the first antenna element 221-1 of the incoming wave from the angle θ direction as L, the wavefront becomes a line indicated by a dotted line.

Therefore, the difference between the arrival distances of the first antenna element 221-1 and the m-th antenna element 221-m is ΔL _m as shown in the figure. If the antenna element interval is d, the path difference between the mth and m + 1th antenna elements is d · sin θ. However, here, ΔL _m is used to define the path length difference with reference to the first antenna element 221-1 in order to generalize the case including arrangements other than the linear array. Here, if the received signal of the m-th antenna element 221-m is expressed by being divided into k-th subcarriers φ _m ^(k) (t), the received signal Φ _m (m) of the time t at the m-th antenna element 221-m. t) can be expressed by the following formula (61) as the sum of the subcarrier components.

Here, if the transmission signal component at time t is _represented by S _k (t), the reception signal of the radio frequency f _c + f _k with respect to the center frequency f _c can be represented by the following equation (62).

Here, c _m ^(k) is a coefficient (relative channel information) of a received signal at the m-th antenna element 221-m when the received signal at the first antenna element 221-1 is used as a reference in the k-th subcarrier. Yes, it is given by the following equation (63).

Here, α _m is a constant related to the m-th antenna element 221-m and is defined by the following relational expression (formula (64)).

The right side of the equation (63) is expressed as a product of the powers of two natural logarithms. The former is a constant having no frequency dependency for each subcarrier, and takes the different value for each antenna element, while the latter is a frequency. Indicates dependency. Considering that W is the frequency bandwidth, if x = f _k / W, when α _m is smaller than 1, complex when x varies between −1/2 and +1/2. The phase changes linearly at 2πα _m x. By using this characteristic that the complex phase changes linearly, it is possible to determine the number of subcarriers and perform linear interpolation of the complex phase component for all subcarriers without calculating the reception weight for all subcarriers. It is possible to calculate the receiving weight.

In the above description, the right side of Equation (63) is a complex number with an absolute value of 1, but in actual measurement results, it is not strictly a complex number with an absolute value of 1 due to the influence of noise and reflected waves. In such cases, observed _c ^m a ^{_{(k) c m (k)}} / | c m (k) | with substituted on normalized value and performing the following processes. In addition, when the reception weight is calculated using the complex conjugate value of the channel information standardized in this way, the weight is obtained as the weight of equal gain synthesis instead of the maximum ratio synthesis. Since it is assumed that the sight wave is dominant, it is expected that there will be no significant difference in characteristics between equal gain synthesis and maximum ratio synthesis.

FIG. 52 is a diagram showing an outline of channel estimation using limited subcarriers. In this figure, channel estimation is performed for the m-th antenna element for four subcarriers f _{A to} f _D within the frequency bandwidth W, and normalized relative channel information _cm obtained by measurement. A value obtained by dividing a value obtained by taking a natural logarithm (a logarithm using the base e of the natural logarithm) with respect to ^(fk) by 2πj is defined as Y _m (f _k ) (Expression (65)).

Referring to the equation (65), when f _k / W is x and Y _m (f _k ) is y and plotted on the graph, the right graph of FIG. 52 is plotted against x between −1/2 to +1/2. Thus, Y _m (f _k ) observed in the vicinity of the line y = ax + b in a straight line is plotted. Therefore, the frequency dependence of the channel information can be acquired by fitting this straight line by the least square method. Conventionally, there has been a technique for estimating a channel by using a subcarrier that transmits a training signal as a missing tooth. However, since there was no information about the frequency dependence of the channel information in the entire band, linear interpolation between the two observed subcarriers and the frequency dependence before and after the two points were observed. In consideration of this, it has been limited to processing such as nonlinear interpolation. In the present embodiment, when focusing only on the line-of-sight component, the fact that the frequency dependence is linear in the complex phase in the entire frequency band is utilized, and the linear interpolation utilizing the frequency dependence over the entire band. Is to do. However, it should be noted here that, when taking the natural logarithm of a complex number, even if an integer multiple of 2πj is added to the power of e, Y _m (f _k ) has the same value. The point which must be corrected is mentioned.

For example, in the example shown in FIG. 52, the least squares method uses a regression line and four points (f _A / W, Y _m (f _A )), (f _B / W, Y _m (f _B )), (f _C / W, Y _m (f _C )) and (f _D / W, Y _m (f _D )). In addition to this, for example, with respect to η _A , η _B , η _C , and η _D each given by an offset value of ± 1 or 0, a regression line and four points (f _A / W, Y _m (F _A ) + η _A ), (f _B / W, Y _m (f _B ) + η _B ), (f _C / W, Y _m (f _C ) + η _C ), (f _D / W, Y _m (f _D ) + η _D ) is performed to calculate a and b for each combination of offset values, and the combination of a and b that minimizes the following equation (66) is calculated. It can be regarded as a true regression line.

Here, since the offset values can take values of −1, 0, and +1 in each of the four subcarriers, 3 ⁴ = 81 kinds of least squares are implemented as a combination of all of them. The channel information of the following equation (67) is calculated for an arbitrary frequency f _k using a and b thus obtained, and the reception weight is calculated as this complex conjugate.

In a slight supplement, the coefficient a which is the slope of the regression line is a physical quantity corresponding to α _m in the equation (64). If this is a value sufficiently smaller than 1, for example, the y = ax + b straight line is sufficiently smaller than 1 when x changes between −1/2 and 1/2, and noise and reflected waves Even if the influence of the above is taken into consideration, it is considered that the offset amount is sufficient in the range of ± 1. However, if the value of ΔL _m increases to some extent, the change in y exceeds 1, and in such a case, it can be dealt with by expanding the selection range such as 0, ± 1, ± 2, etc. It becomes.

If the value of [Delta] L _m is small to some extent, on the other hand, for example, _f B / and _Y m _{(f B)} observed in W, is observed at the front and rear of _f A / W and _f C / W _Y m ( f _A ) and Y _m (f _C ) are not so different. In this case, both | Y _m (f _A ) −Y _m (f _B ) + η _B | and | Y _m (f _C ) −Y _m (f _B ) + η _B | are both predetermined values (for example, 0 An offset amount η _{B that} is larger than (such as .5 or 0.3, which is smaller than 1) can be excluded from consideration. Here, for example, Y _m (f _A ), Y _m (f _D ) and the like are observation points at both ends, and therefore, comparison with only one of the observation points may be made instead of comparison with observation points at both ends. In this way, it is possible to limit the search range while taking the offset amount into consideration, and it is possible to suppress an increase in the amount of calculation associated with performing the individual least square method.

  In the above channel estimation, it is only necessary to transmit a training signal with a limited number of subcarriers from a transmitting antenna. To be able to. For example, if the total number of subcarriers is 1000, focusing on four subcarriers allows transmission with 250 times the transmission power compared to transmitting to all subcarriers, and therefore 10 Log 250≈24 [dB]. You can earn line gain. Therefore, the channel estimation accuracy is improved accordingly.

Further, if four subcarriers f _A ′ to f _D ′ of subcarriers different from the above-mentioned subcarriers f _A to f _D are used, it is possible to simultaneously perform the same channel estimation for different antenna elements. It is. For example, even if a 250-element antenna is provided on the transmission side, if each transmits with only four subcarriers, a total of 1000 subcarriers is sufficient. That is, even if a MIMO channel is composed of 250 antenna elements and a large number (for example, 100 elements) of antenna elements, channel information of a 250 × 100 MIMO channel matrix of all subcarriers by one channel estimation. Can be acquired.

  Here, if the number of effective subcarriers (subcarriers used in data communication) is not a multiple of 4, it is necessary to cope with at least the fractional adjustment. For example, for a certain antenna element, a part of the training signal to which only slightly smaller 3 subcarriers are assigned may be used, or adjustment may be performed without assigning a training signal to some effective subcarriers. good. This side depends on the system parameters.

  Here, for example, as shown in FIG. 16, consider a case where a plurality of first signal processing units 304 are provided with a plurality of antenna elements. Assuming that each first signal processing unit 304 includes 50 antenna elements and further includes five first signal processing units 304-1 to 30-5, the total number of antenna elements is 250. . As described above, when channel information in the downlink is acquired using these antennas, all of them can be acquired at once. In addition, when spatial multiplexing is performed using a plurality of virtual transmission lines corresponding to the first singular value, transmission directivity control across different first signal processing units 304 is not performed. When ten signal processing units 304 are provided, the same processing is performed by the first to fifth signal processing units 304, the sixth to tenth first signal processing units 304, and the like. The channel estimation may be performed in two steps.

  What is important is that the channel estimation related to the antenna elements related to the same first signal processing unit 304 is performed at the same time, and the channel estimation is performed collectively for the antenna elements belonging to different first signal processing units 304. do not have to. Incidentally, when the number of subcarriers is 2000, 4 subcarriers × 50 antennas × (5 sets + 5 sets) is 2000, so channel estimation is possible if divided into two. As described above, the total number of subcarriers, the number of subcarriers used for channel estimation once, and the like are parameters in the system design, and this embodiment can be applied to any other values as well.

Here, points to be noted when the transmission side performs channel estimation using a plurality of antenna elements will be described. First, in the above description, as shown in the equation (63), the reception signals of the plurality of antenna elements on the reception side are divided by the reception signals of the reference antenna element (here, the first antenna element is assumed). That is, relative channel information is used as a relative relationship of channel information based on the first antenna element rather than absolute channel information. If this is explicitly shown, the relative channel matrix H _relative obtained by the above-described procedure sets the component of the channel information between the j-th transmission antenna and the i-th reception antenna of the k-th subcarrier as h _ij ^(k) . Then, it is given by the following formula (68).

A feature of the relative channel matrix H _relative is that each component of the row vector of the first row is 1, and all the information is based on the first antenna element on the receiving side. All the relative channel information for each subcarrier obtained by the least square method described above relates to the receiving antenna after the second antenna, and the channel information related to the first antenna element on the receiving side is outside the above discussion. Of course, since the sampling information regarding Φ ₁ (t) is acquired, the absolute channel information can be obtained through FFT processing or the like based on this information over one OFDM symbol. However, as described above, for example, signals transmitted from the first antenna on the transmitting side are only signals on subcarriers f _A to f _D, and signals on other subcarriers f _A ′ to f _D ′ are transmitted on the transmitting side, for example. The signal is transmitted from the second antenna.

That is, only the channel information of the first transmission antenna can be acquired for the same subcarrier f _A , and only the channel information of the second transmission antenna can be acquired for the subcarrier f _A ′. For example, when antenna elements are changed between even-numbered subcarriers and odd-numbered subcarriers, channel information of subcarriers for which no training signal is transmitted can be predicted by linear interpolation. If the training signal is limited to the carrier, linear interpolation between them becomes impossible.

  However, as will be described later, when this matrix is subjected to singular value decomposition, the left singular matrix is affected by the diagonal component of the second term of the product of the right side of Equation (68). A similar left singular matrix is generated.

The reason is as follows. First, when all the components of a matrix are multiplied by a given coefficient, if the matrix is singularly decomposed, all the singular values are multiplied by the given coefficient, and the right singular matrix and the left singular matrix have the same value. It becomes. Next, 1 / h _1j ^(k) of each diagonal component of the second term on the right side of Expression (68) is a line-of-sight wave and has an antenna element in the vicinity thereof, and therefore corresponds to the amplitude of the received signal | h _1j ^(K) | is almost equal.

That is, when the entire matrix is multiplied by a coefficient | h ₁₁ ^(k) |, each diagonal component of the second term on the right side of Equation (68) becomes | h ₁₁ ^(k) | / h _1j ^(k) , All diagonal terms have absolute values of 1 and differ only in complex phase. In this matrix, arbitrary column vectors are orthogonal to each other, and the diagonal term of the matrix taking the complex conjugate of this matrix is the inverse of the original diagonal term, and the product of the complex conjugate matrix multiplied by the original matrix is It is a unit matrix. That is, since this is a unitary matrix, if the matrix of the first term on the right side of Equation (68) is singularly decomposed, the left singular matrix and the matrix having singular values in the diagonal terms are common and the right singular matrix Is only the matrix transformed into the right singular matrix rotated by the unitary matrix described above.

Even if the entire channel matrix, which is the matrix of the first term on the right side of Equation (68), is not known, it is sufficient to know the relative channel matrix H _{relative for} calculating the reception weight vector for reception. By the way, if the reception weight vector is known, the transmission weight vector can also be acquired by calibration processing. As a result, if the channel estimation is performed by the above-described method, both the reception weight vector and the transmission weight vector are received on the side receiving the training signal. , That means both are acquirable.

  Thus, in the above-described method, for example, by performing channel estimation in the downlink, it is possible to obtain a reception weight vector and a transmission weight vector (or a weight matrix formed by combining the vectors) on the terminal station side. However, in order to obtain the reception weight vector and the transmission weight vector (or the weight matrix formed by combining the vectors) on the base station apparatus side, a training signal is similarly transmitted from the terminal station apparatus side, and the uplink Thus, the same processing can be performed.

(Optional function for improving channel estimation accuracy)
Since the system to which the above-described method is applied targets the case where the arrival direction of radio waves exists in a relatively narrow range, it is assumed that an antenna element having a high directivity gain is used. In this case, since the line-of-sight wave is dominant in the line-of-sight environment, the channel information is expected to show a linear behavior on the frequency axis as described above. Therefore, there is a possibility that the behavior deviates from the straight line due to the frequency selective distortion. The degree of this shift depends on the ratio of the power of the reflected wave. However, for example, if regression calculation is performed with only four subcarriers, the accuracy may be lowered. Therefore, it is considered to perform comprehensive regression calculation on the entire channel information of a plurality of antenna elements.

For example, considering a linear array as shown in FIG. 51, it is assumed that the path length difference ΔL _m between the first antenna element and the m-th antenna element is expressed by the following relational expression (69).

That is, α _m in the equation (64) is given by the following relational equation (70).

In this case, the entire channel information may be acquired with the constraint condition of Expression (70). It should be noted that there are several possible variations on how to apply this constraint condition. For example, a and b are obtained individually by the above least square method for all antennas, and since the slope a of this straight line coincides with α _m , a / (m−1) is calculated for m of 2 or more. an average value, an average value ^ alpha ₂ of alpha ₂ of the formula (70) with respect to two or more m. Next, for the m-th antenna, (m−1) × ＾ α ₂ may be substituted for a, and the Y intercept at the slope may be obtained for each antenna element by the least square method.

Examples of other methods include the following methods. When the angle θ of the arrival direction of the radio wave and the antenna element interval d are obtained, since α ₂ can be estimated approximately by d · sin θ × W / c, a plurality of values in the vicinity thereof can be used as test α ₂ values. A = α _m (that is, the slope of the regression line) is given by (m−1) × α ₂ , and by using the a, the least square method in which the offset value η _k is introduced as described above for each antenna. The y-intercept value b _{m of} the m-th antenna and the offset value group {η _k } used at that time are obtained together, and using this α ₂ and {b _m } {η _k } corresponding thereto, the following equation ( 71) F (α ₂ , {b _m }, {η _k }) given in 71) is calculated.

Here, Σ of the sum with respect to _k means the sum of only the subcarriers used in each antenna element, and η _k means the offset value of the kth subcarrier used with the m-th antenna. _{F (α 2, {b m} }, {η k}) for such alpha ₂ in a, _{F (α 2, {b m} }, {η k}) and alpha ₂ when is minimum, The channel information of each subcarrier of each antenna element can be acquired using the set of Y-intercepts {b _m } at that time.

As a supplementary explanation, since the value of this Y intercept is originally given by f _c ΔL _m / c, it should have the periodicity of the equation (69) with respect to the antenna number m. However, considering the case where the center frequency is 80 GHz, for example, even when dividing by the speed of light of 3 × 10 ⁸ , if f _c / c is about 267 and the value of ΔL _m becomes 3.35 mm, the complex phase goes around once. It will be. Since it is observed in a state in which a return offset is given by exceeding 2π as in f _c ΔL _m / c + η, and the offset value η can be a relatively large value, the expression (70) It becomes impossible to grasp the proportional relationship.

  Therefore, a constraint condition such as Expression (70) set to the value b of the regression line for each antenna element cannot be expressed in a simple form, and each has a relationship to be evaluated as an independent Y-intercept value. This is the reason why it is difficult to calculate the exact solution of this problem. However, if the received weight vector is calculated using the channel information obtained by taking into account such effects and improving the accuracy of the above least squares method, the present embodiment can be applied with high accuracy. Is possible.

  In the example described above, an example is shown in which four subcarriers are used. However, the number of subcarriers may be other than four as a matter of course. If the number of subcarriers used here is small, the accuracy of the least squares method will decrease, but conversely, if the number of subcarriers is increased, the transmission power per subcarrier will decrease, so the SNR for channel estimation will decrease. The estimation accuracy is degraded. The accuracy of the least squares regression line slope a can be improved by using a plurality of receiving antennas in combination, but the accuracy of the Y intercept b cannot be improved by the above-described method. . Therefore, in the actual system design, the number of subcarriers used for channel estimation is optimized in a manner that balances these.

In the above description, the channel estimation by transmitting the training signal once has been described. However, as is clear from the equation (63), when acquiring the relative channel information with the reference antenna element, There is no necessity to perform channel estimation in the carrier at the same time. For example, if channel estimation is performed for only subcarriers f _A to f _{D and then} channel estimation is performed for other subcarriers f _A ′ to f _D ′ at different times, eight subcarriers Information can be acquired. If these processes are repeated, the accuracy of regression line fitting can be increased by using more subcarriers.

(Supplementary information on how to obtain a complete channel matrix)
In the above description, for example, in the case of the downlink, each component of the matrix of the second term on the right side of Equation (68) assuming that the terminal station device alone calculates the reception weight and transmission weight on the terminal station device side. Was undefined. However, if it is applied to a fixedly installed wireless entrance or train moving cell, it is possible to aggregate the information acquired on the downlink and the information acquired on the uplink and perform the processing after understanding both It is.

For example, if Equation (68) is downlink information, the component of the first column vector of the relative channel matrix H _relative obtained in the uplink is calibrated. If the component of the channel information between the j-th transmitting antenna and the i-th receiving antenna of the downlink k-th subcarrier converted by this calibration processing is h ′ _ij ^(k) , the above processing for the uplink is performed. h ′ ₁₂ ^(k) / h ′ ₁₁ ^(k) , h ′ ₁₃ ^(k) / h ′ ₁₁ ^(k) ,..., h ′ _1M ^(k) / h ′ ₁₁ ^(k) are obtained. . When each component of the downlink relative channel matrix H _relative is multiplied by the coefficient h ′ ₁₁ ^(k) , the (j, j) diagonal component of the second term matrix on the right side is h ′ ₁₁ ^(k) / h _1j. ^(K) . If this can be approximated by h ′ ₁₁ ^(k) / h ′ _1j ^(k), it can be _{calculated as} h ′ ₁₂ ^(k) / h ′ ₁₁ ^(k) , h ′ ₁₃ ^(k) / h obtained for the uplink. ^{_{'11 (k), ···,}} h' corresponds to the reciprocal of the ^{_{1M (k) / h '11}} (k). Therefore, it is possible to obtain all channel information by using this information.

  As a supplement, the above-described approximate solution of channel estimation is a channel estimation method using a condition in which only the line-of-sight wave is dominant. Of course, the absolute value of the first singular value in the single MIMO channel matrix is naturally used. The absolute value of the singular value below the second singular value with respect to the value is expected to be a very small value. Therefore, the MIMO channel itself is not suitable for spatial multiplexing of a plurality of signal sequences. However, efficient spatial multiplexing transmission is possible if a plurality of different antenna groups are used and a plurality of virtual transmission paths corresponding to the first singular value are used in parallel. If used for such a purpose, such an approximate solution of channel estimation functions effectively.

Furthermore, with regard to “optional function for improving channel estimation accuracy”, the accuracy is conscious of the regularity of the path length difference for each antenna element using the case where the receiving side antenna elements are arranged in a linear array. Signal processing for improvement has been described. Regarding the technology in the preceding stage, the path length difference does not impose any restrictions on the arrangement of the antenna elements as described as general ΔL _m . Since the discussion is based on the premise that the line-of-sight component is dominant, this technology can be applied even if the distance between the antenna elements is somewhat wide if the range of the offset value η _k is allowed to be slightly expanded. Is possible. In particular, when the antenna element interval is widened, the absolute values of the inner product values of the column vectors or row vectors of the relative channel matrix H _relative remain relatively high, but their components change almost irregularly. However, since restrictions on the antenna configuration and arrangement are not particularly limited as application conditions, the present embodiment can be applied under a wide range of conditions as long as the line-of-sight wave is dominant.

(Processing flow of approximate solution for channel matrix acquisition when line-of-sight wave is dominant)
The processing operation of the approximate solution method for obtaining the channel matrix will be described below with reference to the drawings. Here, no distinction is made between uplink and downlink, and either the base station apparatus or the terminal station apparatus transmits a training signal for channel estimation, and the other receives the training signal and acquires a channel matrix. If a similar process is performed in the reverse direction after acquiring the channel matrix in one direction, a bidirectional channel matrix can be acquired. When performing transmission / reception on the virtual transmission line corresponding to the first singular value, a complete channel matrix is not necessarily required, and it is only necessary to obtain a relative channel matrix H _relative as shown in Expression (68). The channel matrix acquisition method will be described. For convenience of explanation, the case where a training signal is transmitted on the downlink will be described as an example.

  Next, with reference to FIG. 53, the processing operation of the approximate solution for obtaining the channel matrix in this embodiment will be described. FIG. 53 is a flowchart showing the processing operation of the approximate solution for channel matrix acquisition in the present embodiment. First, when the base station apparatus transmits a training signal (step S5301), the terminal station apparatus receives the training signal at each antenna element (step S5302). In this reception, the signal is amplified by a low noise amplifier, downconverted from a radio frequency to a baseband signal by a mixer, a signal component outside the band is removed by a filter, and sampling is performed by A / D conversion. This signal is subjected to FFT processing and converted to a signal on the frequency axis.

The training signal here is a training signal in which a transmission signal exists only on a specific subcarrier on the frequency axis in order to increase the transmission power per subcarrier. The subcarriers including the signal component are extracted from the subcarriers and set as channel estimation results of the subcarriers (step S5303). Then, the terminal station apparatus divides this result by the similar channel estimation result of the reference antenna to obtain _cm ( _k ) of Expression (63) (step S5304).

Next, the terminal station apparatus obtains Y _m (f _k ) from Expression (65) (Step S5305), and calculates a and b that minimize Expression (66) by the least square method considering the offset value η _k. (Step S5306). Subsequently, the terminal station apparatus performs channel estimation for each subcarrier by substituting into the equation (67) based on a and b obtained here (step S5307), and further performs calibration to perform bidirectional processing. Relative channel matrix is obtained (step S5308), and this is recorded and managed as a relative channel matrix (step S5309).

  On the other hand, recording management of the transmission / reception weight vector is more important than recording management of the channel matrix. In this case, after the terminal station apparatus obtains the channel estimation result of each subcarrier by Equation (67), Value decomposition is performed to obtain a reception weight vector as the first left singular vector (step S5310). Next, the terminal station apparatus performs a calibration process on the reception weight vector (step S5311), and obtains a transmission weight vector. Then, the terminal station apparatus records and manages the transmission / reception weight vector acquired in this way (step S5312).

The above is a case where channel estimation is performed while closed to a single antenna element, but it is also possible to improve estimation accuracy in consideration of a plurality of antenna elements as described above. FIG. 54 is a flowchart showing the processing operation of the approximate solution of the channel matrix using a plurality of antennas. Similarly to the processing operation shown in FIG. 53, when a training signal is received by each antenna element in the terminal station apparatus, a predetermined reception process is performed (steps S5401-1 to S5401-3), and signal components in discontinuous subcarriers. Are extracted as subcarrier channel estimation results (steps S5402-1 to S5402-3). Subsequently, the terminal station apparatus divides this result by the same channel estimation result of the reference antenna to obtain _cm ( _k ) of Equation (63) (steps S5403-1 to S5403-2), Y _m (f _k ) is obtained for each antenna element by Expression (65) (steps S5404-1 to 5404-2).

Here, the terminal station apparatus sets, for example, a value of α ₂ with a predetermined step size, and a slope of the regression line as a = (m−1) α _{2 with} respect to Y _m (f _k ) of the m-th antenna element. are fixed and determine the b by the least square method in consideration of the offset _{value, F (α 2, {b} m}, {η k}) of the formula (66) determine the alpha ₂ that minimizes (step S5405) Then, using the α ₂ , the least square method considering the offset value in each antenna element is performed using the equation (71) (step S5406), and the equation (67) is obtained based on a and b obtained here. Substituting and performing the channel estimation value of each subcarrier (step S5407), further performing calibration (step S5408), obtaining a bidirectional relative channel matrix, and recording and managing this (step S5409).

  On the other hand, recording management of the transmission / reception weight vector is more important than recording management of the channel matrix. In this case, after the terminal station apparatus obtains the channel estimation result of each subcarrier by Equation (67), Value decomposition is performed to obtain a reception weight vector as the first left singular vector (step S5410). Next, the terminal station apparatus performs a calibration process on the reception weight vector (step S5411) to obtain a transmission weight vector. Then, the terminal station apparatus records and manages the transmission / reception weight vector acquired in this way (step S5412).

  Next, with reference to FIG. 55, processing operation for obtaining an approximate solution of a channel matrix from bidirectional channel estimation results will be described. FIG. 55 is a flowchart showing a processing operation for obtaining an approximate solution of a channel matrix from a bidirectional channel estimation result. First, the downlink will be described. When the base station apparatus transmits a training signal (step S5501), the terminal station apparatus receives the training signal at each antenna element (step S5502).

The training signal here is a training signal in which a transmission signal exists only on a specific subcarrier on the frequency axis in order to increase the transmission power per subcarrier. The subcarriers including the signal component are extracted from the subcarriers and used as channel estimation results for the subcarriers (step S5503). Then, the terminal station apparatus divides this result by the similar channel estimation result of the reference antenna, and obtains c _m ( _k ) in Expression (63) (step S5504).

Next, the terminal station apparatus obtains Y _m (f _k ) from Expression (65) (Step S5505), and calculates a and b that minimize Expression (66) by the least square method considering the offset value η _k. (Step S5506). Next, the terminal station apparatus performs channel estimation of each subcarrier by substituting into the equation (67) based on a and b obtained here (step S5507).

  Next, the uplink will be described. When the terminal station device transmits a training signal (step S5508), the base station device receives the training signal at each antenna element (step S5509).

The training signal here is a training signal with missing teeth on the frequency axis in order to increase the transmission power per subcarrier, and the base station apparatus includes signal components in the discontinuous subcarriers. A subcarrier is extracted and used as a channel estimation result of the subcarrier (step S5510). Then, the base station apparatus divides this result by the similar channel estimation result of the reference antenna, and obtains c _m ( _k ) in Expression (63) (step S5511).

Next, the base station apparatus obtains Y _m (f _k ) from Expression (65) (Step S5512), and calculates a and b that minimize Expression (66) by the least square method considering the offset value η _k. (Step S5513). Subsequently, the base station apparatus performs channel estimation of each subcarrier by substituting into Equation (67) based on a and b obtained here (step S5514).

  Next, an absolute channel matrix is calculated from the uplink and downlink relative channel matrices using equation (68) (step S5515). Then, calibration is performed (step S5516), and this is recorded and managed as a bidirectional channel matrix (step S5517).

  Here, the calibration process in step S5516 is basically performed by the process from step S5502 to step S5507 and the individual process from step S5509 to step S5516. Other than indefiniteness (relative relationship of channel information) can be acquired. For this reason, if the inconsistency of the row vector component of the first row in both uplink and downlink is compensated by matching each information in step S5515, calibration is not performed (that is, step S5516 is omitted). It is also possible to obtain a bidirectional channel matrix.

As described above, the above explanation is a procedure for obtaining the relative channel matrix. However, if information on both directions of the uplink and downlink is known, not only the relative channel matrix but also the absolute value of the channel matrix can be obtained. Can be sought. Similarly to the above description, after receiving the training signal by each antenna element, channel estimation of each subcarrier is performed by Expression (67) by predetermined processing. This is acquired separately on the downlink terminal station apparatus side and the uplink base station apparatus side. Both the obtained results are collected and (h ₁₁ ^(k) / h ₁₁ ^(k) , h ₁₂ ^(k) / h ₁₁ ^(k) , h ₁₃ ^(k) / h ₁₁ ^(k) ,..., H _1M ^(k) / h ₁₁ ^(k) ) is multiplied from the right of the relative channel matrix by multiplying the matrix having the diagonal component by 1 / h ₁₁ ^(k) into the first item matrix on the right side of Equation (68). The multiplied absolute channel matrix can be obtained. If this is subjected to calibration processing, a bidirectional channel matrix can be obtained, and this is recorded and managed.

  As described above, in a line-of-sight environment and when the antenna element spacing is narrow, even if the channel information has frequency dependence, it is expected to show linear frequency dependence in terms of the complex phase of the relative channel information. it can. Moreover, it is possible to improve the channel estimation accuracy before directivity formation by limiting the subcarriers used for the training signal and increasing the transmission power per subcarrier. Using these features, while performing channel estimation with a small number of subcarriers, it is possible to provide a method of estimating with high accuracy by estimating channel information of subcarriers that have not been acquired in a least-squares manner. . Furthermore, if different subcarriers are used for each antenna element, it is possible to simultaneously transmit training signals from a plurality of antenna elements, and it is possible to cope with a plurality of transmission antennas that cannot be solved by implicit feedback. Channel information can be efficiently fed back simultaneously to a plurality of antenna elements or antenna element groups.

  Note that since the actual channel includes reflected waves, the channel information is not linear on the frequency axis and includes many errors, but if least squares processing is performed at a larger number of samples, the line-of-sight component It is expected that features can be extracted.

[Seventh Embodiment]
[Approximate solution of weight vector corresponding to multiple first singular values]
(Outline of the basic principle according to the seventh embodiment)
In the case of normal MIMO transmission, it is considered ideal that the antenna element spacing is as far as possible for the purpose of reducing the correlation between the antenna elements. However, in order to actively use the virtual transmission line corresponding to the first singular value as described above, it is better to narrow the antenna element interval and increase the correlation. Of course, if the antenna element spacing is ½ wavelength or less, the mutual coupling between the antenna elements cannot be ignored. Therefore, it is necessary to set the spacing of about ½ wavelength, but the spacing of that degree is set. If so, it is ideal to increase the correlation by narrowing the antenna element spacing. In the above-mentioned approximate solution method of channel estimation, a method that can be applied without restricting the antenna arrangement has been described. It is possible to calculate the transmission / reception weight by a simpler method.

  Here, although the channel correlation of the channel matrix itself is large, for example, when the channel information between the first transmitting antenna and the first receiving antenna is compared with the channel information of the second antenna with the same first antenna for transmission, Most of the values show completely different values due to the path length difference. In order to perform singular value decomposition, the same calculation is required regardless of whether the channel correlation is strong or weak, so even if only the right singular vector or the left singular vector related to the first singular value is obtained, Similarly, the load required for computation is not light. In particular, since the number of antenna elements is enormous, the matrix size becomes enormous, and it is necessary to calculate the right singular vector and the left singular vector corresponding to the first singular value by a simple calculation.

  Here, considering that the correlation of the antenna is high, an approximate solution of the right singular vector and the left singular vector corresponding to the first singular value is obtained by the following method. First, when the first left singular vector is used as the reception weight vector, a virtual one antenna element is formed by multiplying the reception signal of each reception antenna element by the weight vector. Since this forms a beam tuned to a line-of-sight wave, it is expected that the beam is directed from the receiving antenna group toward the transmitting antenna group. Since the antenna elements on the transmitting side are installed with an interval of at least about ½ wavelength or more, strictly speaking, the first right singular vector for singular value decomposition is formed in consideration of its spread. However, since it is substantially close to a state where one virtual antenna element is arranged in a very narrow area, reception toward one specific antenna installed near the center of gravity of these transmission antenna groups It is expected that the weight vector can be approximated.

Therefore, a reception weight vector directed to one specific transmission antenna element near the center of gravity of the transmission antenna group is obtained, and on the transmission side, a gain when transmitting using the first right singular vector obtained by singular value decomposition, Consider a case where the right and left singular vectors corresponding to the first singular value are used for both the transmission and reception weight vectors. First, as an example, a channel matrix is assumed in which the (i, j) -th component of the matrix is given by the line-of-sight model of Expression (20) with 31 antennas for transmission and reception. The first right singular vector when the singular value decomposition is performed as in Expression (4) is v ₁ , and the first left singular vector is u ₁ . The gain when the transmission / reception weight vector is formed based on these is given by the following equation (72).

On the other hand, the SIMO channel vector (h _1,16 , h _2,16 , h _3,16 ,..., H _31,16 ) ^T between the 16th element at the center of 31 elements of the transmitting antenna group and the receiving antenna group. On the other hand, a vector obtained by standardizing the Hermitian conjugate vector is approximately approximated to the reception weight vector w _rx-app. (Formula (73)).

  The gain in this case is as shown in equation (74) below.

  Furthermore, in addition to one-way, in both directions, a training signal is transmitted from the antenna elements near the center of gravity of the antenna group to obtain both received weight vectors, and a transmission weight vector is obtained therefrom by calibration processing. By doing so, it is possible to simplify the transmission weight vector and the reception weight vector from the same channel information feedback to the transmission / reception weight vector calculation process. The gain at this time is given by the following equation (75).

Here, G _rx-app. -G _ideal and G _app. The result of evaluating the distribution characteristics of -G _ideal is shown in FIG. The evaluation condition is that the 31 elements of the antenna are arranged as a linear array on the xy two-dimensional plane with one wavelength interval in parallel to the y axis, and ± 5 [ m], the installation point was determined with a uniform random number in the range of ± 5 [m] in the y-axis direction, and the CDF distribution of the gain difference was evaluated. As parameters, the case where the frequency is 20 GHz and the distance L in the x direction is 20 m, 30 m, and 100 m is evaluated.

Looking at these results, there is almost no difference between using the approximate weight vector and using the ideal weight vector by singular value decomposition, and the closer the distance between the transmitting and receiving stations, the larger the gain gap, but the distance is It can be seen that even at 20 m, the gain difference is within 0.25 dB. That is, in the present embodiment, when signal transmission is performed using a virtual transmission line corresponding to the first singular value, the reception weight vector w _rx-app. And a calibration process is performed on the transmission weight vector w _tx-app. By performing the signal transmission / reception by obtaining the above, it is possible to perform a communication comparable to the case of using the optimum transmission / reception weight vector obtained by the singular value decomposition. Although the 20 GHz band has been described as an example here as a frequency, when the frequency becomes higher, such as the 80 GHz band, the antenna element interval can be shortened by shortening the wavelength, so that it is physically reduced to almost one point. Since the antenna is arranged in the vicinity, the approximation accuracy is further increased and the gain gap can be ignored.

(Option for approximate solution of weight vector corresponding to multiple first singular values)
The above is the basic principle of a simple weight vector calculation method, but some problems remain as it is. For example, when gain is gained in line design by using a large-scale antenna as described above, the accuracy of channel estimation of one antenna to one antenna cannot be said to be very high. Therefore, the number of subcarriers used for channel estimation is greatly limited, as in the technique used in the “approximate solution for channel matrix acquisition when the line-of-sight wave is dominant” (sixth embodiment). Is applied to improve the channel estimation accuracy for the subcarrier. Furthermore, if the forehead between antenna elements is used, channel estimation with fairly good accuracy can be maintained even if the transmitting antenna for transmitting training signals in channel estimation is slightly away from the vicinity of the center of gravity of the antenna group. Become.

  For example, when the same evaluation as in FIG. 56 is performed, it is considered that a transmission / reception weight vector is obtained using an antenna element slightly apart from the center of gravity without using an antenna near the center of gravity. As an example, in a case where 61 elements are arranged in a linear array at ½ wavelength intervals in the same manner as in the evaluation of FIG. 56 and a transmission / reception weight vector using the 31st antenna element corresponding to the center of gravity is used, When using the element, the gain when using the 61st antenna element at the end of the linear array is evaluated as the amount of deterioration from the case of using an ideal transmission / reception weight using singular value decomposition for both transmission and reception.

  FIG. 57 is a diagram showing gain characteristics in a transmission / reception weight vector approximate solution using an antenna away from the center of gravity. As can be seen from the figure, the median value of the gain gap (CDF 50% value) is 0.5 dB for the 46th antenna element, even for the 61st antenna element, even under severe conditions such as a frequency of 20 GHz and a distance of 20 m. 0.8 dB. If this level of gain reduction is allowed, all 61 antenna elements are used to transmit training signals of different subcarriers, and training signals of all frequency components transmitted from any of the antenna elements are transmitted. The transmission / reception weight vector in the receiving station can be easily obtained.

  For example, if a training signal of 4 subcarriers is transmitted from each antenna element and the number of antenna elements is 64 elements, for example, it is possible to simultaneously perform channel estimation for 256 subcarriers. In the above example, a linear array has been described as an example for the sake of simplicity. However, if a 16 × 16 square array antenna (256 elements in total) having a square lattice with a ½ wavelength interval is used, the above-described FIG. Since similar characteristics are expected, it is possible to estimate the channel of 1024 subcarriers at a time by using 256 elements simultaneously.

  Assuming that the uplink is assumed here, there are cases where not so many antenna elements can be arranged on the terminal station side, but even in that case, it was explained in “Approximate Solution for Channel Matrix Acquisition when Line-of-Sight Wave is Dominant”. Similarly, it is known that the weight value between subcarriers of each component of the transmission / reception weight vector vector (corresponding to individual antenna elements) shows a gradual fluctuation in terms of complex phase. In the complex phase graph, if the offset value is corrected in consideration of the return of 2π period, it is assumed that the weight value is within a linear and gradual change within the entire band. It is also possible to acquire a subcarrier weight value that cannot be acquired by linear interpolation while extending the complex phase of the weight value outside the range of -π to + π. In this case, it is preferable to use relative channel information based on the complex phase of the reference antenna as channel information, and obtain a reception weight from the relative channel information.

  Here, an example of using antenna elements near the center of gravity of the entire antenna will be described with reference to FIG. FIG. 58 is a diagram illustrating an example of an antenna element used for transmission of a training signal when a linear array is used. In this figure, the case of a 31-element linear array is taken as an example, and its center-of-gravity antenna is the 16th antenna in the middle (indicated by a double circle in the figure). Basically, as shown in FIG. 58 (a), only the sixteenth element, which is the antenna element at the center of gravity, is used for transmission of the training signal, but in this case, transmission of the training signal is performed in order to improve transmission power per subcarrier. If the subcarriers used for the above are limited, the number of subcarriers from which channel information can be acquired decreases, and the accuracy of interpolation of channel information during that time decreases. Therefore, for example, as shown in FIG. 58 (b), the training signal may be transmitted within the range of the center of gravity ± 5 elements, or the training signal may be transmitted within the range of ± 10 elements as shown in FIG. 58 (c). Good. As shown in FIG. 57, if there is no problem in channel estimation accuracy, all elements can be used for transmission of training signals as shown in FIG.

  Next, FIG. 59 is a diagram illustrating an example of an antenna element used for transmission of a training signal when a close-packed two-dimensional antenna arrangement is used. In this figure, the case of a two-dimensional array of 37 elements is taken as an example, and symbols a to z and A to K are given for convenience of explanation. The center-of-gravity antenna is the a-th antenna in the middle (indicated by double circles in FIG. 59). Since the antenna element at the center of gravity is the basis as shown in FIG. 59 (a), only the a-th element is used for training signal transmission. However, in order to improve the transmission power per subcarrier, When the subcarriers used for transmission are limited, the number of subcarriers from which channel information can be acquired decreases, and the accuracy of interpolation of channel information during that time decreases. Therefore, for example, as shown in FIG. 59 (b), the training signal is transmitted in the range of the bth to sth elements (indicated by black circles in FIG. 59) in addition to the ath element, and the remaining tth to tth elements are transmitted. The K element antenna is used for communication but not for channel feedback. Or the structure which uses all these antenna elements for both transmission of a training signal and data communication may be sufficient.

  The same idea can be applied when the number of antenna elements that can be used for transmission of training signals is small. For example, when a one-dimensional linear array is used for transmission and reception for some reason, only antenna elements near the center of gravity as shown in FIG. 58B can be used for transmission of training signals in terms of channel estimation accuracy. Separately, an antenna element used only for training signal transmission can be added and operated. FIG. 60 is a diagram illustrating an example in which an antenna element for training signal transmission is added near the center of gravity of the linear array. As shown in this figure, the antenna elements E, D, C, B, A, z, r, g, a, d, l, t, u, v, w, x, and y constitute a linear array. The center of gravity is the a-th element, and the a to s-elements are arranged in a close-packed manner in the center of this element. The antenna elements b, c, e, f, h, i, j, k, m, n, o, p, q, and s that are out of the linear array are not used for data communication. A training signal is transmitted using the a to s elements.

  This is an example. For example, even when a close-packed two-dimensional antenna arrangement as shown in FIG. 59 is used, the training signal is 1 around the a-th element while the close-packing is performed at intervals of 5 wavelengths. A configuration may be employed in which antenna elements are intensively arranged and used in a close-packed manner at an interval of two wavelengths. In this way, the density of the antenna elements may be increased near the center of gravity, and the training signal may be transmitted using only the antenna elements near the center of gravity. In the description of FIG. 60, the antenna elements b, c, e, f, h, i, j, k, m, n, o, p, q, and s deviating from the linear array are not used for data communication. There is no problem even if these antenna elements are used for data communication.

FIG. 61 is a diagram showing a specific example of linear interpolation of approximate weight values. The horizontal axis of the graph represents subcarriers, and the vertical axis represents the complex phase of weight values. For example, it is assumed that the weight values of f ₁ , f ₉ , f ₁₆ , and f ₂₅ are acquired as plotted points 291 to 294 at intervals of 8 subcarriers, respectively. Since this complex phase is given in the range of π to + π, for example, when linear interpolation is performed based on the plot points 291 to 295, the complex phase of the weight value of f ₂₅ exceeds + π, and actual observation The plotted point 294 has a value near −π. However, since the complex phase cannot change close to −2π when the subcarrier changes from f ₁₆ to f ₂₅ , the plot point 295 inevitably obtained by adding the offset of + 2π to the plot point 294 is the actual complex. It can be considered as a phase.

If the complex phase value corrected by adding an offset in this way is used, the complex phase between them can be obtained by linear interpolation. For example, regarding the subcarriers f _{2 to} f _8, it is also possible to connect the plot points 291 and 292 with a straight line and perform interpolation. Similarly, if the least square method is applied using the plot points 291 to 293 and the plot point 295, a linear regression line can be obtained, and therefore, the weight value of each subcarrier may be given by the regression line. At this time, the acquired weight values of the subcarriers f ₁ , f ₉ , f ₁₆ , and f ₂₅ can be used as they are, but on the other hand, instead of using them as they are, they are replaced with predicted values obtained by linear interpolation. If a carrier weight value is given, an estimation error component such as noise may be suppressed. Furthermore, the amount of change taking into account the 2π offset from the weight value of the neighboring subcarriers is extremely large (for example, the amount of change is ± π / 2 or more) in addition to other plot points. It is also possible to increase accuracy by excluding it from linear interpolation because it has low accuracy.

  Next, with reference to FIG. 62, the operation of the complex phase offset determination process in the linear interpolation of the approximate weight value performed by the base station apparatus or the terminal station apparatus will be described. FIG. 62 is a flowchart showing the operation of a complex phase offset determination process in linear interpolation of approximate weight values. Here, the processing operation shown in FIG. 62 will be described as being performed by the base station apparatus. If the number of subcarriers used for channel estimation can be secured to some extent (including the case where training signals are transmitted from a plurality of antenna elements), the interval between subcarriers on which channel estimation is performed can be made relatively small. The difference in complex phases of channel estimation results of adjacent subcarriers is considered to be sufficiently small.

  Therefore, after performing channel estimation with discrete subcarriers, the base station apparatus reads each component of the transmission / reception weight vector (or channel vector or channel matrix) for complex phase information interpolation (step S6201). Then, the base station apparatus reads the weight values Wk and Wk ′ of successive subcarriers (step S6202). At this time, for each vector or matrix, the acquired channel information is read in order, for example, from the smaller subcarrier number. Then, the base station apparatus obtains a complex phase θ (Wk) of Wk and Wk ′ of consecutive subcarriers acquired (because they are discrete, k and k ′ are not necessarily adjacent to each other). θ (Wk ′) is acquired (step S6203). Subsequently, the base station apparatus adds + 2π if the difference between the complex phases θ (Wk ′) and θ (Wk) is −π or less (steps S6204 and S6205). 2π is subtracted (steps S6206 and S6207) so that the absolute value of the difference between the corrected complex phases θ (Wk) and θ (Wk ′) is π or less.

  Basically, this is sufficient, but when complex phase difference information of ± π / 2 or more is judged to have low credibility, corrected complex phases θ (Wk) and θ (Wk ′) Are compared again (steps S6208 and S6209). If the difference is −π / 2 or less or π / 2 or more, it is determined that the channel information is not reliable, and the weight value Wk ′ is discarded (step S6208). S6210). If all subcarriers that have been acquired have been inspected, the process ends. Implement (step S6211).

  Although correction processing is performed for the complex phase, since the actual weight value is given by Exp (jθ (k), the correction of the complex phase 2π does not cause any difference in weight. It is possible to apply an arbitrary interpolation process on the complex phase corrected in this way, and linearly interpolate from the two adjacent complex phases. Alternatively, linear interpolation may be performed in a least-squares manner from complex points at multiple points, or other non-linear interpolation may be performed. Extrapolation may be used.

  In addition, in the above various explanations, “offset value” was introduced by linear interpolation (regression calculation by least squares method) such as channel vector, channel matrix, transmission / reception weight vector, etc. The correction of the complex phase performed without causing a sudden change in complex phase between the subcarriers is basically equivalent to the introduction of the “offset value”. If the processing of FIG. The introduction of “offset value” (processing such as performing regression calculation by a least square method on a plurality of offset value candidates) is not necessarily required, and can be omitted. The correction process described above can be used in the same way in the process according to the sixth embodiment.

(Processing of approximate solution of weight vector corresponding to a plurality of first singular values)
Next, with reference to FIG. 63, the processing operation of the approximate solution of the weight vector corresponding to the plurality of first singular values will be described. FIG. 63 is a flowchart showing the processing operation of the approximate solution method for weight vectors corresponding to a plurality of first singular values. Here, regardless of uplink or downlink, either the base station apparatus or the terminal station apparatus transmits a training signal for channel estimation, and the other receives the training signal to acquire a channel vector. Is used to calculate the reception weight vector. Further, a transmission weight vector is acquired by the calibration process. If the same operation is performed in the reverse direction after acquiring the transmission / reception weight vector in one direction, it is possible to acquire the bidirectional transmission / reception weight vector. Further, for convenience of explanation, the following explanation will be given with an example in which a training signal is transmitted on the downlink.

  In order to increase the transmission power per subcarrier, the base station apparatus transmits a training signal using some of the subcarriers in the whole (step S6301). At this time, the training signal is transmitted by the antenna element around the center of gravity of the antenna group. The terminal station apparatus receives this training signal and performs a predetermined reception process (step S6302). In this reception process, the signal is amplified by a low noise amplifier, downconverted from a radio frequency to a baseband signal by a mixer, an out-of-band signal component is removed by a filter, and sampling is performed by A / D conversion. For example, the signal is subjected to FFT processing and converted to a signal on the frequency axis. The terminal station apparatus takes a complex conjugate (and includes a normalization process as necessary) for the channel vector thus obtained, and calculates a reception weight vector (step S6303).

  Subsequently, the terminal station apparatus performs the complex phase correction process (the process shown in FIG. 62) here (step S6304). Then, the terminal station apparatus acquires channel vectors of all subcarriers by linear interpolation or the like (step S6305). Subsequently, the terminal station apparatus predicts the reception weight vector of the subcarrier that cannot be acquired by various interpolation processes such as linear interpolation and nonlinear interpolation, and calculates the transmission weight vector by the calibration process after acquiring all the reception weight vectors. (Step S6306). Finally, the terminal station apparatus records and manages the transmission weight vector (step S6307).

  As described above, when acquiring transmission / reception weights for using the virtual transmission line corresponding to the first singular value in the line-of-sight environment, the fact that the antenna element group is limited to a very narrow area is used. The reception weight can be calculated very easily by acquiring the reception signal vector using a single antenna element near the center of gravity of the antenna group and taking the complex conjugate thereof. If the calibration process is performed, the transmission weight can be acquired. If this process is performed in both directions, the base station apparatus and the terminal station apparatus can easily approximate the transmission / reception weight of the virtual transmission path corresponding to the first singular value. Can be requested. In addition, while only the antenna near the center of gravity is used for training signal transmission, it is also possible to improve the line gain and enhance the spatial multiplexing characteristics by using the entire antenna element having a wide aperture for data transmission / reception.

  If the transmission weight vector is known on the transmission side, it is not necessary to transmit the training signal by using only the antenna near the center of gravity and limiting the subcarriers to be used. It is also possible to transmit a training signal with increased gain by performing directivity formation using transmission weight vectors with all antenna elements using carriers. Therefore, this embodiment is used as a method for acquiring the initial value of the transmission weight vector, and after acquiring the transmission weight vector, directivity formation is performed using the transmission weight vector, and a training signal is transmitted. The channel is estimated with the line gain increased at the same time, the received weight vector is obtained with high accuracy by the complex conjugate of the channel vector, and the high-accuracy received weight vector is subjected to calibration processing to achieve high accuracy. A procedure such as obtaining a transmission weight vector may be taken. By sequentially repeating such processing, communication can be performed in a more stable situation in a steady communication state.

[Eighth Embodiment]
[About simultaneous channel estimation in multiple radio stations]
(Outline of the basic principle according to the eighth embodiment)
In the above-mentioned “approximate solution for acquiring channel matrix when line-of-sight wave is dominant” and “approximate solution for weight vector corresponding to a plurality of first singular values”, the number of subcarriers for transmitting training signals is limited to 1 It was shown that the channel estimation accuracy of this single antenna pair can be improved. Also, with these techniques, it is possible to simultaneously transmit training signals with different subcarriers from a plurality of antennas, thereby acquiring channel information regarding different antenna elements.

  In the description of these techniques, for example, a training signal is transmitted from a plurality of antenna elements of the same terminal station apparatus using different subcarriers, or a plurality of first signals are transmitted on the base station apparatus 303 side as shown in FIG. The case where the processing unit 304 is included has been described. In this case, the transmission of the training signal from the antenna element of the first signal processing unit 304 is adjusted by the communication control circuit or the like inside the base station apparatus 303, and the training signal is transmitted all at once with the same timing. It was controlled to do. On the other hand, it may be effective to perform channel estimation of a plurality of antenna elements simultaneously under other conditions. As an example, in addition to applying the present embodiment to a wireless entrance line as described above, it can be used for an access system of a small cell base station apparatus.

  FIG. 64 is a diagram showing a device configuration for performing simultaneous channel estimation between a plurality of small cells existing in the same area. In FIG. 64, for example, the left diagram shows an environment in which a plurality of small cell base station devices 231 to 234 are installed on the wall surface of a building street as an example, and each service area extends around the base station devices 231 to 234. ing. In this service area, there are terminal station devices 235 to 239 that perform wireless communication with the base station devices 231 to 234. The terminal station device 235 is under the control of the small cell base station device 231, the terminal station device 236 is under the control of the small cell base station device 232, the terminal station device 237 is under the control of the small cell base station device 233, and the terminal station The device 238 and the terminal station device 239 are under the small cell base station device 234.

  As described in the background art, the small cell is installed in order to offload the traffic in the area where the traffic is concentrated locally from the macro cell. A small cell in a small area is set for an area where traffic is concentrated, and the small cell base station apparatus performs communication only for terminal stations in the small cell. In the case of a macro cell, it is common to perform station station design strictly and manage the level of mutual interference between macro cells to reuse frequency resources. However, small cells are installed in spots according to requirements such as traffic concentration, and station placement design is not always implemented systematically.

  In such an environment, a small cell configuration technique using the Massive MIMO technique has been attracting attention for the purpose of reducing mutual interference in accordance with the requirement for securing a line gain. In this small cell, in order to reduce mutual interference, transmission power per element is not high, but a huge line gain is secured by directivity gain formed by a large number of antenna elements. However, in order to secure the line gain, the MIMO channel matrix between each transmitting antenna element and the receiving antenna element needs to be known, and a channel feedback technique for this purpose is important.

  Furthermore, in order to obtain this line gain efficiently, the accuracy of channel estimation is also important. However, when a training signal is transmitted for channel estimation, interference waves from neighboring small cells are transmitted on the same subcarrier. In the case of reception, the channel estimation accuracy deteriorates significantly due to the interference signal, and not only a sufficient line gain cannot be secured, but also the radiation in a direction other than the desired direction increases, and thus the risk of increased mutual interference. There is also. This problem is called pilot contamination, and the problem of contamination of the training signal due to mutual interference is an important problem.

  To simply avoid this problem, shift the transmission timing of the training signal for each small cell (if the training signal is transmitted in some nearby small cell, refrain from communication in other small cells) Was considered effective. However, with this method, the overhead due to the channel estimation training signal increases and the transmission efficiency in the MAC layer decreases. In particular, in a small cell in 5G, since it is assumed that a single base station apparatus collectively accommodates a large number of terminal station apparatuses, these terminal station apparatuses frequently transmit training signals, and each small cell If the transmission timing is divided, the MAC efficiency is greatly reduced.

  Therefore, as shown in the right diagram of FIG. 64, among the many subcarriers in the entire frequency bandwidth W, the base station apparatuses 231 to 234 segregate the subcarriers, and at the same time in the downlink Send a training signal in line. Similarly, the terminal stations 235 to 238 of each small cell also segregate subcarriers among a number of subcarriers in the entire frequency bandwidth W, and arrange training signals at the same time on the uplink. Send. For example, the base station device 231 and the terminal station device 235 use a plurality of subcarrier groups 241 in the right diagram, the base station device 232 and the terminal station device 236 use a plurality of subcarrier groups 242, The terminal station apparatus 237 uses a plurality of subcarrier groups 243 shown in the right figure, and the base station apparatus 234 and the terminal station apparatus 238 use a plurality of subcarrier groups 244 shown in the right figure. Since these do not overlap on the frequency axis, there is no interference of the same frequency (the same subcarrier) at the same time, and the problem of Pilot Contamination does not occur.

  Also, regarding the uplink, for example, the terminal station device 238 and the terminal station device 239 subordinate to the base station device 234, for example, the terminal station device 238 uses the subcarrier group 244, while the terminal station device 239 (the base station device 239). 233 and the terminal station device 237 (provided that the subcarrier group 243 does not communicate), a training signal is transmitted using the subcarrier group 243, and channel information of the terminal station device 238 and the terminal station device 239 is estimated simultaneously. It is also possible to configure.

  In general, the base station apparatus is allowed to increase the apparatus scale, while the terminal station apparatus cannot increase the apparatus scale so much. In addition, as shown in FIG. 16, when a plurality of first signal processing units 304 are provided and each has a plurality of antenna elements, the total number of antennas included in one base station device is The total number of antennas is expected to be small. In that case, it is reasonable to simultaneously perform channel estimation with a large number of terminal station apparatuses in the same cell (or may include different cells) as described above.

  Specifically, for example, it is assumed that the base station apparatus includes five first signal processing units 304, each of which includes 50 antenna elements. When a training signal is transmitted on four subcarriers per antenna element, channel estimation of 1000 subcarriers can be performed simultaneously with 4 × 50 × 5 = 1000. Further, if the total number of subcarriers is 2000, for example, base station apparatus 231 and base station apparatus 233 perform channel estimation using even subcarriers, and base station apparatus 232 and base station apparatus 234 use odd subcarriers. Is possible. Although depending on the directivity characteristics of the antenna, etc., if each antenna element on the base station device side has a certain degree of directivity, it may interfere with the adjacent small cell, but In some cases, the interference may be negligible with the small cell. In such a case, the above-described setting is established.

  On the other hand, if the number of antenna elements provided in the terminal station apparatus is 25 elements, 100 subcarriers are used in total for one terminal station apparatus, assuming that 4 subcarriers are used per element. However, since one base station apparatus uses 1000 subcarriers for channel estimation, even if 10 terminal station apparatuses transmit uplink channel training signals simultaneously, the same frequency ( There is no interference in the same subcarrier), and channel estimation can be performed on these simultaneously. Regarding the downlink, when the base station apparatus transmits a training signal once, all the terminal station apparatuses that have received the training signal can simultaneously estimate the downlink channel for the own station. However, with respect to the uplink, each terminal station apparatus must individually transmit a training signal, and how to prevent a decrease in MAC efficiency due to overhead associated with the transmission of the training signal, while increasing the channel estimation accuracy. Is a challenge.

  Here, in order to simultaneously transmit and receive the above channel estimation between different small cells or between different terminal station devices in the same small cell, (1) training to be transmitted by each terminal station device It is important that the subcarrier of the signal can be grasped, and (2) that the timing at which each terminal station apparatus should transmit the training signal can be grasped.

FIG. 65 is a diagram showing an example of a frame configuration in the present embodiment. In FIG. 65, reference numerals 251-1 to 251-3 are slots for the base station apparatus to transmit a training signal in the downlink, and reference numerals 252-1 to 252-3 are data communication between the base station apparatus and the terminal station apparatus. It is a slot to do. Reference numerals 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-6 are used for the terminal station apparatus to transmit a training signal on the uplink. Represents a slot. Also, the time length from slot 251-1 to slot 256-1, the time length from slot 251-2 to slot 256-2, and the time length from slot 251-3 to slot 256-6 are respectively the basic frame period. T _b-frame . Strictly speaking, since switching from the uplink to the downlink is performed between the end of the slot 256-1 and the slot 251-2, between the end of the slot 256-2 and the slot 251-3, and the like, the guard time A predetermined time may be set as

  In the example shown in FIG. 65, a long-period superframe structure in which a basic frame is repeated N times is taken, and slots 253-1 to 253-3, 254-1 to 254-3 in different basic frames in this superframe, 255-1 to 255-3 and 256-1 to 256-3 are assigned to transmit uplink training signals by different terminal station apparatuses. If attention is paid to any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-6, each is divided into groups of subcarriers. Has been. In the above-described example, for a total of 1000 subcarriers, one terminal station apparatus transmits 25 training elements on each of 4 subcarriers. For this reason, for example, subcarriers of the jth group (j is an integer of 1 to 10) use the (10 × m + j) th subcarrier for the integer m of 0 to 99, so that the n′th of the nth basic frame. (In the figure, since 4 slots from 253 to 256 are assigned, n ′ is an integer of 1 to 4) The terminal station apparatus is assigned to use the j-th group of subcarriers in the slot.

  In FIG. 65, each frame and slots 251-1 to 251-3 are allocated at the head of each basic frame. However, one or more slots may be included in the long-period superframe structure. If it is arranged at the head of a predetermined basic frame, it can be handled even if it is not placed at the head of all the basic frames. Similarly, slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, 256-1 to 256-6, and the like are described as the same number of slots in each basic frame. However, the same number is not necessarily required, and there may be a basic frame that does not include one in some cases.

  If the data communication slots 252-1 to 252-3 are premised on time-division TDD assuming that, for example, implicit feedback is performed, the bandwidth request from the terminal station device or the network side Bands in the data communication slots 252-1 to 252-3 are adaptively allocated according to data arriving at the station apparatus. At this time, the uplink and downlink time lengths may be fixedly set, or the allocation may be variable according to traffic conditions. In any case, the present embodiment can be applied. The data communication slots 252-1 to 252-3 are further subdivided to perform transmission / reception in fine slot units.

  Each base station apparatus and terminal station apparatus are assumed to know the start timing of the super frame and the basic frame by, for example, GPS (Global Positioning System) or some other means. Other than the GPS, the macro cell signal may be used, or the timing information may be notified by another wireless system. In addition, a terminal station device that newly enters the area needs to access the base station first by seeking attribution processing, but for that access, it performs downlink channel estimation, and from the estimation result Unless the obtained reception weight vector and uplink transmission weight vector are acquired, it is not possible to secure a line gain by using a large number of antennas.

  In a normal wireless system, a signal that can be used for timing detection is generally given to the head of a wireless packet. However, in this embodiment, signals in slots 256-1 to 256-6 are missing in subcarriers. Since the signal is an incomplete signal and is a broadcast signal that has not been subjected to directivity formation, it is generally not suitable for timing detection. Even if the base station apparatus transmits another timing detection signal, if the terminal station apparatus does not form a directional beam using the received weight vector, the base station apparatus will receive a line gain shortage. The signal for timing detection cannot be received. Therefore, it is preferable that some mechanism for assuming that the heads of the basic frame and the super frame are known is implemented in this embodiment, and in the following description, it is assumed that these timing information is known. This timing information may be grasped by utilizing other wireless systems such as GPS, or may be grasped by utilizing macro cell information (that is, synchronized with the periodicity of macro cell communication).

(Processing flow for simultaneous channel estimation by multiple radio stations)
Here, a processing operation for starting communication between a terminal station apparatus and a base station apparatus newly entering the area will be described. FIG. 66 is a flowchart showing a processing operation for starting communication between a terminal station apparatus and a base station apparatus newly entering the area. When the terminal station apparatus starts communication (for example, when the user turns on the power of the terminal station apparatus), the terminal station apparatus acquires the start timing of the basic frame and the super frame by a predetermined means (step S6601), and the frame head Training signal is received (step S6602). Then, the terminal station apparatus acquires a reception weight vector in the downlink and a transmission weight vector in the uplink from the received slots 251-1 to 251-3 (step S6603).

  Next, the terminal station apparatus selects one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-6, and any of the slots. Within the subcarrier group, a slot for conditions assigned to the system as a slot for random access and a slot for conditions for random access assigned to a combination of subcarriers assigned to the system as a subcarrier for random access. (Step S6604), and a training signal for channel estimation is transmitted there by random access (step S6605). For example, predetermined subcarriers in the slots 256-1 to 256-3 at the end of the basic frame (in the above example, divided into 10 groups, which may be all or some groups) may be random A slot for access may be used, or a slot for random access may be arranged only in a predetermined group of subcarriers in the last slot 256-1 in the first basic frame in the superframe.

  Each time the base station apparatus receives the random access slot (step S6621), the base station apparatus performs FFT processing, and acquires the reception level of a subcarrier that may contain a training signal as the random access slot (step S6622). ). If the subcarrier of the received signal is at a predetermined level (step S6623), the base station apparatus regards it as an initial access from the new terminal station apparatus, and performs uplink channel estimation from the training signal. (Step S6624). Subsequently, the base station apparatus calculates the uplink reception weight vector and acquires the downlink transmission weight vector by calibration (step S6625). Using the transmission weight vector acquired here, the base station apparatus transmits an uplink signal addressed to the terminal station apparatus to one of the subdivided slots among the data communication slots 252-1 to 252-3. Transmission of a slot assignment instruction signal for transmission is performed (step S6626). Here, since the base station device does not grasp the identity of the terminal station device, it is not possible to use the identification number of the terminal station device for the allocation instruction, so as an instruction for initial access to the terminal station device that has been randomly accessed, For example, the terminal station device specifies information such as the slot and subcarrier in which the training signal is transmitted by random access, and informs the terminal station device that it is an assignment to the own station.

  In response to this, the terminal station apparatus forms a directional beam with the acquired reception weight vector after the above-described training signal transmission by random access, and determines a timeout (step S6606). It waits (step S6607). If the slot assignment instruction signal transmitted by the base station apparatus can be received using the above-described reception weight vector (step S6608), control information containing the information of the own station is transmitted using the indicated slot. (Step S6609), an attribution request to the base station is made.

  The base station apparatus waits for signal reception from the terminal station apparatus while determining a timeout in the slot allocated by itself by using the reception weight vector acquired in the previous random access slot (step S6627). The base station apparatus performs signal reception with high quality by utilizing the line gain secured by forming the directional beam in both the base station apparatus (step S6628). The base station apparatus manages the attribution of the apparatus on its own database after the information exchange (step S6629).

  In addition, the base station apparatus includes any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-6, and any of the subs. The carrier group is allocated as a slot for channel feedback with the terminal station apparatus (step S6630), and the contents are similarly notified in the subdivided slots in the data communication slots 252-1 to 252-3. To do. Receiving this, the terminal station apparatus grasps a channel estimation slot for periodically transmitting a training signal to the base station apparatus in the uplink (step S6610). Thereafter, bandwidth allocation and bandwidth request are performed in the subdivided slots in the data communication slots 252-1 to 252-3, and communication is performed while the base station apparatus and the terminal station apparatus appropriately update the transmission / reception weight vector. Continue.

  Although a detailed explanation has been omitted in the above processing, in order to avoid a stagnation due to a code error and normal information exchange cannot be performed, a general timer is set up to determine whether or not there is a response by a timeout. It is also possible to apply a combination of management techniques. In addition, when the base station apparatus includes a plurality of first signal processing units as shown in the first embodiment, the amount of control information such as attribution requests is limited. There is no necessity to perform communication while performing spatial multiplexing using the signal processing unit, and a specific one first signal processing unit is selected and only the transmission weight vector directed to the first signal processing unit is used. The control information may be transmitted. Similarly, when transmitting control information from the base station apparatus, a configuration may be adopted in which one specific first signal processing unit is selected and control information is transmitted without performing spatial multiplexing.

  Next, with reference to FIG. 67, the transmission / reception signal processing operation of the base station apparatus in this embodiment will be described. FIG. 67 is a flowchart showing the transmission / reception signal processing operation of the base station apparatus. In FIG. 67, (a) shows reception processing, and (b) shows transmission processing. First, the reception process will be described. When the base station apparatus acquires the frame timing (step S6701), the base station apparatus recognizes the reception timing of the radio packet from the terminal station grasped due to the base station centralized control in the data communication slot following the slot at the head of the basic frame. (Step S6702). Then, the base station apparatus reads out the reception weight vector in the plurality of first signal processing units 304 of the own station for the terminal station apparatus from the memory (step S6703), and receives the reception signal vector of each first signal processing unit. The weight vector is multiplied (step S6704).

  Subsequently, the base station apparatus aggregates the signals of one system for each virtual transmission path, and based on the channel estimation result for the training signal received in the head region of each wireless packet (step S6705), linear A second reception weight matrix is calculated (step S6706), and signal detection processing is performed using this (step S6707). Then, the base station apparatus repeats the signal detection process in the same manner while the data spans a plurality of symbols (step S6708), and determines whether or not the frame ends after the data ends (step S6709). The above-described processing (signal reception from other terminal station devices) is repeated. At the end of the frame, the base station apparatus receives a slot for transmitting a training signal from the terminal station apparatus side at the end of the frame (step S6710), and calculates a transmission / reception weight vector related to the terminal station apparatus based on the received signal. This is recorded and managed, and the process ends (step S6711).

  Next, transmission processing will be described. When the base station apparatus acquires the frame timing (step S6721), the base station apparatus transmits a training signal for calculating a reception weight vector used by the terminal station apparatus in the first signal processing unit in the slot at the head of the basic frame (step S6722). . Then, when the base station apparatus recognizes the transmission timing of the wireless packet to the terminal station apparatus that is grasped due to the base station centralized control (step S6723), the base station apparatus stores a plurality of first stations of the own station for the terminal station apparatus from the memory. The transmission weight vector in the signal processing unit 304 is read (step S6724).

  Subsequently, after generating a transmission signal (step S6725), the base station apparatus multiplies the first transmission weight vector for each first signal processing unit (step S6726), and performs signal transmission (step S6727). If the data continues, the base station apparatus repeats the above processing (step S6728). If the data ends, the base station apparatus determines whether or not it is the end of the frame (step S6729). If not, the next transmission process or the reception process of FIG. 67A is repeated. If the frame ends, the process ends.

  Next, with reference to FIG. 68, the transmission / reception signal processing operation of the terminal station apparatus in the present embodiment will be described. FIG. 68 is a flowchart showing the transmission / reception signal processing operation of the terminal station apparatus. In FIG. 68, (a) shows reception processing and (b) shows transmission processing. First, the reception process will be described. When the terminal station apparatus acquires the frame timing (step S6801), the terminal station apparatus receives the training signal transmitted by the base station apparatus in the first slot of the basic frame (step S6802), and based on this, the first reception weight vector is virtually It is calculated for each general transmission path (for each of the plurality of first signal processing units) (step S6803). When the terminal station apparatus receives a radio packet from the base station apparatus (step S6804), the terminal station apparatus reads out the reception weight vector in the plurality of first signal processing units 304 of the terminal station apparatus from the memory, and stores each virtual signal in the reception signal vector. The reception weight vector corresponding to the transmission path is multiplied (step S6805).

  Next, the terminal station apparatus aggregates one system of signals for each virtual transmission path, and based on the channel estimation result (step S6806) for the training signal received in the head region of each wireless packet, A second reception weight matrix is calculated (step S6807), and signal detection processing is performed using this (step S6808). The terminal station apparatus repeats the signal detection process in the same manner while the data is over a plurality of symbols (step S6809). Then, after the end of the data, the terminal station apparatus determines whether or not the frame ends (step S6810). If the frame continues, if there is an assignment addressed to itself, the above reception process is repeated. At the end of the frame, the process ends.

  Next, transmission processing will be described. When the terminal station apparatus acquires the frame timing (step S6821), the terminal station apparatus performs a calibration process on the reception weight vector obtained by the above-described reception process, and calculates a first transmission weight vector (step S6822). Then, when the terminal station apparatus recognizes the transmission timing of its own station (step S6823), after generating the transmission signal (step S6824), the terminal station apparatus multiplies the signal for each virtual transmission path by the first transmission weight vector. (Step S6825), the result of multiplication is added and synthesized for each antenna element to perform signal transmission (step S6826).

  Next, when the data continues, the terminal station apparatus repeats the above processing (step S6827). When the data ends, the terminal station apparatus determines whether or not the frame ends (step S6828). If not completed, the next transmission process is repeated if there is an assignment addressed to the own station. If it is the end of the frame, the terminal station device transmits the training signal in the slot and subcarrier allocated to the own station for the training signal at the end of the frame (step S6829), and ends the process.

  Although details are omitted here, the above processing is basically performed for each subcarrier on the frequency axis. The above processing is the description in the case of the configuration shown in FIG. 69 as a precondition. FIG. 69 is a diagram showing a frame configuration as a precondition. In this figure, reference numerals 257-1 to 257-3 represent training signals, and 258-1 to 258-3 represent data payloads. In the description of FIGS. 67 and 68 described above, first, channel estimation for signal separation of MIMO transmission is performed on a signal sequence for each of a plurality of virtual transmission paths aggregated by multiplying by the first reception weight vector. As described above, the training signals 257-1 to 257-3 are used for this. As for the training signals 257-1 to 257-3 and the data payloads 258-1 to 258-3, it is possible to adaptively use the uplink and the downlink in a time-sharing manner so that the uplink or the downlink is not clearly indicated. Is possible. For example, the training signals 257-1 to 257-2 and the data payloads 258-1 to 258-2 are downlink signals (transmitted by the base station), and the training signal 257-3 and the data payload 258-3 are uplink signals. (The terminal station transmits), all may be downlink, and all may be uplink. In any case, the training signals 257-1 to 257-3 are signals with a sufficient line level multiplied by the first transmission weight vector regardless of the uplink or downlink, so that they are high at a sufficient reception level. Channel estimation is possible with accuracy.

  On the other hand, since the training signals 251-1 to 251-3 are all signals that are not subjected to directivity formation by the first transmission weight vector, the transmission power is concentrated on a small number of subcarriers. The estimation accuracy is relatively low. However, the second reception weight matrix can be calculated using the training signals 251-1 to 251-3 if the channel estimation accuracy with no operational problems can be secured. In this case, as shown in FIG. 70 instead of FIG. 69, the training signals 257-1 to 257-3 are omitted, and communication is performed using only the data payloads 258-1 to 258-3. FIG. 70 is a diagram illustrating an example in which the training signal is omitted and communication is performed using only the data payload.

  Note that the training signals 257-1 to 257-3 shown in FIG. 69 are for performing channel estimation for signal separation in MIMO transmission. For example, if the number of spatial multiplexing is large, the training signals that are orthogonal to that amount are necessary. There is a corresponding number of symbol overheads. However, in FIG. 70, when the channel estimation for signal separation of MIMO transmission is performed by making the best use of the training signals 251-1 to 251-3 at the head of the frame, the spatially multiplexed signal series is neatly separated. It is only necessary to perform channel estimation necessary for signal detection / demodulation processing of the independent signal sequence, and all the training signals are transmitted at the same time (for example, one symbol in the case of OFDM). Received signal processing can also be performed as if it were a SISO (Single Input Single Output) signal. In this sense, FIG. 70 may include a training signal for signal detection / demodulation of a SISO signal.

  Next, with reference to FIG. 71, another processing operation for calculating a second reception weight matrix in the present embodiment will be described. FIG. 71 is a flowchart showing another second reception weight matrix calculation processing operation. This processing operation is applicable to both processing of the base station apparatus in the uplink and processing of the terminal station apparatus in the downlink. Further, if it is uplink, it is performed on the training signal transmitted from the terminal station apparatus at the end of the frame, and if it is downlink, it is performed on the training signal transmitted from the base station apparatus at the head of the frame. Or you may implement with respect to the training signals 257-1 to 251-3 shown in FIG. Here, it demonstrates as operation | movement of a base station apparatus.

  First, when receiving a training signal (step S7101), the base station apparatus calculates a first reception weight vector for each virtual transmission path (step S7102). The first reception weight vector is one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-6, and its slot. A reception weight vector calculated using any one of the subcarrier groups may be substituted. Thereafter, the base station device multiplies the training signal used to calculate the first reception weight vector by the first reception weight vector for each virtual transmission path (step S7103). By this multiplication, signal sequences of the number of virtual transmission paths are acquired. In receiving the training signal for each virtual transmission line, since subcarriers are segregated for each virtual transmission line, the crosstalk component between each virtual transmission line can also be individually extracted. The extracted channel estimation result corresponds to each component of the channel matrix of Equation (55). Based on the channel matrix obtained in this way, a second reception weight matrix is calculated (step S7104).

For example, a case where the number of virtual transmission lines is 3 will be described as an example. As an example, the first virtual transmission path has multiples of 3 subcarriers, the second virtual transmission path has multiples of 3 + 1 subcarriers, and the third virtual transmission path has multiples of 3 + 2 subcarriers. Assume that a training signal is transmitted using a carrier. At this time, attention is focused on a signal obtained by multiplying each subcarrier component of the reception signal by the reception weight of the first virtual transmission path. Focusing on 3 multiple subcarriers, since the first receiving weight vector may perform channel estimation of the desired signal after the multiplication, h ₁₁ components of the channel matrix of 3 × 3 can be obtained.

On the other hand, when attention is paid to a subcarrier of a multiple of 3 + 1, a signal component that the signal of the second virtual transmission path leaks into the first virtual transmission path can be extracted, so h in the 3 × 3 channel matrix ₁₂ components can be obtained. Similarly, when attention is paid to a subcarrier of a multiple of 3 + 2, the signal component that the signal of the third virtual transmission path leaks into the first virtual transmission path can be extracted, so h ₁₃ of the 3 × 3 channel matrix Ingredients can be obtained.

Similarly, attention is paid to a signal obtained by multiplying the reception weight of the second virtual transmission path. Focusing on subcarriers 3 multiples +1, since the second receiving weight vector may perform channel estimation of the desired signal after the multiplication, h ₂₂ components of the channel matrix of 3 × 3 can be obtained. On the other hand, when attention is paid to subcarriers that are multiples of 3, the signal component that the signal of the first virtual transmission path leaks into the second virtual transmission path can be extracted, so h ₂₁ of the 3 × 3 channel matrix Ingredients can be obtained. Similarly, when attention is paid to a subcarrier of a multiple of 3 + 2, the signal component that the signal of the third virtual transmission line leaks into the second virtual transmission line can be extracted, so h ₂₃ of the 3 × 3 channel matrix Ingredients can be obtained.

Similarly, attention is paid to a signal obtained by multiplying the reception weight of the third virtual transmission line. Focusing on multiples of 3 + 2 subcarrier, because the third receiving weight vector may perform channel estimation of the desired signal after the multiplication, h ₃₃ components of the channel matrix of 3 × 3 can be obtained. On the other hand, when attention is paid to subcarriers that are multiples of 3, the signal component that the signal of the first virtual transmission path leaks into the third virtual transmission path can be extracted, so h ₃₁ of the 3 × 3 channel matrix. Ingredients can be obtained. Similarly, when attention is paid to a subcarrier of a multiple of 3 + 1, a signal component leaking into the third virtual transmission path can be extracted from the signal of the second virtual transmission path, so h ₃₂ of the 3 × 3 channel matrix. Ingredients can be obtained.

  In this way, all 3 × 3 matrix elements can be acquired. Although the case where the number of virtual transmission paths is 3 has been described as an example here, a channel matrix can be obtained for an arbitrary number of virtual transmission paths. If these channel matrices can be obtained, the processing thereafter is the same as the spatial multiplexing signal processing using a normal MIMO channel. Therefore, in addition to the use of ZF type linear weights and MMSE type linear weights, MLD and QR -Thousand number detection can be performed by arbitrary signal processing including nonlinear signal processing such as MLD. In the description of FIG. 67 and FIG. 68 described above, an example in which a linear reception weight matrix is used has been described, but it is naturally possible to apply signal detection processing of an arbitrary MIMO channel as described above.

  In addition, although the example which aims at orthogonalization by dividing the subcarrier used for a training signal for every virtual transmission line was shown here, if it can be given for every signal series which spatially multiplexes other orthogonalized training signals, It is also possible to perform similar signal processing using the training signals 257-1 to 257-3 shown in FIG. 69 using the orthogonality. For example, in the case of OFDM, if 3 OFDM symbols are used and training signals for each signal series are used while being shifted in time, it is naturally possible to perform channel estimation using all subcarriers.

(Utilization of time axis beam forming)
As described in the section on time axis beam forming described above, generally, when an antenna element exists in a very narrow area, the frequency dependence of the reception weight (that is, each component of the reception weight vector) for each antenna element. Is expected to be relatively small. In order to increase the degree of signal separation between the virtual transmission lines, it is possible to add and synthesize the complex phases of the received signals of the respective antenna elements with the respective subcarriers by using the optimum receiving weight as much as possible. preferable. However, if the crosstalk component between the virtual transmission lines can be separated with high accuracy using the second-stage reception weight matrix in the subsequent stage, even if the accuracy of the first reception weight vector is low to a certain degree, Don't be.

  For example, consider the addition of Sin (θ) and Sin (θ + δ). From the trigonometric formula, the following relational expression (76) is obtained.

  This equation means that when the in-phase synthesis is incomplete and the phase error δ exists, the amplitude at the time of addition is not doubled but doubled by 2 × Cos (δ / 2) due to in-phase synthesis. To do. Taking the case where δ is 30 degrees as an example, the amplitude is Cos (30/2) = 0.9659..., resulting in a gain loss of about 3.4%. That is, if the error is within 30 degrees for all antennas, the gain loss due to this error is only about 0.3 dB. Even if δ is 60 degrees, it is only about 1.2 dB with a gain loss of about 13.4%.

  Therefore, even if F (α) in FIG. 44 is not in a region where it exceeds 30 dB, if it is assumed that signal separation processing on the frequency axis is performed in the subsequent stage, the first signal processing unit 304 performs the processing. This means that there is no problem as long as the time-dependent beam forming can be performed with low accuracy and a limited decrease in gain can be tolerated. That is, after obtaining the reception weight vector on the frequency axis with the training signals 251-1 to 251-3 of FIG. 65, IFFT processing is performed on the reception weight for each frequency for each element of the reception weight vector, so that Convert to weight. Then, a reception weight vector on the time axis is constructed using only the value of this leading component (corresponding to the line-of-sight component). For each sampling value, multiplying the received signal vector of the sampling value composed of the sampling value of each antenna element by the reception weight vector on the time axis virtually transmits the sampling signal with the line gain increased by directivity control It can be acquired for each road. In this way, it is possible to acquire a received signal in a state where timing can be detected.

  If the speed of channel time fluctuation is moderate to some extent with respect to the basic frame period, the relative channel information extracted from a plurality of signals (that is, when receiving training signals 251-1 to 251-3) (that is, If the channel information using the relative complex phase based on the complex phase of the reference antenna is averaged, the noise component can be suppressed and the channel estimation accuracy can be improved. If averaging is not performed, absolute channel information can be used as it is, not relative channel information.

  Next, with reference to FIG. 72, another signal processing operation when time-axis beamforming is applied to the present embodiment will be described. FIG. 72 is a flowchart showing another signal processing operation of time-axis beam forming. In the description of the fifth embodiment, in consideration of the fact that the SNR between one antenna element on the transmission side and one antenna element on the reception side is significantly insufficient in line design, the equation (52 ) And the like have been described. With the techniques shown in the sixth embodiment, the seventh embodiment, and the eighth embodiment, one antenna element on the transmission side is compared with one on the reception side. Channel information between the antenna elements in the time axis can be obtained with high accuracy on the frequency axis, and channel information on the time axis used for time axis beam forming by IFFT processing using the channel information on the frequency axis. It becomes possible to get. Here, the processing of the terminal station apparatus in the downlink is shown as an example, but the same processing is possible in the uplink. First, the base station apparatus transmits a training signal for channel estimation in the frame head region (step S7201). The terminal station apparatus receives this training signal and performs received signal processing (step S7211). As described above, this signal processing assumes a training signal that is slightly longer than the OFDM symbol period that does not include a guard interval, and a sampling signal cut out in an appropriate FFT window even with rough frame timing as a reference. The FFT processing is performed using

  Next, the terminal station apparatus performs channel estimation with reference to a known training signal for the signal subjected to the FFT processing, and performs channel information for each virtual transmission path (or for each virtual transmission path for the reference antenna). Get relative channel information). The channel information obtained here is in a state where information exists only in specific subcarriers for each virtual transmission path. Similarly, a channel estimation result is obtained (step S7212). Further, the reception weight vector on the frequency axis is calculated based on the acquired channel information (step S7213).

  Next, the terminal station apparatus performs IFFT processing on the subcarriers of the weight information of each antenna element (step S7214). Then, the terminal station apparatus extracts the value of the leading component (corresponding to the line-of-sight component) from the weight information on the time axis obtained as a result, calculates the reception weight on the time axis of each antenna element, A reception weight vector having these as vector components is acquired for each virtual transmission path (step S7215). Calibration processing is performed on the reception weight vector on the time axis (step S7216). Subsequently, the terminal station apparatus calculates transmission weight vectors, and records and manages these transmission / reception weight vectors on the first-stage time axis (step S7217).

  The above description is the first-stage time axis weight calculation operation, but the subsequent frequency axis weight calculation is performed as follows. At the stage of calculating the reception weight vector on the time axis (step S7215), the terminal station apparatus uses the sampling value for each antenna element as an element for each sampling value of the above-described reception training signal used for this reception weight calculation. The received signal vector (sampling value) to be multiplied by the reception weight vector for each virtual transmission path is separated for each signal series of the virtual transmission path (step S7221). Subsequently, the terminal station apparatus individually performs FFT processing on the obtained signal series of sampling values for the number of virtual transmission paths, and performs channel estimation with reference to a known training signal (step S7222). ). As for the channel estimation result here, only the subcarrier to which the training signal is assigned for each virtual transmission path is the channel estimation result of the desired virtual transmission path, and the subcarrier without the training signal is from a different virtual transmission path. Represents the crosstalk component of.

  Although the case where the value of the leading component (corresponding to the line-of-sight component) is extracted and used from the weight information on the time axis has been introduced here, the time axis of the delay component of one sample shown in Expression (57) Can be calculated based on the second component subjected to IFFT, and the reception weight on the time axis of the delay component of the two samples shown in the equation (59) can be calculated based on the third component subjected to IFFT. Similarly, it is possible to use more delayed wave components.

Next, if the terminal information that the signal transmitted to the i-th virtual transmission path in the k-th subcarrier leaks into the j-th virtual transmission path is h _ji ^(k) , these are specified. Since the information exists only in the subcarriers, interpolation processing of the missing subcarriers by linear interpolation etc. is performed, and a complete channel matrix including crosstalk components from different virtual transmission paths is obtained. (Step S7223). Then, the terminal station apparatus calculates a reception weight matrix for signal separation based on the reception weight matrix obtained in this way (step S7224), and this is used as a reception weight matrix on the second-stage frequency axis. Is recorded and managed (step S7225).

  The first-stage time axis transmission / reception weight vector and the second-stage frequency axis reception weight matrix acquired as described above can be continuously used in subsequent frames.

  As described above, in the transmission / reception weight acquisition for using the virtual transmission line corresponding to the first singular value in the line-of-sight environment, channel estimation is performed with a small number of subcarriers, the antenna element interval is narrow, and the line-of-sight wave is not generated. If it is dominant, it is possible to estimate the transmission / reception weights of the remaining subcarriers by linear interpolation or the like using the characteristic that the frequency dependence of the channel information becomes small. For this reason, it becomes possible to perform channel estimation simultaneously in a form in which subcarriers do not overlap in a plurality of radio stations. In order to perform this efficiently, the training signal for channel estimation by the base station apparatus and the individual terminal station apparatus in the frame is set as a fixed slot in the frame structure having periodicity, and the same subcarrier is used. In order to avoid interfering with each other, slot allocation is performed by dividing timing and subcarrier, and channel feedback is periodically performed in the slot.

  In general, a training signal for performing feedback of channel information in the prior art assigns an identification number of a transmitting station to a payload portion subsequent to the training signal. In addition, since the frequency dependence of the channel information is generally large, the channel information for each antenna is designed to be estimated with all subcarriers. These overheads are very large and cause a significant loss of MAC efficiency. However, in the eighth embodiment, since this training signal is assigned to a known fixed location (slot and subcarrier) for each base station device or terminal station device, the identification number of the transmitting station, etc. is included in the subsequent payload. If it is not necessary to assign sub-carriers, it is possible to simultaneously perform multiple channel estimation and channel feedback while segregating subcarriers in one slot.

[Ninth Embodiment]
In the above description of the eighth embodiment, the description has focused on the case where channel information or transmission / reception weights have frequency dependence. However, as shown in the fifth embodiment, it is possible to extend even when the channel information or the transmission / reception weight has no frequency dependency.

In the above-described fifth embodiment, channel information is acquired from the correlation value evaluation on the time axis of the received training signal by using, for example, Equation (52). The expression (52) means that if the channel information is not frequency-dependent, S _j (n) S ₁ (n) ^* at each sampling time in Σ of the expression (52) excludes noise components. Therefore, the noise components are averaged with N _FFT sampling values, and as a result, relative channel information (correlation value for each antenna) can be obtained with high channel estimation accuracy.

  However, since signals from different antenna elements on the frequency axis are mixed in the eighth embodiment, if a correlation is made between received signals in which signals from different terminal station apparatuses or different antenna elements are mixed, a specific transmission antenna The correlation value in the state where signals from different terminal station devices and different antenna elements are mixed is acquired instead of the correlation of each signal from the receiving antenna, and the specific terminal station device and the specific The correlation value for the antenna element cannot be extracted. In a state where signals from different terminal station devices and different antenna elements are mixed, it is possible to easily separate the signals by performing FFT. In this case, the obtained channel estimation result of each subcarrier is Therefore, it is required to have sufficiently high channel estimation accuracy (that is, sufficient SNR characteristics).

  In the sixth embodiment, the seventh embodiment, and the eighth embodiment described above, the number of subcarriers used is limited to 4 as an example to improve the channel estimation accuracy. However, these channel estimation results are If there is no frequency dependence, it is possible to further average these channel estimation results and acquire channel information common to all subcarriers by the average value.

Here, c _j (f _k ) represents channel information on the frequency axis in subcarrier f _k . Alternatively, relative channel information based on the complex phase of the reference antenna may be used.

Also, here, taking the case of using 4 subcarriers as an example, a coefficient of ¼ is specified for the purpose of averaging the four used subcarriers f _k, but if the number of other subcarriers, Averaging is performed according to the number. The channel information obtained as a result is equivalent to that obtained by Equation (52), and the reception weight is calculated using Equation (50), Equation (51), and the like.

  Although the same applies to the seventh embodiment in addition to the eighth embodiment, the case where a training signal including four subcarriers is used as a training signal for acquiring channel information has been described as an example. However, the number of subcarriers is not four, and any other number of subcarriers may be used. As the ultimate condition, it is possible to set the number of subcarriers used to one.

  In the above fifth embodiment, the antenna element interval of each first signal processing unit on the base station apparatus side is narrowed to create a situation where the frequency dependence of the channel information or transmission / reception weight is very small, and as a result, all Channel information or transmission / reception weight common to subcarriers is set. Although it is possible to expect suppression of measurement error by using a plurality of subcarriers, there are some advantages to using a single subcarrier as the ultimate condition.

  As a training signal using only this single subcarrier, if the purpose is to acquire relative channel information for each antenna element (that is, if there is no purpose such as timing detection), then it is assumed that OFDM is used. However, it is not necessary to provide a guard interval, and continuous transmission of a simple sine wave (that is, an unmodulated signal) of the frequency of the assigned subcarrier is used. The feature of this signal is that the ratio PAPR (Peak to Average Power Ratio) of the temporal variation between the average transmission power and the instantaneous transmission power is 1 because the signal has a completely constant amplitude. .

  In general, when OFDM is used, even when a modulation method with constant amplitude such as QPSK is used, the amplitude of the signal varies with time in order to synthesize independent signals for each subcarrier. For this reason, the PAPR has a relatively large value, and the peak transmission power is much larger than the average transmission power. Since the linearity of the high power amplifier is limited, when the power at the peak exceeds the linear region of the high power amplifier, nonlinear distortion occurs and communication characteristics deteriorate. Therefore, in OFDM, a so-called back-off margin of transmission power is expected, and signal transmission is performed with the transmission power lowered by the margin.

  From the viewpoint of circuit design, for example, in a system using 1024-point FFT (using about 1000 as an effective subcarrier), it is necessary to expect a backoff of at least about 10 dB. However, since a complete sine wave having a constant amplitude does not require such back-off, it is possible to increase the effective transmission power by the back-off value for transmission. Increasing the transmission power for this back-off value can be used even when a single subcarrier is used. For example, even when using 4 subcarriers, the initial phase of each subcarrier is adjusted to an arbitrary value. By configuring the training signal so that the PAPR in one symbol period is minimized, it is possible to operate with a lower back-off value according to the PAPR value at that time.

  When transmitting a training signal, it is desirable to set the back-off value at the time of training signal transmission to (x / 2) dB or less, where xdB is the minimum back-off value used in other data communication. Specifically, it is desirable to set the back-off value at the time of training signal transmission to 0 dB.

As a result, for example, if transmission power for 1000 subcarriers is focused on one subcarrier, a line gain of 30 dB (= ₁₀ Log ₁₀ 1000) can be obtained there, and a line of about 10 dB for backoff. Channel estimation in a good state with gain added is possible. That is, an improvement of about 40 dB can be expected. Originally, in the high frequency band such as the millimeter wave band, the line design was strict due to the high frequency. For example, if the frequency is increased 10 times, the increase in loss due to the term depending on the frequency of the free space propagation loss is 20 dB. Also, the output of the high power amplifier is relatively lowered.

  Furthermore, in Massive MIMO, since it is necessary to mount 100 or more RF devices, the price of each device is expected to be reduced, and the linearity of the amplifier is also lowered by using inexpensive component materials. As a result, there is a problem that the channel estimation accuracy is lowered unless an additional line gain of 30 dB or more is secured. However, a channel in which transmission power is increased by focusing on 1 subcarrier or 4 subcarrier and suppressing PAPR value. Estimating can solve these problems.

  Note that if channel estimation is performed using only one subcarrier per antenna element, the channel estimation result for each subcarrier is actually channel information based only on line-of-sight components, due to the influence of various reflected wave components. May show different values. However, as shown in the seventh embodiment, a plurality of antenna elements are provided on the wireless station side that transmits the training signal, and unmodulated continuous signals of different subcarriers are transmitted from the plurality of antenna elements. Then, the radio station that receives the training signal receives the training signals of a plurality of subcarriers. If the relative channel information is obtained from the obtained channel information, the influence of the different antenna elements on the transmitting side is relatively small, and the relative channel information obtained by the plurality of antenna elements is approximately constant on the frequency axis. Should be considered. If averaging is performed using this feature, the reflected channel components that are affected differently for each subcarrier are relatively suppressed, and the relative channel information on the time axis in which the line-of-sight component is effectively extracted. And the transmission / reception weight of the time axis can be calculated based on this.

  The averaging process here is basically intended to average the complex phase, but it is expected that the amplitude is generally constant and the fluctuation range of the complex phase is expected to be small. The channel information itself given as a complex number may be simply averaged on the complex plane.

[Tenth embodiment]
[Multiuser MIMO transmission using a virtual transmission line corresponding to the first singular value]
(Outline of the basic principle according to the tenth embodiment)
In the above description, as shown in FIG. 16, for example, each first signal processing unit 304 transmits and receives a single signal sequence between a single terminal station apparatus 302 and a base station apparatus 303, and a plurality of first signal processing units 304 As a whole, spatial multiplex transmission is performed using the signal processing unit 304 of FIG. This corresponds to communication between the base station apparatus 303 and the terminal station apparatus 302 being performed by single user MIMO transmission. However, as a matter of course, the present invention can also be applied to a case where spatial multiplexing transmission (that is, multiuser MIMO) is performed between a plurality of terminal station apparatuses 302 and base station apparatuses 303.

  Normally, in multi-user MIMO, when signals are simultaneously transmitted to a plurality of terminal station apparatuses on the same frequency, nulls are formed in the downlink so that mutual signals do not cause mutual interference between the terminal station apparatuses. The signal was transmitted while performing. However, the null formation may be broken if there is a time variation of the channel, and residual interference may occur. This is because the transmission weight used for transmission directivity control is calculated based on the result of channel feedback performed at a predetermined period, but in order to reduce the overhead associated with this channel feedback, channel feedback is performed so frequently. As a result, the transmission weight must be calculated based on the channel information acquired in the past. Therefore, no matter how highly accurate the transmission weight is calculated, there is no meaning if there is a change in the channel over time.

  When the virtual transmission line corresponding to the first singular value shown in the first embodiment is positively used, the value of the line gain obtained on the transmission line is different on the other transmission line using the reflected wave. It is relatively higher than the gain. In addition, when a large number of antenna elements are used on both the base station apparatus side and the terminal apparatus side, it is considered that an additional line gain corresponding to the product of the number of antenna elements can be obtained. Then, it is possible to obtain a desired SIR characteristic without deliberately forming a complete null control.

  FIG. 73 is a diagram illustrating configurations of a base station device and a terminal station device when multi-user MIMO is applied in the tenth embodiment. 16 differs from FIG. 16 in that the terminal station apparatus is only the terminal station apparatus 302 in FIG. 16, but the terminal station apparatus is added to the terminal station apparatus 302 in addition to the terminal station apparatus 302 in FIG. In each of the first signal processing units 304-1 to 304-4 of the base station device 303, virtual transmission paths corresponding to the first singular values are formed toward the terminal station devices 302, 308, and 309, respectively. Thus, multi-user MIMO transmission is performed.

  Conventionally, since a plurality of first signal processing units 304-1 to 304-4 were not implemented, when a plurality of signal sequences are simultaneously spatially multiplexed for a single terminal station device, One transmission / reception signal processing circuit performs multi-user MIMO signal processing using different transmission / reception weight vectors. Here, it is necessary to perform complete null control on the transmission side so that mutual interference does not occur between the terminal station apparatuses.

On the other hand, in the embodiment of the present invention, unlike the conventional multi-user MIMO, each of the first signal processing units 304-1 to 304-4 uses one transmission / reception weight vector for one terminal station apparatus. Only minute signal processing is performed. For example, one first signal processing unit 304-1 transmits signals for three systems in total, one for each of the terminal station devices 302, 308, and 309. At this time, the transmission weight vector for the virtual transmission path corresponding to the first singular value for each of the terminal stations 302,308,309 and _{v 11, v 12, v 13,} respectively. Since these are transmission weight vectors for the individual single user MIMO described above, they are not orthogonal to each other. In contrast, the transmission weight vectors v ′ ₁₁ , v ′ ₁₂ , and v ′ _{13 that} are actually used for multi-user MIMO are such that the vector v ′ ₁₁ is orthogonal to v ₁₂ and v ₁₃ , and the vector v ′ ₁₂ is v _11. and _v so as to orthogonal to _13, the vector v _'13 to set so that orthogonal to _{v 11} and _{v 12.} Specifically, the transmission weight vector v ′ ₁₁ for the terminal station device 302 is expressed by the following equation (78) using Gramschmitt orthogonalization.

This may be similarly processed for the terminal station devices 308 and 309 to obtain v ′ ₁₂ and v ′ ₁₃ . The above is the description for the first signal processing unit 304-1, but the same processing may be performed for the first signal processing units 304-2 to 304-3. In this way, the transmission weight vectors v ′ _{11 to} v ′ ₁₃ for the first signal processing unit 304-1, the transmission weight vectors v ′ _{21 to} v ′ ₂₃ for the first signal processing unit 304-2, the first The transmission weight vectors v ′ _{31 to} v ′ ₃₃ for the signal processing unit 304-3 and the transmission weight vectors v ′ _{41 to} v ′ ₄₃ for the first signal processing unit 304-4 may be obtained. Through the above processing, each of the first signal processing units 304-1 to 304-4 transmits three signal sequences, and a total of twelve systems of spatial multiplexing transmission can be performed. As a supplement here, unlike the case of the conventional multi-user MIMO, for example, transmission weight vectors v ′ ₁₁ and v ′ 2 _j are not orthogonal to j where j ≠ 1, resulting in complete null formation. Not. However, since these are different combinations of the first signal processing unit 304 of the transmission source and the terminal station apparatus to receive, the mutual interference is relatively small because of the virtual transmission path having a small correlation. As described above, this is based on the fact that the absolute value of the first singular value protrudes and shows a high gain, and the high directivity gain of the pinpoint that comes from the product of many antenna elements of the base station device and the terminal station device. To do. For this reason, it is not necessary to guarantee the orthogonality between the transmission weight vectors v ′ ₁₁ and v ′ _2j .

Similar processing may be performed similarly for processing on the base station apparatus side in the uplink. That is, although the above equation (78) shows the orthogonalization process for the transmission weight vector, the same orthogonal relationship is applied to the reception weight, and the first singularity for each terminal station apparatus 302, 308, 309 is applied. The received weight vectors for the virtual transmission line corresponding to the value are read as v ₁₁ , v ₁₂ , and v ₁₃ , respectively, and the received weight vectors v ′ ₁₁ , v ′ ₁₂ , and v ′ ₁₃ used for multiuser MIMO are converted into the vector v _'11 so as to orthogonal to the _{v 12} and _{v 13,} the vector v' as the ₁₂ orthogonal to the _{v 11} and _{v 13,} it Rure be set so that orthogonal to the vector v _{'13 v 11} and _{v 12} . In this case, equation (78) can be applied as it is to calculate the reception weight. Further, the wireless station apparatus side may be exactly the same processing and circuit configuration as the wireless station apparatus in the first embodiment, without being conscious of multi-user MIMO.

  In this case, the circuit configuration of the base station apparatus 303 needs to be changed as compared with FIG. FIG. 74 shows the first transmission corresponding to the first transmission signal processing units 181-1 to 181-4 of the base station apparatus 70 corresponding to the base station apparatus 303 when multiuser MIMO is applied in the embodiment of the present invention. 3 is a diagram illustrating a circuit configuration of a signal processing unit 182. FIG. In FIG. 74, as multi-user MIMO transmission, as shown in FIG. 73, the number of terminal station apparatuses simultaneously spatially multiplexed is three. The configuration in FIG. 74 is similar to the configuration in FIG. 6, but the first transmission signal processing circuits 113-1 to 113-3 correspond to three terminal station apparatuses that perform spatial multiplexing at the same time. The transmission weight processing unit 150 generates and manages a transmission weight vector for multi-user MIMO used for transmission on a virtual transmission line corresponding to the first singular value for three terminal station apparatuses that simultaneously perform spatial multiplexing. Specifically, in the transmission weight processing unit 150, the channel information acquisition circuit 151 separately acquires the channel information acquired by the reception unit via the communication control circuit 120, and sequentially updates the channel information storage circuit Store in 152. At the time of signal transmission, in accordance with an instruction from the communication control circuit 120, the multiuser MIMO (MU-MIMO) transmission weight calculation circuit 153 reads channel information corresponding to the destination station from the channel information storage circuit 152, and based on the read channel information. The transmission weight vector is calculated as shown in the above equation (78). The point to be noted here is that in the case of single user MIMO, the first right singular vector is calculated and used as it is as a transmission weight vector. In order to orthogonalize transmission weight vectors between the terminal station apparatuses, for example, from the first right singular vector for the terminal station apparatus 302, the first right singular vector component for the terminal station apparatus 308, and the first for the terminal station apparatus 309. The components of the right singular vector are subtracted and signal transmission is performed in a form orthogonal to each other. The multiuser MIMO transmission weight calculation circuit 153 outputs the transmission weight calculated in this way to the first transmission signal processing circuits 113-1 to 113-3.

Next, in FIG. 75, in the tenth embodiment, the first reception signal processing units 185-1 to 185-4 of the base station device 70 corresponding to the base station device 303 when multiuser MIMO is applied. The circuit configuration of the received signal processing unit 186 of FIG. Also in FIG. 75, as multi-user MIMO transmission, as shown in FIG. 73, the number of terminal station apparatuses simultaneously spatially multiplexed is three. The configuration in FIG. 73 is similar to the configuration in FIG. 7, but the first reception signal processing circuits 114-1 to 114-3 correspond to the three-station terminal stations that simultaneously perform spatial multiplexing. The reception weight processing unit 170 generates and manages reception weight vectors for multiuser MIMO used for reception on the virtual transmission path corresponding to the first singular value for the three terminal station apparatuses that simultaneously perform spatial multiplexing. Specifically, in the reception weight processing unit 170, a known signal for channel estimation separated into subcarriers based on information input from the FFT circuit 857 in the channel information estimation circuit 172 in accordance with an instruction from the communication control circuit 120. From the (preamble signal or the like given to the head of the radio packet), the antenna elements of the terminal station devices 60 corresponding to the terminal station devices 302, 308, and 309 and the antennas of the base station device 70 corresponding to the base station device 303 The channel information with the element 851 is estimated for each subcarrier, and the estimation result is output to the multiuser MIMO reception weight calculation circuit 173. Multiuser MIMO reception weight calculation circuit 173 calculates a reception weight vector to be multiplied for each subcarrier based on the input channel information. Here, although the expression (78) has described the processing using the transmission weight vector v in the case of single user MIMO, if this is similarly replaced with the reception weight vector u in the case of single user MIMO, the expression (78) will remain as it is. Is applicable. At this time, the reception weight vector for combining the signals received by the antenna elements 851-1 to 851-N _MT-Ant is different for each terminal station apparatus 60, and the first reception corresponding to the terminal station apparatus 60 to be extracted is performed. The signal processing circuits 114-1 to 114-3 are individually calculated. In the second received signal processing unit 75 at the latter stage of this signal processing, the crosstalk component between the signal sequences from the same radio station apparatus is removed. It is expected that the mutual interference component between them is sufficiently reduced, so there is no need to be conscious of multiuser MIMO when calculating the reception weight matrix here, and MIMO signal detection processing is individually performed for each radio station apparatus Just do it. However, since different MIMO station detection processes are required for different radio station apparatuses, it is necessary to provide the second received signal processing units 75 in parallel for the radio station apparatuses that perform spatial multiplexing simultaneously. .

  In this way, information between the first transmission signal processing unit 181 and the second transmission signal processing unit 71 and information between the first reception signal processing unit 185 and the second reception signal processing unit 75. However, there is a difference that is expanded to a plurality of systems, but spatial multiplexing transmission is performed with a single terminal station apparatus using a plurality of first transmission signal processing units 181 and first reception signal processing units 185 Thus, the configuration is different from that of the prior art.

  As described above, the wireless communication system 90 according to the tenth embodiment includes the base station device 303 (first wireless station device) and a plurality of terminal station devices (second wireless station device) such as the terminal station device 308. With. The base station apparatus 303 according to the tenth embodiment performs a plurality of first signal processing units 304 having a first antenna element group and radio communication signal processing associated with the plurality of first signal processing units. And a second signal processing unit 305 (including strictly other base station apparatus functions such as an interface circuit, a MAC layer processing circuit, and a communication control circuit) to be executed. The terminal station device 308 of the tenth embodiment includes a second antenna element group and a transmission / reception unit that performs wireless communication with the first signal processing unit via the second antenna element group. The same applies to the terminal station apparatus 302 and the terminal station apparatus 309.

  The first signal processing unit 304 of the tenth embodiment uses the transmission weight vector of the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group as the first channel of the MIMO channel matrix. Calculation is based on at least one of the first right singular vector and the first left singular vector corresponding to one singular value, or at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. . The first signal processing unit 304 generates a transmission signal for each terminal station apparatus, and generates a signal vector by multiplying the generated transmission signal by a transmission weight vector. The first signal processing unit 304 adds and synthesizes signal vectors for terminal station apparatuses that are spatially multiplexed at the same time. The first signal processing unit 304 transmits a signal based on the added and combined signal vector from the first antenna element group to the plurality of terminal station devices using the same frequency channel at the same time.

  The first signal processing unit 304 of the tenth embodiment uses the received weight vector of the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group as the MIMO channel matrix. Based on at least one of the first right singular vector and the first left singular vector corresponding to one singular value, or based on at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. For each wireless station device. The first signal processing unit 304 multiplies the signal vector based on the signal received via the first antenna element group by the reception weight vector for each terminal station apparatus, thereby providing one system signal for each terminal station apparatus. Generate a series. The second signal processing unit 305 removes residual interference components from the signal based on the generated signal sequence.

  That is, the first signal processing unit sets the transmission weight vector for the MIMO channel matrix used for radio communication between the first antenna element group and the second antenna element group for each second radio station apparatus to the second At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix of each radio station apparatus, or the approximate solution of the first right singular vector and the first left singular vector Based on at least one of the approximate solutions, at least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the other second radio station apparatus spatially multiplexed at the same time, or Calculation is performed so as to be orthogonal to at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. The first signal processing unit generates a transmission signal for each second radio station apparatus, and generates a transmission signal vector by multiplying the generated transmission signal by a transmission weight vector. The first signal processing unit adds and synthesizes transmission signal vectors for the second radio station apparatus spatially multiplexed at the same time, and a plurality of signals based on the components of the added and synthesized signal vectors are transmitted from the corresponding first antenna element. To the second radio station apparatus.

  Further, the first signal processing unit outputs a reception weight vector for the MIMO channel matrix used for radio communication between the first antenna element group and the second antenna element group for each second radio station apparatus, to the second At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix of each radio station apparatus, or the approximate solution of the first right singular vector and the first left singular vector Based on at least one of the approximate solutions, at least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the other second radio station apparatus spatially multiplexed at the same time, or Calculation is performed so as to be orthogonal to at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. The first signal processing unit multiplies the signal vector based on the signal received via the first antenna element group by the reception weight vector for each second radio station apparatus, for each second radio station apparatus. 1 system signal sequence is generated. The second signal processing unit removes the residual interference component from the signal based on the plurality of signal sequences for each generated first signal processing unit.

  Accordingly, the base station device 70, the terminal station device 60, the wireless communication system 90, and the wireless communication method of the tenth embodiment can increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. The base station device 70, the terminal station device 60, the wireless communication system 90, and the wireless communication method of the tenth embodiment can increase the transmission capacity in an environment where the line-of-sight environment is dominant even when multiuser MIMO is performed. It becomes possible.

[Eleventh embodiment]
[Mobile Fronthaul Configuration Method Using Virtual Transmission Line Corresponding to First Singular Value]
(Outline of the basic principle according to the eleventh embodiment)
In the fifth embodiment, time base beamforming is used to multiply one system of digital baseband sampling data by a transmission weight for each antenna system and each sampling value, thereby providing a virtual corresponding to the first singular value. The directivity formation for transmission on the transmission line was performed. Since transmission weight multiplication here can be performed directly on the sampling data, as with the mobile fronthaul configuration shown in FIG. 12A, the BBU side generates digital baseband sampling data. All signal processing is implemented, and one system of digital baseband sampling data is transmitted on the optical fiber, and the remaining processing can be performed on the RRH side. On the RRH side, transmission weight multiplication processing for time-axis beamforming is required, but this transmission weight information is transferred on an optical fiber, and this transferred transmission weight is converted into digital baseband sampling data on the RRH side. Since only multiplication and transmission are performed, all processes depending on the wireless communication system are basically implemented on the BBU side. The only element that depends on the radio communication system in the RRH is that the transmission directivity control does not have frequency dependence. In effect, communication is performed by forming a very narrow directional beam by a multi-element antenna. Means to do. Ideally, this directional beam should be directed to the communication partner station in the line-of-sight environment, but if a very strong reflected wave arrives even if there is no line of sight, it will be reflected at the reflection point. This can be done by forming a directed directional beam. In a situation where a very small number of reflected waves are gathered together, resulting in a reasonable reception level, the line gain of the virtual transmission line corresponding to the first singular value is degraded compared to the line-of-sight environment. However, there is a tendency to use antenna elements with high directivity gain because the distance attenuation is large and the line design is severe, and there is a tendency to use a large number of multiple reflected waves. Is expected. For this reason, as a mobile fronthaul in a wireless system using a high frequency band in particular, a time-axis beamforming is performed using a virtual transmission line corresponding to the first singular value, so that light can be utilized while utilizing a large-scale antenna. It is possible to reduce the information transmission capacity on the fiber.

  The information capacity when this transmission weight information is transferred on the optical fiber is estimated. For example, considering the case of using OFDM, the number of FFT points at that time is assumed to be 1024, for example. In this case, for example, assuming a guard interval of 256 samples, the number of samples of 1 OFDM is 1280 samples. On the other hand, for example, when 256 antenna elements are used, 256 pieces of transmission weight information are required (because the transmission weight vector is 256-dimensional). If the number of bits of one transmission weight information and the number of bits of one sample are the same, the transmission weight information corresponds to 20% of information. In other words, even when transmission weight information is transferred every time for every OFDM symbol, it is accommodated by 20% more than the mobile fronthaul configuration shown in FIG. This is overwhelmingly smaller than the 256 times the amount of information in the mobile fronthaul configuration shown in FIG. 13A, and is almost the same as the configuration shown in FIG. Furthermore, if the transmission weight information is not transferred every OFDM symbol but is transmitted every several OFDM symbols, the amount of information of the transmission weight information can be further compressed. . Alternatively, the transmission weight information may be transferred only when the transmission weight information is changed. In this way, the signal processing function depending on the radio transmission system is concentrated on the BBU side, and the time axis beam for configuring the virtual transmission path corresponding to the first minimum singular value necessary on the RRH side. It becomes possible to make it the structure which mounts a forming function.

  FIG. 76 is a diagram showing an overview of the function sharing of the mobile fronthaul using the virtual transmission path corresponding to the first singular value in the eleventh embodiment. Here, only the functions related to signal transmission (BBU to RRH direction) related to the direction from the network side to the user are extracted. In the figure, reference numeral 401 denotes a MAC layer processing circuit, reference numeral 402 denotes a transmission signal processing circuit, reference numeral 403 denotes a time axis signal generation circuit, reference numeral 454 denotes an optical interface circuit, reference numeral 455 denotes an optical fiber, reference numeral 456 denotes an optical interface circuit, reference numeral 457 Is a time axis transmission weight multiplication circuit, 427 is a D / A converter, 428 is an RF processing circuit, 429 is an antenna element, 432 is a transmission weight processing circuit, 441-4 is BBU, 442-4 is Represents RRH. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the time axis signal generation circuit 403, and the time axis transmission weight multiplication circuit 457 constitute an area 446-1 and an area 446-2 that perform baseband signal processing related to radio as a whole.

17 and 47 and FIG. 76, the MAC layer processing circuit 78 in FIG. 17 corresponds to the MAC layer processing circuit 401 in FIG. The second transmission signal processing unit 71 in FIG. 17 corresponds to the transmission signal processing circuit 412 in FIG. 47 corresponds to the time-axis signal generation circuit 403. The first transmission signal processing circuit 311 in FIG. 47 corresponds to the time axis transmission weight multiplication circuit 457 in FIG. 47 corresponds to the D / A converter 427 of FIG. 76. The D / A converters 814-1 to 814 -N ′ _BS-Ant of FIG. Mixer _{816-1~816-N 'BS-Ant,} filter 817-1~817-N' in FIG. 47 _BS-Ant, high-power amplifier _{818-1~818-N 'BS-Ant} corresponding to the RF processing circuit 428 To do. Antenna elements 819-1 to 819 -N ′ _BS-Ant correspond to antenna element 429. The first transmission weight processing unit 330 in FIG. 47 corresponds to the transmission weight processing circuit 432 in FIG.

  In FIG. 76, when a signal to be transmitted is input from the network side to the BBU 441-1, the MAC layer processing circuit 401 performs the signal processing of the MAC layer, and the frame format used for transmission / reception in the wireless section and flows through the network side. Data frame format conversion / termination is performed, and a radio packet format signal is input to the transmission signal processing circuit 412. The transmission signal processing circuit 412 performs transmission signal processing of radio signals. Here, the transmission scheme in the radio section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. The signal generated in this manner is converted into a time axis signal by the time axis signal generation circuit 403. For example, taking the case of OFDM as an example, IFFT is performed to convert a frequency axis signal into a time axis signal, and a guard interval is inserted to perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a continuous signal in time series. On the other hand, for example, reception weight information collected in a reception weight processing circuit on the receiving side of the BBU not shown here is transmission weight processing via a communication control circuit or the like not shown here (or directly). Provide to circuit 432. The transmission weight processing circuit 432 performs a calibration process on the reception weight information and the like to calculate a transmission weight. Each sampling value of the digital baseband signal generated by the time axis signal generation circuit 403 and the transmission weight information calculated by the transmission weight processing circuit 432 are converted into a predetermined frame format by the optical interface circuit 454, and the optical signal is converted into an optical signal. It is converted into a signal and output to the optical fiber 455.

  The signal output to the optical fiber 455 is transmitted to the RRH 442-4 side. In the RRH 442-4, the optical interface circuit 456 converts the optical signal into an electrical signal, terminates the signal of a predetermined format, and is a digital baseband signal. As described above, the information sequence of sampling values and the transmission weight information are reproduced. Both the sampling value and the transmission weight information are input to the time axis transmission weight multiplication circuit 457 as the digital baseband signal. The time axis transmission weight multiplication circuit 457 multiplies the sampling value as a digital baseband signal by the transmission weight for each antenna system in sampling value units. Each sampling value multiplied by the transmission weight is input to a D / A converter 427. The D / A converter 427 converts a signal for each antenna system into an analog baseband signal at a predetermined clock rate, and an RF processing circuit 458. Then, each antenna system is converted to a radio frequency signal by an up-converter, and after the out-of-band radiation signal is removed by a filter for each antenna system, it is amplified by a high power amplifier for each antenna system. Send.

  As described above, the functions corresponding to the base station apparatus for wireless communication as a whole are accommodated in the BBU 441-4 and RRH 442-4 that are mediated by optical fibers. The feature here is that the BBU 441-4 provided in the network-side office is integrated with the wireless digital baseband signal processing. There is a merit that the change of the side is enough.

  The above description corresponds to the case where there is only one first transmission signal processing unit 181 in FIG. Is possible. FIG. 77 is a diagram illustrating an outline (downlink) of functional sharing of the mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. Here, only functions related to signal transmission (direction from BBU 441-5 to RRH 442-4a, 442-4b) regarding the direction (downlink) from the network side to the user are extracted. In the figure, reference numeral 461 denotes a MAC layer processing circuit, reference numeral 462 denotes a transmission signal processing circuit, reference numerals 403a to 403b denote time axis signal generation circuits, reference numerals 454a to 454b denote optical interface circuits, reference numerals 455a to 455b denote optical fibers, and reference numeral 456a. 456b is an optical interface circuit, 457a to 457b are time axis transmission weight multiplication circuits, 427a to 427b are D / A converters, 428a to 428b are RF processing circuits, 429a to 429b are antenna elements, and 470 is Transmission weight processing circuit, reference numeral 441-5 represents BBU, and reference numerals 442-4a to 442-4b represent RRH.

  77, the same symbols are assigned to the same components as those in FIG. 77. Since there are a plurality of RRH 442-4a to 442-4b in FIG. 77 that are the same as RRH 442-4 in FIG. 76, suffixes a and b are given for identification. Similarly, the time axis signal generation circuits 403a to 403b, the optical interface circuits 454a to 454b, and the optical fibers 455a to 455b are also used by the number of RRHs 442-4a to 442-4b that are the same as those in FIG. The signal processing is the same as that in FIG. 76, but the MAC layer processing circuit 461 and the transmission signal processing circuit 462 process the signal series for the number of systems of RRH 442-4a to 442-4b. For example, if spatial multiplexing is performed for one terminal station device using MIMO, the MAC layer processing circuit 461 performs MAC layer processing as a signal of one terminal station device, while the transmission signal processing circuit 462 performs spatial processing. In consideration of multiplexing, serial / parallel conversion into signals of a plurality of systems is performed, and transmission signal processing is performed for each. The transmission weight processing circuit 470, for example, receives the reception weight information collected in the reception weight processing circuit on the receiving side of the BBU (not shown here) via a communication control circuit (not shown here) (or directly). To get information). Processing such as calibration is performed based on the reception weight information for the plurality of systems, the respective transmission weights are calculated, and the transmission weight information is transferred to the RRHs 442-4a to 442-4b corresponding to the transmission weights. Note that the transmission weight processing circuit 470 may acquire the transmission weight in the same manner as the first transmission weight processing units 330 and 340 in the fifth embodiment.

  Next, FIG. 78 is a diagram illustrating an outline (uplink) of functional sharing of the mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. Here, only functions related to signal transmission (direction from RRH 444-1a, 444-1b to BBU 443-1) related to the direction (uplink) from the user side to the network are extracted. In the figure, reference numeral 471 is a MAC layer processing circuit, reference numeral 472 is a received signal processing circuit, reference numerals 474a to 474b are optical interface circuits, reference numerals 475a to 475b are optical fibers, reference numerals 476a to 476b are optical interface circuits, and reference numerals 473a to 473b. Is a time axis reception weight multiplication circuit, reference numerals 477a to 477b are A / D converters, reference numerals 478a to 478b are RF processing circuits, reference numerals 479a to 479b are antenna elements, reference numeral 480 is a reception weight processing circuit, and reference numerals 481a to 481b are correlations. Calculation circuit, reference numeral 443-1 represents BBU, 444-1a to 444-1b represent RRH. 78 shows a plurality of systems of RRH 444-1a to 444-1b as corresponding to FIG. 77, but only one system may be used as in FIG.

  In FIG. 78, when the antenna elements 479a to 479b receive a signal, the RF processing circuits 478a to 478b amplify the signal with a low noise amplifier for each antenna system, and multiply the signal by a common local oscillator as a whole with a mixer. Are converted, and an analog baseband signal is generated for each antenna system by removing signals outside the band with a filter. This signal is sampled by A / D converters 477a to 477b, and a data string of sample values of digital baseband signals for each antenna system is generated. On the other hand, the time axis reception weight multiplication circuits 473a to 473b multiply the reception weights, add and synthesize (multiply the reception signal vector by the reception weight vector), and convert it into one system digital baseband signal. On the other hand, the A / D converters 477a to 477b also output a data string of sample values of the digital baseband signal to the correlation calculation circuits 481a to 481b.

The correlation calculation circuit 481A～481b, the sampled values of the received signals of the second to the _{N Ant} antenna elements in the antenna elements 479A～479b, a complex conjugate value of the sampling values of the received signal of the first antenna element Multiply and add the operation results for a predetermined period. Similarly, the sampling value of the received signal of the first to N-th _Ant antenna elements of the antenna elements 479a to 479b is multiplied by the complex conjugate of the sample value, and the calculation result for a predetermined period is added. Based on the above result, the correlation calculation is performed by the equation (52), and the complex conjugate is taken to calculate the reception weight of each antenna element. The correlation calculation circuits 481a to 481b receive timing instruction signals from the reception weight processing circuit 480 on the BBU 443-1 side via the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface circuits 476a to 476b. Is done.

  Each time the correlation calculation circuits 481a to 481b receive the timing signal, the correlation calculation circuits 481a to 481b perform the correlation calculation using the equation (52) and reset the memory value of each addition. The correlation calculation circuits 481a to 481b collect the correlation information acquired here for all the antenna sequences, and transfer them to the reception weight processing circuit 480 via the optical interface circuits 476a to 476b, the optical fibers 475a to 475b, and the optical interface circuits 474a to 474b. To do. The reception weight processing circuit 480 calculates reception weight information based on the correlation information received here, records and manages the calculated reception weight information, and sends the correlation information or reception weight information to the transmission weight processing circuit 470 as necessary. Notification is performed here, and calibration processing or the like is performed for use in transmission weight calculation.

  Also, the reception weight processing circuit 480 grasps the scheduling information of the reception processing of the BBU 443-1 and the RRHs 442-4a to 442-4b (for example, the communication control circuit or MAC layer processing circuit 471 not shown here). The timing information indicating the switching timing is determined as the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface. It outputs to the correlation calculation circuits 481a to 481b via the circuits 476a to 476b. This timing instruction takes into account the time lag caused by passing through the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface circuits 476a to 476b. When the correlation calculation circuits 481a to 481b receive this switching timing instruction, the correlation calculation circuits 481a to 481b output the reception weight information calculated based on the correlation information calculated immediately before to the time axis reception weight multiplication circuits 473a to 473b. The time-axis reception weight multiplication circuits 473a to 473b continue to multiply the reception signal with the reception weight information last specified until this switching instruction is given.

FIG. 79 is a diagram showing an outline of the correlation detection circuit 481 (481a, 481b) in the eleventh embodiment. In the figure, reference numerals 491-2 to 491-N _Ant and reference numerals 495-1 to 495-N _Ant are multipliers, reference numerals 492-2 to 492-N _Ant and reference numerals 496-1 to 496-N _Ant are adders, Reference numerals 493-2 to 493 -N _Ant and reference numerals 497-1 to 497 -N _Ant are memories, reference numerals 498-1 to 498 -N _Ant are square root acquisition circuits, and reference numeral 494 is a correlation calculation control unit. The correlation calculation circuit 481 is connected to the A / D converter 477, the time axis reception weight multiplication circuit 473, and the optical interface circuit 476. Further, it is also connected to a reception weight processing circuit 480 on the BBU 443-1 side via an optical interface circuit 476, an optical fiber 475, and an optical interface circuit 474.

Correlation calculating circuit 481, the sampling values of each antenna system from the A / D converter 477 is input, to the first to sampled values of the received signal of the N _Ant antenna elements in the antenna element 479, multiplied by vessel in _{491-2~491-N Ant,} multiplied by the sampled value of the second to _{N Ant} antenna elements in the first complex conjugate value of the sampling values of the antenna and the antenna element 479 of the antenna element 479, a multiplication result Are output to the adders 492-2 to 492-N _Ant . Adder _{492-2~492-N Ant} adds the value of the input value and the memory _{493-2~493-N Ant,} stored by entering the addition result to the memory _{493-2~493-N Ant} Let By sequentially rotating the calculated values between the adders 492-2 to 492 -N _Ant and the memories 493-2 to 493 -N _Ant , the correlation results are sequentially added and the accumulated value is obtained. When a correlation value reset instruction is input from the reception weight processing circuit 480 on the BBU 443-1 side to the memories 493-2 to 493-N _Ant via the optical interface circuit 474, the optical fiber 475, and the optical interface circuit 476, the memory The 493-2 to 493 -N _Ant outputs the accumulated value to the correlation calculation control unit 494 and resets its value to zero.

In parallel with the above processing, when the sampling value of each antenna system is input from the A / D converter 477, the sampling value of the received signal of the first to N _Ant antenna elements in the antenna element 479 is multiplied by the multiplier. 495-1 to 495-N _Ant , the sampling values of the first to N _Ant antennas of the antenna element 479 and their complex conjugate values are multiplied by multipliers 495-1 to 495-N _Ant , and sampling is performed. The square value of the absolute value of the value is output to the adders 496-1 to 496-N _Ant . Adder _{496-1~496-N Ant} adds the value of the input value and the memory _{497-1~497-N Ant,} stored by entering the addition result in the memory _{497-1~497-N Ant} Let By sequentially rotating the calculated values between the adders 496-1 to 496-N _Ant and the memories 497-1 to 497-N _Ant , the square values of the absolute values of the sampling values are sequentially added, and the accumulated value is obtained. It will be requested.

When a correlation value reset instruction is input from the reception weight processing circuit 480 on the BBU 443-1 side to the memories 497-1 to 497-N _Ant via the optical interface circuit 474, the optical fiber 475, and the optical interface circuit 476, the memory _{497-1~497-N Ant} outputs the accumulated value to the square root acquisition circuit _{498-1~498-N Ant,} reset their values to zero. The square root acquisition circuits 498-1 to 498 -N _Ant determine the square root of the accumulated value and output the value to the correlation calculation control unit 494. The correlation calculation control unit 494 calculates a reception weight by performing a correlation calculation using the equation (52) and further taking the complex conjugate. The reception weight information is a collection of information regarding all antenna systems, and is transferred to the reception weight processing circuit 480 via the optical interface circuit 476, the optical fiber 475, and the optical interface circuit 474. Similarly, this reception weight is also input to the time axis reception weight multiplication circuit 473.

  FIG. 80 is a diagram illustrating an outline (uplink) of function sharing of another mobile fronthaul using virtual transmission lines corresponding to a plurality of first singular values in the eleventh embodiment. Here, only functions related to signal transmission (direction from RRH 444-2a to 444-2b to BBU 443-2) related to the direction (uplink) from the user side to the network are extracted. In the figure, reference numerals 484a to 484b are optical interface circuits, reference numerals 485a to 485b are optical fibers, reference numerals 486a to 486b are optical interface circuits, reference numeral 482 is a reception weight processing circuit, reference numerals 481a to 481b are correlation calculation circuits, and reference numeral 443 is provided. 2 represents BBU, and symbols 444-2a to 444-2b represent RRH.

  In the figure, the same components as those in FIG. 78 are denoted by the same reference numerals. The difference between FIG. 80 and FIG. 78 is that the reception weight input to the time axis reception weight multiplication circuit 473 is changed from the reception weight processing circuit 482 on the BBU 443-2 side to the optical interface circuit 484, the optical fiber 485, and the optical interface circuit 486. It is a point to notify via. For this reason, it is possible to instruct to use the reception weight acquired in the past managed on the reception weight processing circuit 482 side without necessarily using the reception weight calculated by referring to the training signal received immediately before. Yes.

  As another variation, in FIG. 78, the correlation calculation circuits 481a and 481b are provided with a storage / management function for reception weights, and instead of notifying the reception weight information itself via the optical fiber 475 as shown in FIG. The reception weight information managed by the circuit 481 is notified from the BBU 443-1 to the RRH 444-1a and 444-1b, and the correlation calculation circuits 481a and 481b store the reception weight information based on the identification number. May be read out and output to the time axis reception weight multiplication circuit 473.

  Next, an embodiment when the mobile fronthaul configuration in the present embodiment is applied to multi-user MIMO will be described. FIG. 81 is a diagram illustrating an overview (downlink) of functional sharing when mobile fronthaul multi-user MIMO is applied using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. . Here, only functions related to signal transmission (direction from BBU 441-6 to RRH 442-6) regarding the direction (downlink) from the network side to the user are extracted. In the figure, reference numeral 463 indicates a transmission signal processing circuit, reference numeral 464 indicates a transmission weight processing circuit, reference numeral 465 indicates an addition / synthesis circuit, reference numeral 441-6 indicates BBU, and reference numeral 442-6 indicates RRH.

  In the figure, the same components as those in FIG. 77 are denoted by the same reference numerals. In the description so far, basically, a plurality of RRHs may perform spatial multiplexing communication as a single user MIMO with a single terminal station device, or may communicate with different terminal station devices assuming that mutual interference is limited. I've explained that you can do it. On the other hand, FIG. 81 is extended to multi-user MIMO for transmitting signals from one RRH 442-6 to different terminal station apparatuses, and FIG. 74 used for explaining the tenth embodiment related to multi-user MIMO is cited. While explaining. 74 is not necessarily based on time axis beam forming, but here, description will be made based on the example of FIG. 74 in the case of performing time axis beam forming.

In the configuration of the first transmission signal processing unit 181 of the base station apparatus 70 corresponding to the base station apparatus 303 at the time of applying the multiuser MIMO shown in FIG. The transmission signal processing circuit 463 in FIG. 81 performs processing corresponding to the transmission signal processing circuits 113-1 to 113-3. Also, the addition synthesis circuit 465 of FIG. 81 performs processing corresponding to the addition synthesis circuit 812-1 to 812-N ′ _BS-Ant of FIG. Further, the processing corresponding to the transmission weight processing unit 150 of FIG. 74 is configured to be executed by the transmission weight processing circuit 464 of FIG. Further, the outputs of the plurality of RRH 442-4a to 442-b time base transmission weight multiplication circuits 457a to 457b in FIG. The adder / synthesizer 465 adds and synthesizes signals transmitted by the same antenna element, and outputs the addition result to the D / A converter 427a. The addition combining circuit 465 combines the two systems of transmission signals into one signal.

  In the description of FIG. 77, two RRHs 442-5a to 442-b may transmit signals to one terminal station apparatus or may transmit to different terminal station apparatuses. Since it is MIMO, different signal sequences are transmitted to different terminal station apparatuses. Except for the above differences, the same signal processing is performed with basically the same configuration as in FIG. Here, regarding the optical interface circuits 454a to 454b, the optical fibers 455a to 455b, and the optical interface circuits 456a to 456b, the BBU 441-6 and the RRH 442-6 are connected via physically different optical fibers and optical interface circuits. Although an example of the case has been shown, if an optical fiber capable of realizing a large-capacity transmission is used, it is also possible to physically exchange information on the same optical interface circuit and the same optical fiber. In this sense, what is shown in FIG. 81 may be understood as a “logical transmission line”.

  FIG. 82 is a diagram illustrating an overview (uplink) of functional sharing when mobile fronthaul multi-user MIMO is applied using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. . Here, only functions related to signal transmission (direction from RRH 444-3 to BBU 443-3) regarding the direction (uplink) from the user side to the network side are extracted. In the figure, reference numeral 466 represents a received signal processing circuit, reference numeral 467 represents a reception weight processing circuit, reference numeral 443-3 represents BBU, and reference numeral 444-3 represents RRH. In the figure, the same components as those in FIG. Also in this description, the extension to multi-user MIMO that receives signals from different terminal station devices in one RRH will be described with reference to FIG. 75 that explains multi-user MIMO. However, although FIG. 75 is not necessarily based on time-axis beam forming, here, description will be made based on the example of FIG. 75 in the case of performing time-axis beam forming.

  In the configuration of the first received signal processing unit 186 of the base station device 70 corresponding to the base station device 303 when the multiuser MIMO is applied, shown in FIG. The time axis reception weight multiplication circuit 483 in FIG. 82 performs processing corresponding to the reception signal processing circuits 114-1 to 114-3. In the fifth embodiment, the first received signal processing unit 385 of the base station apparatus performs signal processing for extracting a virtual transmission path corresponding to the first singular value of the MIMO channel matrix. Further, as in the tenth embodiment, when multi-user MIMO is applied, the first received signal processing unit 186 performs signal processing for extracting a virtual transmission path corresponding to the first singular value of the MIMO channel matrix. In the first embodiment, interference components that could not be completely removed by this signal processing are removed by the second received signal processing unit 75 in the subsequent stage. On the other hand, in the eleventh embodiment, the second-stage signal processing is performed to remove the interference component that the reception signal processing circuit 466 could not completely remove. By performing the two-stage signal processing in this way, signal separation can be performed efficiently.

  Also, the processing corresponding to the reception weight processing unit 170 in FIG. 75 is configured to be performed by the reception weight processing circuit 467 in FIG. Further, in FIG. 75, a signal is output only to one time axis reception weight multiplication circuit 473 in one RRH 444-1, whereas in FIG. 82, a plurality of outputs from one A / D converter 477a are output. Are output to the time axis reception weight multiplication circuits 483a to 483b.

  The plurality of time axis reception weight multiplication circuits 483a to 483b multiply the same reception signal vector by the time axis reception weight vector corresponding to each different terminal station apparatus. The multiplication results are input to the reception signal processing circuit 466 via the optical interface circuits 486a to 486b, the optical fibers 485a to 485b, and the optical interface circuits 484a to 484b, respectively. Except for the above differences, basically the same signal processing is performed with the same configuration as in FIG. Here, regarding the optical interface circuits 486a to 486b, the optical fibers 485a to 485b, and the optical interface circuits 484a to 484b, the RRH 444-3 and the BBU 443-3 are connected via physically different optical fibers and optical interface circuits. Although an example of the case has been shown, if an optical fiber capable of realizing a large-capacity transmission is used, it is also possible to physically exchange information on the same optical interface circuit and the same optical fiber. In this sense, what is shown in FIG. 82 may be regarded as a “logical transmission path” for understanding.

  In the wireless communication system according to the eleventh embodiment, the BBU includes an optical interface circuit 454 serving as a first transmission unit and a third transmission unit. The optical interface circuit 454 transmits the sampling data sequence generated by the time axis signal generation circuit 403 and transmission weight information (transmission weight vector) to the RRH. The BBU includes reception weight processing circuits 480, 482, and 467 as reception weight acquisition means. The reception weight processing circuit acquires reception weight information (reception weight vector) used by the RRH for reception from the terminal station apparatus. The reception weight information is calculated based on the correlation value calculated in RRH. The reception weight processing circuit instructs RRH to calculate a correlation value and reception weight information.

  In the radio communication system according to the eleventh embodiment, the RRH includes a time axis reception weight multiplication circuit 473 as a second transmission unit. A time axis reception weight multiplication circuit 473 sequentially multiplies the reception sampling signal vector sequence, which is a data sequence of sample values of the digital baseband signal output from the A / D converter 477, for each sampling data, The multiplication results are added and combined to generate a single sampling data string, and a single sampling data series is transmitted to the BBU via the optical interface circuit 456. The RRH includes an optical interface circuit 456 as a fourth transmission unit. The optical interface circuit 456 collects the correlation information calculated by the correlation calculation circuit 481 for all antenna sequences, and transmits it to the BBU via the optical interface circuit 476.

  In the radio communication system according to the eleventh embodiment, the RRH includes optical interface circuits 456 and 476 serving as a first receiving unit and a second receiving unit. The optical interface circuit 456 receives transmission weight information (transmission weight vector) and a sampling data string from the BBU. The optical interface circuit 476 receives reception weight information (reception weight vector) from the BBU. The RRH includes a correlation calculation circuit 481 as a reception weight calculation unit. Correlation calculation circuit 481 uses one of a plurality of antenna elements 479 as a reference antenna element based on a signal when each antenna element 479 receives a known signal transmitted from a terminal station apparatus, and uses the reference antenna element as a reference antenna element. The correlation value over a predetermined number of samples between the known signal received in step S4 and the known signal received by the other antenna element 479 is calculated, and the reception weight vector is calculated based on the calculated correlation value.

  With the above correspondence, according to the radio communication system in the eleventh embodiment, in the downlink from the BBU to the RRH direction, in the uplink from the RRH to the BBU direction, and further including extension to multi-user MIMO. The RRH is equipped with a multi-element antenna, so that the amount of information flowing on the optical fiber is not enormous, and the basic aim of the mobile fronthaul is achieved with a slight increase in the amount of information than before. It is possible to optimize the function distribution as it is maintained, and it is possible to realize band expansion in a situation where the RRH and the terminal station apparatus are in a line-of-sight environment and the direct wave is dominant.

  In the wireless communication system according to the eleventh embodiment, the configuration in which the optical fiber is used as the communication medium between the BBU and the RRH has been described. However, another medium, for example, a medium for transmitting an electrical signal such as a coaxial line or a wireless A line may be used.

[Additional matters regarding the relationship between various embodiments]
First, the embodiment of the present invention is based on the fact that a plurality of signal sequences are spatially multiplexed by utilizing virtual transmission paths corresponding to the plurality of first singular values shown in the first embodiment as the basis. It is a feature. The relationship with specific embodiments is summarized below.

(Combination with the second embodiment of the present invention)
The second embodiment of the present invention is a specific embodiment related to optimization of the arrangement interval of the first signal processing unit on the base station apparatus side in order to efficiently realize the first embodiment of the present invention. It is shown. In the case of the fourth embodiment, since the distance between the base station apparatus and the train changes with time, the optimization condition shown in the second embodiment cannot be used directly, but other embodiments In this case, it is possible to utilize the second embodiment. Specifically, in the third embodiment, since the satellite base station device and the central base station device are fixedly installed, the satellite base station device is regarded as the first signal processing unit, and the central base station device Is assumed to be a terminal station device, and spatial multiplexing transmission is performed between the satellite base station device and the overall base station device using virtual transmission paths corresponding to a plurality of first singular values, the satellite base station device is If it is arranged in a straight line and the arrangement condition is set so as to satisfy a predetermined condition with respect to the average distance between the central base station apparatus and the satellite base station apparatus, the third embodiment and the second implementation Operation combining the form and the first embodiment is possible. Also, in contrast to the example in which the satellite base station apparatus of the third embodiment as shown in FIG. 32 is used, the present technology is applied to the access system purely using the first embodiment instead of the satellite base station apparatus. 83 in the case of applying to the first signal processing shown in the second embodiment with respect to the average distance between the first signal processing unit and the ground terminal station device. By optimizing the arrangement interval of the parts, it is possible to stabilize the communication between each terminal station device and the base station device because it can maintain a generally good orthogonal relationship although it is out of the perfect orthogonal condition. become. For this reason, even when the present technology is applied to an access system, an operation combining the second embodiment and the first embodiment is possible. In terms of application to this access system, the eighth embodiment and the tenth embodiment are also applicable, and the second embodiment and the first embodiment can be further combined with these embodiments. It is.

  In the description of the second embodiment, for example, in FIG. 29, as described with the antenna element on the base station apparatus side replaced with a parabolic antenna, it is assumed that the terminal station apparatus side includes a plurality of antenna elements. However, the base station apparatus side does not necessarily include the multi-element antenna group as shown in the first signal processing unit of the first embodiment, and is substantially composed of one antenna element with an ultrahigh gain. The terminal station apparatus side can configure a virtual transmission line corresponding to the first singular value in combination with a plurality of antenna elements.

(Combination with the third embodiment of the present invention)
As described above, in the case of the fourth embodiment, since it is specialized in direct communication between the terminal station device on the train side and the base station device, the third embodiment of the present invention that performs 2-hop relay transmission The combination with the form is difficult to think, but the combination of the third embodiment and the second embodiment of the present invention is possible, and in the other operating conditions of the fourth embodiment, It is applicable also about a combination. For example, not only the first embodiment is performed by single user MIMO transmission, but also the tenth embodiment (the tenth embodiment) can be expanded to multi-user MIMO transmission as shown in the tenth embodiment. It is also possible to realize multi-user MIMO while combining the satellite base station apparatus of the third embodiment by combining the first and second embodiments).

(Combination with the fourth embodiment of the present invention)
As described above, the fourth embodiment of the present invention is a kind of unique usage, but here, the signal processing for directivity formation on the time axis shown in the fifth embodiment may be performed in combination. The channel estimation process can be used in combination with the sixth and seventh embodiments of the present invention. Here, only one-to-one communication between one terminal station device on the train side and the base station device is described as an example, and in this sense, single-user MIMO transmission is fundamental, In a case where multiple trains exist simultaneously in the same section, in addition to segregating by changing the frequency channel for each train, if multi-user MIMO shown in the tenth embodiment is used in combination, Communication with a plurality of trains can be realized in parallel on the same channel, which makes it possible to effectively use frequency resources.

(Combination with the fifth embodiment of the present invention)
The technology intended by the fifth embodiment of the present invention can be applied to a system of any condition such as a single carrier transmission system as a wireless communication system or a multicarrier system such as OFDM. In the case of a system that presupposes signal processing on the frequency axis such as OFDM, the direct merit of applying the fifth embodiment of the present invention is small (however, the number of mounted FFT circuits and IFFT circuits can be suppressed). However, when a large-scale antenna is mounted on the RRH side in the mobile fronthaul as in the eleventh embodiment, even if it is a multi-carrier system such as OFDM, it is transmitted on the optical fiber. In order to expand the information capacity to a large antenna while compressing the information capacity, a signal on the time axis is transmitted on the optical fiber, and the time axis beam forming shown in the fifth embodiment is used on the RRH side. In this sense, the eleventh embodiment is premised on the combined use of the fifth embodiment (and the first embodiment).

(Combination with the sixth and seventh embodiments of the present invention)
The sixth and seventh embodiments of the present invention are embodiments showing means for feedback of channel information used for directivity formation, and can be used in combination in all the embodiments of the present invention. This can be used also in a typical fixed installation type terminal station apparatus used in the description of the first embodiment, and can also be applied in an environment in which a channel varies with time in an access system. In the eighth embodiment of the present invention, a specific configuration method for simultaneously performing channel estimation between a plurality of adjacent cells or a plurality of terminal station apparatuses in the same cell, assuming use in an access system in particular. It is shown and becomes embodiment which assumed the combination of this invention 6th Embodiment and 7th Embodiment. Furthermore, if the channel gain in the channel estimation is usually insufficient, the channel gain between the terminal station device on the train side and the base station device shown in the fourth embodiment can be used to improve the channel gain. It is also possible to operate by combining the sixth embodiment and the seventh embodiment (and the first embodiment) of the present invention with the fourth embodiment.

  In particular, as a specific embodiment in which the sixth embodiment and the seventh embodiment are directly combined, training is performed on discrete subcarriers by avoiding overlapping of subcarriers between antenna elements of a terminal station device or a base station device. When assigning to signals, it is assumed that each antenna element of the terminal station apparatus or base station apparatus is transmitted from the same virtual antenna element, and relative channel information acquired for a plurality of subcarriers from a plurality of different antenna elements On the other hand, the relative channel information including subcarriers to which no training signal is assigned is obtained by using the least square method or the like, and the reception weight (and the cancellation of the complex phase difference of the relative channel information based on the relative channel information (and It is possible to calculate the transmission weight by calibration processing).

(Combination with the eleventh embodiment of the present invention)
The eleventh embodiment of the present invention is similar to a mode in which the first signal processing unit and the second signal processing unit in the first embodiment of the present invention are isolated and operated between them using an optical fiber. However, in the first embodiment, various types of signal processing for forming directivity for transmission and reception are concentrated in the first signal processing unit, whereas in the eleventh embodiment, it corresponds to the second signal processing unit. It has been consolidated on the BBU side. However, when performing spatial multiplexing transmission by actively utilizing virtual transmission paths corresponding to a plurality of first singular values, the basic principle is shared, and in this respect, the eleventh embodiment. And the operation corresponding to the combination of the fifth embodiment and the first embodiment (the arrangement of the first signal processing unit is strictly different, but in the sense of utilizing the virtual transmission line corresponding to the first singular value) This is considered to correspond to the first embodiment). Further, as shown in the eleventh embodiment, it can be extended to the multiuser MIMO transmission shown in the tenth embodiment. In this case, the eleventh embodiment, the tenth embodiment, and the The operation corresponds to the combination of the fifth embodiment and the first embodiment. As described above, the train moving cell shown in the fourth embodiment also regards the plurality of first signal processing units as RRHs using the eleventh embodiment, and the BBU-side first through the optical fiber. If the information is exchanged with the two signal processing units, the operation corresponding to the combination of the eleventh embodiment, the fourth embodiment, and the first embodiment is also possible. Naturally, if this is combined with multi-user MIMO including terminal equipment of a plurality of trains, the tenth embodiment is combined at the same time.

  As described above, various embodiments of the present invention can be operated in combination with each other.

[Other supplementary items of the embodiment of the present invention]
Below, some supplementary matters regarding the embodiment according to the present invention will be described.

In the description of each embodiment of the present invention, the receiving-side antenna elements 851-1 to 851- (N ′ _BS-Ant ) and the transmitting-side antenna elements 819-1 to 819- (N ′ _BS-Ant ) are Although different explanations are given individually, the antenna elements (851-1 to 851- (N′BS _-Ant )) or the transmission system and the reception system that are commonly used via the TDD-SW or the like are naturally used. 819-1 to 819- (N′BS _-Ant )). In particular, when performing implicit feedback, it is essential to share the antennas of the transmission system and the reception system, and TDD control that switches between the uplink and the downlink in time is fundamental. The switching of the TDD-SW here is managed and controlled by the communication control circuit 120 or the like.

Also, receiving-side antenna elements 851-1 to 851- (N ′ _BS-Ant ) and transmitting-side antenna elements 819-1 to 819- (N ′ _BS-Ant ) related to the base station apparatus, and receiving related to the terminal station apparatus The antenna elements 851-1 to 851- (N _MT-Ant ) on the transmission side and the antenna elements 819-1 to 819- (N _MT-Ant ) on the transmission side use common numbers to simplify the explanation. And explain. However, the requirements such as the antenna element size and directivity gain related to the base station apparatus and the requirements such as the antenna element size and directivity gain of the terminal station apparatus generally do not match and the functions are the same. Although the same numbers are assigned to the antenna elements, antenna elements having different conditions in terms of size, directivity gain, and the like may be used in actual operation. The same applies to other circuits, and a common number is used to simplify the description. For example, high power amplifiers 818-1 to 818-(N ′ _BS-Ant and N _{MT− Ant} ), low noise amplifiers 852-1 to 852- (N'BS _-Ant and NMT _-Ant ), etc., the requirements for the amplifier amplification factor, power consumption, size, allowable heat generation, etc. are generally the same. Since the functions are the same and the functions are the same, the same numbers are assigned. However, circuits under different conditions may be used in actual operation.

  Further, the transmission weight vector in the embodiment of the present invention means each column vector (or a part of column vectors) of the transmission weight matrix, and is a transmission weight expressed as a vector focusing on one of the signal sequences to be spatially multiplexed simultaneously. is there. Specifically, each column vector of a transmission weight matrix when spatially multiplexing a plurality of signal sequences is a vector notation of weights (transmission weight vector components) focused on a certain signal sequence in the plurality of signal sequences. Thus, it is sequentially obtained in the process of generating the entire transmission weight matrix based on the channel vector or channel matrix of the spatially multiplexed terminal station apparatus. Therefore, generation of a transmission weight vector (and the case where words such as “calculation”, “decision”, “multiplication”, and “component” follow) is generally equivalent to generation of a transmission weight matrix, and in particular, When considering the sequence of generating the matrix row vector or column vector in order, it may be labeled as “transmission weight vector generation” in the same meaning as “transmission weight matrix generation”.

  Similarly, the reception weight vector in the embodiment of the present invention means each row vector (or a part of the row vectors) of the reception weight matrix, and is a vector notation that focuses on one of the spatially multiplexed signal sequences at the same time. Receive weight. Specifically, each row vector of the reception weight matrix when a plurality of signal sequences are spatially multiplexed is a vector notation of weights (reception weight vector components) focusing on a certain signal sequence in the plurality of signal sequences. Thus, it is sequentially acquired in the process of generating the entire reception weight matrix based on the channel vector or channel matrix of the spatially multiplexed terminal station apparatus. Therefore, the generation of the reception weight vector (and the case where words such as “calculation”, “decision”, “multiplication”, “component”, etc.) follow is equivalent to the generation of the reception weight matrix as a whole, and in particular, When considering the sequence of generating matrix row vectors or column vectors in order, it may be labeled as “receiving weight vector generation” in the same meaning as “receiving weight matrix generation”.

  Furthermore, “transmission weight vector”, “reception weight vector”, “channel (information) vector”, “transmission signal vector”, “reception signal vector”, etc. These are all vectors corresponding to each antenna element, including “transmission weight”, “reception weight”, “channel (information)”, “transmission signal”, and “reception signal”. Even if there is no explicit description of “vector” in each embodiment, it is clear that those components are those “vectors” in the embodiment. It should be understood by supplementing these contents. Further, “transmission signal” and “reception signal” in “transmission signal vector” and “reception signal vector” are transmission or reception signals corresponding to or based on each antenna element, and are actually transmitted and received. It may be a digitized baseband (or intermediate frequency) signal instead of a radio frequency analog signal. This signal includes both a signal on the frequency axis and a sampling signal on the time axis. Therefore, in the above description of the embodiment, there is a part that is not explicitly described, but these signals are not only signals transmitted / received at a radio frequency but also signals transmitted / received at a radio frequency. Some signal processing may be performed between them.

  In addition, in the description related to channel estimation in the embodiment of the present invention, a case where channel estimation results are averaged in addition to the case where channel information is acquired by one channel estimation is described. For example, if the line gain by utilizing a large-scale antenna element is considered, there is no problem even if the line gain is insufficient in the line design between one antenna and one antenna. An appropriate transmission / reception weight is necessary to increase the performance gain, and the channel information necessary for calculating the transmission / reception weight is basically based on the channel estimation result between the single antenna and the single antenna. If the line gain is insufficient, the subsequent directivity gain cannot be obtained. As a countermeasure for this, Non-Patent Document 2 and the like received a plurality of symbols of training signals, and compensated for a lack of line gain by averaging the received signals. As the averaging of the estimation results of a plurality of times, when a periodic training signal is continuous over several symbols, a method of performing short-time averaging over several symbols using the periodicity, There is a method of performing long-time averaging in which channel estimation results performed at a time are collected and averaged multiple times. Here, in the case of short-time averaging, since a plurality of symbols to be averaged are continuous, it is considered that all symbol timings are preserved due to their periodicity. There was no need to treat it as a relative channel based on complex phase. On the other hand, regarding channel estimation results performed at discrete times, the symbol timing is generally not necessarily the same. Therefore, in order to eliminate the influence of the symbol timing error, relative channel information is used. It was necessary to take a configuration to obtain and average. In this case, the complex phase of the channel information of the reference antenna element is considered to be all zero (that is, take a real value). In calculating the relative channel information, the channel information of all antenna elements is used as a reference in addition to multiplying the channel information of all antenna elements by Exp (−jθ) to the complex phase θ of the reference antenna. You may obtain | require in the form divided by the channel information of an antenna element. Such utilization of relative channel information is generally indispensable for averaging channel estimation results with different symbol timings. However, when the symbol timings are common, the relative channel information is generally used when calculating transmission / reception weights. There is no need to use information. It is only necessary to simply calculate the transmission / reception weight for the acquired channel information using Equation (1) or Equation (7). However, even if the symbol timing is common, there is no problem even if the relative channel information is used. Therefore, as the above explanation, sometimes it is explained as relative channel information or simple channel information is used. In some cases, the description is used as it is. However, the difference is as described above, and it is not essential to use relative channel information.

  Relative channel information can also be handled as channel information obtained by correcting the complex phase with reference to the complex phase of the reference antenna, and the channel information of each antenna element is based on the channel information of the reference antenna including the amplitude. The information may be divided. For example, the amplitude of each relative channel information has a meaning in the case of the maximum ratio combining reception weight represented by Expression (7), but in the case of the equal gain combining transmission / reception weight represented by Expression (1). do not have. Furthermore, in practice, it is expected that the amplitude deviation for each antenna is extremely limited in the line-of-sight environment. In that case, even if all of them are approximated to the same amplitude, there is no significant difference. In the first place, the absolute value of the transmission / reception weight has no significant meaning and needs to be optimized separately so that it becomes an efficient value within the finite number of quantization bits, but it depends on the number of such quantization bits. The discussion is completely different from the embodiment of the present invention, and may be optimized among existing technologies. In that sense, the magnitude (absolute value) of the transmission / reception weight vector as a vector has no special meaning here, and the transmission / reception weight vector multiplied by an arbitrary coefficient is also assumed here to retain its statistical properties. I am explaining.

  By using the above relative channel information, various signal processing can be simplified in the embodiment of the present invention. On the other hand, in the conventional MIMO transmission technique, for example, the reception weight matrix multiplication processing is often described as including processing for directly converting to a state suitable for signal detection. That is, even for a SISO signal, for signal detection processing, the received signal is divided by the channel estimation result, and the signal points on the constellation in which the I and Q axes are correctly set from the received signal subjected to channel distortion. Needed to be converted to In the embodiment of the present invention, the transmission / reception weight multiplication processing performed in the first signal processing unit is mainly for signal processing for ensuring directivity gain and rough signal separation. The reception signal is divided by the channel estimation result, and the I and Q axes are obtained from the reception signal subjected to the channel distortion only by making it possible to handle a large number of antenna elements as one virtual antenna element by weight multiplication. Does not include the process of converting the signal point to a signal point on the constellation set correctly. However, on the receiving side, it is also possible to perform processing such as signal detection in the subsequent stage, and for these processing, it is possible to apply the same signal processing as that performed in conventional MIMO transmission or SISO transmission. is there. In particular, in the reception system of the base station apparatus, these processes are performed in the second signal processing circuit.

  In addition, when transmitting training signals composed of limited subcarrier components from different antenna elements on the transmitting station side while avoiding duplication of subcarriers assigned by each antenna element, there is no duplication on the frequency axis. The nuclear training signal is synthesized in space, and is received as a training signal including more subcarrier components on the receiving station side. In this sense, on the receiving station side, it differs from the training signal transmitted by each antenna element on the transmitting side. Signal processing can be performed. When using such a combined training signal, it is effective to convert channel information into relative channel information that is a relative value with respect to a reference antenna element.

  In the above description, for the sake of simplicity, k (for example, the k-th frequency component) representing a subcarrier is omitted, and further, description regarding individual subcarriers is also omitted. The assumed system is a broadband system, and all signal processing in the channel information, transmission / reception weight, transmission signal, reception signal, etc. is basically the same except for the signal processing on the time axis in the fifth embodiment. Specifically, it should be individually defined and processed for each subcarrier on the frequency axis. Therefore, in order to simplify the explanation, in many explanations, the subscripts that explicitly represent the subcarriers are omitted. However, these explanations are actually performed individually for each subcarrier, and in this case, the explanation can be interpreted strictly by adding a subscript representing the subcarrier. In each signal processing circuit, for example, signal processing up to the previous stage of IFFT processing on the transmission side (for example, assuming an OFDM modulation method, bit string interleaving processing, signal point mapping, signal modulation processing, transmission weight vector The signal processing after FFT processing on the receiving side (also assuming the OFDM modulation method is assumed), reception weight multiplication, signal detection processing, signal de- duction are all performed for each subcarrier. Mapping, deinterleave processing, etc.) are all performed for each subcarrier.

  In terms of circuit configuration, individual circuits may be provided for each subcarrier, and since the same processing is performed, processing is serially performed for each subcarrier, and the circuit is shared with the subcarriers. It is also possible to Furthermore, in the middle, a plurality of circuits may be prepared, subcarriers may be divided as appropriate, and parallel processing may be performed serially by the plurality of circuits. These are common to all embodiments.

  Also, since all subcarriers are used for communication with the same terminal station apparatus in the OFDM modulation scheme, transmission / reception weights (average transmission / reception weight vectors and real-time transmission / reception weight matrices) at that time are common to all subcarriers. The transmission / reception weight for the combined terminal station apparatus is used. However, in OFDMA, since allocation to terminal station apparatuses of different combinations in a patchwork pattern on the time axis and the frequency axis is gathered, they are allocated for each time (OFDM symbol) and frequency (subcarrier). It is necessary to use transmission / reception weights for the terminal station apparatus. However, except for the difference, OFDM and OFDMA can be processed in exactly the same way. In this specification, the description has been focused on OFDM, but the embodiment of the present invention is also applied to OFDMA. can do.

  In addition, there are various operational variations regarding SC-FDE, but the received signal processing after the transmission weight vector is multiplied on the transmission side and the signals transmitted from the respective antenna elements are combined in space, and In any of the reception signal processing after the reception weight is multiplied on the reception side and the signals of the respective antenna elements are added and synthesized, the processing performed in the conventional SC-FDE can be applied as it is in each of the above-described configuration examples. Since it is configured, it can be applied to all variations of SC-FDE. In this case, after performing signal processing on a single carrier instead of signal processing of the OFDM modulation scheme, if it is a downlink, each subcarrier is processed by performing FFT processing on the signal on the time axis of the single carrier. Signal components are generated, these signal components are regarded as subcarrier signals generated by the OFDM modulation scheme, and the transmission weight vector generated by the embodiment of the present invention is multiplied. Similarly, in the case of uplink, a signal obtained by performing FFT processing on the received signal is handled in the same manner as in the case of the OFDM modulation method, and signal separation is performed by multiplying the reception weight vector generated by the present invention. What is necessary is just to convert into the signal of the single carrier on a time axis by performing IFFT processing with respect to the signal of a subcarrier. As described above, although some signal processing is different between the OFDM modulation scheme and SC-FDE, transmission / reception weight generation and multiplication processing are common, and the embodiment of the present invention can be applied to either of these signal schemes. Applicable.

  In addition, a plurality of series of signals are transmitted in parallel in the spatial multiplexing transmission. In the above-described embodiment, an example in which the processing such as error correction performed on these signal series is performed independently for each signal series. Of course, the signal after error correction coding is serially / parallel converted and separated into a signal sequence that is spatially multiplexed on the transmitting side, and the signal before error correction processing is performed on the receiving side. Then, parallel / serial conversion may be performed, and then error correction processing such as one system Viterbi decoding may be performed. Furthermore, any error correction processing may be performed regardless of the essence of the embodiment of the present invention including other variations. In this case, the exchange of signals between the MAC layer processing circuit and the second signal processing unit or the like is not performed in the spatial multiplexing system, but for example, information is exchanged as one system signal, and serial / A function corresponding to parallel conversion, parallel / serial conversion, error correction, and the like are included in the second signal processing unit.

  In the above explanation, the right singular value decomposition right singular vector is explained. However, the left singular vector obtained by singular value decomposition of the transposed matrix of the singular value decomposition target is the singular value decomposition of the matrix not transposed. Equivalent to the right singular vector. Similarly, even if eigenvalue decomposition is performed by multiplying the original channel vector by a Hermite conjugate vector, a right singular vector or a vector equivalent to the left singular vector can be obtained. In this sense, even when a vector mathematically equivalent to the “right singular vector” is used, it is within the range of the “right singular vector” intended by the embodiment of the present invention. Similarly, when a vector mathematically equivalent to the left singular vector is used, the range of the “left singular vector” intended by the embodiment of the present invention is also used.

  Furthermore, in the above description, only the virtual transmission line corresponding to the first singular value is used between each first signal processing unit on the base station side and the terminal station apparatus, and the singular values after the second singular value are used. However, for example, when a shared antenna for a plurality of polarizations such as V polarization and H polarization is used, each of the first signal processing units uses Since a virtual transmission line corresponding to the first singular value is used in the polarization antenna, a virtual transmission line corresponding to two singular values is used in one first signal processor in terms of form. It corresponds to being. In particular, when there is a slight leakage between polarized waves, the first signal processing unit collectively accommodates and processes all the two types of polarization antennas to suppress crosstalk components between the polarized waves. In other words, using a virtual transmission line corresponding to the first singular value and the second singular value is excellent in terms of transmission efficiency. In this case, the mathematical expression corresponds to the use of the first singular value and the second singular value, but in practice, a virtual transmission line corresponding to the first singular value is provided for each polarization antenna group. It corresponds to using. The intention of the present invention is that when a plurality of antenna groups that are expected to have a very low correlation with each other, such as when using such a polarized antenna, are used in terms of signal processing or circuit. Includes the case of using in parallel the virtual transmission path corresponding to the first singular value that effectively corresponds to each antenna group, even when used in the same first signal processing unit, A spatial multiplexing transmission utilizing a virtual transmission line corresponding to the first singular value in an effective sense is transmitted using a plurality of first signal processing units. In addition, the plurality of first signal processing units provided in the base station apparatus basically perform up-conversion and down-conversion processing between the baseband and the radio frequency using individual local oscillator signals. However, it is preferable that the signals of the respective local oscillators behave as semi-synchronously as possible. For this purpose, a reference signal such as a common reference clock may be supplied from the third signal processing unit. Alternatively, a radio signal for performing up-conversion and down-conversion between the baseband and the radio frequency by supplying an intermediate frequency reference signal from the second signal processing unit to the first signal processing unit and multiplying it, for example. It is also possible to generate a frequency signal. Depending on the value of the intermediate frequency used here and the stability of the circuit that performs the multiplication processing, the uncertainty of the complex phase of the radio signal between the plurality of first signal processing units is also limited in the transmission signal processing In this case, it is also possible to perform precoding processing between the plurality of first signal processing units in the second signal processing unit. However, basically, at least the reference signal supplied from the second signal processing unit to the plurality of first signal processing units is transmitted on the cable between the plurality of first signal processing units in the second signal processing unit. In order to reduce the loss of the local signal, the frequency is basically ½ or less of the frequency of the local signal used for the up-conversion and down-conversion processing between the baseband and the radio frequency. Independent between signal processing units. However, when the base station apparatus and the terminal station apparatus are arranged at a short distance, L in Expression (29) becomes small, and as a result, the interval between the first signal processing units can be shortened. Depending on the value of the cable loss, it is possible to directly supply the local oscillator signal from the second signal processing unit to the plurality of first signal processing units.

  In order to realize the third embodiment, the central base station not only manages communication between the satellite base station apparatus and the central base station, but also manages communication between the satellite base station apparatus and the terminal station apparatus. Will do. As described above, when a satellite base station apparatus capable of performing low-correlation communication is selected from among a plurality of satellite base station apparatuses to perform communication, the controlling base station apparatus The satellite base station apparatus to be used is selected as the scheduling process, and the selection result is instructed to the satellite base station apparatus to perform communication. Furthermore, as shown in FIG. 33, especially when communication is performed across a plurality of small cells, the distance from the satellite base station device to be used varies, and there is a gain difference in circuit design. In some cases, management such as transmission power control is required. These management functions are implemented on the central base station apparatus side, and various wireless communication controls are stably realized.

  In the description of the tenth embodiment, the transmission weight processing circuits 432, 464, and 470 in the BBU are configured to notify the transmission weight to the RRH side via the optical fibers 455, 455a, and 455b. Receive weight processing circuits 467, 480, 482 from the receive fibers to the RRH side via the optical fibers 475a, 475b, 485a, 485b, and the RRH side correlation calculation circuits 481a, 481b, 483a, 483b to the optical fibers 475a, 475b. 485a and 485b, the reception weight information or the channel information on the reception side is notified to the BBU side. Here, the information exchanged between the BBU and the RRH is not necessarily digitally represented. It doesn't have to be For example, if the beam pattern used by the RRH side at the time of transmission / reception is menud and coefficients for forming a large number of transmission / reception weights corresponding to each beam pattern are listed in the RRH, the beam pattern It is also possible to manage the number as the identifier of the code book as a code book and exchange the code book value between the BBU and the RRH. In this case, the correlation calculation circuits 481a, 481b, 483a, 483b on the RRH side compare the calculated correlation value for each antenna element with the coefficient on the code book, and the code book value having the smallest error from the coefficient. It is also possible to adopt a configuration for notifying the function of selecting and the selected codebook value.

  Further, particularly when the present invention is used in an access system, the antenna element of the terminal station apparatus and the antenna element of the base station apparatus are in a line-of-sight environment when the antenna installation position of the base station apparatus is installed in a low place. In some cases, it is required to be able to communicate stably even under conditions that are not present. In this case, the superiority of the first singular value over the second singular value is diminished, and as a result, a sufficient gain is obtained due to the effect of division loss caused by signal processing by dividing the entire antenna element into a plurality of groups. There are cases where it cannot be secured. In such a case, for example, it is better to combine all the antenna elements in one place and perform MIMO transmission normally than in the case of dividing into a plurality of first signal processing units. Is expected. When designing a system including such a case, as an example, a part (for example, one) of a plurality of first signal processing units is mounted with a larger number of antenna elements than the other first signal processing units. Depending on the situation, the special first signal processing unit performs MIMO transmission using the second and subsequent singular values and the virtual signal corresponding to the first singular value. It is also possible to operate while switching the mode when performing MIMO transmission using a general transmission path. For example, when it is determined that the number of antenna elements of this special first signal processing unit is 256 elements and the number of antenna elements of the other first signal processing units is 64, and that the line of sight with the terminal device can be generally secured. Performs four multiplexing using four virtual transmission lines corresponding to the first singular value, and when the line of sight cannot be secured, the normal MIMO communication using the 256 element antenna element of the special first signal processing unit May be performed. At this time, the special first signal processing unit is not necessarily installed in the vicinity of the plurality of other first signal processing units. For example, the special first signal processing unit is in a low place and the remaining ones The plurality of first signal processing units may be provided with a physically distinct difference such as being installed at a high place. In this case, if there is a low possibility that a special first signal processing unit installed in a low place has a view of the terminal station device, a virtual transmission line corresponding to the first singular value is used. However, only the first signal processing unit installed at a high place may be used. If these modes are switched as appropriate, communication can always be performed under optimum conditions corresponding to the environment. In addition, the apparatus corresponding to this special 1st signal processing part does not necessarily need to be accommodated in the same 2nd signal processing part. For example, in addition to a base station apparatus that performs MIMO transmission using a virtual transmission path corresponding to the first singular value shown in the embodiment of the present invention, a conventional base station apparatus that uses the second singular value and beyond is also nearby. It is also possible to adopt a configuration in which the terminal station apparatus selects the optimum base station apparatus according to the situation and performs communication while both cooperate with each other.

  In the second embodiment, the first signal processing unit or the directional antenna element on the base station apparatus side is defined on the straight line. In other embodiments, the first signal is processed. The processing units are not necessarily arranged on a straight line, and may be two-dimensionally spread. Particularly when the present invention is used in an access system, the correlation between the first signal processing units is reduced by two-dimensionally arranging the first signal processing units on the wall surface of the building. Is also possible.

  In each of the embodiments described above, the configuration based on the Massive MIMO implicit feedback has been described. However, strictly speaking, if the MAC efficiency is neglected and discussed, there is no problem with the method using explicit feedback using control information.

  Also, in the above description, the transmission / reception signal and transmission / reception weight multiplication processing has been described as being performed on the baseband, but it shows typical signal processing, and equivalent signal processing is performed on the baseband signal. It is of course possible to implement on an intermediate IF (Intermediate Frequency) signal between the RF signal and the radio frequency RF signal. In terms of signal processing, it is simpler to perform processing in a lower frequency band. However, in the fifth generation mobile communication, communication is performed in a high frequency band such as a millimeter wave band with a bandwidth of several hundred to 1 GHz. Even if processing is performed in the IF band of several GHz, the difficulty of processing does not change greatly. In the present specification, a baseband signal or a baseband band is used in a broad sense as a frequency band capable of performing digital signal processing. In this sense, an IF different from a baseband band in a narrow sense. Even in the case of signal processing in a band, the present invention is essentially applicable, and the claims of the present invention include such a broad baseband signal and baseband band.

  Further, in the present invention, the number of antenna elements included in the radio station apparatus is enormous, so even if, for example, a few antenna elements among them are subjected to exceptional signal processing, the majority of the remaining antenna elements It is possible to obtain generally expected characteristics due to the effect. However, even if some of these antenna elements are processed exceptionally, the result of signal processing using the remaining antenna elements will largely determine the characteristics. Even if the exception processing is applied, if the effect is limited without obtaining the extended effect by the exception application, it should be included in the claims of the present invention.

  You may make it implement | achieve the base station apparatus and terminal station apparatus in embodiment mentioned above with a computer. In that case, a program for realizing this function may be recorded on a computer-readable recording medium, and the program recorded on this recording medium may be read into a computer system and executed. Here, the “computer system” includes an OS and hardware such as peripheral devices. The “computer-readable recording medium” refers to a storage device such as a flexible medium, a magneto-optical disk, a portable medium such as a ROM and a CD-ROM, and a hard disk incorporated in a computer system. Furthermore, a “computer-readable recording medium” dynamically holds a program for a short time, like a communication line when transmitting a program via a network such as the Internet or a communication line such as a telephone line. In this case, a volatile memory in a computer system serving as a server or a client in that case may be included, and a program that holds a program for a certain time. Further, the program may be for realizing a part of the functions described above, and may be a program capable of realizing the functions described above in combination with a program already recorded in the computer system. It may be realized using hardware such as PLD (Programmable Logic Device) or FPGA (Field Programmable Gate Array).

  As mentioned above, although embodiment of this invention has been described with reference to drawings, the said embodiment is only the illustration of this invention, and it is clear that this invention is not limited to the said embodiment. is there. Therefore, additions, omissions, substitutions, and other modifications of the components may be made without departing from the technical idea and scope of the present invention.

1 ... base station device,
2 ... Radio station equipment,
3 ... Prospect wave,
4 ... Stable reflected wave by structure,
5 ... Multiple reflected waves near the ground,
6 ... Multiple reflected waves near the ground,
7 ... structure,
11: Transmitting station,
12 ... receiving station,
21 ... High power amplifier (HPA),
22 ... Low noise amplifier (LNA),
23 Time-division switch (TDD-SW),
24 ... Antenna element 25 ... Wireless module,
26: Antenna terminal,
27 ... Coaxial cable,
31 ... Overall base station device,
32 ... Satellite base station equipment,
33 ... terminal station device,
34 ... Overall base station device,
35 ... Satellite base station device,
36. Terminal station device,
37 ... structure,
38 ... structure,
39 ... Cell,
40. Wireless communication system,
41 ... Base station device,
50. Wireless communication system,
51. Terminal station device,
52 ... Wireless communication system,
53. Wireless communication system,
60 ... a terminal station device,
61 ... Transmitter,
65 ... receiving part,
67. Interface circuit,
68 ... MAC layer processing circuit,
70 ... base station device,
71 ... a second transmission signal processing unit,
75 ... a second received signal processing unit,
77. Interface circuit,
78 ... MAC layer processing circuit,
80 ... base station device,
81... Transmitter
85 ... receiving part,
87: Interface circuit,
88 ... MAC layer processing circuit,
90 ... wireless communication system,
111... First transmission signal processing circuit,
113 ... Transmission signal processing circuit,
114... Received signal processing circuit,
120. Communication control circuit,
121. Communication control circuit,
130: a first transmission weight processing unit;
131... First channel information acquisition circuit,
132. First channel information storage circuit,
133... First transmission weight calculation circuit,
140... First transmission weight processing unit,
141 ... Channel information acquisition circuit,
142 ... Channel information storage circuit,
143 ... first transmission weight calculation circuit,
144: First reception weight processing unit,
145 ... Received signal processing circuit,
146 ... first channel information estimation circuit,
147. First reception weight calculation circuit,
148 ... second transmission signal processing circuit,
149 ... transmission weight calculation circuit,
150 ... transmission weight processing unit,
151 ... Channel information acquisition circuit,
152 ... Channel information storage circuit,
153 ... Multi-user MIMO transmission weight calculation circuit,
154: a first reception weight processing unit,
155. First received signal processing circuit,
156 ... first channel information estimation circuit,
157 ... First reception weight calculation circuit,
158... First received signal processing circuit,
159 ... second received signal processing circuit,
160... First reception weight processing unit,
161. First channel information estimation circuit,
162... First reception weight calculation circuit,
170 ... reception weight processing unit,
172 ... Channel information estimation circuit,
173 ... Multi-user MIMO reception weight calculation circuit,
181 ... first transmission signal processing unit,
182: a first transmission signal processing unit,
185: a first received signal processing unit,
186: first received signal processing unit,
190 ... second received signal processing circuit,
191 ... Channel matrix acquisition circuit,
192: Receive weight matrix calculation circuit,
193: Reception weight matrix multiplication circuit,
194: Signal detection circuit,
201 ... Virtual antenna element,
202 ... Virtual antenna element,
203 ... Virtual antenna element,
204 ... Base station apparatus,
205 ... Antenna element group,
206 ... Terminal station device,
221-1 to 5 ... Antenna element,
231, 232, 233, 234 ... base station apparatus,
235, 236, 237, 238, 239 ... terminal station devices,
241, 242, 243, 244 ... subcarriers,
251-1-3, 253-1-3, 254-1-3, 255-1-3, 256-1-3.
252-1 to 3 ... Data communication slots,
257-1 to 3 ... Training signal,
258-1 to 3 ... Data payload,
291-295 ... plot points,
301 ... Base station apparatus,
302 ... Terminal station device,
303 ... Base station device,
304: first signal processing unit;
305 ... Second signal processing unit,
306 ... Base station device,
307 ... Parabolic antenna,
308 ... Terminal station device,
309 ... a terminal station device,
310 ... base station apparatus antenna,
311: a first transmission signal processing circuit;
312 ... Terminal station apparatus antenna element group,
320 ... Terminal station antenna element group,
330 ... first transmission weight processing unit,
331: a first transmission signal processing circuit,
332: a first channel information acquisition circuit,
333 ... First channel information storage circuit,
334... First transmission weight calculation circuit,
340 ... first transmission weight processing unit,
341 ... first channel information acquisition circuit,
342 ... First channel information storage circuit,
343 ... first transmission weight calculation circuit,
348 ... second transmission signal processing circuit,
354 ... First reception weight processing unit,
355 ... First received signal processing circuit,
356. First channel information estimation circuit,
357. First reception weight calculation circuit,
358. First received signal processing circuit,
359 ... Second received signal processing circuit,
360. First reception weight processing unit,
361 ... transmission unit,
362 ... first channel information estimation circuit,
363: a first reception weight calculation circuit,
365: receiving unit,
381... First transmission signal processing unit,
385 ... First received signal processing unit,
401 ... MAC layer processing circuit,
403, 403a, 403b ... time axis signal generation circuit,
412 ... transmission signal processing circuit,
427, 427a, 427b ... D / A converter,
428, 428a, 428b ... RF processing circuit,
429, 429a, 429b ... antenna elements,
432 ... Transmission weight processing circuit,
441-4, 441-5, 441-6 ... BBU,
442-4, 442-4a, 442-4b, 442-6 ... RRH,
443-1, 443-2, 443-3 ... BBU,
444-1a, 444-1b, 444-2a, 444-2b, 444-3... RRH,
446-1, 446-2 ... area,
454, 454a, 454b ... optical interface circuit,
455, 455a, 455b ... optical fiber,
456, 456a, 456b, optical interface circuits 457, 457a, 457b, time axis transmission weight multiplication circuits,
461 ... MAC layer processing circuit,
462... Transmission signal processing circuit,
463 ... transmission signal processing circuit,
464: Transmission weight processing circuit,
465... Addition synthesis circuit,
466... Received signal processing circuit,
467 ... Receive weight processing circuit,
470: Transmission weight processing circuit,
471 ... MAC layer processing circuit,
472 ... received signal processing circuit,
473a, 473b ... time axis reception weight multiplication circuit,
474a, 474b ... Optical interface circuit,
475a, 475b ... optical fiber,
476a, 476b ... Optical interface circuit,
477a, 477b ... A / D converter,
478a, 478b ... RF processing circuit,
479a, 479b ... antenna elements,
480 ... Receive weight processing circuit,
481, 481a, 481b ... correlation calculation circuit,
482 ... Receive weight processing circuit,
483a, 483b ... time axis reception weight multiplication circuit,
484a, 484b ... Optical interface circuit,
485a, 485b ... optical fiber,
486a, 486b ... Optical interface circuit,
491-2, 491-3, 491 -N _Ant ... multiplier,
492-2, 492-3, 492-N _Ant ... adder,
493-2, 493-3, 493-N _Ant ... memory,
494 ... correlation calculation control unit,
495-1, 495-2, 495-3, 495-N _Ant ... a multiplier,
496-1, 496-2, 496-3, 496-N _Ant ... adder,
497-1, 497-2, 497-3, 497-N _Ant ... memory,
498-1, 498-2, 498-3, 498-N _Ant ... square root acquisition circuit,
600 ... base station apparatus,
601, 601-1, 601-2, 601-3, 601-4, 601-5, 601-6, 601-7, 601-8 ... the first signal processing unit,
602-1, 602-2, 602-3, 602-4, 602-5, 602-6, 602-7, 602-8 ... geometric coordinate information,
603 ... Train,
604 ... a terminal station device,
605 ... Camera,
606: Weight matrix / coordinate database,
607: Weight vector / coordinate database,
612 ... Coordinate information acquisition circuit,
613 ... Time information acquisition circuit,
614 ... Communication control circuit,
622 ... coordinate information acquisition circuit,
623 ... time information acquisition circuit,
624 ... Communication control circuit,
700 ... antenna element,
701 ... TDD-SW,
702 ... Low noise amplifier (LNA),
703 ... a mixer,
704 ... filter,
705 ... A / D converter,
706 ... FFT circuit,
707 ... reception weight multiplication circuit,
708 ... transmission weight multiplication circuit,
709 ... IFFT & GI giving circuit,
710 ... D / A converter,
711 ... a mixer,
712 ... filter,
713 ... High power amplifier,
714 ... TDD-SW,
721 ... an antenna element,
722 ... Low noise amplifier (LNA),
723 ... a mixer,
724 ... Filter,
725 ... A / D converter,
726 ... FFT circuit,
727 ... Receive weight multiplication circuit,
728 ... Transmission weight multiplication circuit,
729 ... IFFT & GI giving circuit,
730 ... D / A converter,
731 ... a mixer,
732 ... Filter,
733 ... High power amplifier,
740 ... transmission weight processing unit,
741 ... Local oscillator 742 ... Local oscillator 743 ... Communication control circuit,
781... Scheduling processing circuit,
811 ... transmission signal processing circuit,
812 ... Addition synthesis circuit,
813: IFFT & GI adding circuit,
814 ... D / A converter,
815 ... a local oscillator,
816 ... Mixer,
817: Filter,
818 ... High power amplifier,
819: Antenna element,
820 ... a communication control circuit,
821... First transmission signal processing circuit,
830 ... transmission weight processing unit,
831 ... Channel information acquisition circuit,
832 ... Channel information storage circuit,
833 ... Multi-user MIMO transmission weight calculation circuit,
851 ... Antenna element,
852 ... Low noise amplifier (LNA),
853: Local oscillator,
854 ... Mixer,
855 ... filter,
856 ... A / D converter,
857 ... FFT circuit,
858 ... Received signal processing circuit,
860 ... reception weight processing unit,
861... Channel information estimation circuit,
862 ... Multi-user MIMO reception weight calculation circuit,
881... Scheduling processing circuit

Claims (4)

Α = dWsin θ / c α, where the distance d [m] between the nearest antenna elements, the assumed maximum signal arrival direction θ [rad], the bandwidth W [Hz], and the speed of light c [m / s] A wireless communication device that operates under the condition that the value of is 1 or less,
Channel estimation means for receiving a training signal transmitted from all subcarriers within a communication band limited to some subcarriers, performing channel estimation of the subcarrier based on the received training signal, and acquiring channel information; ,
A regression line is obtained by applying a complex phase folding of 2π period to the complex phase of the relative channel information based on the complex phase of the channel information or the complex phase of the reference antenna and the frequency of the subcarrier, and the regression line A complex phase calculation means for calculating a complex phase of the channel information or relative channel information of each subcarrier based on:
A radio communication apparatus comprising: reception weight calculation means for calculating a reception weight based on the complex phase of the channel information or the relative channel information calculated by the complex phase calculation means. The complex phase calculation means includes
The regression line is obtained by a least square method, and in the least square method, 0 and ± as complex phase offsets with respect to the complex phase of the channel information or the relative channel information in the subcarrier directly subjected to the channel estimation. Assuming a value of 2π and adding the offset, the sub-carrier of each subcarrier obtained based on the information of the regression line obtained as a result of searching for a state where the cumulative value of the square error of the least square method is minimized The wireless communication apparatus according to claim 1, wherein the complex phase of the channel information or the relative channel information is calculated. The complex phase calculation means includes
When the antenna elements are configured in a linear array and used, a constraint condition is set so that the slope of the regression line of the least squares method is proportional to the array order of the antenna array. The radio communication apparatus according to claim 1, wherein a least square method over elements is applied in a lump. Α = dW sin θ / c α, where the element spacing d [m] of the nearest antenna, the assumed maximum signal arrival direction θ [rad], the bandwidth W [Hz], and the speed of light c [m / s] A wireless communication method performed by a wireless communication device that operates under the condition that the value of 1 is 1 or less,
A channel estimation step of receiving a training signal transmitted from all subcarriers within a communication band limited to some subcarriers, performing channel estimation of the subcarrier based on the received training signal, and acquiring channel information; ,
A regression line is obtained by applying a complex phase folding of 2π period to the complex phase of the relative channel information based on the complex phase of the channel information or the complex phase of the reference antenna and the frequency of the subcarrier, and the regression line A complex phase calculation step of calculating a complex phase of the channel information or relative channel information of each subcarrier based on:
A radio communication method comprising: a reception weight calculation step of calculating a reception weight based on the complex phase of the channel information or the relative channel information calculated in the complex phase calculation step.