JP2015012706A - Driving circuit for transistor, semiconductor breaker using the same, and method for controlling cut-off of the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a driving circuit for a transistor, a semiconductor breaker and a method for controlling cut-off of the same which suppress an overcurrent generated when a transistor is cut off to a fixed set electromotive voltage, cut off the overcurrent at high speed, and can also cope with variations in characteristics of an element.SOLUTION: Current-voltage characteristics of a semiconductor switch are determined, and a first gate control voltage V1 at the time of cut off is determined from an overcurrent value Iov and an electromotive voltage upper limit value Vdmax or a power supply voltage value Vdd from a current-voltage characteristic diagram. A second gate control voltage V2 is determined from a current value of 0 and the power supply voltage Vdd or the electromotive voltage upper limit value Vdmax from the current-voltage characteristic diagram. A cut-off time Tf during which the voltage is lowered from the gate control voltage V1 to the gate control voltage V2 is determined by calculation from a circuit inductance L, the electromotive voltage upper limit value Vdmax, the power supply voltage Vdd, and an overcurrent threshold Iov.

Description

  The present invention relates to a transistor drive circuit, a semiconductor circuit breaker using the same, and a circuit breaker control method thereof. More specifically, the present invention limits overcurrent in a short time and occurs when a transistor is cut off without using a snubber circuit. The present invention relates to a transistor drive circuit, a semiconductor circuit breaker, and a circuit breakage control method that can suppress an overvoltage to a set value or less.

  In recent years, development has been made to supply power to a load by DC power supply from the power source side. Although it is an attempt to reduce the number of AC-DC conversion and DC-AC conversion and eliminate the loss in the conversion, the circuit in case of overcurrent due to short circuit accident, ground fault, etc. on the load side in DC power supply Therefore, a different blocking method from that for AC is required. A method using an insulated gate field effect transistor (MOSFET), a bipolar transistor, or the like as a semiconductor serving as a shut-off element has been studied. However, if a circuit is shut off and opened in a state where a direct current flows, circuit inductance on the circuit As a result, there are problems such as generation of high electromotive voltage and oscillation at the drain and collector of the semiconductor element, destruction of the element, and long time for convergence of oscillation.

  In addition, it is necessary to consume energy stored in the circuit inductance somewhere. Usually, various types of snubber circuits are used. When a large current is handled, the elements used for the snubber circuits are enlarged. Other breaker devices include fuses and breaker breakers (MCCB), but the break time during a short circuit is as long as milliseconds. In a semiconductor circuit breaker, this speed is expected to be several tens of microseconds. Further, the fuse needs to be replaced in terms of reusability, and the breaker (MCCB) also has a problem of deterioration of contact resistance due to repeated use. The semiconductor breaker is also excellent in this repeated use.

  A snubber circuit is a protection circuit that absorbs a transient high voltage that occurs in an electric circuit when a switch is cut off. When a circuit current is suddenly cut off, the voltage suddenly rises due to circuit inductance. However, the snubber circuit suppresses this spike-like high voltage, thereby preventing damage to the switch itself and surrounding electronic parts and minimizing electromagnetic noise.

For example, Patent Document 1 discloses a semiconductor circuit breaker having a snubber-less configuration that uses a semiconductor switch and does not use a snubber circuit. An insulated gate field effect transistor (MOSFET) is used as the semiconductor switch.
FIG. 14 is a block configuration diagram of the DC current interrupting device described in Patent Document 1. The conventional DC current interrupting device is provided with an RC snubber circuit in which a resistor R and a capacitor C are connected in parallel with the switch unit 12 (see FIG. 12 of Patent Document 1). In the DC current interrupting device 10 described in Patent Document 1, when an overcurrent flows on the power supply line 17 due to a short circuit accident, the current detection unit 11 detects the overcurrent, and the overcurrent determination unit 14 of the control circuit 13 detects the occurrence of an overcurrent. The gate control voltage for switching the switch from the on state to the off state is supplied from the switch driving unit 15 to the gate of the transistor of the switch unit 12. Although the gate control voltage is gradually and gradually lowered, various gate voltage drop waveform patterns set in advance by the arbitrary waveform generator 16 are used.

For example, in patent document 2, the electrostatic induction transistor (SIT) is used as a semiconductor switch. The static induction transistor is a kind of junction field effect transistor (JFET), and controls the depletion layer width of the pn junction by the gate voltage to turn on / off the conductive channel.
FIG. 15 is a block configuration diagram of the semiconductor interrupting device described in Patent Document 2. The operation is similar to that of Patent Document 1 described above. In the semiconductor circuit breaker 21, the current value is accumulated in the storage unit 54 by the current sensor 32 and the current measuring unit 33, and the control unit 52 determines the occurrence of overcurrent based on this information. Further, the control unit 52 determines control of the gate signal of the semiconductor unit 31 from the current value information and the waveform control parameter set in the setting storage unit 53 from the outside via the input interface 35. Reference numeral 22 denotes a power supply device, 23 denotes a load device, 34 denotes a transistor drive circuit, 41 and 42 denote DC output lines, 43 and 44 denote DC input lines, and 51 denotes a drive unit.

FIG. 16 is a circuit model diagram for explaining the shut-off operation of the semiconductor shut-off device in FIG. FIG. 17 is a diagram showing temporal changes in voltage and current of each part during the shut-off operation in FIG. An overcurrent is detected at time t2, and the subsequent control waveform of the gate control voltage, the drain current of the circuit, the drain-source voltage, and the like are shown.
A gate voltage at which the transistor is turned on is a voltage V1, and a gate voltage at which the transistor is turned off is a voltage V2, which are 0V and -20V, respectively. After detecting the overcurrent, the gate voltage Vgs is lowered to the voltage V5 at time t3, is raised to the voltage V3 after a certain time, and is lowered to the voltage V4 over the time T1. The off voltage V2 is set at time t6. During this time, the drain voltage Vd rises rapidly and gradually converges to the power supply voltage.

In Patent Document 2 described above, since an electrostatic induction transistor is used as a semiconductor switch, unlike the insulated gate field effect transistor (MOSFET), the control voltage width of the gate voltage Vgs for controlling the on-resistance value of the switch. Can be taken widely and has good blocking characteristics, and the blocking is completed in several tens of microseconds.
FIGS. 18A and 18B are explanatory diagrams regarding the determination method of the voltages Va and Vb described in Non-Patent Document 1, and FIG. 19 shows a plurality of samples when the determination method of the voltages Va and Vb is used. It is the figure which showed the result by in a table | surface.

  The voltage V3 = Va is the gate voltage between the on state and the intermediate state of the transistor, and the voltage V4 = Vb is the voltage between the intermediate state and the off voltage. In order to determine the voltage V3 and the voltage V4, a triangular wave gate waveform is applied to the transistor of FIG. 16, and the voltage at which the current starts flowing from the off state is V4. The gate voltage is lowered from the on state and the current starts to decrease. It is proposed to determine the voltage as V3. As described above, FIG. 19 shows the peak voltage value (over voltage) of the drain voltage Vd when driven by the voltage V3 and the voltage V4 of this determination for a plurality of samples. It was stated that overvoltage could be suppressed.

JP 2008-67440 A JP 2010-220325 A

2013 IEEJ National Convention Paper No. 4-138 “Examination of SiC-SIT Gate Voltage Setting Method for DC Power Supply System Circuit Breakers” (issued March 5, 2013)

As described above, in Patent Document 1, an insulated gate field effect transistor (MOSFET) is used as a semiconductor switch. In this insulated gate field effect transistor (MOSFET), the on / off state is rapidly switched around the gate voltage Vgs around the threshold value Vth, and the on-resistance value of the transistor transitions between a small state and a very large value.
In order to make the interruption circuit of this patent document 1 snubberless, the current is consumed by the conduction resistance when this switch transits from the on-state to the off-state. For this purpose, the energy stored in the circuit inductance is used. Accordingly, the consumption time must be sufficiently increased, and the gate control voltage needs to be slowly and gradually lowered. Further, Patent Document 1 considers a gate control time on the order of milliseconds, and has a cutoff response time equivalent to that of a conventional fuse or cutoff breaker (MCCB), and is expected to be several tens of microseconds as a semiconductor breaker. Response is difficult.

  Therefore, there is a possibility that the response time can be shortened if the gate voltage Vgs is kept accurately in the vicinity of the threshold value Vth and the voltage is gradually lowered. However, as is well known in semiconductor transistors, the threshold value Vth varies from element to element. Even if the gate voltage Vgs is set to be constant, the characteristics are greatly changed for each element. For this reason, it is necessary to take measures against threshold variation so that the characteristics are stable even if the element changes.

Further, in Patent Document 2 described above, it is described that a high-speed cutoff characteristic is realized by gate voltage control using the normally-on characteristic of the electrostatic induction transistor. In the gate voltage control described above, the gate voltage V3 is described. No information is disclosed on how to determine the gate voltage V4, gate voltage V5, T1 time, etc. in relation to the element characteristics.
That is, when the gate voltage V3 is close to the off potential, the transistor is turned off rapidly, so that the drain voltage rises rapidly and a high electromotive voltage is generated, which may destroy the element.

When the gate voltage V3 is close to the on-potential, the current value decreases little and the drain voltage may be lower than the power supply voltage.
The allowable increase value of the drain voltage depends on the breakdown voltage of the transistor and the characteristics of the circuit system to which the transistor is connected, and is specified systematically. This defined upper limit electromotive voltage value is defined as an electromotive voltage upper limit value. Further, the overcurrent value when the overcurrent is detected to enter the shut-off operation is regulated systematically because if it becomes too large, the transistor is destroyed and further the circuit system to which the transistor is connected is affected. This defined upper limit overcurrent value is defined as an overcurrent threshold.

Non-Patent Document 1 proposes a method for setting voltage V3 and voltage V4 that was not disclosed in Patent Document 2 described above, and it is said that the electromotive voltage can be suppressed when driven by the setting method. This is not a gate voltage setting method for controlling the electromotive voltage below a certain set voltage. Also, a method for setting the cutoff time is not disclosed.
The present invention has been made in view of such a problem, and an object of the present invention is to suppress an overvoltage generated when a transistor is shut down to a predetermined set electromotive voltage or less, and to shut off the overcurrent at high speed. It is an object of the present invention to provide a transistor drive circuit, a semiconductor circuit breaker, and a circuit breakage control method thereof that can cope with variations in characteristics of the transistor.

  The present invention has been made to achieve such an object. The invention according to claim 1 is directed to a current detector (67) for detecting a current on the source side of a transistor (66), and the transistor ( 66), a control voltage storage unit (70) in which the first gate control voltage value (V1) and the second gate control voltage value (V2) derived based on the current-voltage characteristics are stored, and the transistor ( 66), a reference transition time value (Tf), which is a time until the gate voltage value (Vgs) of (66) is changed from the first gate control voltage value (V1) to the second gate control voltage value (V2), is stored. A transition time storage unit (71), an output of the current detection unit (67), the first gate control voltage value (V1), the second gate control voltage value (V2), and the reference transition time value ( Tf) and It is a drive circuit of the transistor, characterized in that a Njisuta (66) gate control unit (69) for controlling the operation of the. (FIGS. 1 to 3, FIG. 5; Examples 1, 3, and 5)

  According to a second aspect of the present invention, in the first aspect of the present invention, the first gate control voltage value (V1) of the control voltage storage unit (70) is equal to the current − of the transistor (66). In relation to the voltage characteristics, the drain voltage (Vds) is a direct-current power supply voltage (Vdd) and the drain current (Id) is a first gate voltage value (V1a) determined by an overcurrent threshold (Iov). The drain voltage (Vds) is an electromotive voltage upper limit (Vdmax) and the drain current (Id) is an overcurrent threshold (Iov), which is determined by the second gate voltage value (V1b), The gate voltage value (Vgs) is greater than a gate voltage value (V1b) of 2 and less than or equal to the first gate voltage value (V1a). (Fig. 7)

  According to a third aspect of the present invention, in the first or second aspect of the present invention, the second gate control voltage value (V2) of the control voltage storage unit (70) is equal to that of the transistor (66). In relation to the current-voltage characteristics, the drain voltage (Vds) is the power supply voltage (Vdd) and the drain current (Id) is determined to be a third gate voltage value (V2b), The drain voltage (Vds) is an electromotive voltage upper limit value (Vdmax) and the drain current (Id) is zero, or the fourth gate voltage value (V2a) or the fourth gate voltage value (V2a). ), And any one of the gate voltage values (Vgs) which is not more than the third gate voltage value (V2b). (Fig. 7)

According to a fourth aspect of the present invention, in the first, second, or third aspect of the invention, the transistor (66) includes a DC power source (92) and a circuit inductance (L) on the drain side, and A transition time calculation unit (73) that calculates the reference transition time value (Tf) from the circuit inductance (L), the overcurrent threshold value (Iov), the electromotive voltage upper limit value (Vdmax), and the DC power supply voltage (Vdd). ). (Fig. 2, Fig. 5)
In the invention described in claim 5, in the invention described in claim 4, the reference transition time value Tf is a value calculated by a relational expression of Tf = L × Iov / (Vdmax−Vdd). It is characterized by.

The invention according to claim 6 is the invention according to any one of claims 1 to 5, wherein the gate controller (69) sets the gate voltage value (Vgs) of the transistor (66) to the first value. A change time for stepping down from one gate control voltage value (V1) to the second gate control voltage value (V2) is determined based on the reference transition time (Tf).
According to a seventh aspect of the present invention, in the sixth aspect of the present invention, the gate control unit (69) sets the gate voltage value (Vgs) of the transistor (66) to the first gate control voltage value. It is characterized in that the change time for stepping down from (V1) to the second gate control voltage value (V2) is controlled to be longer than the reference transition time (Tf).

The invention according to claim 8 is the invention according to claim 6 or 7, wherein the gate controller (69) sets the gate voltage value (Vgs) of the transistor (66) to the first gate control. The voltage is stepped down linearly from the voltage value (V1) to the second gate control voltage value (V2) along the elapsed time.
The invention according to claim 9 is the invention according to claim 6 or 7, wherein the gate controller (69) sets the gate voltage value (Vgs) of the transistor (66) to the first gate control. When the voltage is lowered from the voltage value (V1) to the second gate control voltage value (V2), the drain voltage value (Vds) is controlled to coincide with the electromotive voltage upper limit value (Vdmax). .

  The invention according to claim 10 is the invention according to claim 9, further comprising a voltage detection unit (74) for detecting the drain voltage value (Vds), wherein the gate control unit (69) includes the transistor. When the gate voltage value (Vgs) of (66) is stepped down from the first gate control voltage value (V1) to the second gate control voltage value (V2), the drain voltage value (Vds) is changed to the electromotive voltage. Control is performed so as to coincide with the upper limit value (Vdmax). (FIG. 3; Example 5)

The invention according to claim 11 is the invention according to claim 6 or 7, wherein the gate control unit (69) is connected to the control voltage storage unit (70) and the transition time storage unit (71). And a gate voltage correction unit (82) for correcting the gate signal of the transistor (66) based on the gate signal output from the linear gate voltage generation unit (81). It is characterized by having. (Fig. 4)
According to a twelfth aspect of the invention, in the invention of the sixth or seventh aspect, the gate controller (69) sets the gate voltage value (Vgs) of the transistor (66) to the first gate control. When the voltage is lowered from the voltage value (V1) to the second gate control voltage value (V2), the drain current (Ids) is controlled to decrease linearly.

  According to a thirteenth aspect of the present invention, in the twelfth aspect of the present invention, the gate control unit (69) sets the gate voltage value (Vgs) of the transistor (66) to the first gate control voltage value. When the voltage is stepped down from (V1) to the second gate control voltage value (V2), the drain current (Ids) is controlled to decrease linearly using the current value of the current detection unit (67). Features.

  The invention according to claim 14 is the invention according to claim 6 or 7, wherein the gate controller (69) sets the gate voltage value (Vgs) of the transistor (66) to the first gate control. When the voltage value (V1) is stepped down to the second gate control voltage value (V2), the gate voltage value (Vgs) when the gate voltage value (Vgs) is stepped down linearly with time, and the drain voltage value (Vds) is controlled so as to take a gate voltage value (Vgs) in an intermediate region from the gate voltage value (Vgs) when the control is performed so as to match the electromotive voltage upper limit value (Vdmax). .

The invention according to claim 15 is the invention according to any one of claims 1 to 14, wherein the transistor (66) is a junction field effect transistor, and the junction field effect transistor is silicon, silicon Carbide or gallium nitride is formed as a semiconductor material.
The invention according to claim 16 is the invention according to claim 15, characterized in that the junction field effect transistor is a normally-on junction field effect transistor.

According to a seventeenth aspect of the present invention, there is provided a transistor (66) provided between the DC power source (62) and the load device (63), and a transistor drive circuit according to any one of the first to sixteenth aspects. It is a semiconductor circuit breaker characterized by including. (FIGS. 1 to 3; Examples 2, 4, and 6)
According to an eighteenth aspect of the present invention, there is provided a transistor drive circuit according to any one of the first to sixteenth aspects, a load device (63) connected to a power supply line from a direct current power source (62), and the direct current. In the circuit breaker control method for the semiconductor circuit breaker (61) provided between the power supply (62), the semiconductor circuit breaker (61) detects a current flowing through the power supply line (67), and A semiconductor switch (66) for controlling a current flowing through the feeder line, a control voltage storage unit (70) for a gate control voltage value related to a current-voltage characteristic of the semiconductor switch (66), and the semiconductor switch (66). A transition time storage unit (71) that stores a reference transition time value (Tf) between two gate control voltages (V1, V2), an output of the current detection unit (67), and the gate control voltage values (V1, V2) ) And the group A gate control unit (69) for determining the driving state of the semiconductor switch (66) from the transition time value (Tf), and the first gate control voltage value (V1) of the control voltage storage unit (70) is In relation to the current-voltage characteristics of the semiconductor switch (66), the drain voltage value (Vds) is a DC power supply voltage value (Vdd), and the drain current value (Ids) is an overcurrent threshold value flowing through the feeder line. The first gate voltage value (V1a) or the drain voltage value (Vds) is an electromotive voltage upper limit value (Vdmax), and the drain current value (Ids) is an overcurrent threshold value (Iov) flowing through the feeder line. The gate voltage value (Vgs) that is greater than the second gate voltage value (V1b) that is greater than the second gate voltage value (V1b) and less than or equal to the first gate voltage value (V1a). In addition, the second gate control voltage value (V2) of the control voltage storage unit (70) is related to the current-voltage characteristics of the semiconductor switch (66), and the drain voltage value (Vds) is the direct current. The third gate voltage value (V2b) determined by the drain current value (Ids) being zero at the power supply voltage (Vdd), or the drain voltage value (Vds) is the electromotive voltage upper limit value (Vdmax). Thus, the fourth gate voltage value (V2a) determined by the drain current (Ids) being zero or greater than the fourth gate voltage value (V2a) and the third gate voltage value (V2b). ) Any one of the following gate voltage values (Vgs), and the reference transition time value (Tf) of the control voltage storage unit (70) is set between the DC power supply voltage value (Vdd) and the semiconductor switch (66). The reference transition time from the circuit inductance (L), the overcurrent threshold (Iov) flowing through the feeder line, the electromotive voltage upper limit (Vdmax) of the drain voltage (Vds), and the DC power supply voltage (Vdd). After calculating the value (Tf) and detecting that the current flowing through the feeder line exceeds the overcurrent threshold (Iov) by the current detection unit (67), the gate voltage value (66) of the semiconductor switch (66) Vgs) is set to the first gate control voltage value (V1), and the voltage is stepped down to the second gate control voltage value (V2) based on the reference transition time value (Tf). (FIGS. 1 to 3, FIG. 5, FIG. 7; Examples 2, 4, 6)

Further, in the invention described in claim 19 (corresponding to claim 5), in the invention described in claim 18, the reference transition time value Tf is calculated by Tf = L × Iov / (Vdmax−Vdd). Features.
The invention according to claim 20 (corresponding to claim 7) is the gate voltage value (Vgs) of the semiconductor switch (66) according to the invention according to claim 18 or 19, wherein the gate controller (69) ) Is decreased from the first gate control voltage value (V1) to the second control voltage value (V2) so as to be longer than the reference transition time value (Tf). .

The invention according to claim 21 (corresponding to claim 8) is the gate voltage value of the semiconductor switch (66) according to the invention according to claim 18, 19 or 20, wherein the gate controller (69) (Vgs) is stepped down linearly along the elapsed time from the first gate control voltage value (V1) to the second gate control voltage value (V2).
The invention according to claim 22 (corresponding to claim 9) is the gate voltage value of the semiconductor switch (66) according to the invention according to claim 18, 19 or 20, wherein the gate controller (69) When dropping (Vgs) from the first gate control voltage value (V1) to the second gate control voltage value (V2), the drain voltage value (Vds) matches the electromotive voltage upper limit value (Vdmax). It controls to do.

  The invention according to claim 23 (corresponding to claim 10) is the invention according to claim 18, 19 or 20, further comprising a voltage detector (74) for detecting the drain voltage value (Vds), When the gate controller (69) steps down the gate voltage value (Vgs) of the semiconductor switch (66) from the first gate control voltage value (V1) to the second gate control voltage value (V2). The drain voltage value (Vds) is controlled to coincide with the electromotive voltage upper limit value (Vdmax).

The invention according to claim 24 (corresponding to claim 11) is the invention according to claim 18, 19 or 20, wherein the gate controller (69) is connected to the control voltage memory (70) and A linear gate voltage generation unit (81) connected to the transition time storage unit (71); and a gate voltage correction unit (82) connected to the linear gate voltage generation unit (81), the gate voltage correction unit (82), the gate signal of the transistor (66) is corrected based on the gate signal output from the linear gate voltage generator (81). (Fig. 4)
The invention according to claim 25 (corresponding to claim 12) is the gate voltage value of the semiconductor switch (66) according to the invention according to claim 18, 19 or 20, wherein the gate controller (69) When the voltage (Vgs) is lowered from the first gate control voltage value (V1) to the second gate control voltage value (V2), the drain current value (Ids) is controlled to decrease linearly. Features.

  The invention according to claim 26 (corresponding to claim 13) is the gate voltage value of the semiconductor switch (66) according to the invention according to claim 18, 19 or 20, wherein the gate controller (69) When the voltage Vgs is lowered from the first gate control voltage value (V1) to the second gate control voltage value (V2), the drain current value is detected using the current detection value of the current detection unit (67). (Ids) is controlled so as to decrease linearly.

  The invention according to claim 27 (corresponding to claim 14) is the gate voltage value of the semiconductor switch (66) according to the invention according to claim 18, 19 or 20, wherein the gate controller (69) When the gate voltage value (Vgs) is stepped down linearly along the elapsed time when (Vgs) is stepped down from the first gate control voltage value (V1) to the second gate control voltage value (V2) The gate voltage value (Vgs) in the intermediate region between the gate voltage value (Vgs) and the gate voltage value (Vgs) when the drain voltage value (Vds) is controlled to match the electromotive voltage upper limit value (Vdmax). Vgs) is controlled.

ADVANTAGE OF THE INVENTION According to this invention, the drive circuit of the transistor which can set the interruption | blocking time used as a favorable interruption | blocking operation | movement corresponding to the set electromotive voltage upper limit and overcurrent threshold value is realizable.
In addition, the semiconductor breaker and the break control method of the present invention can suppress the overvoltage generated in the DC breaker below the upper limit of the electromotive voltage of the system specification, and can be cut off at an optimum time corresponding to the overcurrent threshold. It becomes possible. In addition, the above-described gate control can be performed in consideration of the circuit inductance due to the circuit breaker installation environment and the variation in characteristics of the semiconductor switch elements.

1 is a block configuration diagram for explaining a transistor drive circuit and a semiconductor circuit breaker using the transistor drive circuit according to the present invention. FIG. 1 is a block configuration diagram for explaining a transistor drive circuit in which a transition time calculation unit is added to Embodiment 1 and a semiconductor circuit breaker using the transistor drive circuit. FIG. FIG. 2 is a block configuration diagram for explaining a transistor drive circuit in which a voltage detection unit is added to the first embodiment and a semiconductor circuit breaker using the transistor drive circuit. It is explanatory drawing of the interruption | blocking operation | movement using the drain voltage of the semiconductor circuit breaker based on the present Example 6 shown in FIG. (A), (b) is the interruption | blocking circuit for demonstrating operation | movement of a semiconductor switch, and its gate waveform diagram. (A), (b) is a figure explaining the IV characteristic of a semiconductor switch. It is a figure which shows the example which calculated | required the IV characteristic about the element of the element characteristic 2. FIG. (A), (b) is a figure which shows the example of a measurement of the relationship between the gate voltage V1 and the drain electromotive voltage at the time of interruption | blocking. (A) And (b) is a figure for demonstrating a gate control voltage and interruption | blocking time. It is a figure which shows the IV characteristic in the case of the element characteristic 2. FIG. (A) thru | or (c) is a figure which shows the current waveform and voltage waveform at the time of interruption | blocking of the element characteristic 1. FIG. (A) And (b) is a figure which shows the current waveform and voltage waveform at the time of interruption | blocking of the element characteristic 2. FIG. (A), (b) is a figure which shows the current waveform and voltage waveform at the time of interruption | blocking of the element characteristic 1. FIG. It is a block block diagram of the direct-current interruption device described in patent documents 1. It is a block block diagram of the semiconductor interruption | blocking apparatus described in patent document 2. FIG. It is a circuit model figure for demonstrating interruption | blocking operation | movement of the semiconductor interruption | blocking apparatus in FIG. It is a figure which shows the time change of the voltage of each part at the time of interruption | blocking operation | movement in FIG. (A), (b) is explanatory drawing regarding the determination method of voltage Va, Vb described in the nonpatent literature 1. FIG. It is the figure which showed the result in the several sample at the time of using the determination method of voltage Va and Vb in the table | surface.

Embodiments of the present invention will be described below with reference to the drawings.
FIG. 1 is a block diagram for explaining a transistor drive circuit and a semiconductor circuit breaker using the transistor drive circuit according to the present invention. In the figure, 61 is a semiconductor circuit breaker, 62 is a DC power supply, 63 is a load device, 64 is a positive power supply line, 65 is a negative power supply line, 66 is a semiconductor switch (transistor), 67 is a current detector, and 68 is a current. A conversion unit (current measurement unit), 69 is a gate control unit, 70 is a control voltage storage unit, 71 is a transition time storage unit, and 72 is a system controller. The transistor drive circuit includes a current detection unit 67, a current conversion unit (current measurement unit) 68, a gate control unit 69, a control voltage storage unit 70, and a transition time storage unit 71.

The invention according to the first embodiment is the transistor drive circuit shown in FIG. The transistor drive circuit according to the first embodiment is a drive circuit for the transistor 66 that can limit an overcurrent in a short time and suppress an overvoltage generated when the transistor is shut down to a specified upper limit value or less.
That is, the transistor drive circuit according to the first embodiment includes a current detection unit 67 that detects a current on the source side of the transistor 66, and a first gate control voltage value V1 that is derived based on the current-voltage characteristics of the transistor 66. And the control voltage storage unit 70 in which the second gate control voltage value V2 is stored, and the transition from the first gate control voltage value V1 to the second gate control voltage value V2 of the transistor 66. A transition time storage unit 71 that stores a reference transition time value Tf that is a time, an output of the current detection unit 67, a first gate control voltage value V1, a second gate control voltage value V2, and a reference transition time value Tf. And a gate controller 69 for controlling the operation of the transistor 66 based on the above.

The gate control unit 69 determines a change time for reducing the gate voltage value Vgs of the transistor 66 from the first gate control voltage value V1 to the second gate control voltage value V2 based on the reference transition time Tf. It is.
Further, the gate control unit 69 sets the change time for stepping down the gate voltage value Vgs of the transistor 66 from the first gate control voltage value V1 to the second gate control voltage value V2 longer than the reference transition time Tf. It is controlled so as to be longer than a percent.

The gate control unit 69 linearly steps down the gate voltage value Vgs of the transistor 66 from the first gate control voltage value V1 to the second gate control voltage value V2 along the elapsed time.
Further, when the gate control unit 69 steps down the gate voltage value Vgs of the transistor 66 from the first gate control voltage value V1 to the second gate control voltage value V2, the drain voltage value Vds matches the electromotive voltage upper limit value Vdmax. It controls to do.

Further, the gate control unit 69 controls the drain current Ids to linearly decrease when the gate voltage value Vgs of the transistor 66 is stepped down from the first gate control voltage value V1 to the second gate control voltage value V2. Is.
Further, the gate control unit 69 uses the current value of the current detection unit 67 to drain current when the gate voltage value Vgs of the transistor 66 is stepped down from the first gate control voltage value V1 to the second gate control voltage value V2. Ids is controlled to decrease linearly.

  Further, when the gate control unit 69 steps down the gate voltage value Vgs of the transistor 66 from the first gate control voltage value V1 to the second gate control voltage value V2, the gate control unit 69 steps down the gate voltage value Vgs linearly with time. The gate voltage value Vgs in the intermediate region between the gate voltage value Vgs and the gate voltage value Vgs when the drain voltage value Vgs is controlled to coincide with the electromotive voltage upper limit value Vdmax is controlled.

  The current detector 67 detects a current on the source side of the transistor 6. The current measurement unit 68 outputs the current value detected by the current detection unit 67 to the gate control unit 69 as an analog value or a digital value. The gate control unit 69 compares the measured current value with the overcurrent threshold, and when the measured current value is exceeded, the gate voltage of the transistor 6 is set to two control voltage values in the control voltage storage unit 70 and the two control voltages. The drive state of the transistor 66 is determined from the transition time value of the transition time storage unit 71 that determines the time change interval between the two. By this gate voltage control, the drain voltage of the transistor 6 is controlled to be equal to or lower than the upper limit value of the designated overvoltage, and the drain current decreases in a short time.

  The transistor 66 is preferably a field effect transistor. The field effect transistor is preferably a junction field effect transistor, and the junction field effect transistor is preferably formed using silicon, silicon carbide, or gallium nitride as a semiconductor material. The junction field effect transistor is preferably a normally-on junction field effect transistor.

  As described above, according to the transistor drive circuit of the present invention, it is possible to limit the overcurrent in a short time and to suppress the overvoltage generated when the transistor is shut off to a specified upper limit value or less.

The invention according to the second embodiment is the semiconductor circuit breaker shown in FIG. The semiconductor circuit breaker 61 according to the second embodiment uses the transistor drive circuit according to the first embodiment described above, and is connected between the load device 63 connected to the power supply lines 64 and 65 from the DC power supply 62 and the DC power supply 62. A semiconductor circuit breaker 61 is provided.
That is, the semiconductor circuit breaker 61 includes a transistor 66 provided between the DC power supply 62 and the load device 63, and the transistor drive circuit described above and the transistor drive circuits described in the following embodiments. is there.

The current detection unit 67 detects a current flowing through the feeder line 65. The current measurement unit 68 outputs the current value detected by the current detection unit 67 to the gate control unit 69 as an analog value or a digital value. The semiconductor switch 66 controls the current flowing through the feeder line 64.
Further, the gate control unit 69 compares the measured current value of the power supply line with the overcurrent threshold, and when exceeding this, the gate voltage of the semiconductor switch 66 is compared with the two control voltage values of the control voltage storage unit 70. The driving state of the semiconductor switch 66 is determined by determining the transition time value of the transition time storage unit 71 that determines the time change interval between the two control voltages. By this gate voltage control, the drain voltage of the semiconductor switch 66 is controlled to be equal to or lower than the upper limit value of the designated overvoltage, and the drain current decreases in a short time.

The current detection unit 67 may be a current sensor that is a non-contact type using a Hall element or the like, or a type using a shunt resistor. Alternatively, a silicon Hall element may be used to form a single chip with the current measuring unit 68 or the like.
The gate control unit 69 has an overcurrent threshold storage unit, compares the current value from the current measurement unit 68 with the overcurrent threshold, and determines the current flowing through the semiconductor switch when the current value exceeds the overcurrent threshold. A gate voltage control operation for blocking is entered. In order to turn off the semiconductor switch, the gate voltage is set to the second gate control voltage value after a certain time from the first gate control voltage value. Used, the gate output voltage is changed by the DA converter, and the time change interval between the two control voltage values is measured by a clock circuit, and the input control voltage value to the DA converter can be changed.

  The semiconductor switch 66 is preferably a field effect transistor. The field effect transistor is preferably a junction field effect transistor, and the junction field effect transistor is preferably formed using silicon, silicon carbide, or gallium nitride as a semiconductor material. The junction field effect transistor is preferably a normally-on junction field effect transistor.

As described above, according to the semiconductor circuit breaker of the present invention, it is possible to limit the overcurrent in a short time and to suppress the overvoltage generated when the transistor is shut down to a specified upper limit value or less.
In addition, the semiconductor circuit breaker breaking control method of the present invention uses the transistor drive circuit of the first embodiment described above, and uses a load device 63 connected to the power supply lines 64 and 65 from the DC power supply 62, the DC power supply 62, It is the interruption | blocking control method of the semiconductor circuit breaker 61 provided between these.

  As described above, the semiconductor circuit breaker 61 is related to the current detector 67 that detects the current flowing through the power supply line, the semiconductor switch 66 that controls the current flowing through the power supply line, and the current-voltage characteristics of the semiconductor switch 66. A control voltage storage unit 70 for the gate control voltage value, a transition time storage unit 71 for storing a reference transition time value Tf between the two gate control voltages V1 and V2 of the semiconductor switch 66, an output of the current detection unit 67 and gate control And a gate control unit 69 that determines the driving state of the semiconductor switch 66 from the voltage values V1 and V2 and the reference transition time value Tf.

  The first gate control voltage value V1 of the control voltage storage unit 70 is related to the current-voltage characteristics of the semiconductor switch 66, the drain voltage value Vds is the DC power supply voltage value Vdd, and the drain current value Ids is an excess current flowing through the feeder line. The first gate voltage value V1a determined based on the current threshold value or the drain voltage value Vds is the electromotive voltage upper limit value Vdmax, and the drain current value Ids is the overcurrent threshold value Iov flowing through the feeder line. The second gate voltage value V1b is greater than the second gate voltage value V1b and less than or equal to the first gate voltage value V1a.

  The second gate control voltage value V2 of the control voltage storage unit 70 is related to the current-voltage characteristics of the semiconductor switch 66, the drain voltage value Vds is the DC power supply voltage Vdd, and the drain current value Ids is zero. Or the fourth gate voltage value V2a determined by the drain voltage value Vds being the electromotive voltage upper limit value Vdmax and the drain current Ids being zero. The gate voltage value Vgs is greater than the third gate voltage value V2b and less than or equal to the third gate voltage value V2b.

Further, the reference transition time value Tf of the control voltage storage unit 70 is set such that the DC power supply voltage value Vdd, the circuit inductance L between the semiconductor switches 66, the overcurrent threshold Iov flowing through the feeder line, the electromotive voltage upper limit value Vdmax of the drain voltage Vds, and the DC. The reference transition time value Tf is calculated from the power supply voltage value Vdd.
Then, after the current detector 67 detects that the current flowing through the feeder line has exceeded the overcurrent threshold Iov, the gate voltage value Vgs of the semiconductor switch 66 is set to the first gate control voltage value V1, and the reference transition time value Tf Is reduced to the second gate control voltage value V2. This reference transition time value Tf is calculated by the relational expression Tf = L × Iov / (Vdmax−Vdd) described above.

  FIG. 2 is a block configuration diagram for explaining a transistor drive circuit obtained by adding a transition time calculation unit to the first embodiment and a semiconductor circuit breaker using the transistor drive circuit. Reference numeral 73 in the figure denotes a transition time calculation unit. In addition, the same code | symbol is attached | subjected to the component which has the same function as FIG.

The invention according to Embodiment 3 is the transistor drive circuit shown in FIG. The transistor drive circuit according to the third embodiment includes the transistor drive circuit according to the first embodiment and a transition time calculation unit 73. The transition time calculator 73 obtains information on the circuit inductance L, the overcurrent threshold Iov of the current flowing through the source side of the transistor, the electromotive voltage upper limit value Vdmax of the drain voltage, and the voltage value Vdd of the DC power supply from the system controller 72. In response, the transition time Tf of the following equation (1) is calculated. In addition, the calculation result is stored in the transition time storage unit 71.
Tf = L × Iov / (Vdmax−Vdd) (1)

This calculation can be performed by using a multiplier and an adder inside the transition time calculation unit 73, or can be calculated using only the adder. This calculation is performed on the system controller side in software, and the result is written in the transition time storage unit 71.
That is, the transistor drive circuit of the third embodiment includes the DC power supply 92 and the circuit inductance L on the drain side of the transistor 66, and further includes the circuit inductance L, the overcurrent threshold value Iov, the electromotive voltage upper limit value Vdmax, and the DC power supply voltage. A transition time calculation unit 73 that calculates a reference transition time value Tf from Vdd is provided, and this reference transition time value Tf is a value calculated by the above-described relational expression Tf = L × Iov / (Vdmax−Vdd). .

The invention according to the fourth embodiment is the semiconductor circuit breaker shown in FIG. The semiconductor circuit breaker 61 according to the second embodiment uses the transistor drive circuit according to the third embodiment described above, and is connected between the load device 63 connected to the power supply lines 64 and 65 from the DC power supply 2 and the DC power supply 62. A semiconductor circuit breaker 61 is provided.
In addition, the semiconductor circuit breaker breaking control method of the present invention uses the transistor drive circuit of the third embodiment described above, a load device 63 connected to the power supply lines 64 and 65 from the DC power supply 62, the DC power supply 62, It is the interruption | blocking control method of the semiconductor circuit breaker 61 provided between these.

FIG. 3 is a block configuration diagram for explaining a transistor drive circuit obtained by adding a voltage detection unit to the first embodiment and a semiconductor circuit breaker using the transistor drive circuit. Reference numeral 74 in the figure denotes a voltage detection unit. In addition, the same code | symbol is attached | subjected to the component which has the same function as FIG.
The invention according to the fifth embodiment is a transistor drive circuit. The transistor drive circuit according to the fifth embodiment includes the transistor drive circuit according to the first embodiment and a voltage detection unit 74. The voltage detector 74 detects the drain voltage of the transistor and outputs it to the gate controller 69. When the measured current value exceeds the overcurrent threshold, the gate control unit 69 changes the gate voltage of the transistor 66 between the two control voltage values of the control voltage storage unit 70 and the change time between the two control voltages. Is determined based on the transition time value of the transition time storage unit 71 that determines the interval of the above and the drain voltage from the voltage detection unit 74, and the driving state of the transistor 66 is determined.

  That is, the transistor drive circuit of this embodiment includes a voltage detection unit 74 that detects the drain voltage value Vds, and the gate control unit 69 determines the gate voltage value Vgs of the transistor 66 from the first gate control voltage value V1. When the voltage is lowered to the gate control voltage value V2 of 2, the drain voltage value Vds is controlled to coincide with the electromotive voltage upper limit value Vdmax.

The invention according to the sixth embodiment is the semiconductor circuit breaker shown in FIG. The semiconductor circuit breaker 61 according to the sixth embodiment uses the transistor drive circuit according to the fifth embodiment described above, and is connected between the load device 63 connected to the power supply lines 64 and 65 from the DC power supply 62 and the DC power supply 62. A semiconductor circuit breaker 61 is provided.
Further, the method for controlling the breaker of the semiconductor circuit breaker 61 according to the present invention uses the transistor drive circuit of the fifth embodiment described above, the load device 63 connected to the feeder lines 64 and 65 from the DC power source 62, and the DC power source 62. It is the interruption | blocking control method of the semiconductor circuit breaker 61 provided between these.

FIG. 4 is an explanatory diagram of the breaking operation using the drain voltage of the semiconductor breaker according to the sixth embodiment shown in FIG. In the figure, 81 is a linear gate voltage generator, 82 is a gate voltage corrector, 83 is a DAC code generator, 91 is circuit inductance, and 92 is a DC power source.
The gate control unit 69 includes a linear gate voltage generation unit 81 connected to the control voltage storage unit 70 and the transition time storage unit 71, and a gate signal of the transistor 66 based on the gate signal output from the linear gate voltage generation unit 81. And a gate voltage correction unit 82 for correcting the above.
Then, the gate control unit 69 corrects the gate signal of the transistor 66 based on the gate signal output from the linear gate voltage generation unit 81 by the gate voltage correction unit 82.

  Furthermore, with reference to FIG. 4, the interruption | blocking operation | movement of Example 6 is demonstrated. For the sake of simplicity, the feeder line is only in one direction. The drain side voltage of the semiconductor switch 66 is detected by the voltage detection unit 74 and input to the gate control unit 69 which is a partial functional block of the gate control unit shown in FIG. The gate control unit 69 includes a linear gate voltage generation unit 81 and a gate voltage correction unit 82, and the gate signal output from the linear gate voltage generation unit 81 is corrected by the output of the gate voltage correction circuit 82 and the gate of the semiconductor switch 66. Voltage.

  The gate controller 69 enters a gate voltage control operation for cutting off the current flowing through the semiconductor switch 66 when the value of the current flowing through the feeder line becomes larger than the overcurrent threshold. The linear gate voltage generator 81 linearly changes the output gate voltage value from the first gate control voltage value V1 to the second gate control voltage value V2 at the time interval Tf. The DAC code generator 83 linearly calculates and generates a voltage between the gate control voltage value V1 and the gate control voltage value V2 at the time Tf, encodes the generated voltage value, and inputs the code to the DA converter. Is output. With this controlled gate voltage, it is possible to drive the semiconductor switch as it is to perform the shut-off operation.

  In the sixth embodiment, the gate voltage correction circuit 82 corrects the output gate voltage of the linear gate voltage generator 81 in order to further improve the cutoff characteristic. The gate voltage correction circuit 82 detects the difference in magnitude between the drain voltage value (Vdin) detected by the voltage detector 74 and the electromotive voltage upper limit value (Vref), and the drain voltage value (Vdin) becomes the electromotive voltage upper limit value (Vref). ) Is smaller, the gate voltage of the semiconductor switch 66 is lowered, and feedback is performed so that the drain voltage value approaches the electromotive voltage upper limit value. As the electromotive voltage value of the drain voltage is larger, the time during which the overcurrent is consumed is shorter. Therefore, the control to the allowable electromotive voltage upper limit value is the most efficient control by reducing the current value.

  Further, the gate correction circuit 82 increases the gate voltage of the semiconductor switch 66 when the drain voltage value (Vdin) tends to be larger than the electromotive voltage upper limit value (Vref), and the drain voltage value exceeds the electromotive voltage upper limit value. Work to suppress so as not to. In this feedback operation, the cutoff operation proceeds, and when the current value approaches zero, the drain voltage cannot be controlled to the upper limit of the electromotive voltage in a circuit, and the output of the linear gate voltage generation unit 81 becomes dominant and is cut off. As the operation proceeds, the drain voltage becomes the power supply voltage value.

The gate control unit 69 is configured by a combination of the linear gate voltage generation unit 81 and the gate voltage correction circuit 82. However, the gate control unit 69 only needs to have a circuit configuration that provides feedback so that the drain voltage becomes the upper limit of the electromotive voltage during the cutoff operation. .
The semiconductor switch 66 is preferably a field effect transistor. The field effect transistor is preferably a junction field effect transistor, and the junction field effect transistor is preferably formed using silicon, silicon carbide, or gallium nitride as a semiconductor material. The junction field effect transistor is preferably a normally-on junction field effect transistor.

As described above, according to the semiconductor circuit breaker 61 of the present invention, the overcurrent can be limited in a short time, and the overvoltage generated when the semiconductor circuit breaker is interrupted can be suppressed to a specified upper limit value or less.
Next, the operation of the semiconductor circuit breaker will be described particularly in relation to the control voltage value and the transition time that govern the basic operation of the present invention.

5A and 5B are a cutoff circuit and a gate waveform diagram thereof for explaining the operation of the semiconductor switch.
First, the semiconductor switch operation of the semiconductor circuit breaker will be described based on FIG. The situation in which the circuit breaker detects an overcurrent and performs a circuit break operation is explained in the same way by the circuit for examining circuit breakage in FIG. When the gate voltage Vgs changes from the off voltage Voff to the on voltage Von in the gate signal (waveform) 93 of FIG. 5B, the current value of the semiconductor switch (transistor) 66 increases due to the power supply voltage (VDD) 92 in the on state. To go. When a short circuit accident occurs on the load side, overcurrent flows through the power supply path. When the drain current of the semiconductor switch (transistor) 66 in FIG. 5A is a current flowing through the feeder line, and this current value reaches the overcurrent threshold Iov, the cutoff operation is started. In the gate waveform, the gate voltage is lowered from the Von level to the first gate control voltage value V1 level to change the semiconductor switch (transistor) 66 from the ON state to the intermediate state, and the resistance value between the drain and source of the transistor is changed. Increase to decrease the current value.

In the present invention, as shown in FIG. 5B, the gate waveform is lowered from Von to the first gate control voltage value V1, and further to the second gate control voltage value V2 at a constant time interval, as shown in FIG. Consider the shutoff operation on the premise of control.
When the gate voltage is suddenly lowered at the time of shut-off, the drain voltage rapidly rises due to the circuit inductance of the power supply path on the drain side of the transistor. The generated electromotive voltage has an upper limit value depending on the breakdown voltage of the transistor and the device connected to the power supply path, and this upper limit value is set as an electromotive voltage upper limit value Vdmax. Electrical energy is accumulated in the circuit inductance due to the current flowing through the power supply path during normal operation. At the time of interruption, it takes time for the current to flow in order to consume the energy of the circuit inductance by the resistance of the transistor. However, gate voltage control that consumes this current and shortens the interruption operation is desired.

  6A and 6B are diagrams for explaining the IV characteristics of the semiconductor switch (transistor). Normally, the IV characteristic of a transistor is measured by a curve tracer. However, since the region where the transistor of the present invention operates is a high voltage and high current region, it cannot be measured by a normal curve tracer. The present inventors obtain the IV characteristics based on the drain voltage and drain current values at the time of switching, and are related to the electromotive voltage upper limit value Vdmax and the gate control voltage value at the time of shutoff according to this IV characteristic diagram. I found out.

  FIG. 6A shows changes in drain voltage and drain current when the circuit of FIG. 5A is the gate drive waveform of FIG. 5B and V1 = V2. When the gate voltage is lowered from the ON voltage to the first gate control voltage V1, the drain current is lowered and the drain voltage is also lowered. In this process, the relationship between the drain current Ids and the drain voltage Vds is changed by changing the gate voltage Vgs and the power supply voltage Vdd. FIG. 6 (b) shows the measurement. FIG. 6B shows the case of element characteristic 1, and this IV characteristic changes depending on the structure and process. In the Example of this invention, it implements about the element of the element characteristic 1 and the element characteristic 2. FIG.

FIG. 7 is a diagram illustrating an example in which the IV characteristic is obtained for the element having the element characteristic 2. The plurality of lines correspond to different gate voltages Vgs, and from Vgs1 to Vgs12, the gate voltage difference ΔVgs is 0.2V and becomes smaller.
The electromotive voltage upper limit value Vdmax, the gate control voltage Vgs, etc. in the circuit of FIG. 5A will be described with reference to FIG. When the overcurrent increases in the circuit of FIG. 5A and reaches the overcurrent threshold value Iov, when the current reaches 10 A in this embodiment, the drain voltage is zero in the IV characteristic diagram of FIG. Is at point A on the current axis 10A. Next, the gate control voltage is set to the first gate control voltage value V1. At this time, on the IV characteristic diagram, the drain current is substantially the overcurrent threshold value Iov and remains constant at 10 A, while the drain voltage is constant. The gate voltage Vgs is V1a at the point where the power supply voltage Vdd is reached, the gate voltage Vgs is V1b at the point where the overcurrent threshold value Iov is 10 A and the drain voltage is the upper limit of the electromotive voltage Vdmax. It can be seen that when the gate electromotive voltage Vgs is smaller than V1b, the electromotive voltage is larger than the upper limit value Vdmax.

  That is, the first gate control voltage value V1 of the control voltage storage unit 70 is related to the current-voltage characteristics of the transistor 66, and the drain voltage Vds is the DC power supply voltage Vdd and the drain current Id is the overcurrent threshold Iov. Or the second gate voltage value V1b determined by the drain voltage Vds being the electromotive voltage upper limit value Vdmax and the drain current Id being the overcurrent threshold value Iov, or One of the gate voltage values Vgs that is greater than the second gate voltage value V1b and less than or equal to the first gate voltage value V1a.

An embodiment in which the gate voltage is controlled based on the values of the power supply voltage and the upper limit of the electromotive voltage based on the IV characteristics will be described below.
FIGS. 8A and 8B are diagrams showing measurement examples of the relationship between the gate voltage V1 and the drain electromotive voltage when cut off. FIG. 8A is a current interruption waveform, and FIG. 8B is a diagram showing a current interruption characteristic in a table. The current cutoff waveform shown in FIG. 8A is the current cutoff waveform of the first gate control voltage V1 shown in FIG. 8B when the power supply voltage is 100 V, the electromotive voltage upper limit is 150 V, and the overcurrent threshold is 2 A. The graph shows changes in drain current Ids and drain voltage Vds when changing from (1) to (3) in the characteristic table. The second gate control voltage V2 is constant. In this embodiment, an element having element characteristics 2 is used.

  (2) shown in FIG. 8B is the gate control voltage V1 = −4.4V, and when the gate electromotive voltage is changed to V1, the electromotive voltage upper limit value is 150V. (1) is the gate control voltage V1 = -4.3V, and when the gate electromotive voltage is changed to V1, the drain voltage becomes around the power supply voltage 100V. With the gate control voltage of (1), when the drain voltage Vds rises, the electromotive voltage upper limit value 150V is not reached, but thereafter, the peak voltage of the drain voltage Vds reaches the electromotive voltage upper limit value 150V and further decreases. In (1) and (2), the decreasing tendency of the current value at the start of interruption is larger in (2). (3) shows a case where the gate control voltage is set lower than (2) for comparison. The drain voltage exceeds the upper limit of the electromotive voltage of 150 V and reaches 270 V, but the decrease in the drain current Ids at the start of the cutoff is larger than the others.

  When the gate voltage V1 is greater than (1), the voltage at the rise of the drain voltage Vds is lower than the power supply voltage 100V, which is close to the on state of the transistor. At this time, the drain current is reduced in a direction worse than that in the case of (1), which is disadvantageous. In the description of FIG. 7, the gate drive condition (1) corresponds to the gate voltage V1a, and the gate drive condition (2) corresponds to the gate voltage V1b.

  FIGS. 9A and 9B are diagrams for explaining the gate control voltage and the cutoff time. FIG. 10 is a diagram illustrating the IV characteristic in the case of the element characteristic 2. The time interval Δt between the second gate control voltage V2 and the first gate control voltage V1 will be described based on FIGS. 9A and 9B and FIG. The model circuit shown in FIG. 9A is a diagram in which the transistor portion is replaced with a resistor in the cutoff circuit of FIG. It shows that the transistor is equivalent to the variable resistance value by gate voltage control.

This model circuit satisfies the following differential equation (2).
L * (dI / dt) + R (t) * I = Vdd (2)
Assuming that the overcurrent value Iov becomes zero after Tf due to the interruption operation, the interruption time Tf is expressed by the following expression (3) from Expression (2).

The term R * I in equation (3) is equal to the drain voltage Vds.
The cut-off time Tf becomes smaller as the integration becomes smaller, but this value depends on what locus the drain voltage Vds follows. In FIG. 10, when the case where the drain voltage Vds takes two loci of “a” and “b” from the overcurrent value Iov and the electromotive voltage upper limit value Vdmax is compared, the denominator in the integral of the expression (3) is (Vds −Vdd), it is understood that the convergence time Tf is shorter when following the trajectory of “a”. In the situation where the current value has become sufficiently small, the resistance value of the transistor remains almost constant even when the gate voltage is controlled, and the time required for shutoff is expressed differently from Equation (3). The approximate cutoff time can be approximated by equation (3).

Assuming that the gate voltage can be controlled so that the drain voltage Vds follows the trajectory of “a”, the cutoff time Tf is the smallest when Vds = Vdmax in the integration of Expression (3). At this time, the value during integration is a constant value, L / (Vdmax−Vdd), which can be removed from the integration. After all, the ideal value of the cutoff time Tf is the following equation (4)
Tf = L × Iov / (Vdmax−Vdd) (4)
It becomes. The actual cutoff time is longer than this Tf. However, if there is a parasitic capacitance C in addition to the inductance L and the transistor resistance value R in the equation (2), the cutoff time is shorter than the cutoff time Tf in the equation (4). It may become.

Further, the time change of the current value when the gate voltage can be controlled so that the drain voltage Vds takes the locus of “a” is expressed by the equation (2) as R (t) * I = Vdmax. The following equation (5) is obtained.
dI / dt = (Vdd−Vdmax) / L (5)
Solving for current I, the following equation (6)
I (t) = Iov− (Vdmax−Vdd) / L * t (6)
Thus, the current decreases linearly with time from the overcurrent value Iov. It can be seen that controlling the gate voltage so that the drain current decreases linearly is equivalent to controlling the gate so that the drain voltage is kept constant at Vdmax.

  The second gate control voltage V2 is a voltage between a gate voltage V2b at which the drain voltage Vds becomes the power supply voltage Vdd and a voltage value including the gate voltage V2a at which the drain voltage Vds becomes Vdmax in a state where the current value is zero. Is possible. When the gate voltage Vgs is smaller than V2a, the drain electromotive voltage may exceed Vdmax, and thus cannot be applied. When the gate voltage Vgs is larger than V2b, there is a possibility that the current remains and the drain voltage Vds may be smaller than the power supply voltage Vdd, which is not converged.

  Although the IV characteristic and the threshold value of the semiconductor switch are different for each element, these are limited within the range of the characteristic value described in the specifications of the data sheet of the element. By examining the range in which the IV characteristics vary in advance, it is possible to select the gate voltage Vgs that does not exceed the upper limit of the electromotive voltage and set it as the first gate control voltage. Similarly, the second gate control voltage at the time of current interruption can be selected as a gate voltage at which the current is cut even if the element characteristics vary. As a result, it is possible to eliminate the need to adjust the gate voltage and the breaking time for each semiconductor switch used and for each breaker.

  FIGS. 11A to 11C are diagrams showing current waveforms and voltage waveforms when the element characteristic 1 is cut off. The cut-off waveform is shown when the element having the element characteristic 1 is used and the above-described first gate control voltage V1, second gate control voltage V2, and cut-off time Tf are determined. The time change of the gate voltage from the first gate control voltage V1 to the second gate control voltage V2 is linearly changed by the cutoff time Tf.

When the circuit inductance L of the feeder line is 500 uH, the power supply voltage is 100 V, and the electromotive voltage upper limit value is 150 V, and the overcurrent threshold value is 4 A, the cutoff time Tf is Tf = 40 us and the overcurrent threshold value is 8 A from Equation (4) The cutoff time Tf = 80 us.
FIG. 11A is a cut-off waveform diagram when V1 = −4.68, V2 = −4.85, and Δt = 40 us. The second gate control voltage V2 is a gate voltage Vgs when current = zero and drain voltage = 100V in the IV characteristic diagram.

FIG. 11B shows the case where the gate control voltage is the same, and the cut-off time Tf = 45 us is increased by about 10%. In FIG. 11A, the drain voltage does not converge at the cutoff time Tf = 40 us and oscillates when the gate voltage Vgs is off. However, when the cutoff time Tf is increased by about 10%, the oscillation is suppressed, The voltage is cut off.
FIG. 11C is a cut-off waveform diagram when the overcurrent threshold is changed from 4 A to 8 A and the cut-off time is 90 us. V1 = −4.23 and V2 = −4.85. Since the overcurrent threshold is 8 A, the cutoff time Tf is calculated to be 80 us, but it is extended by 10% to 90 us. Both current and voltage are cut off.

12A and 12B are diagrams showing a current waveform and a voltage waveform when the element characteristic 2 is interrupted. The cut-off waveform when an element having the element characteristic 2 is used and the above-described first gate control voltage V1, second gate control voltage V2, and cut-off time Tf are determined is shown.
When the circuit inductance L of the feeder line is 500 uH, the power source voltage is 100 V, and the electromotive voltage upper limit value is 200 V, and the overcurrent threshold is 4 A, the cutoff time Tf is Tf = 20 us from the equation (4).

  In the gate voltage waveform shown in FIG. 12A, the time interval between the first gate control voltage V1 and the second gate control voltage V2 is constant at 20 us of the reference transition time Tf. Is not linear with respect to the elapsed time, and the drain voltage is changed to be constant at Vdmax. V1 = −4.78 and V2 = −5.48. On the time axis, the voltage V1 is around 18 us and the voltage V2 is around 38 us. Thereafter, the gate voltage is decreased in such a manner as to extend the time change of the gate voltage until then.

  FIG. 12B shows a drain voltage and a current waveform. The drain voltage continues to be fixed at 200 V by the control. Further, the drain current decreases almost linearly up to around 30 us where the drain voltage is kept constant. By controlling the gate voltage between the first gate control voltage V1 and the second gate control voltage V2 so that the drain voltage becomes constant at Vdmax, the drain current and voltage are approximately cut off time Tf = 20 us. It has converged.

  13A and 13B are diagrams showing a current waveform and a voltage waveform when the element characteristic 1 is cut off. A voltage between the gate control voltage V1 and the second gate control voltage V2 is determined by using the element having the element characteristic 1 to determine the first gate control voltage V1, the second gate control voltage V2, and the cutoff time Tf. The gate voltage is changed so that the drain voltage becomes the electromotive voltage upper limit value Vdmax. The same gate voltage control as that of the gate voltage waveform in FIG.

The circuit inductance L of the power supply line is 500 uH, the power supply voltage is 100 V, the electromotive voltage upper limit value is 150 V, and the overcurrent threshold is 8 A. The cutoff time Tf is Tf = 80 us from the equation (4).
FIG. 13A is a gate voltage waveform diagram. The curve “a” is the time waveform of the gate voltage obtained from the IV characteristic diagram so that the drain voltage becomes Vdmax, and “b” is the gate voltage waveform actually created.

  FIG. 13B shows the current, voltage waveform, and gate voltage waveform. Both the current value and the drain voltage value converge at a cutoff time of 80 us. FIG. 11C shows a waveform in which the gate voltage is linearly changed between the gate control voltage V1 and the gate control voltage V2 and converges at 90 us. However, the drain voltage is made to coincide with the upper limit of the electromotive voltage. When the gate voltage is controlled, convergence is improved. From the examples of FIGS. 13A and 13B and FIGS. 11A to 11C, the gate control voltage V1 and the gate control voltage V2 are changed as time changes from the gate control voltage V1 to the gate control voltage V2. It is possible to make the waveform between two waveforms of the gate waveform when the gate voltage is controlled so that the gate voltage and the drain voltage become the upper limit of the electromotive voltage when the time is linearly changed. I understand. Preferably, control is performed so that the drain voltage matches the upper limit of the electromotive voltage.

  As described above, the IV characteristics of the semiconductor switch are obtained, and the first gate control voltage, the second gate control voltage, and the cutoff time are determined from the system specifications such as the power supply voltage, the electromotive voltage upper limit value, and the overcurrent threshold. Thus, the drain voltage can be suppressed below the upper limit of the electromotive voltage at the time of interruption, and the interruption time can be determined appropriately.

DESCRIPTION OF SYMBOLS 10 DC current interruption | blocking apparatus 11 Current detection part 12 Switch part 13 Control circuit 14 Overcurrent determination part 15 Switch drive part 16 Arbitrary waveform generation part 17 Feed line 21 Semiconductor circuit breaker 22 Power supply apparatus 23 Load apparatus 31 Semiconductor part 32 Current sensor 33 Current Measurement unit 34 Transistor drive circuit 35 Input interface 41, 42 DC output lines 43, 44 DC input line 51 Drive unit 52 Control unit 53 Setting storage unit 54 Storage unit 61 Semiconductor circuit breaker 62 DC power supply 63 Load device 64 Positive power supply line 65 Negative side feed line 66 Semiconductor switch (transistor)
67 Current detection unit 68 Current conversion unit (current measurement unit)
69 Gate control unit 70 Control voltage storage unit 71 Transition time storage unit 72 System controller 73 Transition time calculation unit 74 Voltage detection unit 81 Linear gate voltage generation unit 82 Gate voltage correction unit 83 DAC code generator 91 Circuit inductance 92 DC power supply 93 Gate signal

Claims (27)

A current detector for detecting the current on the source side of the transistor;
A control voltage storage unit for storing a first gate control voltage value and a second gate control voltage value derived based on the current-voltage characteristics of the transistor;
A transition time storage unit that stores a reference transition time value that is a time until the gate voltage value of the transistor is changed from the first gate control voltage value to the second gate control voltage value;
A gate control unit that controls the operation of the transistor based on the output of the current detection unit, the first gate control voltage value, the second gate control voltage value, and the reference transition time value. A transistor driving circuit. The first gate control voltage value of the control voltage storage unit is
In relation to the current-voltage characteristics of the transistor,
A first gate voltage value determined by the drain voltage being a DC power supply voltage and the drain current being an overcurrent threshold;
The second gate voltage value determined by the drain voltage being an electromotive voltage upper limit value and the drain current being an overcurrent threshold;
2. The transistor drive circuit according to claim 1, wherein the gate voltage value is greater than the second gate voltage value and less than or equal to the first gate voltage value. The second gate control voltage value of the control voltage storage unit is
In relation to the current-voltage characteristics of the transistor,
A third gate voltage value determined by the drain voltage being the power supply voltage and the drain current being zero;
A fourth gate voltage value determined by the drain voltage being an electromotive voltage upper limit value and the drain current being zero;
3. The transistor drive circuit according to claim 1, wherein the transistor has a gate voltage value that is greater than the fourth gate voltage value and less than or equal to the third gate voltage value. 4.   The transistor includes a DC power supply and a circuit inductance on the drain side of the transistor, and further calculates a reference transition time value based on the circuit inductance, the overcurrent threshold, the electromotive voltage upper limit value, and the DC power supply voltage. The transistor drive circuit according to claim 1, wherein the transistor drive circuit is provided. The reference transition time value Tf is
Tf = L × Iov / (Vdmax−Vdd)
5. The transistor drive circuit according to claim 4, wherein the value is calculated by the relational expression:   The gate control unit determines a change time for stepping down the gate voltage value of the transistor from the first gate control voltage value to the second gate control voltage value based on the reference transition time. A transistor driving circuit according to claim 1.   The gate control unit controls a change time for dropping the gate voltage value of the transistor from the first gate control voltage value to the second gate control voltage value so as to be longer than the reference transition time. The transistor drive circuit according to claim 6, wherein:   The gate control unit linearly steps down the gate voltage value of the transistor from the first gate control voltage value to the second gate control voltage value along an elapsed time. A driving circuit of the transistor according to 1.   When the gate control unit steps down the gate voltage value of the transistor from the first gate control voltage value to the second gate control voltage value, the drain voltage value matches the electromotive voltage upper limit value. 8. The transistor driving circuit according to claim 6, wherein the transistor driving circuit is controlled.   A voltage detection unit configured to detect the drain voltage value; and when the gate control unit steps down the gate voltage value of the transistor from the first gate control voltage value to the second gate control voltage value, The transistor drive circuit according to claim 9, wherein a voltage value is controlled to coincide with the upper limit of the electromotive voltage.   The gate control unit corrects a gate signal of the transistor based on a linear gate voltage generation unit connected to the control voltage storage unit and the transition time storage unit, and a gate signal output from the linear gate voltage generation unit The transistor drive circuit according to claim 6, further comprising: a gate voltage correction unit configured to operate.   The gate control unit controls the drain current to linearly decrease when the gate voltage value of the transistor is lowered from the first gate control voltage value to the second gate control voltage value. The transistor drive circuit according to claim 6 or 7.   When the gate control unit steps down the gate voltage value of the transistor from the first gate control voltage value to the second gate control voltage value, the drain current is linearized using the current value of the current detection unit. 13. The transistor driving circuit according to claim 12, wherein the transistor driving circuit is controlled so as to decrease to a minimum value.   When the gate control unit steps down the gate voltage value of the transistor from the first gate control voltage value to the second gate control voltage value, the gate voltage when stepping down the gate voltage value linearly with time 8. A gate voltage value in a region intermediate between a value and a gate voltage value when the drain voltage value is controlled to coincide with the electromotive voltage upper limit value is controlled. A driving circuit of the transistor according to 1.   15. The transistor according to claim 1, wherein the transistor is a junction field effect transistor, and the junction field effect transistor is formed using silicon, silicon carbide, or gallium nitride as a semiconductor material. Transistor drive circuit.   16. The transistor drive circuit according to claim 15, wherein the junction field effect transistor is a normally-on junction field effect transistor.   17. A semiconductor circuit breaker comprising: a transistor provided between a DC power supply and a load device; and a transistor drive circuit according to claim 1. In the shutoff control method for a semiconductor circuit breaker provided between the transistor drive circuit according to any one of claims 1 to 16, a load device connected to a power supply line from a DC power supply, and the DC power supply.
The semiconductor circuit breaker is
A current detection unit for detecting a current flowing through the power supply line; a semiconductor switch for controlling a current flowing through the power supply line; a control voltage storage unit for a gate control voltage value related to a current-voltage characteristic of the semiconductor switch; A transition time storage unit that stores a reference transition time value between two gate control voltages of the semiconductor switch, and a driving state of the semiconductor switch is determined from the output of the current detection unit, the gate control voltage value, and the reference transition time value And a gate control unit
The first gate control voltage value of the control voltage storage unit is
Related to the current-voltage characteristics of the semiconductor switch,
A drain voltage value is a direct-current power supply voltage value and a drain current value is an overcurrent threshold value flowing through the feeder line, or a first gate voltage value determined by
The drain voltage value is an electromotive voltage upper limit value, and the drain current value is a second gate voltage value determined by being an overcurrent threshold value flowing through a feeder line,
Any one of the gate voltage values that is greater than the second gate voltage value and less than or equal to the first gate voltage value;
A second gate control voltage value of the control voltage storage unit is
Related to the current-voltage characteristics of the semiconductor switch,
A third gate voltage value determined by the drain voltage value being the DC power supply voltage and the drain current value being zero;
The drain voltage value is the upper limit of the electromotive voltage, and the fourth gate voltage value determined by the drain current being zero,
Any one of the gate voltage values that is greater than the fourth gate voltage value and less than or equal to the third gate voltage value;
The reference transition time value of the control voltage storage unit,
The reference transition time value is calculated from the DC power supply voltage value, the circuit inductance between the semiconductor switches, the overcurrent threshold value flowing through the feeder line, the electromotive voltage upper limit value of the drain voltage, and the DC power supply voltage value,
After detecting that the current flowing through the feeder line exceeds the overcurrent threshold, the gate voltage value of the semiconductor switch is set as the first gate control voltage value, and based on the reference transition time value And a step-down control method for the semiconductor circuit breaker, wherein the voltage is stepped down to the second gate control voltage value. The reference transition time value Tf is set to Tf = L × Iov / (Vdmax−Vdd)
19. The circuit breaker control method for a semiconductor circuit breaker according to claim 18, wherein the control method is as follows.   The gate control unit controls the transition time for stepping down the gate voltage value of the semiconductor switch from the first gate control voltage value to the second control voltage value to be longer than the reference transition time value. 20. The circuit breaker control method for a semiconductor circuit breaker according to claim 18 or 19.   The gate control unit linearly steps down the gate voltage value of the semiconductor switch from the first gate control voltage value to the second gate control voltage value along an elapsed time. The circuit breaker control method for a semiconductor circuit breaker according to 19 or 20.   When the gate control unit steps down the gate voltage value of the semiconductor switch from the first gate control voltage value to the second gate control voltage value, the drain voltage value matches the electromotive voltage upper limit value. 21. The circuit breaker control method for a semiconductor circuit breaker according to claim 18, 19 or 20, wherein A voltage detection unit for detecting the drain voltage value;
When the gate control unit steps down the gate voltage value of the semiconductor switch from the first gate control voltage value to the second gate control voltage value, the drain voltage value matches the electromotive voltage upper limit value. 21. The circuit breaker control method for a semiconductor circuit breaker according to claim 18, 19 or 20, wherein   The gate control unit includes a linear gate voltage generation unit connected to the control voltage storage unit and the transition time storage unit, and a gate voltage correction unit connected to the linear gate voltage generation unit, the gate voltage correction 21. The semiconductor breaker interrupt control method according to claim 18, wherein the gate signal of the transistor is corrected based on a gate signal output from the linear gate voltage generator.   The gate control unit controls the drain current value to decrease linearly when the gate voltage value of the semiconductor switch is stepped down from the first gate control voltage value to the second gate control voltage value. The circuit breaker control method for a semiconductor circuit breaker according to claim 18, 19 or 20.   When the gate control unit steps down the gate voltage value of the semiconductor switch from the first gate control voltage value to the second gate control voltage value, the drain current is detected using the current detection value of the current detection unit. 21. The interruption control method for a semiconductor circuit breaker according to claim 18, wherein the value is controlled so as to decrease linearly.   When the gate control unit steps down the gate voltage value of the semiconductor switch from the first gate control voltage value to the second gate control voltage value, the gate control unit linearly steps down the gate voltage value along the elapsed time. And a gate voltage value in a region intermediate between the gate voltage value in the case of controlling the drain voltage value so as to coincide with the upper limit of the electromotive voltage. Item 20. A circuit breaker control method for a semiconductor circuit breaker according to Item 18, 19 or 20.

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