JP2011217469A - Apparatus for control of motor drive device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To achieve an apparatus for control of a motor drive device, capable of properly completing stronger field control in a configuration for executing stronger field control and square wave control, based on a voltage index of a modulation factor, or the like.SOLUTION: A voltage waveform control unit 10 executes PWM control when the voltage index M indicating voltage command values Vd, Vq to a DC voltage Vdc is less than a prescribed square wave threshold, and executes square wave control when the voltage index M is not less than the square wave threshold. A field adjustment unit 8 executes stronger field control on the condition that the voltage index M becomes not less than a prescribed stronger field threshold smaller than the square wave threshold. A mode control unit 5 completes stronger field control by the field adjustment unit 8 on the condition that a field adjustment command value ΔId becomes not less than an adjustment command threshold value ΔIdT decided based on target torque TM and a voltage speed ratio RVω in a direction for strengthening a field magnetic flux.

Description

  The present invention relates to a control device that controls a motor drive device including a DC / AC converter that converts a DC voltage into an AC voltage and supplies the AC voltage to an AC motor.

  2. Description of the Related Art Generally, an electric motor driving device that drives an AC motor by converting a DC voltage from a DC power source into an AC voltage by an inverter is generally used. In such a motor drive device, sinusoidal PWM (pulse width modulation) control based on vector control and maximum frequency are used to efficiently generate torque by supplying a sinusoidal AC voltage to the coils of each phase of the AC motor. A lot of torque control is performed. By the way, the induced voltage of the electric motor increases as the rotational speed increases, and the AC voltage (hereinafter referred to as “required voltage”) required to drive the electric motor also increases. When this necessary voltage exceeds the maximum AC voltage that can be output from the inverter (hereinafter referred to as “maximum output voltage”), it is impossible to flow the necessary current through the coil, and the motor can be controlled appropriately. Can not. In order to reduce the induced voltage, field weakening control is performed to weaken the field magnetic flux of the electric motor. However, when the field weakening control is performed, the maximum torque control cannot be performed, so that the maximum torque that can be output decreases and the efficiency also decreases.

  With respect to such a problem, the following Patent Document 1 discloses an electric motor that shifts from sinusoidal PWM control to overmodulation PWM control and further to rectangular wave control as the rotational speed of the motor increases and the induced voltage increases. The technology of the control device of the drive device is described. Here, regarding the modulation rate which is the ratio of the effective value of the fundamental wave component of the AC voltage waveform to the DC power supply voltage (system voltage), the upper limit of the modulation rate is 0.61 in the sinusoidal PWM control. On the other hand, in the overmodulation PWM control, the modulation rate can be increased to a range of 0.61 to 0.78, and in the rectangular wave control, the modulation rate becomes a maximum of 0.78. Therefore, according to the control device described in Patent Document 1, by increasing the amplitude of the fundamental wave component of the AC voltage waveform supplied to the AC motor by overmodulation PWM control or rectangular wave control (increasing the modulation factor). Compared with a configuration in which only the sine wave PWM control is performed, the rotation speed region in which the maximum torque control can be performed by effectively using the DC voltage is expanded. When the required voltage of the motor is lower than the maximum output voltage, maximum torque control is performed together with sine wave PWM control or overmodulation PWM control. When the required voltage of the motor reaches the maximum output voltage, field weakening control is performed together with rectangular wave control.

  By the way, in the control device described in Patent Document 1, PWM control is performed in an operation region in which maximum torque control can be performed. However, such PWM control has a large number of on / off switching elements constituting the inverter. , Switching loss tends to increase. In order to further improve the efficiency of the electric motor, it is effective to suppress such switching loss. On the other hand, according to the rectangular wave control, the number of on / off times of the switching element can be significantly reduced as compared with the PWM control, so that switching loss can be suppressed. In Patent Document 2 below, a rectangular wave having a modulation rate as a maximum value is determined by determining a field adjustment command value in a direction in which the field magnetic flux of the AC motor is strengthened even in an operation region where PWM control can be performed. It is described that control (one pulse drive) is performed. As a result, the current flowing through the motor increases and the loss in the motor slightly increases, but the switching loss in the inverter can be reduced, and the efficiency of the entire system can be increased.

JP 2006-31770 A JP 2008-079399 A

  By the way, since the modulation factor is maintained at the maximum value during the rectangular wave control, in the configuration in which the rectangular wave control and the PWM control are switched according to the modulation factor, the rectangular wave control is ended even if the operating state of the AC motor changes. I can't. For this reason, even when the rotational speed of the AC motor is reduced or the target torque is reduced, the rectangular wave control is not completed only by increasing the field adjustment command value in the direction of increasing the field magnetic flux. Therefore, there is a possibility that the efficiency decreases due to an increase in the field adjustment command value, or that the output torque of the AC motor is vibrated by performing the rectangular wave control in a region where the rotational speed is low. However, Patent Document 2 does not describe any configuration for properly ending the rectangular wave control and the strong field control in the configuration in which the rectangular wave control is executed by the strong field control that strengthens the field magnetic flux. .

  Therefore, it is desired to realize a control device for an electric motor driving device that can appropriately end the strong field control in the configuration in which the strong field control and the rectangular wave control are executed based on the voltage index such as the modulation factor.

  In order to achieve the above object, according to the present invention, a characteristic configuration of a control device that controls a motor driving device including a DC / AC converter that converts a DC voltage into an AC voltage and supplies the AC voltage to the AC motor is the AC motor. A current command determining unit that determines a basic current command value that is a command value of a current to be supplied from the DC / AC converter to the AC motor based on a target torque of the current, and a field that is an adjustment value of the basic current command value Based on the field adjustment unit for determining the adjustment command value, the adjusted current command value after adjusting the basic current command value by the field adjustment command value, and the rotational speed of the AC motor, the DC-AC conversion A voltage command determining unit that determines a voltage command value that is a voltage command value to be supplied to the AC motor from a unit, and controlling the DC to AC conversion unit based on the voltage command value, and performing pulse width modulation control and rectangular wave control. A voltage waveform control unit that executes voltage waveform control including at least a mode control unit that controls the field adjustment unit and the voltage waveform control unit, and the voltage waveform control unit includes the voltage with respect to the DC voltage. When the voltage index representing the magnitude of the command value is less than a predetermined rectangular wave threshold, the pulse width modulation control is executed, and when the voltage index is equal to or larger than the rectangular wave threshold, the rectangular And the field adjustment unit determines the field adjustment command value so as to adjust the field flux of the AC motor to the basic current command value, and the basic field control. It is configured to execute field control including at least normal field control for determining the field adjustment command value so as not to adjust the current command value, and the voltage index is smaller than the rectangular wave threshold value. The strong field control is executed on condition that a certain strong field threshold value has been reached, and the mode control unit uses the ratio of the DC voltage and the rotational speed of the AC motor as a voltage speed ratio. The field-enhancing field control by the field adjustment unit is performed on condition that the magnetic field adjustment command value is equal to or greater than an adjustment command threshold value determined based on the target torque and the voltage speed ratio in the direction of increasing the field magnetic flux. It is in the point to end.

  According to this characteristic configuration, the strong field control is executed based on the voltage index representing the magnitude of the voltage command value with respect to the DC voltage, and the voltage index is raised by executing the strong field control to control the voltage waveform control as the rectangular wave control. Can be migrated to. Therefore, it is possible to widen the operation range where the rectangular wave control is performed in the AC motor, and it is possible to increase the efficiency by reducing the switching loss in the DC / AC converter. At this time, by appropriately determining the field adjustment command value by the field adjustment unit and changing the degree of the strong field, the torque corresponding to the target torque is appropriately set regardless of the rotational speed of the AC motor. Can be output. Further, according to this characteristic configuration, the field enhancement command control is terminated on the condition that the field adjustment command value is equal to or greater than the adjustment command threshold value determined based on the target torque and the voltage speed ratio in the direction of increasing the field magnetic flux. Therefore, the field control can be properly terminated before the efficiency is lowered due to the increase of the field adjustment command value. In other words, the field control can be terminated appropriately according to the relationship between the increase in loss in the motor due to the increase in the field adjustment command value and the reduction in switching loss due to the execution of the rectangular wave control. It can suppress that the efficiency as the whole system containing an electric motor and an electric motor drive device deteriorates. At this time, by using an adjustment command threshold value determined based on the target torque and the voltage / speed ratio, an appropriate adjustment command threshold value corresponding to the target torque and the voltage / speed ratio can be set.

  Here, the mode control unit decreases the adjustment amount of the field magnetic flux when the strong field control is terminated during the strong field / rectangular wave control mode in which the rectangular wave control is executed together with the strong field control. The normal field control is performed through a strong field / pulse width modulation control mode in which the voltage index is gradually decreased by gradually changing the field adjustment command value in a direction to be performed, and the pulse width modulation control is executed together with the strong field control. At the same time, it is preferable to shift to the normal field / pulse width modulation control mode in which the pulse width modulation control is executed.

  According to this configuration, when the strong field control is terminated from the strong field / rectangular wave control mode, the normal field / pulse width is obtained via the strong field / pulse width modulation control mode in which the pulse width modulation control is executed together with the strong field control. Since the mode is shifted to the modulation control mode, it is possible to suppress a sudden change in the field adjustment command value and the voltage index when the strong field control is terminated. Accordingly, it is possible to suppress a sudden change and overshoot of the current flowing in the coil of the AC motor, and it is possible to suppress the occurrence of vibration of the output torque of the AC motor.

  Further, the loss of the AC motor and the motor driving device when the normal field / pulse width modulation control mode for executing the pulse width modulation control together with the normal field control is executed is defined as a normal time loss, and the rectangular shape is set together with the strong field control. When the strong field / rectangular wave control mode for executing the wave control is executed, the loss of the AC motor and the motor driving device is set as the strong field time loss, and the strong field time loss is smaller than the normal time loss. It is preferable that the upper limit in the direction of increasing the field magnetic flux in the field adjustment command value range is the adjustment command threshold value.

  According to this configuration, in accordance with the strong field time loss and the normal time loss that change according to the target torque and the voltage-speed ratio, the strong field time loss is less than the normal time loss, that is, the strong field / rectangular wave. When the control mode is executed, the field control command should be properly terminated so that the field adjustment command value reaches the upper limit of the range where the loss is smaller than when the normal field / pulse width modulation control mode is executed. An adjustment command threshold can be set. Thus, the strong field control may be terminated when the efficiency reduction due to the increase in the motor loss due to the increase in the field adjustment command value exceeds the efficiency improvement associated with the reduction in the switching loss due to the rectangular wave control. it can. Therefore, the effect of the efficiency improvement accompanying the reduction of the switching loss can be maximized, and the efficiency of the entire system including the AC motor and the motor drive device can be improved.

  Further, the mode control unit satisfies both conditions that the field adjustment command value is equal to or greater than the adjustment command threshold value and that the rotation speed is less than a predetermined rotation speed threshold value. It is preferable to terminate the strong field control when it is determined and at least one of the conditions is satisfied.

  According to this configuration, in addition to the field adjustment command value being equal to or greater than the adjustment command threshold, it is also determined as a condition that the rotation speed is less than the rotation speed threshold, and at least one of the conditions Since the strong field control is terminated when the condition is satisfied, the strong field control can be appropriately terminated before the rotational speed of the AC motor drops below a rotational speed suitable for execution of the rectangular wave control. Therefore, by performing the rectangular wave control in the region where the rotational speed is low, it is possible to suppress the occurrence of vibration or the like in the output torque of the AC motor.

  Here, it is preferable that the rotational speed threshold value is determined based on the target torque and the DC voltage. According to this configuration, it is possible to set an appropriate rotation speed threshold according to the target torque and the DC voltage.

  Further, it is preferable that a rotation speed at which the voltage index becomes the strong field threshold value during the execution of the normal field control according to both values of the target torque and the DC voltage is the rotation speed threshold value. It is.

  According to this configuration, during the execution of the normal field control, in accordance with the voltage index that changes according to the target torque and the DC voltage, the voltage index is substantially less than the strong field threshold. The rotation speed threshold value can be appropriately set so as to end the strong field control. Thereby, the conditions for ending the strong field control can be set so as to match the conditions for starting the strong field control. Further, since the end condition of such strong field control can be determined based on the rotation speed according to both values of the target torque and the DC voltage, the strong field control can be ended easily and appropriately.

  Further, when the target torque of the AC motor is out of a predetermined strong field allowable torque range, the mode control unit is configured to control so that the field adjustment unit does not execute the strong field control. Is preferred.

  Here, in the rectangular wave control, harmonic components other than the fundamental component included in the current flowing through the coil are likely to be large. Therefore, depending on the value of the target torque of the AC motor, it may not be appropriate to shift to rectangular wave control by performing strong field control. According to this configuration, by restricting the torque range that allows the strong field control to be performed, the strong field is performed only in a state where it is appropriate to shift to the rectangular wave control, and the rectangular wave control is appropriately executed. Can do.

  The mode control unit is configured to control the field adjustment unit so as to change the field adjustment command value from a current value toward zero at a constant change rate when ending the strong field control. This is preferable.

  According to this configuration, when the strong field control is terminated, the field adjustment command value is changed so as to decrease toward zero at a constant speed, so that the voltage index can be gradually decreased. As a result, it is possible to appropriately execute the strong field / pulse width modulation control mode while the voltage index gradually decreases from the rectangular wave threshold value. Therefore, the field adjustment command value and the voltage index can be prevented from changing suddenly when the strong field control is finished, and the rapid change and overshoot of the current flowing in the coil of the AC motor can be suppressed, and the output of the AC motor can be controlled. Generation of torque vibration can be suppressed.

  The voltage command determination unit performs feedback control on the adjusted current command value based on an actual current value that is an actual value of the current supplied from the DC / AC conversion unit to the AC motor, and the voltage command It is preferable to determine the value.

  According to this configuration, the voltage command value can be appropriately determined by current feedback control based on the deviation between the actual current value detected by the current sensor or the like and the current command value adjusted by the field adjustment command value. it can.

It is a circuit diagram which shows the structure of the electric motor drive device which concerns on embodiment of this invention. It is a functional block diagram of a control device concerning an embodiment of the present invention. It is a figure which shows the example of the voltage control area | region map which concerns on embodiment of this invention. It is a figure which shows the example of the basic d-axis electric current command value map which concerns on embodiment of this invention. It is a figure which shows the example of the q-axis current command value map which concerns on embodiment of this invention. It is a figure which shows the example of the conversion map used in the integral input adjustment part which concerns on embodiment of this invention. It is a conceptual diagram which shows the derivation | leading-out method of the rotational speed threshold value which concerns on embodiment of this invention. It is a conceptual diagram which shows the derivation | leading-out method of the adjustment command threshold value which concerns on embodiment of this invention. It is a flowchart which shows the flow of operation | movement of the control apparatus which concerns on embodiment of this invention. It is a figure which shows an example of the change of d-axis current command value and q-axis current command value accompanying the change of a target torque and rotational speed in the control apparatus which concerns on embodiment of this invention.

  First, an embodiment of the present invention will be described based on the drawings. As shown in FIG. 1, in the present embodiment, the motor drive device 1 is a synchronous motor 4 (IPMSM, hereinafter simply referred to as “motor 4”) having an embedded magnet structure as an AC motor that operates by three-phase AC. The case where it is comprised as an apparatus to drive is demonstrated as an example. The electric motor 4 is configured to operate as a generator as required. The electric motor 4 is used as a driving force source for an electric vehicle or a hybrid vehicle, for example. The electric motor drive device 1 includes an inverter 6 that converts a direct current voltage Vdc into alternating current and supplies it to the electric motor 4. And in this embodiment, as shown in FIG. 2, the control apparatus 2 controls the electric motor drive apparatus 1, and performs the current feedback control of the electric motor 4 using a vector control method. At this time, the controller 2 is configured to execute pulse width modulation (hereinafter referred to as “PWM”) control and rectangular wave control as voltage waveform control. Further, the control device 2 performs, as field adjustment control, normal field control without adjusting the basic current command values Idb and Iqb determined based on the target torque TM, and a basic current command so as to weaken the field flux of the motor 4. The field weakening control for adjusting the values Idb and Iqb and the field strengthening control for adjusting the basic current command values Idb and Iqb so as to increase the field magnetic flux of the electric motor 4 can be executed. Then, in the configuration in which the control device 2 performs the strong field control and the rectangular wave control based on the modulation factor M as the voltage index, the control device 2 can appropriately end the strong field control according to the operating state of the electric motor 4. It is characterized in that it is constructed. Hereinafter, the electric motor drive device 1 and its control device 2 according to the present embodiment will be described in detail.

1. Configuration of Electric Motor Drive Device First, the configuration of an electric motor drive device 1 according to the present embodiment will be described with reference to FIG. The electric motor drive device 1 includes an inverter 6 that converts a DC voltage Vdc into an AC voltage and supplies the AC voltage to the electric motor 4. In addition, the electric motor drive device 1 includes a DC power source 3 that generates a DC voltage Vdc, and a smoothing capacitor C1 that smoothes the DC voltage Vdc from the DC power source 3. As the DC power source 3, for example, various secondary batteries such as a nickel hydride secondary battery and a lithium ion secondary battery, a capacitor, or a combination thereof is used. The DC voltage Vdc, which is the voltage of the DC power supply 3, is detected by the voltage sensor 41 and output to the control device 2.

  The inverter 6 is a device for converting a direct current direct current voltage Vdc into an alternating current voltage and supplying the alternating current voltage to the electric motor 4, and corresponds to a direct current to alternating current converter in the present invention. The inverter 6 includes a plurality of sets of switching elements E1 to E6 and diodes D1 to D6. Here, the inverter 6 is a pair of switching elements for each of the phases (U phase, V phase, W phase) of the electric motor 4, specifically, a U-phase upper arm element E1 and a U-phase lower phase. The arm element E2, the V-phase upper arm element E3, the V-phase lower arm element E4, the W-phase upper arm element E5, and the W-phase lower arm element E6 are provided. In these examples, IGBTs (insulated gate bipolar transistors) are used as the switching elements E1 to E6. The emitters of the upper arm elements E 1, E 3, E 5 for each phase and the collectors of the lower arm elements E 2, E 4, E 6 are connected to the coils of the respective phases of the electric motor 4. The collectors of the upper arm elements E 1, E 3, E 5 for each phase are connected to the system voltage line 51, and the emitters of the lower arm elements E 2, E 4, E 6 for each phase are connected to the negative line 52. In addition, diodes D1 to D6 that function as freewheeling diodes are connected in parallel to the switching elements E1 to E6, respectively. As the switching elements E1 to E6, power transistors having various structures such as a bipolar type, a field effect type, and a MOS type can be used in addition to the IGBT.

  Each of the switching elements E <b> 1 to E <b> 6 performs an on / off operation according to the switching control signals S <b> 1 to S <b> 6 output from the control device 2. Thereby, the inverter 6 converts the DC voltage Vdc into an AC voltage and supplies it to the electric motor 4 to cause the electric motor 4 to output a torque corresponding to the target torque TM. At this time, each of the switching elements E1 to E6 performs a switching operation according to PWM control or rectangular wave control described later in accordance with the switching control signals S1 to S6. In the present embodiment, the switching control signals S1 to S6 are gate drive signals that drive the gates of the switching elements E1 to E6. On the other hand, when the electric motor 4 functions as a generator, the inverter 6 converts the generated AC voltage into a DC voltage and supplies it to the system voltage line 51. Each phase current flowing through the coils of each phase of the electric motor 4, specifically, the U-phase current Iur, the V-phase current Ivr, and the W-phase current Iwr is detected by the current sensor 42 and output to the control device 2.

  Further, the magnetic pole position θ at each time point of the rotor of the electric motor 4 is detected by the rotation sensor 43 and output to the control device 2. The rotation sensor 43 is configured by, for example, a resolver. Here, the magnetic pole position θ represents the rotation angle of the rotor on the electrical angle. The target torque TM of the electric motor 4 is input to the control device 2 as a request signal from another control device such as a vehicle control device (not shown). That is, the target torque TM is a command value (torque command value) for output torque to the electric motor 4.

2. Configuration of Control Device Next, the configuration of the control device 2 shown in FIG. 1 will be described in detail with reference to FIGS. Each functional unit of the control device 2 described below is based on hardware and / or software (program) or both for performing various processes on input data using a logic circuit such as a microcomputer as a core member. It is configured. As described above, the target torque TM and the magnetic pole position θ are input to the control device 2. Furthermore, the control device 2 also receives a U-phase current Iur, a V-phase current Ivr, and a W-phase current Iwr. Therefore, as shown in FIG. 2, the control device 2 is based on the target torque TM, the magnetic pole position θ, the rotational speed ω of the motor 4 derived from the magnetic pole position θ, and the phase currents Iur, Ivr, and Iwr. Then, current feedback control using a vector control method is performed, and voltage command values Vd and Vq which are command values of voltages supplied to the motor 4 are determined. Then, based on the voltage command values Vd and Vq, switching control signals S1 to S6 for driving the inverter 6 are generated and output, and drive control of the electric motor 4 is performed via the inverter 6.

2-1. Control Mode In the present embodiment, the control device 2 is configured to be able to execute PWM control and rectangular wave control with respect to voltage waveform control performed by controlling the inverter 6 based on the voltage command values Vd and Vq. Further, regarding field control for adjusting the field magnetic flux of the motor 4 by determining the d-axis current adjustment command value ΔId as the field adjustment command value for the basic current command values Idb and Iqb, normal field control, strong field control, and field weakening control. Is configured to run. The control device 2 selectively executes any one of a plurality of control modes by combining the voltage waveform control and the field control.

  In the PWM control, on / off of the switching elements E1 to E6 of the inverter 6 is controlled based on the three-phase AC voltages Vu, Vv, and Vw (see FIG. 2) based on the voltage command values Vd and Vq. Specifically, the PWM waveform that is the output voltage waveform of the inverter 6 of each phase of U, V, and W includes a high level period during which the upper arm elements E1, E3, and E5 are on, and the lower arm elements E2 and E4. The duty ratio of each pulse is controlled so that the fundamental wave component is substantially sinusoidal in a certain period, and is composed of a set of pulses composed of a low level period in which E6 is in the ON state. In the present embodiment, the PWM control includes two control methods of normal PWM control and overmodulation PWM control.

  The normal PWM control is PWM control in which the AC voltage waveforms Vu, Vv, and Vw are less than or equal to the amplitude of the carrier waveform. As such normal PWM control, sine wave PWM control is typical, but in this embodiment, a space vector PWM (applying a neutral point bias voltage to the fundamental wave of each phase of sine wave PWM control. Space Vector PWM (hereinafter referred to as “SVPWM”) control is used. In the SVPWM control, the PWM waveform is directly generated by digital calculation without being compared with the carrier, but even in this case, the AC voltage waveforms Vu, Vv, and Vw are less than the amplitude of the virtual carrier waveform. In the present invention, such a method of generating a PWM waveform without using a carrier is also included in normal PWM control or overmodulation PWM control in comparison with the amplitude of a virtual carrier waveform. Assuming that the ratio of the effective value of the fundamental wave component of the output voltage waveform of the inverter 6 to the DC voltage Vdc is the modulation factor M (see formula (4) described later), in the SVPWM control as the normal PWM control, the modulation factor M is “0”. It can be changed in the range of “˜0.707”.

  The overmodulation PWM control is PWM control in which the amplitude of the AC voltage waveforms Vu, Vv, and Vw exceeds the amplitude of the carrier waveform (triangular wave). In the overmodulation PWM control, the waveform of the fundamental wave component of the output voltage waveform of the inverter 6 is distorted by making the duty ratio of each pulse large on the peak side of the fundamental wave component and smaller on the valley side than in the normal PWM control, Control is performed so that the amplitude is larger than that of the normal PWM control. In the overmodulation PWM control, the modulation factor M can be changed in a range of “0.707 to 0.78”.

  The rectangular wave control is rotation synchronous control in which each switching element E1 to E6 is turned on and off once per electrical angle cycle of the electric motor 4, and one pulse is output per electrical angle half cycle for each phase. is there. In other words, in the rectangular wave control, the output voltage waveform of the inverter 6 of each phase of U, V, and W alternately appears between the high level period and the low level period once per cycle, and these high level periods. And a low-level period are controlled so as to be a rectangular wave of 1: 1. At this time, the output voltage waveforms of the respective phases are outputted with a phase shift of 120 °. Thus, the rectangular wave control causes the inverter 6 to output a rectangular wave voltage. In the rectangular wave control, the modulation factor M is fixed to “0.78” which is the maximum modulation factor Mmax. That is, when the modulation factor M reaches the maximum modulation factor Mmax, rectangular wave control is executed. For this reason, in this embodiment, the rectangular wave threshold value Mb, which is the threshold value of the modulation factor M for executing the rectangular wave control, is set to the maximum modulation factor Mmax.

  As described above, the field control in the present embodiment includes normal field control, strong field control, and weak field control. As will be described later, the current command determination unit 7 determines basic current command values Idb and Iqb, which are command values of the current supplied from the inverter 6 to the motor 4 based on the target torque TM of the motor 4. The field control is a control for adjusting the field magnetic flux of the electric motor 4 by the field adjustment command value (d-axis current adjustment command value ΔId) for adjusting the basic current command values Idb and Iqb determined as described above. Specifically, the current command determination unit 7 determines a basic d-axis current command value Idb and a basic q-axis current command value Iqb as basic current command values based on the target torque TM. Here, in the current vector control method, the d-axis is set in the field magnetic flux direction, and the q-axis is set in a direction advanced by π / 2 in electrical angle with respect to the field direction. Therefore, the field magnetic flux of the electric motor 4 can be adjusted by appropriately determining the d-axis current adjustment command value ΔId for adjusting the basic d-axis current command value Idb as the field adjustment command value.

  As will be described later, the current command determination unit 7 determines the basic current command values Idb and Iqb so as to perform maximum torque control. Here, the maximum torque control is control for adjusting the current phase so that the output torque of the electric motor 4 becomes maximum with respect to the same current. In this maximum torque control, torque can be generated most efficiently with respect to the current flowing through the armature coil of the electric motor 4. The current phase is the phase with respect to the q-axis of the combined vector of the d-axis current command value and the q-axis current command value. The normal field control is field control that does not adjust the basic current command values Idb and Iqb determined by the current command determination unit 7. That is, in the normal field control, the d-axis current adjustment command value ΔId is set to zero (ΔId = 0) so as not to adjust the basic d-axis current command value Idb. Therefore, in the present embodiment, the control device 2 performs the maximum torque control during the execution of the normal field control. In other words, the normal field control according to the present embodiment is maximum torque control.

  The strong field control is field control that adjusts the basic current command values Idb and Iqb so as to increase the field magnetic flux of the electric motor 4 as compared with the normal field control (maximum torque control). That is, the strong field control is a control that adjusts the current phase so that a magnetic flux in a direction that strengthens the field magnetic flux of the electric motor 4 is generated from the armature coil. Here, in the strong field control, the d-axis current adjustment command value ΔId is set so as to delay the current phase compared to the normal field control. Specifically, in the strong field control, the d-axis current adjustment command value ΔId is set to a positive value (ΔId> 0) so as to change (increase) the basic d-axis current command value Idb in the positive direction.

  The field weakening control is field control for adjusting the basic current command values Idb and Iqb so as to weaken the field magnetic flux of the electric motor 4 as compared with the normal field control (maximum torque control). That is, the field weakening control is a control for adjusting the current phase so that the magnetic flux in the direction of weakening the field magnetic flux of the electric motor 4 is generated from the armature coil. Here, in the field weakening control, the d-axis current adjustment command value ΔId is set so that the current phase is advanced as compared with the normal field control. Specifically, in the field weakening control, the d-axis current adjustment command value ΔId is set to a negative value (ΔId <0) so as to change (decrease) the basic d-axis current command value Idb in the negative direction.

  FIG. 3 shows an example of a voltage control region map 34 (see FIG. 2) that defines regions in which each control mode is executed in the operable region of the electric motor 4 defined by the rotational speed ω and the target torque TM. FIG. As shown in this figure, in the present embodiment, the control device 2 includes a normal field / PWM control mode A1 that executes PWM control together with normal field control, and a strong field / PWM control mode that executes PWM control together with strong field control. A2, a strong field / rectangular wave control mode A3 that executes rectangular wave control together with strong field control, and a field weakening / rectangular wave control mode A5 that executes rectangular wave control together with weak field control are configured to be executable. Further, when the control device 2 shifts to the weak field / rectangular wave control mode A5 without going through the strong field / PWM control mode A2 and the strong field / rectangular wave control mode A3, the normal field / Between the PWM control mode A1 and the field weakening / rectangular wave control mode A5, a field weakening / PWM control mode A4 that executes PWM control together with field weakening control is configured to be executable. A region F shown in the map of FIG. 3 is a strong field control region where the strong field control is executed. In the strong field control region F, the strong field / rectangular wave control mode A3 is basically executed, but the strong field / rectangular wave control mode A3 and the strong field are changed during the transition between the strong field / rectangular wave control mode A3 and another mode. The magnetism / PWM control mode A2 is executed.

  Further, as described above, in the present embodiment, two voltage waveform controls of normal PWM control and overmodulation PWM control are executed as PWM control. Therefore, the normal field / PWM control mode A1 includes a normal field / normal PWM control mode A1a that executes normal PWM control together with normal field control, and a normal field / overmodulation PWM control that executes overmodulation PWM control together with normal field control. Mode A1b. On the other hand, the strong field / PWM control mode A2 is a strong field / overmodulation PWM control mode A2b in which overmodulation PWM control is executed together with the strong field control. Further, here, the field weakening / PWM modulation mode A4 is a field weakening / overmodulation PWM control mode A4a in which the overmodulation PWM control is executed together with the field weakening control.

  In the example of the voltage control region shown in FIG. 3, the curves L1 to L3 are all determined by the rotational speed ω of the motor 4 and the target torque TM when the modulation factor M is a certain value during normal field control (maximum torque control). Is a line. A curve L1 is a line at which the modulation factor M during normal field control becomes the maximum modulation factor Mmax (= 0.78). A curve L2 is a line in which the modulation factor M during the normal field control becomes an overmodulation threshold value Mo (= 0.707) set to a boundary value between the normal PWM control and the overmodulation PWM control. In the present embodiment, a strong field threshold value Ms described later is set to coincide with the overmodulation threshold value Mo. A curve L3 is a line in which the modulation factor M during normal field control becomes a value (for example, 0.76) set between the overmodulation threshold Mo and the maximum modulation factor Mmax.

  Incidentally, the induced voltage of the electric motor 4 increases as the rotational speed ω increases, and the AC voltage (hereinafter referred to as “necessary voltage”) required to drive the electric motor 4 also increases. When the necessary voltage exceeds the maximum AC voltage that can be output from the inverter 6 by converting the DC voltage Vdc at that time (hereinafter referred to as “maximum output voltage”), a necessary current flows through the coil. As a result, the electric motor 4 cannot be appropriately controlled. Therefore, the field weakening / rectangular wave control mode A5 is executed in the region on the higher rotation side than the curve L1 where the modulation factor M representing the required voltage of the motor 4 with respect to the maximum output voltage based on the DC voltage Vdc reaches the maximum modulation factor Mmax. The The necessary voltage and the maximum output voltage can be compared with each other as the effective value of the AC voltage.

  Further, in the present embodiment, when a predetermined condition is satisfied even when the modulation factor M is lower than the maximum modulation factor Mmax, the strong field / rectangular wave control mode A3 for executing the rectangular wave control together with the strong field control is executed. To do. Further, at the time of transition between the strong field / rectangular wave control mode A3 and another mode, the current command values Id and Iq after the adjustment change abruptly, so that the current flowing through the coil of the motor 4 changes suddenly or overshoots. In order to suppress the chute and to suppress the vibration of the output torque of the electric motor 4, the strong field / PWM control mode A2 is executed. The strong field control is basically executed to perform the rectangular wave control while outputting the torque corresponding to the target torque TM to the motor 4 in a state where the modulation rate M is lower than the maximum modulation rate Mmax if the normal field control is performed. Is done.

  As shown in FIG. 3, the strong field control region F is set within the strong field allowable torque range TMR defined for the target torque TM. That is, the strong field control region F is within the strong field allowable torque range TMR, and the modulation rate M during the normal field control is the strong field threshold value Ms (here, coincides with the overmodulation threshold value Mo, curve L2). To the maximum modulation rate Mmax (curve L1) (Ms ≦ M <Mmax). When the operating point determined by the rotational speed ω of the motor 4 and the target torque TM moves from the normal field / PWM control mode A1 region and enters the strong field control region F, the control device 2 Control is performed from the PWM control mode A1 to the strong field / rectangular wave control mode A3 via the strong field / PWM control mode A2. On the other hand, when the operating point of the electric motor 4 moves from the strong field control region F and enters the normal field / PWM control mode A1, the control device 2 controls the strong field / rectangular wave control mode A3. To shift to the normal field / PWM control mode A1 through the strong field / PWM control mode A2. When the operating point of the electric motor 4 remains in the strong field control region F, the execution state of the strong field / rectangular wave control mode A3 is continued. By setting such a strong field control region F, rectangular wave control in the operable region of the electric motor 4 is executed as compared with the conventional case of having only the weak field / rectangular wave control mode A5. The area to be expanded is enlarged. In FIG. 3, a broken line that divides the strong field control region F is a region in which the strong field / PWM control mode A2 is executed when the rotation speed ω or the target torque TM of the motor 4 changes at a predetermined change speed. 2 shows an example of a boundary where the strong field / rectangular wave control mode A3 is switched. The position of this boundary differs depending on the rotational speed ω or the change speed of the target torque TM.

  The normal field / normal PWM control mode A1a is executed in the region on the lower rotation side than the curve L2. In addition, outside the strong field allowable torque range TMR, the normal field / overmodulation PWM control mode A1b is executed in a region on the higher rotation side than the curve L2 and on the lower rotation side than the curve L3. Further, outside of the strong field allowable torque range TMR, the field weakening / overmodulation PWM control mode A4a (field weakening / PWM control mode A4) is higher in the region higher than the curve L3 and lower than the curve L1. Is executed. In this field weakening / overmodulation PWM control mode A4a, the current command values Id and Iq after adjustment are changed from the normal field / overmodulation PWM control mode A1b to the state in which the rectangular wave control is performed together with the field weakening control suddenly. It is executed to suppress a rapid change.

2-2. Next, the functional units of the control device 2 will be described based on the functional block diagram of the control device 2 shown in FIG. As shown in FIG. 2, the target torque TM is input to the d-axis current command value deriving unit 21. The d-axis current command value deriving unit 21 derives a basic d-axis current command value Idb based on the input target torque TM. Here, the basic d-axis current command value Idb corresponds to a command value for the d-axis current when maximum torque control is performed. In the present embodiment, the d-axis current command value deriving unit 21 derives a basic d-axis current command value Idb corresponding to the value of the target torque TM using the basic d-axis current command value map shown in FIG. In the illustrated example, when a value of “TM1” is input as the target torque TM, the d-axis current command value deriving unit 21 derives “Id1” as the basic d-axis current command value Idb accordingly. To do. Similarly, when the values “TM3” and “TM5” are input as the target torque TM, the d-axis current command value deriving unit 21 sets “Id3” and “Id5” as the basic d-axis current command value Idb. Derived respectively. The basic d-axis current command value Idb derived in this way is input to the adder 23. The adder 23 is further supplied with a d-axis current adjustment command value ΔId derived by an integrator 32 described later. The adder 23 adds the d-axis current adjustment command value ΔId to the basic d-axis current command value Idb and derives the adjusted d-axis current command value Id as shown in the following formula (1).
Id = Idb + ΔId (1)

  The target torque TM and the d-axis current adjustment command value ΔId are input to the q-axis current command value deriving unit 22. The q-axis current command value deriving unit 22 derives the adjusted q-axis current command value Iq based on the input target torque TM and the d-axis current adjustment command value ΔId. In the present embodiment, the q-axis current command value deriving unit 22 uses the q-axis current command value map shown in FIG. The command value Iq is derived. In FIG. 5, the thin solid line is an equal torque line 61 indicating a combination of values of the d-axis current and the q-axis current for outputting the torques TM1 to TM5, and the thick solid line is for performing maximum torque control. It is the maximum torque control line 62 which shows the value of d-axis current and q-axis current. In FIG. 5, a thick one-dot chain line is a voltage limiting ellipse 63 that indicates a range of values that can be taken by the d-axis current and the q-axis current that are limited by the rotational speed ω of the electric motor 4 and the DC voltage Vdc. The diameter of the voltage limiting ellipse 63 is inversely proportional to the rotational speed ω of the electric motor 4 and proportional to the DC voltage Vdc. When the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq take values on the voltage limit ellipse 63, the modulation factor M becomes the maximum modulation factor Mmax (= 0.78). At this time, the control device 2 causes the voltage waveform control unit 10 to perform rectangular wave control. Further, a strong field control region F shown by hatching in FIG. 5 indicates a region where the strong field / PWM control mode A2 and the strong field / rectangular wave control mode A3 are executed. The upper limit of the strong field control region F is defined by the point where the maximum torque control line 62 intersects the voltage limit ellipse 63. As will be described later, the strong field control starts when the modulation factor M during the normal field control reaches the strong field threshold value Ms, and ends when a predetermined strong field end condition is satisfied. Therefore, the lower limit of the strong field control region F is defined by the strong field threshold value Ms and the strong field end condition.

  In the illustrated example, when a value of “TM1” is input as the target torque TM, the q-axis current command value deriving unit 22 determines whether the equal torque line 61 and the maximum torque control line 62 of the target torque TM = TM1. “Iq1” that is the value of the q-axis current at the intersection is derived as the basic q-axis current command value Iqb. Here, the basic q-axis current command value corresponds to the command value of the q-axis current when maximum torque control is performed. In this case, both the weak field control and the strong field control are not performed, and the d-axis current adjustment command value ΔId input from the integrator 32 described later is zero (ΔId = 0). Therefore, the adjusted q-axis current command value Iq is the same value as the basic q-axis current command value Iqb. At this time, the control device 2 executes the normal field / PWM control mode A1.

  When a value of “TM3” is input as the target torque TM, the q-axis current command value deriving unit 22 determines the q at the intersection of the equal torque line 61 and the maximum torque control line 62 of the target torque TM = TM3. The value of the axis current “Iq3” is derived as the basic q-axis current command value Iqb. At this time, since the basic d-axis current command value Idb and the basic q-axis current command value Iqb are within the strong field control region F, the strong field control is performed. In this case, a positive value as the d-axis current adjustment command value ΔId, here “ΔId1” (ΔId1> 0), is input from the integrator 32 described later. Therefore, the q-axis current command value deriving unit 22 is the value of the q-axis current on the voltage limiting ellipse 63 moved by “ΔId1” along the equal torque line 61 of the target torque TM = TM3 in the positive direction of the d-axis. “Iq4” is derived as an adjusted q-axis current command value Iq. At this time, the control device 2 executes the strong field / rectangular wave control mode A3.

  When a value of “TM5” is input as the target torque TM, the q-axis current command value deriving unit 22 determines the q at the intersection of the equal torque line 61 and the maximum torque control line 62 of the target torque TM = TM5. The value of the axis current “Iq5” is derived as the basic q-axis current command value Iqb. At this time, since the basic d-axis current command value Idb and the basic q-axis current command value Iqb are outside the voltage limit ellipse 63, field weakening control is performed. In this case, a negative value as the d-axis current adjustment command value ΔId, here, “−ΔId2” (−ΔId2 <0) is input from the integrator 32 described later. Therefore, the q-axis current command value deriving unit 22 is the value of the q-axis current on the voltage limiting ellipse 63 moved by “−ΔId2” in the negative direction of the d-axis along the equal torque line 61 of the target torque TM = TM5. A certain “Iq6” is derived as an adjusted q-axis current command value Iq. At this time, the control device 2 executes the field weakening / rectangular wave control mode A5.

  The d-axis current values (Id1, Id3, Id5) corresponding to the basic q-axis current command values Iqb (Iq1, Iq3, Iq5) obtained from the q-axis current command value map of FIG. 5 are the basic values shown in FIG. This coincides with the basic d-axis current command value Idb obtained using the d-axis current command value map. Therefore, the basic d-axis current command value Idb can be obtained from the map shown in FIG. In the present embodiment, a d-axis current command value deriving unit 21 and a q-axis current command value deriving unit 22 that determine the basic d-axis current command value Idb and the basic q-axis current command value Iqb based on the target torque TM of the electric motor 4. However, this constitutes the current command determination unit 7 in the present invention. The basic d-axis current command value Idb and the basic q-axis current command value Iqb are the basic current command values in the present invention, which are command values for the current supplied from the inverter 6 to the motor 4.

  The adjusted d-axis current command value Id and the adjusted q-axis current command value Iq derived as described above are input to the current control unit 24. Further, the current control unit 24 receives the actual d-axis current Idr and the actual q-axis current Iqr from the three-phase to two-phase conversion unit 27, and receives the rotation speed ω of the electric motor 4 from the rotation speed deriving unit 28. The actual d-axis current Idr and the actual q-axis current Iqr correspond to the actual value of the current supplied from the inverter 6 to the electric motor 4, and the U-phase current Iur detected by the current sensor 42 (see FIG. 1), Based on the V-phase current Ivr and the W-phase current Iwr and the magnetic pole position θ detected by the rotation sensor 43 (see FIG. 1), the three-phase to two-phase conversion unit 27 performs three-phase to two-phase conversion. . The rotational speed ω of the electric motor 4 is derived by the rotational speed deriving unit 28 based on the magnetic pole position θ detected by the rotation sensor 43 (see FIG. 1).

  The current control unit 24 performs feedback control on the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq based on the actual d-axis current Idr and the actual q-axis current Iqr, and the voltage command values Vd, Vq To decide. For this purpose, the current control unit 24 calculates a d-axis current deviation δId that is a deviation between the adjusted d-axis current command value Id and the actual d-axis current Idr, and an adjusted q-axis current command value Iq and the actual q-axis current Iqr. Q-axis current deviation δIq, which is a deviation of The current control unit 24 performs a proportional-integral control calculation (PI control calculation) based on the d-axis current deviation δId to derive a basic d-axis voltage command value Vzd, and also performs a proportional integration based on the q-axis current deviation δIq. A control calculation is performed to derive a basic q-axis voltage command value Vzq. Note that it is also preferable to perform a proportional integral derivative control calculation (PID control calculation) instead of these proportional integral control calculations.

Then, as shown in the following equation (2), the current control unit 24 performs adjustment to subtract the q-axis armature reaction Eq from the basic d-axis voltage command value Vzd to derive the d-axis voltage command value Vd. To do.
Vd = Vzd-Eq
= Vzd-ω · Lq · Iqr (2)
As shown in this equation (2), the q-axis armature reaction Eq is derived based on the rotational speed ω of the electric motor 4, the actual q-axis current Iqr, and the q-axis inductance Lq.

Furthermore, the current control unit 24 adds the induced voltage Em caused by the d-axis armature reaction Ed and the armature interlinkage magnetic flux of the permanent magnet to the basic q-axis voltage command value Vzq as shown in the following equation (3). Thus, the q-axis voltage command value Vq is derived.
Vq = Vzq + Ed + Em
= Vzq + ω · Ld · Idr + ω · Mif (3)
As shown in this equation (3), the d-axis armature reaction Ed is derived based on the rotational speed ω of the electric motor 4, the actual d-axis current Idr, and the d-axis inductance Ld. The induced voltage Em is derived based on the induced voltage constant MIf determined by the effective value of the armature linkage flux of the permanent magnet and the rotational speed ω of the motor 4.

  In the present embodiment, the d-axis voltage command value Vd and the q-axis voltage command value Vq correspond to the voltage command value in the present invention. Then, the adjusted current command values Id and Iq after the field adjustment is performed on the basic current command values Idb and Iqb by the d-axis current adjustment command value ΔId, the rotational speed ω of the motor 4, and the actual d-axis current. Based on Idr and actual q-axis current Iqr, voltage command values Vd and Vq are determined. Therefore, the current control unit 24 constitutes the voltage command determination unit 9 in the present invention.

  The voltage waveform control unit 10 controls the inverter 6 based on the voltage command values Vd and Vq, and executes voltage waveform control including at least PWM control and rectangular wave control. In the present embodiment, the voltage waveform controller 10 selectively executes any one of normal PWM control, overmodulation PWM control, and rectangular wave control. In the present embodiment, the voltage waveform controller 10 executes rectangular wave control when the modulation factor M is equal to or greater than the rectangular wave threshold Mb (= 0.78) in accordance with a command from the mode controller 5 described later. To do. Further, when the modulation factor M is less than the rectangular wave threshold Mb, the voltage waveform controller 10 further performs normal PWM control or overmodulation PWM control based on the overmodulation threshold Mo (= 0.707). Execute. The voltage waveform control unit 10 includes a three-phase to two-phase conversion unit 25 and a control signal generation unit 26.

  The d-axis voltage command value Vd and the q-axis voltage command value Vq are input to the two-phase / three-phase conversion unit 25. Also, the magnetic pole position θ detected by the rotation sensor 43 (see FIG. 1) is input to the two-phase / three-phase converter 25. The two-phase / three-phase conversion unit 25 performs two-phase / three-phase conversion on the d-axis voltage command value Vd and the q-axis voltage command value Vq using the magnetic pole position θ to obtain a three-phase AC voltage command value, that is, a U-phase. A voltage command value Vu, a V-phase voltage command value Vv, and a W-phase voltage command value Vw are derived. However, since the waveforms of these AC voltage command values Vu, Vv, and Vw are different for each control mode, the two-phase / three-phase conversion unit 25 has AC voltage command values Vu, Vv, and Vw having different voltage waveforms for each control mode. Is output to the control signal generator 26. Specifically, when the two-phase / three-phase conversion unit 25 receives a normal PWM control execution command from the mode control unit 5 to be described later, the AC voltage command value Vu of the AC voltage waveform corresponding to the normal PWM control. , Vv, Vw are output. Here, since the normal PWM control is SVPWM control, AC voltage command values Vu, Vv, and Vw are output according to the AC voltage waveform for the SVPWM control. In addition, when the two-phase / three-phase conversion unit 25 receives an overmodulation PWM control execution command from the mode control unit 5, the AC voltage command values Vu, Vv, Vw is output. In addition, when the two-phase / three-phase conversion unit 25 receives a rectangular wave control execution command from the mode control unit 5, the two-phase / three-phase conversion unit 25 converts the AC voltage command values Vu, Vv, and Vw of the AC voltage waveform according to the rectangular wave control. Output. Here, the AC voltage command values Vu, Vv, and Vw when executing the rectangular wave control can be set as command values for the on / off switching phases of the switching elements E1 to E6 of the inverter 6. This command value corresponds to the on / off control signal of each of the switching elements E1 to E6, and is a command value that represents the phase of the magnetic pole position θ that represents the timing for switching on or off of each of the switching elements E1 to E6.

  The control signal generator 26 receives the U-phase voltage command value Vu, the V-phase voltage command value Vv, and the W-phase voltage command value Vw generated by the three-phase / two-phase converter 25. The control signal generator 26 generates switching control signals S1 to S6 for controlling the switching elements E1 to E6 of the inverter 6 shown in FIG. 1 according to the AC voltage command values Vu, Vv, and Vw. The inverter 6 performs on / off operations of the switching elements E1 to E6 according to the switching control signals S1 to S6. Thereby, PWM control (normal PWM control or overmodulation PWM control) or rectangular wave control of the electric motor 4 is performed.

The modulation factor deriving unit 29 receives the d-axis voltage command value Vd and the q-axis voltage command value Vq derived by the current control unit 24. Further, the value of the DC voltage Vdc detected by the voltage sensor 41 is input to the modulation factor deriving unit 29. The modulation rate deriving unit 29 derives the modulation rate M based on these values according to the following equation (4).
M = √ (Vd ² + Vq ² ) / Vdc (4)
In the present embodiment, the modulation factor M is the ratio of the effective value of the fundamental wave component of the output voltage waveform of the inverter 6 to the DC voltage Vdc. Here, the effective value of the three-phase line voltage is the value of the DC voltage Vdc. Derived as a divided value. In the present embodiment, the modulation factor M corresponds to a voltage index representing the magnitudes of the voltage command values Vd and Vq with respect to the DC voltage Vdc at that time. As described above, the maximum value of the modulation factor M (maximum modulation factor Mmax) is “0.78” corresponding to the modulation factor M when the rectangular wave control is executed. Here, the maximum modulation rate Mmax is also the rectangular wave threshold value Mb.

The subtracter 30 receives the modulation factor M derived by the modulation factor deriving unit 29 and a predetermined command modulation factor MT. In the present embodiment, the command modulation rate MT is set to the maximum modulation rate Mmax (= 0.78). The subtracter 30 derives a modulation factor deviation ΔM obtained by subtracting the command modulation factor MT from the modulation factor M as shown in the following equation (5).
ΔM = M−MT (5)
In the present embodiment, the modulation factor deviation ΔM represents the degree to which the voltage command values Vd and Vq exceed the maximum AC voltage value that can be output by the DC voltage Vdc at that time. Therefore, the modulation factor deviation ΔM substantially functions as a voltage shortage index that represents the degree of shortage of the DC voltage Vdc.

  A modulation factor deviation ΔM derived by the subtracter 30 is input to the integral input adjustment unit 31. The integral input adjustment unit 31 performs a predetermined adjustment on the value of the modulation factor deviation ΔM, and outputs an adjustment value Y that is the adjusted value to the integrator 32. FIG. 6 is a diagram showing an example of a conversion map used by the integral input adjustment unit 31. As shown in FIG. As shown in this figure, in this embodiment, the integral input adjustment unit 31 is in a state where the modulation factor deviation ΔM is equal to or greater than a predetermined strong field start deviation ΔMs (ΔMs <0) and less than zero (ΔMs ≦ ΔM <0). Outputs a positive adjustment value Y (Y> 0) and outputs a negative adjustment value Y (Y <0) when the modulation factor deviation ΔM is greater than zero (0 <ΔM), increasing the modulation factor deviation ΔM. In a state where the field start deviation ΔMs is less than (ΔM <ΔMs) and a state where the modulation factor deviation ΔM is zero (ΔM = 0), zero (Y = 0) is output as the adjustment value Y. More specifically, the integral input adjustment unit 31 increases as the modulation factor deviation ΔM increases in a state where the modulation factor deviation ΔM is equal to or greater than the strong field start deviation ΔMs and less than the intermediate deviation ΔMsm (ΔMs ≦ ΔM <ΔMsm). The adjustment value Y to be output is output. In this range, the relationship between the modulation factor deviation ΔM and the adjustment value Y can be expressed by a linear function. Thus, by setting the conversion map region in which the adjustment value Y increases as the modulation factor deviation ΔM increases, it is possible to suppress the d-axis current adjustment command value ΔId from rapidly increasing immediately after the strong field control is started. Therefore, the current command values Id and Iq after the adjustment based on the d-axis current adjustment command value ΔId abruptly change, thereby suppressing a sudden change or overshoot of the current flowing in the coil of the motor 4 and the output torque of the motor 4. Can be suppressed.

  Further, the integral input adjustment unit 31 outputs an adjustment value Y that decreases as the modulation factor deviation ΔM increases in a state where the modulation factor deviation ΔM is equal to or greater than the intermediate deviation ΔMsm (ΔMsm ≦ ΔM). In this range, the adjustment value Y is proportional to the modulation factor deviation ΔM, and the proportionality constant is a negative value. Here, the strong field start deviation ΔMs is a threshold value of the modulation factor deviation ΔM for starting the strong field control, and is set to a value less than zero. This strong field start deviation ΔMs constitutes the start condition of the strong field control together with the command modulation rate MT. Therefore, the strong field start deviation ΔMs is set so that the strong field threshold value Ms determined in combination with the command modulation factor MT (= 0.78) becomes an appropriate value. As described above, in this embodiment, the strong field threshold value Ms is set to coincide with the overmodulation threshold value Mo (= 0.707). Therefore, here, the strong field start deviation ΔMs is set to “−0.073” (= 0.707−0.78). The intermediate deviation ΔMsm is set to a value larger than the strong field start deviation ΔMs and less than zero, for example, “−0.035”. Thus, the strong field threshold value Ms is set to a value smaller than the rectangular wave threshold value Mb (equal to the command modulation rate MT and the maximum modulation rate Mmax in this embodiment). The strong field control is executed on condition that the modulation factor M is equal to or greater than the strong field threshold value Ms.

  As shown in FIG. 2, the adjustment value Y derived by the integration input adjustment unit 31 is input to the integrator 32. The integrator 32 integrates the adjustment value Y using a predetermined gain, and derives the integration value as a d-axis current adjustment command value ΔId. In the present embodiment, the d-axis current adjustment command value ΔId is an adjustment value of the basic current command values Idb and Iqb, and corresponds to a field adjustment command value for adjusting the field magnetic flux of the motor 4. The d-axis current adjustment command value ΔId is determined by the modulation factor deriving unit 29, the subtractor 30, the integral input adjusting unit 31, and the integrator 32. Therefore, in the present embodiment, the field adjustment unit 8 is configured by the modulation factor deriving unit 29, the subtractor 30, the integral input adjusting unit 31, and the integrator 32. Then, normal field control (maximum torque control), strong field control, or weak field control is selectively executed according to the d-axis current adjustment command value ΔId. Here, when the d-axis current adjustment command value ΔId is zero (ΔId = 0), normal field control (maximum torque control) is performed. When the d-axis current adjustment command value ΔId has a positive value (ΔId> 0), adjustment is performed to increase the field magnetic flux of the electric motor 4 with respect to the basic current command values Idb and Iqb. That is, when a strong field current that is a positive d-axis current adjustment command value ΔId flows, the field flux of the electric motor 4 is strengthened compared to the normal field control, and the strong field control is performed. When the d-axis current adjustment command value ΔId takes a negative value (ΔId <0), adjustment is performed to weaken the field magnetic flux of the electric motor 4 with respect to the basic current command values Idb and Iqb. That is, when a field weakening current that is a negative d-axis current adjustment command value ΔId flows, the field magnetic flux of the electric motor 4 is weakened compared to the normal field control, and the field weakening control is performed.

  As described above, when the modulation factor deviation ΔM is greater than the strong field start deviation ΔMs and less than zero (ΔMs ≦ ΔM <0), a positive value (Y> 0) is output as the adjustment value Y. The d-axis current adjustment command value ΔId derived by 32 increases (changes in the positive direction), and the d-axis current adjustment command value ΔId changes in the direction of increasing the field magnetic flux of the electric motor 4. In the state where the modulation factor deviation ΔM is greater than zero (0 <ΔM), a negative value (Y <0) is output as the adjustment value Y. Therefore, the d-axis current adjustment command value ΔId derived by the integrator 32 is output. Decreases (changes in the negative direction), and the d-axis current adjustment command value ΔId changes in the direction of weakening the field magnetic flux of the electric motor 4. When the modulation factor deviation ΔM is less than the strong field start deviation ΔMs (ΔM <ΔMs) and the modulation factor deviation ΔM is zero (ΔM = 0), zero (Y = 0) is output as the adjustment value Y. The d-axis current adjustment command value ΔId derived by the controller 32 is not changed, and the d-axis current adjustment command value ΔId is determined so as not to change the field magnetic flux of the electric motor 4.

  As described above, in the normal field control according to the present embodiment, the maximum torque control is performed to adjust the current phase so that the output torque of the motor 4 is maximized with respect to the same current. Therefore, as the d-axis current adjustment command value ΔId changes from the reference value (ΔId = 0) of the d-axis current adjustment command value ΔId for executing the normal field control in the direction in which the field magnetic flux of the motor 4 is increased, the same torque The adjusted current command values Id and Iq required for outputting the voltage increase, and the voltage command values Vd and Vq and the modulation factor M derived based on the current command values Id and Iq increase. In other words, the voltage command determination unit 9 increases the voltage command values Vd and Vq as the d-axis current adjustment command value ΔId increases (changes in the positive direction) from the reference value (ΔId = 0). The modulation factor deriving unit 29 increases the modulation factor M as the d-axis current adjustment command value ΔId increases from the reference value (ΔId = 0) (changes in the positive direction).

  The mode control unit 5 determines a control mode to be executed from among a plurality of control modes based on the operating state of the electric motor 4 including the rotational speed ω and the target torque TM and the DC voltage Vdc, and the field control is performed according to the control mode. Each part of the control apparatus 2 including the adjustment part 8 and the voltage waveform control part 10 is controlled. Further, the mode control unit 5 determines the strong field end condition during the execution of the strong field control, and also performs the strong field end control to end the strong field control when the strong field end condition is satisfied. . Here, as shown in FIG. 2, the rotational speed ω, the target torque TM, the DC voltage Vdc, the modulation factor M, and the d-axis current adjustment command value ΔId are input to the mode control unit 5, and the mode is based on these. The control operation of the control unit 5 is performed. In the present embodiment, the mode control unit 5 basically determines the control mode based on the voltage control region map 34 shown in FIG. Further, the mode control unit 5 determines the end of the strong field control based on the strong field end condition map shown in FIGS. 7 and 8 during execution of the strong field control. Details of the mode control unit 5 will be described below.

2-3. 3. Details of Mode Control Unit As shown in FIG. 3, the mode control unit 5 except the strong field control region F, the normal field / normal PWM control mode A1a as the rotational speed ω and the target torque TM of the motor 4 increase. The control mode is shifted in the order of the normal field / overmodulation PWM control mode A1b, the field weakening / overmodulation PWM control mode A4a, and the field weakening / rectangular wave control mode A5. As described above, the boundaries (curves L1, L2, and L3) between these control modes are set at positions where the modulation factor M is constant during normal field control (maximum torque control). In this, the curve L1 is set at a position where the modulation factor M during the normal field control becomes the maximum modulation factor Mmax (= 0.78), and the normal field control should be performed based on the rotational speed ω and the target torque TM. In the state where the derived modulation factor M exceeds the maximum modulation factor Mmax, the control device 2 executes the field weakening / rectangular wave control mode A5.

The strong field control region F is set within the strong field allowable torque range TMR defined for the target torque TM. The strong field control region F is a modulation rate when it is assumed that the normal field control is performed without performing the strong field control over the entire region excluding the weak field control region (the region in which the weak field / rectangular wave control mode A5 is executed). M is set to a region (Ms ≦ M <Mmax) from the strong field threshold value Ms (curve L2) to the maximum modulation rate Mmax (curve L1). Here, the strong field threshold value Ms is determined by setting both the command modulation factor MT and the strong field start deviation ΔMs. That is, in a situation where the modulation factor M gradually increases and approaches the command modulation factor MT, the integral input adjustment unit 31 has the modulation factor deviation ΔM greater than the field start deviation ΔMs (ΔMs <0) and less than zero as described above. A positive adjustment value Y (Y> 0) is output in the state (ΔMs ≦ ΔM <0). The modulation factor deviation ΔM is obtained by subtracting the command modulation factor MT from the modulation factor M, as shown in the above equation (5). Therefore, the strong field threshold value Ms, which is the value of the modulation factor M when starting the strong field control, is obtained by adding the strong field start deviation ΔMs to the command modulation factor MT as shown in the following equation (6). Is required.
Ms = MT + ΔMs (6)
In this embodiment, since the command modulation rate MT is set to “0.78” and the strong field start deviation ΔMs is set to “−0.073”, the strong field threshold value Ms is overmodulated. It becomes “0.707” which is equal to the threshold value Mo. Therefore, when the modulation factor M exceeds the strong field threshold value Ms during execution of the normal field / normal PWM control mode A1a in a state where the target torque TM is within the strong field allowable torque range TMR, that is, the electric motor. When the operating point No. 4 enters the strong field control region F, the field adjustment unit 8 starts the strong field control.

  In addition, the mode control unit 5 causes the voltage waveform control unit 10 to execute rectangular wave control when the modulation factor M is equal to or higher than the rectangular wave threshold value Mb (maximum modulation factor Mmax), and the modulation factor M is equal to the rectangular wave threshold value. In a state of less than Mb, the voltage waveform control unit 10 is caused to execute PWM control. Furthermore, in this embodiment, since the PWM control includes two types of normal PWM control and overmodulation PWM control, the mode control unit 5 is in a state where the modulation factor M is less than the rectangular wave threshold Mb, and the overmodulation is performed. The voltage waveform control unit 10 executes normal PWM control in a state below the threshold Mo (= 0.707), and in the state greater than the overmodulation threshold Mo (= 0.007), the voltage waveform control unit 10 Modulation PWM control is executed. As described above, the voltage waveform control unit 10 includes the three-phase / two-phase conversion unit 25 and the control signal generation unit 26, and voltage waveform control including PWM control and rectangular wave control is executed by these.

  When the operating point of the motor 4 determined by the rotational speed ω and the target torque TM enters the strong field control region F, the integral input adjusting unit 31 is set by setting the command modulation rate MT and the strong field start deviation ΔMs as described above. A positive adjustment value Y is output from, and a positive d-axis current adjustment command value ΔId is output by the integrator 32. Thereby, the strong field control is started. As described above, the strong field threshold value Ms (curve L2) that defines the strong field control region F is determined by the command modulation rate MT (= 0.78) and the strong field start deviation ΔMs (= −0.073). In this example, it coincides with the overmodulation threshold Mo (= 0.707). The mode control unit 5 first causes the voltage waveform control unit 10 to execute PWM control after starting the strong field control. In this example, since the modulation factor M at the start of the strong field control is the overmodulation threshold Mo, the mode control unit 5 causes the voltage waveform control unit 10 to perform overmodulation PWM control. That is, when the strong field control is started, the mode control unit 5 first executes the strong field / overmodulation PWM control mode A2b. Thereafter, the modulation factor M gradually increases due to the strong field control, and finally reaches the rectangular wave threshold value Mb. After the modulation factor M reaches the rectangular wave threshold value Mb, the mode control unit 5 causes the voltage waveform control unit 10 to execute rectangular wave control. Thereby, the strong field / rectangular wave control mode A3 is executed.

  By the way, after the modulation rate M exceeds the strong field threshold value Ms and the field control unit 8 starts the strong field control, the field control unit 8 sets the d-axis current adjustment command value ΔId so that the modulation rate M matches the command modulation rate MT. adjust. Here, the command modulation rate MT is set to the maximum modulation rate Mmax (= 0.78) value as with the rectangular wave threshold value Mb. Therefore, after starting the strong field control, the modulation factor M finally converges to the maximum modulation factor Mmax. After the modulation factor M reaches the maximum modulation factor Mmax, which is the rectangular wave threshold value Mb, the mode control unit 5 causes the voltage waveform control unit 10 to execute rectangular wave control. Further, from this state, when the modulation factor M changes as the target torque TM or the rotational speed ω of the electric motor 4 changes, the modulation factor deviation ΔM also changes in accordance with the change of the modulation factor M. In the part 8, the d-axis current adjustment command value ΔId is appropriately changed in the direction of increasing or decreasing the field magnetic flux. As a result, the d-axis current adjustment command value ΔId appropriately changes from a positive value at which the strong field control is performed to a negative value at which the weak field control is performed. In the state where the d-axis current adjustment command value ΔId is a negative value, field weakening control is executed. Regardless of whether the strong field control or the weak field control is performed, the modulation factor M converges to the maximum modulation factor Mmax, which is the rectangular wave threshold value Mb, and the state in which the rectangular wave control is executed is maintained.

2-4. As described above, in the control device 2 according to the present embodiment, the d-axis is set so that the modulation factor M is maintained at the maximum modulation factor Mmax, which is the rectangular wave threshold value Mb, during execution of the rectangular wave control. The current adjustment command value ΔId is determined, and the strong field control and the weak field control are executed. Therefore, in the configuration in which the rectangular wave control and the PWM control are switched only by the modulation factor M, the rectangular wave control does not end even when the operating state of the electric motor 4 changes. That is, when one or both of the rotational speed ω and the target torque TM decrease and the operating point of the electric motor 4 enters the region of the normal field / normal PWM control mode A1a on the left side of the curve L2 in FIG. However, only the d-axis current adjustment command value ΔId increases in the direction of increasing the field magnetic flux, and the rectangular wave control and the strong field control are not completed. For this reason, when the d-axis current adjustment command value ΔId increases, the efficiency decreases, or when the rectangular wave control is performed in a region where the rotational speed ω is low, vibration or the like may occur in the output torque of the electric motor 4. Therefore, the mode control unit 5 performs the strong field termination control so that the rectangular wave control can be terminated by appropriately terminating the strong field control in such a case.

That is, the mode control unit 5 determines a strong field end condition, which is a condition for ending the strong field control, based on the target torque TM, the DC voltage Vdc, and the d-axis current adjustment command value ΔId. When the strong field end condition is satisfied, the mode control unit 5 ends the strong field control by the field adjustment unit 8. In the present embodiment, the strong field end condition satisfies one of the following three conditions (A), (B), and (C).
(A) Rotational speed ω of motor 4 <rotational speed threshold value ωT
(B) d-axis current adjustment command value ΔId ≧ adjustment command threshold value ΔIdT
(C) The target torque TM is outside the strong field allowable torque range TMR In the present embodiment, as shown in the above condition (C), the target torque TM is outside the strong field allowable torque range TMR. By determining that this is included in the strong field end condition, the strong field control is restricted to be performed only within the strong field allowable torque range TMR. Hereinafter, the strong field end condition and the end operation of the strong field control will be described in detail.

2-4-1. Strong field end condition (A): End condition based on rotational speed ω As described above, the mode control unit 5 uses an end condition based on the rotational speed ω of the electric motor 4 as the strong field end condition (A). That is, the mode controller 5 adjusts the field on the condition that the rotational speed ω of the electric motor 4 is less than the rotational speed threshold value ωT determined based on the target torque TM and the DC voltage Vdc (ω <ωT). The strong field control by the unit 8 is terminated. In the present embodiment, the electric motor 4 in which the modulation factor M becomes the above-described strong field threshold value Ms (= 0.707) during the execution of the normal field control according to the values of both the target torque TM and the DC voltage Vdc. The rotation speed ω is set as a rotation speed threshold value ωT.

  The control device 2 uses a rotation speed threshold value map 35A (see FIG. 7C) that defines an appropriate rotation speed threshold value ωT in association with the target torque TM and the DC voltage Vdc, as a strong field end condition map 35 ( 1). The mode control unit 5 derives an appropriate rotation speed threshold value ωT corresponding to the target torque TM and the DC voltage Vdc based on the rotation speed threshold value map 35A. FIG. 7 is a conceptual diagram showing a method for deriving the rotation speed threshold value ωT, in other words, a method for creating the rotation speed threshold value map 35A.

  The appropriate rotation speed threshold value ωT can be obtained experimentally using the actual control device 2. For example, as shown in FIG. 7A, first, an arbitrary voltage, “Vdc1” in this case, is selected from the range of the DC voltage Vdc that the DC power supply 3 can take (Vdc = Vdc1). Further, an arbitrary torque, for example, “TM1” is selected from the range of the target torque TM that the electric motor 4 can take (TM = TM1). Next, the selected DC voltage Vdc = Vdc1 and the target torque TM = TM1 are input to the control device 2, and the inverter 6 is caused to execute PWM control (here, normal PWM control), so that the rotational speed ω of the motor 4 gradually increases from zero. Raise. Then, the d-axis current adjustment command value ΔId corresponding to the rotation speed ω is measured, and the rotation speed ω at the moment when the d-axis current adjustment command value ΔId changes from zero to a positive value is measured. As described above, the field adjustment unit 8 is configured to output a positive d-axis current adjustment command value ΔId when the modulation factor M exceeds the strong field threshold value Ms. Therefore, by monitoring the d-axis current adjustment command value ΔId, it is possible to measure the rotational speed ω when the modulation factor M becomes the strong field threshold value Ms. In the example shown in FIG. 7A, the rotational speed ω at this time is “ω11”. The rotational speed ω = ω1 obtained in this way is set as the rotational speed threshold ωT at the DC voltage Vdc = Vdc1 and the target torque TM = TM1. That is, this rotational speed threshold value ωT = ω1 becomes the value of the rotational speed threshold value map 35A (see FIG. 7C) corresponding to the DC voltage Vdc = Vdc1 and the target torque TM = TM1 as arguments.

  After that, while maintaining the DC voltage Vdc = Vdc1, various torques are selected within the range of the target torque TM that can be taken by the electric motor 4, and similarly, the rotational speed ω of the electric motor 4 is changed from zero while executing the PWM control. The rotational speed ω at the moment when the d-axis current adjustment command value ΔId gradually changes from zero to a positive value is measured. In the example of FIG. 7A, the rotational speed ω at the target torque TM = TM2 is “ω12”, and the rotational speed ω at the target torque TM = TM3 is “ω13”. By selecting a large number of torques and measuring the rotational speed ω for each torque, modulation is performed during normal field control in the state where the DC voltage Vdc is “Vdc1” as shown by a curve LωT in FIG. The relationship between the target torque TM at which the rate M becomes the strong field threshold value Ms (= 0.707) and the rotational speed ω can be obtained. This curve LωT theoretically coincides with the curve L2 in which the modulation factor M is the overmodulation threshold value Mo (= 0.707). As shown in FIG. 7B, the relationship (curve LωT) between the target torque TM and the rotational speed ω thus obtained is a map of the rotational speed threshold ωT for the DC voltage Vdc = Vdc1. Thereafter, various voltages are selected within the range of the DC voltage Vdc that the DC power supply 3 can take, such as DC voltage Vdc = Vdc2, DC voltage Vdc = Vdc3,. The relationship between the target torque TM at which the magnetic threshold value Ms (= 0.707) is reached and the rotational speed ω is obtained. Then, the relationship between the target torque TM and the rotational speed ω obtained for each DC voltage Vdc is registered in the rotational speed threshold map 35A as a map of the rotational speed threshold ωT for each DC voltage Vdc.

  As described above, as shown in FIG. 7C, the rotation speed threshold value map 35A defining the appropriate rotation speed threshold value ωT in association with the target torque TM and the DC voltage Vdc can be created. The control device 2 includes the rotational speed threshold value map 35A as described above as a part of the strong field end condition map 35 shown in FIG.

2-4-2. Strong field end condition (B): End condition based on d-axis current adjustment command value ΔId As described above, mode controller 5 uses d-axis as a field adjustment command value as strong field end condition (B). An end condition based on the current adjustment command value ΔId is used. That is, the mode control unit 5 determines that the d-axis current adjustment command value ΔId is equal to or greater than the adjustment command threshold value ΔIdT determined based on the target torque TM and the voltage speed ratio RVω in the direction of increasing the field magnetic flux (ΔId). On the condition of ≧ ΔIdT), the strong field control by the field adjusting unit 8 is terminated. Here, the voltage speed ratio RVω is a ratio between the DC voltage Vdc and the rotational speed ω of the electric motor 4. In the present embodiment, the switching loss reduction effect in the inverter 6 obtained by executing the rectangular wave control together with the strong field control, and the efficiency deterioration by increasing the d-axis current adjustment command value ΔId in the direction of increasing the field magnetic flux. Pay attention to the relationship. Specifically, when the normal field / PWM control mode A1 (here, the normal field / normal PWM control mode A1a) is executed, the loss of the motor 4 and the motor drive device 1 is assumed to be the normal loss Loss1, and the strong field When the rectangular wave control mode A3 is executed, the loss of the electric motor 4 and the electric motor driving device 1 is set as the strong field loss Loss2, and the efficiency improvement by executing the strong field control is the loss difference ΔLoss (= Loss1-Loss2). To do. The upper limit in the direction of increasing the field magnetic flux in the range of the d-axis current adjustment command value ΔId in which the strong field loss Loss2 is smaller than the normal loss Loss1, that is, the loss difference ΔLoss is positive (ΔLoss> 0). The adjustment command threshold value ΔIdT.

  The control device 2 uses the adjustment command threshold value map 35B (see FIG. 8C) that defines an appropriate adjustment command threshold value ΔIdT in association with the target torque TM and the voltage speed ratio RVω to strengthen the field end condition map 35. (See FIG. 1). The mode control unit 5 derives an appropriate adjustment command threshold value ΔIdT according to the target torque TM and the voltage speed ratio RVω based on the adjustment command threshold value map 35B. FIG. 8 is a conceptual diagram showing a method for deriving the adjustment command threshold value ΔIdT, in other words, a method for creating the adjustment command threshold value map 35B.

  An appropriate adjustment command threshold value ΔIdT can be obtained experimentally using the actual control device 2. For example, as shown in FIG. 8A, first, an arbitrary voltage is selected from the range of the DC voltage Vdc that the DC power supply 3 can take, and an arbitrary torque from the range of the target torque TM that the motor 4 can take. . Here, as an example, “Vdc1” is selected as the DC voltage Vdc, and “TM3” is selected as the target torque TM (Vdc = Vdc1, TM = TM3). Then, at the selected DC voltage Vdc = Vdc1 and target torque TM = TM3, a PWM possible upper limit speed ωU that is a rotational speed ω capable of executing PWM control is derived. This PWM possible upper limit speed ωU is the maximum torque control line 62 and the equal torque line 61 of the target torque TM = TM3 when the DC voltage Vdc = Vdc1 on the Id-Iq plane as shown in FIG. The rotation speed ω of the voltage limiting ellipse 63U that passes through the intersection can be obtained. Next, the selected DC voltage Vdc = Vdc1 and target torque TM = TM3 are input to the control device 2, and the inverter 6 is caused to execute PWM control (here, normal PWM control), so that the rotational speed ω of the motor 4 can be set to the PWM upper limit The speed is gradually decreased from the speed ωU. Then, the relationship between the d-axis current adjustment command value ΔId that changes in accordance with the rotation speed ω and the normal loss Loss1 is measured. Further, the rectangular wave control is executed by the inverter 6 under the conditions of the same DC voltage Vdc and the target torque TM, and the rotational speed ω of the electric motor 4 is gradually decreased from the PWM possible upper limit speed ωU. Then, the relationship between the d-axis current adjustment command value ΔId that changes according to the rotational speed ω and the strong field loss Loss 2 is measured. Here, the normal loss Loss1 and the strong field loss Loss2 include a copper loss and an iron loss in the electric motor 4 and a switching loss in the electric motor driving device 1 in the PWM control or the rectangular wave control, respectively. 3 is obtained from the difference between the electric power supplied from 3 to the electric motor driving device 1 and the output obtained by the electric motor 4.

  From the above, the relationship among the d-axis current adjustment command value ΔId that changes according to the rotational speed ω, the normal loss Loss1, and the strong field loss Loss2 is obtained. Therefore, as shown in FIG. 8A, the rotational speed is calculated from the difference (Loss1−Loss2) between the normal loss Loss1 and the strong field loss Loss2 at each d-axis current adjustment command value ΔId (rotational speed ω). The relationship between the d-axis current adjustment command value ΔId that changes according to ω and the loss difference ΔLoss is derived. Then, based on the relationship between the d-axis current adjustment command value ΔId and the loss difference ΔLoss, the rotational speed ω and the d-axis current adjustment at the moment when the loss difference ΔLoss changes from positive to negative (the moment when the loss difference ΔLoss = 0). The command value ΔId is measured. In the example shown in FIG. 8A, the rotational speed ω at this time is “ω1”, and the d-axis current adjustment command value ΔId is “ΔId31”. The ratio between the rotational speed ω = ω1 and the DC voltage Vdc = Vdc1 obtained in this way is the voltage speed ratio RVω1 at this time. The d-axis current adjustment command value ΔId = ΔId31 obtained in this way is set as an adjustment command threshold value ΔIdT at the voltage speed ratio RVω = RVω1 and the target torque TM = TM3. That is, this adjustment command threshold value ΔIdT = ΔId31 becomes a value of the adjustment command threshold value map 35B (see FIG. 8C) corresponding to the voltage speed ratio RVω = RVω1 and the target torque TM = TM3 as arguments. .

  Thereafter, as shown in FIG. 8B, while maintaining the DC voltage Vdc = Vdc1, the target torque TM that the motor 4 can take such as the target torque TM = TM1, the target torque TM = TM2,. Various torques are selected within the range, and similarly, the relationship between the d-axis current adjustment command value ΔId that changes in accordance with the rotational speed ω and the loss difference ΔLoss is derived. For each target torque TM, the rotational speed ω and the d-axis current adjustment command value ΔId at the moment when the loss difference ΔLoss changes from positive to negative are obtained, and the ratio between the rotational speed ω and the DC voltage Vdc is determined as the voltage at this time. As the speed ratio RVω, the d-axis current adjustment command value ΔId thus obtained is set as the adjustment command threshold value ΔIdT at the voltage speed ratio RVω and the target torque TM. Further, various voltages are selected within the range of DC voltage Vdc that the DC power supply 3 can take, such as DC voltage Vdc = Vdc2, DC voltage Vdc = Vdc3,... Then, the target torque TM is selected in various ways, and the relationship between the d-axis current adjustment command value ΔId that changes according to the rotational speed ω and the loss difference ΔLoss is derived. Then, for each combination of the DC voltage Vdc and the target torque TM, the rotational speed ω and the d-axis current adjustment command value ΔId at the moment when the loss difference ΔLoss changes from positive to negative are obtained, and the rotational speed ω and the DC voltage Vdc are obtained. The voltage / velocity ratio RVω at this time is used as the ratio, and the d-axis current adjustment command value ΔId thus obtained is used as the adjustment command threshold value ΔIdT at the voltage / speed ratio RVω and the target torque TM. Then, the relationship among the voltage speed ratio RVω, the target torque TM, and the adjustment command threshold value ΔIdT is registered in the adjustment command threshold value map 35B as a map of the adjustment command threshold value ΔIdT.

  As described above, as shown in FIG. 8C, the adjustment command threshold value map 35B defining the appropriate adjustment command threshold value ΔIdT in association with the target torque TM and the voltage speed ratio RVω can be created. The control device 2 includes the adjustment command threshold value map 35B as described above that can be referred from the mode control unit 5 as a part of the strong field end condition map 35 shown in FIG. In the above method, the voltage speed ratio RVω as an argument of the adjustment command threshold value ΔIdT is obtained based on the rotational speed ω at the moment when the loss difference ΔLoss changes from positive to negative. Therefore, the voltage speed ratio RVω constituting the vertical axis of the adjustment command threshold map 35B may not be the same value for each target torque TM. In that case, it is preferable to obtain and map the adjustment command threshold value ΔIdT when the voltage-speed ratio RVω is set to a predetermined value by linear interpolation or the like.

2-4-3. Strong field end condition (C): End condition based on strong field permissible torque range TMR Further, in the present embodiment, the mode control unit 5 uses the strong field permissible torque range TMR as the strong field end condition (C). An end condition based on is used. That is, the mode control unit 5 ends the strong field control so that the field adjustment unit 8 does not execute the strong field control when the target torque TM of the electric motor 4 is out of the predetermined strong field allowable torque range TMR. Let That is, the mode control unit 5 sets the upper limit of the strong field allowable torque range TMR as the allowable torque upper limit TMRH, sets the lower limit as the allowable torque lower limit TMRL, and sets target torque TM <allowable torque lower limit TMRL or target torque TM> allowable torque upper limit TMRH. At this time, the strong field control is terminated. Here, the allowable torque upper limit TMRH is, for example, the current flowing through the armature coil of the motor 4 when the rectangular wave control in which harmonic components other than the fundamental wave component of the alternating current flowing through the motor 4 are likely to be large is performed. It is preferable that the current limit value allowed for the electric motor 4 is not exceeded. For example, it is preferable that the allowable torque lower limit TMRL is set so as to exclude the torque range that is not suitable for performing the rectangular wave control because the output torque is too small and exclude it from the field allowable torque range TMR.

2-4-4. End operation of the strong field control The mode control unit 5 performs control to set the d-axis current adjustment command value ΔId to zero when any of the above-described strong field end conditions (A) to (C) is satisfied. . That is, if the strong field end condition is satisfied, the mode control unit 5 outputs a command for setting the d-axis current adjustment command value ΔId to zero to the integrator 32, and the d-axis current adjustment command output by the integrator 32. The value ΔId is set to zero. At this time, the mode control unit 5 controls the field adjustment unit 8 so as to change the d-axis current adjustment command value ΔId from the current value toward zero at a constant change rate. That is, since the d-axis current adjustment command value ΔId is a positive value during the execution of the strong field control, the mode control unit 5 sets the d-axis current adjustment command value ΔId to the time when ending the strong field control. As the time elapses, the current value gradually decreases (decreases) from zero to zero. The mode control unit 5 performs control to gradually decrease the modulation factor M by gradually changing the d-axis current adjustment command value ΔId in a direction to decrease the adjustment amount of the field magnetic flux when the strong field control is finished in this way. Do. Thereby, the modulation factor M is gradually decreased from the rectangular wave threshold value Mb (maximum modulation factor Mmax = 0.78) at which the rectangular wave control mode is executed until the d-axis current adjustment command value ΔId becomes zero. The strong field / overmodulation PWM control mode A2b (strong field / PWM control mode A2) is executed until the modulation rate M reaches the overmodulation threshold Mo (= 0.707). . When the d-axis current adjustment command value ΔId becomes zero and the modulation factor M becomes less than the overmodulation threshold Mo, the normal field / normal PWM control mode A1a (normal field / PWM control mode A1) is set. Transition.

  Therefore, in this embodiment, when ending the strong field control, the mode control unit 5 goes from the strong field / rectangular wave control mode A3 to the strong field / PWM control mode A2 to the normal field / PWM control mode A1. To migrate. As a result, when the strong field control is terminated, it is possible to prevent the current command values Id and Iq after the adjustment based on the d-axis current adjustment command value ΔId from changing suddenly and the modulation factor M from changing suddenly. As a result, it is possible to suppress a sudden change and overshoot of the current flowing through the coil, and to suppress the vibration of the output torque of the electric motor 4 from occurring. The mode control unit 5 forcibly sets the d-axis current adjustment command value ΔId to zero when all of the strong field termination conditions (A), (B), and (C) are not satisfied. Stop the end operation. Thereby, the control in which the integrator 32 integrates the adjustment value Y to derive the d-axis current adjustment command value ΔId is resumed.

3. Operation of Control Device Next, the operation of each part of the control device 2 will be described in detail with reference to FIGS. 9 and 10. FIG. 9 is a flowchart showing the flow of operation of each part until the voltage command values Vd and Vq are derived in the control device 2 according to the present embodiment.

  As shown in FIG. 8, the control device 2 first derives the modulation factor M by the modulation factor deriving unit 29 (step # 01). Next, the subtractor 30 derives a modulation factor deviation ΔM (= M−MT) obtained by subtracting the command modulation factor MT (maximum modulation factor Mmax = 0.78) from the modulation factor M (step # 02). Thereafter, the control device 2 determines whether or not the d-axis current adjustment command value ΔId is greater than zero (ΔId> 0) (step # 03). This determination is to determine whether or not the control device 2 is performing strong field control at that time. If the d-axis current adjustment command value ΔId is less than or equal to zero (ΔId ≦ 0) (step # 03: No), it can be determined that the control device 2 is in normal field control or weak field control. Therefore, it is next determined whether or not the modulation factor deviation ΔM is less than zero (ΔM <0) (step # 04). This determination is to determine whether or not the modulation factor M is less than the command modulation factor MT. If the modulation factor deviation ΔM is greater than or equal to zero (ΔM ≧ 0) (step # 04: No), the process proceeds to step # 06, and is output from the integral input adjustment unit 31 based on the modulation factor deviation ΔM. The adjustment value Y below zero (see FIG. 6) is integrated by the integrator 32 to derive the d-axis current adjustment command value ΔId (step # 06). As a result, the d-axis current adjustment command value ΔId changes in the negative direction, that is, in the direction in which the field magnetic flux of the electric motor 4 is weakened. At this time, the field weakening control is started if the normal field control is being performed, and the degree of the field weakening is increased if the field weakening control is being performed.

  If the modulation factor deviation ΔM is less than zero (ΔM <0) (step # 04: Yes), then, whether the modulation factor deviation ΔM is greater than or equal to the strong field start deviation ΔMs (ΔM ≧ ΔMs). Is determined (step # 05). When the modulation factor deviation ΔM is less than the strong field start deviation ΔMs (ΔM <ΔMs) (step # 05: No), the integral input adjustment unit 31 outputs zero as the adjustment value Y (see FIG. 6). . Therefore, the adjustment value Y is not integrated by the integrator 32, and the process proceeds to step # 07. Therefore, the d-axis current adjustment command value ΔId does not change. At this time, if the normal field control is being performed, the normal field control is continued, and if the weak field control is being performed, the weak field control is continued. When the modulation factor deviation ΔM is equal to or greater than the strong field start deviation ΔMs (ΔM ≧ ΔMs) (step # 05: Yes), the integral input adjustment unit 31 outputs a positive value as the adjustment value Y (FIG. 6). reference). Therefore, the integrator 32 integrates the positive adjustment value Y to derive the d-axis current adjustment command value ΔId (step # 06). As a result, the d-axis current adjustment command value ΔId changes in the positive direction, that is, the direction in which the field magnetic flux of the electric motor 4 is strengthened. At this time, if the normal field control is being performed, the strong field control is started, and if the weak field control is being performed, the degree of the weak field is reduced or the process proceeds to the strong field control.

  On the other hand, when the d-axis current adjustment command value ΔId is larger than zero (ΔId> 0) (step # 03: Yes), it can be determined that the control device 2 is in the strong field control. Therefore, next, the mode control unit 5 determines the above-described strong field end conditions (A) to (C). Specifically, condition (A): whether or not the rotational speed ω of the electric motor 4 is less than a rotational speed threshold value ωT determined based on the target torque TM and the DC voltage Vdc (ω <ωT) (step # 10). ), Condition (B): Whether or not the d-axis current adjustment command value ΔId is equal to or greater than an adjustment command threshold value ΔIdT determined based on the target torque TM and the voltage speed ratio RVω (ΔId ≧ ΔIdT) (step # 11). Condition (C): It is determined whether or not the target torque TM of the electric motor 4 is outside the predetermined strong field allowable torque range TMR (step # 12). When any of these strong field end conditions (A) to (C) is satisfied (step # 10: Yes, step # 11: Yes, or step # 12: Yes), the mode control unit 5 performs strong field control. Perform the end operation. That is, the mode control unit 5 sets the d-axis current adjustment command value ΔId to zero at a constant change rate in order to end the strong field control (step # 13). As a result, the strong field control is terminated and the normal field control is executed. When none of the above strong field end conditions (A) to (C) is satisfied (step # 10: No, step # 11: No, and step # 12: No), the strong field control is to be continued. The process proceeds to step # 06. Therefore, the adjustment value Y output from the integral input adjustment unit 31 according to the modulation factor deviation ΔM is integrated by the integrator 32 to derive the d-axis current adjustment command value ΔId (step # 06). Thereby, even during the strong field control, the d-axis current adjustment command value ΔId is appropriately adjusted according to the modulation factor deviation ΔM. At this time, the d-axis current adjustment command value ΔId may change in the negative direction and shift from the strong field control to the weak field control.

  Thereafter, the basic d-axis current command value Idb derived by the d-axis current command value deriving unit 21 and the d-axis current adjustment command value ΔId derived by the integrator 32 are added to obtain an adjusted d-axis current command value Id. Derived (step # 07). Further, the q-axis current command value deriving unit 22 derives the adjusted q-axis current command value Iq (step # 08). Based on the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq, the current control unit 24 derives voltage command values Vd and Vq (step # 18). The process ends here.

  Next, a specific example of the operation of the control device 2 according to the flowchart shown in FIG. 9 will be described with reference to FIGS. 3 and 10. FIG. 10 shows the time when the operating point of the electric motor 4 is changed sequentially from the point t0 to the point t6 shown in FIG. 3 as the time T passes, and then the operating point of the electric motor 4 is changed sequentially from the point t7 to the point t13. FIG. 6 is a diagram showing an example of changes in current command values Id and Iq after adjustment based on target torque TM, rotation speed ω, and d-axis current adjustment command value ΔId. Specifically, FIG. 10A shows the change in the target torque TM along the time axis T, FIG. 10B shows the change in the rotational speed ω, and FIG. 10C shows the adjusted d-axis at that time. Changes in the current command value Id and the adjusted q-axis current command value Iq are shown.

  In this example, at time t0 to t1, the rotational speed ω is increased from zero to ω1 with the target torque TM being zero. At this time, the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq remain zero. From time t1 to t2, the target torque TM is increased from zero to TM6 with the rotational speed ω kept constant at ω1. At this time, the adjusted d-axis current command value Id decreases to Id8 in proportion to the target torque TM, and the adjusted q-axis current command value Iq increases to Iq8 in proportion to the target torque TM. From time t2 to t6, the rotational speed ω is increased from ω1 to ω2 while the target torque TM is kept constant at TM6. At this time, the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq are kept constant at time points t2 to t3 until the operating point of the electric motor 4 enters the strong field control region F. At time points t0 to t3, the normal field / PWM control mode A1 (normal field / normal PWM control mode A1a) is executed. At time points t3 to t4 after the operating point of the electric motor 4 enters the strong field control region F, the strong field control is executed by increasing the d-axis current adjustment command value ΔId, and the adjusted d-axis current command value Id is changed from Id8. The adjusted q-axis current command value Iq increases from Iq8 to Iq9. At this time, the strong field / PWM control mode A2 is executed until the modulation factor M reaches the rectangular wave threshold value Mb (time points t3 to t4).

  Thereafter, since the diameter of the voltage limiting ellipse 63 shown in FIG. 5 is reduced by increasing the rotational speed ω from time t4 to time t5, the adjusted d-axis set on the voltage limiting ellipse 63 during rectangular wave control. Both the current command value Id and the adjusted q-axis current command value Iq decrease. Specifically, the adjusted d-axis current command value Id decreases from Id9 to Id8, and the adjusted q-axis current command value Iq decreases from Iq9 to Iq8. At this time, the d-axis current adjustment command value ΔId also decreases. From time t4 to t5, the strong field / rectangular wave control mode A3 is executed. At time t5, the d-axis current adjustment command value ΔId becomes zero, and the strong field control ends. At time points t5 to t6 after exiting the strong field control region F, the d-axis current adjustment command value ΔId further decreases to become a negative value, whereby field-weakening control is executed, and the adjusted d-axis current command value Id is Id8. The q-axis current command value Iq after adjustment decreases from Iq8 to Iq7. From time t6 to t7, both the rotational speed ω and the target torque TM are maintained constant, so that both the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq do not change.

  From time t7 to t11, the rotational speed ω is decreased from ω2 to ω1 with the target torque TM being constant at TM6. At this time, from time t7 to time t8 until the operating point of the electric motor 4 enters the strong field control region F, the d-axis current adjustment command value ΔId gradually increases while the field weakening control is executed, and the adjusted d-axis current command value Id is The adjusted q-axis current command value Iq increases from Iq7 to Iq8 from Id7 to Id8. At time t8, the d-axis current adjustment command value ΔId becomes zero and field weakening control ends. At time points t5 to t8, field weakening / PWM control mode A4 is executed. At the time t8 to t9 after the operating point of the electric motor 4 enters the strong field control region F, the diameter of the voltage limiting ellipse 63 shown in FIG. Both the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq set on the limit ellipse 63 increase. Specifically, the adjusted d-axis current command value Id increases from Id8 to Id9, and the adjusted q-axis current command value Iq increases from Iq8 to Iq9. At this time, the d-axis current adjustment command value ΔId also increases. From time t8 to t9, the strong field / rectangular wave control mode A3 is executed. In this example, any one of the strong field end conditions (A) to (C) is satisfied at time t9, and thereafter, the d-axis current adjustment command value ΔId is changed at a constant change rate (decrease) until time t10. Set the speed to zero. As a result, the adjusted d-axis current command value Id decreases from Id9 to Id8, and the adjusted q-axis current command value Iq decreases from Iq9 to Iq8. Since the rate of decrease of the d-axis current adjustment command value ΔId is thus restricted, the rate of decrease of the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq after adjustment by the d-axis current adjustment command value ΔId. It is also regulated and increases to draw a gentle curve. As a result, the rate of change (lowering rate) of the modulation factor M is regulated, and a predetermined time is ensured until the modulation factor M reaches the strong field threshold value Ms (curve L2 in FIG. 3). The strong field / PWM control mode A2 is executed at (time points t9 to t10).

  At times t10 to t11 after the operating point of the electric motor 4 comes out of the strong field control region F, the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq are kept constant. From time t11 to t12, the target torque TM is decreased from TM6 to zero with the rotation speed ω kept constant at ω1. At this time, the adjusted d-axis current command value Id increases from Id8 to zero in proportion to the target torque TM, and the adjusted q-axis current command value Iq decreases from Iq8 to zero in proportion to the target torque TM. From time t12 to t13, the rotational speed ω is decreased from ω1 to zero while the target torque TM is zero. At this time, the adjusted d-axis current command value Id and the adjusted q-axis current command value Iq remain zero. From time t10 to t13, the normal field / PWM control mode A1 (normal field / normal PWM control mode A1a) is executed.

4). Other Embodiments (1) In the above embodiment, the case where a value determined based on the target torque TM and the DC voltage Vdc is used as the rotation speed threshold value ωT used in the strong field end condition (A) will be described as an example. did. However, the embodiment of the present invention is not limited to this. For example, setting the rotation speed threshold ωT to a constant value regardless of the target torque TM and the DC voltage Vdc is one of the preferred embodiments of the present invention. It is also a preferred embodiment of the present invention to set the rotation speed threshold value ωT to a value determined based on either the target torque TM or the DC voltage Vdc. Further, the rotational speed threshold value ωT is preferably a value calculated by a predetermined formula based on the target torque TM, the DC voltage Vdc, the adjustment command threshold value ΔIdT, and the like. When the rotational speed threshold value ωT is determined in this way, the strong field end condition (A) based on the rotational speed threshold value ωT and the strong field end condition (B) based on the adjustment command threshold value ΔIdT described above. It is particularly preferable that the strong field control is terminated when both of the above are satisfied. In this case, the mode control unit 5 indicates that the d-axis current adjustment command value ΔId is equal to or greater than the adjustment command threshold value ΔIdT in the direction of increasing the field magnetic flux, and the rotation speed ω is less than the rotation speed threshold value ωT. As a condition, the strong field control is terminated. Even in this case, the strong field end condition (C) based on the strong field allowable torque range TMR is a selective condition, and either the condition (A), the condition (B), or the condition (C) is satisfied. In such a case, it is more preferable to end the strong field control.

(2) In the above embodiment, (A) the rotational speed ω of the electric motor 4 <the rotational speed threshold value ωT, (B) the d-axis current adjustment command value ΔId ≧ the adjustment command threshold value ΔIdT, and (C) the target torque. An example has been described in which the strong field control is terminated when one of the three strong field end conditions, that is, TM is outside the strong field allowable torque range TMR, is satisfied. However, the embodiment of the present invention is not limited to this. For example, the mode control unit 5 may be configured to determine only the strong field end condition (B), and may be configured to perform the strong field end control only when the condition (B) is satisfied. This is one of the preferred embodiments. Further, the mode control unit 5 determines the strong field end conditions (B) and (A) or the strong field end conditions (B) and (C), and any of these strong field end conditions is satisfied. It is also a preferred embodiment of the present invention that the strong field termination control is performed only when it is performed.

(3) In the above embodiment, the rotation of the electric motor 4 in which the modulation factor M becomes the strong field threshold value Ms (= 0.707) as the rotation speed threshold value ωT used in the strong field end condition (A). The case where the speed ω is used has been described. However, the embodiment of the present invention is not limited to this. It is also possible to set the rotation speed threshold value ωT to the rotation speed ω when the modulation factor M becomes a constant value other than the strong field threshold value Ms. Therefore, the rotation speed threshold value ωT is rotated when the modulation factor M is smaller than the strong field threshold value Ms (for example, M = 0.7, M = 0.65, M = 0.5, etc.). It is possible to set the rotational speed ω when the speed ω is set, or when the modulation factor M is larger than the strong field threshold value Ms (for example, M = 0.72, M = 0.75, etc.). It is one of the preferred embodiments of the invention. Further, the rotation speed ω is not limited to a value at which the modulation factor M becomes a constant value, and a predetermined rotation speed ω determined based on the target torque TM and the DC voltage Vdc can be set as the rotation speed threshold value ωT. It is. For example, the rotational speed ω that satisfies TM = −αω + β (α and β are constants) is set for each value of the DC voltage Vdc, and is set as the rotational speed threshold value ωT in the preferred embodiment of the present invention. One.

(4) In the above embodiment, the adjustment command threshold value ΔIdT used in the strong field termination condition (B) is set so that the loss difference ΔLoss (= Loss1−Loss2), which is an improvement in efficiency by executing the strong field control, is positive. The case where the upper limit of the range of the d-axis current adjustment command value ΔId is set as an example has been described. However, the embodiment of the present invention is not limited to this. For example, the adjustment command threshold value ΔIdT is set to an arbitrary value within the range of the d-axis current adjustment command value ΔId where the loss difference ΔLoss is positive, or the d-axis current adjustment command value ΔId where the loss difference ΔLoss is negative. It is also possible to set within the range. Further, the adjustment command threshold value ΔIdT can be set as the adjustment command threshold value ΔIdT based on the target torque TM and the voltage / speed ratio RVω regardless of the loss difference ΔLoss.

(5) In the above embodiment, when the strong field control is terminated during the strong field / rectangular wave control mode, the modulation factor M is obtained by gradually decreasing the d-axis current adjustment command value ΔId at a constant change rate. An example has been described in which control is performed such that the control is shifted to the normal field / pulse width modulation control mode via the strong field / pulse width modulation control mode. However, the embodiment of the present invention is not limited to this. For example, the change time until the d-axis current adjustment command value ΔId becomes zero from the current value is constant regardless of the current value of the d-axis current adjustment command value ΔId when the strong field control ends. In another preferred embodiment of the present invention, the d-axis current adjustment command value ΔId is gradually decreased. Also in this case, since the time until the d-axis current adjustment command value ΔId becomes zero is secured, when the transition from the strong field / rectangular wave control mode to the normal field / pulse width modulation control mode is made The field / pulse width modulation control mode can be executed.

(6) In the above embodiment, the case where the strong field threshold value Ms is set to coincide with the overmodulation threshold value Mo (= 0.707) has been described as an example. However, the embodiment of the present invention is not limited to this. The strong field threshold Ms is set to a value smaller than the overmodulation threshold Mo (for example, M = 0.7, M = 0.65, M = 0.5, etc.), or the overmodulation threshold Mo Setting a larger value (for example, M = 0.72, M = 0.75, etc.) is also one preferred embodiment of the present invention. When the strong field threshold Ms is set to a value larger than the overmodulation threshold Mo, the normal field / PWM control mode A1 is set as the normal field / PWM control mode A1 before the strong field control is started. The modulation PWM control mode A1b is executed.

(7) In the above embodiment, the case where the motor driving device 1 is configured to supply the DC voltage Vdc from the DC power source 3 to the inverter 6 has been described as an example. However, the embodiment of the present invention is not limited to this. For example, a voltage conversion unit such as a DC-DC converter that converts a power supply voltage from the DC power supply 3 to generate a system voltage having a desired value is provided, and the system voltage generated by the voltage conversion unit is used as a DC / AC conversion unit. A configuration in which the power is supplied to the inverter 6 is also a preferred embodiment of the present invention. In this case, the voltage conversion unit can be a boost converter that boosts the power supply voltage, a step-down converter that steps down the power supply voltage, or a step-up / step-down converter that both boosts and steps down the power supply voltage. it can.

(8) In the above embodiment, the case where the AC motor 4 is a synchronous motor (IPMSM) having an embedded magnet structure that operates by three-phase AC has been described as an example. However, the embodiment of the present invention is not limited to this. For example, a synchronous motor (SPMSM) having a surface magnet structure can be used as the AC motor 4, or other than the synchronous motor, for example, induction An electric motor or the like can also be used. Moreover, as an alternating current supplied to such an alternating current motor, a single-phase other than three phases, a two-phase, or a polyphase alternating current having four or more phases can be used.

(9) In the above embodiment, the case where the electric motor 4 is used as a driving force source for an electric vehicle, a hybrid vehicle, or the like has been described as an example. However, the use of the electric motor 4 according to the present embodiment is not limited to this, and the present invention can be applied to electric motors of all uses.

  INDUSTRIAL APPLICABILITY The present invention can be suitably used for a control device that controls a motor driving device including a DC / AC converter that converts a DC voltage into an AC voltage and supplies the AC voltage to an AC motor.

1: Motor drive device 2: Control device 4: AC motor 5: Mode control unit 6: Inverter (DC / AC conversion unit)
7: Current command determination unit 8: Field adjustment unit 9: Voltage command determination unit 10: Voltage waveform control unit Vdc: DC voltage TM: Target torque ω: Rotational speed Idb: Basic d-axis current command value (basic current command value)
Id: Adjusted d-axis current command value (adjusted current command value)
Iqb: Basic q-axis current command value (basic current command value)
Iq: q-axis current command value after adjustment (current command value after adjustment)
ΔId: d-axis current adjustment command value (field adjustment command value)
Vd: d-axis voltage command value (voltage command value)
Vq: q-axis voltage command value (voltage command value)
M: Modulation rate (voltage index)
Mb: rectangular wave threshold value Ms: strong field threshold value RVω: voltage speed ratio ωT: rotation speed threshold value ΔIdT: adjustment command threshold value TMR: strong field allowable torque range A1: normal field / PWM control Mode A2: Strong field / PWM control mode A3: Strong field / rectangular wave control mode

Claims (9)

A control device that controls a motor drive device including a DC / AC converter that converts a DC voltage into an AC voltage and supplies the AC voltage to an AC motor,
Based on the target torque of the AC motor, a current command determination unit that determines a basic current command value that is a command value of a current supplied from the DC / AC conversion unit to the AC motor;
A field adjustment unit for determining a field adjustment command value that is an adjustment value of the basic current command value;
Based on the adjusted current command value after adjusting the basic current command value by the field adjustment command value and the rotation speed of the AC motor, the command value of the voltage supplied from the DC / AC converter to the AC motor A voltage command determination unit for determining a voltage command value,
A voltage waveform controller that controls the DC-AC converter based on the voltage command value, and executes voltage waveform control including at least pulse width modulation control and rectangular wave control;
A mode control unit for controlling the field adjustment unit and the voltage waveform control unit,
The voltage waveform control unit executes the pulse width modulation control when a voltage index representing the magnitude of the voltage command value with respect to the DC voltage is less than a predetermined rectangular wave threshold value, and the voltage index is If the rectangular wave threshold is exceeded, execute the rectangular wave control,
The field adjustment unit adjusts the basic current command value and the strong field control for determining the field adjustment command value so as to adjust the magnetic flux of the AC motor to the basic current command value. It is configured to execute field control including at least normal field control for determining the field adjustment command value so as not to be performed, and the voltage index is equal to or greater than a predetermined strong field threshold value smaller than the rectangular wave threshold value. The strong field control is executed on the condition that
The mode control unit uses the ratio of the DC voltage and the rotational speed of the AC motor as a voltage speed ratio, and the field adjustment command value increases the field magnetic flux in the direction of increasing the target torque and the voltage speed ratio. A control device for an electric motor drive device that terminates the strong field control by the field adjusting unit on condition that the adjustment command threshold value determined based on   In the strong field / rectangular wave control mode for executing the rectangular wave control together with the strong field control, the mode control unit reduces the adjustment amount of the field magnetic flux when ending the strong field control. The voltage index is gradually decreased by gradually changing the field adjustment command value, and the pulse and the pulse width modulation control mode for executing the pulse width modulation control together with the strong field control and the pulse along with the normal field control. The control device for an electric motor drive device according to claim 1, wherein the control is shifted to a normal field / pulse width modulation control mode for executing width modulation control. When the normal field / pulse width modulation control mode for executing the pulse width modulation control together with the normal field control is executed, the loss of the AC motor and the motor driving device is set as a normal time loss, and the rectangular wave control is performed together with the strong field control. As a field time loss to strengthen the loss of the AC motor and the motor drive device when executing the strong field / rectangular wave control mode
The upper limit in the direction in which the field magnetic flux is strengthened in the field adjustment command value range in which the strong field time loss is smaller than the normal loss is defined as the adjustment command threshold value. Control device for motor drive device.   The mode control unit determines both conditions that the field adjustment command value is equal to or greater than the adjustment command threshold value and that the rotation speed is less than a predetermined rotation speed threshold value. The control device for an electric motor drive device according to any one of claims 1 to 3, wherein the strong field control is terminated when at least one of the conditions is satisfied.   The motor drive apparatus control device according to claim 4, wherein the rotation speed threshold value is determined based on the target torque and the DC voltage.   The rotational speed at which the voltage index becomes the strong field threshold during execution of the normal field control according to both values of the target torque and the DC voltage is set as the rotational speed threshold. The control apparatus of the electric motor drive device of description.   The mode control unit performs control so that the field adjustment unit does not execute the strong field control when a target torque of the AC motor is out of a predetermined strong field allowable torque range. The control apparatus of the electric motor drive device as described in any one of these.   2. The mode control unit controls the field adjustment unit to change the field adjustment command value from a current value toward zero at a constant change rate when ending the strong field control. The control apparatus of the electric motor drive device as described in any one of 1 to 7.   The voltage command determination unit performs feedback control on the adjusted current command value based on an actual current value that is an actual value of a current supplied from the DC / AC conversion unit to the AC motor, and determines the voltage command value. The control device for an electric motor drive device according to any one of claims 1 to 8, which is determined.

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