RU2463699C1 - Power conversion device for electric motor excitation - Google Patents

Abstract

FIELD: electricity. ^ SUBSTANCE: in the power conversion device the second control unit (100) includes the current control command shaping unit (10) shaping the electric motor (6) current control command based on the torque control command T*, the voltage amplitude index calculation unit (150) that calculates the voltage amplitude index (PMF-modulation factor) based on the current control command, the current control commands adjustment unit (80) shaping the value of current control commands adjustment dV based on the PMF-modulation factor and the electric motor (6) frequency FINV and the unit (50) for shaping pulse duration modulation signals/voltage control commands including the ripple suppression signal shaping unit shaping the ripple suppression signal based on DC voltage EFC to suppress the ripple component of the power source 2f-component for shaping a gate signal ( pulse duration modulation signal) into the inverter. ^ EFFECT: ensuring suppression of the power source 2f-component combined with simultaneous suppression of current overload shaping or excessive torque ripple in an AC electric motor wherein a single-pulse mode is used. ^ 20 cl, 14 dwg

Description

Technical field

The present invention relates to a power conversion device for driving an electric motor suitable for controlling an AC electric motor.

State of the art

In recent years, an AC electric motor has been used as a power source in the fields of engineering of industrial machines and household appliances, as well as in the fields of automobile transport, electric vehicles, etc. To drive an AC motor, a DC power supply or an AC power source is required. In general, a power conversion device for driving an electric motor for which a DC power source is used as an input power source has a configuration in which a power conversion device receives an input power of a DC voltage supplied from a DC power source, generates an AC voltage having arbitrary frequency using the inverter circuit and excites an AC motor. In general, a power conversion device for which an AC power source is used as an input power source has a configuration in which a power conversion device includes a converter circuit on an input side, once converts an AC voltage received by a converter circuit into a voltage direct current and supplies this direct current voltage to the inverter circuit to drive an alternating current motor.

Configuration, etc. a power conversion device for driving an electric motor is explained with respect to a power conversion device for driving a motor used for an alternating current electric railway, as an example. The voltage in the wires as an AC power source is a single-phase alternating voltage of 20-25 kV. This single-phase AC voltage is reduced to approximately 1-2 kV by a transformer and then introduced into a converter circuit of a power conversion device for driving an electric motor. The converter circuit receives a single-phase voltage input power of 1-2 kV AC, converts a single-phase AC voltage to a DC voltage of approximately 1500-3000 V, and outputs a DC voltage to the inverter circuit.

It is known that the DC voltage as the output for the converter circuit includes a ripple of the frequency component, twice the frequency of the AC power source (hereinafter referred to as the "2f-component of the power source"). When the frequency of the AC motor is located near this 2f component of the power source, it is likely that the electric current of the AC motor changes to overcurrent, or a large ripple occurs in the torque of the AC motor, which prevents safe operation.

Patent Document 1 discloses that such a 2f component of a power source included in the DC voltage is extracted, and the pulse width of the pulse width modulation of the inverter circuit is adjusted to suppress the influence of the 2f component of the power source.

Patent Document 1: Japanese Laid-Open Patent Application No. S56-49693

Summary of the invention

However, the control for suppressing the 2f component of the power source disclosed in Patent Document 1 cannot be applied to all examples of applications. For example, in order to maximize the applied voltage to an AC motor, it is difficult to apply control to a vehicle with an electric motor and the like, which selects and uses the so-called single-pulse mode as a switching state of the inverter circuit.

A single-pulse mode is a mode for using a switching state in which the number of pulses included in the half-period of the inverter output line voltage is one. However, it is not possible to adjust the pulse width in the work area in this single-pulse mode. If the technology of Patent Document 1 is applied to a vehicle with an electric motor and the like that selects and uses single-pulse mode, the problem arises that the AC electric motor generates current overload, or an excessively large torque ripple occurs. Therefore, it is difficult to apply the technology of patent document 1, the main aspect of which is to adjust the pulse width of the pulse width modulation for the inverter circuit, to a vehicle with an electric motor and the like, which selects and uses a single-pulse mode.

An object of the present invention is to provide a power conversion device for driving an electric motor that provides suppression control for a 2f component of a power source while suppressing the formation of current overload or an excessively large ripple of torque in an alternating current motor in an example application in which single-pulse mode is selected and used as the switching state of the inverter circuit.

In order to solve the above problems and achieve the above task, the power conversion device for driving the electric motor according to one aspect of the present invention is configured to include: a first power conversion unit that is connected to an AC power source and converts the AC voltage from the AC power source current to DC voltage; a second power conversion unit, which is connected to the first power conversion unit and converts the DC voltage to AC voltage, and outputs the AC voltage to an AC electric motor; a first control unit that controls the first power conversion unit; and a second control unit that controls the second power conversion unit, wherein the second control unit includes: a current control command generation unit that generates, based on at least a torque control command, a current control command for the alternating current electric motor; a voltage amplitude index calculation unit that calculates, based on a current control command, a voltage amplitude index to be applied to the alternating current motor; a control unit for controlling current control commands, which generates, based on at least an index of the amplitude of the voltage and frequency of the AC motor, the amount of regulation of current control commands for adjusting the current control command; and a ripple suppression signal generating unit that generates a ripple suppression signal based on the DC voltage, and the second control unit generates, based on the control amount of the current control, controlled by the current control amount and ripple suppression signal, a pulse width pulse modulation to a second power conversion unit and outputs a pulse width modulation signal.

Further, a power conversion device for driving an electric motor according to another aspect of the present invention is configured to include: a first power conversion unit that is connected to an AC power source and converts an AC voltage from an AC power source to a DC voltage; a second power conversion unit, which is connected to the first power conversion unit and converts the DC voltage to AC voltage and outputs the AC voltage to an AC electric motor; a first control unit that controls the first power conversion unit; and a second control unit that controls the second power conversion unit, wherein the second control unit includes: a current control command generation unit that generates, based on at least a torque control command, a current control command for the alternating current electric motor; and a voltage amplitude index calculation unit that calculates, based on the current control command, a voltage amplitude index to be applied to the alternating current motor, and the first control unit includes: a direct current voltage control command generation unit that generates a direct current voltage control command which is the target value of the DC voltage; and a DC voltage control unit that performs control to match the DC voltage and the DC voltage control command with each other, and when the frequency of the AC motor is present in a predetermined range and the output voltage of the second power conversion unit is set to a predetermined value smaller the maximum voltage that can be output in accordance with the DC voltage, block forming anda controlling DC voltage generates and outputs a control command DC voltage to provide a coincidence output voltage of the second power converting unit with a predetermined value.

Using a power conversion device for driving an electric motor according to the present invention, a pulse width modulation signal for a second power conversion unit is generated according to a control signal including a current control command adjustable by a current command amount for adjusting a current control command and a ripple signal for controlling a component ripple of the 2f component of the power source. Therefore, such an advantage is provided that suppression control can be performed for the 2f component of the power supply while suppressing the formation of overcurrent or excessively large ripple of the torque in the AC motor in an example application where the single-pulse mode is selected and used as a switching state of the circuit inverter.

Brief Description of the Drawings

The invention is further explained in the description of the preferred embodiments with reference to the accompanying drawings, in which:

1 is a diagram of an example configuration of a power conversion device for driving a motor in a first embodiment of the present invention.

FIG. 2 is a diagram of an example of a detailed configuration of a current control command generation unit shown in FIG. 1.

FIG. 3 is a diagram of an example of a detailed configuration of a pulse width modulation signal generation unit / voltage control command shown in FIG. 1.

FIG. 4 is a diagram of an example of a detailed configuration of a ripple suppression signal calculating unit shown in FIG. 3.

5 is a diagram of an example of an internal state of a ripple suppression signal calculating unit in the first embodiment.

FIG. 6 is a diagram of an example of a detailed configuration of a current command control unit shown in FIG. 1.

Fig.7 depicts an enlarged diagram of a unit for generating commands for controlling the modulation coefficient shown in Fig.6.

Fig. 8 is a diagram for explaining the relationship between the inverter output frequency FINV and the modulation factor transition PMF, the pulse mode transition, and the selection switch operation (see Fig. 3) in the first embodiment.

Fig. 9 is a diagram of general control characteristics of permanent magnet synchronous motors in a first embodiment of the present invention and a prior art example.

10 is a diagram for explaining control modes in the first embodiment of the present invention.

11 is a diagram of an example configuration of a power conversion device for driving a motor in a second embodiment of the present invention.

12 is a diagram of a first configuration example of a DC voltage command generation unit in the second embodiment shown in FIG. 11.

FIG. 13 is a diagram of a second configuration example of a DC voltage command generation unit in the second embodiment shown in FIG. 11.

14 is a diagram for explaining operating modes in an example in the prior art.

Description of preferred embodiments of the invention

Embodiments of a power conversion device for driving an electric motor according to the present invention are explained in detail below with reference to the accompanying drawings. The present invention is not limited to the embodiments explained below.

First Embodiment

1 is a diagram of an example configuration of a power conversion device for driving a motor in a first embodiment of the present invention. Figure 1 shows an example configuration of a power conversion device for driving an electric motor that controls a permanent magnet synchronous electric motor as an alternating current electric motor.

1, a power conversion device 300 for driving an electric motor in a first embodiment includes a converter 220 serving as a first power conversion unit that receives a single-phase AC voltage from an AC power source 230 and converts a single-phase AC voltage to a DC voltage a current, a capacitor 1 serving as a DC power source, an inverter 2 serving as a second power conversion unit, which converts the direct current voltage from the capacitor 1 to an alternating current voltage having an arbitrary frequency, and an alternating current motor 6 (hereinafter simply referred to as the "electric motor"). As a converter 220, a single-phase two-level PWM converter, a single-phase three-level PWM converter, etc. is suitable for use. As the inverter 2, an inverter based on the type of voltage, such as a three-phase two-level PWM inverter or a three-phase three-level PWM inverter, is suitable for use. Since the configurations of the power circuits of both of the converter 220 and the inverter 2 are well known, a detailed explanation of the converter 220 and the inverter 2 is omitted.

The AC power source 230 is a power source that outputs, for example, a single-phase voltage of 1-2 kV AC. Converter 220 is a voltage converter unit that takes a single-phase AC voltage as input, converts a single-phase AC voltage, for example, to a DC voltage of approximately 1500-3000 V, and outputs a DC voltage to capacitor 1. DC voltage (capacitor voltage 1) as the output of the converter 220 includes approximately 5% of the ripple of the frequency component, twice the frequency of the power source of the AC power source 230 th current (hereinafter referred to as "power supply 2f-component").

Converter 220 as a first power conversion unit receives a single-phase AC voltage from AC power source 230, converts a single-phase AC voltage to DC voltage and outputs a DC voltage to capacitor 1. As a converter 220, a so-called PWM converter can be used, which Converts from AC to DC using a switching element (not shown), such as IGBT (Isolation Bipolar Transistor) shutter). Since the configuration of the power circuit of the converter 220 is well known, a detailed explanation of the converter 220 is omitted.

A current sensor 214, which detects an input current from an AC power source 230, is housed in a power conversion device 300 for driving an electric motor. An input current IS detected by the current sensor 214 is inputted to the first control unit 200. A control signal CG for controlling the switching element of the converter 220 is generated by the first control unit 200 and output to the converter 220.

A voltage sensor 8 that detects a voltage (hereinafter referred to as "voltage across the capacitor") of the EFC of the capacitor 1 is located in the power conversion device 300 for driving the electric motor. On the output line connecting the inverter 2 and the electric motor 6, current sensors 3, 4 and 5 are placed which detect electric currents iu, iv and iw flowing to the output line. The rotation sensor 7, which detects a signal (mechanical angle θm of the rotor) representing the state of rotation of the rotor, is located in the electric motor 6. These detection signals of the current sensors 3, 4 and 5 and the rotation sensor 7 are input to the second control unit 100.

A sensorless rotation system can be used that calculates a position signal from a detected or estimated voltage value, current value, and the like. electric motor 6 instead of a signal (position signal) obtained from the rotation sensor 7. In this case, the rotation sensor 7 is optional. In other words, the receipt of the rotation state signal is not limited to using the rotation sensor 7.

The current sensors 3, 4 and 5 must be installed in at least only two phases. In this case, the current in the remaining one phase can be obtained by calculating on the basis of the given output signals of the current sensors in two phases. The output current of the inverter 2 can be reproduced and obtained using the direct current of the inverter 2.

The gate signals U, V, W, X, Y, and Z generated by the second control unit 100 are input to the inverter 2. The switching element included in the inverter 2 is subjected to PWM control. As an inverter 2, a PWM voltage inverter is suitable for use. Since the configuration of the inverter 2 is well known, a detailed explanation of the inverter 2 is omitted.

The configuration of the second control unit 100 is explained below. As shown in FIG. 1, a torque control command T * is input to the second control unit 100 from an external control unit not shown. This second control unit 100 is a component having a control function of the inverter 2 so that the generated torque T of the electric motor 6 coincides with the input torque control command T *. The second control unit 100 includes a current control command generation unit 10, a voltage amplitude index calculation unit 150, a phase angle control calculating unit 40, a pulse width modulation signal / voltage command generating unit 50, a current control command adjusting unit 80, block 69 calculating the angular frequencies of the inverter, block 95 calculating the reference phase angles and block 90 converting three-phase coordinates along dq-axes. The voltage amplitude index calculation unit 150 includes a d-axis current control unit 20, a q-axis isolation calculation unit 21, a d-axis noise immunity calculation unit 22, a q-axis current control unit 23, and a modulation coefficient calculation unit 30.

The reference phase angle calculation unit 95 calculates the phase reference angle θe from the mechanical rotor angle θm. The dq-axis three-phase coordinate transformation unit 90 generates a d-axis id current and a d-axis current iq from a three-phase current iu, iv and iw detected by current sensors 3, 4 and 5 and a phase reference angle θe. The inverter angular frequency calculating unit 69 calculates the output angular frequency ω of the inverter from the reference phase angle θe. The current control command generation unit 10 generates the d-axis current control command id * and the q-axis current control command iq * from the torque control command T * input from the outside and the control value dV of the current control command.

The d-axis current control unit 20 subjects the current id deviation between the d-axis current control command id * and the d-axis current id to proportional-integral control and generates an error pde on the d-axis current. The q-axis isolation unit 21 calculates the forward q-axis voltage vqFF from the d-axis current command id * and the inverter output angular frequency ω. The d-axis noise immunity calculation unit 22 calculates a forward d-axis voltage vdFF from the q-axis current command iq * and the inverter output angular frequency ω. The q-axis current control unit 23 subjects the current diq deviation between the q-axis current control command ip * and the q-axis current iq to proportional-integral control and generates an error pqe with respect to the q-axis current. The modulation coefficient calculation unit 30 calculates the modulation coefficient PMF from the d-axis voltage control command vd *, which is the sum of the dpe error of the d-axis current and the d-axis forward voltage vdFF, the q-axis voltage control command vq *, which is the sum of the error pqe on the q-axis current and forward q-axis voltage vqFF, the phase reference angle θe and the capacitor EFC.

The control phase angle calculation unit 40 calculates the control phase angle θ from the d-axis voltage control command vd *, the q-axis voltage control command vq * and the phase reference angle θe. The current control command adjusting unit 80 generates a control value dV of the current control command from the modulation factor PMF and the inverter output frequency FINV. The pulse width modulation signal / voltage control command generation unit 50 generates gate signals U, V, W, X, Y, and Z from the inverter 2 from the PMF modulation coefficient, the control phase angle θ, and the output frequency FINV of the inverter.

According to the functions of the components configured as explained above, the voltage amplitude index calculation unit 150 generates a modulation coefficient PMF, a d-axis voltage control command vd * and a q-axis voltage control command vq * using the current did deviation, forward voltage vqFF q- axis, forward voltage vdFF of the d-axis, deviation of the current diq, voltage EFC on the capacitor and the reference phase angle θe, outputs the modulation coefficient PMF to the pulse-width modulation signals generation / voltage control unit 50 and outputs A team management d-axis voltage vd * a command vq *, and the control voltage in q-axis control unit 40 calculate the phase angle.

According to the functions of the components configured as explained above, the second control unit 100 generates gate signals U, V, W, X, Y, and Z using the rotor mechanical angle θm, three-phase currents iu, iv, and iw, torque command T * and the EFC voltage across the capacitor and outputs the gate signals U, V, W, X, Y, and Z to inverter 2.

The detailed configurations and operation of the control units explained above are explained below. Firstly, the reference phase angle calculation unit 95 calculates, based on the following formula, the reference phase angle θe as the electric angle from the rotor mechanical angle θm.

θe = θm * PP (one),

where PP represents the number of pole pairs of the motor 6.

The dq-axis three-phase coordinate transformation unit 90 generates, based on the following formula, the d-axis current id and q-axis current iq from the three-phase currents iu, iv and iw and the reference phase angle θe.

(2)

The inverter angular frequency calculating unit 69 calculates, based on the following formula, the inverter output angular frequency ω by differentiating the reference phase angle θe.

ω = dθe / dt (3)

When the inverter output angular frequency ω is calculated, the inverter output frequency FINV obtained by dividing the inverter output angular frequency ω by 2π is also calculated.

The detailed configuration and operation of the current control command generation unit 10 is explained below with respect to FIG. FIG. 2 is a diagram of an example of a detailed configuration of a current control command generation unit 10 shown in FIG.

The current control command generation unit 10 is a component having a generation function based on the torque control command T * input from the outside, the d-axis current control command id * and the q-axis current control command iq *. The current control command generating unit 10 includes a d-axis base current command generating unit 11, a q-axis current command generating unit 15 and an adder 14. Examples of a method for generating a d-axis current control command id * and a current control command iq * The q-axes include a maximum torque / current control method for generating maximum torque with a specific electric current, and a maximum efficiency control method for maintaining electric motor efficiency at maximum. These optimal control methods are methods for performing control using the motor speed, the absolute value of the output torque, and the like. as parameters in such a way that the actual current of the electric motor 6 coincides with a predetermined calculation formula or an optimal torque component current control command (q-axis current control command iq *) and a magnetic flux component current control command (current control command id * d- axis) obtained by saving to the table in advance.

In the current control command generation unit 10 according to this embodiment, as shown in FIG. 2, the torque control command T * is input to the d-axis base current command generation unit 11, and the d-axis basic current control command id1 * is generated as The first d-axis current control command. As a method for generating the d-axis base current control command id1 *, a method for controlling maximum torque is known by which the electric motor 6 can generate the required torque with minimum current. For example, there is provided a method for obtaining, based on a torque control command T *, an optimal d-axis basis current control command id1 * for accessing the map and a method for obtaining an optimal d-axis basis current control command id1 * for arithmetic formula. In both methods, the d-axis basic current command generating unit 11 can be configured using various well-known technologies. Therefore, a more detailed explanation is omitted.

The d-axis base current control command id1 * generated by the d-axis base current control command generation unit 11 is input to the adder 14 and added to the control value dV of the current control command, whereby the d-axis current control command id * is generated as the second d-axis current control command. The control value dV of the current control command mainly takes a negative value and gives a correction in the negative direction for the d-axis base current control command id1 *. Explaining in more detail, the regulating value dV of the current control command acts as a control output signal for performing so-called magnetic flux control with field weakening to increase the d-axis current control command id * in the negative direction, forming the magnetic flux in the direction in which the magnetic flux formed by the permanent magnet included in the electric motor 6 is suppressed, and weakening the flux linkage of the electric motor 6 to lower the electric motor voltage 6. The control value dV of the current control command is a control output signal generated by the current control command adjusting unit 80. A detailed configuration of the current command control unit 80 is explained below.

The d-axis current control command id * is output to the voltage amplitude index calculating unit 150 as an output signal of the current control command generating unit 10 and, on the other hand, is input to the q-axis current command generating unit 15. In the q-axis current control command generation unit 15, the q-axis current control command iq * as the first q-axis current control command is generated from the d-axis current control command id * and the torque control command T *. As a method for generating the q-axis current control command iq *, similarly to the method for generating the d-axis base current control command id1 *, a method is provided for obtaining the optimal q-axis current control command iq * for accessing the card and a method for obtaining the optimal iq * command q-axis current control according to the calculation formula. In both methods, the q-axis current command generation unit 15 can be configured using various well-known technologies. Therefore, a more detailed explanation is omitted.

The operation of the voltage amplitude index calculation unit 150 is explained below. Referring again to FIG. 1, the q-axis current control unit 23 generates, on the basis of formula (4), the q-axis current error pqe obtained by proportionally-integrated amplification of the difference between the q-axis current control command iq * and the q-current iq axis. The d-axis current control unit 20 generates, on the basis of formula (5), an error pde in the d-axis current obtained by proportionally-integrated amplification of the difference between the d-axis current control command id * and the d-axis current id.

pqe = (K1 + K2 / s) • (iq * -iq) (four) pde = (K3 + K4 / s) • (id * -id) (5)

In the above formulas, K1 and K3 represent proportional gains, and K2 and K4 represent integral gains.

According to need, the voltage amplitude index calculation unit 150 may be a control system that can select whether pqe and pde are used or not for control (i.e., whether pqe and pde are set to zero or not).

The d-axis noise immunity calculation unit 22 calculates the forward voltage vdFF of the d-axis based on formula (6). The q-axis isolation unit 21 calculates the forward voltage vqFF of the q-axis based on formula (7).

vdFF = (R1 + s • Ld) • id * -ω • Lq • iq * (6) vqFF = (R1 + s • Lq) • iq * + ω • (Ld • id * + ϕa) (7)

In the above formulas, R1 represents the primary resistance (Ω) of the motor 6, Ld represents the inductance along the d-axis (H), Lq represents the inductance along the q-axis (H), ϕ a represents the magnetic flux of the permanent magnet (Wb) and s represents the operator differentiation.

The modulation coefficient calculation unit 30 calculates, based on the following formula, the modulation coefficient PMF as the voltage amplitude index from the d-axis voltage control command vd *, which is the sum of the pde error in the d-axis current and the d-axis forward voltage vdFF, the voltage control command vq * q-axis, which is the sum of the error pqe in the current of the q-axis and the forward voltage vqFF of the q-axis, the reference phase angle θe and the voltage EFC on the capacitor.

PMF = VM * / VMmax (8)

VMmax and VM * in the formula (8) are represented by the following formulas:

/π)•EFCVMmax = (

/ π) • EFC (9) MV * = sqrt (vd * ² + vq * ² ) (10).

The modulation factor PMF indicates the absolute value MV * of the vector of the inverter output voltage control command as a ratio to the maximum voltage VMmax (defined by formula (9)) that can be output by the inverter. For example, in the case of PMF = 1.0, the absolute value VM * of the inverter output control command vector is equal to the maximum voltage VMmax that can be output by the inverter.

As can be understood from formulas (2) - (10), the modulation coefficient PMF has a characteristic that the modulation coefficient PMF changes according to the d-axis current control command id * and the q-axis current control command iq * generated by the control command generating unit 10 electric current.

The control phase angle calculation unit 40 calculates, on the basis of the following formula (11), the control phase angle θ from the d-axis voltage control command vd *, which is the sum of the error pde in the d-axis current and the d-axis forward voltage vdFF, the control command vq * q-axis voltage, which is the sum of the pqe error in the q-axis current and the forward q-axis voltage vqFF, and the phase reference angle θe.

θ = θe + π + THV (eleven)

The THV in formula (11) is represented by the following formula:

THV = tan ^-1 (vd * / vq *) (12).

The configuration and operation of the pulse width modulation signal generation / voltage control unit 50 is illustrated in relation to FIG. FIG. 3 is a diagram of an example of a detailed configuration of a pulse width modulation signal / voltage control command generation unit 50 shown in FIG. 1.

As shown in FIG. 3, the pulse width modulation signal / voltage control command generation unit 50 includes a ripple suppression signal calculating unit 71 that receives an EFC voltage across the capacitor as an input and generates a ripple suppression signal BTPMFCMP. The pulse width modulation signal / voltage control command generation unit 50 multiplies the PMF modulation coefficient by a ripple suppression signal BTPMFCMP to generate a PMFM, which is an amplitude control command signal of the voltage control commands. The configuration of the ripple suppression signal calculating unit 71 is explained below.

The voltage control command calculating unit 55 generates, based on the following formula, the U-phase voltage control command Vu *, the V-phase voltage control command Vv * and the W-phase voltage control command Vw *, which are three-phase voltage control commands, from the PMFM signal and the control signal phase angle θ.

Vu * = PMFM • sin θ (13) Vv * = PMFM • sin (θ- (2 • π / 3)) (fourteen) Vw * = PMFM • sin (θ- (4 • π / 3)) (fifteen)

The absolute values of the U-phase voltage control command Vu *, the V-phase voltage control command Vv * and the W-phase voltage control command Vw * generated by the voltage control command calculating unit 55 are compared with the carrier signal CAR by comparison modules 61-63. The gate signals U, V and W and the inverted gate signals X, Y and Z are generated, inverted through the inverting circuits 64-66.

The carrier signal CAR is one of the signals selected in the selection switch 59 by the pulse mode processor 60 serving as the pulse mode switching unit. Any of an asynchronous multi-pulse (generally approximately 1 kHz) carrier signal A generated by an asynchronous multi-pulse carrier signal generating unit 57, a synchronous three-pulse carrier signal B generated by a synchronous three-pulse carrier generating unit 58, and a zero value C selected in a synchronous single-pulse mode, is selected via the select switch 59. The asynchronous multi-pulse carrier signal A and the synchronous three-pulse carrier signal B take values from -1 to 1, centered around zero.

The pulse mode switching processor 60 switches the selection switch 59 according to the values of the modulation coefficient PMF and the control phase angle θ. In particular, in the area in which the modulation coefficient PMF is low (equal to or less than 0.785), the selection switch 59 switches to the side of the asynchronous multi-pulse carrier signal A to select the asynchronous multi-pulse mode. When the modulation PMF is greater than 0.785 and less than 1.0, the selection switch 59 switches to the synchronous three-pulse carrier signal B side to select the synchronous pulse mode. When the modulation coefficient PMF reaches approximately 1.0 (the modulation coefficient PMF may be 0.99 or the like, and not only 1.0), the selection switch 59 switches to the zero value side C. With this configuration, at a time when the modulation coefficient PMF is approximately 1.0, you can automatically switch the pulse mode to synchronous single-pulse mode. Conversely, when the PMF modulation coefficient is less than about 1.0, the pulse mode can be automatically switched to the synchronous three-pulse mode. In other words, it is possible to easily transfer the output voltage of the inverter 2 from minimum to maximum.

When switching the pulse mode, the signal addressed by the pulse switching processor 60 is preferably a modulation factor PMF, which is a signal before the ripple suppression signal BTPMFCMP, explained below, is reflected. By performing the configuration in which the modulation factor PMF is referred to, instability of the switching operation of the pulse mode by the pulse switching processor 60 can be prevented.

A three-pulse synchronous mode is a pulse mode necessary for outputting a voltage having a PMF modulation factor equal to or greater than 0.785, which cannot be output in an asynchronous multi-pulse mode. If the overmodulation method is used in asynchronous multi-pulse mode, synchronous five-pulse mode, synchronous nine-pulse mode, etc., it is possible to output a voltage equivalent to the voltage in synchronous three-pulse mode. However, when this method is used, the modulation factor PMF and the output voltage of the inverter 2 are extremely nonlinear. Therefore, it is necessary to correct this nonlinearity. There is an inconvenience that the configuration is complex.

In the above explanation, the threshold of the modulation coefficient PMF for switching the asynchronous multi-pulse carrier signal and the synchronous three-pulse carrier signal is set to 0.785. However, a threshold value other than 0.785 may be used.

As explained below, the carrier signal CAR in comparison with the voltage control commands has at least an asynchronous multi-pulse carrier signal and a synchronous carrier signal. The carrier signal CAR may be selected according to the pulse mode selected by the pulse mode processor 60 serving as the pulse mode control unit.

An asynchronous multi-pulse carrier signal is a carrier signal that determines the frequency regardless of the inverter output frequency FINV. The frequency is approximately 1000 Hz.

The frequency of a synchronous carrier signal, such as a three-pulse synchronous carrier signal, is determined as a function of the inverter output frequency FINV so that the number of pulses and the position of the pulses included in the inverter output voltage are identical in the half-cycle of the positive side and the half-cycle of the negative side of the inverter output voltage. In this embodiment, an example is explained in which only a synchronous three-pulse carrier signal is used as a synchronous carrier signal. However, the synchronous carrier signal may be, for example, a synchronous five-pulse carrier signal and the like, in addition to a synchronous three-pulse carrier signal. A plurality of synchronous carrier signals may be prepared and included as needed.

In the state in which the asynchronous multipulse mode is selected, when the inverter output frequency FINV is near the frequency of the asynchronous multipulse carrier signal, the number of pulses included in the half-period of the inverter output voltage is reduced. The frequency of the asynchronous multi-pulse carrier signal is a value determined independently of the inverter output frequency FINV. Therefore, when the electric motor 6 is excited in this state, the number of pulses and the position of the pulses, respectively included in the positive half cycle and the negative half cycle of the inverter output voltage, are unbalanced or temporarily oscillate, and the positive and negative symmetry of the voltage applied to the electric motor 6 is broken. The oscillation of the current and ripple of the torque occur in the electric motor 6, which causes noise and vibrations.

On the other hand, when a synchronous carrier signal is used, the number of pulses and the position of the pulses respectively included in the positive half cycle and the negative half cycle of the inverter output voltage are equal, and the positive and negative symmetry of the voltage applied to the electric motor 6 is ensured. Therefore, the occurrence of current fluctuation can be prevented and ripple torque in the electric motor 6 and stably excite the electric motor 6.

Regarding the synchronous single-pulse mode, the number of pulses included in the half-cycle of the inverter output voltage is always equal to one and is fixed without a temporary change. Therefore, the number of pulses and the position of the pulses are identical in the positive half cycle and negative half cycle of the inverter output voltage. Positive and negative symmetry of the voltage applied to the electric motor 6 can be ensured. Therefore, there is no fear that current fluctuation and torque ripple occur in the electric motor 6.

A configuration may be added to precisely control the switching time distribution for the pulse mode according to the control phase angle θ. Such an advantage is provided that it is possible to suppress the ripple of the motor current during the switching of the pulse mode.

The configuration and operation of the ripple suppression signal calculating unit 71, which acts as the ripple suppression signal generating unit, is explained with reference to FIG. FIG. 4 is a diagram of an example of a detailed configuration of a ripple suppression signal calculating unit 71 shown in FIG.

In the ripple suppression signal calculating unit 71, as shown in FIG. 4, the capacitor EFC is input to a band-pass filter (hereinafter referred to as “BPF”) 72. The capacitor EFC is filtered by BPF 72, and the signal EFCBP1 is generated. BPF 72 is set so that the 2f component of the frequency of the power source of the AC power source 230 can be effectively extracted.

In the adder 73, the signal EFCBP2 is generated as the sum of the generated signal EFCBP1 and the capacitor voltage control command EFC *, which is the voltage control command for the capacitor 1. The capacitor voltage control command EFC * is the target value of the capacitor voltage EFC at the time that the converter 220 executes control for converting the AC voltage of the AC power source 230 to a DC voltage (= capacitor EFC voltage). Typically, the capacitor voltage control command EFC * takes a value of approximately 1500-3000 V.

Instead of the capacitor voltage control command EFC *, a signal can be generated by passing the voltage EFC on the capacitor through an LPF (not shown) and removing the variable current component to leave only the constant current component. This signal may be added to the signal EFCBP1 by the adder 73 to generate the signal EFCBP2.

The capacitor voltage control command EFC * and the signal EFCBP2, which is the output of the adder 73, are input to the divider 74. In the divider 74, the capacitor voltage control command EFC * is divided by the signal EFCBP2. The division result is output as a ripple suppression signal BTPMFCMP.

Similarly to the signal EFCBP2, instead of the capacitor voltage control command EFC *, a signal can be generated by passing the voltage EFC on the capacitor through an LPF (not shown) and removing the AC component to leave only the DC component of the current. This signal can be divided into EFCBP2 by a divider 74 to generate a ripple suppression signal BTPMFCMP.

The ripple suppression signal BTPMFCMP thus obtained indicates the inverse of the ratio of the voltage EFCBP2 on the capacitor including the ripple component of the 2f component of the power supply to the DC component of the EFC voltage on the capacitor.

5 is a diagram of an example of an internal state of a ripple suppression signal calculating unit 71 in the first embodiment. 5, an internal state in which the center value of the capacitor voltage EFC at 3000 V is shown as an example.

As shown in FIG. 5, the EFC voltage across the capacitor includes, together with the 2f component of the power source, a ripple component formed by the switching operation of the converter 220 and having a frequency higher than the frequency of the 2f component of the power source (see the waveform in the upper step figures).

The EFCBP1 signal is a signal from which the ripple component is removed by means of the BPF 72 function and includes only the 2f component of the power supply (see the waveform in the upper middle part of the drawing).

The signal EFCBP2 is the value obtained by adding EFC *, which is the capacitor voltage control command, to the signal EFCBP1. Only the 2f component of the power source is included in the EFCBP2 signal as a component of the oscillation (see the wave-like shape in the middle lower step part of the figure).

You can see that the ripple suppression signal BTPMFCMP indicates the reciprocal of the capacitor EFCBP2 voltage, which includes the ripple component of the 2f component of the power supply, relative to the DC component of the EFC voltage on the capacitor (see the wave-like shape in the middle lower step part of the figure).

The ripple suppression signal BTPMFCMP, which is the output of the ripple suppression signal calculating unit 71, is input to the multiplier 70 of the pulse width modulation signal / voltage control command generation unit 50 and multiplied by the modulation factor PMF (see FIG. 3). By multiplying the ripple suppression signal BTPMFCMP by the modulation factor PMF, the amplitude control command signal PMFM can be generated to suppress the ripple component of the 2f component of the capacitor EFC voltage power supply.

As shown in FIG. 3, the output voltage control commands to the inverter 2 are generated based on the signal PMFM of the amplitude control command of the voltage control commands. Thus, it is possible to adjust the width of the voltage pulse output by the inverter to suppress the 2f component of the power source. Therefore, it is possible to solve the problem in that the AC motor generates current overload, and an excessively large ripple of the torque occurs in the region in which the inverter output frequency FINV and the frequency of the 2f component of the power source are close to each other.

The configuration and operation of the current command control unit 80 is illustrated with respect to FIG. 6. FIG. 6 is a diagram of an example of a detailed configuration of a current control command adjusting unit 80 shown in FIG. 1.

The current command control unit 80 is a component having a function of generating, based on the output frequency FINV of the inverter, the control value dV of the current command. The current command control block 80 includes, as shown in FIG. 6, a modulation factor control command block 85, a subtraction block 84, a limit block 81, and an amplifier 82 (having a gain K).

The modulation factor control command generation unit 85 functions as a voltage amplitude control command generation command generation unit and generates a modulation factor control command PMF *, which is a voltage amplitude control target command, based on the inverter output frequency FINV. The subtracting unit 84 outputs a value obtained by subtracting the modulation factor PMF from the modulation coefficient control command PMF *. The restriction unit 81 receives the output of the subtracting unit 84 as an input signal. When the sign of the input signal is a plus, the restriction unit 81 sets the output to zero. When the sign of the input signal is minus, the restriction unit 81 directly outputs the input signal. An amplifier 82 (having a gain of K) amplifies the output signal and outputs the amplified signal as a control value dV of the current control command. The control value dV of the current control command is represented as indicated by the following formula:

dV = LIM (PMF * -PMF) • K (16),

where LIM () represents a function for restricting the upper and lower limits of a value in parentheses according to the method explained above.

The configuration and operation of the modulation coefficient control command generation unit 85 is explained with respect to FIG. FIG. 7 is an enlarged diagram of a modulation coefficient control instruction generation unit 85 shown in FIG. 6.

As explained above, the modulation factor control command generation unit 85 generates a modulation coefficient control command PMF * based on the output frequency FINV of the input inverter. As shown in FIG. 7, the modulation factor control command PMF * is set, for example, to 0.95 in a region in which the inverter output frequency FINV has a value within 120 Hz (120 Hz ± 30 Hz), and is set to 1.0 in other areas.

By configuring the modulation factor control command generation unit 85 so that when the inverter output frequency FINV is in the region of approximately 120 Hz (120 Hz ± 30 Hz), the control value dV of the current control command can be generated and controlled such that the modulation coefficient PMF is 0 , 95.

The region in which the PMF modulation factor is set to 0.95 is explained as the region in which the inverter output frequency FINV has a value within 120 Hz (120 Hz ± 30 Hz). However, this is an example in which the frequency of the AC power source 230 is 60 Hz. This is because 120 Hz is equivalent to the 2f component at 60 Hz. On the other hand, when the frequency of the AC power source 230 is 50 Hz, since the 2f component is 100 Hz, the region in which the modulation factor PMF is set to 0.95 is the region in which the inverter output frequency FINV is 100 Hz (100 Hz ± 30 Hz).

In the configuration based on FIGS. 6 and 7, at the point where the modulation factor PMF exceeds a predetermined modulation factor control command PMF *, the input to the restriction unit 81 is reduced to be equal to or less than zero, and a negative control value may be generated dV current control commands. Therefore, it is possible to control the magnetic flux with attenuation to ensure that the output voltage of the inverter 2 matches the value set by the modulation factor control command PMF *.

In particular, when the voltage control command has a margin relative to the maximum output voltage of the inverter 2, the control value dV of the current control command is not output. At the point when the modulation factor PMF exceeds the modulation factor control command PMF * (the point when the voltage control command exceeds the maximum voltage specified by the modulation factor control command PMF *), a negative value occurs in the output of the restriction unit 81, and the control value dV of the command is output current control. Therefore, an optional d-axis id current is not supplied, and the electric current of the electric motor 6 can be minimized.

FIG. 8 is a diagram for explaining the relationship between the inverter output frequency FINV and the modulation factor transition PMF, the pulse mode transition, and the operation transition of the selection switch 59 (see FIG. 3) in the first embodiment. The acceleration of a vehicle with an electric motor from a stopped state is explained as an example below.

As shown in FIG. 8, when a vehicle with an electric motor moves at a low speed, i.e. when the inverter output frequency FINV is low, the modulation coefficient PMF is small, the pulse mode is an asynchronous multi-pulse mode, and A is selected as a select switch 59. On the other hand, when the speed of the vehicle with an electric motor increases, and the modulation coefficient PMF increases so as to be equal to or greater than 0.785, the output voltage of the inverter 2 is saturated in asynchronous multi-pulse mode. Therefore, the select switch 59 switches to B, and the pulse mode switches to the synchronous three-pulse mode. When the speed of the vehicle with the electric motor is further increased, and the modulation factor PMF reaches 1.0, the select switch 59 switches to C, and the pulse mode switches to the synchronous single-pulse mode.

Slowing a vehicle with an electric motor through the application of regenerative braking is not shown in the drawing. However, according to an order opposite to that explained above, the pulse mode changes from a synchronous single-pulse mode to a synchronous three-pulse mode and an asynchronous multi-pulse mode. The selection switch 59 is switched in the order of C, B, and A.

The advantages of a power conversion device for driving an electric motor according to this embodiment are explained with respect to control operations for the components explained above.

9 is a diagram of general control characteristics of permanent magnet synchronous motors in a first embodiment of the present invention and an example in the prior art. The control characteristics shown in FIG. 9 are control characteristics with respect to permanent magnet synchronous motors designed for electric vehicles. It is assumed that the maximum output torque is 1,500 Newton meters, and the input voltage of the EFC is 3,000 V. A permanent magnet synchronous motor operating in accordance with other parameters allows similar characteristics.

In Fig. 9, the abscissa represents the current id of the d-axis, and the ordinate represents the current iq of the q-axis. The many curves (solid lines) present from the part from the top right to the part from the bottom left of the drawing are fixed torque curves and are curves indicating the relationship between the d-axis id current and the q-axis current iq at the corresponding torques T described in left end of the drawing (relation between current vectors). The curve (dashed line) from the part from the top left to the part from the bottom right in the drawing is a curve indicating the minimum current mode, and is a curve on which the motor current is minimized when a certain torque T is output. In other words, the curve is a curve indicating the mode, in which the so-called maximum torque / current control is possible to generate maximum torque with minimum current.

If the current vector is controlled to the intersection of the curve indicating the minimum current mode and the fixed torque curve, it is possible to obtain a torque T with minimum current. By performing such control, such an advantage is provided that it is possible to minimize winding losses and losses in the inverter of the electric motor 6 when a certain torque T is obtained, and the size and weight of the electric motor 6 and the inverter 2 can be reduced. For example, when it is necessary to output the torque T to 1000 newton meters, if the current control is performed by inverter 2 in such a way that the d-axis id current is close to -125 A and the q-axis current iq is close to 225 A (point P1 shown in the drawing), 1000 newton meters m They can be formed by means of a minimum current.

In the drawing, the curves (lines with alternating long and short dashes) drawn in the form of an elevation are limit voltage curves, which are fixed induced voltage curves, and are curves indicating the relationship between the d-axis id current and q-axis iq current, wherein the voltage at the contact terminals of the electric motor 6 is maximized at a certain output frequency FINV of the inverter (the ratio between current vectors). The drawing shows the limit voltage curves in five cases (60 Hz, 90 Hz, 120 Hz, 150 Hz and 180 Hz), in which the output frequency FINV of the inverter is set as a parameter, provided that the input voltage EFC of the inverter 2 is set to 3000 V.

The combination of the d-axis current id and q-axis current iq, which can be logically selected (current vector), is located on the inside of the voltage limit curves (in the lower side of the curves). When the motor 6 operates with current vectors present on the lines of the limit voltage curves, the linear voltage of the motor 6 is maximized (i.e., the state in which the modulation coefficient PMF of inverter 2 is 1.0 and the maximum voltage is output). The torque T that can be output at this stage is the torque T at the intersection of the limit voltage curve and the fixed torque curve.

When the motor 6 operates with current vectors present on the inner side (lower side) of the voltage limit curves, the linear voltage of the motor 6 takes a value equal to or greater than zero and less than the maximum value (i.e., the modulation coefficient PMF of inverter 2 is less than 1.0) .

Current vectors present on the outer side of the limit voltage curves (upper side of the curves) cannot be selected, since the current vectors are in the region exceeding the maximum output voltage of the inverter 2.

As can be understood from the limiting voltage curves at the output frequencies of the FINV inverter (60 Hz, 90 Hz, 120 Hz, 150 Hz and 180 Hz) in these five cases, shown in Fig. 9, as the speed of the electric motor 6 increases and the output frequency The inverter FINV increases, the voltage limit curves move to the lower side of the drawing, the current vectors that can be selected are limited, and the maximum value of the torque T that can be output is reduced. As the inverter output frequency FINV increases, the torque T, which can be generated on a curve indicating the minimum current mode, also decreases.

When the EFC voltage across the capacitor rises, the voltage limit curve at the identical inverter output frequency FINV moves to the upper side in the drawing. When the EFC voltage across the capacitor drops, the voltage limit curve at the identical inverter output frequency FINV moves to the lower side in the drawing.

For example, when the inverter output frequency FINV of the inverter is 60 Hz, an operating point satisfying the minimum current mode with a maximum torque of 1500 Newton meters (near d-axis id current = -175 A, near q-axis current iq = 295 A, point A in the drawing), is a point sufficiently remote to the lower side of the limit voltage curve.

On the other hand, when the inverter output frequency FINV is 150 Hz, the maximum torque that can be generated is approximately 1200 Newton meters (point P2 in the drawing) near the current id d-axis = -245 A and near the current iq q- axis = 200 A on the limit voltage curve. Similarly, the maximum torque that can be generated in the minimum current mode is approximately 930 Newton meters (point P3 in the drawing) near the current id d-axis = -120 A and near the current iq q-axis = 210 A on the limit curve voltage. Undercurrent operation is not possible in the range from 930 Newton meters to 1200 Newton meters. This is an area in which operation is possible by performing magnetic flux control with attenuation to negatively increase the current id of the d-axis.

However, as the attenuation of the magnetic flux is performed deeper (as the current id of the d-axis decreases negatively), the current vector generated by the current id of the d-axis and current q of the q-axis increases, and the electric the current of the electric motor 6 increases.

In particular, in order to minimize losses in the winding of the electric motor 6 and losses of the inverter 2, it is desirable to control the inverter 2 so as to select a current vector (a combination of the id id of the d-axis and the current of the q-axis iq) on the curve of the minimum current mode as much as possible and instruct electric motor 6 to generate the required torque. When the output frequency FINV of the inverter increases according to an increase in the rotational speed of the electric motor 6, in the region where the required torque cannot be output on the minimum current mode curve due to the limitation of the limit voltage curve, in general, the d-axis id current is negatively increased, and the magnetic flow weakening.

In addition to the control in the minimum current mode explained above (maximum torque / current control), it is also possible to control the current vector on the maximum efficiency curve (not shown), for which losses of the electric motor 6, including losses in the core of the electric motor 6, are minimized, and apply control based on maximum efficiency to control the electric motor 6.

An operational characteristic for the present invention is explained, performed in the area in which the control switches to synchronous single-pulse mode (i.e., the area in which the modulation coefficient PMF takes a value close to 1.0) or when the inverter output frequency FINV is near 2f- component of the power source during operation in synchronous single-pulse mode.

First, the control operation in the above example is explained in order to clarify the details of the problem. Then, the problem solving means in the first embodiment of the present invention is explained with reference to FIG. 14 is a diagram for explaining an operating mode in an example in the prior art, in which MPM, 3PM and 1PM mean asynchronous multi-pulse mode, synchronous three-pulse mode and synchronous single-pulse mode, respectively. An example of control in which the electric motor 6 is started and accelerated from a state in which the electric motor 6 is stopped is shown in FIG. The operating points A, B, C1, D and E shown in Fig. 14 respectively correspond to the operating points A, B, C1, D and E shown in Fig. 9.

On Fig, firstly, the inverter 2 starts at zero time scale, the torque command T is set to 1500 Newton meters, and the voltage is applied to the electric motor 6 to start acceleration. Here, the PMF modulation factor increases from zero in proportion to the inverter output frequency FINV. Until the PMF modulation coefficient reaches 0.785 from zero of the time scale, the asynchronous multi-pulse mode is selected as the pulse mode of the inverter 2, and the torque T is fixed at 1500 Newton meters. Therefore, the electric motor 6 is linearly accelerated, and the inverter output frequency FINV is linearly increased.

At the point when the modulation coefficient reaches 0.785, the pulse mode switches to synchronous three-pulse mode. Between point A and point B, since the modulation coefficient PMF reaches a maximum value of 1.0, the pulse mode switches from a synchronous three-pulse mode to a synchronous single-pulse mode. Between point A (inverter output frequency FINV = 60 Hz) and point B (inverter output frequency FINV = 90 Hz), the torque command T decreases from 1,500 Newton meters in inverse proportion to the inverter output frequency FINV. After the modulation factor PMF reaches 1.0, the generated control command amount dV of the current control increases to the negative side according to the increase in the inverter output frequency FINV. Therefore, since the d-axis current control command id * is negatively increased, magnetic flux control with attenuation is performed. Therefore, the d-axis current control command id * is adjusted so that the modulation factor PMF coincides with the modulation coefficient control command PMF * (= 1.0).

The path of the current vector in the control mode explained above is explained with respect to FIG. 9. 9, as explained above, since the operating point A is on the lower side of the voltage limit curve and far from the voltage limit curve, the modulation coefficient PMF is less than 1.0, and the output voltage of the inverter 2 is a value smaller than the maximum voltage that can be output .

At operating point B, the torque command T is 1400 Newton meters, and the current vector is controlled to a point where the d-axis current control command id * is approximately -170 A and the q-axis current control command iq * is approximately 277 A At this operating point B, the current vector is also maintained on the limit voltage curve at FINV = 90 Hz. The control value dV of the current control command is generated and controlled in such a way that the modulation factor PMF is 1.0.

At operating point C1, the torque command T is 1200 Newton meters, and the current vector is controlled to a point where the d-axis current control command id * is -160 A and the q-axis current control command iq * is approximately 243 A. At this operating point C1, the current vector is also maintained on the limit voltage curve at FINV = 120 Hz. The control value dV of the current control command is generated and controlled in such a way that the modulation factor PMF is 1.0.

At operating point D, the torque command T is 1100 Newton-meters, and the current vector is controlled to a point where the d-axis current control command id * is -177 A and the q-axis current control command iq * is approximately 220 A. At this operating point D, the current vector is also maintained on the limit voltage curve at FINV = 150 Hz. The control value dV of the current control command is generated and controlled in such a way that the modulation factor PMF is 1.0.

At operating point E, the torque command T is 1000 Newton meters, and the current vector is controlled to a point where the d-axis current control command id * is -180 A and the q-axis current control command iq * is approximately 195 A. At this operating point E, the current vector is also maintained on the limit voltage curve at FINV = 180 Hz. The control value dV of the current control command is generated and controlled in such a way that the modulation factor PMF is 1.0.

Thus, in the control example in the prior art, the control operating point moves from operating point A to operating points B, C1, D and E. After the modulation factor PMF reaches 1.0, a control value dV of the current control command is generated to maintain the output voltage of inverter 2 at the maximum value that can be output while outputting the torque T (to maintain the modulation factor PMF = 1.0). Weakened magnetic flux control is performed according to the d-axis current control command id *, which includes a control value dV of the current control command.

According to the control, in the control example in the prior art, after the modulation coefficient reaches 1.0 to maintain the modulation coefficient PMF equal to 1.0 and keep the applied voltage to the AC motor at maximum, the synchronous single-pulse mode is selected in the switching state of the inverter circuit. In the working area of this synchronous single-pulse mode, as explained above, since the pulse width cannot be controlled, control to suppress the 2f component of the power source cannot be performed, in particular, in the region in which the inverter output frequency FINV is located near the 2f component of the source nutrition. Therefore, the problem arises that the AC motor generates current overload, and an excessively large torque ripple occurs.

The control operation in the first embodiment for solving the problem is explained with respect to FIG. 10. 10 is a diagram for explaining control modes in the first embodiment of the present invention. An example of control performed when the electric motor 6 is started and accelerated from a state in which the electric motor 6 is stopped is shown. The operating points A, B, C, D, and E shown in FIG. 10 respectively correspond to the operating points A, B, C, D, and E shown in FIG. 9.

In FIG. 10, firstly, the inverter 2 starts at zero time scale, the torque command T is set to 1500 Newton meters, and the voltage is applied to the electric motor 6 to start acceleration. The PMF modulation factor increases from zero in proportion to the inverter output frequency FINV. Until the PMF modulation coefficient reaches 0.785 from zero of the time scale, the asynchronous multi-pulse mode is selected as the pulse mode of the inverter 2, and the torque T is fixed at 1500 Newton meters. Therefore, the electric motor 6 is linearly accelerated, and the inverter output frequency FINV is linearly increased.

At the point when the modulation coefficient reaches 0.785, the pulse mode switches to synchronous three-pulse mode. Between point A and point B, since the modulation coefficient PMF reaches a maximum value of 1.0, the pulse mode switches from a synchronous three-pulse mode to a synchronous single-pulse mode. Between point A (inverter output frequency FINV = 60 Hz) and point B (inverter output frequency FINV = 90 Hz), the torque command T decreases from 1,500 Newton meters in inverse proportion to the inverter output frequency FINV. After the modulation factor PMF reaches 1.0, the generated control command amount dV is negatively increased according to the increase in the output frequency FINV of the inverter. Therefore, since the d-axis current control command id * is negatively increased, magnetic flux control with attenuation is performed. Therefore, the d-axis current control command id * is adjusted so that the modulation factor PMF coincides with the modulation coefficient control command PMF * (= 1.0). The above control operation is equivalent to the operation in the example in the prior art.

On the other hand, the region between the operating point B and the operating point D is the region in which the output frequency FINV of the inverter and the 2f component of the power supply are located next to each other. Operating point C is the operating point at which the inverter output frequency FINV is 120 Hz, and is the point at which the frequency of the 2f component of the power supply at a time when the frequency of the AC power source 230 is 60 Hz and the inverter output frequency FINV are exactly the same together.

Therefore, in this embodiment, in the range from the operating point B to the operating point D, which is a range in which the output frequency FINV of the inverter and the 2f component of the power source are close to each other or coincide with each other, the coefficient control command PMF * modulation decreases from 1.0 to 0.95. According to this control, a deviation occurs between the modulation coefficient PMF and the modulation coefficient control command PMF *. Therefore, the control value dV of the current control command is generated based on this deviation. The generated d-axis current control command id * operates on the basis of the control value dV of the current control command to further increase in the negative direction. Therefore, the d-axis current control command id * and the q-axis current control command iq * are formed as a current vector located on a fixed torque curve corresponding to the torque command T and on the inner side (lower side) of the limit voltage curve near FINV = 120 Hz. According to the d-axis current control command id * and the q-axis current control command iq * thus formed, the attenuation magnetic flux control applied to the electric motor 6 becomes deeper, and the induced voltage of the electric motor 6 further drops. Consequently, the PMF modulation coefficient also decreases. The modulation factor PMF is controlled to match the modulation factor control command PMF *.

In this area, a power conversion device for driving an electric motor according to this embodiment reduces the modulation coefficient PMF so that it is less than a conventional modulation coefficient in order to switch the pulse mode to a synchronous three-pulse mode, which is a synchronous pulse mode. Therefore, according to the ripple suppression signal BTPMFCMP, which is the output of the ripple suppression signal calculating unit 71, it is possible to adjust the pulse width for the output voltage output by the inverter 2. Control can be performed to suppress the 2f component of the power supply. According to this control, the problem can be solved in that the AC motor generates current overload and an excessively large torque ripple occurs.

Since the synchronous three-pulse mode, which is a synchronous pulse mode, is selected as the pulse mode, the number of pulses and the position of the pulses, respectively, included in the positive half cycle and negative half cycle of the inverter 2 are equal, and positive and negative symmetry of the voltage applied to the electric motor 6 is ensured. Therefore , it is possible to prevent the occurrence of current fluctuations and ripple of the torque in the electric motor 6, to prevent the occurrence of noise and fluctuations in the consequence of current fluctuations and ripple of the torque and to perform stable excitation of the electric motor 6. The operations after the operating point D are identical to the operations in the example in the prior art explained above.

Thus, in the control according to this embodiment, the operating point changes in the order of operating points A, B, C, D and E. The torque commands T at operating points A, B, C, D and E, respectively, are 1,500 Newton meters, 1400 Newton meters, 1200 Newton meters, 1100 Newton meters and 1000 Newton meters. The torque commands T are identical to the commands at the operating points A, B, C1, D and E of the example in the prior art. In other words, in this embodiment, while the control for suppressing the 2f component of the power source is executed, the output characteristic of the torque T is not affected at the operating points including the operating point C.

On the other hand, since the operating point C1 in the example in the prior art and the operating point C in this embodiment are on an identical fixed torque curve (at 1200 Newton meters) corresponding to a predetermined value of the torque control command, the torque output by the electric motor 6, is identical at both the operating point C1 and the operating point C. In other words, in this embodiment, the modulation factor PMF can be reduced, for example, to 0.95 while keeping the output torque of the motor 6 constant, lowering the induced voltage of the motor 6 and lowering the output voltage of the inverter 2. Since the modulation coefficient PMF decreases so as to be lower than the normal modulation coefficient, and the pulse mode switches to a synchronous three-pulse mode, which is a synchronous pulse mode, you can adjust the width of the voltage pulse output through the inverter 2, according to the output of block 71 calculation of the suppression signals ripple and perform control to suppress the 2f component of the power source. Therefore, the problem in the prior art is solved in that the AC motor generates current overload and an excessively large torque ripple occurs.

Current control commands are generated (d-axis current control command id * and q-axis current control command iq *) at which the modulation factor PMF is the same as the modulation factor control command PMF *. Therefore, it is possible to reduce the modulation factor PMF, for example, to 0.95 while maintaining the output torque of the electric motor 6 at a predetermined value, lowering the induced voltage of the electric motor 6 and lowering the output voltage of the inverter 2.

In the example explained above, the motor 6 is accelerated from a stopped state. However, the configuration explained in this embodiment can also be applied when the motor 6 undergoes a recovery operation and stops during high-speed rotation.

Second Embodiment

In a first embodiment, a configuration is disclosed in which the amount of regulation of the current control commands for adjusting the current control command for the power conversion device for driving the electric motor is controlled appropriately, or the switching of the pulse mode is controlled appropriately to provide control of the width of the voltage pulse output by the inverter 2, and provide efficient control execution to suppress the 2f component of the power source included in yhodnoe voltage of the inverter 2. In the second embodiment, a configuration in which converter the voltage control command to control the inverter 220 is further formed appropriately to provide effective reduction of the electric current supplied to the motor 6.

11 is a diagram of an example configuration of a power conversion device for driving an electric motor according to a second embodiment of the present invention. A more detailed configuration of the converter 220, which is the first power conversion unit, in the configuration of the power conversion device for driving the electric motor shown in FIG. 1, is explained below. Of the components shown in FIG. 11, components identical to those shown in FIG. 11 have already been explained. Therefore, mainly, components associated with the second embodiment are explained.

As shown in FIG. 11, the modulation coefficient PMF and the inverter output frequency FINV generated by the second control unit 100, the capacitor EFC detected by the voltage sensor 8, and the input current IS detected by the current sensor 214 are input to the first unit 200 management. This first control unit 200 is a component having an output voltage (DC voltage) control function of the converter 220, and includes a DC voltage command generation unit 210 and a DC voltage control unit 280.

The DC voltage control command generation unit 210 generates a DC voltage control command EFC *, which is a target value of a capacitor voltage and is a capacitor voltage control command EFC *. The voltage control unit 211 receives the input of the DC voltage command EFC * and the EFC voltage on the capacitor and generates a current control command IS * on the capacitor EFC * and the voltage EFC on the capacitor and outputs the current control command IS *. The current control unit 212 receives the input of the current control command IS * and the input current IS detected by the current sensor 214, and generates a converter voltage control command VC * based on the deviation between the current control command IS * and the input current IS. The pulse width modulation signal generating unit 213 receives the input of the converter voltage control command VC * and generates an on / off signal (pulse width modulation signal) CG for the switching element (not shown) of the converter 220 to ensure that the voltage on the input side (power supply side) matches AC) of converter 220 with a voltage control command VC *.

Using the functions of the voltage control unit 211, the current control unit 212, and the pulse width modulation signal generating unit 213, configured as explained above, the DC voltage control unit 280 generates a pulse width modulation signal CG using the constant voltage control command EFC * current, voltage EFC on the capacitor and input current IS and outputs a pulse width modulation signal CG to the converter 220.

The detailed configuration and operation of the DC voltage control command generation unit 210 is explained below with respect to FIG. 12. FIG. 12 is a diagram of a first configuration example of a DC voltage command generation unit 210 in a second embodiment shown in FIG. 11.

As shown in FIG. 12, the DC voltage control command generation unit 210, which is a first configuration example, includes a DC voltage control command table 240. The DC voltage control command table 240 generates, based on the inverter output frequency FINV, the DC voltage control command EFC * and outputs the DC voltage control command EFC *.

When the inverter output frequency FINV is not located near the frequency of the 2f component of the power supply, the DC voltage control command table 240 outputs the voltage during the estimated time as the DC voltage control command EFC *. On the other hand, when the inverter output frequency FINV is located near the frequency of the 2f component of the power supply, the DC voltage control command table 240 outputs a voltage control command above the voltage during the estimated time as the DC voltage control command EFC *.

For example, when the frequency of the AC power supply is 60 Hz, if the inverter output frequency FINV is not in the range of about 90-150 Hz, centered around 120 Hz, which is the frequency of the 2f component of the power supply, the DC voltage control command table 240 outputs for example 3000 V as an EFC * DC voltage command. If the inverter output frequency FINV is in the range of approximately 90-150 Hz, the DC voltage control command table 240 outputs, for example, 3300 V, which is obtained by increasing the voltage during the estimated time by approximately 5-10%, as the voltage control command EFC * direct current.

By configuring the DC voltage control command generation unit 210 so that when the inverter output frequency FINV is in a region near the frequency of the 2f component of the power source, for example, the 90-150 Hz region, the capacitor EFC can be controlled to be high, and, therefore, increase the maximum voltage that can be output through the inverter 2. According to this control, you can reduce the required magnitude of the magnetic flux with the weakening. As a result, it is also possible to reduce the dV value of the current command control and reduce the absolute value of the d-axis current control command id *. Therefore, it is possible to reduce the electric current of the electric motor 6 in comparison with the electric current output when the configuration of the first control unit 200 according to the second embodiment is not applied.

The DC voltage control command generation unit 210 is not limited to the configuration shown in FIG. 12, and may be configured, for example, as shown in FIG. 13. FIG. 13 is a diagram of a second configuration example of a DC voltage command command generation unit 210 in the second embodiment shown in FIG. 11.

The DC voltage control command generation unit 210 shown in FIG. 13 is a component that generates, based on the inverter output frequency FINV and the modulation factor control command PMF * as the target value of the modulation coefficient PMF (voltage amplitude index), the DC voltage control command EFC * current and includes a table 250 of commands for controlling the modulation coefficient, a subtraction unit 251, a restriction unit 252, a proportional integration unit 253, and an adder 254.

The modulation coefficient control command table 250 generates, based on the inverter output frequency FINV, the modulation coefficient control command PMF *. The subtraction unit 251 receives the input of the modulation factor PMF and the modulation factor control command PMF *, generates a deviation signal obtained by subtracting the modulation factor control command PMF * from the modulation factor PMF, and outputs the deviation signal to the limiting unit 252. When the sign of the input signal is a plus, the restriction unit 252 directly outputs the input signal. When the sign of the input signal is minus, the block 252 restriction outputs zero regardless of the value of the input signal. The proportional integration unit 253 outputs a value obtained by calculating the proportional integral output signal by the restriction unit 252. An adder 254 sums the output of the proportional-integration unit 253 and the DC basic voltage control command EFC0 * (e.g., 3000 V) and outputs the added signal as the DC voltage control command EFC *.

For example, when the frequency of the AC power supply is 60 Hz, if the inverter output frequency FINV is not in the range of about 90-150 Hz, centered around 120 Hz, which is the frequency of the 2f component of the power supply, the modulation factor control command table 250 outputs, for example 1.0 as a modulation factor control command PMF *. On the other hand, if the inverter output frequency FINV is in the range of about 90-150 Hz, the modulation coefficient control command table 250 outputs, for example, 0.95 as the modulation coefficient control command PMF *.

By configuring the DC voltage control command generation unit 210 so that when the inverter output frequency FINV is, for example, in the range of 90-150 Hz, the capacitor EFC can be increased so that the modulation factor PMF of inverter 2 is, for example, 0, 95. Therefore, it is possible to increase the maximum voltage that the inverter 2 can output. According to this control, the necessary magnitude of the magnetic flux can be reduced with attenuation. As a result, it is also possible to reduce the dV value of the current command control and reduce the absolute value of the d-axis current control command id *. Therefore, it is possible to reduce the electric current of the electric motor 6 in comparison with the electric current output when the configuration of the first control unit 200 according to the second embodiment is not applied.

With the configuration of the first embodiment explained above, it is possible to reduce the modulation factor PMF, for example, to 0.95 while maintaining the output torque of the electric motor 6 at a predetermined value of the control command, lowering the induced voltage of the electric motor 6 and lowering the output voltage of the inverter 2 in the region, in which the output frequency FINV of the inverter is located near the frequency of the 2f component of the power source. Therefore, since the pulse mode switches to the synchronous three-pulse mode, which is the synchronous pulse mode, it is possible to adjust the width of the voltage pulse outputted by the inverter 2 according to the ripple suppression signal BTPMFCMP, which is the output of the ripple suppression signal calculating unit 71, and perform control to suppress 2f component of the power source. Therefore, the problem is solved in that the AC motor generates current overload and an excessively large torque ripple occurs.

In the configuration of the first embodiment, current control commands (d-axis current control command id * and q-axis current control command iq *) are generated, in which the modulation coefficient PMF coincides with the modulation coefficient control command PMF *. Therefore, it is possible to reduce the modulation factor PMF, for example, to 0.95 while maintaining the output torque of the electric motor 6 at a predetermined value, lowering the induced voltage of the electric motor 6 and lowering the output voltage of the inverter 2.

In the configuration of the first embodiment, since the synchronous three-pulse mode, which is the synchronous pulse mode, is selected as the pulse mode, the number of pulses and the position of the pulses respectively included in the positive half cycle and the negative half cycle of inverter 2 are equal, and positive and negative voltage symmetry are ensured, applied to the electric motor 6. Therefore, it is possible to prevent the occurrence of current fluctuations and ripple of the torque in the electric motor 6, to prevent the occurrence of noise and oscillation due to current oscillations and torque ripples, and to perform a stable excitation of the motor 6.

Further, in the configuration of the second embodiment, the advantage of reducing the electric current of the electric motor 6 is large compared with the configuration of the first embodiment to which the second embodiment does not apply. Since the electric current of the electric motor 6 can be further reduced, the losses of the inverter 2 and the electric motor 6 can be further reduced.

In the explanation of the embodiments, the purpose of the explanation is a power conversion device for driving an electric motor that controls a permanent magnet synchronous electric motor. However, the control method explained above can be applied to power conversion devices for driving an electric motor that control so as to drive other types of electric motors.

The configurations explained in the embodiments are examples of the contents of the present invention. It goes without saying that the configurations can be combined with other well-known technologies and can be changed, for example, by excluding the part without departing from the essence of the present invention.

Additionally, this detailed description mainly explains the application to a power conversion device for driving an electric motor for a vehicle with an electric motor. However, the scope is not limited to this. It goes without saying that it is also possible to apply to other areas of industrial application.

Industrial applicability

As explained above, the power conversion device for driving the electric motor according to the present invention is useful as an invention that allows suppression control for the 2f component of the power source while suppressing the occurrence of overcurrent and excessively large ripple of the torque in the AC motor.

Explanation of Reference Designations

1 - capacitor

2 - second power conversion unit (inverter)

3, 4, 5 - current sensor

6 - electric motor

7 - rotation sensor

8 - voltage sensor

10 - block forming the current control commands

11 is a block generating commands for controlling the base current of the d-axis

14 - adder

15 - q-axis current command generation unit

20 - d-axis current control unit

21 - block calculation of isolation on the q-axis

22 - block calculation noise immunity d-axis

23 - q-axis current control unit

30 - block calculation of modulation coefficients

40 - control phase angle calculation unit

50 - a unit for generating voltage control commands / pulse width modulation signals

55 - voltage control command calculation unit

57 - block forming an asynchronous multi-pulse carrier signal

58 - block forming synchronous three-pulse carriers

59 - select switch

60 - pulse switching processor

61-63 - comparator

64-66 - inverting circuit

69 - block calculating the angular frequencies of the inverter

70 - multiplier

71 is a unit for calculating ripple suppression signals

72 - bandpass filter (BPF)

73 - adder

74 - divider

80 - control unit current control commands

81 - block restrictions

82 - amplifier

84 - subtraction block

85 - unit for generating modulation coefficient control commands

90 - block transform three-phase coordinates in the dq-axis

95 - block calculation of the reference phase angles

100 - second control unit

150 - block amplitude voltage index calculation

200 - first control unit

210 - unit for generating direct current voltage control commands

211 - voltage control unit

212 - current control unit

213 - pulse-width modulation signal generating unit

214 - current sensor

220 - first power conversion unit (converter)

230 - AC power source

240 - DC voltage control command table

250 - table modulation coefficient control commands

251 - subtraction block

252 - restriction block

253 - proportional integration unit

254 - adder

280 - DC voltage control unit

300 is a power conversion device for driving an electric motor.

Claims (20)

1. A power conversion device for driving an electric motor, comprising:
a first power conversion unit (220) that is connected to an AC power source (230) and converts the AC voltage from the AC power source to a DC voltage;
a second power conversion unit (2), which is connected to the first power conversion unit (220) and converts the DC voltage to AC voltage and outputs the AC voltage to the AC motor (6);
a first control unit (200) that controls the first power conversion unit (220); and
a second control unit (100) that controls the second power conversion unit (2), wherein:
the second control unit (100) includes:
a unit (10) for generating current control commands, which generates, based on at least a torque control command, a current control command for an alternating current electric motor;
a voltage amplitude index calculation unit (150) that calculates, based on a current control command, a voltage amplitude index to be applied to the alternating current motor (6);
a unit (80) for regulating current control commands, which generates, based on at least an index of the amplitude of the voltage and frequency of the AC motor, a control amount of the current control command for adjusting the current control command; and
a ripple suppression signal generating unit (71) that generates a ripple suppression signal (BTPMFCMP) based on the DC voltage, wherein
the second control unit (100) generates, based on a control signal including a current control command, adjustable by means of (dv) regulation of current control commands and a ripple suppression signal (BTPMFCMP), a pulse-width modulation signal into a second power conversion unit (2) and outputs a pulse width modulation signal. 2. The device according to claim 1, in which the second control unit (100) controls, when the frequency of the AC motor is in a predetermined range, by the voltage output by the second power conversion unit equal to a predetermined value less than the maximum voltage, which can be outputted in accordance with DC voltage. 3. The device according to claim 2, in which the second control unit (100) selects when the control to ensure that the torque provided by the AC motor matches the torque control command is performed by the current control command generation unit (10) and the control unit (80 ) regulation of current control commands, a current control command on a fixed torque line based on the torque control command and the inside of the limit voltage line. 4. The device according to claim 1, in which the block (80) of regulation of the current control commands includes a block (85) for generating target commands for controlling the voltage amplitude, which generates, based on the frequency of the AC motor, a target command for controlling the voltage amplitude indicating the maximum voltage amplitude index value. 5. The device according to claim 4, in which the unit (80) for regulating the current control commands generates, based on the deviation between the target voltage amplitude control command and the voltage amplitude index, the amount of regulation of the current control command. 6. The device according to claim 4, in which the unit (85) for generating the target commands for controlling the voltage amplitude generates, when the frequency of the AC motor is in a predetermined range, the target command for controlling the voltage amplitude for setting the output voltage of the second power conversion unit (2), equal to a predetermined value less than the maximum voltage that can be output in accordance with the DC voltage. 7. The device according to claim 1, in which:
the current control command generation unit (10) generates the first d-axis current control command, which is the current of the magnetic flux component of the AC motor, from the torque control command, adjusts the first d-axis current control command according to the amount of regulation of the current control command to generate the second d-axis current control command, and generates, on the basis of the torque control command and the second d-axis current control command, the first q-axis current control command, which is I component of the current torque, and
the voltage amplitude index calculation unit (150) calculates, based on the second d-axis current control command and the first q-axis current control command, the voltage amplitude index. 8. The device according to claim 1, in which:
the second control unit (100) includes:
a pulse mode switching unit (60) that switches the pulse mode of the second power conversion unit (2); and
a pulse mode selection unit (59) that selects, according to the control of the pulse mode switching unit (60), at least one of a plurality of pulse modes including an asynchronous pulse mode for generating a pulse width modulation signal asynchronously with the frequency of the AC motor and a synchronous pulse mode for generating a pulse width modulation signal synchronously with the frequency of the AC motor, and
the second control unit selects, when the frequency of the AC motor is present in a predetermined range centered around a frequency twice the frequency of the AC power source, the synchronous pulse mode as the pulse mode. 9. The device according to claim 1, in which:
the second control unit (100) includes:
a pulse mode switching unit (60) that switches the pulse mode of the second power conversion unit (2); and
a pulse mode selection unit (59) that selects, according to the control of the pulse mode switching unit, at least one of a plurality of pulse modes including an asynchronous pulse mode for generating a pulse width modulation signal asynchronously with the frequency of the AC motor and a synchronous three-pulse mode for generating a pulse-width modulation signal, the number of pulses of which in the half-period of voltage is three, generated synchronously with the frequency of the motor alternating current, and
the second control unit (100) selects, when the frequency of the AC motor is in a predetermined range centered around a frequency two times the frequency of the AC power source, the synchronous three-pulse mode as the pulse mode. 10. The device according to claim 8, in which the pulse mode selection unit (59) selects, based on at least the voltage amplitude index, not including the ripple suppression signal, the pulse mode. 11. The device according to claim 9, in which the pulse mode selection unit (59) selects, based on at least the voltage amplitude index, not including the ripple suppression signal, the pulse mode. 12. The device according to claim 2, in which the predefined range is a range centered around a frequency two times the frequency of the AC power source. 13. The device according to claim 2, in which the predetermined value is a value equal to or greater than 90% and less than 100% of the maximum voltage that can be output at constant voltage by the output voltage of the second power conversion unit (2). 14. A power conversion device for driving an electric motor, comprising:
a first power conversion unit (220) that is connected to an AC power source (230) and converts the AC voltage from the AC power source (6) to a DC voltage;
a second power conversion unit (2), which is connected to the first power conversion unit (220) and converts the DC voltage to AC voltage, and outputs the AC voltage to an AC electric motor;
a first control unit (200) that controls the first power conversion unit; and
a second control unit (100) that controls the second power conversion unit (2), wherein:
the second control unit (100) includes:
a unit (10) for generating current control commands, which generates, based on at least a torque control command, a current control command for an alternating current electric motor (6); and
a voltage amplitude index calculator (150) that calculates, based on a current control command, a voltage amplitude index to be applied to the alternating current motor, and
the first control unit (200) includes:
a DC voltage command generation unit (210) that generates a DC voltage control command, which is a target value of the DC voltage; and
a voltage control unit (280) that generates a control signal so that the DC voltage and the DC voltage control command coincide with each other, and
a DC voltage control command when the frequency of the AC motor is in a predetermined range is formed larger than a DC voltage control command generated when the frequency of the AC motor is not in a predetermined range. 15. The device according to 14, in which the block (210) generating commands for controlling the DC voltage generates, based on the frequency of the AC motor, a command for controlling the DC voltage. 16. The device according to 14, in which the block (210) generating commands for controlling the DC voltage generates a command for controlling the DC voltage based on the voltage amplitude index. 17. The device according to 14, in which the unit (210) for generating direct current voltage control commands generates a direct current voltage control command based on the frequency of the alternating current motor and the voltage amplitude index. 18. The device according to 14, in which the block (210) generating commands for controlling the DC voltage generates, based on the frequency of the AC motor, a target value indicating the maximum of the voltage amplitude index, and generates based on the target value indicating the maximum of the voltage amplitude index and index voltage amplitudes DC voltage control command. 19. The device according to 14, in which the predefined range is a range centered around a frequency two times the frequency of the AC power source. 20. The device according to 14, in which the predetermined value is a value equal to or greater than 90% and less than 100% of the maximum voltage that can be output at constant voltage by the output voltage of the second power conversion unit (2).

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