JP2019161748A - Inverter control method and inverter control device - Google Patents

Abstract

To provide an inverter control method or an inverter control device which can prevent occurrence of a control failure.SOLUTION: The control method is for controlling an inverter having a switching element by means of a controller. The method includes calculating a modulation wave for controlling output from the inverter, changing the carrier frequency of a carrier signal so as to match a target carrier frequency, generating a control command for controlling the switching element by comparing the modulation wave and the carrier signal, and changing a parameter for determining the target carrier frequency and/or a parameter for determining the modulation wave so as not to cause a failure in control of the inverter if the current carrier frequency is changed in a case where control of the inverter fails.SELECTED DRAWING: Figure 4

Description

  The present invention relates to an inverter control method and an inverter control device.

  In an electric vehicle motor control device, an inverter control device is known that lowers the carrier frequency when the motor rotation speed is low and the torque is high, and increases the carrier frequency when the rotation speed is high (Patent Document 1). ).

JP 2002-10668 A

  However, there is a problem that when the carrier frequency is changed, the modulation rate increases and the possibility of control failure is high.

  The problem to be solved by the present invention is to provide an inverter control method or an inverter control device capable of preventing the occurrence of control failure.

  The present invention changes at least one of the parameters for determining the target carrier frequency or the modulation wave so that the control of the inverter does not fail when the control of the inverter fails when the current carrier frequency is changed. This solves the above problem.

  According to the present invention, it is possible to prevent the occurrence of control failure.

FIG. 1 is a block diagram of a vehicle drive system according to the present embodiment. FIG. 2 is a block diagram of the drive system according to the present embodiment. FIG. 3 is a block diagram of the inverter control apparatus according to the present embodiment. FIG. 4 is a flowchart showing a control flow of the controller. FIG. 5 is a graph showing the relationship between time and frequency when the limit change amount (Δf_d) is large. FIG. 6 is a graph showing the relationship between time and frequency when the limit change amount (Δf_d) is small. FIG. 7 is a graph for explaining the temporal transition of the carrier signal, the vehicle parameter, and the modulation rate. FIG. 8 is a graph for explaining the temporal transition of the carrier signal, the vehicle parameter, and the modulation rate. FIG. 9 is a graph showing the relationship between the frequency before and after the change and the time when the difference (Δf) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft) is larger than the limit change amount (Δf_d). is there. FIG. 10 is a graph showing the relationship between the frequency before and after the change and the time when the difference (Δf) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft) is larger than the limit change amount (Δf_d). is there. FIG. 11 is a graph for explaining the temporal transition of the carrier signal, the vehicle parameter, and the modulation rate.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

<< First Embodiment >>
An inverter control device and an inverter control method according to an embodiment of the present invention will be described. FIG. 1 is a block diagram showing a vehicle drive system including an inverter control device according to an embodiment of the present invention. Hereinafter, the inverter control device of the present example will be described with reference to an example in which the inverter control device is provided to an electric vehicle. The inverter control device of the present example is a hybrid including at least a motor, such as a parallel type hybrid vehicle and a series type hybrid vehicle. It can also be applied to vehicles.

  As shown in FIG. 1, the vehicle control system including the inverter control device of this example includes a battery 1, an inverter 2, a motor 3, a speed reducer 4, drive wheels 5, and a controller 100. The vehicle drive system is not limited to the configuration shown in FIG. 1 and may include other configurations such as auxiliary devices.

  The battery 1 is a power source of a vehicle and is a battery group in which a plurality of secondary batteries are connected in series or in parallel. A lithium ion battery etc. are used for a secondary battery. The inverter 2 includes a plurality of switching elements such as IGBTs, and converts AC power output from the battery 1 into DC power by switching on and off the switching elements according to a switching signal from the controller 100. Further, the inverter 2 converts AC power generated by the regenerative operation of the motor 3 into DC power, and outputs the DC power to the battery 1. A current sensor is connected between the inverter 2 and the motor 3. The current sensor detects the current flowing through the motor 3 and outputs the detected value to the controller 100.

  The motor 3 is connected to a drive shaft of the vehicle and is a vehicle drive source that is driven by AC power from the inverter 2. The motor 3 is an electric motor such as a permanent magnet synchronous motor. Further, a rotation angle sensor is connected to the motor 3, and a detection value of the rotation angle sensor is output to the controller 100. The output shaft of the motor 3 is connected to the left and right drive wheels 5 via the speed reducer 4 and the left and right drive shafts. The motor 3 regenerates energy by generating a regenerative driving force by the rotation of the drive wheels 5.

  The controller 100 determines the vehicle state such as the accelerator opening, the vehicle speed and the gradient, the SOC of the battery 1, the chargeable / dischargeable power of the battery 1, the generated power of the motor 3, etc. A drive torque (necessary torque) for outputting torque in response to a driver's request is commanded to the inverter 2. The driver's request is determined by the accelerator operation and the brake operation.

  The controller 100 controls the motor 3 via the inverter 2 while optimizing the efficiency of the drive system of the vehicle according to the driving state of the vehicle and the state of the battery 1. In FIG. 1, one controller 100 is shown as a control part for controlling the vehicle, but the controller 100 may be a plurality of controllers such as a motor controller and an integrated controller. Various controllers are connected by a CAN communication network. The controller 100 includes a memory 110, a CPU 120, and the like.

  Next, a vehicle drive system will be described with reference to FIG. FIG. 2 is a block diagram of the drive system.

  The inverter 2 includes upper arm elements 41, 43, and 45 that form an upper arm circuit, lower arm elements 42, 44, and 46 that form a lower knitting circuit, a smoothing capacitor 47, a discharge resistor 48, and a discharge switch 49. And a drive circuit 50.

  The upper arm elements 41, 43, and 45 mainly have a circuit in which switching elements Q1, Q3, and Q5 as power devices and diodes D1, D3, and D5 are connected in parallel. The collector terminal of the switching element Q1 and the cathode terminal of the diode D1 are connected, and the collector terminal of the switching element Q1 and the anode terminal of the diode D1 are connected. Similarly, the lower arm elements 42, 44, and 46 are mainly configured by a circuit in which switching elements Q2, Q4, and Q6 as power devices and diodes D2, D4, and D6 are connected in parallel. The connection between switching element Q2 and switching element Q6 and diodes D2 to D6 is the same as the connection between switching element Q1 and diode D1.

  In the present embodiment, three pairs of circuits in which two switching elements Q1 to Q6 are connected in series are electrically connected to the battery 1 by being connected between the power supply line P and the power supply line N. Each connection point for connecting the switching elements is electrically connected to the three-phase output portion of the three-phase motor 3. The power line P is connected to the positive side of the battery 1, and the power line N is connected to the negative side of the battery 1.

  The connection point between the emitter terminal of the switching element Q1 and the collector terminal of the switching element Q2 is a U-phase output, and the connection point between the emitter terminal of the switching element Q3 and the collector terminal of the switching element Q4 is a V-phase output. The connection point between the emitter terminal of Q5 and the collector terminal of the switching element Q6 is a W-phase output and is connected to the three-phase wiring of the motor 3. The upper arm elements 41, 43, 45 and the lower arm elements 42, 44, 46 constitute a two-level three-phase inverter circuit 20.

  The smoothing capacitor 47 is an element that is connected between the inverter circuit 20 and the battery 1 and smoothes the electric power from the battery 1. The smoothing capacitor 47 is connected between the power supply lines P and N.

  The discharge resistor 47 and the discharge switch 49 are connected in series, and the series circuit of the discharge resistor 47 and the discharge switch 49 is connected between the power supply lines P and N. The discharge resistor 47 discharges the electric charge charged in the smoothing capacitor 47. The controller 100 controls on / off of the discharge switch 49. When the discharge switch 49 is turned on, the smoothing capacitor 47 and the discharge resistor 47 are brought into conduction, and discharge is performed.

  The drive circuit 50 has a function of switching on and off the switching elements S1 to S6 based on a switching signal transmitted from the controller 100.

  The motor 3 is connected to the connection point of the switching elements Q1 and Q2, the connection point of the switching elements Q3 and Q4, and the connection point of the switching elements Q5 and Q6 in each phase of the inverter circuit.

  The relay switch 7 is connected between the battery 1 and the smoothing capacitor 47 of the inverter 2.

  The controller 100 is a controller for controlling the drive circuit 50. Based on the torque command value input from the outside, the phase current of the motor 3, and the rotation speed (rotation speed) of the motor 3, the controller 100 outputs the required torque of the torque command value from the motor 3. Calculate the current command value. The phase current of the motor 3 is detected by a current sensor 8 connected between the inverter circuit 20 and the motor 3, and the rotational speed of the motor 3 is calculated from the detected value of the resolver 9 provided in the motor 3. The

  Then, the controller 100 generates a switching signal for supplying electric power required by the motor 3 and outputs it to the drive circuit 50. And the drive circuit 50 switches on / off of each switching element Q1-Q6 based on the said switching signal. Thereby, the controller 100 is controlling the inverter 2 by PWM control.

  Next, a control block related to the control of the inverter in the controller 100 will be described with reference to FIG. FIG. 3 is a block diagram of the inverter 2, the motor 3, and the controller 100. The controller 100 includes a rotation speed controller 21, a current command value calculator 22, a current controller 23, a non-interference controller 24, a two-phase three-phase voltage converter 25, a rotation speed calculator 26, and a three-phase two-phase current converter. 27. A switching signal (SW signal) controller 30 is provided.

ただし、Ｋ_ｐは比例ゲインを、Ｋ_Ｉは積分ゲインを、Ｋ_Ｄは微分ゲインを、Ｔ_Ｄは近似微分の時定数を、ｓはラプラス演算子を、ω_Ｇは回転数検出値を、ω_Ｇ ^＊は回転数指令値を示す。回転数指令値（ω_Ｇ ^＊）は、コントローラ１００により演算される目標値である。コントローラ１００は、ユーザの要求に応じた目標トルクを車両状態に応じて演算し、目標トルクを出力するために必要な回転数を、回転数指令値として演算している。 The rotation speed controller 21 matches the rotation speed detection value (ω _G : motor rotation speed) output from the rotation speed calculator 26 with the rotation speed command value (ω _G ^* ) of the motor 3 input from the outside. Thus, it is a PID controller that calculates the torque command value (T ^* ) of the motor 3. The rotation speed controller 21 receives the rotation speed command value (ω _G ^* ) and the rotation speed detection value (ω _G ), calculates a torque command value (T ^* ) by the following equation (1), and obtains a current command value. The result is output to the calculator 22.

However, the K _p is a proportional gain, the K _I is an integral gain, the _the K _D derivative gain, the T _D is the time constant of the approximate differentiation, s is a Laplace operator, the omega _G the rotation speed detection value, omega _G ^* indicates the rotational speed command value. The rotational speed command value (ω _G ^* ) is a target value calculated by the controller 100. The controller 100 calculates the target torque according to the user's request according to the vehicle state, and calculates the rotation speed necessary for outputting the target torque as the rotation speed command value.

The current command value calculator 22 inputs the torque command value (T ^* ), the voltage (V _dc ) of the battery 1, and the rotation speed detection value (ω _G ) indicating the angular frequency of the motor 3. The dq-axis current command value (I _d ^* , I _q ^* ) is calculated and output to the current controller 23. The dq axis represents the axis of the rotational coordinate system by the magnetic flux axis of the magnet and the axis orthogonal to the magnet axis. The current command value calculator 22 receives the dq-axis current command values (I _d ^* , I _q ^* ) using the torque command value (T ^* ), the rotation speed detection value (ω _G ), and the voltage (V _dc ) as indices. A map for output is stored in the memory 110. The map is an optimum command value that minimizes the loss of the motor 3 and the loss of the inverter 2 with respect to the input of the torque command value (T ^* ), the rotation speed detection value (ω _G ), and the voltage (V _dc ). Is output. Then, the current command value calculator 22 calculates a dq-axis current command value (I _d ^* , I _q ^* ) with reference to the map.

The current command value calculator 22 includes a dq axis current based on the detected value of the current sensor 8 in addition to the torque command value (T ^* ), the voltage (V _dc ) of the battery 1 and the rotation speed detection value (ω _G ). (I _d , I _q ) and chargeable / dischargeable power (P _in , P _out ) of the battery 1 are input, and the current command value calculator 22 calculates the dq axis current command value. The dq-axis current command value is a target value of the current of the motor 3, and includes an excitation current command value and a torque current command value.

ただし、Ｋ_ｐｄ、Ｋ_ｐｑは比例ゲインを、Ｋ_ｉｄ、Ｋ_ｉｑは積分ゲインを示す。 The current controller 23 receives the dq-axis current command value (I _d ^* , I _q ^* ) and the dq-axis current (I _d , I _q ) as inputs, performs control calculation using the following equation (2), The dq axis voltage command value (v _d ^* , v _q ^* ) is output. The dq-axis voltage command values (v _d ^* , v _q ^* ) are target values of the motor 3 voltage, and include an excitation voltage command value and a torque voltage command value.

Here, K _pd and K _pq indicate proportional gains, and K _id and K _iq indicate integral gains.

The current controller 23 may calculate the dq axis voltage command values (v _d ^* , v _q ^* ) with reference to the map corresponding to the above equation (2).

ただし、Ｌ_ｄはｄ軸インダクタンスを、Ｌ_ｑはｑ軸インダクタンスを、Ｒ_ａはモータ３の巻線抵抗を、ω_ｒｅは電気角速度を、φ_ａは磁束密度（トルク定数）を、ｐは微分演算子を示す。 The non-interference controller 24 calculates dq-axis non-interference voltages (v _ddcpl , v _qdcpl ) for canceling the generated interference voltage when current flows in the d-axis and q-axis of the motor 3. The voltage equation of the motor 3 is generally expressed by the following equation (3) when expressed in dq coordinates.

Where L _d is the d-axis inductance, L _q is the q-axis inductance, _Ra is the winding resistance of the motor 3, ω _re is the electrical angular velocity, φ _a is the magnetic flux density (torque constant), and p is the differential Indicates an operator.

When equation (3) is transformed by Laplace transform for each component, it is expressed by the following equation.

However, each of the current response models Gp is represented by the following equation.

As shown in Equation (3), there is a speed electromotive force that interferes between dq axes, and in order to cancel this, the non-interference controller 24 uses a non-interference voltage (v expressed by the following Equation (6). _ddcpl , _vqdcpl ).

A subtractor is provided on the output side of the current controller 23 and the non-interference controller 24. In the subtracter, the non-interference voltage (v _d ^* , v _q ^* ) represented by the equation (6) ( By subtracting v _ddcpl , v _qdcpl ), the interference term in equation (4) is canceled, and the dq-axis current is expressed by the following equation (7).

The two-phase three-phase voltage converter 25 receives the dq-axis voltage command value (v _d ^* , v _q ^* ) and the detected value θ of the resolver 9 as input, and uses the following equation (8) to Convert dq-axis voltage command values (v _d ^* , v _q ^* ) to u, v, w-axis voltage command values (v _u ^* , v _v ^* , v _w ^* ) in the fixed coordinate system, and switch SW controller Output to. The voltage command values (v _u ^* , v _v ^* , v _w ^* ) correspond to modulated waves in PWM control.

The three-phase two-phase current converter 27 is a control unit that performs three-phase to two-phase conversion. The phase current (I _u , I _v , I _w ) and the detected value θ of the magnetic pole position detector 52 are input as fixed coordinates. The phase currents (I _u , I _v , I _w ) of the system are converted into phase currents (I _d , I _q ) of the rotating coordinate system, and the current command value calculator 22, the current controller 23, and the non-interacting controller 34 are converted. Output to.

The current sensor 51 is provided for each of the U phase and the V phase, detects the motor current (I _u , I _v ), and outputs it to the three-phase two-phase current converter 27. The w-phase current is not detected by the current sensor 8. Instead, the three-phase two-phase current converter 27 calculates the w-phase current based on the input phase currents (I _u , I _v ). .

The resolver (rotation sensor) 9 is a detector that is provided in the motor 3 and detects the position of the magnetic pole of the motor 3, and outputs a detection value (θ) to the rotation speed calculator 26. The resolver 9 detects the rotation state of the motor 3 at a predetermined cycle. The rotation speed calculator 26 calculates a rotation speed detection value (ω _G ) that is an angular frequency of the motor 3 from the detection value (θ) of the resolver 9 and outputs it to the rotation speed controller 21 and the current command value calculator 22. .

Then, when the phase current (I _d , I _q ) is input to the current controller 23, the control device is controlled by a current control loop having a predetermined gain.

The SW signal controller 30 compares the voltage command values (v ^* _u , v ^* _v , v ^* _w ) with the carrier frequency, generates a PWM control signal based on the comparison result, and outputs the PWM control signal to the inverter 2 is output to the drive circuit 50 included in 2. The PWM control signal is a switching signal for switching on and off of the switching element, and is represented by a rectangular wave.

  The SW signal controller 30 includes a carrier frequency setting unit 31, a limit change amount calculation unit 32, and an SSW signal generation unit 33. The carrier frequency setting unit 31 selects one target carrier frequency from among a plurality of target carrier frequencies, and changes the carrier frequency so that the current carrier frequency matches the target carrier frequency. The inverter control device according to the present embodiment changes the carrier frequency in order to suppress noise. The carrier frequency setting unit 31 changes the carrier frequency at a predetermined cycle. Note that the carrier frequency setting unit 31 may change the carrier frequency at an arbitrary timing.

  The limit change amount calculation unit 32 calculates the limit change amount of the carrier frequency based on the vehicle parameter. The limit change amount indicates the upper limit value of the change amount of the carrier frequency, and indicates the change amount of the carrier frequency that does not cause a control failure when the carrier frequency is increased or decreased. For example, when the current carrier frequency is increased, if the carrier frequency is increased by a change amount larger than the limit change amount, the modulation rate becomes higher than 1 and a control failure occurs. Therefore, in the present embodiment, when changing the carrier frequency, the carrier frequency is changed so that control failure does not occur.

The vehicle parameters are represented by torque, magnetic flux density, phase current, inductance value, PID gain, and the like. The torque is an output torque of the motor 3 and is represented by a torque command value. The magnetic flux density corresponds to the magnetic flux density phi _a included in the formula (3). The phase current corresponds to a motor current, and is a current command value (I _d ^* , I _q ^* , I _u ^* , I _v ^* , I _w ^* ) or a voltage command value (v _d ^* , v _q ^* , v _u ^*). , V _v ^* , v _w ^* ). The inductance value is indicated by an inductance value (,) included in Equation (3). The PID gain corresponds to the gain (K _p , K _I , K _D ) included in Equation (1). The limit change amount is a value that changes according to the vehicle parameter. For example, when parameters other than torque are made constant among vehicle parameters, the limit change amount decreases as the torque increases. The limit change amount calculation unit 32 calculates the limit change amount within one cycle of the carrier frequency. Further, since the vehicle parameter is also used for calculating the modulated wave, the modulated wave changes when the vehicle parameter changes. That is, the vehicle parameter corresponds to a parameter for determining the modulated wave.

  The SW signal generation unit 33 compares the voltage command value output from the two-phase three-phase voltage converter 25 with the carrier signal. The carrier signal is a triangular wave having a carrier frequency set by the carrier frequency setting unit 31. The SW signal generation unit 33 sets a period during which the voltage command value (corresponding to the amplitude of the modulation wave) is larger than the amplitude of the carrier signal as an ON period of the switching element, and sets a period during which the voltage command value is equal to or less than the amplitude of the carrier signal. The switching signal is generated so as to be the ON period.

  Next, control of the controller 100 will be described with reference to FIG. FIG. 4 is a flowchart showing a control flow of the controller 100. The control flow shown in FIG. 4 is repeatedly executed at a predetermined cycle.

In step S1, the controller 100 acquires a rotation speed command value (ω _G ^* ) that changes in accordance with the driver's accelerator operation. In step S <b> 2, the controller 100 acquires the current motor speed from the magnetic pole position detector 52 based on the detection value θ, and acquires the current motor current from the current sensor 51.

  In step S3, the controller 100 calculates a modulated wave. The modulation wave is a sine wave indicated by a current command value or a voltage command value. The control process in step S3 corresponds to the calculation process of the current command value calculator 22, the current controller 23, and the two-phase three-phase voltage converter 25 shown in FIG.

  In step S <b> 4, the controller 100 determines whether to change the carrier frequency by the carrier frequency setting unit 31. When changing the carrier frequency periodically, the controller 100 determines to change the carrier frequency when the current time reaches the timing for changing the carrier frequency. When changing the carrier frequency, the control flow proceeds to step S5. On the other hand, when the carrier frequency is not changed, the control flow proceeds to step S10. The control processing from step S4 to step S8 is control processing executed before actually changing the carrier frequency, and the control processing of step S4 is executed before the time for actually changing the carrier frequency. .

  In step S5, the controller 100 calculates a limit change amount (Δf_d) based on the vehicle parameter. In step S6, the controller 100 calculates a difference (Δf) between the current carrier frequency and the target carrier frequency. For example, when the carrier frequency is lowered, the target carrier frequency is a frequency lower than the current carrier frequency.

  In step S7, the controller 100 compares the calculated difference (Δf) with the limit change amount (Δf_d), and determines whether or not the difference (Δf) is larger than the limit change amount (Δf_d). When the difference (Δf) is larger than the limit change amount (Δf_d), the control flow proceeds to step S8. If the difference (Δf) is equal to or less than the limit change amount (Δf_d), the control flow proceeds to step S9.

Here, the relationship between the carrier frequency and the control failure will be described with reference to FIGS. 5 and 6. FIG. 5 is a graph showing the relationship between time and frequency when the limit change amount (Δf_d) is large. FIG. 6 is a graph showing the relationship between time and frequency when the limit change amount (Δf_d) is small. 5 and FIG. 6, t ₀ denotes the current time, F ₀ represents the current carrier frequency. t ₁ indicates a time one cycle after the current carrier signal from the current time. Ft is the target carrier frequency.

As shown in FIG. 5, when the difference (Δf) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft) is equal to or less than the limit change amount (Δf_d), the current carrier frequency (F ₀ ) Even if is changed to the target carrier frequency (Ft), the control failure of the inverter 2 does not occur.

On the other hand, as shown in FIG. 6, when the difference (Δf) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft) is larger than the limit change amount (Δf_d), the current carrier frequency (F _0). ) Is changed to the target carrier frequency (Ft), the control failure of the inverter 2 occurs. Therefore, in this embodiment, when the current carrier frequency is changed in the control processing after step S8, when the inverter control fails, the vehicle parameter is changed so that the inverter control does not fail, and the inverter 2 is I have control.

  In step S8, the controller 100 changes the vehicle parameter so that the difference (Δf) between the current carrier frequency and the target carrier frequency is equal to or less than the limit change amount (Δf_d). When the vehicle parameter is changed, the amplitude of the modulated wave also changes. For example, when the torque is changed among the vehicle parameters and the parameters other than the torque are made constant, the control failure frequency of the inverter 2 increases as the torque is reduced. The control failure frequency is an upper limit value of the carrier frequency at which the control failure of the inverter 2 occurs when the carrier frequency is changed under the current driving state of the inverter 2. The difference between the current frequency and the control failure frequency corresponds to the limit change amount (Δf_d). That is, when the torque is decreased and the control failure frequency is increased, the limit change amount (Δf_d) increases. Therefore, by changing the vehicle parameter, the difference (Δf) can be made equal to or less than the limit change amount (Δf_d) without changing the target carrier frequency.

  As the vehicle parameter (torque) changes, the output torque of the motor decreases. For example, when the torque (phase current) is momentarily reduced while the motor is driven at a high rotation, the inertial force is high, and therefore the influence (behavior change) on the vehicle due to the torque reduction is small. Thus, in the present embodiment, when the current carrier frequency is changed to coincide with the target carrier frequency, the influence on the vehicle due to the change of the vehicle parameter is minimized while preventing the control failure of the inverter 2. Change the vehicle parameters to The controller 100 may appropriately select the vehicle parameter to be changed according to the current traveling state of the vehicle.

  In step S9, the controller 100 changes the carrier frequency so that the current carrier frequency matches the target carrier frequency. In step S10, controller 100 compares the carrier frequency and the modulated wave, and generates a switching signal (SW signal) based on the comparison result. Then, the control flow of the controller 100 ends.

The control of the controller 100 will be described in time series with reference to FIG. FIG. 7 is a graph for explaining the temporal transition of the carrier signal, the vehicle parameter, and the modulation rate. In the graph of FIG. 7, the time (t = 0) corresponds to the current time, and the time (t = 1) is the time when one cycle of the carrier signal has elapsed from the time (t = 0). This corresponds to the timing of changing the frequency. _Ath indicates the upper limit value of the modulation rate. When the modulation rate becomes higher than _Ath , the control failure of the inverter 2 occurs.

  The controller 100 executes the control processing of steps S1 to S8 from the timing (t = 0) one cycle before the current carrier signal (t = 1), and at the next time (t = 1). A target carrier frequency and a modulation wave to be applied are calculated. Further, the controller 100 calculates the difference (Δf) and the limit change amount (Δf_d) from the time (t = 0) to the time (t = 1), and calculates the difference (Δf) and the limit change amount (Δf_d). Compare

  When the difference (Δf) is equal to or less than the limit change amount (Δf_d), the controller 100 changes the carrier frequency at time (t = 1). On the other hand, when the difference (Δf) is larger than the limit change amount (Δf_d), the controller 100 causes the difference (Δf) to be equal to or less than the limit change amount (Δf_d) before the time (t = 1). The vehicle parameter is calculated, and the current vehicle parameter is changed to the calculated vehicle parameter. As the vehicle parameter is changed, the control failure frequency increases, so that the limit change amount (Δf_d) also increases (modulation rate decreases), and the difference (Δf) becomes equal to or less than the limit change amount (Δf_d).

Unlike the present embodiment, when the difference (Δf) is larger than the limit change amount (Δf_d), when the current carrier frequency is changed to the target carrier frequency without changing the vehicle parameter, the modulation rate of FIG. As shown by the dotted line graph, the modulation rate becomes larger than the upper limit value (A _th ) of the modulation rate, and control failure occurs. On the other hand, in this embodiment, since the vehicle parameter is changed so that the difference (Δf) is equal to or less than the limit change amount (Δf_d), the modulation rate is the modulation rate as shown in the solid line graph of the modulation rate in FIG. Since it is below the upper limit (A _th ), no control failure occurs.

  FIG. 8 is a graph for explaining the temporal transition of the carrier signal, the vehicle parameter, and the modulation rate. In FIG. 8, (а) shows a graph when the motor is rotating at a high speed, and (b) shows a graph when the motor is at a low speed. In the example of FIG. 8, the controller 100 increases the limit change amount (Δf_d) by reducing the phase current among the vehicle parameters.

  When the motor speed is high, the limit change amount (Δf_d) is small. Therefore, as shown in FIG. 8A, the amount of decrease in the phase current (vehicle parameter) is increased so that the difference (Δf) is equal to or less than the limit change amount (Δf_d). On the other hand, when the motor speed is low, the limit change amount (Δf_d) is large. Therefore, as shown in FIG. 8B, the decrease amount of the phase current (vehicle parameter) is reduced so that the difference (Δf) is equal to or less than the limit change amount (Δf_d). As described above, the controller 100 adjusts the change amount of the vehicle parameter according to the vehicle state so that the difference (Δf) is equal to or less than the limit change amount (Δf_d).

  As described above, in this embodiment, when the current carrier frequency is changed, when the control of the inverter 2 fails, the parameter (vehicle parameter) for determining the modulation wave so that the control of the inverter 2 does not fail To change. Thereby, it is possible to prevent control failure when changing the carrier frequency.

  In the present embodiment, the limit change amount of the carrier frequency is calculated based on the vehicle parameter, and if the difference between the current carrier frequency and the target carrier frequency is larger than the limit change amount, the difference is equal to or less than the limit change amount. Thus, the vehicle parameter is changed, and the modulated wave is calculated based on the changed vehicle parameter. Thereby, when changing the carrier frequency, it is possible to change the carrier frequency so that the control of the inverter 2 does not fail while obtaining a condition (control failure limit) that causes the control failure by calculation.

  In the present embodiment, the change period when changing the carrier frequency may be shorter than one period of the modulated wave. That is, the controller 100 changes the carrier frequency a plurality of times per cycle of the modulated wave. Thereby, noise can be suppressed.

  In the above description, the case where the carrier frequency is increased has been described. However, the control according to the present embodiment may be applied when the carrier frequency is decreased.

<< Second Embodiment >>
An inverter control device according to another embodiment of the present invention will be described. The present embodiment is different from the first embodiment in that the controller 100 changes the carrier frequency so that the difference (Δf) is equal to or less than the limit change amount (Δf_d). Other configurations and control methods are the same as those in the first embodiment described above, and the description thereof is incorporated.

As in the first embodiment, the controller 100 calculates the difference (Δf) and the limit change amount (Δf_d) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft), and the difference (Δf) is the limit. It is determined whether or not it is larger than the change amount (Δf_d). In the first embodiment, when the difference (Δf) is larger than the limit change amount (Δf_d), the vehicle parameter is changed without changing the target carrier frequency. On the other hand, in the second embodiment, when the difference (Δf) is larger than the limit change amount (Δf_d), the target carrier frequency is changed without changing the vehicle parameter. That is, the controller 100 changes the target carrier frequency so that the difference (Δf) is equal to or less than the limit change amount (Δf_d).

FIG. 9 is a graph showing the relationship between the frequency before and after the change and the time when the difference (Δf) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft) is larger than the limit change amount (Δf_d). is there. Ft represents the target carrier frequency before the change, and Fp represents the target carrier frequency after the change.

As shown in FIG. 9, when the difference (Δf) is larger than the limit change amount (Δf_d), the controller lowers the target carrier frequency (Ft) before the change and decreases the target carrier frequency (Fp) after the change. New calculation. At this time, the reduction range of the target carrier frequency is set in consideration of the influence on the vehicle behavior while taking into account the abnormal vibration and the ripple voltage effect at the time of changing the carrier. The controller 100 may calculate the target carrier frequency (Fp) by adding a change amount smaller than the limit change amount (Δf_d) or the limit change amount (Δf_d) to the current carrier frequency. Then, the controller 100 changes the carrier frequency so that the current carrier frequency (F ₀ ) matches the target carrier frequency (Ft). Then, the controller 100 generates a switching signal based on the changed carrier frequency (Fp) and the modulated wave.

  As described above, in the present embodiment, when the control of the inverter 2 fails when the current carrier frequency is changed, the target carrier frequency is changed so that the control of the inverter 2 does not fail. Thereby, it is possible to prevent control failure when changing the carrier frequency.

  In the present embodiment, the limit change amount of the carrier frequency is calculated based on the vehicle parameter, and if the difference between the current carrier frequency and the target carrier frequency is larger than the limit change amount, the difference is equal to or less than the limit change amount. So that the target carrier frequency is changed. Thereby, when changing the carrier frequency, it is possible to change the carrier frequency so that the control of the inverter 2 does not fail while obtaining a condition (control failure limit condition) that causes the control failure by calculation.

  In the above description, the case where the carrier frequency is increased has been described. However, the control according to the present embodiment may be applied when the carrier frequency is decreased.

  As a modification of the present embodiment, the controller 100 may adjust the change amount of the carrier frequency according to the magnitude of the limit change amount when changing the carrier frequency.

  Further, as a modification of the present embodiment, when the difference between the current carrier frequency and the target carrier frequency is larger than the limit change amount, the controller 100 sets the target carrier frequency and the vehicle so that the difference becomes equal to or less than the limit change amount. You may change the parameters. That is, in a modification, you may combine control which changes a vehicle parameter like 1st Embodiment, and control which changes target carrier frequency like 2nd Embodiment.

<< Third Embodiment >>
An inverter control device according to another embodiment of the present invention will be described. The present embodiment is different from the second embodiment in that the controller 100 changes the carrier frequency by increasing it stepwise when changing the carrier frequency. Other configurations and control methods are the same as those in the second embodiment described above, and the description thereof is incorporated.

  The controller 100 calculates a limit change amount of the carrier frequency based on the vehicle parameter. The limit change amount represents a change amount (change amount per unit time) of a carrier frequency that does not cause control failure when the carrier frequency is changed per unit time. The unit time represents a time interval when the carrier frequency is changed by one step.

The controller 100 calculates the difference (Δf) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft), and determines whether or not the difference (Δf) is larger than the limit change amount (Δf_d). When the difference (Δf) is larger than the limit change amount (Δf_d), the controller 100 limits the change amount of the carrier frequency so that the current carrier frequency (F ₀ ) matches the target carrier frequency (Ft). The carrier frequency is changed in a stepwise manner while keeping it below the amount (Δf_d). The amount of change in frequency per unit time is set in consideration of the influence on the vehicle behavior while taking into account the abnormal vibration at the time of changing the carrier, the ripple voltage effect, and the like.

FIG. 10 is a graph showing the relationship between the frequency before and after the change and the time when the difference (Δf) between the current carrier frequency (F ₀ ) and the target carrier frequency (Ft) is larger than the limit change amount (Δf_d). is there. Tp corresponds to a period when the carrier frequency is increased stepwise.

As shown in FIG. 10, the controller 100, while suppressing the variation of the carrier frequency less than the target carrier frequency (Ft), from the current carrier frequency (F ₀₎ to the target carrier frequency (Ft), stepwise the carrier frequency Has been raised.

  FIG. 11 is a graph for explaining the temporal transition of the carrier signal, the vehicle parameter, and the modulation rate. In FIG. 11, (а) shows a graph when the motor rotates at high speed, and (b) shows a graph when the motor rotates at low speed.

  When the motor speed is high, the limit change amount (Δf_d) is small. For this reason, as shown in FIG. 11A, the number of steps when increasing the carrier frequency is increased to increase the carrier frequency. On the other hand, when the motor speed is low, the output torque of the motor is small and the limit change amount (Δf_d) is large. Therefore, as shown in FIG. 11B, the number of steps when increasing the carrier frequency is reduced to increase the carrier frequency. That is, the controller 100 sets the number of steps when changing the carrier frequency step by step according to the parameter.

  As described above, in the present embodiment, when the carrier frequency is changed from the current carrier frequency (corresponding to the first carrier frequency of the present invention) to the target carrier frequency (corresponding to the second carrier frequency), When the difference between the carrier frequency and the target carrier frequency is larger than the limit change amount (Δf_d), the carrier frequency is changed stepwise from the current carrier frequency to the target carrier frequency. Thereby, it is possible to prevent control failure when changing the carrier frequency while suppressing noise generation.

  In this embodiment, the number of steps when changing the carrier frequency stepwise is set according to the parameter. Thereby, it is possible to prevent control failure when changing the carrier frequency while suppressing noise generation.

  In the above description, the case where the carrier frequency is increased stepwise has been described. However, the control according to the present embodiment may be applied when the carrier frequency is decreased stepwise.

DESCRIPTION OF SYMBOLS 1 ... Battery 2 ... Inverter 3 ... Motor 4 ... Reduction gear 5 ... Drive wheel 7 ... Relay switch 8 ... Current sensor 9 ... Resolver (rotation sensor)
DESCRIPTION OF SYMBOLS 21 ... Speed controller 22 ... Current command value calculator 23 ... Current controller 24 ... Non-interference controller 25 ... Two phase three phase voltage converter 26 ... Speed calculator 27 ... Three phase two phase current converter 30 ... Switching signal (SW signal) controller 31... Carrier frequency setting unit 32... Limit change calculation unit 33... Signal generation units 41, 43, 45.
42, 44, 46 ... Lower arm element (switching element)
DESCRIPTION OF SYMBOLS 50 ... Drive circuit 51 ... Current sensor 52 ... Magnetic pole position detector 100 ... Controller

Claims (8)

A control method for controlling an inverter having a switching element by a controller,
Calculate the modulation wave that controls the output from the inverter,
Change the carrier frequency of the carrier signal to match the target carrier frequency,
By comparing the modulation wave and the carrier signal, a control command for controlling the switching element is generated,
If the control of the inverter fails when the current carrier frequency is changed, at least one of the parameters for determining the target carrier frequency or the modulation wave is changed so that the control of the inverter does not fail Inverter control method. An inverter control method according to claim 1,
Based on the parameter, the limit change amount of the carrier frequency is calculated,
If the difference between the current carrier frequency and the target carrier frequency is greater than the limit change amount, change the parameter so that the difference is less than or equal to the limit change amount,
An inverter control method for calculating the modulated wave based on the changed parameter. An inverter control method according to claim 1,
Based on the parameter, the limit change amount of the carrier frequency is calculated,
When the difference between the current carrier frequency and the target carrier frequency is larger than the limit change amount, the inverter control method of changing the target carrier frequency so that the difference becomes equal to or less than the limit change amount. An inverter control method according to claim 1,
Based on the parameter, the limit change amount of the carrier frequency is calculated,
When the carrier frequency is changed from the first carrier frequency to the second carrier frequency, when the difference between the first carrier frequency and the second carrier frequency is larger than the limit change amount, the first carrier frequency is changed from the first carrier frequency to the second carrier frequency. An inverter control method for changing the carrier frequency stepwise up to a second carrier frequency. An inverter control method according to claim 4,
An inverter control method for setting the number of steps when changing the carrier frequency stepwise according to the parameter. An inverter control method according to any one of claims 1 to 5,
The inverter control method, wherein the parameter indicates at least one of a rotation speed of a motor driven by an output from the inverter, a torque of the motor, a magnetic flux density of the motor, and a control gain of the inverter. An inverter control method according to any one of claims 1 to 6,
An inverter control method for changing the carrier frequency a plurality of times per cycle of the modulated wave. A controller for controlling an inverter having a switching element;
A sensor for detecting a state of a load connected to the inverter;
The controller is
Based on the output command value of the load and the detection value of the sensor, the modulation wave that controls the output from the inverter is calculated,
Change the carrier frequency of the carrier signal to match the target carrier frequency,
By comparing the modulation wave and the carrier signal, a control command for controlling the switching element is generated,
If control of the inverter fails when the current carrier frequency is changed, change at least one of the target carrier frequency or the parameter for determining the modulation wave so that the control of the inverter does not fail Inverter control device.

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