WO2018116668A1 - Motor control device and electric vehicle - Google Patents

Motor control device and electric vehicle Download PDF

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Publication number
WO2018116668A1
WO2018116668A1 PCT/JP2017/040034 JP2017040034W WO2018116668A1 WO 2018116668 A1 WO2018116668 A1 WO 2018116668A1 JP 2017040034 W JP2017040034 W JP 2017040034W WO 2018116668 A1 WO2018116668 A1 WO 2018116668A1
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WIPO (PCT)
Prior art keywords
zero
phase
current
command value
control device
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PCT/JP2017/040034
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French (fr)
Japanese (ja)
Inventor
隆宏 荒木
利貞 三井
宮崎　英樹
矩也 中尾
Original Assignee
日立オートモティブシステムズ株式会社
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Application filed by 日立オートモティブシステムズ株式会社 filed Critical 日立オートモティブシステムズ株式会社
Priority to CN201780078740.9A priority Critical patent/CN110089022B/en
Publication of WO2018116668A1 publication Critical patent/WO2018116668A1/en

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    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L15/00—Methods, circuits, or devices for controlling the traction-motor speed of electrically-propelled vehicles
    • B60L15/20—Methods, circuits, or devices for controlling the traction-motor speed of electrically-propelled vehicles for control of the vehicle or its driving motor to achieve a desired performance, e.g. speed, torque, programmed variation of speed
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L9/00—Electric propulsion with power supply external to the vehicle
    • B60L9/16—Electric propulsion with power supply external to the vehicle using ac induction motors
    • B60L9/18—Electric propulsion with power supply external to the vehicle using ac induction motors fed from dc supply lines
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/10—Arrangements for controlling torque ripple, e.g. providing reduced torque ripple
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/64—Electric machine technologies in electromobility
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility

Definitions

  • the present invention relates to a motor control device and an electric vehicle.
  • Hybrid motors and electric vehicles are required to improve motor output from the viewpoint of improving driving force.
  • an effective current value is increased by using a pseudo-rectangular wave current as a driving current of a motor, thereby improving an output.
  • the motor drive current is always a pseudo-rectangular wave, which may increase torque ripple.
  • a motor control device controls a drive current of a motor having windings wound independently between phases based on a torque command value, a motor angular velocity, and a rotor position.
  • the ratio of the zero-phase current in the drive current is changed according to the magnitude of the torque command value.
  • FIG. 1 is a diagram showing a schematic configuration of a hybrid electric vehicle equipped with a motor according to an embodiment of the present invention.
  • FIG. 2 is a block diagram illustrating a configuration of the power conversion apparatus.
  • FIG. 3 is a block diagram showing details of the motor control device.
  • FIG. 4 is a flowchart illustrating an example of a current command calculation process.
  • FIG. 5 is a diagram illustrating the drive current.
  • FIG. 6 is a block diagram illustrating details of the zero-phase counter electromotive force compensation unit.
  • FIG. 7 is a flowchart for explaining the first modification.
  • FIG. 8 is a diagram illustrating a drive current, a sine wave component of the drive current, and a zero-phase current in the first modification.
  • FIG. 9 is a block diagram illustrating a motor control device according to a second modification.
  • FIG. 1 is a diagram showing a schematic configuration of a hybrid electric vehicle equipped with a motor according to an embodiment of the present invention.
  • the vehicle 100 is equipped with an engine 120, a motor 200, and a battery 180.
  • the battery 180 supplies DC power to the motor 200 via the power converter 600 when the driving force by the motor 200 is necessary, and receives DC power from the motor 200 during regenerative travel. Transfer of direct-current power between the battery 180 and the motor 200 is performed via the power converter 600.
  • the vehicle 100 is equipped with a battery for supplying low voltage power (for example, 14 volt system power), and is used as a power source for a control system, for example.
  • low voltage power for example, 14 volt system power
  • Rotational torque generated by the engine 120 and the motor 200 is transmitted to the front wheels 110 via the transmission 130 and the differential gear 160.
  • the transmission 130 is controlled by a transmission control device 134.
  • the engine 120 is controlled by the engine control device 124.
  • the battery 180 is controlled by the battery control device 184.
  • Transmission control device 134, engine control device 124, battery control device 184, power conversion device 600 and integrated control device 170 are connected by communication line 174.
  • the integrated control device 170 is a higher-level control device than the transmission control device 134, the engine control device 124, the power conversion device 600, and the battery control device 184.
  • the integrated control device 170 receives information representing the states of the transmission control device 134, the engine control device 124, the power conversion device 600, and the battery control device 184 from each of them via the communication line 174.
  • the integrated control device 170 calculates a control command for each control device based on the acquired information. The calculated control command is transmitted to each control device via the communication line 174.
  • the high voltage battery 180 is composed of a secondary battery such as a lithium ion battery or a nickel metal hydride battery, and outputs a high voltage DC power of 250 to 600 volts or more.
  • the battery control device 184 outputs the charge / discharge status of the battery 180 and the state of each unit cell battery constituting the battery 180 to the integrated control device 170 via the communication line 174.
  • the integrated control device 170 determines that the battery 180 needs to be charged based on the information from the battery control device 184, the integrated control device 170 instructs the power conversion device 600 to perform a power generation operation. Further, the integrated control device 170 mainly performs management processing of the output torque of the engine 120 and the motor 200, calculation processing of the total torque and torque distribution ratio between the output torque of the engine 120 and the output torque of the motor 200, and the calculation processing. A control command based on the result is transmitted to transmission control device 134, engine control device 124, and power conversion device 600. Based on the torque command from the integrated control device 170, the power conversion device 600 controls the motor 200 so that a torque output or generated power according to the command is generated.
  • the power conversion device 600 includes an inverter for operating the motor 200 and a motor control device that generates a switching signal to the inverter.
  • the power conversion device 600 controls the inverter based on a command from the integrated control device 170 to operate the motor 200 as an electric motor or a generator.
  • FIG. 2 is a block diagram showing the configuration of the power conversion apparatus 600.
  • the power conversion device 600 includes a motor control device 610, an inverter 620, and a current sensor 220.
  • the motor 200 is constituted by an embedded magnet synchronous motor or the like to which no neutral point is connected.
  • the motor 200 is provided with a position sensor 210 that detects the position of the rotor and outputs the detected rotor position ⁇ .
  • Current sensor 220 detects currents flowing through U-phase winding 201, V-phase winding 202, and W-phase winding 203 wound around the stator of motor 200, and outputs the detected three-phase currents iu, iv, iw. To do.
  • the inverter 620 is provided with a U-phase full-bridge inverter 621, a V-phase full-bridge inverter 622, and a W-phase full-bridge inverter 623, which are connected in parallel to a battery 180 that is a DC power source.
  • the U phase winding 201 of the motor 200 is connected to the output terminal of the U phase full bridge inverter 621
  • the V phase winding 202 is connected to the output terminal of the V phase full bridge inverter 622
  • the W phase winding 203 is W phase full.
  • the motor 200 is not connected to the neutral point, and can control the current flowing through the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 independently.
  • the U-phase full bridge inverter 621 includes switching elements 621a to 621d.
  • Switching element 621a is disposed on the U-phase left leg upper arm.
  • Switching element 621b is arranged on the U-phase left leg lower arm.
  • Switching element 621c is arranged on the U-phase right leg upper arm.
  • Switching element 621d is arranged on the U-phase right leg lower arm.
  • the V-phase full bridge inverter 622 includes switching elements 622a to 622d.
  • Switching element 622a is disposed on the V-phase left leg upper arm.
  • Switching element 622b is disposed on the V-phase left leg lower arm.
  • Switching element 622c is arranged on the V-phase right leg upper arm.
  • Switching element 622d is arranged on the lower arm of the V-phase right leg.
  • the W-phase full bridge inverter 623 includes switching elements 623a to 623d.
  • Switching element 623a is disposed on the W-phase left leg upper arm.
  • Switching element 623b is arranged on the lower arm of the W-phase left leg.
  • Switching element 623c is disposed on the W-phase right leg upper arm.
  • Switching element 623d is disposed on the W-phase right leg lower arm.
  • the switching elements 621a to 621d, the switching elements 622a to 622d, and the switching elements 623a to 623d are configured by combining a metal oxide film field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), and the like with a diode.
  • MOSFET metal oxide film field effect transistor
  • IGBT insulated gate bipolar transistor
  • the inverter 620 is applied from the battery 180 by turning on or off the switching elements 621a to 621d, the switching elements 622a to 622d, and the switching elements 623a to 623d based on the switching signal generated by the motor control device 610.
  • DC voltage is converted to AC voltage.
  • the converted AC voltage is applied to the three-phase windings 201 to 203 wound around the stator of the motor 200 to generate a three-phase AC current.
  • the three-phase alternating current generates a rotating magnetic field in the motor 200, and the rotor of the motor 200 rotates.
  • the motor control device 610 is based on the torque command value T * from the integrated control device 170, the three-phase currents i u , i v , i w detected by the current sensor 220, and the rotor position ⁇ detected by the position sensor 210.
  • the inverter 620 is PWM controlled.
  • FIG. 3 is a block diagram showing details of the motor control device 610.
  • the motor control device 610 includes a current command calculation unit 10, a dq axis current control unit 20, a switching signal generation unit 30, a dq conversion unit 40, a zero phase current calculation unit 50, a zero phase current control unit 60, a speed conversion unit 70, and a zero.
  • a phase back electromotive force compensation unit 80 is provided.
  • the dq converter 40 Based on the three-phase currents i u , i v , i w detected by the current sensor 220 and the rotor position ⁇ detected by the position sensor 210, the dq converter 40 detects the dq-axis current detection value i d , i q is output.
  • the speed conversion unit 70 outputs the angular velocity ⁇ of the rotor based on the rotor position ⁇ detected by the position sensor 210.
  • Zero-phase current calculation section 50 the three-phase currents i u entered, i v, calculates a zero-phase current i z based on i w.
  • the zero-phase current iz is calculated as in the following formula (1).
  • i z i u / ⁇ 3 + i v / ⁇ 3 + i w / ⁇ 3 (1)
  • the current command calculation unit 10 determines the dq-axis current command values i d * and i q * and the zero-phase current command value i z * based on the input torque command value T * , the angular velocity ⁇ , and the rotor position ⁇ . Is calculated. In the present embodiment, there is a feature in the calculation processing in the current command calculation unit 10, and detailed processing will be described later.
  • the dq-axis current control unit 20 determines the dq-axis current command values i d * and i q * input from the current command calculation unit 10 and the dq-axis current detection values i d and i q input from the dq conversion unit 40. Based on this, the dq-axis voltage command values v d * and v q * are output using proportional control, integral control, or the like.
  • the zero phase voltage command value v z * is output using control or the like.
  • the zero-phase voltage command value v z * output from the zero-phase current control unit 60 is added to the zero-phase counter-electromotive voltage compensation value v z ** output from the zero-phase counter-electromotive voltage compensation unit 80 to obtain a signal (v z * + v z **) is input to the switching signal generation unit 30.
  • the zero-phase counter electromotive force compensation value v z ** is for reducing the difference between the zero phase current command value i z * and the detected zero phase current i z, and cancels the zero phase component of the counter electromotive voltage.
  • the zero phase voltage command value v z * is compensated.
  • Detailed processing of the zero-phase back electromotive force compensation unit 80 will be described later.
  • the switching signal generator 30 is the sum of the dq-axis voltage command values v d * , v q * , the zero-phase voltage command value v z *, and the zero-phase counter electromotive voltage compensation value v z ** (v z * + V z ** ) is input. Based on these values, the switching signal generator 30 generates switching signals for turning on or off the switching elements 621a to 621d, 622a to 622d, and 623a to 623d.
  • the switching signal is input to the inverter 620, and the motor drive current flows through the three-phase windings 201 to 203 of the motor 200 by the on / off operation of the switching elements 621a to 621d, 622a to 622d, and 623a to 623d.
  • the motor driving current is controlled to a sine wave, and a rotating magnetic field necessary for driving is generated.
  • the effective value of the drive current cannot be increased after the maximum value of the sine wave reaches a predetermined current, and the output cannot be improved.
  • torque command value T * of magnitude torque command value
  • FIG. 4 is a flowchart illustrating an example of processing of the current command calculation unit 10.
  • step S1 dq-axis current command values i d * and i q * are calculated based on the torque command value T * , the angular velocity ⁇ , and the rotor position ⁇ .
  • a method for calculating the dq-axis current command values i d * and i q * there are a maximum torque current control, a field weakening control, and the like.
  • a table set in advance may be used for calculating the dq-axis current command values i d * and i q * .
  • UVW phase current command values i u * , i v * , i w * are calculated based on the dq axis current command values i d * , i q * and the detected rotor position ⁇ .
  • step S3 among the UVW phase current command values i u * , i v * and i w * calculated in step S2, the current command value having the largest absolute value is set as the maximum phase current command value i max * , and the amplitude Is the minimum phase current command value i min * , and the rest is the intermediate phase current command value i mid * .
  • step S4 it is determined whether or not the absolute value of the maximum phase current command value i max * is equal to or greater than a predetermined current value i rated .
  • the predetermined current value i rated means a maximum current value set to prevent failure of the inverter 620 and the motor 200.
  • the motor drive current is controlled to be a predetermined current value or less.
  • step S4 If it is determined in step S4 that
  • sgn (i max *) in formula (2) represents the sign of i max *, takes a negative or positive sign according to the positive or negative of sgn (i max *).
  • imax ** sgn ( imax * ) * irated (2)
  • step S6 the intermediate phase current command value i mid ** is recalculated by the following equation (3).
  • i mid ** i mid * ⁇ (i max * ⁇ i rated ) (3)
  • step S7 the minimum phase current command value i min ** is recalculated by the following equation (4).
  • i min ** i min * ⁇ (i max * ⁇ i rated ) (4)
  • FIG. 5 illustrates i max ** , i mid ** , and i min ** obtained by the processing from step S3 to step S7.
  • the sine wave curve indicated by the thin line is the U-phase current command value, the V-phase current command value, and the W-phase current command value calculated in step S2.
  • the magnitude relationship of the absolute value of the amplitude in the UVW phase current command value is
  • the maximum phase current command value i max ** , the intermediate phase current command value i mid **, and the minimum phase current command value i min ** are calculated in steps S5 to S7
  • the maximum phase current command value is determined in step S8.
  • i max ** intermediate phase current command value i mid **, and minimum phase current command value i min **
  • dq axis current command values i d * , i q *, and zero phase current command value i z * are calculated. .
  • the calculated dq-axis current command values i d * and i q * are output to the dq-axis current control unit 20, and the zero-phase current command value i z * is output to the zero-phase current control unit 60.
  • step S4 determines whether the absolute value of the maximum phase current command value imax * calculated in step S3 is smaller than the predetermined current value irated . If it is determined in step S4 that the absolute value of the maximum phase current command value imax * calculated in step S3 is smaller than the predetermined current value irated , the process proceeds to step S9 and the zero-phase current command value is set.
  • Set i z * to i z * 0.
  • FIG. 6 is a block diagram illustrating details of the zero-phase counter electromotive force compensation unit 80.
  • the difference between the zero-phase current command value i z * and the detected zero-phase current value i z increases due to the zero-phase component of the counter electromotive voltage generated when the motor is driven.
  • torque ripple may increase.
  • zero-phase was calculated by the zero-phase back electromotive force compensation section 80 back EMF compensation value v z * By * , the zero phase voltage command value v z * is compensated so as to cancel the zero phase component of the counter electromotive voltage.
  • the zero-phase counter electromotive voltage calculating section 81 based on the zero-phase current detection value i z and the rotor position theta, and a voltage drop due to the winding resistance R, the voltage drop due to the z-axis inductance L z, due to the magnetic flux [psi z
  • v Zz that is the sum of the zero-phase induced voltage and the following equation (11) is calculated.
  • motor parameters such as the z-axis inductance L z and the magnet magnetic flux ⁇ z vary depending on the rotor position ⁇ , the driving current and temperature of the motor 200, and so may be calculated using a preset table or approximate expression.
  • v Zz Ri z + L z (di z / dt) + d ⁇ z / dt ... (11)
  • the d-axis interference voltage calculation unit 82 calculates the d-axis interference voltage v dz generated due to the dz -axis interference inductance L dz based on the detected d-axis current value i d and the rotor position ⁇ .
  • the q-axis interference voltage calculation unit 83 calculates the q-axis interference voltage v qz generated due to the qz -axis interference inductance L qz based on the q-axis current detection value i q and the rotor position ⁇ .
  • the q-axis interference voltage v qz is calculated by the following equation (13).
  • v qz L qz (di q / dt) (13)
  • the zero-phase counter electromotive voltage v Zz calculated by the zero-phase counter electromotive voltage calculation unit 81 is added to the d-axis interference voltage v dz output from the d-axis interference voltage calculation unit 82.
  • the zero-phase voltage command value v z * output from the zero-phase current control unit 60 is added with the zero-phase counter electromotive force compensation value v z ** instead of the zero-phase voltage command value v z *.
  • the signal is input to the signal generator 30. That is, a larger zero-phase current iz is generated by the amount corresponding to the zero-phase counter-electromotive force compensation value v z ** so as to cancel the zero-phase component of the counter-electromotive voltage induced in the windings 201 to 203.
  • the zero phase voltage command value v z * is adjusted.
  • the motor control device 610 uses the torque command value T * , the angular velocity ⁇ and the rotation of the motor 200 as the drive current of the motor 200 having the windings 201 to 203 wound independently between phases. Control is performed based on the child position ⁇ . Then, the driving current (i u, i v, i w) ratio of zero-phase current i z is varied depending on the magnitude of the torque command value T * in. That is, the larger the torque command value T * .DELTA.i in FIG 5 is increased, thereby improving the ratio increases the output of the zero-phase current i z accordingly.
  • Modification 1 In the example described in FIGS. 3 and 4, whether to include a zero-phase current i z drive current, the absolute value of the maximum phase current command value i max * is in whether or not a predetermined current value i rated higher was determined, as in the example shown in the flowchart of FIG. 7, may be performed determines whether to include a zero-phase current i z on the drive current based on the magnitude of the torque command value T *.
  • step S101 as in step S1 of FIG. 4, the dq-axis current command values i d * and i q * are calculated based on the torque command value T * , the angular velocity ⁇ , and the rotor position ⁇ . calculate.
  • step S102 it is determined whether or not the torque command value T * is equal to or greater than a predetermined torque Tth.
  • the predetermined torque Tth means a maximum torque value set to prevent failure of the inverter 620 and the motor 200.
  • step S102 If it is determined in step S102 that T * ⁇ Tth, the process proceeds to step S103.
  • step S103 as in the case of step S2 described above, based on the dq-axis current command values i d * and i q * calculated in step S101 and the detected rotor position ⁇ , the UVW phase current command value i u. * , I v * , i w * are calculated.
  • a zero-phase current command value i z * is calculated based on the UVW phase current command values i u * , i v * , i w * and the rotor position ⁇ .
  • Zero phase current command value i z * is calculated by the following equation (14).
  • i z * A ⁇ sin (3 ⁇ + 3 ⁇ ) (14)
  • A means a current amplitude value necessary for reducing the maximum value of the UVW phase current command values i u * , i v * , i w * to be equal to or less than a predetermined current value i rated
  • ⁇ means a current phase obtained from the UVW phase current command values i u * , i v * , i w * and the rotor position ⁇ .
  • UVW phase current command value the frequency of the zero-phase current command value i z * i u *, i v *, was three times the i w *, may not be 3-fold.
  • FIG. 8 is a diagram showing a drive current (U-phase current i u ) when the zero-phase current command value i z * of the equation (14) is used.
  • a curve L1 shows a sine wave component contained in the U-phase current i u
  • the curve L2 represents the zero-phase current i z contained in U-phase current i u.
  • step S102 determines whether the torque command value T * is smaller than the predetermined torque Tth.
  • Predetermined torque Tth in the case of the process shown in FIG. 7 corresponds to a predetermined current value i rated in the case of FIG. 4, switches to state comprising zero-phase current i z at substantially the same timing.
  • the ratio of the zero-phase current i z (curve L2) to the sine wave component (curve L1) increases as the torque command value T * increases. The same operational effects as in the case of the embodiment can be achieved.
  • the zero-phase counter electromotive force compensation unit 80 applies windings 201 to 203 to the windings 201 to 203 based on the drive current ( id , iq , iz ) and the rotor position ⁇ .
  • a zero-phase voltage of the induced voltage (zero-phase counter electromotive force compensation value v z ** ) is calculated, and the zero-phase voltage command value i based on the torque command value T * , the angular velocity ⁇ of the motor 200, and the rotor position ⁇ . It is preferable to control the zero-phase current i z by adding the zero-phase counter electromotive force compensation value v z ** to z * .
  • Torque ripple may increase.
  • the zero-phase component of the counter electromotive voltage induced in the windings 201 to 203 is obtained by adding the zero-phase counter electromotive voltage compensation value v z ** to the zero phase voltage command value i z *.
  • the zero phase voltage command value v z * is adjusted so as to cancel out. As a result, torque ripple can be further reduced.
  • FIG. 9 is a diagram showing a second modification and is a block diagram showing details of the motor control device 610.
  • a UVW conversion unit 90 is added to the configuration of the motor control device 610 shown in FIG.
  • the UVW converter 90 outputs three-phase voltage command values i u * , i v * , i w * based on the dq-axis current command values i d * , i q * and the rotor position ⁇ .
  • the switching signal generation unit 30 the three-phase voltage command values i u *, i v *, i w * on the zero-phase voltage command value v z * and the zero-phase back electromotive force compensation value v z sum of ** (v z * + v z **) obtained by adding a is input. That, i u * + (v z * + v z **), i v * + (v z * + v z **) and i w * + 3 one signal (v z * + v z ** ) is input.
  • the zero-phase back electromotive voltage compensation section 80 dq-axis current detection value i d, i q and instead of the zero-phase current detection value i z, dq axis current command value i d *, i q * and the zero-phase current Zero-phase counter electromotive force compensation value v z ** is calculated using command value i z * .

Abstract

The present invention addresses the problem of providing a motor control device capable of improving output and reducing torque ripple. The motor control device 610 controls the drive current of a motor 200 having windings 201 through 203 independently wound for respective phases on the basis of a torque command value T* and the angular velocity ω and rotor position θ of the motor 200. In addition, the ratio of a zero-phase current in the drive current (iu, iv, iw) is changed in accordance with the magnitude of the torque command value T*. Consequently, output can be improved, and torque ripple can be suppressed due to reduction in the ratio of the zero-phase current in the drive current with decrease in the torque command value T*.

Description

モータ制御装置および電動車両Motor control device and electric vehicle
　本発明はモータ制御装置および電動車両に関する。 The present invention relates to a motor control device and an electric vehicle.
　ハイブリッド自動車や電気自動車は、駆動力向上の観点からモータ出力の向上が要求されている。例えば、特許文献１に記載の発明では、モータの駆動電流に擬似矩形波電流を用いることにより電流実効値を増加させ、出力を向上させるようにしている。 Hybrid motors and electric vehicles are required to improve motor output from the viewpoint of improving driving force. For example, in the invention described in Patent Document 1, an effective current value is increased by using a pseudo-rectangular wave current as a driving current of a motor, thereby improving an output.
特開２００６－１３６１４４号公報JP 2006-136144 A
　しかしながら、特許文献１に記載の方法では、モータの駆動電流を常に疑似矩形波としているため、トルクリプルの増大を招くおそれがある。 However, in the method described in Patent Document 1, the motor drive current is always a pseudo-rectangular wave, which may increase torque ripple.
　本発明の一態様によれば、モータ制御装置は、相間で独立して巻かれた巻線を有するモータの駆動電流を、トルク指令値とモータの角速度および回転子位置とに基づいて制御するモータ制御装置であって、前記駆動電流における零相電流の比率を前記トルク指令値の大きさに応じて変化させる。 According to one aspect of the present invention, a motor control device controls a drive current of a motor having windings wound independently between phases based on a torque command value, a motor angular velocity, and a rotor position. In the control device, the ratio of the zero-phase current in the drive current is changed according to the magnitude of the torque command value.
　本発明によれば、出力の向上およびトルクリプルの低減を図ることができる。 According to the present invention, it is possible to improve the output and reduce the torque ripple.
図１は、本発明の一実施形態のモータを搭載したハイブリッド型電気自動車の概略構成を示す図である。FIG. 1 is a diagram showing a schematic configuration of a hybrid electric vehicle equipped with a motor according to an embodiment of the present invention. 図２は、電力変換装置の構成を示すブロック図である。FIG. 2 is a block diagram illustrating a configuration of the power conversion apparatus. 図３は、モータ制御装置の詳細を示すブロック図である。FIG. 3 is a block diagram showing details of the motor control device. 図４は、電流指令演算処理の一例を示すフローチャートである。FIG. 4 is a flowchart illustrating an example of a current command calculation process. 図５は、駆動電流を説明する図である。FIG. 5 is a diagram illustrating the drive current. 図６は、零相逆起電圧補償部の詳細を説明するブロック図である。FIG. 6 is a block diagram illustrating details of the zero-phase counter electromotive force compensation unit. 図７は、変形例１を説明するフローチャートである。FIG. 7 is a flowchart for explaining the first modification. 図８は、変形例１における駆動電流、駆動電流の正弦波成分および零相電流を示す図である。FIG. 8 is a diagram illustrating a drive current, a sine wave component of the drive current, and a zero-phase current in the first modification. 図９は、変形例２のモータ制御装置を説明するブロック図である。FIG. 9 is a block diagram illustrating a motor control device according to a second modification.
　以下、図を参照して本発明を実施するための形態について説明する。なお、本発明は以下の実施形態に限定されることなく、本発明の技術的な概念の中で種々の変形例や応用例をもその範囲に含むものである。 Hereinafter, embodiments for carrying out the present invention will be described with reference to the drawings. The present invention is not limited to the following embodiments, and includes various modifications and application examples within the scope of the technical concept of the present invention.
　図１は、本発明の一実施形態のモータを搭載したハイブリッド型電気自動車の概略構成を示す図である。車両１００には、エンジン１２０とモータ２００とバッテリ１８０とが搭載されている。バッテリ１８０は、モータ２００による駆動力が必要な場合には電力変換装置６００を介してモータ２００に直流電力を供給し、回生走行時にはモータ２００から直流電力を受ける。バッテリ１８０とモータ２００との間の直流電力の授受は、電力変換装置６００を介して行われる。また、図示していないが、車両１００には低電圧電力（例えば、１４ボルト系電力）を供給するバッテリが搭載されており、例えば、制御系の電源として用いられる。 FIG. 1 is a diagram showing a schematic configuration of a hybrid electric vehicle equipped with a motor according to an embodiment of the present invention. The vehicle 100 is equipped with an engine 120, a motor 200, and a battery 180. The battery 180 supplies DC power to the motor 200 via the power converter 600 when the driving force by the motor 200 is necessary, and receives DC power from the motor 200 during regenerative travel. Transfer of direct-current power between the battery 180 and the motor 200 is performed via the power converter 600. Although not shown, the vehicle 100 is equipped with a battery for supplying low voltage power (for example, 14 volt system power), and is used as a power source for a control system, for example.
　エンジン１２０およびモータ２００による回転トルクは、変速機１３０とデファレンシャルギア１６０を介して前輪１１０に伝達される。変速機１３０は変速機制御装置１３４により制御される。エンジン１２０はエンジン制御装置１２４により制御される。バッテリ１８０は、バッテリ制御装置１８４により制御される。変速機制御装置１３４、エンジン制御装置１２４、バッテリ制御装置１８４、電力変換装置６００および統合制御装置１７０は、通信回線１７４によって接続されている。 Rotational torque generated by the engine 120 and the motor 200 is transmitted to the front wheels 110 via the transmission 130 and the differential gear 160. The transmission 130 is controlled by a transmission control device 134. The engine 120 is controlled by the engine control device 124. The battery 180 is controlled by the battery control device 184. Transmission control device 134, engine control device 124, battery control device 184, power conversion device 600 and integrated control device 170 are connected by communication line 174.
　統合制御装置１７０は、変速機制御装置１３４、エンジン制御装置１２４、電力変換装置６００およびバッテリ制御装置１８４よりも上位の制御装置である。統合制御装置１７０は、変速機制御装置１３４，エンジン制御装置１２４，電力変換装置６００およびバッテリ制御装置１８４の各状態を表す情報を、通信回線１７４を介してそれらからそれぞれ受け取る。統合制御装置１７０は、取得したそれらの情報に基づき各制御装置の制御指令を演算する。演算された制御指令は通信回線１７４を介してそれぞれの制御装置へ送信される。 The integrated control device 170 is a higher-level control device than the transmission control device 134, the engine control device 124, the power conversion device 600, and the battery control device 184. The integrated control device 170 receives information representing the states of the transmission control device 134, the engine control device 124, the power conversion device 600, and the battery control device 184 from each of them via the communication line 174. The integrated control device 170 calculates a control command for each control device based on the acquired information. The calculated control command is transmitted to each control device via the communication line 174.
　高電圧のバッテリ１８０はリチウムイオン電池あるいはニッケル水素電池などの２次電池で構成され、２５０ボルトから６００ボルト、あるいはそれ以上の高電圧の直流電力を出力する。バッテリ制御装置１８４は、バッテリ１８０の充放電状況やバッテリ１８０を構成する各単位セル電池の状態を、通信回線１７４を介して統合制御装置１７０に出力する。 The high voltage battery 180 is composed of a secondary battery such as a lithium ion battery or a nickel metal hydride battery, and outputs a high voltage DC power of 250 to 600 volts or more. The battery control device 184 outputs the charge / discharge status of the battery 180 and the state of each unit cell battery constituting the battery 180 to the integrated control device 170 via the communication line 174.
　統合制御装置１７０は、バッテリ制御装置１８４からの情報に基づいてバッテリ１８０の充電が必要と判断すると、電力変換装置６００に発電運転の指示を出す。また、統合制御装置１７０は、主に、エンジン１２０およびモータ２００の出力トルクの管理、エンジン１２０の出力トルクとモータ２００の出力トルクとの総合トルクやトルク分配比の演算処理を行い、その演算処理結果に基づく制御指令を、変速機制御装置１３４，エンジン制御装置１２４および電力変換装置６００へ送信する。電力変換装置６００は、統合制御装置１７０からのトルク指令に基づき、指令通りのトルク出力あるいは発電電力が発生するようにモータ２００を制御する。 If the integrated control device 170 determines that the battery 180 needs to be charged based on the information from the battery control device 184, the integrated control device 170 instructs the power conversion device 600 to perform a power generation operation. Further, the integrated control device 170 mainly performs management processing of the output torque of the engine 120 and the motor 200, calculation processing of the total torque and torque distribution ratio between the output torque of the engine 120 and the output torque of the motor 200, and the calculation processing. A control command based on the result is transmitted to transmission control device 134, engine control device 124, and power conversion device 600. Based on the torque command from the integrated control device 170, the power conversion device 600 controls the motor 200 so that a torque output or generated power according to the command is generated.
　電力変換装置６００には、後述するように、モータ２００を運転するためのインバータや、インバータへのスイッチング信号を生成するモータ制御装置を備えている。電力変換装置６００は、統合制御装置１７０からの指令に基づきインバータを制御することで、モータ２００を電動機としてあるいは発電機として動作させる。 As will be described later, the power conversion device 600 includes an inverter for operating the motor 200 and a motor control device that generates a switching signal to the inverter. The power conversion device 600 controls the inverter based on a command from the integrated control device 170 to operate the motor 200 as an electric motor or a generator.
　図２は、電力変換装置６００の構成を示すブロック図である。電力変換装置６００は、モータ制御装置６１０と、インバータ６２０と、電流センサ２２０とを備えている。モータ２００は中性点が接続されていない埋込磁石同期モータなどにより構成される。モータ２００には、回転子の位置を検出し、検出した回転子位置θを出力する位置センサ２１０が設けられている。電流センサ２２０は、モータ２００の固定子に巻かれたＵ相巻線２０１、Ｖ相巻線２０２およびＷ相巻線２０３に流れる電流を検出し、検出した三相電流ｉｕ、ｉｖ、ｉｗを出力する。 FIG. 2 is a block diagram showing the configuration of the power conversion apparatus 600. As shown in FIG. The power conversion device 600 includes a motor control device 610, an inverter 620, and a current sensor 220. The motor 200 is constituted by an embedded magnet synchronous motor or the like to which no neutral point is connected. The motor 200 is provided with a position sensor 210 that detects the position of the rotor and outputs the detected rotor position θ. Current sensor 220 detects currents flowing through U-phase winding 201, V-phase winding 202, and W-phase winding 203 wound around the stator of motor 200, and outputs the detected three-phase currents iu, iv, iw. To do.
　インバータ６２０には、Ｕ相フルブリッジインバータ６２１、Ｖ相フルブリッジインバータ６２２およびＷ相フルブリッジインバータ６２３が設けられており、それらは直流電源であるバッテリ１８０に並列接続されている。モータ２００のＵ相巻線２０１はＵ相フルブリッジインバータ６２１の出力端子に接続され、Ｖ相巻線２０２はＶ相フルブリッジインバータ６２２の出力端子に接続され、Ｗ相巻線２０３はＷ相フルブリッジインバータ６２３の出力端子に接続される。モータ２００は中性点が接続されておらず、Ｕ相巻線２０１、Ｖ相巻線２０２およびＷ相巻線２０３に流れる電流をそれぞれ独立に制御することができる。 The inverter 620 is provided with a U-phase full-bridge inverter 621, a V-phase full-bridge inverter 622, and a W-phase full-bridge inverter 623, which are connected in parallel to a battery 180 that is a DC power source. The U phase winding 201 of the motor 200 is connected to the output terminal of the U phase full bridge inverter 621, the V phase winding 202 is connected to the output terminal of the V phase full bridge inverter 622, and the W phase winding 203 is W phase full. Connected to the output terminal of the bridge inverter 623. The motor 200 is not connected to the neutral point, and can control the current flowing through the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 independently.
　Ｕ相フルブリッジインバータ６２１は、スイッチング素子６２１ａ～６２１ｄにより構成される。スイッチング素子６２１ａはＵ相左レグ上アームに配置される。スイッチング素子６２１ｂはＵ相左レグ下アームに配置される。スイッチング素子６２１ｃはＵ相右レグ上アームに配置される。スイッチング素子６２１ｄはＵ相右レグ下アームに配置される。 The U-phase full bridge inverter 621 includes switching elements 621a to 621d. Switching element 621a is disposed on the U-phase left leg upper arm. Switching element 621b is arranged on the U-phase left leg lower arm. Switching element 621c is arranged on the U-phase right leg upper arm. Switching element 621d is arranged on the U-phase right leg lower arm.
　Ｖ相フルブリッジインバータ６２２は、スイッチング素子６２２ａ～６２２ｄにより構成される。スイッチング素子６２２ａはＶ相左レグ上アームに配置される。スイッチング素子６２２ｂはＶ相左レグ下アームに配置される。スイッチング素子６２２ｃはＶ相右レグ上アームに配置される。スイッチング素子６２２ｄはＶ相右レグ下アームに配置される。 The V-phase full bridge inverter 622 includes switching elements 622a to 622d. Switching element 622a is disposed on the V-phase left leg upper arm. Switching element 622b is disposed on the V-phase left leg lower arm. Switching element 622c is arranged on the V-phase right leg upper arm. Switching element 622d is arranged on the lower arm of the V-phase right leg.
　Ｗ相フルブリッジインバータ６２３は、スイッチング素子６２３ａ～６２３ｄにより構成される。スイッチング素子６２３ａはＷ相左レグ上アームに配置される。スイッチング素子６２３ｂはＷ相左レグ下アームに配置される。スイッチング素子６２３ｃはＷ相右レグ上アームに配置される。スイッチング素子６２３ｄはＷ相右レグ下アームに配置される。 The W-phase full bridge inverter 623 includes switching elements 623a to 623d. Switching element 623a is disposed on the W-phase left leg upper arm. Switching element 623b is arranged on the lower arm of the W-phase left leg. Switching element 623c is disposed on the W-phase right leg upper arm. Switching element 623d is disposed on the W-phase right leg lower arm.
　スイッチング素子６２１ａ～６２１ｄと、スイッチング素子６２２ａ～６２２ｄと、スイッチング素子６２３ａ～６２３ｄは、金属酸化膜型電界効果トランジスタ（ＭＯＳＦＥＴ）や絶縁ゲートバイポーラトランジスタ（ＩＧＢＴ）などと、ダイオードを組み合わせて構成される。本実施形態では、ＭＯＳＦＥＴとダイオードを用いる構成で説明する。 The switching elements 621a to 621d, the switching elements 622a to 622d, and the switching elements 623a to 623d are configured by combining a metal oxide film field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), and the like with a diode. In the present embodiment, a configuration using MOSFETs and diodes will be described.
　スイッチング素子６２１ａ～６２１ｄと、スイッチング素子６２２ａ～６２２ｄと、スイッチング素子６２３ａ～６２３ｄとをモータ制御装置６１０で生成されたスイッチング信号に基づいてオンもしくはオフすることで、インバータ６２０は、バッテリ１８０から印加された直流電圧を交流電圧に変換する。変換された交流電圧は、モータ２００の固定子に巻かれた３相巻線２０１～２０３に印加され、３相交流電流を発生させる。この３相交流電流がモータ２００に回転磁界を発生させ、モータ２００の回転子が回転する。モータ制御装置６１０は、統合制御装置１７０からのトルク指令値Ｔ＊、電流センサ２２０で検出された三相電流ｉｕ、ｉｖ、ｉｗ、位置センサ２１０で検出された回転子位置θに基づいてインバータ６２０をＰＷＭ制御する。 The inverter 620 is applied from the battery 180 by turning on or off the switching elements 621a to 621d, the switching elements 622a to 622d, and the switching elements 623a to 623d based on the switching signal generated by the motor control device 610. DC voltage is converted to AC voltage. The converted AC voltage is applied to the three-phase windings 201 to 203 wound around the stator of the motor 200 to generate a three-phase AC current. The three-phase alternating current generates a rotating magnetic field in the motor 200, and the rotor of the motor 200 rotates. The motor control device 610 is based on the torque command value T * from the integrated control device 170, the three-phase currents i u , i v , i w detected by the current sensor 220, and the rotor position θ detected by the position sensor 210. The inverter 620 is PWM controlled.
　図３は、モータ制御装置６１０の詳細を示すブロック図である。モータ制御装置６１０は、電流指令演算部１０、ｄｑ軸電流制御部２０、スイッチング信号生成部３０、ｄｑ変換部４０、零相電流算出部５０、零相電流制御部６０、速度変換部７０および零相逆起電圧補償部８０を備えている。 FIG. 3 is a block diagram showing details of the motor control device 610. The motor control device 610 includes a current command calculation unit 10, a dq axis current control unit 20, a switching signal generation unit 30, a dq conversion unit 40, a zero phase current calculation unit 50, a zero phase current control unit 60, a speed conversion unit 70, and a zero. A phase back electromotive force compensation unit 80 is provided.
　ｄｑ変換部４０は、電流センサ２２０で検出された三相電流ｉｕ、ｉｖ、ｉｗと、位置センサ２１０で検出された回転子位置θとに基づいて、ｄｑ軸電流検出値ｉｄ、ｉｑを出力する。速度変換部７０は、位置センサ２１０で検出された回転子位置θに基づいて、回転子の角速度ωを出力する。零相電流算出部５０は、入力された三相電流ｉｕ、ｉｖ、ｉｗに基づいて零相電流ｉｚを算出する。零相電流ｉｚは次式（１）のように算出される。
　　ｉｚ＝ｉｕ／√３＋ｉｖ／√３＋ｉｗ／√３　　　…（１） Based on the three-phase currents i u , i v , i w detected by the current sensor 220 and the rotor position θ detected by the position sensor 210, the dq converter 40 detects the dq-axis current detection value i d , i q is output. The speed conversion unit 70 outputs the angular velocity ω of the rotor based on the rotor position θ detected by the position sensor 210. Zero-phase current calculation section 50, the three-phase currents i u entered, i v, calculates a zero-phase current i z based on i w. The zero-phase current iz is calculated as in the following formula (1).
i z = i u / √3 + i v / √3 + i w / √3 (1)
　電流指令演算部１０は、入力されたトルク指令値Ｔ＊と、角速度ωと、回転子位置θとに基づき、ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊および零相電流指令値ｉｚ ＊を算出する。本実施の形態では、電流指令演算部１０における演算処理に特徴があり、詳細処理については後述する。 The current command calculation unit 10 determines the dq-axis current command values i d * and i q * and the zero-phase current command value i z * based on the input torque command value T * , the angular velocity ω, and the rotor position θ . Is calculated. In the present embodiment, there is a feature in the calculation processing in the current command calculation unit 10, and detailed processing will be described later.
　ｄｑ軸電流制御部２０は、電流指令演算部１０から入力されたｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊と、ｄｑ変換部４０から入力されたｄｑ軸電流検出値ｉｄ、ｉｑに基づき、比例制御や積分制御などを用いてｄｑ軸電圧指令値ｖｄ ＊、ｖｑ ＊を出力する。零相電流制御部６０は、電流指令演算部１０から入力された零相電流指令値ｉｚ ＊と、零相電流算出部５０で算出された零相電流ｉｚとに基づき、比例制御や積分制御などを用いて零相電圧指令値ｖｚ ＊を出力する。 The dq-axis current control unit 20 determines the dq-axis current command values i d * and i q * input from the current command calculation unit 10 and the dq-axis current detection values i d and i q input from the dq conversion unit 40. Based on this, the dq-axis voltage command values v d * and v q * are output using proportional control, integral control, or the like. Zero-phase current controller 60, the zero-phase current command value i z * input from the current command calculating section 10, based on the zero-phase current i z calculated in the zero-phase current calculator 50, the proportional control and integral The zero phase voltage command value v z * is output using control or the like.
　零相電流制御部６０から出力された零相電圧指令値ｖｚ ＊は、零相逆起電圧補償部８０から出力された零相逆起電圧補償値ｖｚ ＊＊と足し合わされ、信号（ｖｚ ＊＋ｖｚ ＊＊）がスイッチング信号生成部３０に入力される。零相逆起電圧補償値ｖｚ ＊＊は零相電流指令値ｉｚ ＊と検出された零相電流ｉｚとの乖離を低減するためのものであり、逆起電圧の零相成分を打ち消すように零相電圧指令値ｖｚ ＊を補償する。零相逆起電圧補償部８０の詳細処理については後述する。 The zero-phase voltage command value v z * output from the zero-phase current control unit 60 is added to the zero-phase counter-electromotive voltage compensation value v z ** output from the zero-phase counter-electromotive voltage compensation unit 80 to obtain a signal (v z * + v z **) is input to the switching signal generation unit 30. The zero-phase counter electromotive force compensation value v z ** is for reducing the difference between the zero phase current command value i z * and the detected zero phase current i z, and cancels the zero phase component of the counter electromotive voltage. Thus, the zero phase voltage command value v z * is compensated. Detailed processing of the zero-phase back electromotive force compensation unit 80 will be described later.
　スイッチング信号生成部３０には、ｄｑ軸電圧指令値ｖｄ ＊、ｖｑ ＊と、零相電圧指令値ｖｚ ＊と零相逆起電圧補償値ｖｚ ＊＊との和である（ｖｚ ＊＋ｖｚ ＊＊）とが入力される。スイッチング信号生成部３０は、これらの値に基づいてスイッチング素子６２１ａ～６２１ｄ、６２２ａ～６２２ｄおよび６２３ａ～６２３ｄを、オンもしくはオフするスイッチング信号を生成する。スイッチング信号はインバータ６２０に入力され、スイッチング素子６２１ａ～６２１ｄ、６２２ａ～６２２ｄおよび６２３ａ～６２３ｄのオンオフ動作によって、モータ駆動電流がモータ２００の３相巻線２０１～２０３に流れる。 The switching signal generator 30 is the sum of the dq-axis voltage command values v d * , v q * , the zero-phase voltage command value v z *, and the zero-phase counter electromotive voltage compensation value v z ** (v z * + V z ** ) is input. Based on these values, the switching signal generator 30 generates switching signals for turning on or off the switching elements 621a to 621d, 622a to 622d, and 623a to 623d. The switching signal is input to the inverter 620, and the motor drive current flows through the three-phase windings 201 to 203 of the motor 200 by the on / off operation of the switching elements 621a to 621d, 622a to 622d, and 623a to 623d.
　一般的に、モータ駆動電流は正弦波に制御され、駆動に必要な回転磁界を生成している。しかし、駆動電流を正弦波に制御している場合は、正弦波の最大値が所定の電流に達した後は駆動電流の実効値を増加させることができず、出力を向上させることができない。本発明では、以下に説明するようにモータの運転条件（トルク指令値Ｔ＊の大きさ）に応じて零相電流ｉｚを流すことで出力向上を図ると共に、トルクリプルの増加を抑制するようにした。 Generally, the motor driving current is controlled to a sine wave, and a rotating magnetic field necessary for driving is generated. However, when the drive current is controlled to a sine wave, the effective value of the drive current cannot be increased after the maximum value of the sine wave reaches a predetermined current, and the output cannot be improved. In the present invention, together with improve the output by passing a zero-phase current i z according to the motor operating conditions (torque command value T * of magnitude) as described below, so as to suppress an increase in torque ripple did.
（電流指令演算部１０の説明）
　図４は、電流指令演算部１０の処理の一例を示すフローチャートである。ステップＳ１では、トルク指令値Ｔ＊、角速度ωおよび回転子位置θに基づき、ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊を算出する。ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊の計算方法としては、最大トルク電流制御や弱め界磁制御などがあるが、周知のため説明を省略する。なお、ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊の計算には、予め設定したテーブルを使用してもよい。 (Description of the current command calculation unit 10)
FIG. 4 is a flowchart illustrating an example of processing of the current command calculation unit 10. In step S1, dq-axis current command values i d * and i q * are calculated based on the torque command value T * , the angular velocity ω, and the rotor position θ. As a method for calculating the dq-axis current command values i d * and i q * , there are a maximum torque current control, a field weakening control, and the like. A table set in advance may be used for calculating the dq-axis current command values i d * and i q * .
　ステップＳ２では、ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊と検出された回転子位置θとに基づき、ＵＶＷ相電流指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊を算出する。 In step S2, UVW phase current command values i u * , i v * , i w * are calculated based on the dq axis current command values i d * , i q * and the detected rotor position θ.
　ステップＳ３では、ステップＳ２で算出したＵＶＷ相電流指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊の内、振幅の絶対値が最も大きい電流指令値を最大相電流指令値ｉｍａｘ ＊とし、振幅の絶対値が最も小さい電流指令値を最小相電流指令値ｉｍｉｎ ＊とし、残りを中間相電流指令値ｉｍｉｄ ＊とする。 In step S3, among the UVW phase current command values i u * , i v * and i w * calculated in step S2, the current command value having the largest absolute value is set as the maximum phase current command value i max * , and the amplitude Is the minimum phase current command value i min * , and the rest is the intermediate phase current command value i mid * .
　ステップＳ４では、最大相電流指令値ｉｍａｘ ＊の絶対値が所定電流値ｉｒａｔｅｄ以上か否かを判定する。ここで、所定電流値ｉｒａｔｅｄとは、インバータ６２０およびモータ２００の故障を防止するために設定された最大電流値を意味する。本実施の形態では、モータ駆動電流は所定電流値以下に制御される。 In step S4, it is determined whether or not the absolute value of the maximum phase current command value i max * is equal to or greater than a predetermined current value i rated . Here, the predetermined current value i rated means a maximum current value set to prevent failure of the inverter 620 and the motor 200. In the present embodiment, the motor drive current is controlled to be a predetermined current value or less.
　ステップＳ４で|ｉｍａｘ ＊|≧ｉｒａｔｅｄと判定された場合には、ステップＳ５に進んで、次式（２）式により最大相電流指令値ｉｍａｘ ＊＊を再計算する。なお、式（２）においてsgn(ｉｍａｘ ＊)はｉｍａｘ ＊の正負を表しており、sgn(ｉｍａｘ ＊)の正負に応じてマイナス又はプラスの符号を取る。
　　ｉｍａｘ ＊＊＝sgn(ｉｍａｘ ＊)×ｉｒａｔｅｄ　　　…（２） If it is determined in step S4 that | i max * | ≧ i rated , the process proceeds to step S5, and the maximum phase current command value i max ** is recalculated by the following equation (2). Incidentally, sgn (i max *) in formula (2) represents the sign of i max *, takes a negative or positive sign according to the positive or negative of sgn (i max *).
imax ** = sgn ( imax * ) * irated (2)
　ステップＳ６では、中間相電流指令値ｉｍｉｄ ＊＊を次式（３）により再計算する。
　　ｉｍｉｄ ＊＊＝ｉｍｉｄ ＊－（ｉｍａｘ ＊－ｉｒａｔｅｄ）　　　…（３） In step S6, the intermediate phase current command value i mid ** is recalculated by the following equation (3).
i mid ** = i mid * − (i max * −i rated ) (3)
　ステップＳ７では、最小相電流指令値ｉｍｉｎ ＊＊を次式（４）により再計算する。
　　ｉｍｉｎ ＊＊＝ｉｍｉｎ ＊－（ｉｍａｘ ＊－ｉｒａｔｅｄ）　　　…（４） In step S7, the minimum phase current command value i min ** is recalculated by the following equation (4).
i min ** = i min * − (i max * −i rated ) (4)
　図５は、ステップＳ３からステップＳ７までの処理により得られるｉｍａｘ ＊＊、ｉｍｉｄ ＊＊、ｉｍｉｎ ＊＊を図示したものである。細線で示す正弦波曲線はステップＳ２で算出されるＵ相電流指令値、Ｖ相電流指令値およびＷ相電流指令値である。回転子位置θ１では、ＵＶＷ相電流指令値における振幅の絶対値の大小関係は|ｉｕ ＊|＞|ｉｗ ＊|＞|ｉｖ ＊|となっているので、ｉｕ ＊が最大相電流指令値ｉｍａｘ ＊となり、ｉｗ ＊が中間相電流指令値ｉｍｉｄ ＊となり、ｉｖ ＊が最小相電流指令値ｉｍｉｎ ＊となる。このとき、ｉｕ ＊＊，ｉｖ ＊＊，ｉｗ ＊＊は次式（５）～（７）のように表される。
　　ｉｕ ＊＊＝ｉｍａｘ ＊＊＝ｉｒａｔｅｄ　　　…（５）
　　ｉｖ ＊＊＝ｉｍｉｎ ＊＊＝ｉｖ ＊－（ｉｕ ＊－ｉｒａｔｅｄ）　　　…（６）
　　ｉｗ ＊＊＝ｉｍｉｄ ＊＊＝ｉｗ ＊－（ｉｕ ＊－ｉｒａｔｅｄ）　　　…（７） FIG. 5 illustrates i max ** , i mid ** , and i min ** obtained by the processing from step S3 to step S7. The sine wave curve indicated by the thin line is the U-phase current command value, the V-phase current command value, and the W-phase current command value calculated in step S2. At the rotor position θ1, the magnitude relationship of the absolute value of the amplitude in the UVW phase current command value is | i u * |> | i w * |> | i v * |. Therefore, i u * is the maximum phase current. The command value becomes i max * , i w * becomes the intermediate phase current command value i mid * , and i v * becomes the minimum phase current command value i min * . At this time, i u ** , i v ** , and i w ** are expressed by the following equations (5) to (7).
i u ** = i max ** = i rated (5)
i v ** = i min ** = i v * - (i u * -i rated) ... (6)
i w ** = i mid ** = i w * − (i u * −i rated ) (7)
　一方、回転子位置θ２のタイミングでは、ＵＶＷ相電流指令値における振幅の絶対値の大小関係は|ｉｗ ＊|＞|ｉｕ ＊|＞|ｉｖ ＊|となっているので、ｉｗ ＊が最大相電流指令値ｉｍａｘ ＊となり、ｉｕ ＊が中間相電流指令値ｉｍｉｄ ＊となり、ｉｖ ＊が最小相電流指令値ｉｍｉｎ ＊となる。この場合には、ｉｕ ＊＊，ｉｖ ＊＊，ｉｗ ＊＊は次式（８）～（１０）のように表される。
　　ｉｕ ＊＊＝ｉｍｉｄ ＊＊＝ｉｕ ＊－（ｉｗ ＊－ｉｒａｔｅｄ）
　　　　　　　　　　＝ｉｕ ＊＋（ｉｒａｔｅｄ－ｉｗ ＊）　　　…（８）
　　ｉｖ ＊＊＝ｉｍｉｎ ＊＊＝ｉｖ ＊－（ｉｗ ＊－ｉｒａｔｅｄ）
　　　　　　　　　　＝ｉｖ ＊＋（ｉｒａｔｅｄ－ｉｗ ＊）　　　…（９）
　　ｉｗ ＊＊＝ｉｍａｘ ＊＊＝－ｉｒａｔｅｄ　　　…（１０） On the other hand, the timing of the rotor position .theta.2, the magnitude relationship of the absolute value of the amplitude in the UVW phase current command value | i w * |> | i u * |> | i v * | since the turned, i w * Becomes the maximum phase current command value i max * , i u * becomes the intermediate phase current command value i mid * , and i v * becomes the minimum phase current command value i min * . In this case, i u ** , i v ** , and i w ** are expressed by the following equations (8) to (10).
i u ** = i mid ** = i u * -(i w * -i rated )
= I u * + (i rated −i w * ) (8)
i v ** = i min ** = i v * - (i w * -i rated)
= I v * + (i rated −i w * ) (9)
i w ** = i max ** = −i rated (10)
　ステップＳ５～ステップＳ７で最大相電流指令値ｉｍａｘ ＊＊、中間相電流指令値ｉｍｉｄ ＊＊および最小相電流指令値ｉｍｉｎ ＊＊を算出したならば、ステップＳ８において、最大相電流指令値ｉｍａｘ ＊＊、中間相電流指令値ｉｍｉｄ ＊＊および最小相電流指令値ｉｍｉｎ ＊＊に基づきｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊および零相電流指令値ｉｚ ＊を算出する。そして、算出されたｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊をｄｑ軸電流制御部２０へ出力し、零相電流指令値ｉｚ ＊を零相電流制御部６０へ出力する。 If the maximum phase current command value i max ** , the intermediate phase current command value i mid **, and the minimum phase current command value i min ** are calculated in steps S5 to S7, the maximum phase current command value is determined in step S8. Based on i max ** , intermediate phase current command value i mid **, and minimum phase current command value i min ** , dq axis current command values i d * , i q *, and zero phase current command value i z * are calculated. . Then, the calculated dq-axis current command values i d * and i q * are output to the dq-axis current control unit 20, and the zero-phase current command value i z * is output to the zero-phase current control unit 60.
　一方、ステップＳ４において、ステップＳ３で算出された最大相電流指令値ｉｍａｘ ＊の絶対値が所定電流値ｉｒａｔｅｄよりも小さいと判定された場合には、ステップＳ９へ進んで零相電流指令値ｉｚ ＊をｉｚ ＊＝０に設定する。この場合、ステップＳ１で算出されたｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊がｄｑ軸電流制御部２０へ出力され、ステップＳ９で算出された零相電流指令値ｉｚ ＊＝０が零相電流制御部６０へ出力される。 On the other hand, if it is determined in step S4 that the absolute value of the maximum phase current command value imax * calculated in step S3 is smaller than the predetermined current value irated , the process proceeds to step S9 and the zero-phase current command value is set. Set i z * to i z * = 0. In this case, the dq-axis current command values i d * and i q * calculated in step S1 are output to the dq-axis current control unit 20, and the zero-phase current command value i z * = 0 calculated in step S9 is zero. It is output to the phase current control unit 60.
（零相逆起電圧補償部８０の説明）
　図６は、零相逆起電圧補償部８０の詳細を説明するブロック図である。図２に示す構成において零相電流を制御する際、モータ駆動時に発生する逆起電圧の零相成分により零相電流指令値ｉｚ ＊と検出された零相電流値ｉｚとの差が増大し、トルクリプルが増大するおそれがある。本実施の形態では、零相電流指令値ｉｚ ＊と零相電流ｉｚとの乖離を低減するために、零相逆起電圧補償部８０で算出した零相逆起電圧補償値ｖｚ ＊＊により、逆起電圧の零相成分を打ち消すように零相電圧指令値ｖｚ ＊を補償する。 (Description of Zero Phase Back Electromotive Voltage Compensation Unit 80)
FIG. 6 is a block diagram illustrating details of the zero-phase counter electromotive force compensation unit 80. When the zero-phase current is controlled in the configuration shown in FIG. 2, the difference between the zero-phase current command value i z * and the detected zero-phase current value i z increases due to the zero-phase component of the counter electromotive voltage generated when the motor is driven. However, torque ripple may increase. In this embodiment, in order to reduce the deviation between the zero-phase current command value i z * and the zero-phase current i z, zero-phase was calculated by the zero-phase back electromotive force compensation section 80 back EMF compensation value v z * By * , the zero phase voltage command value v z * is compensated so as to cancel the zero phase component of the counter electromotive voltage.
　零相逆起電圧算出部８１では、零相電流検出値ｉｚと回転子位置θとに基づき、巻線抵抗Ｒによる電圧降下と、ｚ軸インダクタンスＬｚによる電圧降下と、磁石磁束ψｚによる零相誘起電圧との和である零相逆起電圧ｖＺｚを、次式（１１）のように算出する。なお、ｚ軸インダクタンスＬｚや磁石磁束ψｚなどのモータパラメータは、回転子位置θやモータ２００の駆動電流および温度等によって変化するため、予め設定したテーブルや近似式を用いて算出してもよい。
　　ｖＺｚ＝Ｒｉｚ＋Ｌｚ（ｄｉｚ／ｄｔ)＋ｄψｚ／ｄｔ　　　…（１１） In the zero-phase counter electromotive voltage calculating section 81, based on the zero-phase current detection value i z and the rotor position theta, and a voltage drop due to the winding resistance R, the voltage drop due to the z-axis inductance L z, due to the magnetic flux [psi z A zero-phase counter electromotive voltage v Zz that is the sum of the zero-phase induced voltage and the following equation (11) is calculated. Note that motor parameters such as the z-axis inductance L z and the magnet magnetic flux ψ z vary depending on the rotor position θ, the driving current and temperature of the motor 200, and so may be calculated using a preset table or approximate expression. Good.
v Zz = Ri z + L z (di z / dt) + dψ z / dt ... (11)
　ｄ軸干渉電圧算出部８２では、ｄ軸電流検出値ｉｄと回転子位置θに基づき、ｄ－ｚ軸間干渉インダクタンスＬｄｚに起因して発生するｄ軸干渉電圧ｖｄｚを算出する。ｄ軸干渉電圧ｖｄｚは、次式（１２）により算出される。
　　ｖｄｚ＝Ｌｄｚ（ｄｉｄ／ｄｔ)　　　…（１２） The d-axis interference voltage calculation unit 82 calculates the d-axis interference voltage v dz generated due to the dz -axis interference inductance L dz based on the detected d-axis current value i d and the rotor position θ. The d-axis interference voltage v dz is calculated by the following equation (12).
v dz = L dz (di d / dt) (12)
　ｑ軸干渉電圧算出部８３では、ｑ軸電流検出値ｉｑと回転子位置θとに基づき、ｑ－ｚ軸間干渉インダクタンスＬｑｚに起因して発生するｑ軸干渉電圧ｖｑｚを算出する。ｑ軸干渉電圧ｖｑｚは、次式（１３）により算出される。
　　ｖｑｚ＝Ｌｑｚ（ｄｉｑ／ｄｔ)　　　…（１３） The q-axis interference voltage calculation unit 83 calculates the q-axis interference voltage v qz generated due to the qz -axis interference inductance L qz based on the q-axis current detection value i q and the rotor position θ. The q-axis interference voltage v qz is calculated by the following equation (13).
v qz = L qz (di q / dt) (13)
　なお、式（１１）～（１３）には表されていない非線形要素を予め設定したテーブルを用いて考慮することにより、零相電流指令値ｉｚ ＊と零相電流検出値ｉｚとの乖離をさらに低減することが可能である。 Note that the difference between the zero-phase current command value i z * and the zero-phase current detection value iz is taken into consideration by using a table in which nonlinear elements not represented in the equations (11) to (13) are set in advance. Can be further reduced.
　零相逆起電圧補償部８０からは、零相逆起電圧算出部８１で算出された零相逆起電圧ｖＺｚに、ｄ軸干渉電圧算出部８２から出力されたｄ軸干渉電圧ｖｄｚとｑ軸干渉電圧算出部８３から出力されたｑ軸干渉電圧ｖｑｚとを加算したものが、零相逆起電圧補償値ｖｚ ＊＊（＝ｖＺｚ＋ｖｄｚ＋ｖｑｚ）として出力される。そして、零相電流制御部６０から出力された零相電圧指令値ｖｚ ＊に零相逆起電圧補償値ｖｚ ＊＊を加算したものが、零相電圧指令値ｖｚ ＊の代わりにスイッチング信号生成部３０に入力される。すなわち、巻線２０１～２０３に誘起された逆起電圧の零相成分を打ち消すように、零相逆起電圧補償値ｖｚ ＊＊の分だけより大きな零相電流ｉｚが生成されるように零相電圧指令値ｖｚ ＊が調整される。 From the zero-phase counter electromotive voltage compensation unit 80, the zero-phase counter electromotive voltage v Zz calculated by the zero-phase counter electromotive voltage calculation unit 81 is added to the d-axis interference voltage v dz output from the d-axis interference voltage calculation unit 82. A sum of the q-axis interference voltage v qz output from the q-axis interference voltage calculation unit 83 is output as a zero-phase counter electromotive voltage compensation value v z ** (= v Zz + v dz + v qz ). The zero-phase voltage command value v z * output from the zero-phase current control unit 60 is added with the zero-phase counter electromotive force compensation value v z ** instead of the zero-phase voltage command value v z *. The signal is input to the signal generator 30. That is, a larger zero-phase current iz is generated by the amount corresponding to the zero-phase counter-electromotive force compensation value v z ** so as to cancel the zero-phase component of the counter-electromotive voltage induced in the windings 201 to 203. The zero phase voltage command value v z * is adjusted.
（Ｃ１）以上説明したように、モータ制御装置６１０は、相間で独立して巻かれた巻線２０１～２０３を有するモータ２００の駆動電流を、トルク指令値Ｔ＊とモータ２００の角速度ωおよび回転子位置θとに基づいて制御する。そして、駆動電流（ｉｕ、ｉｖ、ｉｗ）における零相電流ｉｚの比率をトルク指令値Ｔ＊の大きさに応じて変化させる。すなわち、トルク指令値Ｔ＊が大きいほど図５のΔｉは大きくなり、それに応じて零相電流ｉｚの比率が大きくなって出力の向上が図れる。逆に、トルク指令値Ｔ＊が小さい場合には、Δｉが小さくなるので零相電流ｉｚの比率も小さくなり、上述した特許文献１の発明のように疑似矩形波の駆動電流で制御する場合に比べてトルクリプルの低減を図ることができる。 (C1) As described above, the motor control device 610 uses the torque command value T * , the angular velocity ω and the rotation of the motor 200 as the drive current of the motor 200 having the windings 201 to 203 wound independently between phases. Control is performed based on the child position θ. Then, the driving current (i u, i v, i w) ratio of zero-phase current i z is varied depending on the magnitude of the torque command value T * in. That is, the larger the torque command value T * .DELTA.i in FIG 5 is increased, thereby improving the ratio increases the output of the zero-phase current i z accordingly. Conversely, if the torque command value T * small, .DELTA.i is also reduced ratio of zero-phase current i z becomes smaller, when controlling the driving current of the pseudo rectangular wave as in the invention of Patent Document 1 described above Torque ripple can be reduced as compared with the above.
（Ｃ２）このように駆動電流における零相電流の比率を変化させる際には、図３に示すように、零相電圧指令値ｖｚ ＊を調整して各相電圧の総和である零相電圧を制御することにより比率を変化させる、 (C2) When changing the ratio of the zero-phase current in the drive current in this way, as shown in FIG. 3, the zero-phase voltage command value v z * is adjusted and the zero-phase voltage that is the sum of the phase voltages is adjusted. Change the ratio by controlling
（Ｃ３）また、駆動電流が所定電流値（所定電流値ｉｒａｔｅｄ）以下となるように零相電流の比率を変化させることで、過剰電流によるモータ２００やインバータ６２０の故障を防止することができる。 (C3) In addition, by changing the ratio of the zero-phase current so that the drive current becomes equal to or less than a predetermined current value (predetermined current value i rated ), it is possible to prevent the motor 200 and the inverter 620 from being damaged due to excessive current. .
（変形例１）
　なお、図３，４で説明した例では、駆動電流に零相電流ｉｚを含ませるか否かを、最大相電流指令値ｉｍａｘ ＊の絶対値が所定電流値ｉｒａｔｅｄ以上か否かで判定したが、図７のフローチャートに示す例のように、トルク指令値Ｔ＊の大きさに基づいて駆動電流に零相電流ｉｚを含ませるか否かの判定を行っても良い。 (Modification 1)
In the example described in FIGS. 3 and 4, whether to include a zero-phase current i z drive current, the absolute value of the maximum phase current command value i max * is in whether or not a predetermined current value i rated higher was determined, as in the example shown in the flowchart of FIG. 7, may be performed determines whether to include a zero-phase current i z on the drive current based on the magnitude of the torque command value T *.
　図７に示すフローチャートでは、まず、ステップＳ１０１において図４のステップＳ１と同様に、トルク指令値Ｔ＊、角速度ωおよび回転子位置θに基づき、ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊を算出する。 In the flowchart shown in FIG. 7, first, in step S101, as in step S1 of FIG. 4, the dq-axis current command values i d * and i q * are calculated based on the torque command value T * , the angular velocity ω, and the rotor position θ. calculate.
　ステップＳ１０２では、トルク指令値Ｔ＊が所定トルクＴth以上か否かを判定する。ここで所定トルクＴthとは、インバータ６２０およびモータ２００の故障を防止するために設定された最大トルク値を意味する。 In step S102, it is determined whether or not the torque command value T * is equal to or greater than a predetermined torque Tth. Here, the predetermined torque Tth means a maximum torque value set to prevent failure of the inverter 620 and the motor 200.
　ステップＳ１０２においてＴ＊≧Ｔthと判定されると、ステップＳ１０３に進む。ステップＳ１０３では、上述したステップＳ２の場合と同様に、ステップＳ１０１で算出したｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊と検出された回転子位置θとに基づき、ＵＶＷ相電流指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊を算出する。 If it is determined in step S102 that T * ≧ Tth, the process proceeds to step S103. In step S103, as in the case of step S2 described above, based on the dq-axis current command values i d * and i q * calculated in step S101 and the detected rotor position θ, the UVW phase current command value i u. * , I v * , i w * are calculated.
　ステップＳ１０４では、Ｂ４では、ＵＶＷ相電流指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊と回転子位置θとに基づき、零相電流指令値ｉｚ ＊を算出する。零相電流指令値ｉｚ ＊は次式（１４）で算出される。
　　ｉｚ ＊＝Ａ・sin(３θ＋３α)　　　…（１４） In step S104, in B4, a zero-phase current command value i z * is calculated based on the UVW phase current command values i u * , i v * , i w * and the rotor position θ. Zero phase current command value i z * is calculated by the following equation (14).
i z * = A · sin (3θ + 3α) (14)
　式（１４）において、ＡはＵＶＷ相電流指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊の最大値を所定電流値ｉｒａｔｅｄ以下に低減するために必要な電流振幅値を意味しており、αはＵＶＷ相電流指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊と回転子位置θから求められる電流位相を意味する。なお、式（１４）では、零相電流指令値ｉｚ ＊の周波数をＵＶＷ相電流指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊の３倍としたが、３倍でなくても良い。 In the formula (14), A means a current amplitude value necessary for reducing the maximum value of the UVW phase current command values i u * , i v * , i w * to be equal to or less than a predetermined current value i rated , α means a current phase obtained from the UVW phase current command values i u * , i v * , i w * and the rotor position θ. In formula (14), UVW phase current command value the frequency of the zero-phase current command value i z * i u *, i v *, was three times the i w *, may not be 3-fold.
　図８は、式（１４）の零相電流指令値ｉｚ ＊を用いた場合の駆動電流（Ｕ相電流ｉｕ）を示す図である。なお、曲線Ｌ１はＵ相電流ｉｕに含まれる正弦波成分を示し、曲線Ｌ２はＵ相電流ｉｕに含まれる零相電流ｉｚを示している。このように、零相電流ｉｚを含ませることで、駆動電流を所定電流値ｉｒａｔｅｄ以内に納めつつ出力の向上を図ることができる。 FIG. 8 is a diagram showing a drive current (U-phase current i u ) when the zero-phase current command value i z * of the equation (14) is used. A curve L1 shows a sine wave component contained in the U-phase current i u, the curve L2 represents the zero-phase current i z contained in U-phase current i u. Thus, it is possible by including a zero-phase current i z, the output improving while pay drive current within the predetermined current value i rated.
　一方、ステップＳ１０２において、トルク指令値Ｔ＊が所定トルクＴthより小さいと判定された場合には、ステップＳ１０５へ進んで零相電流指令値ｉｚ ＊をｉｚ ＊＝０に設定する。図７で示す処理の場合の所定トルクＴthは図４の場合における所定電流値ｉｒａｔｅｄに対応しており、ほぼ同一のタイミングで零相電流ｉｚを含む状態へと切り替わる。また、図８における駆動電流において、正弦波成分（曲線Ｌ１）に対する零相電流ｉｚ（曲線Ｌ２）の比率は、トルク指令値Ｔ＊が大きくなるほど大きくなり、変形例１においても上述した実施の形態の場合と同様の作用効果を奏することができる。 On the other hand, if it is determined in step S102 that the torque command value T * is smaller than the predetermined torque Tth, the process proceeds to step S105, and the zero-phase current command value i z * is set to i z * = 0. Predetermined torque Tth in the case of the process shown in FIG. 7 corresponds to a predetermined current value i rated in the case of FIG. 4, switches to state comprising zero-phase current i z at substantially the same timing. In the drive current in FIG. 8, the ratio of the zero-phase current i z (curve L2) to the sine wave component (curve L1) increases as the torque command value T * increases. The same operational effects as in the case of the embodiment can be achieved.
（Ｃ４）さらに、図４，７に示す制御では、トルク指令値Ｔ＊が所定トルク閾値（所定トルクＴth）よりも小さい場合、または、トルク指令値Ｔ＊に基づく最大相電流指令値ｉｍａｘ ＊の絶対値が所定電流値ｉｒａｔｅｄよりも小さい場合には零相電流ｉｚをゼロとしているので、そのような運転状況においては零相電流ｉｚに起因するトルクリプルの発生を防止することができる。 (C4) Further, in the control shown in FIGS. 4 and 7, when the torque command value T * is smaller than the predetermined torque threshold (predetermined torque Tth), or the maximum phase current command value i max * based on the torque command value T * . because when the absolute value is smaller than the predetermined current value i rated has a zero-phase current i z is zero, in such operating conditions it is possible to prevent the occurrence of torque ripple caused by the zero-phase current i z .
（Ｃ５）さらに、図６に示すように、零相逆起電圧補償部８０は、駆動電流（ｉｄ、ｉｑ、ｉｚ）と回転子位置θとに基づいて、巻線２０１～２０３に誘起される電圧の零相電圧（零相逆起電圧補償値ｖｚ ＊＊）を算出し、トルク指令値Ｔ＊とモータ２００の角速度ωおよび回転子位置θとに基づく零相電圧指令値ｉｚ ＊に零相逆起電圧補償値ｖｚ ＊＊を加算したもので零相電流ｉｚを制御するのが好ましい。 (C5) Further, as shown in FIG. 6, the zero-phase counter electromotive force compensation unit 80 applies windings 201 to 203 to the windings 201 to 203 based on the drive current ( id , iq , iz ) and the rotor position θ. A zero-phase voltage of the induced voltage (zero-phase counter electromotive force compensation value v z ** ) is calculated, and the zero-phase voltage command value i based on the torque command value T * , the angular velocity ω of the motor 200, and the rotor position θ. It is preferable to control the zero-phase current i z by adding the zero-phase counter electromotive force compensation value v z ** to z * .
　駆動電流に零相電流ｉｚを含ませる場合、モータ駆動時に発生する逆起電圧の零相成分により、零相電流指令値ｉｚ ＊と零相電流検出値ｉｚとの差が増大し、トルクリプルが増大するおそれがある。しかし、上述のように、零相電圧指令値ｉｚ ＊に零相逆起電圧補償値ｖｚ ＊＊を加算することにより、巻線２０１～２０３に誘起された逆起電圧の零相成分を打ち消すように零相電圧指令値ｖｚ ＊が調整される。その結果、トルクリプルのさらなる低減を図ることができる。 If the inclusion of zero-phase current i z drive current, the zero-phase component of the back electromotive voltage generated when the motor drive, the difference between the zero-phase current command value i z * and the zero-phase current detection value i z increases, Torque ripple may increase. However, as described above, the zero-phase component of the counter electromotive voltage induced in the windings 201 to 203 is obtained by adding the zero-phase counter electromotive voltage compensation value v z ** to the zero phase voltage command value i z *. The zero phase voltage command value v z * is adjusted so as to cancel out. As a result, torque ripple can be further reduced.
（変形例２）
　図９は変形例２を示す図であり、モータ制御装置６１０の詳細を示すブロック図である。図９では、図３に示すモータ制御装置６１０の構成にＵＶＷ変換部９０が追加されている。ＵＶＷ変換部９０は、ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊と回転子位置θとに基づき三相電圧指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊を出力する。 (Modification 2)
FIG. 9 is a diagram showing a second modification and is a block diagram showing details of the motor control device 610. In FIG. 9, a UVW conversion unit 90 is added to the configuration of the motor control device 610 shown in FIG. The UVW converter 90 outputs three-phase voltage command values i u * , i v * , i w * based on the dq-axis current command values i d * , i q * and the rotor position θ.
　スイッチング信号生成部３０には、三相電圧指令値ｉｕ ＊、ｉｖ ＊、ｉｗ ＊に零相電圧指令値ｖｚ ＊と零相逆起電圧補償値ｖｚ ＊＊との和（ｖｚ ＊＋ｖｚ ＊＊）を加算したものが入力される。すなわち、ｉｕ ＊＋（ｖｚ ＊＋ｖｚ ＊＊）、ｉｖ ＊＋（ｖｚ ＊＋ｖｚ ＊＊）およびｉｗ ＊＋（ｖｚ ＊＋ｖｚ ＊＊）の３つの信号が入力される。 The switching signal generation unit 30, the three-phase voltage command values i u *, i v *, i w * on the zero-phase voltage command value v z * and the zero-phase back electromotive force compensation value v z sum of ** (v z * + v z **) obtained by adding a is input. That, i u * + (v z * + v z **), i v * + (v z * + v z **) and i w * + 3 one signal (v z * + v z ** ) is input The
　また、零相逆起電圧補償部８０では、ｄｑ軸電流検出値ｉｄ、ｉｑと零相電流検出値ｉｚの代わりに、ｄｑ軸電流指令値ｉｄ ＊、ｉｑ ＊と零相電流指令値ｉｚ ＊を用いて零相逆起電圧補償値ｖｚ ＊＊が算出される。 Also, the zero-phase back electromotive voltage compensation section 80, dq-axis current detection value i d, i q and instead of the zero-phase current detection value i z, dq axis current command value i d *, i q * and the zero-phase current Zero-phase counter electromotive force compensation value v z ** is calculated using command value i z * .
　このような構成においても、図３に示す構成の場合と同様の作用効果を奏することができる。すなわち、駆動電流における零相電流の比率をトルク指令値の大きさに応じて変化させることにより、出力の向上を図れると共に、トルクリプルの低減を図ることができる。 Even in such a configuration, the same operational effects as in the configuration shown in FIG. 3 can be obtained. That is, by changing the ratio of the zero-phase current in the drive current according to the magnitude of the torque command value, the output can be improved and the torque ripple can be reduced.
　１００…車両、１０…電流指令演算部、２０…ｄｑ軸電流制御部、３０…スイッチング信号生成部、４０…ｄｑ変換部、５０…零相電流算出部、６０…零相電流制御部、７０…速度変換部、８０…零相逆起電圧補償部、２００…モータ、２２０…電流センサ、６００…電力変換装置、６１０…モータ制御装置、６２０…インバータ、６２１…Ｕ相フルブリッジインバータ、６２２…Ｖ相フルブリッジインバータ、６２３…Ｗ相フルブリッジインバータ DESCRIPTION OF SYMBOLS 100 ... Vehicle, 10 ... Current command calculating part, 20 ... dq axis current control part, 30 ... Switching signal production | generation part, 40 ... dq conversion part, 50 ... Zero phase current calculation part, 60 ... Zero phase current control part, 70 ... Speed converter, 80 ... Zero-phase back electromotive force compensation unit, 200 ... Motor, 220 ... Current sensor, 600 ... Power converter, 610 ... Motor controller, 620 ... Inverter, 621 ... U-phase full bridge inverter, 622 ... V Phase full bridge inverter, 623 ... W phase full bridge inverter

Claims (6)

  1. 　相間で独立して巻かれた巻線を有するモータの駆動電流を、トルク指令値とモータの角速度および回転子位置とに基づいて制御するモータ制御装置であって、
    　前記駆動電流における零相電流の比率を前記トルク指令値の大きさに応じて変化させる、モータ制御装置。     A motor control device that controls a drive current of a motor having a winding wound independently between phases based on a torque command value, an angular velocity of the motor, and a rotor position,
    The motor control apparatus which changes the ratio of the zero phase current in the said drive current according to the magnitude | size of the said torque command value.
  2. 　請求項１に記載のモータ制御装置において、
    　各相電圧の総和である零相電圧を制御することにより前記比率を変化させる、モータ制御装置。     The motor control device according to claim 1,
    The motor control apparatus which changes the said ratio by controlling the zero phase voltage which is the sum total of each phase voltage.
  3. 　請求項１または２に記載のモータ制御装置において、
    　前記駆動電流が所定電流値以下となるように前記零相電流の比率を変化させる、モータ制御装置。     The motor control device according to claim 1 or 2,
    The motor control apparatus which changes the ratio of the said zero phase current so that the said drive current may become below a predetermined electric current value.
  4. 　請求項１に記載のモータ制御装置において、
    　前記トルク指令値が所定トルク閾値よりも小さい場合、または、前記トルク指令値に基づく駆動電流指令値が所定電流閾値よりも小さい場合には、前記駆動電流における零相電流をゼロとする、モータ制御装置。     The motor control device according to claim 1,
    When the torque command value is smaller than a predetermined torque threshold value, or when the drive current command value based on the torque command value is smaller than the predetermined current threshold value, the zero phase current in the drive current is set to zero. apparatus.
  5. 　請求項１に記載のモータ制御装置において、
    　前記駆動電流と前記回転子位置とに基づいて、前記巻線に誘起される電圧の零相電圧を算出し、
    　前記トルク指令値と前記モータの角速度および回転子位置とに基づく零相電圧指令値に前記零相電圧を加算したもので前記零相電流を制御する、モータ制御装置。     The motor control device according to claim 1,
    Based on the drive current and the rotor position, calculate a zero-phase voltage of the voltage induced in the winding,
    A motor control device that controls the zero-phase current by adding the zero-phase voltage to a zero-phase voltage command value based on the torque command value and the angular velocity and rotor position of the motor.
  6. 　相間で独立して巻かれた巻線を有するモータと、
    　請求項１から請求項４までのいずれか一項に記載のモータ制御装置とを備える電動車両。     A motor having windings wound independently between phases;
    An electric vehicle comprising the motor control device according to any one of claims 1 to 4.
PCT/JP2017/040034 2016-12-21 2017-11-07 Motor control device and electric vehicle WO2018116668A1 (en)

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