JP3674323B2 - Power converter control device - Google Patents

Description

【００２５】
【数２】

【００２６】
一方、速度検出器によって検出された速度検出値ωｍは、速度制御器１１において速度指令値ωｒと比較され、その値に基づいて電動機に必要なトルク電流指令が演算される。また、電動機の磁束電流指令も、速度に応じて変化させる場合もある。通常は、トルク電流指令をＩｑｒ、励磁電流指令をＩｄｒとして、電流制御器へ出力する。同時に、Ｉｑｒ，Ｉｄｒに基づいて、電動機の一次角周波数ω１を演算する。
【００２７】
【数３】

【００２８】
上式において、ｒ２ならびにＬ２は、電動機の定数である。
【００２９】
電流制御器では、Ｉｄｒ，Ｉｑｒと、検出値Ｉｄ，Ｉｑに基づいて、ｄｑ座標軸上の電圧指令Ｖｄ０，Ｖｑ０が演算される。Ｖｄ０，Ｖｑ０の演算には、通常は、ＰＩ制御（比例積分補償）が使用される。最近では、電流応答の改善のため、非干渉補償を付加する場合が多いが、本実施例では何れの制御方法であっても適用できる。Ｖｄ０，Ｖｑ０は、ｄｑ逆変換器１５において、数４，数５に従って、三相交流電圧指令Ｖｕ０，Ｖｖ０，Ｖｗ０に変換される。
【００３０】
【数４】

【００３１】
【数５】

【００３２】
上式におけるＶｕ０，Ｖｖ０，Ｖｗ０は、定常状態では歪みのない正弦波になる。この交流電圧指令値でも誘導電動機を駆動することができるが、近年では、零相加算器２を付加して、全ての相に共通の信号（零相電圧）を加算し、インバータの出力電圧最大値を向上させることがほぼ常識的になっている（参考文献２：電気学会論文集Ｄ，１０９巻１１号，Ｐ．８０９−８１６，平成元年）。零相加算の詳細については後述する。
【００３３】
零相を加算された電圧指令Ｖｕ１，Ｖｖ１，Ｖｗ１は、ＡＣ電圧リミッタにおいて最大値、ならびに最小値が制限され、最終的にはＶｕ２，Ｖｖ２，Ｖｗ２として出力される。各相の制限値は、電圧制限値演算器４により演算される。
【００３４】
ＰＷＭ制御器５では、Ｖｕ２，Ｖｖ２，Ｖｗ２の値に従い、三角波比較方式等によりＰＷＭ信号を発生し、インバータのスイッチング素子をドライブする。インバータは、Ｖｕ２〜Ｖｗ２に相当する電圧を出力し、誘導電動機を駆動する。
次に電圧制限値演算器４の動作を、図２を用いて説明する。
【００３５】
図２の最大電圧設定器４３において、インバータの出力可能な最大値Ｖmax が設定される。Ｖmax は、
【００３６】
【数６】

【００３７】
の関係にある。上式において、Ｖｄｃはインバータの直流電圧、Ｍmax は、インバータの最大変調率である（Ｍmaxは、０≦Ｍmax≦１）。Ｍmax は、インバータに使用されるスイッチングデバイスにより、０.９〜１.０程度に設定される。
【００３８】
交流電圧指令と、Ｖmax の関係を図３に示す。インバータの直流電圧Ｖｄｃの１／２の値が、理想的には最大電圧になるが、スイッチング素子の最小オフ時間を確保するため、ならびに、アーム短絡防止期間（デッドタイム）の影響による電圧降下分を補償するため、ＶmaxはＶｄｃ／２よりも数％低めに設定される。
絶対値演算器４１では、Ｖｕ１，Ｖｖ１，Ｖｗ１のそれぞれの絶対値を演算し、その中から大きさが最大のものを選択して（これをＶuvwmaxとする）、次式に示す係数Ｋlim を演算する。
【００３９】
【数７】

【００４０】
Ｋlimは、Ｖｕ１〜Ｖｗ１の少なくとも１相が、Ｖmaxを超えた場合に１より小さな値になる。よって、Ｋlimの値と１とを比較器４５で比較し、Ｋlim＜１の場合にＳｗ信号を１にする（Ｋlim＞１の時は、Ｓｗ＝０）。
【００４１】
ＡＣ電圧リミッタでは、Ｓｗ信号が「１」となった場合に、スイッチ３２を 「１」側に切り替えて、電圧指令を入力に対して一定比率に制御（制限）する。その場合、三相全ての電圧指令がＫlim倍に制限されることになる。Ｋlimは、三相の中で最も大きな値の相と、Ｖmax の比であるため、すべての相の電圧指令をＫlim倍することで、三相すべてが±Ｖmaxの範囲内に必ず制限されることになる。また、三相とも共通の比率で制限されるため、電圧指令の位相ずれは生じない。
【００４２】
次に、本発明の動作原理を従来方式と比較して説明する。図４（ａ）の３Ａが、一般的に使用されている従来方式の電圧リミッタの構成図である。図中の３２，４３，４５は、図２の同一番号のものと同じものである。Ｖｕ１が、Ｖmaxの値を超えた場合には、出力がＶmaxに制限され、−Ｖmax より下がった場合には−Ｖmax に制限される。この電圧リミッタの入力電圧Ｖｕ１と出力電圧Ｖｕ２の関係は、図４(ｂ)のようになる。仮に、図４(ｃ)のように、三相の中で一相の交流電圧指令が制限値を超えた場合を考えてみる。図において、Ｖｕ１がＶmax を超えた期間は、Ｖｕ２は図のようにＶmax に制限される。しかし、この間Ｖｖ１とＶｗ１は制限されることはなく、そのまま電圧リミッタを素通りしてしまう。この関係をαβ軸上で見てみると、図５の様になる。αβ変換は、
【００４３】
【数８】

【００４４】
によって定義される。図４（ｃ）のＶｕ１がリミッタにかかっている期間は、
Ｖｖ１，Ｖｗ１が変化しないため、数８よりＶβは変化しない。よって、図５ （ａ）のように、ベクトルとして電圧指令をみた場合、位相ずれが生じることになる（図のΔθ）。
【００４５】
一方、本発明による電圧リミッタ方式では、三相すべての電圧指令を等しい比率で制限するため、Ｖα，Ｖβとも同時に等しい比率で制限される。結果として、図５（ｂ）のように、位相ずれは生じない。よって、位相ずれによる電動機トルクの過不足がなくなり、高応答で安定な電動機制御が実現できる。
【００４６】
次に、零相加算の原理と、零相加算を行った場合の本実施例の動作について説明する。
【００４７】
図１における零相加算器２の構成を図６に示す。２１は、三相交流電圧指令 Ｖｕ０，Ｖｖ０，Ｖｗ０の値に基づいて、零相電圧Ｖ０を演算して出力する零相発生器である。Ｖ０は三相共通の値であるため、どのような波形であっても線間電圧への影響は原理的に生じない。何通りものＶ０が考えられるが、その代表例について説明する。
【００４８】
図７（ａ）が、零相加算を行う前の交流電圧指令である。この波形を式で表わすと次のようになる。
【００４９】
【数９】

【００５０】
次に、三相の電圧指令の中で、大きさが中間である相の１／２を零相電圧とした場合の波形を図７（ｂ）に示す。図７（ａ）において、θ１の区間では、Ｗ相が中間の大きさである。この時の零相電圧Ｖ０１は、
【００５１】
【数１０】

【００５２】
となる。次の９０°から１２０°の区間ではＶ相が中間相となる。このように、６０°毎に零相電圧の関数が変化して行く。Ｖ０１を加算すると、それぞれの電圧指令は図７（ｂ）のようになり、正弦波のピーク付近が下がった波形になる。この零相加算により、インバータの出力電圧最大値は、約１.１５倍（２／√３ 倍）に増加する（参考文献２：電気学会論文集Ｄ，１０９巻１１号，Ｐ．８０９−８１６，平成元年）。尚、この「１.１５倍 」という値は、電圧指令のピーク値がＶmax に一致する時点での比率（波形が歪まない条件での出力電圧最大値の比率）である。
【００５３】
図７（ｃ）は、三相の電圧指令の中で、絶対値の最も大きなものと、Ｖｄｃ／２、あるいは−Ｖｄｃ／２との差を取り、その値を零相電圧として各相に加算したものである。図７のθ２の期間においては、
【００５４】
【数１１】

【００５５】
が零相電圧になる。図７（ｃ）からわかるように、この零相加算を行うと、交流電圧指令の最大値，最小値が±Ｖmax を超えて±Ｖｄｃ／２の値に継続的に張り付いている期間が生じる。電圧指令が±Ｖｄｃ／２となる期間は、その相のスイッチング動作を行わないことを意味している。よって、数１１の零相加算を行うと、スイッチング回数が１／３減少する。また、この方式においても、図７(ａ)の正弦波に比べて、インバータの出力電圧は約１.１５倍(２／√３倍）に増加する（参考文献２：電気学会論文集Ｄ，１０９巻１１号，Ｐ．８０９−８１６，平成元年）。図７（ｃ）のように、三相の内２つの相のみがスイッチングする方式を「二相スイッチング」と呼ぶ。零相加算の方式としては、大きく別けて図７ （ｂ）のような三相スイッチングになるものと、同図（ｃ）のような二相スイッチングになるものに分類できる。
【００５６】
図１に示した電圧リミッタ方式は、図７（ｂ）に示す零相加算方式（三相スイッチング方式）にはそのまま適用できる。三相の電圧指令の中で、２つの相が同時にＶmaxと−Ｖmaxを超えることになるが、問題はない。
【００５７】
図７（ｃ）のような二相スイッチング方式の場合は工夫が必要である。
【００５８】
図８に、二相スイッチング方式を考慮した場合の実施例を示す。図１と比べると、零相加算器２とＡＣ電圧リミッタ３の処理が入れ替わっている。電圧制限値演算器４Ｂでは、Ｖmax がＫ０倍されている。
【００５９】
Ｖｕ０〜Ｖｗ０は、純粋な正弦波である。よって、零相加算の前に予めリミッタ処理を施してしまうと、零相加算による出力電圧の向上効果が得られなくなる。そこで、零相加算を考慮し、Ｖmaxの値を大きく(Ｋ０倍）設定しておく。Ｋ０を、
【００６０】
【数１２】

【００６１】
とすれば、零相加算の効果を活かし、位相ずれをなくすリミッタが実現できる。また、この図８の方式は、図７（ｂ）の零相加算に対しても、そのまま適用できる。
【００６２】
次に、線間電圧を使って、交流電圧指令を制限する方法を示す。これまでの実施例では、各相電圧の絶対値を演算し、その中から最大のものを選択し、係数 Ｋlim を演算していたが、線間電圧を使用しても同じ効果が得られる。図９に、その電圧制限値演算器４Ｃの構成を示す。図の４２〜４６は、図２における同一の番号のものと同じものである。４７は、３つの入力から一番小さな電圧を選択し、出力する最小値選択器である。この出力と、最大値選択器４２の出力の差を取ることで、線間電圧の最大値ＶＬmax が得られる。線間電圧が、インバータの直流電圧と一致する条件が、インバータ出力電圧の最大値であるから、ＶＬmax を１／２し、Ｖmaxとの比を計算することでＫlimが求められる。
【００６３】
尚、係数Ｋ０は、零相加算を行う場合はＫ０＝１とし、行わない場合はＫ０＝√３／２に設定する。これにより、図１，図８と同様に、位相を変えずに電圧指令を制限することができるようになる。
【００６４】
次に、交流電圧指令の位相情報のみを使用して、交流電圧指令を制限する実施例を説明する。図１０にその構成図を示す。
【００６５】
図１０において、１０は、Ｖｄ０，Ｖｑ０とｄｑ座標軸の位相φから、交流電圧指令の位相θを演算する電圧位相演算器、４Ｄは、θの値のみに依存して決定される各相の出力最大値、ならびに最小値を演算する電圧制限値演算器、３Ｄは、各相の最大値ならびに最小値に基づいて、電圧指令を制限するＡＣ電圧リミッタである。他の符号は、図１の発明における同一符号のものと同じものである。
次に、図１１を用いて、図１０の実施例の動作原理について説明する。図１１において、電圧位相演算器１０では、Ｖｄ０，Ｖｑ０の値を用いて、
【００６６】
【数１３】

【００６７】
が演算され、さらに、
【００６８】
【数１４】
θ＝φ＋δｄｑ …（数１４）
として、電圧位相θを計算する。電圧制限値演算器４Ｄでは、θで定まる各相の上限値（Ｖｕmax〜Ｖｗmax）と下限値（Ｖｕmin〜Ｖｗmin）を出力する。ＡＣ電圧リミッタでは、これらの制限値と電圧指令を各相毎に比較し、指令値が上限値を超えた場合には上限値に、下限値を下回った場合は下限値に切り替えて出力する。本実施例においては、図２の実施例のように、すべての相をＫlim 倍に制限するような処理は行わず、図４（ａ）の従来例におけるＶmax（−Ｖmax）の値を、位相θの値に応じて変化させることに相当する。
【００６９】
次に、これらの制限値とθの関係について説明する。
【００７０】
図１２（ａ）は、図１、あるいは図８の実施例において、一部の区間で交流電圧指令が過大になった場合のリミッタ処理後の波形である。この時、零相加算は数１０のもの（図７（ｂ））を使用している。交流電圧指令の振幅をさらに大きくしていくと、最終的には、図１２（ｂ）のようになる。これ以上、振幅を増やしても、リミット後の波形は図１２（ｂ）のままである。
【００７１】
三相交流の場合、位相θが定まると、三相電圧の瞬時値の比率は一義的に定まる。よって、ＡＣ電圧リミッタにより電圧指令値を制限したとしても、すべての相を等しい比率で制限する限り、リミット後の最終的な波形も一義的に定まることになる。この関係を定式化すれば、位相から制限値が求められることになる。
図１２の波形のθ１の区間においては、Ｕ相の上限値はＶmax 、Ｖ相の下限値は−Ｖmax となり、中間相のＷ相のみが複雑な関数になる。
【００７２】
Ｗ相の制限値は以下のようになる。
【００７３】
θ１の区間(３０°≦θ≦９０°)においては、零相加算により、指令値Ｖｗ１は、
【００７４】
【数１５】

【００７５】
となる。また、Ｕ相は、
【００７６】
【数１６】

【００７７】
である。この期間では、Ｕ相とＷ相が同時にＶmaxと−Ｖmaxに到達する。一方を代表して、Ｕ相で係数Ｋlimを計算すると、
【００７８】
【数１７】

【００７９】
となる。これに数１５を掛けたものが、Ｗ相の制限値である。上限値Ｖｗmax は、
【００８０】
【数１８】

【００８１】
下限値Ｖｗmin は、
【００８２】
【数１９】

【００８３】
となる。よって、Ｖｐ（振幅）には全く依存せず、θのみの関数であることがわかる。
【００８４】
ここで、数１８（数１９）を一般化し、
【００８５】
【数２０】

【００８６】
として、各位相区間における各相の制限値を計算すると、図１３のようになる。図１３からわかるように、電圧位相が定まると、各相ともＶmaxかＶminの何れか一方のみが規定されるが、残りの制限値は規定されない（定義不要）。
【００８７】
この図１３の関係を、電圧制限値演算器４Ｄにテーブルとして保存し、与えられたθに対して各制限値を出力すればよい。
【００８８】
尚、本実施例の場合、すべての相が必ず同時にリミットされることになるので、図１１における比較器４５は、一つの相だけに備え、出力を各相共通とすることができる。
【００８９】
以上、本発明の実施例として、交流電圧指令に対してリミット処理を行う手法を説明した。前述したように、図１、ならびに図８の実施例を用いて、電圧指令の振幅を増加させていくと、最終的には図１２（ｂ）の波形となる。よって、図１、ならび図８の実施例は、図１０のものと効果は等しい。
【００９０】
図１２（ｂ）の波形の状態で誘導電動機を駆動すると、波形の歪みは大きくなるが、インバータの出力電圧最大値は平均的に大きくなる。また、この時の位相ずれは生じていない。よって、過渡時等、一時的に大きな電圧が必要な場合に、本発明は極めて有効である。
【００９１】
従来方式（図４）において電圧指令の振幅を増加させていくと、振幅の増加に伴って、中間相（図１２のθ１の区間ではＷ相に相当）の値が変化し続け、最終的には中間相の値もＶmaxか−Ｖmaxに張り付く形になる。この場合にも、インバータの出力電圧最大値は大きくはなるものの、位相ずれは最大で±３０°に達する。よって、軸ずれが発生し、過渡応答が劣化する。
【００９２】
次に本発明の他の実施例について、図１４を用いて説明する。
【００９３】
図１４において、１Ｅは、誘導電動機に印加される電圧指令を演算する電圧指令演算器、３Ｅは、電圧指令Ｖｄ０，Ｖｑ０の大きさを制限するＤＣ電圧リミッタ、４Ｅは、電圧位相θに基づいて、Ｖｄ０，Ｖｑ０の制限値を演算する電圧制限値演算器、１５Ｅは、ｄｑ座標軸上の電圧指令をαβ軸上の電圧指令に変換するｄｑ−αβ座標変換器、５Ｅは、αβ軸上の電圧指令に基づいて、ＰＷＭ信号を発生させるＰＷＭ制御器である。他の符号は、図１、ならびに図１０の実施例の同一符号のものと同じものである。
【００９４】
本実施例では、電圧指令を三相交流に座標変換していない点に特徴がある。その代り、電圧指令のリミッタ処理を、ｄｑ軸上の電圧指令に対して行う。この際、電圧位相θにより、電圧制限値を変化させている。また、ＰＷＭ制御器５Ｅにおいては、αβ軸の電圧指令に基づいて、直接ＰＷＭ信号を作成している。このようなＰＷＭ制御は、空間ベクトル方式と呼ばれており、インバータの出力電圧を二軸の空間に表現して、パルスの出力時間を計算するものである。詳細は後述する。
【００９５】
図１５に、ＤＣ電圧リミッタ３Ｅ、ならびに電圧制限値演算器４Ｅの構成を示す。図の４３Ｅは、ｄｑ軸上の電圧制限値の設定器、４８は、ルート演算器、 ４９は、θに対して所定の関数を発生させる関数発生器である。他の部品は、これまでの実施形態における同一の符号のものと同じものである。
【００９６】
次に、本発明の動作原理を説明する。電圧制限値設定器４３Ｅの値Ｖdqlimit は、
【００９７】
【数２１】

【００９８】
のように設定する。この値は、線間電圧の最大値をＶｄｃ・Ｍmax として（つまり数９において、Ｖｐ＝Ｖｄｃ・Ｍmax／√３ として）電圧をｄｑ変換した時の、ｄｑ座標上におけるベクトルの大きさに相当する。すなわち、正弦波を出力するための最大値に一致する。
【００９９】
ＶdqlimitとＫｖを掛け算し、電圧制限値Ｖlimitを得ている。Ｋｖは、電圧位相θの関数になっており、その関数を図１６に示す。Ｋｖの値は６０°周期で変化する。
【０１００】
一方、Ｖｄ０とＶｑ０は、それぞれ二乗された後に加算され、ルート演算器 ４８によりベクトルの大きさ（｜Ｖ｜）が出力される。電圧を制限する係数
Ｋlimは、｜Ｖ｜とＶlimitの比から求められる。
【０１０１】
ＤＣ電圧リミッタ３Ｅの処理は、ＡＣ電圧リミッタと同様に、ｄｑそれぞれの電圧指令を、Ｋlimが１より小となった場合にＫlim倍して制限する。
【０１０２】
次に本実施例の動作原理を説明するが、まず初めに二軸上でのインバータの出力電圧表現について説明する。
【０１０３】
図１７は、三相フルブリッジ形の電圧形インバータの構成図である。図において、Ｕ〜Ｗ相の各相のスイッチング素子は、上側か下側の必ずどちらか一方のみがオンしている。仮に、Ｕ相は上側(Ｓｕｐ)がオン、Ｖ相，Ｗ相は下側（ＳｖｎとＳｗｎ）がオンしているものとすると、各相電圧Ｖｕｉ〜Ｖｗｉは、
【０１０４】
【数２２】

【０１０５】
となる。これを数８に従い、αβ軸上の電圧にすると、
【０１０６】
【数２３】

【０１０７】
となる。数２３をベクトルＶ１として、αβ空間に図示すると、図１８のようになる。同様に、
【０１０８】
【数２４】

【０１０９】
とすると、
【０１１０】
【数２５】

【０１１１】
となる。このようにして空間ベクトルを計算すると、インバータの出力できる電圧は、図１８に示す８つのベクトル（内、零ベクトルが二つ）になる。
【０１１２】
ＰＷＭ信号を作成するには、電圧指令も空間ベクトルで表現する。図１８に示すように、電圧指令のベクトルＶｒが、Ｖ０（零ベクトル）、Ｖ１，Ｖ２で囲まれる三角形の空間内部にあるものとする。この場合、微小時間Ｔ内において、ベクトルＶ０，Ｖ１，Ｖ２をそれぞれｔ０，ｔ１，ｔ２時間だけ出力し、近似的にＶｒを出力する。電圧指令と、各ベクトルの出力時間の関係は、
【０１１３】
【数２６】
Ｖｒ＝ｔ１・Ｖ１＋ｔ２・Ｖ２＋ｔ０・Ｖ０ …（数２６）
【０１１４】
【数２７】
Ｔ＝ｔ１＋ｔ２＋ｔ０ …（数２７）
の二つであり、この２式より、ｔ０〜ｔ２が求められる。つまり、空間ベクトルでＰＷＭ制御を行う場合は、電圧指令を空間ベクトルで表現し、それを近似するためのインバータのベクトルを選択し、それぞれの出力時間を演算して、最終的にＰＷＭパルスへ変換する。
【０１１５】
次に、空間ベクトルでインバータを制御する場合の出力電圧範囲を考察してみる。電圧指令が正弦波である場合、Ｖｒは、円を描いて一定速度(角周波数速度)で回転する。よって、インバータの出力電圧を正弦波に限定するのであれば、インバータの出力電圧範囲は図１８に示す円の内側になる。この円内に電圧ベクトルを制限すると、従来技術（特開平9−149699 号）の方式になる。本実施例においても、図１５における関数発生器４９の出力Ｋｖを、常に「１」とすると参考文献１と同等になる。
【０１１６】
本実施例においては、インバータの出力電圧を最大に利用し、高応答化を実現するため、Ｋｖをθに応じて１以上に可変させている。出力電圧を正弦波に限定しなければ、インバータの出力電圧範囲は、図１８に示す六角形の内側になる。すなわち、正弦波に限定した場合の出力範囲よりも、広い範囲の電圧が出力可能である。この六角形の領域すべてを利用するため、Ｋｖの値をθに応じて可変としている。
【０１１７】
次に、θとＫｖの関係を図１９を用いて説明する。図において、出力し得る最大のベクトルの大きさをｘとする。図１９の関係より、
【０１１８】
【数２８】
ｙ＝ｘcos（３０°−θ） …（数２８）
となる。よって、
【０１１９】
【数２９】

【０１２０】
となる。ここで、ｙは正弦波出力範囲の最大値に一致するので、ｙ＝Ｖdqlimit である。よって、
【０１２１】
【数３０】

【０１２２】
となる。ただし、上式は、０≦θ≦６０°の範囲で成立する。他のθの区間でも同様にＫｖの値を求めることができる。結果として、図１６に示す関数となる。
以上、図１４の実施例により、ｄｑ座標軸上において、インバータ出力電圧を最大限に利用し、かつ、ベクトル位相を変化させずに指令値を制限することができるようになる。
【０１２３】
次にｄｑ座標軸上の電圧指令を制限する他の実施例について説明する。
【０１２４】
図２０における電圧制限値演算器４Ｆが、本実施例の特徴である。この電圧制限値演算器４Ｆを、図１４における電圧制限値演算器４Ｅの代りに適用し、さらに図１４の電圧位相演算器を排除して使用する。他は図１４と同じである。
【０１２５】
次に、本実施例の動作原理について説明する。図２０において、電圧制限値演算器４Ｆでは、電圧指令Ｖｄ０，Ｖｑ０を入力し、ｄｑ座標軸位相φで一旦三相交流電圧に座標変換する。その後、図９の実施例と同様に、線間電圧の最大値を計算し、その値の１／２とＶmaxの値から係数Ｋlimを演算する。前述したように、線間電圧とインバータの直流電圧が一致した場合が、インバータの最大出力範囲になるため（図１８の六角形に相当）、この演算により、出力電圧範囲は最大限まで活用され、かつ、電圧位相は変化しないことになる。本発明は、図１４の発明と等価になるが、ルート演算が不要であるため、演算時間が短くて済むようになる。
【０１２６】
次に本発明による順変換装置の実施例について、図２１を用いて説明する。
【０１２７】
図２１において、１Ｇは、コンバータ６Ｇ（順変換装置）の入力電圧指令を演算する電圧指令演算器、８Ｇは、三相交流電源、９Ｇは、三相交流電源の電源位相を検出する位相検出器、５０は、コンバータ６Ｇの出力電圧を平滑する平滑コンデンサ、５１は、コンバータ６Ｇの出力電圧Ｖｄｃの電圧を検出する電圧検出器、５２は、コンバータに接続された負荷装置、１７は、コンバータ出力電圧指令Ｖｄｃｒと、検出値Ｖｄｃの値から、コンバータの入力電流指令を演算する電圧制御器である。その他の部品は、これまでの実施例における部品と同一のものである。
【０１２８】
コンバータの制御においては、交流電源の位相φを検出し、電源電圧と同位相の電流成分をＩｑ、９０°位相のずれた成分をＩｄとして、入力電流の制御を行う。この時、Ｉｄ＝０となるように制御することで、常に電源力率１が達成できる。また、直流電圧Ｖｄｃの制御は、Ｉｑを用いて行うようにする。Ｉｑｒには、負荷装置５２が必要とする電流をフィードフォワードして加える場合もある。本実施例においても、電圧制限値演算器４、並びにＡＣ電圧リミッタ３を付加することで、コンバータの入力電圧を最大限に利用することができるようになる。これまで説明したインバータの制御に関する各実施例が、そのまま適用できる。
【０１２９】
現在広く用いられているコンバータの構成を、図２２に示す。仮に、ＰＷＭ制御器の出力であるＰＷＭパルスをすべてオフして、コンバータを構成するすべてのスイッチング素子のゲート信号をオフしたとする。すると、入力電流は、各スイッチング素子に逆並列に接続されたフリーホイールダイオードを介して流れることになり、ダイオード整流器として動作する。
【０１３０】
ダイオード整流器は、コンバータのようにスイッチング動作が伴わないために、効率が非常によい。負荷装置の消費電力が大きくなると、入力電流波形のひずみが大きくなり、力率も悪くなるが、軽負荷状態であれば問題はない。
【０１３１】
よって、負荷装置の消費電力に応じて、ダイオード整流器と、コンバータ動作とを切り替えれば、総合的に効率の良い順変換システムが実現できる。
【０１３２】
しかし、実際にダイオード整流器動作からコンバータ動作へ切り替えると、従来のコンバータでは、図２３（ａ）に示すような突入電流が発生する。これは、ダイオード整流器の出力電圧が、コンバータの出力電圧の最低値よりも低く、図のようにΔＶの差があることに問題がある。図２２の構成のコンバータは昇圧形の変換器であるため、コンバータ入力電圧と、出力直流電圧との関係は反比例になる。つまり、直流電圧を最小にするには、コンバータ入力電圧を最大にする必要がある。しかし、従来のコンバータでは、入力電圧範囲が図１８の円内であったため、電圧マージンを考慮すると、最大電圧を得ることは困難であった。
【０１３３】
しかし、本発明による順変換装置においては、コンバータの入力電圧範囲を図１８の六角形にまで拡大することができるので、入力電圧の平均値を増加することができ、図２３（ｂ）のように、Ｖｄｃをスムーズに立ち上げることが可能になる。よって、切り替わりに伴う突入電流も少なくなる。
【０１３４】
尚、上記の実施例におけるインバータ、およびコンバータは、三相フルブリッジのものを使用したが、３レベル変換器や、その他の多重変換器についても同様に適用できる。
【０１３５】
また、電動機駆動システムとして誘導電動機の例を説明したが、同期電動機等、他のすべての交流電動機に対しても適用可能である。
【０１３６】
【発明の効果】
本発明によれば、インバータの出力電圧を最大限にまで活用し、かつ、電圧位相の誤差を零に制御することができるようになる。
【０１３７】
本発明をコンバータ（順変換装置）に適用した場合、コンバータの出力電圧範囲が拡大し、順変換システムとして、装置の高効率化が実現できるようになる。
【図面の簡単な説明】
【図１】本発明の一実施例である交流電動機駆動装置の構成図。
【図２】ＡＣ電圧リミッタと電圧制限値演算器の構成図。
【図３】交流電圧指令と電圧制限値との関係を示す図。
【図４】従来の電圧リミッタの構成図。
【図５】従来方式と本発明による方式の効果を示す図。
【図６】零相加算器の構成図。
【図７】零相加算の原理を示す図。
【図８】本発明の一実施例である交流電動機駆動装置の構成図。
【図９】電圧制限値演算器の構成図。
【図１０】本発明の一実施例である交流電動機駆動装置の構成図。
【図１１】電圧位相演算器，電圧制限値演算器とＡＣ電圧リミッタの構成図。
【図１２】図１０の実施例の動作原理を説明する図。
【図１３】位相θに対する電圧制限値を示す図。
【図１４】本発明の一実施例である交流電動機駆動装置の構成図。
【図１５】電圧制限値演算器とＤＣ電圧リミッタの構成図。
【図１６】図１４の実施例において使用される関数を表わす図。
【図１７】三相フルブリッジ電圧形インバータの主回路構成図。
【図１８】インバータの出力電圧、ならびに電圧指令の空間ベクトル表示を示す図。
【図１９】図１４の実施例の原理を示す図。
【図２０】ＤＣ電圧リミッタと電圧制限値演算器の構成図。
【図２１】本発明の一実施例である順変換装置の構成図。
【図２２】三相フルブリッジ電圧形コンバータの主回路構成図。
【図２３】従来方式と本発明によるコンバータの動作を示す図。
【符号の説明】
１…交流電圧指令演算器、２…零相加算器、３…ＡＣ電圧リミッタ、４…電圧制限値演算器、５…ＰＷＭ制御器、６…インバータ、７…電流検出器、８…誘導電動機、９…速度検出器、１１…速度制御器、１２…積分器、１３…ｄｑ座標変換器、１４…電流制御器。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an AC motor drive system that drives a three-phase AC motor using a power converter such as an inverter, and a forward converter that converts AC power into DC, and in particular, a voltage utilization rate by improving a voltage limiter. And stabilization of the control system during the transition.
[0002]
[Prior art]
In the case of driving an AC motor using an inverter, vector control has been widely used in recent years. Vector control controls the magnitude and phase of the current flowing through the AC motor, and linearizes the generated torque of the motor with respect to the command torque. Vector control can be realized by separating an alternating current into a magnetic flux component and a torque component and controlling each component. As a result of these current controls, the inverter voltage is output in a predetermined phase. With this technology, AC motors are able to achieve performance equal to or better than DC motors.
[0003]
In the converter (forward converter), similarly to the AC motor, the magnitude and phase of the current input to the converter are controlled. This is a technique equivalent to vector control of an AC motor, and a power factor of 1 can be realized simultaneously with control of DC voltage.
The output (input) voltage of the inverter (converter) has a limit, and is usually used in a region where the voltage is not saturated. The limit value of the voltage is determined by the DC voltage of the converter and the characteristics of the switching elements constituting the converter. Since it is necessary to secure the minimum pulse width (on width, off width) in the switching element, it is necessary to ensure that amount as a voltage margin. Further, in order to prevent a short circuit due to a delay in the switching operation, it is necessary to provide an arm short circuit prevention period. Since the arm short circuit prevention period causes current distortion, a compensation voltage is usually applied. This compensation voltage must also be secured as a voltage margin. In order to ensure these voltage margins, the limit value of the voltage used for the control itself is about several percent lower than the DC voltage.
[0004]
Therefore, a voltage limiter is inserted on the AC side or on the rotation coordinate axis (on the dq coordinate axis) so as not to exceed the above limit value.
[0005]
[Problems to be solved by the invention]
In AC motor control, the voltage command may become large during acceleration or when a load disturbance occurs. For this reason, the voltage value of the inverter output in the steady state is set low so as not to be applied to the limiter. In order to increase the inverter capacity at a constant time, the voltage value in the steady state may be increased, but the voltage limiter is likely to be applied during the transition.
[0006]
As described above, in the vector control that is widely used at present, it is necessary to control the magnitude and phase of the current to predetermined values. When the voltage is applied to the limiter, these relationships are lost, and the transient response is deteriorated and the torque is lost. May occur. In particular, the phase shift causes a difference between the magnetic flux axis for control and the actual magnetic flux axis of the electric motor, and the vector control principle itself is broken.
[0007]
The phase shift occurs because the voltage limiter is inserted independently for each phase. For example, when only one phase exceeds the limit value and the other phases are output as they are, the balance of the three phases changes and the phase relationship is lost. The same applies when the limiter process is performed on the dq coordinate axes.
[0008]
An example of these countermeasures is document 1 (Japanese Patent Laid-Open No. 9-149699). This is a method of correcting the voltage command of each axis so that the voltage commands on the dq coordinate axes are vector-synthesized and only the magnitude is limited without changing the vector phase. However, since the limit value of the voltage vector is constant, the range of the output voltage of the inverter viewed on the dq axis is a circle with the limit value as the radius.
[0009]
The output voltage range of a normal power converter is a hexagon on the dq axis. In the method of Document 1, the circle inscribed in the hexagon is the output voltage range of the inverter, and the unused portion remains. Since the portion between the hexagon and the circle is not used, the transient response may be delayed when the voltage command actually reaches the limit value.
[0010]
In addition to the above problems, the forward conversion device has the following problems. In the voltage type converter, a diode is connected in antiparallel to each switching element, and functions as a diode rectifier by turning off all the switching elements. Therefore, the converter operation and the diode rectifier operation can be switched and used according to the voltage required by the load, and the efficiency of the forward conversion system can be improved. However, the DC voltage value when operating as a diode rectifier is lower than the DC voltage value (minimum value) during converter operation, and it is difficult to smoothly switch between them. This occurs because the aforementioned voltage limit value is several percent less than the DC voltage value. Therefore, if switching between the converter operation and the diode rectifier operation is simply performed, an excessive current accompanying a switching shock may occur and the switching element may be destroyed.
[0011]
[Means for Solving the Problems]
In the control device for the power converter according to the present invention, when the AC voltage command is limited, the voltage commands of all other phases are simultaneously limited when at least one phase exceeds the limit value. If the limiting ratio is made equal for each phase, only the amplitude is suppressed without changing the voltage phase. Thereby, phase shift is suppressed and it can control that torque loss and excessive current occur. Furthermore, the output voltage range extends to the maximum of the converter.
[0012]
Moreover, since the limit value of the final AC voltage command is formulated according to the phase of the voltage command, the AC voltage command is limited (limit processing) using these formulated tables (or arithmetic expressions). However, the same effect can be obtained.
[0013]
Furthermore, when limiting the voltage command value on the dq coordinate axis, these voltage commands are once converted into an AC voltage command, compared with a predetermined limit value, and when at least one phase exceeds the limit value The two voltage commands on the dq coordinate axes may be limited at an equal ratio.
[0014]
In addition, since the final limit value of the voltage command on the dq coordinate is formulated by the phase of the voltage command, the voltage command on the dq coordinate axis can be expressed using these formulated tables (or arithmetic expressions). Limit it.
[0015]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 shows a first embodiment of the present invention.
[0016]
FIG. 1 shows an induction motor driving apparatus that drives a three-phase induction motor by a PWM inverter.
[0017]
In FIG. 1, 1 is an AC voltage command calculator for calculating a voltage command applied to the induction motor, 2 is a zero-phase voltage for improving the maximum output voltage of the inverter, and is added to all three phases. The zero-phase adder 3 performs an AC voltage limiter that limits the magnitude of the three-phase AC voltage command according to the limit values (maximum value and minimum value), and 4 indicates a voltage limit value calculator that calculates the limit value 3 5 is a PWM controller that generates a PWM (Pulse Width Modulation) signal based on an AC voltage command, 6 is an inverter driven by the PWM signal, 7 is a current detector that detects the current flowing through the induction motor, 8 is an induction motor driven by an inverter, 9 is a speed detector for detecting the rotational speed of the induction motor, and 11 is a current index of the induction motor based on the rotational speed ωm of the induction motor and the rotational speed command ωr. A speed controller for calculating Idr, Iqr and the primary angular frequency ω1, 12 an integrator for integrating ω1 to calculate a phase angle φ, and 13 a current detection value of the induction motor based on the phase angle φ. A dq coordinate converter 14 for converting the dq coordinate axis to the current detection values Id, Iq on the dq coordinate axis, and the current controller for calculating the voltage commands Vd0, Vq0 on the dq coordinate axis based on the Idr, Iqr. It is.
[0018]
The configurations of the AC voltage limiter and the voltage limit value calculator in FIG. 1 are shown in FIG. A part number 31 is a multiplier that multiplies the AC voltage commands Vu1, Vv1, and Vw1 and a coefficient Klim output from the voltage limit value calculator, 32 is a switch that switches between two input signals, and the control signal Sw is “1”. "" Is switched to the 1 side, and "0" is switched to the 0 side. 41 is an absolute value calculator that calculates the absolute values of Vu1, Vv1, and Vw1, 42 is a maximum value selector that selects and outputs the largest one of the three input signals, and 43 Is a maximum voltage setter that sets the maximum value Vmax that one phase of the inverter can output, 44 is a divider that divides Vmax by the output of 42, 45 is a "+" side comparing two input signals This is a comparator that outputs “1” when the value of is greater than the value on the “−” side, and outputs “0” in the opposite case.
[0019]
The operation of FIG. 1 will be briefly described as follows. The AC voltage command calculator 1 is a controller for vector control of the induction motor 8. The rotational speed command of the induction motor, the rotational speed, and the current value flowing through the induction motor are read, and the AC voltage commands (Vu0, Vv0, Vw0) required by the induction motor are calculated and output based on these values. The inverter is controlled based on these AC voltage commands, and a voltage according to the command is applied to the induction motor to realize vector control. In order to realize the vector control with high accuracy, calculation by the AC voltage command calculator 1 is important, but it is also important to reduce the error from the AC voltage command to the actual applied voltage as much as possible.
[0020]
The amplitude value of the AC voltage command calculated by the AC voltage command calculator 1 increases in proportion to the speed in a steady state, but when the speed command of the induction motor changes suddenly, or the load of the induction motor changes suddenly. In a transient state, the amplitude value suddenly increases or decreases. At this time, the voltage range that can be output from the inverter may be exceeded, which is a problem.
[0021]
The zero-phase adder 2 is a voltage command corrector for improving the output voltage of the inverter, and is one of the techniques for reducing the above problems. In recent years, it is generally used (details will be described later). Reference numerals 3 and 4 denote voltage limiters which are characteristic portions of the present invention. Even when the voltage changes suddenly and exceeds the allowable value of the inverter, a voltage value limit process is performed so that vector control is established as much as possible.
[0022]
Next, details of the operation of FIG. 1 will be described.
[0023]
In FIG. 1, the alternating currents Iu, Iv, Iw detected by the current detector 7 are converted into currents Id, Iq on the dq coordinate axis according to the following equations 1 and 2 in the dq converter.
[0024]
[Expression 1]

[0025]
[Expression 2]

[0026]
On the other hand, the speed detection value ωm detected by the speed detector is compared with the speed command value ωr in the speed controller 11, and a torque current command necessary for the motor is calculated based on the value. Further, the magnetic flux current command of the electric motor may be changed according to the speed. Normally, the torque current command is set to Iqr and the excitation current command is set to Idr, and then output to the current controller. At the same time, the primary angular frequency ω1 of the electric motor is calculated based on Iqr and Idr.
[0027]
[Equation 3]

[0028]
In the above equation, r2 and L2 are electric motor constants.
[0029]
In the current controller, voltage commands Vd0 and Vq0 on the dq coordinate axis are calculated based on Idr and Iqr and detected values Id and Iq. In general, PI control (proportional integration compensation) is used for the calculation of Vd0 and Vq0. Recently, non-interference compensation is often added to improve the current response, but any control method can be applied in this embodiment. Vd0 and Vq0 are converted into three-phase AC voltage commands Vu0, Vv0 and Vw0 in accordance with Equations 4 and 5 in the dq inverse converter 15.
[0030]
[Expression 4]

[0031]
[Equation 5]

[0032]
Vu0, Vv0, and Vw0 in the above equation are sine waves without distortion in a steady state. Induction motors can be driven with this AC voltage command value, but in recent years, a zero-phase adder 2 is added to add a common signal (zero-phase voltage) to all phases, and the inverter output voltage maximum Increasing the value has become almost common sense (Reference 2: IEEJ Proceedings D, Vol. 109, No. 11, P.809-816, 1989). Details of the zero-phase addition will be described later.
[0033]
The voltage command Vu1, Vv1, Vw1 to which the zero phase is added is limited in the maximum value and the minimum value in the AC voltage limiter, and is finally output as Vu2, Vv2, Vw2. The limit value of each phase is calculated by the voltage limit value calculator 4.
[0034]
The PWM controller 5 generates a PWM signal by a triangular wave comparison method or the like according to the values of Vu2, Vv2, and Vw2, and drives the switching element of the inverter. The inverter outputs a voltage corresponding to Vu2 to Vw2 and drives the induction motor.
Next, the operation of the voltage limit value calculator 4 will be described with reference to FIG.
[0035]
In the maximum voltage setting unit 43 of FIG. 2, a maximum value Vmax that can be output from the inverter is set. Vmax is
[0036]
[Formula 6]

[0037]
Are in a relationship. In the above equation, Vdc is the DC voltage of the inverter, and Mmax is the maximum modulation rate of the inverter (Mmax is 0 ≦ Mmax ≦ 1). Mmax is set to about 0.9 to 1.0 depending on the switching device used for the inverter.
[0038]
FIG. 3 shows the relationship between the AC voltage command and Vmax. The value of ½ of the DC voltage Vdc of the inverter is ideally the maximum voltage. However, in order to ensure the minimum off time of the switching element and the voltage drop due to the influence of the arm short circuit prevention period (dead time) In order to compensate for Vmax, Vmax is set to be several percent lower than Vdc / 2.
The absolute value calculator 41 calculates the absolute value of each of Vu1, Vv1, and Vw1, selects the one with the largest magnitude (referred to as Vuvwmax), and calculates the coefficient Klim shown in the following equation. To do.
[0039]
[Expression 7]

[0040]
Klim becomes a value smaller than 1 when at least one phase of Vu1 to Vw1 exceeds Vmax. Therefore, the value of Klim is compared with 1 by the comparator 45, and the Sw signal is set to 1 when Klim <1 (Sw = 0 when Klim> 1).
[0041]
In the AC voltage limiter, when the Sw signal becomes “1”, the switch 32 is switched to the “1” side to control (limit) the voltage command at a constant ratio with respect to the input. In that case, the voltage commands for all three phases are limited to Klim times. Since Klim is the ratio of the largest value of the three phases to Vmax, the voltage command for all phases must be multiplied by Klim to ensure that all three phases are limited within the range of ± Vmax. become. Further, since the three phases are limited by a common ratio, there is no phase shift of the voltage command.
[0042]
Next, the operation principle of the present invention will be described in comparison with the conventional method. 3A in FIG. 4A is a configuration diagram of a voltage limiter of a conventional method that is generally used. 32, 43, and 45 in the figure are the same as those having the same numbers in FIG. When Vu1 exceeds the value of Vmax, the output is limited to Vmax, and when it falls below -Vmax, it is limited to -Vmax. The relationship between the input voltage Vu1 and the output voltage Vu2 of this voltage limiter is as shown in FIG. Consider the case where the AC voltage command of one phase exceeds the limit value in the three phases as shown in FIG. In the figure, during the period when Vu1 exceeds Vmax, Vu2 is limited to Vmax as shown in the figure. However, Vv1 and Vw1 are not limited during this period, and pass through the voltage limiter as they are. Looking at this relationship on the αβ axis, it becomes as shown in FIG. αβ conversion is
[0043]
[Equation 8]

[0044]
Defined by The period during which Vu1 in FIG.
Since Vv1 and Vw1 do not change, Vβ does not change from Equation 8. Therefore, when the voltage command is viewed as a vector as shown in FIG. 5A, a phase shift occurs (Δθ in the figure).
[0045]
On the other hand, in the voltage limiter system according to the present invention, since all three phase voltage commands are limited at an equal ratio, both Vα and Vβ are simultaneously limited at an equal ratio. As a result, there is no phase shift as shown in FIG. Therefore, excess and deficiency of the motor torque due to the phase shift is eliminated, and stable motor control with high response can be realized.
[0046]
Next, the principle of zero-phase addition and the operation of this embodiment when zero-phase addition is performed will be described.
[0047]
The configuration of the zero-phase adder 2 in FIG. 1 is shown in FIG. Reference numeral 21 denotes a zero-phase generator that calculates and outputs the zero-phase voltage V0 based on the values of the three-phase AC voltage commands Vu0, Vv0, and Vw0. Since V0 is a value common to three phases, no influence on the line voltage occurs in principle regardless of the waveform. A number of V0s are conceivable. A typical example will be described.
[0048]
FIG. 7A shows an AC voltage command before zero-phase addition. This waveform is expressed as follows.
[0049]
[Equation 9]

[0050]
Next, in the three-phase voltage command, a waveform in the case where ½ of the phase having an intermediate magnitude is set as a zero-phase voltage is shown in FIG. In FIG. 7A, the W phase has an intermediate size in the section θ1. The zero-phase voltage V01 at this time is
[0051]
[Expression 10]

[0052]
It becomes. In the next 90 ° to 120 ° interval, the V phase is an intermediate phase. In this way, the function of the zero phase voltage changes every 60 °. When V01 is added, each voltage command becomes as shown in FIG. 7B, and the waveform near the peak of the sine wave is lowered. By this zero-phase addition, the maximum output voltage value of the inverter is increased by about 1.15 times (2 / √3 times) (Reference 2: IEEJ Transaction D, Vol. 109, No. 11, P.809-816). , 1989). The value of “1.15 times” is a ratio when the peak value of the voltage command coincides with Vmax (the ratio of the maximum value of the output voltage under the condition that the waveform is not distorted).
[0053]
Fig. 7 (c) shows the difference between the largest absolute value of the three-phase voltage commands and Vdc / 2 or -Vdc / 2, and adds that value to each phase as a zero-phase voltage. It is a thing. In the period of θ2 in FIG.
[0054]
[Expression 11]

[0055]
Becomes zero phase voltage. As can be seen from FIG. 7 (c), when this zero-phase addition is performed, there occurs a period in which the maximum value and the minimum value of the AC voltage command exceed ± Vmax and are continuously stuck to the value of ± Vdc / 2. . The period during which the voltage command is ± Vdc / 2 means that the switching operation of the phase is not performed. Therefore, when the zero-phase addition of Equation 11 is performed, the number of switching operations is reduced by 1/3. Also in this method, the output voltage of the inverter increases by about 1.15 times (2 / √3 times) compared to the sine wave of FIG. 7A (Reference 2: IEEJ Transactions D, 109, 11, p.809-816, 1989). As shown in FIG. 7C, a system in which only two of the three phases are switched is called “two-phase switching”. The zero-phase addition methods can be broadly divided into those that become three-phase switching as shown in FIG. 7B and those that become two-phase switching as shown in FIG.
[0056]
The voltage limiter method shown in FIG. 1 can be applied as it is to the zero-phase addition method (three-phase switching method) shown in FIG. Of the three-phase voltage commands, two phases will exceed Vmax and -Vmax at the same time, but there is no problem.
[0057]
In the case of the two-phase switching method as shown in FIG.
[0058]
FIG. 8 shows an embodiment in which a two-phase switching method is considered. Compared with FIG. 1, the processing of the zero-phase adder 2 and the AC voltage limiter 3 is interchanged. In the voltage limit value calculator 4B, Vmax is multiplied by K0.
[0059]
Vu0 to Vw0 are pure sine waves. Therefore, if the limiter process is performed in advance before the zero phase addition, the effect of improving the output voltage by the zero phase addition cannot be obtained. Therefore, in consideration of zero-phase addition, the value of Vmax is set large (K0 times). K0,
[0060]
[Expression 12]

[0061]
Then, a limiter that eliminates the phase shift can be realized by utilizing the effect of the zero-phase addition. Further, the method of FIG. 8 can be applied as it is to the zero-phase addition of FIG.
[0062]
Next, a method for limiting the AC voltage command using the line voltage will be described. In the embodiments so far, the absolute value of each phase voltage is calculated, the largest one is selected, and the coefficient Klim is calculated. However, the same effect can be obtained even if the line voltage is used. FIG. 9 shows the configuration of the voltage limit value calculator 4C. 42 to 46 in the figure are the same as those in FIG. Reference numeral 47 denotes a minimum value selector that selects and outputs the smallest voltage from three inputs. By taking the difference between this output and the output of the maximum value selector 42, the maximum value VLmax of the line voltage can be obtained. Since the condition that the line voltage matches the DC voltage of the inverter is the maximum value of the inverter output voltage, Klim can be obtained by halving VLmax and calculating the ratio with Vmax.
[0063]
The coefficient K0 is set to K0 = 1 when zero-phase addition is performed, and is set to K0 = √3 / 2 when it is not performed. As a result, as in FIGS. 1 and 8, the voltage command can be limited without changing the phase.
[0064]
Next, an embodiment in which the AC voltage command is limited using only the phase information of the AC voltage command will be described. FIG. 10 shows a configuration diagram thereof.
[0065]
In FIG. 10, 10 is a voltage phase calculator for calculating the phase θ of the AC voltage command from Vd0, Vq0 and the phase φ of the dq coordinate axis, and 4D is an output of each phase determined depending only on the value of θ. The voltage limit value calculator 3D that calculates the maximum value and the minimum value is an AC voltage limiter that limits the voltage command based on the maximum value and the minimum value of each phase. The other reference numerals are the same as those in the invention of FIG.
Next, the operation principle of the embodiment shown in FIG. 10 will be described with reference to FIG. In FIG. 11, the voltage phase calculator 10 uses the values of Vd0 and Vq0,
[0066]
[Formula 13]

[0067]
Is calculated, and
[0068]
[Expression 14]
θ = φ + δdq (Expression 14)
As a result, the voltage phase θ is calculated. The voltage limit value calculator 4D outputs an upper limit value (Vumax to Vwmax) and a lower limit value (Vumin to Vwmin) of each phase determined by θ. In the AC voltage limiter, the limit value and the voltage command are compared for each phase, and when the command value exceeds the upper limit value, the limit value is switched to the upper limit value, and when the command value falls below the lower limit value, the lower limit value is switched. In the present embodiment, unlike the embodiment of FIG. 2, processing for limiting all phases to Klim times is not performed, and the value of Vmax (−Vmax) in the conventional example of FIG. This corresponds to changing according to the value of θ.
[0069]
Next, the relationship between these limit values and θ will be described.
[0070]
FIG. 12A shows a waveform after the limiter process when the AC voltage command becomes excessive in some sections in the embodiment of FIG. 1 or FIG. At this time, the zero-phase addition is of the number 10 (FIG. 7B). When the amplitude of the AC voltage command is further increased, the final result is as shown in FIG. Even if the amplitude is further increased, the waveform after the limit remains as shown in FIG.
[0071]
In the case of three-phase alternating current, when the phase θ is determined, the ratio of instantaneous values of the three-phase voltage is uniquely determined. Therefore, even if the voltage command value is limited by the AC voltage limiter, as long as all phases are limited at an equal ratio, the final waveform after the limit is uniquely determined. If this relationship is formulated, the limit value is obtained from the phase.
12, the upper limit value of the U phase is Vmax and the lower limit value of the V phase is −Vmax, and only the intermediate W phase is a complex function.
[0072]
The limit values for the W phase are as follows.
[0073]
In the section of θ1 (30 ° ≦ θ ≦ 90 °), the command value Vw1 is
[0074]
[Expression 15]

[0075]
It becomes. The U phase is
[0076]
[Expression 16]

[0077]
It is. In this period, the U phase and the W phase simultaneously reach Vmax and -Vmax. Representing one side, calculating the coefficient Klim in the U phase,
[0078]
[Expression 17]

[0079]
It becomes. Multiplying this by 15 gives the limit value for the W phase. The upper limit value Vwmax is
[0080]
[Expression 18]

[0081]
The lower limit value Vwmin is
[0082]
[Equation 19]

[0083]
It becomes. Therefore, it is understood that it is a function of only θ without depending on Vp (amplitude) at all.
[0084]
Here, Formula 18 (Formula 19) is generalized,
[0085]
[Expression 20]

[0086]
As shown in FIG. 13, the limit value of each phase in each phase interval is calculated. As can be seen from FIG. 13, when the voltage phase is determined, only one of Vmax and Vmin is defined for each phase, but the remaining limit value is not defined (definition is not necessary).
[0087]
The relationship of FIG. 13 may be stored as a table in the voltage limit value calculator 4D and each limit value may be output for a given θ.
[0088]
In the case of this embodiment, since all the phases are always limited at the same time, the comparator 45 in FIG. 11 can be provided for only one phase, and the output can be common to each phase.
[0089]
As described above, as an embodiment of the present invention, the method for performing the limit process on the AC voltage command has been described. As described above, when the amplitude of the voltage command is increased using the embodiment shown in FIGS. 1 and 8, the waveform shown in FIG. 12B is finally obtained. Therefore, the embodiment of FIG. 1 and FIG. 8 are the same as those of FIG.
[0090]
When the induction motor is driven in the state of the waveform of FIG. 12B, the waveform distortion increases, but the maximum output voltage value of the inverter increases on average. Further, no phase shift occurs at this time. Therefore, the present invention is extremely effective when a large voltage is required temporarily, such as during a transition.
[0091]
When the amplitude of the voltage command is increased in the conventional method (FIG. 4), the value of the intermediate phase (corresponding to the W phase in the section of θ1 in FIG. 12) continues to change as the amplitude increases. The intermediate phase value also sticks to Vmax or -Vmax. Also in this case, the maximum value of the output voltage of the inverter becomes large, but the phase shift reaches ± 30 ° at the maximum. Therefore, an axis deviation occurs and the transient response is deteriorated.
[0092]
Next, another embodiment of the present invention will be described with reference to FIG.
[0093]
In FIG. 14, 1E is a voltage command calculator for calculating a voltage command applied to the induction motor, 3E is a DC voltage limiter for limiting the magnitude of the voltage commands Vd0 and Vq0, and 4E is based on the voltage phase θ. , Vd0, Vq0, a voltage limit value calculator 15E for converting a voltage command on the dq coordinate axis to a voltage command on the αβ axis, and 5E for a voltage on the αβ axis. This is a PWM controller that generates a PWM signal based on a command. Other reference numerals are the same as those in the embodiment shown in FIG. 1 and FIG.
[0094]
This embodiment is characterized in that the voltage command is not coordinate-converted into three-phase alternating current. Instead, a voltage command limiter process is performed on the voltage command on the dq axis. At this time, the voltage limit value is changed by the voltage phase θ. Further, the PWM controller 5E directly creates a PWM signal based on the αβ axis voltage command. Such PWM control is called a space vector method, and expresses the output voltage of the inverter in a biaxial space to calculate the pulse output time. Details will be described later.
[0095]
FIG. 15 shows the configuration of the DC voltage limiter 3E and the voltage limit value calculator 4E. In the figure, 43E is a voltage limit value setter on the dq axis, 48 is a route calculator, and 49 is a function generator that generates a predetermined function for θ. Other parts are the same as those of the same reference numerals in the previous embodiments.
[0096]
Next, the operation principle of the present invention will be described. The value Vdqlimit of the voltage limit value setting unit 43E is
[0097]
[Expression 21]

[0098]
Set as follows. This value corresponds to the magnitude of the vector on the dq coordinate when the maximum value of the line voltage is Vdc · Mmax (that is, Vp = Vdc · Mmax / √3 in Equation 9) and the voltage is dq transformed. . That is, it matches the maximum value for outputting a sine wave.
[0099]
The voltage limit value Vlimit is obtained by multiplying Vdqlimit and Kv. Kv is a function of the voltage phase θ, and this function is shown in FIG. The value of Kv changes with a period of 60 °.
[0100]
On the other hand, Vd0 and Vq0 are added after being squared, and the route calculator 48 outputs the vector magnitude (| V |). Factor limiting the voltage
Klim is obtained from the ratio of | V | and Vlimit.
[0101]
In the process of the DC voltage limiter 3E, similarly to the AC voltage limiter, the voltage command for each of dq is limited by multiplying by Klim when Klim is smaller than 1.
[0102]
Next, the operating principle of the present embodiment will be described. First, the output voltage expression of the inverter on two axes will be described.
[0103]
FIG. 17 is a configuration diagram of a three-phase full-bridge voltage source inverter. In the figure, only one of the upper and lower switching elements of the U to W phases is always turned on. Assuming that the upper side (Sup) is on for the U phase and the lower side (Svn and Swn) is on for the V phase and W phase, the phase voltages Vui to Vwi are:
[0104]
[Expression 22]

[0105]
It becomes. When this is converted to a voltage on the αβ axis according to Equation 8,
[0106]
[Expression 23]

[0107]
It becomes. When Expression 23 is represented as a vector V1 in the αβ space, it is as shown in FIG. Similarly,
[0108]
[Expression 24]

[0109]
Then,
[0110]
[Expression 25]

[0111]
It becomes. When the space vector is calculated in this way, the voltages that can be output from the inverter are eight vectors (including two zero vectors) shown in FIG.
[0112]
To create a PWM signal, a voltage command is also expressed by a space vector. As shown in FIG. 18, it is assumed that the voltage command vector Vr is inside a triangular space surrounded by V0 (zero vector), V1, and V2. In this case, within the minute time T, the vectors V0, V1, and V2 are output for the time t0, t1, and t2, respectively, and Vr is approximately output. The relationship between the voltage command and the output time of each vector is
[0113]
[Equation 26]
Vr = t 1 · V 1 + t 2 · V 2 + t 0 · V 0 (Equation 26)
[0114]
[Expression 27]
T = t1 + t2 + t0 (Equation 27)
From these two formulas, t0 to t2 are obtained. In other words, when PWM control is performed using a space vector, the voltage command is expressed by a space vector, an inverter vector for approximating it is selected, each output time is calculated, and finally converted to a PWM pulse. To do.
[0115]
Next, consider the output voltage range when the inverter is controlled by a space vector. When the voltage command is a sine wave, Vr rotates in a constant speed (angular frequency speed) in a circle. Therefore, if the output voltage of the inverter is limited to a sine wave, the output voltage range of the inverter is inside the circle shown in FIG. If the voltage vector is limited within this circle, the method of the prior art (Japanese Patent Laid-Open No. 9-149699) is obtained. Also in the present embodiment, if the output Kv of the function generator 49 in FIG.
[0116]
In this embodiment, in order to maximize the output voltage of the inverter and achieve high response, Kv is varied to 1 or more according to θ. Unless the output voltage is limited to a sine wave, the output voltage range of the inverter is inside the hexagon shown in FIG. That is, it is possible to output a voltage in a wider range than the output range when limited to a sine wave. In order to use all of the hexagonal area, the value of Kv is variable according to θ.
[0117]
Next, the relationship between θ and Kv will be described with reference to FIG. In the figure, the size of the maximum vector that can be output is assumed to be x. From the relationship of FIG.
[0118]
[Expression 28]
y = x cos (30 ° −θ) (Equation 28)
It becomes. Therefore,
[0119]
[Expression 29]

[0120]
It becomes. Here, y is equal to the maximum value of the sine wave output range, so y = Vdqlimit. Therefore,
[0121]
[30]

[0122]
It becomes. However, the above equation holds in the range of 0 ≦ θ ≦ 60 °. Similarly, the value of Kv can be obtained in other sections of θ. As a result, the function shown in FIG. 16 is obtained.
As described above, according to the embodiment shown in FIG. 14, the command value can be limited on the dq coordinate axis by making maximum use of the inverter output voltage and without changing the vector phase.
[0123]
Next, another embodiment for limiting the voltage command on the dq coordinate axis will be described.
[0124]
The voltage limit value calculator 4F in FIG. 20 is a feature of this embodiment. The voltage limit value calculator 4F is applied in place of the voltage limit value calculator 4E in FIG. 14, and the voltage phase calculator in FIG. 14 is excluded and used. Others are the same as FIG.
[0125]
Next, the operation principle of this embodiment will be described. In FIG. 20, a voltage limit value calculator 4F receives voltage commands Vd0 and Vq0 and temporarily converts them into a three-phase AC voltage with a dq coordinate axis phase φ. Thereafter, as in the embodiment of FIG. 9, the maximum value of the line voltage is calculated, and the coefficient Klim is calculated from 1/2 of the value and the value of Vmax. As described above, when the line voltage and the DC voltage of the inverter match, the maximum output range of the inverter is obtained (corresponding to the hexagon in FIG. 18), so that the output voltage range is utilized to the maximum by this calculation. And the voltage phase does not change. The present invention is equivalent to the invention of FIG. 14, but since the route calculation is unnecessary, the calculation time can be shortened.
[0126]
Next, an embodiment of the forward conversion apparatus according to the present invention will be described with reference to FIG.
[0127]
In FIG. 21, 1G is a voltage command calculator that calculates an input voltage command of a converter 6G (forward converter), 8G is a three-phase AC power supply, and 9G is a phase detector that detects the power supply phase of the three-phase AC power supply. , 50 is a smoothing capacitor for smoothing the output voltage of the converter 6G, 51 is a voltage detector for detecting the voltage of the output voltage Vdc of the converter 6G, 52 is a load device connected to the converter, and 17 is a converter output voltage. This is a voltage controller that calculates the input current command of the converter from the command Vdcr and the detected value Vdc. The other parts are the same as those in the previous embodiments.
[0128]
In the control of the converter, the phase φ of the AC power supply is detected, and the input current is controlled by setting the current component in phase with the power supply voltage as Iq and the component shifted by 90 ° as Id. At this time, by controlling so that Id = 0, the power factor 1 can always be achieved. Further, the DC voltage Vdc is controlled using Iq. In some cases, the current required by the load device 52 is fed forward to Iqr. Also in this embodiment, by adding the voltage limit value calculator 4 and the AC voltage limiter 3, the input voltage of the converter can be utilized to the maximum. Each embodiment related to the control of the inverter described so far can be applied as it is.
[0129]
FIG. 22 shows the configuration of a converter that is currently widely used. Suppose that all PWM pulses, which are output from the PWM controller, are turned off, and the gate signals of all switching elements constituting the converter are turned off. Then, the input current flows through a free wheel diode connected in antiparallel to each switching element, and operates as a diode rectifier.
[0130]
The diode rectifier is very efficient because it does not involve a switching operation like a converter. As the power consumption of the load device increases, the distortion of the input current waveform increases and the power factor also deteriorates. However, there is no problem in a light load state.
[0131]
Therefore, if the diode rectifier and the converter operation are switched according to the power consumption of the load device, a comprehensive conversion system with high efficiency can be realized.
[0132]
However, when the diode rectifier operation is actually switched to the converter operation, an inrush current as shown in FIG. 23A is generated in the conventional converter. This is problematic in that the output voltage of the diode rectifier is lower than the minimum value of the output voltage of the converter and there is a difference of ΔV as shown in the figure. Since the converter having the configuration shown in FIG. 22 is a step-up converter, the relationship between the converter input voltage and the output DC voltage is inversely proportional. That is, in order to minimize the DC voltage, it is necessary to maximize the converter input voltage. However, in the conventional converter, since the input voltage range is within the circle of FIG. 18, it is difficult to obtain the maximum voltage in consideration of the voltage margin.
[0133]
However, in the forward conversion device according to the present invention, the input voltage range of the converter can be expanded to the hexagonal shape of FIG. 18, so that the average value of the input voltage can be increased as shown in FIG. In addition, Vdc can be started up smoothly. Therefore, the inrush current accompanying switching is reduced.
[0134]
In addition, although the inverter and converter in said Example used the thing of a three-phase full bridge, it is applicable similarly also to a 3 level converter and another multiple converter.
[0135]
Further, although an example of an induction motor has been described as an electric motor drive system, the present invention can also be applied to all other AC motors such as a synchronous motor.
[0136]
【The invention's effect】
According to the present invention, the output voltage of the inverter can be utilized to the maximum and the voltage phase error can be controlled to zero.
[0137]
When the present invention is applied to a converter (forward conversion device), the output voltage range of the converter is expanded, and the high efficiency of the device can be realized as a forward conversion system.
[Brief description of the drawings]
FIG. 1 is a configuration diagram of an AC motor driving apparatus according to an embodiment of the present invention.
FIG. 2 is a configuration diagram of an AC voltage limiter and a voltage limit value calculator.
FIG. 3 is a diagram showing a relationship between an AC voltage command and a voltage limit value.
FIG. 4 is a configuration diagram of a conventional voltage limiter.
FIG. 5 is a diagram showing the effects of the conventional method and the method according to the present invention.
FIG. 6 is a configuration diagram of a zero-phase adder.
FIG. 7 is a diagram showing the principle of zero-phase addition.
FIG. 8 is a configuration diagram of an AC motor driving apparatus according to an embodiment of the present invention.
FIG. 9 is a configuration diagram of a voltage limit value calculator.
FIG. 10 is a configuration diagram of an AC motor drive device according to an embodiment of the present invention.
FIG. 11 is a configuration diagram of a voltage phase calculator, a voltage limit value calculator, and an AC voltage limiter.
12 is a diagram for explaining the operation principle of the embodiment of FIG. 10;
FIG. 13 is a diagram illustrating a voltage limit value with respect to a phase θ.
FIG. 14 is a configuration diagram of an AC motor driving device according to an embodiment of the present invention.
FIG. 15 is a configuration diagram of a voltage limit value calculator and a DC voltage limiter.
FIG. 16 is a diagram representing functions used in the embodiment of FIG. 14;
FIG. 17 is a main circuit configuration diagram of a three-phase full-bridge voltage source inverter.
FIG. 18 is a diagram showing an output voltage of an inverter and a space vector display of a voltage command.
19 is a diagram showing the principle of the embodiment of FIG.
FIG. 20 is a configuration diagram of a DC voltage limiter and a voltage limit value calculator.
FIG. 21 is a configuration diagram of a forward conversion apparatus according to an embodiment of the present invention.
FIG. 22 is a main circuit configuration diagram of a three-phase full-bridge voltage source converter.
FIG. 23 is a diagram showing the operation of the converter according to the conventional method and the present invention.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 ... AC voltage command calculator, 2 ... Zero phase adder, 3 ... AC voltage limiter, 4 ... Voltage limit value calculator, 5 ... PWM controller, 6 ... Inverter, 7 ... Current detector, 8 ... Induction motor, DESCRIPTION OF SYMBOLS 9 ... Speed detector, 11 ... Speed controller, 12 ... Integrator, 13 ... dq coordinate converter, 14 ... Current controller.

Claims (10)

An AC voltage command generator for generating a voltage command on the AC side of the power converter, an AC voltage limiter for limiting the AC voltage command value, and generating a PWM pulse based on the output of the AC voltage limiter, the power conversion A PWM controller for controlling the device,
When at least one phase voltage or line voltage of the AC voltage command exceeds a predetermined value set in advance, the AC voltage command of all phases is limited at the same ratio, and the output value of the AC voltage limiter The control apparatus of the power converter characterized by changing this. In claim 1,
Means for calculating or detecting the voltage phase of the AC voltage command is defined by the voltage phase when at least one phase voltage or line voltage of the AC voltage command exceeds a predetermined value. A control device for a power converter, wherein an output value of the AC voltage limiter is changed based on a function to be performed. A dq voltage command generator that generates a voltage command on the AC side of the power converter on a rotating coordinate axis (dq coordinate) that rotates in synchronization with a three-phase AC, and dq that limits the value of the voltage command on the dq coordinate axis. A voltage limiter, and a PWM controller that generates a PWM pulse based on the output of the dq voltage limiter and controls the power converter,
When the voltage command is converted into an AC voltage command (dq reverse conversion), and at least one phase voltage or line voltage of the AC voltage command exceeds a preset predetermined value, An apparatus for controlling a power converter, wherein the output value of the dq voltage limiter is changed by limiting two voltage command values at the same ratio. In claim 3,
A means for calculating or detecting a voltage phase on the AC side based on the voltage command on the dq coordinate axis and the phase of the dq coordinate axis, and calculating a vector sum of the voltage command on the dq axis, A control device for a power converter, wherein an output value of the dq voltage limiter is changed based on a function defined by the voltage phase when a predetermined value set in advance is exceeded. In claim 4,
The function in the voltage limiter is a function defined by the following formula for a section of voltage phase θ = 0 to 60 °, and the other sections are functions that repeat the following formula every 60 °. Control device for power converter.

A power converter that converts direct current to three-phase alternating current, an AC motor driven by the power converter, means for detecting or estimating the rotational speed of the AC motor, a speed command, and a detected value of the speed or Based on the estimated value, an AC voltage command generator for generating a three-phase AC voltage command for the AC motor, an AC voltage limiter for limiting the value of the AC voltage command, and a voltage command limited by the AC voltage limiter. A PWM controller for generating a PWM pulse on the basis of and controlling the power converter,
When at least one phase voltage or line voltage of the AC voltage command exceeds a preset predetermined value, the AC voltage command of all phases is limited at the same ratio, and the output value of the AC voltage limiter is set. AC motor system characterized by changing. On a rotating coordinate axis (dq coordinate axis), a power converter that converts direct current into three-phase alternating current, an AC motor driven by the power converter, a means for detecting or estimating the rotational speed of the AC motor, A dq voltage command generator that generates a voltage command on the AC side of the AC motor based on a speed command, a detected value or an estimated value of the speed, a dq voltage limiter that limits the value of the voltage command, and the dq A PWM controller that generates a PWM pulse based on a voltage command limited by a voltage limiter and controls the power converter, and
When the voltage command is converted into an AC voltage command (dq reverse conversion), and at least one phase voltage or line voltage of the AC voltage command exceeds a preset predetermined value, An AC motor system characterized in that the output value of the voltage limiter is changed by limiting two voltage command values at the same ratio. A power converter that converts three-phase AC to DC, a voltage detector that detects a DC voltage of the power converter, a DC voltage command, and a voltage command that calculates a three-phase AC voltage command based on the DC voltage An arithmetic unit, an AC voltage limiter that limits the value of the AC voltage command, and a PWM controller that generates a PWM pulse based on the voltage command limited by the AC voltage limiter and controls the power converter, Prepared,
When at least one phase voltage or line voltage of the AC voltage command exceeds a predetermined value set in advance, the AC voltage command of all phases is limited at the same ratio, and the output value of the AC voltage limiter A forward conversion device characterized by changing the value. On the basis of the DC voltage command and the DC voltage on the power converter that converts three-phase AC to DC, the voltage detector that detects the DC voltage of the power converter, and the rotation coordinate axis (dq coordinate axis), A voltage command generator for generating a voltage command on the AC side of the power converter, a dq voltage limiter for limiting the value of the voltage command, and generating a PWM pulse based on the voltage command limited by the dq voltage limiter; A PWM controller for controlling the power converter,
When the voltage command is converted into an AC voltage command (dq reverse conversion), and at least one phase voltage or line voltage of the AC voltage command exceeds a preset predetermined value, A forward conversion device characterized in that the output value of the dq voltage limiter is changed by limiting two voltage command values at the same ratio.   10. The PWM pulse output from the PWM controller is turned off when the power consumption of the load device is less than a predetermined value or when the input voltage required by the load device is less than a predetermined value. And a forward converter that drives the power converter as a diode rectifier.

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