JP3259805B2 - Control device for synchronous motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for a synchronous motor, and more particularly to a control device capable of controlling a permanent magnet type synchronous motor having a distorted induced voltage waveform at a high power factor without torque pulsation.

[0002]

2. Description of the Related Art In general, when the torque or speed of a synchronous motor is controlled by a power converter, it is desirable that the induced voltage waveform and the armature current waveform of the synchronous motor are sine waves in order to reduce torque pulsation. Have been. However, in order to make the induced voltage waveform a sine wave strictly, the design of the motor is restricted, and it is difficult to manufacture the motor, resulting in an increase in cost and a reduction in the output of the motor.

In order to cope with such a problem, the applicant has previously filed Japanese Patent Application No. 4-246823 (filed on Sep. 16, 1992), and referred to "Permanent Magnet Motor Having Distorted Induced Electromotive Force". Osawa et al.
IEEJ Transactions on Electronics, Vol. 113, No. 10, 1993, p.
1200 to 1208 have been announced (hereinafter collectively referred to as a conventional example).

FIG. 16 is a configuration diagram showing a conventional example. That is, in the conventional example, the reference induced voltage waveform of the motor is stored in the memory 31.
3n, and read out the contents of this memory in accordance with the position θ of the magnetic pole to obtain instantaneous values e _{01 to} e _on of the reference induced voltage corresponding to each phase. Then, in the current command value (target value) calculator 4, the instantaneous values e _{01 to} e _{on of} the reference induced voltages read from the memories 31 to 3n and the torque target value T ^* are used to calculate each phase based on the following equation. Current target value i
Seek ₁ ^* _{~i n} ^*. For example, the k-th target current value i _k ^{* _{is, i k * = T * e}} 0k / given by ^{_{(e 01 2 + e 02 2}} + ... + e on 2), the instantaneous value of each phase of the reference induced voltage The amount is inversely proportional to the sum of squares and proportional to the instantaneous value of the torque target value and the k-th reference induced voltage.

[0005] 51~5n is a current regulator (ACR), the current target value i ₁ ^* ~i _n ^* and the current detection value i ₁ through i _n
And amplify the deviation. The outputs of the ACRs 51 to 5n are subjected to pulse width modulation (PWM), and gate pulses are generated from pulse generators (PG) 61 'to 6n' for each of the single-phase inverters 11 to 1n, so that on / off control is performed. It was made. As a result, the synchronous motor having a distorted induction voltage has a small torque pulsation and a maximum output with a predetermined effective current value.

[0006]

However, in the conventional example described above, the utilization rate of the motor is improved to the maximum, but as a result of examination, as the load increases, the terminal voltage peak value of the motor increases. It has been found that there remains a problem that the required converter capacity increases. Therefore, an object of the present invention is to provide a synchronous motor in which an induced voltage is distorted,
In particular, it is an object of the present invention to improve the converter utilization without causing torque pulsation in the permanent magnet synchronous motor and without significantly reducing the utilization of the motor.

[0007]

In order to solve this problem, according to the present invention, in a control device for a multi-phase (n-phase) synchronous motor supplied with electric power via a power converter, an induction voltage of each phase is provided. the e _1, e _2, ... and e _n, at least a function whose phase is shifted by φ of the fundamental wave component e ₁ to these induced voltages when ', e _2', ... and e _n ', any k-th of the armature current target value ^{_{i k *, i k * =}} k · e k '/ (e 1 e 1' + e 2 e 2 '+ ... e
_n e _n ') (K: is a calculating means for calculating a coefficient) becomes a function, characterized by comprising a current control means for controlling the armature current based on the calculation result.

[0008] In the present invention of claim 1, the induced voltage e ₁ of the phase, e _2, ... a first storage means for storing e _n, at least the fundamental wave component phase relative to the phase of the induced voltage There φ shifted function e ₁ ', e _2', ... is provided and a second storage means for storing e _n ', read in accordance with these first, the contents of the second storage means to a common magnetic pole position signal Te can obtain a target value of the armature current (the invention of claim 2), or, the induced voltage e _1, e ₂ of the respective phases, provided storage means for storing ... e _n, predetermined in the storage means While the magnetic pole position signal is given to read out the induced voltage of each phase, a function in which at least the phase of the fundamental wave component is shifted by φ with respect to the induced voltage of each phase, a phase φ is added to the magnetic pole position signal. The target value of the armature current can be obtained by reading the value (the invention of claim 3).

[0009] In the invention of the first to third aspects, the phase φ
Can be related to a torque command, and φ can be increased with an increase in the torque command (the invention of claim 4), or the phase φ can be related to the rotation speed of the electric motor, and φ = 0 at medium and low speeds, At high speed, φ ≠ 0 can be established (the invention of claim 5), or the phase φ is associated with the torque command and the rotation speed of the motor. Φ can be increased as the speed increases (the invention of claim 6). In the invention of claims 1 to 6, the e ₁ ′, e ₂ ′,
The ... e _n ', can be a function that does not include the zero-phase component (the invention of claim 7), and in the invention of claim 7, the function can be a sine wave (of claims 8 invention).

[0010]

[Function] The induced voltages of the n-phase synchronous motor are represented by e ₁ , e ₂ ,.
e _n , the phase of the fundamental wave component is φ
Shifted by function _{e 1 ', e 2',} ..., when the e _n ', the target value i _k for any k-th of the armature current, the K as proportionality factor, i _k ^* = K · e _{k '/ (e 1 e 1} ' + e 2 e 2 '+ ... + e n e n') ... (1) consisting calculated as a function, it generates a torque pulsation by controlling the armature current based on the calculation result Without letting
In addition, the increase in the motor terminal voltage due to the increase in the load is suppressed to reduce the necessary converter capacity, thereby enabling high power factor and high efficiency control.

Hereinafter, the principle of the present invention will be described while sequentially showing the derivation process of equation (1). Now, the n-phase permanent magnet synchronous motor, the instantaneous value e _1, e ₂ of the induced voltage of each phase by the permanent magnets, ..., a vector of e _n as in the following equation (2), e (→) = _{(e 1, e 2, ...} , e n) and (2). Note that a vector amount is represented by adding a symbol (→), and so on. Further, the instantaneous value i _1, i ₂ of each phase of the armature current, ..., a vector of _{i n, i (→) =} (i 1, i 2, ..., i n) and (3). At this time, the output of the electric motor becomes _{P = e 1 i 1 + e} 2 i 2 + ... + e n i n ... (4).

As described above, the output P is given by the inner product of the vectors e (→) and i (→), and is expressed by the following equation. P = e (→) · i (→) = | e (→) || i (→) | cosφ 0 ... (5) here, | e (→) | = (e 1 2 + e 2 2 + ... + e _{^{n 2) 1/2 ... (5-1)}} | i (→) | = (i 1 2 + i 2 2 + ... + i n 2) is ^1/2 ... (5-2), ^φ ₀ is e (→ ) And i (→).

[0013] In the conventional example described above, to minimize the resistance loss for a given output, to maximize output for a given current effective value other _{words, cosφ 0 = 1 (φ 0} =
0), that is, a control method in which i (→) = ae (→) (6) holds. However, this method reduces the power converter utilization, as described above.

Therefore, in the present invention, i (→) = ae ′ (→) (7) where e (→) and e ′ (→) are such that the power factor angle of the fundamental wave is φ
Assume that there is only a phase difference. Substituting equation (7) into equation (5) to obtain a, a = P / e (→) · e ′ (→) (8) Substituting equation (8) into equation (7) and replacing P with its target value P ^* , | i (→) | = P ^* e ′ (→) / e (→) · e ′ (→) (9) This equation (9) represents a vector in which the output is constant when P ^* is constant, that is, a vector in which torque pulsation does not occur. The above equation (1) is a generalization of the equation (9) by expressing each component of the vector.

Next, a modification of equation (1) or (9) will be attempted. Since the vector e (→) of the induced voltage is proportional to the rotational angular velocity ω, the vector e as the reference of the induced voltage given by e (→) = ωe ₀ (→) (10)
Consider ₀ (→). Similarly, the vector e ′ (→) is set to e ′ (→) = ωe ₀ ′ (→) (11). In addition, since the output P can be expressed by the product of the ω and the torque T, as (9), the target value of the torque ^{T *, | i (→)} | = T * e 0 '(→) / e 0 (→) · e ₀ ′ (→) (12)

When the equation (12) is represented by each component of the vector, the following equation is obtained. _{^{i k * = T * · e}} 0k '/ (e 01 e 01' + e 02 e 02 '+ ... e 0n e 0n') ... (13)

[0017]

FIG. 1 is a block diagram showing one embodiment of the present invention, and FIG. 2 is a block diagram of a main circuit of a power converter to which the present invention is applied. As shown in FIG. 2, the power converter 1 has 11 to 11 here.
The synchronous motor 2 is a permanent magnet type n-phase synchronous motor having n sets of independent windings 21 to 2n. 2 ′ is a magnetic pole position sensor for detecting the position of the magnetic pole.

Referring to FIG. E described earlier₀
Each component e of (→)₀₁, E₀₂… E_0nIs a function of only the magnetic pole position
Therefore, this is stored in the memory 3A of FIG.
The contents of this memory 3A according to the position θ of
Gives e₀₁~ E_onCan be obtained. e₀’(→)
Is the same, and e depends on θ.₀’(→)
Minute e₀₁’, E₀₂’… E _0n'From the memory 3B.
Numeral 4 denotes a calculator for calculating a current command value (target value).
The signals read from the rims 3A and 3B and the torque target value T
^*And the current target value i for each phase based on equation (13) above.
₁ ^*~ I_n ^*Ask for.

[0019] 51~5n is a current regulator (ACR), the current target value i ₁ ^* ~i _n ^* and the current detection value i ₁ through i _n
And amplify the deviation. 61 to 6n are, for example, ACR51 to
This is a drive circuit that performs pulse width modulation (PWM) on the output of 5n and supplies a base current to each of the single-phase inverters 11 to 1n.

FIG. 3 shows a modification of FIG. In FIG. 3, the memory 3 is shared, and an adder (AD
D) 7 to provided, e ₀₁ to e _on the corresponding to the magnetic pole position θ and phi = 0 reads the contents of the memory 3, e ₀₁ '~e _0n' is obtained by adding the phi to the magnetic pole position θ in the adder 7 Value (θ + φ)
To read the contents of the memory 3. That is, in the same shape the waveform of the e ₀₁ to e _on the e ₀₁ In the example 'to e _0n', is a function whose phase is shifted by phi.

Here, the effect of the present invention will be described with reference to FIGS.
This will be described with reference to FIG. For example, as shown in FIG. 4, the windings of the permanent magnet synchronous motor are divided into two sets of three coils having a phase difference of 30 degrees from each other.
Assuming that consists phase winding, each phase of the induced voltage waveform of this case, namely the waveform of e ₀₁ to e _on is as shown in FIG. 5, for example. The waveforms of e ₀₁ ′ to e _0n ′ are e
_{It is} assumed that the waveforms have the same shape as the waveforms of _{01 to} e _on , for example, advance by 10 degrees (φ = 10 °).

FIG. 6 shows the calculation results of the current, voltage and torque waveforms according to the present invention in such a case. As can be seen from the figure, the terminal voltage peak value of the motor is about 1.3 times the induced voltage peak value, the torque is constant and there is no pulsation. On the other hand, in the case of the conventional example,
That is, when φ = 0, as shown in FIG.
Is the same as However, the peak value of the terminal voltage is about 1.7 times the peak value of the induced voltage, which is 30 times that of the present invention.
%To increase. Further, the effective current value is reduced by 2% as compared with the case of the present invention, and the ratio is minute. As described above, according to the present invention, for a predetermined output or torque, the effective current value is slightly increased as compared with the conventional control method, but the peak value of the terminal voltage is greatly reduced. Can be reduced.

The terminal voltage is mainly given by the sum of the induced voltage by the permanent magnet and the reactance voltage drop by the synchronous reactance of the synchronous motor. Since the reactance voltage drop is proportional to the armature current, the terminal voltage increases as the armature current increases. That is, the terminal voltage at light load is almost equal to the induced voltage, and the terminal voltage peak value is small. FIG.
Is an example in consideration of this point.

That is, a function generator 8A for obtaining a phase output φ corresponding to a torque target (command) value T ^* is provided, and a switch 9
Switch between "0" and "φ" by the adder (ADD)
7, the content of the memory 3 is read out by adding to the magnetic pole position signal θ. Thus, at light loads with small terminal voltage peak values, φ is reduced to improve the efficiency of the synchronous motor, and at heavy loads with large terminal voltage peak values, φ is increased to increase the terminal voltage peak values. It is possible to suppress and improve the utilization rate of the power converter.

Since the magnitude of the induced voltage is proportional to the rotation speed N of the motor, the peak value of the terminal voltage does not become large at low and medium speeds even if the load is heavy.
Therefore, as shown in FIG. 9, a function generator 8B for providing the phase output φ as a function of the rotation speed N is provided, and φ = 0 at medium and low speeds, and φ ≠ 0 at high speed. Function generator from 8A to 8
8 is the same as that in FIG.

FIG. 10 is a combination of FIG. 8 and FIG. 9 and shows a function generator 8A that uses the phase output φ as a function of the torque target value T ^* and a function generator that uses the phase output φ as a function of the rotation speed N of the motor. A generator 8B and a multiplier 1 for multiplying both outputs
0 at the time of medium and low speeds, φ = 0, and the torque command T
^* , Φ is increased with an increase in the rotation speed N of the electric motor.

By the way, a three-phase inverter generally applied widely is as shown in FIG.
The sum of the three-phase currents must be zero, that is, the zero-phase current must be zero. However, the current waveform shown in FIG. 6 includes harmonics that are multiples of three. Since the harmonic of a multiple of 3 is a zero-phase component, the current obtained by combining the three-phase currents is not zero.

Therefore, in the present invention, the components e ₀₁ ′, e ₀₂ ′,... E _0n ′ whose phases are shifted by φ with respect to the induced voltage do not include the zero-phase component. That is, the following relationship is provided. The configuration of the control device is shown in FIGS.
10 may be used. e ₀₁ '+ e ₀₂ ' + e ₀₃ '= 0 (14) FIG. 12 shows a waveform example of each component in this case.
In this case, the current target value obtained by Expression (13) is as follows: i ₁ ^* + i ₂ ^* + i ₃ ^* = 0 (15) That is, the zero-phase current is zero, and a target current can be supplied to the synchronous motor via the three-phase inverter. FIG. 13 shows an example of the current waveform in that case.

In FIG. 13, the current peak value with respect to the current effective value is large. When the current peak value is large, it may be necessary to reduce the allowable effective current value due to the maximum current constraint of the self-extinguishing device constituting the inverter. In order to cope with such a problem, in the present invention, the components e ₀₁ ′, e ₀₂ ′,.
_{Let 0n} 'be a sine wave. FIG. 14 shows an example of a target current waveform in this case. That is, the current waveform becomes a sinusoidal waveform containing some harmonic components, and the peak value can be reduced as compared with the case of FIG.

FIG. 15 shows an example of a configuration in which two sets of three-phase inverters are used.
14 to 16 constitute the other three-phase inverter. In this _{^{case, i 1 * + i 2 *}} + i 3 * = 0 i 4 * + i 5 * + i 6 * = 0 ... it is necessary is (16). At this time, e ₀₁ ′ to e
₀₆ ′ is set so that _e01 ′ + _e02 ′ + _e03 ′ = 0 _e04 ′ + _e05 ′ + _e06 ′ = 0 (17).

[0031]

According to the present invention, each phase current of the motor is controlled by the function of the above equation (1) according to the waveform of the induced voltage, so that there is no torque pulsation and the load can be increased. As a result, it is possible to suppress an increase in the terminal voltage of the electric motor, to reduce the required capacity of the converter, and to obtain an advantage that a high power factor and highly efficient control can be performed.

[Brief description of the drawings]

FIG. 1 is a configuration diagram showing a first embodiment of the present invention.

FIG. 2 is a schematic diagram for explaining a relationship between a power converter to which the present invention is applied and a synchronous motor.

FIG. 3 is a configuration diagram showing a modification of FIG. 1;

FIG. 4 is an explanatory diagram for explaining a winding example of a permanent magnet type synchronous motor.

FIG. 5 is a waveform diagram showing an example of a waveform of each phase induced voltage in the case of FIG. 4;

FIG. 6 is a waveform diagram showing a current waveform and an output waveform of each phase when control according to the present invention is performed.

FIG. 7 is a waveform diagram showing a current waveform and an output waveform of each phase according to a conventional control method when control according to the present invention is not performed.

FIG. 8 is a main part configuration diagram showing a second embodiment of the present invention.

FIG. 9 is a main part configuration diagram showing a third embodiment of the present invention.

FIG. 10 is a main part configuration diagram showing a fourth embodiment of the present invention.

FIG. 11 is a circuit diagram showing a general three-phase inverter.

12 is a waveform diagram showing an example of a voltage waveform having a constant phase difference with respect to an induced voltage in the case of FIG.

13 is a waveform chart showing an example of a current waveform to be supplied to the three-phase inverter of FIG.

FIG. 14 shows a case where the voltage waveform example of FIG. 12 is a sine wave;
FIG. 6 is a waveform diagram showing an example of a current waveform to be supplied to a phase inverter.

FIG. 15 is a circuit diagram showing two sets of three-phase inverters.

FIG. 16 is a configuration diagram showing a conventional example.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... Power converter, 11-16 ... Single phase inverter, 2 ... Synchronous motor (synchronous machine), 21-2n ... Synchronous motor winding,
2 ′: magnetic pole position sensor, 3A, 3B, 31 to 3n: memory, 4: current command calculator, 51 to 5n: current controller (A
CR), 61 to 6n: Drive circuit (PWM), 7: Adder (ADD), 8A, 8B: Function generator, 9: Switcher, 10: Multiplier.

Continuation of the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02P 5/408-5/412 H02P 7/628-7/632 H02P 21/00

Claims (8)

(57) [Claims] 1. A multi-phase (n) power supply via a power converter
Phase) The control apparatus for a synchronous motor, e _1, e ₂ of each phase of the induced voltage, ... and e _n, e ₁ function whose phase is shifted by φ of at least the fundamental wave component with respect to those induced voltage ', e ₂ ', ... e _n' when the, target value i _k ^* for any k-th of the armature ^{_{current, i k * = K · e}} k '/ (e 1 e 1' + e 2 e 2 '+ … E
_n e _n ') (K: coefficient) and calculating means for calculating as a function that is, the control apparatus for a synchronous motor, characterized by comprising a current control means for controlling the armature current based on the calculation result. 2. The induced voltages e ₁ , e _{2 ,.}
And a function e in which the phase of at least the fundamental wave component is shifted by φ with respect to the induced voltage of each phase.
₁ ', e _2', ... is provided and a second storage means for storing e _n ', these first, the target value of the armature current is read in accordance with the common pole position signal the contents of the second storage means The control device for a synchronous motor according to claim 1, wherein 3. The induced voltages e ₁ , e _{2 ,.}
Is provided, and a predetermined magnetic pole position signal is given to the storage means to read out the induced voltage of each phase.
A function in which the phase of at least the fundamental wave component is shifted by φ with respect to the induced voltage of each phase is read out as a value obtained by adding the phase φ to the magnetic pole position signal to obtain a target value of the armature current. The control device for a synchronous motor according to claim 1. 4. The synchronous motor control device according to claim 1, wherein the phase φ is associated with a torque command, and φ is increased with an increase in the torque command. 5. The synchronous motor according to claim 1, wherein the phase φ is related to a rotation speed of the motor, and φ = 0 at medium and low speeds, and φ ≠ 0 at high speeds. Control device. 6. The phase φ is associated with a torque command and a rotation speed of a motor, φ = 0 at medium and low speeds, and φ is increased with an increase in the torque command and the rotation speed of the motor. A control device for a synchronous motor according to any one of claims 1 to 3. Wherein said _{e 1 ', e 2',} ... and e _n ', the control system for a synchronous motor according to any one of claims 1 to 6, characterized in that a function without the zero-phase component . 8. The control device according to claim 7, wherein the function is a sine wave.