JP6667425B2 - Motor control device and electric vehicle - Google Patents

Description

  The present invention relates to a motor control device and an electric vehicle.

  Hybrid vehicles and electric vehicles are required to improve motor output from the viewpoint of improving driving force. For example, in the invention described in Patent Literature 1, a pseudo-rectangular wave current is used as a drive current of a motor to increase an effective current value and improve output.

JP 2006-136144 A

  However, in the method described in Patent Document 1, since the drive current of the motor is always a quasi-rectangular wave, the torque ripple may be increased.

According to one aspect of the present invention, a motor control device controls a drive current of a motor having windings independently wound between phases based on a torque command value, an angular velocity of the motor, and a rotor position. A control device, wherein a ratio of a zero-phase current in the drive current is changed according to the magnitude of the torque command value, and a voltage induced in the winding based on the drive current and the rotor position. of calculating a zero-phase voltage, that controls the zero-phase current obtained by adding the zero-phase voltage to the zero-phase voltage command value based on the angular velocity and position of the rotor of the said torque command value motor.

  According to the present invention, it is possible to improve output and reduce torque ripple.

FIG. 1 is a diagram showing a schematic configuration of a hybrid electric vehicle equipped with a motor according to an embodiment of the present invention. FIG. 2 is a block diagram illustrating a configuration of the power conversion device. FIG. 3 is a block diagram showing details of the motor control device. FIG. 4 is a flowchart illustrating an example of the current command calculation process. FIG. 5 is a diagram illustrating the drive current. FIG. 6 is a block diagram illustrating details of the zero-phase back electromotive force compensator. FIG. 7 is a flowchart illustrating the first modification. FIG. 8 is a diagram illustrating a drive current, a sine wave component of the drive current, and a zero-phase current according to the first modification. FIG. 9 is a block diagram illustrating a motor control device according to a second modification.

  Hereinafter, embodiments for carrying out the present invention will be described with reference to the drawings. The present invention is not limited to the following embodiments, but includes various modifications and application examples within the technical concept of the present invention.

  FIG. 1 is a diagram showing a schematic configuration of a hybrid electric vehicle equipped with a motor according to an embodiment of the present invention. The vehicle 100 has an engine 120, a motor 200, and a battery 180 mounted thereon. Battery 180 supplies DC power to motor 200 via power conversion device 600 when driving power from motor 200 is required, and receives DC power from motor 200 during regenerative running. Transfer of DC power between the battery 180 and the motor 200 is performed via the power converter 600. Although not shown, the vehicle 100 is equipped with a battery that supplies low-voltage power (for example, 14-volt power), and is used, for example, as a power supply for a control system.

  Rotational torque from engine 120 and motor 200 is transmitted to front wheels 110 via transmission 130 and differential gear 160. The transmission 130 is controlled by a transmission control device 134. Engine 120 is controlled by engine control device 124. Battery 180 is controlled by battery control device 184. The transmission control device 134, the engine control device 124, the battery control device 184, the power conversion device 600, and the integrated control device 170 are connected by a communication line 174.

  The integrated control device 170 is a higher-level control device than the transmission control device 134, the engine control device 124, the power conversion device 600, and the battery control device 184. The integrated control device 170 receives information representing each state of the transmission control device 134, the engine control device 124, the power conversion device 600, and the battery control device 184 from the respective devices via the communication line 174. The integrated control device 170 calculates a control command for each control device based on the acquired information. The calculated control command is transmitted to each control device via the communication line 174.

  The high-voltage battery 180 is constituted by a secondary battery such as a lithium ion battery or a nickel hydride battery, and outputs high-voltage DC power of 250 volts to 600 volts or more. The battery control device 184 outputs the charge / discharge status of the battery 180 and the status of each unit cell battery constituting the battery 180 to the integrated control device 170 via the communication line 174.

  When the integrated control device 170 determines that the battery 180 needs to be charged based on the information from the battery control device 184, the integrated control device 170 issues a power generation operation instruction to the power conversion device 600. Further, the integrated control device 170 mainly manages the output torque of the engine 120 and the motor 200, calculates the total torque of the output torque of the engine 120 and the output torque of the motor 200, and calculates the torque distribution ratio. A control command based on the result is transmitted to transmission control device 134, engine control device 124, and power conversion device 600. The power conversion device 600 controls the motor 200 based on the torque command from the integrated control device 170 so as to generate a torque output or a generated power according to the command.

  The power conversion device 600 includes an inverter for driving the motor 200 and a motor control device for generating a switching signal to the inverter, as described later. The power conversion device 600 controls the inverter based on a command from the integrated control device 170 to operate the motor 200 as a motor or a generator.

  FIG. 2 is a block diagram showing a configuration of the power conversion device 600. The power conversion device 600 includes a motor control device 610, an inverter 620, and a current sensor 220. The motor 200 is constituted by an embedded magnet synchronous motor to which no neutral point is connected. The motor 200 is provided with a position sensor 210 that detects the position of the rotor and outputs the detected rotor position θ. The current sensor 220 detects a current flowing through the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 wound on the stator of the motor 200, and outputs the detected three-phase currents iu, iv, iw. I do.

  The inverter 620 is provided with a U-phase full-bridge inverter 621, a V-phase full-bridge inverter 622, and a W-phase full-bridge inverter 623, which are connected in parallel to a battery 180 as a DC power supply. The U-phase winding 201 of the motor 200 is connected to the output terminal of the U-phase full bridge inverter 621, the V-phase winding 202 is connected to the output terminal of the V-phase full bridge inverter 622, and the W-phase winding 203 is connected to the W-phase full bridge. Connected to the output terminal of bridge inverter 623. The motor 200 has no neutral point connected, and can independently control the current flowing through the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203.

  The U-phase full-bridge inverter 621 includes switching elements 621a to 621d. Switching element 621a is arranged on the upper arm of the U-phase left leg. Switching element 621b is arranged on the lower arm of the U-phase left leg. Switching element 621c is arranged on the upper arm of the U-phase right leg. Switching element 621d is arranged on the lower arm of the U-phase right leg.

  V-phase full-bridge inverter 622 includes switching elements 622a to 622d. Switching element 622a is arranged on the V-phase left leg upper arm. Switching element 622b is arranged on the lower arm of the V-phase left leg. Switching element 622c is arranged on the V-phase right leg upper arm. Switching element 622d is arranged on the lower arm of the V-phase right leg.

  The W-phase full-bridge inverter 623 includes switching elements 623a to 623d. Switching element 623a is arranged on the upper arm of the W-phase left leg. Switching element 623b is arranged on the lower arm of the W-phase left leg. Switching element 623c is arranged on the upper arm of the W-phase right leg. Switching element 623d is arranged on the lower arm of the W-phase right leg.

  The switching elements 621a to 621d, the switching elements 622a to 622d, and the switching elements 623a to 623d are configured by combining a diode with a metal oxide film type field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), or the like. In the present embodiment, a configuration using a MOSFET and a diode will be described.

By turning on or off the switching elements 621a to 621d, the switching elements 622a to 622d, and the switching elements 623a to 623d based on the switching signal generated by the motor control device 610, the inverter 620 is applied from the battery 180. The DC voltage is converted to an AC voltage. The converted AC voltage is applied to three-phase windings 201 to 203 wound around the stator of motor 200, and generates a three-phase AC current. The three-phase alternating current generates a rotating magnetic field in the motor 200, and the rotor of the motor 200 rotates. Motor control device 610 is based on torque command value T ^* from integrated control device 170, three-phase current i _u , i _v , i _w detected by current sensor 220, and rotor position θ detected by position sensor 210. PWM control of the inverter 620.

  FIG. 3 is a block diagram showing details of the motor control device 610. The motor control device 610 includes a current command calculation unit 10, a dq axis current control unit 20, a switching signal generation unit 30, a dq conversion unit 40, a zero-phase current calculation unit 50, a zero-phase current control unit 60, a speed conversion unit 70, A phase back electromotive voltage compensator 80 is provided.

dq converter 40, a three-phase current _i u is detected by the current sensor _220, i v, _{i w} and, on the basis of the rotor position θ detected by the position sensor 210, dq axis current detection value _i d, Output _iq . The speed converter 70 outputs the angular velocity ω of the rotor based on the rotor position θ detected by the position sensor 210. Zero-phase current calculation section 50, the three-phase currents _i u _entered, i v, calculates a zero-phase current _{i z} based on _{i w.} The zero-phase current _iz is calculated as in the following equation (1).
_{i z = i u / √3 +} i v / √3 + i w / √3 ... (1)

Current calculation unit 10 includes an input torque command value T *, the angular velocity ω and, based on the rotor position theta, dq axis current command value i d *, i q * and zero-phase current command value i z * Is calculated. The present embodiment is characterized by the calculation processing in the current command calculation unit 10, and the detailed processing will be described later.

dq-axis current control unit 20, dq-axis current command value is input from the current command calculating unit 10 _i ^d _*, and _i ^{q *,} dq axis current detection value input from the dq conversion section 40 _i d, the _{i q} Then, the dq-axis voltage command values v _d ^* and v _q ^* are output using proportional control, integral control, or the like. Zero-phase current controller 60, the zero-phase current command value i _z ^* input from the current command calculating section 10, based on the zero-phase current i _z calculated in the zero-phase current calculator 50, the proportional control and integral The zero-phase voltage command value _vz ^* is output using control or the like.

The zero-phase voltage command value v _z ^* output from the zero-phase current control unit 60 is added to the zero-phase counter electromotive voltage compensation value v _z ^** output from the zero-phase back electromotive voltage compensation unit 80, and the signal (v _z ^* + _vz ^** ) are input to the switching signal generator 30. Zero-phase back electromotive force compensation value v _z ^** are intended to reduce the deviation between the zero-phase current i _z detected with the zero-phase current command value i _z ^*, cancel the zero-phase component of the back EMF to compensate for the zero-phase voltage command value v _z ^* as. The detailed processing of the zero-phase back electromotive force compensator 80 will be described later.

The switching signal generation unit 30 provides the sum of the dq-axis voltage command values v _d ^* , v _q ^* , the zero-phase voltage command value v _z ^*, and the zero-phase back electromotive force compensation value v _z ^** (v _z ^* + _Vz ^** ) is input. The switching signal generation unit 30 generates a switching signal for turning on or off the switching elements 621a to 621d, 622a to 622d, and 623a to 623d based on these values. The switching signal is input to the inverter 620, and the motor drive current flows through the three-phase windings 201 to 203 of the motor 200 by the on / off operation of the switching elements 621a to 621d, 622a to 622d, and 623a to 623d.

Generally, the motor drive current is controlled to a sine wave to generate a rotating magnetic field required for drive. However, when the drive current is controlled to a sine wave, the effective value of the drive current cannot be increased after the maximum value of the sine wave reaches a predetermined current, and the output cannot be improved. In the present invention, together with improve the output by passing a zero-phase current i _z according to the motor operating conditions (torque command value T ^* of magnitude) as described below, so as to suppress an increase in torque ripple did.

(Description of the current command calculation unit 10)
FIG. 4 is a flowchart illustrating an example of the process of the current command calculation unit 10. In step S1, the torque command value ^{T *,} based on the angular velocity ω and the rotor position theta, dq axis current command value _i ^d _*, calculates the _i ^{q *.} dq-axis current command value i _{d ^*,} the i _q ^* method of calculating is and maximum torque current control and the field-weakening control is omitted because the known. Incidentally, dq axis current command value _i ^d _*, the _i ^{q *} calculations may be used a table set in advance.

In step S2, dq axis current command value _i ^d _*, based on the _i ^{q *} and the detected rotor position theta, UVW phase current command values ^{_i u} _{*, i} v _*, and calculates a _i ^{w *.}

In step S3, of the UVW-phase current command values i _u ^* , _iv ^* , i _w ^* calculated in step S2, the current command value having the largest absolute value is defined as the maximum phase current command value i _max ^* , and the amplitude Is the minimum phase current command value i _min ^* , and the rest is the intermediate phase current command value i _mid ^* .

In step S4, it is determined whether or not the absolute value of the maximum phase current command value i _max ^* is equal to or greater than a predetermined current value i _rated . Here, the predetermined current value i _rated means a maximum current value set to prevent the failure of the inverter 620 and the motor 200. In the present embodiment, the motor drive current is controlled to a predetermined current value or less.

If it is determined in step S4 that | i _max ^* | ≧ i _rated , the process proceeds to step S5, and the maximum phase current command value i _max ^** is recalculated by the following equation (2). In Equation (2), sgn (i _max ^* ) represents the sign of i _max ^* , and takes a minus or plus sign according to the sign of sgn (i _max ^* ).
i _max ^** = sgn (i _max ^* ) × i _rated (2)

In step S6, the intermediate phase current command value i _mid ^** is recalculated by the following equation (3).
_imid ^** = _imid ^* -( _imax ^* _-irated ) (3)

In step S7, the minimum phase current command value i _min ^** is recalculated by the following equation (4).
_imin ^** = _imin ^* -( _imax ^* _-irated ) (4)

FIG. 5 illustrates i _max ^** , i _mid ^** , and i _min ^** obtained by the processing from step S3 to step S7. The sinusoidal curves indicated by the thin lines are the U-phase current command value, the V-phase current command value, and the W-phase current command value calculated in step S2. At the rotor position θ1, the magnitude relation of the absolute value of the amplitude in the UVW phase current command value is | i _u ^* |> | i _w ^* |> | _iv ^* |, so i _u ^* is the maximum phase current. The command value is i _max ^* , i _w ^* is the intermediate phase current command value i _mid ^* , and _iv ^* is the minimum phase current command value i _min ^* . At this time, _iu ^** , _iv ^** , and _iw ^** are represented as in the following equations (5) to (7).
i _u ^** = i _max ^** = i _rated (5)
^{_{i v ** = i min ** =}} i v * - (i u * -i rated) ... (6)
^{_{i w ** = i mid ** =}} i w * - (i u * -i rated) ... (7)

On the other hand, the timing of the rotor position .theta.2, the magnitude relationship of the absolute value of the amplitude in the UVW phase current command value ^{_{| i w * |> | i}} u * |> | i v * | since the _{turned, i} ^{w *} There the maximum phase current command value _{i max} ^* _{next, i} ^{u *} is the middle phase current command value _{i mid} ^* _{next, i} ^{v *} is the minimum phase current command value _{i min} ^*. In this case, _iu ^** , _iv ^** , _iw ^** are represented as in the following equations (8) to (10).
^{_{i u ** = i mid ** =}} i u * - (i w * -i rated)
= I _u ^* + (i _rated- i _w ^* ) (8)
^{_{i v ** = i min ** =}} i v * - (i w * -i rated)
^{_{= I v * + (i rated}} -i w *) ... (9)
^{_{i w ** = i max ** =}} -i rated ... (10)

Once the maximum phase current command value i _max ^** , the intermediate phase current command value i _mid ^**, and the minimum phase current command value i _min ^** have been calculated in steps S5 to S7, in step S8, the maximum phase current command value i _max ^**, intermediate phase current command value _i ^{mid **} and the minimum phase current command value _i ^min dq-axis current command on the basis of the ^** value _i ^d _*, calculates the _i ^{q *} and zero-phase current command value _i ^{z *} . The calculated dq-axis current command value _i ^d _*, and outputs _i ^{q *} to the dq-axis current control unit 20 outputs a zero-phase current command value _i ^{z *} to zero-phase current control unit 60.

On the other hand, if it is determined in step S4 that the absolute value of the maximum phase current command value i _max ^* calculated in step S3 is smaller than the predetermined current value i _rated , the process proceeds to step S9, where the zero-phase current command value Set _iz ^* to _iz ^* = 0. In this case, dq axis current command value _i ^d * is calculated in step _{S1, i} ^{q *} is output to the dq-axis current controller 20, the zero-phase current command value _i ^z * = 0 calculated in step S9 is zero Output to phase current control section 60.

(Explanation of the zero-phase back electromotive force compensator 80)
FIG. 6 is a block diagram illustrating details of the zero-phase back electromotive force compensator 80. When controlling the zero-phase current in the configuration shown in FIG. 2, the difference between the zero-phase current value i _z detected with the zero-phase current command value i _z ^* by the zero-phase component of the back electromotive voltage generated when the motor drive is increased However, the torque ripple may increase. In this embodiment, in order to reduce the deviation between the zero-phase current command value i _z ^* and the zero-phase current i _z, zero-phase was calculated by the zero-phase back electromotive force compensation section 80 back EMF compensation value v _z ^{* By *} , the zero-phase voltage command value v _z ^* is compensated so as to cancel the zero-phase component of the back electromotive voltage.

In the zero-phase counter electromotive voltage calculating section 81, based on the zero-phase current detection value i _z and the rotor position theta, and a voltage drop due to the winding resistance R, the voltage drop due to the z-axis inductance L _z, due to the magnetic flux [psi _z The zero-phase back electromotive voltage _vZz , which is the sum with the zero-phase induced voltage, is calculated as in the following equation (11). The motor parameters such as z-axis inductance L _z and magnet flux [psi _z, in order to vary the drive current and temperature of the rotor position θ and the motor 200, be calculated using preset table or approximate expression Good.
_{v Zz = Ri z + L z} (di z / dt) + dψ z / dt ... (11)

The d-axis interference voltage calculator 82 calculates a d-axis interference voltage v _dz generated due to the d-z axis interference inductance L _dz based on the d-axis current detection value _id and the rotor position θ. The d-axis interference voltage v _dz is calculated by the following equation (12).
_{v dz = L dz (di d} / dt) ... (12)

In the q-axis interference voltage calculation unit 83, based on the q-axis current detection value i _q and the rotor position theta, and calculates the q-axis interference voltage v _qz caused by the qz-axis interference inductance L _qz. The q-axis interference voltage v _qz is calculated by the following equation (13).
_{v qz = L qz (di q} / dt) ... (13)

The difference between the zero-phase current command value _iz ^* and the zero-phase current detection value _iz is considered by considering a non-linear element not represented in the equations (11) to (13) using a preset table. Can be further reduced.

From the zero-phase back electromotive voltage compensator 80, the zero-phase back electromotive voltage v _Zz calculated by the zero-phase back electromotive voltage calculator 81 is _added to the d-axis interference voltage v _dz output from the d-axis interference voltage calculator 82. A value _obtained by adding the q-axis interference voltage v _qz output from the q-axis interference voltage calculation unit 83 is output as a zero-phase back electromotive force compensation value v _z ^** (= v _Zz + v _dz + v _qz ). Then, a value obtained by adding the zero-phase back electromotive force compensation value v _z ^** to the zero-phase voltage command value v _z ^* output from the zero-phase current control unit 60 is used instead of the zero-phase voltage command value v _z ^* for switching. The signal is input to the signal generator 30. In other words, so as to cancel the zero-phase component of the induced back emf in the winding 201 to 203, as zero-phase back electromotive force compensation value v _z amount corresponding larger zero-phase current i _z of ^** is produced Zero-phase voltage command value v _z ^* is adjusted.

(C1) As described above, the motor control device 610 converts the drive current of the motor 200 having the windings 201 to 203 wound independently between the phases into the torque command value T ^* , the angular speed ω of the motor 200, and the rotation speed. Control based on the child position θ. Then, the driving current _{(i u, i v, i} w) ratio of zero-phase current _{i z} is varied depending on the magnitude of the torque command value ^{T *} in. That is, the larger the torque command value T ^* .DELTA.i in FIG 5 is increased, thereby improving the ratio increases the output of the zero-phase current i _z accordingly. Conversely, if the torque command value T ^* small, .DELTA.i is also reduced ratio of zero-phase current i _z becomes smaller, when controlling the driving current of the pseudo rectangular wave as in the invention of Patent Document 1 described above As a result, torque ripple can be reduced.

(C2) When changing the ratio of the zero-phase current in the drive current in this way, as shown in FIG. 3, the zero-phase voltage command value v _z ^* is adjusted and the zero-phase voltage, which is the sum of the respective phase voltages, is adjusted. To change the ratio by controlling

(C3) Further, by changing the ratio of the zero-phase current so that the drive current is equal to or less than the predetermined current value (predetermined current value i _rated ), it is possible to prevent the motor 200 and the inverter 620 from failing due to excessive current. .

(Modification 1)
In the example described in FIGS. 3 and 4, whether to include a zero-phase current i _z drive current, the absolute value of the maximum phase current command value i _max ^* is in whether or not a predetermined current value i _rated higher was determined, as in the example shown in the flowchart of FIG. 7, it may be performed determines whether to include a zero-phase current i _z on the drive current based on the magnitude of the torque command value T ^*.

In the flowchart shown in FIG. 7, first, as in step S1 of FIG. 4 in step S101, the torque command value ^{T *,} based on the angular velocity ω and the rotor position theta, dq axis current command value _i ^d _*, the _i ^{q *} calculate.

In step S102, it is determined whether the torque command value T ^* is equal to or greater than a predetermined torque Tth. Here, the predetermined torque Tth means a maximum torque value set for preventing the inverter 620 and the motor 200 from malfunctioning.

If it is determined in step S102 that T ^* ≧ Tth, the process proceeds to step S103. In step S103, as in the case of step S2 described above, dq axis current command value _i ^d * computed in step _S101, based on the _i ^{q *} and the detected rotor position theta, UVW phase current command value _{i u} ^* , _Iv ^* , _iw ^* are calculated.

In step S104, in B4, a zero-phase current command value _iz ^* is calculated based on the UVW-phase current command values _iu ^* , _iv ^* , _iw ^* and the rotor position θ. The zero-phase current command value _iz ^* is calculated by the following equation (14).
^{_{i z * = A · sin (}} 3θ + 3α) ... (14)

In the equation (14), A means a current amplitude value necessary to reduce the maximum value of the UVW-phase current command values i _u ^* , _iv ^* , i _w ^* to a predetermined current value i _rated or less, α means a current phase obtained from the UVW phase current command values i _u ^* , _iv ^* , i _w ^* and the rotor position θ. In formula (14), UVW phase current command value the frequency of the zero-phase current command value ^{_{i z * i u *, i}} v *, was three times the _i ^{w *,} may not be 3-fold.

FIG. 8 is a diagram illustrating a drive current (U-phase current i _u ) when the zero-phase current command value _iz ^* of Expression (14) is used. A curve L1 shows a sine wave component contained in the U-phase current i _u, the curve L2 represents the zero-phase current i _z contained in U-phase current i _u. Thus, it is possible by including a zero-phase current i _z, the output improving while pay drive current within the predetermined current value i _rated.

On the other hand, if it is determined in step S102 that the torque command value T ^* is smaller than the predetermined torque Tth, the process proceeds to step S105, where the zero-phase current command value _iz ^* is set to _iz ^* = 0. Predetermined torque Tth in the case of the process shown in FIG. 7 corresponds to a predetermined current value i _rated in the case of FIG. 4, switches to state comprising zero-phase current i _z at substantially the same timing. Further, in the drive current in FIG. 8, the ratio of the zero-phase current _iz (curve L2) to the sine wave component (curve L1) increases as the torque command value T ^* increases, and the above-described embodiment is also applied to the first modification. The same operation and effect as those of the embodiment can be obtained.

(C4) Further, in the control shown in FIGS. 4 and 7, when the torque command value T ^* is smaller than a predetermined torque threshold value (predetermined torque Tth), or the maximum phase current command value i _max ^* based on the torque command value T ^{* .} because when the absolute value is smaller than the predetermined current value i _rated has a zero-phase current i _z is zero, in such operating conditions it is possible to prevent the occurrence of torque ripple caused by the zero-phase current i _z .

(C5) Furthermore, as shown in FIG. 6, the zero-phase back electromotive force compensation section 80, the driving current _{(i d, i q, i} z) and on the basis of the rotor position theta, the windings 201 to 203 The zero-phase voltage of the induced voltage (zero-phase counter-electromotive voltage compensation value v _z ^** ) is calculated, and the zero-phase voltage command value i based on the torque command value T ^* , the angular velocity ω of the motor 200, and the rotor position θ. _z ^* to preferably control the zero-phase current i _z in which the sum of the zero-phase back electromotive force compensation value v _z ^**.

If the inclusion of zero-phase current i _z drive current, the zero-phase component of the back electromotive voltage generated when the motor drive, the difference between the zero-phase current command value i _z ^* and the zero-phase current detection value i _z increases, The torque ripple may increase. However, as described above, by adding the zero-phase back electromotive force compensation value v _z ^** to zero-phase voltage command value i _z ^*, the zero-phase component of the induced back emf in the winding 201 to 203 zero-phase voltage command value v _z ^* is adjusted so as to cancel. As a result, the torque ripple can be further reduced.

(Modification 2)
FIG. 9 is a diagram illustrating Modification Example 2, and is a block diagram illustrating details of the motor control device 610. In FIG. 9, a UVW converter 90 is added to the configuration of the motor control device 610 shown in FIG. UVW conversion unit 90, dq-axis current command value i d *, i q * and the rotor position θ and the basis three-phase voltage command values i u *, i v *, and outputs the i w *.

The switching signal generation unit 30, the three-phase voltage command values ^{_{i u *, i v *,}} i w * on the zero-phase voltage command value _v ^{z *} and the zero-phase back electromotive force compensation value _v ^z sum of ^** (v _z ^* + _vz ^** ) is input. ^{_{That, i u * + (v z}} * + v z **), i v * + (v z * + v z **) and _i ^w * ₊ 3 one signal ^{_{(v z * + v z **}} ) is input You.

Also, the zero-phase back electromotive voltage compensation section 80, dq-axis current detection value i d, i q and instead of the zero-phase current detection value i z, dq axis current command value i d *, i q * and the zero-phase current Using the command value iz * , a zero-phase back electromotive force compensation value vz ** is calculated.

  In such a configuration, the same operation and effect as in the case of the configuration shown in FIG. 3 can be obtained. That is, by changing the ratio of the zero-phase current in the drive current according to the magnitude of the torque command value, the output can be improved and the torque ripple can be reduced.

  100 vehicle, 10 current command calculator, 20 dq-axis current controller, 30 switching signal generator, 40 dq converter, 50 zero-phase current calculator, 60 zero-phase current controller, 70 Speed converter, 80: zero-phase back electromotive force compensator, 200: motor, 220: current sensor, 600: power converter, 610: motor controller, 620: inverter, 621: U-phase full bridge inverter, 622: V Phase full bridge inverter, 623 ... W phase full bridge inverter

Claims (2)

A motor control device that controls a drive current of a motor having windings wound independently between phases based on a torque command value, an angular velocity of the motor, and a rotor position,
Changing the ratio of the zero-phase current in the drive current according to the magnitude of the torque command value ,
Based on the drive current and the rotor position, calculate a zero-phase voltage of a voltage induced in the winding,
A motor control device that controls the zero-sequence current by adding the zero-sequence voltage to a zero-sequence voltage instruction value based on the torque command value and the angular velocity and the rotor position of the motor. A motor having windings independently wound between phases;
An electric vehicle comprising: the motor control device according to claim 1 .

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