WO2017199641A1 - Electric motor control device and electric vehicle equipped with same - Google Patents

Electric motor control device and electric vehicle equipped with same Download PDF

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Publication number
WO2017199641A1
WO2017199641A1 PCT/JP2017/014758 JP2017014758W WO2017199641A1 WO 2017199641 A1 WO2017199641 A1 WO 2017199641A1 JP 2017014758 W JP2017014758 W JP 2017014758W WO 2017199641 A1 WO2017199641 A1 WO 2017199641A1
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WIPO (PCT)
Prior art keywords
phase
zero
pulse width
control device
motor
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PCT/JP2017/014758
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French (fr)
Japanese (ja)
Inventor
隆宏 荒木
利貞 三井
宮崎　英樹
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日立オートモティブシステムズ株式会社
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Publication of WO2017199641A1 publication Critical patent/WO2017199641A1/en

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    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L9/00—Electric propulsion with power supply external to the vehicle
    • B60L9/16—Electric propulsion with power supply external to the vehicle using ac induction motors
    • B60L9/18—Electric propulsion with power supply external to the vehicle using ac induction motors fed from dc supply lines
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility

Definitions

  • the present invention relates to a motor control device, and more particularly to a vehicle-mounted motor control device.
  • Hybrid vehicles and electric vehicles are required to have improved reliability from the viewpoint of preventing failure during vehicle travel and to improve output torque from the viewpoint of vehicle weight reduction.
  • a three-phase six-wire drive device has been considered, but since a motor with no neutral point connected is used, a zero-phase current is superimposed on the drive current that drives the motor, resulting in copper loss. There was a problem that losses such as increased.
  • Patent Document 1 JP-A-63-224693.
  • Patent Document 1 JP-A-63-224693.
  • This publication describes that “a current-carrying mode of the first and second voltage source inverters that reduces the current value of the zero-phase current is selected from a plurality of current-carrying modes”. As a result, the inverter operates so as to reduce the zero-phase current, so that the loss due to the zero-phase current can be reduced.
  • An object of the present invention is to reduce the number of times of switching when reducing the zero-phase current.
  • a motor control device controls a motor in which windings of each phase are independently connected, and controls a voltage to be applied to the motor based on an input torque command.
  • a motor control device that outputs a PWM control signal for the purpose, a pulse width of a zero-phase voltage consisting of the sum of output voltages based on a zero-phase voltage command for reducing the zero-phase current obtained from the AC current of the motor And controlling the amplitude.
  • the inverter control device can reduce the number of times of switching when the zero-phase current is reduced.
  • FIG. 4 is a diagram illustrating a flowchart of a switching signal generation unit 30.
  • FIG. It is a figure which shows the process of step 2 in the flowchart of FIG. 5, and has shown the hexagon part shown with the thick dotted line in the vector diagram of FIG.
  • FIG. 5 shows the process of step 4 in the flowchart of FIG. 5, and has shown the hexagon part shown with the thick dotted line in the vector diagram of FIG.
  • FIG. 1 is a diagram showing a configuration of a motor drive device according to an embodiment of the present invention.
  • the motor driving device includes a motor 200, a position sensor 210, a current sensor 220, an inverter 100, and the inverter control device 1.
  • the motor drive device functions as a motor drive system that drives the motor.
  • the motor 200 is constituted by an embedded magnet synchronous motor or the like to which a neutral point is not connected.
  • the U-phase winding 201 wound around the stator of the motor 200 is connected to the output terminal of the U-phase full bridge inverter 110.
  • the V-phase winding 202 wound around the stator of the motor 200 is connected to the output terminal of the V-phase full bridge inverter 111.
  • W phase winding 203 wound around the stator of motor 200 is connected to the output terminal of W phase full bridge inverter 112.
  • the current flowing in the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 can be controlled independently.
  • the neutral point of the motor 200 since the neutral point of the motor 200 is not connected, the drive current flowing through the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 is not included in Patent Document 1, as described in Patent Document 1. , Zero phase current is included.
  • the position sensor 210 detects the position of the rotor of the motor 200 and outputs the detected rotor position ⁇ .
  • the current sensor 220 detects currents flowing in the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 wound around the stator of the motor 200, and detects the detected three-phase currents iu, iv, iw is output.
  • the inverter 100 includes a U-phase full-bridge inverter 110, a V-phase full-bridge inverter 111, and a W-phase full-bridge inverter 112.
  • U-phase full-bridge inverter 110, V-phase full-bridge inverter 111, and W-phase full-bridge inverter 112 are connected in parallel to a DC power supply (not shown).
  • the U-phase full bridge inverter 110 is composed of switching elements 110a to 110d.
  • Switching element 110a is arranged on the U-phase left leg upper arm.
  • Switching element 110b is arranged on the U-phase left leg lower arm.
  • Switching element 110c is arranged on the U-phase right leg upper arm.
  • Switching element 110d is disposed on the U-phase right leg lower arm.
  • the V-phase full bridge inverter 111 is composed of switching elements 111a to 111d.
  • Switching element 111a is arranged on the V-phase left leg upper arm.
  • Switching element 111b is arranged on the lower arm of the V-phase left leg.
  • Switching element 111c is arranged on the V-phase right leg upper arm.
  • Switching element 111d is arranged on the lower arm of the V-phase right leg.
  • the W-phase full bridge inverter 112 is composed of switching elements 112a to 112d.
  • Switching element 112a is arranged on the W-phase left leg upper arm.
  • Switching element 112b is arranged on the lower arm of the W-phase left leg.
  • Switching element 112c is arranged on the W-phase right leg upper arm.
  • Switching element 112d is arranged on the lower arm of the W-phase right leg.
  • the inverter 100 By turning on or off the switching elements 110a to 110d, the switching elements 111a to 111d, and the switching elements 112a to 112d based on the switching signal generated by the inverter control device 1, the inverter 100 is connected to a DC power supply (not shown).
  • the applied DC voltage is converted into an AC voltage.
  • the converted AC voltage is applied to the U-phase winding 201, the V-phase winding 202 and the W-phase winding 203 wound around the stator of the motor 200 to generate a three-phase AC current. This three-phase alternating current generates a rotating magnetic field in the motor 200, and the rotor rotates.
  • Switching elements 110a to 110d, switching elements 111a to 111d, and switching elements 112a to 112d are configured by combining a metal oxide film type field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), and the like with a diode.
  • MOSFET metal oxide film type field effect transistor
  • IGBT insulated gate bipolar transistor
  • the inverter control device 1 performs PWM control on the inverter 100 based on the torque command value T * from the outside, the three-phase currents iu, iv, iw detected by the current sensor 220, and the rotor position ⁇ detected by the position sensor 210. To do.
  • FIG. 2 is a diagram showing an example of an output voltage waveform of the motor drive device according to the present embodiment.
  • the U phase output voltage Vu is an output voltage of the U phase full bridge inverter 110.
  • V-phase output voltage Vv is an output voltage of V-phase full-bridge inverter 111.
  • W-phase output voltage Vw is an output voltage of W-phase full-bridge inverter 112.
  • the zero-phase output voltage V0 is obtained by the equation (1) from the U-phase output voltage Vu, the V-phase output voltage Vv, and the W-phase output voltage Vw.
  • the U-phase full-bridge inverter 110, the V-phase full-bridge inverter 111, and the W-phase full-bridge inverter 112 output either a positive power supply voltage Vdc, a negative power supply voltage ⁇ Vdc, or zero voltage. .
  • the amplitude of the zero-phase output voltage V0 is one of the following depending on the combination.
  • FIG. 3 is an output voltage vector diagram of the motor drive device according to the present embodiment.
  • the components of the output voltage vector are shown in the order of the U phase, the V phase, and the W phase, and the magnitudes are + when the output voltage is a positive power supply voltage Vdc, and negative power supply voltage ⁇
  • Vdc positive power supply voltage
  • ⁇ negative power supply voltage
  • FIG. 4 is a control block diagram for explaining the first embodiment.
  • the current command calculation unit 10 calculates the d-axis current command value id * and the q-axis current command value iq * based on the input torque command value T * and the angular velocity ⁇ .
  • a method for calculating the d-axis current command value id * and the q-axis current command value iq * there are a maximum torque current control, a field weakening control, and the like. Note that a preset table may be used for calculating the d-axis current command value id * and the q-axis current command value iq *.
  • the dq-axis current control unit 20 receives the d-axis current command value id *, the q-axis current command value iq *, the d-axis current detection value id, and the q-axis current detection value iq, and performs proportional control and integral control. Used to output the d-axis voltage command value Vd * and the q-axis voltage command value Vq *.
  • the d-axis voltage command value Vd *, the q-axis voltage command value Vq *, and the zero-phase voltage command value V0 * are input to the switching signal generation unit 30, and the switching elements 110a to 110d, the switching elements 111a to 111d, A switching signal for turning on or off the switching elements 112a to 112d is generated.
  • the switching signal is input to the inverter 100, and the motor is operated by the above operation.
  • the dq converter 40 receives the three-phase currents iu, iv, iw detected by the current sensor 220 and the rotor position ⁇ detected by the position sensor 210, and detects the d-axis current detection value id, the q-axis current detection.
  • the value iq is output.
  • the zero-phase current calculation unit 50 receives the three-phase currents iu, iv, iw detected by the current sensor 220 and the rotor position ⁇ detected by the position sensor 210, and outputs the zero-phase current i0.
  • a formula for calculating the zero-phase current i0 is shown in Formula (3).
  • the zero phase current i0 may be calculated in consideration of the zero phase current value estimated from the angular velocity ⁇ .
  • the zero-phase current control unit 60 receives the zero-phase current i0 and outputs a zero-phase voltage command value V0 * using proportional control, integral control, or the like.
  • the speed converter 70 receives the rotor position ⁇ detected by the position sensor 210 and outputs an angular speed ⁇ .
  • FIG. 5 is a diagram showing a flowchart of the switching signal generation unit 30.
  • step 1 the switching signal generator 30 determines the polarity of the zero-phase voltage command value V0 * output from the zero-phase voltage controller 60. If the zero-phase voltage command value V0 * is positive, step 2 If the phase voltage command value V0 * is negative, the process of step 9 is performed.
  • Step 2 when the zero-phase voltage command value V0 * is positive, the switching signal generator 30 calculates the pulse width of the P1 mode.
  • step 3 it is determined whether the pulse width of the P1 mode calculated in step 2 is within the carrier period. If the pulse width of the P1 mode is within the carrier period, the process is completed, and the pulse width of the P1 mode is If the period is exceeded, the process of step 4 is performed.
  • step 4 when the pulse width of the P1 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the P2 mode.
  • step 5 it is determined whether the pulse width of the P2 mode calculated in step 4 is within the carrier period. If the pulse width of the P2 mode is within the carrier period, the process is completed, and the pulse width of the P2 mode is If the period is exceeded, the process of step 6 is performed.
  • step 6 when the pulse width of the P2 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the P3 mode.
  • step 7 it is determined whether the pulse width of the P3 mode calculated in step 6 is within the carrier period. If the pulse width of the P3 mode is within the carrier period, the process is completed, and the pulse width of the P3 mode is equal to the carrier period. If the period is exceeded, the process of step 8 is performed.
  • step 8 when the pulse width of the P3 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the P4 mode.
  • step 9 when the zero-phase voltage command value V0 * is negative, the switching signal generator 30 calculates the pulse width of the N1 mode. Thereafter, in step 10, it is determined whether the pulse width of the N1 mode calculated in step 9 is within the carrier period. If the pulse width of the N1 mode is within the carrier period, the process is completed, and the pulse width of the N1 mode is If the period is exceeded, the process of step 11 is performed.
  • step 11 when the pulse width of the N1 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the N2 mode.
  • step 12 it is determined whether the pulse width of the N2 mode calculated in step 11 is within the carrier period. If the pulse width of the N2 mode is within the carrier period, the process is completed, and the pulse width of the N2 mode is If the period is exceeded, the process of step 13 is performed.
  • step 13 when the pulse width of the N2 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the N3 mode.
  • step 14 it is determined whether the pulse width of the N3 mode calculated in step 13 is within the carrier period. If the pulse width of the N3 mode is within the carrier period, the process is completed, and the pulse width of the N3 mode is equal to the carrier period. If the period is exceeded, the process of step 15 is performed.
  • step 15 when the pulse width of the N3 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the N4 mode.
  • FIG. 6 is a diagram showing the process of step 2 in the flowchart of FIG. 5, and shows a hexagonal portion indicated by a thick dotted line in the vector diagram of FIG.
  • the calculation can be performed by rotating the vector diagram of FIG. 6 by 60 ° and inverting the polarity.
  • Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • FIG. 7 is a diagram showing the processing of step 4 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
  • calculation can be performed by rotating the vector diagram of FIG. 7 by 60 ° and inverting the polarity.
  • Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • FIG. 8 is a diagram showing the processing of step 6 and step 8 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
  • the calculation can be performed by rotating the vector diagram of FIG. 8 by 60 ° and inverting the polarity.
  • Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • step 8 t1, t2 and t5 are obtained from the dq-axis voltage command values Vd * and Vq * and the zero-phase voltage command value V0 * by the equation (7).
  • Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • FIG. 9 is a diagram showing the processing of step 9 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
  • calculation can be performed by rotating the vector diagram of FIG. 9 by 60 ° and inverting the polarity.
  • Va and Vb are voltage command values obtained by converting the dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • FIG. 10 is a diagram showing the processing of step 11 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
  • the calculation can be performed by rotating the vector diagram of FIG. 10 by 60 ° and inverting the polarity.
  • Va and Vb are voltage command values obtained by converting the dq axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • FIG. 11 is a diagram showing the processing of step 13 and step 15 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
  • the calculation can be performed by rotating the vector diagram of FIG. 11 by 60 ° and inverting the polarity.
  • Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • step 15 t1, t2, and t8 are dq axis voltage command values Vd *, Vq *, It is obtained from the zero-phase voltage command value V0 * by the equation (11).
  • Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
  • the zero-phase voltage having a small amplitude can be obtained within a range in which the pulse width of the voltage output from the inverter of each phase does not exceed the carrier cycle.
  • the switching loss can be reduced by reducing the number of times of switching.
  • switching element 112 ... W-phase full bridge inverter, 112a ... switching element, 112b ... switching element, 112c ... switching element, 112d ... switching element , 20 ... Motor, 201 ... U phase winding, 202 ... V phase winding, 203 ... W phase winding, 210 ... Position sensor, 220 ... Current sensor, iu ... U phase current, iv ... V phase current, iw ... W phase Current, id * ... d-axis current command value, iq * ... q-axis current command value, i d ... d-axis current detection value, i q ... q-axis current detection value, i0 ...

Abstract

The purpose of the present invention is to reduce the number of times of switching to be performed when reducing a zero-phase current. This electric motor control device controls an electric motor, the phases of which are connected independently of one another, and has a control means which outputs a PWM control signal for controlling a voltage to be applied to the electric motor on the basis of an input torque command. The control means controls, on the basis of a zero-phase voltage command which is obtained from an alternating current of the electric motor and used for reducing a zero-phase current, the pulse width and amplitude of a zero-phase voltage consisting of the sum of output voltages.

Description

電動機の制御装置及びそれを備えた電動車両Electric motor control device and electric vehicle equipped with the same
　本発明は電動機の制御装置に関し、特に車載用の電動機の制御装置に関する。 The present invention relates to a motor control device, and more particularly to a vehicle-mounted motor control device.
　ハイブリッド自動車や電気自動車は、車両走行中の故障発生防止の観点から信頼性の向上と、車両軽量化の観点から出力トルクの向上が要求される。これらの要求に対して三相６線式の駆動装置が考えられているが、中性点が接続されていない電動機を用いるため、電動機を駆動する駆動電流に零相電流が重畳し、銅損などの損失が増加するという課題があった。 Hybrid vehicles and electric vehicles are required to have improved reliability from the viewpoint of preventing failure during vehicle travel and to improve output torque from the viewpoint of vehicle weight reduction. In response to these requirements, a three-phase six-wire drive device has been considered, but since a motor with no neutral point connected is used, a zero-phase current is superimposed on the drive current that drives the motor, resulting in copper loss. There was a problem that losses such as increased.
　本技術分野の背景技術として、特開昭６３－２２４６９３号公報（特許文献１）がある。この公報には、「複数通電モードの中から零相電流の現在値を減少させるような第１および第２の電圧形インバータの通電モードを選択する」と記載されている。これにより零相電流を減少させるようにインバータが動作するため、零相電流による損失を低減することができる。 As background art in this technical field, there is JP-A-63-224693 (Patent Document 1). This publication describes that “a current-carrying mode of the first and second voltage source inverters that reduces the current value of the zero-phase current is selected from a plurality of current-carrying modes”. As a result, the inverter operates so as to reduce the zero-phase current, so that the loss due to the zero-phase current can be reduced.
特開昭６３－２２４６９３号公報Japanese Unexamined Patent Publication No. 63-224693
　特許文献１に記載の方法では、零相電流を減少させるためにスイッチング回数が増加する恐れがある。 In the method described in Patent Document 1, there is a possibility that the number of times of switching increases in order to reduce the zero-phase current.
　本発明の目的は、零相電流を減少させる際のスイッチング回数を低減することである。 An object of the present invention is to reduce the number of times of switching when reducing the zero-phase current.
　上記の課題を解決するため、本発明に係る電動機の制御装置は、各相の巻線が独立して結線された電動機を制御するとともに、入力されたトルク指令に基づき電動機へ印加する電圧を制御するためのＰＷＭ制御信号を出力する電動機の制御装置において、前記電動機の交流電流から求められる零相電流を低減するための零相電圧指令に基づき、　出力電圧の総和からなる零相電圧のパルス幅及び振幅を制御する。 In order to solve the above problems, a motor control device according to the present invention controls a motor in which windings of each phase are independently connected, and controls a voltage to be applied to the motor based on an input torque command. In a motor control device that outputs a PWM control signal for the purpose, a pulse width of a zero-phase voltage consisting of the sum of output voltages based on a zero-phase voltage command for reducing the zero-phase current obtained from the AC current of the motor And controlling the amplitude.
　本発明に係るインバータ制御装置によれば、零相電流を減少させる際のスイッチング回数を低減することができる。 The inverter control device according to the present invention can reduce the number of times of switching when the zero-phase current is reduced.
本実施形態に係るモータ駆動装置の構成を示す図である。It is a figure which shows the structure of the motor drive device which concerns on this embodiment. 本実施形態に係るモータ駆動装置の出力電圧波形例を示す図である。It is a figure which shows the output voltage waveform example of the motor drive device which concerns on this embodiment. 本実施形態に係るモータ駆動装置の出力電圧ベクトル図である。It is an output voltage vector diagram of the motor drive device concerning this embodiment. 第１の実施例を説明する制御ブロック図である。It is a control block diagram explaining a 1st example. スイッチング信号生成部３０のフローチャートを示す図である。4 is a diagram illustrating a flowchart of a switching signal generation unit 30. FIG. 図５のフローチャートにおけるステップ２の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。It is a figure which shows the process of step 2 in the flowchart of FIG. 5, and has shown the hexagon part shown with the thick dotted line in the vector diagram of FIG. 図５のフローチャートにおけるステップ４の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。It is a figure which shows the process of step 4 in the flowchart of FIG. 5, and has shown the hexagon part shown with the thick dotted line in the vector diagram of FIG. 図５のフローチャートにおけるステップ６およびステップ８の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。It is a figure which shows the process of step 6 and step 8 in the flowchart of FIG. 5, and has shown the hexagonal part shown with the thick dotted line in the vector diagram of FIG. 図５のフローチャートにおけるステップ９の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。It is a figure which shows the process of step 9 in the flowchart of FIG. 5, and has shown the hexagon part shown with the thick dotted line in the vector diagram of FIG. 図５のフローチャートにおけるステップ１１の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。It is a figure which shows the process of step 11 in the flowchart of FIG. 5, and has shown the hexagon part shown with the thick dotted line in the vector diagram of FIG. 図５のフローチャートにおけるステップ１３およびステップ１５の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。It is a figure which shows the process of step 13 and step 15 in the flowchart of FIG. 5, and has shown the hexagonal part shown with the thick dotted line in the vector diagram of FIG.
　以下、図面を参照して本発明の実施形態を説明する。ただし、本発明は下記の実施形態に限定解釈されるものではなく、公知の他の構成要素を組み合わせて本発明の技術思想を実現してもよい。なお、各図において同一要素については同一の符号を記し、重複する説明は省略する。 Hereinafter, embodiments of the present invention will be described with reference to the drawings. However, the present invention is not limited to the following embodiments, and the technical idea of the present invention may be realized by combining other known constituent elements. In addition, in each figure, the same code | symbol is described about the same element and the overlapping description is abbreviate | omitted.
　図１は、本発明の実施形態に係るモータ駆動装置の構成を示す図である。 FIG. 1 is a diagram showing a configuration of a motor drive device according to an embodiment of the present invention.
　モータ駆動装置は、モータ２００と、位置センサ２１０と、電流センサ２２０と、インバータ１００と、インバータ制御装置１と、を有する。モータ駆動装置は、モータを駆動するモータ駆動システムとして機能する・
　モータ２００は、中性点が接続されていない埋込磁石同期電動機などにより構成される。 The motor driving device includes a motor 200, a position sensor 210, a current sensor 220, an inverter 100, and the inverter control device 1. The motor drive device functions as a motor drive system that drives the motor.
The motor 200 is constituted by an embedded magnet synchronous motor or the like to which a neutral point is not connected.
　モータ２００の固定子に巻かれたＵ相巻線２０１は、Ｕ相フルブリッジインバータ１１０の出力端子に接続される。 The U-phase winding 201 wound around the stator of the motor 200 is connected to the output terminal of the U-phase full bridge inverter 110.
　モータ２００の固定子に巻かれたＶ相巻線２０２は、Ｖ相フルブリッジインバータ１１１の出力端子に接続される。 The V-phase winding 202 wound around the stator of the motor 200 is connected to the output terminal of the V-phase full bridge inverter 111.
　モータ２００の固定子に巻かれたＷ相巻線２０３は、Ｗ相フルブリッジインバータ１１２の出力端子に接続される。 W phase winding 203 wound around the stator of motor 200 is connected to the output terminal of W phase full bridge inverter 112.
　モータ２００は中性点が接続されていないことにより、Ｕ相巻線２０１と、Ｖ相巻線２０２と、Ｗ相巻線２０３に流れる電流をそれぞれ独立に制御することができる。ただし、モータ２００は中性点が接続されていないため、特許文献１に記載されている通り、Ｕ相巻線２０１と、Ｖ相巻線２０２と、Ｗ相巻線２０３に流れる駆動電流には、零相電流が含まれる。 Since the motor 200 is not connected to the neutral point, the current flowing in the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 can be controlled independently. However, since the neutral point of the motor 200 is not connected, the drive current flowing through the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 is not included in Patent Document 1, as described in Patent Document 1. , Zero phase current is included.
　位置センサ２１０は、モータ２００の回転子の位置を検出し、検出した回転子位置θを出力する。 The position sensor 210 detects the position of the rotor of the motor 200 and outputs the detected rotor position θ.
　電流センサ２２０は、モータ２００の固定子に巻かれた、Ｕ相巻線２０１と、Ｖ相巻線２０２と、Ｗ相巻線２０３に流れる電流を検出し、検出した三相電流ｉｕ、ｉｖ、ｉｗを出力する。 The current sensor 220 detects currents flowing in the U-phase winding 201, the V-phase winding 202, and the W-phase winding 203 wound around the stator of the motor 200, and detects the detected three-phase currents iu, iv, iw is output.
　インバータ１００は、Ｕ相フルブリッジインバータ１１０と、Ｖ相フルブリッジインバータ１１１と、Ｗ相フルブリッジインバータ１１２と、により構成される。Ｕ相フルブリッジインバータ１１０と、Ｖ相フルブリッジインバータ１１１と、Ｗ相フルブリッジインバータ１１２と、は、図略の直流電源に並列接続される。 The inverter 100 includes a U-phase full-bridge inverter 110, a V-phase full-bridge inverter 111, and a W-phase full-bridge inverter 112. U-phase full-bridge inverter 110, V-phase full-bridge inverter 111, and W-phase full-bridge inverter 112 are connected in parallel to a DC power supply (not shown).
　Ｕ相フルブリッジインバータ１１０は、スイッチング素子１１０ａ～１１０ｄにより構成される。スイッチング素子１１０ａはＵ相左レグ上アームに配置される。スイッチング素子１１０ｂはＵ相左レグ下アームに配置される。スイッチング素子１１０ｃはＵ相右レグ上アームに配置される。 The U-phase full bridge inverter 110 is composed of switching elements 110a to 110d. Switching element 110a is arranged on the U-phase left leg upper arm. Switching element 110b is arranged on the U-phase left leg lower arm. Switching element 110c is arranged on the U-phase right leg upper arm.
　スイッチング素子１１０ｄはＵ相右レグ下アームに配置される。 Switching element 110d is disposed on the U-phase right leg lower arm.
　Ｖ相フルブリッジインバータ１１１は、スイッチング素子１１１ａ～１１１ｄにより構成される。スイッチング素子１１１ａはＶ相左レグ上アームに配置される。スイッチング素子１１１ｂはＶ相左レグ下アームに配置される。スイッチング素子１１１ｃはＶ相右レグ上アームに配置される。スイッチング素子１１１ｄはＶ相右レグ下アームに配置される。 The V-phase full bridge inverter 111 is composed of switching elements 111a to 111d. Switching element 111a is arranged on the V-phase left leg upper arm. Switching element 111b is arranged on the lower arm of the V-phase left leg. Switching element 111c is arranged on the V-phase right leg upper arm. Switching element 111d is arranged on the lower arm of the V-phase right leg.
　Ｗ相フルブリッジインバータ１１２は、スイッチング素子１１２ａ～１１２ｄにより構成される。スイッチング素子１１２ａはＷ相左レグ上アームに配置される。スイッチング素子１１２ｂはＷ相左レグ下アームに配置される。スイッチング素子１１２ｃはＷ相右レグ上アームに配置される。スイッチング素子１１２ｄはＷ相右レグ下アームに配置される。 The W-phase full bridge inverter 112 is composed of switching elements 112a to 112d. Switching element 112a is arranged on the W-phase left leg upper arm. Switching element 112b is arranged on the lower arm of the W-phase left leg. Switching element 112c is arranged on the W-phase right leg upper arm. Switching element 112d is arranged on the lower arm of the W-phase right leg.
　スイッチング素子１１０ａ～１１０ｄと、スイッチング素子１１１ａ～１１１ｄと、スイッチング素子１１２ａ～１１２ｄをインバータ制御装置１で生成されたスイッチング信号に基づいてオンもしくはオフすることで、インバータ１００は、図略の直流電源から印加された直流電圧を交流電圧に変換する。変換された交流電圧は、モータ２００の固定子に巻かれたＵ相巻線２０１とＶ相巻線２０２とＷ相巻線２０３に印加され、３相交流電流を発生させる。この３相交流電流がモータ２００に回転磁界を発生させ、回転子が回転する。 By turning on or off the switching elements 110a to 110d, the switching elements 111a to 111d, and the switching elements 112a to 112d based on the switching signal generated by the inverter control device 1, the inverter 100 is connected to a DC power supply (not shown). The applied DC voltage is converted into an AC voltage. The converted AC voltage is applied to the U-phase winding 201, the V-phase winding 202 and the W-phase winding 203 wound around the stator of the motor 200 to generate a three-phase AC current. This three-phase alternating current generates a rotating magnetic field in the motor 200, and the rotor rotates.
　スイッチング素子１１０ａ～１１０ｄと、スイッチング素子１１１ａ～１１１ｄと、スイッチング素子１１２ａ～１１２ｄは、金属酸化膜型電界効果トランジスタ（ＭＯＳＦＥＴ）や絶縁ゲートバイポーラトランジスタ（ＩＧＢＴ）などと、ダイオードを組み合わせて構成される。本実施形態では、ＭＯＳＦＥＴとダイオードを用いる構成で説明する。 Switching elements 110a to 110d, switching elements 111a to 111d, and switching elements 112a to 112d are configured by combining a metal oxide film type field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), and the like with a diode. In the present embodiment, a configuration using MOSFETs and diodes will be described.
　インバータ制御装置１は、外部からのトルク指令値Ｔ＊、電流センサ２２０で検出された三相電流ｉｕ、ｉｖ、ｉｗ、位置センサ２１０で検出された回転子位置θに基づいてインバータ１００をＰＷＭ制御する。 The inverter control device 1 performs PWM control on the inverter 100 based on the torque command value T * from the outside, the three-phase currents iu, iv, iw detected by the current sensor 220, and the rotor position θ detected by the position sensor 210. To do.
　図２は、本実施形態に係るモータ駆動装置の出力電圧波形例を示す図である。 FIG. 2 is a diagram showing an example of an output voltage waveform of the motor drive device according to the present embodiment.
　Ｕ相出力電圧Ｖｕは、Ｕ相フルブリッジインバータ１１０の出力電圧である。Ｖ相出力電圧Ｖｖは、Ｖ相フルブリッジインバータ１１１の出力電圧である。Ｗ相出力電圧Ｖｗは、Ｗ相フルブリッジインバータ１１２の出力電圧である。零相出力電圧Ｖ0は、Ｕ相出力電圧Ｖｕと、Ｖ相出力電圧Ｖｖと、Ｗ相出力電圧Ｖｗから（１）式で求められる。 The U phase output voltage Vu is an output voltage of the U phase full bridge inverter 110. V-phase output voltage Vv is an output voltage of V-phase full-bridge inverter 111. W-phase output voltage Vw is an output voltage of W-phase full-bridge inverter 112. The zero-phase output voltage V0 is obtained by the equation (1) from the U-phase output voltage Vu, the V-phase output voltage Vv, and the W-phase output voltage Vw.
Figure JPOXMLDOC01-appb-M000001
Figure JPOXMLDOC01-appb-M000001
　Ｕ相フルブリッジインバータ１１０と、Ｖ相フルブリッジインバータ１１１と、Ｗ相フルブリッジインバータ１１２は、正極性の電源電圧Ｖｄｃか、負極性の電源電圧－Ｖｄｃか、０のいずれかの電圧を出力する。 The U-phase full-bridge inverter 110, the V-phase full-bridge inverter 111, and the W-phase full-bridge inverter 112 output either a positive power supply voltage Vdc, a negative power supply voltage −Vdc, or zero voltage. .
　そのため、零相出力電圧Ｖ0の振幅は、その組み合わせによって、以下のいずれかとなる。 Therefore, the amplitude of the zero-phase output voltage V0 is one of the following depending on the combination.
Figure JPOXMLDOC01-appb-M000002
Figure JPOXMLDOC01-appb-M000002
　図３は、本実施形態に係るモータ駆動装置の出力電圧ベクトル図である。 FIG. 3 is an output voltage vector diagram of the motor drive device according to the present embodiment.
　本ベクトル図において、出力電圧ベクトルの成分は、Ｕ相、Ｖ相、Ｗ相の順に示され、その大きさは、出力電圧が正極性の電源電圧Ｖｄｃのときは＋、負極性の電源電圧－Ｖｄｃのときは－、０のときは０と表現されている。零相電圧Ｖｚはｚ軸成分として扱い、点の形状によりその大きさを表現した。以降、太線で示した六角形の内接円を出力電圧の最大値とした場合を扱う。 In this vector diagram, the components of the output voltage vector are shown in the order of the U phase, the V phase, and the W phase, and the magnitudes are + when the output voltage is a positive power supply voltage Vdc, and negative power supply voltage − When it is Vdc, it is expressed as-, and when it is 0, it is expressed as 0. The zero-phase voltage Vz is treated as a z-axis component, and the magnitude is expressed by the shape of a point. In the following, the case where the hexagonal inscribed circle indicated by the bold line is the maximum value of the output voltage will be treated.
　図４は、第１の実施例を説明する制御ブロック図である。 FIG. 4 is a control block diagram for explaining the first embodiment.
　電流指令演算部１０は、入力されたトルク指令値Ｔ＊と、角速度ωに基づき、ｄ軸電流指令値ｉｄ＊及びｑ軸電流指令値ｉｑ＊を計算する。ｄ軸電流指令値ｉｄ＊及びｑ軸電流指令値ｉｑ＊の計算方法としては、最大トルク電流制御や弱め界磁制御などがあるが、周知のため説明を省略する。なお、ｄ軸電流指令値ｉｄ＊及びｑ軸電流指令値ｉｑ＊の計算には、予め設定したテーブルを使用してもよい。 The current command calculation unit 10 calculates the d-axis current command value id * and the q-axis current command value iq * based on the input torque command value T * and the angular velocity ω. As a method for calculating the d-axis current command value id * and the q-axis current command value iq *, there are a maximum torque current control, a field weakening control, and the like. Note that a preset table may be used for calculating the d-axis current command value id * and the q-axis current command value iq *.
　ｄｑ軸電流制御部２０には、ｄ軸電流指令値ｉｄ＊及びｑ軸電流指令値ｉｑ＊と、ｄ軸電流検出値ｉｄ、ｑ軸電流検出値ｉｑが入力され、比例制御や積分制御などを用いてｄ軸電圧指令値Ｖｄ＊及びｑ軸電圧指令値Ｖｑ＊を出力する。 The dq-axis current control unit 20 receives the d-axis current command value id *, the q-axis current command value iq *, the d-axis current detection value id, and the q-axis current detection value iq, and performs proportional control and integral control. Used to output the d-axis voltage command value Vd * and the q-axis voltage command value Vq *.
　スイッチング信号生成部３０には、ｄ軸電圧指令値Ｖｄ＊、ｑ軸電圧指令値Ｖｑ＊と、零相電圧指令値Ｖ0＊が入力され、スイッチング素子１１０ａ～１１０ｄと、スイッチング素子１１１ａ～１１１ｄと、スイッチング素子１１２ａ～１１２ｄと、をオンもしくはオフするスイッチング信号を生成する。 The d-axis voltage command value Vd *, the q-axis voltage command value Vq *, and the zero-phase voltage command value V0 * are input to the switching signal generation unit 30, and the switching elements 110a to 110d, the switching elements 111a to 111d, A switching signal for turning on or off the switching elements 112a to 112d is generated.
　インバータ１００には、スイッチング信号が入力され、前記動作によりモータを運転する。 The switching signal is input to the inverter 100, and the motor is operated by the above operation.
　ｄｑ変換部４０には、電流センサ２２０で検出された三相電流ｉｕ、ｉｖ、ｉｗと、位置センサ２１０で検出された回転子位置θが入力され、ｄ軸電流検出値ｉｄ、ｑ軸電流検出値ｉｑを出力する。 The dq converter 40 receives the three-phase currents iu, iv, iw detected by the current sensor 220 and the rotor position θ detected by the position sensor 210, and detects the d-axis current detection value id, the q-axis current detection. The value iq is output.
　零相電流算出部５０には、電流センサ２２０で検出された三相電流ｉｕ、ｉｖ、ｉｗと、位置センサ２１０で検出された回転子位置θが入力され、零相電流ｉ0が出力される。
前記零相電流ｉ0の計算式を（３）式に示す。 The zero-phase current calculation unit 50 receives the three-phase currents iu, iv, iw detected by the current sensor 220 and the rotor position θ detected by the position sensor 210, and outputs the zero-phase current i0.
A formula for calculating the zero-phase current i0 is shown in Formula (3).
Figure JPOXMLDOC01-appb-M000003
Figure JPOXMLDOC01-appb-M000003
　なお、零相電流ｉ0はモータ２００の回転速度により変化するため、角速度ωから推定した零相電流値を考慮して算出してもよい。 Since the zero phase current i0 varies depending on the rotational speed of the motor 200, the zero phase current i0 may be calculated in consideration of the zero phase current value estimated from the angular velocity ω.
　零相電流制御部６０には、零相電流ｉ0が入力され、比例制御や積分制御などを用いて零相電圧指令値Ｖ0＊が出力する。 The zero-phase current control unit 60 receives the zero-phase current i0 and outputs a zero-phase voltage command value V0 * using proportional control, integral control, or the like.
　速度変換部７０には、位置センサ２１０で検出された回転子位置θが入力され、角速度ωが出力する。 The speed converter 70 receives the rotor position θ detected by the position sensor 210 and outputs an angular speed ω.
　図５はスイッチング信号生成部３０のフローチャートを示す図である。 FIG. 5 is a diagram showing a flowchart of the switching signal generation unit 30.
　ステップ１において、スイッチング信号生成部３０は、零相電圧制御部６０から出力される零相電圧指令値Ｖ0＊の極性を判別し、零相電圧指令値Ｖ0＊が正の場合はステップ２、零相電圧指令値Ｖ0＊が負の場合はステップ９の処理を行う。 In step 1, the switching signal generator 30 determines the polarity of the zero-phase voltage command value V0 * output from the zero-phase voltage controller 60. If the zero-phase voltage command value V0 * is positive, step 2 If the phase voltage command value V0 * is negative, the process of step 9 is performed.
　ステップ２において、零相電圧指令値Ｖ0＊が正の場合、スイッチング信号生成部３０は、Ｐ１モードのパルス幅を算出する。 In Step 2, when the zero-phase voltage command value V0 * is positive, the switching signal generator 30 calculates the pulse width of the P1 mode.
　その後、ステップ３において、ステップ２で算出されたＰ１モードのパルス幅がキャリア周期内かを判別し、Ｐ１モードのパルス幅がキャリア周期内の場合は処理を完了し、Ｐ１モードのパルス幅がキャリア周期を超える場合はステップ４の処理を行う。 Thereafter, in step 3, it is determined whether the pulse width of the P1 mode calculated in step 2 is within the carrier period. If the pulse width of the P1 mode is within the carrier period, the process is completed, and the pulse width of the P1 mode is If the period is exceeded, the process of step 4 is performed.
　ステップ４において、Ｐ１モードのパルス幅がキャリア周期を超えた場合、スイッチング信号生成部３０は、Ｐ２モードのパルス幅を算出する。 In step 4, when the pulse width of the P1 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the P2 mode.
　その後、ステップ５において、ステップ４で算出されたＰ２モードのパルス幅がキャリア周期内かを判別し、Ｐ２モードのパルス幅がキャリア周期内の場合は処理を完了し、Ｐ２モードのパルス幅がキャリア周期を超える場合はステップ６の処理を行う。 Thereafter, in step 5, it is determined whether the pulse width of the P2 mode calculated in step 4 is within the carrier period. If the pulse width of the P2 mode is within the carrier period, the process is completed, and the pulse width of the P2 mode is If the period is exceeded, the process of step 6 is performed.
　ステップ６において、Ｐ２モードのパルス幅がキャリア周期を超えた場合、スイッチング信号生成部３０は、Ｐ３モードのパルス幅を算出する。 In step 6, when the pulse width of the P2 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the P3 mode.
　その後、ステップ７において、ステップ６で算出されたＰ３モードのパルス幅がキャリア周期内かを判別し、Ｐ３モードのパルス幅がキャリア周期内の場合は処理を完了し、Ｐ３モードのパルス幅がキャリア周期を超える場合はステップ８の処理を行う。 Thereafter, in step 7, it is determined whether the pulse width of the P3 mode calculated in step 6 is within the carrier period. If the pulse width of the P3 mode is within the carrier period, the process is completed, and the pulse width of the P3 mode is equal to the carrier period. If the period is exceeded, the process of step 8 is performed.
　ステップ８において、Ｐ３モードのパルス幅がキャリア周期を超えた場合、スイッチング信号生成部３０は、Ｐ４モードのパルス幅を算出する。 In step 8, when the pulse width of the P3 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the P4 mode.
　ステップ９において、零相電圧指令値Ｖ0＊が負の場合、スイッチング信号生成部３０は、Ｎ１モードのパルス幅を算出する。その後、ステップ１０において、ステップ９で算出されたＮ１モードのパルス幅がキャリア周期内かを判別し、Ｎ１モードのパルス幅がキャリア周期内の場合は処理を完了し、Ｎ１モードのパルス幅がキャリア周期を超える場合はステップ１１の処理を行う。 In step 9, when the zero-phase voltage command value V0 * is negative, the switching signal generator 30 calculates the pulse width of the N1 mode. Thereafter, in step 10, it is determined whether the pulse width of the N1 mode calculated in step 9 is within the carrier period. If the pulse width of the N1 mode is within the carrier period, the process is completed, and the pulse width of the N1 mode is If the period is exceeded, the process of step 11 is performed.
　ステップ１１において、Ｎ１モードのパルス幅がキャリア周期を超えた場合、スイッチング信号生成部３０は、Ｎ２モードのパルス幅を算出する。 In step 11, when the pulse width of the N1 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the N2 mode.
　その後、ステップ１２において、ステップ１１で算出されたＮ２モードのパルス幅がキャリア周期内かを判別し、Ｎ２モードのパルス幅がキャリア周期内の場合は処理を完了し、Ｎ２モードのパルス幅がキャリア周期を超える場合はステップ１３の処理を行う。 Thereafter, in step 12, it is determined whether the pulse width of the N2 mode calculated in step 11 is within the carrier period. If the pulse width of the N2 mode is within the carrier period, the process is completed, and the pulse width of the N2 mode is If the period is exceeded, the process of step 13 is performed.
　ステップ１３において、Ｎ２モードのパルス幅がキャリア周期を超えた場合、スイッチング信号生成部３０は、Ｎ３モードのパルス幅を算出する。 In step 13, when the pulse width of the N2 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the N3 mode.
　その後、ステップ１４において、ステップ１３で算出されたＮ３モードのパルス幅がキャリア周期内かを判別し、Ｎ３モードのパルス幅がキャリア周期内の場合は処理を完了し、Ｎ３モードのパルス幅がキャリア周期を超える場合はステップ１５の処理を行う。 Thereafter, in step 14, it is determined whether the pulse width of the N3 mode calculated in step 13 is within the carrier period. If the pulse width of the N3 mode is within the carrier period, the process is completed, and the pulse width of the N3 mode is equal to the carrier period. If the period is exceeded, the process of step 15 is performed.
　ステップ１５において、Ｎ３モードのパルス幅がキャリア周期を超えた場合、スイッチング信号生成部３０は、Ｎ４モードのパルス幅を算出する。 In step 15, when the pulse width of the N3 mode exceeds the carrier period, the switching signal generation unit 30 calculates the pulse width of the N4 mode.
　図６は、図５のフローチャートにおけるステップ２の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。 FIG. 6 is a diagram showing the process of step 2 in the flowchart of FIG. 5, and shows a hexagonal portion indicated by a thick dotted line in the vector diagram of FIG.
　以下、本領域においてＰ１モードのパルス幅を算出する方法を述べる。なお、他の領域においては図６のベクトル図を６０°回転させ、極性を反転させることによって算出することができる。 Hereinafter, a method for calculating the pulse width of the P1 mode in this area will be described. In other regions, the calculation can be performed by rotating the vector diagram of FIG. 6 by 60 ° and inverting the polarity.
　図６において、Ｖ1を出力する期間をｔ1、Ｖ2を出力する期間をｔ2、Ｖ3を出力する期間をｔ3、とすると、ｔ1とｔ2とｔ3は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、零相電圧指令値Ｖ0＊から（４）式で求められる。 In FIG. 6, if the period for outputting V1 is t1, the period for outputting V2 is t2, and the period for outputting V3 is t3, t1, t2 and t3 are dq-axis voltage command values Vd *, Vq *, It can be obtained from the zero-phase voltage command value V0 * by the equation (4).
Figure JPOXMLDOC01-appb-M000004
Figure JPOXMLDOC01-appb-M000004
　なお、ＶaとVbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図６の固定座標に変換した電圧指令値である。 Note that Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　図７は、図５のフローチャートにおけるステップ４の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。 FIG. 7 is a diagram showing the processing of step 4 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
　以下、本領域においてＰ２モードのパルス幅を算出する方法を述べる。なお、他の領域においては図７のベクトル図を６０°回転させ、極性を反転させることによって算出することができる。 Hereinafter, a method for calculating the pulse width of the P2 mode in this area will be described. In other regions, calculation can be performed by rotating the vector diagram of FIG. 7 by 60 ° and inverting the polarity.
　図７において、Ｖ1を出力する期間をｔ1、Ｖ2を出力する期間をｔ2、V4を出力する期間をｔ4、とすると、ｔ1とｔ2とｔ4は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、零相電圧指令値Ｖ0＊から（５）式で求められる。 In FIG. 7, if the period for outputting V1 is t1, the period for outputting V2 is t2, and the period for outputting V4 is t4, t1, t2 and t4 are dq axis voltage command values Vd *, Vq *, It can be obtained from the zero-phase voltage command value V0 * by the equation (5).
Figure JPOXMLDOC01-appb-M000005
Figure JPOXMLDOC01-appb-M000005
　なお、ＶaとＶbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図７の固定座標に変換した電圧指令値である。 Note that Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　図８は、図５のフローチャートにおけるステップ６およびステップ８の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。 FIG. 8 is a diagram showing the processing of step 6 and step 8 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
　以下、本領域においてＰ３モードとＰ４モードのパルス幅を算出する方法を述べる。なお、他の領域においては図８のベクトル図を６０°回転させ、極性を反転させることによって算出することができる。 Hereinafter, a method for calculating the pulse widths of the P3 mode and the P4 mode in this area will be described. In other regions, the calculation can be performed by rotating the vector diagram of FIG. 8 by 60 ° and inverting the polarity.
　図８において、Ｖ1を出力する期間をｔ1、Ｖ2を出力する期間をｔ2、V5を出力する期間をｔ5、とすると、ステップ６においてｔ1とｔ2とｔ5は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、零相電圧指令値Ｖ0＊から（６）式で求められる。 In FIG. 8, when the period for outputting V1 is t1, the period for outputting V2 is t2, and the period for outputting V5 is t5, in step 6, t1, t2, and t5 are dq axis voltage command values Vd *, Vq. * And the zero-phase voltage command value V0 *.
Figure JPOXMLDOC01-appb-M000006
Figure JPOXMLDOC01-appb-M000006
　なお、ＶaとＶbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図８の固定座標に変換した電圧指令値である。 Note that Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　また、ステップ８においてｔ1とｔ2とｔ5は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、零相電圧指令値Ｖ0＊から（７）式で求められる。 In step 8, t1, t2 and t5 are obtained from the dq-axis voltage command values Vd * and Vq * and the zero-phase voltage command value V0 * by the equation (7).
Figure JPOXMLDOC01-appb-M000007
Figure JPOXMLDOC01-appb-M000007
　なお、ＶaとＶbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図８の固定座標に変換した電圧指令値である。 Note that Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　図９は、図５のフローチャートにおけるステップ９の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。 FIG. 9 is a diagram showing the processing of step 9 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
　以下、本領域においてＮ１モードのパルス幅を算出する方法を述べる。なお、他の領域においては図９のベクトル図を６０°回転させ、極性を反転させることによって算出することができる。 Hereinafter, a method for calculating the pulse width of the N1 mode in this region will be described. In other regions, calculation can be performed by rotating the vector diagram of FIG. 9 by 60 ° and inverting the polarity.
　図９において、Ｖ1を出力する期間をｔ1、Ｖ2を出力する期間をｔ2、V6を出力する期間をｔ6、とすると、ｔ1とｔ2とｔ6は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、零相電圧指令値Ｖ0＊から（８）式で求められる。 In FIG. 9, if the period for outputting V1 is t1, the period for outputting V2 is t2, and the period for outputting V6 is t6, t1, t2 and t6 are dq axis voltage command values Vd *, Vq *, It can be obtained from the zero phase voltage command value V0 * by the equation (8).
Figure JPOXMLDOC01-appb-M000008
Figure JPOXMLDOC01-appb-M000008
　なお、ＶaとVbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図９の固定座標に変換した電圧令値である。 Note that Va and Vb are voltage command values obtained by converting the dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　図１０は、図５のフローチャートにおけるステップ１１の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。 FIG. 10 is a diagram showing the processing of step 11 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
　以下、本領域においてＮ２モードのパルス幅を算出する方法を述べる。なお、他の領域においては図１０のベクトル図を６０°回転させ、極性を反転させることによって算出することができる。 Hereinafter, a method for calculating the pulse width of the N2 mode in this region will be described. In other regions, the calculation can be performed by rotating the vector diagram of FIG. 10 by 60 ° and inverting the polarity.
　図１０において、Ｖ1を出力する期間をｔ1、Ｖ2を出力する期間をｔ2、V7を出力する期間をｔ7、とすると、ｔ1とｔ2とｔ7は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、零相電圧指令値Ｖ0＊から（９）式で求められる。 In FIG. 10, if the period for outputting V1 is t1, the period for outputting V2 is t2, and the period for outputting V7 is t7, t1, t2 and t7 are dq axis voltage command values Vd *, Vq *, It can be obtained from the zero phase voltage command value V0 * by the equation (9).
Figure JPOXMLDOC01-appb-M000009
Figure JPOXMLDOC01-appb-M000009
　なお、ＶaとVbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図１０の固定座標に変換した電圧指令値である。 Note that Va and Vb are voltage command values obtained by converting the dq axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　図１１は、図５のフローチャートにおけるステップ１３およびステップ１５の処理を示す図であり、図３のベクトル図において太点線で示した六角形部分を示している。 FIG. 11 is a diagram showing the processing of step 13 and step 15 in the flowchart of FIG. 5, and shows the hexagonal portion indicated by the thick dotted line in the vector diagram of FIG.
　以下、本領域においてＮ３モードとＮ４モードのパルス幅を算出する方法を述べる。なお、他の領域においては図１１のベクトル図を６０°回転させ、極性を反転させることによって算出することができる。 Hereinafter, a method for calculating the pulse widths of the N3 mode and the N4 mode in this area will be described. In other areas, the calculation can be performed by rotating the vector diagram of FIG. 11 by 60 ° and inverting the polarity.
　図１１において、Ｖ1を出力する期間をｔ1、Ｖ2を出力する期間をｔ2、V8を出力する期間をｔ8、とすると、ステップ１３においてｔ1とｔ2とｔ8は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、零相電圧指令値Ｖ0＊から（１０）式で求められる。 In FIG. 11, if the period for outputting V1 is t1, the period for outputting V2 is t2, and the period for outputting V8 is t8, t1, t2 and t8 in step 13 are the dq axis voltage command values Vd * and Vq. * And the zero-phase voltage command value V0 *.
Figure JPOXMLDOC01-appb-M000010
Figure JPOXMLDOC01-appb-M000010
　なお、ＶaとVbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図１１の固定座標に変換した電圧指令値である。 Note that Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　また、ステップ１５においてｔ1とｔ2とｔ8は、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊と、
零相電圧指令値Ｖ0＊から（１１）式で求められる。 In step 15, t1, t2, and t8 are dq axis voltage command values Vd *, Vq *,
It is obtained from the zero-phase voltage command value V0 * by the equation (11).
Figure JPOXMLDOC01-appb-M000011
Figure JPOXMLDOC01-appb-M000011
　なお、ＶaとVbは、ｄｑ軸電圧指令値Ｖｄ＊、Ｖｑ＊を図１１の固定座標に変換した電圧指令値である。 Note that Va and Vb are voltage command values obtained by converting dq-axis voltage command values Vd * and Vq * into fixed coordinates in FIG.
　以上の通り、本実施形態においては、零相電圧の振幅とパルス幅を制御することで、各相のインバータが出力する電圧のパルス幅がキャリア周期を超えない範囲で振幅の小さい零相電圧を出力し、スッチング回数の低減によりスイッチング損失が低減される、という効果が得られる。 As described above, in the present embodiment, by controlling the amplitude and pulse width of the zero-phase voltage, the zero-phase voltage having a small amplitude can be obtained within a range in which the pulse width of the voltage output from the inverter of each phase does not exceed the carrier cycle. As a result, the switching loss can be reduced by reducing the number of times of switching.
１…インバータ制御装置、１０…電流指令演算部、２０…ｄｑ軸電流制御部、３０…三相変換部、４０…スイッチング信号生成部、５０…ｄｑ変換部、６０…零相電流算出部、７０…零相電流制御部、８０…速度変換部、１００…インバータ、１１０…Ｕ相フルブリッジインバータ、１１０ａ…スイッチング素子、１１０ｂ…スイッチング素子、１１０ｃ…スイッチング素子、１１０ｄ…スイッチング素子、１１１…Ｖ相フルブリッジインバータ、１１１ａ…スイッチング素子、１１１ｂ…スイッチング素子、１１１ｃ…スイッチング素子、１１１ｄ…スイッチング素子、１１２…Ｗ相フルブリッジインバータ、１１２ａ…スイッチング素子、１１２ｂ…スイッチング素子、１１２ｃ…スイッチング素子、１１２ｄ…スイッチング素子、２００…モータ、２０１…Ｕ相巻線、２０２…Ｖ相巻線、２０３…Ｗ相巻線、２１０…位置センサ、２２０…電流センサ、ｉｕ…Ｕ相電流、ｉｖ…Ｖ相電流、ｉｗ…Ｗ相電流、ｉｄ＊…ｄ軸電流指令値、ｉｑ＊…ｑ軸電流指令値、ｉｄ…ｄ軸電流検出値、ｉｑ…ｑ軸電流検出値、ｉ0…零相電流、Ｔ＊…トルク指令値、Ｖｄｃ…直流電源電圧、Ｖｕ…Ｕ相出力電圧、Ｖｖ…Ｖ相出力電圧、Ｖｗ…Ｗ相出力電圧、Ｖ0…零相出力電圧、Ｖｄ＊…ｄ軸電圧指令値、Ｖｑ＊…ｑ軸電圧指令値、Ｖ0＊…零相電圧指令値、ω…角速度、θ…回転子位置 DESCRIPTION OF SYMBOLS 1 ... Inverter control apparatus, 10 ... Current command calculating part, 20 ... dq axis current control part, 30 ... Three phase conversion part, 40 ... Switching signal production | generation part, 50 ... dq conversion part, 60 ... Zero phase current calculation part, 70 ... Zero-phase current control unit, 80 ... Speed converter, 100 ... Inverter, 110 ... U-phase full bridge inverter, 110a ... Switching element, 110b ... Switching element, 110c ... Switching element, 110d ... Switching element, 111 ... V-phase full Bridge inverter, 111a ... switching element, 111b ... switching element, 111c ... switching element, 111d ... switching element, 112 ... W-phase full bridge inverter, 112a ... switching element, 112b ... switching element, 112c ... switching element, 112d ... switching element , 20 ... Motor, 201 ... U phase winding, 202 ... V phase winding, 203 ... W phase winding, 210 ... Position sensor, 220 ... Current sensor, iu ... U phase current, iv ... V phase current, iw ... W phase Current, id * ... d-axis current command value, iq * ... q-axis current command value, i d ... d-axis current detection value, i q ... q-axis current detection value, i0 ... zero-phase current, T * ... torque command value , Vdc ... DC power supply voltage, Vu ... U-phase output voltage, Vv ... V-phase output voltage, Vw ... W-phase output voltage, V0 ... zero-phase output voltage, Vd * ... d-axis voltage command value, Vq * ... q-axis voltage Command value, V0 *: Zero-phase voltage command value, ω: Angular velocity, θ: Rotor position

Claims (4)

  1. 　各相の巻線が独立して結線された電動機を制御するとともに、入力されたトルク指令に基づき電動機へ印加する電圧を制御するためのＰＷＭ制御信号を出力する電動機の制御装置において、
    　前記電動機の交流電流から求められる零相電流を低減するための零相電圧指令に基づき、
    　出力電圧の総和からなる零相電圧のパルス幅及び振幅を制御する電動機の制御装置。     In the motor control device that controls the motor in which the windings of each phase are independently connected, and outputs a PWM control signal for controlling the voltage applied to the motor based on the input torque command,
    Based on the zero phase voltage command for reducing the zero phase current obtained from the alternating current of the motor,
    A motor control device that controls the pulse width and amplitude of a zero-phase voltage composed of the sum of output voltages.
  2. 　請求項１に記載の電動機の制御装置であって、
    　零相電圧の振幅が小さくなるように零相電圧のパルス幅及び振幅を制御する制御装置。     The motor control device according to claim 1,
    A control device that controls the pulse width and amplitude of the zero-phase voltage so that the amplitude of the zero-phase voltage is reduced.
  3. 　請求項１に記載の電動機の制御装置であって、
    　零相電圧出力時のスイッチング回数を低減する所定のパルス幅と振幅の零相電圧を出力する制御装置。     The motor control device according to claim 1,
    A control device that outputs a zero-phase voltage having a predetermined pulse width and amplitude that reduces the number of switching times when outputting a zero-phase voltage.
  4. 　請求項１乃至３のいずれかに記載の電動機の制御装置を備えた電動車両。 An electric vehicle comprising the motor control device according to any one of claims 1 to 3.
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CN101902860A (en) * 2009-11-21 2010-12-01 英飞特电子（杭州）有限公司 Multipath constant-current driving circuit
CN102064558A (en) * 2010-11-30 2011-05-18 张家港市沙洲特种变压器制造有限公司 Three-phase electromagnetic balanced electricity-saving appliance
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