JP2010154562A - Ad converter, solid state imaging device, and semiconductor device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide: an AD converter employing a DA converter capable of controlling a slope of a reference signal for single-slope integration AD conversion with high accuracy; a solid state imaging device; and a semiconductor device. <P>SOLUTION: A gain control is applied to a first DA conversion section 302 for achieving a reference signal based on a DA output current I<SB>gain</SB>achieved by a second DA conversion section 304. Each of current source cells 353, 355 within a current cell section 350 has an own operation current defined by a bias voltage V<SB>bias</SB>corresponding to the DA output current I<SB>gain</SB>. In the first DA conversion section 302, a lower bit control section 330 performs a frequency-division operation and selects a low-order current source cell 533 of a weighted current value using a (1/2^k) frequency-divided clock signal. An upper bit control section 340 employs, as a shift clock signal, a predetermined frequency-divided clock signal produced through the frequency-division operation of the lower bit control section 330, sequentially changes a shift output of each shift register in a shift register section 342 into H, and, using this shift output, sequentially selects an upper order current source cell 355 of the same weighted current. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to an AD conversion device including a DA (Digital to Analogue) conversion device that converts a digital signal into an analog signal, and a semiconductor device such as an imaging device. For example, the present invention relates to a DA conversion mechanism using a current source cell matrix including current source cells having a cell arrangement structure arranged in a matrix.

  In electronic devices, various DA converters that convert digital signals into analog signals are used.

  For example, a physical quantity distribution detecting semiconductor device in which a plurality of unit components (for example, pixels) that are sensitive to electromagnetic waves input from outside such as light and radiation are arranged in a line or matrix form in various fields. It is used.

  For example, in the field of video equipment, CCD (Charge Coupled Device) type, MOS (Metal Oxide Semiconductor) type or CMOS (Complementary Metal-oxide Semiconductor) type solid-state imaging devices that detect light (an example of electromagnetic waves) in physical quantities are used. It is used. These read out, as an electrical signal, a physical quantity distribution converted into an electrical signal by a unit component (a pixel in a solid-state imaging device). Here, “solid” means made of semiconductor.

  Further, in some solid-state imaging devices, an amplifying solid-state imaging device (APS; Active Pixel Sensor) that has a driving transistor for amplification in a pixel signal generation unit that generates a pixel signal corresponding to the signal charge generated in the charge generation unit. There is an amplification type solid-state imaging device including a pixel having a configuration (also called a gain cell). For example, many CMOS solid-state imaging devices have such a configuration.

  In such an amplification type solid-state imaging device, in order to read out a pixel signal to the outside, address control is performed on a pixel unit in which a plurality of unit pixels are arranged, and signals from individual unit pixels are arbitrarily selected. I am trying to read it out. That is, the amplification type solid-state imaging device is an example of an address control type solid-state imaging device.

  For example, an amplification type solid-state imaging device which is a kind of XY address type solid-state imaging device in which unit pixels are arranged in a matrix form an active element (MOS transistor) such as a MOS structure in order to give the pixel itself an amplification function. ) To form a pixel. That is, signal charges (photoelectrons) accumulated in a photodiode which is a photoelectric conversion element are amplified by the active element and read out as image information.

  In this type of XY address type solid-state imaging device, for example, a plurality of pixel transistors are arranged in a two-dimensional matrix to form a pixel unit, and a signal charge corresponding to incident light for each line (row) or each pixel. Accumulation is started, and a current or voltage signal based on the accumulated signal charge is sequentially read out from each pixel by addressing. Here, in the MOS (including CMOS) type, as an example of address control, a method of simultaneously accessing one row, reading out pixel signals from the pixel unit in units of rows, and outputting them to the outside is often used. In addition, there is a case where an analog pixel signal read from the pixel unit is converted into a digital signal by an analog-digital converter (AD converter; Analog Digital Converter) and then output to the outside ( For example, see Patent Documents 1 and 2).

JP 2000-152082 A JP 2002-232291 A

  As described in Patent Documents 1 and 2 above, there are various AD conversion methods from the viewpoint of circuit scale, processing speed, resolution, and the like. Among them, conversion into analog unit signals and digital signals is performed. A so-called single slope that obtains a digital signal of a unit signal based on a count value at the time when the comparison processing is completed. There is an AD conversion method called an integral type or a ramp signal comparison type, and a DA converter is sometimes used to generate a monotonically changing reference signal.

  On the other hand, in the AD conversion method referred to as a single slope integration type, digital data can be obtained as a result of applying gain adjustment to an analog signal subject to AD conversion processing by changing the slope of the reference signal supplied to the comparison circuit. it can. Since the adjustment accuracy of the slope of the reference signal is related to the adjustment accuracy of the gain adjustment with respect to the analog signal, a high accuracy is required to adjust the change rate of the reference signal when a high-accuracy gain adjustment is required. Become.

  In adjusting the change rate of the reference signal, for example, it may be possible to deal with the reference signal generated at the standard change rate as the DA conversion output by analog gain adjustment by gain adjustment for the analog amplifier. . However, it is difficult to adjust the analog gain in terms of accuracy. For example, it is difficult to make fine adjustment with accuracy of 12 bits or more or to reduce variations in the reference potential.

  The present invention has been made in view of the above circumstances, and an object thereof is to provide a mechanism capable of adjusting the gain of a DA conversion output with high accuracy. For example, it provides a mechanism that can adjust the slope of a reference signal with high accuracy when a reference signal that changes monotonically and in a certain direction used in the so-called single slope integration type AD conversion method is obtained as a DA conversion output. The purpose is to do.

According to the present invention, a reference signal generation unit that generates a reference signal for converting an analog signal into a digital signal, a comparison unit that compares the analog signal and the reference signal generated by the reference signal generation unit, In parallel with the comparison process in the comparison unit, the AD conversion apparatus includes a counter unit that performs a count process with a predetermined count clock and holds a count value when the comparison process in the comparison unit is completed, The reference signal generator includes a first DA converter that obtains the reference signal according to the value of the digital input signal, and a second DA that obtains an analog gain control output signal according to the value of the digital gain control input signal. A reference signal generated by the first DA converter based on the gain control output signal generated by the second DA converter. AD conversion apparatus characterized by adjusting the emission is provided.
According to the present invention, there is provided a solid-state imaging device including a plurality of pixels and an AD converter that converts an analog pixel signal read from the plurality of pixels into a digital signal. A reference signal generation unit that generates a reference signal for converting an analog signal into a digital signal, a comparison unit that compares the analog signal and the reference signal generated by the reference signal generation unit, and a comparison unit In parallel with the comparison process, the counter unit performs a count process with a predetermined count clock and holds a count value at the time when the comparison process in the comparison unit is completed, and the reference signal generation unit includes a digital input signal A first D / A converter that obtains the reference signal according to a value, and a second D that obtains an analog gain control output signal according to the value of the digital gain control input signal A solid-state imaging device comprising: a converter; and adjusting a gain of the reference signal generated by the first DA converter based on a gain control output signal generated by the second DA converter An apparatus is provided.
Furthermore, according to the present invention, a charge generation unit that generates a charge corresponding to an incident electromagnetic wave, and a unit signal generation unit that generates an analog unit signal corresponding to the charge generated by the charge generation unit are included in the unit component. And a reference signal generation unit that generates a reference signal for converting the unit signal into a digital signal, and the unit signal and the reference signal as a functional element that converts the unit signal into a digital signal. A comparison unit that compares the reference signal generated by the generation unit, and a count value at the time when the comparison process in the comparison unit is completed by performing a count process with a predetermined count clock in parallel with the comparison process in the comparison unit And a counter unit for holding a physical quantity distribution, wherein the reference signal generation unit converts the value of the digital input signal to A first DA converter that obtains the reference signal and a second DA converter that obtains an analog gain control output signal corresponding to the value of the digital gain control input signal, and the second DA converter. The semiconductor device is characterized in that the gain of the reference signal generated by the first DA converter is adjusted based on the gain control output signal generated by step (1).

  That is, gain control is applied to the first DA converter that obtains the analog output signal based on the gain control output signal obtained by the second DA converter. Since the gain of the analog output signal generated by the first DA converter is adjusted by digital control, it is possible to easily achieve high accuracy of adjustment. When the reference signal used in the single slope integration type AD conversion method is generated as the analog output signal by the first DA conversion unit, the slope of the reference signal can be adjusted. On the other hand, gain adjustment can be added.

  Preferably, both the first and second DA converters adopt a current output type DA converter circuit configuration.

  In particular, the first DA converter may be configured to select and control the current source cell by dividing the digital input into upper bits and lower bits. For this reason, first, the upper current source cell unit that is selectively controlled by the upper bit control unit is provided with a plurality of upper current source cells that output the same constant current, and the lower current source that is selectively controlled by the lower bit control unit. The cell portion is provided with a lower constant current source cell that outputs a bit weighting current.

  Here, the lower bit control unit uses a frequency divider that performs a frequency dividing operation based on the input count clock to generate a frequency-divided clock that is a power of 2 as bit data. A peripheral clock (that is, bit data) is used as a selection control signal to select a lower current source cell superimposed on a corresponding current value. In addition, the upper bit control unit sequentially activates the shift register shift output using a signal indicating a carry or a carry in the frequency division operation of the lower bit control unit as a shift clock, and selectively controls the shift output. The upper current source cells are sequentially selected by using them as signals.

  The lower bit control unit performs a frequency division operation based on the count clock, and the upper bit control unit uses a signal indicating a carry or a carry generated accompanying the frequency division operation as a shift clock to perform a shift operation. Therefore, the lower bit control unit and the upper bit control unit perform linked (synchronized) operations instead of independent operations, and the upper bit control unit reliably performs the current corresponding to the next bit data. The source cell can be selected.

  According to the present invention, gain control is applied to the first DA converter that obtains the analog output signal based on the gain control output signal obtained by the second DA converter. The accuracy of the gain control output signal generated by the second DA conversion unit depends on the accuracy of the digital gain control input signal, but can be controlled by a digital value. For example, the number of bits of the digital gain control input signal is increased. Therefore, high accuracy can be easily realized.

It is a schematic block diagram of the CMOS solid-state imaging device which is one Embodiment of the semiconductor device which concerns on this invention. It is a figure explaining the function of the DA converter circuit (DAC) of the reference signal production | generation part used in a solid-state imaging device. 3 is a timing chart for explaining signal acquisition difference processing, which is a basic operation in the column AD circuit of the solid-state imaging device shown in FIG. 1. It is a figure which shows the basic structural example of a reference signal production | generation part. It is a figure which shows the specific structural example of a 1st DA conversion part. It is a figure which shows the specific structural example of a 2nd DA conversion part. It is a figure explaining the relationship between an external input code and a current source cell. It is the figure which showed the basic structural example (conceptual figure of a basic current source cell) of each current source cell provided in a current source cell part. It is a figure explaining the detail of the connection mode of a 1st DA converter and a 2nd DA converter. It is a figure explaining the whole operation | movement in a DA converter. It is FIG. (1) explaining the example of the gain adjustment at the time of AD conversion process using a 2nd DA converter. It is FIG. (2) explaining the example of the gain adjustment at the time of AD conversion process using a 2nd DA converter.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the following, a case where a CMOS image sensor, which is an example of an XY address type solid-state imaging device, is used as a device will be described as an example. The CMOS image sensor will be described on the assumption that all pixels are made of NMOS or PMOS.

  However, this is merely an example, and the target device is not limited to a MOS imaging device. All the semiconductor device for physical quantity distribution detection in which a plurality of unit components that are sensitive to electromagnetic waves input from outside such as light and radiation are arranged in a line or matrix form, and all implementations described later. Forms are applicable as well.

<Configuration of solid-state imaging device>
FIG. 1 is a schematic configuration diagram of a CMOS solid-state imaging device (CMOS image sensor) which is an embodiment of a semiconductor device according to the present invention. This CMOS solid-state imaging device is also an aspect of electronic equipment.

  The solid-state imaging device 1 has a pixel unit in which a plurality of pixels including a light receiving element (an example of a charge generation unit) that outputs a signal corresponding to an incident light amount is arranged in rows and columns (that is, in a two-dimensional matrix form). The signal output from each pixel is a voltage signal, and a CDS (Correlated Double Sampling) processing function unit, a digital conversion unit (ADC), etc. are provided in parallel in a column. It is.

  “The CDS processing function unit and the digital conversion unit are provided in parallel with the column” means that a plurality of CDS processing function units and digital conversion units are provided substantially in parallel with the vertical signal line 19 in the vertical column. Means that Each of the plurality of functional units is arranged only on one edge side in the column direction with respect to the pixel unit (imaging unit) 10 (an output side arranged on the lower side of the drawing) when the device is viewed in plan view. Or one end side in the column direction with respect to the pixel portion 10 (the output side disposed on the lower side of the figure) and the other end side on the opposite side (see FIG. May be arranged separately on the upper side). In the latter case, it is preferable that the horizontal scanning unit that performs readout scanning (horizontal scanning) in the row direction is also arranged separately on each edge side so that each can operate independently.

  For example, as a typical example in which a CDS processing function unit and a digital conversion unit are provided in parallel in a column, a CDS processing function unit and a digital conversion unit are arranged for each vertical column in a portion called a column area provided on the output side of the imaging unit. And is a column type that sequentially reads out to the output side. In addition to the column type, a configuration in which one CDS processing function unit or digital conversion unit is assigned to a plurality of adjacent (for example, two) vertical signal lines 19 (vertical columns), or every N (N is It is also possible to adopt a form in which one CDS processing function unit or digital conversion unit is assigned to N vertical signal lines 19 (vertical columns) of a positive integer (with N−1 lines in between).

  Except for the column type, in any form, since a plurality of vertical signal lines 19 (vertical columns) commonly use one CDS processing function unit and digital conversion unit, they are supplied from the pixel unit 10 side. A switching circuit (switch) that supplies pixel signals for a plurality of columns to one CDS processing function unit or digital conversion unit is provided. Depending on the subsequent processing, it is necessary to take measures such as providing a memory for holding the output signal.

  In any case, the signal processing of each pixel signal is read out in units of pixel columns by adopting a form in which one CDS processing function unit or digital conversion unit is assigned to a plurality of vertical signal lines 19 (vertical columns). By performing the processing later, the configuration in each unit pixel can be simplified and the number of pixels of the image sensor can be reduced, the size can be reduced, and the cost can be reduced as compared with the case where the same signal processing is performed in each unit pixel.

  In addition, since a plurality of signal processing units arranged in parallel in a column can simultaneously process pixel signals for one row, one CDS processing function unit or digital conversion unit is provided on the output circuit side or outside the device. Therefore, the signal processing unit can be operated at a low speed as compared with the case where processing is performed, which is advantageous in terms of power consumption, bandwidth performance, noise, and the like. In other words, when the power consumption and bandwidth performance are the same, the entire sensor can be operated at high speed.

  In the case of a column type configuration, it can be operated at a low speed, which is advantageous in terms of power consumption, bandwidth performance, noise, and the like, and has an advantage that a switching circuit (switch) is unnecessary. In the following embodiments, this column type will be described unless otherwise specified.

  As shown in FIG. 1, the solid-state imaging device 1 includes a pixel unit (imaging unit) 10 in which a plurality of unit pixels 3 having a substantially square pixel shape are arranged in rows and columns (that is, a square lattice shape), and a pixel unit. 10 includes a drive control unit 7, a column processing unit 26, a reference signal generation unit 27 that supplies a reference voltage for AD conversion to the column processing unit 26, and an output circuit 28.

  In addition, an AGC (Auto Gain Control) circuit having a signal amplification function or the like can be provided in the same semiconductor region as the column processing unit 26 as needed before or after the column processing unit 26. When AGC is performed before the column processing unit 26, analog amplification is performed. When AGC is performed after the column processing unit 26, digital amplification is performed. If the n-bit digital data is simply amplified, the gradation may be lost. Therefore, it is preferable to perform digital conversion after amplification by analog.

  The drive control unit 7 has a control circuit function for sequentially reading signals from the pixel unit 10. For example, the drive control unit 7 includes a horizontal scanning circuit (column scanning circuit) 12 that controls column addresses and column scanning, a vertical scanning circuit (row scanning circuit) 14 that controls row addresses and row scanning, and an internal clock. And a communication / timing control unit 20 having a function such as generation.

  In the drawing, as shown by a dotted line in the vicinity of the communication / timing control unit 20, the clock conversion unit 23 is an example of a high-speed clock generation unit that generates a pulse having a clock frequency higher than the input clock frequency. May be provided. The communication / timing control unit 20 generates an internal clock based on the input lock (master clock) CLK0 input via the terminal 5a or the high-speed clock generated by the clock conversion unit 23.

  By using a signal derived from the high-speed clock generated by the clock converter 23, AD conversion processing and the like can be operated at high speed. Also, motion extraction and compression processing requiring high-speed calculation can be performed using a high-speed clock. Also, the parallel data output from the column processing unit 26 can be converted into serial data and the video data D1 can be output outside the device. By doing so, it is possible to adopt a configuration in which high-speed operation output is performed with a smaller number of terminals than the number of bits of AD-converted digital data.

  The clock converter 23 includes a multiplier circuit that generates a pulse having a clock frequency faster than the input clock frequency. The clock conversion unit 23 receives the low-speed clock CLK2 from the communication / timing control unit 20, and generates a clock having a frequency twice or more higher based on the low-speed clock CLK2. As a multiplication circuit of the clock converter 23, a k1 multiplication circuit may be provided when k1 is a multiple of the frequency of the low-speed clock CLK2, and various known circuits can be used.

  In FIG. 1, for simplification, some of the rows and columns are omitted, but in reality, in each row and each column, dozens to thousands of unit pixels 3 are arranged, and the pixel unit 10 includes Composed. The unit pixel 3 is typically composed of a photodiode as a light receiving element (charge generation unit) and an in-pixel amplifier having an amplifying semiconductor element (for example, a transistor).

  As the intra-pixel amplifier, for example, a floating diffusion amplifier configuration is used. As an example, with respect to the charge generation unit, a read selection transistor that is an example of a charge readout unit (transfer gate unit / read gate unit), a reset transistor that is an example of a reset gate unit, a vertical selection transistor, and a floating diffusion As a CMOS sensor having a source follower-amplifying transistor, which is an example of a detection element for detecting a change in potential, a sensor composed of four general-purpose transistors can be used.

  Alternatively, as described in Japanese Patent No. 2708455, an amplifying transistor connected to a drain line (DRN) for amplifying a signal voltage corresponding to the signal charge generated by the charge generating unit, and the charge generating unit It is also possible to use a transistor composed of three transistors, each having a reset transistor for resetting and a read selection transistor (transfer gate portion) scanned from a vertical shift register via a transfer wiring (TRF). .

  In addition to an effective image region (effective portion) that is an effective region for capturing an image, the pixel unit 10 includes a reference pixel region that provides optical black arranged around the effective image region. As an example, reference pixels that give optical black for several rows (for example, 1 to 10 rows) are arranged above and below in the vertical column direction, and several pixels to several tens in the horizontal direction including the effective image region 10a. Reference pixels that provide optical black for pixels (for example, 3 to 40 pixels) are arranged.

  The reference pixel for providing optical black is shielded on the light receiving surface side so that light does not enter a charge generation unit made of a photodiode or the like. The pixel signal from this reference pixel is used for the black reference of the video signal.

  Further, in the solid-state imaging device 1, the pixel unit 10 is adapted for color imaging. That is, color separation composed of a combination of a plurality of color filters for capturing a color image on a light receiving surface on which electromagnetic waves (light in this example) of each charge generation unit (such as a photodiode) in the pixel unit 10 is incident. Any color filter of the filter is provided.

  The illustrated example uses a basic color filter of a so-called Bayer array, and unit pixels 3 arranged in a square lattice are three color colors of red (R), green (G), and blue (B). In order to correspond to the filter, the repeating unit of the color separation filter is arranged by 2 pixels × 2 pixels to constitute the pixel unit 10.

  For example, a first color pixel for sensing a first color (red; R) is arranged in an odd-numbered row and an odd-numbered column, and a second color (green; G; ) Is arranged, and the third color pixel for sensing the third color (blue; B) is arranged in the even-numbered row and the even-numbered column, and is different for each row. Further, two color pixels of R / G or G / B are arranged in a checkered pattern.

  In the color arrangement of the basic color filter in such a Bayer arrangement, two colors of R / G or G / B are repeated every two in both the row direction and the column direction.

  Further, as other components of the drive control unit 7, a horizontal scanning circuit 12, a vertical scanning circuit 14, and a communication / timing control unit 20 are provided. The horizontal scanning circuit 12 has a function of a reading scanning unit that reads a count value from the column processing unit 26. Each element of these drive control units 7 is formed integrally with a pixel unit 10 in a semiconductor region such as single crystal silicon using a technique similar to a semiconductor integrated circuit manufacturing technique, and is a solid-state imaging which is an example of a semiconductor system It is configured as an element (imaging device).

  The unit pixel 3 includes a column processing unit 26 in which a vertical scanning circuit 14 is provided via a row control line 15 for row selection, and a column AD circuit 25 is provided for each vertical column via a vertical signal line 19; Each is connected. Here, the row control line 15 indicates the entire wiring that enters the pixel from the vertical scanning circuit 14.

  The horizontal scanning circuit 12 and the vertical scanning circuit 14 include a decoder as will be described later, and start a shift operation (scanning) in response to control signals CN1 and CN2 given from the communication / timing control unit 20. It has become. Therefore, the row control line 15 includes various pulse signals (for example, a reset pulse RST, a transfer pulse TRF, a DRN control pulse DRN, etc.) for driving the unit pixel 3.

  Although not shown, the communication / timing control unit 20 is a master via a functional block of a timing generator TG (an example of a read address control device) that supplies a clock signal necessary for the operation of each unit and a pulse signal of a predetermined timing, and a terminal 5a. A communication interface functional block that receives the clock CLK0, receives data DATA for instructing an operation mode and the like via the terminal 5b, and outputs data including information of the solid-state imaging device 1;

  For example, the horizontal address signal is output to the horizontal decoder 12a and the vertical address signal is output to the vertical decoder 14a, and each decoder 12a, 14a receives it and selects a corresponding row or column.

  At this time, since the unit pixels 3 are arranged in a two-dimensional matrix, analog pixel signals generated by the pixel signal generation unit 5 and output in the column direction via the vertical signal lines 19 are arranged in a row unit (column parallel). (In) Scan (access) to read (vertical) scan, and then access the row direction, which is the arrangement direction of vertical columns, and read out pixel signals (in this example, digitized pixel data) to the output side (horizontal) scan By performing reading, it is preferable to speed up reading of pixel signals and pixel data. Of course, not only scanning reading but also random access for reading out only the information of the necessary unit pixel 3 is possible by directly addressing the unit pixel 3 to be read out.

  In the communication / timing control unit 20, a clock CLK1 having the same frequency as the master clock (master clock) CLK0 input via the terminal 5a, a clock obtained by dividing the clock CLK1, or a low-speed clock obtained by further dividing the device are used as devices. For example, the horizontal scanning circuit 12, the vertical scanning circuit 14, and the column processing unit 26. Hereinafter, the clocks divided by two and the clocks having a frequency lower than that are collectively referred to as a low-speed clock CLK2.

  The vertical scanning circuit 14 selects a row of the pixel unit 10 and supplies a necessary pulse to the row. For example, a vertical decoder 14a that defines a readout row in the vertical direction (selects a row of the pixel unit 10), and a row control line 15 for the unit pixel 3 on the readout address (in the row direction) defined by the vertical decoder 14a. And a vertical drive circuit 14b for driving by supplying a pulse. Note that the vertical decoder 14a selects a row for electronic shutter, in addition to a row from which a signal is read.

  The horizontal scanning circuit 12 sequentially selects the column AD circuit 25 of the column processing unit 26 in synchronization with the low-speed clock CLK2, and guides the signal to a horizontal signal line (horizontal output line) 18. For example, a horizontal decoder 12a that defines a horizontal readout column (selects each column AD circuit 25 in the column processor 26), and each of the column processors 26 according to a read address defined by the horizontal decoder 12a. A horizontal drive circuit 12b for guiding a signal to the horizontal signal line 18. For example, if the number of horizontal signal lines 18 is n (n is a positive integer) handled by the column AD circuit 25, for example, 10 (= n) bits, 10 horizontal signal lines 18 are arranged corresponding to the number of bits. .

  In the solid-state imaging device 1 having such a configuration, the pixel signal output from the unit pixel 3 is supplied to the column AD circuit 25 of the column processing unit 26 via the vertical signal line 19 for each vertical column.

  Each column AD circuit 25 of the column processing unit 26 receives a pixel signal for one column and processes the signal. For example, each column AD circuit 25 has an ADC (Analog Digital Converter) circuit that converts an analog signal into, for example, 10-bit digital data using, for example, a low-speed clock CLK2.

  As the AD conversion processing in the column processing unit 26, a method is adopted in which analog signals held in parallel in units of rows are subjected to AD conversion in parallel for each row using the column AD circuit 25 provided for each column. In this case, for example, Japanese Patent Publication No. 2532374 and academic literature “CMOS image sensor equipped with column AD converter without inter-column FPN” (film science technique, IPU2000-57, pp. 79-84), etc. A single slope integration type (or ramp signal comparison type) AD conversion method shown in FIG. Since this method can realize an AD converter with a simple configuration, it has a feature that the circuit scale does not increase even if it is provided in parallel.

  Although the details of the configuration of the ADC circuit will be described later, the analog processing target signal is converted into digital data based on the time from the start of conversion until the reference voltage RAMP matches the processing target signal voltage. As a mechanism for this, in principle, the ramp-shaped reference voltage RAMP is supplied to the comparator (voltage comparator), and at the same time, counting with the clock signal is started and input via the vertical signal line 19. By comparing the analog pixel signal with the reference voltage RAMP and counting until a pulse signal is obtained, AD conversion is performed.

  At this time, by devising the circuit configuration, the signal level (noise level) immediately after the pixel reset and true (for the voltage mode pixel signal input via the vertical signal line 19 as well as AD conversion) are true ( It is possible to perform processing for obtaining a difference from the signal level Vsig (in accordance with the amount of received light). Thereby, it is possible to remove a noise signal component called fixed pattern noise (FPN) or reset noise.

  The pixel data digitized by the column AD circuit 25 is transmitted to the horizontal signal line 18 through a horizontal selection switch (not shown) driven by a horizontal selection signal from the horizontal scanning circuit 12, and further input to the output circuit 28. The Note that 10 bits is an example, and other bit numbers such as less than 10 bits (for example, 8 bits) and more than 10 bits (for example, 14 bits) may be used.

  With such a configuration, pixel signals are sequentially output for each vertical column for each row from the pixel unit 10 in which light receiving elements as charge generation units are arranged in a matrix. Then, one image corresponding to the pixel unit 10 in which the light receiving elements are arranged in a matrix, that is, a frame image, is shown as a set of pixel signals of the entire pixel unit 10.

<Details of column AD circuit and reference signal generator>
The reference signal generation unit 27 is a DA conversion circuit (DAC; Digital Analog) that is a functional element that generates a reference signal for AD conversion in accordance with the color type and arrangement of the color filters constituting the color separation filter in the pixel unit 10. Converter) is provided separately. Although details will be described later, the reference signal generation unit 27 of the present embodiment employs a current output type DA converter circuit.

  When the pixel unit 10 (device) to be used is determined, the color type and arrangement of the color filters in the color separation filter are determined, and the number of the color filter at an arbitrary position in the two-dimensional grid position is uniquely specified. Can do. Each repetition cycle in the row direction and the column direction of the color filter is also uniquely determined by the arrangement, and one column to be processed by each column AD circuit 25 provided in parallel with the column is used in the color separation filter. There are only pixel signals of a predetermined combination of colors that are less than the total number of colors that are determined and that are determined by the repetition cycle.

  In this embodiment, paying attention to this property, when configuring an AD conversion circuit with a comparison circuit and a counter, individual reference corresponding to color, which is a functional element that generates a reference signal for AD conversion supplied to the comparison circuit A DA conversion circuit as an example of a signal generation output unit is not provided for all the colors used in the color separation filter, but first, a predetermined number that exists in the repetition cycle of the color filter with respect to the row direction as a pixel signal readout unit. Only a few minutes according to the combination of color filters is used, so that the total number of color filters existing in the two-dimensional color filter repetition cycle is reduced. For example, if there is only x (x is a positive integer greater than or equal to 2) colors in any row to be processed, a reference signal for each color corresponding to the x color is supplied to the comparison circuit. It suffices to prepare x DA conversion circuits.

  Note that it is necessary to deal with switching of the processing target row from the viewpoint of supplying individual reference signals having color-related change characteristics and initial values to the comparison circuit. For this purpose, it is preferable to provide a switching mechanism for supplying a reference signal for the current processing color in each of the x DA conversion circuits in the column direction orthogonal to the row direction.

  In other words, with respect to a different direction that is different from the row direction according to the readout unit, that is, the vertical column direction, a change characteristic (specifically, a slope) corresponding to the color characteristic of the color pixel, a black reference, and a circuit offset component A color-corresponding reference signal generator that changes with an initial value defined in terms of non-color characteristics different from color characteristics such as the color characteristics of a color filter of a predetermined color that exists in the repetition cycle of the color filter in the vertical column direction. Provided for each individual DA converter circuit (reference signal generation / output unit) by the number corresponding to the combination, and selects one of the reference signals generated by the reference signal generation / output unit corresponding to the color. Thus, a selection unit for supplying to the comparison circuit is provided.

  In this case, for example, when the same color filter exists in the two-dimensional color filter repetition cycle as in the Bayer array, an individual DA conversion circuit (reference signal generation output unit) is used for the same color filter. Each can also be configured to share (shared) one color-corresponding reference signal generation unit.

  In any configuration, each DA conversion circuit, which is an example of the reference signal generation output unit, performs DA conversion in response to switching of a predetermined color combination existing in the processing target row when the processing target row is switched. The change characteristic (specifically, the slope) of the reference signal (analog reference voltage) generated by the circuit is switched and output according to the characteristic of the color filter, that is, the analog pixel signal. The initial value is set based on a viewpoint different from the color characteristics such as black reference and circuit offset component.

  By doing so, the number of reference voltage generators (corresponding to DA conversion circuits in this example) and wiring from the reference voltage generators can be made smaller than the number of color filters constituting the color separation filter. Further, it is necessary when a reference voltage generator is prepared for each color filter (see Patent Document 1), and an analog reference voltage (corresponding to the reference signal in this example) from each reference voltage generator is selectively used. Since the selection means (multiplexer) for each vertical column to be output is not required, the circuit scale can be reduced. The number of signal lines for transmitting reference signals corresponding to the color pixels to the input side of the comparator can be made smaller than the number of color components of the color filter for capturing a color image.

  Although not employed in the present embodiment, each time the processing target row is switched for each individual DA conversion circuit (reference signal generation output unit), a repeating unit of the color filter array associated with the switching is configured. The initial value based on a different viewpoint from the color characteristics such as the black characteristics and the circuit offset component, and the change characteristics corresponding to the color characteristics of the corresponding color pixel (specifically, the slope) The communication / timing control unit 20 may be set. By doing so, it is not necessary to provide a selection unit for selecting either the color-corresponding reference signal generation unit or the color-corresponding reference signal generation unit in each of the individual DA conversion circuits (reference signal generation output units).

  In other words, the idea is to change the change characteristics (specifically, the slope) and the initial value according to the change in the combination of colors that constitute the repeating unit of the color filter array that accompanies the switching each time the processing target row is switched. Thus, if the DA conversion circuit is set, there is no need to provide a selection unit that switches between the color-corresponding reference signal generation unit corresponding to each color filter and the color-corresponding reference signal generation unit according to the processing target row. The scale of the overall configuration of the signal generation unit 27 can be further reduced. However, in this case, the processing of the control system of the reference signal generator 27 may be complicated.

In this example, as the solid-state imaging device 1, a Bayer-type basic array is used, and as described above, the color filter is repeated every two rows and two columns. The pixel signal is read out in units of rows, and the pixel signal is input to each column AD circuit 25 provided in parallel for each vertical signal line 19. Therefore, one processing target row includes R / G or G / B. There are pixel signals of only two colors. Therefore, in this example, a DA conversion circuit 27a corresponding to an odd number column and a DA conversion circuit 27b corresponding to an even number column are provided.

  Further, the reference signals RAMPa and RAMPb output independently from each DA converter circuit are transmitted to the voltage comparison unit 252 through independent common reference signal lines 251a and 251b (collectively referred to as 251). A plurality of voltage comparison units 252a (in odd-numbered columns) and voltage comparison units 252b (in even-numbered columns) are connected to the common reference signal lines 251a and 251b, respectively.

  In this case, it is configured so as to be transmitted substantially directly to the plurality of voltage comparison units 252a and 252b corresponding to the color filters having common color characteristics via the independent common reference signal lines 251a and 251b. . Transmitting substantially directly through the common reference signal lines 251a and 251b means between the common reference signal lines 251a and 251b and the voltage comparison units 252a and 252b (each of which has a plurality of columns) corresponding to the common reference signal lines 251a and 251b. Means that there is no selection means such as a multiplexer. This is because the reference signal output from each analog reference voltage generator is transmitted to the input side of the comparator provided for each vertical column, and the reference from each analog reference voltage generator immediately before the input side of each comparator. This is greatly different from the configuration of Patent Document 1 in which selection means (multiplexer) for selectively outputting any one of the signals is provided.

  Each DA converter circuit 27a, 27b starts counting clocks CKdaca, CKdacb (count clock CK0 and count clock CK0 from the communication / timing control unit 20 from an initial value indicated by control data CN4 (CN4a, CN4b) from the communication / timing control unit 20. The stepped sawtooth wave (ramp voltage) is generated in synchronization with each other, and the generated sawtooth wave is converted into an AD conversion signal to each column AD circuit 25 corresponding to the column processing unit 26. Reference signals (ADC reference signals) RAMPa and RAMPb are supplied. Although illustration is omitted, a filter for preventing noise may be provided.

  The DA conversion circuits 27a and 27b have functions unique to the present embodiment, when performing AD conversion processing on the signal component Vsig in the pixel signal Vx at a predetermined position using the voltage comparison unit 252 and the counter unit 254, respectively. The initial voltages of the reference signals RAMPa and RAMPb to be emitted are set to values different from those at the time of AD conversion processing for the reset component ΔV, reflecting pixel characteristics and circuit variation, and the pixel is considered in consideration of the arrangement of color filters. It is characterized in that the respective inclinations βa and βb are set so as to match the characteristics.

  Specifically, first, it is assumed that the initial voltages Vas and Vbs of the reference signals RAMPa and RAMPb for the signal component Vsig are calculated based on signals obtained from pixels that generate an arbitrary plurality of black references. The pixel that generates the black reference is a pixel having a light shielding layer on a photodiode or the like as a photoelectric conversion element that forms the charge generation unit 32 arranged outside the color pixel. The arrangement form such as the arrangement location and the number of arrangements and the light shielding means are not particularly limited, and a known mechanism can be adopted.

  The initial voltage includes inherent variation components caused by the characteristics of the DA converter circuits 27a and 27b. Normally, the initial voltages Vas and Vbs are made lower by offsets OFFa and OFFb than the initial voltages Var and Vbr of the reference signals RAMPa and RAMPb for the reset component ΔV, respectively.

  Even if the initial voltages Var and Vbr of the reference signals RAMPa and RAMPb for the reset component ΔV are the same, the offsets OFFa and OFFb usually have different values, so that the initial values of the reference signals RAMPa and RAMPb for the signal component Vsig are different. The voltages Vas and Vbs are different.

  Note that the initial voltages Vas and Vbs of the reference signals RAMPa and RAMPb for the signal component Vsig may include an arbitrary offset in addition to the signal obtained from the pixel generating the black reference.

  The control for the offsets OFFa and OFFb performed by the DA conversion circuits 27a and 27b of the reference signal generation unit 27 has a function of calculating an initial voltage based on signals obtained from, for example, any reference pixel that generates a plurality of black references. The communication / timing control unit 20 may be provided, and may be performed based on an initial value indicated by the control data CN4 from the communication / timing control unit 20. Of course, the DA conversion circuits 27a and 27b may have a function of calculating the initial voltage and calculate the initial voltage by themselves.

  Alternatively, the communication / timing control unit 20 and the DA conversion circuits 27a and 27b in the chip do not have a function of calculating an initial voltage of the reference voltage, but are obtained from a reference pixel that generates a black reference by an external system outside the chip. The initial voltage is calculated based on the received signal, information indicating the initial voltage is notified to the communication / timing controller 20 as a part of the operation mode via the terminal 5b, and the control data CN4 from the communication / timing controller 20 is transmitted. May be notified to the reference signal generator 27.

  Note that the stepped reference signal generated by the reference signal generation unit 27, specifically, the reference signal RAMPa generated by the DA conversion circuit 27a and the reference signal RAMPb generated by the DA conversion circuit 27b are a high-speed clock from the clock conversion unit 23, for example, a multiplier circuit. By generating based on the multiplied clock generated in step (1), it is possible to change the speed faster than that based on the master clock CLK0 input via the terminal 5a.

  The control data CN4a and CN4b supplied from the communication / timing control unit 20 to the DA conversion circuit 27a of the reference signal generation unit 27 also include information for instructing the ramp voltage gradient (degree of change; amount of time change) for each comparison process. It is out.

  The column AD circuit 25 includes the reference signal RAMP generated by the DA conversion circuit 27a of the reference signal generation unit 27 and the vertical signal line 19 (H1, H2) from the unit pixel 3 for each row control line 15 (V1, V2,...). ,...), And a counter unit 254 that counts the time until the voltage comparison unit 252 completes the comparison process and holds the result. And has an n-bit AD conversion function.

  The communication / timing control unit 20 functions as a control unit that switches the count processing mode in the counter unit 254 according to which of the reset component ΔV and the signal component Vsig of the pixel signal the voltage comparison unit 252 is performing comparison processing. have. A control signal CN5 for instructing whether the counter unit 254 operates in the down count mode or the up count mode is input from the communication / timing control unit 20 to the counter unit 254 of each column AD circuit 25. Yes.

  One input terminal RAMP of the voltage comparison unit 252 receives the step-like reference signal RAMP generated by the reference signal generation unit 27 in common with the input terminal RAMP of the other voltage comparison unit 252, and inputs to the other input terminal. Are connected to the vertical signal lines 19 of the corresponding vertical columns, and the pixel signal voltages from the pixel unit 10 are individually inputted. The output signal of the voltage comparison unit 252 is supplied to the counter unit 254.

  The count clock CK0 from the communication / timing control unit 20 is input to the clock terminal CK of the counter unit 254 in common with the clock terminals CK of the other counter units 254.

  The counter unit 254 is omitted from the illustration of the configuration, but can be realized by changing the wiring form of the data storage unit constituted by the latch to the synchronous counter form. Counting is to be performed. Similarly to the stepped voltage waveform, the count clock CK0 is generated based on a high-speed clock (for example, a multiplied clock) from the clock conversion unit 23, so that the count clock CK0 is faster than the master clock CLK0 input through the terminal 5a. be able to.

  An n-bit counter unit 254 can be realized by a combination of n latches, which is halved with respect to the circuit scale of a data storage unit composed of two systems of n latches. In addition, since a counter unit for each column is not necessary, the overall size is significantly reduced.

  Here, although details will be described later, the counter unit 254 uses a common up / down counter (U / D CNT) regardless of the count mode to switch between the down count operation and the up count operation (specifically, It is characterized in that it can be configured to perform count processing (alternately). The counter unit 254 uses a synchronous counter in which the count output value is output in synchronization with the count clock CK0.

  In the case of a synchronous counter, the operations of all flip-flops (counter basic elements) are limited by the count clock CK0. Therefore, when higher frequency operation is required, the counter unit 254 uses an asynchronous counter suitable for high speed operation because its operation limit frequency is determined only by the limit frequency of the first flip-flop (counter basic element). Is more preferable.

  A control pulse is input to the counter unit 254 from the horizontal scanning circuit 12 through the control line 12c. The counter unit 254 has a latch function for holding the count result, and holds the counter output value until an instruction by a control pulse through the control line 12c is given.

  As described above, the column AD circuit 25 having such a configuration is arranged for each of the vertical signal lines 19 (H1, H2,...), And constitutes a column processing unit 26 that is an ADC block having a column parallel configuration. The

  The output side of each column AD circuit 25 is connected to the horizontal signal line 18. As described above, the horizontal signal line 18 has a signal line corresponding to the n-bit width which is the bit width of the column AD circuit 25, and passes through n sense circuits corresponding to the respective output lines (not shown). Then, the output circuit 28 is connected.

  In such a configuration, the column AD circuit 25 performs a count operation in the pixel signal readout period corresponding to the horizontal blanking period, and outputs a count result at a predetermined timing. That is, first, the voltage comparison unit 252 compares the ramp waveform voltage from the reference signal generation unit 27 with the pixel signal voltage input via the vertical signal line 19, and if both voltages are the same, the voltage comparison The comparator output of the unit 252 is inverted (in this example, transition from H level to L level).

  The counter unit 254 starts the count operation in the down-count mode or the up-count mode in synchronization with the ramp waveform voltage generated from the reference signal generation unit 27, and the counter unit 254 is notified of the inverted information of the comparator output. Then, the count operation is stopped and the AD conversion is completed by latching (holding / storing) the count value at that time as pixel data.

  Thereafter, the counter unit 254 sequentially stores the stored and held pixel data based on the shift operation by the horizontal selection signal CH (i) input from the horizontal scanning circuit 12 via the control line 12c at a predetermined timing. The data is output from the output terminal 5 c to the outside of the column processing unit 26 and the chip having the pixel unit 10.

  Although not specifically illustrated because it is not directly related to the description of the present embodiment, other various signal processing circuits may be included in the components of the solid-state imaging device 1.

<Functional description of reference signal generator>
FIG. 2 is a diagram for explaining the function of the DA conversion circuit (DAC) of the reference signal generation unit 27 used in the solid-state imaging device 1.

  The DA conversion circuits 27a and 27b are supplied with the DAC count clock CKdac from the communication / timing control unit 20, and are synchronized with the count clocks CKdaca and CKdacb, for example, in a stepwise sawtooth wave (ramp). Waveform) is generated, and the generated sawtooth wave is supplied to the voltage comparison unit 252 of the column AD circuit 25 as a reference voltage (ADC standard signal) for AD conversion.

  Here, the DA conversion circuits 27a and 27b first set the initial voltage based on the information indicating the initial value of the lamp voltage for each comparison process included in the control data CN4, and are included in the control data CN4. Based on the information indicating the slope (rate of change) of the ramp voltage for each comparison process, the voltage change ΔRAMP per clock is set, and the count value is changed by 1 for each unit time (count clock CKdac). To do. Actually, it is only necessary to set the maximum voltage width for the maximum count number (for example, 1024 in 10 bits) of the count clock CKdac. Any circuit configuration may be used for setting the initial voltage.

  Thus, the DA conversion circuits 27a and 27b decrease the voltage by ΔRAMP for each count clock CKdaca and CKdacb from the voltage (for example, 3.0V) indicating the initial value included in the control data CN4.

  When setting the coefficient for the pixel signal (specifically, the true signal component) from the unit pixel 3, the communication / timing control unit 20 is 1 / m with respect to the reference cycle of the count clock CKdac1 for setting the coefficient 1. The divided count clock CKdacm is supplied to the DA converter circuit 27a. The DA conversion circuit 27a reduces the voltage by ΔRAMP for each count clock CKdacm from the voltage (for example, 3.0V) indicating the initial value included in the control data CN4.

  As a result, the slopes of the reference signals RAMPa and RAMPb supplied to the voltage comparison unit 252 are 1 / m times that when the reference signals RAMPa and RAMPb are generated with the count clock CKdac1 (= CK0). In the unit 254, the count value becomes m times the same pixel voltage, that is, m can be set as a coefficient.

  That is, the slopes of the reference signals RAMPa and RAMPb can be changed by adjusting the periods of the count clocks CKdaca and CKdacb. For example, if a clock divided by 1 / m with respect to the reference is used, the slope becomes 1 / m. If the count clocks CK0 in the counter unit 254 are the same, the counter unit 254 can set the count value to m times the same pixel voltage, that is, m can be set as a coefficient. That is, by changing the slopes of the reference signals RAMPa and RAMPb, it is possible to adjust the coefficient at the time of differential processing described later.

  As can be seen from FIG. 2, the larger the inclination of the reference signals RAMPa and RAMPb, the smaller the coefficient applied to the amount of information stored in the unit pixel 3, and the smaller the inclination, the larger the coefficient. For example, the coefficient can be set to “2” by giving the count clock CKdac2 divided by ½ with respect to the reference period of the count clock CKdac1, and the coefficient can be set by giving the count clock CKdac4 divided by ¼. Can be set to “4”. The coefficient can be set to m / n by giving the count clock CKdacnm divided by n / m.

  By adjusting the inclination, the analog gain can be controlled during AD conversion. In other words, the slope of the reference voltage can be adjusted by adjusting the coefficient, but the amplitude of the reference signal per unit time can be adjusted by changing the slope of the reference voltage, which is compared to the pixel signal to be compared. It can function as all gain adjustments. Further, by dynamically switching this inclination during DA conversion, AD conversion processing can be performed in a form in which γ (gamma) correction is added to the analog signal.

  As described above, the coefficient is set easily and accurately by adjusting the cycle of the count clock CKdacnm supplied to the reference signal generation unit 27 while changing the voltage by ΔRAMP (decrease in this example) for each count clock CKdacm. can do. Note that the sign (+/−) of the coefficient can be designated by adjusting the count processing mode for the signal component Vsig of the pixel signal.

  The coefficient setting method using the slopes of the reference signals RAMPa and RAMPb shown here is merely an example, and the present invention is not limited to such a method. For example, if the period of the count clocks CKdaca and CKdacb supplied to the reference signal generator 27 is constant, the counter output value is x and the slope (change rate) β of the ramp voltage included in the control data CN4 is y = α ( (Initial value) By the information (that is, gain control signal) indicating the slope (rate of change) of the lamp voltage included in the control data CN4, such as outputting a control potential for gain setting calculated by -β * x, Arbitrary circuits can be used, such as adjusting the voltage change ΔRAMP for each count clock CKdac.

  For example, the slope of the ramp voltage, that is, the slope β of the RAMP slope can be adjusted by adjusting ΔRAMP per clock by changing the amount of current of the unit current source in addition to changing the number of clocks, for example.

  The setting method of α (initial value) that can provide an offset and β (coefficient) that can provide a slope may be set in accordance with a circuit configuration that generates a ramp waveform that gradually changes voltage for each of the count clocks CKdaca and CKdacb. . As an example, when a circuit that generates a ramp waveform is composed of a combination of constant current sources and a selection circuit that selects any one of the constant current sources (one or a plurality of arbitrary numbers), α giving an offset Both (initial value) and β (coefficient) that gives a slope can be realized by using a constant current source and adjusting the current flowing through the constant current source. This point will be described in detail later.

  Regardless of the method of generating the reference signal, the reference signal has an inclination corresponding to the color characteristic of the color pixel and has an initial value based on a viewpoint different from the color characteristic, such as a black reference or an offset component of the circuit. By doing so, it becomes possible to perform AD conversion processing using a reference signal suitable for both the viewpoint of color characteristics and the viewpoint different from the color characteristics.

<Operation of solid-state imaging device>
FIG. 3 is a timing chart for explaining signal acquisition difference processing which is a basic operation in the column AD circuit 25 of the solid-state imaging device 1 shown in FIG.

  As a mechanism for converting an analog pixel signal sensed by each unit pixel 3 of the pixel unit 10 into a digital signal, for example, a ramp-wave reference signal RAMP descending with a predetermined inclination and a pixel signal from the unit pixel 3 are used. Find the point where the voltage of the reference component or signal component matches, and from the time of generation of the reference signal RAMP used in this comparison process, the point of time when the electrical signal corresponding to the reference component or signal component in the pixel signal matches the reference signal A method of obtaining a count value corresponding to each size of the reference component and the signal component by counting (counting) up to the count clock.

  Here, the pixel signal output from the vertical signal line 19 is such that the signal component Vsig appears after the reset component ΔV including the noise of the pixel signal as a reference component as a time series. When the first process is performed on the reference component (reset component ΔV), the second process is performed on a signal obtained by adding the signal component Vsig to the reference component (reset component ΔV). This will be specifically described below.

  For the first reading, the communication / timing control unit 20 first sets the mode control signal CN5 to the low level to set the counter unit 254 to the down-count mode and activates the reset control signal CN6 for a predetermined period (high in this example). Level) and the count value of the counter unit 254 is reset to the initial value “0” (t9). Then, after the first reading from the unit pixel 3 of the arbitrary row Vα to the vertical signal lines 19 (H1, H2,...) Is stabilized, the communication / timing control unit 20 moves toward the reference signal generation unit 27. Control data CN4a and CN4b for generating reference signals RAMPa and RAMPb are supplied.

  In response to this, the reference signal generation unit 27 first has a slope βa that matches the color pixel characteristics of one color (R or G in the odd number column) existing on the Vα row, and has a sawtooth shape (RAMP shape) as a whole. ) To generate a reference signal RAMPa having a stepped waveform (RAMP waveform) that is time-varying in the DA converter circuit 27a, and one input terminal RAMP of the voltage comparison unit 252 of the column AD circuit 25 corresponding to the odd number column. Is supplied as a comparison voltage.

  Similarly, a stepped waveform (RAMP) that has a slope βb that matches the color pixel characteristics of the other color (G or B in the even column) existing on the Vα row and is time-changed in a sawtooth shape (RAMP shape) as a whole. A reference signal RAMPb having a waveform) is generated by the DA conversion circuit 27b and supplied as a comparison voltage to one input terminal RAMP of the voltage comparison unit 252 of the column AD circuit 25 corresponding to the even column.

  The voltage comparison unit 252 compares the RAMP waveform comparison voltage with the pixel signal voltage of an arbitrary vertical signal line 19 (Hα) supplied from the pixel unit 10.

  Further, simultaneously with the input of the reference signals RAMPa and RAMPb to the input terminal RAMP of the voltage comparator 252, a reference signal generator is used to measure the comparison time in the voltage comparator 252 with the counter unit 254 arranged for each row. In synchronization with the ramp waveform voltage emitted from the counter 27 (t10), the count clock CK0 is input from the communication / timing control unit 20 to the clock terminal of the counter unit 254, and the initial count value “0” is decreased as the first count operation Start counting. That is, the count process is started in the negative direction.

  The voltage comparison unit 252 compares the ramp-shaped reference signal RAMP from the reference signal generation unit 27 with the pixel signal voltage Vx input via the vertical signal line 19, and when both voltages become the same, The comparator output is inverted from H level to L level (t12). That is, the voltage signal corresponding to the reset component Vrst is compared with the reference signal RAMP, and an active-low (L) pulse signal is generated after a time corresponding to the magnitude of the reset component Vrst and supplied to the counter unit 254. To do.

  In response to this result, the counter unit 254 stops the counting operation almost simultaneously with the inversion of the comparator output, and latches (holds / stores) the count value at that time as pixel data, thereby completing the AD conversion (t12). . In other words, down-counting is started with the generation of the ramp-shaped reference signal RAMP supplied to the voltage comparison unit 252, and counting (counting) with the clock CK0 until an active-low (L) pulse signal is obtained by the comparison process, A count value corresponding to the magnitude of the reset component Vrst is obtained.

  When a predetermined down-count period has elapsed (t14), the communication / timing control unit 20 stops supplying control data to the voltage comparison unit 252 and supply of the count clock CK0 to the counter unit 254. As a result, the voltage comparison unit 252 stops generating the ramp-shaped reference signal RAMP.

  In the first reading, the reset level Vrst in the pixel signal voltage Vx is detected by the voltage comparison unit 252 and the count operation is performed. Therefore, the reset component ΔV of the unit pixel 3 is read.

  The reset component ΔV includes noise that varies for each unit pixel 3 as an offset. However, since the variation of the reset component ΔV is generally small and the reset level Vrst is generally common to all pixels, the output value of the reset component ΔV in the pixel signal voltage Vx of the arbitrary vertical signal line 19 is approximately known.

  Therefore, at the time of reading the reset component ΔV for the first time, it is possible to shorten the down-count period (t10 to t14; comparison period) by adjusting the RAMP voltage. In this embodiment, the comparison of the reset component ΔV is performed by setting the longest period of the comparison process for the reset component ΔV to a count period (128 clocks) of 7 bits.

In the subsequent second reading, in addition to the reset component ΔV, the electric signal component Vsig corresponding to the amount of incident light for each unit pixel 3 is read, and the same operation as the first reading is performed. That is, first, the communication / timing control unit 20 sets the mode control signal CN5 to high level and sets the counter unit 254 to the up-count mode (t18). After the second reading from the unit pixel 3 in the arbitrary row Vα to the vertical signal lines 19 (H1, H2,...) Is stabilized, the communication / timing control unit 20 performs AD conversion processing on the signal component Vsig. Therefore, the control data CN4a for generating the reference signal RAMPa (here, including the offset OFFa and the inclination βa) is supplied to the DA conversion circuit 27a, and the control data CN4b for generating the reference signal RAMPb (here, the offset OFFb and the inclination βb is set). Are supplied to the DA converter circuit 27b.

  In response to this, the reference signal generation unit 27 first has a slope βa that matches the color pixel characteristics of one color (R or G in the odd number column) existing on the Vα row, and has a sawtooth shape (RAMP shape) as a whole. ), A reference signal RAMPa having a step-like waveform (RAMP waveform) changed with time and reduced by the offset OFFa with respect to the initial value Var for the reset component ΔV is generated by the DA converter circuit 27a, and the odd-numbered columns are generated. A comparison voltage is supplied to one input terminal RAMP of the voltage comparison unit 252 of the corresponding column AD circuit 25.

  Similarly, a stepped waveform (RAMP) that has a slope βb that matches the color pixel characteristics of the other color (G or B in the even column) existing on the Vα row and is time-changed in a sawtooth shape (RAMP shape) as a whole. And a reference signal RAMPb that is lower than the initial value Vbr for the reset component ΔV by the offset OFFb is generated by the DA conversion circuit 27b, and the reference signal RAMPb of the column AD circuit 25 corresponding to the even number column A comparison voltage is supplied to one input terminal RAMP.

  The voltage comparison unit 252 compares the RAMP waveform comparison voltage with the pixel signal voltage of an arbitrary vertical signal line 19 (Vx) supplied from the pixel unit 10.

  As described above, the initial voltage of each reference voltage at this time is calculated based on a signal obtained from a pixel that generates an arbitrary plurality of black standards, and is generated from the DA conversion circuit 27a. Different values (offset OFFa and offset OFFb) including inherent variation components respectively generated by the reference signal RAMPa and the reference signal RAMPb emitted from the DA conversion circuit 27b are obtained. In addition, the initial voltage of the reference voltage may include an arbitrary offset in addition to the signal obtained from the pixel that generates the black reference.

  Simultaneously with the input of the reference signals RAMPa and RAMPa to the input terminal RAMP of the voltage comparison unit 252, in order to measure the comparison time in the voltage comparison unit 252 with the counter unit 254 arranged for each row, from the reference signal generation unit 27 In synchronization with the generated ramp waveform voltage (t20), the count clock CK0 is input from the communication / timing control unit 20 to the clock terminal of the counter unit 254, and the unit acquired at the time of the first reading as the second counting operation From the count value corresponding to the reset component ΔV of the pixel 3, up-counting is started contrary to the first time. That is, the count process starts in the positive direction.

  The voltage comparison unit 252 compares the ramp-shaped reference signal RAMP from the reference signal generation unit 27 with the pixel signal voltage Vx input via the vertical signal line 19, and when both voltages become the same, The comparator output is inverted from H level to L level (t22). That is, the voltage signal corresponding to the signal component Vsig is compared with the reference signal RAMP, and an active low (L) pulse signal is generated after a time corresponding to the magnitude of the signal component Vsig, and supplied to the counter unit 254. To do.

  In response to this result, the counter unit 254 stops the counting operation almost simultaneously with the inversion of the comparator output, and latches (holds / stores) the count value at that time as pixel data, thereby completing the AD conversion (t22). . In other words, down-counting is started with the generation of the ramp-shaped reference signal RAMP supplied to the voltage comparison unit 252, and counting (counting) with the clock CK0 until an active-low (L) pulse signal is obtained by the comparison process, A count value corresponding to the magnitude of the signal component Vsig is obtained.

  When the predetermined down-count period has elapsed (t24), the communication / timing control unit 20 stops the supply of control data to the voltage comparison unit 252 and the supply of the count clock CK0 to the counter unit 254. As a result, the voltage comparison unit 252 stops generating the ramp-shaped reference signal RAMP.

  At the time of the second reading, the signal component Vsig in the pixel signal voltage Vx is detected by the voltage comparison unit 252 and the counting operation is performed. Therefore, the signal component Vsig of the unit pixel 3 is read out.

  Here, in the present embodiment, the counting operation in the counter unit 254 is down-counting at the first reading, and up-counting at the second reading, and therefore, automatically in the counter unit 254, using equation (1) The count value corresponding to the subtraction result is held in the counter unit 254.

  Here, equation (1) can be transformed into equation (2). As a result, the count value held in the counter unit 254 is in accordance with the signal component Vsig.

  That is, as described above, each unit pixel 3 is subtracted in the counter unit 254 by two readings and counting processes, such as down-counting at the first reading and up-counting at the second reading. The reset component ΔV including variations and the offset component for each column AD circuit 25 can be removed, and the digital signal about the signal obtained by correcting the black reference component to the signal component Vsig corresponding to the amount of incident light for each unit pixel 3. Only data can be retrieved with a simple configuration. At this time, there is an advantage that circuit variations and reset noise can be removed.

  Therefore, the column AD circuit 25 of the present embodiment operates not only as a digital conversion unit that converts an analog pixel signal into digital pixel data but also as a CDS (Correlated Double Sampling) processing function unit. It will be.

  In addition, since the pixel data indicated by the count value obtained by Expression (2) indicates a positive signal voltage, a complement calculation or the like is unnecessary, and the compatibility with existing systems is high.

  Here, at the time of the second reading, the signal component Vsig corresponding to the amount of incident light is read out. Therefore, in order to determine the amount of light in a wide range, a wide up-count period (t20 to t24; comparison period) is taken, and the voltage It is necessary to change the lamp voltage supplied to the comparison unit 252 greatly.

  Therefore, in the present embodiment, the comparison of the signal component Vsig is performed by setting the longest comparison period for the signal component Vsig to a count period of 10 bits (1024 clocks). That is, the longest period of the comparison process for the reset component ΔV (reference component) is made shorter than the longest period of the comparison process for the signal component Vsig. The longest period of comparison processing of both the reset component ΔV (reference component) and the signal component Vsig, that is, the maximum value of the AD conversion period is not made the same, but the longest period of comparison processing for the reset component ΔV (reference component) is signaled. By making it shorter than the longest period of the comparison process for the component Vsig, the total AD conversion period of two times is devised.

  In this case, although the number of comparison bits is different between the first time and the second time, control data is supplied from the communication / timing control unit 20 to the reference signal generation unit 27, and the reference signal generation unit 27 based on the control data By generating the ramp voltage, the slope of the ramp voltage, that is, the rate of change of the reference signal RAMP is made the same for the first time and the second time. Since the ramp voltage is generated by digital control, it is easy to make the slope of the ramp voltage the same at the first time and the second time. As a result, the precision of AD conversion can be made equal, and the subtraction result represented by the expression (1) by the up / down counter can be obtained correctly.

  At a predetermined timing after completion of the second count process (t28), the communication / timing control unit 20 instructs the horizontal scanning circuit 12 to read out pixel data. In response, the horizontal scanning circuit 12 sequentially shifts the horizontal selection signal CH (i) supplied to the counter unit 254 via the control line 12c.

  In this way, the count value represented by the expression (2) stored and held in the counter unit 254, that is, the pixel data represented by n-bit digital data is sequentially transferred via the n horizontal signal lines 18. Video data D1 representing a two-dimensional image is obtained by outputting from the output terminal 5c to the outside of the column processing unit 26 or the chip having the pixel unit 10 and then repeating the same operation for each row.

  As described above, according to the solid-state imaging device, the count process is performed twice by switching the processing mode while using the up / down counter. Further, in the configuration in which the unit pixels 3 are arranged in a matrix, the column AD circuit 25 is configured by a column parallel column AD circuit provided for each vertical column.

  Here, when the AD conversion circuit is configured by the comparison circuit and the counter, the DA conversion circuit, which is a functional element that generates a reference signal for AD conversion supplied to the comparison circuit, is used in a color separation filter used for color image capturing. Instead of preparing all the colors of the color filter, only the amount corresponding to the combination of the predetermined colors according to the color repetition cycle determined by the type and arrangement of the colors is provided. In addition, a change characteristic (specifically, slope) of the reference signal (analog reference voltage) generated by the DA converter circuit in response to switching of a predetermined color combination existing in the processing target line by switching the processing target line. The initial value is switched according to the characteristics of the color filter, that is, the analog pixel signal.

  As a result, the DA converter circuit functioning as a reference voltage generator and the wiring from the reference voltage generator can be made smaller than the number of color filters constituting the color separation filter, and the reference voltage generator is provided for each color filter. A multiplexer that selectively outputs an analog reference voltage (reference signal) required when the circuit is prepared is not required, so that the circuit scale can be greatly reduced.

  In addition, since the change characteristic (specifically, the slope) of the reference signal generated by the DA converter circuit is switched in accordance with switching of the combination of the predetermined colors existing in the processing target row, the pixel unit 10 is configured. When converting analog pixel signals output from unit pixels into digital data by generating different reference voltages according to the characteristics of each color pixel and performing comparison processing, reference is made according to each color. By adjusting the slope of the signal, the characteristics of each color can be precisely controlled.

  In addition, since the initial value of the reference signal generated by the DA converter circuit is switched and set according to the inherent variation component and black reference component generated in the DA conversion circuit, the circuit variation can be corrected and the black reference component can be corrected. AD conversion can be performed with a simple configuration only for the signal to which the signal is added.

  Further, a subtraction process between the reference component (reset component) and the signal component can be directly obtained for each vertical column as the second count result, and a memory device that holds the respective count results of the reference component and the signal component is provided. This can be realized by a latch function provided in the counter unit, and it is not necessary to prepare a dedicated memory device for holding AD converted data separately from the counter.

  In addition, a special subtracter for taking the difference between the reference component and the signal component becomes unnecessary. Therefore, the circuit scale and circuit area can be reduced as compared with the conventional configuration, and in addition, an increase in noise and an increase in current or power consumption can be eliminated.

  In addition, since the column AD circuit (AD conversion unit) is configured by the comparison unit and the counter unit, the count process can be controlled by one count clock for operating the counter unit and a control line for switching the count mode regardless of the number of bits. A signal line for guiding the count value of the counter unit required in the conventional configuration to the memory device becomes unnecessary, and an increase in noise and an increase in power consumption can be solved.

  That is, in the solid-state imaging device 1 in which the AD conversion device is mounted on the same chip, the voltage comparison unit 252 and the counter unit 254 are paired to configure the column AD circuit 25 as an AD conversion unit, and the operation of the counter unit 254 By using a combination of down-counting and up-counting as a digital data, the difference between the basic component of the signal to be processed (the reset component in this embodiment) and the signal component is converted into digital data. Alternatively, problems such as the number of interface wirings with other functional units, noise and current consumption due to the wirings can be solved.

  Although not shown, a data storage unit as an n-bit memory device that holds the count result held by the counter unit 254 may be provided at the subsequent stage of the counter unit 254. A control pulse is input to the data storage unit from the horizontal scanning circuit 12 through the control line 12c. The data storage unit holds the count value fetched from the counter unit 254 until there is an instruction by the control pulse through the control line 12c. The horizontal scanning circuit 12 reads the count values held by the data storage units in parallel with the voltage comparison units 252 and the counter units 254 of the column processing unit 26 performing the processes that they are responsible for. Has the function of the scanning unit. With this configuration, pipeline processing can be realized.

  That is, before the operation of the counter unit 254 (t6), based on the memory transfer instruction pulse CN8 from the communication / timing control unit 20, the count result at the time of processing of the previous row Hx-1 is transferred to the data storage unit.

  In the operation shown in FIG. 3, the pixel data cannot be output to the outside of the column processing unit 26 until the second reading process for the pixel signal to be processed, that is, the AD conversion process is completed. However, if a data storage unit is provided after the counter unit 254, a count indicating the result of the previous subtraction process prior to the first reading process (AD conversion process) for the pixel signal to be processed is provided. The value can be transferred to the data storage unit, and the reading process is not limited.

  Therefore, by adopting such a configuration, it is possible to independently control the counting operation of the counter unit 254, that is, the AD conversion processing, and the reading operation of the count result to the horizontal signal line 18, and the AD conversion processing and the external ( First, a pipeline operation can be realized in which the signal reading operation to the horizontal signal line 18) is performed independently and in parallel.

<Basic configuration of reference signal generator>
FIG. 4 is a diagram illustrating a basic configuration example of the reference signal generation unit 27. The reference signal generation unit 27 of the present embodiment employs a current output type DA converter circuit as its basic configuration, and the reference signal (ramp voltage) supplied to the voltage comparison unit 252 of the column AD circuit 25. As a mechanism for adjusting the slope, that is, the slope β of the RAMP slope, in addition to changing the count clock number (that is, the clock frequency) used in the DA conversion process, the current amount of the unit current source is changed by digital control, so that It is characterized in that it has a mechanism for adjusting ΔRAMP. The current output type DA converter circuit is of a current source cell matrix type having a cell arrangement arranged in a matrix.

  Conventionally, this current output type DA converter circuit uses a plurality of current source cells superimposed on a predetermined current value to generate the same constant current, and a multi-bit digital input is selected from these current source cells. An analog current output corresponding to a digital input signal value is obtained by selecting a current source cell corresponding to the signal data value and adding the constant current output of the selected current source cell. ing.

  As a circuit method for selecting a current source cell, a number of methods are adopted mainly for a decoding method, a binary method, and a composite method combining both of them. A method of converting a digital input signal into an analog signal in two stages is widely known (see, for example, Japanese Patent Laid-Open No. 11-17545).

  However, in the conventional composite system configuration, the input digital signal is decoded in the decimal system on the upper bit side and divided into the binary system on the lower bit side. The constant current source cells arranged in (1) are selected by decoding and latching. Therefore, when the input digital signal becomes high speed, it becomes difficult to simultaneously operate the devices divided by the decoding method and the binary method, and to select the cell by operating the decoding and latching at high speed and surely. As a result, the cause of occurrence of glitches and miscodes is generated, making it difficult to realize stable operation.

  Further, for example, when supplying a color-corresponding reference signal corresponding to the arrangement of the color separation filters, it is necessary to supply a reference signal that changes at a change rate (for example, a decrease rate) corresponding to the color to the comparator. In this case, high accuracy is required for setting the rate of change in order to adapt to each color.

  Further, in order to ensure a dynamic range, it is necessary to obtain digital data after adjusting the gain (analog gain) of an imaging signal when imaging in a dark part with little input light. Also in this case, high accuracy is required for setting the rate of change in order to adapt to the input light.

  In adjusting the rate of change of the reference signal, for example, it is conceivable to deal with the reference signal generated at the standard rate of change by adjusting its gain using an analog amplifier. However, it is difficult to adjust the analog gain in terms of accuracy. For example, it is difficult to make fine adjustment with accuracy of 12 bits or more or to reduce variations in the reference potential.

  Therefore, in the present embodiment, a mechanism that can solve such a problem is adopted. This will be specifically described below.

  As shown in FIG. 4A, the DA converter 300 of this embodiment, which forms the reference signal generator 27 (DA converter circuits 27a and 27b), generates a reference signal RAMP having a predetermined slope and an initial value. A first DA converter 302 that is a functional part that acts independently, and a second DA converter 304 that is a functional part that acts mainly on the control (gain control) of the slope of the reference signal generated in the first DA converter 302. And. The first DA conversion unit 302 is a main function unit as a slope (slope type DA conversion unit) that generates a slope-like reference voltage, and the second DA conversion unit 304 is a first DA conversion unit. This is a main function unit as a programmable gain amplifier (PGA) unit that controls the unit 302.

  First and second DA converters 302 and 304 control current source cell units 350 and 750 provided with a plurality of current source cells that generate predetermined weighted output current values, and current source cell selection operations, respectively. DAC control units 310 and 710 for controlling the current source cell are selected by digital processing, and DA conversion is performed by adding the current output from the selected current source cell. It has a DA conversion circuit configuration.

  Here, on the output side of the second DA converter 304, a current source cell for adjusting the amount of current (hereinafter also referred to as a gain adjusting current source cell) 308 is provided. In the second DA converter 304, each current source cell is selected and controlled by digital processing, the current is added at the DAC output terminal DACgain based on one of the differential current outputs, and this added current Igain is gain-adjusted current source. Supply to cell 308. The gain adjustment current source cell 308 supplies a control voltage (gain control output signal) Vbais based on the addition current Igain to each current source cell of the first DA converter 302 via the current control line 592. By controlling the control voltage Vbais, the slope of the reference signal RAMP can be controlled. That is, the slope of the reference signal RAMP used in the single slope integration AD conversion method generated by the first DA converter 302 is adjusted by digital control by the second DA converter 304.

  Here, the first DA converter 302 and the second DA converter 304 are connected by a current mirror via a gain adjustment current source cell 308. That is, the gain adjustment current source cell 308 forms a current mirror with each current source cell in the current source cell unit 350.

  For example, as shown in FIG. 4B, the gain adjustment current source cell 308 includes an NMOS transistor 790 in which a drain and a gate to which an addition current Igain is supplied are connected and a source is connected to an analog ground line. ing. A bias voltage Vbais corresponding to the added current is generated at the gate (also drain) of the NMOS transistor 790. This bias voltage Vbais is supplied to a current control line 592 that is commonly connected to the gates of NMOS transistors constituting each current source cell arranged on the first DA converter 302 side.

  In FIG. 4B, an NMOS transistor 790 is used as the gain adjustment current source cell 308. This is because each current source cell constituting the second DA converter 304 is a PMOS transistor. As shown in FIG. 4C, when each current source cell constituting the second DA converter 304 is an NMOS transistor, first, the PMOS transistor 792 receives the addition current Igain. become.

  Further, when the transistors constituting the gain adjustment current source cell 308 and the transistors constituting each current source cell arranged on the first DA converter 302 side have different polarities (channels), the first DA converter A current / current conversion unit is provided so that a current mirror can be configured with the transistors constituting each current source cell arranged on the 302 side.

  For example, in FIG. 4C, a PMOS transistor 794 that is current-mirror connected to the transistor 792 and an NMOS transistor 796 that receives current output from the transistor 794 are provided. Connect in the same way.

  That is, in the transistor 796, an addition current is supplied to the drain, the drain and the gate are connected, and the source is connected to the analog ground line. The gate (also drain) of the transistor 796 is connected in common with the gate of the NMOS transistor constituting each current source cell arranged on the first DA converter 302 side. A bias voltage Vbais corresponding to the added current is generated at the gate (also drain) of the NMOS transistor 796. This bias voltage Vbais is supplied to a current control line 592 that is commonly connected to the gates of the NMOS transistors constituting each current source cell arranged on the first DA converter 302 side.

  4D, an NMOS transistor 798 that is current-mirror connected to the transistor 790 and a PMOS transistor 799 that receives current output from the transistor 798 are provided. An additional current is supplied from the transistor 798 to the drain of the transistor 799, the drain and the gate are connected, and the source is connected to the power supply line. The gate (also drain) of the transistor 799 is connected in common with the gate of the PMOS transistor constituting each current source cell arranged on the first DA converter 302 side. A bias voltage Vbais corresponding to the added current is generated at the gate (also drain) of the PMOS transistor 799. This bias voltage Vbais is supplied to a current control line 592 that is commonly connected to the gate of the PMOS transistor that constitutes each current source cell arranged on the first DA converter 302 side.

  In any operation, the gain adjustment current source cell 308 converts the addition current Igain generated on the second DA converter 304 side into a voltage signal (bias voltage Vbais in the above example), and the converted voltage. It functions as a current / voltage converter that controls the operating current value of each current source cell on the first DA converter 302 side based on the signal.

  As shown in FIG. 9 to be described later, the first DA converter 302 as the slope section is configured by NMOS, the second DA converter 304 as the PGA section is configured by PMOS, and the bias potential thereof is the PMOS configuration. The configuration in which the circuit output is folded is advantageous in the following points. First, in order to create the gate potential of NMOS, it must be output from PMOS.

  Second, it is resistant to variation. The second point will be described in further detail as follows. For example, when the temperature rises and the capability of the transistor decreases, the output from the amplifier is always constant, but the deep gate-source potential Vgs is applied to allow the same current to flow, and this output is input to the turn-back configured by NMOS. As a result, the same output as at room temperature can be maintained. Further, even if there is a difference in capability between the NMOS and PMOS transistors due to process variations, a stable output can be obtained with the same logic as described above.

  The output current Igain of the second DA conversion unit 304 added (synthesized) at the DAC output terminal DACgain is received by the gain adjustment current source cell 308 functioning as a current / voltage conversion unit. A circuit configuration in which a transistor 790 (or a transistor 792) is provided and the gate and drain thereof are connected is employed.

  The first DA conversion unit 302 has a configuration of a current output type DA conversion circuit, and the output voltage of the gain adjustment current source cell 308 functioning as a current / voltage conversion unit is connected to the gate which is a control input terminal of the current source cell. The bias voltage Vbais) is supplied.

  As a whole, the PMOS transistor that constitutes the current source cell of the second DA converter 304 and the NMOS transistor that constitutes the current source cell of the first DA converter 302 are folded back and used, thereby making the variation strong.

  Thus, by setting the operating current of each current source cell in the current source cell unit 350 by digital control, a DA converter circuit that operates with an arbitrarily set gain can be realized, and each current source cell By adjusting the operating current, for example, the gain can be varied from -3 dB to 20 dB. By controlling the current source cell unit 350 with a digital signal, the DA conversion unit 300 can operate as a DA conversion circuit capable of analog output of a desired input digital signal Din with a desired gain.

  Although details will be described later, in particular, the second DA converter 304 of the present embodiment is for adjusting the amount of current for controlling the operating current amount of each current source cell which is an example of a unit current source provided in the current source cell unit 750. This current source cell is driven by digital control, and has a great feature in that the slope per clock (ΔRAMP) in DA conversion in the first DA converter 302 is adjusted.

  By performing digital control with, for example, 12-bit accuracy, as a result, the amount of current addition in the current source cell unit 350 can be adjusted with 12-bit accuracy. This means that when performing single slope integration type AD conversion, digital data is acquired after adjusting the analog gain with high accuracy of 12 bits for the pixel signal to be processed.

  In addition, the operation current control for each current source cell of the current source cell unit 350 by the second DA conversion unit 304 can be uniformly controlled using the current mirror with the gain adjustment current source cell 308. By controlling the position (here, the gain adjustment current source cell 308), uniform gain adjustment can be performed for each current source cell of the current source cell unit 350.

  The accuracy in the amplitude control of the reference voltage is determined by the digital adjustment accuracy of the second DA converter 304 as a gain adjustment DA converter circuit by digital input. Needless to say, the gain setting digital input (input code) Dgain to the second DA converter 304 can be directly input from the outside.

<Configuration of first DA converter>
FIG. 5 is a diagram illustrating a specific configuration example of the first DA converter 302. As shown in the figure, the first DA converter 302 controls the entire first DA converter 302 and has a function of a constant current source selection controller, and a plurality of current source cells (constant current sources). And a current source cell unit 350.

  The DAC control unit 310 includes a block control unit 320 that controls the operation of each unit in the DAC control unit 310, and a lower bit control unit 330 that is disposed in the input stage of the input digital signal Din to be processed and is in charge of lower bit control processing. And an upper bit control unit 340 that is disposed in the input stage of the input digital signal Din to be processed and is in charge of the upper bit control process.

  The current source cell unit 350 and the lower current source cell unit 352 having a plurality of types of lower current source cells 353 that output a predetermined weight current are arranged in a matrix and each output the same (uniform) predetermined weight current. And an upper current source cell unit 354 having an upper current source cell 355.

  The lower-order current source cell unit 352 is provided with j lower-order current source cells 353 that are responsible for the lower-order j bits corresponding to the lower-order bits of the digital input signal one by one. The output terminal of the lower current source cell 353 is connected to a selection output line 396 for synthesizing the output currents of the lower current source cells 353. The selection output line 396 is connected to the DAC output terminal DACout.

The weighting of the current value of each lower current source cell 353 of the lower current source cell unit 352 is 1/2, 1/4, and the current value of each upper current source cell 355 of the upper current source cell unit 354, respectively. ..., 1/2 ^ j. Each lower current source cell 353 is individually selected by the lower bit control unit 330 according to the logical value (“1” or “0”) of the bit corresponding to each of the lower j bits of the input digital signal Din. It has become. The output current of the lower-order current source cell 353 selected by the lower-order bit control unit 330 based on the lower-order j bits of the input digital signal Din is output from the DAC output terminal DACout by one of the reference resistors 398 connected to the reference voltage Vref. Addition output.

  The upper current source cell unit 354 is provided with at least 2 ^ i (preferably (2 ^ i) + y (y is optional)) upper current source cells 355 in charge of the upper i bits in a two-dimensional matrix. It is done. Around this, a matrix selection line 358 corresponding to the matrix arrangement of the upper current source cells 355 is provided to select each upper current source cell 355.

  In the upper current source cell unit 354, the upper bit control unit 340 selects the number of upper current source cells 355 corresponding to the upper bit data value of the digital input signal. The output current of the selected upper current source cell 355 is added and output at the DAC output terminal DACout, similarly to the lower current source cell unit 352. The product of this added current Idac and the resistance value Rref of the reference resistor 398 defines the output voltage of the DAC output terminal DACout.

  The block control unit 320 controls the upper bit control unit 340 and the lower bit control unit 330 based on the multi-bit digital signal Din supplied from the communication / timing control unit 20. As an example, the upper i bits are decoded into a decimal number, and the shift register in the shift register unit 342 is controlled based on the decoded value.

  The block control unit 320 is supplied with various control signals J320 for controlling the DAC mode, the DAC start, or the DAC resolution, and the input digital signal Din (for example, a decoded value) and the control thereof. Based on the signal J320, a control signal J330 for controlling the lower bit control unit 330 is supplied to the lower bit control unit 330, and a control signal J340 for controlling the upper bit control unit 340 is supplied to the upper bit control unit 340.

  As the control signal J330, for example, there is a reset signal for setting the output of the frequency division processing unit 332 to a certain logic level (here, a clear value). As the control signal J340, for example, there are a reset signal and a set signal for setting the output of the shift register unit 342 to a certain logic level (here, a clear value or a full value).

  Further, the DA control count clock CKdac is input from the communication / timing control unit 20 to the block control unit 320 as an external clock, and the count clock CKdac is shaped and used as the internal count clock CKcnt. The data is supplied to the bit control unit 330.

  Further, the block control unit 320 defines the digital resolution of DA conversion (for example, “i” of i bits) by defining the number used by the upper current source cells 355 of the upper current source cell unit 354. A control signal J342 for control is also supplied to the upper bit control unit 340. As this control signal J342, for example, an enable signal for controlling activation of each shift register can be used. When the enable signal supplied to the output enable (OE) terminal of each shift register is activated, the output of each shift register becomes valid, but the enable signal is made inactive to invalidate the output of each shift register (for example, By maintaining the non-inverting output terminal Q at the L level and the inverting output terminal xQ at the H level), the activation of each shift register can be controlled.

  For example, in the upper current source cell unit 354 having the upper resolution current source cells 355 for i bits (2 ^ i) as the maximum resolution, for example, if the resolution is to be controlled purely in bit units, x (x ≦ i When using at a bit resolution, control is performed so that only 2 ^ x upper current source cells 355 are activated. In this case, there may be i (control signals J342_1 to _i) as the control signal J342.

  The control signal J342_1 controls the activation of the first upper current source cell 355, the control signal J342_2 controls the activation of the second to 2 ^ 2 upper current source cells 355, and so on. The control signals J342_1 to _x are used to control activation of the upper current source cells 355 from the 1 + 2 ^ (x-1) th to 2 ^ xth. That is, the total number 2 ^ i (which may be + y) of upper current source cells 355 is divided into x blocks so that only upper current source cells 355 corresponding to a desired digital resolution are used in advance. Select it.

The lower bit control unit 330 includes a frequency division processing unit 332 including a counter (that is, a frequency divider that divides the clock) that counts the internal count clock CKcnt supplied from the block control unit 320, and a first DA conversion unit. And a glitch suppression processing unit 336 that includes a plurality of glitch suppression circuits that suppress glitches that may appear in the output 302. The same number (j) of glitch suppression circuits as the number of lower current source cells 353 provided in the lower current source cell unit 352 are provided.

  The frequency division processing unit 332 is provided so that DA conversion equivalent to 1 LSB can be executed in one clock period. Specifically, a D-type flip-flop (latched; hereinafter referred to as D-FF) is a basic element. Having (j−1) frequency dividers that generate a power of two, and divided by a frequency divider and a frequency divider of the actual count clock CKcnt itself (j− 1) A low-order bit binary output of bits, that is, a divided clock of 1/2, 1/4,..., 1/2 ^ (j−1) is used as a selection control signal to the corresponding glitch suppression circuit of the glitch suppression processing unit 336. Supply. That is, a 1 / 2１ ／ ２k (k is from 0 to j−1) frequency-divided clock that is an output corresponding to the lower-order j-bit binary data is supplied to the glitch suppression processing unit 336_j-k. The frequency division processing unit 332 functions as a selector for the j lower current source cells 353 provided in the lower current source cell unit 352.

  The glitch suppression processing unit 336 performs a glitch suppression process on each frequency-divided clock by each glitch suppression circuit, and then 1 / k with respect to the frequency-divided clock of 1/2 ^ k (k is 0 to (j-1)). The current is supplied to the corresponding lower current source cell 353 of the lower current source cell unit 352 such as a current source cell having a current value of 2 ^ (j−k). For example, it is connected to the lower current source cell 353_j having a current value of 1/2 ^ j for the divided clock of 1 and 1/2 ^ (j-1) of current for the divided clock of 1/2. Is connected to the lower current source cell 353_j-1 having a value, and is connected to the lower current source cell 353_j-2 having a current value of 1/2 ^ (j-2) with respect to the quarter-divided clock. Similarly, the 1/2 ^ (j-2) divided clock is connected to the lower current source cell 353_2 having a current value of 1/4, and the 1/2 ^ (j-1) divided clock is Is connected to the lower current source cell 353_1 having a current value of 1/2.

  Here, although not shown, the glitch suppression processing unit 336 includes delay means having a logic inversion (phase inversion) function for an input signal and a delay function of a predetermined delay amount Δt. The complementary input terminals (here, transistors 524) of the differential switches constituting the current source cells 353 and 355 using the input signal not delayed by the delay means and the inverted signal delayed by the delay means as complementary signals. 526 gate terminal).

  When a complementary signal is simply supplied to each complementary input terminal of the differential switch, it operates as a differential switch forming current source cells 353 and 355 due to temporal variations (delay difference) of complementary inputs to the differential switch. When both the inputs to the transistors 524 and 526 are at the L level and both are turned off, the transistor 524 related to the DA conversion output in the selected output line 396 is turned on from the time when both are turned off. Gliding is likely to occur. This is because when the transistors 524 and 526 are both off, the output current from the current source cell is completely zero, and the current source cell is activated from such a state to suddenly generate the output current. This is considered to be the cause.

  On the other hand, if the transistors 524 and 526 operating as differential switches of the current source cells 353 and 355 are controlled via the glitch suppression processing unit 336, the gate input of the transistor 526 is activated (high level) after the delay Δt period. Or inactive (low level), the transistor 524 is activated and the transistor 526 is deactivated after the delay Δt. In the process where the transistors 524 and 526 are both off, the transistor 524 is turned on. Can be reliably avoided, and the occurrence of glitches can be suppressed (refer to FIG. 9 for details).

  Further, the lower bit control unit 330 has a selection line 338 corresponding to the number of lower current source cells 353 and controls the selection line 338 in order to select each lower current source cell 353 of the lower current source cell unit 352. Thus, the lower current source cell 353 corresponding to the lower bit data value of the digital input signal is selected. In the case of this configuration example, the glitch suppression processing unit 336 is provided on the selection line 338 between the frequency division processing unit 332 and the lower current source cell unit 352.

  Although the details will be described later, the glitch suppression processing unit 336 outputs the positive logic output Q and the negative logic output xQ from each output terminal substantially simultaneously based on the frequency-divided clock from the frequency-dividing processing unit 332 (hereinafter referred to as complementary output). As the selection line 338, two selection lines are connected to each lower current source cell 353 (specifically, its differential switch input terminal) accordingly. It has become.

  Further, the lower bit control unit 330 includes a shift control unit 333 that supplies a signal indicating a carry time or a carry time to the shift register unit 342 of the upper bit control unit 340 as a shift clock CKsr. For example, the shift control unit 333 logically inverts the 1/2 ^ (j-1) frequency-divided clock in the low-order bit binary output of the frequency division processing unit 332 in order to generate the shift clock CKsr indicating the carry time. An inverter 334 having a buffer function is provided. The frequency division processing unit 332 converts the 1/2 ^ (j-1) frequency-divided clock to the reverse phase via the inverter 334 and uses one of the change edges, so that the upper bit control unit 340 is used as the shift clock CKsr. To supply.

  Of course, such a configuration of the shift control unit 333 is an example. For example, a carry pulse that can be generated during the up-count operation can be used as the shift clock CKsr indicating the carry time, or the down-count operation. A Borrow pulse that can sometimes be generated can be used as the shift clock CKsr indicating the time of the carry-down.

  The upper bit control unit 340 includes a shift register unit 342 including upper i bits (2 ^ i) of shift registers, and a plurality of glitch suppression circuits that suppress glitches that may appear in the output of the first DA converter 302. And a glitch suppression processing unit 346. As many glitch suppression circuits as the number of upper current source cells 355 provided in the upper current source cell unit 354 are provided (2 ^ i). The shift register unit 342 is supplied with a shift clock CKsr from the lower bit control unit 330.

  The shift register unit 342 is provided so as to be able to execute DA conversion corresponding to each upper-order i-bit data value. Specifically, the shift-register unit 342 is cascade-connected so as to sequentially correspond to the upper-order i-bit digital signal. A shift register is provided, and based on the shift clock CKsr supplied from the lower bit control unit 330, the shift output terminals are sequentially activated (activated) in a predetermined direction, whereby the upper i bits of the input digital signal are converted into decimal numbers. The decoded data value is output to the shift output terminal.

  Each shift output of the shift register is supplied to the corresponding glitch suppression circuit of the glitch suppression processing unit 346 as a selection control signal. The glitch suppression processing unit 346 supplies each shift output to the corresponding upper current source cell 355 of the upper current source cell unit 354 after performing the glitch suppression processing by each glitch suppression circuit.

  In addition, the upper bit control unit 340 selects the upper current source cells 355 of the upper current source cell unit 354 so as to select a matrix selection line 348 corresponding to the number of upper current source cells 355 (corresponding to the matrix selection line 358). ) And the matrix selection line 348 is controlled to select the upper current source cell 355 corresponding to the upper bit data value of the digital input signal. In the case of this configuration example, the glitch suppression processing unit 346 is provided on the matrix selection line 348 between the shift register unit 342 and the upper current source cell unit 354.

  Similar to the glitch suppression processing unit 336, the glitch suppression processing unit 346 is configured to perform complementary output based on the shift output output from the shift register unit 342, and is arranged around the upper current source cell unit 354. As the matrix selection line 358, two selection lines are connected to each upper current source cell 355 (specifically, its differential switch input terminal) in accordance with each output Q, xQ.

  Note that the shift register unit 342 receives the digital resolution control control signals J342_1 to _i supplied from the block control unit 320 to the output enable terminal of the shift register whose basic element is D-FF, for example, x (x ≦ i) When using at a bit resolution, only 2 ^ x shift registers are activated in advance.

  The control signal J342_1 controls the activation of the first shift register, the control signal J342_2 controls the activation of the second to 2 ^ 2 shift registers, and similarly, the control signals J342_1 to _x Is used to control the activation of the shift registers from the 1 + 2 ^ (x-1) th to the 2 ^ xth. That is, by dividing the total number of 2 ^ i (which may be + y) shift registers into x blocks, a selection is made in advance so that only shift registers corresponding to the desired digital resolution are used. Thus, only the upper current source cell 355 corresponding to the desired digital resolution is used.

  Of course, the digital resolution is not limited to being controlled in bit units, but can be controlled by an arbitrary value z. For example, if the upper i bits are 7 bits, an arbitrary y is 8, and z is 34 divided by 4 shift registers, the control signal corresponding to the desired digital resolution set in advance is sent to the shift register for each z. Entered.

  The upper bit control unit 340 performs DA conversion for the upper bit data by selecting the number of upper current source cells 355 corresponding to the upper bit data value of the digital input signal. At this time, by controlling so that only the number of shift registers and upper current source cells 355 corresponding to the digital resolution are activated, the DA conversion is practically not performed after the set digital resolution is reached. To. In other words, the upper bit control unit 340 starts DA conversion based on the upper bit data based on the shift clock CKsr supplied from the lower bit control unit 330 and automatically reaches a desired digital resolution. DA conversion stops.

  Based on the shift clock CKsr supplied from the lower bit control unit 330, the shift register unit 342 sequentially activates (activates) the shift output terminals of the cascaded shift registers in a predetermined direction. Here, the “predetermined direction” means that when the lower bit control unit 330 is in a carry operation, a shift clock CKsr indicating a shift up is supplied, so that one upper current source cell 355 is further activated. Shift operation in the direction. On the other hand, when the lower bit control unit 330 is in the carry-down operation, the shift clock CKsr indicating the downshift is supplied, so that the upper current source cell 355 in the final stage that is inactivated at that time is inactivated. The shift operation is performed in the direction to be performed.

  By performing such an operation in a chain for shift registers corresponding to the number set (activated) to be usable in advance, an analog voltage corresponding to a desired digital resolution can be generated. For the shift register that is not activated, the non-inverting output terminal Q is set to the L level and the inverting output regardless of the state of the non-inverting output terminal Q in the previous stage even if the active edge of the shift clock CKsr is input to the clock input terminal CK. The terminal xQ is maintained at the H level. For this reason, DA conversion can be virtually prevented from being performed after the set digital resolution is reached, and in effect, DA conversion for the upper bits is stopped when the desired digital resolution is reached. .

  For example, when the digital resolution is 7 bits, DA conversion by the upper bit control unit 340 and the upper current source cell unit 354 stops when the 2 ^ 7th shift register is turned on. Also, the DA conversion for the upper bits can be stopped by stopping the shift clock CKsr from the lower bit control unit 330 when the upper bit data of the desired input digital value (digital code) is reached.

  If the output currents of the current source cells 353 and 355 appearing on the selected output line 396 until the DA conversion is stopped are combined and converted into a voltage by the reference resistor 398, the voltage level of the DAC output terminal DACout gradually changes monotonously. The reference signal to be generated can be generated. This is a DA converter suitable for generating a reference signal voltage used in so-called single slope integration type (or ramp signal comparison type) AD conversion.

  In addition, the DA converter 300 according to the present embodiment is not limited to being applied as a DA converter for generating a reference signal voltage used for single slope integration type AD conversion, but may be used as a general DA converter. it can. For example, if the voltage level of the DAC output terminal DACout in a state where the DA conversion processing is stopped when reaching a value corresponding to the input digital signal is used, an analog voltage corresponding to the input digital signal can be obtained. As described above, DA conversion can be performed on a multi-bit input digital signal.

  In addition, since the setting of the stop timing in the previous example has been described only for the upper bit side, the accuracy for the lower bits cannot be compensated as it is. However, for the lower bits, when the upper bit side has reached the data to be stopped, the internal count clock CKcnt is stopped when the value corresponding to the lower j bits of the input digital signal is reached. The DA conversion process can be stopped. In this case, the analog level corresponding to the multi-bit input digital signal is accurately obtained by using the voltage level of the DAC output terminal DACout in a state where the DA conversion processing is stopped when the value corresponding to the multi-bit input digital signal is reached. A voltage can be obtained, and as a result, high-precision DA conversion can be performed on a multi-bit input digital signal.

<Configuration of second DA converter>
6 and 7 are diagrams for explaining a specific configuration example of the second DA converter 304. Here, FIG. 6 is a diagram illustrating a configuration example of the second DA converter 304, and FIG. 7 is a diagram illustrating a relationship between the external input code and the current source cell.

  As shown in FIG. 6, the second DA converter 304, which is a characteristic part of the present embodiment, controls the entire second DA converter 304 and also has a DAC controller 710 having a function of a constant current source selection controller. And a current source cell unit 750 having the same configuration as the current source cell unit 350 including a plurality of current source cells (constant current sources). The DAC control unit 710 has a function as a current setting unit that sets the operating current of each current source cell in the current source cell unit 350 related to the amplitude of the reference signal generated by the first DA conversion unit 302. It is characterized in that an m-bit digital signal is controlled by being divided into upper s bits and lower t bits (m = s + t).

  For example, the DAC control unit 710 is arranged in the input stage of the predecoder 720 that controls the operation of each unit in the DAC control unit 710 and the gain setting input code (digital gain control input signal) Dgain to be processed. A lower bit control unit 730 that is in charge of control processing and an upper bit control unit 740 that is arranged in the input stage of the gain setting input code Dgain to be processed and is in charge of control processing of upper s bits.

  The current source cell unit 750 is in the current source cell unit 350, the lower current source cell unit 752 is in the lower current source cell unit 352, the lower current source cell 753 is in the lower current source cell 353, and the upper current source cell unit 754 is in the upper current. The source cell unit 354 and the upper current source cell 755 correspond to the upper current source cell 355, respectively. The configuration of the current source cell unit 350 and the distribution of output current weighting are the same as those of the current source cell unit 350, and n is m , I may be replaced with s, j.

  The output terminal of the lower current source cell 753 is connected to a selection output line 796 for synthesizing the output current of each lower current source cell 753. The selected output line 796 is connected to the DAC output terminal DACgain, and the output current of the lower current source cell 753 selected in the lower current source cell unit 752 is added by the selected output line 796 and output. In addition, the output current of the selected upper current source cell 755 in the upper current source cell unit 754 is added with the current through the selection output line 796 and output as in the lower current source cell unit 752. As shown in FIG. 5, the gain adjustment current source cell 308 is connected to the DAC output terminal DACgain.

  The predecoder 720 is supplied with a control signal J720 including a DAC mode switching signal from the communication / timing controller 20 and a gain setting input code Dgain for gain setting. The predecoder 720 divides the input m-bit gain setting input code Dgain into upper s-bit information and lower t-bit information. As a result, the pre-decoder 720 outputs the lower t-bit information in the m-bit digital signal to the lower-bit control unit 730, and the upper s-bit information in the m-bit digital signal is output to the upper-bit control unit 740. The

  The lower bit control unit 730 includes a decoder 732 that generates a selection control signal for selecting each lower current source cell 753 of the lower current source cell unit 752 based on the lower t bit information supplied from the predecoder 720. A glitch suppression processing unit 736 including a plurality of glitch suppression circuits that suppress glitches that may appear in the output of the second DA conversion unit 304. The glitch suppression circuit is the same as that in the glitch suppression processing unit 336, and the same number (t) of current source cells 753 as that of the lower current source cell unit 752 are provided.

  The decoder 732 outputs the low-order t-bit information passed from the pre-decoder 720 as it is to provide a selection control signal for selecting each low-order current source cell 753 of the low-order current source cell unit 752. This is because a current source cell (binary cell) weighted by 2 bits is prepared for the lower bit side.

  That is, the decoder 732 is provided so as to be able to execute DA conversion corresponding to 1 LSB based on t-bit data supplied from the predecoder 720. Specifically, the decoder 732 has a lower order of (t−1) bits. A binary control method for supplying bit binary output, that is, bit data of 1/2, 1/4,..., 1/2 ^ (t-1) as a selection control signal to a corresponding glitch suppression circuit of the glitch suppression processing unit 736. Adopted. The decoder 732 functions as a selector for the t lower current source cells 753 provided in the lower current source cell unit 752.

  The upper bit control unit 740 includes a decoder 742 that generates a selection control signal for selecting each upper current source cell 755 of the upper current source cell unit 754 based on the upper s-bit information supplied from the predecoder 720. And a glitch suppression processing unit 746 that includes a plurality of glitch suppression circuits that suppress glitches that may appear in the output of the second DA conversion unit 304. The glitch suppression circuit is the same as that in the glitch suppression processing unit 346, and is provided in the same number (2 ^ t) as the number of upper current source cells 755 provided in the upper current source cell unit 754.

  The decoder 742 fully decodes (data converts) the upper s-bit multi-bit digital signal supplied from the predecoder 720 into a decimal number, and outputs the decoded value to each upper current source cell 755 of the upper current source cell unit 754. Output as a selection control signal for selection. The decoder 742 functions as a selector for 2 ^ s upper current source cells 755 provided in the upper current source cell unit 754.

  Each output of the decoder 742 is supplied to the corresponding glitch suppression circuit of the glitch suppression processing unit 746 as a selection control signal. The glitch suppression processing unit 746 supplies each output to the corresponding upper current source cell 755 of the upper current source cell unit 754 after performing the glitch suppression process by each glitch suppression circuit.

  Specifically, as shown in FIG. 7, of the m-bit digital code set from the outside, the upper s bits are decoded into a thermo-type by a decoder 742, and a current output type DA converter (second DA conversion unit) 304) the upper current source cell section 754. In the thermo type, for example, when the input code is 7 bits, the decoded output is connected to 2 ^ 7-1 = 127 current source cells. When the input is “0000001”, one current source cell is turned on, when “0000100” is turned on, eight current source cells are turned on, and when “11111111” is turned on, 127 current source cells are turned on. ing. On the other hand, as described above, the lower t bits are directly input to the lower current source cell unit 752 having a binary cell configuration. Therefore, a current value proportional to the external input code is output as the output current.

  The second DA converter 304 having such a configuration divides the m-bit digital signal into upper s bits and lower t bits (m = s + t), and the upper current source cell 755 has the same weight in the upper s bits. An upper current source cell unit 754 having a matrix type current source cell configuration in which the current values are uniformly assigned in accordance with the upper digital signal is configured, and the upper current source cell unit 754 is converted into a decimal number by the upper bit control unit 740. The lower-order current source cell unit of the parallel-type current source cell, which is controlled in accordance with the decoding method of the above and weighted so as to generate a current value having a power of a fraction of 2 with respect to the lower-order current source cell 753 752 is configured, and the lower-order current source cell unit 752 is controlled by the lower-order bit control unit 730 in a binary manner.

  That is, the DA converter 300 including the first DA converter 302 and the second DA converter 304 operates as a clock counter as the first DA converter 302 in order to count bits reliably with high speed. By using the frequency division processing unit 332 and the upper bit control unit 340, the necessary number of bits is divided into upper i bits and lower j bits. Regarding the lower j bits, the frequency division processing unit 332 generates a power of 2 and functions as a selector for the j lower current source cells 353 provided in the lower current source cell unit 352. The frequency division processing unit 332 selectively drives the lower current source cell 353 of the lower current source cell unit 352. For the upper i bits, the shift register of the upper bit control unit 340 is activated / inactivated in a predetermined direction for each clock based on the shift clock CKsr from the frequency division processing unit 332, thereby generating an upper current source. The upper current source cell 355 of the cell unit 354 is selectively driven.

  By doing so, the operation of the lower current source cell unit 352 controlled by the binary counter method and the upper current source cell unit 354 controlled by the decoding method are performed in conjunction with each other, and the input digital signal becomes faster. However, the current source cell section 350 divided by the binary method and the decoding method can be operated almost simultaneously, and as a result, the current source cells 353 and 355 corresponding to the input digital signal are selected at high speed and reliably. To be able to do. As a result, even during high-speed operation, it is possible to prevent the occurrence of glitches and miscodes, and a stable DA conversion operation can be realized.

  In other words, the first DA converter 302 of the present embodiment divides the n-bit digital signal into upper i bits and lower j bits (n = i + j), and the upper bit has the same weight for the upper current source cell 355. An upper current source cell unit 354 having a matrix type current source cell configuration in which current values are uniformly assigned in accordance with an upper digital signal is configured, and the upper current source cell unit 354 is decoded by an upper bit control unit 340. The lower-order current source cell unit 352 of the parallel-type current source cell that is weighted so as to generate a current value having a power that is a power of two to the lower-order current source cell 353 in the lower-order bits is configured The lower-order current source cell unit 352 is controlled by the lower-order bit control unit 330 using a binary counter method.

  Then, the upper bit control unit 340 shifts the built-in shift register in the direction corresponding to the carry or the carry in conjunction with the carry or the carry in the lower bit control unit 330, thereby corresponding to the input digital value. The main feature is that the selection operation of the lower upper current source cell 353 and the upper upper current source cell 355 is performed almost simultaneously. By doing so, it is possible to prevent the occurrence of glitches and miscodes, obtain a stable analog reference signal, and improve the single slope integration type AD conversion accuracy using this analog reference signal.

  In addition, when single-slope integration AD conversion is performed using the reference signal generated by the first DA converter 302, each current source cell 753 of the current source cell unit 750 is converted by the second DA converter 304. , 755 is digitally controlled to obtain digital data after adjusting the analog gain with high accuracy for the pixel signal to be processed.

  In particular, the current source cell of the first DA converter 302 as a slope type DA converter is configured with NMOS, and the current source cell of the second DA converter 304 as a PGA unit is configured with PMOS, so that output is performed on the power supply side. The accuracy can be increased. This is on the low illuminance side when considered as a pixel signal. Therefore, the accuracy of the slope-type DA converter composed of NMOS current source cells is ensured on the low illuminance side and is suitable for a solid state device. The reason why the output accuracy is high on the power supply side is that the transistors constituting the operation switch and the current source cell operate stably because the power supply side is always in a saturated state. Further, in order to prevent the output current of the first DA converter 302 formed of NMOS current source cells from being influenced by process variations and temperature, the potential generated by the PMOS current source cell in the second DA converter 304. Interpolation relationship can be constructed by giving That is, in order to ensure accuracy on the low illuminance side and to be strong against variations, it is highly advantageous to configure the current source cell of the first DA converter 302 with NMOS.

  Hereinafter, details of the main functional units related to the second DA conversion unit 304 and details of the operations will be specifically described.

<Basic configuration of current source cell>
FIG. 8 is a diagram showing a basic configuration example (conceptual diagram of the basic current source cell) of each of the current source cells 353, 355, 753, and 755 provided in the current source cell units 350 and 750. 8A shows a basic current source cell 500 corresponding to the current source cells 353 and 355, and FIG. 8B shows a basic current source cell 800 corresponding to the current source cells 753 and 755. .

  The basic current source cell 500 operates by receiving a complementary signal in accordance with the lower bit control unit 330 and the upper bit control unit 340 which are complementary output types.

  For example, as shown in FIG. 8A, the basic current source cell 500 includes a unit current source 510 and a changeover switch 520 that switches the output current of the unit current source 510.

  The unit current source 510 includes an NMOS transistor 512 that functions as a reference current source. The changeover switch 520 is configured by differentially connecting two NMOS transistors 524 and 526. The transistor 524 functions as the differential switch 1, and the transistor 526 functions as the differential switch 2.

  The transistor 512 has a source terminal connected to the analog ground line 590 and a drain terminal commonly connected to the source terminals of the transistors 524 and 526. In addition, a bias voltage Vbais applied in common to all the cells is applied to the gate terminal from the voltage amplitude control unit 760 through the current control line 592 in a current mirror system, and the current that the transistor 512 flows by the potential difference Vgs. The value of depends on.

  The two transistors 524 and 526 constituting the changeover switch 520 are supplied with respective complementary control signals Qin and xQin (where x represents a logic inversion signal) at their gate terminals, respectively, and lead lines 594 at their drain terminals. , 596 are connected. For example, the active H control signal (non-inverted input) Qin is input to the gate terminal of the transistor 524, the drain terminal thereof is connected to the lead line 594, and the lead line 594 is used as the selection output line 396 for current output. The On the other hand, a control signal (inverted input) xQin is input to the gate terminal of the transistor 526, its drain terminal is connected to the lead line 596, and this lead line 596 is connected to the power supply Vdd. As a whole, the active output is input as the control signal (non-inverted input) Qin to the selection output line 396 related to the DA conversion, and the transistor 524 is turned on so that the current source cell is turned on.

  On the other hand, similarly to the basic current source cell 500, the basic current source cell 800 operates by receiving complementary signals in accordance with the lower bit control unit 730 and the upper bit control unit 740 that are complementary output types. ing.

  For example, as shown in FIG. 8B, the basic current source cell 800 has a configuration in which the channel of the transistor in the basic current source cell 500 is reversed. Specifically, first, the unit current source 810 and And a changeover switch 820 for switching the output current of the unit current source 810.

  The unit current source 810 includes a PMOS transistor 812 that functions as a reference current source. The changeover switch 820 is configured by differentially connecting two PMOS transistors 824 and 826. The transistor 824 functions as the differential switch 1, and the transistor 826 functions as the differential switch 2.

  The transistor 812 has a source terminal connected to the power supply line 891 and a drain terminal commonly connected to the source terminals of the transistors 824 and 826. A bias voltage Vbaisgain applied to all the cells is applied to the gate terminal via the current control line 892, and the value of the current flowing through the transistor 812 is influenced by the potential difference Vgs.

The two transistors 824 and 826 constituting the changeover switch 820 receive complementary control signals Qin and xQin (x represents a logic inversion signal) at their gate terminals, respectively, and lead lines 894 at their drain terminals, respectively. , 896 are connected. For example, an active-L control signal (non-inverted input) Qin is input to the gate terminal of the transistor 824, its drain terminal is connected to the lead line 894, and the lead line 894 is used as the selection output line 796 for current output. Connected to DAC output terminal DACgain. On the other hand, a control signal (inverted input) xQin is input to the gate terminal of the transistor 826, its drain terminal is connected to the lead line 896, and this lead line 896 is connected to the analog ground line 890. As a whole, the active output is input as the control signal (non-inverted input) Qin to the selection output line 796 related to the DA conversion, and the transistor 824 is turned on so that the current source cell is turned on.

<Details of connection mode between first DA converter and second DA converter>
FIG. 9 is a diagram for explaining the details of the connection mode between the first DA converter 302 and the second DA converter 304. As described with reference to FIG. 4, on the output side of the second DA converter 304, the combined current added at the DAC output terminal DACgain of the second DA converter 304 is converted into a voltage signal (bias voltage Vbais). Based on the converted voltage signal, a gain adjustment current source cell 308 that functions as a current / voltage conversion unit that controls the operating current value of each current source cell 353, 355 on the first DA conversion unit 302 side is provided. The conversion unit 302 and the second DA conversion unit 304 are connected by a current mirror via a gain adjustment current source cell 308.

  Each of the current source cells 353 and 355 in the current source cell unit 350 employs the basic current source cell 500 shown in FIG. 8A, but the operating current supplied to each current source cell and the number of combinations thereof are adjusted. Thus, the current source cells 353 and 355 for generating the bit corresponding weight current values are configured. Basically, by adjusting the potential applied to the gate of the transistor 512, a basic current source having a current value superimposed on a predetermined output current is prepared, and a gate of the same potential corresponding to the basic current source is prepared. A current source cell capable of outputting a current proportional to an input power of 2 or 1 / power of 2 is provided.

  Specifically, first, the lower-order current source cell 353_1 having a current value of 1/2 corresponding to the 1/2 ^ (j-1) frequency-divided clock is changed to the basic current source cell 500 shown in FIG. Form with the structure itself. The magnitude of the output current (1/2 current value) depends on the potential input to the gate of the transistor 512 and is controlled by the voltage amplitude controller 360 as described above.

  By providing two cells similar to the lower current source cell 353_1 in parallel, the upper current source cell 355 having a current value of “1” provided in the upper current source cell unit 354 is provided. The upper current source cell unit 354 is configured by preparing i upper current source cells 355 (or y may be further added).

  For the lower current source cells 353_2,..., 355_j weighted to 1/4,..., 1/2 ^ j, first, the output current of the lower current source cell 353_1 having 1/2 output current is used as a reference. Then, a current source (in particular, a relay current source cell) that outputs a current weighted to 1/2 ^ j is provided by dividing the current mirror into 2 ^ (j-1). Then, by providing the basic current source cells 500 having the configuration shown in FIG. 8 (A) operating at the current value of 1/2 ^ j in parallel by the amount corresponding to the predetermined weight, 1/4,... , 1/2 ^ j, lower current source cells 353_2,..., 355_j that generate constant currents are formed. In this way, it is possible to configure a current source cell that outputs currents having different weights with high accuracy using only elements of the same size that are easy to achieve relative ratio accuracy.

  Although the relay current source cell is not shown in the figure, for example, an NMOS type transistor in which the same potential is supplied to the gate of the lower current source cell 353_1 and the upper current source cell 355 which is a basic current source, Two PMOS transistors that are arranged on the output side (drain terminal side) and are current mirror connected, and two that are arranged on one output side (drain terminal side) of the PMOS transistor and have a gate and a drain connected to each other. ^ (J-1) NMOS transistors (referred to as final stage transistors).

  With this configuration, the final stage transistor of the relay current source cell functions as a current source that outputs a current weighted to 1/2 ^ j. The gates of MOS-type final stage transistors connected in parallel with each other are controlled with the same reference voltage to operate at a constant current, and a branch path of a current source cell is formed by a plurality of final stage transistors having the same characteristics, thereby achieving high accuracy. A current branch can be formed.

  Further, one of the final stage transistors and the individual transistors 512 in the lower-order current source cells 353_2,..., 355_j are configured in a current mirror configuration, and the basic current source cell 500 having the configuration shown in FIG. For example, the lower current source cell 353_2 outputs a current having a weight of 1/4 (= 1/2 ^ 2), and the lower current source cell 353_3 A current having a weight of 1/8 (= 1/2 ^ 3) is output, and the lower current source cell 353_4 outputs a current having a weight of 1/16 (= 1/2 ^ 4). 353_j (j = 5) outputs a current having a weight of 1/32 (= 1/2 ^ 5).

  According to such a configuration, an element whose size and shape are extremely deformed is not used, and the number of elements in the upper current source cell unit 354 corresponding to the upper bits of the digital input signal is not significantly increased. Using only the same-sized elements that are easy to obtain relative ratio accuracy, current source cells with different weights can be configured with high accuracy, and the resolution of DA conversion can be increased without increasing the circuit scale so much. .

  The upper current source cell unit 354 selects a number corresponding to the data value (decimal number) of the upper bit of the digital input signal, and the lower current source cell unit 352 selects the bit value (bit) of the lower bit of the input signal. The bit resolution can be increased by adding a lower-order current source cell.

  In addition, as the lower-order current source cell unit 352, a lower-order current source that generates a current value having a weight of 1/2 with respect to a basic current corresponding to a digit value of a predetermined digit (the weight “1” in the sixth bit in the previous example). The cell 353_1 and the basic current are shunted to a branching path (corresponding to the transistor 532 in the previous example) equal to a power of 2 (1/2 ^ j), so that 2 By taking an exponential current as an output current, a highly accurate current source cell can be obtained with elements of the same size.

  In addition, the current source cell is configured by a constant current circuit using MOS transistors, and the current weighting of the constant current circuit is configured by the number of parallel connection of a plurality of MOS transistors having the same characteristics, thereby affecting the influence of manufacturing variations. High relative accuracy can be obtained without receiving.

  In this example, the upper current source cell 355 and the lower current source cell 353_1 are configured with a current source cell that outputs a weighted current value of 1/2 as a basic element, and the operating current value of the lower current source cell 353_1 is 2 A relay current source cell that outputs a weight current value of 1/2 ^ j (1/32 because j = 5 in the previous example) by dividing into ^ (j-1) is formed. The remaining lower current source cells 353_2 to 355_j in the lower current source cell unit 352 are configured with the current source cell that outputs the weighted current value as a basic element. However, this is only an example and is a power of two. Any specific configuration may be used as long as it can output a current weighted with the. However, as in the above description, it should be noted that currents having different weights can be output with high accuracy.

  Each of the current source cells 353 and 355 is controlled by a signal output from the inverting output terminal xQ of the D-FF 610 constituting the frequency division processing unit 332 or the shift register unit 342. Specifically, the inverting output terminal xQ of the D-FF 610 is input to the corresponding glitch suppression processing units 336 and 346 configured by cascade-connected inverters 382 and 384, and undergoes logic inversion and delay processing. A glitch suppression process is performed.

  The glitch suppression processing units 336 and 346 having such a configuration include the selection control signal for the transistor 524 (differential switch 1) not delayed by the inverter 384 and the transistor 526 (differential switch 2) delayed by the inverter 384. ) And the inverted selection control signal are supplied as complementary signals to transistors 524 and 526 forming a differential switch.

  If the transistors 524 and 526 that operate as differential switches of the current source cells 353 and 355 are controlled via the glitch suppression processing units 336 and 346, the gate input of the transistor 526 that is the output of the inverter 384 is greater than the output of the inverter 382. Since the transistor 524 is activated and the transistor 526 is deactivated after the delay Δt0 because the transistor 524 is activated (high level) or deactivated (low level) after the delay Δt0 period, both the transistors 524 and 526 are turned off. From this time, the state where the transistor 524 is turned on can be surely avoided, and the occurrence of glitch can be suppressed. This is because the output from the current source cell has a structure that can always output current through one of the switches (transistors 524 and 526).

  On the other hand, each of the current source cells 753 and 755 in the current source cell unit 750 employs the basic current source cell 800 shown in FIG. 8B, but the operating current supplied to each of them and the number of combinations thereof are adjusted. Thus, the current source cells 753 and 755 for generating the bit-related weighting current value are configured. Basically, by adjusting the potential applied to the gate of the transistor 812, a basic current source having a current value superimposed on a predetermined output current is prepared, and a gate having the same potential corresponding to the basic current source is prepared. A current source cell capable of outputting a current proportional to an input power of 2 or 1 / power of 2 is provided. Such a configuration and the effects obtained thereby are the same as those of the current source cells 353 and 355 constituting the current source cell unit 350. Here, the same explanation is omitted.

  Preferably, the current source cells 353 and 355 constituting the current source cell unit 350 of the first DA converter 302 and the current source cells 753 and 755 constituting the current source cell unit 750 of the second DA converter 304 are combined. It is preferable to use transistors of the same size. This is due to the following reason.

  That is, the potential generated by the PMOS current source cell in the second DA converter 304 so that the output current of the first DA converter 302 composed of the NMOS current source cell is not affected by process variations or temperature. In order to suppress the variation by constructing the interpolating relationship, it is necessary to consider the difference between the inherent capabilities of the NMOS and the PMOS.

  For example, first, the operation including the differential switch only when the bias setting of the transistor of the first DA converter 302 as the slope type DA converter composed of NMOS transistors is between Vgs = Vth + 0.3V and Vgs = Vth + 0.7V. Operates stably (= saturated). If the output current of one current source cell of the slope DA conversion unit (first DA conversion unit 302) of Vgs = Vth + 0.7V is 30 uA and the output current of the second DA conversion unit 304 as the PGA unit is 300 uA, it will be folded. The ratio is 10. At this time, if the capability of the NMOS transistor is twice the capability of the PMOS transistor, if the folding ratio of the slope DA converter (first DA converter 302) and the PGA unit (second DA converter 304) is set to 5, A complementary relationship is established and a configuration that is resistant to variation can be adopted.

  Each current source cell 753, 755 is controlled by a signal output from the inverting output terminal xQ of the decoder 732 or the decoder 742. The inverted output terminal xQ is input to the corresponding glitch suppression processing units 736 and 746 configured by cascade-connected inverters 382 and 384, and undergoes logic inversion and delay processing to perform glitch suppression processing. The operations of the glitch suppression processing units 736 and 746 are the same as the operations of the glitch suppression processing units 336 and 346.

  Specifically, the gain adjustment current source cell 308 functioning as a current / voltage conversion unit arranged between the first DA conversion unit 302 and the second DA conversion unit 304 is a predetermined number (here, k pieces). A combination of parallel connections of transistors (particularly referred to as a current distribution transistor / transistor 790 in the figure). Then, one of the current distribution transistors and the individual transistors 512 in the lower current source cells 353_2,..., 355_j are connected in a current mirror configuration. By doing so, the current Igain added and output at the DAC output terminal DACgain of the second DA converter 304 can be divided into k.

  With this configuration, each of the k current distribution transistors constituting the gain adjustment current source cell 308 functions as a current source that outputs a current weighted by Igain / k. By controlling the gates of MOS-type current distribution transistors connected in parallel to each other with the same reference voltage and operating at a constant current, and forming a branch path of a current source with a plurality of current distribution transistors having the same characteristics, a highly accurate current A branch path can be formed.

  Further, one of the current distribution transistors and the individual transistors 512 in the lower-order current source cells 353_2,..., 355_j are configured in a current mirror configuration, and the basic current source cell 500 having the configuration shown in FIG. , They are provided in parallel corresponding to the respective weight current values.

  According to such a configuration, by adjusting the k number, it is possible to adjust the current / voltage conversion ratio in the gain adjustment current source cell 308 functioning as a current / voltage conversion unit.

<Overview of overall operation>
FIG. 10 is a diagram for explaining the overall operation of the DA converter 300 as described above. Here, in particular, an overall operation outline of the first DA converter 302 centering on the lower bit controller 330 and the upper bit controller 340 will be described using a timing chart. Here, the case where the lower j bits are 5 and the upper current source cell unit 354 is provided with 2 ^ i upper current source cells 355 is shown.

  Assume that the current source cells 353 and 355 are turned on when the outputs of the frequency division processing unit 332 and the shift register unit 342 are at a low level. Therefore, at the initial value, the outputs of the frequency division processing unit 332 and the shift register unit 342 are at the H level, and all the current source cells 353 and 355 are turned off.

  First, the first DA converter 302 operates under the control of the count clock CKdac, the input digital signal Din, and the control signal J320 supplied from the communication / timing controller 20 to the block controller 320. As a preparatory stage, the block control unit 320 first decodes the input digital signal Din and stops supplying the internal count clock CKcnt to the lower bit control unit 330. Further, the block control unit 320 controls output of DA conversion by operating a set signal and a reset signal supplied to the upper bit control unit 340 and the lower bit control unit 330.

  At this time, the second DA conversion unit 304 performs DA conversion processing according to the input gain setting input code Dgain, and applies the bias voltage Vbais corresponding to the added current in the current source cell unit 750 corresponding to the gain setting input code Dgain. This is supplied to the current control line 592 of the 1DA converter 302. As a result, the first DA converter 302 generates a reference signal that changes with an inclination in accordance with the gain setting controlled by the second DA converter 304.

  Specifically, first, on the first DA conversion unit 302 side, in the initial state, the block control unit 320 cuts off the internal count clock CKcnt to the lower bit control unit 330 (fixed to the H level). All D-FFs functioning as frequency dividers in the frequency processing unit 332 and all shift registers (actually D-FFs) in the shift register unit 342 are reset.

  Next, the block control unit 320 supplies the internal count clock CKcnt to the lower bit control unit 330. As a result, the frequency division processing unit 332 starts counting operation in synchronization with the internal count clock CKcnt, and the output of each frequency divider (the non-inverted output Q of the D-FF 610) is supplied to the glitch suppression processing unit 336. The phase is inverted by the inverter 382 of the glitch suppression processing unit 336, and the timing is controlled by the inverter 384 functioning as a delay element. Then, two clocks Q and xQ which are complementary signals are input to a differential switch (transistors 524 and 526) incorporated in the corresponding lower current source cell 353, and 1/2 ^ k (k is from j to 1). The lower current source cell 353 superimposed on is turned on.

  For example, the switching operation (L → H or H → L) of the 1/2 ^ k frequency divider in the frequency division processing unit 332 is performed at the rising edge every 2 ^ (k−1) of the input clock. ^ K input clocks complete one cycle. This is the ground for 1/2 ^ k division.

  At the same time, the frequency-divided clock divided by ½ (j−1) by the frequency division processing unit 332 is shaped into the shift clock CKsr by the shift control unit 333, and then the frequency division processing unit 332 of the upper bit control unit 340. And is used to switch the uniformly weighted lower current source cell 353.

  Here, in the shift register unit 342, a shift operation is performed using the shift clock CKsr generated based on the 1/16 frequency-divided clock output from the 1/16 frequency divider of the frequency division processing unit 332. Every 16 clocks, the output of the shift register is activated sequentially. For example, the non-inverted output of the shift register transitions to the H level and the inverted output transitions to the L level. First, the first upper current source cell 355_1 is turned on. Subsequently, every time the divided clock 16 changes to H level, the non-inverted output of the shift register sequentially changes to H level and the inverted output changes to L level, and the corresponding k-th lower current source cell 353_k is turned on. As long as the control signals J330 and J340 do not output a reset signal, the operation continues until the last 2 ^ i-th upper current source cell 355_2 ^ i is turned on.

  As described above, the DA conversion unit 300 performs DA conversion for reliably engraving the gradation in synchronization with the internal count clock CKcnt by the cooperative operation based on the internal count clock CKcnt of the lower bit control unit 330 and the upper bit control unit 340. The reference voltage that functions as a circuit and changes in a direction that gradually increases with a slope according to the gain setting set by the second DA converter 304 can be generated with high accuracy.

  Here, in order to generate a reference voltage that changes in a gradually increasing direction, the current source cells 353 and 355 are increased in the direction in which the added current by the selected current source cells 353 and 355 in the current source cell unit 350 increases. Although the selection is controlled, if the selection of the current source cells 353 and 355 is controlled in the direction in which the added current by the selected current source cells 353 and 355 in the current source cell unit 350 is decreased, the selection is gradually decreased. A changing reference voltage can be generated.

<Example of gain adjustment by the second DA converter>
11 and 12 are diagrams for explaining an example of gain adjustment at the time of AD conversion processing using the second DA converter 304. FIG. Here, FIG. 11A is a diagram illustrating a state of the output current Igain of the second DA conversion unit 304 with respect to the gain setting input code Dgain which is an external digital input value. FIG. 11B is a diagram illustrating a state of the gain control bias voltage Vbais output from the gain adjustment current source cell 308 with respect to the gain setting input code Dgain which is an external digital input value. FIG. 11C is a diagram illustrating a state of the reference signal RAMP at a certain gain setting value generated by the first DA conversion unit 302 for the input digital signal Din. FIG. 11D is a diagram showing the state of the reference signal RAMP when the gain setting is variously changed by the second DA conversion unit 304.

  Here, the output current Igain of the second DA converter 304 that is uniquely determined for a certain input n-bit value is input to the gain adjustment current source cell 308 that functions as a current-voltage converter. The output current Igain of the second DA converter 304 is expressed by the following equation (3), where Icell_gain is the output current of each of the current source cells 753 and 755 (for the upper current source cell 755, the combined component corresponding to bit data). As shown in FIG. 11A, it changes linearly.

  When this output current Igain is supplied to the folding transistor 790 (or 792) constituting the gain adjustment current source cell 308, the bias voltage Vbais is uniquely determined as shown by the following formula (4). Here, Vth is a threshold voltage, and K is a constant unique to the transistor. As can be seen from the equation (4), the bias potential Vbais is proportional to √ times the output current Igain, and outputs a voltage value multiplied by √ as shown in FIG.

  The first DA converter 302 operates with an operating current defined by the bias potential Vbais. The output voltage Vout is defined by the product of the addition current Idac output from the first DA converter 302 and the resistance value Rref of the reference resistor 398, and is expressed by the following equation (5).

  Here, the added current Idac output from the first DA converter 302 is uniquely determined by the bias potential Vbais generated in the gain adjustment current source cell 308 based on the added current Igain output from the second DA converter 304. When the output current of each of the current source cells 353 and 355 (the combined component corresponding to the bit data with respect to the upper current source cell 355) is Icell_dac, the following expression (6) is obtained.

  As can be seen from the equation (6), the addition current Idac is controlled by the bias potential Vbais generated by the gain adjustment current source cell 308. Further, as can be seen from the equation (5), the amplitude of the output voltage Vout is increased by the addition current Idac. Is determined. That is, the gain setting input code Dgain uniquely determines the gain control bias voltage Vbais, which is the gate voltage of the current source cells 353 and 355 of the first DA converter 302, and uniquely determines the amplitude of the output voltage Vout. . As a result, as shown in FIG. 11C, the slope of the reference signal RAMP is uniquely determined with respect to the gain setting input code Dgain supplied to the second DA converter 304.

  Therefore, as shown in FIG. 11D, if the bias potential Vbais is controlled by the second DA converter 304 by adjusting the gain setting input code Dgain supplied to the second DA converter 304, the amplitude of the output voltage Vout, that is, The slope of the reference signal RAMP that changes linearly can be arbitrarily adjusted.

  Changing the slope of the reference signal RAMP means gain adjustment with respect to the reference signal RAMP, but as described above, the single slope integration type AD conversion process using the reference signal RAMP is a comparison target. It functions as a gain adjustment for the pixel signal. Therefore, the gain adjustment can be performed on the pixel signal with the bit accuracy of the gain setting input code Dgain, and the digital data after the gain adjustment is obtained as the AD conversion result in the column AD circuit 25. . If the gain setting input code Dgain is digitally controlled with high accuracy, as a result, gain adjustment can be performed with high accuracy for the analog signal to be subjected to AD conversion processing.

  In the above description, the constant gain setting input code Dgain is supplied to the second DA converter 304 during the AD conversion process, but this gain setting input code Dgain is dynamically adjusted during the AD conversion process. Then, the change characteristic (so-called inclination) of the DA output in the first DA converter 302 can be dynamically adjusted, and an AD conversion result with gamma correction added can be obtained.

  For example, in the diagram shown in FIG. 12, when the DAC mode switching signal, which is one of the control signals J720 supplied from the communication / timing controller 20 to the second DA converter 304, is at L level, the predecoder 720 DA conversion processing is performed based on one gain setting input code Dgain (indicated by 10b in the figure) designated by the timing control unit 20, and the bias voltage Vbais corresponding to the gain setting input code Dgain is converted to the first DA conversion unit 302. To supply.

  On the other hand, when the DAC mode switching signal is at the H level, the three gain setting input codes Dgain designated by the communication / timing control unit 20 are dynamically changed during the AD conversion processing in the column AD circuit 25. For example, DA conversion processing is initially performed based on the minimum value Dgain_min among the three gain setting input codes Dgain_max, Dgain_mid, and Dgain_min, and the minimum bias voltage Vbais corresponding to the minimum gain setting input code Dgain_min is set as the first DA conversion unit. 302 is supplied. As a result, the DA output from the first DA converter 302 (that is, the reference signal RAMP) changes more slowly than when the DAC mode switching signal = L (the inclination is small; indicated by 11b in the figure). When the DA output is used as a reference signal for AD conversion, the AD output data becomes larger when the inclination is small. This means that the analog gain is increased to perform AD conversion.

  Thereafter, the predecoder 720 of the second DA converter 304 switches the gain setting input code Dgain to be used to the intermediate value Dgain_mid when the DAC code in the second DA converter 304 reaches the predetermined value Da. At this time, the DA output by the first DA converter 302 changes with the same slope (indicated by 10b in the figure) as when the DAC mode switching signal = L.

  Further, when the DAC code reaches the predetermined value Db, the predecoder 720 of the second DA converter 304 switches the gain setting input code Dgain to be used to the maximum value Dgain_max. For this reason, the DA output by the first DA converter 302 changes faster than when the DAC mode switching signal = L (the slope is large; indicated by 9b in the figure)). When DA output is used as a reference signal for AD conversion, AD output data becomes smaller when the slope is large. This means that AD conversion is performed with a reduced analog gain.

  In this example, the DA output (reference signal RAMP in this example) by the first DA converter 302 is changed stepwise while having linearity, but the gain setting input code Dgain is further increased in multiple steps. If it is divided and finely changed, it can be changed gradually and gradually according to a higher-order function such as a quadratic function.

  As a result, for example, when capturing an image in a dark part with little input light, the dynamic range can be ensured by obtaining digital data after adjusting the gain (analog gain) of the image signal to be a high gain. it can. On the other hand, when imaging in bright areas where there is a lot of input light, the gain (analog gain) of the imaging signal is adjusted to a low gain, and digital data is obtained, thereby reducing the dynamic range while preventing saturation. Can be secured. In addition to realizing a wide dynamic range, gamma correction is applied to the sensitivity characteristics, and more natural sensor characteristics can be realized.

  As mentioned above, although this invention was demonstrated using embodiment, the technical scope of this invention is not limited to the range as described in the said embodiment. Various changes or improvements can be added to the above-described embodiment without departing from the gist of the invention, and embodiments to which such changes or improvements are added are also included in the technical scope of the present invention.

  Further, the above embodiments do not limit the invention according to the claims (claims), and all combinations of features described in the embodiments are not necessarily essential to the solution means of the invention. Absent. The embodiments described above include inventions at various stages, and various inventions can be extracted by appropriately combining a plurality of disclosed constituent elements. Even if some constituent requirements are deleted from all the constituent requirements shown in the embodiment, as long as an effect is obtained, a configuration from which these some constituent requirements are deleted can be extracted as an invention.

  For example, the specific configurations of the first DA converter 302 and the second DA converter 304 employed in the above embodiment are merely examples, and various other configurations can be employed. The slope of the reference signal used in the single slope integration type AD conversion method generated by the first DA conversion unit 302 may be adjusted by digital control by the second DA conversion unit 304. This modification is also possible with the present invention.

  In the above-described embodiment, the AD conversion function unit is provided in the column region located on the reading side of the pixel unit 10. However, the AD conversion function unit may be provided in other locations. For example, a configuration may be adopted in which pixel signals are output in analog up to the horizontal signal line 18, and then AD conversion is performed and passed to the output circuit 28.

  In the above-described embodiment, the count process is started from the final count value before the switching at the time of the count process after the mode switching. However, the synchronous output in which the count output value is output in synchronization with the count clock CK0 is used. When an up / down counter is used, this can be realized without requiring any special measures at the time of mode switching.

  However, when using an asynchronous up / down counter that has an advantage that the operation limiting frequency is determined only by the limiting frequency of the first flip-flop (counter basic element) and is suitable for high-speed operation, the count value is changed when the count mode is switched. Is destroyed, and the normal count operation cannot be performed continuously while maintaining the value before and after switching. Therefore, it is preferable to provide an adjustment processing unit that can start the count processing after the mode switching from the count value before the mode switching. Note that the details of the adjustment processing unit are omitted here. When addition processing is performed between a plurality of signals, it is only necessary to set the count modes of the preceding stage and the subsequent stage to be the same, and such a countermeasure is unnecessary.

  In the above-described embodiment, the pixel signal has a signal component Vsig after the reset component ΔV (reference component) for the same pixel as a time series, and the processing unit in the subsequent stage has a positive polarity (a larger signal level indicates a positive value). When a true signal component is obtained in response to what is processed for a large signal), as a first process, a comparison process and a down-count process are performed for the reset component ΔV (reference component), and a second process is performed. Although the comparison process and the up-count process are performed on the signal component Vsig, the combination and processing order of the target signal component and the count mode are arbitrary regardless of the time series in which the reference component and the signal component appear. Depending on the processing procedure, the digital data obtained in the second processing may become a negative value. In such a case, it is sufficient to take measures such as sign inversion or correction calculation.

  Of course, as the device architecture of the pixel unit 10, the reset component ΔV (reference component) must be read after the signal component Vsig, and when the subsequent processing unit processes positive signals, the first time It is efficient to perform the comparison process and the down-count process for the signal component Vsig as the above process, and the comparison process and the up-count process for the reset component ΔV (reference component) as the second process.

  Further, in the above-described embodiment, assuming that the signal component Vsig appears after the reset component ΔV (reference component) for the same pixel as a time series in the above embodiment, the addition calculation is performed between a plurality of pixel signals having different accumulation times. In doing so, the difference processing for obtaining the true signal component is performed for each pixel signal. However, if only the signal component Vsig may be targeted, for example, the reset component ΔV (reference component) can be ignored, the true processing is performed. It is possible to omit the difference processing for obtaining the signal component.

  Further, in the above embodiment, the up / down counter is used in common regardless of the operation mode, and the count process is performed by switching the processing mode. However, the count process is performed by combining the down count mode and the up count mode. However, the present invention is not limited to a configuration using an up / down counter capable of mode switching.

  In the above-described embodiment, a sensor configured by arranging unit pixels composed of NMOS or PMOS in a matrix is described as an example. However, the present invention is not limited to this, and the present invention is also applicable to a line sensor arranged in a row. It is possible to enjoy the same actions and effects as described in the above embodiment.

  In the above-described embodiment, as an example of a solid-state imaging device that can arbitrarily read signals from individual unit pixels by address control, a CMOS sensor including a pixel unit that generates signal charges by receiving light However, the generation of signal charges is not limited to light, and can be applied to electromagnetic waves such as infrared rays, ultraviolet rays, or X-rays in general, and outputs an analog signal corresponding to the amount of the received electromagnetic waves. The matters described in the above embodiments can be applied to a semiconductor device including unit components in which a large number of elements are arranged.

  In addition, the AD conversion circuit described as an example in the above embodiment is not limited to be provided by being incorporated in a solid-state imaging device or other electronic devices, and for example, an IC (Integrated Circuit), an AD conversion module, or data It may be provided as a single device, such as a processing module.

  In this case, an AD conversion device (or data processing device) including a comparison unit and a counter unit may provide the reference signal generation unit that generates a reference signal for AD conversion and supplies the reference signal to the comparison unit, or a counter The control unit that controls the count processing mode in the unit may also be provided by being incorporated in a module that is a combination of an IC (integrated circuit) or an individual chip disposed on the same semiconductor substrate.

By incorporating and providing these, it is possible to collectively handle the functional units necessary for controlling the operations of the comparison unit and the counter unit in realizing the function of generating the image signal, and handling and management of the members can be performed. It becomes simple. In addition, since elements necessary for AD conversion processing are integrated (integrated) as an IC or a module, it becomes easy to manufacture a finished product of a solid-state imaging device or other electronic devices.
In the above embodiment, the CMOS type solid-state imaging device that is sensitive to electromagnetic waves input from the outside such as light and radiation is exemplified. However, in any of the above embodiments, any device that detects a change in physical quantity can be used. The described mechanism can be applied, and is not limited to light or the like. For example, a fingerprint authentication device that detects fingerprint images based on changes in electrical characteristics or optical characteristics based on pressure for information related to fingerprints (Japanese Patent Application Laid-Open No. 2002-7984). In other mechanisms for detecting physical changes such as Japanese Patent Application Laid-Open No. 2001-125734 and the like, the above embodiment can be similarly applied as a mechanism for converting an analog signal into a digital signal.

  DESCRIPTION OF SYMBOLS 1 ... Solid-state imaging device, 3 ... Unit pixel, 7 ... Drive control part, 10 ... Pixel part, 12 ... Horizontal scanning circuit, 14 ... Vertical scanning circuit, 15 ... Row control line, 18 ... Horizontal signal line, 19 ... Vertical signal 20 ... communication / timing control unit 24 ... counter unit 25 ... column AD circuit 26 ... column processing unit 27 ... reference signal generation unit 27a, 27b ... DA conversion circuit (reference signal generation output unit) 28 ... Output circuit, 300 ... DA converter, 302 ... first DA converter, 304 ... second DA converter, 308 ... gain adjustment current source cell (current / voltage converter), 310,710 ... DAC controller (constant current source) Selection control unit), 320 ... block control unit, 330 ... lower bit control unit, 332 ... frequency division processing unit, 333 ... shift control unit, 336, 346 ... glitch suppression processing unit, 340 ... upper bit control unit, 342 ... Register, 350, 750 ... current source cell, 352, 752 ... lower current source cell, 353, 753 ... lower current source cell, 354, 754 ... upper current source cell, 355, 755 ... upper current source cell 720: Predecoder, 732, 742: Decoder

Claims (9)

A reference signal generation unit that generates a reference signal for converting an analog signal into a digital signal, a comparison unit that compares the analog signal and the reference signal generated by the reference signal generation unit,
In parallel with the comparison process in the comparison unit, the AD conversion apparatus includes a counter unit that performs a count process with a predetermined count clock and holds a count value when the comparison process in the comparison unit is completed,
The reference signal generator is
A first DA converter that obtains the reference signal according to a value of a digital input signal;
A second DA converter for obtaining an analog gain control output signal corresponding to the value of the digital gain control input signal,
An AD converter characterized by adjusting a gain of the reference signal generated by the first DA converter based on a gain control output signal generated by the second DA converter. The first DA converter includes a plurality of current source cells capable of outputting a constant current having a predetermined weight according to information supplied to a control input terminal, and the constant current output of the selected current source cell Output current corresponding to the value of the digital input signal is generated as the analog output signal.
The second DA converter includes a plurality of current source cells capable of outputting a constant current having a predetermined weight, and adds the constant current output of the selected current source cell to output the digital gain control input. A second DA converter that obtains an output current according to the value of the signal,
The AD converter according to claim 1, wherein the gain control output signal based on the output current generated by the second DA converter is supplied to the control input terminal. The AD converter includes a current / voltage converter that generates a voltage signal corresponding to the output current generated by the second DA converter,
The AD converter according to claim 2, wherein a voltage signal generated by the current / voltage converter is supplied to the control input terminal. A plurality of pixels;
A solid-state imaging device comprising: an analog-to-digital conversion device that converts analog pixel signals read from the plurality of pixels into digital signals,
The AD converter is
A reference signal generation unit that generates a reference signal for converting an analog signal into a digital signal;
A comparison unit that compares the analog signal with the reference signal generated by the reference signal generation unit;
In parallel with the comparison process in the comparison unit, the counter unit performs a count process with a predetermined count clock, and includes a counter unit that holds a count value when the comparison process in the comparison unit is completed,
The reference signal generator is
A first DA converter that obtains the reference signal according to a value of a digital input signal;
A second DA converter for obtaining an analog gain control output signal corresponding to the value of the digital gain control input signal,
A solid-state imaging device, wherein a gain of the reference signal generated by the first DA converter is adjusted based on a gain control output signal generated by the second DA converter. The first DA converter includes a plurality of current source cells capable of outputting a constant current having a predetermined weight according to information supplied to a control input terminal, and the constant current output of the selected current source cell Output current corresponding to the value of the digital input signal is generated as the analog output signal.
The second DA converter includes a plurality of current source cells capable of outputting a constant current having a predetermined weight, and adds the constant current output of the selected current source cell to output the digital gain control input. A second DA converter that obtains an output current according to the value of the signal,
The solid-state imaging device according to claim 4, wherein the gain control output signal based on the output current generated by the second DA converter is supplied to the control input terminal. The AD converter includes a current / voltage converter that generates a voltage signal corresponding to the output current generated by the second DA converter,
The solid-state imaging device according to claim 5, wherein a voltage signal generated by the current / voltage conversion unit is supplied to the control input terminal. A charge generation unit that generates a charge corresponding to an incident electromagnetic wave, and an effective region including a unit signal generation unit that generates an analog unit signal corresponding to the charge generated by the charge generation unit in a unit component, And as a functional element that converts the unit signal into a digital signal, a reference signal generation unit that generates a reference signal for converting the unit signal into a digital signal, and a reference generated by the unit signal and the reference signal generation unit A comparison unit that compares the signal, and a counter unit that performs a count process with a predetermined count clock in parallel with the comparison process in the comparison unit and holds a count value at the time when the comparison process in the comparison unit is completed A semiconductor device for physical quantity distribution detection provided,
The reference signal generator is
A first DA converter that obtains the reference signal according to a value of a digital input signal;
A second DA converter that obtains an analog gain control output signal corresponding to the value of the digital gain control input signal,
A semiconductor device comprising: adjusting a gain of the reference signal generated by the first DA converter based on a gain control output signal generated by the second DA converter. The first DA converter includes a plurality of current source cells capable of outputting a constant current having a predetermined weight according to information supplied to a control input terminal, and the constant current output of the selected current source cell Output current corresponding to the value of the digital input signal is generated as the analog output signal.
The second DA converter includes a plurality of current source cells capable of outputting a constant current having a predetermined weight, and adds the constant current output of the selected current source cell to output the digital gain control input. A second DA converter that obtains an output current according to the value of the signal,
The semiconductor device according to claim 7, wherein the gain control output signal based on the output current generated by the second DA converter is supplied to the control input terminal. The AD converter includes a current / voltage converter that generates a voltage signal corresponding to the output current generated by the second DA converter,
The semiconductor device according to claim 8, wherein a voltage signal generated by the current / voltage conversion unit is supplied to the control input terminal.

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