JP2009261060A - Driving circuit for power inverter circuit, and power conversion system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To solve the problem that it is difficult to suppress a surge caused by a recovery current. <P>SOLUTION: The voltage of a driving IC32 is applied to the gate of a power switching element Sw through a resistor 42, a parallel circuit composed of a coil 34 and a capacitor 36, and a coil 38a. When a voltage is applied by the driving IC 32, the gate voltage fluctuates, having three steps of rise, drop, and rise by a resonant phenomena of the coils 34 and 38a and a parasitic capacitor 40. Here, the drop period of the gate voltage and the decrease period of the recovery current are synchronized. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention includes a series connection body of a pair of switching elements on a high potential side and a low potential side, and a diode connected in antiparallel to each of the pair of switching elements, and the connection of the pair of switching elements For a power conversion circuit having an inductive load connected to a point, a drive circuit for a power conversion circuit including voltage applying means for applying a voltage to a conduction control terminal to turn on the switching element, and power including the drive circuit Concerning the conversion system.

  As this type of power conversion circuit, for example, an inverter including a series connection body of a pair of switching elements for connecting each phase of a three-phase motor to the positive electrode side and the negative electrode side of a DC power supply is well known. Here, a free wheel diode is connected in antiparallel to each of the pair of switching elements. Here, when a forward current flows through the freewheeling diode connected to one of the pair of switching elements, when the other switching element is turned on, the freewheeling diode is moved from the cathode side to the anode side. Recovery current flows. The recovery current increases and then decreases and eventually becomes zero, but when starting to decrease, a surge voltage is superimposed on both ends of the freewheel diode. This surge voltage becomes a noise source. Further, when the recovery current flows, a through current flows between the free wheel diode through which the recovery current flows and the other switching element, so that the power loss of the free wheel diode and the other switching element increases.

  Therefore, conventionally, as seen in Patent Document 1 below, when turning on the other switching element (IGBT), a capacitor is connected in parallel between the conduction control terminal (gate) and the emitter to charge the conduction control terminal. It has also been proposed to disconnect the capacitor along the way. Thereby, the recovery current can be reduced by reducing the charging speed in the first half of the charging period of the gate, and the surge can be suppressed.

  Conventionally, as seen in Patent Document 2 below, when a recovery current flows, it has also been proposed to place a switching element connected to a freewheel diode through which the recovery current flows. Thereby, the impedance between the cathode and the anode of the freewheel diode can be made low, and as a result, surge can be suppressed.

Further, conventionally, as can be seen in Patent Document 3 below, there has been proposed one that increases the voltage applied to the gate of the other switching element (IGBT) in two stages. Thus, by limiting the current by starting up the IGBT with a voltage at which the IGBT operates, the rate of change of the recovery current can be reduced, and thus the surge voltage can be suppressed.
JP 2006-324794 A JP 2007-267560 A Japanese Patent No. 2760590

  By the way, in the technique described in the above-mentioned patent document 1, since the rising speed of the gate voltage is reduced by connecting the capacitor, the time until the rising of the gate voltage is prolonged, and the power loss of the switching element may increase. There is. In the technique described in Patent Document 2, since one switching element is turned on, a short-circuit current flows between the other switching elements, which may increase the conduction loss of both switching elements. is there. In the technique described in Patent Document 3, the collector current is reduced in the other switching element due to the reduction in the recovery current, and the voltage between the collector and the emitter of this switching element is lowered. It is difficult to sufficiently suppress the voltage.

  The present invention has been made in order to solve the above-described problems. The object of the present invention is to connect a series connection body of a pair of high-potential side and low-potential side switching elements, and to each of the pair of switching elements in antiparallel. A power conversion circuit driving circuit capable of suitably suppressing a surge caused by a recovery current in a power conversion circuit including a connected diode and having an inductive load connected to a connection point of the pair of switching elements And providing a power conversion system including the drive circuit.

  Hereinafter, means for solving the above-described problems and the operation and effects thereof will be described.

  The invention according to claim 1 includes a series connection body of a pair of switching elements on a high potential side and a low potential side, and a diode connected in antiparallel to each of the pair of switching elements, and the pair of switching elements In the power conversion circuit in which an inductive load is connected to the connection point of the switching element, in the drive circuit of the power conversion circuit including voltage applying means for applying a voltage to the conduction control terminal to turn on the switching element, the voltage A parallel circuit of an inductor and a capacitor is provided between the applying means and the conduction control terminal.

  In the above invention, in order to provide a parallel circuit of an inductor and a capacitor, a voltage determined according to the capacitance of a capacitor component (such as a parasitic capacitor) between the conduction control terminal and the output terminal of the switching element and the capacitance of the capacitor of the parallel circuit Until then, the voltage of the conduction control terminal of the switching element changes at high speed. For this reason, the loss of the switching element at the time of switching a switching state from an OFF state to an ON state can be reduced. The voltage at the conduction control terminal of the switching element further changes due to the resonance phenomenon using the inductor. For this reason, when a recovery current flows through a diode connected in reverse parallel to one switching element, the current flowing through the switching element is limited compared to when the voltage at the conduction control terminal of the other switching element changes rapidly. Thus, the recovery current can be reduced. For this reason, the surge voltage resulting from the recovery current can be suitably suppressed.

  Furthermore, a parasitic inductor component usually exists between the voltage applying means and the conduction control terminal. Depending on how the resonance phenomenon occurs due to the parasitic inductor, the voltage of the conduction control terminal can be changed to the side where the switching element is turned off during the period in which the recovery current decreases. In this case, since the voltage between the input / output terminals of the switching element that is turned on can be maintained high during this period, the surge voltage caused by the recovery current can be more suitably suppressed.

  The invention according to claim 2 is the invention according to claim 1, further comprising an inductor connected in series to the parallel circuit between the voltage applying means and the conduction control terminal.

  When a recovery current flows through a diode connected to one switching element, the other switching element is turned on. Here, due to the resonance phenomenon using the inductors connected in series, the voltage between the conduction control terminal and the output terminal of the switching element that is turned on can be changed in the opposite direction to that during the turning on operation. As a result, the voltage between the input and output terminals of the switching element that is turned on can be maintained high during the period in which the recovery current decreases, and the surge voltage caused by the recovery current can be suitably suppressed.

  According to a third aspect of the present invention, in the second aspect of the present invention, the inductance of the inductor connected in series to the parallel circuit causes a forward current to flow in a diode connected in reverse parallel to one of the pair of switching elements. When the other switching element is turned on, the voltage at the conduction control terminal of the other switching element is reduced during a period in which the recovery current of the diode through which the forward current flows increases and then decreases. Is set so that it can be changed to an off-operation side.

  In the above invention, by setting the inductance of the inductors connected in series, the voltage between the input and output terminals of the switching element that is turned on can be kept high during the period in which the recovery current is reduced. Surge voltage to be suppressed can be suitably suppressed.

  According to a fourth aspect of the present invention, in the third aspect of the present invention, the inductance of the inductor connected in series to the parallel circuit has the conduction control terminal in the period when the current flowing through the switching element is equal to or less than a specified value. The voltage is set so that the voltage can be changed to the side of turning off the switching element.

  The surge that occurs when switching the switching element from the OFF state to the ON state tends to gradually increase and then gradually decrease as the current flowing between the input and output terminals increases. For this reason, the surge becomes most noticeable not when the current flowing between the input / output terminals of the switching element is maximum but when the current is relatively small. In this regard, in the above-described invention, when the surge becomes most prominent, it can be suitably reduced.

  The invention according to claim 5 is the invention according to any one of claims 1 to 4, further comprising a resistor between the voltage applying means and the conduction control terminal.

  When voltage is applied to the conduction control terminal of the switching element by voltage application means, the resonance phenomenon using the inductor causes the voltage of the conduction control terminal to be the voltage on the side that turns on the switching element and the side that turns off the switching element. Oscillates between voltages. Here, if this vibration does not attenuate, the switching element reluctantly switches between the on state and the off state even during a period in which it is desired to keep the switching element in the on state, that is, a period in which voltage is applied by the voltage applying means. May repeat. In this regard, in the above-described invention, by providing the resistor, the vibration of the voltage can be appropriately attenuated, and thus the above problem can be avoided.

  The invention according to claim 6 is the invention according to any one of claims 1 to 5, wherein between the voltage application means and the conduction control terminal, from the conduction control terminal side of the switching element to the voltage application means side. It further comprises a protective diode whose forward direction is the forward direction.

  According to the resonance phenomenon using the inductor, the voltage between the conduction control terminal and the output terminal of the switching element may exceed the voltage applied by the voltage applying unit. In this case, the voltage between the conduction control terminal and the output terminal of the switching element may exceed the withstand voltage. In this regard, in the above-described invention, since the protective diode is provided, it is possible to suitably avoid that the voltage between the conduction control terminal and the output terminal of the switching element greatly exceeds the applied voltage of the voltage application unit, and thus the conduction. It can be suitably avoided that the voltage between the control terminal and the output terminal exceeds the withstand voltage.

  The invention according to claim 7 is a power conversion system including the drive circuit for the power conversion circuit according to any one of claims 1 to 6 and the power conversion circuit.

  Since the power conversion system includes the drive circuit according to any one of claims 1 to 6, the power conversion system is a system capable of reducing a surge with a simple configuration.

(First embodiment)
Hereinafter, an embodiment in which a drive circuit of a power conversion circuit according to the present invention is applied to a drive circuit of a power conversion circuit of a hybrid vehicle will be described with reference to the drawings.

  FIG. 1 shows the system configuration of this embodiment. As shown in the figure, a motor generator 10 as an in-vehicle rotating machine is connected to a high voltage battery 12 via an inverter IV. The inverter IV is configured by connecting three series-connected bodies of a high-potential side power switching element Swp and a low-potential side power switching element Swn in parallel. The connection points of these series connection bodies are connected to the respective phases of the motor generator 10. Between the input / output terminals (between collector and emitter) of the high potential side power switching element Swp and the low potential side power switching element Swn, there is a high potential side freewheel diode FDp and a low potential side freewheel diode. The cathode and anode of FDn are connected.

  A drive circuit DC is connected to the conduction control terminals (gates) of the power switching elements Swp and Swn constituting the inverter IV. Thereby, the power switching elements Swp and Swn are driven by the microcomputer (microcomputer 20) using the low voltage battery 16 as a power source via the drive circuit DC and the interface 14. Here, the interface 14 includes an insulating means such as a photocoupler that insulates the high voltage system including the inverter IV and the converter CV from the low voltage system including the microcomputer 20. The microcomputer 20 controls operation signals gup, gvp, and gwp for operating the power switching element Swp for each of the U phase, V phase, and W phase of the inverter IV based on detection values of various sensors (not shown), and power switching. Operation signals gn, gvn, and gwn for operating the element Swn are generated and output. Thereby, the switching elements Swp and Swn are operated by the microcomputer 20 via the drive circuit DC. The high-potential side operation signals gup, gvp, and gwp of each phase and the low-potential side operation signals gun, gvn, and gwn are respectively a high-potential side switching element Swp and a low-potential side switching element. Swn may be driven complementarily to each other. In other words, during the period when one of the operation signals is a signal for turning on, the other operation signal may be a signal for turning off.

  Each of the power switching elements Swp and Swn is a switching element that has an input terminal and an output terminal that are uniquely defined and prevents a current from flowing from the output terminal to the input terminal. Specifically, these are constituted by insulated gate bipolar transistors (IGBT). For this reason, in a situation where a current can flow through the power switching element Swp on the high potential side, by turning it off, the current does not flow through the power switching element Swn on the low potential side, and this is antiparallel. A current flows through the freewheeling diode FDn connected to. Further, in a situation where a current can flow through the low potential side power switching element Swn, by turning it off, no current flows through the high potential side power switching element Swp, and this is in antiparallel. A current flows through the connected freewheeling diode FDp. In this case, when the power switching element Swp (Swn) is turned on again, a recovery current flows through the freewheel diode FDn (FDp). This recovery current causes a surge voltage when the power switching elements Swp and Swn are turned on. Hereinafter, this will be further described with reference to FIG.

  FIG. 2 illustrates a situation where a current can flow through the power switching element Swp on the high potential side. Under such circumstances, as shown in FIG. 2A, when the high-potential side power switching element Swp is in the OFF state, the collector current ic is zero, and the forward current flows through the low-potential-side freewheel diode FDn. id flows. At this time, since the voltage vd across the free wheel diode FDn is substantially zero, the voltage Vce between the emitter and the collector of the power switching element Swp on the high potential side is about the voltage Vdc of the high voltage battery 12.

  Here, when the power switching element Swp on the high potential side is turned on at time t1, the collector current ic gradually increases as shown in FIG. Along with this, the current id flowing through the free wheel diode FDn gradually decreases to zero. At this time, the voltage Vce between the collector and the emitter is obtained by subtracting the product of the current change rate of the collector current ic and the inductor component existing in the wiring of the inverter IV (schematically represented as wiring inductor Lp in the figure) from the power supply voltage Vdc. It becomes the value.

  When the current flowing through the free wheel diode FDn becomes zero, a recovery current that is a reverse current starts to flow due to the carrier accumulation effect of the free wheel diode FDn. At this time, a recovery current flows between the collector and the emitter of the switching element Swp on the high potential side in addition to the current (load current) exchanged with the motor generator 10. After that, the carrier at the junction of the free wheel diode FDn decreases, and the recovery current gradually changes from increasing to zero and eventually becomes zero. Here, when the recovery current changes from gradual increase to gradual decrease, a surge voltage is generated by the current change rate of the recovery current and the voltage component generated by the inductor component existing in the wiring of the inverter IV.

  FIG. 3 shows ideal waveforms for reducing the surge with respect to the gate voltages of the power switching elements Swp and Swn that are turned on. Specifically, FIG. 3A shows the current of the freewheel diode FDn (FDp) shown in FIG. 1, and FIG. 3B shows the power switching element Swp (Swn) shown in FIG. The transition of the voltage between the gate and emitter (gate voltage Vge) is shown. In the following, description will be given by taking as an example a situation in which a current can flow through the freewheeling diode FDn and the power switching element Swp.

  As shown in the figure, in the period T1, the gate voltage Vge is increased at a stretch to reduce the voltage between the input / output terminals (collector and emitter) of the power switching element Swp, and the conduction loss of the power switching element Swp is reduced. . At this time, using the wiring inductor Lp, the voltage Vdc of the high-voltage battery 12 and the collector current ic, the voltage Vce between the collector and the emitter becomes “Vce = Vdc−Lp · dic / dt”. Here, when the gate voltage Vge is increased at a stroke as described above, since the collector current ic increases rapidly, the current change rate dic / dt increases, and the voltage Vce decreases.

  In the subsequent period T2, the recovery current starts to flow. For this reason, the gate voltage Vge is held in a state lower than the period T1. As a result, the current flowing between the collector and emitter of the power switching element Swp is limited. For this reason, the peak value of the recovery current can be reduced, and the loss between the power switching element Swp and the free wheel diode FDn can be reduced.

  The gate voltage Vge is further reduced in the period T3 from when the recovery current starts to gradually decrease until it reaches zero. Here, when the recovery current of the free wheel diode FDn decreases, the collector current of the power switching element Swp also decreases. Here, when the gate voltage Vge is held, the voltage Vce between the collector and the emitter of the power switching element Swp is reduced as the collector current ic decreases, as shown by the one-dot chain line and the x mark in FIG. Will fall. On the other hand, the voltage Vd applied to the freewheeling diode FDn in the period T3 is “Vd = Vdc + Ldic / dt−Vce”, and therefore decreases as the voltage Vce increases. For this reason, the voltage Vd applied to the freewheel diode FDn increases as the voltage Vce decreases. Here, as shown in FIG. 3, by reducing the gate voltage Vge in the period T3, the voltage Vce can be kept high, as indicated by Δ in FIG. For this reason, the surge voltage applied to the free wheel diode FDn can be reduced. In addition, since the voltage applied to the wiring inductor can be reduced, the current change rate when the recovery current is reduced can be reduced, and the voltage Vd can be sufficiently reduced.

  Then, in the period T4 after the recovery current becomes zero, the gate voltage Vge is rapidly increased, so that the power switching element Swp is completely turned on to prevent an increase in switching element loss.

  Note that, in the period T1 shown in FIG. 3, as shown by the alternate long and short dash line, even if the gate voltage Vge is the same as that in the period T2, it is considered that the same effect as described above can be obtained.

  In the present embodiment, a voltage that approximates the ideal gate voltage waveform shown in FIG. 3 is applied to the gates of the power switching elements Swp and Swn by devising the charging path of the gate in the drive circuit DC. FIG. 5 shows a circuit configuration of the drive circuit DC according to the present embodiment. In FIG. 5, the power switching elements Swp and Swn are collectively expressed as a power switching element Sw, the free wheel diodes FDp and FDn are collectively expressed as a free wheel diode FD, and the operation signals gup, gvp, gwp , Gun, gvn, and gwn are expressed as operation signals g.

  The drive IC 32 uses the power supply 30 with the voltage value Vcc as a power supply source, and outputs a voltage with the voltage value Vcc as the operation signal g rises. A parallel circuit of a coil 34 and a capacitor 36 is connected between the power switching element Sw and the driving IC 32. For this reason, in the drive IC 32, the voltage of the voltage value Vcc is applied to the gate of the power switching element Sw through the parallel circuit with the rising of the operation signal g. In FIG. 5, the parasitic inductor 38 existing between the driving IC 32 and the gate and the parasitic capacitor 40 between the gate and the emitter are indicated by broken lines.

  FIG. 6 shows a simulation result of the waveform of the gate voltage Vge by the drive circuit DC. 6A, the inductance Lg1 of the coil 34 is “0.1 μH”, and the inductance Lg2 of the parasitic inductor 38 between the drive IC 32 and the gate is “0H”. In this case, the gate voltage Vge quickly rises to “Vcc · C / (C + Cp)” using the electrostatic capacitance C of the capacitor 36 and the electrostatic capacitance Cp of the parasitic capacitor 40. Thereafter, the gate voltage Vge repeatedly increases and decreases gradually due to the resonance phenomenon of the coil 34 and the parasitic capacitor 40. However, since the time scale (about “100 ns”) shown in FIG. 6A is shorter than the period of gradual increase and decrease, only a part of the gradual increase period is shown. The gate voltage waveform over a period longer than the time scale shown in FIG. 6A is the one shown by the one-dot chain line in the period T1, except for the period T3 in the gate voltage waveform shown in FIG. The waveform approximates this waveform.

  FIG. 6B shows a case where the inductance Lg2 of the parasitic inductor 38 has a finite value and is smaller than the inductance Lg1 of the coil 34. Specifically, here, the inductance Lg2 is set to about “20 to 30 nH”. As illustrated, in this case, the gate voltage Vge repeatedly rises, falls, and rises. Here, the period Tr from when the voltage rises to when it rises again and starts to rise again is determined depending on the inductance Lg2 of the parasitic inductor 38. On the other hand, the waveform of the gate voltage Vge having three steps of increase, decrease, and increase simulates the periods T1 to T4 in FIG. Therefore, if the period Tr in FIG. 6B is about the sum of the periods T1 to T3 in FIG. 3B, a waveform approximating the gate voltage waveform shown in FIG. 3B is realized. I think it can be done. In FIG. 6B, after the gate voltage Vge rises again, it greatly decreases again. This phenomenon is undesirable from the viewpoint of keeping the power switching element Sw in the on state. However, since the actual charging path between the drive IC 32 and the gate of the power switching element Sw includes a resistance component, the vibration of the actual gate voltage waveform is attenuated as compared with that shown in FIG. It will be a thing. For this reason, it is possible to keep the gate voltage Vge appropriate for keeping the power switching element Sw on.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) A parallel circuit of a coil 34 and a capacitor 36 is provided between the gate of the power switching element Sw and the driving IC 32. Thereby, the gate of the power switching element Sw can be charged at high speed to a voltage determined according to the capacitor component (parasitic capacitor 40) between the gate and emitter of the power switching element Sw and the capacitance of the capacitor 36. For this reason, the loss of the switching element at the time of switching a power switching state from an OFF state to an ON state can be reduced. Furthermore, depending on how the resonance phenomenon occurs due to the parasitic inductor 38 between the gate of the power switching element Sw and the driving IC 32, it is possible to reduce the voltage of the parasitic capacitor during the period in which the recovery current decreases. In this case, since the voltage Vce between the collector and the emitter of the power switching element Sw that is turned on can be kept high during this period, the surge voltage applied to the freewheel diode due to the recovery current is further increased. It can suppress suitably.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 7 shows a circuit configuration of the drive circuit DC according to the present embodiment. In FIG. 7, members corresponding to those shown in FIG. 5 are given the same reference numerals for convenience.

  As illustrated, in the present embodiment, a coil 38 a is provided in series with a parallel circuit of the coil 34 and the capacitor 36 in the charging path between the gate of the power switching element Sw and the drive IC 32. Thereby, the inductance Lg2 of the coil 38a can be freely adjusted. For this reason, the period Tr shown in FIG. 6B can be freely adjusted. As a result, after the gate voltage Vge once rises and then decreases and becomes near the minimum value, and after the recovery current gradually increases. The gradually decreasing timing can be synchronized with high accuracy.

  Specifically, the inductance Lg2 of the coil 38a is adjusted so that the synchronization can be achieved when the collector current of the power switching element Sw is equal to or less than the specified current. This is a setting in view of the fact that the surge voltage generated when switching the power switching element Sw from the off state to the on state tends to gradually increase and then gradually decrease as the collector current increases. Thereby, this can be reduced under the situation where the surge voltage becomes most prominent.

  According to this embodiment described above, in addition to the effect according to the above (1) of the first embodiment, the following effect can be obtained.

  (2) In the charging path between the gate of the power switching element Sw and the driving IC 32, the coil 38a is provided in series with the parallel circuit of the coil 34 and the capacitor 36. By adjusting the inductance Lg2 of the coil 38a, the voltage between the input and output terminals of the power switching element Sw can be maintained high during the period in which the recovery current is reduced, and the surge voltage caused by the recovery current is suitably increased. Can be suppressed.

  (3) The inductance Lg2 of the coil 38a is set so that the charge of the gate of the power switching element Sw can be extracted during a period in which the recovery current of the freewheel diode FD increases and then decreases. Thereby, the surge voltage resulting from the recovery current can be suitably suppressed.

  (4) The inductance Lg2 of the coil 38a is set so that charges can be extracted during the period when the current flowing through the power switching element Sw is equal to or less than the specified value. Thereby, when a surge voltage becomes the most remarkable, this can be reduced suitably.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 8 shows a circuit configuration of the drive circuit DC according to the present embodiment. In FIG. 8, members corresponding to those shown in FIG. 5 are given the same reference numerals for the sake of convenience.

  As illustrated, in the present embodiment, a resistor 42 is provided in series with a parallel circuit of a coil 34 and a capacitor 36 in a charging path between the gate of the power switching element Sw and the driving IC 32. The resistor 42 is a well-known linear element.

  FIG. 9 shows a simulation result of the waveform of the gate voltage Vge by the drive circuit DC. In FIG. 9A, the inductance Lg1 of the coil 34 is set to “0.1 μH”, and the inductance Lg2 of the parasitic inductor 38 between the drive IC 32 and the gate is set to “0H”. In this case, the gate voltage Vge quickly rises to a voltage determined according to the capacitance C of the capacitor 36 and the capacitance Cp of the parasitic capacitor 40. Although this rising speed is lower than that in the first embodiment, it is set sufficiently large so that the switching loss can be reduced. Thereafter, the gate voltage Vge repeatedly increases and decreases gradually due to the resonance phenomenon of the coil 34 and the parasitic capacitor 40. However, since the time scale (about “100 ns”) shown in FIG. 9A is shorter than the cycle of gradual increase and decrease, only a part of the gradual increase period is shown. Here, the difference between FIG. 9A and FIG. 6A is whether or not the vibration of the gate voltage Vge caused by the resistor 42 is attenuated.

  FIG. 9B shows a case where the inductance Lg2 of the parasitic inductor 38 has a finite value and is smaller than the inductance Lg1 of the coil 34. Specifically, here, the inductance Lg2 is set to about “20 to 30 nH”. In this case, the vibration of the gate voltage Vge is appropriately damped as compared with that shown in FIG. For this reason, the gate voltage Vge has three steps of increase, decrease, and increase, but no remarkable decrease thereafter. For this reason, the ideal gate voltage waveform shown in FIG. 3B can be simulated more appropriately.

  According to this embodiment described above, in addition to the effect according to the above (1) of the first embodiment, the following effect can be obtained.

  (5) A resistor 42 is provided in the charging path between the gate of the power switching element Sw and the driving IC 32. Thereby, the vibration of the gate voltage Vge caused by the resonance phenomenon using the inductor component (the coil 34 and the parasitic inductor 38) in the charging path can be appropriately attenuated.

(Fourth embodiment)
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the third embodiment.

  FIG. 10 shows a circuit configuration of the drive circuit DC according to the present embodiment. In FIG. 10, members corresponding to those shown in FIG. 8 are given the same reference numerals for convenience.

  As illustrated, in the present embodiment, a coil 38 a is provided in series with a parallel circuit of the coil 34 and the capacitor 36 in the charging path between the gate of the power switching element Sw and the drive IC 32.

  FIG. 11A shows the waveform of the gate voltage Vge according to the present embodiment, and FIG. 11B shows the transition of the current of the free wheel diode FD according to the present embodiment. Specifically, in the figure, the solid line shows the case where the drive circuit DC according to the present embodiment is used, and the one-dot chain line shows the case where the inductance Lg2 of the coil 38a is “0H”, and the broken line The conventional example in which the gate of the power switching element Sw and the driving IC 32 are connected by a resistor having a resistance value “2Ω” is shown.

  As shown in the figure, the maximum value of the recovery current can be reduced when the parallel circuit of the coil 34 and the capacitor 36 and the resistor 42 are used as compared with the case where the gate resistance is used (broken line). . Furthermore, as in the present embodiment, by further including the coil 38a, the voltage Vce between the input and output terminals of the power switching element Sw during the period in which the recovery current gradually decreases can be increased, and thus the surge can be appropriately reduced. be able to.

  According to this embodiment described above, the effect according to the effect (1) of the previous first embodiment, the effect (2) to (4) of the previous second embodiment, The effect (5) of the third embodiment can be obtained.

(Fifth embodiment)
Hereinafter, the fifth embodiment will be described with reference to the drawings with a focus on differences from the third embodiment.

  FIG. 12 shows a circuit configuration of the drive circuit DC according to the present embodiment. In FIG. 12, members corresponding to those shown in FIG. 8 are given the same reference numerals for the sake of convenience.

  As shown in the figure, the present embodiment includes a protective diode 44 whose forward direction is the direction from the gate side of the power switching element Sw to the output terminal side of the drive IC 32. The protective diode 44 is for avoiding an excessive increase in the voltage Vce between the gate and the emitter of the power switching element Sw. That is, when the voltage between the gate and the emitter of the power switching element Sw is higher than the output voltage of the driving IC 32 (more precisely, when the voltage drop amount of the protective diode 44 is higher than the output voltage), the protection A forward current flows through the diode 44. As a result, the gate voltage Vge is output under the situation where the gate voltage Vge can be higher than the output voltage of the drive IC 32 due to the resonance phenomenon using the capacitor component and the inductor component between the gate of the power switching element Sw and the drive IC 32. Exceeding the voltage excessively can be suitably avoided. For this reason, dielectric breakdown between the gate and the emitter of the power switching element Sw caused by the resonance phenomenon can be suitably avoided.

  According to this embodiment described above, in addition to the effect according to the effect (1) of the first embodiment and the effect (5) of the third embodiment, the following effect is further obtained. Can be obtained.

  (6) A protective diode 44 is provided between the gate of the power switching element Sw and the driving IC 32. Thereby, it is possible to suitably avoid the voltage between the gate and the emitter of the power switching element Sw exceeding the breakdown voltage.

(Sixth embodiment)
Hereinafter, the sixth embodiment will be described with reference to the drawings with a focus on differences from the fifth embodiment.

  FIG. 13 shows a circuit configuration of the drive circuit DC according to the present embodiment. In FIG. 13, members corresponding to those shown in FIG. 12 are given the same reference numerals for convenience.

  As illustrated, in the present embodiment, a coil 38 a is provided in series with a parallel circuit of the coil 34 and the capacitor 36 in the charging path between the gate of the power switching element Sw and the drive IC 32.

  FIG. 14 shows an actual measurement result of the surge voltage according to the present embodiment using an actual machine. Specifically, FIG. 14A shows the voltage Vce between the collector and the emitter of the power switching element Sw and the voltage Vd across the freewheel diode FD, and FIG. 14B shows the current id and the freewheel diode FD. The transition of the change rate did / dt is shown.

  As shown in the drawing, in the present embodiment, the voltage Vce between the collector and the emitter is kept high during the period when the recovery current (id <0) decreases from the peak value. For this reason, a surge can be reduced.

  FIG. 15 shows the actual measurement results of the present embodiment and the conventional example using surge voltage actual devices. Specifically, FIGS. 15A1 and 15A2 show the transition of the current id of the freewheel diode FD and its change rate did / dt, and FIGS. 15B1 and 15B2 show the freewheel. The transition of the voltage Vd applied to the diode FD is shown. However, FIGS. 15A1 and 15B1 show the case where the drive circuit DC according to the present embodiment is used, and FIGS. 15A2 and 15B2 show the gate of the power switching element Sw and the drive IC. This is a case of a conventional example in which a gate resistor is connected between them. As illustrated, according to the present embodiment, the peak value of the recovery current change rate did / dt of the freewheeling diode FD is decreased. For this reason, the voltage Vd applied to both ends of the freewheel diode FD can be reduced. Furthermore, since the peak value of the recovery current of the free wheel diode FD is reduced, the loss between the power switching element and the free wheel diode can be reduced.

  According to the present embodiment described above, the effect according to the effect (1) of the previous first embodiment, the effect (2) to (4) of the previous second embodiment, The effect (5) of the third embodiment and the effect (6) of the previous fifth embodiment can be obtained.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  In the second, fourth, and sixth embodiments, the coil 38 a may be connected between the parallel connection body of the coil 34 and the capacitor 36 and the drive IC 32.

  In the third to sixth embodiments, the resistor 42 may be connected between the parallel connection body of the coil 34 and the capacitor 36 and the gate of the switching element Sw. Furthermore, in the fourth and sixth embodiments, the resistor 42 may be connected between the gate of the switching element Sw and the coil 38a.

  The circuit for charging the conduction control terminal (gate) of the switching element Sw is not limited to those exemplified in the above embodiments. For example, as shown in FIG. 16, a diode 44 may be connected in parallel with the coil 34 and the capacitor 36. In FIG. 16, the coil 38 a is not provided, but the coil 38 a may be provided between the coil 34 and the drive IC 32, or between the gate of the switching element Sw and the coil 34. In FIG. 16, the resistor 42 is not provided. However, the resistor 42 may be provided between the coil 34 and the drive IC 32, or between the gate of the switching element Sw and the coil 34.

  In addition, as shown in FIG. 17, in a configuration that does not include the resistor 42, a diode 44 having a forward direction from the gate side of the switching element Sw to the drive IC 32 side may be provided. In FIG. 17, the coil 38 a is provided. However, the parasitic inductor 38 may be used without providing the coil 38 a.

  Further, as shown in FIG. 18, a resistor 42 a may be connected in series to the coil 34, and a resistor 42 b may be connected in series to the capacitor 36, and these may be connected in parallel. In this case, the degree of freedom for adjusting the behavior of the gate voltage can be improved by making the resistance values of the resistors 42a and 42b different.

  Instead of using the parasitic capacitor 40 as a capacitor component between the conduction control terminal (gate) and the output terminal (emitter) of the switching element Sw, a capacitor may be connected. According to this, the capacitance between the conduction control terminal and the output terminal of the switching element Sw can be set with higher accuracy.

  The switching element Sw is not limited to the IGBT but may be a MOSFET, for example.

  In the above embodiment, the present invention is applied to the drive circuit of the series connection body of the pair of switching elements Swp and Swn on the high potential side and the low potential side constituting the inverter IV, but is not limited thereto. For example, the present invention may be applied to the drive circuit for the pair of switching elements with respect to a booster circuit that connects a connection point of a series connection body of the pair of switching elements to a DC power source via a coil. Here, if such a power conversion circuit is an in-vehicle system, it is a power device that handles a large amount of power, and therefore the influence of surge voltage or the like is likely to be serious, so that the application of the present invention is particularly effective.

The figure which shows the system configuration | structure concerning 1st Embodiment. The time chart for demonstrating the production | generation of the surge resulting from a recovery current. The time chart which shows the gate voltage waveform ideal in this embodiment. The figure for demonstrating the effect of the said gate voltage waveform. The circuit diagram which shows the circuit structure of the drive circuit concerning the said embodiment. The time chart which shows the simulation result of the gate voltage waveform in the embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning 2nd Embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning 3rd Embodiment. The time chart which shows the simulation result of the gate voltage waveform in the embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning 4th Embodiment. The time chart which shows the effect of the embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning 5th Embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning 6th Embodiment. The time chart which shows the effect of the embodiment. The time chart which shows the effect of the embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning the modification of each said embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning the modification of each said embodiment. The circuit diagram which shows the circuit structure of the drive circuit concerning the modification of each said embodiment.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 10 ... Motor generator (one embodiment of inductive load), 12 ... High voltage battery, 34 ... Coil, 36 ... Capacitor, 40 ... Parasitic capacitor, 32 ... Driving IC (one embodiment of voltage application means), IV ... Inverter, DC ... Drive circuit.

Claims (7)

A series connection body of a pair of switching elements on a high potential side and a low potential side, and a diode connected in antiparallel to each of the pair of switching elements, and an inductive load at a connection point of the pair of switching elements In the drive circuit of the power conversion circuit comprising voltage application means for applying a voltage to the conduction control terminal so as to turn on the switching element,
A drive circuit for a power conversion circuit, comprising a parallel circuit of an inductor and a capacitor between the voltage application means and the conduction control terminal.   2. The drive circuit for a power conversion circuit according to claim 1, further comprising an inductor connected in series to the parallel circuit between the voltage application unit and the conduction control terminal.   Inductance of the inductor connected in series to the parallel circuit is such that when a forward current flows through a diode connected in reverse parallel to one of the pair of switching elements, the other switching element is turned on. It is set so that the voltage of the conduction control terminal of the other switching element can be changed to the side for turning off the switching element in a period in which the recovery current of the diode in which the forward current has flowed increases and then decreases. The drive circuit for the power conversion circuit according to claim 2, wherein:   The inductance of the inductor connected in series to the parallel circuit may change the voltage of the conduction control terminal to the side for turning off the switching element during the period when the current flowing through the switching element is equal to or less than a specified value. 4. The drive circuit for a power conversion circuit according to claim 3, wherein the drive circuit is set so as to be able to do so.   The drive circuit of the power conversion circuit according to any one of claims 1 to 4, further comprising a resistor between the voltage application unit and the conduction control terminal.   The protective diode having a forward direction from the conduction control terminal side of the switching element to the voltage application means side is further provided between the voltage application unit and the conduction control terminal. 6. The drive circuit for the power conversion circuit according to any one of 5 above. A drive circuit for the power conversion circuit according to any one of claims 1 to 6,
A power conversion system comprising the power conversion circuit.

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