JP2007221915A - Dc-dc converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a DC-DC converter which has high conversion efficiency even at a small output and is capable of reducing the number of parts. <P>SOLUTION: The DC-DC converter includes a voltage resonance circuit into which electric power from a low-voltage DC power supply is inputted for DC-AC conversion by zero-voltage switching. The AC-converted electric power is supplied to an insulated high-frequency transformer current-resonated by a current resonance circuit positioned on the secondary side of the transformer, and converted into a double voltage by a double voltage rectification circuit and a switching circuit. The converted double voltage is smoothed by a smoothing circuit and outputted. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

    The present invention relates to a DC-DC converter, and more particularly to an insulated DC-DC converter for a distributed power source that converts power from a distributed DC power source into medium-power capacity power of 1 kW or less, and the DC-DC converter. It relates to the connected inverter used.

    A distributed power source that converts power from a distributed DC power source, for example, a household fuel cell, a solar power generation system, or a wind power generation system into a medium power capacity (0.3 kW to 10 kW) power, includes a power conversion device such as an inverter. In this power converter, insulation between the input (primary side) and the system (secondary side) is desired. Even if a high-frequency insulation type converter is used in such a power conversion device, there is a problem that efficiency is deteriorated as compared with a non-insulation type converter.

Regarding DC-DC converters, Patent Document 1, Patent Document 2, and Patent Document 3 are known. In Patent Document 1, Patent Document 2 and Patent Document 3, a switching circuit is used to switch an input signal and input to the primary side of the transformer, and the output from the secondary side of the transformer is rectified to be DC. A power supply circuit that outputs a voltage is disclosed.
2001-128452 JP-A-9-163734 Japanese Patent Application No. 11-306724

  In addition, since a power source such as a fuel cell inevitably increases the frequency of operation at an output less than the rated value, not only the efficiency is improved at the rated output as described above, but also the small power of 50% or less of the rated output is small. Improving efficiency during output operation is an important issue.

  In response to such a request, the power supply circuits disclosed in Patent Document 1, Patent Document 2 and Patent Document 3 are not efficient because the input DC voltage is simply switched and output through the transformer. It has been pointed out that there is a problem that the output is not stable.

  From such a background, the inventors have already proposed a DC-DC converter capable of improving the efficiency at the time of small output operation in Japanese Patent Application No. 2005-077788. Although it has been found that the DC-DC converter according to this proposal can improve the conversion efficiency satisfactorily, further reduction of the number of parts constituting the circuit and improvement of the efficiency are desired.

  The present invention has been made to solve the above problems, and an object of the present invention is to provide a DC-DC converter having high conversion efficiency even at a small output and having a reduced number of parts. .

According to this invention,
A first voltage resonance circuit that receives DC power from a low-voltage DC power supply whose output voltage fluctuates, performs DC-AC conversion, and outputs;
A first insulated high-frequency transformer having a primary side and a secondary side, into which an output voltage from the first voltage resonance circuit is input;
A first current resonance circuit comprising a choke coil connected to the first terminal on the secondary side of the first transformer and first and second capacitors connected in series to the choke coil;
A first diode connection in which the first and second diodes are connected in series, a third capacitor connected in parallel to the first diode connection, and a second diode in which the second and third diodes are connected in series And a fourth capacitor connected in parallel to the second diode connection, and a connection point between the first and second diode connections and a connection point between the third and fourth capacitors are the first and second capacitors. Connected to a second terminal on the secondary side of one transformer, the third capacitor is connected to a connection point between the first and second diodes, and the fourth capacitor is connected between the third and fourth diodes. A voltage doubler connected to the connection point of
A switching circuit for switching one voltage of the third and fourth capacitors or a voltage in series connection of the third and fourth capacitors;
A smoothing circuit for smoothing and outputting the output from the switching circuit;
A DC-DC converter is provided.

  According to the DC-DC converter of the present invention, the number of components is reduced, the resonance point is prevented from being shifted based on the difference in characteristics of circuit components, and as a result, high-efficiency conversion without switching loss is realized. be able to.

  Hereinafter, a DC-DC converter according to an embodiment of the present invention will be described with reference to the drawings as necessary.

  A converter unit (DC-DC converter) according to an embodiment of the present invention is applied to an interconnected inverter to constitute a distributed power supply system. In a distributed power system, a DC power supply with fluctuation in output, for example, a fuel cell, a solar battery, or an output (DC power) from wind power generation is input to a connected inverter as a power conditioner, DC-DC conversion is performed in the converter unit, and the converted DC output is converted into an AC output and a relatively small output (for example, about 0.3 kW to several tens kW) by the inverter unit that performs DC-AC conversion, and is applied to the load. For example, it is output as a commercial voltage (system voltage) to a load in the home. Here, the commercial voltage (system voltage) corresponds to 101V or 202V (in the case of single-phase three-wire connection) in Japan, and corresponds to 115V or 230V in the United States.

  In the fuel cell system, an input voltage of 80 V or less is input to the converter unit, and currently, a voltage of 20 V to 60 V is input. The output voltage Vout is the highest when there is no load, and the voltage is 25% as the load increases. It has a characteristic of decreasing by about 30%. Moreover, in a solar power generation system provided with a solar cell module, the voltage of 17-21V is output by one solar cell module, and 170V-350V is output as a system. The output voltage Vout varies in the range of 120V to 450V. Furthermore, in the wind power generation system, an output voltage Vout of about 50V is generated, but when the blades are rotating, the output varies within a range of 30V to 50V.

  FIG. 1 shows a circuit configuration of a converter unit according to an embodiment of the present invention that can be applied to the above-described interconnected inverter.

  The converter unit shown in FIG. 1 is a high-frequency insulation type DC-DC converter, and includes a voltage resonance circuit 11 to which a voltage from a direct current power supply 3 that varies in output is input. A high frequency voltage is output from the voltage resonance circuit 11, and this high frequency voltage is input to the primary side of the high frequency transformer 12. A current resonance circuit 13 is disposed on the secondary side of the high-frequency transformer 12, and the current supplied from the high-frequency transformer 12 is current-resonated by the current resonance circuit 13. The output voltage of the current resonance circuit 13 is supplied to the voltage doubler rectifier circuit 14, and the output voltage from the voltage doubler rectifier circuit 14 is switched by the switching circuit 15, so that the output voltage of the current resonance circuit 13 is converted to a voltage doubler. And output through the smoothing circuit 16. Therefore, a voltage whose output is varied is input to the converter unit shown in FIG. 1, and a smoothed output voltage is output from the converter unit.

  The switching circuit 15 includes a pulse width modulation control circuit 18 (PWM control circuit: Pulse Width Modulation control circuit) that controls the switching circuit 15 according to the output voltage Vout from the smoothing circuit 16, and is stable according to the output voltage Vout. And applied to the smoothing circuit 15. Therefore, a stable output voltage is output from the smoothing circuit 16.

  Since the DC-DC converter shown in FIG. 1 is applied to the power source 3 having a relatively low voltage, the voltage resonance circuit 11 is arranged on the primary side of the high-frequency transformer 12 and the high-frequency transformer 12 that outputs a high voltage is used. A current resonance circuit 13 is arranged on the next side. The voltage resonance circuit 11 disposed on the primary side includes a switching element such as an FET (field effect transistor) or an IGBT (insulated gate / bipolar transistor), and between the source and drain of the switching element (in the case of IGBT, an emitter-collector). A capacitor is connected between the voltage resonance circuit 11 and the voltage resonance circuit 11 is configured to perform voltage resonance. The current resonance circuit 13 arranged on the secondary side is configured to resonate by series resonance. Therefore, a highly efficient DC-DC converter can be realized.

The voltage resonance circuit 11 shown in FIG.
(1) Full circuit (2) Half bridge (3) Push-pull Three circuit configurations can be adopted. Specific circuit examples of these voltage resonance circuits are shown in FIGS.

  The current resonance circuit 13 shown in FIG. 1 can employ the circuit configuration of the voltage doubler rectifier circuit 14. A circuit example of the voltage resonance circuit 11 will be described with reference to FIGS. 2 to 4, an ordinary electrolytic capacitor is used as the storage capacitor C1, but the description is omitted because it is common in each circuit. A case where an FET is used as the switching element will be described.

  FIG. 2 shows a first circuit example in which the voltage resonance circuit 11 is configured by a full bridge circuit. In the voltage resonance circuit shown in FIG. 2, a storage capacitor C1 is connected between the positive and negative sides of a DC power supply, a switching element Q1 and a switching element Q2 are connected in series, and a switching element Q3 and a switching element Q4 are connected in series. It is connected. Capacitors C2 to C5 are connected in parallel between the sources and drains of the switching elements Q1 to Q4 to the switching elements Q1 to Q4, respectively. The series circuit of the switching elements Q1 and Q2 and the series circuit of the switching elements Q3 and Q4 are respectively connected in parallel to the DC power supply on the input side so as to form a full bridge circuit. Therefore, the full bridge circuit and the capacitor C1 are connected in parallel to the DC power source. That is, the drains of the switching elements Q1 and Q3 are connected to the positive side of the power supply, and the sources of the switching elements Q2 and Q4 are connected to the negative side of the power supply. Further, a connection part between the switching element Q1 and the switching element Q2 is connected to one end part of the transformer T1 on the output side, and a connection part between the switching element Q3 and the switching element Q4 is connected to the other end part of the transformer T1.

  In the full bridge circuit shown in FIG. 2, a switching control unit 17 is provided to turn on and off the switching elements Q1 to Q4 at a predetermined timing. The switching control unit 17 includes drivers DR1, DR2, and MCU (micro control unit) 18. In the switching control unit 17, a gate pulse having a frequency suitable for the current resonance circuit is given to the switching elements Q1 to Q4.

  FIG. 3 shows a second circuit example in which the voltage resonance circuit 11 is configured by a half-bridge circuit. In FIG. 3, the same circuit components and the same parts as those in FIG.

  In the voltage resonance circuit shown in FIG. 3, a switching element Q1 and a switching element Q2 are connected in series, and capacitors C2 and C3 are connected in parallel between the source and drain of the switching element, respectively. Yes. In addition, in the series circuit of the switching elements Q1 and Q2, capacitors C6 and C7 connected in series are connected in parallel to form a half bridge circuit.

  A connection portion between the switching elements Q1 and Q2 is connected to one end portion of the transformer T1, and a connection portion between the capacitors C6 and C7 is connected to the other end portion of the transformer T1.

  In the half bridge circuit shown in FIG. 3, a driver DR1 is provided to turn on and off the switching elements Q1 and Q2 at a predetermined timing. Similarly, a gate pulse having a frequency suitable for the current resonance circuit is applied to the switching elements Q1 and Q2.

  FIG. 4 shows a third circuit example in which the voltage resonance circuit 11 is configured as a push-pull type. FIG. 4 shows a push-pull type voltage resonance circuit. 4, the same circuit parts and the same parts as those in FIG.

  In FIG. 4, the drain of the switching element Q1 is connected to one end of the transformer T1, the drain of the switching element Q2 is connected to the other end of the transformer T1, and the sources of the switching elements Q1 and Q2 are connected to the negative side of the DC power supply. It is connected. Further, the positive side of the DC power supply is connected to an intermediate portion between one end and the other end of the transformer T1.

  The push-pull type voltage resonance circuit 11 shown in FIG. 4 is provided with a driver DR1 for turning on and off the switching elements Q1 and Q2 at a predetermined timing. Similarly, the switching elements Q1, Q2 are given a gate pulse having a frequency suitable for the current resonance circuit.

  The current resonance circuit 13, the voltage doubler rectifier circuit 14, the switching circuit 15 and the smoothing circuit 16 shown in FIG. 1 are configured as shown in FIG. 5 as an example. In the circuit shown in FIG. 5, the choke coil CH1 is connected to the secondary high-voltage terminal of the high-frequency transformer 12, and the secondary high-voltage terminal is bifurcated to be connected to capacitors C8 and C9, respectively. Thus, the current resonance circuit 13 is configured. That is, the secondary high-voltage terminal of the high-frequency transformer 12 includes a first series resonance circuit composed of the choke coil CH1 and the capacitor C8 and a second series resonance circuit composed of the choke coil CH1 and the capacitor C9. It is connected.

  The first and second series resonant circuits are respectively connected to a connection point between the diodes D1 and D2 and a connection point between the diodes D3 and D4 constituting the voltage doubler rectifier circuit 14. In the voltage doubler rectifier circuit 14, a series circuit of diodes D3 and D4 is connected to a series circuit of diodes D1 and D2, and a secondary low voltage terminal of the high-frequency transformer 12 is connected between the series circuits. In addition, a capacitor C10 is connected in parallel to the series circuit of the diodes D1 and D2, and a capacitor C12 is connected in parallel to the series circuit of the diodes D3 and D4, so that a voltage doubler rectifier circuit 14 is configured.

  A series circuit of the flywheel diode D5 and the switching transistor Q7 is connected in parallel to the series circuit of the diodes D1 and D2, and the switching circuit 15 is configured. In the switching circuit 15, when the switching transistor Q7 is turned off, the flywheel diode D5 is connected to the secondary low voltage terminal of the high frequency transformer 12, and a low potential is applied to the choke coil CH2 from the cathode of the flywheel diode D5. It is done. On the other hand, when the switching transistor Q7 is turned on, the secondary high voltage potential of the high frequency transformer 12 is applied to the capacitor C10 via the capacitor C8 and the diode D1. Therefore, a potential difference of the double voltage is generated in the series circuit of the capacitors C10 and C12, and a potential corresponding to the double voltage is output from the cathode of the flywheel diode D5.

  A choke coil CH2 is connected to the cathode of the flywheel diode D5, and a series circuit of the choke coil CH2 and the capacitor C14 is connected in parallel to the series circuit of the capacitors C10 and C12 via the switching transistor Q7. Both sides of the capacitor C14 are connected to an output terminal to constitute a smoothing circuit 16. In other words, in the circuit shown in FIG. 5, there is no potential difference and a double voltage is alternately generated in the series circuit of the capacitors C10 and C12 according to the ON and OFF cycles of the switching transistor Q7, and an AC voltage having a double voltage amplitude. Is applied to the smoothing circuit 16 and the smoothed voltage doubled is output from the smoothing circuit 16.

  The switching circuit 15 including a series circuit of the flywheel diode D5 and the switching transistor Q7 shown in FIG. 5 may be connected in parallel to the capacitor C12 as shown in FIG. That is, in the circuit shown in FIG. 6, when the switching transistor Q7 is turned off, the flywheel diode D5 is connected to the secondary low voltage terminal of the high frequency transformer 12, and a low potential is applied to the anode of the flywheel diode D5. The anode side of the diode D1 is also maintained at a low potential. On the other hand, when the switching transistor Q7 is turned on, the potential at the connection point between the capacitors C10 and C12 is raised to the high-voltage side potential on the secondary side of the high-frequency transformer 12 via the capacitor C9 and the diode D4. . Further, since the secondary high-voltage potential of the high-frequency transformer 12 is further applied to the capacitor C10 via the capacitor C8 and the diode D1, a potential difference of double voltage is generated in the series circuit of the capacitors C10 and C12. Is applied to the choke coil CH2.

  In the DC-DC converter, a PWM control signal is given to the switching transistor Q7 shown in FIG.

  The operation of the DC-DC converter in the rated output mode in which the DC power supply 3 generates an output voltage (target voltage Vout) at a rated value will be described with reference to FIGS. 2 and 8A to 8H. In the no-load mode and the small output mode, the operation is the same as the operation in the rated mode with only the voltage and current waveform of each part being different, so that the description thereof is omitted.

  When the DC-DC converter is connected to the DC power source 3 via a switch (not shown), charging of the capacitor C1 is started. Similarly, charging of the series circuit of capacitors C2 and C3 and the series circuit of capacitors C4 and C5 connected in parallel to the capacitor C1 is also started.

  At a certain time t1, a control pulse signal is given to the driver circuits DR1 and DR2, and the driver circuits DR1 and DR2 are operated. At this time t1, the first and fourth gate signals shown in FIG. 8E are switched from the high level to the low level in synchronization with the control pulse signal. Therefore, as shown in FIG. 8A, the FETs Q1 and Q4 to which the first and fourth gate pulses have been applied are kept off.

  After the time t1, the voltage between the source and drain of the FETs Q2 and Q3 starts to decrease as shown in FIG. 8B due to the exciting current of the transformer, and as shown in FIG. The voltage across the drain begins to rise. Further, as shown in FIG. 8C, the primary side voltage of the high-frequency transformer T1 also starts to drop.

  When reaching a time point t2 after a predetermined time Δt has elapsed from the time point t1, the gates of the FETs Q2 and Q3 are supplied with the second and third gate signals shown in FIG. As shown in B), the source-drain voltage of the FETs Q2 and Q3 is reduced to zero, and the FETs Q2 and Q3 are maintained in the on state. Further, the source-drain voltages of the FETs Q1 and Q4 that are kept off reach the input voltage as shown in FIG. Therefore, as shown in FIG. 8C, the primary side voltage of the high-frequency transformer T1 also reaches a predetermined voltage, current is supplied to the FETs Q2 and Q3, and the drain current is increased as shown in FIG. 8F. The This current is supplied as an exciting current to the primary side of the high-frequency transformer T1, and as a result, an induced voltage is generated on the secondary side.

  Since the impedance of the current resonance circuit connected to the secondary side of the high frequency transformer T1 is high immediately after the FETs Q2 and Q3 are turned on, the drain currents of the FETs Q2 and Q3 are gradually increased from zero. Further, at time t2 to time t3, a drain current having a half-wave sine wave is generated according to the resonance frequency of the current resonance circuit connected to the secondary side of the high-frequency transformer T1.

  At time t3, when the second and third gate signals applied to the FETs Q2 and Q3 are turned off, the FETs Q2 and Q3 are turned off and the drain current becomes zero as shown in FIG. Accordingly, the supply of energy to the secondary side of the high-frequency transformer T1 is stopped. Further, the source-drain voltage of the FETs Q2, Q3 turned off as shown in FIG. 8B is gradually increased, and the source-drain voltage of the FETs Q2, Q3 turned off as shown in FIG. 8A. Is gradually raised. As the voltage between the source and drain of the FETs Q2 and Q3 increases, the voltage between the source and drain of the FETs Q1 and Q4 decreases. Accordingly, the primary voltage of the high-frequency transformer T1 is also gradually reduced.

  When reaching a time point t4 after a predetermined time Δt has elapsed from the time point t3, the gates of the FETs Q1 and Q4 are given the first and fourth gate signals shown in FIG. Conduction is performed as shown in A), and the source-drain voltages of the FETs Q1 and Q4 are reduced to zero. From time t4 to time t5, the FETs Q1 and Q4 are maintained in the on state. Further, the source-drain voltages of the FETs Q2 and Q3 that are kept off reach the input voltage as shown in FIG. 8B. Therefore, as shown in FIG. 8C, the primary voltage of the high-frequency transformer T1 also reaches a predetermined voltage on the negative side, and current is supplied from the capacitors C1, C2, and C3 to the FETs Q2 and Q3 that are conducted, and the drain current thereof. Is increased as shown in FIG. This current is supplied as an exciting current to the primary side of the high-frequency transformer T1, and as a result, an induced voltage is generated on the secondary side.

  Here, from time t3 to time t4, the capacitors C2 and C5 connected in parallel to the FETs Q1 and Q4 are slowly discharged, and therefore the source-drain voltages of the FETs Q1 and Q4 are also gradually lowered. Thereafter, the FETs Q1 and Q4 are turned on at time t4, but the change in the voltage between the source and drain of the FETs Q1 and Q4 at the moment of switching is extremely small, and a substantial zero voltage resonance switching (ZVS) is realized.

  From time t5, the same operation as at times t1 to t4 is repeated again to generate an induced voltage on the secondary side of the high-frequency transformer T1. Here, time points t5, t6, t7, and t8 correspond to time points t1, t2, t3, and t4, respectively, and refer to the description of the corresponding time points.

  Here, from time t5 to t6, the capacitors C2 and C5 connected in parallel to the FETs Q1 and Q4 are similarly slowly charged, and accordingly, the source-drain voltages of the FETs Q1 and Q4 are also gradually increased. . Thereafter, the FETs Q2 and Q3 are turned on at the time point t6, but the change in the voltage between the source and drain of the FETs Q2 and Q3 at the moment of switching is extremely small, and a substantial zero voltage resonance switching (ZVS) is realized.

  By operating the voltage resonance circuit as described above, the voltage waveform and the current waveform as shown in FIGS. 9A and 9B are output to the secondary side of the high-frequency transformer T1. That is, a trapezoidal wave voltage appears on the secondary side of the high-frequency transformer T1, as shown in FIG. 9A, corresponding to the voltage waveform on the primary side of the high-frequency transformer T1 shown in FIG. Corresponding to the current waveform on the primary side of the high-frequency transformer T1 shown in FIG. 8H, a sinusoidal current appears on the secondary side of the high-frequency transformer T1 as shown in FIG. 9B.

  The operation of the half-bridge voltage resonance circuit 11 shown in FIG. 3 and the push-pull type voltage resonance circuit 11 shown in FIG. 4 is similarly operated by zero voltage resonance switching (ZVS), and the full-bridge voltage resonance circuit 11 of FIG. Since those skilled in the art can easily understand the description, the description is omitted.

  Next, the operation of the circuit shown in FIG. 5 will be described with reference to FIGS. 5 and 10A to 10K.

  The circuit shown in FIG. 6 is also operated in substantially the same manner as the circuit shown in FIG.

  When the voltage VT1 and the current iT1 as shown in FIGS. 10A and 10B are generated on the secondary side of the transformer 12, the current iT1 flows into the choke coil CH1 as the current iL1 and passes through the choke coil CH1. To the capacitors C8 and C9. Therefore, AC voltages vC8 and vC9 as shown in FIGS. 10C and 10E are generated at both ends of the capacitors C8 and C9, respectively, and from the capacitors C8 and C9, respectively, FIG. 10D and FIG. AC currents iC8 and iC9 as shown in FIG. 10 (F) are supplied to the connection points of the diodes D1 and D2 and the connection points of the diodes D3 and D4. Capacitor C10 is charged with current iC8 via diode D1, and capacitor C12 is charged with current iC9 via diode D3. Therefore, the potential of the diode D1 is maintained at the potential Va as shown in FIG. Here, the potential Va corresponds to a substantially quadruple voltage obtained by adding the voltage of the capacitor C10 to the potential between the capacitors C10 and C12.

  As shown in FIG. 10 (H), when the switching signal SQ1 is given as the PWM control signal to the switching element Q7, the potential Vb is generated at the anode of the flywheel diode D5 in accordance with the switching element Q7 being turned on and off. When the switching element Q7 is off, the potential Va is not applied to the cathode of the flywheel diode D5 via the switching element SQ7, and corresponds to the potential Vb2 between the capacitors C10 and C12 as shown in FIG. 10 (I). A potential Vb is generated at the anode of the flywheel diode D5. On the other hand, when switching element Q7 is on, potential Va is applied to the cathode of flywheel diode D5 via switching element SQ7, and potential Vb1 corresponding to potential Va as shown in FIG. 10 (I). Is applied to the cathode of the flywheel diode D5. Therefore, the potential Vb on the cathode side of the flywheel diode D5 varies as a rectangular wave between the potential Vb1 and the potential Vb2 with reference to the potential VOUT in accordance with the on / off of the switching element Q7. The rectangular wave potential Vb shown in FIG. 10 (I) is applied to the smoothing circuit 16 including the choke coil CH2 and the capacitor C14. Therefore, as shown in FIG. 10J, in this smoothing circuit, the capacitor C14 is charged with the smoothing current iC0, and the output voltage VOUT shown in FIG. 10K is output as the voltage across the capacitor C14.

  When the capacitors C8 and C9 form a resonance circuit with the choke coil CH1, the capacitors C8 and C9 simultaneously function as a charge pump to realize voltage doubler rectification. At this time, the voltage Va can be expressed as follows.

Va = 2 × vT10 + 2 × vT12
vT10 is a positive terminal voltage of the high-frequency transformer 12, and vT12 is a negative terminal voltage of the high-frequency transformer 12.

  In PWM control, if the duty for turning on the switching element Q7 is d, the output voltage VOUT can be expressed by the following equation.

VOUT = 2 × vT10 × d + 2 × vT12
In the converter circuit described above, the smoothed output voltage VOUT is output from the smoothing circuit 16 as shown in FIG. 10 (K) in accordance with the pulse width of the PWM signal. Here, if the pulse width of the PWM signal is large, the output voltage VOUT from the smoothing circuit 16 increases. If the pulse width of the PWM signal is small, the output voltage VOUT from the smoothing circuit 16 decreases. Therefore, the output voltage from the smoothing circuit 16 is detected by the PWM signal generator 18, and the output of the smoothing circuit 16 can be made constant by selecting an appropriate pulse width.

  In the circuit shown in FIG. 5, only a single circuit is provided on the secondary side of the transformer instead of two circuits. Normally, when two secondary windings are provided on the secondary side in one transformer, the resonance frequency shifts due to a difference in leakage inductance. Two transformers are required to minimize the difference in leakage inductance. However, since the secondary side of the transformer is composed of a single circuit, it is possible to provide a DC-DC converter having high conversion efficiency even at a small output.

  As described above, it is possible to provide a DC-DC converter having high conversion efficiency not only at the rated output but also at the small output.

  The present invention is not limited to the above-described embodiments, and various modifications can be made without departing from the spirit of the invention at the stage of implementation. Further, the above embodiments include inventions at various stages, and various inventions can be extracted by appropriately combining a plurality of disclosed constituent elements.

  In addition, for example, even if some structural requirements are deleted from all the structural requirements shown in each embodiment, the problem described in the column of the problem to be solved by the invention can be solved, and the effect described in the effect of the invention Can be obtained as an invention.

It is a block diagram which shows the circuit structure of the DC-DC converter part concerning one Embodiment of this invention. FIG. 2 is a circuit diagram illustrating an example of a voltage resonance circuit illustrated in FIG. 1. FIG. 3 is a circuit diagram showing another example of the voltage resonance circuit shown in FIG. 1. FIG. 6 is a circuit diagram showing still another example of the voltage resonance circuit shown in FIG. 1. It is a circuit diagram which shows the circuit example of the current resonance circuit, voltage doubler circuit, switching circuit, and smoothing circuit which are shown in FIG. FIG. 3 is a circuit diagram showing another circuit example of the current resonance circuit, voltage doubler circuit, switching circuit, and smoothing circuit shown in FIG. 1. It is a control part block diagram which shows the function of MCU of the switching control circuit shown in FIG. (A)-(H) is a wave form diagram which shows the waveform of each part in the voltage resonance circuit shown in FIG. (A) And (B) is a wave form diagram which shows a voltage and an electric current output from the high frequency transformer shown in FIG. (A)-(K) is a wave form diagram which shows the waveform of each part in the circuit shown in FIG.

Explanation of symbols

  C1 to C14 ... capacitors, Q1 to Q7 ... switching elements, DR1, DR2 ... drivers, D1 to D5 ... diodes, CH1, CH2 ... choke coils, 11 ... voltage resonance circuit, 12 ... high frequency transformer, 13 ... current resonance circuit, 14 Rectifier circuit, 15 switching circuit, 16 smoothing circuit, 18 PWM control circuit, 17 switching control circuit

Claims (4)

A first voltage resonance circuit that receives DC power from a low-voltage DC power supply whose output voltage fluctuates, performs DC-AC conversion, and outputs;
A first insulated high-frequency transformer having a primary side and a secondary side, into which an output voltage from the first voltage resonance circuit is input;
A first current resonance circuit comprising a choke coil connected to the first terminal on the secondary side of the first transformer and first and second capacitors connected in series to the choke coil;
A first diode connection in which the first and second diodes are connected in series, a third capacitor connected in parallel to the first diode connection, and a second diode in which the second and third diodes are connected in series And a fourth capacitor connected in parallel to the second diode connection, and a connection point between the first and second diode connections and a connection point between the third and fourth capacitors are the first and second capacitors. Connected to a second terminal on the secondary side of one transformer, the third capacitor is connected to a connection point between the first and second diodes, and the fourth capacitor is connected between the third and fourth diodes. A voltage doubler connected to the connection point of
A switching circuit for switching one voltage of the third and fourth capacitors or a voltage in series connection of the third and fourth capacitors;
A smoothing circuit for smoothing and outputting the output from the switching circuit;
A DC-DC converter comprising:     2. The DC-DC converter according to claim 1, wherein each of the first and second voltage resonance circuits includes a switching element, and performs DC-AC conversion by zero voltage switching to output a high-frequency voltage.     The DC-DC converter according to claim 2, wherein the first and second voltage resonance circuits are either a bridge type or a push-pull type.     The switching circuit includes a series circuit of a diode and a switching element connected in parallel to one of the third and fourth capacitors, and the smoothing circuit is connected in parallel to a series connection of the third and fourth capacitors. 2. The DC-DC converter according to claim 1, wherein the DC-DC converter includes a choke coil and a fifth capacitor, and a voltage across the fifth capacitor is output.

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