JP2005328688A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To minimize an increase in circuit scale and cost, and to improve a power factor by a power regenerating system. <P>SOLUTION: A control transformer PRT is provided with a high-frequency inductor as controlled wiring NR. The inductance of the controlled wiring NR is variably controlled by a first amplifying circuit system including transistors Q12, Q13, in response to the level of a rectified and smoothed voltage Ei (an AC level of a commercial AC power supply), and also variably controlled by a second amplification circuit system including Q11 in response to the load current level i.e. load fluctuations. The power factor value improved by a power factor improving means can be controlled so as to be constant with respect to fluctuations of a load and an AC input voltage VAC. The magnetic flux density of an isolated converter transformer is set to a predetermined value or lower. A secondary-side rectification operation is always brought into a continuous mode for preventing the discontinue period of a secondary-side rectification current regardless of the fluctuation of a secondary-side DC output voltage i.e. level fluctuations of the load and the commercial AC power (AC input voltage). <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit including a power factor correction circuit.

The present applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. Various power supply circuits configured with a power factor correction circuit for improving the power factor of the resonant converter have also been proposed.
FIG. 22 is a circuit diagram showing an example of a switching power supply circuit having a power factor correction function, which is configured based on the invention previously filed by the present applicant.

  The power supply circuit of FIG. 22 employs a configuration in which a power factor correction circuit 20 is provided for a separately excited type current resonance type switching converter. The switching converter in this power supply circuit is a switching converter that combines a current resonance type converter using a half-bridge coupling method and a partial voltage resonance circuit that performs voltage resonance only when the semiconductor switch (switching element) is turned off.

In the power supply circuit shown in FIG. 22, first, a common mode noise filter composed of two sets of filter capacitors CL and CL and one set of common mode choke coil CMC is connected to commercial AC power supply AC.
As a rectifying / smoothing circuit for generating a DC input voltage from the commercial AC power supply AC, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci is provided for the subsequent stage of the common mode noise filter.
In this case, the rectified output obtained by the full-wave rectification operation of the bridge rectifier circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 20. As a result, a rectified and smoothed voltage Ei (DC input voltage) corresponding to a level equal to the AC input voltage VAC is obtained at both ends of the smoothing capacitor Ci.
The power factor correction circuit 20 will be described later.

  As shown in the figure, as the current resonance type converter for switching by inputting the DC input voltage, two switching elements Q1, Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 formed of body diodes are connected in parallel with each other between the drains and sources of the switching elements Q1 and Q2, respectively, in the direction shown in the drawing.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, in order to switch the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 using a general-purpose IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. Then, a drive signal (gate voltage) having a required frequency is applied to each gate of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, thereby obtaining a switching output.
The other end of the primary winding N1 is connected to the connection point between the cathode of the fast recovery type switching diode D1 and the high frequency inductor L10 in the power factor correction circuit 20 through the series resonant capacitor C1. Yes.

  In this case, the series resonant capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonant capacitor C1 and the leakage inductance of the insulating converter transformer PIT including the primary winding N1 (series resonant winding) ( The primary side series resonance circuit for making the operation of the switching converter into a current resonance type is formed by the leakage inductance component L1.

According to the above description, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the part by the partial voltage resonance circuit (Cp // L1) described above. A voltage resonance operation is obtained.
That is, the power supply circuit shown in this figure adopts a form in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit. In this specification, such a switching converter is referred to as a composite resonance type converter.

In this case, the low-voltage winding N4 is wound on the primary side of the insulating converter transformer PIT. An alternating voltage induced by the primary winding N1 is generated in the low voltage winding N4. A half-wave rectifier circuit composed of a diode D4 and a capacitor C4 is connected to the low-voltage winding N4. This half-wave rectifier circuit performs a rectifying / smoothing operation to generate a low-voltage DC voltage E4 as a voltage across the capacitor C4. Generate.
The oscillation / drive circuit 2 is supplied with operating power obtained by setting the low-voltage DC voltage E4 to a predetermined level by the resistor R4. The Zener diode ZD4 inserted between the resistor R4 and the power input terminal of the oscillation / drive circuit 2 and the primary side ground has a voltage of the operating power supplied to the oscillation / drive circuit 2 above a certain level. A protection circuit is formed so as not to exceed.

Although illustration explanation here is omitted, the structure of the insulating converter transformer PIT includes, for example, an EE type core combined with an E type core made of a ferrite material. And after dividing | segmenting a winding site | part by the primary side and a secondary side, primary winding N1 (and low voltage | pressure winding N4) and secondary winding N2 are made with respect to the center magnetic leg of EE type | mold core. Wrapped.
Further, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT. Thus, a coupling coefficient of about 0.85 is obtained by the primary winding N1 and the secondary winding N2.

A secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT.
In this case, the secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and a double-wave rectifier circuit comprising rectifier diodes Do1 and Do2 and a smoothing capacitor Co is provided. Connected. Thereby, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the control circuit 1.

In this case, the control circuit 1 outputs, for example, a control signal having a level corresponding to the fluctuation of the secondary side DC output voltage Eo to the oscillation / drive circuit 2. Based on the control signal supplied from the control circuit 1, the oscillation / drive circuit 2 changes the frequency of the switching drive signal supplied from the oscillation / drive circuit 2 to the gates of the switching elements Q1 and Q2, thereby making the switching frequency variable. Like to do. In this way, the switching frequency of the switching elements Q1 and Q2 is variably controlled according to the level of the secondary side DC output voltage Eo, so that the resonance impedance of the primary side series resonance circuit changes to change the primary side series resonance circuit. The energy transmitted from the primary winding N1 forming the secondary side to the secondary side is also varied, and the level of the secondary side DC output voltage Eo is variably controlled. Thereby, constant voltage control of the secondary side DC output voltage Eo is achieved.
Hereinafter, the constant voltage control method for stabilizing the DC voltage by variably controlling the switching frequency as described above will be referred to as a “switching frequency control method”.

Next, the configuration of the power factor correction circuit 20 will be described.
The power factor correction circuit 20 employs a configuration as a power regeneration system using a magnetic coupling type.
In the power factor correction circuit 20, a switching diode D1 and a high frequency inductor L10 are connected in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci.
The filter capacitor CN is inserted between the anode side of the switching diode D1 and the positive terminal of the smoothing capacitor Ci, thereby functioning as a filter that absorbs normal mode noise (alternating component of the switching period).

  The power factor correction circuit 20 is connected to the above-described primary side series resonance circuit (C1-N1 (L1)) at the connection point between the cathode of the switching diode D1 and the high frequency inductor L10. The switching output obtained in the circuit is fed back.

In the power factor correction circuit 20 having such a configuration, the primary side series resonance current flowing in the primary side series resonance circuit (C1-N1 (L1)) is regenerated as electric power in accordance with the switching output of the switching elements Q1, Q2. Thus, the feedback is made to the smoothing capacitor Ci through the high-frequency inductor L10. As a result, when the positive and negative absolute values of the AC input voltage VAC are 1/2 or more, the fast recovery type diode D1 is operated so that the rectified output voltage level is higher than the voltage across the smoothing capacitor Ci. The charging current to the smoothing capacitor Ci is allowed to flow even during the low period.
As a result, the average waveform of the AC input current IAC approaches the waveform of the AC input voltage, and the conduction angle of the AC input current IAC is expanded. As a result, the power factor is improved.

  When the rectifying / smoothing circuit for rectifying the commercial AC power supply AC is a voltage doubler rectifier circuit, the power factor correction circuit 20 of the type shown in FIG. 22 has a power factor correction circuit as shown in FIG. It can be configured as 20A. Note that the same portions as those in FIG. 22 are denoted by the same reference numerals and description thereof is omitted.

  A power factor correction circuit 20A shown in FIG. 23 includes rectifier diodes D11 and D12, smoothing capacitors Ci1 and Ci2, a high-frequency inductor L10, and a filter capacitor CN. The rectifier diodes D11 and D12 and the smoothing capacitors Ci1 and Ci2 are circuit components that form the voltage doubler rectifier circuit. Therefore, the power factor improving circuit 20A has a configuration combined with the voltage doubler rectifier circuit.

  In this case, the common mode noise filter includes a set of common mode choke coils CMC and a filter capacitor CL, and is connected to the commercial AC power supply AC as shown in the figure.

In the power factor correction circuit 20A, one end of the high-frequency inductor L10 is connected to one line of the commercial AC power supply AC that is the subsequent stage of the common mode noise filter. The other end of the high frequency inductor L10 is connected to a connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12.
The cathode of the rectifier diode D11 is connected to the positive terminal of the smoothing capacitor Ci1, and the anode of the rectifier diode D12 is connected to the primary side ground.
Two sets of smoothing capacitors Ci1 and Ci2 are connected in series. In addition, the positive terminal of the smoothing capacitor Ci1 is connected to the cathode of the rectifier diode D11 as described above, and is also connected to the drain side of the switching element Q1. The negative terminal of the smoothing capacitor Ci2 is connected to the primary side ground. A connection point between the negative electrode terminal of the smoothing capacitor Ci1 and the positive electrode terminal of the smoothing capacitor Ci2 is connected to another line of the commercial AC power supply AC, which is a subsequent stage of the common mode noise filter. Here, fast recovery diodes are selected as the rectifier diodes D11 and D12.
The filter capacitor CN is inserted between a connection point between the high-frequency inductor L10 and the line of the commercial AC power supply AC and a connection point between the smoothing capacitor Ci1 and the smoothing capacitor Ci2.
The end of the primary side series resonance circuit (C1-N1 (L1)) is connected to a connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12.

  In the voltage doubler rectifier circuit formed in the power factor correction circuit 20A constituted by the above-described connection form, the rectifier diode D11 is used in the commercial AC power supply AC (AC input voltage VAC) in one half cycle. The AC power source AC is rectified, and the smoothing capacitor Ci1 smoothes the rectified output, thereby generating a rectified and smoothed voltage corresponding to the commercial AC power source AC as the voltage across the smoothing capacitor Ci1. Similarly, in a period in which the commercial AC power supply AC (AC input voltage VAC) is in the other half cycle, the rectifier diode D12 rectifies the commercial AC power supply AC, and the smoothing capacitor Ci2 smoothes the rectified output. As a voltage across the smoothing capacitor Ci2, a rectified smoothing voltage corresponding to the same size as the commercial AC power supply AC is generated. As a result, the rectified and smoothed voltage Ei having a level corresponding to twice the level of the commercial AC power supply AC is obtained as the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. That is, the voltage doubler rectification operation is performed, and this rectified and smoothed voltage Ei is supplied as a DC input voltage to the subsequent switching converter.

In the power factor correction circuit 20A, the end of the primary side series resonance circuit (C1-N1 (L1)) is connected to the connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12. For this reason, the power factor improving circuit 20A also regenerates the primary side series resonance current flowing through the primary side series resonance circuit (C1-N1 (L1)) as electric power, and passes through the high frequency inductor L10 so as to pass the smoothing capacitor Ci. It is possible to obtain an operation of returning to
As a result, in the power factor correction circuit 20A, the fast recovery rectifier diodes D11 and D12 have an absolute value of the AC input voltage VAC of 1/2 in each half cycle period in which the AC input voltage VAC is positive / negative. At the above time, the rectification current is intermittently operated by performing switching. That is, in this circuit, the rectifier diodes D11 and D12 serve as the functions of the rectifier for rectifying the commercial AC power supply AC and the power factor improving switching elements Q1 and Q2 for switching the rectified current. As a result, similarly to the power factor correction circuit 20 of FIG. 22, the charging current to the smoothing capacitor Ci is allowed to flow even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. The conduction angle of the input current IAC is expanded and the power factor is improved.

By the way, in the configuration of the power factor improvement circuits 20 and 20A shown in FIGS. 22 and 23, the high frequency inductor L10 and the power factor improvement circuit that are assumed to be in the rectification current path of the rectification smoothing circuit that generates the rectification smoothing voltage Ei. The primary side series resonance circuit is directly connected to the connection point of the switching elements (D1, D11, D12). That is, the switching outputs of the switching elements Q1 and Q2 are directly regenerated and fed back to the rectified current path of the rectifying and smoothing circuit that generates the rectified and smoothed voltage Ei. As a result, the ripple of the commercial AC power cycle superimposed on the primary side series resonant current flowing in the primary side series resonant circuit increases. As a result, the ripple voltage of the commercial power cycle of the secondary side DC output voltage Eo is also increased. , It will increase compared to the case without the power factor improvement configuration.
For example, the power circuit shown in FIGS. 22 and 23 configured to have a power factor of about 0.8 or more at the maximum load power, and the power factor improving circuits 20 and 20A are omitted from the power circuits shown in FIGS. When compared with the power supply circuit having the above-described configuration, the ripple voltage of the secondary side DC output voltage Eo in the former power supply circuit increases 5 to 6 times the ripple voltage of the latter power supply circuit.

  As a countermeasure, the capacitance of the smoothing capacitor Co for generating the secondary side DC output voltage must be increased 5 to 6 times. That is, even if the gain of the control circuit 1 is improved as much as possible, in order to obtain the same ripple voltage level as that of the circuit before the power factor improvement, it is necessary to increase the capacitance of the smoothing capacitor Co 5 to 6 times. The cost will be greatly increased, and practical application will be impractical. Since such a countermeasure is not practical, it is difficult to employ the power supply circuit as shown in FIGS. 22 and 23 in a device that strictly requires, for example, a ripple voltage to be a certain level or less.

  In view of this, the present applicant has previously proposed that a power supply circuit having a power factor improvement function based on a power regeneration system be configured as the power factor correction circuit illustrated in FIGS. 24 to 27, the same portions as those in FIGS. 22 and 23 are denoted by the same reference numerals, and the description thereof is omitted.

First, the power supply circuit shown in FIG. 24 will be described.
The basic configuration of the power supply circuit shown in FIG. 24 is the same as that shown in FIG. 22, and is a full wave consisting of a bridge rectifier circuit Di and one smoothing capacitor Ci as a rectifying / smoothing circuit for generating a rectified smoothing voltage Ei from the commercial AC power supply AC. A rectifier circuit is provided. In addition, the switching converter adopts a composite resonance type converter in which a half-bridge coupling type current resonance type converter by separate excitation and a primary side partial voltage resonance circuit are combined.

24, first, the tertiary winding N3 is wound around the primary side of the insulation converter transformer PIT. Then, as shown in the figure, a series connection circuit of a switching diode D1, a high-frequency inductor L10, and a tertiary winding N3 is formed. Insert for the line between. A line between the positive electrode output terminal of the bridge rectifier circuit Di and the positive electrode terminal of the smoothing capacitor Ci becomes a part of a rectified current path formed by the full-wave rectifier circuit including the bridge rectifier circuit Di and the smoothing capacitor Ci.
The filter capacitor CN is inserted between the connection point between the anode of the switching diode D1 and the positive output terminal of the bridge rectifier circuit Di and the connection point between the tertiary winding N3 and the positive terminal of the smoothing capacitor Ci. In other words, the switching diode D1, the high-frequency inductor L10, and the tertiary winding N3 are connected in parallel.

In such a configuration of the power factor correction circuit 21, an alternating voltage is induced in the tertiary winding N3 according to the switching output (primary side series resonance current) obtained in the primary winding N1 forming the primary side series resonance circuit. Is done. The tertiary winding N3 is inserted in series with the rectification current path of the commercial AC power supply AC together with the switching diode D1 and the high-frequency inductor L10. The voltage will be superimposed. That is, in this case, the primary side series resonance current is regenerated as power through the magnetic coupling of the primary winding N1 and the tertiary winding N3, and is fed back to the rectification current path.
The switching diode D1 is applied with the rectified output voltage on which the alternating voltage component is superimposed as described above, so that the bridge rectifier circuit Di can be obtained even when the absolute value of the AC input voltage VAC is ½ or more. The rectified current obtained by the rectifying operation is switched (interrupted). As a result, the power factor correction circuit 21 also allows the charging current to flow to the smoothing capacitor Ci even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, and the AC input current IAC is conducted. The corners are expanded to improve the power factor.

Next, the power supply circuit shown in FIG. 25 will be described.
In the power supply circuit of FIG. 25, the rectifying / smoothing circuit for generating the rectified smoothing voltage Ei is a voltage doubler rectifying circuit as in FIG. The power factor correction circuit 21A shown in FIG. 25 has a configuration in which the form of the power factor improvement circuit 21 shown in FIG. 24 is applied to a voltage doubler rectifier circuit.
In this power factor correction circuit 21A, a voltage doubler rectifier circuit is formed by rectifier diodes D11 and D12 and two smoothing capacitors Ci1 and Ci2 connected in series as in the case of FIG. Further, since the rectifier diodes D11 and D12 also function as switching elements for power factor improvement, a high-speed recovery type is selected.

A high frequency inductor L10 is connected to the positive line of the commercial AC power supply AC, which is a subsequent stage of the common mode noise filter (CMC, CL), and a tertiary winding N3 is connected in series to the high frequency inductor L10. . The end of the tertiary winding N3 on the side not connected to the high frequency inductor is connected to a connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12.
The cathode of the rectifier diode D11 is connected to the positive terminal of the smoothing capacitor Ci1, and the anode of the rectifier diode D12 is connected to the primary side ground.
The connection point of the smoothing capacitors Ci1 and Ci2 connected in series is connected to the positive line of the commercial AC power supply AC that is the subsequent stage of the common mode noise filter (CMC, CL). The negative terminal of the smoothing capacitor Ci2 is grounded to the primary side ground.
The filter capacitor CN is inserted between the connection point of the positive line of the commercial AC power supply AC and the high frequency inductor L10 and the connection point of the smoothing capacitors Ci1 and Ci2.

The operation of the voltage doubler rectifier circuit in the power factor correction circuit 21A configured in this way is as follows. During the period when the commercial AC power supply AC is in one half cycle (positive polarity), the commercial AC power supply AC → (CMC winding). Line) → high frequency inductor L10 → tertiary winding N3 → rectifier diode D11 → smoothing capacitor Ci1 → (winding of CMC) → commutation current path of commercial AC power supply AC is formed, and rectification diode D11 rectifies commercial AC power supply AC The smoothing capacitor Ci1 can smooth the rectified output.
Further, during the period in which the commercial AC power source AC is in the other half cycle (negative polarity), the commercial AC power source AC → (CMC winding) → smoothing capacitor Ci2 → rectifier diode D12 → tertiary winding N3 → high frequency inductor L10 → (CMC) → A rectified current path of the commercial AC power supply AC is formed, the rectifier diode D12 rectifies the commercial AC power supply AC, and the smoothing capacitor Ci2 smoothes the rectified output.
As a result, a rectified and smoothed voltage having a level corresponding to the same size as that of the commercial AC power supply AC is obtained as the voltage across the smoothing capacitors Ci1 and Ci2. Therefore, the rectified and smoothed voltage Ei having a level corresponding to twice the level of the commercial AC power supply AC is obtained as the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. That is, a voltage doubler rectification operation is obtained.

  Further, according to the rectified current path described above, a series connection circuit of the high frequency inductor L10, the tertiary winding N3, and the rectifier diode D11 is formed during the half-cycle period in which the commercial AC power supply AC is positive. In a half-cycle period in which AC is negative, a series connection circuit of rectifier diode D12 → tertiary winding N3 → high frequency inductor L10 is formed. As a result, in each period in which the commercial AC power supply AC is positive / negative, the rectifier diodes D11 and D12 are switched on / off so that the rectifier currents are switched by the superimposed alternating voltage induced in the tertiary winding N3. Will be obtained. As a result, as understood from the above description, the power factor is improved.

In the case of the power factor correction circuits 21 and 21A shown in FIGS. 24 and 25, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit (C1−N1 (L1)) is obtained from the insulation converter transformer PIT. Via the magnetic coupling, the voltage is induced in the tertiary winding N3 as a voltage and fed back to the rectification current path. In other words, as in the case of the power factor correction circuits 20 and 20A in FIGS. 22 and 23, the switching output of the primary side series resonance circuit is directly fed back to the rectified current path. Not.
Accordingly, in the power supply circuit shown in FIGS. 24 and 25, the ripple voltage of the commercial power supply cycle of the DC output voltage Eo on the secondary side is significantly reduced as compared with the power supply circuits shown in FIGS. .

  Next, the power supply circuit shown in FIG. 26 employs the same basic configuration as the power supply circuits of FIGS. 22 and 24. In other words, the rectifying / smoothing circuit for generating the rectifying / smoothing voltage Ei is a full-wave rectifying circuit composed of a bridge rectifying circuit Di and a smoothing capacitor Ci, and a half-bridge coupled current resonance converter using a separately excited type and a partial voltage on the primary side. A composite resonance type converter combined with a resonance circuit is provided.

  In addition, a power factor improving transformer VFT is provided in the power factor improving circuit 22 shown in FIG. This power factor improving transformer VFT is configured to have a predetermined coupling coefficient that is loosely coupled to the primary winding N11 and the secondary winding N12.

In this case, the primary winding N1 of the power factor improving transformer VFT is inserted in series between the primary winding N1 and the primary side series resonance capacitor C1. That is, it is provided so as to be included in the inductance component of the primary side series resonance circuit.
The secondary winding N12 of the power factor improving transformer VFT is connected in series with the switching diode D1. A series connection circuit composed of the switching diode D1 and the secondary winding N12 is inserted into the line of the rectified current path between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci.
In this case, the secondary winding N12 of the power factor improving transformer VFT is regarded as a combination of the functions of the tertiary winding N3 and the high frequency inductor L10 in the power factor improving circuit 21 shown in FIG. Can do. That is, it has a function as an inductor that regenerates the primary series resonance current by magnetic coupling and feeds it back as a voltage, and an inductor that receives an alternating voltage as the feedback output.

The power factor correction circuit 22 configured as described above is included in the primary side series resonance circuit in response to the switching output (primary side series resonance current) being obtained in the primary side series resonance circuit. A current as a switching output flows through the primary winding N11 of the power factor improving transformer VFT. In the power factor improving transformer VFT, an alternating voltage is induced in the secondary winding N12 in accordance with the alternating current flowing in the primary winding N11.
In this case, the secondary winding N12 of the power factor improving transformer VFT is inserted in the rectified current path of the commercial AC power supply AC in a form connected in series with the switching diode D1. For this reason, the alternating voltage induced in the secondary winding N12 is superimposed on the rectified output voltage. That is, the primary side series resonance current is voltage fed back to the rectification current path through the magnetic coupling of the power factor improving transformer VFT. The switching diode D1 operates so as to switch the rectified current by the superimposed amount of the alternating voltage. As a result of such an operation, the conduction angle of the AC input current IAC is expanded and the power factor is improved.

FIG. 27 shows a configuration in which the power factor improving configuration including the power factor improving transformer VFT is applied when the rectifying / smoothing circuit for generating the rectifying / smoothing voltage Ei is a voltage doubler rectifying circuit. In this figure, the same parts as those in FIG.
In the power factor improvement circuit 22A shown in FIG. 27, the secondary winding N12 of the loosely coupled power factor improvement transformer VFT is connected to the positive line of the commercial AC power source in the rectified current path of the commercial AC power source AC and the fast recovery type. The rectifier diode D11 is connected to the connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12. That is, also in this case, the secondary winding N12 of the power factor improving transformer VFT can be viewed as an equivalent of the tertiary winding N3 and the high-frequency inductor L10 previously shown in FIG. it can.

  Therefore, also in the power factor correction circuit 22A, in the same manner as the power factor improvement circuit 22 shown in FIG. 26, the primary side series resonance current is regenerated by the power factor improvement transformer VFT to generate a voltage with respect to the rectification current path. As a result, the power factor is improved.

In the power factor correction circuits 22 and 22A shown in FIGS. 26 and 27, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit (C1−N1 (L1)) is a loose coupling force. The voltage is fed back to the rectification current path through the magnetic coupling of the rate improving transformer VFT. That is, even with this configuration, as in the case of the power factor correction circuits 20 and 20A of FIGS. 22 and 23, the configuration in which the switching output of the primary side series resonant circuit is directly applied to the rectified current path. Not as.
Thereby, also in the power supply circuit shown in FIGS. 24 and 25, the ripple voltage of the commercial power supply cycle of the DC output voltage Eo on the secondary side is reduced as compared with the power supply circuits shown in FIGS.

JP 2003-189615 A JP 2003-189627 A

As described above, in the power supply circuit shown in FIGS. 24 to 27, the configuration of power factor improvement by the power regeneration system including the tertiary winding N3 of the insulating converter transformer PIT or the loosely coupled power factor improvement transformer VFT. As a result, the switching output from the primary side series resonance circuit to the rectification current path of the commercial AC power supply AC is not made direct, thereby reducing the ripple of the secondary side DC output voltage Eo. Yes.
However, in the power supply circuit having such a configuration, the power factor that is improved by the operation of the power factor correction circuit as the input level of the commercial AC power supply AC increases and the level of the AC input voltage VAC tends to increase. Has been found to decrease.
Further, in these power supply circuits, it is known that the value of the power factor decreases even when the load power Po of the load connected to the secondary side DC output voltage Eo becomes a light load condition.
For example, assuming that the power circuits shown in FIGS. 24 to 27 are set so that a power factor of about 0.8 or more can be obtained at the maximum load power, for example, the load power having a value that is ½ of the maximum load power. Under the conditions, the power factor falls below 0.75. When the power factor decreases like this, the power factor may not be sufficient depending on conditions.

In view of the above problems, the present invention is configured as a switching power supply circuit as follows.
That is, a rectifying / smoothing means for generating a rectified and smoothed voltage by inputting a commercial AC power supply, a switching means formed by including a switching element for performing switching by inputting the rectified and smoothed voltage as a DC input voltage, and switching driving the switching element. Switching driving means.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. An insulating converter transformer is formed.
Further, the primary side series is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series with the primary winding, and the operation of the switching means is a current resonance type. A resonance circuit is provided.
Also, a secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage, A constant voltage configured to perform constant voltage control on the secondary side DC output voltage by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. Control means are provided.
Moreover, a power factor improvement means is provided.
The power factor improving means is a series connection circuit comprising a power factor improving switching element and a high frequency inductor, and the power factor improving series connection inserted in series with respect to the rectified current path formed by the rectifying and smoothing means. Provide a circuit. Further, a saturable reactor in which a high-frequency inductor is used as a controlled winding, a detection winding wound in a tightly coupled state with respect to the controlled winding, and a control winding is provided. A control transformer is provided that operates so as to vary the inductance of the controlled winding in accordance with the level of the control current that is a direct current flowing through the winding.
A first control current generating circuit that operates to flow the control current at a level corresponding to the level of the commercial AC power supply; and a level according to a voltage level induced in the detection winding from the controlled winding A second control current generation circuit that operates so as to flow the control current.
Then, the end of the primary side series resonance circuit is connected to the connection point between the power factor improving switching element and the high frequency inductor.
For the insulating converter transformer, the secondary side rectified current is set to the continuous mode regardless of the fluctuation of the secondary side DC output voltage, and the magnetic flux density is set to be a predetermined value or less.

The switching power supply circuit according to the present invention having the above configuration includes a current resonance type converter as the primary side switching converter. Also, power factor improvement is achieved by connecting the end of the primary side series resonant circuit to the connection point between the power factor improving switching element and the high frequency inductor, thereby obtaining the primary side series resonant circuit obtained in the primary side series resonant circuit. A configuration is adopted in which the current is regenerated as power and is performed by a power regeneration system that feeds back to the rectified current path.
In addition, the power factor improving means is provided with a control transformer having a high-frequency inductor as a controlled winding. First, the first control current generating circuit described above controls the power transformer according to the level of the commercial AC power supply (AC input voltage). The level of the control current flowing through the control winding of the control transformer is made variable. That is, the inductance of the high-frequency inductor (controlled winding) is varied by the operation of the first control current generation circuit.
In this way, by changing the inductance of the high-frequency inductor (controlled winding) in the power factor improving means, the feedback amount of the switching output obtained in the power factor improving means is changed according to the fluctuation of the AC input voltage level. As a result, it is possible to control the power factor value improved by the power factor improving means so as to be constant with respect to the fluctuation of the AC input voltage level.
At this time, as the high-frequency inductor, the end of the primary side series resonant circuit is connected as described above to form the inductance of the primary side series resonant circuit. The inductance of the primary side series resonant circuit also changes according to the voltage level. And by changing the inductance of the primary side series resonance circuit in this way, it also changes as the resonance frequency of the primary side series resonance circuit, and accordingly, the primary side series resonance current is changed according to the level of the AC input voltage. The level is controlled, and as a result, the level of the primary side series resonance current is controlled to be substantially constant regardless of the fluctuation of the AC input voltage level. If the level of the primary side series resonance current becomes constant with respect to the fluctuation of the AC input voltage level, the power transmitted to the secondary side can be made constant regardless of the AC input voltage level. A wide-range configuration that can operate with a 200V system can be realized.
Further, the control transformer is further provided with a detection winding, and the control current flowing through the control winding is also controlled by the second control current generation circuit according to the voltage level induced in the detection winding. The level is made variable. The voltage induced in the detection winding corresponds to the current flowing through the high-frequency inductor (controlled winding). As described above, since the primary side series resonance circuit is connected to the high frequency inductor (controlled winding), the primary side series resonance current flows. Since the primary side series resonance current is the switching output of the current resonance type converter, it shows a level change corresponding to the load current level. Therefore, in this case, depending on the operation of the second control current generation circuit, the inductance of the high-frequency inductor (controlled winding) can be varied depending on the load current level.
Thus, if the inductance of the high-frequency inductor (controlled winding) is varied even in accordance with the load current level, the power factor value improved by the power factor improving means becomes constant with respect to the load fluctuation. To be.
Furthermore, in the present invention, by setting the magnetic flux density of the insulating converter transformer to a predetermined value or less, the secondary side DC output voltage changes, that is, regardless of load fluctuations or commercial AC power supply (AC input voltage) level fluctuations. As the side rectification operation, a continuous mode in which a period in which the secondary side rectification current is discontinuous does not occur is always set. When the secondary side rectification operation is in the continuous mode, the ripple of the commercial AC power cycle superimposed on the primary side series resonant circuit is reduced when the power factor is improved by the power regeneration method. The ripple voltage of the commercial AC power supply period superimposed on the secondary side DC output voltage is suppressed.

As described above, according to the present invention, as a switching power supply circuit for improving the power factor by the power regeneration method based on the current resonance type switching converter, the power factor for the fluctuation of the AC input voltage (commercial AC power supply) level and the load fluctuation is changed. This solves both the problem of reduction and the problem of the ripple of the commercial AC power supply period being superimposed on the secondary side DC output voltage.
In order to obtain such an effect, the power regeneration improvement configuration by the power regeneration method is to directly connect the end of the primary side series resonance circuit to the connection point of the high frequency inductance and the switching element for power factor improvement. The simplest configuration is possible. In addition, a control transformer and a control current generation circuit are added, and for the isolation converter transformer, the continuous mode of the secondary side rectification operation is maintained regardless of the fluctuation of the secondary side DC output voltage, and the magnetic flux below the required level is maintained. It tries to be density. In other words, the additional parts are only parts corresponding to the control transformer and the control current generation circuit, and the insulation converter transformer can be changed for setting the magnetic flux density, and the circuit scale and cost increase are minimized. It can be said that it is in place.

  Further, by configuring so that the inductance of the high-frequency inductor (controlled winding) is variably controlled according to the level of the commercial AC power supply (AC input voltage) as in the present invention, the level of the primary side series resonance current is reduced. It can be controlled to be constant with respect to fluctuations in the AC input voltage, thereby realizing a configuration compatible with a wide range.

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to a first embodiment as the best mode for carrying out the present invention (hereinafter also referred to as an embodiment). The power supply circuit shown in this figure employs a configuration in which a partial voltage resonance circuit is combined with a separately excited current resonance converter using a half-bridge coupling method as a basic configuration on the primary side.
In addition, the power supply circuit of the first embodiment shown in FIG. 1 has a so-called wide-range configuration that can operate in response to both AC100V and AC200V inputs.

In the power supply circuit shown in FIG. 1, first, a common mode noise filter including filter capacitors CL and CL and a common mode choke coil CMC is formed for a commercial AC power supply AC.
A full-wave rectifying / smoothing circuit including a bridge rectifier circuit Di (rectifier circuit unit) and a single smoothing capacitor Ci is connected to the commercial AC power supply AC that is the subsequent stage of the noise filter. This full-wave rectifier circuit connects the positive line of the commercial AC power supply AC to the positive input terminal of the bridge rectifier circuit Di, and connects the negative line of the commercial AC power supply AC to the negative input terminal of the bridge rectifier circuit Di. ing. The positive output terminal of the bridge rectifier circuit is connected to the positive terminal of the smoothing capacitor Ci through a series connection of the switching diode D1 and the controlled winding (high frequency inductor) NR in the power factor correction circuit 10.
The configuration and operation of the power factor correction circuit 10 will be described later.

  The full-wave rectifying / smoothing circuit receives the commercial AC power supply AC and performs full-wave rectifying operation, so that the rectified and smoothed voltage Ei (DC input voltage) obtained by smoothing the rectified output of the bridge rectifying circuit Di is applied to both ends of the smoothing capacitor Ci. ) Is obtained. In this case, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC. In this case, a low speed recovery type is selected for the four rectifier diodes forming the bridge rectifier circuit Di.

  As shown in the figure, the current resonance type converter for switching (intermittently) by inputting the DC input voltage includes a switching circuit in which two switching elements Q1, Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, an oscillation / drive circuit 2 is provided to drive the switching elements Q1, Q2 in a switching manner. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and for example, a general-purpose IC can be used. Then, a drive signal (gate voltage) having a required frequency is applied to the gates of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit in the oscillation / drive circuit 2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side.
One end of the primary winding N1 of the insulation transformer PIT is connected to a controlled winding NR (high frequency inductor) in the power factor correction circuit 10 via a series connection of a primary side series resonant capacitor C1, and a high speed type (high speed). A recovery type switching diode D1 is connected to the connection point with the anode. The other end of the primary winding N1 is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, so that the switching output is transmitted. Yes.

  Here, the insulating converter transformer PIT generates a required leakage inductance L1 in the primary winding N1 of the insulating converter transformer PIT by a structure described later. A primary side series resonant circuit is formed by the capacitance of the primary side series resonant capacitor C1 and the leakage inductance L1. According to the connection mode described above, the switching outputs of the switching elements Q1, Q2 are transmitted to the primary side series resonance circuit. The primary side series resonance circuit performs a resonance operation by the transmitted switching output, thereby making the operation of the primary side switching converter a current resonance type.

According to the above description, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the part by the partial voltage resonance circuit (Cp // L1) described above. A voltage resonance operation is obtained.
That is, the power supply circuit shown in this figure has a configuration as a complex resonance type converter in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit.

A low voltage winding N4 is wound around the primary side of the insulating converter transformer PIT. An alternating voltage induced by the primary winding N1 is generated in the low voltage winding N4.
A half-wave rectifier circuit composed of a diode D4 and a capacitor C4 is connected to the low-voltage winding N4. This half-wave rectifier circuit performs a rectifying / smoothing operation to generate a low-voltage DC voltage E4 as a voltage across the capacitor C4. Generate.
The oscillation / drive circuit 2 is supplied with operating power obtained by setting the low-voltage DC voltage E4 to a predetermined level by the resistor R4. The Zener diode ZD4 inserted between the resistor R4 and the power input terminal of the oscillation / drive circuit 2 and the primary side ground has a voltage of the operating power supplied to the oscillation / drive circuit 2 above a certain level. A protection circuit is formed so as not to exceed.

An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding N2 of the insulating converter transformer PIT.
In this case, the secondary winding N2 of the insulating converter transformer PIT is divided into the secondary winding N2A and the secondary winding N2B with the center tap as a boundary by providing a center tap as shown in the figure. . These secondary winding N2A and secondary winding N2B are wound with the same predetermined number of turns.

The center tap of the secondary winding N2 is connected to the secondary side ground.
The secondary winding N2 is connected to a double-wave rectifier circuit comprising rectifier diodes Do1 and Do2 and a smoothing capacitor Co as shown in the figure. Here, the anode of the rectifier diode Do1 is connected to the end of the secondary winding N2A, and the anode of the rectifier diode Do2 is connected to the end of the secondary winding N2B. The cathodes of the rectifier diodes Do1 and Do2 are connected to the positive terminal of the smoothing capacitor Co, and the negative terminal side of the smoothing capacitor Co is grounded to the secondary side ground. The rectifier diodes Do1 and Do2 are selected as high-speed Schottky diodes according to the rectification of the alternating voltage of the switching period.
A secondary side DC output voltage Eo is obtained as a voltage across the smoothing capacitor Co by the secondary-side double-wave rectifier circuit thus formed. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the control circuit 1 described below.

The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage Eo to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. By changing the switching frequency of the switching elements Q1 and Q2, the resonance impedance of the primary side series resonance circuit is changed and transmitted from the primary winding N1 of the insulating converter transformer PIT to the secondary winding N2 (N2A, N2B) side. Although the amount of electric power to be changed also changes, this operates so as to stabilize the level of the secondary side DC output voltage Eo.
For example, when the secondary-side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. However, this reduces the resonant impedance. The secondary side DC output voltage Eo is raised. On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the resonance impedance is increased by controlling the switching frequency to be increased, and the secondary side The DC output voltage Eo is reduced.

Next, the configuration of the power factor correction circuit 10 will be described.
As described above, the power factor correction circuit 10 is provided so as to be inserted into the rectification current path in the rectification smoothing circuit for obtaining the DC input voltage (Ei) from the commercial AC power supply AC. The magnetic coupling type is adopted as the power regeneration system.

  In the power factor correction circuit 10, the anode of a fast recovery type switching diode (power factor correction switching element) D1 is connected to the positive output terminal of the bridge rectifier circuit Di. The anode of the switching diode D1 is connected to the positive terminal of the smoothing capacitor Ci via a series connection of the controlled winding NR of the control transformer PRT. In other words, in this case, in the rectified current path for generating the rectified and smoothed voltage Ei, the switching diode D1−the controlled winding is applied to the line between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. A series connection circuit of the line NR is inserted. Here, the controlled winding NR has a function as a high-frequency inductor that receives feedback of the primary-side series resonance current that has been regenerated in the power factor correction circuit using the power regenerative method.

  The control transformer PRT provided in the power factor correction circuit 10 winds the controlled winding NR, and the control winding Nc so that the winding direction is orthogonal to the controlled winding NR, for example. Is configured to function as a saturable reactor, whereby the leakage inductance in itself is varied according to the level of the direct current (control current Ic) flowing through the control winding Nc.

In such a control transformer PRT, the level of the direct current flowing through the control winding Nc depends on the amplified output of the first amplifier circuit, which is composed of the transistors Q12 and Q13 and resistors R13 to R18 shown in the figure. Variable.
In this first amplifier circuit, first, the resistor R17 and the resistor R18 are divided resistors, and the series connection circuit of these resistors R17-R18 has a connection point between the positive terminal of the smoothing capacitor Ci and the drain of the switching element Q1. It is inserted in series between the primary side ground.
The base of the transistor Q13 is connected to the connection point between the resistor R17 and the resistor R18. The emitter of the transistor Q13 is connected to the primary side ground via a resistor R16. The collector of the transistor Q13 is connected to the low-voltage DC voltage E4 described above via a series connection of a resistor R13 and a resistor R5 as shown. Note that a Zener diode ZD5 inserted between the connection point of the resistors R13 and R5 and the primary side ground is provided to protect the control current Ic from an unacceptable level.

A series connection circuit consisting of a resistor R14 and a resistor R15 is inserted between the collector of the transistor Q13 and the primary side ground as shown in the figure. Further, a connection point between these resistors R14 and R15 is connected to the collector of the transistor Q13. The base of the transistor Q12 is connected.
The emitter of this transistor Q12 is also connected to the primary side ground. The collector of the transistor Q12 is connected to the low-voltage DC voltage E4 through the series connection of the control winding Nc and the resistor R5.

  In the first amplifier circuit having such a configuration, a direct current corresponding to the level of the rectified and smoothed voltage (DC input voltage) Ei is obtained by the resistors R17 to R18. An amplification operation is performed by the transistors Q13 and Q12 according to the DC current obtained in this way, and the collector current of the transistor Q12 as the amplification output is passed from the low-voltage DC voltage E4 through the resistor R5 and the control winding Nc. It will flow. In other words, the first amplifier circuit operates so that a control current Ic having a level corresponding to the level of the DC input voltage Ei flows through the control winding Nc.

In this case, the level of the control current Ic flowing through the control winding Nc of the control transformer PRT is also variable according to the alternating voltage level obtained in the illustrated detection winding ND wound around the control transformer PRT. The
The detection winding ND is wound around the control transformer PRT so as to be tightly coupled to the controlled winding NR in the same winding direction as the controlled winding NR.

  One end of the detection winding ND is connected to the primary side ground, and the other end is connected to the anode of the rectifying diode D5. The cathode of the diode D5 is connected to the positive terminal of the capacitor C5, and the negative terminal of the capacitor C5 is connected to the primary side ground. That is, a half-wave rectifier circuit is formed by the diode D5 and the capacitor C5. Depending on the half-wave rectifier circuit, a voltage VD1 obtained by rectifying and smoothing the alternating voltage VD obtained in the detection winding ND and converting it to a direct current is obtained as a voltage across the capacitor C5.

Then, this voltage VD1 is input to a second amplifier circuit made up of a transistor Q11, a resistor R10, a resistor R11, and a resistor R12 as shown.
In the second amplifier circuit, the resistor R10 is a base resistor and is inserted between the positive terminal of the capacitor C5 and the base of the transistor Q10. The resistor R11 is a base bias resistor and is connected between the base of the transistor Q11 and the primary side ground. The collector of the transistor Q11 is connected to the low-voltage DC voltage E4 via a series connection of the control winding Nc of the control transformer PRT and the resistor R5. As a result, the collector current of the transistor Q11, which is the output of the second amplifier circuit, flows from the low-voltage DC voltage E4 through the resistor R5 and the control winding Nc.
That is, with this configuration, in the second amplifier circuit, the control current Ic as a direct current of a level corresponding to the alternating voltage level induced in the detection winding ND is passed through the control winding Nc.
The operation accompanying the level control of the control current Ic by the first and second amplifier circuits will be described later.

  In the power factor correction circuit 10, one end of the primary winding N1 is connected to the connection point between the switching diode D1 and the controlled winding NR as described above. That is, the end of the primary side series resonance circuit (C1-N1 (L1)) is connected.

  The filter capacitor CN is provided to suppress the normal mode noise by absorbing the alternating component of the switching period generated by the switching operation of the switching diode D1, and in this case, the switching diode D1-controlled winding N R. Are connected in parallel to the series connection circuit.

  In such a circuit configuration of the power factor correction circuit 10, the switching output obtained in the primary side series resonance circuit (C 1 -N 1 (L 1)) is applied to the controlled winding NR that functions as a high frequency inductor in the power factor correction circuit 10. In contrast, it is communicated directly. Thereby, in the power factor correction circuit 10, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit (C1-N1 (L1)) is regenerated as electric power, and the controlled winding which is a high frequency inductor. It can be said that the operation of returning to the smoothing capacitor Ci through the magnetic coupling by the line NR is obtained.

2 and 3 are waveform diagrams corresponding to the power factor correction operation of the power factor correction circuit 10. Here, as will be described in detail later, the power supply circuit shown in FIG. 1 adopts a configuration corresponding to the wide range corresponding to the input of the commercial AC power supply AC of 100V system and 200V system. FIG. 3 shows an operation waveform when the input voltage VAC = 100V, and FIG. 3 shows an operation waveform when the AC input voltage VAC = 230V.
Note that the operation waveforms shown in these figures show the results when the load power Po = 100 W and the conditions are constant.

Here, as shown in FIGS. 2 and 3, it is assumed that the switching elements Q1 and Q2 perform switching under a state where the AC input voltage VAC is input. Accordingly, as described above, power regeneration is performed from the primary side series resonance circuit to the power factor correction circuit 10 side.
Accordingly, the voltage V1 that is the potential between the connection point of the switching diode D1 and the controlled winding NR and the primary side ground in the power factor correction circuit 10 is, for example, a rectified smoothing voltage as shown in FIGS. A waveform obtained by superimposing an alternating voltage component obtained on the controlled winding NR, which is a high-frequency inductor, on Ei is obtained. Further, during the period in which the AC input current IAC is conducted and rectification is performed by the rectifier diode of the bridge rectifier circuit Di, the level of the voltage V1 decreases due to the rectified output voltage.

  The voltage V1 having the above waveform is applied to the fast recovery type switching diode D1, so that the switching diode D1 is turned on / off (switching). Since the switching diode D1 is inserted in the rectification current path, for example, even when the peak value of the positive / negative absolute value of the AC input voltage VAC is about ½ or more, the switching diode D1 causes the bridge rectifier circuit. An operation of intermittently flowing a rectified current that flows from the positive output terminal of Di is obtained.

  The conduction period of the rectified current obtained by the switching diode D1 performing the switching operation as described above is a period in which the rectified output voltage level output from the bridge rectifier circuit Di is lower than the voltage across the smoothing capacitor Ci. The conduction period of the AC input current IAC shown in FIG. 2 and FIG. 3 is also substantially coincident with the conduction period of the rectified output current. That is, the conduction angle of the AC input current IAC is larger than that without the power factor correction circuit, and the waveform of the AC input current IAC approaches the waveform of the AC input voltage VAC. Yes. That is, power factor improvement is achieved.

Here, the power factor correction circuit 10 in this case includes the control transformer PRT and the first amplifier circuit described above, and converts the power factor value improved by the operation as described above into the AC input voltage. Control is performed so as to be substantially constant with respect to the fluctuation of VAC. Hereinafter, this point will be described.
First, as described above, in the first amplifier circuit, a DC current corresponding to the level of the DC input voltage Ei is input via the resistors R17 and R18. Then, a collector current corresponding to the direct current flows through the transistor Q13.
At this time, since the collector of the transistor Q13 is also connected to the base of the transistor Q12 via the connection point of the resistors R14-R15 according to the above description, the collector current level of the transistor Q13 is increased. Along with this, the base current level of the transistor Q12 decreases, so that the level of the collector current of the transistor Q12 also decreases. Further, as the collector current level of the transistor Q13 decreases, the base current level of the transistor Q12 increases, and as a result, the collector current level of Q12 tends to increase.
That is, in this case, when the level of the DC input voltage Ei increases and the collector current of the transistor Q13 increases, the base current of the transistor Q12 decreases and the collector current of the transistor Q12 that is the output of the first amplifier circuit decreases. To be done. Further, when the level of the DC input voltage Ei decreases and the collector current of the transistor Q13 also decreases, the base current of the transistor Q12 increases and the collector current of the transistor Q12 also increases.

As a result, the first amplifying circuit operates so as to decrease the level of the control current Ic according to the increase in the level of the DC input voltage Ei, and the control current according to the decrease in the level of the DC input voltage Ei. It operates to increase the level of Ic.
At this time, the level of the DC input voltage Ei varies according to the level of the AC input voltage VAC (that is, the commercial AC power supply AC). In this case, the control current is increased according to the increase of the AC input voltage VAC. The operation is performed to increase the level of the control current Ic in response to the decrease in the level of Ic and the decrease in the AC input voltage VAC.

The control transformer PRT, which is a saturable reactor, operates so as to increase the inductance of the controlled winding NR in accordance with a decrease in the level of the control current Ic flowing through the control winding Nc.
According to this, the fact that the level of the control current Ic is decreased in accordance with the increase in the level of the AC input voltage VAC (DC input voltage Ei) as described above is that Control is performed so that the inductance of the control winding NR increases.

The increase in the inductance of the controlled winding NR that functions as a high-frequency inductor in the power factor correction circuit 10 means that the controlled winding is controlled according to a certain amount of power fed back from the primary side series resonant circuit. The energy stored in the inductor as the line NR also increases. That is, the inductance of the controlled winding NR is increased in accordance with the increase in the AC input voltage VAC as described above, which means that the amount of power feedback for power factor improvement is increased in accordance with the increase in the AC input voltage VAC. Control is performed so as to increase.
As a result of such control, the power factor tends to decrease as the AC input voltage VAC increases, but this tends to improve as the power feedback amount increases. As a result, the power factor can be controlled to be substantially constant with respect to the fluctuation of the AC input voltage VAC.

By the way, for the controlled winding NR in which the inductance control is performed as described above, as described above, the primary side series N1 of the insulating converter transformer PIT and the primary side series resonance capacitor C1 are connected. A resonant circuit is connected. As a result, the primary side series resonance current flowing in the primary side series resonance circuit is caused to flow to the smoothing capacitor Ci via the controlled winding NR.
According to such a connection form, it can be considered that the primary side series resonance circuit in this case is formed from the switching output point of the switching elements Q1, Q2 to the positive terminal of the smoothing capacitor Ci. When the primary side series resonance circuit is thus formed, the controlled winding NR is inserted in series with the primary winding N1 in the primary side series resonance circuit. Therefore, it can be considered that the inductance component of the primary side series resonance circuit is formed.

In view of this, the fact that the inductance of the controlled winding NR increases as the level of the DC input voltage Ei (that is, the AC input voltage VAC) increases as described above means that the AC input voltage VAC increases. Accordingly, the control is performed so as to increase the inductance value of the primary side series resonance circuit.
As the inductance of the primary side series resonant circuit is increased in this way, the resonance frequency of the primary side series resonant circuit is also changed, whereby the primary side series resonant current flowing through the primary side series resonant circuit is changed. The level will also change.

In the embodiment, the configuration corresponding to the wide range is realized by controlling the primary side series resonance current level accompanying the variable control of the inductance of the controlled winding NR.
That is, by using the variable inductance control operation of the controlled winding NR for the power factor stabilization control with respect to the level fluctuation of the AC input voltage VAC described above, the operation corresponding to the wide range is realized at the same time. It is what.

FIG. 4 and FIG. 5 are operation waveform diagrams for explaining the wide range compatible operation using such variable inductance control of the controlled winding NR.
In these figures, the voltage VQ2 generated at both ends of the switching element Q2 shown in FIG. 1 and the waveform of the switching current IQ2 of the switching element Q2 are shown, respectively. In FIG. FIG. 5 shows each waveform when the AC input voltage VAC = 230V. In these figures, the load power is constant at Po = 100W.

First, the switching period of the switching element Q2 is represented by the waveform of the voltage VQ2 shown in each figure. That is, the period during which the voltage VQ2 is at the 0 level represents the period during which the switching element Q2 is on, and the period during which the peak level is at the switching element Q2 is off. In this case, since the switching element Q2 and the switching Q1 are alternately turned on / off as described above, the voltage VQ1 (not shown) generated on the switching element Q1 side is the voltage VQ2 Thus, the waveform is obtained by reversing the positive and negative signs.
It is assumed that the switching periods of the switching elements Q1 and Q2 are substantially equal when the AC input voltage VAC = 100 and VAC = 230V shown in FIGS.

Further, the illustrated switching current IQ2 has a waveform indicating a switching output by the switching element Q2. If the switching output of one switching element Q1 is a switching current IQ1 (not shown), the waveform of the switching current IQ1 is 180 ° out of phase with respect to the waveform of the switching current IQ2, and is positive or negative. It will be obtained as an inverted waveform.
From this, the primary side series resonance current Io, which is a composite component of the switching current IQ1 and the switching current IQ2, is a sinusoidal waveform having a peak level PQ2 (as an absolute value) shown as a positive / negative peak level. It will be obtained as a waveform. In other words, the waveform of the switching current IQ2 shown in the drawing shows the waveform of such a half-wave period of the primary side series resonance current Io.

  As can be seen with reference to the waveforms of the switching current IQ2 and the voltage VQ2 shown in FIG. 4, the primary side series resonance current Io at the AC input voltage VAC = 100 V is a half-wave waveform as the switching current IQ2. However, it is made to fall within the period when the switching element Q2 is turned on. For this reason, in the power supply circuit in this case, corresponding to the AC 100 V system, the constants of the primary side series resonance circuit (L1, C1, and the circuit to be covered are set so that the switching period and the resonance frequency of the primary side series resonance circuit substantially coincide. It can be understood that the inductance of the control winding NR is set.

In this way, it is assumed that the AC input voltage VAC = 230 V is input under the condition that the switching cycle and the resonance frequency of the primary side series resonance circuit are substantially matched in the AC 100 V system.
First, as described above, control is performed so that the inductance forming the primary side series resonance circuit increases as the AC input voltage VAC increases due to the input corresponding to the AC 200V system.
Here, as the inductance of the primary side series resonance circuit increases, the resonance frequency of the primary side series resonance circuit tends to decrease. That is, as the AC input voltage VAC increases, control is performed so that the resonance frequency of the primary side series resonance circuit decreases.
As the resonance frequency is reduced in this way, the resonance waveform obtained in the primary side series resonance circuit (that is, the waveform that forms the basis of the waveform of the primary side series resonance current Io) is higher than that in the AC100V system. The period is made larger.
As the AC input voltage VAC increases as described above, the peak level of the primary side series resonance current Io should also increase.

However, in this case, assuming that the switching frequency is substantially equal between the AC 100V system and the 200V system as described above, the primary side actually obtained in the AC 200V system is corresponding to this case. As can be seen from the waveform of the series resonance current Io with reference to the waveform of the switching current IQ2 shown in FIG. 5, the half wave portion of the sine wave as the resonance waveform does not fall within the period when the switching element is turned on. It is obtained as a substantially sawtooth waveform that falls to 0 level before reaching the peak level of the half wave.
In this way, a sawtooth waveform that falls to 0 level before reaching the peak level of the sine wave based on the resonance waveform is used, so that the peak level PQ2 of the switching current IQ2 in this case (that is, the primary side series resonance current The peak level of Io is set lower than the level that should be originally obtained. That is, in this case, in response to the increase in the AC input voltage VAC, the control is performed such that the increase in the peak level of the primary side series resonance current Io is suppressed.
If the increase of the peak level of the primary side series resonance current Io is suppressed in accordance with the increase of the AC input voltage VAC in this way, the primary side series resonance current Io can be reduced even when the level of the AC input voltage VAC varies. The level can be controlled to be substantially constant.

Thus, if the level of the primary side series resonance current Io can be made substantially constant regardless of the level fluctuation of the AC input voltage VAC, the secondary side is excited according to the primary side series resonance current Io. The power level to be applied can be made substantially constant irrespective of the level fluctuation of the AC input voltage VAC.
In other words, as a result, the power level excited to the secondary side can be made equal between the AC100V system and the AC200V system, and it can operate corresponding to the input of the AC100V system and the AC200V system. Is realized.

As can be understood from the above description, in this case, in order to make the power factor constant, the power factor decreases with respect to the increase of the AC input voltage VAC, whereas the AC input voltage VAC increases. Accordingly, control is performed so as to increase the inductance of the controlled winding NR.
Even in the case of wide range compatibility, the inductance of the controlled winding NR is increased with respect to the increase of the AC input voltage VAC, whereas the primary side series resonance current level tends to increase as the AC input voltage VAC increases. Increased control should be performed.
In this way, since the control to be performed for the realization of both has the same tendency, in this case, each control can be performed by a common control system that varies the inductance of the controlled winding NR. It becomes possible. That is, in other words, both the constant power factor and the wide range are realized by configuring only one control system that varies the inductance of the controlled winding NR in accordance with the AC input voltage VAC. It is something that can be done.

Further, in the power factor correction circuit 10 shown in FIG. 1, in addition to the control according to the fluctuation of the AC input voltage VAC as described above, the load fluctuation is caused by the operation of the second amplifier circuit and the control transformer PRT. Control is performed so as to make the power factor substantially constant even in response to. This will be described.
First, as described above, in the control transformer PRT, the detection winding ND is wound so as to be tightly coupled to the controlled winding NR. An alternating voltage VD of a level corresponding to the alternating voltage generated in the controlled winding NR is induced. As described above, the controlled winding NR is an inductor inserted between the primary winding N1 and the smoothing capacitor Ci, so that a current of a level corresponding to the primary side series resonance current flows, and therefore the controlled winding NR. A voltage having a level corresponding to the primary side series resonance current is obtained at both ends of the control winding NR.
The level of the primary side series resonance current changes according to the load current level flowing through the load connected to the secondary side DC output voltage Eo. In other words, the level of the voltage VD induced in the detection winding ND thereby indicates the load current level. In the present embodiment, since the secondary side DC output voltage Eo is stabilized, the load current level is obtained as it is as the load power value. That is, it can be said that the load power is detected depending on the detection winding ND. Hereinafter, the voltage VD across the detection winding ND is also referred to as a detection voltage VD.

  As described above, the detection voltage VD is converted into a direct current by the half-wave rectifier circuit (D5, C5) and input to the second amplifier circuit including the transistor Q11. The second amplifier circuit is configured to flow a direct current as an amplified output through the control winding Nc as a control current Ic.

  The load current level decreases as the load power decreases (lightens). Accordingly, the level of the detection voltage VD obtained by the detection winding ND is also reduced because the level of the primary side series resonance current indicating the level change corresponding to the load current decreases as the load power decreases. As the value decreases, it decreases. For this reason, the level of the control current Ic, which is a direct current flowing through the control winding Nc, also tends to decrease as the load power decreases.

If the level of the control current Ic flowing through the control winding Nc decreases, the control transformer PRT that is a saturable reactor operates so as to increase the inductance of the controlled winding NR. That is, in this case, the control is performed so that the inductance of the controlled winding NR increases as the load power decreases.
As described above, when the inductance of the controlled winding NR increases, the feedback amount of the power fed back from the primary side series resonance circuit tends to increase accordingly, so that the load power decreases. Is controlled so that the power factor tends to improve.
In other words, the power factor tends to decrease as the load power becomes lighter due to this, and the control is performed so as to improve the power factor. It can be controlled to be constant.

Thus, the power factor stabilization control with respect to the load fluctuation is also performed by variable control of the inductance of the controlled winding NR provided in the power factor correction circuit 10.
For this reason, according to the configuration of the power supply circuit of the embodiment, the power factor stabilization control with respect to the fluctuation of the AC input voltage VAC described above, and the primary side series resonance current with respect to the fluctuation of the AC input voltage VAC are stabilized. For the control (that is, wide range compatible) and the power factor stabilization control with respect to the fluctuation of the load power, a variable inductance element required for realizing each of them is set as one of the controlled windings NR. It is understood that they can be shared.
Such commonization can be realized by a control transformer in which a primary side series resonance current is directly fed back as a power factor correction circuit, and a high frequency inductor receiving the feedback is wound as a controlled winding NR. This is because the PRT is provided.

  6 and 7 show control current Ic and power factor change characteristics as characteristics of the power supply circuit shown in FIG. FIG. 6 shows each characteristic with respect to fluctuations in the AC input voltage VAC = 85V to 288V, and FIG. 7 shows each characteristic with respect to fluctuations in the load power Po = 100 W to 0 W. In FIG. 6, the characteristic when the load power Po = 100 W is indicated by a solid line, and the characteristic when the load power Po = 25 W is indicated by a broken line. Further, in FIG. 7, the characteristics when the AC input voltage VAC = 100 V are indicated by solid lines, and the characteristics when the AC input voltage VAC = 230 V are indicated by broken lines.

First, according to FIG. 6, as the AC input voltage VAC increases, the level of the control current IC decreases both when the load power Po = 100 W and when Po = 25 W. It is shown that the power factor is controlled to be constant as a result of an increase in the inductance of the controlled winding NR as the control current level decreases.
From this, it is understood that the fluctuation of the AC input voltage VAC is such that the power factor PF is controlled to be constant by the operation of the control system including the first amplifier circuit described above.
Note that C2 indicated by an arrow in the figure indicates the difference in level of the control current Ic obtained when the load power Po = 100 W and Po = 25 W, respectively, when the AC input voltage VAC is the same value. Since the control current Ic level (control current IC1) is not changed by the control system including the first amplifier circuit if the AC input voltage VAC is the same value, this C2 is the control current Ic in the range of Po = 100 W to 25 W. This indicates the variable width of the level (control current IC2).

Further, according to FIG. 7, when the AC input voltage VAC = 100V and VAC = 230V, the level of the control current Ic is linear with respect to the fluctuation of the load power Po = 100W to the load power Po = 0W (no load). It can be seen that it has dropped. As a result of the operation of the control system including the second amplifier circuit that reduces the control current Ic level as the load power Po decreases in this way, the inductance of the controlled winding NR increases. The power factor PF is made constant with respect to load fluctuations as shown in the figure.
According to the experiment, the power factor PF in this case was constant at a power factor PF = 0.83 in the range of the load power Po = 100 W to 10 W when the AC input voltage VAC = 100 V. . Further, when the AC input voltage VAC = 230 V, the power factor PF = 0.83 was also constant in the range of the load power Po = 100 W to 25 W.
With such a characteristic, a sufficient power factor can be obtained even from the maximum to the minimum load power fluctuation range. For example, the power supply harmonic distortion regulation value must be cleared with a sufficient margin. Can do.

In this case, C1 indicated by an arrow in the figure indicates a difference in level of the control current Ic obtained when the load power Po is the same value when the AC input voltage VAC = 100V and VAC = 230V. . That is, as C1, the variable width of the control current Ic level (control current IC1) in the range of AC input voltage VAC = 100V to 230V is shown.
In this case, the variable width of the control current Ic itself is 0 to 160 mA as shown in the figure.

Here, as a reference, an example of the structure of the control transformer PRT provided in the power factor correction circuit 10 described so far will be described with reference to FIGS.
First, the insulating converter transformer PIT shown in FIG. 8 includes two double U-shaped cores CR1 and CR2 each having four magnetic legs. A solid core is formed by joining the ends of the magnetic legs of the double U-shaped cores CR11 and CR12. In this case, the double U-shaped cores CR11 and CR12 having the same size can be used.
When the three-dimensional core is formed in this way, there are four joint portions of the double U-shaped cores CR11 and CR12 corresponding to each of the four magnetic legs. In this case, gaps G and G having a predetermined length are formed for two adjacent joints in these four joints, and no gap is formed for the remaining two joints.

In the three-dimensional core formed in this way, first, the controlled winding NR is set in a predetermined manner so as to be wound around, for example, two magnetic legs that do not form the gap G on the side of the double U-shaped core CR11. Wind the number of turns. In this case, the detection winding ND is also wound with a predetermined number of turns so as to be tightly coupled to the controlled winding NR.
As shown in the figure, the control winding Nc is divided into a magnetic leg portion in which the gap G is formed and a magnetic leg portion in which the gap G adjacent to the magnetic leg is not formed on the double U-shaped core CR12 side. A predetermined number of turns are wound so as to be wound.

By winding the controlled winding NR (detection winding ND) and the control winding Nc as described above, the winding direction of the controlled winding NR (detection winding ND) is the same as that of the control winding Nc. It will be orthogonal to the winding direction. That is, a structure as a so-called orthogonal transformer is obtained.
With such an orthogonal transformer structure, the control transformer PRT is configured as a saturable reactor that is in a magnetic saturation state due to an increase in the direct current (control current Ic) flowing through the control winding Nc. Then, by using the saturable reactor, the inductance of the controlled winding NR changes according to the level of the control current Ic flowing through the control winding Nc.

In addition, as another structure of the control transformer PRT, as shown in FIG. 9, with respect to the three-dimensional core, one core remains a double U-shaped core CR12 having four magnetic legs. The core may be formed in combination as a single U-shaped core CR13 having an arbitrary U-shaped cross section instead of the double U-shaped core CR11. Also in this core structure, two gaps G and G are formed by the same positional relationship as the control transformer PRT of FIG.
In contrast to this core structure, the controlled winding NR (and the detection winding ND) is wound around the magnetic leg on the side where the gap is not formed in the single U-shaped core CR13. The winding N1 is wound around the magnetic leg on the side where no gap is formed in the single U-shaped core CR3.
For the control winding Nc, as shown in the drawing, a magnetic leg portion where a gap G is formed at a position on the magnetic leg end side of the double U-shaped core CR12 and a gap G adjacent to the magnetic leg are It winds so that it may wind around the magnetic leg part which is not formed.
Also in this case, the control winding Nc and the set of the controlled winding NR (and the detection winding ND) are set so that their winding directions are orthogonal to each other, and a configuration as an orthogonal transformer is obtained.

Further, the control transformer PRT may have a structure using a core having the shape shown in FIG. 10 in addition to using the above-described double U-shaped core or single U-shaped core.
First, in the control transformer PRT shown in FIG. 10, two half-eye-shaped cores CR21 and CR22 are prepared and combined so that the magnetic legs of these cores face each other, so that one planar-type core-shaped core is obtained. Form. In addition, in the square core, a total of four magnetic legs, two outside and two inside, face each other. Of these, each surface facing the two inside magnetic legs has a predetermined length. Gaps G and G are formed.
The controlled winding NR (and the detection winding ND) is wound a predetermined number of turns so as to straddle the two inner magnetic legs in one half-eye-shaped core CR21.
The control winding Nc is wound with a predetermined number of turns so as to straddle one outer magnetic leg in the other half eye-shaped core CR22 and one inner magnetic leg adjacent to the outer magnetic leg. To do.

  In such a structure of the control transformer PRT, the outer magnetic leg around which the controlled winding NR (and the detection winding ND) is wound is different from the outer magnetic leg around which the control winding Nc is wound. However, this relationship is equivalent to making the winding directions orthogonal as shown in FIGS. Therefore, even with the structure shown in FIG. 10, the control transformer PRT functions as a saturable reactor.

Here, when the control transformer PRT shown in FIG. 8 is taken as an example, the control transformer PRT is configured by selecting each part specifically as follows.
First, as the core size, the size of a × b × c shown in FIG. 8 is 16 mm × 16 mm × 23 mm. Thus, the size of the control transformer PRT is considerably reduced. Then, the gap G = 25 μm, the controlled winding NR = 25T, the detection winding ND = 1T, and the control winding Nc = 1000T are selected. The characteristics shown in FIG. 6 and FIG. 7 are obtained when the constants of the component parts of the insulating converter transformer PIT are selected as described later, and the control transformer PRT is set to the above-described selected values of each part. It is obtained.

  By the way, as a basic configuration of the power factor improvement circuit 10 of the power supply circuit shown in FIG. 1, a power regeneration system based on a magnetic coupling type is used, and the power factor improvement circuit 20 (20A) previously shown in FIG. 22 (and FIG. 23) is used. The same structure is adopted. In other words, the end of the primary side series resonant circuit (C1-N1 (L1)) is directly connected to the end of the controlled winding NR that is a high-frequency inductor to form a magnetic coupling type. It is designed to improve power factor. In such a configuration, as described above, the ripple of the commercial AC power supply period superimposed on the primary side series resonance current results in the ripple of the commercial AC power supply period superimposed on the secondary side DC output voltage Eo. growing. In the present embodiment, this point is dealt with as follows.

In the power supply circuits of FIGS. 22 to 27 shown as the prior art, the coupling coefficient k between the primary side and the secondary side in the insulating converter transformer PIT is set to be k = 0.85. It can be said that this coupling coefficient has a relatively high degree of coupling although it is not tightly coupled. Further, in the power supply circuits of FIGS. 22 to 27, the secondary side DC output voltage is stabilized by variable control of the switching frequency of the primary side switching element.
In the case of adopting such a configuration, for example, in a state where the load tends to be light, stabilization is achieved by controlling the switching frequency to be high. In this state, the secondary-side rectifier circuit operates in a so-called continuous mode in which the period during which the secondary-side rectified current flows through the secondary-side smoothing capacitor is continuous and there is no period of pause.
On the other hand, if the primary side switching frequency is controlled to decrease as the secondary side DC output voltage decreases due to a heavy load tendency, the secondary side smoothing capacitor will The so-called discontinuous mode is entered, in which the secondary side rectified current stops flowing continuously and a current discontinuous period occurs. That is, there is a state in which the discontinuous mode is set according to the load variation as the secondary-side double-wave rectification operation.
The secondary side DC output voltage varies depending on the commercial AC power supply AC (AC input voltage VAC), and a constant voltage control operation corresponding to this is also performed, so that the secondary side DC output voltage also depends on the level of the AC input voltage VAC. There will be a state of discontinuous mode.

In the power supply circuit shown in FIG. 22 (and FIG. 23), the ripple superimposed on the secondary side rectified and smoothed voltage Eo is increased because a relatively high coupling coefficient is set for the insulating converter transformer PIT as described above. Thus, the root cause is the environment where the secondary side rectification operation becomes a discontinuous mode according to the fluctuation of the secondary side DC output voltage. Under this environment, the increase in the ripple is not negligible because the end of the primary side series resonance circuit is directly connected to the high frequency inductor as in the power factor correction circuits 20 and 20A. It will be big.
In other words, the cause is eliminated if the continuous mode is maintained as the secondary side rectification operation regardless of the load fluctuation and the fluctuation of the AC input voltage VAC. This means that an increase in the ripple voltage of the commercial AC power supply cycle superimposed on the side DC output voltage can be effectively suppressed.

Therefore, in this embodiment, the power supply circuit shown in FIG. 1 has the following configuration.
FIG. 11 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 (and the low-voltage winding N4) is wound around one winding portion of the bobbin B. Further, the secondary winding N2 (N2A, N2B) is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap G having a gap length of about 1.6 mm is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with k = 0.8 or less is obtained. Note that k = 0.75 was set as the actual coupling coefficient k. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

Further, the induced voltage level per 1T (turn) of the secondary winding is also lower than that of the power supply circuit shown in FIGS. Set the number of windings (turns). For example, by setting the primary winding N1 = 45T and the secondary winding N2A = N2B = 10T, the induced voltage level per 1T (turn) of the secondary winding is 2.5 V / T or less.
In this case, as the EE type core of the insulating converter transformer PIT, the EER-35 type is selected, and the leakage inductance of the insulating converter transformer PIT and the primary side series resonance capacitor C1 forming the primary side series resonance circuit are as follows. 0.027 μF is selected.

  By setting the structure of the insulating converter transformer PIT and the number of windings of the primary winding N1 and the secondary winding (N2A, N2B), the magnetic flux density in the core of the insulating converter transformer PIT in this case decreases. As a result, the leakage inductance in the insulating converter transformer PIT of the power supply circuit shown in FIGS.

  Incidentally, the insulation converter transformer PIT of the power supply circuit as the prior art shown in FIGS. 22 to 27 has an EE core gap of 1 mm or less, and an induced voltage level per 1 T (turn) of the secondary winding. The number of turns of the primary winding N1 and the secondary winding N2 has been set so as to be 5V, thereby obtaining a degree of coupling of a coupling coefficient k = 0.85. It is a thing.

The operation of the circuit shown in FIG. 1 provided with the insulating converter transformer PIT having the configuration shown in FIG. 11 will be described with reference to the waveform diagram of FIG.
The waveform diagram of FIG. 12 shows the voltage VQ2 across the switching element Q2 in the primary side switching converter and the rectified current Is that flows into the smoothing capacitor Co in the secondary side double wave rectifier circuit. The line on which the rectified current Is is shown is a line between the connection point of each cathode of the secondary side rectifier diodes Do1 and Do2 forming the double-wave rectifier circuit and the positive terminal of the smoothing capacitor Co. Therefore, the secondary side In the rectified current path, the alternating voltage excited by the secondary winding N2 becomes a line through which the rectified current flows in both the positive and negative periods.

As described above, the power supply circuit shown in FIG. 1 is stabilized by the switching frequency control method. However, as the secondary side DC output voltage Eo decreases under heavy load conditions, the switching frequency is lowered. Control to do. In the waveform diagram of FIG. 12, the load state corresponds to the operation in the state of the maximum load power. That is, the switching frequency is almost the lowest in the control range.
The voltage VQ2 shown in FIG. 12 is a sawtooth waveform, and the period of the sawtooth waveform corresponds to the on / off timing of the switching element Q2 as described above. In other words, the ON period of the switching element Q2 is a period in which the sawtooth wave is 0 level, and the period clamped at one predetermined level is the OFF period in which the switching element Q2 is OFF. Further, the switching element Q1 is turned on / off at an alternate timing with respect to the switching element Q2, and the voltage between both ends thereof is obtained by shifting the voltage VQ2 by 180 °.

The waveform of the secondary side rectified current Is is a waveform in which a half-wave sine wave on the positive polarity side continues in accordance with the cycle timing of the voltage VQ2 across the switching element Q2. Such a waveform is obtained, for example, in one half cycle of the alternating voltage excited in the secondary winding N2, in which the rectified current rectified by the rectifier diode Do1 flows in a positive polarity, and rectified by the rectifier diode Do2 in the other half cycle. It is obtained by repeating the operation that the rectified current flows in a positive polarity.
In this case, as the rectified current Is flowing as described above, it can be seen that there is no current discontinuous period in which the zero level continues between the adjacent half-wave sine waves. That is, the rectified current Is is continuously flowing.
In this way, in this embodiment, for example, the secondary side DC output voltage Eo is lowered in response to a heavy load or a drop in the AC input voltage VAC, so that the switching frequency is controlled to be lowered. The continuous mode is obtained as the secondary side rectified current even when the

  As described above, the continuous mode is obtained even under heavy load (and low AC input voltage) conditions, as can be understood from the above description, the coupling coefficient of the insulating converter transformer PIT is determined by setting the gap length. Is reduced to a required value to achieve a more loosely coupled state, and the primary winding N1 and the secondary windings N2A, N2B are set so that, for example, the induced voltage level per turn of the secondary winding is also lower than the required level. The number of turns (number of turns) is set, and thereby the magnetic flux density generated in the core of the insulating converter transformer PIT is reduced to a required level or less.

In this way, the continuous mode is obtained even in the state of a heavy load and a low AC input voltage, regardless of the fluctuation of the secondary side DC output voltage Eo due to load fluctuation, AC input voltage fluctuation or the like. This means that the secondary side rectification operation is always performed in the continuous mode.
As a result, as in this embodiment, the switching output of the primary side series resonance circuit is directly supplied to the high-frequency inductor (controlled winding NR) as a power factor correction circuit using a power regeneration system. Even in the case of adopting such a feedback path configuration, an increase in the ripple of the commercial AC power source period superimposed on the primary side series resonance current is greatly suppressed.
This is mainly due to the suppression of the peak level of the secondary side rectification current by the secondary side rectification operation being in the continuous mode. In addition, as described above with reference to FIG. 11, since the magnetic flux density (coupling coefficient) of the insulating converter transformer PIT is set below the required value, a change occurs in the power transmission from the primary side to the secondary side, Accordingly, the transmission intensity to the secondary side of the ripple component of the commercial AC power supply period generated in the primary side series resonance current is also reduced.

As a result, the level of the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage is also reduced, and it is not necessary to increase the capacitance of the secondary side smoothing capacitor to the same level as in the prior art. That is, the practical use of the switching power supply circuit including the power factor correction circuit based on the power regeneration method can be easily realized. For example, in the power supply circuit shown in FIG. 22 and FIG. 23, the ripple voltage has increased 5 to 6 times compared to the case where the power factor correction circuit is not provided. It can be suppressed to less than. If the ripple is suppressed to this level, there will be no problem in practical use.
Specifically, in the configuration in which the power factor correction circuit 10 is omitted based on the power supply circuit shown in FIG. 1, the ripple voltage was measured as 15 mV, whereas the power factor improvement circuit 10 as shown in FIG. 20 mV was measured in a power supply circuit having a configuration including:
In this embodiment, such a ripple is suppressed by, for example, the structure of the insulating converter transformer PIT itself without winding the tertiary winding N3 around the insulating converter transformer PIT or providing the power factor improving transformer VFT. It can be said that this is realized. That is, no circuit parts are added, and it can be said that the ripple suppressing effect is obtained without particularly expanding the circuit and increasing the cost.

  Although illustration of the circuit is omitted, as described above, the secondary-side rectification operation is set to the continuous mode regardless of load fluctuations and AC input voltage VAC level fluctuations. As a circuit, it is possible to effectively reduce power loss on the secondary side by providing a winding voltage detection type synchronous rectifier circuit using a low on-resistance MOS-FET as a rectifier. This point will be described.

As described above, when a power supply circuit adopting the switching frequency control system is set to a conventional degree of coupling with a coupling coefficient k = 0.85 as an insulating converter transformer PIT, for example, a tendency of heavy load may occur. Alternatively, when the AC input voltage VAC tends to decrease, a discontinuous mode in which the secondary rectified current flowing in the secondary-side double-wave rectifier circuit does not continuously flow is set.
Such a discontinuous mode state can be said to be a state in which the secondary side rectified current flows in a period shorter than the period in which the primary side series resonance current flows. And since the rectified current flows in a short period in this way, the peak level of the rectified current at this time becomes relatively high, and accordingly, the conduction loss of each rectifier diode on the secondary side is relatively large. turn into.
Such a discontinuous mode is adopted in a configuration in which a normal double-wave rectifier circuit including a high-speed rectifier diode element such as a Schottky diode is used as the secondary rectifier circuit after adopting the switching frequency control method. Due to the continuity loss of the rectifier diode, a corresponding power loss has occurred on the secondary side.

  Therefore, as one of the techniques for reducing the secondary-side power loss due to the conduction loss of such a rectifier diode, a synchronous circuit using a low-on-resistance MOS-FET as a rectifier for the secondary-side double-wave rectifier circuit. It is known as a rectifier circuit. For example, compared to a Schottky diode or the like, the on-resistance is much smaller in a MOS-FET or the like having a trench structure. Therefore, by making the secondary side rectifier circuit a synchronous rectifier circuit, it is possible to reduce conduction loss in the rectifier element and reduce power loss on the secondary side.

  As such a synchronous rectifier circuit, for example, a resistance element for detecting an alternating voltage obtained at the secondary winding N2 (secondary winding N2A, N2B) of the insulating converter transformer PIT is provided, and the detected voltage serves as a rectifier element. The MOS-FET is turned on / off. This is also called a winding voltage detection method. This winding voltage detection system has an advantage that the circuit configuration is simplified because the drive circuit basically includes a resistance element that detects the current flowing through the secondary winding.

However, in the discontinuous mode state, even after the charging current to the smoothing capacitor becomes 0 level, the primary side series resonance current having the same polarity flows in the primary winding N1 during the discontinuous period. The polarity of the induced voltage of the winding N2 is not reversed, and during this period, the MOS-FET is not turned off and is kept on.
Since the MOS-FET is turned on even after the charging current to the smoothing capacitor becomes 0 level in this way, a reverse current flows as a rectified current during this period, and the invalidity due to the reverse current is lost. Electricity is generated.
From this, in the case of the synchronous rectifier circuit employing the winding voltage detection method, although the conduction loss in the rectifier element is reduced, the generation of reactive power due to the reverse current as described above is effective for the overall power conversion efficiency. It is difficult to improve.

  Therefore, as a technique for solving the problem of generation of reactive power due to the reverse rectified current as described above, a synchronous rectifier circuit based on a rectified current detection method is known. This rectified current detection method is a technique for turning off the MOS-FET before the rectified current charged in the secondary-side smoothing capacitor Co becomes 0 level.

As a circuit configuration for this purpose, for example, a current (rectified current) flowing through the secondary winding N2 is detected by a current transformer or the like. The current detected by the current transformer is output as a voltage (detection voltage), and this detection voltage is compared with a predetermined reference voltage by a comparator.
Here, when the rectified current starts flowing so as to charge the smoothing capacitor Co, the rectified current is detected by the current transformer, and a detection voltage corresponding to the rectified current level is input to the comparator. The comparator compares the reference voltage with the detection voltage, and outputs an H level when the detection voltage exceeds the reference voltage, for example. This H level output is applied as an on-voltage from the buffer to the gate of the MOS-FET, which is a rectifying element, to turn on the MOS-FET. As a result, the rectified current flows in the source-to-drain direction of the MOS-FET.
Then, the level of the rectified current decreases with time, and the comparator inverts the output when the detection voltage, which is the output of the current transformer, becomes lower than the reference voltage. By outputting this inverted output through the buffer, the gate capacitance of the MOS-FET which is a rectifying element is discharged, and the MOS-FET is turned off.

  With such an operation, the MOS-FET which is a rectifying element is turned off at a timing before the rectified current becomes 0 level. As a result, like the synchronous rectification circuit using the winding voltage detection method, the reverse current does not flow through the MOS-FET in the period in which the rectification current is discontinuous, and no reactive power is generated. Conversion efficiency is increased.

However, in the synchronous rectification circuit of the above-described rectification current detection system, as can be seen from the above description, at least one set of current transformers and one MOS-FET are output by the output of the current transformer, corresponding to one MOS-FET. A relatively complicated drive circuit system for driving is required. As a result, the circuit configuration becomes complicated, resulting in inconveniences such as reduced manufacturing efficiency, increased costs, and increased circuit board size.
In particular, when the configuration of a current resonance type switching converter is basically used like the power supply circuit shown in FIG. Therefore, two sets of the above-described current transformer and driving circuit system are required corresponding to each half-wave period, and the above-described problem is further increased.

  Considering the adoption of a synchronous rectifier circuit on the premise that there is a state of discontinuous mode as the secondary side rectification operation in this way, the merits of the winding voltage detection method and the rectification current detection method are Must be in a trade-off relationship. That is, the winding voltage detection method is disadvantageous in terms of power conversion efficiency, but the circuit configuration is simplified. In contrast, the rectified current detection method is advantageous in terms of power conversion efficiency because reactive power is not generated, but the circuit configuration is complicated. In other words, as long as there is a condition for the discontinuous mode as the secondary side rectification operation, for example, even if an attempt is made to adopt a synchronous rectification circuit in consideration of power conversion efficiency, a rectification current detection method must be adopted. As a result, the circuit configuration becomes complicated.

However, in this embodiment, the configuration of the insulating converter transformer PIT described above makes the secondary-side double-wave rectification operation a continuous mode regardless of load fluctuations and AC input voltage fluctuations. .
Therefore, even if a synchronous rectifier circuit based on a winding voltage detection system is provided as a secondary side rectifier circuit of the power supply circuit shown in FIG. 1, a continuous mode is always obtained as described above, so that a current discontinuity is obtained. This means that no reactive power will be generated for the period.
That is, in the case of the power supply circuit shown in FIG. 1, if the secondary side rectifier circuit is provided with a synchronous rectifier circuit of the winding voltage detection system, the circuit scale can be suppressed as a simple circuit configuration, and the cost can be further reduced. While avoiding the increase, the problem of a decrease in power conversion efficiency due to the reactive power in the current discontinuous period is effectively solved.

Next, FIG. 13 shows a configuration example of a switching power supply circuit as a second embodiment of the present invention.
The power supply circuit shown in this figure is the same as the power factor correction circuit 10 provided in the power supply circuit of the first embodiment when the rectifying and smoothing circuit for generating the rectified and smoothed voltage Ei is a voltage doubler rectifying circuit. It is the applied configuration. In this figure, the same parts as those in FIG.

In the power factor correction circuit 11 shown in this figure, rectifier diodes D11 and D12 (first rectifier and second rectifier) and smoothing capacitors Ci1 and Ci2 (first) are used to form a voltage doubler rectifier circuit. Smoothing capacitor, second smoothing capacitor).
In order to improve the power factor, a control transformer PRT and a filter capacitor CN are provided. Further, a first amplifier circuit (Q12, Q13, R13 to R18, R5, ZD5) for obtaining a control current Ic (control current Ic1) corresponding to the level of the rectified smoothing voltage Ei (AC input voltage VAC), detection winding A half-wave rectifier circuit (D5, C5) and a second amplifier circuit (Q11, R10 to R12) for obtaining a control current Ic corresponding to the voltage across the line ND (detection voltage VD) are provided. The power factor improving switching element includes the rectifier diodes D11 and D12. That is, the rectifier diodes D11 and D12 in this case serve both as a rectifier diode for obtaining a rectified output in the voltage doubler rectifier circuit and as a power factor improving switching element for switching the rectified current. Since the rectifier diodes D11 and D12 switch the rectified current in the switching cycle of the switching elements Q1 and Q2, the high-speed recovery type is selected as the high-speed type diode element.
The control transformer PRT has the same core and winding structure (see FIGS. 3 to 5) as in the first embodiment, so that the control winding NR, the detection winding ND, and the control winding Nc Configured as a wound saturable reactor. Also in this case, the control winding NR and the detection winding ND are tightly coupled.

  Note that the power supply circuit of the second embodiment is based on the configuration of the voltage doubler rectifier circuit as described above, and therefore a configuration corresponding to the wide range as in the case of the first embodiment is adopted. The following explanation will be continued as if it were not.

  In this case, the common mode noise filter provided for the commercial AC power supply AC is composed of a set of each of the common mode choke coil CMC and the filter capacitor CL. Connected to.

In the power factor correction circuit 11, the controlled winding N R of the control transformer PRT is provided with a tap at a predetermined winding position, so that the winding N R1 (first inductor: first winding) , N R2 (second inductor portion: second winding portion).
The end of the controlled winding NR on the winding portion NR1 side is connected to one line of the commercial AC power supply AC that is the subsequent stage of the common mode noise filter. The tap of the controlled winding NR (which becomes a connection point between the winding portions NR1 and NR2) is connected to the connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12.
The cathode of the rectifier diode D11 is connected to the positive terminal of the smoothing capacitor Ci1, and the anode of the rectifier diode D12 is connected to the primary side ground.

Two sets of smoothing capacitors Ci1 and Ci2 are connected in series. In addition, the positive terminal of the smoothing capacitor Ci1 is connected to the cathode of the rectifier diode D11 as described above, and is also connected to the drain side of the switching element Q1. The negative terminal of the smoothing capacitor Ci2 is connected to the primary side ground. A connection point between the negative electrode terminal of the smoothing capacitor Ci1 and the positive electrode terminal of the smoothing capacitor Ci2 is connected to the other line of the commercial AC power supply AC, which is a subsequent stage of the common mode noise filter.
The filter capacitor CN is inserted between the connection point between the end of the controlled winding NR on the winding part NR1 side and the line of the commercial AC power supply AC and the connection point between the smoothing capacitor Ci1 and the smoothing capacitor Ci2. .
The end of the primary side series resonance circuit (C1-N1 (L1)) is connected to the connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12 via the winding part NR2.

The voltage doubler rectifier circuit formed in the power factor correction circuit 11 constituted by the above-described connection form is connected to the commercial AC power supply AC → during a period in which the commercial AC power supply AC (AC input voltage VAC) is in one half cycle. (CMC winding) → winding portion NR1 → rectifier diode D11 → smoothing capacitor Ci1 → (CMC winding) → commutation current flows through the path of commercial AC power supply AC. In other words, the rectifier diode D11 rectifies the commercial AC power supply AC, and the smoothing capacitor Ci1 smoothes the rectified output, so that the rectified and smoothed voltage corresponding to the same size as the commercial AC power supply AC is obtained as the voltage across the smoothing capacitor Ci1. Generate.
Similarly, during a period in which the commercial AC power supply AC (AC input voltage VAC) is in the other half cycle, the commercial AC power supply AC → (winding of CMC) → smoothing capacitor Ci2 → rectifier diode D12 → winding portion N R1 → ( CMC winding) → Rectified current flows through the path of commercial AC power supply AC. That is, the rectifier diode D12 rectifies the commercial AC power supply AC, and the smoothing capacitor Ci2 smoothes the rectified output, so that the rectified and smoothed voltage corresponding to the same size as the commercial AC power supply AC is obtained as the voltage across the smoothing capacitor Ci2. Generate.

As a result, a rectified and smoothed voltage Ei having a level corresponding to twice the level of the commercial AC power supply AC is obtained as the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. The rectified and smoothed voltage Ei is supplied as a DC input voltage to the subsequent switching converter.
Further, according to the above description, the winding part NR1 is inserted in a common line in the rectification current path formed every half cycle of the commercial AC power supply AC (AC input voltage VAC). Become.

In the power factor correction circuit 11, the primary series resonant circuit (C1-N1 (L1)) is connected to the connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12 via the winding NR2. The ends are connected. As a result, the power factor correction circuit 11 also regenerates the primary side series resonance current flowing in the primary side series resonance circuit (C1-N1 (L1)) as electric power, and the controlled winding NR (NR1) that is a high frequency inductor. The operation of returning to the smoothing capacitor Ci via the rectification current path of the rectifier diode D11 or the rectifier diode D12 via the magnetic coupling is obtained.
In other words, the power factor improving circuit 11 also obtains the power factor improving operation by the magnetic coupling type power regeneration system.

  Also in the power factor correction circuit 11, a control system comprising a first transformer, a second amplifier, and a control transformer PRT around which a detection winding ND is wound, as in the case of the circuit of FIG. As in the case of the power factor correction circuit 10, the power factor is controlled to be constant with respect to fluctuations in both the AC input voltage VAC and the load power.

  Further, according to the configuration of the power factor correction circuit 11 shown in FIG. 13, in the controlled winding NR, the operation of inputting the switching output of the primary side series resonance circuit on the winding portion NR2 side, and the winding portion On the NR1 side, an operation in which a commercial AC power supply is superimposed on the switching output generated by power regeneration is obtained in a composite manner. Since the winding portions NR1 and NR2 are divided by forming taps on the windings of the inductor that is originally configured as one component element, the winding portions NR1 and NR2 are in a serial connection relationship. Are tightly coupled to each other. Therefore, the operations of the winding portions NR1 and NR2 described above are performed in a state where the winding portions NR1 and NR2 are tightly coupled.

As a result of obtaining such an operation, it has been experimentally confirmed that the rectified current that flows in the power factor correction circuit 11 including the controlled winding NR approaches a more perfect sine wave shape. As the rectified current becomes sinusoidal in this way, the AC input current IAC also approaches an almost perfect sinusoidal shape in the same manner. Depending on this, higher order and odd order distortion levels such as 9th order, 11th order, 13th order, etc., which are assumed to be generated in the waveform of the AC input current IAC, are reduced.
For example, high odd-order distortion as described above may cause a disadvantage that the margin for the regulation value defined by the power supply harmonic distortion regulation is reduced. Since the waveform of the AC input current IAC becomes a more complete sine wave, it is possible to obtain a sufficient margin for the regulation value defined by the power supply harmonic distortion regulation as described above. In order to obtain such an effect, a tap is provided for the inductor as the controlled winding NR of the control transformer PRT without adding a component element as a configuration for improving the power factor. As shown in FIG. 13, connection may be made in the power factor correction circuit 11.

Waveform diagrams corresponding to the power factor correction operation of the power factor correction circuit 11 according to the above description are shown in FIGS.
In this case, unlike the power supply circuit shown in FIG. 1, the power supply circuit shown in FIG. 13 corresponds to the condition of AC input voltage VAC = 85 V to 144 V and load power Po = 0 W to 150 W. In addition, FIG. 14 shows an operation when the maximum load power Po = 150 W, and FIG. 15 shows an operation under a relatively light load condition when the load power Po = 25 W.
Here again, it is assumed that the switching elements Q1 and Q2 perform switching under the state in which the AC input voltage VAC is input as shown in FIGS. Accordingly, an operation of regenerating and returning the switching output from the primary side series resonance circuit to the power factor correction circuit 11 is obtained.
Accordingly, as shown in FIGS. 14 and 15, the voltage V1 across the rectifier diode D12 is superimposed with the alternating voltage of the switching period during the period when the rectifier diode D12 is non-conductive (switching stop period). In the conduction period of the rectifier diode D12 (period in which the rectified current is switched), the waveform is clamped at a predetermined level. Note that the voltage across the rectifier diode D11 is substantially similar to the voltage V1 across the rectifier diode D12 and has a waveform that is shifted by 180 °.

As the AC input current IAC, the rectifier diodes D11 and D12 flow in accordance with the switching period of the rectified current. As understood from the above description, the AC input current IAC The conduction angle is enlarged compared to the case where the power factor correction circuit is not provided, and the power factor is improved.
Furthermore, as described above, the actual detailed waveform of the AC input current IAC is compared with the case where the controlled winding NR is not divided into the power factor correction circuit and is completely inserted into the rectified current path. Thus, the amplitude of the odd harmonic component is suppressed, and a waveform shape closer to a half-wave sine wave is obtained.
In order to obtain a circuit configuration in which the controlled winding NR is not divided in the power factor correction circuit 11, the connection point between the cathode of the rectifier diode D11 and the anode of the rectifier diode D12 is not a tap of the controlled winding NR. What is necessary is just to connect with the series resonance capacitor C1.

16 and 17 show the stabilization characteristics of the power factor PF in the power factor correction circuit 11. FIG. 16 shows change characteristics of the power factor PF and the control current Ic with respect to fluctuations in the AC input voltage VAC (the load power Po is fixed). FIG. 17 shows change characteristics of the power factor PF and the control current Ic with respect to load power fluctuation (fixed at AC input voltage VAC = 100 V). In FIG. 16, each characteristic shows the result when the solid line is constant at a load power Po = 150 W, and the broken line is the result when the load power Po = 25 W is constant.
First, as shown in FIG. 16, the level of the control current Ic is linear at both load power Po = 150 W and Po = 25 W with respect to the fluctuation of the AC input voltage VAC = 85V to 144V. It has dropped to. As a result of the control current Ic being variably controlled in this way, the inductance of the controlled winding NR changes. As a result, the power factor PF also becomes constant regardless of the fluctuation of the AC input voltage VAC. It can be seen that
For such fluctuations in the AC input voltage VAC, a practically sufficient power factor PF = 0.80 in the range of VAC = 85 to 144 V at both load power Po = 150 W and Po = 25 W. Results were obtained that could be constant in rate.

In addition, when the AC input voltage VAC = 100 V in FIG. 17, the level of the control current Ic decreases linearly in this case with respect to the fluctuation from the maximum load power Po = 150 W to the minimum load power Po = 0 W (no load). To do. In accordance with such variable control of the control current Ic, the power factor PF is controlled so as to be constant with respect to the load fluctuation. In this case, the load power Po = 150 W to 25 W, and similarly PF = 0. A constant result is obtained at .80.
In this case, the variable range of the control current Ic level is 5 mA to 75 mA.

  From such a result, in the power supply circuit of the second embodiment, the power supply circuit is effective against a relatively wide range of fluctuation between the range of the AC input voltage VAC = 85 to 144 V and the range of the load power Po = 150 W to 25 W. It can be seen that the rate can be controlled to be constant at a practically sufficient level.

In addition, the power supply circuit shown in FIG. 13 also includes the insulating converter transformer PIT having the configuration described with reference to FIG. As a result, the ripple superimposed on the secondary side DC output voltage Eo is reduced to less than twice. In the power supply circuit in which the power factor correction circuit 11 is omitted based on the circuit shown in FIG. 20 mV was measured.
In addition, since the continuous mode is maintained as the secondary side rectification operation regardless of the fluctuation of the secondary side DC output voltage Eo, it is simplified if a synchronous rectification circuit using a winding voltage detection system is provided. It is possible to effectively reduce the power loss on the secondary side.

In the power supply circuit shown in FIG. 13, for example, constants are set for the main parts as follows. The results shown in FIGS. 14 to 17 are obtained by experimenting with the following constants set.
For the insulating converter transformer PIT, EER-40 was selected for the core, and the gap G was 1.6 mm. The primary winding N1 = 60T and the secondary winding N2A = N2B = 10T. Thereby, the coupling coefficient k = 0.75 is obtained.
In addition, 0.018 μF was selected for the primary side series resonant capacitor C1.
When the configuration of FIG. 8 is adopted as the control transformer PRT, the size of a × b × c shown in FIG. It is possible to select 20 mm, gap G = 25 μm, detection winding ND = 1T, and control winding Nc = 1000T. In addition, the controlled winding NR is divided so that the winding portion NR1 = 13T and the winding portion NR2 = 5T.

By the way, as a modification of the power factor correction circuit 11 shown in FIG. 13, description by illustration is omitted. However, instead of dividing the controlled winding N R, the primary winding N 1 is divided and the same as described above. Thus, it is possible to obtain a better power factor improvement result by making the waveform of the AC input current IAC closer to a perfect sine wave.
In such a configuration, first, a predetermined winding position of the primary winding N1 is tapped to be divided into primary winding portions N1A and N1B. In this case, the controlled winding NR is not divided. The tap is connected to a connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12. Further, the end of the primary winding N1 on the primary winding portion N1B side is connected to the controlled winding NR. Thus, in one half cycle of the AC input voltage VAC in the rectified current path of the voltage doubler rectifier circuit, a series connection circuit of the rectifier diode D11-primary winding portion N1B-controlled winding NR is inserted, and the AC input voltage In the other half cycle of VAC, a series connection circuit of rectifier diode D12-primary winding portion N1B-controlled winding NR is inserted. That is, the series connection circuit of the primary winding N1B and the controlled winding NR is inserted into a common line in the rectified current path formed every half cycle of the commercial AC power supply AC (AC input voltage VAC).
In this case, as the primary side series resonance circuit, the end of the primary winding N1A is connected to the connection point between the anode of the rectifier diode D11 and the cathode of the rectifier diode D12. The resonant capacitor C1 is inserted between the end of the primary winding N1A of the primary winding N1 and the switching output point.

  In the case of the power factor correction circuit 11 shown in FIG. 13, of the winding portions NR1 (first inductor portion) and NR2 (second inductor portion) formed by tapping the controlled winding NR, the winding portion NR2 Is not inserted in the rectified current path, but is connected in series with the primary winding N1 via the series resonant capacitor C1. Therefore, it functions as an inductance component forming the primary side series resonance circuit. The winding portion NR2 transmits the switching output of the primary side series resonance circuit to the winding portion NR1 which is assumed to be in a tightly coupled relationship. Thus, the power regeneration operation in which the commercial AC power is superimposed on the switching output is obtained.

On the other hand, even when the primary winding N1 is divided and the primary windings N1A and N1B are formed as described above as the configuration of the power factor correction circuit 11, the primary winding N1B is the primary side series resonance circuit. Is connected in series with the controlled winding NR, which is a high-frequency inductor, so that it can be almost tightly coupled to the controlled winding NR. This means that the inductance of the primary winding N1B may be regarded as a part of the inductance component that forms a high-frequency inductor for power factor improvement. Then, with such a relationship being obtained, the switching output is transmitted from the primary winding portion N1B to the controlled winding NR, which is similar to the power factor correction circuit 11 as shown in FIG. Power regeneration operation can be obtained.
From such an operation, the relationship between the winding part NR1 and the winding part NR2 in the power factor correction circuit 11 as shown in FIG. 13 is that the controlled winding NR (first stage) when the primary winding N1 is divided. It can be said that this is equivalent to the relationship between the first inductor portion) and the primary winding portion N1B (second inductor portion).

As described above, the rectified current approaches a more complete sine waveform by the operation of the winding part NR1 and the winding part NR2 in the power factor correction circuit 11, and accordingly, the AC input current IAC is also complete. It approaches a sine waveform. Even in the configuration in which the primary winding N1 is divided, such an operation can be obtained by the controlled winding NR and the primary winding portion N1B. Therefore, the level of distortion of odd harmonics is suppressed. Will approach a more complete sine wave.
The configuration of the power factor correction circuit that divides the controlled winding NR or the configuration of the power factor improvement circuit that uses the winding obtained by dividing the primary winding N1 is shown in FIG. The present invention can also be applied to the power supply circuit of the first embodiment.

FIG. 18 is a circuit diagram showing a configuration of a switching power supply circuit according to the third embodiment of the present invention.
In the power supply circuit according to the third embodiment, the rectifying / smoothing circuit for the commercial AC power supply AC is a full-wave rectifier circuit as in the circuit of FIG. Then, as the power factor improving circuit 12 shown in the figure, in the power factor improving circuit 10 of the circuit of FIG. 1, in the series connection circuit of the controlled winding N R of the switching diode D 1 and the control transformer PRT, one end of the controlled winding N R Is connected to the positive terminal of the smoothing capacitor Ci, as shown in the figure, the cathode of the switching diode D1 is connected to the positive terminal of the smoothing capacitor Ci.
According to such a connection form, as a path for feeding back the primary side series resonance current Io, a path flowing into the smoothing capacitor Ci through the switching diode D1, and a smoothing through the controlled winding NR → filter capacitor Cn. Two paths are formed, the path flowing into the capacitor Ci.
The power regeneration in this case is also performed via a diode, and here, such a configuration is referred to as a diode coupling method.
In FIG. 18, the first amplifier circuit system for controlling the control current Ic level according to the level of the rectified and smoothed voltage Ei obtained in the resistors R17 to R18 shown in FIG. 1, and the detection voltage of the detection winding ND. A second amplifier circuit system that controls the control current Ic level according to the level is shown as a power factor control circuit 13.
Further, other parts in the circuit of FIG. 18 are the same as those of FIG. 1, and therefore, the same reference numerals are given here and description thereof is omitted.

A waveform diagram corresponding to the power factor correction operation by the power factor correction circuit 12 in this case is shown in FIG.
In order to obtain the results shown in this figure, each part of the circuit of FIG. 18 was selected as follows.
Insulating converter transformer PIT: EER-35 type core, gap G = 1.6 mm, coupling coefficient k = 0.75
Primary winding N1 = 45T, secondary winding N2A = N2B = 10T
・ Primary side series resonant capacitor C1 = 0.027μF
Control transformer PRT: 16 mm × 16 mm × 23 mm for a × b × c size shown in FIG. 8, gap G = 25 μm, control winding Nc = 1000T, controlled winding N R = 18T

In FIG. 19, the switching elements Q1 and Q2 are also switched under the condition where the AC input voltage VAC is input as shown in FIG. In response to this, an operation of regenerating and returning the switching output from the primary side series resonance circuit to the power factor correction circuit 12 is obtained.
The voltage V1 obtained between the connection point of the controlled winding NR and the anode of the switching diode D1 and the primary side ground is an alternating voltage corresponding to the switching output with respect to the rectified output voltage component by the bridge rectifier circuit Di. A waveform in which is superimposed is obtained. The peak level of the voltage V1 in this case is clamped at the level of the rectified and smoothed voltage Ei as shown.

  By applying such a voltage V1, in this case as well, the switching diode D1 performs the switching operation such that the rectified current is interrupted. As a result, the current I1 having the waveform shown in the figure flows through the switching diode D1.

In this figure, the waveform of the current IR flowing through the controlled winding NR is also shown.
Here, in this case, as described above, as a path for feeding back the primary side series resonance current Io, there are two paths: a path via the switching diode D1 and a path via the controlled winding NR. Is formed.
According to this, during the period in which the switching operation of the switching diode D1 is not performed and the current I1 does not flow, an alternating waveform according to the switching period is obtained in the current IR as shown in the figure. The primary side series resonance current Io flows only through the path via N R. When the switching diode D1 starts the switching operation, it can be seen that the primary side series resonance current Io gradually diverts to the switching diode D1 side as can be seen by comparing the currents I1 and IR.
In this way, the primary side series resonance current Io in this case flows in a divided manner between the path on the switching diode D1 side and the path on the controlled winding NR side according to the period during which the switching diode D1 performs the switching operation. It has become a thing.
In this case, the same 14A is obtained as the peak level (Ap) of the current I1 and the level (Ap-p) between the positive and negative peak levels of the current IR shown in the figure. From this, it can be understood that the current I1 and the current IR are obtained by diverting the same primary side series resonance current Io.

In the waveform of the current IR obtained as described above, the component of the primary side series resonance current Io indicated by the alternating waveform flows into the smoothing capacitor Ci by flowing through the filter capacitor Cn. Therefore, in this case, as the rectified current flowing through the bridge rectifier circuit Di, a waveform substantially corresponding to the waveform obtained by removing the alternating waveform from the current IR is obtained.
When such a rectified current waveform is obtained, the AC input current IAC also flows with a waveform as shown in the figure. In other words, the conduction period of the AC input current IAC is a period that substantially matches the conduction period of the portion obtained by removing the alternating waveform from the current IR.
As shown in the figure, the peak level of the AC input current IAC in this case is 9 Ap.

  In this case as well, the switching diode D1 performs a switching operation during a period in which the level of the AC input voltage VAC is equal to or higher than ½ of the peak level, so that the rectified current flows. The conduction angle is expanded as compared with the case where the power factor correction circuit is not provided. If the conduction angle of the rectified current is increased in this way, the conduction angle of the AC input current IAC is also increased, thereby improving the power factor in the circuit of FIG.

As shown in FIG. 19, in the case of the third embodiment adopting a diode-coupled configuration, the voltage V1 applied to the switching diode D1 is switched to the rectified output voltage of the bridge rectifier circuit Di. Therefore, the peak level is clamped at the level of the rectified and smoothed voltage Ei.
On the other hand, in the case of the magnetic coupling type shown in FIG. 1, the voltage V1 applied to the switching diode D1 is a switching output with respect to the rectified and smoothed voltage Ei as described in FIGS. Since the corresponding alternating voltage component is superimposed, the peak level exceeds the level of the rectified and smoothed voltage Ei.
The voltage V1 in the case of the circuit of FIG. 18 can be understood with reference to the waveform of FIG. As the alternating voltage component superimposed on the rectified output voltage, the peak level of the switching output by the switching elements Q1 and Q2 is clamped at the level of the rectified and smoothed voltage Ei. Never exceed. Therefore, in the case of the circuit of FIG. 18, the peak level of the voltage V1 does not exceed the level of the rectified and smoothed voltage Ei.

Therefore, if the diode-coupled configuration shown in FIG. 18 is adopted, the level of the voltage V1 applied to the switching diode D1 is set to be higher than that in the magnetic-coupled configuration shown in FIG. Can be reduced.
According to this, a low breakdown voltage product can be selected as the switching diode D1 as compared with the case of the circuit of FIG. 1, and the circuit manufacturing cost can be reduced accordingly. Specifically, in the case of the circuit of FIG. 1, a 600V withstand voltage product is used, but in the case of the circuit of FIG. 18, a 400V withstand voltage product can be selected.

20 and 21 show rectified smoothing voltage Ei, control current Ic, and power factor variation characteristics as power factor improvement characteristics of the power supply circuit shown in FIG.
As in the case of the circuit of FIG. 1, the circuit of FIG. 18 has a configuration corresponding to fluctuations in the AC input voltage VAC = 85 V to 288 V and fluctuations in the load power Po = 100 W to 0 W. In FIG. 20, each characteristic with respect to the fluctuation | variation of alternating current input voltage VAC = 85V-288V at the time of load electric power Po = 100W is shown. FIG. 21 shows characteristics with respect to fluctuations in the load power Po = 100 W to 0 W. The characteristics when the AC input voltage VAC is 100 V are indicated by solid lines, and the characteristics when the AC input voltage VAC is 230 V are indicated by broken lines.
The experimental results shown in these figures are also obtained by selecting each part of the circuit in FIG. 18 in the same manner as described in FIG.

  First, in FIG. 20, in this case as well, the level of the control current Ic decreases as the AC input voltage VAC increases. It is shown that the power factor PF is controlled to be constant as a result of an increase in the inductance of the controlled winding NR as the control current level decreases. In other words, the control is performed so that the control current Ic level is decreased in accordance with the level increase of the rectified smoothing voltage Ei accompanying the increase of the AC input voltage VAC. It is what has been broken.

Further, according to FIG. 21, also in this case, when the AC input voltage VAC = 100V and VAC = 230V, the level of the control current Ic is varied from the load power Po = 100W to the load power Po = 0W (no load). Also decreases. And it turns out that the power factor PF is made constant according to the fall of the control current Ic level accompanying the fall of such load electric power Po. In this case, the control range of the control current Ic is 70 mA to 0 A.
According to the experiment, the power factor PF in this case was constant at the power factor PF = 0.80 in the range of the load power Po = 100 W to 5 W when the AC input voltage VAC = 100V. Further, when the AC input voltage VAC = 230 V, the power factor PF = 0.80 was also constant in the range of the load power Po = 100 W to 20 W.
With such a characteristic, a sufficient power factor can be obtained even from the maximum to the minimum load power fluctuation range. For example, the power supply harmonic distortion regulation value must be cleared with a sufficient margin. Can do.
In FIG. 21, the rectified and smoothed voltage Ei has a tendency to slightly increase with respect to a decrease from around Po = 25 W to no load, and is almost constant with respect to an increase to 100 W.

Further, although explanation by illustration is omitted, according to experiments, the circuit of FIG. 18 adopting the diode coupling type has an AC → DC power conversion efficiency of 0.4 than the case of the magnetic coupling type circuit of FIG. As a result, it is improved by about%.
This is because, in the case of the diode coupled type, the primary-side series resonance current Io can be branched and flowed as described above, and accordingly, power regeneration can be efficiently performed on the smoothing capacitor Ci. Conceivable. Incidentally, the AC → DC power conversion efficiency of the circuit of FIG. 18 is ηAC → DC = 90.0% when the load power Po = 100 W and the AC input voltage VAC = 100 V, and ηAC → DC = 86 when the AC input voltage VAC = 230. .8% was obtained.

The present invention should not be limited to the embodiments described so far.
For example, as for the insulating converter transformer PIT, the structure thereof including the core type may be appropriately changed as long as the magnetic flux density is less than the required one.
In addition, for example, the switching converter according to each of the above embodiments is based on a separately excited current resonance converter, but may be configured to include a self-excited current resonance converter. In this case, for example, a bipolar transistor can be selected as the switching element. Furthermore, the present invention can also be applied to a current resonance type converter in which four stone switching elements are full-bridge coupled. For example, as the primary side switching elements (Q1, Q2) of the switching converter, an element other than a MOS-FET may be employed as long as it is an element that can be used by another excitation type, such as an IGBT (Insulated Gate Bipolar Transistor). Absent. Also, the constants of the component elements described above may be changed according to actual conditions.
Furthermore, the power factor correction circuit is not limited to the one shown in the embodiment, and is based on circuit forms based on various power regeneration (feedback) systems that the applicant has proposed so far. It is also possible to adopt.

1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a first embodiment of the present invention. It is a wave form diagram which shows the operation waveform at the time of alternating input voltage VAC = 100V as an operation waveform of the principal part in the power supply circuit of 1st Embodiment. It is a wave form diagram which shows the operation waveform at the time of alternating input voltage VAC = 230V as an operation waveform of the principal part in the power supply circuit of 1st Embodiment. FIG. 3 is a characteristic diagram illustrating a power factor variation characteristic with respect to a variation in AC input voltage of the power supply circuit according to the first embodiment and a variation characteristic of a control current level flowing in a control winding. It is the characteristic view shown about each variation characteristic of the power factor with respect to the load variation of the power supply circuit of 1st Embodiment, and a control current. FIG. 4 is a waveform diagram showing an operation waveform at the time of an AC input voltage VAC = 100 V as a waveform diagram for explaining an operation corresponding to a wide range obtained in the power supply circuit of the first embodiment. FIG. 4 is a waveform diagram showing an operation waveform at the time of an AC input voltage VAC = 230 V as a waveform diagram for explaining an operation corresponding to a wide range obtained in the power supply circuit according to the first embodiment. It is a figure which shows the structural example of the control transformer with which the power supply circuit of embodiment is equipped. It is a figure which shows the structural example of the control transformer with which the power supply circuit of embodiment is equipped. It is a figure which shows the structural example of the control transformer with which the power supply circuit of embodiment is equipped. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a wave form diagram which shows the rectification operation | movement of the secondary side double wave rectifier circuit at the time of heavy load in the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit as 2nd Embodiment. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement (at the time of maximum load electric power) of the power supply circuit of 2nd Embodiment. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement (at the time of light load) of the power supply circuit of 2nd Embodiment. It is a figure which shows the characteristic of the control current and power factor with respect to alternating current input voltage fluctuation | variation as a power factor stabilization characteristic about the power supply circuit of 2nd Embodiment. It is a figure which shows the characteristic of the control current and power factor with respect to load electric power fluctuation | variation as a power factor stabilization characteristic about the power supply circuit of 2nd Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit as 3rd Embodiment. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the power supply circuit of 3rd Embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage, control current, and power factor with respect to alternating current input voltage fluctuation | variation as a characteristic view about the power supply circuit of 3rd Embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage, control current, and power factor with respect to load electric power fluctuation | variation as a characteristic view about the power supply circuit of 3rd Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Control circuit, 2 Oscillation drive circuit 10, 11, 12 Power factor improvement circuit, 13 Power factor control circuit, Di bridge rectifier circuit, D11, D12 rectifier diode (fast recovery type), Ci, Ci1, Ci2 smoothing capacitor, Q1, Q2 switching element, PIT isolation converter transformer, C1 primary side series resonance capacitor, Cp partial resonance capacitor, N1 primary winding, N2 (N2A, N2B) secondary winding, CN filter capacitor, D1 switching diode, PRT control transformer , NR controlled winding (high frequency inductor), NR1, NR2 winding, ND detection winding, control winding Nc, Q11, Q12, Q13 transistor

Claims (7)

Rectifying and smoothing means for inputting a commercial AC power source and generating a rectified and smoothed voltage;
Switching means formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An isolated converter transformer,
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonant capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage;
The switching drive means is controlled according to the level of the secondary side DC output voltage, and the switching frequency of the switching means is varied to perform constant voltage control on the secondary side DC output voltage. Constant voltage control means;
Power factor improvement means,
The power factor improving means is
A series connection circuit comprising a power factor improving switching element and a high frequency inductor, the power factor improving series connection circuit inserted in series with respect to the rectified current path formed by the rectifying and smoothing means;
A saturable reactor in which the high-frequency inductor is a controlled winding, a detection winding wound in a tightly coupled state with respect to the controlled winding, and a control winding. A control transformer that operates so as to vary the inductance of the controlled winding according to the level of the control current that is a direct current flowing in the winding;
A first control current generating circuit that operates to flow the control current at a level corresponding to the level of the commercial AC power supply;
A second control current generation circuit that operates to flow the control current at a level corresponding to a voltage level induced in the detection winding from the controlled winding;
An end of the primary side series resonance circuit is connected to a connection point between the power factor improving switching element and the high frequency inductor,
The insulation converter transformer is
Regardless of the fluctuation of the secondary side DC output voltage, the secondary side rectified current is set to the continuous mode, and the magnetic flux density is set to be a predetermined value or less.
A switching power supply circuit. In the series connection circuit for power factor improvement, one end of the high-frequency inductor as the controlled winding is connected to a positive terminal of a smoothing capacitor provided in the rectifying and smoothing means.
The switching power supply circuit according to claim 1. The power factor improving series connection circuit includes a high-speed diode element as the power factor improving switching element, and a cathode of the diode element is connected to a positive terminal of a smoothing capacitor included in the rectifying and smoothing means. ,
The switching power supply circuit according to claim 1. The rectifying and smoothing means is
A first rectifying element that rectifies the commercial AC power supply corresponding to one half cycle of the commercial AC power supply, and a first smoothing capacitor that generates a DC voltage by smoothing the rectified output of the first rectifying element; A second rectifying element for rectifying the commercial AC power supply corresponding to the other half cycle of the commercial AC power supply, and a second smoothing capacitor for smoothing the rectified output of the second rectifying element to generate a DC voltage And with
A voltage doubler rectifier circuit configured to generate a voltage across the series connection of the first smoothing capacitor and the second smoothing capacitor as the rectified smoothing voltage;
The power factor improving means is
The power factor improving switching element includes the first rectifying element and the second rectifying element by a high-speed diode element,
The high-frequency inductor includes a rectifying current path of the rectifying / smoothing means formed corresponding to one half cycle of the commercial AC power source and the rectifying / smoothing formed corresponding to the other half cycle of the commercial AC power source. Inserted into a common line in the rectified current path of the means,
The switching power supply circuit according to claim 1. The power factor improving means is
As an inductance component of the high frequency inductor,
The first inductor portion provided so as to form a series connection circuit with the power factor improving switching element, and the first inductor portion have a connection relationship connected in series, and the primary side series resonance circuit is A second inductor portion inserted so as to be included in an inductance component for forming,
The switching power supply circuit according to claim 1. The first inductor portion and the second inductor portion are each formed by providing a tap for one high-frequency inductor winding so that the winding of the high-frequency inductor is divided by the tap. A winding part and a second winding part,
The switching power supply circuit according to claim 5. The first inductor portion is provided as one inductor element,
The second inductor portion is one of winding portions formed so as to divide the primary winding by the tap by providing a tap for the primary winding of the insulating converter transformer.
The switching power supply circuit according to claim 5.

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